Ideal Flyback Topology

Ionel Jitaru1, Andrei Savu2, Bogdan Jitariu2 1 Rompower Energy Systems Inc., United States of America 2 Rompower International SRL, Romania

Abstract Two simple circuit modification which are designed to correct the main drawbacks of the Flyback topology are presented. The flyback topology it is the simplest topology used in power conversion field and as result has the largest utilization, especially in low and medium power applications. The first drawback of the flyback topology is the hard switching operation. This leads to high switching losses and also leads to large voltage spikes across the secondary rectifier, at the time when the primary switch turns on. Another drawback of the conventional flyback topology is the lack of utilization of the leakage inductance energy from the transformer which creates voltage spikes across the primary switch when the primary switch turns off. In this paper are presented two simple modification of the conventional flyback topology which converts the flyback topology in a soft switching topology, wherein the primary switch turns on at zero voltage in any operating conditions, and the energy from the leakage inductance which is not recycled in the secondary is used to discharge the parasitic capacitance across the main switch creating zero voltage switching conditions.

I. Introduction The flyback topology is, arguably, one of the most used circuit topologies in the field of power conversion, especially in lower to medium power application (such as AC-DC adapters, for example). The reason for such a high level of utilization of the flyback topology is rooted in its simplicity and low cost of implementation, as well as in the fact that it can operate efficiency over a very large range of input voltage. The AC-DC adapters with power under 70W, in order to gain the capability of operating all over the world, requires an operation from an alternating input voltage ranging from 90Vac to 264Vac. To meet this requirement the DC-DC converter should be able to operate efficiently in a range of 3:1, from 127Vdc to 375Vdc.

In addition to that, the new standards for power delivery requires that the AC-DC adapters provide an output voltage ranging from 3.3V to 20V which is a range of 6:1. The forward-derived topologies (such as, for example, half-bridge topology, two-transistor forward topology, full bridge topology, to name just a few) are not able to

operate efficiently over such large input - output voltage range.

The trend for miniaturization of portable equipment (for example, portable computing devices such as laptops and tablets) extends this demand even further, as a result, the AC-DC adapters are also under the pressure of miniaturization. Presently, most of the laptops and tablets require, for operation, power ranging from 30W to 65W. The ability to reduce the size of the AC-DC adapters while maintaining the convection-based cooling methodology used today requires some significant improvement in efficiency of the adapters as well as a decrease in the size of the magnetic and capacitive storage elements.

Over the years, the efficiency of the AC-DC adapters has been increased from about 70% to about 89-90% (in the most recent products such as the Apple 30W adapter, for example), mostly due to the significant progress in semiconductor industry, intelligent controllers and a better understanding of magnetic technology.

II. The Impact of the Leakage Inductance

One of the major drawbacks of Flyback topology is the fact that the energy of transformer’ leakage inductance between the primary and the secondary winding in the flyback it is not naturally recycled like in other topologies, such as the two-transistor forward topology, or full bridge phase shifted topology. In Figure 1A is presented the simplified schematic of a flyback topology wherein the leakage inductance, Llk is depicted as a discreate element of the transformer Tr1.

Fig. 1A: Simplified Schematic of a Flyback topology.

At the time wherein the main switch M1 turns off the energy stored in the magnetizing inductance of the transformer Tr1 will be transferred to the secondary via the synchronized rectifier SR. At the time wherein the main switch M1 turns off the leakage inductance Llk starts resonating with the parasitic capacitance of the main switch Ceq, see Fig. 1A. This leads to very large voltage spikes which may exceed the voltage rating of the main switch M1, shown in Fig. 1B.

Fig. 1B: Voltage across M1 due to the leakage inductance

In conclusion there is a need to find a solution to prevent the voltage across the main switch to increase and exceed the voltage rating of the main

switch M1. Over the years there were many attempts to address this problem, such as different types of lossless snubbers and even active clamp, though these solutions were inefficient, complex and expensive. For a better understanding of the advantages of Rompower high efficiency passive clamp I will start with a conventional solution which is known in the field of power conversion as the RDC snubber. This solution it is the most popular because it does limit the voltage spikes across the main switch, and it is relatively simple and low cost. Such a RDC snubber is shown in Fig. 2, it consists of a diode D1, a capacitor C1, a discharge resistor R2 and a snubber resistor R1. The characteristics of the diode D1 plays a key role in the functionality

of this type of snubber, and more specific the

Fig. 2: The Flyback topology with RDC Snubber

reverse recovery characteristic of the diode D1.

In Figure 3 is presented the reverse recovery characteristic of a diode and the key parameters which characterizes the reverse recovery. When a current is flowing through a diode and a reverse voltage Vr is placed across the diode as shown in Fig. 3, the current starts to decay with the slope proportionate with the parasitic inductor element L. At t1 the current through the diode reaches zero and further the charge present in the junction allows further conduction even with negative current until t2 when the reverse current through the diode reaches its highest amplitude Irrm. Between t1 to t2 the charge present in the junction is depleted and the diode becomes a high impedance device at t2. The negative voltage across the diode reaches the level Vrm, amplitude which is given by the formula from upper corner of Fig. 3. The slope of the current during the Tb portion of the reverse recovery time, dI(rec)/dt ,differentiates the diodes in soft reverse recovery type diodes and snappy reverse recovery type diodes. A key characteristic of the diode for application in passive snubber is “ta”, wherein there is still charge in the junction. A larger “ta”, will allow a reverse current through the diode which will

discharge the clamp capacitor, and that charge will be transferred to the output winding, and further to the output capacitor Co.

Fig. 3: Reverse recovery characteristics of a diode

The key waveforms of the passive snubber shown in Fig. 4 are: IdM1 which is the current through the main switch M1; ID1 is the current through the clamp diode, D1; Vds(M1) which is the voltage across M1. At t0, the main switch M1 turns on and the current through the transformer primary winding, L1, builds up to a peak level Ipk, level which is reached at t1. At t1 the main switch M1 turns off and the magnetizing current in the transformer Tr starts flowing into the secondary

Fig. 4: The key waveforms of standard RDC clamp

winding L2 via the synchronized rectifier SR. The current through the leakage inductance will flow through the passive clamp, via R1 and via D1 into the clamp capacitor C1, which was slightly discharge by R2 prior t1. A resonant circuit is formed by the leakage inductance of the

transformer reflected in the primary and the capacitor C1, wherein R1, acts as a dumping resistor. Between t1 and t2 the current through D1 has a positive polarity and during this time C1 is charged. The charge which will flow into C1, it is Qa depicted by the shaded area between t1 and t2, from Fig. 4. As presented in Fig. 3 the current through a diode like D1 can flow in reverse due to the charge stored in the junction of D1.

At t3 the charge in the junction of D1 is depleted and the diode D1 becomes a high impedance device and the reverse recovery current through D1 reaches ID1(rrm) which is the peak reverse current, as depicted in Fig. 3. To comply with the preservation of charge, Qa=Qb, which means that the charge which was injected in C1 between t1 to t2 is transferred back to the secondary winding and further to Co during t2 to t3. The larger the reverse recovery time, especially Ta portion as depicted in

Fig. 3 of the diode D1, the more charge is recycled to the secondary winding through SR, and to the secondary. After t4 the resonant circuit formed by the leakage inductance of the transformer and C1 changes. This is due to the diode D1 which becomes a high impedance device when the charge in the junction of D1 is depleted. The new resonant circuit is formed now by the leakage inductance reflected in the primary of transformer Tr, and the parasitic capacitance of diode D1. The capacitor C1 it is still in the circuit, but its value is much larger than the parasitic capacitance of D1 as a result its impact in the resonant circuit is negligible. The high frequency ringing (HF ringing) caused by the resonance between the leakage inductance reflected in the primary of the transformer Tr and the parasitic capacitance of D1 in parallel with the Ceq across the primary switch will create a ringing across the primary switch M1 as depicted in Fig. 5 wherein is presented the

Fig. 5: Voltage across main switch with a standard RDC clamp

voltage across the main switch in a flyback topology with a DC input voltage of 327Vdc, a turn

ratio in the transformer of 5.5 , an output voltage of 20V, and a leakage inductance in the transformer of 2.5uH, and S3N as D1. The snubber resistor R1 can have a strong impact in the attenuation of the ringing across the main switch.

In Figure 6 is presented the voltage across the main switch using the same transformer and the same parts at the same input and output conditions with the snubber resistor of 51Ohm. Though the ringing across the main switch at the time wherein M1 turns off it is no longer present, the peak voltage across the main switch in the same conditions went from 500V in Fig. 5 to 560V in Fig. 6. In addition to that there is a decay in efficiency because a snubber using a resistor in series with the diode is dissipative. The additional voltage across the main switch is due to voltage across the snubber resistor R1, caused by the peak current through D1, creating ΔV, as depicted in Fig. 4. In conclusion the passive snubber solution does limit the voltage stress on the main switch and function of the reverse recovery characteristics of D1, some of the leakage inductance energy is recycled to the secondary by the reverse conduction through the diode due to diode’ reverse recovery characteristic.

Fig. 6: Measured Waveforms of VCM1 with a large R1

III. High Efficiency passive clamp

in Flyback topology

In Figure 7 is presented a modification of the

traditional passive snubber, wherein the snubber capacitor Cr is in series with a circuit formed by two low voltage diodes, D1 and D2 and a voltage source Vb. The sub-circuit, 100, formed by the two diodes and the voltage source Vb is referred in this paper as “energy extraction circuit” and it plays a key role in converting a RDC snubber into an high efficiency passive snubber having the characteristics of an active clamp.

Fig. 7: Schematic of Rompower snubber.

At the time when the primary switch M1 turns off, the equivalent circuit for the leakage inductance is the one depicted in Fig. 8A.

In Fig. 8B is presented the current through the leakage inductance versus time for different values of the Vb. The larger the value of Vb the faster the current through the leakage inductance Llk will decay towards zero. The area in between the current and the time axes, such as ACh, represents the charge injected into the clamp capacitor Cr. A larger value for Vb will decrease

Fig. 8A: Equivalent Circuit

Fig. 8B: Leakage inductance

the charge injected by the leakage inductance in the clamp capacitor Cr. In compliance with the law of “conservation of charge”, the charge which will be extracted from Cr during the time when D2

conducts in reverse will also decrease. For a given value of the leakage inductance in the transformer, a given amplitude of the peak current through the primary winding and a given Vb level, the charge injected and also the charge extracted from Cr can become comparable with the reverse recovery charge of D1. The “energy extraction circuit” from Fig. 7, which decreases the charge injected in Cr, does allow a full active clamp function while using a simple diode by employing the reverse recovery characteristic of the diode. For many years the power conversion engineers have looked for a solution of using a passive clamp and be able to emulate the function of an active clamp, wherein the leakage inductance energy is fully utilized, some of it being transferred to the secondary and the remaining energy used for other purposes. However, this cannot be done without the “energy extraction circuit”

The introduction of the “energy extraction circuit” does decrease significantly the charge injected and retrieved from Cr to make this concept work over the entire operation conditions of the flyback converter using commercially available diodes wherein the reverse recovery time varies over a large range. The key wave forms representing the operation of the circuit depicted in Fig. 7 are depicted in Fig. 9. The key waveforms include, a) the control signal VcM1 of the main switch M1; b) the voltage (VdsM1) across the main switch M1; c) the current IL1 through the primary winding L1 of the transformer Tr; d) the current ICr through the clamp capacitor Cr; e) the current ISR through the synchronous rectifier (SR, 28). Referring further to Fig. 9, in the period between t0 and t1, the main switch M1 is configured to conduct, and the magnetizing current build up the transformer Tr. At the time period t1, the main switch M1 turns off and the magnetizing current starts flowing towards the secondary winding L2 and through the synchronous rectifier SR. The current flowing through the leakage inductance in the primary side start flowing though the passive clamp circuit formed by Dc1 and the clamp capacitor Cr, and then through the first rectifier D1 towards the auxiliary energy storage, depicted as voltage source, Vb. By design the time interval between t2 to t3 is smaller than of the reverse recovery time of diode Dc1. The design of the high efficiency passive clamp wherein the passive clamp operates in the same way as an active clamp without the need of an additional active clamp switch, has to comply with some boundaries. The reverse recovery characteristics of the diode Dc1 has to be larger than the time interval between t2

to t3, in all the operating conditions of the flyback converter. That will require that the leakage inductance of the transformer Tr, should be under a certain value for a given power level (ex. 3-4uH), a given turns ratio of the transformer Tr and a given value for Vb. This concept does also apply to active clamp as presented in reference #2, wherein the RMS current through the active clamp decreases several times allowing the use of a smaller device for the active clamp. Injecting the current through D1 into a voltage source Vb leads to transfer of energy to the voltage source VB and produces a dumping effect, substantially reducing the ringing in the passive clamp portion of the circuitry. In this technology, a portion of the leakage inductance energy is transferred to Vb. The voltage across Vb is reflected across the voltage across the main switch, as depicted in the Vds M1 of Fig. 9 an “overshoot” portion of the curve. The reduction of the duration of the time interval (t1 to t2) leads to a considerable decrease of the RMS current through clamp circuit formed by Dc1 and Cr.

Fig. 9: Characteristic Waveforms of Rompower snubber circuit

In Figure 10 is depicted the voltage across the main switch M1, for a 65W AC-DC converter at Vin=230vac. The “overshoot” which is the reflection of the Vb voltage plus the voltage ripple on Cr, is approximately 40V. The energy from the leakage inductance which is harvested in the voltage source Vb can be used to provide the bias

power in the flyback converter circuitry and it can be also used to power the Current Injection circuit in order to provide ZVS for the main switch, concept which will be presented in the next chapter or transferred to the output if the diode from the bias winding is substituted by a small Mosfet. In conclusion the modification of the clamp circuit from the conventional circuit from Figure 2 to the circuit from Figure 7 decreases the charge injected in the clamp capacitor Cr significantly and as a consequence reduces accordingly the reverse charge through Cr, which becomes smaller than the reverse recovery time of Dc1. In addition to this the circuit from Figure 7 transfers also a portion of the energy from the leakage inductance to Vb acting in this way as a lossless snubber eliminating the ringing associated with the passive snubber. By employing the “energy extraction circuit” the passive clamp acts as an active clamp and recycles and uses the energy contained in the leakage inductance. The “energy extraction circuit” can be used as snubber in many other application as presented in reference #2.

Fig. 10: Voltage across the main switch of 65W AC to DC Converter

IV. ZVS using Current Injection

technology

In Figure 11A and Figure 11B is presented current injection technology in flyback topology as a simple and effective solution to obtain ZVS in any operating conditions. Rompower current injection technology can apply to any topology as is presented in Reference #1. In Fig. 11A is depicted a flyback power train having a transformer with a primary winding L1 and a secondary winding L2, a primary switch M1 having a parasitic capacitance Ceq which represents the total parasitic capacitance reflected across M1, including the Coss of Mosfet M1,and in addition to it the parasitic capacitances reflected across the main switch from the transformer and other parasitic capacitances reflected from the secondary and

even layout. In addition to the primary and secondary winding in the transformer is added an auxiliary winding, referred in this paper as current injection winding. In most of the applications the current injection winding has several turns, like one or two turns. A low power and low voltage Mosfet, Minj, is also added, in most of the applications 100V device. The current injection circuit is powered up by Vinj which is Vb from previous chapter, wherein

Fig. 11A: ZVS Current Injection Circuit

Fig. 11B: Key waveforms of the current injection circuit

Vb represents the energy extracted from the leakage inductance as depicted in Fig. 7. The Vinj can be generated also through other means, as presented in reference #1. In Figure 11B are depicted the key waveforms of the circuit from

Figure 11A such as: VdsM1 represents the voltage across the primary switch M1; VcMinj is the control signal for the current injection switch; VcM1 is the control signal for the main switch M1; and Iinj, the current injection, which flows through the Linj, and the primary winding of the transformer, L1 in order to discharge Ceq towards zero. At t0, the Minj switch is turned on and the current starts flowing from Vinj via Dinj and the winding Linj via the leakage inductance between the primary winding L1 and the current injection winding Linj. The current injection

will flow as depicted in Fig. 11A discharging the parasitic capacitance across the main switch M1, creating Zero Voltage Switching conditions for M1.

ω =1

Ni√𝐿𝐶𝑒𝑞 𝑁𝑖 =

𝑁(𝐿1)

𝑁(𝐿𝑖𝑛𝑗) 𝑍𝐶 = √

𝐿

𝐶𝑒𝑞 (1)

𝐼𝑖𝑛𝑗 =[(𝑉𝑖𝑛𝑗 ∗ 𝑁𝑖 − 𝑉𝑖𝑛) + 𝑉𝑖]

𝑍𝑐sin 𝜔𝑡 (2)

Equation (1) and Equation (2) are depicting the key formulas for the current injection. L represents the leakage inductance between the primary winding L1 and the current injection winding Linj. Ceq is the parasitic capacitance reflected across the primary switch. N(L1) represents the number of turns in the primary winding. N(Linj) represents the number of turns in the current injection winding Linj. Ni represents the turns ratio between the primary winding and current injection winding. In Equation (2) is presented the formula for the peak current injection measured in the current injection winding. The current injection is proportionate with [(Vinj*Ni-Vin)+Vi], wherein Vin is the input voltage, Vinj is the voltage applied to the current injection winding and Vi is the voltage across the main switch at the time wherein the current injection switch Minj is turned on. Observed from Eq. (2) is that the larger the voltage across the main switch (Vi), the larger the current injection amplitude. This is a great feature of Rompower current injection, feature referred in this paper as self-adjusting feature. Another aspect of the self-adjusting feature is that the current injection amplitude is proportionate also with the value of the Ceq. From Eq. (2), the current injection is proportionate with √𝐶𝑒𝑞 which is contained in the 𝑍𝑐 = √𝐿 𝐶𝑒𝑞⁄ . For example, if the value of Ceq varies due to tolerances, the amplitude of the current injection self-adjust accordingly to ensure ZVS. Unlike other solutions such as the reference #3, #5 and #8,

Rompower current injection solution, through the current is shaped as half sinusoidal, it is not a resonant current injection, which has to be adjusted and tailored just right. Rompower current injection solution as is described in detail in reference #1, it does not use a resonant capacitor with limited energy to deliver the Iinj in a resonant manner. Rompower current injection, via Vinj, can deliver the necessary energy to ensure ZVS not only in discontinuous mode of operation but also in continuous mode of operation, regardless of the value of Ceq.

The self-adjusting feature of Rompower current injection is visible in Figure 12A and Figure 12B. In these pictures are depicted three key waveforms; the upper trace is the voltage across the primary switch, the middle trace represents the current injection current, the lower trace is the gate voltage for the primary switch. In Figure 12A the current injection is activated at higher voltage level across the primary switch, close to the hill of the ringing during the dead time. The current injection amplitude self-adjusts to be higher in

Fig. 12A: Self-adjusting feature the of current injection

Fig. 12B: Self-adjusting feature the of current injection

order to ensure ZVS. In Figure 12B the current injection is activated on the valley at a lower voltage level across the primary switch and as depicted, the current injection self-adjusts its amplitude to ensure ZVS with minimum energy injection. The key features of Rompower current injection are detailed in reference #1.

In Figure 13A and 13B are presented two key waveforms in the flyback topology, the voltage across the primary switch at a scale of 200V/div and the voltage across the synchronized rectifier at a scale of 50V/div. Figure 13A depicts the key waveforms without the use of Rompower passive clamp and without Rompower current injection. Figure 13B depicts the same key waveforms wherein Rompower technologies are implemented. It is visible that the spikes and ringing across the primary switch are eliminated by the use of Rompower passive clamp and the spikes and ringing across the synchronized rectifier are totally eliminated due ZVS across the primary switch. This applies regardless of the operating conditions, including during the burst mode of operation at no load.

Fig. 13A: Key waveforms of classical Flyback Topology

Fig. 13B: Key waveforms of the Ideal Flyback Topology

V. Experimental Prototype In Figure 14 is presented the efficiency of a 65W adapter using Rompower technologies presented above. This is done by using a 650V CoolMOSª,C7,Power Transistor from Infineon,

IPL65R195C7 and the XDPS 2110 controller from Infineon. The efficiency is above 94% starting from 100Vac and peaks at 94.6%. Our prototypes using GaNs well exceed 95% most due to a lower driving power and a lower energy required for current injection to obtain ZVS. The main advantage of Rompower technologies is that it harvests a portion of the leakage inductance energy, otherwise dissipated, in order to discharge the parasitic capacitance reflected across the primary switch to obtain ZVS, eliminating the switching losses, and eliminating the spikes and ringing across the SR and the need for a dissipative snubber.

Fig. 14: Rompower 65W Adapter (27W/inc3)

An implementation of Rompower technologies is presented in Figure 15, wherein the AC-DC adapter has a power density above 27W/inc3 including the case. The operation frequency is approximately 100Khz and it does comply with class B, EMI requirement and 10uA leakage current. The measurements are done with USB-PD, wherein the output voltage profile is from 3.3V to 20V. As visible from Figure 15, the 65W AC-DC

Fig. 15: Rompower Adapter Efficiency

adapter has a similar size to the AC plug. This high power density is possible due to the high efficiency of the AC-DC converter which is employing Rompower technologies presented in this paper.

VI. Conclusion

An Ideal Flyback is presented wherein ZVS is obtained in all the operating conditions using energy from the leakage inductance of the transformer, via a passive clamp, eliminating in this way the spikes and ringing across the primary switch and across the SR. The key features of the flyback topology are preserved, such as simplicity, low cost and the use of a passive snubber. This Ideal Flyback can operate at constant frequency in discontinuous and continuous mode while ZVS feature is always present. Other solutions for ZVS flyback use a complex active clamp and do operate in critical mode with modulation in frequency and train of pulses, solutions which do not reach the “Ideal Flyback” efficiency performances. The Ideal Flyback operates exactly as the conventional QR flyback but in ZVS mode and without spikes and ringing across the primary and secondary switching elements, allowing the use of an optimal voltage rating for the SR and eliminating a dissipative snubber. The simplicity and low cost of the ideal flyback makes it the best solution for high efficiency and high-power density AC-DC adapter with Power Delivery. The Ideal Flyback Efficiency performance are obtained by using popular silicon Mosfets are the best in the industry, and if GaNs are used the efficiency well exceeds 95%.

References

[1] Ionel Jitaru “Self-Adjusting Current Injection Technology”, US Patent 10,574,148 B1, February 25, 2020.

[2] Ionel Jitaru “Energy Recovery from the Leakage Inductance of the Transformer”, US Patent 10,651,748 B2, May 12, 2020.

[3] Henry-Hung Chung et al. “Zero-current switching PWM Flyback Converter with a simple Auxiliary Switch” IEE Transactions on Power Electronics, Vol,14,NO2, March 1999.

[4] Ionel Jitaru & Alexandru Ivascu” Very High Power Density Flyback Converter”, PCIM 1998, page 97-103.

[5] E. Adib, H. Farzanehfard “New Zero Voltage Switching PWM flyback Converter”, 1th Power Electronic Conference , 2010 IEEE

[6] Ionel Jitaru “High Efficiency Flyback Converter” US Patent 7,764,518 B2 July, 27, 2010

[7] Ionel Jitaru “Soft Switching High Efficiency Flyback Converter” US Patent 7,450,402 B2, November 11, 2008

[8] Hong Mao “Zero Voltage Switching DC-DC Converters with Synchronous Rectifiers” US Patent 7,548,435 B2