Nuclear Physics B (Proc. Suppl.) 32 (1993) 487-505

North-Holland

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PROCEEDINGS SUPPLEMENTS

The state-of-the-art o f low noise design in particle physics

P. F. Manfredi and V. Speziali

Dipartimento di Elettronica - Universita' di Pavia, Via Abbiategrasso, 209 - 27100 Pavia, Italy Istituto Nazionale di Fisica Nucleare, Via Celoria, 16 - 20133 Milano, Italy

This paper discusses the limits reached so far in the design of monolithic front-end systems and the perspectives opened-up by the research aiming at integrating detector and front-end electronics on the same chip. It considers, as an issue of the greatest importance, the radiation hardening of the front-end electronics, without which the design would be unsuitable for the applications at the future accelerators of high luminosity.

1. I N T R O D U C T I O N

The use of sensors with a high spatial density of signal electrodes, like the microstrip detectors, under the constraints set by collider experiments has stimulated, about ten years ago, a significant breakthrough in the development of electronics for radiation detectors, that is, the design of front-end systems in monolithic form [1-4].

Since then, a considerable improvement has taken place in the monolithic preamplifier systems to be bonded to the detector, with the introduction of front-end devices with better and better noise performances and the development of effective f'dters on the monolithic chips [5-8].

The monolithic approach to front-end design is being gradually extended to a broader range of detector applications, including calorimetry. In this case the need for closely spaced preamplifier systems and monolithic low noise circuits of accordingly large density is less stringent. However, the monolithic design provides the circuit compactness required to have preamplifiers and other analog circuits on a single chip able, for instance in the case of calorimetry with cryogenic liquids, to operate at low temperatures [9-11].

In the detector applications quoted so far the capacitances of the signal sources are large enough to be affected by the strays capacitances of the bonding wires but to a negligible extent.

This means that a solution, where the front-end system is realised on a separate monolithic chip and the input of each preamplifier is wire-bonded to the relevant electrode on the detector chip, is adequate. In this case the two processes, the one employed for the detector and the one for the front-end, are not required to be compatible.

In the detector applications, notably with solid- state drift chambers or pixel assemblies, where the

capacitances of the signal electrodes may be well below 1 pF, the need arises of integrating the front- end electronics on the detector chip, as suggested by the requirement of preserving the intrinsically high signal-to-noise ratio made possible by these detectors [12-16].

The realisation of the front-end electronics on the detector chip requires either that the process employed for the front-end be compatible with the detector's, or alternatively, that detector and front- end be implemented in regions that, though belonging to the same chip, have different electrical characteristics because they are isolated from each other.

The previous considerations about the requirements set to the front-end system in different detector applications are summarised in Table 1, where the present status is outlined.

As pointed out before, the development of front- end systems for microstrip detectors was, historically, the first to adopt the monolithic approach and adequate solutions are now available [17]. Some of them were upgraded to meet the requirements arising from fixed target or from experiments at colliders with bunch crossing rates higher than LEP. A solution designed to be employed at HERA will be described at this conference [18].

In the domain of monolithic preamplifiers for applications with large detector capacitances, among which calorimetry, working examples have already been produced [19, 20].

The integration of front-end electronics on the detector chip, besides supplying at least one readout system for pixel detectors, has produced results of a high qualitative value, by demonstrating the feasibility of charge measurements with noise dispersions down to a very few electrons [21-25]. The question which comes next is how the background

0920-5632/93~06.00 © 1993 - Elsevier Science Publishers B.V. All rights reserved.

488 P.E Manfredi, V. Speziali / The state-of-the-art of low noise design bt particle physics

Table 1

Solution Considerations S tatu s

Front-end electronics on the detector chip

Mandatory for SSC drift chambers

and

pixel detectors

A readout system integrated on the pixel detector chip is available

JFETs and preamplifiers realised on the high resistivity silicon of a drift chamber chip are available

Other examples of detector-front- end integration exist

Monolithic high density front-end chips to be bonded to the detector

Mandatory for microstrip detectors employed in a collider experiment

May provide a viable approach for pixel detectors until the fully integrated solut ions are accomplished

Several solutions tailored to microstrip applications are avalailable: CMOS, JFET- CMOS, bipolar

Solutions tailored to pixel detector applications axe being tested

Monolithic preamplifier chips Useful in applications with calorimeters and other detectors of comparatevely large capacitance

Solutions based on alI-NJFET and Bi-CMOS processes are available

achieved so far in the realisation of monolithic front- end systems and in the integration of the front-end electronics on the detector chip can be extended to meet the requirements set by the forthcoming generation of hadron colliders of high luminosity. Two fundamental points have to be borne in mind in designing front-end systems for applications with the future colliders.

One is related to the considerable increase in bunch-crossing rates over the existing ones, which is expected to bring the time available to process the detector signals down to the 10 ns range.

The other one has to do with the considerable integrated doses of ionising radiation and neutron fluences to which the front-end electronics will be exposed during its lifetime. This implies that the design has to cope with the constraint that the circuits be still functional with specified characteristics after integrated doses of up to 100 Mrad of gamma rays and 1014 neutrons/cm 2. For some applications one further requirement may arise, that is, the operation at cryogenic

temperatures. The present paper, besides reviewing the more

significant results achieved so far in the area of low- noise monolithic front-end systems, aims at discussing the open problems in view of the experiments at the future hadron colliders and illuslrating the steps which are being taken now in order to make the front-end systems suitable for those experiments.

2 . N O I S E L I M I T S I N D I F F E R E N T D E T E C T O R A P P L I C A T I O N S

Some basic concepts will be revisited here, in order to discuss the limits set by noise to the dispersion in charge measurements carried out in the framework of different detector applications and to analyse the ENC degradation in a front-end system exposed to radiation.

In what follows, the input noise in the front-end system will be described with two uncorrelated

P.E Manfredt~ V. Speziali / The state-of-the-art of low noise design bt panicle physics 489

sources: a series voltage source of spectral density

de 2 af a t df = aw +~-[+~" (1)

and a parallel current source of constant (white) density:

m

di 2 - - = bw (2) df

The spectral densities are intended to be the mathematical ones, with f ranging from -oo to +oo.

The aw term in eq. (1) describes the channel thermal noise in field-effect devices or the collector shot-noise in bipolar transistors. The second term arises from the 1/f-noise in the drain or collector current. The third term in eq. (1) accounts for the noise due to carrier trapping by point-like defects in the channel. The spectral density (2) results from all the parallel noise contributions, among which the shot noise in the detector leakage current, the shot noise in the input current of the front-end device and the thermal noise in the bias resistors of detector and front-end device. The parallel noise term related to dielectric losses has been neglected.

In this paper, where the focus is mainly on the applications at the short processing times that will be employed in the future hadron colliders, the interest in the low-frequency noise behaviour is explained by the following remark. Evidence from irradiation tests carried out so far, mainly with gamma rays, has shown that the low frequency noise terms are largely affected by the ionising radiation.

So, if it is true that at the beginning of the front-end lifetime these components, along with the parallel noise, are overwhelmed by aw in their impact on ENC, it is also true that the radiation- induced increase in the 1/f and 1/f 2 terms ofeq. (1) may explain the ENC degradation observed respectively in irradiated MOS and JFET preamplifiers [26-30].

If the front-end preamplifier described by the input noise densities (1) and (2) is associated to a filter whose weighting function has peaking time tp and shape coefficients A 1, A 2, A 3 respectively for the t °, fq and f-2 series densities, the resulting ENC 2 is:

ENC 2 = aw(C~ +Ci) 2 . t ~ + 2/gaf(CD +Ci) 2. A2+

[bw + 4/l:2at (CD +Ci)21"A3tp (3)

where CD results from the sum of detector capacitance CD, strays and integrating capacitance Cf if the preamplifier is of the charge-sensitive type. Ci is the input capacitance of the front-end device.

It is useful to rewrite eq. (3) by introducing the intrinsic noise parameters, aw = aw • C i, Hf = arC i,

= arC i and the mismatch coefficient m = CD/C i.

I 1 1 12 ENC 2=a~" m ~+m -~ - c ~ . A I +

tp

I 1 1) 2 +2trill. mT+m -~ .C~.A2+

+ bw +4~2a~ • m 7 +m -7 -C~ .A3I p (4)

which points out the importance of the capacitive- matched operation (m = 1) to minimise the ENC: contributions brought about by the series noise sources.

• . i ¢ ,

The quantmes a~, H e, a t provide for the three series noise contributions a description which depends on the process parameters and the gate length of the front-end device, but is independent of its gate width W.

ENC: can be rewritten in a more significant way by making appear explicitely in its expression

2 ENC,,,__ 0, that is, the value of ENC 2 achievable,

in absence of 1/f noise, with the matched filter of infinite duration and tp,op t, the value of tp which brings the ENC: given by (4) to its minimum value.

According to the matched filter theory, ENC: is given by:

2 ENC.,a,=0 = 1 II =2. a~. bw+4tr2at • m2+m -~ 'CD •

I ! 1 • (5)

490 PE Manfredi, V. Speziali / The state-of-the-art of low noise design M particle physics

while tp,op t is obtained by differentiating (4) as: 1

( ± _1,~2 1F a : ' [ m 2 + m 2J "CD [ . (A l l½

tp'°pt = ( 1 __1"~ 2 [ bw+47r2a : . (m2+m 2 j .C~ l t 'A3)

(6)

If eqs.(5) and (6) are taken into account, from (4) ENC 2 can be rewritten as:

1 ENC 2 = ENC~,a,=0. (AI-A3)2-"

(ill 1 tP'°Pt + + 2

"2" tp tp,opt P ' ENC~,at=O (7)

which identifies the ENC 2 degradation over ENC 2 as being due to the use of a suboptimal filter, degradation quantitatively described by the factor (A1A3) 1/2, to a time mismatch off tp,op t and to the presence of 1/f-noise. In eq. (7) the parameter p =

" ~ I A 3 / A 2 has been introduced. Its values, for the currently employed filters, range between 1 and 1.36 ['.31].

The minimum ENC values attainable in different detector applications and the degradations brought about by the use of a suboptimal filter at tp = tp,op t, by the addition of 1/f-noise and by time mismatch tp ~ tp,op t are discussed in the Table 2.

The three columns in Table 2 refer to the

following three situations: C~ in the .1 to 1 pF range as in the case of solid-state drift chambers and pixel detectors, C D in the 10 to 30 pF range where

microstrip applications are located and C~ from 100 pF to 1000 pF to describe the case of dE/dx measurements and the calorimetry applications.

One example is fully analysed in each column.

A CD values of lpF has been assumed as descriptive of the pixel detector case, which may refer to a monolithic preamplifier system realised on an independent chip and wire-bonded to the detector.

The value CD = 10 pF is considered

representative of the microstrip applications, although larger values are expected to occur in the long microstrip detectors that are being planned for use in some vertex systems. In the third column the value CD = 100 pF has to be considered the lower limit in calorimetry and other energy measurement applications.

The data represented in Table 2 have been obtained by introducing into eq. (1) the A 1, A2, A 3 coefficients relevant to a symmetric, piecewise quadratic weighting function of peaking time tp, that is:

A 1 = 2.67 A 2 = 1.15 A3= 0.77.

In all examined cases, the ENC evaluations refer to the capacitively matched operation, m = 1.

As to the noise description of the front-end devices, the a~ parameter has been expressed as:

F a* = 2 k T - - (8)

CO T

where, k is Boltzmann's constant, T is the absolute temperature, ro T is the transition angular frequency of the input device and F a coefficient which is equal to .5 in a bipolar transistor, to 2/3 in Si JFETs and MOS and which exceeds 1 in short channel Silicon and GaAs devices.

An roT value of 109 rad/s has been introduced in the calculations for all the devices considered. It may be argued that such a value underestimates the possibilities offered by the more advanced bipolar transistors or by the GaAs MESFETs. However, it is also true that it overrates the actual behaviour of ordinary JFETs and MOS operating at low current values in high density monolithic circuits, so it can be accepted as a reasonable reference value.

As to the description of the front-end devices from the standpoint of low-frequency series noise, the following options have been considered.

For the situation Co = 1 pF the front-end device has been assumed to be a P-channel MOS with an intrinsic 1/f-noise parameter H e of 4 • 10 -25 J, value which falls in the range spanned by the currently available CMOS processes.

For the cases Co = 10 pF, and CD = 100 pF the front-end device has been assumed to be an epitaxial channel JFET, part of the monolithic buried-layer process, with intrinsic parameters of 1/f and 1/f 2 noises respectively H e = 5" 10 .28 J and ~* = 3 -

PE Manfred~ V Speziali / The state-of-the-art of low noise design in particle physics 491

Table 2 Pixel Microstrip

C D range .1 ÷ 1 pF 10- 30 pF

assumed CI~ value 1 pF l0 pF

parallel noise 10"28 -~- 10"28 A--~-2Hz

Grad Grad 0~ r 1 s 1 s

tp,op t 0.9 Its 3 Its

ENC~,at--0 61 rms e 109 rmse

ENC2,a, 0A2R 73 rms e 130 rms e

fl( ENC2'af=0 + a2) A2p 82 rms e 130 rms e

tp 20 ns 20 ns

~/( ENC2'a,=00 + a2) A2p 350 rms e 1100 rmse

tp 100 ns lOOns

~( ENC2'a'=°O + a2) A2p 160 rmse 492 rmse

Calorimetry

100÷ 1000 pF

100 pF

10.28 A 2 Hz

1 Grad S

9 Its

194 rms e

232 rms e

232 rm se

20 ns

3477 rms e

100 ns

1551 rms e

2 ~ tp tp,op t ) a 2 =

I I 12 27rHfCD. m ~+ m -~

2 p ' ENC~,af =0

10 -28 W. For reference purposes, the spectral density of parallel noise has been assumed in all cases equal to 1.6 10 -28 A2/Hz, corresponding to a total leakage current of 1 nA. It may be worth commenting some of the numerical results presented in Table 2.

In the case process, CD = lpF, the use of the matched filter of infinite duration, would yield, in absence of 1/f-noise, an equivalent noise charge of about 60 rmse. Owing to the previously discussed degradation terms, ENC would rise to 160 rms e at tp = 100 ns and 350 rms e, at tp -- 20 ns. The latter value would correspond to a signal-to-noise ratio of 70 to 1 in the case of a minimum ionising particle crossing a 300 ~tm thick Si pixel detector.

At C D = 10 pF, the 109 rms e achievable in absence of 1/f-noise with a matched filter of

infinite duration would go up to 490 rms e at tp 100 ns and 1100 rms e at I1, 20 ns when the effect of all degrading terms is taken into account. The last value corresponds to a 22 to 1 signal-to-noise ratio for a minimum ionising particle crossing a 100 lam thick Si detector.

$ In the case CD = 100 pF, the ENC of 194 rms

e obtainable in absence of 1/f-noise with the matched filter of infinite duration increases, once all the degradation terms are introduced to 1550 rms e at tp = 100 ns and 3477 rms e at tp = 20 ns. The last value corresponds to a 6.7 to 1 signal-to- noise ratio referred to a minimum ionising particle across a 300 gm thick Si detector.

These values are given with the aim of providing an oversimplified picture of what the situation may look like at such short values of tp. It can be done better, but it can also be done worse

492 P.E Manfredi, V. Speziali / The state-of-the-art of low noise design b~ particle physics

than this. At the short tp values that are going to be

employed in experiments at high luminosity colliders, it has to be expected that, according to eq. (4), the dominant ENC 2 contribution is the one related to the aw spectral density, that is:

ENC22,Tr I 1 1/2 ('OT C D " m~ + m- i . A_~p~ (9)

which clearly emphasises the importance of employing front-end devices of large transition frequency. According to (9), doubling tL~ r would result in a 40% reduction in ENC.

The finite gain-bandwidth product of the front- end device is just one of the aspects that will affect the dispersion in charge measurements at the high luminosity colliders. The other one is related to the radiation-induced increase in the low frequency noise. The low frequency noise terms affect the ENC values at short tp' s to a negligible extent in the nonirradiated device as apparent from columns 2 and 3 of Table 2. As the dose of absorbed radiation increases, they may invade regions of the frequency spectrum that are of interest in the actual charge measurement, thereby degrading ENC [28].

Such a consideration can be understood with reference to figs. 1 and 2. A model of the ENC degradation at short processing times which may be brought about by the experimentally proven

SPECTRAL DENS1T~

INCREASING

NOISE

, , , , , ,

10 3 10 4 10 5 10 5 10 7 10 S

LHC-SSC

Figure 1. Model of ENC degradation at short I1, values related to the radiation-induced increase in 1/f-noise.

radiation-induced increase in 1/f-noise of MOS devices is sketched in fig. 1. According to fig. 1, the 1/f-noise has a little impact on ENC at the processing times to be employed in the high luminosity colliders (frequency region of interest 107 + 108 Hz ) in the non irradiated device. Ionising radiation, in particular "prays, has a negligible effect on the white noise term of the spectral density, while it considerably affects the 1/f-noise.

The latter may become important at comparatively high frequencies, thereby affecting ENC at the Iv values that are of interest for applications with high luminosity colliders.

The noise degradation in an irradiated JFET is described in fig. 2, where the spectra of the series noise voltage before irradiation and after exposure to increasing doses of 6o Co ~/-rays are plotted.

The process of noise increase at higher frequencies and therefore of ENC degradation at short peaking times is again related to the low- frequency noise mechanism.

-Z :=

>

"O

102

101.

l O o .

I I I t l l l l l I I I I I I I I I I ~ I l l ~ l t l I l l ~ { l l ~ l I 1 1 1 1 1

40 Mrad

dose 0

10 -1 ~ I t I [ i t l l i i I I i i i l l i i i I i t l l ~ i I I I I I I I I i i I l l l l

101 102 103 104 105 106

f [Hz]

Figure 2. Spectra of series noise voltage in a JFET before irradiation and after irradiation steps at different dose levels of 6o Co T-rays.

According to the spectra af fig. 2, the absorbed radiation seems to be responsible for the appearence of Lorentzian noise terms which may invade the frequency range of interest for the actual applications.

In choosing a device for low noise front-end design, due attention has to be devoted to the radiation-induced effects with the aim of keeping the ENC degradation aring from the enhanced low frequency noise terms under strict control.

P.E Manfred~ V. Speziali / The state-of-the-art of low noise design b~ panicle physics 493

3. FRONT-END DEVICES ON THE DETECTOR CHIP

developed with the aim of realising front-end devices on a silicon drift chamber [21,22].

Since the early attempts to use CCD's with a built-in active elements in vertex detectors for elementary particle physics it was clear that the way to minimum ENC had to go through the integration of the front-end device on the detector chip. Although the MOS source followers that are present in CCD's are usually not purposely designed for low-noise applications, the extremely small input capacitance resulting from this integration explains the low ENC values achievable in this way [32].

The need to integrate in the simplest instance an active device and in a more advanced perspective an entire preamplifier on the detector chip arose when detectors featuring extremely small eleclrode capacitance, like the solid state drift chambers, the totally depleted CCD's and the pixel detectors made thier appearance [12, 13, 15].

The compatibility of a solid-state detector which is manufactured on a high resistivity-bulk and the active devices which ordinarily require a low resistivity material is a difficult problem. Some solutions that have been provided in very recent times were proven of very great qualitative importance, as they demonslrated the feasibility of charge measurements with noise dispersions down to a very few electrons, with detector capacitances in the 100 fF region [33].

Perspectively, amplifiers with very low noise can be integrated with a detector to obtain an equivalent noise charge well below 1 rms e [33].

An example of integrated configuration featuring ENC values of about 14 e-rms at room temperature with a semigaussian shaper of 3 ~ts tp and 5 electrons at cryogenic operation with larger peaking times is shown in fig. 3 [24].

The schematic of the preamplifier circuit is shown in fig. 4, where the dotted line encloses the two JFETs (J1, J2) that are realised on the detector chip.As shown in fig. 4, the part of the front-end preamplifier integrated on the detector chip consists of a source followed 01) and a charge reset switch (J2).

The off-the chip JFET J3 which provides an active load for the source of J1 and the following charge-sensitive preamplifier complete the front-end system.

The JFETs J1, J2 integrated on the detector chip are based on a process which was originally

[ Firsffet [ I Reseffet I

Figure 3. Front-end devices integrated with the output section of a totally depleted CCD.

- u ° + t J 0 Reset , . . . . . . . .

CTes~ ~ J 3 R Test in _~_

L..f o -- _U k

°-4 ks

-U S

ut

Figure 4. Schematic of the preamplifier circuit.

Alternative solutions are being tested with the aim of providing a feasible approach to the iintegration of a front-end system on a pixel detector. The DEPMOS idea looks to be very promising in this respect [16].

494 P.E Manfredi, V. Speziali / Tire state-of-the-art o flow noise design in particle physics

An integrated structure of pixels and associated readout PMOS has been developed and succesfully tested on the beam [25, 34].

4 . L O W - N O I S E D E S I G N E M P L O Y I N G M O S AS F R O N T - E N D D E V I C E S

The monolithic preamplifiers systems realised so far for applications at intermediate speed in experiments at LEP, CDF and CERN colliders are based upon CMOS processes. The low noise design in CMOS technology seems to be settled now on using a P-channel rather than an N-channel MOS as a front-end element. Such a choice is dictated by the fact that the P-channel MOS has a considerably smaller 1/f-noise than the N-channel MOS. The increased channel thermal noise arising from the lower carrier mobility may be an acceptable limitation as long as weighting functions with peaking times around lIts or longer are employed [10, 17, 35].

The analysis of the published noise characteristics of preamplifiers employing a P- MOS at the input has led to the values of the Hf parameters given in Table 3,which shows that they

TaNe 3

Preamplifier He [J]

1 4.5 10 -25

2 4.2 10 .25

3 2.2 10 -24

4 1.7 10 .25

5 3.4 10 .25

are scattered over not more than an order of magnitude. Just to give a numerical example, a PMOS featuring Hf = 10 .24 J, of a size suitable to match a 10 pF detector capacitance, would yield an 1/f-noise-related ENC contribution of about 70 rms

e and a dENC/d C~ sensitivity due to 1/f-noise of about 3.5 e/pf.

The available preamplifier systems employing a PMOS as a front-end device show a substantial alignement in their noise performances, as apparent

from the results quoted below. For instance, the preamplifier employed in the

AMPLEX chip has a front-end PMOS with W = 5000 Itm and L = 3 Bm. With a semigaussian weighting function of 750 ns peaking time, it

features an ENC at C D = 0 of 300 rms e and

dENC/dC~ sensitivity of 30 e/pF [10]. Another preamplifier of very recent design,

based on an input PMOS with W = 4300 Itm and L = 1.5 Itm, tested with a semigaussian shaper of 1.5 Its peaking time was found to comply with the relationship:

ENC = 160 + 12. C~ (10)

where ENC is expressed in electrons and C~ in pF [35].

A third example, the R.A.L.-Orsay preamplifier developed for the upgraded version of Delphi microstrip readout is described by the circuit diagram of fig. 5 [17].

I j+

vu,

Figure 5. CMOS preamplifier using a PMOS as a front-end element and a switch actuated reset of the integrating capacitor.

In the more recent version, where a substantial reduction in the gate spreading resistance has been achieved as a result of a redesigned gate geometry, the input PMOS has W= 3500 Bm, L = 3 Bm. The value of ENC, measured with a semigaussian weighting function of 1.5 Its and a 28 pF externally added capacitance is about 900 rms e, inclusive of the contribution brought about by a 270 ~ series protection resistor.

P.E ManfredJ, V. Speziali / The state-of-the-an of low pwise design hi particle physics 495

Interesting perspectives for applications at short tp values are being opened-up by short channel CMOS processes (L between .8 and 1,2 I.tm)

Some of these processes employ a thinner gate oxide (20 nm against 40 nm) than the ordinary ones, which goes in the right direction toward radiation hardening.

An appreciable evaluation work has been carried out on some of these processes, oriented to determine the values of the multiplying coefficient "~ in the spectral density of channel thermal noise:

m

de 2 2, 2/3 (11) =2kT. -~-

df

and the 1/f-noise parameters [29, 30, 36]. Values of "{ between 0.93 and 1.15 for the

PMOS and around 1.2 for the NMOS have been measured on the 1.2 ~tm processes. As in CMOS of longer gate, the 1/f-noise is smaller in the P- channel device, where values of Hf of about 10 .24 J have been found.

In one of these CMOS processes, which is claimed to be radiation hard, absorbtion of a 5 Mrad dose "y-rays was found to increase the spectral voltage density of series noise by about a factor 2, over the whole frequency spectrum in the NMOS.

The same dose on the PMOS is responsible for an increase in the spectral voltage density of a factor 1.7 at 10 kHz and 1.2 at 6 MHz, which is consistent with the usually experienced result of a higher radiation-sensitivity of the noise in the lower frequency region. The comparatively small variation of noise at 6 MHz, a frequency which approaches the region of interest for applications at high luminosity colliders seems to point out that this process looks to be well promising for the realisation of front-end systems exposed to high irradiation doses.

5. L O W N O I S E D E S I G N E M P L O Y I N G JFETS AS FRONT-END ELEMENTS.

The interest toward JFETs as front-end elements has to be put in relation with their excellent noise characteristics, and with their being intrinsically radiation hard devices and suitable for operation at cryogenic temperatures.

Two monolithic JFET processes are already available and sufficiently well mastered, while

introduction of JFETs into other processes is presently under way [37-39].

Of the two established technologies one is a JFET-CMOS process where N-channel JFETs are employed as front-end devices and more generally used to implement all functions for which low noise requirements have to be met. The MOS are available for all other functions.

The second process is based on epitaxial- channel N-JFETs and is the extension to monolithic structures of the technology employed in the realisation of discrete JFETs.

The way in which JFETs are realised in the two processes is described in fig. 6.

S G D S G D

uI N-WEI.I. j k,,,. N-WELL . , /

P-substrate

a) P+ topside gate finger P+ topside gate finger

/ l e l ° ' " l I ~÷ U,,ri~a'layf ~÷ b~i~t layf

N-subs~ate

b) Figure 6. JFET realisation in two different processes. a) CMOS-compatible JFETs b) Epitaxial channel, buried-layer process.

In the CMOS-compatible process, fig. 6 a), the implanted N-well provides the channel region for the NJFET. The topside gate is a P+ implant, while the substrate, which is shared by all the devices in the circuits, acts as a backside gate to the NJFET. Obviously topside and backside gates are isolated from each other, so that the CMOS- compatible JFET is actually a tetrode-like structure.

496 P.E Manfredi, V Speziali / The state-of-the-art of low noise &sign b~ particle physics

In the epitaxial channel, buffed-layer process the JFETs are Mode-like structures realised in isolation tubs. The tubs are defined by the P+ buried layers diffused into the N substrate and by the vertical P+ isolations.

The JFETs in the CMOS-compatible process can be manufactured with a minimum gate length of 1.6 lam. The gate length in the buried-layer JFETs is presently 5 l.tm, but upgrading of the process to bring it down to 2.5 ~tm is under way.

The JFETs in the two processes have rather similar noise characteristcs in the low frequency region, as pointed out in Table 4, where the values of Hf and a t are given.

Table 4

Hf[J] a*[W] JFET-CMOS compatible process

2.4. 10 -27 1.5. 10 -27

Epitaxial channel JFET, buried layer process

5" 10 -28 3" 10 .28

Differences, instead, are observed in the behaviour of the white components of the series noise spectral density. In the CMOS-compatible JFETs the white spectral density is larger than expected on the base of the theoretical gm- dependence because of contributions brought about by the gate spreading resistance and substrate series resistance.

In the epitaxial-channel JFETs obtained with the buried-layer process, the white term in the series density closely tracks the values that are theoretically expected for the thermal-noise in the channel.

Thermal noise contributions from spreading resistances are negligible, as a result of the triode- like structure featuring a very high conductivity in all P+ regions that build-up the gate.

From the standpoint of applications, the CMOS-compatible JFET process is employed to realise high-density front-end systems for multielectrode detectors with capacitances in the 10 pF-regions. The availability of CMOS offers to the design a unique versatility.

The epitaxial channel buried layer-process is employed whenever the ultimate noise performances are required at larger detector

capacitances, preferably in the 100 pF region. The basic structure of a charge-sensitive

preamplifier obtained with the CMOS-compatible process is shown in fig. 7.

v A o

%J214 VB O I q J3 J5

VDD

O OUT

Vss

Figure 7. Charge-sensitive preamplifier based on the CMOS-compatible process.

The circuit in fig. 7 consists of a complementary cascode made of an N-JFET and a PMOS. In the version employing an input JFET with 400 ktm width and 1.6 ~tm gate length, it behaves noisewise as described by the plots in fig. 8. The data displayed in fig. 8 were obtained with the preamplifier followed by a spectroscopy

1200

1000 ...................... ~ ...~..: i ......................... i . . . . . . . . . . . . .

800 _%-k il ~ 600

400 =0 ~

200

0 + l I I I I I I l ,~, , , , i , I I I L l [

0.1 t [gts] 10 100 p

Figure 8. Equivalent noise charge as a function of the peaking time tp with the detector capacitance as a parameter.

amplifier of current production with a semigaussian weighting function.

P.E Manfredi, V. Speziali / The state-of-the-art of low noise design in particle physics 497

The ENC degradation in the same preamplifier exposed to increasing doses of 6°Co ),-rays is described in fig. 9.

Cn* = 20 pF 1400

. . . . . . . . i . . . . . . . . . . . . . .

10001200 iiiii ii i i i i i i ~ i9~ii~;~..

,~ 800

200 ......................................................... d?se0

0 I t i i i i l l i i I t i i ] 1 1 1 i . . . . . . .

0.1 1 10 100 tp [~]

Figure 9. ENC as a function of the peaking time of the semigaussian weighting function for the JFET- MOS preamplifier exposed to ~-rays. The integrated dose appears as a parameter.

In the epitaxial channel, buffed-layer process there is no P-channel device to ensure the complementarity, so the design has to rely only on

+ 1 2 V

J4

J3

J7

V ~ - - 6 , - - I I--------

J9

C~ RF ~ R1

Out

- 6 V

Figure 10. Charge sensitive preamplifier obtained with the buried-layer process.

the N-channel devices. The configuration of a monolithic preamplifier

based on the buffed-layer process is given in fig. 10.

The preamplifier of fig. 10 consists of a cascode 01, J2) whose output signal current is injected across the high dynamic impedance presented by the active load 03, J4, J7). Besides buffeting the signals from the high impedance point, J7 has also a bootstrapping action on the gate of J4. The output signals are transmitted to the load by J9.

In the circuit of fig. 10 the feedback resistor Re and capacitor Cf as well as R1, R2 are off-the-chip components.

In the present version the preamplifier of fig. 10 employs an input JFET with W = 11000 ~tm, working at a standing current of about 10 mA and featuring a transconductance of 48.5 mA/V.

The standing current in J1 actually exceeds the design value by more than a factor 2 as a result of a poor control of the pinch-off voltage in the process, not yet thoroughly tuned to the realisations.

Analysis of the noise behaviour of the circuit has shown that the noise density referred to the input is close to the theoretical value of the channel thermal noise in J1.

The ENC dependence on the externally added CD capacitance is plotted in fig. 11 at different values of the peaking time tp.

4oo . . . . . . . . i . . . . . . . . . i . . . . . . . .

- 2 0 0 0 - . . . . . . . i . . . . . . . . i . . . . . . . . . . . i . . . . . . . : . . . . . . . . . . . i ....... -:i s

1000 . . . . .=-. ,_

0 , , , t . . . . i . . . . t . . . . t . . . . t . . . . I , , ,

0 100 200 300 400 500 600 700 C* D = [pFI

Figure 11. ENC as a function of the externally ~ Co capacitance with the peaking time tp as a parameter.

To obtain the curves of fig. 11, the preamplifier was associated to the same amplifier with gaussian weighting function used for the measurements to

498 P.E Manfre~ V. Speziali / The state-of-the-art o flow iwise design in particle physics

which fig. 8 refers.The radiation hardness of the preamplifier was tested by exposing it to increasing doses of 6°Co "/-rays and monitoring ENC after each irradiation step.The radiation-induced degradation in ENC is apparent in fig. 12. For values of integrated dose up to 10 Mrad there is limited increase in ENC at the lower end of the explored tp range. The degradation becomes more pronounced at 40 Mrad.

The ENC vs tp curve relevant to the highest dose value shows the behaviour which is expected in presence of Lorentzian terms in the spectral density.

"6"

i

7000

6000

5000

4000

3000

2000

1000

0 0.1

CD* = 500 pF

:i ..................... ........ : : ........... T . . . . . . . . . . . . . . . . . . . - . . . . . . . . . . . . . . . ~ - . . . . . . . ~ . . . . . . . ~ . . . . . ~ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . . . . . . . . . i . . . . . . . .

1 10 100 tp [~t~l

Figure 12. ENC vs tp dependence in the preamplifier exposed to increasing doses of 6°Co ~- rays. The integrated dose appears as a parameter.

The ENC behaviour in the irradiated preamplifier was not explored in the region of tp values below 100 ns, which is the one of higher interest for the applications at the future hadron colliders. From the trend shown in fig. 12 and from the nature of the noise which is enhanced by the radiation, it is expected that the degradation in ENC should be less noticeable at shorter tp. This may mean that the preamplifier is sufficiently radiation tolerant at least after exposure to 6°Co "/-rays.

More radiation tests on the buried-layer preamplifier are in progress. They aim at determining its behaviour under neutron irradiation at room temperature and neutron and ~, irradiation at cryogenic temperatures.

P-channel JFETs are available in a new process where several device types are made compatible by the use of isolating layers [39].

Results of preliminary noise tests show that a

small device (W = 48 Ixm, L = 2 pan) features about

30 n V H - ~ at I00 Hz and a channel thermal noise

of 3.2 nV/~f-~ at 200 IxA-Io.

. L O W N O I S E D E S I G N U S I N G BIPOLAR TRANSISTORS AS F R O N T - E N D E L E M E N T S

At the values of tp to be used with the high luminosity colliders the parallel noise in the bipolar transistors is of limited importance. Conversely, the high transition frequencies of several bipolar monolithic processes make the bipolar transistor a suitable choice for these applications.

Because of these considerations some favour is being conveyed on the bipolar transistor as a front- end element and examples of monolithic circuits based on it are already available.

A peculiar feature of the bipolar transistors in detector charge measurements is that at a given (C~), tp) combination, a condition of minimum ENC can be attained by acting on the collector standing current [40].

The minimum ENC is determined by the base spreading resistance rBB', the transition frequency fT and the current gain hFE

Several bipolar or BiCMOS processes feature fT values well above 1 GHz at reasonable current levels and hFE around 100 or more. The base spreading resistance, by suitable design techniques, can be brought down to values in the 10 ~ region.

Some interesting preamplifiers have been developed in bipolar technology, matching different capacitance ranges.

Several preamplifier configurations of either charge-sensitive or current-sensitive type have been proposed to cover the region of detector capacitance values around 10 pF, which includes the microstrip case, and monolithic circuits have been implemented. The preamplifiers appeared in the literature work at power levels scattered on a broad range, from very few mW up to a few tens of mW [41-43].

The feasibility of preamplifiers featuring values of ENC around 1000 electrons at 10 pF externally added capacitance and dENC/d C~ sensitivities of 30 e/pF with semigaussian weighting functions of 10 ns peaking time has been proven.

P.E Manfredi, V. Speziali / The state-of-the-an of low noise design in panicle physics 499

A microstrip readout system made of channels that consist of a preamplifier and a hit detecting comparator is available and is going to be employed at Hera. More detail about this system will be presented at this conference [44].

A solution tailored to calorimetry applications that is, detector capacitances of hundreds of pF has been implemented on the ground of a BiCMOS process [20]. The preamplifier, a charge-sensitive structure is based on a complementary cascode made of a bipolar transistor in the common emitter configuration and a P-MOS as a common gate element, as shown in fig. 13

<

1, %

Figure 13. BiCMOS charge-sensitive preamplifier.

The front-end transistor has an rBB' of about 14 f~. An ENC value of 4700 rms electrons at 150 pF of external capacitance with RC-CR shaping of 20 ns time constant characterise the noise behaviour of the circuit.

Circuit improvement, consisting in the replacement of the P-MOS on the signal path by a dielectrically isolated lateral PNP is under way.

The previous discussion points out that bipolar processes may provide suitable solutions to low noise amplification at values of I1, in the region of interest for experiments at high luminosity colliders.

Monolithic cxircuits based upon bipolar transistor design with transition frequencies well above 10 GHz are expected to become available s o o n .

However, the question about the radiation hardness of existing bipolar processes hasn't been thoroughly answered yet.

Some data have been provided about hFE variations in bipolar transistors exposed to proton

irradation. An h~ reduction by about a factor 2 has been observed in an NPN part of a monolithic process after exposure to 1014 P/cm2[30].

Such a reduction in h~ may be still acceptable in applications at short tp's where the parallel noise is not of dominant importance.

However, in the PNP parts of the same process after the same exposure, h~ has been found to drop to values that would make their use critical.

7 . L O W N O I S E D E S I G N B A S E D ON G a A s M E S F E T s

The GaAs MESFETs are to be taken into serious consideration as front-end elements for applications at short processing times by virtue of the very high values of transition frequency they feature.

GaAs MESFETs have a large amount of 1/f- noise associated with the drain current. Values of Hfup to 10 -22 J have been measured at room temperature. As an interesting feature, the 1/f- noise in GaAs devices decreases steadily as the temperature is reduced. In some devices Hf can become as small as 10 -25 J or even less at 4*K [45].

In spite of their large 1/f-noise, GaAs MESFETs may provide adequately large signal-to- noise ratios in the preamplification of signals from microstrip detectors at peaking times of 10 ns or less, as it was proven by a discrete preamplifier using a GaAs MESFET as a front-end element [46].

The use of GaAs MESFET at cryogenic temperatures is far more interesting. Preamplifiers entirely based on GaAs devices were developed for operation at the temperature of the cryogenic liquid in liquid Argon calorimetry.

A hybrid version followed by a semigaussian shaper with 18 ns peaking time features 1200 rms e at 100 pF of externally added capacitance and a

dENC/d C~ sensitivity of about 23 e/pF [48]. To the writers' knowledge, the development of

monolithic charge-sensitive preamplifiers is presently under way [48, 49]. As an intermediate solution, a charge-sensitive preamplifier was developed on the ground of monolithic arrays of GaAs MESFETs mounted on a hybrid substrate.

Of the truly monolithic GaAs preamplifiers, one is intended for detector capacitances around 10

500 PE Manfrech; V Speziali / The state-of-the-art of low noise design in particle physics

pF. It is based upon a classical design approach with only N-channel devices, and consists of an input cascode, an active load and an output buffer. It employes as a front-end device a MESFET with 1200 Itm gate width and 1.2 Itm length [49].

Another project oriented to realising a monolithic GaAs charge-sensitive preamplifiers assumes, as a preliminary step, the optimisation from the noise standpoint of the channel length in the front-end device [48].

Some information has become recently available about the noise behaviour in irradiated GaAs MESFET parts [48].

ENC tests at processing times in the 20 ns region have shown low sensitivity to neutron irradiation, about 5% ENC increase at a neutron fluence of 1014 cm 2.

Conversely, a sensitivity to ~, has been noticed, because of which a sizable increase in ENC has been recorded at a comparatively small dose, 1 Mrad of 6°C0 T-rays.

8 . F I L T E R I N G IN M O N O L I T H I C F R O N T - E N D S Y S T E M S

The need to perform the filtering function on the monolithic chip has brought to different solutions, based on either time-invariant or sampled-data concepts.

The first monolithic preamplifier systems were conceived for microstrip detectors in experiments at LEP. If full advantage has to be taken of the comparatively long time spacing between bunch crossing of this accelerator in order to reduce the ENC contribution due to series white noise, values of Iv in the Its region ought to be employed. The time-invariant approach to such long weighting functions is not readily feasible on a monolithic chip.

The sampled data solution, made possible in the switched-capacitor form by the availability of MOS devices on most of realised front-end systems, alleviates the problem of the time- constants.

It doesn't remove it, though, for the switched- capacitor filter must be preceded by a time-invariant low-pass one. The latter has the function of smoothing down the transitions in the weighting functions, that otherwise would bring about large ENC contributions from series white noise.

An example of weighting function obtained

from the association of a switched capacitor filter and a time-invariant RC integrator is provided by the correlated double sampling operation implemented in the most recent version of Delphi microstrip front-end, fig. 14 [17].

f

• ' s T

C H A R G g S N N RESISTOR CONTROL LOOP

a)

,~ t 2- t 1 ~

b) Figure 14. a) Filter implemented in Delphi micros~p front-end. b) Resulting weighting functions.

To an ordinary correlated double sampling filter, made of switches S 1, S 2, capacitors C 1, C 2, and difference amplifier A, a series bandlimiting resistor has been added, whose value R is controlled by an external voltage. By acting on this control voltage, the RC time constant can be varied. Depending on the ratio (t 2 - tl)/RC where t 1 is the sampling instant preceding the detector signal and t 2 the one which follows, different shapes of the weighting function can be obtained, as shown in fig. 14 b).

A second example, the switched capacitor filter employed in both CMOS and JFET-CMOS Aleph front-end, is less demanding in terms of time- invariant prefilter. It builts-up the following linear combinations of two sets of equal numbers of samples of the preamplifier output baseline v 0 (t), one set being taken prior to the detector signal and one after it:

PE Manfredi, I4. Speziali / The state-of-the-art of low noise design h: particle physics 501

~[A, "v0(t'i)-B," v0(t~)] i=l- - -

f

In the relationship above t i and t i are the sampling instants belonging to the two sets. Ai,B i, the weighting coefficients, must obey the

r l

condition ~ ( A i - B i ) = 0 to ensure that the

weighting function is insensitive to the preamplifier output offset. A broad variety of weighting functions can be obtained by changing the { A i }, { B i } sets or the time-sequences of

sampling instants { t l } , {ti' }.

In the actual implementation the time-invariant prefiltering relies upon the limitations in gain- bandwidth product of the charge-sensitive preamplifier which is purposely reduced below its intrinsic value. For instance, fig. 15 a) shows the weighting functions that are obtained, with n = 4, employing constant coefficient A i = B i = k and equally spaced sampling instants within the two sets. The different weighting functions correspond to different values of the ratio between the sampling period and the time-constant limiting the gain-bandwith product in the preamplifier.

An example of cusp realisation with n = 4 is shown in fig. 15 b). The cusp is obtained with a suitable scaling of the Ai, Bi coefficients and some minor modifications in the sampling sequences.

The described discrete-time weighting function find their best application in the operating conditions of colliding-beam machines, where the detector signals are bound to occur at a predictable time location. In the switched-capacitor realisations they may require a comparatively small silicon area, so they turn out to be suitable in the design of high-density monolithic chips when CMOS are available.

The extension of the multiple sampling principle to the time-scale of the forthcoming hadron colliders is not so easy to be figured out, as it would require devices with switching times in the ns region. This is just one of the reasons of the increasing favour being attributed to the time- invariant solutions.

Some examples of time-invariant filters implemented in the multichannel preamplifier systems already exist. For instance, in AMPLEX, a CMOS signal processor for multi-element detectors, a semigaussian time-invariant filter is

present in each amplification channel [10]. The filters consists of four RC integrators

and one differentiator and is based upon feedback

16

12

8

4

0

Jamex weighting functions mmm jm

mm mmK'

I/./I",61 1 I //,A// I

V_ II 0 1 2 3 4

t[l.t s]

Z"~_'N \ -~x,x!

5 6 7 8

a)

Ideal truncated cusp vs. best discrete-time cusp

12 J

8

4

0

,4

0 4 8 12 t[~s]

16

b)

Fig. 15 - a) Weighting functions obtained with equal A, B coefficient and different values of the ratio between sampling period and dominant time- constant in the preamplifier. b) Discrete-time cusp realisation (a) compared to its time-invariant equivalent (b).

con f igu ra t i ons d e s i g n e d a round two transconductance operational amplifiers, one employed as input charge-sensitive loop, the other one as a second stage integrating loop.

In the a present version the filter response to a 6-impulse detector signal has a peaking time of about 750 ns.

An alternative approach to time-invariant filtering is based upon the circuit of fig. 16, which defines the following transfer function:

502 P.E Manfred~ V. Speziali / The state-of-the-art of low noise design ht particle physics

Vo(S) = sCnrp (12)

1 1 where rp = - - , r n - , and gm,p, gm,n are the

gm,p gmga

transconductances of the P and N-channel MOS respectively [7].

IIp

v~ (s) [

Fig. 16 - Balanced filtering structure.

"vbi~

v (s) O

The circuit of fig. 16 is an example of a filter where resistors are obtained as the inverse of transconductances of active devices. The values of the time constants can be adjusted by acting upon the standing currents I in the two branches.

The filter of fig. 16 is an open-loop structure. However, some cancellation of nonlinearity results from its balanced nature.

In the bipolar technology based upon the modem semicostum processes of large fT, the availability of resistors makes the time-invariant way to f'dtering suitable, especially when the time- constants are not required to be large.

The design of a semigaussian filter whose weighting function has a peaking time of about 10 ns has been described.

The filter consists of an input cell which

implements pole-zero cancellation and the first integration, followed by two more integration cells.

The solution is very neat and close to the concepts employed in the design of filters realised in discrete or hybrid form. The question, however, arises whether or not the time-invariant filters based on integrated resistors compatible with the bipolar technology are still a viable solution in going to high density monolithic chips.

Thinking of the filtering operation for the future hadron colliders, it can be stated that the time-invariant solution based on the 1/g m realisation of resistors for the CMOS circuits and the one employing integrated resistors available in bipolar processes may suit the time-scale requirements set by those accelerators.

The discrete-time, switched capacitor approach may have to be reconsidered once monolithic design based either on submicrometric channel N- MOS or GaAs MESFET becomes available.

9 . D I R E C T I O N S P U R S U I N G

W O R T H

It seem quite obvious to predict that during the next few years a good amount of design effort will be poured into the integration of the front-end electronics on the detector chip and this trend is stimulated basically by the need of making pixel detector and silicon drift chambers usable in the experimental conditions set by high luminosity colliders.

Design of front-end electronics on a separate chip still needs a considerable effort before being fully adequate to match the requirement of a high speed of operation and to be radiation-hard to a satisfactory extent.

Short channel radiation-hard CMOS processes look to be very promising. To judge how closely they may meet the requirements set by experiments at high luminosity colliders, more investigation about the limits speed in monolithic preamplifier systems and about the characteristics of radiation hardness at doses well above 10 nrad has to be carried out.

JFET-based monolithic processes seem to be suitably radiation hard, but the transition frequencies are not yet high enough. Steps are now being taken in the direction of shrinking the channel. The JFET-CMOS process described in sect. 5 already employs a 1.6 Pm gate NJFET. The

P.E Manfreda; V. Speziali / The state-of-the-art of low noise desigl in panicle physics 503

potentiality of this device has not yet been put in its Irue perspective.

Short channel parts (2.5 I~m-gate) have been recently developed in the buried-layer process and proven to be successful in upgrading the gm/Ci ratio considerably. Preliminary noise tests have shown that the thermal noise in the channel complies quite accurately with the theoretical gin" dependence, while their low-frequency noise behaviour is somewhat worse than for the corresponding units of longer channel [50].

Monolithic processes based on bipolar transistors, besides meeting the speed requirements set by experiments at high luminosity hadron colliders, have as an additional feature the availablity of resistors which makes the implementation of time-invariant filters possible. As to the radiation-hardness feature of bipolar processes more thorough investigation is required.

GaAs MESFETs, at the level of discrete devices or circuits obtained from arrays, were proven to be suitable for utilisation at short processing times in applications with small detector capacitances at room temperature and more proficient in calorimetry applications at cryogenic temperatures. Development of monolithic preamplifiers, now in progress, will help to answer the question about possible degradations in the noise behaviour of GaAs MESFETs when they are realised on the same substrate. One more question on GaAs refers to the aspect of radiation hardness, to settle which more investigation effort is required.

Mixed technological processes, making devices of different nature available on the same substrate, like BiCMOS and JFET-CMOS may be appreciable for the design versatility they offer provided that the CMOS components in the processes be made adequately radiation-hard.

A remarkable level of confidence is presently being conveyed on processes employing dielectric isolation. Dielectric isolation is a clearly effective step towards the achievement of device compatibility. Besides, it may bring about advantages in terms of gain-bandwidth product by reducing strays and substrate-related capacitances [39, 51]. The extent to which dielectric isolation may improve the radiation hardness characteristics, however, is still to be thoroughly investigated. Processes where dielectric isolation is adopted to arrive at circuits based on CMOS, PJFET, NPN bipolar transistors, each of them residing on an isolation island are being proposed for detector application [39].

An example of front-end preamplifier realised in SOS technology has been implemented and the preliminary test result seem to be encouraging [52].

C O N C L U S I O N S

The paper has reviewed the present status of integrated systems of detector and front-end electronics and monolithic front-end systems residing on a substrate different from the detector's. Suggestions to arrive at front-end solutions suitable for applications in experiments at high luminosity colliders have been given. Transition frequencies, radiation hardness and radiation-sensitivity of the noise appear issues of highest importance in the evaluation of how closely a monolithic process can match the requirements set by the future experiments of elementary particle physics.

A C K N O W L E D G E M E N T S

The authors acknowledge the contribution to the subjects discussed in this paper given by E. Gatti, V. Radeka, R.L. Chase, A. Longoni, S. Rescia, A. Hrisoho, C. de la Taille, P. Jarron, E. Beuville, M. Dentan, D. Camin.

The authors are grateful to J. Millaud and people of the microelectronics group who provided advanced information about activities going on at LBL.

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