rectangular dielectric rod antenna fed by air-substrate parallel strip...

6308 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 67, NO. 10, OCTOBER 2019

Rectangular Dielectric Rod Antenna Fed byAir-Substrate Parallel Strip Line

Yuefeng Hou , Student Member, IEEE, Yue Li , Senior Member, IEEE,

Zhijun Zhang , Fellow, IEEE, and Quan Xue , Fellow, IEEE

Abstract— In this paper, a rectangular dielectric rod antennafed by an air-substrate parallel strip line is proposed. Thedielectric rod has a compact and regular structure, which is easyto fabricate. The matching of the antenna is mainly determinedby the parallel strip line, which simplifies the design process.Fed by an air-substrate parallel strip line, the antenna obtainsa relatively uniform magnitude distribution on the surface anda moderate phase constant. With the length of 6 wavelengths atthe center frequency of 6 GHz, the antenna generates a highmeasured gain of 16.7 dBi. The measured 1 dB gain bandwidthis about 18.3%. Over the entire operating band from 5.4 to6.6 GHz, a stable endfire radiation pattern and a good matchingare achieved by the antenna.

Index Terms— Air substrate, parallel strip line, rectangulardielectric rod antenna.

I. INTRODUCTION

D IELECTRIC rod antennas have received extensive atten-tion for many years. With the merits of wide bandwidth,high radiation efficiency and stable radiation pattern, dielectricrod antennas have been widely applied to high-speed wirelesscommunication, radar, and satellite communication systems.During recent years, it is found that dielectric rod antennashave the advantages of low cost, lightweight, easy fabrication,and easy integration with semiconductor devices in millimeter-wave (mm-wave) band. Therefore, dielectric rod antennasare promising candidates for variable mm-wave applications,such as automobile collision avoidance radar, chip-to-chipcommunication, and mm-wave imaging.

The configuration of the dielectric rod antennas could beflexibly designed. To obtain a good impedance matching tofree space and decrease the sidelobe level, the taper dielectricrods [1]–[3] and dielectric rods loading with air holes [4], [5]were adopted in a large number of dielectric rod antennas.

Manuscript received December 23, 2018; revised May 6, 2019; acceptedMay 19, 2019. Date of publication June 6, 2019; date of current versionOctober 4, 2019. This work was supported by the National Natural ScienceFoundation of China under Contract 61525104. (Corresponding author:Zhijun Zhang.)

Y. Hou, Y. Li, and Z. Zhang are with the Department of ElectronicEngineering, Beijing National Research Center for Information Science andTechnology (BNRist), Tsinghua University, Beijing 100084, China (e-mail:[email protected]).

Q. Xue is with the School of Electronic and Information Engineering,South China University of Technology, Guanzhou 510006, China (e-mail:[email protected]).

Color versions of one or more of the figures in this article are availableonline at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TAP.2019.2920269

To further increase the fundamental-mode operating band-width, the dielectric rods with multiple layers of differentmaterials were applied to several dielectric rod antennas[6]–[8]. For enhancing the gains of the dielectric rod antennas,the dielectric rods could be designed with a large cross-section [9], [10] or a nonlinear profile optimized by geneticalgorithm [11]. By adding periodical elements on the dielectricrod, the impedance matching and the gain could be increasedsimultaneously [12]. Nevertheless, due to the irregular andcomplicated structures of the dielectric rods, the dielectric rodantennas might suffer from the complex design process, highfabrication cost, and low mechanical strength.

Dielectric rod antennas could be excited by several types offeed structures. For the ultra-wideband dielectric rod antennas,dipole antennas with reflector [13], [14], Vivaldi antennas[15]–[17], and conical or V-shaped waveguides [18]–[20] wereadopted as the feed structures. With the merits of small dimen-sion and broadband [21], standard open-end waveguides werepopularly utilized to feed the dielectric rod antennas [22], [23].When the end of open-end waveguides were added bylaunching horns [2], [3] or antipodal linearly tapered slotantennas [10], the unguided radiation from the feed struc-ture could be suppressed. The open-end waveguides couldbe designed using substrate integrated waveguide technique[24], [25], which reduces the integration complexity andfabrication cost of the dielectric rod antennas. For the feedstructures mentioned above, when they are employed as thefeed structure of the dielectric rod antennas, the dielectricrods are excited from one side with one feed point. Dueto the different modal field configurations between the feedstructures and the dielectric rods, the magnitude distributionson the surface of the dielectric rod are not uniform, whichmight have negative influences on the radiation patterns andgains [1].

This paper presents a rectangular dielectric rod antenna fedby an air-substrate parallel strip line. The parallel strip line isa continuous feed structure. It is placed in the middle of thedielectric rod. The length of the parallel strip line is the sameas the dielectric rod. As presented in Fig. 1, the structure ofthe proposed antenna is different from the typical dielectricrod antenna. By adopting the parallel strip line with air mediaas the feed structure, the proposed antenna has the followingfeatures

1) The dielectric rod has a regular and compact structure,which is easy to fabricate.

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https://orcid.org/0000-0003-4771-0619https://orcid.org/0000-0001-9562-3136https://orcid.org/0000-0002-8421-2419https://orcid.org/0000-0002-4226-2127

HOU et al.: RECTANGULAR DIELECTRIC ROD ANTENNA FED BY AIR-SUBSTRATE PARALLEL STRIP LINE 6309

Fig. 1. Comparison of the geometries between the (a) typical dielectric rodantenna [1] and the (b) proposed dielectric rod antenna.

2) The reflection magnitude of the antenna is insensitive tothe material and dimension of the dielectric rod and ismainly determined by the matching of the parallel stripline, which simplifies the design process.

3) The dielectric rod could be readily designed with a longstructure and a relatively uniform surface magnitudedistribution, which is helpful to realize a high gain.

4) The antenna has a moderate phase constant over a broadbandwidth, leading to a stable endfire radiation pattern.

5) With a matching port (Port 2), the undesired energycould be absorbed to realize a high front-to-back ratio.

Based on the performance mentioned above, the antenna iseasy to design and fabricate, and it could realize a stableendfire radiation pattern with high gain in a broad bandwidth.

In Section II, the detailed configuration, working principle,design considerations, and performance analysis are systemat-ically described. In Section III, a fabricated prototype of theproposed antenna is presented to verify the design strategy.

II. ANTENNA DESIGN AND ANALYSIS

A. Antenna Geometry

The geometry of the proposed rectangular dielectric rodantenna is plotted in Fig. 2. Without loss of generality,the center frequency of the antenna is selected as 6 GHz. Theantenna consists of a parallel strip line and a dielectric rod. Theparallel strip line is the feed structure, and the dielectric rodis the radiating structure of the antenna. To properly energizethe antenna, the two ends of the parallel strip line are bothconnected to a conversion transformer and a coaxial line.

With a simple structure and reasonable operating bandwidth,the conversion transformer shown in [26, Fig. 4] is adopted inthe proposed antenna. As depicted in Fig. 2(a), the classicalconversion transformer in [26] is composed of a taper, a closedexpanded shield, and a coaxial line extended shield. Becausethe dielectric rod should be directly fed by the parallel stripline, the expanded shield surrounding the parallel stripline,which is used for suppressing the undesired radiation from theparallel strip line, is removed in the conversion transformer ofthe proposed antenna. As plotted in Fig. 2(b), the conversiontransformer of the proposed antenna consisted of a taper, anda balun is presented. The taper structures are used to mitigate

Fig. 2. Geometry of the proposed rectangular dielectric rod antenna. (a) Crosssection of the classical conversion transformer in [26]. (b) Cross section ofthe conversion transformer in the proposed antenna. (c) 3-D view of the feedstructure. (d) 3-D view of the proposed antenna.

the effect of the discontinuities. The balun is formed by thecoaxial line extended shield and the other part of the expandedshield. It is utilized to reduce the electric current on the outside


TABLE I

DETAILED DIMENSION OF THE PROPOSED ANTENNA

Fig. 3. Simulated cutoff frequency fc of the TE1 and TM2 modes versusthe height H and the relative permittivity εd of the dielectric rod.

of the coaxial line and maintain the electric currents on thetwo types of transmission line in their proper relations [26].

The air-substrate parallel strip line operates on TEM mode.With a very low profile FH of 2 mm (0.04 wavelength atthe center frequency of 6 GHz), the parallel strip line has aweak radiating capability and good transmission performance.As plotted in Fig. 2(b), the parallel strip line is located in themiddle of the dielectric rod. The dielectric rod is designedby Teflon (εd = 2.05, tanσ = 0.002). It is composed of twoidentical rectangular dielectric structures. The width W of thedielectric rod is the same as the parallel strip line. With arectangular dielectric rod and an embedded parallel strip line,the structure of the antenna is compact and regular, which iseasy to design and fabricate. The detailed dimensions of theantenna are shown in Table I.

B. Design Considerations

In the design of the proposed dielectric rod antenna,the dimension and the material of the dielectric rod are themost crucial parameters. In this section, the field configurationand parameters of the dielectric rod are investigated elabo-rately.

First of all, to obtain a stable performance, the dielectricrod should work on the fundamental operating mode overthe entire operating bandwidth. The fundamental operatingmode of the dielectric rod is the transverse magnetic zero(TM0) mode. As shown in Fig. 2(b), due to the effect of theparallel strip line, the proposed dielectric rod antenna could

Fig. 4. Schematic vector electric field distribution of the (a) parallel stripline and the (b) dielectric rod on the yoz plane. Schematic vector magneticfield distribution of the (c) parallel strip line and the (d) dielectric rod on thexoy plane. The parallel strip line works on TEM mode and the dielectric rodworks on TM0 mode. To simplify the explanation, only the field distributionsinside the dielectric rod are plotted in (b) and (d).

be regarded as a combination of two half-mode dielectric rodsmounted on the conducting plane. Owing to the boundary con-dition, the dielectric rod of the proposed antenna only supportsthe operating modes of TMm odd and transverse electric (TEm)even modes [27]. Therefore, the lowest operating three modesof the proposed antenna are TM0, TE1, and TM2.

Based on (1), the cutoff frequency fc of the TE1 and TM2operating modes are determined by the height H and therelative permittivity εd of the dielectric rod [27]. As plottedin Fig. 3, when the height H or the relative permittivity εd ofthe dielectric rod increase, the cutoff frequency fc of higherorder modes decreases. To suppress the higher order modes inthe dielectric rod, the material of low relative permittivity εdis a better choice for the dielectric rod

( fc)m =m

4H√

μdεd − μ0ε0m= 0, 2, 4, . . . ,T Mz(odd)m= 1, 3, 5, . . . ,T Ez(even) (1)

Next, the feasibility of the dielectric rod fed by a parallelstrip line is evaluated from the perspective of electric andmagnetic field configurations. The field configurations of thedielectric rod working on TM0 mode and the air-substrateparallel strip line working on TEM mode are presented inFig. 4. From Fig. 4, the vector electric and magnetic field dis-tributions outside the parallel strip line and inside the dielectricrod are similar to each other. It indicates that the energyin the parallel strip line could be coupled by the dielectric


Fig. 5. Simulated S-parameters of the feed structure and the proposedantenna. The configurations of the feed structure and the proposed antennaare exhibited in Fig. 2.

rod when the parallel strip line is placed in the middle ofthe dielectric rod. However, as shown in Fig. 4(a) and (c),the energy of the parallel strip line mainly concentrates on thecenter. For feeding the dielectric rod effectively, the parallelstrip line should be long enough and work as a continuousfeed structure. For the proposed antenna, the length L of theparallel strip line is designed similarly as the dielectric rod.

Due to the similar modal field configurations plotted inFig. 4, when the parallel strip line is loaded by dielectricrod antenna, the reflection magnitude of the parallel strip lineis not affected seriously. As illustrated in Fig. 5, over theentire operating bandwidth, the reflection magnitude of theparallel strip line in Fig. 2(a) and the proposed dielectricrod antenna in Fig. 2(b) are similar to each other, whichare both lower than −15 dB. However, since the energy inthe parallel strip line is coupled effectively to the dielectricrod, the transmission magnitude of the proposed dielectricrod antenna drops dramatically around the center frequencycompared with the parallel strip line alone.

Then, the influence of the dielectric rod’s material onthe performance of the proposed antenna is investigated.The reflection magnitudes and the phase constants βz alongthe propagation direction (z-axis) of the proposed antennaversus different relative permittivity εd of the dielectric rodare exhibited in Fig. 6. As a comparison, the phase constantβH−W satisfies Hansen–Wordyard (H–W) condition [28] andphase constant β0 in free space are added together. In H–Wcondition, the phase difference between the surface wave andthe free-space wave is approximately 180◦, which is presentedin the following equation:

LβH_W − Lβ0 ≈ π. (2)For a dielectric rod working on TM0 mode, the phase

constant βz could be represented by (3)–(5) and βyd, ay0 andβz must all be real, where ay0 is the attenuation constant alongy-axis in free space and βyd is the phase constant along y-axisin dielectric rod [27]. By solving (3)–(5) to achieve the phaseconstant βz, it is found that the dielectric rod with a higher

Fig. 6. Simulated reflection magnitude and normalized phase constant of theproposed antenna at 6 GHz versus relative permittivity εd of the dielectricrod. At a different relative permittivity εd , the dimension of the dielectric rodis optimized to make the antenna obtain the maximum leakage constant.

Fig. 7. Simulated reflection magnitude and normalized leakage constant ofthe proposed antenna at 6 GHz versus the height H of the dielectric rod. Therelative permittivity εd of the dielectric rod is 2.05.

dielectric constant has a large dispersion versus frequency [27]

β2z = β20 + α2y0 = ω2μ0ε0 + α2y0 (3)β2z = β2d − β2yd = ω2μdεd − β2yd (4)

βyd tan(βydh) = εdαy0. (5)As shown in Fig. 6, with the relative permittivity εd

increases from 2.0 to 6.0, the phase constant βz of the proposedantenna raises slightly and keeps close to βH−W. Consideringthe influence of the dispersion and the higher order modes,the dielectric constant of the dielectric rod should be low.Owing to the merits of low cost and easy fabrication, Teflon isadopted to make up the dielectric rod of the proposed antenna.

Finally, the influence of the dielectric rod’s dimension on thecoupling between the dielectric rod and the parallel strip lineis studied. To simplify the design process, the width W of thedielectric rod is the same as the parallel strip line. As shownin Fig. 7, when the material of dielectric rod is specified, theleakage constant α is adjusted with the change of the heightH . When the height H of the dielectric rod is 15 mm, about


99.5% of the energy in the parallel strip line is coupled todielectric rod based on the following equation [1]. From Fig. 7,with suitable width W and height H of the dielectric rod, theparallel strip line could feed the dielectric rod effectively

Percentage of power coupling = 100×(

1 − e −4παLk0λ0)

. (6)

In conclusion, the design process of the proposed antennais relatively simple. As exhibited in Figs. 5–7, the reflectionmagnitude of the proposed antenna is insensitive to the mate-rial and dimension of the dielectric rod, and the reflectionmagnitude is only determined by the matching of the parallelstrip line. Therefore, in the design process of the proposedantenna, after designing a parallel strip line with a goodmatching, only the material and the dimension of the dielectricrod need to be considered to couple the energy effectively fromthe parallel strip line.

C. Antenna Performance Analysis

The performance of the proposed antenna is investigated inthis section, which further explains the operating principle ofthe proposed antenna.

For the typical dielectric rod antennas, they might have anonuniform energy distribution on the surface of the dielectricrod, which is caused by the single feed point and the dif-ferent field configurations between the feed structure and thedielectric rod [1]. Fed by a parallel strip line, the charactersof the proposed antenna has some different from the typicaldielectric rod antennas. As shown in Fig. 8(a), when port 1of the proposed antenna is excited, and port 2 has a goodmatching, the energy in the parallel strip line is graduallycoupled by the dielectric rod, and the energy in the dielectricrod disperses along the propagation direction (z-axis).

The electric field magnitude distributions at different posi-tions are presented in Fig. 9 to evaluate the energy distributionin the dielectric rod. As shown in Fig. 9(a), when the positionis close to the feed port (Port 1), the energy is mainly gatheredaround the parallel strip line. At the position close to theterminal port (Port 2), the energy in the dielectric rod isrelatively uniform. As shown in Fig. 9(b), influenced by theparallel strip line, the energy distribution on the surface of thedielectric rod is still relatively uniform, although the energyin the parallel strip line reduces gradually.

Fig. 10 presents the normalized leakage constant α/k0 andphase constant βz/k0 versus frequency. Around the centerfrequency of 6 GHz, the coupling between the dielectric rodand the parallel strip line is higher, achieving a higher leakageconstant. Based on (6), the value of power coupling is higherthan 68.7% from 5 to 7 GHz with the maximum value of99.7%. With the frequency increasing, the normalized phaseconstant βz/k0 of the proposed dielectric rod antenna risesat the same time, which is similar to the performance ofthe dielectric waveguide antennas [29], [30]. From Fig. 10,the value of the normalized phase constant βz/k0 is moderateand close to the phase constant of H–W condition over theentire operating bandwidth, which indicates that the proposedantenna satisfies the requirement of the endfire radiation.

Fig. 8. Simulated electric field magnitude distribution of the proposedantenna at 6 GHz on the (a) reference plane A and (b) reference B,respectively.

With a relatively uniform energy distribution on the sur-face of the dielectric rod and a moderate phase constant,the proposed antenna could generate a large effective radiatingaperture as plotted in Fig. 8(b), resulting in a high endfire gain.

III. EXPERIMENTAL VERIFICATION

A prototype is fabricated and tested to provide verifica-tion of the proposed design strategy. The configuration ofthe proposed dielectric rod antenna is exhibited in Fig. 11.The height FT of the copper strip is 0.5 mm. A foam with thethickness FH of 2 mm is filled between the two copper stripsas support. The foam and the two copper strips form theair-substrate parallel strip line. The two dielectric sticks arelocated on the two sides of the parallel strip line, respectively.The expanded shield of the balun is realized by a hollowmetallic cylinder. The proposed antenna is fed from Port 1, andPort 2 is terminated with a matching load. By using a N5071Bvector network analyzer (300 kHz–9 GHz), the measuredS-parameters was obtained. In a far-field anechoic chamber,the gains and radiation patterns of the proposed antenna weremeasured.

The simulated and measured radiation pattern of the antennaat the center frequency is presented in Fig. 12. The simulatedand the measured half-power beam widths of the radiationpattern on E-plane are 26◦ and 25◦, respectively. The simulatedand measured half-power beam widths of the radiation pattern


Fig. 9. Simulated electric field magnitude distribution of the proposedantenna at 6 GHz. (a) Electric field magnitude distribution along the y-axis ondifferent positions of the top-half dielectric rod. (b) Electric field magnitudedistribution along the z-axis on the top surface of the dielectric rod and themiddle of the parallel strip line, respectively.

Fig. 10. Simulated normalized leakage constant and phase constant of theproposed antenna versus frequency. As a comparison, the phase constants infree space and theoretical Hansen–Woodyard condition are added in the figure.

on H-plane are 28◦ and 26◦, respectively. From Fig. 12,the cross-polarizations are lower than −30 dB at the mainbeam direction. As shown in Fig. 13, from 5 to 7 GHz, the

Fig. 11. Photograph of the fabricated proposed antenna.

Fig. 12. Simulated and measured normalized radiation patterns of theproposed antenna at the center frequency of 6 GHz. (a) E-plane. (b) H-plane.

simulated and measured reflection magnitude of the antennaare both lower than −15 dB, which indicates that the antennahas a good matching. At the center frequency of 6 GHz,the simulated and measured transmission magnitude are bothlower than −26 dB. The simulated transmission magnitude


TABLE II

COMPARISON OF DIMENSIONS, GAINS, AND OPERATING BANDWIDTH (NG: NOT GIVEN)

Fig. 13. Simulated and measured S-parameters of the proposed antenna.

is lower than −10 dB from 5.5 to 6.6 GHz. The measuredtransmission magnitude is lower than −10 dB from 5.3 to6.7 GHz. According to Fig. 13, most of the energy in theair-substrate parallel strip line is coupled by the dielectric rodantenna over a wide bandwidth around the center frequency.The simulated and measured endfire and backfire gains areillustrated in Fig. 14. Over the entire operating bandwidthfrom 5.4 to 6.6 GHz, the simulated and measured endfiregains are both higher than 15 dBi, and the simulated andmeasured backfire gains are both lower than 1.5 dBi. Thesimulated and measured gains at the center frequency are16.8 and 16.7 dBi, respectively. The maximum simulated gainis 17.4 dBi at 6.5 GHz, and the maximum measured gainis 17.8 dBi at 6.4 GHz. The simulated and measured 1 dBgain bandwidths are about 20.0% and 18.3%, respectively. Thesimulated efficiency is also presented in Fig. 14. It is notedthat most of the energy is radiated around the center frequency.From 5.4 to 6.6 GHz, the efficiency of the antenna is higherthan 86.2%. The slight difference between the simulatedand measured results may be attributed to the fabricationand assembly errors. However, it still indicates that goodagreement between the simulated and measured results.

Table II exhibits a comparison between the proposedantenna and the dielectric rod antennas in the open literature.The dimensions and gains of the antennas are calculated at the

Fig. 14. Simulated and measured endfire gains, and simulated efficiency ofthe proposed antenna.

center frequency. The gain-to-length ratio is calculated basedon (7), which is defined in [1]. The length in (7) is the physicallength of the antenna. As presented in Table II, comparingwith most of the dielectric rod antennas, such as [6] and [23],the proposed antenna has a relatively high gain-to-length ratioand could be designed with a longer structure to achieve ahigher gain. Comparing with the dielectric rod antenna withhigh gain-to-length ratio [10], [11], the proposed antenna has awider operating bandwidth or a narrower structure. Therefore,the proposed antenna has the merits of compact structure, highgain, high gain-to-length ratio, and broadband simultaneously.Moreover, as mentioned earlier, the proposed antenna is easyto design and fabricate

Gain_to_length ratio ∼= G×λ0L

. (7)

IV. CONCLUSION

This paper proposes a dielectric rectangular dielectric rodantenna fed by an air-substrate parallel strip line. Due to thesimilar modal field configurations, the dielectric rod could cou-ple the energy from the parallel strip line with little effect onthe reflection magnitude of the parallel strip line. The materialand the dimension of the dielectric rod have an influence on theoperating mode of the dielectric rod and the coupling abilitybetween the parallel strip line and dielectric rod. The energy


coupled from the parallel strip line disperses gradually in thedielectric rod along the propagation direction (z-axis) leadingto a relatively uniform magnitude distribution. Designed withthe material of low dielectric constant, the dielectric rodworking on TM0 mode has a moderated phase constant overthe entire operating bandwidth.

With the advantages of high gain, high gain-to-length ratio,stable radiation pattern, broadband, and easy design, the pro-posed antenna is a good candidate for various high gain andbroadband applications. However, it is found that when thefrequency deviates from the center frequency, the transmissionmagnitude increases, which leads to the 1 and 3 dB gain band-width reduction. For this reason, further studies are requiredto improve the gain-bandwidth performance of the proposedantenna.

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Yuefeng Hou (S’16) received the B.S. degree fromthe University of Electronic Science and Technologyof China, Chengdu, China, in 2015. He is currentlypursuing the Ph.D. degree in electrical engineeringwith Tsinghua University, Beijing, China.

His current research interests include leaky-waveendfire antenna array, reduced massive MIMOantenna array, and metamaterial.

Yue Li (S’11–M’12–SM’17) received the B.S.degree in telecommunication engineering fromZhejiang University, Hangzhou, China, in 2007, andthe Ph.D. degree in electronic engineering fromTsinghua University, Beijing, China, in 2012.

In 2012, he joined the Department of Elec-tronic Engineering, Tsinghua University, as a Post-Doctoral Fellow. In 2013, he joined the Departmentof Electrical and Systems Engineering, University ofPennsylvania, Philadelphia, PA, USA, as a ResearchScholar. He was a Visiting Scholar with the Insti-

tute for Infocomm Research (I2R), A*STAR, Singapore, in 2010, and theHawaii Center of Advanced Communication (HCAC), University of Hawaiiat Manoa, Honolulu, HI, USA, in 2012. Since 2016, he has been withTsinghua University, where he was an Assistant Professor and is currentlyan Associate Professor with the Department of Electronic Engineering. Hehas authored or coauthored over 110 journal papers and 45 internationalconference papers. He holds 17 granted Chinese patents. His current researchinterests include metamaterials, plasmonics, electromagnetics, nanocircuits,mobile and handset antennas, MIMO and diversity antennas, and millimeter-wave antennas and arrays.

Dr. Li was a recipient of the Issac Koga Gold Medal from URSI GeneralAssembly in 2017, the Second Prize of the Science and Technology Award ofthe China Institute of Communications in 2017, the Young Scientist Awardsfrom the conferences of ACES 2018, AT-RASC 2018, AP-RASC 2016,


EMTS 2016, and URSI GASS 2014, the Best Paper Awards from theconferences of CSQRWC 2018, NCMMW 2018 and 2017, APCAP 2017,NCANT 2017, ISAPE 2016, and ICMMT 2016, the Outstanding DoctoralDissertation of Beijing Municipality in 2013, and the Principal Scholarship ofTsinghua University in 2011. He is serving as an Associate Editor for the IEEETRANSACTIONS ON ANTENNAS AND PROPAGATION, the IEEE ANTENNASAND WIRELESS PROPAGATION LETTERS, and Computer Applications inEngineering Education, and also on the Editorial Board of Scientific Report.

Zhijun Zhang (M’00–SM’04–F’15) received theB.S. and M.S. degrees from the University of Elec-tronic Science and Technology of China, Chengdu,China, in 1992 and 1995, respectively, and the Ph.D.degree from Tsinghua University, Beijing, China,in 1999.

In 1999, he joined the Department of ElectricalEngineering, The University of Utah, Salt Lake City,UT, USA, as a Post-Doctoral Fellow, where he wasappointed as a Research Assistant Professor in 2001.In 2002, he joined the University of Hawaii at

Manoa, Honolulu, HI, USA, as an Assistant Researcher. In 2002, he joinedAmphenol T&M Antennas, Vernon Hills, IL, USA, as a Senior Staff AntennaDevelopment Engineer and was then promoted to the position of AntennaEngineer Manager. In 2004, he joined Nokia Inc., San Diego, CA, USA, as aSenior Antenna Design Engineer. In 2006, he joined Apple Inc., Cupertino,CA, USA, as a Senior Antenna Design Engineer and was then promotedto the position of Principal Antenna Engineer. Since 2007, he has been withTsinghua University, where he is currently a Professor with the Department ofElectronic Engineering. He has authored the book Antenna Design for MobileDevices (Wiley, 1st ed. 2011, 2nd ed. 2017).

Dr. Zhang served as an Associate Editor for the IEEE TRANSACTIONS ONANTENNAS AND PROPAGATION from 2010 to 2014 and the IEEE ANTENNASAND WIRELESS PROPAGATION LETTERS from 2009 to 2015.

Quan Xue (M’02–SM’04–F’11) received the B.S.,M.S., and Ph.D. degrees in electronic engineeringfrom the University of Electronic Science andTechnology of China (UESTC), Chengdu, China,in 1988, 1991, and 1993, respectively.

In 1993, he joined UESTC, as a Lecturer, wherehe became a Professor in 1997. From 1997 to1998, he was a Research Associate and then aResearch Fellow with the Chinese University ofHong Kong, Hong Kong. In 1999, he joined theCity University of Hong Kong, Hong Kong, as a

Chair Professor of microwave engineering, where he was the AssociateVice President (Innovation Advancement and China Office) from 2011 to2015, the Director of the Information and Communication TechnologyCenter, Shenzhen, China, and the Deputy Director of the State KeyLaboratory of Millimeter Waves. In 2017, he joined the South ChinaUniversity of Technology, Guangzhou, China, where he is currently aProfessor and also the Dean of the School of Electronic and InformationEngineering. He has authored or coauthored over 300 internationallyrefereed journal papers and over 130 international conference papers. Heholds five granted Chinese patents, 15 granted U.S. patents, and 26 filedpatents. His current research interests include microwave/millimeter-wave/tetrahertz passive components, active components, antenna,microwave monolithic integrated circuits, and radio frequency integratedcircuits.

Dr. Xue served as an AdCom Member of IEEE MTT-S from 2011 to 2013,an Associate Editor for the IEEE TRANSACTIONS ON MICROWAVE THEORYAND TECHNIQUES from 2010 to 2013, an Editor of the International Journalof Antennas and Propagation from 2010 to 2013, and an Associate Editorfor the IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS from 2010 to2015. He has been an Associate Editor of the IEEE TRANSACTIONS ONANTENNAS AND PROPAGATION since 2016. He was a recipient of the 2017H. A. Wheeler Applications Prize Paper Award.

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