fast turn off

8/2/2019 Fast Turn Off

1/16

Application Note 22Issue 1 May 1996

AN22 - 1

High Frequency DC-DC Conversion using HighCurrent Bipolar Transistors

Neil ChaddertonDino Rosaldi

Introduction

DC-DC conversion is one of thefundamental circuit functions within theelectronics industry and addresses awide f ie ld o f market sectors ,applications and supply requirements. Ageneral trend, (to pursue cost, size andreliability advantages) has been toreduce the size of the DC-DC converter

systems, and as the inductive elementsof such a system are quite often thebulkiest components, increases inswitching frequency provide onemethod to a l low th is . Switch ingfrequencies therefore have increasedfrom tens of kHz to hundreds of kHz,which has forced a revision of theenabling switching devices.

It is an often assumed maxim thatbipolar transistors are useful for DC-DCconversion functions up to perhaps50kHz, and for service beyond thisfrequency, that MOSFETs provide theonly solution. The purpose of thisapplication note is to demonstrate thatbipolar transistors possessing superiorgeometrys - thereby providing a highercurrent capability per die area, can andare operated to reasonably h igh

switching frequencies with minimalloss. This often allows a more compact

design (due to the higher sil iconefficiency of bipolar technology) andlower cost.

Background

To obtain the most efficient performancefrom bipolar t ransis tors a t h igh

switching frequencies, an appreciationof the basic switching mechanisms andbase charge analysis is useful. AppendixA provides an introduction to bipolartransistor switching behaviour, andappendix B provides references formathematical treatments.

There are various methods forincreasing the maximum switchingspeed of bipolar transistors. Some of

these rely on preventing saturation ofthe device, thus vastly reducing thestored charge, while other methodsremove the charge at turn-off. Figures 1,2 and 5 show a number of commonspeed-up networks. Figures 1a through1d show various options for preventingsaturation, and effectively reduce/limitbase drive when the collector terminalhas fallen to a specific level. (Figure 1a is

often termed a Schottky transistor, andis often used for IC transistors, and

400kHz Operation with Optimised Geometry Devices


2/16

Figures 1c and 1d are variants of the socalled Baker clamp circuit).

These methods (the Baker clamp beingpreferred for high power applications) ofcourse do not allow the transistorscollector-emitter voltage to saturate to avery low level, and so the resultanton-state loss may be high and thereforeprohibit the use of smaller packaged,though adequately current capabledevices. Figures 2 and 5 show twomethods that allow true saturation bypermitting sufficient forward base drive,while removing charge quickly at turn-off.Design of such networks is non-critical,and by suitable choice of componentsallow high levels of base overdrive with nopenalty to turn-off time duration.

Figure 2 shows a method of speeding upPNP devices to enable replacement oflarge P-Channel MOSFETs - this particularcircuit being designed for fast charging ofbattery packs by BenchmarqMicroelectronics. Figures 3 and 4 show therelevant waveforms, for a standardpassive turn-off method (base-emitterpull-up resistor) and the speed-up circuitof Figure 2 - the combined storage and falltimes being reduced from 1.2s to 80ns.Importantly, the major switching losscontributor - the fall time, has beensignificantly reduced to 40ns, which iscomparable to, or better than P-ChannelMOSFET performance. This allows costeffective replacement of large packageddevices.


AN22 - 2

Q2

Q1

Q1

Q1

R1

R2

R3C1

D1

D2

D3

Q1

a b

c d

Figure 1.

Non-saturating Bipolar Transistor Speed-up Methods.


3/16


AN22 - 3

Q12N3904

1K

L110uH

2N3904

1N5820

L2

200uH

100uF

+Bat

1N4148

Q3

ZTX789AInput

Drive

Feedback

Q2

330

Figure 2.PNP Transistor Speed-up Circuit to Allow Replacement of P-Channel MOSFETs withinHigh Current Converters. (The circuit shown is suitable for output currents up to 1.5A;other variants are capable of operation to 5A output).

Figure 3.Turn-off Waveforms for PNP Step-downConverter using Passive Turn-off.Upper Trace - Collector-to-0V, 5V/div;Middle Trace - PNP base-to-0V;Lower Trace - PWM IC Output, 5v/div.Timebase at 2s/div.

Figure 4.Turn-off Waveforms for PNP Step-downConverter (of Figure 2) using ActiveTurn-off.Upper Trace - Collector-to-0V, 5V/div;Middle Trace - PNP base-to-0V;Lower Trace - PWM IC Output, 5v/div.

Timebase at 2s/div.


4/16


5/16


6/16


FMMT718

22pF

LL5818

220uH

30K

470uF

1000uF

150

1.8n

PWM IC

+ve

-ve

Osc

Gnd

E

CILim

Vin

Vin

0V

+5V/+3.3V

0V

0.05

220

470K

220pF 13K7K5/

(Set Vout)

Figure 9.Basic Step-Down DC-DC Converter using the LM3578 and the FMMT718.(With values shown, Fop=50kHz, IB=43mA, peak efficiency=88%).

0.01 10Output Current (A)

Efficiency v Output Current

10

100

Efficiency(%)

0.1 1

20

40

60

80

1

2

3

5 15

Efficiency(%)

100

0

Input Voltage (V)

Efficiency v Input Voltage

20

40

60

80

7 9 11 13

Figure 10.Efficiency against Load Current for theConverter of Figure 9, with IB as aparameter.Curve 1 - IB=9.4mA; Curve 2 - IB=43mA;Curve 3 - IB=170mA. Fop=50kHz; Vin=7V;Vout=5V.

Figure 11.Efficiency against Input Voltage for theConverter of Figure 9. (IB=43mA; Iout=1A).

AN22 - 6


7/16


AN22 - 7

Optimisation, and some scope for higherfrequency operation is possible with thisbasic converter. Figure 12 shows theeffect of varying the inductors value,with the switching frequency set to90kHz. Turn-on time was recorded at55ns, and turn-off at 1.5s.

To investigate options for higherswitching frequencies requires a changeto the PWM controller IC. Figure 13shows a circuit based on the TI5001device. This IC is capable of operation upto a switching frequency of 400kHz, andcan sink a maximum of 20mA into the VOpin.

The initial circuit was configured tooperate at 150kHz, with the FMMT718base current set to 9mA by a 560 baseresistor (R1) - effectively a forced gain of

222 at 2A. Turn-on and turn-off timeswere measured at 200ns and 1.44s

0.01 10

Output Current (A)


10

100

Efficiency(%)

0.1 1

20

40

60

80

1

2

Figure 12.Efficiency against Load Current for theConverter of Figure 9.IB=43mA; Fop=90kHz; Vin=7V; Vout=5V.

Effect of Variation of Inductor Value.Curve 1 - 25H, Curve 2 - 180H.

TL5001

Vcc

Vo

Comp

F/b

Gnd

SCP

DTC

RT

ZTX718

4.7nF

R1

560

1000uF

470uF

L1 40uH

D1

1N5818

Rf

22k

2k2

5k6

5k6

Vin

Vout

10nF

0.1uF

Figure 13.Basic Step-Down DC-DC Converter using the TI5001 and the FMMT718. (With values

shown, Fop=150kHz, IB=9mA, peak efficiency=81%).


8/16


0.01 10

Efficiency(%)

100

0

Output Current (A)


20

40

60

80

0.1 1

1

2

Figure 14.Efficiency against Load Current for theConverters of Figure 13 and 16.

Q1

R1

Q2

R2

C1

D1

PWM IC

Figure 15.Bipolar Transistor Turn-off Circuit to allowCapacitive Turn-off Charge Neutralisationwith PWM Controller IC.

TL5001

Vcc

Vo

Comp

F/b

Gnd

SCP

DTC

RT

4.7nF

Rf

2k2

5k6

5k6

VinVout

0.1uF

10nF

560

FMMT718

390

2.2nF

1N4148

470uF

D1

1N5818

L1 40uH

FMMT3904

1000uF

22k

Figure 16.Basic Step-down DC-DC Converter using the TI5001/FMMT718 Combination with

Capacitive Turn-off Circuit.

AN22 - 8


9/16


AN22 - 9

respectively. The chart shown in Figure14 (curve 1) illustrates the resultant

efficiency versus load profile. Thecircuits conversion efficiency peaks at81% at 23mA output, but falls thereafterto 71% at 2A, due mainly to switchinglosses, though in some part toincreasing VCE(sat) at higher currents. Thelatter factor could be addressed in partby considering the load requirements,and opt imis ing for the re levantcondition, but the switching losseswould remain.

By considering the base turn-off chargeand effecting a suitable turn-off circuit(Figure 15) the circuit shown in Figure 16was produced. This shows a significantimprovement in conversion efficiency atmedium to higher currents; as shown in

Figure 17.Switching Waveforms for the circuit ofFigure 16, operating at 150kHz. Uppertrace; 2V/div, Collector-to-0V. Lowertrace; 2V/div, Base-to-0V. Vin=7V,Vout=5V, IL=220mA.

0.01 10

Efficiency(%)

100

0

Output Current (A)

20

40

60

80

0.1 1

1

2 4

3

5

Figure 18.Efficiency against Load Current for the Converter of Figure 16. Curve 1 - F op=150kHz;curve 2 - 220kHz; curve 3 - 300kHz; curve 4 - 400kHz. I B=10mA. Curve 5 - 400kHz andIB=5mA.Vin=7V, Vout=5V.


10/16

Figure 14, curve 2. This shows a peak

efficiency of 92%. Figure 17 shows theswitching waveforms, including thecollector-to-0V waveform - note therapid turn-off edge; this was measuredto be 25ns. The efficiency at lowercurrents has of course reduced, due tothe extra current taken by the turn-offcircuit. This efficiency profile can bemodified for specific applications bysacrificing high current efficiency infavour of low current performance or

vice-versa.

Further modifications were effected toassess performance at still higherswitching frequencies. Figure 18 showsthe efficiency/load current profiles forswitching frequencies from 150kHz to400kHz. Figure 19 shows the efficiencyagainst input voltage for the 220kHz


155

Efficiency(%)

100

0

Input Voltage (V)

Efficiency v Input Voltage

20

40

60

80

7 9 11 13

Figure 19.Efficiency against Input Voltage for theConverter of Figure 16.Fop=220kHz; Vout=5V; load current=1A.

Figure 20.Switching Waveforms for the Converterof Figure 16 operating at 400kHz.Turn-on edges. Risetime= 20ns.Upper trace - Base-to-0V, 2V/div.Lower trace - Collector-to-0V, 2V/div.Timebase at 50ns/div.

Figure 21.Switching Waveforms for the Converterof Figure 16 operating at 400kHz.Turn-off edges. Falltime=30ns.Upper trace - Base-to-0V, 2V/div.

Lower trace - Collector-to-0V, 2V/div.Timebase at 50ns/div.

AN22 - 10


11/16


AN22 - 11

version - the curve varying little over the

measured range. Figure 18 curve 5, isagain for a switching frequency of400kHz, but with lower base drive(RB=680) and a reduction to 1.5nF forthe turn-off capacitor. The oscillographsshown in Figures 20 and 21 show theturn-on and turn-of f swi tch ingwaveforms for the 400kHz version.These show collector rise and fall timesof 20ns and 30ns respectively.

Conclusions

This application note has demonstratedthat with due attention to the base andcollector charge phenomena in bipolartransistors, the operating switchingfrequency of those parts can beextended well beyond the currentlyaccepted notional maximum of 40kHz, toseveral hundred kHz. Various speed-up

methods have been summarised, andexamples of those particularly suitableand complementary to high currentcapable transistors examined, andcircuit examples presented.


12/16


13/16


14/16

charge necessary for small signal typesZTX107, ZTX300 and ZTX500, andswitching transistors ZTX310 andZTX510.


Figure 25.Trigger Waveforms and Effect of VaryingTrigger Capacitance. Upper trace - Inputwaveform; lower traces - output atcollector, (a) Variable capacitor C1 belowcritical value, (b) C1 at critical value withcollector voltage rising to 90% of final value,(c) C1 above critical value. Horizontalscale=200ns/div, Vertical scale=2V/div.

ZTX300, IB=1mA, IC=10mA.

Q1

RCRB

+5V

To Oscilloscope

0VInput

C1

Figure 24.Circuit to Determine Minimum Value of Trigger Charge for Bipolar Transistor.

AN22 - 14

100u 1m 10mIB - Base Current (A)

Min. Trigger Charge v IB for Small Signal

and Switching Transistors

10p

10n

MinimumT

riggerCharge(C)

ZTX310100p

1n

abc

cba

ZTX107ZTX300ZTX500

ZTX510

Figure 26.Minimum Trigger Charge for Small Signaland Switching Transistors. Forced Gains(IC/IB) of : a)10; b)20; c)40.No significant difference observed fordifferent forced gains on the ZTX310 series.


15/16


AN22 - 15

Appendix BReferences

(References 1 , 2 and 3 conta inmathematical treatment of chargeanalysis with respect to switchingcharacteristics of bipolar transistors -similar background can be found in mostsemiconductor physics books).

1. Switching Transistor Handbook,chapter five - Transient Characteristicsof Transis tors . Motoro la Inc . ,

Technical Editor William D. Roehr.

2 . Pu lse , Dig i ta l and Switch ingWaveforms, Chapter 20 - Transient

Switching Characteristics, Millman andTaub. McGraw-Hill Book Company.

3. Semiconductor Device Technology,section 2.2.26 BJT Modelling - PhysicalBJT Models, subsection 2) ChargeControl BJT Model. Malcolm E. Goodge.MacMillan Press.

4. Switching and Linear Power Supply,Power Converter Design. Abraham I.

Pressman. Hayden Press.

Appendix C

Characterisation Charts showing typical stored charge as a function of load currentand bias level.

0.1 1 10

IC - Collector Current (A)

Stored Charge v IC for ZTX869

1n

10n

100n

StoredCha

rge(C) IC/IB=50

IC/IB=100

IC/IB=150

ZTX869

IC/IB=150IC/IB=100

IC/IB=50

ZTX949

StoredCha

rge(C)

10n

1n

0.1

100p

1 10IC - Collector Current (A)


StoredCharge(C)

100n

100p0.1 1 10



IC/IB=150IC/IB=200

IC/IB=100IC/IB=50

ZTX968

1n

10n

100n

1n

0.1IC - Collector Current (A)

Stored Charge v IC for ZTX1048A

1 10

StoredCharge(C)

ZTX1048A

IC/IB=50

IC/IB=100

IC/IB=150

IC/IB=20010n

IC/IB=200


16/16


0.1 1 10

IC/IB=50

StoredCh

arge(C)

100n

100p



10n

1n

100p

10n

1n

100p

100u 1m 10m

IB - Base Current (A)

Min. Trigger Charge v IB for Small Signaland Switching Transistors

10p

10n

Minim

umT

riggerCharge(C)

ZTX310100p

1n

abc

cba

StoredCharge(C)

10n

1n

100p

10p

100u 1m 10m

IB - Base Current (A)

Stored Charge v IB for Small Signal and

Switching Transistors

ZTX310

abc

ba

c

1n

10n IC/IB=100IC/IB=150

IC/IB=200

ZTX618

StoredCh

arge(C)

ZTX717 IC/IB=50

IC/IB=100

IC/IB=150

IC/IB=200



0.1 1 10

StoredCharge(C)

0.1 1 10

IC/IB=200

IC/IB=150

IC/IB=100

IC/IB=50

ZTX718



100n

10n

1n

ZTX849

StoredCharge(C)



IC/IB=150

IC/IB=100

IC/IB=50

0.1 1 10

ZTX500ZTX300ZTX107

ZTX510

ZTX107ZTX300ZTX500

ZTX510

Forced gains (IC/IB) of a:)10, b)20, C) 40.

No significant difference observed for the different forced gains for the ZTX310 series

AN22 - 16

Appendix C (continued)

fast turn off

Documents