design of a dual band bandpass filter having
TRANSCRIPT
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P a g e | 1
1 | P a g e
DESIGN OF A DUAL BAND BANDPASS FILTER HAVING CUTOFF FREQUENCY 2.4 GHZ & 5.4 GHZ
A PROJECT REPORT A compact DBBPF of 2.4/5.4 GHz cutoff frequency for WLAN application with good frequency selectivity and independently controllable bandwidth for each passband is proposed using an interdigital capacitor loop step impedance resonator (ICLSIR) and short-circuited stub loaded resonator (SSLR)
2012
ANKITSINGHAL (11533003) ASHWINI SAWANT(11533004)
MTECH I YEAR (RF & MICROWAVE ENGG.) Indian Institute of Technology, Roorkee
4/26/2012
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DESIGN OF A DUAL BAND BANDPASS FILTER HAVING
CUTOFF FREQUENCY 2.4 GHZ & 5.4 GHZ
A Project
Submitted in partial fulfllment of the
Coursework of subject
RF RECEIVER CIRCUIT DESIGN
of
MASTER OF TECHNOLOGY
in
ELECTRONICS & COMMUNICATION ENGINEERING
(Specialization in RF & Microwave Engineering)
By
ANKIT SINGHAL(11533003)
ASHWINI SAWANT(11533004)
Department of Electronics & Computer Engineering
Indian Institute of Technology Roorkee
Roorkee 247667, Uttarakhand, India
MAY 2012
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ABSTRACT
As the demand for the wireless application products increases, low cost
and highly integration have become the major issues in radio frequency
(RF) circuit design. To meet various application requirements, the new
WLAN standards such as IEEE 802.11a and IEEE 802.11g
specifications have been developed. However, the rapid growth of the
WLAN standards has brought significant challenges, especially in the
design of the RF front-ends. It is critical for RF system that can offer
multistandard operations and maintain a competitive hardware cost. In
recent years, dual-band filters have been proposed and exploited
extensively as a key circuit block in dual-band wireless communication
systems.
a compact DBBPF for 2.4/5.4 GHz WLAN application with good
frequency selectivity and independently controllable bandwidth for each
passband is proposed using an interdigital capacitor loop step
impedance resonator (ICLSIR) and short-circuited stub loaded resonator
(SSLR) .This DBBPF configuration is similar to but fewer resonators are
used.
ADS is a sophisticated circuit simulator. ADS can run on a variety
of operating systems. The current version runs on a UNIX machine and
windows XP. We have used ADS for simulating the desired DUAL BAND
BANDPASS FILTER design . Momentum based simulation is performed
by us which is based upon the method of moment technique.
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Contents INTRODUCTION:- .................................................................................................................. 6
LITERATURE SURVEY:- .......................................................................................................... 8
MICROWAVE FILTERS:- ....................................................................................................... 10
PROTOTYPE FILTER.......................................................................................................... 11
BUTTERWORTH LOW PASS PROTOTYPE FILTER:.......................................................... 11
BUTTERWORTH (MAXIMALLY FLAT) RESPONSE .......................................................... 12
CHEBYSHEV LOW PASS PROTOTYPE FILTERS: ............................................................. 13
CHEBYSHEV OR EQUAL RIPPLE RESPONSE ................................................................... 14
IMPEDANCE AND FREQUENCY SCALING ......................................................................... 15
FILTER TRANSFORMATIONS: ........................................................................................... 16
RESONANT CIRCUITS ....................................................................................................... 18
QUALITY FACTOR ............................................................................................................ 19
RF AND MICROWAVE RESONATORS ............................................................................... 19
RESONATORS:- ................................................................................................................ 21
A. ELECTRIC COUPLING ............................................................................................. 21
B. MAGNETIC COUPLING ............................................................................................. 23
HAIRPIN RESONATOR:- ................................................................................................... 26
DUAL BAND FILTER DESIGNING:- .................................................................................... 28
LAYOUT:- ......................................................................................................................... 35
SIMULATED RESULT:- ...................................................................................................... 36
CONCLUSION:.................................................................................................................. 37
REFERENCES:- .................................................................................................................. 37
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Figure 1 Proposed Dual Band Filter Based on OLRRs ............................................................ 8 Figure 2 prototype of filter ................................................................................................. 11 Figure 3 butterworth response ........................................................................................... 12 Figure 4 chebyshev response .............................................................................................. 14 Figure 5 Low pass to high pass transformation ................................................................. 16 Figure 6 : Low pass to band pass transformation ............................................................... 17 Figure 7 Low pass to band stop transformation ................................................................. 17 Figure 8 :The perfect filter response ................................................................................... 18 Figure 9 A practical filter response ..................................................................................... 18 Figure 10 Typical I/O coupling structures for coupled resonator filters. (a) Tapped-line coupling (b) coupled line coupling. .................................................................................... 21 Figure 11 Asynchronously tuned coupled resonator circuits with electric coupling. .......... 22 Figure 12 Typical coupling structures of coupled resonators with electric coupling .......... 22 Figure 13 Asynchronously tuned coupled resonator circuits with magnetic coupling. ....... 24 Figure 14 Typical coupling structures of coupled resonators with magnetic coupling. ..... 24 Figure 15 Equivalent circuits of λg/2 hairpin resonator at resonance frequency a) Odd mode b) Even mode............................................................................................................ 27 Figure 16 Equivalent circuits of ICLSIR at resonance frequency a) Odd mode b) Even mode ........................................................................................................................................... 29 Figure 17 Simulated |S21| with different l3 and Cic .......................................................... 30 Figure 18 Equivalent odd and even mode resonators. (a) Odd mode and (b) even mode . 31 Figure 19 Schematic diagram of proposed DBBPF .............................................................. 33 Figure 20 Simulated frequency responses .......................................................................... 34 Figure 21 layout .................................................................................................................. 35 Figure 22 variation of S21 with respect to frequency ......................................................... 36
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INTRODUCTION:- The continual evolution of telecommunication systems leads to increase the number of
frequency bands they use. In this way, each constitutive element (antennas, filters, amplifiers)
of such a system needs to be designed in order to present a multiband behaviour. low cost and
highly integration have become the major issues in radio frequency (RF) circuit design. To
meet various application requirements, the new WLAN standards such as IEEE 802.11a and
IEEE 802.11g specifications have been developed. In recent years, dual-band filters have
been proposed and exploited extensively as a key circuit block in dual-band wireless
communication systems.
Microstrip filter is a key component rejecting unwanted signals in radio wireless
communication systems. Recent microwave communication system has presented new
challenges to produce microstrip dual band filters with low cost, light weight and simple
design. The dual-band bandpass filter finds variety of applications such as global system for
mobile communications GSM, wireless local area networks and wireless code division
multiple access networks.
There are different approaches for designing of dual band bandpass filters.
First one is the direct methods for a DBBPF. In this two filters operating at different
frequency bands are combined with common input/output ports. filter is easy to design
because the two passbands can be designed independently using traditional single-band filter
design procedure. However, it requires a large circuit board. But, large size and complexity in
design have become the key problematic issues in this approach.
In second approach transmission zeros are used to separate the single-passband filter
into a multi-passband. Therefore a wideband filter is divided into a dual-band filter with two
narrower passbands. The major advantage of this approach is that the stopband response is
properly considered. Such a dualband filter can be used when the two passbands are
relatively close to each other because a wide stopband cannot easily be achieved. It generally
occupies a large area of circuit board because it has a large number of resonators. these
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methods increase the complexity of the filter design process and the overall size of a filter
circuit and getting a good rejection between the two bandpasses when they are very close is
difficult.
In third approach dual-mode resonator, which has a controllable first-harmonic resonant
frequency, is used to create the second passband. This type of DBBPF can only determine the
central frequencies of two passbands. It uses modified resonators, including stepped
impedance resonators (SIRs) stub-loaded open-loop resonators or split-ring resonators.
Among these, SIRs may be the most popular. For this type
of dual-band filter, the total length and impedance ratio of the resonators are used to control
the first harmonic frequency while the fundamental frequency is fixed; thus the central
frequencies of the two bands are controlled simultaneously. The major advantage of this
method is the
small size of the filter. In addition, they have difficulty regulating the bandwidths of the
passbands to achieve various specifications for each passband. However, these filters have
limited controllable bandwidth resulting from the common coupling path and require a
complex design procedure.
The design of a DBBPF with high-performance characteristics is still challenging.
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LITERATURE SURVEY:-
In [1] the novel microstrip dual passband filter based on open-loop ring resonators
(OLRRs) shown in Fig. 1 is proposed. The filter is comprised of two uniform microstrip lines
and two pairs of OLLRs. The OLRRs are placed between two microstrip lines and each has a
perimeter about a half-wavelength. Each of the fold half-wavelength ring resonators has the
maximum electric field density near the open ends of the line, and the maximum magnetic
field density around the center valley of the line at resonance. The proposed dual-band filter
using respective coupling mechanism has advantages of low insertion loss, wide tunable
range of either passband, transmission zeros, easy to design, and without external impedance-
matching block. Finally,
two high performance filters with dual-bands at 2.4/5.2 GHz and
2.4/5.7 GHz are optimally designed and fabricated.
Figure 1 Proposed Dual Band Filter Based on OLRRs
In [2] proposes an original topology derived from Dual Behaviour Resonator (DBR)
filter .This approach allows the direct design of dual-band resonator. In fact, DBR is based on
the parallel association of two bandstop structures. The electrical response of the resulting
structure is composed of one bandpass between the two transmission zeros associated to their
bandstop structures. In this way, as shown in
Fig. I, a simple solution to create two bandpasses consists in adding another bandstop
structure to the basic DBR. this concept can be applied to realize multi-band filters through
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stub addition. However, the increasing complexity of the junctions discontinuities requires
specific investigations.
Wu utilised two sets of resonators with different resonance frequencies, except the
common taped input/output resonator which has stepped impedance. This structure has two
sets of control parameters.[3]
A new miniaturized micro strip filter using 3 stepped impedance resonators and
micro strip tapped feed line is designed for dual band applications at the frequencies 2.5 and
6 GHz . The microstrip line is loaded with stepped impedance hairpin resonators through
coupling lines . To construct band pass filter parallel and series resonance characteristics of
stepped impedance hairpin resonator is utilized. The stepped impedance resonator acts as a
loading impedance and creates two pass bands. Three attenuation poles are created at 0.5, 3.8
and 6.8 GHz respectively. The designed dual band bandpass filter achieves the insertion loss
less than 1dB and the return loss is -17dB and -25 dB at 2.5 and 6 GHz respectively .This
proposed filter structure is suitable for satellite, mobile and other wireless communication
systems.[4]
In this paper[5], a novel design of a dual-band bandpass filter (BPF) with a broad
bandwidth and low insertion loss is presented. The proposed BPF consists of short-stub
composite right/left handed transmission lines and dual-band admittance inverters. Both
theory and experiment have been presented in order to validate the proposed structure. The
proposed dual-band BPF has the 3-dB fractional bandwidths (FBWs) of 50.24% and 20.20%
at center frequencies of 2.40 GHz and 5.20 GHz, respectively, useful for wireless local area
networks applications. From the measured performance, the dualband BPF has the low
insertion losses of 0.28 dB and 0.46 dB for each passband and high isolation of 39.9 dB in
between two passbands.
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MICROWAVE FILTERS:- Periodic structures generally exhibit passband and stopband characteristics in various bands
of wave number determined by the nature of the structure. This was originally studied in the
case of waves in crystalline lattice structures, but the results are more general. The presence
or absence of propagating wave can be determined by inspection of the k-β or ω-β diagram.
For our purposes it's enough to know the generality that periodic structures give rise to bands
that are passed and bands that are stopped.
To construct specific filters, we'd like to be able to relate the desired frequency characteristics
to the parameters of the filter structure. The general synthesis of filters proceeds from
tabulated low-pass prototypes. Ideally, we can relate the distributed parameters to the
corresponding parameters of lumped element prototypes. As we will see, the various forms of
filter passband and stopband can be realized in distributed filters as well as in lumped
element filters.
Over the years, lumped element filters have been developed that are non-minimum phase;
that is, the phase characteristics are not uniquely determined by the amplitude characteristics.
This technique permits the design of filters for communications systems that could not be
constructed using only minimum phase filter concepts. This generally
requires coupling between multiple sections, and can be extended to distributed filters.
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PROTOTYPE FILTER
BUTTERWORTH LOW PASS PROTOTYPE FILTER: For Butterworth or maximally flat lowpass prototype filters having a an insertion loss LAr =
3.01 dB at the cutoff frequency of 1 HZ, the element values as referring to Figure may be
computed by
Figure 2 prototype of filter
For Butter worth response
퐿 (휔 ) = 10log (1 + ) dB
The g notation used in figure signifies roots of an nth order transfer function that governs its
characteristics. These represents the normalized values of filter elements with a cut off
frequency of Ω = 1(rad/s).
Element values for normalized Butterworth low pass prototype filter are:
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g =1
g = 2*sin (( ) ) , k =1, 2 ….n
g =1
To determine the order of a Butterworth low pass prototype filter, the specification that
usually given is the stop band attenuation(퐿 in dB) at Ω= Ω for Ω >1.
n ≥ ( )
BUTTERWORTH (MAXIMALLY FLAT) RESPONSE
Figure 3 butterworth response
Since quantity in square bracket in eqn has 2n zero derivatives at 휔 =0, hence its name is
“maximally flat”.
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CHEBYSHEV LOW PASS PROTOTYPE FILTERS: For chebyshev low pass prototype filters having a transfer function given in
ripple constant = ε = 10 − 1
Where 퐿 is the pass band ripple in dB.
The element values of the two port network of may be computed using the following
formulas:
g =1.0
g = sin( )
g = ( ) ∗ ( )
(( ) ) For i = 2, 3...n
g = 1.0 For n odd
coth (β) For n even
β = ln[coth.
]
γ = sinh( β )
for the required pass band ripple ,minimum stop band attenuation the order of the chebyshev
lowpass filter is found by
n ≥
.
.
Ω
Where L is the stop band attenuation at Ω
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For ‘n’ odd, we have equal source and load impedances, but for ‘n’ even we have unequal
source and load impedances. If design leads to ‘n’=even and we need equal source and load
impedances increase either ‘n’ by one or put a impedance matching network at the load end.
CHEBYSHEV OR EQUAL RIPPLE RESPONSE
Figure 4 chebyshev response
L Is the maximum attenuation (dB) in the passband
= Equal ripple band edge
For 휔 < 휔
L (휔 ) =10log (1 + ϵ cos (n cos ( )) and
휔 > 휔
L (휔 ) =10log (1 + ϵ cosh (n cos ( ))
For both maximally flat as well as chebyshev type response, “n” is the order or number of
reactive elements present in the circuit.
For a lowpass chebyshev response, if n=even, there are ( ) frequencies where L =0 while if
n= odd there will be ( ) where L = 0
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15
IMPEDANCE AND FREQUENCY SCALING Impedance scaling:
In the prototype design, the source and load resistances are unity .A source resistance of R
can be obtained by multiplying the impedances of the prototype design by R .Let prime
denotes impedance scaled quantities. New filter component values are given by
L =R L
C =
R = R
R = R R
L, C and R are the component values for the original prototype filter.
Frequency scaling:
For changing the cutoff frequency of a low pass prototype from unity to Ω requires the
scaling of frequency by the factor of 1ωc
which is accomplished by replacing ω by .
Thus new elements values are obtained by applying the substitution of ω to the
series reactances and shunt susceptance of the prototype filter.
Thus
jX = j L푘 = jωL
jB = j 퐶푘 = jωC
When both impedance and frequency scaling is done then the new element values then
obtained are
L =
C =
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FILTER TRANSFORMATIONS:
Low pass to high pass transformation:
The frequency substitution where ω can be used to convert a low pass response
to a high pass response .so the basic elements of the low pass elements like inductor is
converted to capacitor and capacitor is converted into inductor in high pass filter
transformation.
Figure 5 Low pass to high pass transformation
Low pass to band pass transformation:
The frequency substitution where ω ( − ) = ∆ ( − )
Where ∆ is fractional bandwidth of the pass band ω and ω denote the edges of the pass
band and the centre frequency ω =√ω1 ∗ ω2
In this transformation series inductor of low pass filter is converted to series LC circuit and
shunt capacitor of low pass filter is converted to parallel LC circuit
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Figure 6 : Low pass to band pass transformation
Low pass to band stop transformation:
The frequency substitution where ω ∆( − ) is the general transformation
for low pass to band stop transformation .In this transformation series inductor is replaced
by parallel LC circuit and shunt C is replaced by series LC circuit
Figure 7 Low pass to band stop transformation
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RESONANT CIRCUITS Resonant circuits are used in practically every transmitter , receiver to selectively pass a
certain frequency or group of frequencies from a source to a load while attenuating all other
frequencies outside of this passband.The perfect resonant circuit pass band would appear as
Figure 8 :The perfect filter response
Figure 9 A practical filter response
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QUALITY FACTOR The quality factor of a resonator (Q) is a figure of merit that determines the energy
dissipated by a resonant circuit. The general definition for the Q-factor is given by,
Q = ω
At Resonance Q- factor is
Q =ω Wm+WePloss
= ω
Where ω is the resonant frequency and W , W are the average Electric and Magnetic
energies stored in the resonant structures and P is the power loss. At the resonant
frequency both the Magnetic and Electric energies are equal.
In practice, for a resonator to be used in a system, it has to be loaded by external Circuitry.
The losses associated with the external circuitry add to the losses in the resonant structure and
affects the Q -factor measurement of the resonant structure. In Order to measure the Q-factor
of the resonant structure exactly, the Q -factor associated with the external circuitry (Q )
has to be known. The Q-factor of the resonant Structure and the external circuitry is called
the loaded Q-factor (Q ). The Q-factor associated with the resonant structure alone is called
the unloaded Q-factor (Q ) of the structure,Q , Q and Q are related as
= +
The unloaded Q-factor for the end-coupled and tapped resonator filters are approximately
140-150. This reasonably high value of Q factor is a result of low loss in the filter and
justifies the use of microstrip designs for filters in front-end receiver electronics. The loss can
be further decreased by housing the filter in a metal casing such that radiation losses are
minimized thus increasing unloaded Q-factor of the filter. This is especially important in
microstrip circuits since they tend to have high radiation losses due to fringing fields at the
edges of the microstrip line.
RF AND MICROWAVE RESONATORS RF and microwave resonators are lumped element networks or distributed circuit structures
that exhibit minimum or maximum real impedance at a single frequency or at multiple
frequencies. The resonant frequency f is the frequency at which the input impedance or
admittance is real. The resonant frequency may be further defined in terms of series or shunt
mode of resonance. The series mode is associated with small values of input resistance at the
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resonant frequency, while the shunt mode is associated with large values of resistance at the
resonant frequency. Some typical lumped and distributed resonators are shown in
Resonators may be characterized by their unloaded quality factor Q which is the ratio of the
energy stored to the energy dissipated per cycle of the resonant frequency .Resonators are
also characterized with respect to their reactance (α) or susceptance (β) slope
parameters,which are defined respectively as
α = ( ) at ω = ω and β = ( ) at ω = ω
These are important resonator parameters because they influency Q and the coupling factor
between resonators in multiple resonator filters. The following table provides the reactance
and susceptance slope parameters of some common lumped element and distributed
resonators.
Resonator type Reactance slope (α) Susceptance slope ( β )
Series LC ω L or ω
Shunt LC ω C or ω
Shunt π 2ω C
Line (short) π푍2
Line (open) πY2
Line (short + C) (cotθ + θ csc θ )
Line (short) πY4
Q = may also be defined in terms of the reactance or susceptance slope parameters as
Q = α = βR where R = Resonator series resistance ,R = Resonator shunt
resistance.Together these resistances represent the resonator loss.
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RESONATORS:-
Two typical input/output (I/O) coupling structures for coupled microstrip resonator
filters, namely the tapped line and the coupled line structures, are shown with the microstrip
open-loop resonator, though other types of resonators may be used (see Figure 5). For the
tapped line coupling, usually a 50 ohm feed line is directly tapped onto the I/O resonator, and
the coupling or the external quality factor is controlled by the tapping position t, as indicated
in Figure 5(a). For example, the smaller the t, the closer is the tapped line to a virtual
grounding of the resonator, which results in a weaker coupling or a larger external quality
factor. The coupling of the coupled line stricture in Figure 5(b) can be found from the
coupling gap g and the line width w. Normally, a smaller gap and a narrower line result in a
stronger I/O coupling or a smaller external quality factor of the resonator.
Figure 10 Typical I/O coupling structures for coupled resonator filters. (a) Tapped-line coupling (b) coupled line coupling.
A. ELECTRIC COUPLING
In the case, when we are only concerned with the electric coupling, an equivalent lumped-
element circuit, as shown in Figure 8.7, may be employed to represent the coupled resonators.
The two resonators may resonate at different frequencies of W01 = (L1C1)–1/2 and W02=
(L2C2)–1/2, respectively, and are coupled to each other electrically through mutual
capacitance Cm. For natural resonance of the circuit of Figure 8.7, the condition is
ZL = –ZR ..............(1)
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Figure 11 Asynchronously tuned coupled resonator circuits with electric coupling.
Figure 12 Typical coupling structures of coupled resonators with electric coupling
where ZL and ZR are the input impedances when we look at the left and the right of
reference plane T–T_ of Figure 6. The resonant condition of (1) leads to an
eigenequation
...........(2)
After some manipulations the equation of (2) can be written as
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........................(3)
We note that the equation of (3) is a biquadratic equation having four solutions
or eigenvalues. Among those four, we are only interested in the two positive real
ones that represent the resonant frequencies that are measurable, namely,
................(4) The other two eigenvalues may be seen as their image frequencies. Define a parameter
.........................(5)
where w2 > w1 is assumed. Since w01 = (L1C1)–1/2 and w02 = (L2C2)–1/2 we have by
substitution
.........................(6)
Now, define the electric coupling coefficient
.....................(7)
in accordance with the ratio of the coupled electric energy to the average stored energy,
where the positive sign should be chosen if a positive mutual capacitance Cm is
defined.
B. MAGNETIC COUPLING Shown in Figure 8.8 is a lumped-element circuit model of asynchronously tuned
resonators that are coupled magnetically, denoted by mutual inductance Lm. The two
resonant frequencies of uncoupled resonators are w01 = (L1C1)–1/2 and
w02 = (L2C2)–1/2 respectively. The condition for natural resonance of the circuit of Figure
7 is
YL = –YR. ......................... (8)
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Figure 13 Asynchronously tuned coupled resonator circuits with magnetic coupling.
Figure 14 Typical coupling structures of coupled resonators with magnetic coupling.
where YL and YR are the pair of admittances on the left and the right of reference
plane T–T_ of Figure 8.8. This resonant condition leads to
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.......(9)
The eigenequation (9) can be expanded as
.........................(10)
This biquadratic equation has four eigenvalues, and the two positive real values of interest are
.............................(11)
To extract the magnetic coupling coefficient we define a parameter
...................(12)
Assume w2 > w1 in (8.59) and recall w01 = (L1C1)–1/2 and w02 = (L2C2)–1/2 so that
...............................(13)
Defining the magnetic coupling coefficient as the ratio of the coupled magnetic energy
to the average stored energy, we have
............(14)
The choice of a sign depends on the definition of the mutual inductance, which is
26
normally allowed to be either positive or negative, corresponding to the same or opposite
direction of the two loop currents.
These coupled structures result from different orientations of a pair of open-loop resonators,
which are separated by a spacing s. It is obvious that any coupling in those structures is
proximity coupling, which is, basically, through fringe fields. The nature and the extent of the
fringe fields determine the nature and the strength of the coupling. It can be shown that at
resonance of the fundamental mode, each of the open-loop resonators has the maximum
electric field density at the side with an open gap, and the maximum magnetic field density at
the opposite side. Because the fringe field exhibits an exponentially decaying character
outside the region, the electric fringe field is stronger near the side having the maximum
electric field distribution, whereas the magnetic fringe field is stronger near the side having
the maximum magnetic field distribution. It follows that the electric coupling can be obtained
if the open sides of two coupled resonators are proximately placed, as Figure 6(a) shows, and
the magnetic coupling can be obtained if the sides with the maximum magnetic field of two
coupled resonators are proximately placed, as Figure7(a) shows.
HAIRPIN RESONATOR:- The stepped impedance hairpin resonator has U shaped structure and possesses inner coupled
lines utilizing both open ends for miniaturization. Fig.1 shows the equivalent circuit of λg/2
hairpin resonator at resonance. The hairpin resonator with open ends possesses maximum
electric field distribution near both open circuited ends at resonance. The SIR with internal
coupled lines creates an odd-mode field distribution at fundamental frequency f0. For higher
resonance mode electric field distribution is even. Even and odd mode impedance of parallel
coupled lines are represented by Z0e , Z0o . An SIR with coupled lines can be expressed as a
conventional SIR composed of two single transmission lines. The resonance condition
equations are
27
Figure 15 Equivalent circuits of λg/2 hairpin resonator at resonance frequency a) Odd mode b) Even mode
28
DUAL BAND FILTER DESIGNING:- a compact DBBPF for 2.4/5.2 GHz WLAN application with good frequency
selectivity and independently controllable bandwidth for each passband is proposed using an
interdigital capacitor loop step impedance resonator (ICLSIR) and short-circuited stub
loaded resonator (SSLR) [7]. This DBBPF configuration is similar to [1], but fewer
resonators are used. Five transmission zeros (TZs) can be allocated. Frequency selectivity can
be improved using these TZs.
The proposed dual-mode resonator for lower band of certer frequency 2.4 GHZ is
constructed on a half-wave stepped impedance resonator by adding interdigital capacitor and
a short-circuited stub. This symmetric structure can support two modes, i.e., an even mode
and odd mode. Each resonance condition is analyzed. This proposed dual-mode resonator
generates two additional TZs. Furthermore, with a cross-coupling between the input/output
(I/O) feed lines, one tunable TZ occurs in the lower stopband.
The interdigital capacitor is approximated to simplify the analysis, as shown Figure 9.
Equivalent interdigital capacitances Cic and CP are calculated with EM simulated Y-
parameters of the specific interdigital capacitor. Cic and CP are given by
CP = Im(Y11 + Y21)/x, Cic = Im(Y21)/x, where x is the angular frequency .
29
Figure 16 Equivalent circuits of ICLSIR at resonance frequency a) Odd mode b) Even mode
Two transmission zeros are generated in the stopband, effected by self cross couplings. The
inserted capacitance can be considered as electric couplings (E) and the applied shunt stub
can be considered as a magnetic coupling (M). The position of the first TZ is determined as
follows. In the case, where M is dominant, i.e., fodd is higher than feven, the TZ would be on
the upper band. In the case, where E is dominant, i.e., fodd is lower than feven, the TZ would
be on the lower band. The second TZ is generated by harmonic effects. If only M or E exists,
i.e., Cic is 0 pF or l3 is 0 mm, only one TZ is generated, as shown in Figure 2.
30
Figure 17 Simulated |S21| with different l3 and Cic
(l1 =5.85 mm, w1 = 0.2 mm, l2=2.3 mm, w2= 0.2 mm, w3 = 0.4 mm, and Cp = 0.17 pF): (a) different stub lengths l3 with Cic = 0.24 pF and (b) different series capacitance Cic with l3 =2.5 mm.
31
The loaded capacitor can be converted into equivalent microstrip lengths lc to simplify the
resonator design.
where CL is loaded capacitance, Y2 is admittance of Z2 Consider the lc< , then
where vp is phase velocity of propagation in the stub. The even and odd mode resonators are
reshown, as seen in Figure 11, where lco and lce are equivalent microstrip lengths of Codd and
Ceven, respectively. Both odd and even mode resonators are identical to
the capacitively loaded quarterwave lengths SIR. Let Yodd = 0
Figure 18 Equivalent odd and even mode resonators. (a) Odd mode and (b) even mode
then the odd mode resonance condition can be obtained, as follows
32
and similarly, for the even mode resonator
where h1, h2, h3, hco, and hce refer to the electrical lengths of the sections of lengths
respectively.
Fig. 12 shows the layout of the proposed DBBPF. Two dual-mode resonators are coupled
with common I/O feedlines. The I/O feedlines have small cross-coupling paths to generate
extra transmission zeros. This structure can enable each passband to be independently
designed. Fig. 2 shows the simulated responses of the dual-mode for each passband and
combined DBBPF with the same feedline structure. A dual-mode filter is designed using an
ICLSIR for the 2.4 GHz passband with three transmission zeros. Two transmission
zeros are generated by the ICLSIR and one is generated by crosscouplings between I/O ports
[7]. A dual-mode filter is designed using the SSLR for a 5.2 GHz passband with two
transmission zeros by cross-couplings between I/O ports. The characteristics of each
passband are reflected in the combined DBBPF. The SSLR is little affected by the
ICLSIR but the ICLSIR is not affected by the SSLR. Therefore, the second passband is easily
tuned without effects on the first passband.
33
Figure 19 Schematic diagram of proposed DBBPF
The dimensions (units: mm) of the fabricated filter are chosen as follows:
l1 =2.3,
l2= 3,
l3=5.6,
l4 = 0.9,
l5 = 5.1,
l6 =7.1,
ls1 = 1.8,
ls2 = 0.7,
w1 = 0.2,
w2 =0.2,
w3 = 0.2.
filter is centred at 2.45 and 5.25 GHz, have insertion loss of , 1.4 and , 1.3 dB, and
have return loss of . 14.1 and . 22.3 dB, respectively. In addition, the filter has five
transmission zeros at 1.18 GHz with 76.5 dB rejection, 2.85 GHz with 49.5 dB rejection, 4.42
GHz with 47 dB rejection, 6.13 GHz with 38 dB rejection and 9.49 GHz with 44.2 dB
rejection, respectively. These transmission zeros improve the rejection level in the stopband
and much improve passband selectivity.
34
Figure 20 Simulated frequency responses
35
LAYOUT:-
Figure 21 layout
36
SIMULATED RESULT:-
Figure 22 variation of S21 with respect to frequency
1 2 3 4 50 6
-70
-60
-50
-40
-30
-20
-10
-80
0
freq, GHz
dB(S(2,1))
37
CONCLUSION:
A novel compact DBBPF is proposed with controllable bandwidth. It operates at
2.4/5.2 GHz with two dual-mode resonators. The proposed DBBPF filter also has five
transmission zeros.
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