ultra-wideband multifunctional communications/radar system

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 55, NO. 7, JULY 2007 1431

Ultra-Wideband MultifunctionalCommunications/Radar System

George N. Saddik, Student Member, IEEE, Rahul S. Singh, Member, IEEE, and Elliott R. Brown, Fellow, IEEE

Abstract—We have designed, simulated, fabricated, and testedan ultra-wideband (UWB) multifunctional communication andradar system utilizing a single shared transmitting antenna aper-ture. Two surface acoustic wave bandpass chirp filters were used tomodulate the radar and communications pulses, generating linearfrequency modulation waveforms with opposite slope factors. Thesystem operates at a center frequency of 750 MHz with 500 MHzof instantaneous bandwidth. The measured range resolution is63 cm (25 in) using targets with a radar cross section of 2.7 m2.The probability of detection was measured to be 99%, and theprobability of false alarm was 7% with the communication andradar systems operating simultaneously. The bit error rate forsimultaneous communication at 1 Mb/s, and radar at 150 kHzpulse repetition frequency and 1.5-ns pulsewidth is 2 3. OurUWB multifunctional system demonstrates the ability to simulta-neously interrogate the environment and communicate througha shared transmitting antenna aperture, while realizing a simplesystem architecture with low output power and not employingtime-division multiplexing.

Index Terms—Chirp, communications, radar, RF.

I. INTRODUCTION

TODAY, multifunctional systems are used in our daily lifefrom personal digital assistants (PDAs) to cell phones.

Among the advantages of having a multifunctional system arelow cost and reduced size. The military has taken advantage ofmultifunctional systems by developing broadband RF aperturesthat are capable of simultaneously operating communication,radar and electronic warfare [1]–[4], and multifunctionalunattended ground sensor networks that can be optimized de-pending on location [5], [6]. To achieve simultaneous operation,we have proposed [7], [8] using linear frequency modulated(LFM) signals of opposite slopes for the communications andradar transmitted pulses. The quasi-orthogonality of oppositeslope LFM signals is being exploited to allow for simultaneouscommunication and radar operation through a shared antennaaperture.

The quasi-orthogonal concept has been adapted from orthog-onal signal design in communication theory. The opposite slopeexpanded LFM radar and communication signals, combined and

Manuscript received September 8, 2006; revised March 23, 2007. This workwas supported by the U.S. Army Research Office under Multiuniversity Re-search Initiative Grant “Multifunctional, Adaptive Radio, Radar and Sensors.”

The authors are with the Electrical and Computer Engineering De-partment, University of California at Santa Barbara, Santa Barbara, CA93016 USA (e-mail: [email protected]; [email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TMTT.2007.900343

transmitted through a shared aperture, are not completely or-thogonal, but are near orthogonal and, hence, affect each otherin their respective receivers. The combined radar and communi-cations signals are inputs to the matched chirp filter in the appro-priate radar and communications receivers; the signal matchedto the filter is compressed and the unmatched signal is furtherexpanded, reduced in amplitude, and is seen as additional noiseat the output of the matched filter. If the signals were completelyorthogonal, then the unmatched signals would not be seen at theoutput. In Section III, we simulate this behavior and show thatthe two signals are not completely orthogonal to each other, andquantify in terms of signal-to-noise ratio (SNR) the quasi-or-thogonal term.

While traditionally LFM signals have been used for radar toimprove range resolution while maintaining adequate averagetransmitting power [9], in 1962 Winkler proposed the use ofLFM signals for analog communications [10]. Soon thereafter,digital communication applications were proposed and imple-mented using LFM signals [11]–[16]. Recently, an indoor chirpspread-spectrum wireless communication system [17], [18]was demonstrated using chirped -differential quadraturephase-shift keying ( -DQPSK) to achieve high data rates.The primary advantage found in using chirp waveforms was themitigation of multipath fading without the use of complicatedsignal processing [19]. Another advantage is using passivesurface acoustic wave (SAW) chirp filters, as done in [17], [18]and in this study; they bring much greater hardware simplicityand reduction in power in the transmitter and receiver designs.

Simulations were conducted previously to establish fea-sibility [7], and the pursuit of hardware realization wasundertaken. This paper reports on the successful design, imple-mentation, and demonstration of simultaneous communicationsand radar operation using the quasi-orthogonality of the up-and down-chirp waveforms, and without the constraint oftime-division multiplexing.

II. ULTRA-WIDEBAND (UWB) SYSTEM OVERVIEW

A. System Architecture

The communication and radar transmitters use LFM sig-nals that are identical in frequency range, but have oppositechirp slopes: positive (up-chirp) and negative (down-chirp),respectively (Fig. 1). Both the expander and compressor chirpfilters used in our implementation have down-chirp slopes; toachieve the up-chirp, the spectrum was inverted using a mixerwith MHz either after the expanderfilter or before the compressor filter followed by a low-passBessel filter to suppress the upper sideband [20]. The centerfrequency of the chirp filters is 750 MHz with a bandwidth of

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1432 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 55, NO. 7, JULY 2007

Fig. 1. Multifunctional communications/radar system block diagram. The transmitter is a down-chirped gated continuous wave (CW) radar signal that is powercombined with an up-chirped BPSK communication signal and transmitted through an RHCP helical antenna. The radar echo is received by an LHCP helical an-tenna that is pulse compressed through a matched filter (up-chirp filter). The communication signal is received by an RHCP helical antenna that is pulse compressedusing a down-chirp filter.

500 MHz, chirp duration of 500 ns, chirp rate of 1 GHz/ s,time bandwidth product of 250, and a fractional bandwidth of0.66, calculated using the UWB definition [21].

UWB signals are further classified based upon the form ofsignal used: step functions with very short rise times, a verynarrow pulse (i.e., impulse), and a single cycle sinusoid pulse.Another classification for UWB signals are those with a band-width inversely proportional to the pulsewidth [21]. Addition-ally, the Federal Communications Commission (FCC) desig-nated frequency bands below 960 MHz for ground penetratingradar, and from 3.1 to 10.6 GHz for other applications [22], suchas communication. UWB signals are uniquely different fromnarrowband signals in that they do not suffer from Rayleighfading as narrowband systems do, but rather multipath disper-sion [23]. Secondly, UWB is known for its penetration capabil-ities. Thirdly, it can create images with less clutter. The secondand third properties are independent of the system architecture;however, the first property is dependent on the system architec-ture [21]. For these reasons, UWB signals have found applica-tions in military and commercial radar, as well as both militaryand commercial communication systems.

The obtained compressive filter has an advantage for thecommunication side of the system; it has a raised cosinewindow, which provides sidelobe suppression, for reductionof intersymbol interference (ISI). However, this advantagecomes at the expense of a broadening of the received com-pressed pulse, which, in terms of radar operation, degrades therange resolution. When is measured on the expansive andcompressive filter, it shows an insertion loss of approximately35 dB. Although this is a major disadvantage of the currentSAW chirp filter, the high center frequency, the large band-width, and time-bandwidth product of 250 (processing gain of24 dB) more than compensates for the insertion loss penalty.

The communication system uses a binary phase-shift keying(BPSK) modulation scheme at a data rate of 1 Mb/s. The mod-ulated data signal is injected into a single-pole single-throw(SPST) switch, which reduces the bit time from 1 s to 10 ns.The reason for the bit time reduction is to make use of the ex-pansive and compressive properties of the chirp filters, the bitwidth must have 1–2 cycles of the 750-MHz center frequencyof the chirp filters. This puts a limit on the minimum and max-imum width that a communication bit can have: 1.3 and 2.6 ns,

SADDIK et al.: UWB MULTIFUNCTIONAL COMMUNICATIONS/RADAR SYSTEM 1433

respectively. Having a bit width greater than the 2.6-ns limit re-duces the ability of the chirp filter to expand and compress thebit. The modulated and bit reduced data signal is fed into the ex-pander down-chirp filter, which is followed by a mixer to invertthe spectrum.

The radar signal is a single sinusoid pulsed waveform witha 750-MHz carrier, pulse duration of 1.5 ns, and a pulse repe-tition frequency (PRF) of 150 kHz. The waveform is amplifiedand fed into the down-chirp filter followed by a gain stage. Thetwo opposite chirp signals, radar and communication, are thensimply combined through a Wilkinson power combiner, ampli-fied ( 27-dBm peak output power), and transmitted through asingle wideband shared antenna aperture. The radar and com-munication receivers employ separate antennas for reception.The radar receive antenna is located adjacent to the radar trans-mitter antenna (bistatic configuration), and the communicationreceiver antenna is located several meters down range from thetransmitter.

The first stage of the communication and radar receiver is again stage followed by a chirp filter (compressor) having the ap-propriate chirp slope. The output from the communication com-pression filter is connected to gain stage and followed by a car-rier recovery circuit. The recovered carrier is used as the localoscillator (LO) into a down-conversion mixer, which demodu-lates the RF signal to baseband. The down converted signal isthen directed into a clock and data recovery (CDR) circuit.

The output from the radar compression filter is fed into a gainstage, which is coupled to a square law detector and the outputis displayed on the Agilent 54846B Infiniium digital samplingoscilloscope.

B. Antenna Design and Link Analysis

Two right-handed circularly polarized (RHCP) and one left-handed circularly polarized (LHCP) antennas were designed forthe system [24]. The helical antenna was chosen for its ease ofconstruction, large fractional bandwidth (from 1.7 to 1) [25],and circular polarization. The designed antenna medium direc-tivity was 10.7 dBi and the fractional bandwidth was matchedto that of the chirp filters, with 7.5 dB of return loss acrossthe operating band with reference to 50 . Finally, circular po-larization is beneficial due to the inherent immunity to multiplereflections and multipath.

With the antennas and system architecture chosen, a simpleanalysis was performed to evaluate the expected performanceusing Friis’s radar equation. The maximum range was calcu-lated to be 19.2 m using the radar equation

(1)

where ( 27 dBm) is the power transmitted, and(10.7 dB) are the gain of the transmitting and receiving an-tennas, respectively, (40 cm) is the free-space wavelength,(2.7 m ) is the radar cross section (RCS) for the targets used inSection IV, and (10 dB) is the minimum detectable signal.The measured maximum range with an output SNR of 10 dBat the receiver is 15.2 m, which is within close agreement ofthe calculated maximum range with the same SNR requirement.

The measured range above is significantly outside the near-fieldrange (i.e., in the far field), thus permitting the use of Friis’sequation. The free-space link loss was calculated to be 87.9 dBusing

(2)

where is the free-space wavelength, is the RCS, and isthe maximum range. The receiver RF gain was calculated to be40.5 dB. The receiver bandwidth is 500 MHz, and the calculatednoise figure is 2 dB. The minimum detectable signal was calcu-lated to be 75 dB with an of 10 dB.

III. SIMULATIONS AND ANALYSIS

To better understand the simultaneous operation of thecommunications and the radar systems, an analysis and sim-ulation was performed on the communications receiver. Theobjective was to quantify the interference of the radar signal onthe communications signal through the matched filter and thequasi-orthogonality between the two signals. The simulationwas performed in MATLAB by filtering a communication andradar signal through the communication matched down-chirpfilter. The analysis was performed to verify the results and toprovide a closed-form expression for each signal through thecommunication’s matched filter.

The communications and radar signals are given as

(3)

(4)

where ( MHz ns) is the modulation rate, ( MHz) is thecenter frequency of operation, and ( ns) is the pulsewidth.The unitary energy communications matched filter is

(5)

where ( ) is the time-bandwidth product. The time-do-main output signals through the matched filter were computedby multiplying the signals and the filter in the frequency domainand then taking the inverse Fourier transform [26]. The resultingmatched filter signals for the communications and radar signalsas inputs is then found to be

(6)

(7)

(8)

is the expected compression waveform of an up-chirpwaveform, while is the output of a down-chirp signalthrough a down-chirp compressor. Using MATLAB to numer-ically evaluate , it is found to be approximately 2 and istreated as a constant to evaluate the performance of the system.


Fig. 2. Spectrum and phase plots of simulated signals. (a) Communicationsup-chirp in the frequency domain. (b) Communications (up) and radar (down)chirp phases.

From the expression, it is observed that the output is not com-pressed, but is rather a down-chirp signal that is twice as long asthe input with the modulation rate halved, and the overall powerhalved. This leads to being able to state a peak power commu-nication SNR for our system due to signal interference from theradar signal

(9)

The transmitted communications up-chirp signal (3) andradar down-chirp signal (4) were convolved with the com-munication down-chirp matched filter (5) in MATLAB. Thissimulated the output of the communications receiver stimu-lated by a matched communication and unmatched radar inputwaveforms. The input waveforms are plotted in Fig. 2, and thesimulated output waveforms of the communication receiver isplotted in Fig. 3. The simulated output plots (solid line) and theevaluated analytic (diamonds) expression (6)–(8) are plottedtogether to show agreement between the simulated results andanalytical expression. Finally, the output plots show for theinput communications signal that the peak output voltage ofthe communications receiver is equal to (15.8 V), andfor the input radar signal, that the peak output voltage of thecommunications receiver is equal to V (the input signalswere 1 ). The peak voltage outputs support the above powerSNR expression (9).

IV. EXPERIMENTAL RESULTS

A. Radar Results

The radar experimental results were collected in a large openparking lot where the interference from other objects was mini-mized. The targets used in our experiment were two rectangular

Fig. 3. Simulated and analytical expression (6)–(8) results are displayed via thesolid line and diamonds, respectively. (a) Expected pulse compression output ofthe matched filter (down-chirp) to the input communication signal (up-chirp).(b) Input radar signal (down-chirp) being further expanded by the communica-tion matched filter (down-chirp), i.e., the signal being expanded twice.

Fig. 4. Measured range resolution of two targets placed 10 m from the trans-mitting and receiving antennas.

sheets of metal measuring 45 cm 40 cm with a correspondingRCS of 2.7 m .


Fig. 5. Measured BER versus PRF in log–log scale with data rate of 2.5 Mb/sand radar pulse 1.5 ns, and exponential curve fit.

Fig. 6. Measured communications eye diagram with: (a) the radar off and(b) the radar on. (a) Communication system operating independently hasvery little noise, which is attributed to the electronics. (b) In comparison, asignificant rise in the noise level due to the radar operating simultaneously, butstill maintaining two distinct voltage levels (i.e., the eye is still open).

As an example of the radar operation (Fig. 4), the two targetswere placed approximately 10 m away from the system and sep-arated from each other in range by 63 cm.

The probability of detection was measured to be 99% withthe communication system off using a logic analyzer. With thecommunication system operating at a data rate of 1 Mb/s, theprobability of detection was 99%, and the probability of falsealarm was 7%.

B. Communications Results

To achieve benchmark results for the system, the bit error rate(BER) of the system was experimentally measured under thefollowing conditions.

1) The transmitter output was connected directly to the re-ceiver input with a 60-dB attenuator, the same attenuationas expected through 30 m of free space.

2) The carrier signal was hardwired from the transmitter tothe receiver down-conversion mixer as the LO.

3) The data clock was hardwired to the receiver re-timing cir-cuit.

The BER was measured by comparing 0.1 Mbits of transmittedbinary data to the received data. The BER for the communica-tions system only (radar was off) was measured to be less than10 at 1 Mb/s, and with both systems operating simultane-ously, the BER was with the data rate at 1 Mb/s, andthe radar operating at a PRF of 150 kHz. Under the same con-ditions, the BER was measured while sweeping the PRF of theradar from 100 to 1000 kHz. Fig. 5 shows on a log–log scalean exponential increase in the BER as the PRF is increased. Tofurther illustrate the simultaneous operation of radar and com-munications, an eye diagram of the communications waveform(with the radar off) at 1 Mb/s and an eye diagram with the radaron with a PRF of 150 kHz are given in Fig. 6. As seen fromFig. 6(a) and (b), the eye diagram closes in the vertical and hor-izontal directions by 6 and 4.7 dB, respectively, which is withinclose agreement of the simulated results discussed above.

V. CONCLUSION

A multifunctional UWB communication and radar systemsharing the same antenna aperture, with a simple architecture,and low power has been implemented and tested. The simula-tions and implemented system have demonstrated the feasibilityof simultaneous operation of radar and communications using ashared transmitting antenna aperture. The range resolution wasdemonstrated at 10 m to be 63 cm and the probability of detec-tion and probability of false alarm were 99% and 7%, respec-tively. The best case BER was measured to be at a datarate of 1 Mb/s, with the radar system on simultaneously and aradar pulsewidth and PRF of 1.5 ns and 150 kHz, respectively.

In future systems, to improve system performance such asmaximum range, range resolution, and the BER, several designchanges to the system hardware and signal processing could bemade. The first significant change to improve system operationwould be to increase the time-bandwidth product by increasingthe bandwidth and/or the dispersion time in the chirp filter; thiswould increase the SNR for the system, as shown in (9). Oneapproach to increasing the time-bandwidth product is to designa SAW chirp filter at a higher frequency with larger bandwidthand dispersion time than the chirp filters used in the system. Theauthors have simulated in [8] a SAW filter using AlN-on-SiC ata 10-GHz frequency and bandwidth near 7.5 GHz. We believethat fabricating a SAW filter at 10 GHz and a bandwidth near7.5 GHz is feasible. It has been shown in the past that SAW fil-ters in the 10-GHz range on lithium niobate can be fabricatedusing electron beam lithography, proper transducer design, andelectrode material [27]. SAW filters utilizing AlN-on-SiC havealso been demonstrated to operate in the 30-GHz range [28].Thus, with the current advancement in electron beam lithog-raphy, the high quality of III–V material available at the Univer-sity of California at Santa Barbara (UCSB), and the proper de-sign of the transducer, an AlN-on-SiC SAW filter in the 10-GHz


range and bandwidth near 7.5 GHz is feasible. Further, we be-lieve that by scaling up higher in frequency ( 40 GHz), the in-sertion loss of the SAW filter will significantly increase since, asthe metal electrodes are reduced in thickness and width, losseswill arise due to skin effects.

In addition, the system operation can be improved by usingUWB antennas specifically designed for the receiver and trans-mitter, thus improving the phase linearity of the system, direc-tivity, and gain. This would improve the dispersion and mis-match loss with respect to the antenna used in the system, inturn improving the maximum range, range resolution, and BER.Finally, improving the various system components, such as thelow-noise amplifier (LNA) sensitivity in the receiver, and in-creasing the transmit power would further improve the systemperformance.

ACKNOWLEDGMENT

The authors wish to thank R. Bernardo, Integrated CircuitSystems Inc., Worcester, MA, for his generous donation of twoexpansive and two compressive matched SAW chirp filters. Theauthors additionally wish to thank Dr. D. Palmer, U.S. Army Re-search Office AMSRD-ARL-RO-EL, Research Triangle Park,NC, for program management.

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IRE, vol. 36, no. 10, pp. 1236–1242, Oct. 1948.[26] C. E. Cook, “Pulse compression—Key to more efficient transmission,”

Proc. IRE, vol. 48, no. 3, pp. 310–316, Mar. 1960.[27] K. Yamanouchi, “Generation, propagation, and attenuation of 10 GHz-

range SAW in LiNbO3,” in IEEE Ultrason. Symp., 1998, pp. 57–62.[28] Y. Takagaki, T. Hesjedal, O. Brandt, and K. H. Ploog, “Sur-

face-acoustic-wave transducers for the extremely-high-frequencyrange using AlN/SiC(0001),” Semiconduct. Sci. Technol., vol. 19, pp.256–259, 2004.

George N. Saddik (S’97) received the Bachelorof Science degree in electrical engineering fromCalifornia State Polytechnic University, Pomona,in 1998, and is currently working toward the M.S.and Ph.D. degrees at the University of California atSanta Barbara (UCSB).

He was a Product Engineer with the Watkins-Johnson Company, Palo Alto, CA, where he wasinvolved with microwave transceivers, and was lateran Application Engineer involved with RF semicon-ductors for commercial applications. His research

interests include design and fabrication of FBAR filters in AlN and BST andtheir integration with GaN-based HEMT technology, and their application tocommunication systems.

Rahul S. Singh (M’06) received the B.Scs. degreein electrical engineering and mathematics fromSouthern Methodist University, Dallas, TX, in1997, and the M.S. and Ph.D. degrees in electricalengineering from the University of California at LosAngeles (UCLA), in 1999 and 2005, respectively.

He is currently a Post-Doctoral Researcher withthe Electrical and Computer Engineering Depart-ment, University of California at Santa Barbara,where he conducts research in RF systems andintegration, biomedical ultrasound including both

hard tissues (dental) and soft tissues, terahertz imaging, and gigasonics. Hisresearch interests also include piezoelectric materials (perovskites and nitrides),transducer design, analog and digital signal processing, radar applications, andsystem research.


Elliott R. Brown (M’92–SM’97–F’00) received thePh.D. degree in applied physics from the CaliforniaInstitute of Technology, Pasadena, in 1985

He is currently a Professor of electrical and com-puter engineering with the Electrical and ComputerEngineering Department, University of Californiaat Santa Barbara (UCSB), where he teaches coursesin solid-state engineering, RF sensors, and terahertzscience. His research concerns the terahertz field inseveral areas, including ultra-low-noise rectifiers,photomixing sources, the terahertz phenomenology

of biomaterials, and terahertz remote sensor and imager design and simulation.

Other areas of his research include multifunctional RF electronics and systemsand biomedical ultrasonic imaging in hard tissue, particularly imaging inhuman teeth in collaboration with the University of California at Los Angeles(UCLA) Dental School. Prior to UCSB, he was a Professor of electricalengineering with UCLA. Prior to that, he was a Program Manager with theElectronics Technology Office, Defense Advanced Research Projects Agency(DARPA), Arlington, VA. He performed his post-doctoral research with theLincoln Laboratory, Massachusetts Institute of Technology (MIT).

Dr. Brown is a member of the American Physical Society. He was the recip-ient of a 1998 Award for Outstanding Achievement presented by the U.S. Officeof the Secretary of Defense.

ultra-wideband multifunctional communications/radar system

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