soft switching power converter

ABB Corporate Research, ABB Semiconductors AG, ABB Industrie AG IGCTs In Soft Switching Power Converters

EPE of 11 Lausanne, 1999

IGCTs in Soft Switching Power Converters

Steffen BernetABB Corporate Research

Speyerer Strasse 4D-69115 Heidelberg

[email protected]

Matthias LüscherABB Semiconductors

Tu 4CH-5300 Turgi

Switzerlandmatthias.luescher

@chind.mail.abb.com

Peter K. SteimerABB Industrie

IA-TCH-5300 Turgi

Switzerlandpeter.steimer

@chind.mail.abb.com

KeywordsSemiconductor devices, Soft switching, ZVS converter, ZCS converter

AbstractThis paper investigates the behavior of Integrated Gate Commutated Thyristors (IGCTs) at soft switching. Initially softswitching Voltage Source Inverters are presented which are promising candidates for high power industrial or tractionapplications (S≥500kVA). A test circuit is derived, which enables an experimental investigation of 51mm (4500V; 650A)reverse conducting IGCTs as Zero Voltage Switch, Zero Current Switch and at hard switching. The occurring IGCT and diodeswitching transients are analyzed and measured switching losses are discussed. Additionally the impact of soft switching on thegate drive is considered. The results achieved enable a first evaluation of the potential of IGCTs at soft switching.

IntroductionThe development of new high power semiconductors such as3.3 to 4.5kV Insulated Gate Bipolar Transistors (IGBTs)(e.g. [1], [2], [3]) and 4.5 to 5.5kV Integrated GateCommutated Thyristors (IGCTs) (e.g. [4], [5], [6]) pushedthe development of snubberless Pulse Width Modulated(PWM) Voltage Source Coverters (VSC) in medium voltageapplications. Depending on the direction of the flow ofenergy these converters possess a passive front end(Structure of topology: diode bridge-dc voltage link-PWMVoltage Source Inverter (VSI)) or an active front end(Structure of topology: PWM Voltage Source Rectifier(VSR)-dc voltage link- PWM VSI) [7], [8]. Meanwhile theseconverters, ranging from 0.5MVA to 10MVA, are becomingprice competitive against conventional three-phase rectifiersand cycloconverters on the basis of conventional thyristors.Advantages of PWM-VSCs with active front end likereduced line harmonics, a better power factor and a highersystem efficiency enable a cost reduction of the system indifferent applications like for instance rolling mills and highvoltage DC transmission.Despite a price reduction of Gate Turn Off thyristors(GTOs) by a factor of two to three over the last five years[9] also conventional GTO Voltage Source Inverters andCurrent Source Inverters (CSI) are increasingly replaced byPWM VSIs on the basis of IGCTs and IGBTs due toexpensive and bulky snubber circuits as well as the complexgate drive of conventional GTOs.A detailed comparison of a (3300V; 1200A) IGBT moduleand (4500V; 1560A/3120A) IGCTs in a 1.14MVA PWM-inverter showed, that the considered IGBT offers interestingfeatures like active control of dv/dt and di/dt, activeclamping, short circuit current limitation, and active

protection [10]. However, in comparison to IGCTsespecially the higher on-state and total losses, a substantiallysmaller utilization of the active silicon area, an open circuitafter destruction, and reliability concerns aredisadvantageous characteristics of currently available highvoltage IGBT modules.Low total losses at current densities which are about 1.5-2.9times higher than in up to date high voltage IGBT modules,a small part count of the Gate Commutated Thyristor (GCT),the reliable press pack in a compact mechanical arrangementwhich can be easily assembled enable the design of low cost,compact, reliable, highly efficient, and 100% explosion freeIGCT converters [6], [10]. A further reduction of the size,weight and costs of passive components (e.g. output filters inmedium voltage drives) and an improvement of the staticand dynamic characteristics of medium voltage convertersrequires the increase of the switching frequency which islimited in up to date IGCT converters to about 1kHz by theoccurring switching losses. However, the switchingfrequency of IGCTs can be increased substantially, if IGCTsare used as Zero Voltage Switches (ZVS) and Zero CurrentSwitches (ZCS) in soft switching topologies respectively.Since IGCTs are designed, tested and specified for thesnubberless hard switching operation and the behavior ofIGCTs at soft switching has not been investigated yet, thispaper discusses the behavior of IGCTs as ZVS and ZCS.Initially advantageous soft switching VSIs are presentedwhich are promising candidates for high power industrial ortraction applications (S≥500kVA). A test circuit is derived,which enables an experimental investigation of 51mm(4500V; 650A) reverse conducting (RC) IGCTs in differentsoft switching topologies and at hard switching. Thebehavior of IGCTs as ZVS and ZCS is analyzed anddescribed in detail for different operating ranges and test



conditions. The impact of soft switching on the gate drive isdiscussed. The results achieved enable a first evaluation ofthe potential of IGCTs at soft switching.

Soft Switching Voltage Source Inverters

In the last years a large number of diverse soft switchingthree-phase VSIs, CSIs or matrix converters have beenproposed in the literature (e.g. [12], [13], [14], [15], [16]).All soft switching topologies require additional active and/orpassive components to create conditions of zero voltageswitching and/or zero current switching for the mainswitches. In general additional components increase costsand complexity and reduce reliability. Up to now thesedisadvantages have outweighed the advantages of softswitching converters like lower total losses, a higherutilization of the power semiconductors, reduced size andweight of filters due to an increase of the switchingfrequency and reduced noises in basically all high powerforced commutated converters.The main requirement for an attractive soft switchingconverter is therefore a low additional expense of active andpassive components. Considering high power VSIs both thelimited switching frequency of some kHz even at softswitching and the excellent characteristics of availablecontrol schemes for different loads (e.g. induction andsynchronous machines) require additionally the use of PWMand/or space vector control schemes. The attraction of a softswitching VSI increases essentially, if the VSI limits thesteepness of its voltage slopes to about dv/dt Inv≤1000-1500V/µs avoiding the use of additional output filters toprotect the insulation of the connected interface cables,motors or transformers.Two interesting soft switching VSIs for high powerapplications are depicted in the figures 1 and 2.

Vdc

CC Cr

Lr

Sr

SC

Sl1 Sl3 Sl5

Sl6Sl4Sl2

v03

v02

v01

i03

i02

i01vl1

vl2

vl3

Fig. 1: Circuit configuration of an Actively ClampedQuasi-Resonant DC link Voltage Source

Inverter (ACQRLVSI)

Fig. 1 shows the circuit configuration of an actively clampedquasi-resonant dc link VSI (ACQRLVSI) [16], [17]. Theauxiliary decoupling network oscillates the commutationvoltage of the inverter to zero before each commutation ofthe inverter switches. Thus all commutations of the mainswitches take place at zero voltage and the switching lossesare reduced significantly. The switches of the auxiliarynetwork are operated as ZCS and ZVS respectively. Theadvantages of this soft switching VSI are:

• the possibility to apply PWM- or space vector controlschemes,

• the use of a resonant inductor between inverter and dclink capacitor, which can be used to limit the short circuitcurrent of an IGCT VSI,

• the limitation of the dv/dt at the inverter output and• the posssible extension of the operating principle to

three-level neutral point clamped VSIs (3L-NPC VSI).Disadvantages are:• the fact, that the auxiliary switches have a 6 times higher

switching frequency compared to the main switches sincethe decoupling network is used at every commutation ofthe main switches,

• the resulting high switching losses in the auxiliarynetwork,

• the difficult design of the resonant inductor since theinductor has to be designed for the sum of load andresonant current at stationary operation and the shortcircuit current in the failure mode respectively and

• the slightly increased control complexity. The Auxiliary Resonant Commutated Pole Voltage SourceInverter (ARCPVSI), depicted in Fig. 2, is one of the bestsuited topologies for high power applications (e.g. [11],[22]).

vo1

vo2

vo3

io1

io2

io3

vl1

vl2

vl3

SAS1

VDC/2

VDC/2

SAS2

SAS3

Lr1

Lr2

Lr3

Sl2 Sl4 Sl6

Sl1 Sl3 Sl5

Cr1 Cr3 Cr5

Cr2 Cr4 Cr6

Fig. 2: Circuit configuration of an Auxiliary CommutatedPole Voltage Source Inverter (ARCPVSI)

In contrast to hard switching converters an additionalcommutation unit, consisting of one bi-directional auxiliaryswitch and one resonant inductor, per phase and parallelresonant capacitors across the main switches enable zerovoltage switching of the main switches and zero currentswitching of the auxiliary switches. Advantageouscharacteristics of this inverter are:• the possibility to apply any PWM- or space vector

control scheme,• the decoupled operation of all three inverter phases,• the fact, that the auxiliary switches of one phase operate

at the same switching frequency as the correspondingmain switches of one inverter phase since the auxiliarycommutation unit is not applied, when capacitive(forced) commutations with a negative gradient of powertake place,

• the substantial reduction of switching and total lossescompared to snubberless operating converters,

• the possible limitation of the dv/dt at the inverterterminals and

• the posssible extension of the operating principle tothree-level neutral point clamped VSIs.



Drawbacks are:• the relatively high expense of additional active and

passive components (three bi-directional switches, threeresonant inductors, six resonant capacitors, voltageclamps of the auxiliary branches),

• the varying dv/dt at the converter output,• the increased control complexity and• the necessary change of the protection scheme (e.g. to

fuses) if IGCTs are applied.Since available IGCT data sheets describe the behavior ofIGCTs at snubberless operation only, the investigation ofzero voltage switching and zero current switching IGCTs isa necessary condition to evaluate the potential and risks ofboth soft switching power converters and conventionalsnubber circuits with IGCTs.

Test Circuit for the Investigation of IGCTsas ZVS and ZCS

General Considerations

In contrast to hard switching converters, where the switchescarry out inductive (natural) commutations with a positivegradient of power (initiated by an active turn on transient ofan IGCT) and capacitive (forced) commutations with anegative gradient of power (initiated by an active turn offtransient of an IGCT) alternately, the switches in softswitching converters usually realize continuos inductive(ZCS) or capacitive (ZVS) switching transients [18], [19],[21]. The waveforms of ZVS and ZCS differ from each otherin different soft switching topologies, since the necessarypulsation or polarity change of the instantaneous value of thepower between two successive soft switching transients isusually realized by different means in different circuits.However, the fundamental stress of both ZVS and ZCS isbasically the same in all soft switching power converters[18], [21].The switching behavior of a ZVS is characterized by apassive turn on transient at zero voltage and an active turnoff transient in parallel to a capacitor which limits the dv/dtand therefore the occuring turn off losses. The losses of thezero voltage turn on transient are usually very low. They aredetermined by the space charge modulation of a diodeand/or switch if the gate drive operates properly [20]. Thesubstantial switching losses of the active turn off transientsat a given commutation current, commutation voltage,junction temperature and gate drive are basically determinedby the value of the parallel capacitors – that means the dv/dtacross the turning off switches- and the design of thesemiconductors [18], [20].Typical switching transients of ZCS are the active turn ontransient in series to an inductor which limits the rate ofcurrent rise and therefore the turn on losses, the turn offtransient of an inverse or series connected diode during theinterruption of the reverse recovery current and the behaviorof the active semiconductor, when it is stressed with forwardblocking voltage after it turned off at zero current and zerovoltage [18], [20], [21]. While the losses of the active turnon transients, caused by the space charge modulation, areusually very low, the reverse recovery losses of the inverse

and/or series connected diode and the zero current turn offlosses of the active semiconductor when it takes overforward blocking voltage are substantially higher [18], [20],[21].Considering the fundamental similarity of soft switchingtransients in different topologies it is obviously sufficient toinvestigate the fundamental behavior of a semiconductor asZVS and ZCS in one test circuit. If the adjustable parametersof the test circuit (e.g. commutation voltage, commutationcurrent, resonant elements, hold off time, gate driveconditions) are varied in a sufficient large range, the resultsachieved can also be used for a first evaluation of thepotential of the investigated semiconductors in different softswitching topologies.

Circuit Configuration and Function of the TestCircuit

Lload

Lr

Vdc/2 CS1

CS2

iload

iAS1

Vdc/2CC

DC1 DC2

DC3 DC4

SAS2

S1

Lσ/2

Lσ/2

S2

vAS1

iS1

iS2

iCS1

iCS2

vS2

vS1

Vload

RC

SAS1

Fig. 3: ARCP test circuit for the investigation of IGCTs at softswitching

Fig. 3 shows the test circuit applied to investigate thebehavior of IGCTs at soft switching. The circuit consistsbasically of one phase leg of an ARCPVSI [11], [22]. Themain switches are operated as ZVS and the auxiliary switch,consisting of the series connection of one reverse conductingIGCT and one diode, is operated as ZCS.The capacitive rectifier consisting of the diodes DC1-DC4, thecapacitor CC and the resistor RC operates as voltage clamp toprotect the auxiliary switch from overvoltages caused by theinterruption of the reverse recovery current of the seriesdiode of SAS1 and the polarity change of the voltage acrossthe auxiliary branch. This clamp is distinctly more efficientthan a RC snubber, since the clamp capacitor does not haveto be recharged during a polarity change of the voltageacross the auxiliary switch SAS1. To enable the investigationof the semiconductors at any desired junction temperature,the circuit was designed to operate in single shot operation.



t

t

t

t

t

t

t

t

vS1, iS1

vS2, iS2

vAS1, iAS1

vload, iload

vG1

vG2

vAG1

vAG2Start

iS1 vS1

iS2

vS2

vAS1

iAS1

vload

iloadiTest

1 2 3 4 5 6 7 8 9

active iload=iTest vS≥0

vS≤0 iS≤0

iS≤0

vS≤0 is ≤0

tH

irr

iload

iload+irrVdc/2Zr

Fig. 4: Principle waveforms of the ARCP test circuit

Fig. 4 shows the principle waveforms of the circuitconfiguration. The turn on transient of the switch SAS2 startsthe operation. The load current increases until it reaches thedesired test value (iload=iTest). Than a capacitive commutationis initiated by an active turn off transient of the switch S1.The load current commutates into the capacitors CS1 and CS2

and recharges them linearly. The capacitive commutation iscompleted, when the inverse diode of the switch S2 turns onat zero voltage taking over the load current iload. Fig. 5 showsmeasured waveforms of the capacitive commutation. Thecurrent IS2 oscillates only slightly damped due to a resonancebetween the snubber capacitors and the stray inductances ofthe snubber and the dc link. Simulations have shown, thatthese undesired oscillations, which increase the rms currentof the capacitors CS (CS=CS1=CS2) can be damped by a smallresistance of some 10 mΩ in series to the resonant capacitorsCS , if the stray inductances of the parallel connectedcapacitors and the dc link can be kept small by a propermechanical arrangement. After this commutation the loadcurrent decreases, until the turn on transient of the auxiliaryswitch SAS1 initiates the ARCP commutation by an activeturn on transient. Thus the current iAS1 increases as fast asthe absolute value of the current iS2 decreases until theinverse diode of S2 turns off during the interruption of thereverse recovery current. The initiated oscillation betweenthe resonant inductor Lr and the capacitors CS recharges theresonant capacitors until VS2 reaches the dc link voltage andS1 turns on at zero voltage. Since the voltage across theauxiliary branch has changed its polarity during thisoscillation, the current iAS1 decreases linearly after the turnon transient of S1.

0

1

2

3kV

0

0.4

0.8

1.2kA

0

1

2

3kV

0

0.4

0.8

1.2

kA

2 4 6 8 10 12 16µs

VS1IS1

IS2

VS2

Tcap

Fig. 5: Waveforms of the IGCT-ZVS during the capacitive(forced) commutation

The ARCP commutation is completed, when the switch SAS1

turns off at zero current during the interruption of the reverserecovery current of its series diode.The clamp network across the auxiliary switch limits theoccurring diode reverse blocking voltage. Fig. 6 showsprinciple waveforms of the ARCP commutation.

0

1.0

2.0

3.0kV

0

0

0

1

0

1.0

2.0

3.0kV

0

0

0

1

-2.0

-1.0

0

1.0

2.0kV

-0

0

0

1

1k

70 75 80 85 90 95 100 110µs

VS1 IS1

IS2VS2

IAS1VAS1

tARCP

Fig 6: Waveforms of the IGCT-ZVS and the IGCT-ZCS duringthe ARCP commutation

It should be noticed, that the reverse recovery current of theinverse diode of the switch S2 was always high enough, toenable an oscillation of the voltage VS1 to zero despite theresistive losses of the resonant circuit of the ARCPcommutation. Thus an additional boost current of the switchS2 was not required. After an adjustable time interval anothercapacitive commutation is started by an active turn offtransient of the switch S1. Therefore the switch SAS1 isstressed with forward blocking voltage after the hold offtime tH (Fig. 4). After this capacitive commutation the loadcurrent goes to zero and the single shot test is completed.



Data of the Test Circuit

The test circuit was designed using the reverse conducting51mm IGCTs 5SGX0845F0001 (VDRM=4500V,Vdc link=2700V, ITGQM=650A) as switches S1, S2 and SAS1.The 38mm diode 5SDF0345D0006 (VRRM=4500V,Vdc link=2700V, IF=650A ) served as series diode of theswitch SAS1. The resonant capacitors CS1 and CS2 were lowinductive (Lstray<10nH) metallised polypropylene capacitors(CS: 0.25/ 0.5/1/2 µF) and both load and resonant inductorwere air coils (Lr: 4/ 7.5 µF). The dc link voltage (Vdc:675V, 1350V, 2700V), the load current (Iload: 163A, 325A,488A, 650A, 1000A) and the hold off time tH (3µs≤tH≤65µs)were varied in a wide range to enable the use of themeasured waveforms and losses in different soft switchingtopologies. The junction temperature of the semiconductorswere Tj=25°C and Tj=115°C respectively.

Behavior of IGCTs as Zero VoltageSwitches

Capacitive Commutation

When the load current reaches the test value (iload=iTest) theactive turn off transient of the switch S1 starts theinvestigated capacitive commutation taking place in timeinterval 2 of Fig. 4. The commutation is completed by thezero voltage turn on transient of the inverse diode of theswitch S2. While the external capacities CS1 and CS2 limit thedv/dt across the switches at soft switching, measurements ofsnubberless hard switching transients were carried outwithout the capacities CS1 and CS2 for comparison purposes.

Active turn off transients

Fig. 7 and Fig. 8 show waveforms and losses of active turnoff transients of the considered 51mm 4.5kV IGCTs atsnubberless operation (hard switching, CS=0µF) and at softswitching (CS=0.5µF). Obviously the turn off losses can bereduced by about 50%, if the rate of voltage rise of theturning off switch is limited to about 500V/µs. Themaximum instantaneous power losses are reduced by afactor of 7 from 1.8MW at snubberless switching (Fig. 7) to250kW at soft switching (Fig. 8).

The measured turn off losses as a function of thecommutation current iload for different junction temperaturesare to be seen in the figures 9 and 10. Depending on thevalue of the capacitors CS soft switching transients cause40%-80% fewer switching losses than the correspondingsnubberless turn off transients. The normalized switchinglosses Eoff/CS>0 / Eoff/CS=0 are 5-10% lower at Tj=25°C than atTj=115°C using the same parallel capacitors.

VS1

IS1

0

1.0

2.0

3.0

kV

0

0.4

0.8

1.2

kA

ES1

PS10

0.4

0.8

1.2

1.6

2.0

J

0

0.4

0.8

1.2

1.6

2.0

MW

-2 0 2 4 6 8 10 12 14 18µs

Fig 7: Waveforms (VS1, IS1) and losses (power: PS1; energy: ES1)of a snubberless hard turn off transient of an ICGT (Vdc= 2.7kV; I load = 650A; Tj = 115°C; 51mm 4.5kVRCIGCT;Eoff = 2.1J)

VS1

IS1

-0.5

0

0.5

1.0

1.5

2.0

2.5

3.5

kV

-0.2

0

0.2

0.4

0.6

0.8

1.0

1.4

kA

ES1

PS1

0

0.2

0.4

0.6

0.8

1.0

J

0

100

200

300

400

500

kW

-2 0 2 4 6 8 10 12 14 18µs

Fig 8: Waveforms (VS1, IS1) and losses power: PS1; energy: ES1)of a soft (ZVS) turn off transient of an IGCT (Vdc=2.7kV;Iload=650A; Tj=115°C; CS=0.5µF; 51mm 4.5kV RCIGCT;Eoff=1.05J)



0

0.5

1

1.5

2

2.5

0 100 200 300 400 500 600 700

Iload [A]

Eo

ff [

J]

0uF

0.25uF

0.5uF

1uF

2uF

Fig. 9: Turn off losses of IGCTs as a function of thecommutation current (Vdc=2.7kV; Tj= 25°C; 51mm4.5kV RCIGCT; ZVS and snubberless [CS = 0 µF]operation

0

0.5

1

1.5

2

2.5

0 100 200 300 400 500 600 700

Iload [A]

Eo

ff [

J]

0uF

0.25uF

0.5uF

1uF

2uF

Fig.10: Turn off losses of IGCTs as a function of thecommutation current (Vdc=2.7kV; Tj=115°C; 51mm4.5kV RCIGCT; ZVS and snubberless [Cs = 0µF]operation)

The diagram in Fig. 11 indicates, that even relatively smallparallel capacities (CS≤0.5µF) reduce the switching lossescompared to the snubberless operation substantially. Afurther increase of the capacities (0.5µF≤CS≤2µF), causingextensive commutation durations, leads only to a slightfurther reduction of the switching losses.

0

0.5

1

1.5

2

2.5

0 0.5 1 1.5 2

CS [uF]

Eo

ff [

J]

163A

325A

488A

650A

Fig. 11: Turn off losses of IGCTs as a function of the parallelcapacitors CS=CS1=CS2 (Vdc=2.7kV; Tj=115°C; 51mm4.5kV RCIGCT; ZVS and snubberless [CS=0µF]operation)

Fig. 12 shows the turn off losses as a function of thecommutation voltage.

0

0.5

1

1.5

2

2.5

0 500 1000 1500 2000 2500 3000

Vdc [V]

Eo

ff [

J]

0.0uF

0.5uF

2uF

Fig. 12: Turn off losses of IGCTs as a function of the dc linkvoltage (Iload=650A; Tj=115°C; 51mm 4.5kV RCIGCT;ZVS and snubberless [CS=0µF] operation)

The losses at snubberless operation, where the switch currentis equivalent to the load current during the rise of the switchvoltage, increase clearly faster with increasing voltage thanthe losses at soft switching, where the switch current isequivalent to the tail current during the voltage rise of theswitch voltage (Fig. 7, Fig. 8). If the tail current goes to zerobefore the voltage rise is completed, the switching lossesbecome independent of the commutation voltage (CS=2µF inFig. 12). Fig. 13 shows a turn off transient of a 51mm(4500V/650A) RCIGCT at a current of iload=1000A applyingparallel capacitors of CS1=CS2=1µF (dv/dt=420V/µs).Obviously IGCTs can be operated at substantially highercurrents in ZVS operation compared to the rated current atsnubberless operation, if the gate drive is designed to handlethe increased negative gate current during the active turn offtransient.

VS1

IS1

0

0.5

1.0

1.5

2.0

2.5

3.5

kV

0

0.2

0.4

0.6

0.8

1.0

1.4

kA

2 4 6 8 10 12 14 16 20µs

Fig. 13: ZVS turn off waveforms of an IGCT at 154% rated turnoff current (Vdc=2.7kV; Iload=1kA; Tj=115°C; CS=1µF;51mm 4.5kV; Eoff=1.1J)

Passive turn on transients at zero voltage

Fig. 14 shows, that the turn on losses caused by the spacecharge modulation of the turning on inverse diode of S2 arevery small and basically independent of the resonantcapacitors CS.



0

20

40

60

80

100

120

140

0 100 200 300 400 500 600 700

Iload [A]

Eo

n [m

J]

0.0 uF

0.25 uF

0.5 uF

1.0 uF

2.0 uF

Fig 14: Turn on losses of the IGCT inverse diode as a functionof the commutation current (51mm 4.5kV RCIGCT;Vdc=2.7kV; Tj=115°C; ZVS and snubberless [CS=0µF]operation)

Auxiliary Resonant Commutated PoleCommutation

After time interval 3 with a duration of about 90µs in whichthe diode of the switch S2 reaches its stationary chargedistribution, the investigated ARCP commutation takes placein the time intervals 4, 5 and 6 of Fig. 4.

Passive turn off transients

The inverse diode of S2 turns off during the interruption ofthe reverse recovery current at the end of the time interval 4(Fig. 4, Fig. 6). The dv/dt and therefore the switching lossesof the turning off diode are limited by the occurringoscillation between the resonant inductor Lr and the parallelconnection of the capacitors CS1 and CS2. It can be takenfrom the switching losses depicted in Fig.15, that theswitching losses decrease substantially with increasing valueof the capacitors CS1 and CS2 due to the recombination ofexcessive charge carriers in the diode during the rise of theswitch voltage.

0

50

100

150

200

250

300

350

400

450

500

0 100 200 300 400 500 600 700

Iload [A]

Eo

ff [m

J]

0.25 uF

0.5 uF

1.0 uF

2.0 uF

Fig. 15: Turn off losses of the IGCT inverse diodeas a function of the commutation current (Vdc=2.7kV;Tj=115°C; Lr=7.5µH; 51mm4.5kV RCIGCT; ZVS operation)

Active turn on transients at zero voltage

When the voltage across the switch S1 reaches zero, the gateunit delivers a positive gate current which turns on the IGCT(Fig. 4, Fig. 6) . Thus the occurring losses are caused by thespace charge modulation. Both the losses and the behaviorof the IGCT depend on the gate current, the rate of currentrise (diS1/dt) and the stationary value of the impressed switchcurrent. The measured turn on losses in Fig. 16 indicate, thatthe switching losses are basically independent of the value ofthe capacitors CS.

0

20

40

60

80

100

120

140

160

180

0 100 200 300 400 500 600 700

Iload [A]

Eo

n [m

J]

0.5 uF

1.0 uF

Fig. 16: Turn on losses of IGCTs at zero voltage as a function of thecommutation current.(Vdc=2.7kV; Tj=115°C; 4.5kV RCIGCT; ZVS operation)

Gate Drive Considerations

To determine the requirements of an IGCT gate unit at zerovoltage switching, the influence of the soft turn off transientson the gate charge flowing into the gate unit wasinvestigated. To measure the gate current in the extremelylow inductive gate current path a special coaxial shunt wasdeveloped and inserted in both gate connections. The shuntswere constructed as compact as possible to avoid additionalstray inductance. The figures 17 and 18 show waveforms ofthe turn off transients of the investigated 51mm 4.5kV IGCTincluding the measured gate currents. The evaluation of themeasured gate currents at different load currents andjunction temperatures showed, that the gate charge of turnoff transients is reduced by about 10%-15% at zero voltageturn off transients (CS1=CS2=0.5µF; dv/dt=500V/ µs)compared to the snubberless operation due to the increasedrecombination of charge carriers during the reduced rate ofrise of the switch voltage at zero voltage switching. Thepeak gate current is slightly higher than the anode current tobe turned off at hard and soft switching since the IGCT ishard driven in both cases. The influence of the gate driveconditions (diG/dt, peak value and duration of the pulse ofthe positive gate current) on the IGCTs turning on at zerovoltage has not been investigated yet. However, it isexpected that the turn on gate current (peak value andduration) can be reduced in comparison to the snubberlessoperation.



IGS1

IS1

VS1

-100

0

100

200

300

400

500

600

800

A

-0.5

0.0

0.5

1.0

1.5

2.0

2.5

3.0

4.0

kV

0 1 2 3 4 5 6 7 8 10µsFig. 17: Measured anode current IS1, gate current IGS1and anode-

cathode voltage VS1 of an IGCT at a snubberless turn offtransient (Vdc=2.7kV; Tj=115°C; Iload=650A; 51mm 4.5kVRCIGCT)

IGS1

IS1VS1

-100

0

100

200

300

400

500

600

800

A

-0.5

0

0.5

1.0

1.5

2.0

2.5

3.0

4.0

kV

0 1 2 3 4 5 6 7 8 10µsFig. 18: Measured anode current IS1, gate current IGS1 and anode-

cathode voltage VS1 of an IGCT at a ZVS turn off transient(Vdc=2.7kV; Tj=115°C; Iload=650A; Lr=7.5µH; CS=0.5µF;51mm 4.5kV RCIGCT)

Behavior of IGCTs as Zero Current

-0.4

0

0.4

0.8

1.2

kA

0

0.5

1.0

1.5

kV

75 80 85 90 95 100 105 110 120µs

tH

IAS1

VAS1,IG

tx

IGAS1

IGAS1

IAS1

Fig. 19: Measured anode current IAS1, gate current IGAS1 ZCS (Vdc=2.7kV; Iload=650A; Tj=115°C; tH=12µs;CS=0.25µF; Lr=7.5µH; 51mm 4.5kV RCIGCT)

The principle waveforms of the auxiliary switchconfiguration, consisting of a RCIGCT and a seriesconnected diode, can be taken from Fig. 4 and Fig. 6respectively. The use of the auxiliary switch is only requiredduring the ARCP commutation. Obviously the active turn ontransient of the IGCT, the turn off transient of the seriesdiode during the reverse recovery process and the behaviorof the IGCT when it is stressed with forward blocking

voltage after the end of the hold off time are the importantswitching transients of the considered ZCS. Fig. 19 showsthe corresponding measured waveforms of the IGCT of theauxiliary switch SAS1.

Auxiliary Resonant Commutated PoleCommutation

Active turn on transients

The active turn on transient of the IGCT of SAS1 initiates theARCP commutation. The IGCT turns on at half of the dclink voltage and the rate of the current rise is limited by the

resonant inductor to r

dcAS

L

V

dt

di

⋅=

2

1 . Thus most of the

commutation voltage of the auxiliary switch drops off acrossthe resonant inductor and the turn on losses of both IGCTand series connected diode are very low at moderate rates ofcurrent rise of diAS1/dt = 180-340A/µs (Lr=7.5 / 4µH) atVdc=2700V. The turn on process of the IGCT is basicallyequivalent to that in an up to date IGCT converter and theoccurring low turn on losses (Eon≤100mJ) of the auxiliaryswitch are caused by the space charge modulation of theIGCT and the series diode.

Passive turn off transients

After the current iAS1 reaches zero at the end of time interval6 the series diode of SAS1 turns off taking over reverseblocking voltage during the interruption of the reverserecovery current (Fig. 4, Fig. 6). The turn off losses of thediode are determined by the excessive charge carriers insidethe diode after the zero crossing of the diode current. Thusthe turn off losses depend on the time interval of the ARCPcommutation (sum of time intervals 4,5 and 6):

⋅⋅⋅

⋅⋅⋅⋅+

⋅⋅⋅+

⋅⋅⋅=

S

rR

dcSr

dc

Rr

dc

loadrARCP

C

LI

VCL

V

IL

V

iLt

2ˆ2

arctan22ˆ2

22

2 (1)

where $IR : Peak Reverse Recovery Current of the inverse

diode of S2 during the ARCP-commutation

and the occurring peak current

S

r

dcRloadAS

C

L

VIii

⋅⋅

++=

22

ˆˆ1 (2)

at a given dc link voltage. The diode turn off losses depictedin Fig. 20 show, that the losses just slightly depend on theload current. The losses increase with increasing value of thecapacitors CS due to an increasing peak current îAS1

according to equation 2.



0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

0 100 200 300 400 500 600 700

Iload [A]

Eo

ff [

J]

0.25 uF

0.5 uF

1.0 uF

2.0 uF

Fig. 20: Turn off losses of the series diode of theIGCT-ZCS as a function of the commutationcurrent (Vdc=2.7kV; Tj=115°C; Lr=7.5µH;38mm 4.5kV diode)

Capacitive Commutation

Behavior of the IGCT at a polarity change of theswitch voltage

When the current iAS1 reaches zero at the end of time interval6 and the series diode of SAS1 turns off taking over reverseblocking voltage, the IGCT turns off at zero current and zerovoltage (Fig. 19). However, there are still excessive chargecarriers inside the IGCT when the current iAS1 reaches zero,if the current iAS1 falls with typical rates of some 100A/µs.The reverse recovery current of the series diode flowsthrough the IGCT inverse diode and the IGCT evacuating apart of its excessive storage charge. If the gate is connectedto a negative voltage after the time interval tx after the zerocrossing of the current iAS1 the negative gate currentevacuates both the gate cathode junction as well as themiddle pn junction of the IGCT via the inverse diode [18].Thus the excessive charge carriers of the IGCT after the zerocrossing of the switch current are evacuated byrecombination, the negative gate current and the reverserecovery current of the series diode.

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0 10 20 30 40 50 60 70

tH [us]

Eo

ff [

J]

163A

325A

650A

Fig. 21: ZCS turn off losses of the IGCT–ZCS as afunction of the hold off time (Vdc=2.7kV;Tj=115°C; Lr=7.5µH; CS=0.25µF; 51mm4.5kV RCIGCT)

If there are still excessive charge carriers when the IGCT isstressed with forward blocking voltage at the end of smallhold off times (0µs≤tH ≤35µs), a substantial so called″Forward Recovery Current″ evacuates the remaining chargecarriers during the rise of the switch voltage causingessential switching losses of IGCTs in the ZCS mode (Fig.19).These forward recovery losses, which are proportional to theforward recovery charge, are depicted in Fig. 21 as afunction of the hold off time. Obviously the lossesdrastically decrease with increasing value of the hold offtime due to the occurring recombination of charge carriers.The forward recovery losses are reduced by about 60% at ahold off time of tH=15µs in comparison to the minimum holdoff time of tH=3µs. After about 60µs the excessive chargecarriers are recombined completely, totally avoiding the ZCSturn off losses of IGCTs. Like expected, measurements atvarying time intervals tx (0µs≤tx≤3µs) showed, that the exacttime point of turning on the negative gate voltage does notinfluence the occurring ZCS turn off losses. A parasitic turnon transient of the IGCT could not be observed in the entiremeasured hold off time range, since the majority of theforward recovery charge flows into the gate unit.

Gate Drive Considerations

A substantial simplification of the IGCT gate unit iscertainly one important advantage if IGCTs are operated asZCS. The IGCT gate charge as a function of the hold offtime tH is depicted in Fig. 22.

AS1 turned off tX after zero crossing of current

0

200

400

600

800

1000

0 10 20 30tH [us]

QG [

uA

s]

25°C, tX=1us115°C, tX=1us25°C, tX=0us115°C, tX=0us

Fig. 22: Gate charge of the auxiliary switch IGCT-ZCS as a function of the hold off time(Vdc=2.7kV; Iload=650A; Lr=7.5µH;CS=0.25µF; 51mm 4.5kV RCIGCT)

In comparison to the case, where the negative gate voltage isapplied at the zero crossing of the switch current (tx=0µs) adelay of 1µs of turning on the negative gate voltage reducesthe IGCT gate charge by 15%-20%. It can be taken fromTable I that both the turn off gate charge and the negativepeak gate current of the auxiliary switch (IGCT-ZCS) issubstantially reduced in comparison to the snubber-lessoperation and zero voltage switching. Since the turn ontransient of an IGCT ZCS is basically equivalent to that ofan IGCT in a conventional IGCT converter the required turnon gate current is basically the same in both applications.



TABLE I: Gate charge and peak gate current during turn off transients(Vdc=2700V; Iload=650A; ZVS/ZCS:CS=0.5µF; Lr=7.5µH; tH=5µs; tx=1µs; Tj=115°C)

Snubberlessoperation

IGCT-ZVS IGCT-ZCS

QG1430 µAs(100%)

1280 µAs(89%)

818 µAs(57%)

IGmax740 A

(100%)730 A(99%)

175 A(23%)

Comparison of IGCTs at Zero VoltageSwitching and Zero Current Switching

Table II summarizes the switching losses of the considered51mm 4.5kV RCIGCTs at snubberless operation, zerovoltage switching and zero current switching for areasonable design of the resonant elements. Since the IGCT-ZCS operates only at half the dc link voltage in the ARCPtest circuit, Table II contains a column with the originalmeasured ZCS switching losses with a commutation voltageof VC=1350V and a column in which the ZCS switchinglosses are scaled to a commutation voltage of VC=2700V.

TABLE II: Switching losses of 51mm 4.5kV RCIGCTs (Vdc=2700V;Iload=650A; ZVS/ZCS: CS=0.5µF;Lr=7.5µH; Tj=115°C)

SwitchingTransient

SnubberlessOperation

IGCT-ZVS

IGCT-ZCS(VC=1350V)

IGCT-ZCS(VC=2700V,

scaled)

IGCTturn on

[mJ]

248[100%]

160[65%]

100[40%]

248[100%]

IGCTturn off

[mJ]

2100[100%]

1050[50%]

650 (tH=5µs)[31%]

410 (tH=10µs)[20%]

1300 (tH=5µs)[62%]

820 (tH=10µs)[40%]

Diodeturn off

[mJ]

1490[100%]

340[23%]

670[45%]

1340[90%]

Compared to snubberless operation the switching losses ofIGCTs and diodes can be reduced by about 50% (IGCTs)and 75% (diodes) at zero voltage switching using resonantcapacitors of CS=0.5µF which corresponds to a rate ofvoltage rise of dv/dt =500V/µs across the commutatingswitches. In the original test circuit the IGCT switchinglosses at zero current switching are reduced by about 70%(tH=5µs) to 80% (tH=10µs) and the diode turn off losses aredecreased by about 55%. Considering the scaled ZCSswitching losses at a commutation voltage of VC=2700V theIGCT switching losses are reduced by about 35% (tH=5µs)to 55% (tH=10µs). The turn off losses of the ZCS seriesdiode are in the same range like those of a correspondinginverse diode in a conventional IGCT converter since theturn off transients are very similar in both cases.

Summarizing it should be noticed, that zero voltageswitching enables a reduction of the IGCT switching lossesby 50% at moderate dv/dt’s (CS=0.5µF; dv/dt=500V/µs @Iload=650A) to 80% % at low dv/dt’s (CS=2µF; dv/dt=162.5V/µs @ Iload=650A). While the gate turn off charge isslightly reduced by about 10%-15% the peak turn off gatecurrent at zero voltage switching is equal to that atsnubberless operation. However, it is expected, that the turnon gate current pulse can be reduced at zero voltage turn ontransients compared to the active turn on transients inconventional snubberless IGCT converters. The soft turn offtransients of the IGCT inverse diodes and the substantialreduction of the diode switching losses are additionaladvantages of zero voltage switching.The IGCT switching losses at zero current switching can bereduced by about 35% at small hold off times (tH=5µs) toabout 95% at large hold off times (tH=30µs) at acommutation voltage of VC=2700V. Since the active turn ontransient of an IGCT ZCS is basically equivalent to that ofan IGCT in a conventional IGCT converter, the turn on gatecurrent pulse should be similar in both applications.However, both the peak turn off gate current and the gateturn off charge is substantially reduced at ZCS operationcompared to ZVS or snubberless operation of IGCTs. Therelatively high diode turn off losses and the necessity of aclamp circuit to limit the occurring overvoltages during theinterruption of the reverse recovery current aredisadvantages of the ZCS operation.

Potential of IGCTs at Soft Switching

Assuming an overall reduction of the switching losses byabout 50%-60% in soft switching IGCTs an increase of theswitching frequency by a factor of two is possible incomparison to up to date snubberless IGCT converters. Thusswitching frequencies of fs=1-3kHz are realistic in highpower converters. Alternatively the decreased switchinglosses could also be used to increase the output current of asoft switching IGCT converter in comparison to aconventional IGCT converter using the same devices. If thesoft switching IGCT converter is designed to limit its rate ofvoltage rise at the converter terminals to dv/dt≤1000-1500V/µs an additional output filter, which is usuallyapplied in IGCT converters to protect the insulation of theconnected interface cables and motors from the steepdv/dt’s, can be avoided. Further investigations will show, ifthe increased silicon utilization and the saved output filterovercompensate the costs for the additional expense ofactive and passive components of a soft switching IGCTconverter. The optimization of IGCTs and diodes for softswitching could further increase the potential of softswitching IGCT converters in high power applications.Only minor modifications of a standard IGCT gate unit werenecessary to manufacture and test a 3kHz IGCT gate unit.Both the bill of materials and the production process proof,that the costs of a high switching frequency IGCT gate unitare just slightly higher in comparison to that of a standardIGCT gate unit.



Conclusions

The paper has shown, that the switching losses ofcommercially available 4.5kV IGCTs can be reducedsubstantially in soft switching topologies. Extensivemeasurements at soft switching and snubberless operation inan ARCP test circuit showed a reduction of the IGCTswitching losses by 50% (ZVS operation) to 80% (ZCSoperation) compared to the snubberless operation applying areasonable design of the resonant elements. The diode turnoff losses could be reduced by 75% (ZVS operation) to 55%(ZCS operation) in the same circuit. The experimentalinvestigation of a turn off transient of a 51mm (4500V,650A) IGCT at Vdc=2700Vand Iload=1000A proofed, thatIGCTs can handle essentially increased maximum turn offcurrents at soft switching. Further measurements showed anessential reduction of the gate charge by 40%-50% and ofthe negative peak gate current by about 75% if IGCTs areused as ZCS. Only minor modifications of a standard IGCTgate unit were necessary, to manufacture a low cost 3kHzgate unit.The excellent behavior of IGCTs at soft switching is anencouraging intermediate result for the potential use ofIGCTs in soft switching converters for industrial and tractionapplications.

References

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soft switching power converter

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