single- and dual-polarized tunable slot-ring antennas

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 57, NO. 1, JANUARY 2009 19

Single- and Dual-PolarizedTunable Slot-Ring Antennas

Carson R. White, Member, IEEE, and Gabriel M. Rebeiz, Fellow, IEEE

Abstract—Single- and dual-polarized slot-ring antennas withwideband tuning using varactor diodes have been demonstrated.The single-polarized antenna tunes from 0.95 to 1.8 GHz withbetter than �� dB return loss. Both polarizations of the dual-po-larized antenna tune from 0.93 to 1.6 GHz independently withbetter than �� dB return loss and ��dB port-to-port isolationover most of the tuning range. The capacitance of the varactordiodes varies from 0.45 to 2.5 pF, and the antennas are printedon 70 70 0.787 mm� substrates with � � � �. The dual-po-larized slot-ring antenna can either be made both frequency-and polarization-agile simultaneously, or can operate at twoindependent frequencies on two orthogonal polarizations. To ourknowledge, this is the first dual-polarized tunable antenna withindependent control of both polarizations over a 1.7:1 frequencyrange.

Index Terms—Annular slot antennas, dual-polarized antennas,reconfigurable antennas, slot antennas, slot-ring antennas, tunableantennas.

I. INTRODUCTION

F REQUENCY and polarization agility are necessary formany advanced communication systems. Frequency Hop-

ping Spread Spectrum systems switch between many differentfrequencies at a high rate to provide immunity to jamming,narrow-band interferers, and multi-path fading. Software De-fined Radios can be reconfigured to communicate using manydifferent protocols at different frequencies and polarizations.Cognitive Radios, in which the frequency and data rate are au-tomatically determined depending on the available spectrum atruntime, are currently being explored [1]. Frequency agile sys-tems must be able to receive signals over a large frequencyrange, and therefore, require either wide-band or tunable an-tennas. The instantaneous bandwidth of efficient antennas islimited as they become small with respect to the wavelength [2],and tunable narrow-band antennas can be advantageous if smallefficient antennas are required to cover a large frequency range.In addition, tunable narrow-band antennas provide frequencyselectivity—relaxing the requirements of the receive filters.

Resonant slot antennas are good candidates for frequencytuning because their resonance frequency can be changed easilywith varactors or switches. Peroulis et al. demonstrated a slot an-tenna that can be switched to four different frequency bands over

Manuscript received January 01, 2008; revised June 04, 2008. Current versionpublished March 04, 2009.

C. R. White is with HRL Laboratories, LLC., Malibu, CA 90265 USA(e-mail: [email protected]).

G. M. Rebeiz is with the Department of Electrical and Computer Engineering,University of California, San Diego, La Jolla, CA 92093 USA.

Digital Object Identifier 10.1109/TAP.2008.2009664

a 1.7:1 bandwidth using PIN diodes [3]. Behdad et al. demon-strated a single-polarized dual-band slot antenna where both ofthe bands can be tuned independently, covering the 1.1–1.34 and1.74–2.94 GHz bands [4].

An excellent candidate for a dual-polarized tunable antennais the slot-ring antenna. Slot-ring antennas that are tunable orreconfigurable over a small frequency range have been pre-sented using either capacitive loading [5]–[7] or reconfigurablematching networks [8], and a slot-ring antenna that tunesfrom 2–5 GHz was demonstrated using fixed capacitors [9]. Afixed-frequency dual-polarized slot-ring antenna was demon-strated by Raman et al. [10], and a dual-polarized slot-ringantenna where both polarizations can be tuned independentlyfrom 1.92–4.4 GHz was demonstrated by White et al. usingfixed capacitors [11]. Although single- and dual-polarizedslot-ring antennas with excellent tunability were demonstratedin [9] and [11], true tuning elements and their biasing schemeswere not included.

In this paper, single- and dual-polarized slot-ring antennasare tuned with varactor diodes from 0.95–1.8 GHz and0.93–1.6 GHz, respectively. Each polarization of the dual-po-larized antenna is tuned independently, and therefore, it canoperate at two different frequencies on orthogonal polariza-tions. Furthermore, frequency-agile polarization diversity andagility can be implemented if the two polarizations are tunedto the same frequency. The relevant resonance frequenciesof the capacitively loaded slot-ring antenna are discussed inSection II, followed by the design and measured results of thesingle- and dual-polarized antennas in Sections III and IV,respectively.

II. CAPACITIVELY LOADED SLOT-RING

RESONANCE FREQUENCIES

If the slot width , is much smaller than the wavelength, the slot can be considered as a slot-line transmission line.

The characteristic impedance , propagation constant , andtransverse electric field distribution , can be calculated bymethods described in [12]. The fundamental resonance of theunloaded slot-ring occurs when the circumference of the ring isone guided wavelength. At this frequency, there are two orthog-onal voltage modes and , whose radiation patterns are -and -polarized, respectively. has even and odd symmetrieswith respect to the and planes, respectively, and

is odd and even with respect to the and planes,respectively.

Capacitive loading reduces the resonance frequency of allmodes whose voltage is not zero at the locations of the loadingcapacitors. The loaded resonance frequencies of the slot-ringcan be approximated by assuming that it is lossless and using

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20 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 57, NO. 1, JANUARY 2009

Fig. 1. The slot-ring antenna loaded along the x and y axes and the even- andodd-mode equivalent circuits with respect to the x-z plane when � � � �

� . (a) Capacitively loaded slot-ring and (b) equivalent circuits.

the transverse resonance method, as in [4] for the linear slot an-tenna. Resonance occurs when

(1)

where and are the admittances looking left and right, re-spectively, from an arbitrary point on the resonator.

Consider the slot-ring that is loaded with shunt capacitancesand along the axis, and and along the

axis [Fig. 1(a)]. Symmetry about the and planes ispreserved if and , enablingindependent tuning of and . The important resonancesoccur for even-odd and even-even symmetries.

A. Even-Odd Symmetry

Let the voltage be even with respect to the plane andodd with respect to the plane (the same symmetries as

); the equivalent circuit is shown in Fig. 1(b). The resonancecondition is found by applying (1) at , and is given by

(2)

The fundamental mode, , occurs at a frequency, , when, and therefore, is a tuned version of .

If the voltage is even with respect to the plane and oddwith respect to the plane, then (the tuned version of ),

Fig. 2. Calculated even-odd and even-even resonance frequencies, � and� , when � � � � � � � � �� mm, � � � mm, and the slot-ring isprinted on a 0.787-mm-thick substrate with � � 2.2.

and its resonance frequency, , can be found. The resonancecondition is

(3)

Therefore, if and , there are two resonantmodes, and , that can be tuned independently.

B. Even-Even Symmetry

The even-even modes are dependent on both and , andare found by applying (1) to the even mode circuit in Fig. 1(b).Resonance occurs when , or equivalently

(4)

The resonance frequency , of the lowest order mode ,is higher in frequency than both and . This can be seenfor by the following: occurs when the capacitively ter-minated stubs are at their first series resonance ,but occurs when they are inductive (above the first seriesresonance) and resonate with . The same is shown forby exchanging and . and are compared in Fig. 2for mm.

III. SINGLE-POLARIZED SLOT-RING ANTENNA

A. Design

The single-polarized slot-ring antenna geometry is shownin Fig. 3 with a width of 3 mm. The antenna is capaci-tively loaded along the -axis with varactor diodes (M/A-ComMA46H071-1056, 2.5–0.45 pF from 0–20 V) [13], and thereare also varactor-mounting pads along the -axis so that thesame slot-ring resonator can be used for both the single-and dual-polarized antennas. The antenna is printed on a70 70 0.787 mm Taconic TLY-5 substrate (

at 10 GHz).The tuning characteristics were investigated by simulating the

slot-ring antenna with gap ports along the -axis with IE3D[14] [Fig. 4(a)]. The input impedance of the capacitively-loadedantenna, , was calculated as shown in Fig. 4(b). The realpart of , is plotted in Fig. 5(a) for tuning capacitances

from 0 to 2.5 pF assuming an infinite ground

WHITE AND REBEIZ: SINGLE- AND DUAL-POLARIZED TUNABLE SLOT-RING ANTENNAS 21

Fig. 3. Layout of (a) the single-polarized slot-ring antenna and (b) microstrip-to-slot antenna transition and biasing scheme. All dimensions are in millimeters.

Fig. 4. Calculation of the antenna impedance, � , of the capacitively-loadedslot-ring antenna. � and � are gap ports. (a) Slot-ring geometry and (b) ca-pacitively loaded antenna.

plane and ideal capacitors. The resonant input resistance in-creases from 300 in the unloaded state to more than 3 kwhen loaded with 2.5 pF, and therefore, it is difficult to matchthe antenna at resonance over a large tuning range. However,the slot-ring antenna can be matched directly to the character-istic impedance, , at a frequency, , that is below resonanceby placing a reactive element in series with the antenna. If thepoint on each resistance curve is chosen such that ,the antenna reactance at , must then be cancelled.

Both the substrate size and the varactor series resistanceaffect the impedance matching. for is com-pared in Fig. 5(b) for the following cases: infinite substrate and

, infinite substrate and , finite substrate and, and finite substrate and . In the case of an

infinite substrate and varies as approximately, which can be cancelled with a series capacitor. However,

Fig. 5. Simulated (a) antenna resistance � for different loading capacitancevalues �� and (b) antenna reactance at � � � �� , (where� � �� ) as the loading capacitance changes from 0 to 2.5 pF.

the nonidealities of the varactor change the shape of .At low frequencies, the quality factor of the antenna becomescomparable to the of the varactors, and the required reactanceto match the antenna decreases. Whereas affects atthe low frequencies, the finite substrate has an effect over thewhole frequency range; it has standing wave currents that in-crease the effective size of the antenna, reducing the . In thiscase, the required reactance is nearly constant at the high endof the tuning range, but rises sharply at lower frequencies. Theeffects of both the finite substrate and on cancel outsufficiently for the antenna to be matched to better than dBover the tuning range.

It is convenient to feed the slot-ring antenna with a microstripline, and the analysis of the microstrip-to-slot antenna transitioncan be found in [12]. The microstrip feed structure is shownin detail in Fig. 3(b), and the equivalent circuit of the antennaand transition is shown in Fig. 6. The slot-ring antenna is cou-pled in series with the microstrip line through a transformer, andtherefore

(5)

where is the shunt capacitance of ev-erything on the right-hand side of the slot—two 83- open-cir-cuited stubs and the 50 fF parasitic capacitance of the bias re-sistor, is slightly less than 1, and k is highenough to be neglected. The microstrip line would logicallycross the slot along the x-axis, but this is directly under one ofthe loading varactors. The effect of the varactor’s package on


Fig. 6. Equivalent circuit of the microstrip-fed single-polarized tunable slot-ring antenna.

Fig. 7. Measured and simulated return loss of the single-polarized antenna. Thereference plane is the input port.

the microstrip-to-slot fields is not known, so the 50 microstripline is split into two 100 lines and crosses on either side of thevaractor. This maintains the symmetry about the plane andkeeps the majority of the coupling away from the varactor.

The varactor diode biasing is achieved by placing a voltage,, on the center of the slot-ring antenna. is externally applied

to the RF line with a Bias-Tee, and is connected to the center ofthe slot-ring through a 100 k resistor and a via hole (Fig. 3).

B. Results and Discussion

1) Impedance: The input impedance was measured at dif-ferent bias states with an Agilent E5071B vector network ana-lyzer at dBm source power. Ferrite beads [15] were placedon the measurement cable at the input port to suppress the cou-pling to the cable shield, and the antenna was placed on a foamblock about 1 m from any scatterers. The center frequency variesfrom 0.95–1.8 GHz as varies from 0 to 20 V with dBreturn loss.

Before the varactor diodes were installed on the antenna,their equivalent series capacitance and resistance were mea-sured versus both frequency and bias voltage using an AgilentE4991A impedance analyzer. The return loss was then simu-lated using the measured varactor impedance, and is comparedto the measured return loss in Fig. 7. The measured centerfrequency is slightly higher for every bias state except 20 V.This difference is most likely due to coupling to the feed cablebecause the effect of the cable was not simulated. It will beseen in Section III-B.2 that although ferrite beads were used,the coupling to the cable was significant.

The instantaneous 10-dB bandwidth, , is plotted inFig. 8 as a function of , and is 6% at 1.8 GHz and 1% at

Fig. 8. Measured and simulated bias voltage and instantaneous 10-dB band-width over the tuning range.

Fig. 9. Measured and simulated antenna gain and efficiency of the single-po-larized antenna.

0.95 GHz. Although the antenna (including the substrate) isnot electrically small [2], the decrease in bandwidth can be ex-plained as follows. The antenna is operating in a parallel res-onance mode, and the radiation conductance decreases as thecapacitive loading increases [Fig. 5(a)]. The quality factor, ,of a parallel resonator is

(6)

where , and are the equivalent capacitance, inductance,and conductance. decreases as the antenna is loaded withhigher capacitance because increases and decreases at thesame time.

2) Radiation Patterns and Efficiency: The radiation patternsof the antenna were measured using a Satimo Stargate 32spherical near-field chamber ( dB) [16],and the measured and simulated gain and efficiency are plottedin Fig. 9, where efficiency is defined as the ratio of antennagain to directivity. The measurement cable was coated withabsorbing material to reduce scattering, and ferrite beads wereplaced near the input port to reduce the coupling to the cableshield. Both the measured and simulated efficiencies increasewith frequency because the radiation of the antenna decreaseswith frequency and the of the varactor diodes increases asthe capacitance decreases. The measured efficiency is lowerthan simulated for all frequencies, and the difference is greatestat the low frequencies.


Fig. 10. Co- and cross-polarized antenna gain patterns in the �� and � � �

planes. (a) � � �� GHz (� � � V) and (b) � � �� GHz (� � �� V).

The difference in gain is most likely caused by coupling tothe shield of the measurement cable, which was not includedin the simulation. The substrate is smaller than at all fre-quencies, and therefore, the standing-wave current on the fi-nite ground plane is significant. Ferrite beads reduce the cou-pling from the antenna to the cable shield by presenting a highimpedance to the current that would otherwise flow on the cableshield, but the place where the cable is connected is an open-cir-cuit point of the slot-ring-finite-substrate resonator (as can beseen by symmetry), and therefore, current still flows and theefficiency suffers. Some of the lost power is dissipated in theferrite beads and the resistive coating on the cable, and the restradiates from the cable—causing ripple in the radiation pattern.The beads used in these measurements have an impedance peakat 2.45 GHz, which is consistent with the fact that the differencebetween the measured and simulated efficiencies is smaller atthe higher frequencies. In future designs, the cable should beattached along the -axis, which is a short-circuit point of theslot-ring-finite-substrate resonator and is normal to the -polar-ized radiation, and therefore, will result in less coupling to thecable shield.

The radiation patterns are similar for all tuning states, and themeasured and simulated patterns at 0.95 and 1.8 GHz are shownin Fig. 10. The -plane (H-plane) pattern is nearly constant,and as expected for a slot antenna in a small finite ground plane,there are nulls in the plane (E-plane) in the plane of theantenna. The simulated cross-polarization is zero in theplane (due to symmetry), and better than dB in theplane at all bias states, but the measured cross-polarization is ashigh as dB at 0.95 GHz. The high cross-polarization level,

Fig. 11. Measurement setup for IIP3. The setup for � was the same, exceptthat a single source was used at � , and therefore, no power combiner was used.

Fig. 12. Measured input � and IIP3 in the transmit mode.

as well as the ripple in the -plane pattern, is most likelydue to radiation and scattering from the measurement cable.

3) Nonlinearities: The varactor-tuned slot-ring antenna isnonlinear because the junction capacitance of the varactordiodes depends on the voltage across it

(7)

where is the zero bias junction capacitance, is thebuilt-in potential, and is the power law exponent of thejunction capacitance. The nonlinearities are most severe atthe lower frequencies for two reasons: first, the C-V curve issteeper at lower bias voltages (lower resonance frequencies),and second, the voltage swing across the diodes is larger at thelower resonance frequencies for the same power level (higher

).The 1-dB compression point, , and the input third-order

intercept point, IIP3, were measured in the transmit mode(Fig. 11), and the results are shown in Fig. 12. was mea-sured by feeding the slot-ring antenna with a continuous-wave(CW) source and measuring the signal at one angle with aridge-horn antenna and a spectrum analyzer (Agilent E4407B).For the IIP3 measurement, two CW sources at 100 kHz offsetwere added with a power combiner. It is assumed that theantenna pattern does not change at power levels in and near thelinear region and over a 300 kHz bandwidth.

A considerable contribution to the gain compression of theslot-ring antenna at is reflection loss due to a shift in the reso-nance frequency. This can be seen in the measured return loss at2 V bias with different power levels (Fig. 13). The same effect isobserved for 0 and 1 V bias. The return loss at could not bemeasured for higher bias voltages because the power levels weretoo high for the network analyzer. The resonance frequencydecreases with increased power because as the voltage swingacross the diodes increases, the effective capacitance increases.


Fig. 13. Measured return loss when � � � V at three power levels: thesmall signal region (�� dBm) and the 1 and 3 dB compression points (1.2and 5.4 dBm, respectively).

This suggests that can be improved by increasing the biasvoltage as the power level is increased.

IV. DUAL-POLARIZED SLOT-RING ANTENNA

A. Design

A dual-polarized slot-ring antenna can be made by feedingthe slot-ring resonator along both the and axes as shown inFig. 14. The microstrip lines continue along the and axes toport 1 and port 2, respectively, at the edges of the 70 70 mmsubstrate where SMA connectors are attached. The same slot-ring and substrate is used for both the single- and dual-polarizeddesigns, so the dimensions that are not specified in Fig. 14 canbe found in Fig. 3.

It was shown in Section II that there are two orthogonal res-onant modes, and , that can be tuned independently ifthe antenna is symmetric about both the and axes. The mi-crostrip feeds destroy this symmetry by adding reactance toand , and therefore, microstrip reactive loads were placedopposite the feeds to restore the symmetry. If symmetry werenot restored, port 1 would couple to a mode that is not zero alongthe axis, and therefore, is affected by and . For thesame reason, there would be significant coupling between ports1 and 2. This port-to-port coupling, and the effectiveness of themicrostrip reactive loads in reducing it, were demonstrated in[11], where the isolation at the center frequency was improvedfrom 11 dB to dB with the use of the reactive loads. A sideeffect of the reactive loads is that they reduce the resonance fre-quency for a given capacitance value.

The reactive loads have the same shape as the microstrip-to-slot-antenna transitions except for the width of the end stub,which was tuned using IE3D to result in maximum isolation at

V (Figs. 14(b) and 15). They have a similarfrequency response to the feed transition, and therefore, providewideband decoupling of ports 1 and 2.

Capacitive loading is achieved by placing varactor diodes(M/A-Com MA46H071-1056, 2.5–0.45 pF from 0–20 V) acrossthe slot as shown in Fig. 14. Independent tuning is achieved withthe following conditions:

Fig. 14. Layout of (a) the slot-ring antenna and (b) the microstrip-to-slot an-tenna transition and microstrip reactive loads. All dimensions are in millimeters;dimensions that are not specified are the same as in Fig. 3.

where and are the - and -polarized bias voltages. A150 m gap is cut in the center part of the slot-ring in orderto decouple and , and 22 pF capacitors (AVX ACCU-P,0603, 1.8 GHz SRF) are placed across the gaps to allow the RFcurrent to pass. The bias voltages are applied to the center partof the slot-ring from the RF line through 100 k resistors andvia holes.

B. Results and Discussion

The two-port S-parameters of the dual-polarized slot-ring an-tenna were measured using an Agilent E5071B network ana-lyzer at all combinations of and in .Ferrite beads were placed on the cables at the connections tothe antenna to reduce the coupling to the cable shields, and theantenna was placed on a foam block about 1 m away from anyscatterers. Fig. 15 shows the S-parameters of four different com-binations of and . As is held at 20 V and variesfrom 20 to 0 V, the -polarized center frequency remainsnearly constant and the -polarized center frequency , varies


Fig. 15. Measured S-parameters when (from top to bottom) � � � �

�� V; � � �� V and � � � V; � � �� V and � � � V; and� � � V and � � � V.

from 1.6–0.93 GHz with better than dB isolation. Aschanges, changes accordingly without changing .

It is apparent that there is a second resonance in which ports1 and 2 are moderately coupled. This resonance , occurs ata frequency [see (4)] that is higher than both and ,and is dependent on both and . When is small andis large, is very close to and the impedance match andisolation are degraded.

The S-parameters of all 36 bias-voltage combinations aresummarized in Figs. 16 and 17. Both and the -polarized10-dB impedance bandwidth , are nearly constant as

varies from 0 to 20 V, except for the points where islarge and is small. Fig. 17 shows the minimum port-to-portisolation within centered about . The isolation isbetter than 20 dB over most of the tuning range, and is 14 dBwhen GHz and GHz (the maximumfrequency difference). If and are interchanged, the

-polarized center frequency, bandwidth, and isolation arenearly identical to the -polarized characteristics shown inFigs. 16 and 17, as expected by symmetry.

The -polarized radiation patterns were measured at severalbias voltage combinations. Port 1 was connected to the RFsource through a bazooka balun (centered at either 0.8 or1.6 GHz), and a coaxial cable with ferrite beads was connectedto port 2 in order to set . The -polarized antenna patternswere not measured, but should be nearly identical to the -po-larized patterns due to symmetry.

As expected, the radiation patterns and efficiencies are verysimilar to those of the single-polarized antenna. The measured

Fig. 16. Measured ��-polarized (a) center frequency and (b) 10-dB bandwidthat different � and � .

Fig. 17. Measured port-to-port isolation (in decibels) within� centeredabout � .

and simulated efficiencies at 0.93 GHz are 20% and 25%, re-spectively; at 1.6 GHz, they are 85% and 93% respectively. Theagreement between the measured and simulated efficiencies at1.6 GHz, which is at the center frequency of the balun, suggestthat the measured efficiency at 0.93 GHz would be closer to 25%if a 0.93 GHz balun were used.

The measured radiation patterns at 0.93 and 1.56 GHz areshown in Fig. 18. The -plane (H-plane for port 1) patternsare smooth despite the fact that a coaxial cable was attachedto port 2 during the measurement. This supports the theorythat placing the feed cable of the single-polarized antenna inthe H-plane will improve the efficiency and make the patternsmoother. The simulated cross-polarization levels are below

dB in the principle planes and, even with the scatteringfrom the two cables feeding the antenna at ports 1 and 2, the


Fig. 18. Measured ��-polarized (port 1) radiation patterns of the dual-polarizedantenna at 0.93 GHz (� � �V and � � �V) and 1.56 GHz (� � � �

�� V) in the � � � and � � � planes.

measured cross-polarization levels are at least dB at mostangles at 1.43 (not shown) and 1.56 GHz.

V. CONCLUSION

This paper shows that the slot-ring antenna is an excellentcandidate for both single- and dual-polarized frequency tuning.A single-polarized antenna that tunes from 0.95 GHz withbetter than dB return loss has been demonstrated usingtwo varactor diodes, and a dual-polarized antenna has beendemonstrated whose orthogonal polarizations can be tunedfrom 0.93–1.6 GHz independently using four varactor diodes.The impedance match is maintained by operating at frequen-cies below resonance where the resistance of the antenna is50 , and cancelling the reactance with an open-circuited-stubcapacitor. The use of reactive loads enables dB isolationover most of the tuning range of the dual-polarized antenna.The efficiency and linearity at the lower frequencies can beimproved significantly by using MEMS varactors, and the effi-ciency of the single-polarized antenna can be improved furtherby moving the input port so that the cable is perpendicular tothe electric field.

ACKNOWLEDGMENT

The authors would like to thank Qualcomm, Inc., San Diego,CA, for the use of their antenna pattern measurement system,and M/A-Com, Lowell, MA, for providing the varactor diodes.

REFERENCES

[1] J. Mitola and G. Q. Maguire, “Cognitive radio: Making software ra-dios more personal,” IEEE Personal Commun. Mag., vol. 6, no. 4, pp.13–18, Aug. 1999.

[2] R. C. Hansen, “Fundamental limitations in antennas,” Proc. IEEE, vol.69, no. 2, pp. 170–182, Feb. 1981.

[3] D. Peroulis, K. Sarabandi, and L. P. B. Katehi, “Design of reconfig-urable slot antennas,” IEEE Trans. Antennas Propag., vol. 53, no. 2,pp. 645–654, Feb. 2005.

[4] N. Behdad and K. Sarabandi, “Dual-band reconfigurable antenna witha very wide tunability range,” IEEE Trans. Antennas Propag., vol. 54,no. 2, pp. 409–416, Feb. 2006.

[5] C. Hong, “Small annular slot antenna with capacitor loading,” Electron.Lett., vol. 36, no. 2, pp. 110–111, Jan. 2000.

[6] I. Carrasquillo-Rivera et al., “Tunable and dual-band rectangular slot-ring antenna,” in Proc. IEEE Antennas Propag. Symp., Jun. 2004, vol.4, pp. 4308–4311.

[7] E. Erdil et al., “Reconfigurable cpw-fed dual-frequency rectangularslot antenna using RF MEMS technology,” in Proc. IEEE AntennasPropag. Symp., Jul. 2005, vol. 2A, pp. 392–395.

[8] S. Nikolaou et al., “Pattern and frequency reconfigurable annular slotantenna using PIN diodes,” IEEE Trans. Antennas Propag., vol. 54, no.2, pp. 439–448, Feb. 2006.

[9] C. R. White and G. M. Rebeiz, “A slot-ring antenna with an octave oftunability,” in Proc. IEEE Ant. Prop. Symp., Jun. 2007, pp. 5841–5844.

[10] S. Raman and G. M. Rebeiz, “Single- and dual-polarized millimeter-wave slot-ring antennas,” IEEE Trans. Antennas Propag., vol. 44, no.11, pp. 1438–1444, Nov. 1996.

[11] C. R. White and G. M. Rebeiz, “A dual-polarized slot-ring antennawith independent tuning,” in Proc. IEEE Antennas Propag. Symp., Jun.2007, pp. 113–116.

[12] R. Garg et al., Microstrip Antenna Design Handbook. Boston, MA:Artech House, 2001, ch. 7.

[13] MA46H071-1056 M/A-Com. Lowell, MA: M/A-COM/Tyco Ele-cotronics.

[14] IE3D 11. Fremont, CA: Zeland Software Inc.[15] Microwave Bluetooth Ferrites. Tampa, FL: Ferrishield, Inc.[16] SG 32 Aug. 2008, Satimo. Courtaboeuf, France, http://www.

satimo.fr/eng/index.php? categoryid=127.

Carson White (S’03–M’08) received the B.S.degree in electrical engineering from the Universityof Washington, Seattle, in 2002, and the M.S. andPh.D. degrees in electrical engineering from theUniversity of Michigan, Ann Arbor, in 2004 and2008, respectively.

He is currently a Research Staff Member with HRLLaboratories, LLC., Malibu, CA, where he works inthe field of applied electromagnetics and antennas.

Gabriel M. Rebeiz (F’97) received the Ph.D.degree from the California Institute of Technology,Pasadena.

He is currently a Professor of electrical and com-puter engineering at the University of California,San Diego. Prior to this appointment, he was atthe University of Michigan from 1988 to 2004.He leads a group of 18 Ph.D. students and threePostdoctoral Fellows in the areas of millimeter-waveRFIC, microwaves circuits, RF MEMS, planarmillimeter-wave antennas and terahertz systems. He

is the Director of the UCSD/DARPA Science and Technology Center on RFMEMS Reliability and Design Fundamentals. He is the author of the book RFMEMS: Theory, Design and Technology (Wiley 2003). He has contributed toplanar millimeter-wave and THz antennas and imaging arrays from 1988 to1996, and his group has optimized the dielectric-lens antennas, which is themost widely used antenna to date at millimeter-wave and THz frequencies.He is also the inventor of the Integrated Horn Antennas and was the firstto demonstrate antennas on thin dielectric membranes. His group recentlydeveloped 6–18 GHz and 30–50 GHz 8 and 16-element phased arrays on asingle chip, making them one of the most complex RFICs at this frequencyrange.

Prof. Rebeiz is an IEEE Fellow, an NSF Presidential Young Investigator, anURSI Koga Gold Medal Recipient, an IEEE MTT Distinguished Young Engi-neer (2003), and is the recipient of the IEEE MTT 2000 Microwave Prize. Healso received the 1998 Eta Kappa Nu Professor of the Year Award, and the 1998Amoco Teaching Award given to the best undergraduate teacher. He has beenan Associate Editor of the IEEE TRANSACTIONS ON MICROWAVE THEORY AND

TECHNIQUES and a Distinguished Lecturer for IEEE MTT and IEEE AP.

single- and dual-polarized tunable slot-ring antennas

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