simplified approach for predicting dynamic thermal

Simplified Approach for Predicting Dynamic Thermal Behavior ofSwitching Electronic Devices

A. Lakhsasi (a) , Y. Hamri (a) and A. Skorek (b)

(a) Department of Computer Sciences, Université du Québec à Hull, Hull,(PQ) J8X-3X7, Canada.,(b) Industrial Electroheat Laboratory, Université du Québec à Trois-Rivières, Trois-Rivières,(PQ) G9A 5H7, Canada.

Abstract: Thermal investigation is a fundamental issue in power converter design since IGBT’s (Insulate Gate Bipolar Transistor) are widelyused in the application of motor drivers, switching supplies and other power conversion systems. Accurate thermal prediction of junctiontemperature of semiconductor devices has become the major issue with the increase of the current density and high switching frequency ofadvanced power devices. The paper presents a simplified approach for predicting working temperature of switching devices. As an exampleIGBT PWM (Pulse-Width Modulation) inverter is given to demonstrate the application of this approach. Based on the measurement of IGBT'sdynamic characteristics, the estimation of power loss considering the junction temperature is introduced. Then the finite element analysis is usedto accurate peak junction temperature prediction needed during dynamic operating conditions. In addition, the effect of switching frequenciesduring transient thermal response is investigated. The new approach developed can be used for accurate rating semiconductor devices or heatsink systems in power circuit design. Results comparison between proposed approach and commercial simulator shows that this approach iseffective as a designing step.

Key words: Transient thermal analysis, Finite element model, IGBT, dynamic thermal behavior, power device rating.

IPACK2001-15595

I. INTRODUCTION

Dynamic thermal junction analysis is of crucial importance inthe design of modern converters due to the increasing currentdensity and higher power requirement. Simplicity of gatedrives; ease of protection, snubberless operation, and lowswitching times makes IGBT converters very attractive. In thecase of power converters based on PWM (Pulse-WidthModulation) high switching frequencies (i.e. > 1 kHz) thedevices dissipate a considerable amount of heat. Regardless ofthe electrical quality of the semiconductor devices, the deviceswill fail if the heat cannot be removed effectively. Therefore,thermal analysis is a fundamental issue in power converterdesign, since the most convenient choice of the switchingpower device and associated heat sink are both strictly relatedto the basic requirement of keeping the junction temperaturebelow the maximum admissible value. Due to the lack ofprecise dynamic thermal model of switching semiconductordevices much of the effort is spent in performing experiments,by indirect method [1], on line thermal measurement [2], tomeasure junction temperature generated in the switchingpower devices. This works require complex dynamic controland takes expensive amounts of time and money. However,due to the complexity of these problems, no simple and precisetool has been developed to solve them. The multilevelstructure of power device increases the difficulty of spatialthermal analysis since the behavior of structure is not a simpleaverage of single-layer behavior [3].Nowadays the practical and available approach for dynamicthermal analysis is to use a thermal network provided bydevice manufacturers. Figure 1 shows an example of thermalequivalent circuit for IGBT used for predicting dynamicthermal behavior of junction temperature [4]. Each thermalmodel represents the thermal impedance (transient response)

1

and resistance (steady state) of a particular part. Using singleRC model and lumping all of the device components together(silicon, solder and package) approximate thermal analysiswill be possible but will give a poor representation of thejunction temperature for anything but very low (<100Hz)frequencies [3,5]. High switching and/or conductiondissipation levels can result in junction temperaturesexceeding device's rating even at moderate frequencies.Therefore, single RC representing the power device will notrespond properly; it will indicate a lower junction temperatureor virtually no temperature rise. Hence, in these cases wherethe main amount of power is dissipated during short time thetransient finite element computations are stronglyrecommended.The purpose of this paper is to present a simple approach toestimate the power loss and working temperature in switchingdevices. Based on partially coupled approach the new methoddeveloped incorporate power loss is in an NISA finite elementprogram for device dynamic thermal analysis. It shows thatpartially electro-thermal coupling is the alternative to fullelectro-thermal simulation [5]. Furthermore, the newapproach developed gives easy evaluation of devicestemperature during system startup comparatively to themethod that combine on-line thermal measurement withclosed-loop control to reduce switching component thermalstress during transient thermal response of silicon chipdevices.

II. SWITCHING DEVICES THERMAL RATING

Electronic devices and heat sink rating for the design of powerconverter circuits necessarily involves a thermal analysis. Tothis aim, in practice the simple dynamic thermal model shownin Fig. 1, which exactly corresponds to the thermalcharacterization provided by most device manufacturers, isused traditionally for device thermal rating.

Copyright © 2001 by ASME

In this case, thermal fluctuations of the junction temperatureare assumed to be negligibly small within period offundamental (16.67 ms for 60 Hz). This model represents thejunction temperature Tj in term of the static junction-to-casethermal resistance Rjc, and the thermal time constant τth = Rthj

Cthj. Clearly, this is a simplified model because it does not takeinto account the non-uniform internal device temperature andswitching frequency. For low frequencies, device selection andcircuit design are normally carried out exclusively on dataprovided by manufacturers, this is the only scheme that can bepractically used. Hence, when using this simple model,dynamic thermal analysis of pulse-width modulation (PWM)controlled DC/AC converters is not a trivial task, owing to thecomplex shape and spectrum of electrical variables. Up tonow, this problem has been faced by using full electro-thermalsimulation techniques [5], that may provide sufficientlyaccurate results, but are not suitable for a first tentative designprocedure owing to their complexity.

In present study high frequencies PWM operating conditionsare used. Hence, thermal analysis of IGBT devices, even whenusing the simple model in Fig. 1, is quite complicated owingto both the dynamics involved and the complex spectrum ofthe electric variables coupled with thermal diffusion equation.In this case, the thermal analysis can be greatly simplifiedwhen switching losses and thermal equations are partiallycoupled.

III- SIMPLIFIED APPROACH TO THERMAL ANALYSIS

The new approach proposed in this paper can be convenientlyused in the thermal design of PWM- controlled DC/AC powerconverters. Therefore, the upper limits on operating conditions(i.e., load current, converter frequency, etc.) can be directlyexpressed in terms of a given upper limit on die junctiontemperature. Figure 2 shows electro-thermal simulationscheme used for partially coupling dynamic thermal analysisproblems. Hence, the new approach is based on switching lossmeasurement combined with transient finite element analysisto predict spatial dynamic thermal behavior at differentlocation in the device structure.

Figure 1: Illustration of IGBT’s traditional thermalequivalent circuit (RC) used for predictingdynamic thermal behavior of device

2

The thermal feedback Tj in Fig. 2 is needed since power lossmodels are temperature dependent. Hence, the partiallycoupled approach to thermal analysis described above involvesseveral, mildly pessimistic approximations (e.g., conductionand switching losses linearly dependent on i(t), averagedpower dissipation is used for predicting steady state). In orderto verify that such approximations do not lead to anexcessively pessimistic estimate of Tj, the thermal diffusionequation (15) have been numerically solved by directlyconsidering the actual voltage and current waveforms in thesame inverter driving an asynchronous motor RL equivalent.The full coupled electro-thermal simulation was carried outusing Saber simulator [6].

IV. THERMAL BEHAVIOR IN SWITCHING DEVICES

Sometimes, experimental data provided by devicemanufacturers (e.g., graphs providing SAT

ceV and switching loss

energies as functions of current, voltage and temperature) canbe directly used as look-up table models for finite elementthermal investigation. By using these available models theaverage transistor dissipated power swP can be computed [1, 7-

12] and, consequently, the average junction-to-casetemperature drop jcT can be estimated using the proposed

approach. On this basis, for converter operating conditionsthat involve non-negligible "ripple" in the junctiontemperature, a correct thermally safe design (i.e. combinedheat sink and power transistor rating) must be carried out [4,5]. This happens when output fundamental frequencies aremuch higher than the cut-off frequency associated to thermaldynamics (e.g., in DC/DC or DC/AC converters withrelatively high output frequencies and fast switching devices).In such conditions, the peak value peak

jcT of the junction-to-case

temperature, which can be much greater than its mean valuemeanjcT . Therefore, mean

jcT must be correctly estimated for thermally

safe design.

V. SWITCHING DEVICES LOSS CHARACTERIZATION

The power structure of a Voltage Source Inverter (VSI) of astandard industrial electrical drive (for power range above3kW typically) consists of 6 IGBT devices and 6 anti-parallelFree Wheel Diodes (FWD).

Figure 2: Measurement-based simulation scheme used forpartially coupling dynamic thermal analysis.

ElectricalWaveforms

Measurement

PowerLosses

Evaluation

FiniteElementThermal Model

( )tT j

( )tTa

( )tTc

( )tvL

( )tiL


The topology of the power converter considered in this paperis shown in Fig. 3, this circuit was adopted to measure theprecise transient switching processes of IXGK 50N60AU1,600V,75A, IGBT power device, although many variations ofthis basic inverter are used in practice. The proposed approachcan be applied to either a three-phase or to a single-phaseconverter. The same commutation law, except for theassociated time-shifts, is assumed for all the "legs" of theconverter. The previous hypotheses simplify the thermalanalysis since, under such conditions, all the 'legs" are subjectto the same thermal stresses. For this reason only one "leg" ofthe converter is considered. In the same way, we assume thatboth transistors of each "leg" operate under the same electricaland thermal stresses. Thus, only one transistor for each 'leg" isconsidered for thermal analysis (in the following S1 will bechosen).

- Switching loss measurement:

A number of loss models, including both conduction andswitching losses (i.e., turn-on and turn-off loss energies) havebeen proposed in the literature [7-14]. The common way toestimate the power loss of devices is based on the exact currentand voltage waveforms of device [7]. Obviously, it is verydifficult to get the waveforms from simulating each pulse ofPWM exactly with the variation of the voltage and the current.

Figure 4: IGBT ‘s power loss characterization duringON-OFF pulse including free wheel diode.

Figure 3: Experimental circuit adopted to measure IGBT’sprecise transient switching losses .

3

The accumulated error of loss would be generated with theerror of such waveforms. Another difficulty is that some of theprecise parameters of power semiconductor devices which arereferred in the model [7-12] are unavailable. In the powerswitching devices, the critical factor to care most is thejunction temperature that has the peculiar safe operating area.Usually the power loss is calculated under the constantjunction temperature [3]. However the power loss does dependon the junction temperature during transient switchingoperation. Therefore the power loss estimation and thejunction temperature computation should be combinedtogether to find out the devices working temperature. Thepower loss of each switching operation of IGBT is divided intothree portions illustrated in Fig.4. Hence, it can be assumedthat the transient power loss of turn-on or turn-off depends onthe DC-link voltage V and collector current i and junctiontemperature Tj of IGBT.

Therefore, for the three parameters ( Tj , V, i ) we candefine two following functions:

( )( ) )2( ,

)1( ,

..

..

iTfV

iTfV

jsatdsatd

jsatSsatS

=

=

Total power loss during each pulse of IGBT is the sum ofturn-on loss, turn-off loss and saturation loss and function ofthree parameters (Tj , V, i) which are expressed in thefollowing equations :

( )( )

( )( ) (6) ,,

(5) ,,

(4) ,,

(3) ,,

..

..

..

..

iVTfP

iVTfP

iVTfP

iVTfP

joffdoffd

jondond

joffSoffS

jonSonS

=

=

=

=

Therefore, total heat dissipated is the sum over n transientphase (ON_OFF) and n saturation. Also the losses of anti-paralleled diode are included.

∑

∑++=

++=

nd

nS

nPnPnPP

nPnPnPP

satdoffdond

satSoffSonS

(8)

(7)

...

...

(9) dStotal PPP +=

Hence the loss is the sum of transient losses (ON-OFF) andsaturated loss of all IGBTs and diodes. For the PWM invertersthe collector currents of devices vary according to thecontrolling command. Therefore, junction temperatures alsorise up and fall down with the changes of the nature of load(impedance). The average power of IGBT # 1 can beestimated with the following equations (see Fig. 6):

(10) 2

...

2

0

ontgFdtttgt tvP on

satonαα == ∫

where tI

on

Mtg =α current speed, depending to inductive load

value (L): LE

dt

dI L =


Hence, we calculate : E/L = 320 / 7.5 = 43.14 A 1−sµ and

we measure : tg α = 42 / 1U = 42 A 1−sµfor satV = 2.7 V ; onP = 1.2 watt

The power dissipated during swintching off can berepresented by the following expression

(11) ...1

exp. CEVFdtCRR

EVFP sat

cCsatoff =

−= ∫

in the present measurement: offP = 35 W and condP = 22.6

W.

However, the power dissipation calculation method is exposedand described in detail in ref. [14]. Fig. 5 shows experimental

Time 20 ns/div

Time 0.25 us/div

Figure 6: The collector current (ICE) and the emittercollector voltage (VCE) measurement duringturnOFF-turnON cycle.

Signal 1 : VL , 2 : IL

Figure 5: Measured load current IL and voltage VL for 900Hz switching (15 A /div, 300 V /div and 3 ms/div).

4

results: (1) load voltage ( )tVL, and (2) load current ( )tiL

.

Hence, fig. 6 shows current and voltage measurement duringswitching (ON-OFF) and conduction.

The corresponding dissipated power per cycle is summarizedin Table I .

TABLE I: Averaged IGBT’s switching power lossesSwitching-OFF power dissipation 35 WSwitching-ON power dissipation 1.2 WConduction power loss 22.6 W IGBT’s total power dissipation 58.80 W

- Average power dissipation

The locally averaged power (averaged over one PWM period

TPWM = 1/f pwm ) on IGBT swP , and on FWD dP follows from:

( ) (12) ∫ ++= tofftonsatpwmsw EEdtVIfP

( ) (13) ∫ ++= rrtonddpwmd EEdtVIfP

where integration is done over a PWM period. The I and Vsat

are current and voltage of the IGBT during conduction, Id andVd are current and voltage of the FWD during conduction.Finally, Eton and Etoff are switching energy losses of the IGBTand Err is energy recovery losses of the FWD. Note that in (12)and (13) voltage and energy losses are expressed as functionsof instantaneous value of current. Locally averaged powerfrom (12) and (13) due to high PWM frequency can be, forpractical purposes, viewed as the instantaneous power.

It can be noted that switching losses are almost linearlycurrent dependent and mildly temperature dependent [14].Moreover, SAT

ceV shows only a mild dependence on both current

and temperature.

Electrical part: simulation and experiment

Experiment was done for 900 Hz switching frequency. Theschematic of Fig, 7 consists of the switching control logic for

IGBT gate drivers. The logic circuit of the network uses event-driven logic elements to implement the open-loop, sine-

Figure 7: Logic control and IGBT gate-drive circuit used toproduce the open-loop sine-triangle, PWM.


triangle, PWM control system for the switching devices. Fig. 5and 8 shows a comparison between measurement result andsimulation under the same conditions. However, fig, 8 showsthe simulation results using Saber package [6] (a) the sine-triangle control signals for the inverter of Fig. 3, (b) logicinput signals to the IGBT gate drivers (S1 and S3), and (c) loadcurrent ( )tiL

. The simulation of Fig, 8 is for a relatively low

IGBT switching frequency (900 Hz) to illustrate the behaviorof the circuit. In the PWM inverter of Fig. 3, the duty cycle ofthe input signal to the IGBT gate drivers is varied using thesine-triangle comparison technique to produce a 60-Hzsinusoidal variation of the load current. Finally, fig. 5 showsexperimental results: (1) load voltage ( )tVL

, and (2) load

current ( )tiL. These results are compared with those given in

figure 8. Therefore, the experimental results obtained are infull agreement with Saber simulation.

VI- Device thermal analysis using finite element method

A major feature of the thermal problem is the need to simulatea very large region of the device and substrate, in fact thepackage geometry often needs to be taken into account. Thesimultaneous solution of the three-dimensional (3-D) electro-thermal problem is therefore difficult due to the need for veryfine meshing of the device equations at the junctions and aneed for a large simulation region to produce an accuratethermal simulation. However, the thermal behavior of thedevice and package can be simulated without solving thedevice physical equations if the power generation region isknown a priori. Knowing the power generation position theheat equation can be solved independently to calculate thedevice temperature.NISA is a sophisticated, commercially available 3-D FEMsimulator [15] that has the ability to simulate time-dependentand steady state 3-D nonlinear thermal systems. NISA is anintegrated environment, in which a 2-D or 3-D model can beconstructed and meshed, boundary conditions set and theresulting nonlinear set of equations solved. The tool includes asophisticated DISPLAY4 to aid in building the model anddisplaying the results.

- Thermal boundary conditions

A wide variety of boundary conditions can be applied usingNISA. However, the boundary condition on the vertical sidesof the simulation region is somewhat problematic. Placing afixed boundary condition on these surfaces produces adramatically incorrect result, unless a very large simulationregion is used at the expense of very long simulation runtimes. A more natural boundary condition is a zero flowcondition across these surfaces (adiabatic boundaryconditions). The remaining boundary condition to be definedis on the bottom surface of the simulation representing thechip/package interface. The simplest approach is to fix thebottom surface at a constant temperature representing thepackage temperature.

5

To introduce pulse heating this approach uses a point-by-pointuser-defined function available in NISA, to model the dynamiccharacteristics of systems. Since there is no device physicsmodeling required. The proposed approach is applicable toboth static and dynamic modeling, on a cycle-by-cycle basis,which is important for dynamic power dissipation and thermalanalysis. The simulation includes IGBT turn-on and turn-offtransients, IGBT saturation. Using power loss measurementthe simulation results are verified by comparison with theSaber commercial simulator.A 2-D transient and steady-state thermal analysis of thestructure (0.63 mm thick and 0.8 mm wide) shown in figure 9has been implemented in the NISA simulator which usesHEAT2 to solve the heat flow equation. Furthermore, thermalproperties of each material have been supposed to bedependent of the temperature. This model is quiterepresentative of a real power device. Hence, heat is dissipatedat the junction surface (equivalent to small region with

Figure 8: Simulation of electrical circuits of Fig. 3 and 5for 900 Hz:, graph (a) shows sine-triangle PWM, graph (b)shows logic input signal to IGBT 1 and 3, and graph (c)shows load current.

(a)

(b)

(C)

Ta

TC

Tj

Figure 9: 2-D finite element model used to thermalanalysis of the structure (0.63mm thick and0.8 mm wide) implemented in NISA [15].


dimensions 0.25 x 0.1 mm2) as chown in fig. 9. Finally, thesimulated material thermal properties are listed in table II.

Table II: Thermal properties of principal junction materialsused in finite element model.

IGBT’smaterial

Density3/ cmKg

Specific heatCp ( ckgJ o./ )

Thermalconductivity

( cmW o/ )Aluminum 25.02 25.07 25.2Copper 25.9 27.3 35.6Silicon 26.5 30.2 41.3Bonding 30 39.5 64.7

- Heat diffusion equations

The conservation of energy law gives the heat conductionequation by considering the heat flow equilibrium inside thedevice body:

( ) ( ) ( ) (14) tT

CqzT

TKzy

TTK

yxT

TKx pzyx ∂

∂=+

∂∂

∂∂

+

∂∂

∂∂

+

∂∂

∂∂

ρ

(9)

where Kx, Ky, Kz are the same as Kxx, Kyy, Kzz , respectively,P(t) is the rate of heat generated per unit volume, t is the time,ρ is the masse density of the material and CP is the specificheat (where in the case of conduction in solids, no distinction[16] is made between the specific heat at constant pressure,CP, and the specific heat at constant volume, CV).

Equation (14) is the general equation for heatconduction in solids and in the case of power devices, allmaterials are isotropic with constant conductivity (Kx= Ky=Kz)are, K, therefore, the equation reduces to

)15( ))((tT

CTTK p ∂∂

ρ=∇⋅∇

In this model, the device substrate were approximated as heatsink at constant ambient temperature (Dirichlet boundaryconditions).

VII- SIMULATION RESULTS AND DISCUSSION

- Transient thermal investigation

The single heating cycle (on/conduction/off) response isshown in Fig. 10 to illustrate the dynamic behavior of IGBTduring startup. It shows the evolution of the temperature atthe three different device region. As shown, a non-uniformtemperature profile exists through the device during ON-OFFcycle. After the end of ON-pulse heating, there is subsequentrelaxation on temperature gradients through the structureleading to an essentially uniform temperature variation.However, after the end of OFF-pulse heating, the temperaturedecays exponentially with a time constant determined by theconduction of heat from the junction interface into the wall

6

Figure 10: single heating cycle (on/conduction/off)response temperature waveforms at the silicon-chip surface Tj, chip-package interface Tc, andpackage heat-sink interface Ta for IGBT 1.

ON

OFF

Conduction

Figure 11: temperature waveforms at the silicon-chipsurface Tj, chip-package interface Tc, andpackage heat-sink interface Ta for IGBT 1operating at 900 Hz switching frequency.

device. The cooling of the device is essentially controlled byconduction through to the device package. Therefore, themaximum thermal variation, for one cycle ON-OFF, ∆T is

0.01 oC. It is observed that the maximum temperature occursat the completion of the OFF-pulse.

Fig. 11 shows the temperature waveforms for three 60 Hzcycle at the silicon-chip surface Tj, chip-package interface Tc,and package heat-sink interface Ta for IGBT 1 operating at900 Hz switching frequency. Notice that the silicon-chipsurface-temperature waveform of IGBT 1 has spikes at a 900Hz rate during the phase that the device is switching due tothe switching energy losses. Also, notice that the chip-surfacetemperature cools after the peak in load current and during the

phase in which the device is off. The 60- Hz variations of thechip-surface temperature waveform are due to the sinusoidalload-current variation and due to cooling during the half of the60-Hz cycle in which the device is off all of the time.

From the temperature waveforms of Fig. 11, it is evident thatthe thermal response of the silicon chip determines the IGBT


temperature variations during the device 900 Hz switchingcycle because the temperature at the package heat-sink Ta doesmildly change during the 900 Hz cycles. In this case, peakthermal gradient is approximately 0.1 oC.

To illustrate the complexity of heat flow in device junctionregion Fig. 12 shows device in-plane spatial temperaturedistribution. The heat is shown as being generated in a well-defined region at the device junction. A portion of heat flowsinto the substrate spreading in two dimensions as it flows

Figure 12: spatial temperature distribution for IGBT 1 atthe completion of the third 60 Hz cycle.

Figure 13: temperature waveforms at Tj, Tc, and Ta forIGBT 1 operating at 20 kHz switching frequency.

Figure 14: simulated temperature response (Tj, Tc, andTa) at startup using averaged power dissipation in IGBT1.

7

toward the bottom of the substrate, which is a device heat sink.

PWM inverters are typically operated with switchingfrequencies in the range of 20 kHz in order to push theswitching frequency beyond the audible frequency range. Aconsiderable amount of heat is dissipated in the IGBT's due tothe on-state losses for the large load current and switchinglosses that occur for the typical 20 kHz device switchingfrequency. Figure 13 shows the temperature waveforms at Tj,Tc, and Ta for IGBT 1 operating at 20 kHz switchingfrequency. Hence, in this case, instantaneous peak thermalgradient is approximately 5 oC, which can significantlycontribute to thermal stress level by reducing device life.Therefore, thermal stress study can be conducted forpredicting stress and distortion of junction interface forenhanced device rating.

- Steady state thermal analysis

For the inverter of Fig. 3, approximately 1hour of real time isrequired for the device temperatures to reach a steady-statecondition. Thus, a simulation of the complete thermal startupcondition would take an inordinate amount of simulation timefor the 20 kHz inverter operation. Therefore, average powercan be calculated using dissipation in the IGBT 1 over a single60 Hz cycle of the inverter.Because the IGBT electrical characteristics change withtemperature, the average power-dissipation level changes asthe heat-sink temperature rises. To account for this, after eachcomplete 60 Hz cycle update of power dissipation level is thenperformed for the next cycle.Fig. 14 shows simulated response at startup using averagedpower dissipation in IGBT 1. Therefore, for steady state, thethermal response of the device is determined by the total heatcapacity of the device package and Dirichlet boundaryconditions with ambient environment. Table III summarizesthermal drop at different location for the considered IGBT (S1)after 3 cycle of 60 Hz.

Table III: Thermal drop at different location for the IGBT(S1)

Fsw Ta(oC) Tc(

oC) Tj (oC) ∆TjMax(

oC)900 Hz 25.02 25.07 25.2 0.25 kHz 25.9 27.3 35.6 2.210 kHz 26.5 30.2 41.3 3.320 kHz 30 39.5 64.7 4.8

In summary, the modeling approach produces a wealth of newinsights in transient thermal behavior of switching devices andpromises to simplify the dynamic thermal analysis forpredicting working junction temperature.

In fact, thermal fluctuations are significantly higher than thejunction temperature predicted by the conventional steadystate model. Therefore, evaluation of the dynamic thermalbehavior of packaged semiconductor components is needed.Hence, during design of electronic system, precise thermal


Proceedings of IPACK'01 The Pacific Rim/ASME International Electronic Packaging

Technical Conference and Exhibition

cycle evaluation is important for control of the internal heatflow path of a chip package.

To avoid overheating causing device failure (Yield Stress) andto have optimal utilization of electronic devices in the wholeoperating range of the drives, a dynamical thermal analysis isnecessary. The new approach developed to accurate predictionof device junction temperature to maintain yield stress belowcritical values. However, dynamic thermal analysis willbecome a more useful design approach when device materialsthermophysical properties are available in component librariesfor all of the standard thermal components used for typicalsystem design.

VIII- CONCLUSION

Partially coupled approach for analyzing dynamic thermalbehavior is an indispensable tool for thermal rating ofswitching devices. In this paper, new approach is presented toprovide, in general, a methodology to evaluate dynamicthermal behavior in switching electronic devices. This is animportant capability because the cost of the thermalmanagement of electronic systems depends heavily upon theefficiency of the circuit design and because the self-heating ofthe semiconductor devices can affect the operation of theelectronic circuit. It is shown that partially electro-thermalcoupling is the alternative to full electro-thermal simulation.Certainly, the new approach developed gives easy evaluationof device temperature during system startup comparatively tothe method combining on-line thermal measurement withclosed-loop control.Modern computational techniques such as finite elementsimulations are useful tools for the package design engineerand help to prevent unexpected pitfalls. These simulationsallow an accurate prediction of the temperature distribution ina package and help to keep localized thermal peak occurringin different packaging designs to a minimum. Consequentlythe design engineer is able to optimize the geometry and thechoice of materials of a package even before prototypes arebuilt. This greatly reduces development time and increases thequality of the product. Furthermore, the new approach shedslight on the phenomenon of heat transfer in multileveldevices. To demonstrate the application of this approach, amultilevel device such as IGBT is selected for present study. Itis shown that the new approach can be easy applied forprediction of transient thermal stress of multilevel structuressubject to repetitive pulse heating.

REFERENCES

[1] Szekely, V, "THERMODEL: A tool for compact dynamicthermal model generation", Microelectronics Journal,1998, v.29, n.4, p.257, 12p.

[2] Szekely, V, "A new evaluation method of thermaltransient measurement results", MicroelectronicsJournal, 1997, v.28, n.3, p.277, 16p.

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[3] Kraus, R; Mattausch, H J, "Status and Trends of PowerSemiconductor Device Models for Circuit Simulation" ,IEEE Transactions on Power Electronics, 1998, v.13,n.3, p.452, 14p.

[4] Sofia, J W , "Analysis of Thermal Transient Data withSynthesized Dynamic Models for SemiconductorDevices", IEEE Transactions on Components Packagingand Manufacturing Technology - Part A, 1995, v.18, n.1,p.39, 9p.

[5] A. R. Hefner Jr. "A Dynamic Electro.-Thermal Modelfor the IGBT" IEEE Transactions on IndustryApplications Vol 30 No 2 March/April 1994.

[6] Saber Simulator, Analogy Inc., P.O. Box.1669,Beaverton, OR 97075-1669, USA.

[7] Mantooth, H A; Hefner Jr, A R, "ElectrothermalSimulation of an IGBT PWM Inverter", IEEETransactions on Power Electronics, 1997, v.12, n.3,p.474, 11p.

[8] Hefner Jr, A R , "Modeling Buffer Layer IGBT's forCircuit Simulation", IEEE Transactions on PowerElectronics, 1995, v.10, n.2, p.111, 13p.

[9] Hefner, A R; Blackburn, D L, "Thermal ComponentModels for Electrothermal Network Simulation",Transactions on Components Packaging andManufacturing Technology - Part A, 1994, v.17, n.3,p.413, 12p.

[10] Hefner, A R; Blackburn, D L, "Simulating the DynamicElectrothermal Behavior of Power Electronic Circuitsand Systems", IEEE Transactions on Power Electronics,1993, v.8, n.4, p.376, 10p.

[11] Hefner Jr, Allen R, "A Dynamic Electro-Thermal Modelfor the IGBT", IEEE Transactions on IndustryApplications, 1994, v.30, n.2, p.394, 12p.

[12] Mitter, Chang Su; Hefner, Allen R; Chen, Dan Y; Lee,Fred C, "Power Electronics Devices and ComponentsCommittee: Insulated Gate Bipolar Transistor (IGBT)Modeling Using IG-Spice", IEEE Transactions onIndustry Applications, 1994, v.30, n.1, p.24, 10p.

[13] Clemente, S, "Transient Thermal Response of PowerSemiconductors to Short Power Pulses", IEEETransactions on Power Electronics, 1993, v.8, n.4,p.337, 5p.

[14] Filicori, F; Bianco, C Guarino Lo, "A SimplifiedThermal Analysis Approach for Power Transistor Ratingin PWM-Controlled DC/AC Converters", IEEETransactions on Circuits and Systems - Part I -Fundamental Theory and Applications, 1998, v.45, n.5,p.557, 10p.

[15] NISA II “user's manual “ EMRC (EngineeringMechanics Research Corporation) Troy, Michigan,USA (1998).

[16] M. Plischke and B. Bergersen, “Equilibrium StatisticalPhysics”, Prentice Hall, Englewood Cliffs, New Jersey07632 (1989).

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