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Scattering Parametersmeasurement setup

for PA transistors

RIEHL Christophe

Internship at EADS-Telecom

June, 7th 2004 – November, 12th 2004

Scattering Parameters measurement setup for PA transistors

Summary

I. EADS Telekom 4

1. The EADS group...................................................................................4 2. EADS Telekom..................................................................................... 6 3. Tetrapol standard...................................................................................7 4. Research & Developpement Division...................................................8 5. The Superstar project............................................................................ 8

II. Introduction - Methodology 9

1. Pa design problems............................................................................... 9 2. Low Z0 measuring................................................................................ 9 3. High power measuring........................................................................ 10 4. Precision and error mode.................................................................... 10 5. Connections.........................................................................................10 6. Methodology....................................................................................... 10

III. Low-Z S-Parameter measurement tests with transformers 12

1. Transformed choice.............................................................................12 2. Tests.................................................................................................... 12 3. Results.................................................................................................13

IV. High Power S-Parameter measurement tests 16

1. Test setup............................................................................................ 16 2. Amplifier.............................................................................................16 3. Isolation...............................................................................................17 4. Coupler................................................................................................18 5. RF switch............................................................................................ 18 6. Test Setup............................................................................................18 7. D.U.T: device under test..................................................................... 19 8. Results.................................................................................................19

V. Transmission Line Transformers 21

1. Choice of TLTs................................................................................... 21 2. Basic principle of TLTs...................................................................... 21 3. Frequency & Bandwidth..................................................................... 22 4. Input matching.................................................................................... 23 5. Transformer ratio................................................................................ 23 6. Using TLTs for PA design.................................................................. 23

RIEHL Christophe ENSEIRB – EADS – 2004 2/52


VI. Adapter 24

1. Principle.............................................................................................. 24 2. 1-100 MHz adapter............................................................................ 25

VII. Transistor biasing circuit 28

1. Principle of operation..........................................................................28

VIII. Measuring & Procedures 30

1. Connecting :........................................................................................ 30 2. Oscillation checking : .........................................................................30 3. VNA settings for S-parameters measurement.....................................30 4. Calibration...........................................................................................31 5. Power sweep measurements................................................................31

IX. Results 32

1. Accuracy..............................................................................................32 2. Conversion of S-parameters to other Z0 and validity......................... 33 3. Power sweep....................................................................................... 33

X. Conclusion 35

1. Feasibility............................................................................................35 2. Accuracy..............................................................................................35 3. Design extension................................................................................. 35

XI. Bibliography 36

XII. Annexes 37



I. EADS Telekom

1. The EADS group

With defence, service and aeronautic activities, EADS ( European Aeronautic Defence ans SpaceCompany ) is the biggest aerospace company in Europe, and the second worldwide. More than 110000 people work in 70 production centers, essentially in Germany, France, Spain and Great Britain.There are 35 external buros in the world for maintaining contact with all the customers.

EADS was founded June, 1st 200, from a fusion of Daimler Chrysler AG , Matra-Aerospatiale andCASA. A part of 60% of the capital belongs to Sogeade and Daimler Chrysler.With the spanishSEEPI, this 3 infrastructures hold 66.1% of the total capital in a Contractual Partnership(netherlands right). The last 33.9% are floating, with 3.5% belonging to EADS employees.

The Fig. 1

Fig. 1Capital parts (July '04)



EADS is composed of 5 major divisions :

Fig. 2

– Airbus

Second worldwide Airliners producer. Over 45 000 employees in France, Germany, Great Britainand Spain work on modern aircrafts:

– A320 family: A318, A319, A320, A321 (107 to 185 seats)

– A300/310 : 220 to 266 seats

– A330/340 : 253 to 380 seats

– A380 Family (double deck, over 500 seats) to come in 2006

– Military Transport Aircraft



This division designs light and mid-weight military transport aircrafts. Modernisation of Avionikelectronics, and variants of Airbus series aircrafts are in production.

– Aeronautics

This division manages all aeronautic activities that are not comprised in Airbus and MTA. 23400people work on productions like :

– Military and civilian helicopters

– Regional Airliners

– Sport and travel aircrafts

– repair, modification, and other oprations on aircrafts.

– Space systems

This represents 10% of EADS activities. 12 300 people work on space activities :

– Space transportation and infrastructures

– Military and civilian satellites

– design of rocket structures

– optical and laser-systems

– Defence & Security Systems

This division is the youngest, and has been created July, 1st 2003. This division designs systemsfor the new war techniques, and for other security applications. The biggest applications are :

– Military aircrafts (eurofighter typhoon for example)

– Missiles navigation systems

– Aeronaval defence systems

– Communication systems, with the EADS telecom subdivision

2. EADS Telecom

EADS Telecom was founded March 2001, from the formaly telecommunication division of DASA,Aerospatiale, and Matra-Nortel. The aim was to make a common structure for all thetelecommunication activities in EADS. 3500 people (600 ingineers) work in 12 places. The mainproduction sites are located in France, Germany, Great Britain, and USA.

The main activities are military, and governement services, but private telecommunications arecommon too :

– Defense :

– Crypted telephony

– Tactical Networks



– Secure IP Backbones

– Public security :

– Digital radio networks and call centers, for Police, Firemen, airports, ambulances ...

– Information systems

– Civilian applications

– IP telephony

– Complete solutions for passenger transportation communication networks

3. Tetrapol standard

EADS develops communication systems using the tetrapol standard. This system is a cellularmobile communication system. Each cell has one base station and mobile transceivers can move inand between cells.

Technical details :

– TDMA canal repatition

– GMSK modulation

– 10 or 12.5 kHz Canal bandwit

– 8 kbps bitrate

– Cryptologic security, end-to-end

– -119dBm sensitivity

Other facts :

– Huge bandwith economy

– Good range

– Hierarchical network

– interfaces to other networks, and trunked systems

– Central or partitionned network management

– Simplex communications without network structures



4. Research & Developpement Division

In Ulm, the R&D part is sub-divided in laboratories. I worked in the RD4 lab. This one is called“RADIO HARDWARE”, and is parted into 3 laboratories :

– RD 41 : Digital products

– RD 42 : Analog products

– RD 43 : Industrialization

I worked in the RD 41 lab, which is composed of 9 people, including a electrotechnical assistant,and 8 Telecom-engineers. The average age in this division is 35 years.

The lab designs almost only hardware for non-military communication systems. The maincustomers are big governemental structures like Police, Firemens, ambulances, train companies likeDeutsche Bahn, or airports, for internal communications. The tetrapol standard is used on almost alldevices, even analog ones.

Even if the lab is calle R&D lab, most time people work on commercial products, especially testingof specifications, CEM tests, problems and bugs identification, and so on. Research is not alwaysthe main topic.

5. The Superstar project

The superstar project is a new concept of base station. The aim is to replace a multi-channel GMSK(FM) transmitter/receiver in classical base stations with an unique receiver whih can digitally handlemany channels.

This technique is derived from GSM-BS-technologies, thus using common components, andproviding a low-cost and miniaturized basestation. The project is only in the demonstator designphase. Such a transceiver needs wideband components, and has big problems with receiverdynamics, PA linearities, and so on. Fig. 3 Shows the principle of this basestation. Instead multiplereceivers, an entire spectrum is digitized in an IF frequency, and the receivers are digital.


Fig. 3Superstar Basestation principle


II. Introduction - Methodology

1. Pa design problems

To achieve a good PA design, the procedure is like this:

– selection of architecture (Push pull, single transistor, array, ...)

– selection of bias class (A, AB, B, ...)

– selection of transistor model

– selection of matching network types

– design of bias circuitry

– design and simulation of the matching networks

– thermal and mechanical design

– tests, and adjustments of the matching networks

As you can see, the matching networks are the main part who is adjusted experimentally. This ismainly due to the uncertainty of the transistor characteristics at high signal levels. The transistor isalmost always used near saturation, to achieve an acceptable efficiency. So the measured smallsignal S-parameters are not valid, and the mismatch is compensated experimentally. This approachis very time consuming and costly, especially if the matching network is made in a microstriptechnology.

Thus, being able to measure the transistor characteristics under his working condition could lead toa better simulation. With a conventional VNA, two main difficulties make this measure impossible:low impedance of the transistor, compared with the 50 Ohms of a VNA, and the maximal poweroutput of most VNA's are limited to 0dBm.

2. Low Z0 measuring

This can only be done using a real low-Z source and load (transformer). The source and load match



must be of a sufficient low impedance, or too much power will be lost with the mismatch.

There are no transmission lines, or connectors for low impedances, and no couplers, or otherdevices, so the impedance transformation has to be done as near as possible of the DUT.

3. High power measuring

S-parameter measurements with high power require all the output RF stage of the VNA to bedesigned for enough power. Some HP analyzers ( HP8753ES option 14 ) provide a convenient wayto connect extern components to perform such test configurations. This can be done with otheranalyzers, provided you can access the signals before the couplers. I did this with the Rohde &Schwartz ZVRE.

4. Precision and error mode

The error model of modern VNA's is able to compensate all type of parasite linear networks. Theonly things to take care of is not to have more than 360 phase rotation between open or short, forexample. A VNA can measure phases borned to 360°, and the algorithms work with such a limit.However, if there is too much phase rotation, the linear interpolation between the calibrated valueswill be absolutely false. So the matching network mustn't have too much poles and zeros, thusmakes measuring with a conventional matching network uncertain.

5. Connections

The low-Z interconnect is a problem, since there exists no connectors, nor coax lines. Theconnections should be done with direct soldering, and taking care of all additional little lengths,who can add significants inductance in comparison of Z0.

Transmission lines could be made of microstrips, but low-Z microstrips are very large. Thinsubstrates are a solution. Lengths should be reduced to a minimum, to avoid huge losses.

Connection repeatability should be tested, they could add serious errors in measurements.

6. Methodology



I first designed the Bias circuitry. This circuit is necessary to handle the polarization of the DUT.

Then I tested the method of calibrating the VNA after a low-power transformer. The results werevery good, so the concept of the measurement was validated.

After testing High-power VNA configuration, I developed the TLTs. These two concepts are themain frame of my work.

Finally I designed the adapter circuit, and then I used all the stuff to measure S-parameters.



III. Low-Z S-Parameter measurementtests with transformers

We would try to calibrate the VNA with a low-Z calibration plane who is located after thetransformer. The aim was to test if the S-parameter calibration model is able to deal with the hugeerrors an impedance transformation could induce. Rohde & Schwartz ZVRE documentation speaksfrom a “golden device” calibration, to compare a device line to a reference device used incalibration. But this is apparently not used for phase measurements, just Through amplitude.

1. Transformed choice

I choose the Micro-circuits ADT16-1-1 SMD transformer because of availability of enoughtransformers in the RD41 Lab. This transformer has a bandwidth of 1.5 – 160 MHz, and aimpedance ratio of 1:16. No choice over frequency, but 160MHz will be sufficient for this basictest.

2. Tests

With low-Z measurments, every little impedance in series is non negligible in regard with the lowZ0, and we can't use 50 Ohm connectors. The procedure is to use ONE transformer per experimentwith the secondary test or cal circuit soldered directly on the pins, and primary soldered to a SMAconnector. The calibration plane is located just after the transformer pins.

So I build a 50Ohm SMA cal kit, and a 3,125 Ohm cal kit with transformers. The DUT is a simple



RC network 2.7 Ohm / 470pF. (see Fig. 4)

Note the grounding of the one of the secondary transformer connections, this showed to beabsolutely necessary to prevent unwanted transformer resonances.

3. Results

The result seems pretty good within the transformer bandwidth. After cal, the cal kit impedance iscorrect, and the DUT characteristic seems good, better than the 50Ohm measures of the same DUT(see Fig.5 & 6 ) Perhaps a mis calibration in the 50 Ohm because 50 Ohm measures of a 3 Ohmimpedance can be very sensitive to little changes.

So the measurement with an impedance matching network is possible, until we stay in the matchingconditions (transformer bandwidth). This measuring method could be more precise that thetraditional 50 Ohm mismatching, given some conditions.



Fig. 4



Fig. 5 Measurement with matching transformer, Z0=3,125

Fig. 6 Measurement with DUT directly soldered to SMA Z0=50



IV. High Power S-Parametermeasurement tests

This aims to test the possibility of high power S-parameter tests in near 400 MHz with about 10Wpower.

1. Test setup

For this kind of measurement, we cannot use the S-parameter bridge of an conventional VNA,because the couplers, switches, terminations can't handle the amount power. So we need to built theoutput bridge of the VNA with the appropriate components.

We need: a linear amplifier, isolation, couplers, and attenuators.

2. Amplifier

I selected two linear amplifiers that I found in the lab:

BrandPower

WBandwidth

MHzGaindB

Misc

SCO -Nucletudes 20W (43 dBm) 500 - 1000 45 OK for 400MHz

E.I.N. 9.5W (40 dBm) 1.7 - 500 45

We have to select the right amp taking account of the measurement frequency band we need.



3. Isolation

The linear PA must be isolated from the wave that could return from the DUT. This return could be100%, specially when calibrating. If The amplifier is unconditionally stable, we could miss thisisolation, but the problem is that the a1 wave reference of our VNA is taken internally, before thePA. So the PA must work on the same conditions as when calibrating, and the return wave must notreach the PA. In a standard VNA, this is achieved by amplifying, and then attenuating the signal.The attenuator ensures sufficient isolation of the PA. With our high power, this approach isn't reallyfeasible.

So we have to use an isolator. The problem is these kind of devices have usually a small bandwidth.After many searches, I found some Octave-bandwidth isolators. One VALVO for 225-400MHz, anda PHILIPS for 400-470 MHz are available in the lab. The bandwidth exceeds a bit those values, ifwe consider a 15dB isolation, that is sufficient, and 3dB loss. For example, the philips isolator has abandwidth of 350 - 550 MHz (fig. 17)

Fig. 7 Philips isolator model 2722 162 01572 frequency response



4. Coupler

For the coupler, we need a device with a very good directivity. The best coupler available was aHP778D measuring coupler, with back and forward coupling, 20dB coupling and directivity betterthan 45dB.

This coupler is able to handle up to 50W power.

5. RF switch

A switch was inserted before the P.A. Assuring RF power to be applied only during measurements.A conventional VNA with 0 dBm output is not dangerous, and interference is limited, but with 20Woutput power, care must be taken on RF leakage, operator protection and so on. The RF switch takesplace before the PA because it is not able to handle big amounts of power.

6. Test Setup

This is the setup I used for this test is basically a boosted VNA. Reference data was the same DUTtested with a conventional VNA setup.

Fig. 8 Power VNA test setup



7. D.U.T: device under test

For this test, we need a device that has a linear response for a power as high as 20W and a very bigdynamic in the measurable bandwidth, so I choose a 400 MHz cavity filter. With this device it'seasy to compare the classical and high-power S-parameters, which are identical (no saturation at20W).

Fig. 9 D.U.T.

8. Results

The measurements in transmitted amplitude were very similar than the reference measurements,made with the VNA normal ports (without power) The two curves could hardly be distinguished,even at the limit of the passband of the isolator, and with a great dynamic. See Fig.10 .

Phase measuring results are not bad too. A little error is visible on the smith diagram, but this isusable. There seems to be a constant phase error and an amplitude one on the maximal reflexion.Perhaps a thermal misalignment due to power differences (see Fig. 11). We can see the absorptionat each of the four cavities of this filter set. S11 seems to be bad for the cavities that's mainlybecause there aren't enough points near the resonance frequency, that's why we see a triangle insteadof a circle. Note that the two measurements are very near anyway.



Fig. 10 Scalar mesurement: S21

Fig. 11 Vector S11 reflexion mesurment



V. Transmission Line Transformers

1. Choice of TLTs

The PA transistors have a very low characteristic impedance, input and output. Zin= ( 2 + 1j) is atypical value for LDMOS or bipolar PA transistors. Achieving broadband match to 50 Ohm forsuch low impedances is very hard in practice, and much structures could be (or couldn't be) used.

Aiming broadband performance, the best way to match very different impedances is the use oftransformers. Resonant circuit approaches are only valid for a very narrow bandwidth.

Because of the parasitic impedance of the end connections, or even a little length of wire, the mostconvenient way to use transformers was transmission line transformers (TLT). With transmissionlines, even small length changes don't affect impedance match.

2. Basic principle of TLTs

Transmission line transformers work using the constant impedance of a transmission line. A signalentering one line comes out the same, if care is taken about input and output match.

The output of the line is seen like a Z0 generator, and the input as a Z0 load. So we are likely to putmore of these generators in series, and loads in parallel (see Fig. 12). The problem is the output andinput of one line aren't isolated. So we cannot, in theory connect the lines such a way. In fact we canexploit a few HF-effects to perform this isolation. For example, if the line is exactly ¼ wavelengthlong, then the impedance of the in/out coupling is nearly infinite. But this is only true on onefrequency.



Fig. 12. TLT example, ratio 1: N²

Extending the bandwidth can be done by wounding the lines around ferrite tores, or in air cores, orto pass the lines into ferrite beads. All this tends to add inductance in the common mode impedanceof the transmission line. This is to say, the impedance of the input to output direct coupling.

This impedance has to be as high as possible, but Z >> Z0 is mostly enough to approach theoreticaloperation.

The transformers design I saw used a few techniques used to increase the common modeimpedance:

– ferrite tores

– air winding of the lines

– immersion of the lines in ferrite powder

– resonance attenuation through resistors (flattens the frequency response)

3. Frequency & Bandwidth

The first transformer I tried was a 1:4 for 250 MHz. I tried different places and numbers of theferrite beads and toroidal cores. The transmission lines are made of 25 ohm semi-rigid coax.

The length of the lines, basically ¼ λ, has to be adjusted experimentally. The transformer withoutferrite and TL windings is narrow band. Wounding and ferrites extends the low cut off frequency.Thus it's pretty easy to achieve a 1-100 MHz bandwidth. The 400MHz upper limit is tricky to obtainbecause of the ferrites losses in UHF, so special ferrites are used. This ferrites, as the low-Z semi-rigid coax cables were difficult to obtain, because of the rarity of such components.



4. Input matching

The best possible input matching is essential for the measuring transformer because if thetransformer itself reflects a large amount of power, the measuring return loss will be blended. But atoo good transformer is not worth too, because most of the case the measured impedance is farenough from Z0, so the reflected power from the transformer is negligible in comparison with thisof the DUT.

5. Transformer ratio

The simple transformation ratios are in n², and we need an impedance in order of a few ohms. Thebest choice could be 1/25 (5 coax cables of 10 ohms impedance), but I first made a 1/16 transformerbecause of the availability of coaxes. This results in a Z0= 3,125 Ohms.

6. Using TLTs for PA design

TLTs can be used for a PA, and this approach has a few advantages over a conventional matchingnetwork solution. A large bandwith, and excellent repeatability can be optained, in addition of aneasy obtained low-z match.

Fig. 13 Shows a photograph on one of my early TLT prototypes. This unit is built with differentialmicrostrips. This transformer could easily be built on a conventional thin PCB substrate, perhapswith some milling between traces, and fa ferrite powder immersion of a part of the PCB. The sametransformers and transmission lines could be used for power distribution in PAs with multipletransistors. Bypass capacitors can be places on the TLT lines, featuring a standard SMD solderingprocess for almost all the PA board.

This could lead to a very low-cost PA design, but for design there is a strong need to have a goodsimulation program capable of 2-3D RF simulation. Without this, many prototypes will beconstructed before optimal performances. The arrow shows the position of a little SMD resistor,used to attenuate some resonance, thus flattening the wideband response.

Fig. 13TLT prototype with microstrips



VI. Adapter

1. Principle

To measure Scatter parameters under low-z and high powers, we have to transform impedances.For active devices, like transistors, a DC way must be provided. For a 1-way measurement, the testsetup Fig.14 Is the base line for the interfaces. The transformer permits to inject the RF energy andto measure S11. R3 provides a load, an attenuated output on 50 Ohm (this matching isunidirectional and the damping is 12dB). C1 and C3 provide DC isolation for the DUT. The biasinginjections are done via L1, L2, C2, and C4. The calibration is done directly on the calibration plane.For the open, we just don't solder the DUT. The short is done soldering the cal plane 1 to GND, theload with a SMD resistor. Trough is done soldering the output and the input of the DUT.

Fig. 14 Principle of 1 way -2 port low-Z mesurement

We could imagine a 2 way adapter, using two transformers, but this requires power switching



somewhere in the VNA, so we didn't plan it. Flipping the DUT is a simpler solution. Thetransformer approach on the second port could make the load resistor simpler to construct (50 ohm),but the matching is a little worse.

2. 1-100 MHz adapter

I built an adapter on the previous principle, for 1- 100 MHz. Two octave-bandwidth is achieved bycarefully designing the TLT, and adjusting the number and type of ferrites (See the component listfor more details).

Fig. 15 Shows a photograph of the test setup. Note that the GND plane is not continous at thetransistor, allowing through coupling for calibration.

The output attenuation is 12 dB. This is due to the resistive match. You have to take this in accountwhen power calibrating the test adapter.

Fig. 15 Photo of the 100 MHz test setup


TLT coax lines

Ferrites

Input DC stuff

TransistorBypass capacity Microstrip


Fig. 16 1-100 MHz test setup schematic



The big problems were the low frequency resonances in the biasing network. L1 and C1form aparallel LC resonant circuit, because the TLT is a short circuit in LF. So the transistor saw a hi-Z atthe resonant frequency (measured sometimes higher than 50 Ohm). With such an impedance, theDUT can oscillate. This problem was solved adding a damping resistor R1 across L1. This lowersthe Q of this parasitic resonance, and thus lowering the impedance seen by the transistor. To test ifthere is no resonance, I first checked the impedance seen from the Gate connection, soldering aSMA at this place. When applying DC power to the DUT, I first checked if no spontaneous wavecomes out on port 2, even LF. If no, you can connect to the analyzer.

The microstrips have a very low impedance, so they are very large. I used a thin substrate FR4(~0,4mm ) who was available in the stock. Because of the lack of etching machines, I simply cut thedouble sided PCB and soldered it on a copper-plate (See Fig.17). This is not a perfect microstrip,but with such a large strip, but the side-effects of the lack of dielectric can be neglected.

The microstrip is 21 mm large for a characteristic impedance of 3,125 Ohm, with the selectedmaterials.

Fig. 17 Microstrip cross section showing construction details

Fig. 18 Material list


HoleSolder Cu-plate (GND plane)

} Double sided PCB

CuFR4, 0,4mm thick

Designator DescriptionR3 15 * 50 Ohm/ 1/4W paralellC4 4,7nF ceramic // 1uF tantalC3 5*1nF// 22nF // 100nF // 1uFC2 4,7nF ceramic // 1uF tantalC1 5*1nF// 22nF // 100nF // 1uFL2B 10 turns 0,4mm wire air wound diameter 10mmL1B 11 turns 0,4mm wire air wound diameter 10mmL1A 15 turns, 0,4mm wire, ferrite core 2323136010L2A 16 turns, 0,4mm wire, ferrite tore 2323136010M3 3,125 Ohm microstrip, any lengthM2 3,125 Ohm microstrip, any lengthM4 3,125 Ohm microstrip, any lengthM1 3,125 Ohm microstrip, any lengthFERRITES 11 * ferrite tore 2323136010 R1 2,2 Ohm 1/4WR2 2,2 Ohm 1/4WCx2 2* 25 Ohm flexible coax 540mm (+ 30mm for soldering)Cx1 3* 25 Ohm flexible coax 540mm (+ 30mm for soldering)Cx4 4* 25 Ohm flexible coax 540mm (+ 30mm for soldering)Cx3 5* 25 Ohm flexible coax 540mm (+ 30mm for soldering)


VII. Transistor biasing circuit

1. Principle of operation

Fig. 19 Bias circuit principle

The basic function block of the DC biasing circuit is shown on Fig. 19. There are two regulationloops, and the most limiting loop wins. The under loop controls the Gate voltage (or Base current ifNPN). The other loop controls Drain current. The diodes make it possible to have a limit for both,but only one loop at time is closed, the other is "comparing" the reference with the actual value,while the corrector saturates.

All the measured and maximal values are available on 4mm connectors. The currents are convertedwith a ratio of 1V/A.

The schematic sheets are provided in the annexes. See the version 3 for the used schematics (V2 is



the etched board who as to be a little modified)

On sheet 1 you can see the main schematic, with the DUT, power and monitor connections, and theDrain current loop.

There is a security for enabling the regulation only if the proper power supplies are applied. Thisensures not to damage the fragile DUT while powering. Supply voltages are compared to a safereference. (see sheet 7)

The on/off switching is also smoothed, and the potentiometers values are filtered, in order to avoidcontact defaults reaching the DUT.



VIII. Measuring & Procedures

To measure efficiently, and to achieve a reasonable accuracy, many calibration and measuring issueshave to be taken in account.

1. Connecting :

See previous chapters for details

– Connect all the test setup like in Fig.8.– Build and connect the transistor test setup (Fig. 16)– Connect DC biasing circuit (Fig. 9) to the transistor test setup

2. Oscillation checking :

The measuring procedure has to take account of the oscillation risk. Do not connect the VNA(especially without attenuators) if you have not checked the oscillations of you test setup.

– Solder all the test setup in measuring position (ie. DUT in the middle), without connecting VNA.Place a 50 Ohm at the input and a Spectrum analyzer at the output

– Apply DC power slowly while checking for oscillations in the analyzer (wide range, 1 kHz upto Ft). If the circuits oscillates, remove power, find the reason for the oscillations.

– Connect the test setup to the VNA.

3. VNA settings for S-parameters measurement

– Set your VNA to single sweep.(SWEEP > → > SINGLE ) Switch on the RF just when needed !Press RESTART to sweep, then switch off.

– Set your View to S11 and S21. Don't try other parameters, the test power VNA setup is only able



to measure 1 path style. Set the graph type to Smith chart for both, and screen split.

– Set the VNA to Extern mode (MODE > EXTERNAL)– Set the start and stop frequencies. Limit this only to the measured isolator bandwidth. Measuring

without a circulator on large BWs can be done inserting an 3 or 6 dB attenuator, but you needmore power at the PA.

– Set your VNA output power (SOURCE > POWER and SOURCE > ATT a1). You have to takeaccount all the test setup gain and loss. The best to do is to solder the setup to through, and tomeasure the output power, + 12 dB.

4. Calibration

– remove DC power and Solder out the DUT and calibrate: 1 path – 2 port S.O.L.T. (Short, open,load, through) at the DUT plane ! Do not apply DC power while calibrating, ensure that Vds isoff too.

– Solder the test setup to the DUT again.

– Set Z0 to the correct value (Measure > complex convers > Set Z0)

– Apply DC power

– Switch on RF.

– Now you can measure, under different bias situations

5. Power sweep measurements

– Set the analyzer in power sweep mode (MODE > SWEEP MODE > POWER SWEEP)

– set single sweep mode (SWEEP > → > SINGLE)

– set the sweep mode to reverse sweep. This is better to limit the time the maximal power isapplied since the sweep ends and stays on the minimal power (SWEEP > → > SWEEPDIRECTION > REV)

– set your output power to a minimal value.

– Solder the test set to Through, and measure the power available at the output. Use this value tooffset your measurement and to set power output. (Loss of test set = 12 dB)

– Set attenuators and power output (SOURCE > POWER and SOURCE > ATT a1)

– you can apply the output offset like this : (SCALE > ADD CONST ) for referring your Y-axis tothe real power.

– Measure



IX. Results

1. Accuracy

The fig. 20 Gives some comparisons between different measurement methods. Note that the S-parameter measurements are close to the datasheet, and the conventional values. This measurementshave been done at low-power to test the accuracy of the low-z measuring system. Residual errors arenot negligible, especially for S11, but acceptable. Note that the datasheet specs are sometimes closerto low-Z results than to the classical measurement system. A power sweep could probably be veryprecise because of supposed linearity of errors.

Note although the negative resistance found with a 3,125 Ohm load on the Drain of the mosfet. Thisshows the tendency of this transistor to instability with low-z loads, which has been experienced ona previous 80 MHz PA project with the same transistor.

Data sheet spec Measurement

@ Z0= 50 Ohm

Measurement

@Z0= 3,125 Ohm

(normed to 50 Ohms)

Z in (50 Ohm load out) 5 – 7,84 j 4,97 – 5,4 j 3,6 – 7,6 j

Z in (3 Ohm load out) -2,6 – 20 j

S11 (conv. to 50 Ohms) 0,837

-162°

0,8

-166°

0,91

-161°

S21 (conv. to 50 Ohms) 13,3

89°

11,8

86°

12

80,5°

S12 (conv. to 50 Ohms) 0,018

-1°

0,018

2,5°

0,017

-2°

S22 (conv. to 50 Ohms) 0,780

-168°

0,74

-169°

0,808

-165°

Fig. 20 Characteristics of the PDS5503S @12V 500mA @ 50 MHz -20 dBm



2. Conversion of S-parameters to other Z0 and validity

In order to convert S-parameters to another Z0, We need to have all four parameters completelycharacterized. Since the aim of the project is to measure the s-parameters in the non-linear region ofthe DUT, this can lead to mis-interpreting the measurements. S21 and S11 are effectively measuredin the desired power. But S12 and S22 don't have any meaning in high-power measurement. So it isrecommended to use the low-Z parameters directly, for example in a simulation software, and to tryto work at that Z0, or to introduce perfect transformers at the input ad output, to convert themeasured S-parameters to the 50 Ohm ones. Another solution to imagine is to use the imperfect-but-approaching S22 and S12.

Converting the S-parameters to another Z0 is done with the formulas Fig. 21, where r is animagination reflexion coefficient, and [I] the unity-matrix.

Given r=Z ' 0−Z 0

Z ' 0Z 0 and [S ]=[S 11 S 12

S 21 S 22] and considering [ I ]=[1 0 0 1 ]

The transformation of the S matrix is made with the following formula:

[S ' ]=[S ' 11 S ' 12

S ' 21 S ' 22]=[S ]−r [ I ]∗[ I ]−r [S ]−1

Which gives, developed:

[S ' ]= 1 1−rS 111−rS 22−r 2 S 12 S 21 [S 11−r 1−rS 22−rS 12 S 21 S 121−r2

S 211−r2 S 22−r 1−rS 11−rS12 S 21]

Fig. 21 Transformation formulas for changing Z0 of S-parameter matrixes

These formulas were used for conversion of the low-Z results Fig. 20 Up to 50 Ohms.

3. Power sweep

Power sweeps can be very accurate, Calibrating is not possible on the ZVRE analyzer, but simplesubstraction of an offset is. So we have to masure the input and output offset.



Fig. 22 Power sweep of the PDS5503S: input offset 50 dB

Fig.22 Shows a power sweep example of a power sweep. The 1dB compression point for thePDS5503S is for 10,2 dBm at the input (added an input offset of 50 dB from the measurng chain)

The two peaks visible in the saturation region are not an transcient error, but stable measurement.These could be the rectifying limit of an harmonic, or another non-linear effect.



X. Conclusion

1. Feasibility

The results showed that it is perfectly possible to acquire amplitudes and phases on actual PAtransistors in their working conditions. Due to late material arrivals for the 400 MHz version, Icould test only up to 100 MHz.

2. Accuracy

The accuracy of the measurements is quite high enough to use the values in a simulation model.Careful interpretation and validation should always be a matter, or the results could be very wrong.

3. Design extension

The test setup described here is in fact a working PA. With simple and low-cost design of the TLTs,one could achieve a very compact, effective, and wide band PA. The TLTs can be made of PCBtraces, with perhaps some ferrite inserts. These inserts can probably be avoided, if careful design ismade. So new PA technologies could be designed.



XI. Bibliography

– A simple analysis of the broadband TLTs (High frequency electronics magazine, feb 2004) :http://www.highfrequencyelectronics.com/Archives/Feb04/HFE0204_Sevick.pdf

– Document of the LGE laboratory (universite de Pau) on using TLTs for high voltage pulsegenerators : http://www.univ-pau.fr/RECHERCHE/LGE/Chapitre3.pdf

– ARRL book: Transmission Line Transformers.

– Zveitor-Analyse mit Leistungswellen, Editions Teubner Studienbücher

– PDS5503S Datasheet, ST semiconductors

– Rohde & Schwartz ZVRE analyzer documentation

– HP 8753 analyzer datasheet



XII. Annexes

Annexe 1 Test setup photograph 33

Annexe 2 High-power VNA setup 34

Annexe 3 various TLT prototypes photos 35

Annexe 4 Various TLT characteristics 36

Annexe 5 Adapter schematic 39

Annexe 6 Biasing circuit schematics 40


Annexe 1 Test setup photograph

Annexe 2 High-power VNA setup

Annexe 3 various TLT prototypes photos

Annexe 4 Various TLT characteristics

Without any ferrites

Wound on ferrite cores,beads added (vs previous measurement)

Optimum for 80 MHz band measuring

Annexe 5 Adapter schematic

Annexe 6 Biasing circuit schematics

scattering parameters measurement setup for pa transistorsf4eru.free.fr/rapport_eads.pdf ·...

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