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Chapter 13 Output Stages and Power Amplifiers
13.1 General Considerations 13.2 Emitter Follower as Power Amplifier 13.3 Push-Pull Stage 13.4 Improved Push-Pull Stage 13.5 Large-Signal Considerations 13.6 Short Circuit Protection 13.7 Heat Dissipation 13.8 Efficiency 13.9 Power Amplifier Classes
1
Why Power Amplifiers?
Drive a load with high power.
Cell phone needs 1W of power at the antenna.
Audio system needs tens to hundreds Watts of power.
Ordinary Voltage/Current amplifiers are not equipped for such applications
2CH 13 Output Stages and Power Amplifiers
Power Amplifier Characteristics
Experiences small load resistance.
Delivers large current levels.
Requires large voltage swings.
Draws a large amount of power from supply.
Dissipates a large amount of power, therefore gets “hot”.
4CH 13 Output Stages and Power Amplifiers
Power Amplifier Performance Metrics
Linearity
Power Efficiency
Voltage Rating
5CH 13 Output Stages and Power Amplifiers
Emitter Follower Large-Signal Behavior I
As Vin increases Vout also follows and Q1 provides more current.
6CH 13 Output Stages and Power Amplifiers
Emitter Follower Large-Signal Behavior II
However as Vin decreases, Vout also decreases, shutting off Q1 and resulting in a constant Vout.
7CH 13 Output Stages and Power Amplifiers
Example: Emitter Follower
11ln
0.5 211
outin T out
L S
in out
VV V I V
R I
V V V mV
11 1
1 1
ln
0.01 390
Cin T C L
S
C in
IV V I I R
II I V mV
8CH 13 Output Stages and Power Amplifiers
Linearity of an Emitter Follower
As Vin decreases the output waveform will be clipped, introducing nonlinearity in I/O characteristics.
9CH 13 Output Stages and Power Amplifiers
Push-Pull Stage
As Vin increases, Q1 is on and pushes a current into RL. As Vin decreases, Q2 is on and pulls a current out of RL.
10CH 13 Output Stages and Power Amplifiers
I/O Characteristics for Large Vin
For positive Vin, Q1 shifts the output down and for negative Vin, Q2 shifts the output up.
Vout=Vin-VBE1 for large +Vin
Vout=Vin+|VBE2| for large -Vin
11CH 13 Output Stages and Power Amplifiers
Overall I/O Characteristics of Push-Pull Stage
However, for small Vin, there is a dead zone (both Q1 and Q2 are off) in the I/O characteristic, resulting in gross nonlinearity.
12CH 13 Output Stages and Power Amplifiers
Small-Signal Gain of Push-Pull Stage
The push-pull stage exhibits a gain that tends to unity when either Q1 or Q2 is on.
When Vin is very small, the gain drops to zero.13CH 13 Output Stages and Power Amplifiers
Sinusoidal Response of Push-Pull Stage
For large Vin, the output follows the input with a fixed DC offset, however as Vin becomes small the output drops to zero and causes “Crossover Distortion.”
14CH 13 Output Stages and Power Amplifiers
Improved Push-Pull Stage
With a battery of VB inserted between the bases of Q1 and Q2, the dead zone is eliminated.
VB=VBE1+|VBE2|
15CH 13 Output Stages and Power Amplifiers
Implementation of VB
Since VB=VBE1+|VBE2|, a natural choice would be two diodes in series.
I1 in figure (b) is used to bias the diodes and Q1.16CH 13 Output Stages and Power Amplifiers
Example: Current Flow I
1 1 2in B BI I I I
Iin
If Vout=0 & β1=β2>>1=> IB1=IB2
17CH 13 Output Stages and Power Amplifiers
Example: Current Flow II
VD1≈VBE → Vout≈Vin
If I1=I2 & IB1≈IB2
→ Iin=0 when Vout=0
18CH 13 Output Stages and Power Amplifiers
Addition of CE Stage
A CE stage (Q4) is added to provide voltage gain from the input to the bases of Q1 and Q2.
19CH 13 Output Stages and Power Amplifiers
Bias Point Analysis
For bias point analysis, the circuit can be simplified to the one on the right, which resembles a current mirror.
The relationship of IC1 and IQ3 is shown above.
VA=0 Vout=0
IC1=[IS,Q1/IS,D1]×[IC3]
20CH 13 Output Stages and Power Amplifiers
Small-Signal Analysis
Assuming 2rD is small and (gm1+gm2)RL is much greater than 1, the circuit has a voltage gain shown above.
AV=-gm4(rπ1||r π2)(gm1+gm2)RL
21CH 13 Output Stages and Power Amplifiers
Output Resistance Analysis
3 4
1 2 1 2 1 2
||1( )( || )
O Oout
m m m m
r rRg g g g r r
If β is low, the second term of the output resistance will rise, which will be problematic when driving a small resistance.
22CH 13 Output Stages and Power Amplifiers
Example: Biasing
CE AV=5Output Stage AV=0.8
RL=8Ωβnpn= 2βpnp=100
IC1≈IC2
1 2
11 2
1 2
1 2
3 4
12
4
6.5|| 133
195
m m
m m
C C
C C
g g
g g
I I mAr rI I A
23CH 13 Output Stages and Power Amplifiers
Problem of Base Current
195 µA of base current in Q1 can only support 19.5 mA of collector current, insufficient for high current operation (hundreds of mA).
24CH 13 Output Stages and Power Amplifiers
Modification of the PNP Emitter Follower
Instead of having a single PNP as the emitter-follower, it is now combined with an NPN (Q2), providing a lower output resistance.
2 3
11out
mR
g
25CH 13 Output Stages and Power Amplifiers
Example: Input Resistance
3
2 3
3 2 3
111
( 1)
Lin in in
Lm
in L
Ri v vr R
g
r R r
26CH 13 Output Stages and Power Amplifiers
Additional Bias Current
I1 is added to the base of Q2 to provide an additional bias current to Q3 so the capacitance at the base of Q2 can be charged/discharged quickly.
27CH 13 Output Stages and Power Amplifiers
Example: Minimum Vin
Min Vin≈0Vout≈|VEB2|
Min Vin≈VBE2
Vout≈|VEB3|+VBE2
28CH 13 Output Stages and Power Amplifiers
HiFi Design
Using negative feedback, linearity is improved, providing higher fidelity.
29CH 13 Output Stages and Power Amplifiers
Short-Circuit Protection
Qs and r are used to “steal” some base current away from Q1 when the output is accidentally shorted to ground, preventing short-circuit damage.
30CH 13 Output Stages and Power Amplifiers
Emitter Follower Power Rating
1 2P
av CCVP I V
Maximum power dissipated across Q1 occurs in the absence of a signal.
,max 1av CCP TV
31CH 13 Output Stages and Power Amplifiers
Example: Power Dissipation
1 10
1 1
1 sinT
I p EE
I EE
P I V t V dtT
P I V
Avg Power Dissipated in I1
32CH 13 Output Stages and Power Amplifiers
Push-Pull Stage Power Rating
4CCP P
avL
VV VPR
2
,max 2CC
avL
VP
R
Maximum power occurs between Vp=0 and 4Vcc/π.33CH 13 Output Stages and Power Amplifiers
Example: Push-Pull Pav
4CCP P
avL
VV VPR
If Vp = 4VCC/π → Pav=0
Impossible since Vp cannot goabove supply (VCC)
34CH 13 Output Stages and Power Amplifiers
Heat Sink
Heat sink, provides large surface area to dissipate heat from the chip.
35CH 13 Output Stages and Power Amplifiers
Thermal Runaway Mitigation
1 21 2
, 1 , 2 , 1 , 2
C CD D
S D S D S Q S Q
I II II I I I
Using diode biasing prevents thermal runaway since the currents in Q1 and Q2 will track those of D1 and D2 as long as theie Is’s track with temperature.
36CH 13 Output Stages and Power Amplifiers
Efficiency
out
out ckt
PP P
Efficiency is defined as the average power delivered to the load divided by the power drawn from the supply
Emitter Follower Push-Pull Stage
2
21
22 2 2
4
P LEF
P L CC P
PEF
CC
V RV R I V V
VV
I1=VP/RL
2
21
22 2 / 4
4
P LPP
P L CC P
PP P CC
V RV R I V V
V V
I1=VP/RL
37CH 13 Output Stages and Power Amplifiers
Example: Efficiency
Emitter FollowerVP=VCC/2
115
Push-PullI1=(VP/RL)/β
14
P
CC P
VV V
38CH 13 Output Stages and Power Amplifiers
Power Amplifier Classes
Class A: High linearity, low efficiency
Class B: High efficiency, low linearity
Class AB: Compromise between Class A and B
39CH 13 Output Stages and Power Amplifiers
Chapter 14 Analog Filters
14.1 General Considerations 14.2 First-Order Filters 14.3 Second-Order Filters 14.4 Active Filters 14.5 Approximation of Filter Response
40
Why We Need Filters
In order to eliminate the unwanted interference that accompanies a signal, a filter is needed.
42CH 14 Analog Filters
Filter Characteristics
Ideally, a filter needs to have a flat pass band and a sharp roll-off in its transition band.
Realistically, it has a rippling pass/stop band and a transition band.
43CH 14 Analog Filters
Example: Filter I
Design goal: Signal to Interference ratio of 15 dB
Solution: A filter with stop band of 40 dB
Given: Adjacent channel Interference is 25 dB above the signal
44CH 14 Analog Filters
Example: Filter II
Given: Adjacent channel Interference is 40 dB above the signal
Design goal: Signal to Interference ratio of 20 dB
Solution: A filter with stop band of 60 dB at 60 Hz
45CH 14 Analog Filters
Example: Filter III
A bandpass filter around 1.5 GHz is needed to reject the adjacent Cellular and PCS signals.
46CH 14 Analog Filters
Filter Transfer Function
Filter a) has a transfer function with -20dB/dec roll-off Filter b) has a transfer function with -40dB/dec roll-off, better
selectivity.
A B
51CH 14 Analog Filters
General Transfer Function
Pn =n’th pole
Zm=m’th zero
1 2
1 2( ) m
m
s Z s Z s ZH s
s P s P s P
52CH 14 Analog Filters
Position of the Poles
Poles on the RHPUnstable (no good)
Poles on the jω axisOscillatory(no good)
Poles on the LHPDecaying
(good)
54CH 14 Analog Filters
Sensitivity
PC
dP dCSP C
Sensitivity measures the variation of a filter parameter due to variation of a filter component.
P=Parameter
C=Component
56CH 14 Analog Filters
First-Order Filters
1
1( ) s zH s
s p
First-order filters are represented by the transfer function shown above.
Low/high pass filters can be realized by changing the relative positions of poles and zeros.
58CH 14 Analog Filters
Second-Order Filters
2
2 2( )
nn
s sH ss s
Q
1,2 211
2 4n
np jQ Q
Second-order filters are characterized by the “biquadratic” equation with two complex poles shown above.
61CH 14 Analog Filters
LC Realization of Second-Order Filters
11 2
1 1 1L sZ
L C s
An LC tank realizes a second-order band-pass filter with two imaginary poles at ±j/(L1C1)1/2 , which implies infinite impedance at ω=1/(L1C1)1/2.
67CH 14 Analog Filters
Example: Tank
ω=0, the inductor acts as a short. ω=∞, the capacitor acts as a short.
68CH 14 Analog Filters
RLC Realization of Second-Order Filters
1 12 2
1 1 1 1 1
R L sZR L C s L s R
11,2 2
1 1 1 11 1
1 1 12 4
Lp jR C R CL C
With a resistor, the poles are no longer pure imaginary which implies there will be no infinite impedance at any ω.
69CH 14 Analog Filters
Voltage Divider Using General Impedances
( )out P
in S P
V ZsV Z Z
Low-pass High-pass Band-pass
70CH 14 Analog Filters
Low-pass Filter Implementation with Voltage Divider
12
1 1 1 1 1
out
in
V RsV R C L s L s R
71CH 14 Analog Filters
Example: Frequency Peaking
12
1 1 1 1 1
out
in
V RsV R C L s L s R
12
Q Peaking existsVoltage gain larger than unity
72CH 14 Analog Filters
Low Pass Circuit Comparison
The circuit on the left has a sharper roll-off at high frequency than the circuit on the right.
Good Bad
73CH 14 Analog Filters
High-pass Filter Implementation with Voltage Divider
2
1 1 12
1 1 1 1 1
out
in
V L C R ssV R C L s L s R
74CH 14 Analog Filters
Band-pass Filter Implementation with Voltage Divider
2
12
1 1 1 1 1
out
in
V L ssV R C L s L s R
75CH 14 Analog Filters
Sallen and Key (SK) Filter: Low-Pass
2
1 2 1 2 1 2 2
11
out
in
Vs
V R R C C s R R C s
11 2
1 2 2
1 CQ R RR R C
1 2 1 2
1n R R C C
Sallen and Key filters are examples of active filters. This particular filter implements a low-pass, second-order transfer function.
76CH 14 Analog Filters
Sallen and Key (SK) Filter: Band-pass
3
4
2 31 2 1 2 1 2 2 2 1 1
4
1
1
out
in
RV RsV RR R C C s R C R C R C s
R
77CH 14 Analog Filters
Sensitivity in Band-Pass SK Filter
1 2 1 2
12
n n n nR R C CS S S S
1 2
2 2
1 1
12
Q QR R
R CS S QR C
1 2
2 2 1 2
1 1 2 1
12
Q QC C
R C R CS S QR C R C
1 1
2 2
QK
R CS QKR C
K=1+R3/R4
79CH 14 Analog Filters
Example: SK Filter Sensitivity I
1 2
1 2
1 12 31 22 3
3
Q QR R
Q QC C
QK
S SK
S SK
KSK
1 2
1 2
R R RC C C
80CH 14 Analog Filters
Example: SK Filter Sensitivity II
22
QK
1
1
2 2
1 1
1 1
2 2
3 14
1 58 4
81.5
QR
QC
QK
R C SR C
R C SR C
S
81CH 14 Analog Filters
Integrator-Based Biquads
2
2 2
out
ninn
V ssV s s
Q
2
21.n n
out in out outV s V s V s V sQ s s
It is possible to use integrators to implement biquadratic transfer functions.
The block-diagram above illustrates how. 82CH 14 Analog Filters
KHN Biquads
2
21.n n
out in out outV s V s V s V sQ s s
5 6
4 5 31
R RR R R
4
4 5 1 1
1.n RQ R R R C
2 6
3 1 2 1 2
1.nRR R R C C
83CH 14 Analog Filters
Versatility of KHN Biquads
2
2 2
out
ninn
V ssV s s
Q
High-Pass
2
2 2 1 1
1.X
ninn
V ssV R C ss s
Q
Band-Pass
2
22 2 1 2 1 2
1.Y
ninn
V ssV R R C C ss s
Q
Low-Pass
84CH 14 Analog Filters
Sensitivity in KHN Biquads
1 2 1 2 4 5 3 6, , , , , , , 0.5nR R C C R R R RS
1 2 1 2, , , 0.5QR R C CS
3 6
3 6 2 2,
5 3 6 1 1
4
2 1
QR R
R R R CQS R R R R CR
4 5
5,
4 51Q
R RR
SR R
85CH 14 Analog Filters
Tow-Thomas Biquad
3 42
1 2 3 4 1 2 2 4 2 3
1.Y
in
R RVV R R R R C C s R R C s R
2 3 4 22
1 2 3 4 1 2 2 4 2 3.out
in
V R R R C sV R R R R C C s R R C s R
Band-Pass Low-Pass
86CH 14 Analog Filters
Example: Tow-Thomas Biquad
2 4 1 2
1n R R C C
Adjusted by R2 or R4
2 4 2
3 1
1 R R CQR C
Adjusted by R3
87CH 14 Analog Filters
Differential Tow-Thomas Biquads
By using differential integrators, the inverting stage is eliminated.
88CH 14 Analog Filters
Simulated Inductor (SI)
1 35
2 4in
Z ZZ Z
Z Z
It is possible to simulate the behavior of an inductor by using active circuits in feedback with properly chosen passive elements.
89CH 14 Analog Filters
Example: Simulated Inductor I
By proper choices of Z1-Z4, Zin has become an impedance that increases with frequency, simulating inductive effect.
2in X YZ R R Cs
90CH 14 Analog Filters
High-Pass Filter with SI
With the inductor simulated at the output, the transfer function resembles a second-order high-pass filter.
2
12
1 1 1 1 1
out
in
V L ssV R C L s L s R
92CH 14 Analog Filters
Example: High-Pass Filter with SI
4 1 Yout
X
RV VR
Node 4 is also an output node
93CH 14 Analog Filters
Low-Pass Filter with Super Capacitor
1
1inX
ZCs R Cs
2 21 1 1
11
out in
in in X
V ZV Z R R R C s R Cs
Low-Pass
94CH 14 Analog Filters
Example: Poor Low Pass Filter
4 2out XV V R Cs
Node 4 is no longer a scaled version of the Vout. Therefore the output can only be sensed at node 1, suffering from a high impedance.
95CH 14 Analog Filters
Frequency Response Template
With all the specifications on pass/stop band ripples and transition band slope, one can create a filter template that will lend itself to transfer function approximation.
96CH 14 Analog Filters
Butterworth Response
2
0
1( )
1n
H j
ω-3dB=ω0, for all n
The Butterworth response completely avoids ripples in the pass/stop bands at the expense of the transition band slope.
97CH 14 Analog Filters
Poles of the Butterworth Response
02 1exp exp , 1,2, ,
2 2kj kp j k n
n
2nd-Order nth-Order98CH 14 Analog Filters
Example: Butterworth Order
The Butterworth order of three is needed to satisfy the filter response on the left.
n=3
22
164.2
nff
2 12f f
99CH 14 Analog Filters
Example: Butterworth Response
12 22 *(1.45 )* cos sin3 3
p MHz j
32 22 *(1.45 )* cos sin3 3
p MHz j 2 2 *(1.45 )p MHz
RC section
2nd-Order SK
100CH 14 Analog Filters
Chebyshev Response
2 2
0
1
1 n
H j
C
The Chebyshev response provides an “equiripple” pass/stop band response.
Chebyshev Polynomial
101CH 14 Analog Filters
Chebyshev Polynomial
Chebyshev Polynomial for n=1,2,3
Resulting Transfer function forn=2,3
10
0 0cos cos ,nC n
10
0cosh cosh ,n
102CH 14 Analog Filters
Example: Chebyshev Attenuation
23
2
0 0
1
1 0.329 4 3
H j
ω0=2π X (2MHz)
A third-order Chebyshev response provides an attenuation of -18.7 dB a 2MHz.
103CH 14 Analog Filters
Example: Chebyshev Order
Passband Ripple: 1 dB Bandwidth: 5 MHz Attenuation at 10 MHz: 30 dB What’s the order?
2 2 1
1 0.03161 0.509 cosh cosh 2n
n>3.66
104CH 14 Analog Filters
Example: Chebyshev Response
1 10 0
2 1 2 11 1 1 1sin sinh sinh cos cosh sinh2 2k
k kp j
n n n n
K=1,2,3,42,3 0 00.337 0.407p j
SK21,4 0 00.140 0.983p j
SK1
105CH 14 Analog Filters
Chapter 15 Digital CMOS Circuits
15.1 General Considerations 15.2 CMOS Inverter 15.3 CMOS NOR and NAND Gates
106
Inverter Characteristic
_X A
An inverter outputs a logical “1” when the input is a logical “0” and vice versa.
108CH 15 Digital CMOS Circuits
NMOS Inverter
The CS stage resembles a voltage divider between RD and Ron1 when M1 is in deep triode region. It produces VDD when M1 is off.
11
( )on
n ox DD TH
RWC V VL
109CH 15 Digital CMOS Circuits
Transition Region Gain
Ideally, the VTC of an inverter has infinite transition region gain. However, practically the gain is finite.
Infinite Transition Region Gain Finite Transition Region Gain
110CH 15 Digital CMOS Circuits
Example: Transition Gain
Transition Region: 50 mV
Supply voltage: 1.8V
1.8 360.05vA V0 – V2: Transition Region
111CH 15 Digital CMOS Circuits
Logical Level Degradation
Since real power buses have losses, the power supply levels at two different locations will be different. This will result in logical level degradation.
112CH 15 Digital CMOS Circuits
Example: Logic Level Degradation
5 25 125V A m mV
Supply A=1.8VSupply B=1.675V
113CH 15 Digital CMOS Circuits
The Effects of Level Degradation and Finite Gain
In conjunction with finite transition gain, logical level degradation in succeeding gates will reduce the output swings of gates.
114CH 15 Digital CMOS Circuits
Small-Signal Gain Variation of NMOS Inverter
As it can be seen, the small-signal gain is the largest in the transition region.
115CH 15 Digital CMOS Circuits
Above Unity Small-Signal Gain
The magnitude of the small-signal gain in the transition region can be above 1.
116CH 15 Digital CMOS Circuits
Noise Margin
Noise margin is the amount of input logic level degradation that a gate can handle before the small-signal gain becomes -1.
117CH 15 Digital CMOS Circuits
Example: NMOS Inverter Noise Margin
21 22out DD n ox D in TH out out
WV V C R V V V VL
Vin=VIH122
in THout
n ox D
V VV
WC RL
1L IL TH
n ox D
NM V VWC RL
1:
H DD IHNM V V 2: 118CH 15 Digital CMOS Circuits
Example: Minimum Vout
To guarantee an output low level that is below 0.05VDD, RD is chosen above.
19
D
n ox DD TH
RWC V VL
119CH 15 Digital CMOS Circuits
Dynamic Behavior of NMOS Inverter Gates
Since digital circuits operate with large signals and experience nonlinearity, the concept of transfer function is no longer meaningful. Therefore, we must resort to time-domain analysis to evaluate the speed of a gate.
It usually takes 3 time constants for the output to transition.120CH 15 Digital CMOS Circuits
Example: Time Constant
219
D Xn DD TH
LR CV V
Assuming a 5% degradation in output low level, the time constant at node X is shown above.
122CH 15 Digital CMOS Circuits
Example: Interconnect Capacitance
Wire Capacitance per Mircon: 50x10-18 F/µm
Total Interconnect Capacitance: 15000X50x10-18 =750 fF
Equivalent to 640 MOS FETs with W=0.5µm, L=0.18µm, Cox =13.5fF/µm2
123CH 15 Digital CMOS Circuits
Power-Delay Product
2DD XPDP V C
The power delay product of an NMOS Inverter can be loosely thought of as the amount of energy the gate uses in each switching event.
124CH 15 Digital CMOS Circuits
Example: Power-Delay Product
23 DD oxPDP V WLC
3
3PLH D X
DD DD D X
T R CPDP I V R C
125CH 15 Digital CMOS Circuits
Drawbacks of the NMOS Inverter
Because of constant RD, NMOS inverter consumes static power even when there is no switching.
RD presents a tradeoff between speed and power dissipation.
126CH 15 Digital CMOS Circuits
Improved Inverter Topology
A better alternative would probably have been an “intelligent” pullup device that turns on when M1 is off and vice versa.
127CH 15 Digital CMOS Circuits
Improved Falltime
This improved inverter topology decreases falltime since all of the current from M1 is available to discharge the capacitor.
128CH 15 Digital CMOS Circuits
CMOS Inverter
A circuit realization of this improved inverter topology is the CMOS inverter shown above.
The NMOS/PMOS pair complement each other to produce the desired effects.
129CH 15 Digital CMOS Circuits
CMOS Inverter Small-Signal Model
1 2 1 2||outm m O O
in
vg g r r
v
When both M1 and M2 are in saturation, the small-signal gain is shown above.
130CH 15 Digital CMOS Circuits
Switching Threshold
The switching threshold (VinT) or the “trip point” of the inverter is when Vout equals Vin.
If VinT =Vdd/2, then W2/W1=µn/µp
131CH 15 Digital CMOS Circuits
Example: VTC
As the PMOS device is made stronger, the VTC is shifted to the right.
W2
133CH 15 Digital CMOS Circuits
Noise Margins
NML =VIL
NMH =Vdd-VIH
1 2 1 2211 3
dd TH TH dd TH THIL
a V V V V aV VV
aa a
1 2 1 2211 1 3
dd TH TH dd TH THIH
a V V V V aV VV
aa a
1
2
n
p
WL
aWL
VIL is the low-level input voltage at which (δVout/ δVin)=-1
VIH is the high-level input voltage at which (δVout/ δVin)=-1
134CH 15 Digital CMOS Circuits
VIL of a Symmetric VTC
1 12 2 3 1
1 3DD TH DD TH
ILa V V a V a V
Va a
Symmetric VTC: a=1
13 18 4IL DD THV V V
135CH 15 Digital CMOS Circuits
Noise Margins of an Ideal Symmetric VTC
, , 2DD
H ideal L idealVNM NM
136CH 15 Digital CMOS Circuits
Floating Output
1
2
/ 2
/ 2TH DD
TH DD
V V
V V
When Vin=VDD/2, M2 and M1 will both be off and the output floats. 137CH 15 Digital CMOS Circuits
Charging Dynamics of CMOS Inverter
As Vout is initially charged high, the charging is linear since M2 is in saturation. However, as M2 enters triode region the charge rate becomes sublinear.
138CH 15 Digital CMOS Circuits
Charging Current Variation with Time
The current of M2 is initially constant as M2 is in saturation. However as M2 enters triode, its current decreases.
139CH 15 Digital CMOS Circuits
Size Variation Effect to Output Transition
As the PMOS size is increased, the output exhibits a faster transition.
140CH 15 Digital CMOS Circuits
Discharging Dynamics of CMOS Inverter
Similar to the charging dynamics, the discharge is linear when M1 is in saturation and becomes sublinear as M1 enters triode region.
141CH 15 Digital CMOS Circuits
Rise/Fall Time Delay
2 2
22
2
2ln 3 4THL TH
PLHDD TH DD
p ox DD TH
VC VTW V V VC V VL
Rise Time Delay
1 1
11
1
2ln 3 4THL TH
PHLDD TH DD
n ox DD TH
VC VTW V V VC V VL
Fall Time Delay
142CH 15 Digital CMOS Circuits
Example: Averaged Rise Time Delay
222
14AVG p ox DD TH
WI C V VL
2
22
22
/ 2. DD DD THL
PLHDD TH
p ox DD TH
V V VCTW V VC V VL
2 243PLH on LT R C
143CH 15 Digital CMOS Circuits
Low Threshold Improves Speed
The sum of the 1st and 2nd terms of the bracket is the smallest when VTH is the smallest, hence low VTH improves speed.
2 /1 2 /1/
2 /1/ 2 /1
2 /1
2ln 3 4TH THL
PLH HLDD TH DD
p n ox DD TH
V VCTW V V VC V VL
1st Term
2nd Term
144CH 15 Digital CMOS Circuits
Example: Increased Fall Time Due to Manufacturing Error
'1 1 1'
11 1
1|| 2on on ON
n ox DD TH
R R RW WC V VL L
Since pull-down resistance is doubled, the fall time is also doubled.145CH 15 Digital CMOS Circuits
Power Dissipation of the CMOS Inverter
2_
12Dissipation PMOS L DD inP C V f
2_
12Dissipation NMOS L DD inP C V f
2supply L DD inP C V f
146CH 15 Digital CMOS Circuits
Example: Energy Calculation
2
2
2
12
12
stored L DD
dissipated L DD
drawn L DD
E C V
E C V
E C V
147CH 15 Digital CMOS Circuits
Power Delay Product
2 21 1
11
1
2ln 3 4THin L DD TH
DD TH DDn ox DD TH
Vf C V VPDPW V V VC V VL
Ron1=Ron2
148CH 15 Digital CMOS Circuits
Crowbar Current
When Vin is between VTH1 and VDD-|VTH2|, both M1 and M2 are on and there will be a current flowing from supply to ground.
150CH 15 Digital CMOS Circuits
NMOS Section of NOR
When either A or B is high or if both A and B are high, the output will be low. Transistors operate as pull-down devices.
151CH 15 Digital CMOS Circuits
Example: Poor NOR
The above circuit fails to act as a NOR because when A is high and B is low, both M4 and M1 are on and produces an ill-defined low.
152CH 15 Digital CMOS Circuits
PMOS Section of NOR
When both A and B are low, the output is high. Transistors operate as pull-up devices.
153CH 15 Digital CMOS Circuits
CMOS NOR
Combing the NMOS and PMOS NOR sections, we have the CMOS NOR.
154CH 15 Digital CMOS Circuits
Example: Three-Input NOR
'outV A B C
Equal Rise & Fall (µn≈2µp)
W1=W2=W3=WW4=W5=W6=6W
155CH 15 Digital CMOS Circuits
Drawback of CMOS NOR
Due to low PMOS mobility, series combination of M3 and M4 suffers from a high resistance, producing a long delay.
The widths of the PMOS transistors can be increased to counter the high resistance, however this would load the preceding stage and the overall delay of the system may not improve.
156CH 15 Digital CMOS Circuits
PMOS NOR Section
When either A or B is low or if both A and B are low, the output is high.
158CH 15 Digital CMOS Circuits
CMOS NAND
Just like the CMOS NOR, the CMOS NAND can be implemented by combining its respective NMOS and PMOS sections, however it has better performance because its PMOS transistors are not in series.
159CH 15 Digital CMOS Circuits
Example: Three-Input NAND
'outV ABC
Equal Rise & Fall (µn≈2µp)
W1=W2=W3=3WW4=W5=W6=2W
160CH 15 Digital CMOS Circuits
NMOS and PMOS Duality
In the CMOS philosophy, the PMOS section can be obtained from the NMOS section by converting series combinations to the parallel combinations and vice versa.
C is in “parallel” with the “series” combination of A and B
C is in “series” with the “parallel” combination of A and B
161CH 15 Digital CMOS Circuits