lte l1 la5.0 algo specifications v1.1

LTE - LA5.0 algorithm specifications

Document number: LTE/BTS/DD/XXXXX Document issue: 01.01 / EN Document status: Draft Date: 12/21/2010

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LTE LA5.0 algorithms specifications, V1. 1

1Passing on or copying of this document, use and communication of its contents not permitted without Alcatel·Lucent written authorization

LTE/BTS/DD/XXXX V01.01/EN Draft 11/05/2010 /449

Mis en forme : Anglais(Royaume-Uni)


Mis en forme : Anglais(Royaume-Uni)PUBLICATION HISTORY 1

12/21/2010 2

Creation 3

Modifications with respect to V1.2 LA4.0.1: 4

1.4 & 3MHz RACH design 5

1.4 & 3MHz PUSCH design 6

1.4 & 3MHz DL design 7

1.4 & 3MHz SRS design 8

9

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10

1. INTRODUCTION .........................................................................................................................10 11

1.1. SCOPE OF THIS DOCUMENT .....................................................................................................10 12

1.2. AUDIENCE FOR THIS DOCUMENT ..............................................................................................10 13

1.3. AUTHOR CONTACTS ...............................................................................................................11 14

2. RELATED DOCUMENTS.................................. ..........................................................................11 15

2.1. REFERENCE DOCUMENTS........................................................................................................11 16

3. SYSTEM PARAMETER ................................... ...........................................................................12 17

3.1. FRAME/SLOT STRUCTURE .....................................................................................................12 18

3.2. SYSTEM PARAMETERS.............................................................................................................12 19

4. OVERVIEW..................................................................................................................................12 20

4.1. RECEIVER STRUCTURE ........................................................................................................12 21

4.1.1 Pilot blocks....................................................................................................................13 22 4.1.2 data blocks....................................................................................................................13 23

4.2. TRANSMITTER STRUCTURE ......................................................................................................14 24

4.3. METHODOLOGY TO DETERMINE BITWIDTHS FOR BLOCK INTERFACES..........................................14 25

5. FRONT END ................................................................................................................................15 26

5.1. INPUT DATA ............................................................................................................................15 27

5.2. RSSI COMPUTATION...............................................................................................................16 28

5.3. ANTENNA-PATH FAILURE DETECTION AND HANDLING .................................................................19 29

5.4. CP REMOVAL .........................................................................................................................20 30

5.5. 7.5KHZ FREQUENCY OFFSET COMPENSATION ..........................................................................21 31

5.6. FFT AND SUB-CARRIER DEMAPPING.........................................................................................22 32

5.7. POST-FFT AGC.....................................................................................................................25 33

6. PUSCH SIMO ..............................................................................................................................29 34

6.1. PILOT CHANNEL ESTIMATION ...................................................................................................29 35

6.1.1 Reference signal compensation ...................................................................................30 36 6.1.2 Reference sequence generation...................................................................................31 37 6.1.2.1 generation of the cyclic shift exponentials ................................................................33 38

6.1.2.2 Generation of MnNnxnr qvu <≤= 0),mod()( RSZC, ..........................................33 39

Computer generated sequences ..............................................................................................33 40 6.2. SYNCHRONIZATION .................................................................................................................34 41

6.2.1 Carrier Frequency Offset (CFO) processing.................................................................34 42 6.2.1.1 Frequency offset estimation ......................................................................................34 43 6.2.1.1.1 algorithm description .................................................................................................34 44 6.2.1.1.2 case of ONE TTI latency (NOT implementED) .........................................................37 45 6.2.1.1.3 case of two TTI latency ( CURRENT implementation)..............................................39 46 6.2.1.1.4 fixed point implementation ........................................................................................42 47 6.2.1.2 Frequency offset compensation ................................................................................52 48 6.2.2 Timing offset estimation................................................................................................57 49

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6.3. MEASUREMENTS ....................................................................................................................74 50

6.3.1 Long term averaging.....................................................................................................74 51 6.3.2 Noise and power estimation .........................................................................................77 52 6.3.2.1 introduction................................................................................................................77 53 6.3.2.2 noise variance estimation for scheduled PRB ..........................................................77 54 6.3.2.3 noise variance estimation for un-scheduled PRB .....................................................82 55 6.3.2.4 instantaneous power estimation................................................................................83 56 6.3.2.5 Noise and power absolute values (ecem only) .........................................................84 57 6.3.3 metrics exchange between FPGA and DSP.................................................................85 58 6.3.4 metrics used in L1 processing for demodulation ..........................................................91 59 6.3.5 Speed estimation ..........................................................................................................92 60 6.3.5.1 Theoretical background.............................................................................................92 61 6.3.5.2 practical implementation ...........................................................................................93 62 6.3.6 DTX detection ...............................................................................................................97 63 6.3.7 Post iDFT SNR estimation............................................................................................98 64 6.3.7.1 Theoretical computation ............................................................................................99 65 6.3.7.2 algorithm simplifications ..........................................................................................101 66 6.3.7.3 fallback solution.......................................................................................................102 67

6.4. CHANNEL ESTIMATION ..........................................................................................................102 68

6.4.1 Frequency domain filtering .........................................................................................102 69 6.4.2 Time domain filtering...................................................................................................105 70 6.4.2.1 Algorithm overview..................................................................................................105 71 6.4.2.2 Filters coefficients computation...............................................................................107 72 6.4.2.3 Filter application ......................................................................................................112 73

6.5. DEMODULATION ....................................................................................................................114 74

6.5.1 Frequency domain equalizer ......................................................................................114 75 6.5.2 iDFT ............................................................................................................................123 76 6.5.3 QAM demapping and LLR computation......................................................................127 77 6.5.4 De-interleaving, de-rate matching and H-ARQ recombination ...................................133 78 6.5.5 Decoder ......................................................................................................................133 79

7. PUCCH PROCESSING: ACK-NACK AND SR.................. .......................................................134 80

7.1. CHANGES WITH RESPECT TO LA2.0 .......................................................................................134 81

7.2. GENERAL CONSIDERATIONS...................................................................................................134 82

7.3. FREQUENCY OFFSET ESTIMATION ..........................................................................................135 83

7.4. FREQUENCY OFFSET COMPENSATION ....................................................................................135 84

7.5. TIMING OFFSET ESTIMATION ..................................................................................................136 85

7.6. TIMING OFFSET COMPENSATION ............................................................................................136 86

7.7. ORDER OF FRONT END PROCESSING.....................................................................................137 87

7.8. CHANNEL ESTIMATION AND ACCUMULATION ON RS.................................................................137 88

7.9. CHANNEL ESTIMATION AND ACCUMULATION ON DATA ..............................................................140 89

7.9.1 data accumulation.......................................................................................................140 90 7.9.2 data Descrambling ......................................................................................................142 91

7.10. ACK-NACK DETECTION..........................................................................................................142 92

7.10.1 algorithm description...................................................................................................142 93 7.11. SR DETECTION.....................................................................................................................145 94

7.12. NOISE ESTIMATION ...............................................................................................................148 95

7.12.1 Noise estimation for Ack-Nack....................................................................................148 96 7.12.2 empty pucch................................................................................................................151 97 7.12.3 combining noise estimates from both pucch ..............................................................152 98 7.12.4 long term averaging of noise estimates ......................................................................152 99

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7.13. POWER ESTIMATION .............................................................................................................153 100

7.14. ANTENNA-PATH FAILURE DETECTION AND HANDLING FOR ACK-NACK ON PUCCH ....................154 101

8. PUCCH PROCESSING: CQI AND CQI&ACK-NACK ............. .................................................155 102

8.1. CHANGES WITH RESPECT TO LA2.0 .......................................................................................155 103

8.2. GENERAL CONSIDERATIONS...................................................................................................155 104

8.3. FREQUENCY OFFSET ESTIMATION ..........................................................................................157 105

8.4. FREQUENCY OFFSET COMPENSATION ....................................................................................157 106

8.5. TIMING OFFSET ESTIMATION ..................................................................................................157 107

8.6. TIMING OFFSET COMPENSATION ............................................................................................157 108

8.7. ORDER OF FRONT END PROCESSING.....................................................................................157 109


8.8.1 Channel Estimation: ACK NACK & CQI Mux..............................................................161 111 8.9. DATA ESTIMATION.................................................................................................................161 112

8.9.1 CQI only ......................................................................................................................161 113 The parameter “ o ” defines the size of the shift chosen in order to get 11 bits at the output....163 114

8.9.2 ACK NACK & CQI.......................................................................................................163 115 8.10. NOISE ESTIMATION ...............................................................................................................165 116

8.10.1 CQI or CQI&ACK/NACK .............................................................................................165 117 8.11. POWER ESTIMATION .............................................................................................................168 118

8.12. DATA DECODING...................................................................................................................168 119

8.12.1 descrambling...............................................................................................................168 120 8.12.2 decoding .....................................................................................................................169 121 8.12.3 Reliability Metrics for CQI ...........................................................................................169 122

9. UPLINK CONTROL CHANNELS ON PUSCH................... .......................................................169 123

9.1. CHANGES WITH RESPECT TO LA2.0.......................................................................................169 124

9.2. NUMBER OF QAM SYMBOLS OCCUPIED BY RI AND ACK-NACK ................................................169 125

9.3. SYMBOL EXTRACTION AND LLR COMPUTATION .......................................................................171 126

9.4. CASE OF ONE BIT TRANSMISSION ...........................................................................................171 127

9.4.1 DTX Detection.............................................................................................................172 128 9.4.2 Ack-nack and RI bit detection .....................................................................................177 129

9.5. CASE OF TWO BITS TRANSMISSION ........................................................................................177 130

9.5.1 coding scheme............................................................................................................177 131 9.5.2 soft detection...............................................................................................................178 132

10. CQI ON PUSCH.........................................................................................................................187 133

10.1. CHANGES WITH RESPECT TO LA2.0.......................................................................................187 134

10.2. SYMBOLS EXTRACTION ..........................................................................................................187 135

10.3. NUMBER OF QAM SYMBOLS OCCUPIED BY CQI ......................................................................187 136

10.3.1 General considerations ...............................................................................................187 137 10.3.2 limitations from the DSP/FPGA B interface ................................................................188 138 10.3.3 Number of CQI bits for the different modes................................................................188 139 10.3.3.1 aperiodic Cqi ...........................................................................................................188 140 10.3.3.2 periodic Cqi .............................................................................................................189 141

10.4. CQI RATE DE-MATCHING .......................................................................................................190 142

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10.5. CQI DECODING : CASE OF 12BITS OR MORE ...........................................................................192 143

10.6. CQI DECODING : CASE OF 11BITS OR LESS.............................................................................193 144

10.6.1 decoding .....................................................................................................................193 145 10.6.2 Reliability Metrics ........................................................................................................196 146

11. SOUNDING REFERENCE SIGNALS: ECEM IMPLEMENTATION .... .....................................197 147

11.1. INTRODUCTION .....................................................................................................................197 148

11.2. SRS STRUCTURE .................................................................................................................198 149

11.2.1 general assumptions...................................................................................................198 150 11.2.2 User multiplexing strategy ..........................................................................................200 151 11.2.3 sequence generation ..................................................................................................200 152 11.2.4 cyclic shift based user indexing ..................................................................................202 153 11.2.5 time domain user distribution......................................................................................202 154

11.3. SRS SEPARATION ................................................................................................................203 155

11.3.1 algorithm description...................................................................................................203 156 11.3.2 CAZAC Compensation................................................................................................205 157 11.3.3 zero padding ...............................................................................................................205 158 11.3.4 IDFT ............................................................................................................................206 159 11.3.5 time domain filtering....................................................................................................207 160 11.3.5.1 reduced filter length.................................................................................................207 161 11.3.5.2 margin for timing offset error ...................................................................................209 162 11.3.6 time domain noise removal.........................................................................................209 163 11.3.7 time domain cyclic shift...............................................................................................213 164 11.3.8 IDFT/DFT scaling........................................................................................................214 165 11.3.9 zero removal ...............................................................................................................214 166

11.4. POST-DFT AGC ..................................................................................................................215 167

11.5. SYNCHRONIZATION ...............................................................................................................215 168

11.6. SNR ESTIMATION .................................................................................................................216 169

11.6.1 noise power estimation ...............................................................................................216 170 11.6.2 Signal power estimation..............................................................................................217 171

11.7. SRS ABSOLUTE POWER ........................................................................................................220 172

11.8. SRS CHANNEL AMPLITUDE ESTIMATION .................................................................................221 173

11.9. SRS FREQUENCY DOMAIN CORRELATION...............................................................................222 174

12. SOUNDING REFERENCE SIGNALS: BCEM IMPLEMENTATION.... .....................................223 175

12.1. INTRODUCTION .....................................................................................................................223 176

12.2. BCEM VS ECEM IMPLEMENTATIONS .......................................................................................223 177

12.3. CAZAC COMPENSATION.................................................................................................223 178

12.4. ZERO PADDING .....................................................................................................................224 179

12.5. IDFT .....................................................................................................................................224 180

12.6. TIME DOMAIN FILTERING........................................................................................................225 181

12.7. TIME DOMAIN NOISE REMOVAL ...............................................................................................226 182

12.8. DFT......................................................................................................................................227 183

12.9. IDFT/DFT SCALING...........................................................................................................227 184

12.10. SYNCHRONIZATION ...........................................................................................................228 185

12.11. POST-DFT AGC ..............................................................................................................229 186

12.12. SIGNAL POWER ESTIMATION ..............................................................................................230 187

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12.13. SRS CHANNEL AMPLITUDE ESTIMATION..............................................................................233 188

12.14. SRS FREQUENCY DOMAIN CORRELATION ...........................................................................234 189

13. RACH PROCESSING................................................................................................................235 190

13.1. OVERVIEW ...........................................................................................................................235 191

13.2. SIGNAL STRUCTURE .............................................................................................................235 192

13.2.1 Random Access Preamble Formats ...........................................................................235 193 13.2.2 Baseband RACH Signal..............................................................................................237 194 13.2.3 Large Cell Operation...................................................................................................238 195 13.2.4 Random Access Burst Configuration..........................................................................238 196 13.2.5 Operation in High Mobility Environment .....................................................................240 197 13.2.6 High Level Requirement .............................................................................................241 198

13.3. RACH PROCESSING ........................................................................................................242 199

13.3.1 RACH Front End Processing ......................................................................................242 200 13.3.2 Inputs to RACH Front End Processing .......................................................................244 201 13.3.3 Frontend Bit Selection.................................................................................................245 202 13.3.4 Max Magnitude CALCULation ....................................................................................246 203 13.3.5 Frequency Shift to Baseband .....................................................................................246 204 13.3.6 Upsampling, Filtering and Decimation ........................................................................247 205 13.3.7 Summary of Filtering Parameters ...............................................................................260 206 13.3.8 Sequence Separation for Format 2.............................................................................265 207 13.3.9 Pre 2048-point FFT Scaling........................................................................................265 208 13.3.10 2048-point Forward FFT .........................................................................................266 209 13.3.11 ZC Sequence Extraction .........................................................................................267 210 13.3.12 Scaling of Extracted ZC Sequence .........................................................................268 211

13.4. RACH BACK END PROCESSING ............................................................................................270 212

13.4.1 Overview of Back End Processing..............................................................................272 213 13.4.2 Generate Frequency Domain Root ZC Sequence......................................................272 214 13.4.3 Multiply RACH Signal with ZC Sequence...................................................................273 215 13.4.4 1024-Point FFT ...........................................................................................................274 216 13.4.5 Rescale/Equalize ........................................................................................................274 217 13.4.5.1 Rescale for N=2 case..............................................................................................274 218 13.4.5.2 Rescale for N=4 case..............................................................................................275 219 13.4.5.3 Rescale for N=8 case..............................................................................................276 220 13.4.6 Compute Metrics.........................................................................................................277 221 13.4.6.1 Metrics for Format 0,1 .............................................................................................277 222 13.4.6.2 Metrics for Format 2 ................................................................................................279 223 13.4.7 Compute Histogram....................................................................................................280 224 13.4.8 Analyze CDF...............................................................................................................281 225 13.4.9 Handle Special cases .................................................................................................282 226 13.4.10 Compute FIRst Quartile ..........................................................................................282 227 13.4.11 Compute Threshold.................................................................................................284 228 13.4.12 Compute Reciprocal of Threshold...........................................................................285 229 13.4.13 Peak Detection and Scaling ....................................................................................286 230 13.4.14 Other Rach Outputs ................................................................................................288 231

13.5. LOW MOBILITY THRESHOLD CALCULATIONS..........................................................................289 232

13.5.1 Theory of Threshold Calculation.................................................................................289 233 13.5.2 The N = 1 Case...........................................................................................................290 234 13.5.3 The N = 2 Case...........................................................................................................291 235 13.5.4 Then N=4 Case..........................................................................................................293 236 13.5.5 The N=8 Case............................................................................................................296 237 13.5.6 Calculation of 1st Quartile Value .................................................................................298 238 13.5.7 Threshold Scaling .......................................................................................................300 239 13.5.8 Summary of Threshold Calculations...........................................................................300 240

13.6. INTERPRETATION OF RACH TIME DELAY OFFSET ..................................................................302 241

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13.6.1 Time Delayed RACH Signal .......................................................................................302 242 13.6.2 Time Delay Implementation Issues.............................................................................303 243 13.6.3 Example of RACH Correlation ....................................................................................304 244 13.6.4 Setting the RACH Search Window .............................................................................305 245 13.6.5 Calculation of Search Window Size............................................................................306 246 13.6.6 Summary of Setting the RACH Search Window.........................................................307 247

13.7. SIGNATURE DETECTION ........................................................................................................308 248

13.8. CALCULATION OF N-AVERAGE DETECTION THRESHOLD .........................................................308 249

14. PUSCH MIMO............................................................................................................................313 250

14.1. CHANGES WITH RESPECT TO PREVIOUS RELEASE ...................................................................313 251

14.2. REFERENCE SIGNALS MULTIPLEXING ......................................................................................313 252

14.3. REFERENCE SIGNALS DEMULTIPLEXING..................................................................................314 253

14.3.1 algorithm Overview .....................................................................................................314 254 14.3.2 zero padding ...............................................................................................................315 255 14.3.3 idft / DFT .....................................................................................................................316 256 14.3.4 time domain filtering....................................................................................................319 257 14.3.5 time domain cyclic shift...............................................................................................320 258 14.3.6 DFT .............................................................................................................................321 259 14.3.7 zero removal ...............................................................................................................321 260

14.4. SYNCHRONIZATION ..............................................................................................................322 261

14.4.1 Carrier Frequency Offset (CFO) processing...............................................................322 262 14.4.1.1 Frequency offset estimation ....................................................................................322 263 14.4.1.2 Frequency offset compensation for pilot blocks......................................................322 264 14.4.2 Timing offset estimation..............................................................................................322 265


14.6. MEASUREMENTS ...................................................................................................................323 267

14.6.1 noise estimation for FDE.............................................................................................323 268 14.6.2 noise estimation for time domain MMSE channel estimation.....................................326 269

14.7. DEMODULATION....................................................................................................................329 270

14.7.1 Joint frequency domain equalization and CFO compensation for data blocks...........329 271

15. L1/L2 INTERFACE .................................... ................................................................................340 272

15.1. L1/L2 INTERFACE FOR DL SCHEDULING .................................................................................341 273

15.2. L1/L2 INTERFACE FOR UL SCHEDULING .................................................................................343 274

16. TRANSMITTER ALGORITHMS............................. ...................................................................346 275

16.1. OVERVIEW ...........................................................................................................................346 276

16.1.1 Chapter organization...................................................................................................346 277 16.1.2 Physical Layer Parameters .........................................................................................346 278 16.1.3 Physical Channels And Signals ..................................................................................347 279

16.2. ELEMENTARY ENGINES .........................................................................................................348 280

16.2.1 CRC Encoding ............................................................................................................348 281 16.2.2 Code block Segmentation...........................................................................................348 282 16.2.3 Channel encoding .......................................................................................................349 283 16.2.4 Rate matching.............................................................................................................349 284 16.2.5 Code block concatenation ..........................................................................................349 285 16.2.6 Scrambling ..................................................................................................................350 286 16.2.7 Modulation ..................................................................................................................350 287 16.2.8 Layer mapping ............................................................................................................352 288 16.2.9 Precoding....................................................................................................................353 289

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16.2.10 Power scaling ..........................................................................................................359 290 16.2.11 IFFT processing ......................................................................................................364 291

16.3. TRANSPORT CHANNELS PROCESSING ....................................................................................365 292

16.3.1 DL-SCH and PCH processing ....................................................................................365 293 16.3.2 DCI processing ...........................................................................................................366 294 16.3.3 BCH processing ..........................................................................................................368 295 16.3.4 CFI processing............................................................................................................368 296 16.3.5 HI processing ..............................................................................................................369 297

16.4. PHYSICAL CHANNELS PROCESSING ........................................................................................369 298

16.4.1 PDSCH processing.....................................................................................................369 299 16.4.2 PDCCH processing.....................................................................................................371 300 16.4.3 PBCH processing........................................................................................................373 301 16.4.4 PCFICH processing ....................................................................................................376 302 16.4.5 PHICH processing ......................................................................................................376 303

16.5. SIGNALS PROCESSING ..........................................................................................................379 304

16.5.1 Reference signals processing.....................................................................................379 305 16.5.2 Positioning Reference signals processing..................................................................379 306 16.5.3 Synchronization signals processing............................................................................379 307

17. OPEN ISSUES & FURTHER STUDIES ....................................................................................384 308

17.1. OPEN ISSUES ...................................................................................................................384 309

17.2. FURTHER STUDIES (POST DROP 1) ..................................................................................384 310

18. ANNEX.......................................................................................................................................384 311

ANNEX 1. CALCULATION OF COFF ...............................................................................................384 312

ANNEX 2. PROOF FOR EQUATION (2).........................................................................................385 313

ANNEX 3. CAZAC SEQUENCES .................................... ...............................................................385 314

ANNEX 4. TABLE OF COMPLEX EXPONENTIAL WITH THE RESIDUAL FREQ UENCY OFFSET402 315

ANNEX 5. PRE STORED FILTER FOR T-MMSE PUSCH.............................................................402 316

ANNEX 6. PRE STORED FILTER FOR T-MMSE PUCCH.............................................................405 317

ANNEX 7. SINCOS TABLE (CFO) FOR SIMO, COMMON FOR ALL BANDWID THS .................409 318

ANNEX 8. SINCOS TABLE (CFO) FOR MIMO, COMMON FOR ALL BANDWID THS.................409 319

ANNEX 9. C(K) , 5 TAPS ...................................... ..........................................................................410 320

ANNEX 10. SYNCHRO SEQUENCES FIX POINT IMPLEMENTATION......... .............................429 321

ANNEX 11. DONWLINK FIX POINT POWER OFFSETS ................... ..........................................431 322

ANNEX 12. LUT FOR 7.5KHZ COMPENSATION ........................ ................................................434 323

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ANNEX 13. POWER COMMANDS MEDIATION IN IBTS................... ..........................................443 324

CASE 3 MHZ / 5 MHZ / 10 MHZ / 15MHZ / 20MHZ BANDWIDTH.....................................................443 325

SUMMARY:........................................... ...............................................................................................446 326

CASE 1.4 MHZ BANDWIDTH ............................................................................................................447 327

19. ABREVIATIONS AND DEFINITIONS ....................... ................................................................448 328

19.1. ABREVIATIONS ......................................................................................................................448 329

1. INTRODUCTION 330

This document specifies the algorithms for the 3GPP Long Term Evolution (LTE) uplink baseband 331 receiver to be implemented in the Channel Element partition of the xCEM and derivatives, together 332 with the downlink baseband transmitter. In uplink, the LTE physical layer uses single-carrier 333 frequency division multiple access (SC-FDMA) technology which applies DFT-precoding to 334 OFDMA to reduce peak to average power ratio (PAPR) in the UE. The SC-FDMA scheme still 335 possesses the properties of uplink orthogonality consistent with OFDMA. In downlink, classical 336 OFDMA is used. 337

Single carrier transmission in uplink precodes the modulated QAM symbols with a DFT. The DFT 338 filtered outputs are then mapped to the subcarriers of the scheduled physical resource blocks 339 similar to OFDMA schemes. Requirements imposed by SC-FDMA are similar to the ones of 340 OFDMA receiver. Frequency-domain equalization at the receiver is required. Correction for timing 341 and frequency offsets are required in some dispersive channels and with imperfect timing and 342 frequency tracking. Overall, the SC-FDMA receiver and baseband-processing algorithm have 343 similar characteristics as those in OFDMA. The receiver and transmitter algorithms are specified 344 in functional blocks. For each of the functional blocks, this document provides the following 345 information: 346

� Functional specification 347 � Block diagram 348 � Interface signals definition 349 � Fixed point implementation 350 351

Also note that the algorithms and the interfaces in this document are designed based on march 09 352 3GPP specifications, including the CRs listed in the Verizon Specifications. 353

354

1.1. SCOPE OF THIS DOCUMENT 355

This document specifies the LTE algorithms that are to be implemented in FPGA/DSP baseband 356

receiver and transmitter at e-NodeB for LA3.0 release. Even if some descriptions concern drop 1 in 357

the document, these descriptions are given for information only and this document is dedicated to 358

drop 2. Another document with specific labeling versions is dedicated to drop 1. 359

1.2. AUDIENCE FOR THIS DOCUMENT 360

This document is intended for FPGA/DSP developers who are responsible for implementing the 361 LTE receiver and transmitter. 362 363

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Supprimé : 1. INTRODUCTION 11¶1.1. SCOPE OF THIS DOCUMENT 11¶1.2. AUDIENCE FOR THIS DOCUMENT 11¶1.3. AUTHOR CONTACTS 12¶2. RELATED DOCUMENTS 12¶2.1. REFERENCE DOCUMENTS 12¶3. SYSTEM PARAMETER 13¶3.1. FRAME/SLOT

STRUCTURE 13¶3.2. SYSTEM PARAMETERS 13¶4. OVERVIEW 13¶4.1. RECEIVER STRUCTURE 13¶4.1.1 Pilot blocks 14¶4.1.2 data blocks 14¶4.2. TRANSMITTER STRUCTURE 15¶4.3. METHODOLOGY TO DETERMINE BITWIDTHS FOR BLOCK INTERFACES 15¶5. FRONT END 16¶5.1. INPUT DATA 16¶5.2. RSSI COMPUTATION 17¶5.3. ANTENNA-PATH FAILURE

DETECTION AND HANDLING 20¶5.4. CP REMOVAL 21¶5.5. 7.5KHZ FREQUENCY

OFFSET COMPENSATION 22¶5.6. FFT AND SUB-CARRIER DEMAPPING 23¶5.7. POST-FFT AGC 26¶6. PUSCH SIMO 30¶6.1. PILOT CHANNEL ESTIMATION 30¶6.1.1 Reference signal compensation 31¶6.1.2 Reference sequence generation 32¶6.1.2.1 generation of the cyclic shift exponentials 33¶6.1.2.2 Generation of

Nnxnr qvu = ),mod()( RSZC,

34¶Computer generated sequences 34¶6.2. SYNCHRONIZATION 35¶6.2.1 Carrier Frequency Offset (CFO) processing 35¶6.2.1.1 Frequency offset estimation 35¶6.2.1.1.1 algorithm description 35¶6.2.1.1.2 case of ONE TTI latency (NOT implementED) 38¶6.2.1.1.3 case of two TTI

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1.3. AUTHOR CONTACTS 364

For specific sections in this document, the following author(s) may be contacted. 365

366

Author Initials Contact Location Sections

Moussa Abdi MA [email protected]

Villarceaux All UL algorithms except RACH

Jiang Hao D JHD [email protected]

Shanghai RACH

Yann Leost YLE [email protected]

Villarceaux Transmitter algorithms

Amira Alloum AA [email protected]

Villarceaux CQI demodulation

Fatma Kharrat-Kammoun

FKK [email protected]

Villarceaux Ack-Nack on PUCCH

Slim Chabbouh SC Slim.Chabbouh @alcatel-lucent.com

Villarceaux All fixed point implementation

Carle Lengoumbi CL [email protected]

Villarceaux PUSCH AGC

PUSCH long term measures

367

2. RELATED DOCUMENTS 368

2.1. REFERENCE DOCUMENTS 369

[R1] 3GPP Technical Specifications TS 36.201 (input document) 370




[R5] SRD-7271: SRD channel element 374

[R6] Xilinx DFT v3.1 DS615 December 2, 2009, 375

[R7] Xilinx FFT v7.0, DS260 June 24, 2009 Product Specification 376

[R8] Xilinx turbodecoder: 3GPP LTE Turbo decoder V1.3 (input only) 377

[R9] LTE RRM Uplink Dynamic scheduler, LTE/SYS/DD/029668 378 379

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Supprimé : 2







3. SYSTEM PARAMETER 380

3.1. FRAME/SLOT STRUCTURE 381

The figure below describes the UL frame structure for FDD. The UL frame with normal CP length is 382

of length 10ms and contains 20 slots of length 0.5ms. On each slot, we have 7 SC-FDMA symbols, 383

and the central symbol is a pilot block. One TTI or sub-frame contains 2 slots, and thus two pilot 384

blocks are located at positions 3 and 10 (starting from 0). For further details, see [R2]. 385

386

387 388

Figure 3-1 UL Frame Structure 389 390

3.2. SYSTEM PARAMETERS 391

Transmission parameters are described in [R2]. All the system parameters (CP lengths, etc…) can 392

be found in this reference. Some parameters are still unknown are let as void. Depending on 3GPP 393

completion time, they will be either proprietary or 3GPP aligned. 394

4. OVERVIEW 395

4.1. RECEIVER STRUCTURE 396

The two schemes below describe the functions embedded in the receiver for both pilot and data 397

blocks, together with the notations described in the document. Notice that the schemes below 398

describe only the SIMO receiver. MIMO receiver will be described in the MIMO section. Notice also 399

that RACH is not described in these figures, but this is described in details in the RACH section. 400

1 Frame=10ms 1 sub -frame= 1TTI = 1ms

Symbol-0

UL slot

Symbol-1 Symbol-2 PILOT Symbol-4 Symbol-5 Symbol-6

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4.1.1 PILOT BLOCKS 401

402 403

Figure 4-1 : General receiver structure for a pilot block 404

405

4.1.2 DATA BLOCKS 406

407

( )mx0

pζ

ε∆

pilotˆmH

pilot~mH

mH UL CQI

computation mW

pilot~mH

From other Rx antennas

mx mX

FFT

Sub-carrier demapping

CP removal

Pilot channel

estimation

residual CFO estimation

CFO compensation

From other TTIs (if time

domain accumulation)

Timing estimation

Noise and SNR

estimation

Speed estimation

Channel estimation:

Frequency domain filtering

Timing advance

command pZ

Time domain MMSE filter

computation for channel

estimation

Time domain channel filtering

(MMSE)

To scheduler

ε obtained from previous TTIs

CFO estimation update

Filter computations for frequency

domain equalizer

from RACH

7.5kHz Offset Removal

Post-FFT AGC

RSSI Comput

ation

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408 Figure 4-2 General receiver structure for a data block. 409

410

4.2. TRANSMITTER STRUCTURE 411

This will be compliant to the latest specification of 3GPP. 412

4.3. METHODOLOGY TO DETERMINE BITWIDTHS FOR BLOCK 413

INTERFACES 414

To choose the right number of bits to encode the information on each interface between 2 blocks of 415

algorithms in the UL receiver chain, for a fixed point implementation, a methodology has been 416

chosen and is explained in the following lines. 417

Each block of one or more signal processing algorithm in the UL receiver chain is briefly described 418

in the 2 diagrams for “Pilot blocks” and “Data blocks” at the previous section. 419

Each interface is represented by the output of one block that is also the input of the next one. The 420

aim is then to find the right number of bits to encode the information going though this interface for 421

a fixed point representation, in terms of quantization and saturation. 422

423

This bitwidth must be the smallest one giving the same performance in terms of block error rate 424

(BLER) versus signal to noise ratio (SNR) as the performance obtained with an UL receiver using 425

only floating point representation. A degradation of about 0.1 dB (margin) is however accepted. 426

mY

mS ms

mx

mX

( )mx0

ε

From CFO estimation

CFO compensation

From other Rx antennas

Frequency domain equalization

(MMSE)

From filter computation

IDFT

LLR computation

H-ARQ

Decoder

Control channels extraction (ACK/NACK, CQI)

From data SNR estimation from pilot blocks (not

necessary if LLR are

approximated)

Channel de-interleaver

Rate De-matching

CRC check

FFT


CP removal

7.5kHz Offset Removal

Post-FFT AGC

RSSI computation

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Two types of simulations have been used to reach this goal: simulations with only one interface 427

under test, and at the end a validation being a simulation with all interfaces quantized and 428

saturated with respect to the particular bitwidth found for each interface. 429

Different fading environments have been simulated: AWGN, Pedestrian A, Pedestrian B, Vehicular 430

A and Typical Urban; at different speeds: 3, 50 and 120 kmph. 431

In terms of modulations, QPSK, 16QAM and 64QAM have been run, with one or more PRBs 432

assigned to 1 or 2 users in SIMO or MIMO configurations, using coding rates going from 1/3 up to 433

7/8 for QPSK and 16QAM, and 5/6 for 64QAM. 434

The results presented in this document are given for QPSK and16QAM, but only for “light” 64QAM 435

(not exhaustive simulations and no critical cases studied for 64QAM). 436

437

2’s complements representation is assumed. 438

Following conventions are used in the document: 439

• sn or n is a signed integer of size n (including the sign bit), 440

• un or n is an unsigned integer of size n, 441

• (nI,nQ) is a complex integer (sign), 442

• Log2(n) is the ceil of the 2 basis logarithm of n, 443

• Satn for signed is the saturation between 2n-1-1 and -2n-1 444

• Satn for unsigned is the saturation to 2n-1 445

• (.)>>n is a function from a k bits to (k-n) bits (by suppressing the last n LSB) including 446 rounding (i.e. addition of 1<<(n-1)) if not otherwise specified. 447

• The product of un and um results in un+m 448

• The product of un and sm results in sn+m 449

• The product of sn and sm results in sn+m-1 with saturation between -2n+m-2 and (2n+m-2 -1). 450

• The square of sn results in u2n-2 451

• Qx.y stands for integer representation of the number z/2^y using (x+y) bits, sign bit 452 included. 453

5. FRONT END 454

5.1. INPUT DATA 455

Let us denote N the total number of samples in a TTI. 456

The input sequence in time ( ) ( ) ( ) ( )[ ]1,,...,1,0 −Nxnxxx aaaa K can be split in OFDM symbols as 457

follows: 458

( ) ( ) ( )( )[ ]( ) ( ) ( )( )[ ]( ) ( ) ( )( )[ ]

++

+

13,13,,13,,,13,0

,...,,,,,,,,0

,...,0,0,,0,,,0,0

χχ

χ

FFTaaa

FFTaaa

FFTaaa

Nxlxx

mmNxmlxmx

Nxlxx

KK

KK

KK

where : 459

FFTN is the FFT size 460

m is an index of OFDM symbol ( 130 ≤≤ m ) 461

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( )mχ is the CP size for OFDM symbol number m 462

a is the antenna index 463 464

We further define ( ) ( ) ( ) ( )[ ]mNxmlxmxmx aaaa ,,,,,,,00KK= so that input data sequence can 465

be also denoted ( ) ( ) ( ) ( )[ ]13,,,...,1,0 0000aaaa xmxxx K . 466

467

INPUT QUANTIZATION 468

We suppose that ( )nxa has been normalized so that 469

( )10

2

10

1)(

Ga nxE = where G is the constant radio AGC gain. 470

We also suppose that ( )nxa has been saturated so that: 471

( ) 0.10.1 +<<− nxa 472

473 The radio provides 15 bits (including sign). 474

This means that we will represent the complex number ( )nxa by two 15-bits signed integers (15I , 475

15Q) where fixed point is Q(1.14), so that [-214 , …, 0, 1,…, 214-1] is mapped to [-1.0 , … 0, 1/214 , 476 … , (1-1/214)]. 477 478 All the fixed point implementations given in this document are based on the assumption that the 479 noise at the frontend of the receiver is coded on 3bits. This means that the square root of the 480 variance of the noise occupies a dynamic of 3 bits. 481

5.2. RSSI COMPUTATION 482

Prior to any processing on received (I, Q) samples, RSSI computation is peformed to estimate the 483 globale received wide band energy. On estimated is computed per TTI by the FPGA and 484 transmitted to the DSP that further performs long term averaging and conversion to dBm. 485 486 From an algorithm point of view, RSSI computation simply consists in an averaging of (I²+ Q²) on a 487 TTI basis. To simplify the design, this is done per sub-blocks of equal sizes. This size is chosen as 488 128 in order to ensure commonality of processing for all bandwidth. 489 490 The algorithm is described below. 491 492 General principle : 493

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494 495 The FPGA processing is described below: 496 497

498 499

…

…

(15I,15Q) |.|2

|.|2

15

28

29

128 samples

36

7>>

29 + X

To DSP rescaling and long term averaging

kx

InstRSSI_

From other 128 size blocks of the same TTI

29

>>(X-1)

30

X=4 for 1.4MHz X=5 for 3MHz X=6 for 5MHz X=7 for 10MHz X=8 for 15&20MHz

All samples of a TTI

128 samples

Average over 128 samples

128 samples

Average over 128 samples

… …

Final scaling inside the DSP

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Rescaling inside the DSP 500 501 Three functions need to be implemented inside the DSP: 502

• Rescaling by the number of size 128 blocks inside a TTI (15/ 30 / 60 / 120 / 180 / 240 for 1.4/ 3 / 5 / 503 10 / 15 / 20MHz respectively) 504

• Long term averaging 505 • Conversion to absolute powers 506

507 The rescaling is not explicitly done and is instead directly taken into account as an additional gain 508 inside the conversion to absolute powers. This is the same for all right shifts except the (>>7) that 509 stands for the normalization over the sub-block size. 510 511 Long term averaging 512 513 The long term averaging is described below: 514 515 516 517 518 519 520 521 522 523 524 525 526 527 528 529 530 531 The averaging factor β sets the length of averaging window. Its default value is 0.99, 532

corresponding roughly to an averaging over a 100ms window. β is coded as 32440 in Q0.15, and 533

this should be changed to the desired value if needed. 534 535 Conversion to absolute powers 536 537 The conversion to absolute powers uses the UL RF antenna gain described below. 538 539 The UL RF Antenna Gain provides the RF gain of the receiver for one antenna. This value 540 indicates the relationship between the absolute level (Panalog, in dBm) at the RF connector of the 541 radio module and the power of the signal carried over the digital interface between the Radio 542 Equipment and the Radio Equipment Control. 543 This gain is computed in the digital domain, in dBFS (dB to Full Scale): 544 545 Pdig(dBFS) = 10*log10(mean(I²+Q²)/ (Imax²))) 546 547 Panalog(dBm) = Pdig(dBFS) – (UL RF receiver gain). 548 549 The UL RF Antenna gain is antenna specific, i.e. one gain for main and one gain for Div. 550 Each RRH will be calibrated in factory and shall provide its own static gain per chain. This 551 parameter is used in place of an AGC gain and is static. The granularity of this gain is equal to 552 0.1dB. 553 554 Since the radio sends samples to the digital interface on a format (15I, 15Q), we have Imax=2^14-1 555 reached when all 14 bits are equal to 1, and Imax² is approximately equal to 2^28. 556 557

u30

>>15

u15 u32

s32 s47 s32

u32

β z-1

-

u32

LTRSSI_ InstRSSI_

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On top of UL RF antenna gain, we also need to consider the digital gain that is applied to rescale 558 the noise at the expected level expressed in number of bits at the end of section Erreur ! Source 559 du renvoi introuvable. . 560 561 The processing described above introduces a right shift of >>(X-1) but we omitted to rescale by the 562 number of size 512 blocks inside a TTI (15 / 30 / 60 / 120 / 180 / 240 for 1.4 / 3 / 5 / 10 / 15 / 563

20MHz respectively). Both factors are taken into account with the scaling gain ScalingG when 564

performing the conversion to dBm of LTRSSI_ . 565

566 The global conversion formula is then given by: 567 568

LTRSSI_ (dBm) = 10log10 ( LTRSSI_ ) - ScalingG - 10*log10(Imax²) – (UL RF receiver gain) – 569

(UL Digital gain) 570 571

572

The values of ScalingG are given below for the different bandwidths: 573

574

dB73.2Scaling =G for 1.4 / 3 / 5 / 10 / 20MHz 575

dB48.1Scaling =G for 15MHz 576

5.3. ANTENNA-PATH FAILURE DETECTION AND HANDLING 577

In the case where the CCM detects that an antenna path is in failure, this will deliver zeros to the 578

modem as (I,Q) samples. At the modem side, an antenna failure detection will be performed using 579

the RSSI as computed above. 580

From a modem point of view, we declare that the antenna has been zeroed out if all instantaneous 581

RSSI values as defined in previous section on a whole radio frame are equal to zero for this 582

specific antenna. 583

Once an antenna is declared as zeroed, this state is unchanged during the whole radio frame 584 following the detection. In all cases, the modem will continue processing the incoming signal as for 585 the nominal configuration (i.e. using the zeroed inputs as if they were valid inputs). This means that 586 the FPGAs does not have to select between antenna in failure or not, and just operates on the 587 whole set of available antennas including antennas delivering zeros. 588 589 However, the thresholds used for PUCCH / SR detection are then adjusted to ensure that the false 590 detection is not increased with respect to the target. This threshold update is described in the 591 concerned sections. 592 593 For LA5.0, the following cases need to be distinguished: 594 595

• 4Rx receiver with one zeroed antenna: since 3Rx receiver is not supported, the modem will 596 continue processing the incoming signal as for the nominal 4Rx configuration. To do so, as 597 explained before, it just continues processing all received samples from all antennas 598 without antenna selection. The thresholds used for PUCCH / SR detection are then 599 adjusted (see concerned sections). 600

601 • 4Rx receiver with two zeroed antennas: since 2Rx receiver is supported, the modem will 602

switch from 4Rx mode to 2Rx mode. The modem just continues processing all received 603 samples from all antennas without antenna selection However, the values of the different 604 parameters describing the fixed point implementation (shifts, bitwidth, saturations, etc…) 605 will be the ones of the two Rx mode. The thresholds used for PUCCH / SR detection are 606 then adjusted (see concerned sections). 607

608

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Supprimé : 2


Supprimé : Erreur ! Source du renvoi introuvable.

Supprimé : LA4.0.1







609 • 4Rx receiver with three zeroed antennas, equivalent to 2Rx receiver with one zeroed 610

antenna: The modem also continues processing all received samples from all antennas 611 without antenna selection. However, the values of the different parameters describing the 612 fixed point implementation (shifts, bitwidth, saturations, etc…) will be either the ones of the 613 single Rx mode if it is supported, or the ones of the initial non-broken antenna configuration 614 (2 or 4) if single Rx mode is not supported.The thresholds used for PUCCH / SR detection 615 are then adjusted (see concerned sections). 616

617 618

619

5.4. CP REMOVAL 620

For OFDM symbol of index m, the sequence length before CP removal can be written: 621

( ) ( )( ) ( ) ( )[ ]mNxmxmmxmx ,1,0,0 FFT0 −−= KKχ 622

Notice that all symbols do not have the same CP length. 623

CP removal consists of removing the first ( )mχ samples of this sequence to keep only the 624

sequence: 625

( ) ( ) ( )[ ]mNxmxmx ,1,0 FFT −= K 626

This is a function that is common to all users and it is done once every OFDM symbol. Since there 627 is one FFT per Rx antenna, CP removal is done once per antenna. 628

629

630

Figure 5-1 CP removal for a given Rx antenna (one such process ing is applied for each Rx antenna) 631 632

CP REMOVAL INTERFACE DEFINITION 633

The interface signals and their specifications for this block are given in the next table below. 634

635

Signal Name Type Format I/O Size Description

N Integer uX (see below)

I 1 Number of samples in one

symbol

InSample Complex Integer

(15I,15Q) I N symbol samples (time domain)

OutSample Complex Integer

(15I,15Q) O NFFT symbol samples (time domain)

636 Table 5-1 :CP removal interface definition 637

Sequence ( )mx0 of

length ( )mNFFT χ+

To 7.5kHz FO compensation

Sequence ( )mx of length FFTN

CP removal

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638

X equals 8 for 1.4MHz, 9 for 3MHz, 10 for 5MHz, 11 for 10MHz and 15MHz, and 12 for 639

20MHz. 640

641

5.5. 7.5KHZ FREQUENCY OFFSET COMPENSATION 642

Since a frequency shift of +7.5kHz is applied at the transmitter, this has to be compensated for at 643 the receiver. This compensation is done prior to scaling. Offset is compensated by multiplying by a 644 complex exponential: 645

646

For 1,,0 −= FFTNk K ,

−=

FFTkk N

kjxx

2

2exp.~ π

647

648 This compensation is common to all users and it is done once every OFDM symbol for each 649

antenna. To simplify the notations, we will keep the notation kx instead of kx~ in the rest of the 650

document. 651

652

7.5KHZ COMPENSATION INTERFACE DEFINITION 653


655


NFFT Integer uX (see below)

I 1 FFT size


(15I,15Q) I NFFT

Sub-frame samples (time domain)


(15I,15Q) O NFFT

Sub-frame samples (time domain)

Table 5-2: 7.5kHz compensation interface definition 656 657

X equals 7 for 1.4MHz, 8 for 3MHz, 9 for 5MHz, 10 for 10MHz, and 11 for 15MHz and 658

20MHz. 659

660 661 Fixed point implementation is described below for the real part of the signal. This is similar for the 662 imaginary part. 663 664

Supprimé : 4.0.1

Supprimé : 2







665 Figure 5-2. 7.5kHz compensation for a given Rx antenna (one such processing is appl ied for each Rx 666

antenna) 667

To obtain the values of the complex exponentials, we have to represent ( )( )Nk 2/2sin π− and 668

( )( )Nk 2/2cos π− for 1,,0 −= FFTNk K . This is done through Lut. Since the 5 Mhz LUT is a 669

subset of the 10 Mhz LUT (i.e. every other 10 Mhz LUT sample value is a 5 Mhz LUT sample 670 value), and since the 10 Mhz LUT is itself a subset of the 20 Mhz LUT (i.e. every other 20 Mhz LUT 671 sample value is a 10 Mhz LUT sample value) a single LUT with each sample separated by 672 2*PI/4096 can be used for bands 1.4, 3, 5, 10, and 20MHz. 673

For the 15MHz case, a specific table shall be used. 674

For all bandwidth, since the values of the cos on the second half of the table are the opposite of the 675

values on the first part of the table, and since they are the same for the sin table, then we can store 676

only the first half of the tables. The corresponding values are given in annex Annex 12. 677

678

5.6. FFT AND SUB-CARRIER DEMAPPING 679

This is common to all UEs. For a given Rx antenna, there is a single FFT window shared by all 680 users (synchronized hypothesis). After FFT, the users are separated in the frequency domain by 681 sub-carrier de-mapping (notice that for MIMO we have two users sharing the same sub-carriers). 682 683 There is one FFT per Rx Antenna. The input of the FFT is the output of the CP removal function. 684 When there is no ambiguity, we omit the OFDM symbol index, as well as the antenna index. Since 685 the 7.5kHz has already been taken into account in section 5.5, the FFT can be written for the ith 686 sample: 687

688

∑−

=

−=

1

0

2exp1 FFTN

k FFTk

FFTi N

kijx

NX π 1,,0 −= FFTNi K 689

690 This should be followed by physical de-mapping in order to recover the right order in the frequency 691 allocation (this is explained more in details below). 692

Here 1−=j . 693

(15I,15Q)

(15I,15Q)

>>14 Sat15 15

29

30

16

FFTN times

( )Nkje 2/2π−

kx

Supprimé : 4.0.1

Supprimé : 2







One shall notice that FFTN

1factor is not preserving energy (unlike

FFTN

1) but appropriate shifts 694

are specified in the fix implementation section, in order to compensate this effect. Furthermore the 695

FFTN

1 factor has not to be explicitly computed in the implementation. 696

After front-end FFT, sub-carrier de-mapping permits to separate the users from each other in the 697 frequency domain. This is done by selecting for each user the samples located on the sub-carriers 698 it has been allocated by MAC scheduler. 699 700 Sub-carrier de-mapping is called once per OFDM symbol, per user, and per antenna. 701 For pilot blocks, the output of sub-carrier de-mapping is the input of the CFO derotation. For data 702 blocks, it is the input of the CFO de-rotation process. 703 The output of the sub-carrier de-mapping on antenna a , OFDM symbol m, for user of index u 704

which is allocated ( )uM sub-carriers is denoted by ( )[ ]amuuM

amu

amu XXX ,,1,,0, −= K . When 705

there is no ambiguity, we may keep only tones indexes and write it as [ ]10 −= MXXX K . 706

707 Figure 5-3 FFT and sub-carrier de-mapping for a given Rx anten na (one such processing is applied for each 708

Rx antenna) 709

FFT INTERFACE DEFINITION 710


712


FFTN Integer uX (see below)

I 1 FFT size

TimeSample Complex Integer

(15I,15Q) I NFFT samples (time domain)

FreqSample Complex Integer

see fix implemen

tation section

O NFFT samples (freq domain)

713 Table 5-3 :FFT interface definition 714

X equals 7 for 1.4MHz, 8 for 3MHz, 9 for 5MHz, 10 for 10MHz, and 11 for 15MHz and 20MHz. 715

716

amUX ,1− to synchronization

procedures

amX ,0 to synchronization procedures

From CP removal : Sequence

( )mx of

length FFTN

Front-end

FFT

Sub-carrier de-

mapping

Samples of UE N° (U-1)

Samples of UE N° 0

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Centré,Retrait : Gauche : 0 cm







FIX IMPLEMENTATION 717

The FFT is decomposed into two steps: 718 • Step 1 : Compute FFT 719 • Step 2 : Perform Physical de-mapping 720

721 These three steps are described below: 722

STEP 1 723

Perform classical FFT: 724

∑−

=

−=

1

0

2exp1 FFTN

k FFTk

FFT

ti N

ikjx

NX π , 1,,0 −= FFTNi K 725

STEP 2 726

Perform physical de-mapping: 727

tN

ii FFT

XX2

+= when 1

2,,0 −= FFTN

i K 728

tN

ii FFT

XX2

−= when 1,,

2−= FFT

FFT NN

i K 729

730 The Xilinx’s FFT block will be used as follows: 731

732 733

Figure 5-4 xilinx FFT 734 735

The output of the Xilinx FFT block is on 1)(2log15 ++ FFTN , where we recall that FFTN is the 736

FFT size (this results in 23 bits output for 1.4 MHz, 24 bits output for 3 MHz, 25 bits output for 5 737 MHz, 26 bits output for 10MHz, and 27 bits output for 15 and 20MHz). The adaptation of this output 738 bandwidth to the bit width used in subsequent blocks is done by the post FFT AGC block 739

740

Here is the exhaustive list of Xilinx FFT block parameters (See R[10]): 741

point size 7,8,9,10, 11 direction 1 (forward)

Xilinx

FFT 0… FFTN -1

∑−

=

−=

1

0

2expFFTN

k FFTk

ti

N

ikjxX π

FFTN

inputs

(15I,15Q)

FFTN outputs

(k I, k Q)

exponent

overflow

…

…

Supprimé : 4.0.1

Supprimé : 2







input_bits 15 twiddle_bits 15 architecture 3 (radix-2 single output) rounding 1 (rounding is used) scaling 0 (no scaling) scaling ch NA channels To be filled by architects

Table 5-4 :FFT parameters 742 743

5.7. POST-FFT AGC 744

The FFT output is on 1)(2log15 ++ FFTN bits and has to be reduced to 12 bits, which is the bit 745

width used by forthcoming processing blocks. 746

The post FFT gain control is basically a right shift by a certain number of bits. This number is 747 specific to one UE or control channel. For control channels, the shift is static and hard coded. For 748

data channels, this shift is dynamically determined based on a signal strength computation. 749

For all the REs that contain control channels (PUCCH/SRS) or unallocated PRB, the processing is 750

simple: 751

• Right shift the FFT output signal by a fixed shift, equal to 0 for PUCCH/unallocated PRB and 752 SRS (this 0 shift for SRS is introduced in drop 2) 753

• Saturate the resulting signal to 12 bits for PUSCH, 16bits for PUCCH and 18bits for SRS (12 754

bits for drop 1).. 755

For REs containing data (PUSCH), the processing is to be done on a user basis. The adaptation is 756 a two step processing: 757

• A signal amplitude computation is performed on all PRB allocated to the UE and all receive 758 antennas. This power computation is to be performed on the first OFDM symbol of the TTI. 759

Depending on the resulting computed amplitude, deduce by threshold comparison the shift to 760

apply. This shift is to be applied for the whole TTI to that user. 761

• Apply the computed shift on FFT output signal, for all PRB allocated to the considered UE and 762

then saturate to 12 bits. 763

The resulting block diagram of the post FFT processing, combining control channels processing 764 and data channel processing the following: 765

Supprimé : 4.0.1

Supprimé : 2







766

767

Figure 5-6 - Post FFT AGC stage 768 769 The shift determination block is described in the following diagram: 770

FFT >>1 Sat 17

SRS 18

PUCCH 16

>>0 Sat 18

12

>> 0m Sat 12

>> km

Shift Computation

12

Sat 12

Shift Computation

15

15+log2( FFTN )+1

22 for 1.4MHz, 23 for 3MHz, 24 for 5MHz, , 25 for 10MHz, 26 for 15MHz and 20MHz

17

…

…

…

Sat 16

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 25 for 10MHz

Supprimé : ,







771

Figure 5-5 - Average signal computation block 772 773

Here is the description of each of the elementary blocks in this diagram: 774

• ABS block computes for all input samples their absolute value 775

• SUM block adds all the input sample’s absolute amplitude values 776

• Threshold comparison block compares the input thresholds with the computed sum of signal 777 amplitudes and deduce the shift to apply. 778

The block diagram of the threshold comparison step is given in the following figure: 779

>>6 ABS SUM

Threshold C

omparison

km

TH1

u14

tonesRx..2 NN

u23

TH2

u23

u23

u23

u23

TH3

TH4

TH5

s17 s11 u10 Sat 23

u24

u23

u23

TH6 u23

TH7

Supprimé : 4.0.1

Supprimé : 2







780

Figure 5-60 - AGC threshold comparison 781 782

The input thresholds are given in Table 5.5 hereafter. 783 784

Values Description

TH1 5 Threshold value to decide between shift 0 and 1



Supprimé : 4.0.1

Supprimé : 2











Table 5-5 :Threshold values 785 786 The table above is common to all bands. actually, the current working view is that the shifts are common to 787 all bands 1.4/3/5/10/15/20MHz. However, it should be taken into account that some further studies are 788 currently on-going to determine if we should apply one set of shifts per band or not. In the case where one 789 set of shifts per band is necessary, the corresponding tables will be given in a further version of the 790 document. 791 792

POST-FFT AGC INTERFACE DEFINITION 793


795


N Integer u12 I 1 Number of samples in one sub-frame (one TTI)


(17I,17Q) I N Sub-frame samples


(12I,12Q) O N Sub-frame samples

e (agc_shift) Integer u3 O 1 AGC shift

796 797

6. PUSCH SIMO 798

6.1. PILOT CHANNEL ESTIMATION 799

This function is called for each UE, on each antenna and only for pilot blocks (i.e. OFDM symbols 800 of indexes m=3 and m=10 of each TTI). 801 802 Since we use a feedback loop based CFO processing algorithm, CFO de-rotation is performed 803 right after sub-carrier de-mapping and before any other processing, so that the inputs of pilot 804 channel estimation are the outputs of CFO de-rotation on pilot blocks, and its outputs are sent to 805 the frequency domain filtering and to all measurements. 806

This algorithm computes an estimation of the channel coefficients on the pilot symbol positions 807 based on the observed signal pilot blocks. This estimation is obtained by a multiplication by the 808

expected conjugate pilot sequence in the frequency domain. The obtained estimation is only a 809

rough one and has to be further filtered. 810

Supprimé : 4.0.1

Supprimé : 2







In a non-MIMO configuration, the UE pilots are separated by FDM, i.e. the pilot symbols of different 811 UEs are sent on different tones. Therefore, all the processing should be performed on a user by 812

user basis after sub-carrier de-mapping. For MIMO, specific processing is described in the chapter 813

devoted to MIMO algorithms. 814

6.1.1 REFERENCE SIGNAL COMPENSATION 815

To obtain an estimation of the channel coefficients on pilot tones, a ZF is performed. 816

For pilot tone ( ) 10 −≤≤ uMi , antenna 1,,0 −= RXNa K , user u=0, …, U-1, and OFDM 817

symbol m=3 or 10 (indexes of pilot symbols), the pilot channel estimation writes: 818 819

( ) pilot,,,

*,,

,,

pilot,,,pilot

,,,ˆ

ˆ~uamiumi

umi

uamiuami Hp

p

HH == 820

where umip ,, denotes the i-th pilot symbol of user u on OFDM symbol m. These symbols are 821 CAZAC or computer generated sequences and defined in 3GPP specifications number 36.211. 822

Reference sequence generation is described in next section. 823

When no ambiguity is possible, we omit the user, block, and antenna indexes to keep only tone 824 index and write: 825 826

( ) pilot*pilot

pilot ˆˆ~

iii

ii Hp

p

HH == 827

This sequence will be written ( )[ ]pilot,,,1

pilot,,,0

pilot

,,

~~~uamuMuamuam HHH −= K , or 828

[ ]pilot1

pilot0

pilot ~~~−= MHHH K when no ambiguity is possible. 829

830 Figure 6-1 Pilot channel estimation for a given UE on a given Rx antenna (one such processing is applied for 831

each Rx antenna, each UE, and each pilot block) 832

INTERFACE DEFINITION 833


835 Signal Name Type Format I/O Size Description

M Integer u11 I 1 Number of subcarriers of the Ue

Pilot_Ref Complex Fractional

(12I,12Q) I M Pilot Reference Sequence

Pilot_PerAnt Complex Integer

(12I,12Q) I M

Pilot symbols for one pilot block of the TTI after CFO

compensation pointing of the

Pilot channel estimation (UE u, antenna a, pilot symbol m)

pilot

,,ˆ

uamH From

CFO de-rotation of both pilot

blocks

pilot

,,

~uamH , to CFO

measurement update and frequency domain

channel estimation

Supprimé : 4.0.1

Supprimé : 2







first PRB of considered user.

H_PerAnt Complex Integer

(12I,12Q) O M

Estimated Channel on the considered antenna, for one

block 836 837

Table 6-1: Interface definition for Pilot channel e stimation 838 839

Fixed point implementation is the following. The CAZAC sequence (or computer generated CAZAC 840 sequence, or exponentials representing cyclic shifts) is stored under signed 12 bits format as 841 real(CAZAC) = Sat12(round(real(p)x(1<<11))), and the same for imaginary part. 842 Next plot is for real part, imaginary part can be straightforward deduced. 843 844

845 Figure 6-2 Fixed point implementation for r eference signal compensation (real part only, imagi nary part is 846

the same). 847 848

6.1.2 REFERENCE SEQUENCE GENERATION 849

In this section we describe the generation of the reference sequences. As described in [R2], in 850

each subframe, Mnnrenr vunj

vu <≤= 0),()( ,)(

,αα should be generated. The index u is the 851

group number, the sequence v is the sequence number inside the group, and α is the cyclic shift. 852 The sequence generation is the same as for LA1.0. We give in this version the cyclic shifts for 853

values 12

2 CSnπα = where 11,,0K=CSn . 854

855

For a number of PRB greater than or equal to 3, the sequence Mnnr vu <≤0),(, is constructed 856

from a CAZAC sequence as: 857

MnNnxnr qvu <≤= 0),mod()( RSZC, 858

where the thq root Zadoff-Chu sequence is defined by 859

(12I,12Q)

(12I,12Q)

>>11 Sat 12

M times

12

23

24

13

-

Supprimé : 4.0.1

Supprimé : 2







( ) 10,)1(

exp RSZCRS

ZC

−≤≤

+−= NmN

mmqjmxq

π 860

and the length RSZCN of the Zadoff-Chu sequence is given by the largest prime number such 861

that MN ≤RSZC . 862

The root sequence q is computed from u and v as below: 863

31)1(

)1(21RSZC

2

+⋅=

−⋅++=

uNq

vqq q

864

For a number of PRB lower or equal to 2, the sequence Mnnr vu <≤0),(, is a computer 865

generated sequence: 866

10,)( RSsc

4)(, −≤≤= Mnenr njvu

πϕ 867

where φ(n) = (1,3,-3,-1) which is equivalent to φ(n) = (1,3,5,7). 868 869

On each TTI, the reference signal generation is done by searching the cyclic shift exponential 870

values nje α

and the values of the sequence Mnnr vu <≤0)(, in the lookup tables, and then 871

by the multiplication described below: 872 873

874

875

876

877

878

879

880

881

882

883

884

885

886

887

Figure 6-3 Fixed point implementation for the mult iplication between nominal CAZAC sequence 888 and cyclic shift 889

In the following sections we describe the construction of both Mnnr vu <≤0)(, and the cyclic 890

shift exponential. 891

(12I,12Q)

>>11 Sat 12

M times

12

23

u10

Supprimé : 4.0.1

Supprimé : 2







6.1.2.1 GENERATION OF THE CYCLIC SHIFT EXPONENTIALS 892

For nje α

, 12

2 CSnπα = where 11,,0K=CSn and 0 ≤ n ≤ M – 1. As a result a 12 entry lookup 893

table can be used to represent nje α

. The real and imaginary parts of the complex exponential are 894 coded on 12 bits as defined in the table below. 895

896

CSn 0 1 2 3 4 5

122cos CSnπ )

2047 1774 1024 0 -1024 -1774

122sin CSnπ

0 1024 1774 2047 1774 1024

897

898

899

CSn 6 7 8 9 10 11

122cos CSnπ

-2048 -1774 -1024 0 1024 1774

122sin CSnπ

0 -1024 -1774 -2048 -1774 -1024

900

6.1.2.2 GENERATION OF MnNnxnr qvu <≤= 0),mod()( RSZC, 901

902

COMPUTER GENERATED SEQUENCES 903

10,)( 4)(, −≤≤= Mnenr njvu

πϕ , φ(n) = (1,3,5,7). 904

905 The real and imaginary parts of the complex exponential are coded on 12 bits as defined in the 906 table below. 907 908 909

φ(n) 1 3 5 7 cos(φ(n)*PI/4) 1448 -1448 -1448 1448

sin(φ(n)*PI/4) 1448 1448 -1448 -1448

910

CAZAC SEQUENCES 911 912

Supprimé : 4.0.1

Supprimé : 2







MnNnxnr qvu <≤= 0),mod()( RSZC, where 913

( ) 10 RSZC

/2

)1(2

)1(RSZC −≤≤==

+−+−

NmeemxRSZCN

mmqj

N

mqmj

q

ππ

914 915

Because of periodicity of the exponential, we can generate a single table per value of RSZCN containing the 916

values of

−

RSZCN

pπ2cos and

−

RSZCN

pπ2sin , 1,,0 RSZC −= Np K . 917

Moreover, because of the symmetry )()1( mxmNx qRSZCq =−− , only the first 12/ +RS

ZCN values of the 918

sequence need to be given. These values are given as Q12 in annex. 919 920

6.2. SYNCHRONIZATION 921

6.2.1 CARRIER FREQUENCY OFFSET (CFO) PROCESSING 922

CFO occurs when the carrier frequencies at both Tx and Rx are not equal to the frequency duplex. 923

This has two causes, first a clock difference between the UE and the base station (i.e. the UE’s 924

clock is not perfectly synchronized with the Base station clock due to the inaccuracy of the local 925 oscillators at the UE and the Base Station) and secondly when the Doppler offset becomes non-926

negligible. In this situation, we have to estimate the frequency offset and then to remove its effect. 927

Next plot gives the functional fixed-point implementation for the CFO processing block. It is 928 composed of two sub-blocks, one for the compensation and other for estimation of the CFO. 929

930

931

932

933

934

935

936

6.2.1.1 FREQUENCY OFFSET ESTIMATION 937

6.2.1.1.1 ALGORITHM DESCRIPTION 938

This function is called once per TTI and per transmitting user. This is common to all antennas. 939

In this section, we omit the user index. 940

The algorithm is a Feedback loop based algorithm which works with the correlations of the channel 941 estimates on both DMRS blocks in a TTI. The chosen strategy is to first de-rotate the pilot channel 942 estimation based on the CFO estimated in the previous TTIs, and then perform an estimation of the 943 residual CFO based on these de-rotated pilot channel estimation. 944

The initial value of the CFO can be obtained from the RACH detection (see chapter [7]). 945

The compensation algorithm is described in section 6.2.1.2. The schematic view for DMRS blocks 946 is described below: 947

CFO Compensation

(12Q,12I)

s8 (phase)

(12Q,12I)

CFO Estimation

(12Q,12I)

Supprimé : 4.0.1

Supprimé : 2







948

Figure 6-2 De-rotation of the pilot channel estimation 949

To further simplify the notations, for a given user on a given antenna, let pilot

,

~apH denote the vector 950

containing the compensated frequency domain channel coefficients on pilot tones (after pilot 951

channel estimation) of block number p, p=0, 1, and antenna 1,,0 −= RXNa K . Notice that with 952

previous notations, p=0 corresponds to m=3 and p=1 corresponds to m=10. We have: 953

[ ]pilot,,1

pilot,,0

pilot

,

~~~apMapap HHH −= K 954

Let us also denote SN the number of time domain samples (including CP) separating both pilot 955

blocks. SN is equal to the difference between the index of the first sample of the second pilot 956

block and the index of the first sample of the first pilot block, which is equal to 3840 for 5MHz and 957 7680 for 10MHz. 958

Under the assumption that the channel varies slowly between both pilot blocks, we can write the 959 approximations: 960

aFFT

Sa X

N

NjX ,3,10

.2exp

≈

επ and pilot

,0

pilot

,1

~.2exp

~a

FFT

Sa H

N

NjH

∆≈ επ 961

Here, ε is the effective frequency offset relative to the sub-carrier spacing for the UE of interest, i.e 962

f

f

∆= δε , where kHz15=∆f is the sub-carrier spacing, and fδ is the value of the offset 963

expressed in Hz. The variable ε is the most up-to-date estimate of ε which is obtained from 964

previous TTIs, and εεε ˆ−=∆ is the residual frequency offset taken with respect to ε . 965

We then have the following relation: 966

( ) ∑∑−

=

−

=

∆≈

1

0

2pilot

,0

1

0

pilot

,1

pilot

,0

~.

1.

.2exp

~.

~.

1 RXRX N

aa

RXFFT

SN

aa

H

aRX

HMNN

NjHH

MN

επ 967

So that we can compute an estimation of the residual CFO by: 968

( ) ( )

=

=∆ ∑ ∑∑−

=

−

=

∗−

=

1

0

1

0

pilot,1,

pilot,0,

1

0

pilot

,1

pilot

,0

~.

~.

1angle

2~

.~

.

1angle

2ˆ

RXRX N

a

M

iaiai

RXS

FFTN

aa

H

a

RXS

FFT HHMNN

NHH

MNN

N

ππε969

970

Here the angle is taken in the interval ] [ππ ,− . 971

Notice that the estimation above is based on the DMRS available on the current TTI only. However, 972 if we do not accumulate the statistics in time, this algorithm may not be accurate enough. In the 973 case of 1PRB allocation for example, the number of observations in a TTI is not sufficient to have 974 an accurate estimation of the residual error. The risk is then that, based on a rough first CFO 975 estimation, we add rough residual corrections so that in the end we cannot reach any accurate 976 estimation. Therefore, we have to add some long term averaging of the correlations to ensure a 977 better estimation. 978

mX from sub-carrier

De-mapping pilot

,ˆ

apH

Frequency Offset de-rotation

Pilot

channel estimation

pilot

,~

apHCFO from frequency offset estimation based on previous TTIs

Supprimé : 4.0.1

Supprimé : 2







The long term statistic that we have to use here is ( )εγ ˆ,,LongTerm qp which stands for the CFO 979

compensated correlations defined as: 980

( ) ( )

Ε= εεγ ˆ

~.

~ˆ,,pilot

,

pilot

,LongTerm aq

H

ap HHqp 981

In the equation above, the expectation is taken for a given value of ε , with respect to the channel 982

values only, i.e. no expectation on ε is considered. 983

We will also denote ( )( )nqp TTI,ˆ,,ST εγ the instantaneous value of the quantity above, computed 984

with observations available during TTI(n). This is thus equal to: 985

( )( ) ( )( ) ( )∑ ∑−

=

−

=

∗=

1

0

1

0

pilot,,

pilot,,ST

~.

~

.

1TTI,ˆ,,

RXN

a

M

iaqiapi

RX

nHnHMN

nqp εγ 986

In both equations above, the channels ( )nH ap

pilot

,

~and ( ) nH aq

pilot

,

~are obtained by CFO compensation 987

at TTI(n) with the considered frequency offset ε . In practice, these quantities will be computed per 988 PRB and per antenna, cf. implementation section. 989

The values of p and q stand for pilot indexes, and can be equal to 0 or 1. Notice that for CFO 990

estimation, only ( )εγ ˆ,1,0LongTerm shall be used. However, in section 6.4.2, the values 991

( )εγ ˆ,0,0LongTerm and ( )εγ ˆ,1,1LongTerm shall also be used for speed estimation, so that we define 992

here the general correlation and describe their updates with any value for p and q. 993

Using approximationpilot

,0

pilot

,1

~.2exp

~a

FFT

Sa H

N

NjH

∆≈ επ , we have: 994

( ) ( )

Ε

∆≈

Ε= εεπεεγ ˆ

~.2expˆ

~.

~ˆ,1,02pilot

,0

pilot

,1

pilot

,0LongTerm a

FFT

Sa

H

a HN

NjHH 995

So that we can thus compute an estimation of the residual CFO by: 996

( ){ } ( )

Ε==∆ επ

εγπ

ε ˆ ~

.~

angle2

ˆ,1,0angle2

ˆpilot

,1

pilot

,0LongTerm a

H

aS

FFT

S

FFT HHN

N

N

N 997

Range of estimation of the residual frequency offset 998

This algorithm permits to estimate residual offsets in the range ] [maxmax, εε ∆∆− with maxε∆ such 999

that ] [ππεπ ,.

2 max −∈∆±FFT

S

N

N, i.e 0.0667

2max ==∆S

FFT

N

Nε , corresponding to a residual CFO of 1000

1000± Hz. Therefore, if the initial frequency offset error is less than 1000± Hz, the convergence 1001 is ensured for speeds smaller than 120kmph@2GHz, corresponding a maximum Doppler shift of 1002 222Hz. 1003

Notice that this range does not depend on the system bandwidth since S

FFT

N

N is constant for the 1004

different bandwidths. 1005

1006

In practice, only approximations ( )εγ ˆ,,ˆLongTerm qp of ( )εγ ˆ,,LongTerm qp are available. We explain 1007

below how to compute recursive estimations of ( )εγ ˆ,,ˆLongTerm qp . 1008

Supprimé : 4.0.1

Supprimé : 2







The computation is done in three steps. First, the correlation for the current TTI is done PRB by 1009 PRB inside the FPGA C. These PRB based correlations are transmitted to the DSP. The DSP 1010

averages the different correlations to obtain one correlation per TTI. Notice that averaging over 1011

antennas is done in the DSP as part of the second step. Finally, exponential averaging is 1012 performed to obtain a long term correlation. 1013

Notice that for the long term averaging of the correlations, and more generally for all the 1014 measurements involving long term averaging (noise, timing offset, etc…), the averaging beyond 1015 1ms is performed in the DSP. This implies a delay, so that the parameter estimation based on the 1016 observation corresponding to TTI(n-1) will be available only at TTI(n+TLat-1). This is described in 1017 the figure below. 1018

1019

1020 Figure 6-3: General principle of a long term measurement update 1021

1022

We distinguish two cases, depending on the latency of long term measurements between FPGA 1023 and DSP. These cases correspond respectively to a latency of one or two TTIs, i.e. TLat =1 and 1024 TLat =2. 1025

The current working version is the second one. 1026

6.2.1.1.2 CASE OF ONE TTI LATENCY (NOT IMPLEMENTED) 1027

Let us assume that we are processing the TTI number n, and denote ( )1ˆ −nε 1028

and ( )( )1ˆ,,ˆLongTerm −nqp εγ the most up-to-date estimates of ε and 1029

( )( )1ˆ,,LongTerm −nqp εγ obtained from all observations until TTI number (n-1). Based on 1030

observations of TTI number n, we have to update ( )1ˆ −nε to ( )nε , and ( )( )1ˆ,,ˆLongTerm −nqp εγ 1031

to ( )( )nqp εγ ˆ,,ˆLongTerm . 1032

For TTI number n, new compensated pilot channel estimates ( )nH a

pilot

,0

~ and ( )nH a

pilot

,1

~ are 1033

available. These estimations have been obtained by a de-rotation of pilot channel estimates with 1034

the corresponding frequency offset ( )1ˆ −nε . We can then update the different statistics with the 1035

steps below: 1036

• Update of ( )( )1ˆ,,ˆLongTerm −nqp εγ with exponential averaging: 1037

…….. TTI(n+ TLat-1) TTI(n-1)

DSP

β×

Current long term averaged

estimation

Instantaneousestimation

To DSP

Processing Time

( )β−× 1

Supprimé : 4.0.1

Supprimé : 2







( )( ) ( ) ( )( ) ( )( ) ( )( ) ( )

−+−←− ∑

−

=

1

0

pilot

,

pilot

,LongTermLongTerm

~.

~.

11ˆ,,ˆ1ˆ,,ˆ

RXN

aaq

H

apRX

nHnHMN

nnqpnnqp

βεγβεγ 1038

• Computation of the residual frequency offset estimation based on the 1039

updated ( )( )1ˆ,1,0ˆLongTerm −nεγ : 1040

( ) ( )( ){ } 1ˆ,1,0ˆangle2

ˆLongTerm −=∆ n

N

Nn

S

FFT εγπ

ε 1041

• Update of ( )1ˆ −nε : 1042

( ) ( ) ( )nnn εεε ˆ1ˆˆ ∆+−= 1043

• Update of ( )( )1ˆ,1,0ˆLongTerm −nεγ to ( )( )nεγ ˆ,1,0ˆLongTerm : 1044

( )( ) ( ) ( )( )1ˆ,1,0ˆˆ.

2expˆ,1,0ˆLongTermLongTerm −

∆−= n

N

nNjn

FFT

S εγεπεγ 1045

Notice that this last update of ( )( )nεγ ˆ,1,0ˆLongTerm is not necessary for ( )( )nεγ ˆ,0,0ˆLongTerm and 1046

( )( )nεγ ˆ,1,1ˆLongTerm update since both corresponding channel estimates are taken from the same 1047

OFDM symbol (in other words, they are powers estimates and are thus not sensitive to 1048 multiplications by complex exponentials). 1049

The parameter ( )nβ is the forgetting factor which drives the compromise between adaptation to 1050

the changing environment and the noise removal. This parameter is UE specific and depends on 1051 both the number of PRBs allocated to the UE of consideration in TTI(n) and scheduling rate. The 1052 way to compute this parameter is described in section 6.3.1. 1053

1054

The whole processing is described in the figure below. 1055

1056

Figure 6-4 : Block diagram for enhanced residual f requency offset estimation for a given UE on TTI 1057 number n 1058

Updated

( )nε for

future subframes

From pilot blocks sub-carrier de-mapping of

TTI(n)

( )1ˆ −nε

from previous

subframes

CFO de-rotation+

pilot channel

estimation

Residual CFO Estimation to

compute ( )nε∆

Processing of TTI(n) : channel estimation, noise

and speed estimation, etc…

( )( )1ˆ,1,0ˆLongTerm −nεγfrom previous

subframes

Update

( )( )1ˆ,1,0ˆLongTerm −nεγ

Compute

( )( )nεγ ˆ,1,0ˆLongTerm

from

( )( )1ˆ,1,0ˆLongTerm −nεγ

Updated

( )( )nεγ ˆ,1,0ˆLongTerm

for future subframes

Supprimé : 4.0.1

Supprimé : 2







1059

Figure 6-5: Residual frequency offset estimation for a given UE on a TTI number n (one such processing is 1060 applied for each UE, and each TTI) 1061

1062

6.2.1.1.3 CASE OF TWO TTI LATENCY ( CURRENT 1063

IMPLEMENTATION) 1064

Impacts with regards to one TTI latency 1065

Although the one TTI latency case can not be implemented, we keep its description as it 1066 provides a general understanding of the residual frequency offset estimation algorithm. 1067

Actually, the fpga/DSP timing constraints are such that a frequency offset is computed by the 1068 DSP at TTI (n-2), and used by the FPGA at TTI (n-1). The pilots cross-correlations are 1069

received by the DSP at TTI (n) for long term measurement updates. 1070

Compared to the previous algorithm the impact is thus the following. At TTI (n), the DSP has 1071 available the long term cross-correlation stored in previous TTI and thus derotated with the 1072

absolute frequency offset )1(ˆ −nε and the short term pilot cross-correlation based on pilots 1073

compensated with )2(ˆ −nε . Before exponential averaging, it is necessary for the short term 1074 correlation to be corrected with the residual frequency offset between )2(ˆ −nε and )1(ˆ −nε 1075 i.e. )1(ˆ −∆ nε . Note that the short term derotation should only be performed in the case 1076

where the UE has been granted in TTI (n-2). 1077

1078

Frequency Offset Granularity difference between FPG A and DSP 1079

Within the DSP a small frequency offset granularity is necessary for long term measurement 1080 accuracy and speed estimation. Within the FPGA however, the frequency offset granularity 1081

must be sufficient so that demodulation is not impacted. The higher the granularity, the 1082

smaller the CFO compensation coefficient table. The granularity ratio between DSP and 1083 FPGA has been chosen to be 1/8. This should be taken into account in the CFO estimation 1084 algorithm. From the frequency offset estimated in the DSP )2(ˆ −nε , a quantized value is 1085

transmitted to the FPGA quant( )2(ˆ −nε ). When the next short-term cross-correlations is 1086 received from the FPGA, it should first be derotated with the quantization error )2(ˆ −nε -1087

quant( )2(ˆ −nε ). 1088

1089

Updated

( )( )nεγ ˆ,1,0ˆ LongTerm

Updated ( )nε

Frequency offset estimation

p ilo t

,

~ap

H

CFO compensated pilot

channel estimates ( )nH ap

pilot

,

~

( )( )1ˆ,1,0ˆLongTerm −nεγ from previous sub-

frames

( )1ˆ −nε from

previous sub-frames

e

Supprimé : 4.0.1

Supprimé : 2







Fixed point accuracy considerations 1090

Although mathematically equivallent, frequency offset derotation should only be performed if 1091

the angle to compensate is not null. This recommendation permits to reach a higher accuracy 1092

in fixed point implementation 1093

1094

Algorithm description 1095

We will denote below 1096

( )( ) ( )( ) ( )( ) ( )∑ ∑−

=

−

=

∗==

1

0

1

0

pilot,1,

pilot,0,STST

~.

~

.

1TTI,ˆ,1,0TTI,ˆ

RXN

a

M

iaiai

RX

nHnHMN

nn εγεγ where the 1097

channels have been derotated with the frequency offset appearing in the equation, and: 1098

( )( ) ( )( )nn εγεγ ˆ,1,0ˆˆˆ LongTermLongTerm = 1099

Notice that if for TTI(n) the user is not scheduled, then ( )nε∆ for that user should be set to 0. 1100

Actually, if a user is not scheduled in TTI(n), and scheduled at a later TTI (n+1) or higher, the 1101 full correction including the ( )1ˆ −∆ nε processing calculated at (n-1) will already be applied to 1102

the FPGA CFO compensation for the next scheduled TTI.. 1103

We thus describe below the processing for a subframe number n. 1104

• On each TTI #n where the UE is allocated (FPGA side ): 1105

1. The FPGA has available ( )1ˆ −nε sent from the DSP at TTI(n-1) or earlier 1106

2. Derotation with ( )1ˆ −nε 1107

3. Computation of ( ) ( )( )nn TTI,1ˆST −εγ and send it to the DSP 1108

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Supprimé : 2

Mis en forme : Retrait :Gauche : 2,11 cm, Avec puces+ Niveau : 1 + Alignement : 1,5 cm + Tabulation après : 2,14 cm + Retrait : 2,14 cm,Tabulations : 2,75 cm,Tabulation de liste + Pas à 2,14 cm







1109 • On each TTI #n if the UE was allocated in the TTI(n -1) (DSP side): 1110 1111

1. Short-term correlation derotation 1112 If the user was allocated at TTI(n-1), the DSP has received at that TTI(n-1) from 1113

the FPGA ( ) ( )( )1TTI,2ˆST −− nnεγ . In this case it should compute at TTI(n) 1114

( ) ( )( )1TTI,1ˆST −− nnεγ from ( ) ( )( )1TTI,2ˆST −− nnεγ , quant_error( )1(ˆ −nε ) 1115

and ( )1ˆ −∆ nε as below: 1116

Derot = quant_error( )1(ˆ −nε ) 1117

IF the user was allocated at TTI(n-2) 1118 THEN 1119

Derot += ( )1ˆ −∆ nε 1120

ENDIF 1121 IF Derot != 0 1122 THEN 1123

( ) ( )( ) ( ) ( )( )1TTI,2ˆˆ.

2exp1TTI,1ˆˆ STST −−

−=−− nn

N

DerotNjnn

FFT

S εγπεγ1124

1125 ELSE 1126

( ) ( )( ) ( ) ( )( )1TTI,2ˆˆ1TTI,1ˆˆ STST −−=−− nnnn εγεγ 1127

ENDIF 1128 1129

2. exponential averaging 1130

If the user was allocated at TTI(n-1), the DSP should update ( )( )1ˆˆLongTerm −nεγ 1131

with ( ) ( )( )1TTI,1ˆˆST −− nnεγ 1132

( )( )1ˆˆLongTerm −nεγ = ( )( ) ( ) ( )( )1TTI,1ˆˆ)).(1(1ˆˆ).( STLongTerm −−−+− nnnnn εγβεγβ 1133

1134 3. residual frequency offset estimation 1135

( ) ( )( ){ } 1ˆˆangle2

ˆ LongTerm −=∆ nN

Nn

S

FFT εγπ

ε 1136

4. absolute frequency offset update 1137

( ) ( ) ( )nnn εεε ˆ1ˆˆ ∆+−= 1138

5. Frequency offset quantization 1139

- compute the quantized angle to compensate quant( )(ˆ nε ) and transmit to FPGA 1140

- memorize the quantization error quant_error( )(ˆ nε ). 1141

1142 6. Long-term correlation derotation 1143

Compute ( )( )nεγ ˆˆLongTerm from ( )( )1ˆˆLongTerm −nεγ and ( )nε∆ as below: 1144

IF ( )nε∆ != 0 1145

THEN 1146

( )( ) ( ) ( )( )1ˆˆˆ.

2expˆˆ LongTermLongTerm −

∆−= n

N

nNjn

FFT

S εγεπεγ 1147

ELSE 1148

( )( ) ( )( )1ˆˆˆˆ LongTermLongTerm −= nn εγεγ 1149

ENDIF 1150

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Supprimé : 2

Mis en forme : Retrait :Gauche : 2,11 cm, Avec puces+ Niveau : 1 + Alignement : 1,5 cm + Tabulation après : 2,14 cm + Retrait : 2,14 cm,Tabulations : 2,75 cm,Tabulation de liste + Pas à 2,14 cm

Mis en forme : Retrait :Gauche : 3,11 cm, Numéros +Niveau : 1 + Style denumérotation : 1, 2, 3, … +Commencer à : 1 + Alignement: Gauche + Alignement : 2,5cm + Tabulation après : 3,13cm + Retrait : 3,13 cm,Tabulations :Pas à 3,13 cm




Mis en forme : Retrait :Suspendu : 0,13 cm, Numéros+ Niveau : 1 + Style denumérotation : 1, 2, 3, … +Commencer à : 1 + Alignement: Gauche + Alignement : 2,5cm + Tabulation après : 3,13cm + Retrait : 3,13 cm

Mis en forme ... [2]







6.2.1.1.4 FIXED POINT IMPLEMENTATION 1151




M Integer uX I 1 Number of subcarriers of the Ue

e Integer u3 I 1 Input scaling factor

H_ant1 Complex Integer

(12I,12Q)

I 2 x M

Estimated Channel on the Main antenna (only antenna in

mode 1Rx) after CFO compensation (for both pilot blocks) pointing of the first PRB of considered user.


(12I,12Q)

I 2 x M

Estimated Channel on the Div antenna after CFO

compensation (for both pilot blocks) pointing of the first PRB of considered user.


(12I,12Q)

I 2 x M

Estimated Channel on the 2nd Div antenna before CFO

compensation (for both pilot blocks) pointing of the first PRB of considered user,

(when applicable)


(12I,12Q)

I 2 x M

Estimated Channel on the 3rd Div antenna before CFO,


(when applicable)

Phase Integer s8 I/O 1 Estimated CFO

Gamma_CFO Complex Integer

(31I, 31Q) I/O 1 Long term statistics

1154 Table 6-2 : Interface definition for residual frequ ency Offset Estimation 1155 1156

X = 7, 8, 9, 10, 10, 11 for 1.4, 3, 5, 10, 15, and 20MHz. 1157

1158


Instantaneous correlation computation 1160

The correlation fixed implementation is given below. Only the real part is shown, the computation 1161 for imaginary part is done the same way. 1162

1163 The figure below describes the fixed point implementation of the correlation over one PRB: 1164

1165

Supprimé : 4.0.1

Supprimé : 2







TD_corr(PRB, a) = ( )∑−

=

1

0

pilot

,1,

*pilot

,0,

0 ~.

~M

iaiai HH . (we recall that 120 =M ). 1166

1167

1168 1169 1170

The DSP averages these values on all PRB of the user and on all antennas to obtain the user 1171 based short term time domain correlation (format s26): 1172

1173

(12I,12Q)

(12I,12Q)

23

24+log2(M0) 24

>>2 DSP

22+log2(M0)=26

23

(12I,12Q)

(12I,12Q)

M0 times

24+log2(M0) 24

>>2

DSP

22+log2(M0)=26

antenna 0

antenna NRX -1 if NRX >1

M0 times

...

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Supprimé : 2







1174 1175 1176

1177

In the DSP we define a table containing 1/N in Q4.12 (used to compute the user based ST and LT 1178

noise power(Y) and channel power (X) ) as described below: 1179

const u16_t coeff_prb_divide[NB_MAX_PRB] = 1180

{4095 2047 1365 1024 819 682 512 455 409 341 273 256 227 205 171 164 152 136 1181

128 114 102 91 85 82 76 68 64 57 55 51 51 46 43 }; 1182

And 1/(N*12) (format Q0.16) =( (coeff_prb_divide[N-1]* 0x0AAB) >> 11) 1183

Where 0x0AAB is 1/12 in format Q1.15 1184

Notice that in the table above, only the values corresponding to authorized PRB sizes have been 1185

given. The corresponding PRB sizes are given as: 1186

N={1 2 3 4 5 6 8 9 10 12 15 16 18 20 24 25 27 30 32 36 40 45 48 50 54 60 64 72 1187

75 80 81 90 96}; 1188

Long term time domain correlation per user (format s31) is computed by the DSP using exponential 1189 averaging as described in the figure below: 1190 1191 The DSP long term measurement update is realized as follows (forgetting factors are under Q0.10 1192 format as described in section 6.3.1): 1193

1194

1195

1196

1197

1198

>> X

>>11

Sat26

)(log25 2 N+

)(log24 2 N+

u16 1/(N*12)

)(log40 2 N+

N: number of allocated PRBs for the

N

TD correlation

>>1

>>1

>>1

Sum over NRx antennas

X=1 if NRX=1 X=2 if NRX =2 X=3 if NRX=4 >>1

>>1

>>1

N

.

.

.

Sum over the PRBs

Sum over the PRBs

number of PRB

number of PRB

Antenna 0

Antenna NRX-1 if NRX >1

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Supprimé : 2







eCEM IMPLEMENTATION 1199

1200

1201

1202

1203

1204

1205

1206

( ) ( )( )1TTI,2ˆˆST −− nnεγ

−

FFT

S

N

DerotNj

.2exp π

( ) ( )( )1TTI,1ˆˆST −− nnεγ

( )( )1ˆˆLongTerm −nεγ

Angle Computation

( )nε∆ ( )

∆−

FFT

S

N

nNj

επˆ.

2exp

( )1ˆ −nε To FPGA C quant( )(ˆ nε )

s25

s32

<<2e >>2

-

>>8 >>2

u10

s37 s30

β

Sat 30

Sat 32 >>1

s32

z-1

Angle Quantization

quant_error( )(ˆ nε )

IF Derot != 0

IF )(ˆ nε∆ != 0

Supprimé : 4.0.1

Supprimé : 2







bCEM IMPLEMENTATION 1207

The new long term block avoids saturations that would occur after (<<2e) since now max(e)=7. 1208

1209

In the previous figure, the block named “short term max left shift computation and application” 1210 replaces the legacy shift (<<2e) by a dynamic shift <<(2e- SST,cor ). SST,cor avoids exceeding 32 bits, 1211 it is equivalent to a right shift after the legacy shift (<<2e). As a consequence, short term and long 1212 term correlation may not be at the same scale. To solve this issue, the block named “shift 1213 computation for short term and long term rescaling” computes the values of two shifts RST and RLT 1214 to be respectively applied to short term and long term correlations. 1215

Angle Computation

To FPGA C quant( )

s25

-

z-1

Angle Quantization

quant_error( )

IF Derot != 0

IF != 0

>>10

u10

s32

SAT 32

s32

s32

s32

-

>>RST

LT right shift update

LT correlation rescaling

>>RLT SAT 32

s32

>>1

ST max left shift

computation and

application

Shift computation

for ST-LT rescaling

( ) ( )( )1TTI,1ˆˆST −− nnεγ

( ) ( )( )1TTI,2ˆˆST −− nnεγ

−

FFT

S

N

DerotNj

.2exp π

( )( )1ˆˆLongTerm −nεγ

( )nε∆( )

∆−FFT

S

N

nNj

επˆ.

2exp

( )1ˆ −nε

)(ˆ nε∆

)(ˆ nε

)(ˆ nε

β

Supprimé : 4.0.1

Supprimé : 2







The last block named “long term right shift update and long term correlation rescaling” computes 1216 SLT,cor , the long term value of SST,cor and ensures that SLT,cor is as small as possible (the long term 1217 correlation then occupies the maximum number of bits on 32). 1218

Hereafter, theses 3 blocks are described in details. 1219

1220

Short Term (ST) max left shift computation and appl ication 1221

1222

It replaces the legacy shift (<<2e) by a dynamic shift of (<<(2e- SST,cor)) . The value SST,cor is 1223 computed to stay on 32 bits. The entry signal is s25, 1224

� if e <= 3 , after (<<2e) the signal does not exceed 32 bits. 1225

� If e>= 3, after (<<2e) the signal can exceed 32 bits and the aim of the block is to avoid it. 1226

ST Block Inputs Initialization ST Block Outputs

STγ

XST = 2e Mask=0x40000000

SST,cor

corSTST Se ,2ˆ −<<γ

1227

1228

Shift computation for Short Term (ST) and Long Term (LT) rescaling 1229

Short Term max left shift computation and application

STγ

Two Outputs:

corSTST Se ,2ˆ −<<γ

corSTS ,

Supprimé : 4.0.1

Supprimé : 2







The long term value of SST,cor computed at TTI n is called SLT,cor while the long term value computed 1230 at TTI n-1 is called oldSLT,cor. 1231

If SST,cor and oldSLT,cor are different then the short term correlation (TTI n) and the long term 1232 correlation (TTI n-1) do not have the same scale. The short term and long term correlation have to 1233 be at the same scale to compute their difference. So, the real and imaginary part of ST correlation 1234 computed at TTI n are shifted by : RST = Max ((oldS LT,cor - SST,cor ), 0) and the real and imaginary part 1235 of LT correlation computed at TTI n-1 are shifted by : RLT = Max ((SST,cor - oldS LT,cor ), 0). Only one of 1236 these two shifts is not zero. 1237

1238

Long Term (LT) right shift update and long term cor relation rescaling 1239

SLT,cor = 0 at UE set up. 1240

The long term shift SLT,cor is memorized between 2 runs with the long term correlation LTγ . 1241

The long term shift SLT,cor is the max value between SST,cor (computed at TTI n) and oldSLT,cor (which 1242 is SLT,cor computed at TTI n-1). 1243 If SLT,cor is not zero and both real and imaginary part of LT correlation are not on 32 bits, LT 1244

correlations (Re, Im) are left shifted and SLT,cor is decreased to avoid the loss of precision caused by 1245 a high right shift on the next short term correlation. 1246

1247

Block Inputs Initialization Block Outputs SST,cor oldSLT,cor

LTγ

Mask=0x40000000 SLT,cor NewLTγ

1248

1249

1250

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Supprimé : 2







Equivalent Block 1251

The global block of long term measurement is equiva lent to 1252

1253

1254

This ends the description specific to BCEM implemen tation. The following is common to 1255 eCEM and bCEM implementation. 1256

Phase computation 1257

ε is a s8 , range [-0.0667 … +0.0667[ is represented as [-128 … +127] 1258

We then convert it into a quantized quant( ε ) is a s5, range [-0.0667 … +0.0667] is represented as 1259 [-8 … +8] ] and keep track of the quantization error so that it can be compensated the next time a 1260 short term correlation is received from the fpga. The angle quantization block is as follows: 1261

1262 1263 1264 1265 1266 1267 1268 1269

1270

1271

1272

1273

1274

For quant( )(ˆ nε ) compensation coefficients, we use one look-up (a sub-part of the table shall be 1275

stored due to symetry properties). See Annex 7 for the corresponding values in the SIMO case. 1276

1277

Exponential multiplication 1278

The corrective multiplication by complex exponential of the residual offset multiplication coded on 1279 Q1.16 format is realized as follows 1280

>> SLT,cor << 2e >> 1

>> 4 quant( )(ˆ nε ) )(ˆ nε

<< 4

+ -

quant_error( )(ˆ nε )

Supprimé : 4.0.1

Supprimé : 2







1281

Again, bear in mind that this multiplication is conditional i.e. only performed if the angle to 1282 compensate is not null. 1283

The values of the complex exponentials ( )

∆

FFT

S

N

nNj

επˆ.

2exp are stored in a table as described 1284

in annex 4. 1285

1286

Angle computation 1287

( ) ( )( )( )( )

−−

=∆ )1ˆ,1,0ˆRe(

)1ˆ,1,0ˆIm(arctan

2ˆ

LongTerm

LongTerm

n

n

N

Nn

S

FFT

εγεγ

πε 1288

1289

( )nε and futhermore ( )nε∆ are s8 numbers, represented as [-128; 127] and equivalent to 1290

[ ]ππ ,− . 1291

So the y = K.arctan(x) function can be simply calculated with a table: with axial symetries, only the 1292 values in the first quarter are needed. A 64 values table is enough. 1293

This table is computed as described below: 1294

To follow the specificities of the y = K.arctan(x) function and to have a harmonious distribution of 1295 the values in the full range [0; 64], this table is calculated with 2 different laws. 1296

Under its quantized format, the function is written Y = arctan(i): 1297

When i [ ]1,0 N∈ )arctan128

(1

⋅=

N

iROUNDY

π saturated to 7 bits 1298

1299

When i [ ]1,1 21 −+∈ NN ( )

)1arctan128

(1

21

+−⋅=

N

NiROUNDY

π 1300

saturated to 7 bits 1301

When i 2N= 64=Y (asymptotic value) 1302

1303

(32I,32Q)

(16I,16Q)

>>7 Sat 32 32

41

40

33

>>6

>>6 >>2

>>2

Supprimé : 4.0.1

Supprimé : 2







where N1 = 32, N2 = 64 1304

0

10

20

30

40

50

60

70

0 32

Y(i)

1305

Table 6-3 : Arctan table 1306 1307

Arctan(.) implementation 1308

Arctan( ) argument i is computed by dividing imaginary part by real part, and taking account the 1309 classical axial considerations (if real part is negative, final angle will become angle−π ; if 1310

imaginary part is negative, angle will become angle− ). 1311

The following block shows the computation of Arctan() argument. 1312

1313

1314

1315

Exception: if real part is nul, angle is equal to 0. 1316

We remind that the order function provides the index of the first ‘1’ of its input argument counting 1317 from the MSB to the LSB. 1318

To find the correct value in the table, an offset to be added to the beginning of the table is 1319 calculated as indicated below: 1320

When i [ ]1,0 N∈ i 1321

When i [ ]1,1 21 −+∈ NN 11 NNi +− 1322

Order

Re

Im s

<< ( 6 – MAX ( 0 , 6 – s ) )

>> MAX ( 0 , 6 – s )

y / x

x

y

>>1 ( )( )1ˆ,1,0ˆLongTerm −nεγ

Supprimé : 4.0.1

Supprimé : 2







When i ∞→ 2N 1323

The arctan table is given below for index range [0;63] 1324

index 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15

Atan() 0 1 3 4 5 6 8 9 10 11 12 13 15 16 17 18

1325

Index 16 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32

Atan() 19 20 21 22 23 24 25 25 26 27 28 29 29 30 31 31 32

1326

Index 33 34 35 36 37 38 39 40 41 42 43 44 45 46 47 48

Atan() 33 34 37 40 43 46 49 51 53 54 56 57 58 58 59 59

1327

Index 49 50 51 52 53 54 55 56 57 58 59 60 61 62 63

Atan() 60 60 61 61 61 61 62 62 62 62 62 62 63 63 63

1328

The square root function is determined with an another table of 33 values (Z = k2 with k going from 1329 0 to 32). The value of Z is located by dichotomie in the table and the found index is the searched 1330 squared root. 1331

1332

6.2.1.2 FREQUENCY OFFSET COMPENSATION 1333

This function is called once per OFDM symbol (data or pilot), per antenna, and per transmitting 1334 user. Its inputs/outputs are as follows: 1335

• For pilot blocks, its inputs are the outputs of pilot sub-carrier de-mapping together with the 1336 frequency offset estimation obtained from previous sub-frames or RACH CFO estimate 1337 output, and its outputs are sent to the pilot channel estimation 1338

• For data blocks, its inputs are the output of sub-carrier de-mapping together with the offset 1339 estimation obtained from previous sub-frames, and its outputs are sent to the frequency 1340 domain equalizer. 1341

Based on the available frequency offset estimation, (see section 6.2.1.1), we have to perform de-1342 rotation for both pilot blocks and data blocks. Notice that it would be more accurate to use the 1343 offset estimated from the current sub-frame, but due to implementation constraints, we cannot 1344 afford it. 1345

To perform CFO de-rotation for a given UE with estimated offset ε , we use two de-rotation 1346 processes that can be performed serially or in a single step: 1347

• First, we have to compensate for the global rotation of the OFDM symbol from the 1348

beginning of the TTI. This is done by multiplying the signal by ( )

Ι−

FFT

ˆ..2exp Nmj επ 1349

where ( )mΙ denotes the number of samples in the TTI that were received before the 1350

current block of index m is being treated ( ( )0Ι =0 for the first block). 1351

• Secondly, we have to remove the self Inter-Carrier Interference created by the offset inside 1352 each OFDM symbol. This is done by multi-taps frequency domain filtering as explained 1353 below. 1354

Supprimé : 4.0.1

Supprimé : 2







If L denotes the number of taps (assumed odd), the filter coefficients for multi-taps frequency 1355 domain filtering are computed as: 1356

( ){ }( ){ }

( ){ }( ){ }

−±±=−

−−

−=2

1, 2, 1, 0,

/ˆsin

ˆsin

/ˆexp

ˆexp1)(

FFTFFTFFT

Lk

Nk

k

Nkj

kj

Nkc K

επεπ

επεπ

1357

In practice, these filters are pre-computed and pre-stored for different quantized values of the 1358

frequency offset. The minimum number of taps required from performance point of view is an 1359

increasing function of ε . In practice, this minimum number of taps is equal to L=1 for 301ˆ <ε 1360

(corresponding to 500Hz), and L=3 taps for higher offset (this is sufficient for speeds targeted by 1361

LA3.0). It is up to the hardware designers to use this minimum number of taps or more taps. More 1362

taps can be used in future versions since higher speeds should be targeted. Notice that single tap 1363

filtering reduces to a simple scalar multiplication by an exponential term, which means that since 1364

c(0)=1, the multiplication by c(0) can be omitted. 1365

1366

ε <0.034 (i.e. CFO<500Hz)

>0.034 (i.e. CFO>500Hz)

L 1 tap 3 taps

1367

Minimum number of taps required from performance point of view as a functi on of the offset 1368

The figure below describes the filtering with L=5 coefficients. 1369

1370 Figure 6-6 : Multi-tap frequency domain filtering for a given OF DM symbol (one such processing is applied for 1371

each UE, on each Rx antenna, and on each OFDM block ) 1372 The values of ( )mΙ are given in Annex 7. 1373

Notice that the filtering operation is applied across PRB boundaries (this is possible since in UL all 1374

PRBs allocated to a UE are contiguous), and that the reduction of the filter size is thus applied only 1375

at the edges of the whole UE band, not on each PRB band. 1376

)0(c

)1(c

)2(c

)0(c

)1(−c

)0(c

)1(−c

)1(c

)2(−c

)2(c

)2(−c

Supprimé : 4.0.1

Supprimé : 2







1377 Figure 6-7 : Overall CFO synchronization procedure (one such pro cessing is applied for each UE, on each Rx 1378

antenna, and on each OFDM block) 1379

1380




M Integer uX (see below)

I 1 Number of subcarriers of the

Ue

amX

Complex Integer

(12I,12Q) I/O 2x 14 x M

Data and pilot blocks to be compensated, for Main et Div

pointing of the first PRB of considered user.

Filter_coeff Complex Integer

(12I,12Q) I L Coefficients c(k) of the

frequency domain filter

L Integer u3 I 1 Length of the CFO compensation filter

Phase Integer s8 I 1 Estimated phase

1383

10X

3X

Apply de-rotation

filter

To pilot channel estimation

ε

Sub-carrier de-

mapping

FFT data

( )

−≤2

1 ,

Lkkc

Apply de-rotation

filter

UE Demodulation (to Freq. Domain

MMSE)

( )

Ι−

Nmj επ ˆ..2exp

Block index m

Supprimé : 4.0.1

Supprimé : 2







Table 6-4 :Interface definition for Frequency Offse t Compensation 1384 1385

X=7, 8, 9, 10, 10, 11 for 1.4, 3, 5, 10, 15, 20MHz. 1386

1387

Omitting again the user index, the outputs of the CFO compensation are denoted by 1388

[ ]pilot,,1

pilot,,0

pilot

,ˆˆˆ

apMapap HHH −= K for pilot block, and by [ ]amM

am

am YYY ,1,0 −= K for data blocks. 1389

1390

Figure 6-8 : CFO compensation procedure for pilot block and data block respectively (one such processing is 1391 applied for each UE, on each Rx antenna,) 1392

1393


Performed for the whole sub-frame and for all antennas, it is basically the concatenation of the 1395 three following blocks: 1396

1397 sincos computation 1398

Use the tables in Annex 7 to compensate for the global rotation of the OFDM symbol from the 1399 beginning of the TTI. The sine and cosine values directly represent the multiply factor 1400

( )

Ι−

FFT

ˆ..2exp Nmj επ since the beginning of the TTI for the current block. The parameter 1401

“quant” is the current phase (from -8 to +8) and the parameter “m” is the number of blocks before 1402 the current one since the beginning of the TTI. 1403

For information, the value of ( )mΙ is also given in this annex. 1404

1405

Exponential multiplication 1406

sincos Exponential multiplication

De-rotation filter

(12I,12Q)

C(k) Estimated offset

(12I,12Q)

(13I,13Q)

amX from FFT + sub-

carrier de-mapping output

Filter coefficients for frequency offset compensation

Frequency offset

compensation

pilot

,ˆ

apH to pilot channel

estimation

Filter coefficients for frequency offset compensation

amX from FFT + sub-

carrier de-mapping output Frequency

offset compensation

amY to frequency domain

equalizer

Supprimé : 4.0.1

Supprimé : 2







Performs the multiplication of c(k) by a complex exponential coded on 13 bits (Q1.12) 1407

1408

1409

De rotation filter 1410

This has to be carried for each sub-carrier and antenna. The coefficients are stored in a LUT. 1411

quant(epsilon^) as a s5, range [-0.0667 ... +0.0667] is represented as [-8 ... +8] 1412

1) When there are 1 tap, only c(0)=1.0 (transparent – other being zeroed) 1413

2) otherwise, c(k) are defined in annexe 9 1414

Scale factor is 12 bits so c(k)*X>>12 is using a 12 bit shift. The used format is Q1.12. 1415

1416

Shift + Saturation 13 bits is modeled as follows : 1417

(13I,13Q)

(13I,13Q)

>>12 Sat13 13

25

26

14

M times

(12I,12Q)

(13I,13Q)

L times

24

25+log2(L)

25 >>12

Sat12

13+log2(L)

12

Supprimé : 4.0.1

Supprimé : 2







Y = CLAMP( X >> 12 , -2^12 , +2^12-1 ) 1418

6.2.2 TIMING OFFSET ESTIMATION 1419

Two stages of time synchronization are performed at the UL receiver. The initial coarse time 1420

synchronization is performed through the RACH preamble detection. The initial timing offset is 1421

estimated and computed for each UE during the RACH detection. The time advance command is 1422

sent to the UE to adjust its transmission time. The time advance feedback will assure the coarse 1423

time alignment of multi-user packet arrival at the UL receiver. The initial timing offset estimation 1424

from RACH preamble detection is covered in section 13. 1425

1426

The 2nd stage of the time synchronization is to maintain the uplink synchronization and to minimize 1427

the timing offset when the data or reference signals are transmitted. The received signal timing 1428

from a UE may change due to the user mobility and variation of channel delay profile. The goal of 1429

2nd stage synchronization is to control the UE UL transmission time to achieve the coarse time 1430

alignment of all UE signals. The UL timing control is achieved through a close loop time advance 1431

function. 1432

1433

There are several proposed timing estimation schemes for OFDM systems. For LA3.0, for sake of 1434

simplicity, we kept the Pilot Phase Based algorithm which is described below. For future releases, 1435

we plan either to deliver an enhanced version of timing offset estimation or to add further 1436

processing to this existing pilot phase based algorithm (e.g. add a time domain filtering algorithm 1437

which detects the different delays in the time domain when the bandwidth of a specific user is large 1438

enough). 1439

1440

CHANGES WITH RESPECT TO LA2.0 1441

Until LA2.0, timing offset estimation was done based on DMRS only. For LA3.0 and beyond, we 1442

use both DMRS and SRS. For SRS, we use the frequency domain correlation as described in 1443

section 11.9 These correlations ( )ausCSRST ,, are per user, antenna and transmitted SRS. These 1444

correlations are long term averaged and combined with the DMRS based correlations before angle 1445

computation. 1446

1447

The DMRS based correlations are computed from the compensated pilot channel estimates pilotapiH ,,

~ 1448

for tone i, pilot p and antenna a. 1449

1450

Supprimé : 4.0.1

Supprimé : 2







TIMING OFFSET ESTIMATION ALGORITHM – PILOT PHASE BASED 1451

ALGORITHM 1452

This algorithm is based on the cross-correlations of pilot sub-carriers (both DMRS and SRS) in a 1453

symbol within space s (and s/2 for SRS because of comb shape allocation) which are computed 1454

and accumulated over length in time. The timing offset is estimated from the accumulated cross-1455

correlation output. 1456

The pilot phase based algorithm performance can be improved significantly if the sub-carriers used 1457

to compute the correlation are not adjacent but spaced with a larger-space. If necessary, the 1458

computation complexity can be reduced if we average only a few pair of cross-product value per 1459

slot for a given user no matter how many RB it is assigned. However, we nevertheless recommend 1460

using all the available pilot tones to perform the correlations since the estimation will be more 1461

accurate. 1462

To compute the pilot tones cross-correlation, we perform cross-correlations by averaging for the 1463

UE of interest. 1464

1465

FO Compensated pi lot Ch Resp. from antena 1

Timing Offset

FO Compensated Pi lot Ch Resp. from antena 2

Accumulation

In Time

Cross -Correlation

Averaging

& Combining

Phase

Angle

Computation

e

1466 Figure 6-9 : Block Diagram of Timing Offset Estimation 1467

1468 In the figure above, the cross-correlation are combined from both SRS and DMRS. 1469

1470

The pilot phase based algorithm uses the frequency offset compensated pilot (both SRS and 1471

DMRS) channel estimates. For a given TTI, the timing offset estimate is derived using the average 1472

of the frequency offset compensated pilot outputs in the following 1473

1474

1475

For DMRS, the instantaneous correlation is computed as 1476

( ) ( )∑ ∑ ∑ ∑−

= =

−

=

−−

=

∗+++=

1

0

1

0

112/

0

1

0

pilot,,

pilot,,

0

00

~~1 RXN

a p

M

j

sM

iapsjMiapjMiT HH

MsC 1477

where s is the interval between the pilot sub-carriers, used to compute the cross-correlation, and M 1478

is used as before to denote the total number of sub-carriers allocated to the UE of interest. In 1479

practice, s=6 is the recommended value. As for Frequency-offset estimation, we have to use long 1480

term statistics to have an accurate estimation. This is done by exponential averaging where the 1481

factor ( )nβ is the same as for CFO estimation. The DMRS and the SRS correlations are long term 1482

averaged separately before combination. 1483

Supprimé : 4.0.1

Supprimé : 2

( )( )

∆= −

)Re(

)Im(tan

1

2

1ˆ 1

sC

sC

fs nT

nT

πτ







1484

We describe below the long term averaging of DMRS based correlations, the same principle is 1485

used for SRS based correlations. 1486

Its computation is described in section 6.3.1. Let us denote ( )sCnT

1− the value of the statistic above 1487

based on the observations of all TTIs until TTI(n-1) included, and ( )nH apipilot

,,

~ the pilot channel 1488

estimates corresponding to TTI number n. Then the update of ( )sCnT is done by: 1489

( ) ( ) ( ) ( )( ) )(11 sCnsCnsC TnT

nT ββ −+= − 1490

The angle computation to invert the tan(.) function is done the same way as for CFO estimation, 1491

see section 6.2.1.1. 1492

The whole timing offset processing including SRS based correlations is described below: 1493

1494

1495

CORRELATION UPDATE AFTER TA TRANSMISSION TO THE UE 1496

In LA2.0, the value of ( )sCnT was re-initialized to zero when a timing offset command is applied by 1497

the UE. This is not the case for LA3.0. Instead, the correlation is updated by a rotation 1498

corresponding to the applied timing offset command. The SRS based and DMRS based 1499

correlations are updated separately with the same complex exponential. 1500

The multiplicative complex exponential is equal to ( )TaCmdscfsj ...2exp ∆π where TaCmdscis 1501

the TA command expressed in seconds. 1502

Update long term averaged value

DMRS reception

DMRS Instantaneous correlation (1TTI)

Long Term averaged DMRS based correlation

Update long term averaged value

SRS reception SRS Instantaneous correlation (1TTI)

Long Term averaged SRS based correlation

Combination

Angle computation

TA Command

Supprimé : 4.0.1

Supprimé : 2







Fixed point implementation : The exponentials can be precomputed for different values of 1503

TaCmdscwith a s16 format. The multiplication is followed by a shift >>15. 1504

1505

UL TIMING CONTROL DESCRIPTION 1506

The timing offset estimation algorithm described above computes an estimation of the position of 1507

the center of gravity of the different paths corresponding to the channel profile of the UE of interest. 1508

The aim of the timing control device is to minimize the amount of the channel’s energy falling 1509

outside the CP based on this timing estimate. To do so, the NodeB sends timing advance 1510

commands to the UE through DL signaling. Upon receiving the timing control command from the 1511

eNodeB, the UE adjusts its UL transmission time. Moreover, the NodeB has to re-initialize the 1512

value of ( )sCnT to zero as soon as the UE applies the new tranmission timing.. The approach used 1513

is to ensure that these commands are such that the center of gravity of the paths of the UE is 1514

located in a fixed position TAτ on the middle of the CP. This means that the timing advance 1515

command is computed by the difference between the timing offset estimation and this chosen 1516

position TAτ . The granularity of the timing control command is equal to 0.52µs, which corresponds 1517

to 16 samples for 20MHz. 1518

In a future release, we can consider placing the center of gravity at a fixed time position TAτ on the 1519

left of the CP, which is motivated by the fact that for most channels, the power profile is more 1520

spread on the right of the center of gravity than on its left. Therefore, if we want to minimize the 1521

amount of energy falling outside the CP, taking into account this unbalanced power distribution with 1522

respect to the center of gravity, we could consider taking a back-off on the left to place the center 1523

of gravity. For current release, the effective value of TAτ will be equal to 2TA

CP=τ .This 1524

corresponds to 80 samples for 20MHz. 1525

Notice that due to real time constraints, in case a timing advance command has to be sent to a UE, 1526

this command is based on timing offset estimation based on pilot information of previous 1527

subframes since timing estimation requires processing over several TTIs that should be done in 1528

the DSP. 1529

This choice of 2TA

CP=τ rather than 4TA

CP=τ as recommended in the previous version of the 1530

document is motivated by the need to have a constant target rather than changing from 2TA

CP=τ 1531

at RACH output to 4TA

CP=τ later as was previously done. 1532

The choice of CP/2 as the constant target is due to the lower granularity of RACH timing estimates: 1533

we recommend to be more conservative when we place the center of gravity of the UE’s channel if 1534

Supprimé : 4.0.1

Supprimé : 2







it based on only RACH delays estimates. Therefore, once the RACH delays are identified, we 1535

recommend to place the center of gravity on position 2

CPRACHTA =τ . 1536

We recommend to ensure convergence of ( )sCnT before sending timing advance command. A 1537

recommended minimum value of 500ms between two consecutive transmissions should be 1538

ensured. 1539

1540 Figure 6-10 Timing advance commands computation 1541

1542

Periodicity of timing advanced command transmission 1543

The timing advance command has to be sent to the UE only if we detect that there is a too 1544

important difference between the estimated timing and the expected value TAτ . However, we have 1545

to pay attention to the fact that if the timing offset estimation is badly estimated, we should not send 1546

a timing advance command but wait until we have reached good convergence. This is in particular 1547

the case if the value of ( )sCnT has just been initialized: in this case, we can observe successive 1548

timing estimations falling on both sides of TAτ with a big timing difference (greater than 0.52µs), 1549

especially for small PRB size transmission. 1550

In order to deal with such initial random behavior, we were using up to LA2.0 a window mechanism 1551

consisting in defining a window spanning 0.52µs on both sides of TAτ , and sending a timing 1552

advance command only if a number N0 of successive estimates fall outside the window and on the 1553

same side. The foreseen value for N0 is N0=5. 1554

CP

Timing advance command sent to UE 2

Timing advance command sent to UE 1

Center of gravity of the channel power profile of UE 1

Center of gravity of the channel power profile of UE 2

Time

TAτ

Supprimé : 4.0.1

Supprimé : 2







This is no longer used in LA3.0 since with the increased number of users, waiting for 5 timing offset 1555

measures may take a while. Instead, we wait for a convergence time of 500ms before sending a 1556

command. Notice that if this simplifies the design, we can make the TTI where the Timing Advance 1557

command is sent coincide with a TTI where an SRS is received (i.e. after expiration of the 500ms 1558

timer, we can wait for the first SRS reception to send the TA command). 1559

1560



1563


M Integer u11 I 1 Number of subcarriers of the Ue


s Integer u3

I 1

Interval between the pilot sub-carriers used to compute the

cross-correlation

H_Main Complex Integer

(12I,12Q) I 2 x M


mode 1Rx) after CFO compensation (for both pilot blocks) pointing of the first PRB of considered user.

CP

Time

TAτ = CP/2

Acceptable timing offset values

If N0 successive timing estimates fall in this zone, send a timing advance command

If N0 successive timing estimates fall in this zone, send a timing advance command 0.52µs

Supprimé : 4.0.1

Supprimé : 2






Mis en forme : Anglais(Royaume-Uni)H_Div

Complex Integer

(12I,12Q) I 2 x M

Estimated Channel on the Div antenna after CFO

compensation (for both pilot blocks) pointing of the first PRB of considered user.

H_Div2 Complex Integer

(12I,12Q) I 2 x M

Estimated Channel on the 2nd Div antenna before CFO


(when applicable)


(12I,12Q) I 2 x M

When applicable, Estimated Channel on the 3rd Div antenna before CFO


(when applicable)

CT_s Complex Integer

(31I, 31Q) I/O 1 Long term statistic

TO Real Integer

s4

O

Per accumulation period

Timing Offset Estimation output (multiple of 0.52µs)

1564 Table 6-5 :Interface definition for Timing Offset A lgorithms 1565

1566 1567


SRS Correlation computation : 1569

This is described in section 11.9. However, we need to add an averaging over the antennas and 1570

over the number of tones equivalent to the averaging present in the DMRS processing. 1571

Averaging over antennas is done by summation followed by ( )RXN2log>> . 1572

For averaging over tones, we need to use a scaling factor equivalent to the DMRS scaling factor. 1573

To do so we need to take into account that: 1574

• DMRS averaging over the tones is done through ( (coeff_prb_divide[N-1]* 0x0AAB) >> 11) 1575

>>11 whereas the floating point representation of 1/(N*12) is in fact ( (coeff_prb_divide[N-1576

1]* 0x0AAB) >> 11) >>16. There is thus the equivalent of an extra multiplication by 32 in 1577

the averaging. 1578

• For SRS, the equivalent of the DMRS averaging is thus equivalent to the process above 1579

but with N*12 replaced by 561 / 417 / 273 / 129 / 57 / 92

2 =

−− sNM SRS

Sk for 1580

1.4 / 3 / 5 / 10 / 15 / 20MHz. 1581

• However, we need to take into account that SRS correlations have already been scaled by 1582

>>2 inside the FPGA 1583

Taking into account the (>>2) inside the FPGA, we only need to divide by

−−2

24

1 sNM SRS

Sk 1584

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 2

−− NM SRSSk







1585

Therefore averaging over tones shall be done the same way as for DMRS but with the following 1586

coefficients for coeff_prb_divide 1/(Nx12) in u16 for the different bandwidths from 1.4 to 20MHz: 1587

coeff_prb_divide =[21843 3449 1524 720 471 350] 1588

1589

1590

1591

1592

DMRS Correlation computation : 1593

As for frequency offset estimation, the computation is done in three steps. First, the correlation for 1594

the current TTI is done PRB by PRB inside the FPGA C. These PRB based correlations are 1595

transmitted to the DSP. The DSP averages the different correlations to obtain one correlation per 1596

TTI. Finally, exponential averaging is performed to obtain a long term correlation. 1597

Frequency domain correlation per PRB per antenna (format s24) is calculated by the FPGA C as: 1598

FD_corr (PRB, α) = ( )*1

0

112

0,,,,

~~∑ ∑

=

−−

=+

p

s

i

pilotapsi

pilotapi HH 1599

The corresponding fixed point implementation is described below (we recall that 120 =M ): 1600

1601 1602

( )0,, =ausCSRST

( )1,, −= RXSRST NausC

>>X u16 1/(Nx12)

>>11 To SRS long term averaging Sat 31

Sat 31

…

X=log2 (NRX)

Supprimé : 4.0.1

Supprimé : 2







1603 1604 1605 The User based short term cross-correlation (format s26) is computed by the DSP by averaging 1606

the different quantities above on the PRBs allocated to the user as described below: 1607

(12I,12Q)

(12I,12Q)

2(M0-s) times

23 24

( )( ) sM −>> 02 2log

24 DSP

antenna 0

M0=12 •

20

(12I,12Q)

(12I,12Q)

23

24

antenna NRX-1 if NRX >1

( )( ) sM −>> 02 2log

2(M0-s) times

24 DSP

…

Supprimé : 4.0.1

Supprimé : 2







The following block has been updated in this version: 1608 1609

1610 1611

1612

1613

1614

1615

1616

1617

1618

1619

1620

1621

1622

1623

1624

1625

N

FD correlation

>>11

Sat26

Sum over NRX antennas

)(log24 2 N+

)(log25 2 NX ++

u16 1/(N*12)

)(log41 2 N+

>>X

X=log2(NRX)

N

sum over the PRB

sum over the PRB

…

Antenna 0

Antenna NRX-1 if NRX >1

number of PRB

number of PRB

Supprimé : 4.0.1

Supprimé : 2







LONG TERM AVERAGING OF THE FREQUENCY DOMAIN CORRELA TIONS 1626 1627

For DMRS based correlations, the user based long term cross-correlation (format s31) is calculated by 1628

DSP by exponential averaging of the quantities above (forgetting factors are under Q0.10 format as 1629

described in section 6.3.1): 1630

eCEM IMPLEMENTATION for long term cross correlation (DMRS) 1631

1632

1633

bCEM IMPLEMENTATION for long term cross correlation (DMRS) 1634

1635

1636

1637

1638

1639

1640

1641

1642

1643

1644

1645

Equivalent block 1646

The global block above is equivalent to the block h ereafter. 1647

1648

1649

>> SLT << 2e >> 1

s26

>>8 >>2

u10

s32

Sat 32

s39 s30 s40 s32

s31

β

Sat 30

<<2e

z-1

-

>>2 >>1

s32

Supprimé : 4.0.1

Supprimé : 2







Details of block: Short Term (ST) max left shift co mputation and application 1650

It replaces the legacy shift (<<2e) by a dynamic shift of (<<(2e-SST)) . The value SST is computed to 1651

stay on 32 bits. The entry signal is s26, 1652



Block Inputs Initialization Block Outputs

STγ

SST = 2e Mask=0x40000000

SST

STST Se−<< 2γ

1655

1656

1657

1658

1659

1660

1661

1662

1663

1664

1665

1666

1667

1668

1669

1670

1671

1672

Details of block : Shift computation for Short Term (ST) and Long Term (LT) rescaling 1673

The long term value of SST computed at TTI n is called SLT while the long term value computed at 1674 TTI n-1 is called oldSLT. 1675

If SST and oldSLT are different then the short term correlation (TTI n) and the long term correlation 1676

(TTI n-1) do not have the same scale. To rescale them, the real and imaginary part of ST 1677

correlation computed at TTI n are shifted by RST = Max ((oldS LT - SST), 0) and the real and 1678

imaginary part of LT correlation computed at TTI n-1 are shifted by RLT = Max ((SST - oldS LT), 0). 1679

Details of block : Long Term (LT) right shift updat e and long term correlation rescaling 1680

SLT = 0 at UE set up. 1681

The long term shift SLT is memorized between 2 runs with the long term correlation LTγ . 1682

Supprimé : 4.0.1

Supprimé : 2







The long term shift SLT is the max value between SST (computed at TTI n) and oldSLT (SLT 1683

computed at TTI n-1). 1684

If SLT is not zero and both real and imaginary part of LT correlation are not on 32 bits, then real and 1685

imaginary parts of LT correlation are left shifted and SLT is decreased. Indeed, a small SLT helps 1686

avoiding a high right shift on the next short term correlation which would result in a loss of 1687

precision. 1688

Block Inputs Initialization Block Outputs SST oldSLT

LTγ

Mask=0x40000000 SLT NewLTγ

1689

1690

1691

1692

1693

1694

1695

1696

1697

1698

1699

1700

1701

1702

1703

1704

1705

The following is common to eCEM and bCEM IMPLEMENTA TION 1706

For SRS based correlations, we do not need to pre-compensate by the AGC for drop 1 since this is 1707

common to all transmissions However, this is no more the case for drop 2 since the AGC shift is 1708

computed dynamically. For drop 1,the AGC compensation was compensated for directly on the long 1709

term estimate. For drop 2, this compensation has to be taken care of at the input of the long term 1710

averaging engine in order to be sure to average correlations with the same scales. 1711

On the other hand, we add a scaling >>8 so that after the front-end shift we are still on 32 signed bits. 1712

This is done to keep the same processing as for drop 1 after the shift. Doing this, all processing after 1713

this shift, including the weights computations and combination, can be kept the same. 1714

The long term estimation is described below: 1715

1716

Supprimé : 4.0.1

Supprimé : 2







1717

1718

1719

1720

1721

1722

1723

1724

1725

1726

1727

Notice that since the SRS allocation is constant and happens at fixed predefined instants, there is no need 1728

to have a dynamically updated SRSβ based on the past allocation. 1729

In order to have proportional weighting between SRS and DMRS, we need to compute SRSβ based on the 1730

same rules as for DMRS. 1731

We use the same notations as in section 6.3.1 for SRS, i.e. nr is the number of PRBs allocated to the user 1732

of interest at TTI n, and nR the averaged value of past allocation sizes nr . Since the SRS is transmitted 1733

with period SRST (expressed in number of TTIs) with the same number of TTI, we have after 1734

convergenceSRS

nn T

rR = . We can then apply the formula ( ) nn

nn rR

R

1

010

10

ββββ

−+=

−

− which gives 1735

( ) SRSSRS T00

0

1

ββββ−+

= . 1736

1737

For 99.00 =β and SRS periods {2,5,10,20,40,80,160,320}, this gives in floating point: 1738

0.2363 0.3822 0.5531 0.7122 0.8319 0.9083 0.9519 0.9802=SRSβ for the different 1739

periods. 1740 1741 In fixed point with coding on u10, this gives: 1742 1743

242 391 566 729 852 930 975 1004=SRSβ 1744

1745 Specific case of SRS dropping: 1746 1747 In some specific cases, it may happen that SRS are dropped. This can happen in the case of management 1748 gap processing, but the statement below is not limited to this case. Since this behavior will occur in most 1749 cases, we recommend to do the design assuming that SRS transmission is not periodic. 1750 If some SRS symbols are dropped, then we cannot consider anymore that SRS allocation is constant and 1751 happens at fixed predefined instants as above. In this case, we need to use the same processing as for 1752

DMRS to compute a dynamic SRSβ that takes into account the variable period in SRS transmission. . 1753

However, on the contrary of DMRS, we will use the number of SRS correlations per TTIs instead of the 1754

number of PRBs to set the dynamic value of SRSβ . This is because for SRS we remove some tones at the 1755

edge so that we need to go to a tone resolution. This means that for each TTI , we will update the value of 1756

s32

>>8 >>2

u10

s32

Sat 32

s31 s30 s40 s32 s31

SRSβ

Sat 30

z-1

-

>>1 >>1

s32

<<(2 e(SRS) – 8)

Supprimé : 4.0.1

Supprimé : 2






Mis en forme : Anglais(Royaume-Uni)SRSβ using either 0 if no transmission or 561 / 417 / 273 / 129 / 57 / 9

22 =

−− sNM SRS

Sk for 1757

1.4/3/5/10/15/20MHz in case of SRS transmission. This value can be coded on a u10. 1758 The corresponding processing is described below. The initialization is the same as for DMRS. 1759 1760 1761 1762

1763

1764

1765

1766

1767

1768

1769

1770

1771

1772

Fixed point implementation for update of Rn (SRS only) 1773 1774

1775 1776

Fixed point implementation for update of nβ (SRS only) 1777

1778 1779

COMBINATION OF DMRS AND SRS FREQUENCY DOMAIN CORREL ATIONS 1780 1781

After long term averaging, DMRS and SRS correlations have to be put at the same scale before being 1782

combined. We need to take into account for the following scaling: 1783

• SRS AGC 1784

u10

>>10

u10

u16

Sat 16

s32 s42 s32

u16

0β

z-1

-

1−nR

nR

<<6 u16

1−z

Rn

<<12

BA

A

B

0β

>>12 Sat 10

nβ

u16

u28

u16

u10

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 2

−− NM SRSSk







• The ( )( ) sM −>> 022log inside the DMRS correlation, not present in SRS correlation 1785

• Internal shift difference for long term averaging between DMRS and SRS : 1786

o For eCEM >>2 for DMRS and >>1 for SRS 1787

o For bCEM equivalent shift >>SLT for DMRS and >>1 for SRS 1788

1789

For LA3.0 drop 1, rescaling was done through ( ) ( )( ) ( ) 44822log2 0 =−=−−<< sMSRSe 1790

For post LA3.0 drop 1, this is exactly the same thanks to the front-end shift <<(2 e(SRS) – 8). Notice that 1791

this implicitly assumes that e(SRS) is lower or equal to 4, which gives a limitation on the SNR range which 1792

can be supporter by SRS (far higher what is needed). 1793

Moreover, since for each occurence the values of the correlations are averaged over the number of 1794

correlations per TTI, we need to put some extra-weights on DMRS and SRS correlations to take into 1795

account these relative number of correlations. Actually, an averaged correlation should have a higher 1796

weight in the combining if it has been averaged over more tones. This is easily understood if we take the 1797

example of a UE with systematic PUSCH transmission over 1 PRB to be combined with the SRS 1798

correlations, we would need to put a high weight on the SRS correlation and a lower weight on the DMRS 1799

correlation. 1800

Before combining, we multiply the Long Term averaged correlation by weights SRSw and DMRSw for SRS 1801

and DMRS correlations respectively. The weights computation is explained below. As stated above, these 1802

weights should be proportional to the number of correlations per TTI. We compute them as: 1803

( )112 >>×= nDMRS Rw , where the computation of nR is described in section 6.3.1. 1804

For SRS weight SRSw we take the output of the long term averaging SRSnR of

−−2

2s

NM SRSSk as 1805

described above shifted by one. 1806

The exact fixed point scheme for the correlations weighting is described later (see figure). 1807

In order to avoid saturation, we have to shift both correlations by a common shift “o” depending on their 1808

MSB positions.This parameter “o” which defines the size of the shift is chosen in order to minimize the loss 1809

of LSB to a minimum as described below. 1810

Let “order” denotes the function computing the number of bits representing the sign before the first 1811

significative bits.For LT averaged DMRS correlation, the function order is applied on 31 bits while it is 1812

applied on 31 bits for LT averaged SRS correlation. Then we compute: 1813

scaling_0 = 31 - order( ( )nCorrelatio SRS averaged LTRe ); 1814

scaling_1 = 31 - order( ( )nCorrelatio SRS averaged LTIm ); 1815

scaling_2 = 31 - order( ( )nCorrelatio DMRS averaged LTRe ); 1816

scaling_3 = 31 - order( ( )nCorrelatio DMRS averaged LTIm ); 1817

After multiplication by the weights, we should have a maximum of 30 bits to avoid saturations, and since 1818

the weights are on 11bits, we should have the correlations occupation equal to 19bits before weights 1819

multiplication. 1820

Supprimé : 4.0.1

Supprimé : 2







o=max(scaling_0, scaling_1, scaling_2, scaling_3)-19 1821

Notice that o is a signed value (which is therefore not equal to zero if the result is negative) 1822

Then all correlations need to be shifted by “o” before weight computation. 1823

1824


1826

1827


1829

1830

1831

The following is common to eCEM and bCEM 1832

Long Term averaged SRS correlation

Long Term averaged DMRS correlation

( )( ) ( )122log8 0 −−−<< sM

To TA computation

o>>

o>>

order computation

5>>

5>>

u15 ( )112 >>× nR

1>>SRSnR

u15

Long Term averaged SRS correlation

Long Term averaged DMRS correlation

( )( ) ( ))1(22log8 0 −−−−<< LTSsM

To TA computation

o>>

o>>

order computation

5>>

5>>

u15 ( )112 >>× nR

1>>SRSnR

u15

s32

s32

s31 s31

Supprimé : 4.0.1

Supprimé : 2







The angle computation is computed the same way as for frequency offset estimation, see section 1833

6.2.1.1. Based on this, the timing advance command is generated as described below. 1834

1835

Timing computation: 1836

1837

1838

Angle is defined in the [-128; +127] interval, corresponding to [-Pi; +Pi]. 1839

To compute the CP value, we take into account the shortest one: 4,69 µs. 1840

CP = 4,69µs * 2 * Pi * s * 15kHz = 107 1841

TAstep = 0,52µs * 2 * Pi * s * 15kHz = 12 1842

Scaling = 2^10 * (2^3 / TAstep) = 683 1843

6.3. MEASUREMENTS 1844

6.3.1 LONG TERM AVERAGING 1845

We describe here the strategy used to compute long term averaging of any aperiodic measurement 1846

when needed. This is mainly valid for: 1847

• Time domain correlations used to compute CFO estimation (see section 6.2.1.1) and 1848

speed estimation (see section 6.3.5.2) 1849

• Frequency domain correlations used to compute Timing Offset estimation (see section 1850

6.2.2). 1851

• Long term user’s power (see section 6.3.4) 1852

Notice that this is not valid for noise since this is a periodic observation (actually, it is measured 1853

on both scheduled and unscheduled PRBs). 1854

The computation of any long term measurement is done through exponential averaging. The 1855

forgetting parameter β is dynamic and depends on the scheduling history of the users and its 1856

current number of allocated PRBs. 1857

u6

Scaling TAτ

> 0

yes

- s8

Angle

u10

>> 13 - x

- abs

TAstep/2 u3

TaCmd

s4

u9

s9 sat4

no

0

Update acc_corr_re Update acc_corr_im

s4

0 or 1

1

Supprimé : 4.0.1

Supprimé : 2







Assuming that at each TTI n we observe an instantaneous value nM of a measure, and that 1858

we want to compute a long term estimation LTM based on these successive instantaneous 1859

values. 1860

Let us denote nr the number of PRBs allocated to the user of interest at TTI n, and nR the 1861

averaged value of past allocation sizes nr . 1862

nR is obtained from nr by an exponential averaging with a constant parameter 0β which is the 1863

same as for long term noise estimation: 1864

( ) nnn rRR 1 010 ββ −+= − 1865

Notice that this update is done for all TTIs, even for those on which the user is not allocated. 1866

This means that when the UE is not allocated, the update below is performed using 0=nr . 1867

Based on that, the value of the exponential parameter ( )nβ used to update the value of LTM 1868

when we receive a new instantaneous measure nM on TTI n writes: 1869

( ) nn

n

n

nn rR

R

R

R

1

010

1010 ββ

βββ−+

==−

−− 1870

The update of LTM then writes: 1871

( ) nnLTnLT MMM ββ −+← 1 1872

1873 1874

nβ is always coded as u10. 1875

1876 1877 1878



R Integer u12

I/O 1 Long term averaged value of

the number of subcarriers allocated of the Ue

r Integer u12

I 1

Current number of subcarriers allocated of the Ue (zero if not

allocated)

Beta_0 Integer u12

I 1 Static averaging parameter

1880 Table 6-6 :Interface definition for update of Rn 1881

1882 1883 1884 1885 1886 1887


R Integer u12

I 1

Long term averaged value of the number of subcarriers allocated of the Ue (before

update)

Supprimé : 4.0.1

Supprimé : 2







r Integer u12 I 1

Current number of subcarriers allocated of the Ue

Beta_0 Integer u10

I 1 Static averaging parameter

Beta_n Integer u10 O

1 Dynamical averaging

parameter

1888

Table 6-7 :Interface definition for update of nβ 1889

FIXED POINT IMPLEMENTATION 1890

1891

1892

1893

1894

1895

1896

1897

1898

1899

1900

Fixed point implementation for update of Rn 1901 1902

1903 1904

Fixed point implementation for update of nβ 1905

1906 Initialization: 1907

u12

>>10

u10

u12

Sat 12

s13 s23 s13

u13 u12

0β

z-1

-

1−nR

nR

<<6 u12

1−z

Rn

<<12

BA

A

B

0β

>>12 Sat 10

nβ

u12

u22

u12

u10

Supprimé : 4.0.1

Supprimé : 2







At the first user transmission, we should have no past so that in theory 1−nR should be equal 1908

to zero. However, from a fixed point implementation perpsective, we will adopt the following 1909 rule: 1910 1911

Whenever 01 =−nR , this value is replaced by the value of nr after the left shift (<<6) of 6. 1912

1913 1914

6.3.2 NOISE AND POWER ESTIMATION 1915

6.3.2.1 INTRODUCTION 1916

There are two SNR estimation functions in the uplink SC-FDMA receiver. The 1st SNR estimation 1917

is for frequency domain equalization of SC-FDMA in the spatial processing and equalization, and 1918

also for time domain channel estimation. The coherent demodulated pilots and the derived pilot raw 1919

channel response are used for this SNR estimation, i.e pilots before and after frequency domain 1920

filtering. The 2nd SNR estimation is the data SNR estimation, and this is described in section 6.3.7. 1921

The function computing the 1st SNR estimation is called on each antenna, for each PRB and for 1922

each TTI. Its inputs are the outputspilot

apH ,

~ of the CFO compensation applied to pilot blocks. Its 1923

output is sent to the time domain filtering part of channel estimation, to frequency domain equalizer. 1924 1925

This SNR estimate for MMSE equalization and channel estimation is obtained from the UE 1926 transmitted power spectrum density over the estimated noise variance. The noise variance 1927 estimation assumes that the noise spectrum is white within the PRB of interest. Since this noise 1928 includes both thermal noise and Intercell Interference, it cannot be considered as white over the 1929 whole bandwidth but only on a PRB, so that the algorithm works on a PRB by PRB basis. 1930

We use a long term averaging of the noise power estimation with a forgetting factor 0Noise ≥β . 1931

This averaging is done on a PRB by PRB basis with the same forgetting factor on all PRBs. Notice 1932

that in this case, this parameter Noiseβ can be chosen constant in time since noise is observed at 1933

each TTI with the same number of observations per PRB. The definitive value of Noiseβ shall be 1934

provided in a future release of the document. Notice that the case where no long term averaging is 1935

performed is the particular case corresponding to 0Noise =β . 1936

6.3.2.2 NOISE VARIANCE ESTIMATION FOR SCHEDULED PRB 1937

We describe here the method used to compute the noise power on a given PRB. All the processing 1938 below is done on a PRB basis. 1939

In order to perform noise estimation, we first perform a frequency domain filtering of the 1940

compensated pilot channel estimates pilot,,

~apiH as in the channel estimation algorithm of 6.4.1, but 1941

with parameters NoiseK and NoiseG specific to the noise estimation and equal to 1Noise =K and 1942

1Noise =G . Notice also that if we use K=1 for frequency domain channel filtering (which is by the 1943

way the recommended value), both filtering operations can be performed jointly without increasing 1944 the complexity by getting this intermediate results. Actually, for each tone, we can filter with 1945

1Noise =K and 1Noise =G , and use the result of this first filtering to continue the second filtering 1946

with 1=K and 1≥G . 1947

Notice that since this noise estimation is done on a PRB basis, we can perform this filtering PRB by 1948 PRB, i.e. we do not have to filter across PRB boundaries as in the frequency domain channel 1949 estimation. Notice also that as described later, we do not have to use the first and last values of the 1950

Supprimé : 4.0.1

Supprimé : 2







filtered channel estimates so that they do not need to be computed. The tone indexes used for 1951

noise estimation are thus: 2,,1 0 −= Mi K where 0M is the number of tones per PRB 1952

(i.e. 120 =M ). 1953

We denote [ ]ai

ai

a

i 1,0, ,ζζζ = the frequency domain filtered pilot channel response vector computed 1954

with 1Noise =K and 1Noise =G for i-th sub-carrier and a-th antenna, corresponding to the two 1955

demodulation reference symbols within the TTI. 1956

1957 The general formula for noise estimation on a single PRB will have the form: 1958 1959

( ) ∑ ∑=

−

=

−=1

0

2

1

2

,pilot

,,2

PRB,

0 ~2

1ˆ

p

M

i

apiapia H

Tζσ 1960

where T is a constant depending on NoiseK , NoiseG and 0M only: 1961

( )3

20

2

112

NoiseNoise0 =

+−−=

GKMT . 1962

1963 If we make the assumption of identical noise power on all antennas, we have 1964 1965

( ) ∑ ∑ ∑−

= =

−

=

−=1

0

1

0

2

1

2

,pilot

,,2

PRB

0 ~2

11ˆ

RXN

a p

M

i

apiapi

RX

HNT

ζσ 1966

1967 Notice that this noise estimator is fully unaffected by fast channel variation in time and particularly 1968 robust against the effect of overestimating noise in case of high SNR and high frequency selectivity 1969 (or timing offset), which is a problem of many other noise estimators. 1970

Notice also that even if the edge tones have not been used for the estimation, this estimation is 1971 also valid for these edge tones. 1972

Supprimé : 4.0.1

Supprimé : 2







1973 Figure 6-11 : Noise estimation on a given PRB 1974

1975 Notice that noise power estimation is used as such for frequency domain equalization in data 1976

blocks but this is divided by ( )GK 2+ when used in the time domain filtering part of channel 1977

estimation, see section 6.4.2. (One has to pay attention to the fact that the parameters NoiseK and 1978

NoiseG are used for noise estimation only but the processing gain is instead to parameters used for 1979

channel estimation which are K and G). 1980 1981 1982 1983 1984 1985 1986 1987

1988

a

1ζ

- +

Frequency domain

filtering with

NoiseK and

NoiseG

pilot

a 0,

~H

a

0ζ

∑i

2

pilot

a 1,

~H

- +

∑i

2

40

3

2

11 =×T

( )2,PRBˆ aσ

( )2,PRBˆ aσ from previous

TTIs

( )Noise1 β−×

Noiseβ×

Sum over the tones of the PRB except the first and last

ones

Per PRB noise estimation

( )2,PRBˆ aσ

Optionnal if

0Noise =β

Supprimé : 4.0.1

Supprimé : 2









pilot

a p,

~H

Complex Integer

(12I,12Q) I

( 0M )x

(Number of pilot blocks)

Pilot channel estimates on the antenna of interest

a

pζ Complex

Integer

(12I,12Q) I ( 0M -2)x

(Number of pilot blocks)

Frequency domain filtered channel estimation with

parameters NoiseK and NoiseG

(is an internal variable if K>1)

Flag boolean u1

I 1 Boolean (K==1) to decide if

a

pζ is internal or external

( )2a PRB,σ Integer

u24 O 1

Estimated noise power on the antenna and PRB of interest

(to be further averaged on the different antennas)

( )aPRBE Integer

u24 O 1 Estimated energy on the

antenna and PRB of interest

1990 Table 6-8 :Interface definition for instantaneous n oise and SNR estimation for a given antenna and 1991

a given PRB 1992 1993

Simplification assumptions 1994

In order to simplify the frequency domain equalization for data demodulation, we can make the 1995 assumption that noise variance is the same for all antennas, and use the average noise variance: 1996

∑−

==

1

0

2,PRB

2PRB ˆ

1ˆ

RXN

aa

RXNσσ 1997

1998

Supprimé : 4.0.1

Supprimé : 2







1999 2000

Figure 6-12 :Noise estimation (one such processing is applied on each Rx antenna, and on each PRB of each 2001 TTI) 2002

The value of 1/T and other parameters are given below. 2003 2004

1Noise =K ,

1Noise =G

T1 - float

0.15

T1 – 10 bits

154

0

1M -10 bits

85

=K

GKi1

1

102 −

−=K

GKMi

10

2005 2006 Next figure provides the fixed point implementation of instantaneous noise estimator for a given PRB and a 2007 given antenna. Then averaging on all antennas has to be performed. 2008

( )2a PRB,σ = noise estimation for

a single PRB occupied by user u

[ ]aupM

aup

a

up ,,1,,0, 0 −= ζζζ K

From frequency domain filtering with

1NoiseNoise == GK

[ ]pilot

,,,1

pilot

,,,0

pilot

,, 0

~~~uapMuapuap HHH −= K

From pilot channel estimation

Noise

estimation for PRB of interest

Instantaneous

PRB based ( )2PRBσ

Long term estimation

Update long term PRB based

Noise estimation (no

update if

0Noise =β )

PRB based updated long term estimation

GK 2

1

+×

Time domain

channel estimation (MMSE)

Data demodulation

NoiseNoise0 , , GKM

( )aPRBE = Energy estimation

for a single PRB occupied by user u ; forced to zero if negative.

e

Supprimé : 4.0.1

Supprimé : 2







2009 2010

2011 2012 2013 2014

6.3.2.3 NOISE VARIANCE ESTIMATION FOR UN-SCHEDULED PRB 2015

Noise estimation is also carried out on unscheduled PRB for Uplink Scheduler processing. 2016 However, the PRBs where RACH transmission can occur do not have to be considered as non-2017 scheduled PRB, and the noise estimation below does not apply for those PRB. 2018 2019 The noise estimation on an un-allocated PRB consists in a simple averaging of the squared 2020 amplitudes of the post-FFT signal: 2021 2022

( ) ∑∑=

−

=

=1

0

1

0

2

,,0

2 PRB,

0

2

1ˆ

p

M

iapia X

Mσ 2023

Notice that on the contrary of allocated PRBs, the sum contains both edge tones. This sum should 2024 be computed for all un-allocated PRBs for each TTI since its output is sent to the scheduler. 2025

The figure below describes the fixed point implementation. 2026

1541 =T

(12I,12Q) |.|2

|.|2

|.|2

|.|2

>>5

>>5

Sat 24

i2-i1 tones

(12I,12Q)

(12I,12Q)

12

13

24 25

u26+log2(i2-i1)

u21+log2(i2-i1)

u32

u24

Pilot(0)

Pilot(1)

u8

-

-

-

-

∑ ∑=

−

=

−1

0

2

1

2

,pilot

,,

0 ~

p

M

i

apiapiH ζ

pilot

,0,~

aiH

ai 0,ζ

pilot

,1,~

aiH

ai 1,ζ

Supprimé : 4.0.1

Supprimé : 2







2027

6.3.2.4 INSTANTANEOUS POWER ESTIMATION 2028

Instantaneous signal power before noise subtraction, per PRB and antenna is computed as 2029 below: 2030

2031

( )

= ∑∑

=

−

=

1

0

1

0

2pilot,,

0

BiasedPRB

0 ~2

1

p

M

iapiH

MaE 2032

2033 The corresponding fixed point implementation is given below. 2034 2035 2036 2037

851

0

=M

(12I,12Q) |.|2

|.|2

|.|2

|.|2

>>3

>>7

Sat 24

M0 tones

u21+log2(M0)

(12I,12Q)

12

22 23

u24+log2(M0)

u22+log2(M0)

u32

u24

Sat 21+ log2(M0)

Pilot(0)

Pilot(1)

u7

∑ ∑=

−

=

1

0

1

0

2

,,

0

p

M

iapiX

aiX ,0,

aiX ,1,

M0 tones

Supprimé : 4.0.1

Supprimé : 2







2038 2039 2040

We can compute an unbiased estimation of the instantaneous useful signal power on a given PRB 2041 by the formula (it has to be forced to zero if it is negative): 2042 2043

( ) ( ) ( )( )0,ˆmax 2,PRB

BiasedPRBPRB aaEaE σ−= 2044

2045 Notice that all the tones of the PRB have been used for this estimation. Eprb(a) is needed to 2046 calculate the instantaneous energy per user Eu(a) which is used for the Filters Coefficients 2047 Computation, for T-MMSE (see section 6.3.4). 2048 2049

6.3.2.5 NOISE AND POWER ABSOLUTE VALUES (ECEM ONLY) 2050

In this section we describe the way to compute the absolute values for power and noise expressed 2051 in dBm based on the output of the MEA module. 2052

The first step is to map the samples received by the L1 to powers in dBm. This is done as per 2053 section 5.2 by using the UL RF antenna gain and the digital gain. 2054 2055 On top of that, we need to include the FFT gain and the channel element gain CE_Gain, i.e. the 2056 scaling introduced by the L1 receiver along the processing. 2057 2058

(12I,12Q) |.|2

|.|2

>>3

>>7

Sat 24

u10

24

(12I,12Q)

M0 tones

12

22

24+log2(M0)

22+log2(M0)

1/M0 = 85

|.|2

|.|2

12

22

25

32

Sat 21+log2(M0)

Pilot(0)

Pilot(1)

Supprimé : 4.0.1

Supprimé : 2







For Short Term Noise and Signal power, the processing introduces a factor 2x coming from the fact 2059 that there is no division by 2 to take into account the two pilot blocks 2060 2061 There is also a (>>2) inside the Long Term averaging loop. Therefore, the global scaling is (>>1). 2062

2063 We need to add on top of that the post-FFT shift scaling which is equal to 1. The resulting shift is 2064 therefore >>2. 2065

2066 Therefore, for both PUSCH noise and power long term values, the CE gain is equal to: CE 2067 gain=0.25 2068

2069

The overall conversion formula is then: 2070

MEA_Power_dBm = 10log10 (MEA_Power) - 10*log10(Imax²) – (UL RF receiver gain) – (UL Digital 2071 gain) - 10*log10(FFT_Gain) - 10*log10(CE_Gain) 2072 2073 The FFT gain is equal to the FFT size since this is an unscaled FFT. 2074

As for section 5.2 we have Imax² = 2^28 2075

2076

The different numerical values that are to be used in this equation are thus given below: 2077

10*log10(Imax²) = 10*log10(2^28) = 84.28 2078

10*log10(CE_Gain) = - 6.02 2079

10*log10(FFT_Gain) = 21.07 / 24.08 / 27.09 / 30.10 / 31.86 / 33.11 respectively for 1.4 / 3 / 5 / 2080 10 / 15 / 20 MHz 2081

Notice that in the formula above, when considering the value given by the DSP, we should not pay 2082 attention to the effective format in QX.Y of the power estimates but only consider the integer value. 2083 Stated differently, in the formula above should be considered as coded in QX.0 format. 2084

2085

6.3.3 METRICS EXCHANGE BETWEEN FPGA AND DSP 2086

The preferred approach consists in providing the scheduler with short term estimates of noise and 2087 signal powers at the lowest granularity so that the scheduler can compute whatever metrics it 2088 wants. The DSP has to compute long term average of noise and signal powers PRB by PRB. In 2089 order to compute these quantities, it is better to compute long term averages of noise and signal 2090 power separately before doing the subtraction since both quantities are not averaged at the same 2091 rate. This means that the FPGA will provide the DSP with the following quantities: 2092 2093

• Instantaneous signal power before noise subtraction, per PRB and antenna: 2094 2095

( )

= ∑∑

=

−

=

1

0

1

0

2pilot

,,0

BiasedPRB

0 ~2

1

p

M

iapiH

MaE 2096

2097 • Instantaneous noise power per allocated PRB and antenna: 2098

2099

( ) ∑ ∑=

−

=

−=1

0

2

1

2

,pilot

,,2

PRB,

0 ~2

1ˆ

p

M

i

apiapia H

Tζσ 2100

2101 2102

• Instantaneous noise power per non-allocated PRB and antenna: 2103 2104

Supprimé : 4.0.1

Supprimé : 2







( ) ∑∑=

−

=

=1

0

1

0

2

,,0

2 PRB,

0

2

1ˆ

p

M

iapia X

Mσ 2105

2106 ENERGY POST PROCESSING INSIDE THE DSP 2107

The DSP then performs a user by user averaging of the biased energy above over all the allocated 2108 PRB to obtain the instantaneous energy per user and per antenna: 2109

( )∑ ∑−

=

=PRB Used

1

0

BiasedPRB

11)(

RXN

aRXPRBST aE

NNuX 2110

The fixed point implementation is described in section 6.3.4. 2111

The DSP finally performs exponential averaging of )(uXST to obtain the long term user based 2112

biased energy )(uX LT . The fixed point implementation is described below: 2113

2114


2116

2117

2118

2119

2120

2121

2122


u24

>>8 >>2

u10

u31

Sat 31

s37

s28 s38 s30

u31 u30

β

Sat 28

<<2e

z-1

-

>>2 >>1

u36

Supprimé : 4.0.1

Supprimé : 2







2124

2125

Equivalent block 2126

The global block above is equivalent to the block h ereafter 2127

2128

2129

Block Short Term (ST) max left shift computation and appl ication 2130

It replaces the legacy shift (<<2e) by a dynamic shift of (<<(2e-SST,pow)) . The value SST,pow is 2131

computed to stay on 32 bits. The entry signal XST is u24, 2132



2135

Block Inputs Initialization Block Outputs

STX

SST = 2e Mask=0x40000000

SST,pow

powSTST SeX ,2 −<<

>> SLT,pow << 2e >> 1

Supprimé : 4.0.1

Supprimé : 2







2136

Block Shift computation for Short Term (ST) and Long Term (LT) rescaling 2137

The long term value of SST,pow computed at TTI n is called SLT,pow while the long term value 2138 computed at TTI n-1 is called oldSLT,pow. 2139

If SST,pow and oldSLT,pow are different then the short term power and the long term power do not have 2140

the same scale. To rescale them, the real and imaginary part of ST power computed at TTI n are 2141

shifted by RST = Max ((oldS LT,pow - SST,pow ), 0) and the real and imaginary part of LT power 2142

computed at TTI n-1 are shifted by RLT = Max ((SST,pow - oldS LT,pow ), 0). Only one of these two 2143

shifts is not zero. 2144

2145

Block Long Term (LT) right shift update and long te rm power rescaling 2146

SLT,pow = 0 at UE set up. 2147

The long term shift SLT,pow is memorized between 2 runs with the long term power LTX . 2148

The long term shift SLT,pow is the max value between SST,pow (computed at TTI n) and oldSLT,pow 2149

(SLT,pow computed at TTI n-1). 2150

If SLT,pow is not zero and LT power is not on 32 bits, LT power is left shifted and SLT,pow is decreased 2151

to avoid the loss of precision caused by a high right shift on the next short term power. 2152

Block Inputs Initialization Block Outputs SST,pow oldSLT,pow

LTX

Mask=0x40000000 SLT,pow NewLTX

2153

Supprimé : 4.0.1

Supprimé : 2







2154

2155

NOISE POST PROCESSING INSIDE THE DSP 2156

o Short term noise power per PRB per antenna is provided to the DSP by the FPGA C. The 2157 format is u24 for simo (see section 6.3.2.2) and un-allocated PRBs (see section 6.3.2.3), u23 2158 for mimo PRBs (see section 14.6). 2159

2160 o Long term noise power per PRB per antenna YLT(PRB,a) is calculated by the DSP as 2161

described below, The format is u30. 2162 2163

2164 eCEM IMPLEMENTATION 2165 2166 2167

2168 2169 2170

u24

>>8 >>2

u10

u31

Sat 31

s38

s28 s38 s30

u31 u30

β

Sat 28

<<2e

z-1

-

>>2 >>1

u37

Supprimé : 4.0.1

Supprimé : 2







2171 bCEM IMPLEMENTATION 2172 2173

2174 2175 o This PRB based long term noise power is then averaged over the PRBs allocated to a user. 2176

The obtained user based long term noise power YLT(u) is calculated by the DSP as described 2177

below, the format is u30. 2178

2179

o Finally, the user based long term useful signal (format u30) is calculated by DSP by 2180

ELT(u) = XLT(u) – YLT(u) which is used to perform speed estimation. 2181

2182

# of PRBs (averaged on NRX antennas)

>>9

>>3

Sat 30

u12

u21+Log2(NPRB)

u33+Log2(NPRB)

u30

1/nPrb(u)

Supprimé : 4.0.1

Supprimé : 2







2183

2184

6.3.4 METRICS USED IN L1 PROCESSING FOR 2185

DEMODULATION 2186

This section refers to the metrics used for Time domain MMSE channel (noise and signal powers) 2187

and the Frequency domain equalizer (noise power only). The current working version is that both 2188

short term and long term metrics usage should be made possible. 2189

Long term metrics 2190

If long term metrics are used, then per user )(uELT and per PRB )(PRBYLT as described in the 2191

previous section and provided by the DSP are used by the FPGA when determining T-MMSE and 2192 F-MMSE filter coefficients. Notice that because of scaling, these quantities are on 30bits. 2193

Short term metrics 2194

If short term metrics are used, the metrics come from the FPGA and not from the DSP. 2195

Per user per antenna energy value used by the FPGA when determining T-MMSE filter coefficients 2196

is as defined below: 2197

( ) ( )∑=userPRBsPRB

aEN

aE PRBu

1

2198

While the energy is per user and not per PRB, noise variance ( )2,PRBˆ aσ is still on PRB basis. For 2199

simplification, the same value ( ) ( )∑−

==

1

0

2,PRB

2PRB ˆ

1ˆ

RXN

aa

RXNσσ is used. 2200

Notice that since these quantities are internal to FPGA and do not come from DSP, they are on 2201 24bits. 2202

Averaging of energy on all prbs and all antennas is implemented like below, with the following 2203 remarks: 2204

- Eprb(a) is used for short term measures only (done by the fpga) 2205

- Eprb is used for long term measures only (done by the dsp) 2206

2207

Current working version 2208

The current working version is to use the short term metrics. 2209

Supprimé : 4.0.1

Supprimé : 2







1 2210

nPrb 1 2 3 4 5 6 8 9 10 12 15

1/nPrb 4095 2048 1365 1024 819 683 512 455 410 341 273

nPrb 16 18 20 24 25 27 30 32 36 40 48

1/nPrb 256 228 205 171 164 152 137 128 114 102 85

nPrb 50 54 60 64 72 75 80 81 90 96

1/nPrb 82 76 68 64 57 55 51 51 46 43

2211

2212

6.3.5 SPEED ESTIMATION 2213

This function is called for each UE, once per TTI. Its inputs are the long term pilot channel 2214

estimates correlations ( )( )1ˆ,,ˆLongTerm −nqp εγ as defined in section 6.2.1 and the noise estimates 2215

of section 6.3.2. Its output is the speed estimate, which is sent to the scheduler and to weight 2216 computations for time domain filtering. 2217

6.3.5.1 THEORETICAL BACKGROUND 2218

From a theoretical point of view, the UE speed depends on the (unknown) correlations: 2219

E prb(a)

All prb, antenna 0

u24

u31

>>9

Sat 24

>>3 Sat 21

>>log2(NRX)

Sat 24

u24

+

1/nPrb(u)

E prb(a)

u24

u31 >>9

Sat 24 >>3 Sat 21

All prb, antenna NRX-1 if NRX >1

1/nPrb(u)

u12

…

Sum over N RX

antennas

+

u12

Supprimé : 4.0.1

Supprimé : 2







( ) ( )( ){ }∗Ε= ideal,1,

ideal,0,0 pilot(1),pilot(0) aiai

a HHγ and 2220

( ) ( )( ){ } ( )pilot(1),pilot(1)pilot(0),pilot(0) 0ideal

,0,ideal

,0,0a

aiaia HH γγ =Ε=

∗ 2221

between pilots of both blocks on antenna a, where ideal,, apiH is the (unknown) ideal channel on pilot 2222

block p, antenna a, and tone i. 2223

If we assume for example that the fading follows the Jakes model, this correlation is equal to 2224

( ) ( )( )dHa ftJE πγ 2.pilot(1),pilot(0) pilot(1),pilot(0)00 ∆= where pilot(1),pilot(0)t∆ denotes the time 2225

difference between both pilot blocks (=0.5ms) and df the maximum Doppler shift. 2226

Notice that this does not depend on the antenna index, so that we can have a common speed 2227 estimation for all antennas. 2228

Using the same notations as before, we have approximately: 2229

aiaiai nHH ,0,ideal

,0,pilot

,0,

~ += where ain ,0, is an AWGN sample, and the same relation for second block, 2230

so that: 2231

( )( ){ } ( ) ( )( )duaiai ftJaEHH π2.~~

pilot(1),pilot(0)0Idealpilot

,1,pilot

,0, ∆=Ε∗

2232

Where ( )aEuIdeal denotes the (unknown) mean energy of the UE u on antenna a. Notice that this 2233

mean energy does not depend on the antenna of consideration, so that we denote it as IdealuE . We 2234

recall that pilot,,

~apiH denotes the output of the pilot channel estimation on pilot blocks p, antenna a, 2235

tone i. 2236

We also have: 2237

( )( ){ } ( )( ){ } 2Idealpilot,1,

pilot,1,

pilot,0,

pilot,0,

~~~~auaiaiaiai EHHHH σ+=Ε=Ε

∗∗ where 2

aσ is the noise power on the 2238

observed tone. 2239

We then obtain the following relation: 2240

( )( )( )( ){ }

( )( ){ }( )∑

∑−

=

∗

−

=

∗

−Ε

Ε=∆ 1

0

2pilot,0,

pilot,0,

1

0

pilot,1,

pilot,0,

pilot(1),pilot(0)0 ~~

~~

2RX

RX

N

aaaiai

N

aaiai

d

HH

HHftJ

σπ 2241

This means that we can estimate the speed from the normalized value of correlations of the pilot 2242 channel estimates 2243

6.3.5.2 PRACTICAL IMPLEMENTATION 2244

Speed estimation is based on normalized value of correlations of the pilot channel estimates 2245

coupled with a look up table where typical correlation values are stored. These correlations have 2246

already been estimated as ( )( )1ˆ,,ˆLongTerm −nqp εγ in section 6.2.1 CFO estimation, and can thus 2247

also be used here. For sake of simplicity, we omit here the frequency offset reference and simply 2248

write ( )qp,ˆLongTermγ . Notice however that all correlations used for speed estimation are real so 2249

that we should work with the real part of the ( )1,0ˆLongTermγ (the other values of ( )qp,ˆLongTermγ are 2250

already real by construction). 2251

Supprimé : 4.0.1

Supprimé : 2







The channel power estimation, corresponding to p=q in the correlation estimation, has to be 2252 unbiased. This is done by subtracting the value of the noise estimate. Notice however that since 2253 the correlations are long term averaged, the noise used to unbiased them has also to be long term 2254 averaged to have consistent subtraction). We have to compute a long term average of the noise 2255

power estimation ( )2LongTermPRBσ at least for speed estimation. This has to be done per PRB, and 2256

since observations are available on all TTIs, we can take a constant forgetting parameter equal to 2257 0.99. 2258

2259

Moreover, since the correlation estimates are UE based, this long term noise estimation has to be 2260 linearly averaged over the PRBs allocated to the UE of interest: 2261

( ) ( )∑=PRBPRBN

2LongTermPRB

2LongTermu ˆ

1ˆ σσ 2262

Speed estimate are then be computed from the ratio: 2263

( )( )( ) ( ){ } ( )2LongTerm

uLongTermLongTerm

LongTerm

ˆ1,1ˆ0,0ˆ2

1

1,0ˆRe

σγγ

γ

−+. 2264

2265

Notice that in practice only the sum ( ) ( )1,1ˆ0,0ˆ LongTermLongTerm γγ + is provided by the DSP to the 2266

FPGA C, and the terms ( )0,0ˆLongTermγ and ( )1,1ˆLongTermγ are not available separately. 2267

The Mean Squared Errors (MSE) between this correlation ratio and pre-stored typical ratios are 2268 computed, and the lowest MSE gives the corresponding values of the speed. In practice, as shown 2269 in the figure below, the division above has not to be explicitly computed since we use only 2270 comparisons to find the right speed in the look-up table. 2271

The inputs of the speed estimation are then simply ( )0,0ˆLongTermγ , ( )1,1ˆLongTermγ , ( )1,0ˆLongTermγ , 2272

and ( ) ( )∑=PRBPRBN

2LongTermPRB

2LongTermu ˆ

1ˆ σσ . 2273

2274

2275 Figure 0-1 Speed estimation 2276

2277 2278

The pre-stored values are stored under Q0.12 format. Values are given in next table (for 2GHz 2279

carrier frequency). 2280

2281

υ

( )1,0ˆLongTermγ

LUT

( )( )1,0ˆRe LongTermγ

( )0,0ˆLongTermγ , ( )1,1ˆLongTermγ

( ) ( )∑=PRBPRBN

2LongTermPRB

2LongTermu ˆ

1ˆ σσ

Supprimé : 4.0.1

Supprimé : 2







Index (t) Detected Doppler frequency /

Corresponding speed @2GHz (kmph)

Normalized correlation threshold (float) - t

Normalized Correlation threshold (Q0.12) - t

Number of MMSE coef. (see section 6.4.2)

0 5.6Hz/ 3 km/h 0.9980 2044 1

1 46.3Hz / 25 km/h 0.9881 2023 2

2 92.6Hz / 50 km/h 0.9672 1981 2

3 138.9Hz / 75 km/h 0.9363 1917 4

4 185.2Hz / 100 km/h 0.8958 1835 4

5 231.5Hz / 125 km/h na na 4

2282


Signal Name

Type I/O Format

Size Description

power Complex Integer

I

u30 1 Long term useful power estimate

SLT,pow integer I

u8 1 Long term power equivalent

right shift used for bCEM implementation

Gamma_01 Complex Integer

I

s31 1 Long term correlation of pilot 0 and 1

SLT,cor integer I

u8 1 Long term correlation

equivalent right shift used for bCEM implementation

Sigma2_U integer I u30

M/12 Average of noise power over all the PRBs allocated to the UE

Speed enum O {…} 1 Speed Estimation

Interface definition for speed estimation 2284

2285 2286 The L1 layer has also to provide the Downlink Scheduler with a high speed flag indicating whether 2287 the detected speed is lower or higher than 8km/h @ 2GHz. 2288 This high speed flag is provided only in the case where the detected speed is equal to 3km/h, since 2289 in all other cases the speed is already know as greater than 8km/h. 2290 This is done by comparing the normalized correlation to a specific threshold given below: 2291 2292 2293

Detected Doppler frequency /

Corresponding speed @2GHz (kmph)

Normalized correlation threshold (float) - t

Normalized Correlation threshold (Q0.12)

14.8Hz/ 8 km/h 0.9995 2046

Supprimé : 4.0.1

Supprimé : 2







A flag equal to 0 means that the speed is less than 8km/h, 1 means that the speed is greater than or 2294 equal to 8km/h, 2295

2296 2297 Next figure provides the fixed point implementation for the combined noise estimation. 1/NPRB is 2298 coded on 12 bits (unsigned). 2299

2300 And the associated implementation is described in the next figure,common to eCEM and bCEM 2301

implementation. Notice that we also give below the number of filter coefficients considered for 2302

channel estimation (see section 6.4.2). 2303

2304

2305

2306

2307

2308

2309

2310

2311

2312

2313

u30

>>9

>>3

Sat 30

u12

# of PRBs (averaged on NRX antennas)

u21+Log2(NPRB)

u33+Log2(NPRB)

u30+Log2(NPRB)

u30

1/NPRB

Supprimé : 4.0.1

Supprimé : 2







The entries P and γ are dependent on eCEM or bCEM implementation as explained below. 2314

2315

2316


2318

2319 2320 2321 bCEM IMPLEMENTATION 2322 2323 2324 2325 2326 2327 2328 2329 The algorithm of rescaling is the following. 2330 2331 If (SLT,pow > SLT,cor) 2332 2333 ( )0,0ˆLongTermγ rescaled = ( )0,0ˆLongTermγ 2334

( )1,1ˆLongTermγ rescaled= ( )1,1ˆLongTermγ 2335

Gamma_01rescaled= Gamma_01 >> (SLT,pow - SLT,cor) 2336

( )2LongTermuσ rescaled = ( )2LongTerm

uσ >> SLT,pow 2337

2338 Else if (SLT,cor > SLT,pow) 2339 2340 Gamma_01rescaled= Gamma_01 2341 ( )0,0ˆLongTermγ rescaled = ( )0,0ˆLongTermγ >> (SLT,cor - SLT,pow) 2342

( )1,1ˆLongTermγ rescaled= ( )1,1ˆLongTermγ >> (SLT,cor - SLT,pow) 2343


uσ >> SLT,cor 2344

Else 2345 Gamma_01rescaled= Gamma_01 2346 ( )0,0ˆLongTermγ rescaled = ( )0,0ˆLongTermγ 2347

( )1,1ˆLongTermγ rescaled= ( )1,1ˆLongTermγ 2348


uσ >> SLT,cor 2349

end 2350 2351 2352 For MMSE filter coefficient selection (see section 6.4.2), the same kind of search is applied for 2353 SNR determination and appropriate filter selection once speed and SNR indexes are found. 2354

2355

6.3.6 DTX DETECTION 2356

Some DTX detection algorithm is necessary to check that the UE has correctly detected the grants 2357

sent by the eNB. Actually, if the UE misses the detection of the grant and transmits nothing, the 2358

Node B should avoid updating the different measurements which are averaged over several TTIs 2359

P =( ( ( )0,0ˆLongTermγ + ( )1,1ˆLongTermγ )>>1) ( )2LongTermuσ−

γ : Gamma_01

P =( ( ( )0,0ˆLongTermγ rescaled+ ( )1,1ˆLongTermγ rescaled)>>1) ( )2LongTermuσ− rescaled

γ : Gamma_01rescaled

Supprimé : 4.0.1

Supprimé : 2







like speed or CFO estimates. However, noise can still be computed using the same algorithm as 2360 before. 2361

The DTX detection is computed by comparing the average SINR of the pilot symbols to a 2362

threshold. Ideally, this threshold should depend on both the MCS and the number of PRBs. 2363 However, for LA2.0, the value of this threshold only depends on the number of PRBs allocated to 2364

the UE of interest. Actually, in case of DTX, the DTX detection probability only depends on the 2365 number of noise samples we accumulate, i.e. on the number of tones allocated to the UE. 2366

In practice, the noise and useful power estimation are computed as in section 6.3.2. We then 2367

average the signal and noise powers on the antennas, PRBs, and both pilots in order to have one 2368

noise and one signal power estimation per TTI. We then compare the corresponding SNR to a 2369 given threshold depending on the number of PRBs to decide if we have a DTX or not (Notice that in 2370

practice, the division corresponding to the SNR computation can be avoided). The value of the 2371

threshold will be given in a future version of the document in the form of a look-up table. 2372

When a DTX is detected, two actions are to be processed: 2373

• Give this information to the scheduler 2374

• Cancel the update of the different measures, e.g. pilot correlations for both timing offset 2375 and frequency offset estimation, speed estimates, CFO estimates, etc… performed when 2376

we have processed the user’s date to evaluate its CRC. 2377

A simple solution for the second action consists in keeping in memory the former values of these 2378 parameters, i.e. before and after update, and point back to the former values if a DTX is detected. 2379

The exhaustive list of measures to be post-corrected is: 2380

• Pilot correlations in time domain: ( )( )nεγ ˆ,1,0ˆLongTerm 2381

• Power measurment 2382

• Pilot correlations in frequency domain: ( )sCnT of section 6.2.2 2383

• Speed estimates 2384

• CFO estimates 2385

Notice that the DTX detection is not used to by-pass the data estimation but only to avoid updating 2386 the different long term measurements with noise. Therefore, even if a DTX is detected, the whole 2387

classical processing (channel estimation, data demodulation and HARQ buffer update and CRC 2388 processing) is carried out independently of the result of the DTX detection. This is because the 2389 eNode B can set a SINR target corresponding to multiple transmissions, so that the first 2390

transmission by itself is likely to be erroneous. 2391

6.3.7 POST IDFT SNR ESTIMATION 2392

The data SNR estimation is the SNR estimate of the data after IDFT. The data SNR estimate 2393 output is used as the input to the de-mapper and LLR computation block if we use the optimal LLR 2394

computation. 2395

This is interesting for HARQ processing when the different retransmissions occurs at very different 2396 SNR, so that the packets having high SNR will have higher LLR weights than the packets having 2397

low SNR. 2398

Supprimé : 4.0.1

Supprimé : 2







Since this feature is an enhancement and not a 3GPP alignement, this is not seen as to be 2399 embedded for current release. We nevertheless describe the algorithm to for future releases. 2400

6.3.7.1 THEORETICAL COMPUTATION 2401

METHODOLOGY OVERVIEW 2402 2403 We describe here the consecutive stages of the methodology used to compute post IDFT SNR. 2404 This post IDFT SNR has to take into account the antenna diversity gain and the channel estimation 2405 loss, as well as the DFT spreading. 2406 2407 We describe in the schematic view below the general processing for post-IDFT SNR computation. 2408 Each block is detailed below. 2409

2410

2411

Pre-processing SNR 2412 2413 The first step is to compute the pre-processing SNR for each tone i and antenna j using the 2414 classical formula: 2415

2416

2417

Here, 2jσ is the frequency domain noise power for antenna j, and Hi,j is the frequency domain 2418

channel attenuation for the receiver antenna j for the sub-carrier i. 2419

The SIR for tone i on antenna j after multiplication by the conjugate of the channel estimates is 2420 equal to: 2421

1++×

=pilotjiji

pilotjijiChEst

ji SNRSNR

SNRSNRSNR

,,

,,, 2422

Pre-processing

SNR computation

Include channel

estimation loss

Include antenna

combining

Include DFT-spreading

effects

One Post-IDFT SNR

per SC-FDMA symbol

To LLR computation

Supprimé : 4.0.1

Supprimé : 2

2

2

,),(

j

jiHjiSNR

σ=







where pilotjiSNR, is the SNR of the channel estimation used to demodulate the i-th sub-carrier of the 2423

j-th antenna. This SNR can be approximated by ( )2

2

,

, 2j

jipilotji

HGKSNR

σ+≈ because of the 2424

processing gain of frequency domain filtering. 2425 2426

The SIR after combination of the signals of alll antennas is given by the summation of the SIR of all 2427 antennas: 2428

∑=

=RXN

j

ChEstji

combi SNRSNR

1, 2429

In DFT-SOFDM, the post DFT SNRs of the M QAM symbols that are spread together with the DFT 2430 are the same and are equal to: 2431

∑

∑

∑ +

+=

−

+

=

−

−

= iCombi

iCombi

Combi

N

iCombi

Combi

N SNR

SNR

SNR

SNR

SNRSINR

11

11

1

1

1

1

0

1

2432

2433 MIMO case (2Rx only) 2434 In MIMO configuration, only the pre-processing SNR is different, the rest of the processing is the 2435 same as for SIMO. 2436 Let us assume the following model for a given tone (tone index is omitted): 2437

nSHX += 2438

With 2439

•

=

2

1

X

XX is the input signal vector from antenna 1 and 2 respectively. 2440

•

=

2221

1211

HH

HHH is the channel matrix for the tone of interest 2441

• 2442

•

=

2

1

S

SS is the vector containing the post-DFT information of user 1 and user 2 2443

respectively. We assume that these symbols have unit energy. 2444

•

=

2

1

n

nn is the noise vector on antenna 1 and 2 respectively, ),0(~ 2

jj Nn σ , for a = 1 and 2. We 2445

further assume that 22

21 σσ = 2446

2447

Then we have (see MIMO section): { } XHIHHS HH 12ˆ −+= σ 2448

Let us further denote

=

u

uu H

HH

2

1 the uth column of H corresponding to user u. Then the 2449

post-processing SNR, i.e. the SINR at the output of the Frequency Domain MMSE, is given for 2450 user u by: 2451

Supprimé : 4.0.1

Supprimé : 2






Mis en forme : Anglais(Royaume-Uni)( )

( )( )222

2*

2

2

MMSEpost σσσ +−=−

v

vuu

H

HHHuSNR 2452

In this formula, v denotes the index of the second user. This means that: 2453

( )( )( )22

22

2

2*

1

2

2

1MMSEpost 1

σσσ +−=−

H

HHHSNR 2454

( )( )( )22

12

2

2*

1

2

2

2MMSEpost 2

σσσ +−=−

H

HHHSNR 2455

6.3.7.2 ALGORITHM SIMPLIFICATIONS 2456

For practical implementation, we can first consider working by PRB instead of working by tone. In 2457 practice, the difference between both has been observed as non-significant. This means that we 2458 will use for pre-processing SNR the values obtained for noise and energy estimation per PRB and 2459

per antenna. ( )2

2

,

, 2j

jipilotji

HGKSNR

σ+≈ 2460

We can use the unbiased estimation of the instantaneous useful signal power on a given PRB by 2461 the formula (it has to be forced to zero if it is negative): 2462 2463

( ) ( ) ( )( )0,ˆmax 2,PRB

BiasedPRBPRB aaEaE σ−= 2464

2465

( )2,PRBˆ aσ 2466

2467

We then have ( ) ( )( )2

,PRB

PRB

ˆ,

a

aEaiSNR

σ≈ and ( ) ( )

( )2,PRB

PRB,

ˆ2

a

pilotai

aEGKSNR

σ+≈ 2468

2469

In order to compute the post-IDFT formula

∑

∑

+

+=

iCombi

iCombi

Combi

SNR

SNR

SNR

SINR

1

11

, we can use a look-2470

up table for the function 1+

→x

xx for x varying in a range of [-5dB, 30dB] with 3dB 2471

granularity. 2472

Moreover, since 11

1

1

1

+=

+ x

x

x, we can use this table for the computation of both

1+Combi

Combi

SNR

SNR 2473

and 1

1

+CombiSNR

. 2474

2475 The exact fixed point implementation will be described in a further version of the document. 2476 2477

Supprimé : 4.0.1

Supprimé : 2







6.3.7.3 FALLBACK SOLUTION 2478

In practice, the computation of the post-IDFT SNR is a heavy task. A simple sub-optimal 2479

alternative for LA2.x is to just use a linear averaging instead of 2480

∑

∑

+

+=

PRB PRB

PRB PRB

PRB

11

1

Comb

Comb

Comb

SNR

SNR

SNR

SINR . This amounts to ∑≈PRB

CombSNRSINR PRBN_PRB

1. 2481

Notice however that the degradation is not negligible if the channel varies significantly in the 2482 frequency domain on the allocated bandwith (e.g. high number of PRB, or frequency selective 2483 channel like ETU). 2484

2485

6.4. CHANNEL ESTIMATION 2486

Channel estimation is used to give an estimate of the channel coefficients for each tone of each 2487 user, on each Rx antenna. This is done with two successive procedures, which are frequency 2488 domain filtering, and time domain filtering. 2489

6.4.1 FREQUENCY DOMAIN FILTERING 2490

This function is called for each UE, on each antenna and for each pilot block of all TTIs. Its inputs 2491

are the outputspilot

apH ,

~ of the pilot channel estimation. Its output is sent to the time domain filtering 2492

part of channel estimation. 2493 Based on the rough estimated channel coefficients obtained from pilot channel estimation, this 2494

function computes refined channel estimation for pilot blocks by filtering the pilot channel estimates 2495

in the frequency domain. The filtering is performed using a block moving average filter, whose 2496 length is derived from the delay spread estimation if available, or set to a default variable instead. 2497 Some enhancement can be further added using a SNR weighting. 2498

This frequency domain filtering is applied over each pilot blocks and over each antenna. For 2499 clarification purpose however, we will consider only the tone index and drop the antenna, user and 2500

OFDM symbol index. 2501

Notice that it would be better to compensate for the timing offset before any frequency domain 2502 averaging in order to have a smoother channel. However, this is not necessary for channel 2503

estimation as described here since the length of the window over which we average is small (the 2504

recommended value for this window is actually 5). Notice that for PUCCH processing, as we 2505 average over the whole PRB, timing offset compensation will be carried out. 2506

The block moving average is described in the figure below. It uses two windows: 2507

• An internal window of size K on which filtered values will be applied (this means in 2508 particular that the estimated channel is constant over consecutive blocks of size K) 2509

• An external window made by adding G coefficients on each side of the internal window. 2510

This external window contains together with the internal one the input of the filter, i.e. the 2511 values of the channels that will be averaged to estimate the single channel value that will 2512

be considered on the corresponding internal window. G is strictly positive. 2513

2514

Supprimé : 4.0.1

Supprimé : 2







2515

2516

2517

2518

2519

2520

2521

2522

2523

2524

2525

2526

2527

2528

2529

2530

Figure 0-2 :Channel estimation: Frequency domain filtering for pilot blocks 2531 2532

As before, the number of pilot symbols of the UE of interest inside an OFDM symbol is denoted by 2533

M (equal to the number of data symbols), so that

K

M is the number of consecutive blocks of size 2534

K inside the UE frequency band, where x stands for the smallest integer greater or equal to x . 2535

The block index r varies from 0 to

K

M-1. For a given such block, the corresponding tone index 2536

varies in the interval ( ) 11,, −+ KrrK K . 2537

The channel coefficients on the block of index r are all equal and given by the formula: 2538

∑

∑=

ii

piloti

ii

k

g

HgZ

~

where the index in the sum varies inside the interval of size K+2G 2539

{ } ( ){ } 1,1min,,0,max −++−= MGKrGrKi K . The indexes k of the tones where this 2540

estimation is applied vary in the interval ( ) 11,, −+= KrrKk K . We recall that r is the block 2541

index and varies in the interval 1,,0 −

=K

Mr K . 2542

The filter coefficients ig are currently all equal to one, but can be modified to obtain more efficient 2543

estimations (we can use an MMSE weighting for example). Notice that the classical moving 2544 average is a particular case of this algorithm corresponding to K=1. 2545

Tones used to compute the averaged channel

K tones over which the averaged channel is applied

G coefficients

Supprimé : 4.0.1

Supprimé : 2

Supprimé : ¶¶¶¶¶¶¶¶¶







The parameters K and G are based on delay spread estimations and CPU constraints imposed by 2546 the number of divisions required for frequency domain equalization. Notice that if we have enough 2547

CPU, we should take 1=K , i.e. we should use the classical Moving Average filter. 2548

Since we do not have an estimation of the delay spread, a relevant value for G is 2, meaning that 2549 the recommended values are K=1 and G=2, so that we filter over 5 tones. 2550

If we want to save CPU (e.g. for UL MIMO implementation) we have to down-sample the channel 2551 estimate, meaning that we have to take K>1. The value of this parameter is then driven by the 2552 required down-sampling rate to reach a CPU limitation. Notice that the effect on link level 2553

performance may not be negligible, especially with high order modulations. 2554

Notice that as for CFO compensation, the filtering operation is applied across PRB boundaries so 2555 that the reduction of the filter size is thus applied only at the edges of the whole UE band, not on 2556

each PRB band. Interface definition 2557


Signal Name Type I/O Format Size Description

M Integer I uX (see below) 1 Total number of

subcarriers of the UE

K Integer I u3 1

Number of sub-carriers over which

the channel is assumed to be

constant

G Integer I u3 1

Half-length of the external extra-coefficients for

frequency domain channel estimation

pilotapH ,

~ Complex Integer I (12I,12Q) M Pilot channel

estimation

a

pζ Complex Integer

O (12I,12Q) M

Frequency domain filtered channel estimation with

parameters NoiseK

and NoiseG

Flag boolean

I u1 1

Boolean (K==1) to

decide if a

pζ is

computed or not

pHi_a Complex Integer O (12I,12Q) M

Instantaneous channel impulse

response estimation on pilot blocks

(before Time Domain Filtering)

Table 0-1 :Interface definition for Frequency Domai n Filtering for a given antenna 2559

2560

X = 7, 8, 9, 10, 10, 11 for 1.4, 3, 5, 10, 15 and 20MHz. 2561

2562

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Text;Body;8pt;Interligne : simple, Retrait :Gauche : 1,25 cm

Supprimé : ¶Saut de page







2563 Figure 0-15: Frequency domain filtering for pilot blocks (one su ch processing is applied for each UE, on each 2564

Rx antenna, and on each pilot block 2565 2566 Next plot gives the fixed point implementation: 2567

2568

Next table gives the values for GK 2

1

+on 10 bits (unsigned). 2569

2570 K=1;G=1 K=1;G=2 K=1;G=3 K=2;G=0 K=2;G=1 K=2;G=2

GK 2

1

+float

0.3333 0.2000 0.1429 0.5000 0.2500 0.1667

GK 2

1

+10 bits

341 205 146 512 256 171

2571

6.4.2 TIME DOMAIN FILTERING 2572

6.4.2.1 ALGORITHM OVERVIEW 2573

This function is called for each UE, on each antenna and for each TTI. Its inputs are the 2574

outputsapZ of the frequency domain filtering of both pilot blocks, the noise and power estimations 2575

per PRB, and the pilot channel correlations of the UE of interest as described in section 6.2.1. Its 2576 output is sent to the frequency domain equalizer. 2577 Based on the refined channel coefficients of both pilot blocks of a TTI, time domain filtering 2578

computes channel estimates for all data blocks of this TTI. The channel coefficients on pilot blocks 2579

>>10

12

10

(K+2G) inputs

(12I,12Q)

(12I,12Q)

1/(K+2G)

12+log2(K+2G)

Sat 12

Frequency domain

filtering

pilot

,

~apH from pilot

channel estimation [ ]apM

ap

ap ZZZ ,1,0 −= K

to time domain filtering.

Flag [ ]apM

ap

a

p ,1,0 −= ζζζ K

to noise estimation.

Supprimé : 4.0.1

Supprimé : 2







are time domain filtered to obtain the channels on all data blocks. A time domain MMSE filter is 2580

used, which coefficients are computed based on speed and SNR estimations. 2581

The time domain filter coefficients only depend on the speed and SNR of the UE of interest. One 2582

filter is applied per PRB and OFDM symbol, and for a given PRB and a given OFDM symbol, the 2583

same filter coefficients are applied on all tones. Notice that this is different from what was described 2584

in issue 01.03 where the filter coefficients depended on the UE of interest only and not on the PRB. 2585

For tone with index i, antenna with index a, and OFDM symbol m, the filtering operation is 2586

computed as: 2587

( ) ( ) ( )ai

ami

ai

amia

i

aia

miami ZWZW

Z

ZWH 1,1,,PRB0,0,,PRB

1,

0,,PRB,

ˆ +=

= 2588

−=−=

===

1,,0

1,,0

omitted) being 10 and 3( 13,,0

0

RXNa

Mi

mmm

K

K

K

2589

( ) ( ) ( )[ ]ami

ami

ami WWW 1,,PRB0,,PRB,PRB ,= is the 1x2 vector containing the weights of the time domain 2590

channel interpolation filtering for m-th symbol and antenna a, corresponding to the tone number i 2591 (notice that these coefficients are constant over the PRB(i) which stands for the PRB containing 2592 tone number i). 2593

Filter coefficients are different from one PRB to another PRB of the same UE since the noise 2594 estimation is different. 2595

amiH ,

ˆ is the channel estimation output of the time domain channel interpolation filtering for m-th 2596

symbol, i-th sub-carrier, and a-th antenna. 2597

The theoretical expression giving the filter coefficients corresponding to the m-th OFDM symbol 2598

writes: 2599

( )

( ) ( )[ ] ( )( )

( )( )( ) ( )

1

2u

2

PRB

0

000

,PRB

.21pilot(1),pilot(0)

pilot(1),pilot(0)1pilot(1),pilot(0),

−

++

=

IaEGK

mm

W

i

ami

σγ

γγγ

2600

2601

Where: 2602

• ( )pilot(0),0 mγ is the correlation between channels of symbol number m and the first pilot 2603

block, and ( )pilot(1),0 mγ is the same for the second pilot block. 2604

• ( )pilot(1),pilot(0)0γ is the correlation between channels of both pilot blocks 2605

• ( )

( )( )2iPRB

u

σaE

is the estimated SNR on antenna a and PRB containing the tone number i as 2606

computed in section 6.3.2 2607

Notice that there is an additional factor (K+2G) in the SNR estimation. This is because the noise 2608

power used to compute the filters is the noise power experienced at the input of the time domain 2609

filtering, i.e. the noise power after frequency domain filtering, which is equal to the noise power 2610

estimated in section 6.3.2 divided by the processing gain obtained from the filtering operation. 2611

Supprimé : 4.0.1

Supprimé : 2

Supprimé :







Actually, making the assumption that the channel is approximately constant over the Moving 2612

Average window of length (K+2G) used for filtering, we can make the approximation that the 2613

Moving Average processing divides the noise power by (K+2G). 2614

Notice that edge effects on each PRB are considered as negligible. If they were taken into account, 2615

we should have a special processing for the edge tones of each PRB, resulting in a more CPU 2616

greedy algorithm. 2617

The relation between speed and correlation coefficients depends on the fading model we consider. 2618 For Jakes fading, we have the relation: 2619

( ) ( )( )dm ftJm πγ 2pilot(0), pilot(0),00 ∆= 2620

Here, cd fc

fυ= is the maximum Doppler shift, cf is the carrier frequency, 810.3=c is the speed 2621

of light, and υ is the mobile speed. pilot(0),mt∆ is the time difference (expressed in seconds) 2622

separating OFDM symbol m with the first pilot block. The same holds for the second pilot block. 2623

6.4.2.2 FILTERS COEFFICIENTS COMPUTATION 2624

In practice, we use an alternative to the mathematical formula of the previous section to compute 2625 the MMSE filters. The filter coefficients are pre-stored for different quantized values of noise, useful 2626 signal powers, and speed. 2627

Additionally, we describe here how to save calculation resources by adjusting the parameter 2628

tMMSE_outpuN . tMMSE_outpuN is the number of estimations per TTI, assumed regularly spaced in time, 2629

i.e. the number of filters amW per TTI for a given PRB. 2630

Ideally, time domain interpolation should compute one such channel filter per OFDM symbol, i.e. 12 2631

estimations per tone and per TTI. However, if necessary, this number can be reduced depending 2632

on the speed and a resource limitation NMMSEmax indicator. This is the case for example if we want 2633

to limit the number of divisions in the frequency domain equalizer, and in particular for MIMO 2634

processing. 2635

The calculation of NMMSEmax is out of the scope of this document and depends on implementations 2636

constraints. 2637

2638

Supprimé : 4.0.1

Supprimé : 2







2639 Figure 0-16: Time domain MMSE filter coefficient computation 2640

2641

Notice that the noise variance and user power can be short term and long term ones, and are 2642 selected as indicated in section 6.3.4. The bit widths of both quantities are equal to 24 bits as short 2643 term metrics are used. 2644

2645

The SNR computation inside the filter selection process is performed using a dichotomic search as 2646 described below with some prestored SNR values which are: 2647

The SNR thresholds are: 2648

Index (s) SNR (dB)

0 -3

1 0

2 3

3 6

4 9

5 12

6 15

7 18

8 21

9 24

10 27

11 30

2649

Estimates of noise variance

( )2PRBσ (per

PRB)

LUT

Apply time domain filtering

Filter coefficients

selection (one per PRB)

( )GK 2

1

+×

Channel estimates

( )auE

Estimates of signal power

Average over all PRBs of the UE

of interest

Speed estimate of the UE of

interest

NMMSEmax

default value = 4

NMMSE_output computation

NMMSEoutput

Supprimé : 4.0.1

Supprimé : 2







The dichotomic search described below works the following way: If the entry of the dichotomic 2650 search is found as being in an interval [S-3, S], then the filter constructed with the SNR S is 2651 selected. 2652

However, we notice that the filter with SNR S has to be selected for effective signal to noise ratio at 2653 the MMSE input in the interval [S-1.5, S+1.5]. 2654

Therefore, the entry at the input of the dichotomic search has to be equal to this effective signal to 2655 noise ratio shifted to the left by 1.5dB, so that an entry in [S-3, S] corresponds to an the effective 2656 signal to noise ratio in [S-1.5, S+1.5]. 2657

This 1.5dB shift to the left can be done by scaling the noise at the input of the dichotomic search. 2658 Since noise has already to be scaled by 1/(K+2G), we can instead multiply it by 2659 (1/(K+2G)).10^(1.5/10), which corresponds to 289 in u10 instead of 205 for (1/(K+2G)) scaling. 2660

Multiplication of noise by (1/(K+2G)).10^(1.5/10), is realized as follows: 2661

2662

As we only have multiple or fraction of 2, the threshold multiplication is realized through shift 2663

>>10

289

u24

u22 Noise for T-MMSE

Noise (prb,a)

Supprimé : 4.0.1

Supprimé : 2







2664

Then the output is the selected filter indexed by the speed index t (see 6.3.5.1) and the SNR index 2665 s. 2666

redMMSE_requiN is the number of estimations per TTI, assumed regularly spaced in time, i.e. the 2667

number of filters amW per TTI required which is required for the speed of the UE of interest. At low 2668

speeds, one time domain estimation per tone and per TTI is sufficient, and at higher speeds, more 2669 estimations are necessary to track the channel variations. 2670

E≥(N<<2)

E≥(N<<4)

E≥(N<<3)

4 5

Y

E≥(N<<7)

E≥N

E≥(N>>1)

3 2 0 1

Y

Y

E≥(N<<1)

Y

Y

E= ( )auE

N= ( )2PRBσ .(1/(K+2G)).x1.4125

E≥(N<<8)

E≥(N<<6) E≥(N<<9)

E≥(N<<5)

6 7 8 9 10 11

Y Y

Y

Y

Y

Y

Supprimé : 4.0.1

Supprimé : 2







2671 Figure 0-17: computation of N MMSEoutput 2672

We can establish a mapping between the speed and the value of redMMSE_requiN . For the speed of 2673

Doppler frequency of 231.5Hz (120km/h @ 2GHz) which is the maximum speed for LA3.0, a 2674

maximum number of 4redMMSE_requi =N (i.e. one MMSE filter per 3 OFDM symbols) should be 2675

sufficient to track the time domain variation. 2676

A mapping between Doppler frequencies and redMMSE_requiN is necessary, and is be constructed as 2677

follows: 2678

• 1redMMSE_requi =N if 1dd ff ≤ 2679

• 2redMMSE_requi =N if 2d

1d fff d << 2680

• 4redMMSE_requi =N if dff ≤2d 2681

Of course, the case 4redMMSE_requi =N is also adapted to any speed targeted by LA3.0 since this is 2682

the most general case. 2683

In practice, see section 6.3.2.3, the possible estimated speeds are quantized to the 5 possible 2684 values: 3km/h, 25km/h, 50km/h, 75km/h, 100km/h, and 125km/h. The number of MMSE 2685 coefficients is equal to: 2686

• 1redMMSE_requi =N for Doppler frequency of 5.5Hz (3km/h @ 2GHz) 2687

• 2redMMSE_requi =N for Doppler frequencies of 46.3Hz and 92.6Hz (25 and 50km/h @ 2688

2GHz) 2689

• 4redMMSE_requi =N for Doppler frequencies of 138.9 and above (75km/h and above 2690

@2GHz) 2691

2692

The following figure describes the mapping between redMMSE_requiN and UE speed: 2693

Speed estimate of the UE of

interest

NMMSEmax

default value = 4

NMMSEoutput

NMMSE required

NMMSE_required

computation

MIN

Supprimé : 4.0.1

Supprimé : 2







2694 Figure 0-18 : Number of MMSE coefficients depending on the speed 2695

2696



Sigma2_PRB integer I u24 (ST) M/12 Noise power on the PRB

UE_Energy integer I u24 (ST)

1 Energy of the UE over its

entire band

df enum I {…} 1 Doppler frequency Estimation

NMMSEmax integer I

1 Default value is 4 for LA3.0

speed < 120 km/h

W Integer O

s12

Defined by estimated speed (1,

2 or 4 per PRB)

Time domain MMSE weigths per PRB

2698

Table 0-2 :Interface definition for MMSE weights co mputation 2699

2700

6.4.2.3 FILTER APPLICATION 2701

2702 Figure 0-19: Time domain filtering to obtain the channel on OFDM symbol m (one such processing is applied 2703

for each UE, on each Rx antenna, and on each data b lock) 2704 2705

130 WW == L

60 WW ==L

1dd ff ≤

136 WW ==L

dff ≤2d

2d

1d fff d <<

[ ]amM

am

a

m HHH ,1,0ˆˆˆ

−= K

m=0, …, 13

Time domain filtering

[ ]apM

ap

ap ZZZ ,1,0 −= K

From frequency domain filtering p=0, 1

Weights a

mPRBW ,

Supprimé : 4.0.1

Supprimé : 2










Hpilot Complex Integer

I (12I,12Q)

(Used subcarriers per pilot blocks) * (Number of pilot

blocks)

Frequency domain filtered channel

estimates

ppHi Complex Integer

O (12I,12Q) Used subcarriers per TTI

Instantaneous channel impulse response

estimation

W Complex Integer

I s12 (Number of data blocks per TTI)

MMSE weights

Table 0-3:Interface definition for Time Domain Filt ering for one antenna/PRB 2708

2709

2710 2711

Next figure gives the fixed point implementation. 2712 2713 Format under which the filters are stored is W=sat12(round(W*(1<<10)))) (see Annex 5 for the details). 2714

2715 Figure 0-20 Time domain filtering application for a given pilot symbol (We need to add the contri bution of 2716

the second pilot to obtain the channel estimate) 2717

(12I,12Q)

LUT

12

Doppler snr

>>11

(12I,12Q)

23 24

12

Sat12 13

Supprimé : 4.0.1

Supprimé : 2







2718

6.5. DEMODULATION 2719

6.5.1 FREQUENCY DOMAIN EQUALIZER 2720

Frequency Domain Equalizer is necessary to cancel the effect of DFT spreading at the 2721

transmission (without DFT, a simple MRC would be sufficient). The MMSE works tone by tone and 2722

combines both observations on the Rx antennas with an MRC combiner, and then divide by a 2723

MMSE coefficient to obtain the desired signal. Notice that we have to take into account the fact that 2724

the noise power is different from one PRB to the other. In this section, we omit the PRB index for 2725

simplicity. 2726

GENERAL FORMULA 2727

The input of the MMSE is the sequence [ ]amM

am

am YYY ,1,0 −= K obtained from data blocks 2728

frequency offset compensation. To derive the general formula of the MMSE processing, we neglect 2729

all indexes except the antenna index, and we thus write aY instead of amiY , . 2730

Neglecting residual ICI, we have: 2731

nsPHY += 2732

Where : 2733

(hereafter the index NRX-1 is considered only when NRX >1) 2734

•

=

−1

0

...

RXNY

Y

Y is the received signal vector from antenna 0 to NRX-1 respectively,. 2735

•

=

−1

0

...

RXNH

H

H is the channel on all antennas for the tone of interest as seen from the pilot blocks. 2736

In practice, we rather have their estimated values

=

−1

0

ˆ

...

ˆ

ˆ

RXNH

H

H , which are obtain as the outputs 2737

of the time domain MMSE filtering. 2738 2739 • s is the transmitted complex scalar information of user of interest (i.e. after DFT). We assume that 2740

these symbols have unit average energy (this is the case if the QAM symbols have unit energy 2741

since the DFT conserves the energy): ( ) 12 =Ε s . In the case where the DFT is not normalized, the 2742

corresponding normalization can be included in the power P. 2743

Supprimé : 4.0.1

Supprimé : 2






Mis en forme : Anglais(Royaume-Uni)•

=

−1

0

...

RXNn

n

n is the noise vector on all antennas respectively, ),0(~ 2aa Nn σ , for a = {0…NRX-1}. 2744

Let us denote:

=

−2

1

21

20

...00

0...00

0...0

0...0

RXN

D

σ

σσ

and ( )∏−

=

=1

0

2RXN

aaσγ . In practice, we rather have 2745

estimated values ( )2ˆ aσ of ( )2aσ with a = {0…NRX-1}, obtained from noise estimation function. They 2746

are short term as described in section 6.3.4. The bit widths are equal to 24 bits. 2747

2748 • P is the transmit power for the user of interest, estimated by the receiver. Notice that this power 2749

can be included in the channel coefficients, so that we can use the model with 1=P without loss 2750 of generality. 2751 2752 2753 Then the transmitted information can be estimated as below: 2754

{ } YDHIHDHs HH 111ˆ −−− += γγγ 2755

( ) ( )

( ) ( ) ∏∑ ∏

∑ ∏−

=

∗−

= ≠

∗−

= ≠

+

=1

0

221

0

2

1

0

2

ˆRXRX

RX

N

aaaa

N

a ajj

aa

N

a ajj

HH

YH

s

σσ

σ 2756

In practice, this formula can be implemented in two successive steps which are described below. 2757 With the hypothesis of identical noise variance on all antennas, and with 2758

( ) ( )∑−

=

=1

0

22 ˆ1

ˆRXN

aa

RXNσσ , this can be simplified to: 2759

2760

( )

( )21

0

2

1

0ˆ

σ+

=

∑

∑−

=

−

=

∗

RX

RX

N

aa

N

aaa

H

YH

s 2761

2762 2763 FILTERS COMPUTATION 2764

This function is called once per block of K consecutive tones and tMMSE_outpu

12

N consecutive 2765

OFDM symbol (the parameters K and tMMSE_outpuN are defined in the channel estimation sections 2766

6.4.2.2). Actually, since the channel estimation is constant over rectangles of size 2767

( ) ( )symbols tones tMMSE_outpuNK × , the filters will be constant over these rectangles and equal to: 2768

( )( )22

0

0

ˆˆ

ˆ

σ+

∗

H

H, if NRX=1 2769

Or to 2770

Supprimé : 4.0.1

Supprimé : 2






Mis en forme : Anglais(Royaume-Uni)( )

( ) ( )

++

∗∗

1022

1

2

0

ˆˆˆˆˆ

1HH

HH σ , if NRX=2 2771

2772 Or to 2773

( )( ) ( ) ( ) ( )

++++

∗∗∗∗

321022

3

2

2

2

1

2

0

ˆˆˆˆˆˆˆˆˆ

1HHHH

HHHH σ if NRX=4 2774

2775 Doing so, a single division per rectangle is used. 2776

2777

2778 2779

Figure 0-21: Filter coefficients computation. One such processin g is applied for each rectangle of size 2780

( ) ( )symbols tones tMMSE_outpuNK × 2781

2782 FILTER APPLICATION 2783 2784 This function is called once per allocated tone and OFDM symbol. The inputs are the filters 2785 computed above and the sequence obtain from data blocks frequency offset compensation. Using 2786 the same notations as above, filter application consist in applying the filters to the input vector as: 2787

( )( )

ˆˆ

*ˆˆ

22

0

00

σ+=

∗

H

YHs , if NRX=1 2788

Or 2789

( )( ) ( )

⋅

++=

∗∗

1

010

22

1

2

0

ˆˆˆˆˆ

1ˆ

Y

YHH

HHs

σ if NRX=2 2790

2791 Or 2792

( )( ) ( ) ( ) ( )

⋅

++++=

3

2

1

0

*

3

*

2

*

1

*

022

3

2

2

2

1

2

0

ˆˆˆˆˆˆˆˆˆ

1ˆ

Y

Y

Y

Y

HHHHHHHH

sσ

if NRX=4 2793

2794 2795

Currently, the 3GPP assumption is that the pilot/data offset is equal to one. This offset has to be 2796 removed as 3GPP has decided that it is constant and equal to 1 in any situation. 2797

2798

Filter coefficients used for frequency domain equalizer

Normalized Noise

estimation ( )2σ

Channel estimation a

mH

from time domain MMSE Filters computation

Supprimé : 4.0.1

Supprimé : 2







2799 Figure 0-22: Sequence estimation obtained by filters application s. One MMSE filtering is applied for each tone 2800

on each data OFDM symbol. 2801 2802 MMSE filter computation fixed implementation is depicted on next graphs. The data/pilot power offset is 2803 equal to 1. 2804 2805



2808

2809


NoiseAv Integer u24 (ST) I 1 Noise power averaged over

Main and Div antennae


H_Main Complex Integer

(12I,12Q) I 0M


mode 1Rx)

H_Div Complex Integer

(12I,12Q) I 0M Estimated Channel on the Div

antenna


(12I,12Q) I 0M

Estimated Channel on the 2nd Div antenna, (when

applicable)


(12I,12Q) I 0M

Estimated Channel on the 3rd Div antenna, (when

applicable) Filter

coefficients main Rx

Complex Integer

(8I,8Q) O

0M Filter coefficients on the Main antenna

Filter coefficients

Div Rx

Complex Integer

(8I,8Q) O

0M Filter coefficients on the Div antenna, (only antenna in

mode 1Rx)

Filter coefficients

Div2 Rx

Complex Integer

(8I,8Q) O

0M Filter coefficients on the 2nd

Div antenna, (when applicable)

Filter coefficients

Div3 Rx

Complex Integer

(8I,8Q) O

0M When applicable, Filter

coefficients on the 3rd Div antenna, (when applicable)

m Integer u4 O 0M Scaling factor (order)

Filter coefficients obtained above

Frequency domain MMSE

equalization Sequence amY obtain from

data blocks frequency offset compensation

Sequence

[ ]mMmm sss ,1,0 ˆˆˆ −= K

of estimated transmitted DFT-spread symbols.

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Couleur depolice : Noir























Mis en forme ... [3]







2810 Table 0-4:Interface definition for filter coefficients calcul ation MMSE SIMO per PRB and per OFDM 2811

symbol 2812 2813

The hereafter function denoted “order” gets the number of bits representing the sign before the first 2814 significative bits. For instance: 2815 s8 i=10000111; order(i)=0; 2816 u8 i=00000111; order(i)=5; 2817 2818 Note that β is equal to 1 (according to 3gpp). 2819 2820 2821 MMSE filter computation fixed implementation for NRX=1 is depicted on next graph. 2822 2823

2824 2825 MMSE filter computation fixed implementation for NRX=2 is depicted on next graph. 2826 2827 2828

|.|2

|.|2

>>8

order

>>(8-min(m,8))

m

(12I,12Q)

23

22

16

Sat 16

Sat 8

>>g*

z=f(x,y)

z=f(x,y)

z y x

Noise estimate After averaging on NRX antennas

Supprimé : 4.0.1

Supprimé : 2







2829 2830 • g=3 if short term measurement (recommended behaviour) and g=2*e+1 if long term measurement. 2831 2832 MMSE filter computation fixed implementation for NRX=4 is depicted on next graph. 2833 2834

|.|2

|.|2

|.|2

|.|2

>>8

(12I,12Q)

order

>>(8-min(m,8))

m

(12I,12Q)

12 22 23

24

22

17

Sat 16

Sat 8

>>g*

z=f(x,y)

z=f(x,y)

z=f(x,y)

z=f(x,y)

z y x

Noise estimate After averaging on NRX antennas

Supprimé : 4.0.1

Supprimé : 2







2835 2836 g=3 if short term measurement (recommended behaviour) and g=2*e+1 if long term measurement 2837 2838 Hereafter, F(x,y) implementation for all NRX values: 2839 2840 2841

|.|2

|.|2

|.|2

|.|2

>>8

(12I,12Q)

order

>>(8-min(m,8))

m

(12I,12Q)

12 22

23

25

22

18

Sat 17

Sat 8

>>g*

z=f(x,y)

z=f(x,y)

z=f(x,y)

z=f(x,y)

z y x

|.|2

|.|2

(12I,12Q)

(12I,12Q)

12 22

z=f(x,y)

z=f(x,y)

z=f(x,y)

z=f(x,y)

23

|.|2

|.|2

(12I,12Q)

(12I,12Q)

12 22

23

23

Noise estimate after averaging on NRX antennas

Supprimé : 4.0.1

Supprimé : 2







2842 2843 2844 2845 2846 Error case: y=0 2847 2848 if(x>0) 2849

z=0x7f 2850 else 2851 z=0x80 2852 2853 MMSE filter application fixed implementation is depicted on next graphs for the real part – imaginary part 2854 can be straightforward deduced: 2855 2856 Case NRX=1 2857

2858 2859 Case NRX=2 2860 2861 2862 2863 2864 2865 2866 2867 2868 2869 2870 2871 2872 2873 2874 2875 2876 2877 2878 2879 2880

8 (12I,12Q)

13

>> (17-min(m,8))) Sat 8

(8I,8Q)

20

<<6 a/y Sat8 x

y

z >>1

a

(12I,12Q)

12

8 (12I,12Q)

13

>> (17-min(m,8)))

Sat 8

(8I,8Q)

(8I,8Q)

19 20

21

Supprimé : 4.0.1

Supprimé : 2







2881 Case NRX=4 2882

2883 2884 2885


m Integer u4 I 0M Scaling factor (order)

DataMain Complex Integer

(12I,12Q) I

0M Received Data bits for the12 data blocks of

the TTI on the Main antenna (only antenna in mode 1Rx).

DataDiv Complex Integer

(12I,12Q) I

0M Received Data bits for the12 data blocks of the TTI on the Div antenna.

DataDiv2 Complex (12I,12Q) I 0M Received Data bits for the12 data blocks of

the TTI on the 2nd Div antenna (when

(12I,12Q)

12

8 (12I,12Q)

13

>> (17-min(m,8)))

Sat 8

(8I,8Q)

(8I,8Q)

19 20

22

(12I,12Q)

12

(8I,8Q)

(12I,12Q)

12

20

20

20

(8I,8Q)

Supprimé : 4.0.1

Supprimé : 2













Integer applicable)

DataDiv3 Complex Integer

(12I,12Q) I

0M Received Data bits for the12 data blocks of

the TTI on the 3rd Div antenna, (when applicable)

Filter coefficientsMain

Complex Integer

(8I,8Q) I

0M Filter coefficients on the Main antenna (only antenna in mode 1Rx).

Filter coefficientsDiv

Complex Integer

(8I,8Q) I

0M Filter coefficients on the Div antenna

Filter coefficientsDiv2

Complex Integer

(8I,8Q) I

0M Filter coefficients on the 2nd Div antenna, (when applicable)

Filter coefficientsDiv3

Complex Integer

(8I,8Q) I

0M Filter coefficients on the 3rd Div antenna, (when applicable)

FilteredData Complex Integer

(8I,8Q) O

0M Filtered Data through MMSE

2886 Table 0-5:Interface definition for filter application MMSE SI MO per PRB and per OFDM symbol 2887

2888

6.5.2 IDFT 2889

This function is called for each UE, once per data OFDM symbol. Its inputs are the outputs ms of 2890

the frequency domain equalization, and its output is sent to the QAM demapper. 2891 This is used to invert the effect of the DFT spreading. 2892

At the transmitter, the sequence of QAM symbols of a given user on OFDM symbol of index m is 2893

[ ]mMmm SSS ,1,0 −= K . Its DFT [ ]mMmm sss ,1,0 −= K is mapped in the frequency domain 2894

and is estimated from the frequency domain equalizer. We ignore the OFDM symbol index in the 2895

following and describe the algorithm to compute the IDFT of the sequence [ ]10 −= Msss K . 2896

∑−

=

=1

0

2exp1 M

kki M

ikjs

MS π 2897

2898

One shall notice that M

1factor preserves the energy. Appropriate shifts to include this factor in 2899

the non-normalized Xilinx DFT are specified in the fix implementation section. 2900 2901

RADIX DECOMPOSITION 2902

Since the IDFT size is not a power of two, we cannot apply the FFT algorithm. A radix algorithm 2903

based on prime decomposition is used instead. 2904

Basically, it consists of a projection onto the Fourier basis of size M for each user (we recall that 2905 M denotes the number of QAM symbols per OFDM block and user). It can be performed partly 2906 using the same principle as for FFTs. 2907 2908

Supprimé : 4.0.1

Supprimé : 2

























Notice that since as explained in TS 36.211, a scaling byM

1 is applied at the transmitter when 2909

performing the DFT pre-coding, we have to apply a scaling at the receiver to keep the same 2910 amplitudes for QAM symbols. This means that the DFT definition used here writes: 2911 2912

∑−

=

=1

0

2exp1 M

kki M

ikjs

MS π 2913

2914 2915 The algorithm uses prime radix: 2916 2917 First, let us write the DFT of a size M vector where qpM ×= (we describe below the algorithm 2918

for the classical scaling by M but this is the same for the square root). 2919

( )

∑ ∑

∑∑∑−

=

−

=+

−

=

−

=+

−

=

=

+=

=

1

0

1

0

1

0

1

0

1

0

2exp1

2exp1

2exp1

2exp1

p

k

q

lklp

p

k

q

lklp

M

kki

q

iljs

qpq

ikj

p

pq

klpijs

Mn

ikjs

MS

ππ

ππ 2920

2921 Furthermore S has q periodicity: 2922

∑

∑ ∑−

=

+

−

=

−

=++

=

=

1

0

1

0

1

0

1

2exp1

2exp2exp1

p

k

ki

rkqikpq

p

k

q

lklprqi

Swp

q

iljs

qp

rkj

pq

ikj

pS πππ

2923

2924

where kiS is the i-th element of the IDFT of vector ( )pqkpkk sss )1(,....,, −++ . 2925

2926 So the DFT of the pxq vector can be decomposed into p DFTs of size q. One can now iterate and 2927 reproduce the same algorithm for the q size DFTs. 2928 2929 So, let us consider the prime factor decomposition of n = n1xn2x…np-1. We thus can perform the 2930 IDFT with recursive implementation using n time the function. 2931 2932 For example for 553222600 ×××××==M , the implementation will require 6 stages of 2933 respective sizes 2, 2, 2, 3, 5 and 5. 2934 2935 The IDFT is performed using a look up table lut containing order pq unit root and providing as 2936

function argument the size and its prime factor decomposition. 2937

2938 2939

Figure 0-23:QAM symbols estimation obtained by IDFT. One IDFT i s applied for each user on each data OFDM 2940 symbol. 2941

Sequence

[ ]mMmm SSS ,1,0ˆˆˆ

−= K

of estimated QAM symbols, input of QAM demapper

IDFT

Sequence

[ ]mMmm sss ,1,0 ˆˆˆ −= K

obtained from frequency domain equalization

Supprimé : 4.0.1

Supprimé : 2







IDFT INTERFACE DEFINITION 2942


2944


M Integer u11 I 1 Number of subcarriers of the UE


(8I, 8Q) I 2 x M samples (time domain)


(8I, 8Q) O 2 x M samples (freq domain)

DShift Integer u2 O 1 Shift used for QAM demodulation scaling

2945 Table 0-6 : iDFT interface definition 2946

2947 2948 2949 2950 2951 2952 The Xilinx’s DFT block v1.0 will be used as follows: 2953 2954

2955 0-24: iDFT fix implementation 2956

2957

2958

COMPUTATION OF R 2959

r is used to right shift the output data 2960

Input: s,M and table below w(M) 2961

Output : r 2962

Algorithm : 2963

Xilinx (18 bits) iDFT

∑−

=

=1

0

2expM

kki M

ikjsS π

(18I,18Q)

M outputs

(18I,18Q)

Shift s

>> r saturation rounding right shift

…

…

(8I,8Q)

…

M inputs

(8I,8Q)

…

<< 10

calculates r

r

DShift

calculates DShift

Supprimé : 4.0.1

Supprimé : 2







IF(w(M) > s ) 2964

r=w(M)-s 2965

ELSE 2966

r=0 (no shift, no rounding) 2967

ENDIF 2968

COMPUTATION OF DSHIFT 2969

dshift is a scale indication used by the next stage (QAM demapping). 2970

Input: s,M and table below w(M) 2971

Output : d 2972

Algorithm : 2973

IF( w(M) >= s ) 2974

dshift=0 2975

ELSE 2976

dshift =(s-w(M)) 2977

ENDIF 2978

PARAMETERS 2979

Here are the Xilinx’s block parameters : 2980

n2 Power of 2 prime decomposition n3 Power of 3 prime decomposition n5 Power of 5 prime decomposition direction 0 (inverse)

2981 Here is additional information relative to the scaling: 2982

v Indicates how the scaling is done w Related to the right shift of output data

2983 The parameters depending on the size M are : 2984

M input size

prime decomposition n2

n3

n5 Input bw

Output bw

w In/out scale v

M=12 12=2x2x3 2 1 0 8 8 12 3.0000 M=24 24=2x2x2x3 3 1 0 8 8 13 3.0000 M=36 36=2x2x3x3 2 2 0 8 8 13 4.5000 M=48 48=2x2x2x2x3 4 1 0 8 8 13 6.0000 M=60 60=2x2x3x5 2 1 1 8 8 13 7.5000 M=72 72=2x2x2x3x3 3 2 0 8 8 14 4.5000 M=96 96=2x2x2x2x2x3 5 1 0 8 8 14 6.0000 M=108 108=2x2x3x3x3 4 3 0 8 8 14 6.7500 M=120 120=2x2x2x3x5 3 1 1 8 8 14 7.5000 M=144 144=2x2x2x2x3x3 4 2 0 8 8 14 9.0000 M=180 180=2x2x3x3x5 2 2 1 8 8 14 11.2500 M=192 192=2x2x2x2x2x2x3 6 1 0 8 8 14 12.0000 M=216 216=2x2x2x3x3x3 3 3 0 8 8 14 13.5000 M=240 240=2x2x2x2x3x5 4 1 1 8 8 14 15.0000 M=288 288=2x2x2x2x2x3x3 5 2 0 8 8 15 9.0000 M=300 300=2x2x3x5x5 2 1 2 8 8 15 9.3750

Supprimé : 4.0.1

Supprimé : 2







M=324 324=2x2x3x3x3x3 2 4 0 8 8 15 10.1250 M=360 360=2x2x2x3x3x5 3 2 1 8 8 15 11.2500 M=384 384=2x2x2x2x2x2x2x3 7 1 0 8 8 15 12.0000 M=432 432=2x2x2x2x3x3x3 4 3 0 8 8 15 13.5000 M=480 480=2x2x2x2x2x3x5 5 1 1 8 8 15 15.0000 M=540 540=2x2x3x3x3x5 2 3 1 8 8 15 16.8750 M=576 576=2x2x2x2x2x2x3x3 6 2 0 8 8 15 18.0000 M=600 600=2x2x2x3x5x5 3 1 2 8 8 15 18.7500 M=648 648=2x2x2x3x3x3x3 3 4 0 8 8 15 20.2500 M=720 720=2x2x2x2x3x3x5 4 2 1 8 8 15 22.5000 M=768 768=2x2x2x2x2x2x2x2x3 8 1 0 8 8 15 24.0000 M=864 864=2x2x2x2x2x3x3x3 5 3 0 8 8 15 27.0000 M=900 900=2x2x3x3x5x5 2 2 2 8 8 15 28.1250 M=960 960=2x2x2x2x2x2x3x5 6 1 1 8 8 15 30.0000 M=972 972=2x2x3x3x3x3x3 2 5 0 8 8 15 30.3750 M=1080 1080=2x2x2x3x3x3x5 3 3 1 8 8 16 16.8750 M=1152 1152=2x2x2x2x2x2x2x3x3 7 2 0 8 8 16 18.0000 M=1200 1200=2x2x2x2x3x5x5 4 1 2 8 8 16 18.75

Table 0-7: iDFT fix implementation parameters 2985 2986

The column v is here for information only, it shall not be implemented explicitly. The value v 2987 indicates how the scaling is done. Let’s denote the input integers representing the real 2988

values [ ]mMmm sss ,1,0 −= K as integers [ ]mMmm nnn ,1,0 −= K . Let’s denote the output 2989

integers representing the real values [ ]mMmm SSS ,1,0 −= K as integers 2990

[ ]mMmm NNN ,1,0 −= K . Let’s suppose that inputs are represented as Q7.0 so that 2991

[ ] [ ]mMmmMm nnss ,1,0,1,0 −− = KK . Then (neglecting the rounding effect), 2992

dshift

mkmk

v

NS

2,

, ⋅≈ 2993

This scaling factor dshiftv 2⋅ is further used by the QAM demodulation (LLR). 2994

w has been calculated as follows : w=ceil(log2(sqrt(M)))+p where p=10 is the left shift. 2995

v has been calculated as follows : v=M./(2.^(w-p)) 2996

Maximum value of s is 15, minimum value of w is 12 so dshift is {0,1,2,3} and dshiftmax=3. 2997

6.5.3 QAM DEMAPPING AND LLR COMPUTATION 2998

After IDFT, we have an estimation of the transmitted QAM symbols that need to be converted into 2999

Log-Likelihood Ratios (LLR) for each bit. This is done using LLR approximations to avoid high 3000

complexity computations. 3001

The special case of LLR corresponding to Ack-Nack positions in case of simultaneous transmission 3002

of PUSCH and Ack-Nack is described in section 9. Notice that this includes the LLR corresponding 3003

to CQI transmission which are thus processed the same way as the data LLR. 3004

3005

The definition of the LLR for a given bit adopted below is the following: 3006

3007

Supprimé : 4.0.1

Supprimé : 2







( ) ( )( )

===

0

1log

bP

bPbLLR 3008

This definition is compliant with the definition of LLRs used in the Xilinx Turbo-Decoder. 3009

Data 3010

QPSK 3011

Saturated soft output of IDFT directly provides the soft bits that fed the decoder. However, since 3012 the QPSK mapping is done as 0->1, 1-> -1, we need to reverse the sign of the IDFT output before 3013

feeding the decoder. 3014

3015

3016

16QAM 3017

3018 Figure 0-25: 16QAM mapping 3019

3020

Demapping is performed as followed: let us denote S one soft output of iDFT. 3021

The sign of its real and imaginary parts provide the quarter to which the symbol belongs to and is 3022

given by: ( ) ( )0)Im(0)Re(2 ≤+≤= SSq . 3023

Then we define ( )( ) ( )( )0)Im(21..0)Re(21. ≥−+≥−+= SdjSdSz where d is the distance between 3024

two adjacent symbols ( 10/2=d for the considered constellation when normalized). 3025

8

8

( )1−×

S12=1100 S4=0100 S6=0110 S14=1110

S13=1101 S5=0101 S7=0111 S15=1111

S8=1000 S0=0000 S2=0010 S10=1010

S9=1001 S1=0001 S3=0011 S11=1011

2

0

1 3 d

(Re(S), Im(S)) (- Re(S), - Im(S))

Supprimé : 4.0.1

Supprimé : 2







Finally, the symbol index inside the quarter is:

( ) ( )( ) ( )( ) ( )( ) ( )

≤+≤==

≥+≤==

≤+≥==

≥+≥==

0)Im(0)Re(2,3

0)Im(0)Re(2,2

0)Im(0)Re(2,1

0)Im(0)Re(2,0

zzrq

zzrq

zzrq

zzrq

3026

3027

And the estimated symbol is rqsS += 4ˆ where the subscript “Hard” is for hard decision. 3028

The i-th soft bit of symbol k is given by the LLR: 3029

( )( )( )( )( )

−−

−−

+−=+∑

∑

−=∈

=∈

)(ˆ)(16

2

)(ˆ)(16

2

exp

exp

log)4(ˆ4

ibibQAMs

postm

ibibQAMs

postm

sSSNR

sSSNR

ikbikLLR 3030

where b(k) is the k-th bit of symbol s=[b(0) b(1) b(2) b(3)], and ( )postmSNR is the post IDFT SNR for 3031

the considered long block of index m. Notice that there is a negative sign because of the definition 3032 of the LLR above. 3033 3034

Since this exhaustive computation is complex and involves many divisions, we can use the max-3035

log-MAP approximated version: 3036

( ) postm

ibibQAMsSNRsSSSikbikLLR .minˆ)4(ˆ4

2

)(ˆ)(16

2

−+−−+−=+

−=∈ 3037

In practice, the loss is visible only at the high SNR regions, and this region is not a critical area for 3038

the performances point of view. 3039

Moreover, for further simplification, we can make the assumption of constant post-IDFT SNR along 3040

the different blocks of a TTI. This assumption is of course only true in the low speed configuration. 3041

With this simplification, the multiplicative term postmSNR is constant over the whole TTI. Since we 3042

use the max-log-MAP turbo-decoder, the decoding process is linear, which means that the 3043

performance will not be modified if all the inputs LLRs are scaled by the same factor postSNR1 . 3044

Notice that if we use other turbo-decoding algorithm (e.g. the log-MAP) instead of the max-log-3045 MAP, the decoder output is a nonlinear function of its input, and therefore the scaling factor 3046

postmSNR cannot be discarded, even at low speed. 3047

Some preliminary simulation results showed that using max-log-MAP with constant SNR 3048

approximation, we observe negligible loss at 50km/h and a loss of 0.5dB at 120km/h. With this 3049

simplification, the multiplicative term postmSNR can thus be ignored, leading to: 3050

−+−−+−=+

−=∈

2

)(ˆ)(16

2

minˆ)4(ˆ)4( sSSSikbikLLRibibQAMs

3051

No computation of the post-IDFT SNR is thus necessary. This will also be the case for MIMO, 3052

where the post-IDFT SNR computation would have been even more CPU greedy. 3053

The maximum search does not need to be performed exhaustively at each symbol as the closest 3054

neighbor to a given symbol differing from one bit only can be pre stored. This is described in next 3055

plot. 3056

Supprimé : 4.0.1

Supprimé : 2







3057

So the soft bit is given by:

−+−−+−=+

22ˆ)4(ˆ)4(ki

sSSSikbikLLR 3058

Looking further at the constellation, it appears that for the bits indexed 0 and 2 we have 3059 ( ) ( )

kisiS Re)(ˆRe = whereas for the bits indexed 1 and 3 we have ( ) ( )

kisiS Im)(ˆIm = 3060

This leads to the following relationships: 3061

( )

( )

+−−

−−=

dSd

Sd

dSd

LLR

)Re(22

)Re(2

)Re(22

)0( ,if

−≤<<−

≥

dS

dSd

dS

)Re(

)Re(

)Re(

3062

3063

( )

( )

+−−

−−=

dSd

Sd

dSd

LLR

)Im(22

)Im(2

)Im(22

)1( if

−≤<<−

≥

dS

dSd

dS

)Im(

)Im(

)Im(

3064

( ))Re(2)2( SddLLR −−= 3065

3066

( ))Im(2)3( SddLLR −−= 3067

3068 3069 It is assumed that no scrambling is applied at the UE side. It may be changed once 3gpp has 3070

frozen this part. 3071

COMPUTATION OF THRESHOLD D 3072

d is a threshold for delimitating the 16-QAM symbols iS . When iDFT is preserving energy, we 3073

compare: 3074

∑−

=

=1

0

2exp1 M

kki M

ikjs

MS π and

10

2=d 3075

Since the implemented iDFT differs from that, d shall be defined as : 3076

∑−

=

=1

0

2exp1 M

kki M

ikjs

MS π ,

10

2

⋅=

Md 3077

3078

Supprimé : 4.0.1

Supprimé : 2







Using iN , which is the integer representation of dshifti

iv

NS

2⋅≈ , where dshift is one of the output of 3079

the iDFT, f is the scale factor of the output of MMSE.: 3080

10

2

⋅≥

MSi 3081

10

222

⋅⋅⋅≥⋅⋅

M

vSv

dshift

is 3082

10

22

2 ⋅⋅⋅≥

M

vN dshift

fi 3083

so we get dshiftf

M

vd 2

10

2 1

⋅⋅

⋅=+

to be compared to iN 3084

Let’s define 10

2 max1

⋅⋅=

++

M

vd

fdshift

so that dshiftdshift

dd −=

max2 where dshiftmax=3. 3085

3086

d is tabulated below (right column) : 3087

3088

DFT input size

f MMSE scale factor

v d

u8

M=12 5 3.0000 140 M=24 5 3.0000 99 M=36 5 4.5000 121 M=48 5 6.0000 140 M=60 5 7.5000 157 M=72 5 4.5000 86 M=96 5 6.0000 99 M=108 5 6.7500 105 M=120 5 7.5000 111 M=144 5 9.0000 121 M=180 5 11.2500 136 M=192 5 12.0000 140 M=216 5 13.5000 149 M=240 5 15.0000 157 M=288 5 9.0000 86 M=300 5 9.3750 88 M=324 5 10.1250 91 M=360 5 11.2500 96 M=384 5 12.0000 99 M=432 5 13.5000 105 M=480 5 15.0000 111 M=540 5 16.8750 118 M=576 5 18.0000 121 M=600 5 18.7500 124 M=648 5 20.2500 129 M=720 5 22.5000 136 M=768 5 24.0000 140 M=864 5 27.0000 149 M=900 5 28.1250 152 M=960 5 30.0000 157 M=972 5 30.3750 158

Supprimé : 4.0.1

Supprimé : 2







M=1080 5 16.8750 83 M=1152 5 18.0000 86 M=1200 5 18.75 88

Table 0-8: QAM threshold fix values 3089


d is then calculated as follows : 3091

3092 Figure 0-26: d calculation 3093

3094

There are 12 DShift values for the 12xM data complex samples, where each individual DShift value 3095 is applied on a group of M data complex samples which is related with one OFDM symbol. Notice 3096 that for LA1.0, with the IDFT Xilinx IP used, DShift is always equal to 0. 3097

3098

For soft bits 0 and 1, implementation is given on next plot 3099

3100

3101

For soft bits 2 and 3, we have 3102

(.)≥d <<1

d

(.)≤-d

Sat 8

<<1

9

8

9 8

d

-

S ( )1−×

Sat 8 ( )1−×

Sat 8 ( )1−×

d

d Look up

table

dshift

N_prb

8 bits

11 bits

>> dshiftmax -dshift

for each Ofdm-symbol

Supprimé : 4.0.1

Supprimé : 2







3103

3104

3105


3107


Mod Type enum {QPSK,16QAM} I 1 UE’s used modulation

M Integer u11 I 1 Number of sub-carriers of the Ue

First_Prb Integer u10 I 1 Position of the 1st PRB of the Ue

Data Complex Integer (8I,8Q) I 12 x M DFT output

dshift Integer u2 I 12 DFT output scaling factor

Out Signed integer u8 O 12 x M x modulation size softbit

3108 Table 0-9:Interface definition for LLR 3109

3110

6.5.4 DE-INTERLEAVING, DE-RATE MATCHING AND H-ARQ 3111

RECOMBINATION 3112

Natural order operations is: once we have the soft-bits streams obtained from LLR computation, we 3113

have to perform de-interleaving and de-rate matching before recombining the obtain LLR vector 3114

with LLR vectors from previous transmission. 3115

But in order to reduce the number of bits to store in RAM, we will perform the SNR scaling first, 3116

then de-interleaving / de-rate matching, and then the H-ARQ Recombining. 3117

The HARQ buffer bitwidth is 8 (signed). Saturation is performed when combining buffer’s value with 3118

new LLR outputs. 3119

This will be compliant with 3GPP Rel8 (TS36.212). 3120

6.5.5 DECODER 3121

After H-ARQ recombination, the soft-bit stream is sent to the Turbo-decoder. After decoding, we 3122

have the CRC check to decide if the block is correctly received or not. 3123

-|.| 8

7

d

8

S ( )1−×

Supprimé : 4.0.1

Supprimé : 2







The decoder will be compliant with 3GPP Rel8 (TS36.212). Note that a new Turbo code internal 3124

interleaver, contention-free QPP interleavers, was adopted (as compared to Rel6). 3125

For datas, the Xilinx’s turbo decoder that will be used is described in [R11]. TCC decoder algo type 3126

is "Max" 3127

3128

7. PUCCH PROCESSING: ACK-NACK AND SR 3129

7.1. CHANGES WITH RESPECT TO LA2.0 3130

Unlike LA1.0 and LA2.0, CQI and SR processing for PUCCH are supported in LA3.0. 3131

For Ack-Nack, we use the same Ack-Nack detection method that is described in LA2.0. 3132

7.2. GENERAL CONSIDERATIONS 3133

In addition to transmitted data, the base station has to demodulate the control symbols of all UEs. 3134

The UL control channel decoding is designed to have a code division multiple (CDM) access 3135

scheme with frequency hopping between two slots in a sub-frame for autonomous feedback. The 3136

slot format of the Ack/Nack and CQI only UL control channel is in the following figures. The green 3137

symbols in the figures are pilot symbols. The UL control channel carries data and non-data 3138

associated information. 3139

3140 3141

Data and RS accumulation for

ACK/NACK

ACK/NACK estimation

DTX

N

21−Nσ

Supprimé : 4.0.1

Supprimé : 2







3142 3143

Figure 7-1: ACK/NACK transmission 3144 3145

7.3. FREQUENCY OFFSET ESTIMATION 3146

None is performed on PUCCH. 3147

7.4. FREQUENCY OFFSET COMPENSATION 3148

Each user is frequency compensated with the latest available estimated offset obtained from 3149

PUSCH with L=1 as per 6.2.1.2. It means that we have as many inputs as users each with their 3150

own frequency compensation. 3151

For OFDM symbol p and user u, the frequency offset compensation is done by a multiplication by 3152

the complex exponential: 3153

( ) ( )

−= maxFFT

)(ˆ..2exp,N

upKjupCFO επ 3154

Where, for all bandwidth, 2048maxFFT =N and the values of ( )pK correspond to a 3155

compensation with respect to the middle of the symbol and are given below: 3156

p 0 1 2 3 4 5 6 7 8 9 10 11 12 13

( )pK 1184 3376 5568 7760 9952 12144 14336 16544 18736

20928 23120 25312 27504 29696

3157

frequency spreading

time spreading

ACK/NACK symbol

pilot frequency sequence

time spreading

1 PRB

1 slot

N sub carriers

Supprimé : 4.0.1

Supprimé : 2







7.5. TIMING OFFSET ESTIMATION 3158

None is carried out on PUCCH. 3159

7.6. TIMING OFFSET COMPENSATION 3160

From the frequency compensated samples available along the pilot sequence, time compensation 3161 with constant time shift is done (corresponding to the chosen timing for the implementation ie 3162

2TA

CP==ττ in the equations below, cf. section 6.2.2 – note it is common and constant for all 3163

users). This is done in all processing below by a multiplication by the complex exponential 3164

( )( ) 1,,0 ,..2exp 0 −=∆ MifiTj Kτπ . The used format for the exponential is Q1.15. 3165

The tables ( ) 1,,0 , 0 −= MiiT K depend on both the slot number (0, 1) inside the subframe and 3166

the system bandwidth and are given below for a PUCCH resource m. In LA3.0, the current working 3167 version is that only three PUCCH will be used. These PUCCH are described below with their 3168 resource number m=0, m=1 and m=2. 3169

3170 3171

3172

We give below the fomulas to compute the tables ( ) 1,,0 , 0 −= MiiT K in the general case of 3173

any PUCCH resource number m. 3174

We define the two indexes: 3175

000

1 22MM

mMNI

ULRB −

−= 3176

00

2 22M

mMNNI

ULRB

FFT

+−= 3177

Here, 120 =M is the number of tones per PRB, ULRBN is the total number of PRBs in UL (100 for 3178

20MHz) and FFTN is the FFT size (=2048). 3179

Then the tables are given below for even and odd values of m: 3180

• If mod(m, 2)=0, then the values of T(i) are given by: 3181

m=0

m=1

1 slot

Time

PRB 0

PRB 99 m=0

m=1

m=2

m=2

m=3

m=3

Supprimé : 4.0.1

Supprimé : 2







Slot 0: [ ]11101 2222 +++= IIIIT K 3182

Slot 1: [ ]11101 1111 +++= IIIIT K 3183

• If mod(m, 2)=1, then the values of T(i) are given by: 3184

Slot 0: [ ]11101 1111 +++= IIIIT K 3185

Slot 1: [ ]11101 2222 +++= IIIIT K 3186

7.7. ORDER OF FRONT END PROCESSING 3187

For sake of optimization, we perform the front-end operations in the following order for a given user u, 3188

OFDM symbol p and tone i (notice that this order could be modified in the future in the case of algorithm 3189

change): 3190

• Compute the term by term product of the timing offset exponential ( ) ( )( )τπ fijTiCTO ∆= ..2exp as 3191

defined in section 7.6 and the user specific frequency CAZAC sequence conjugate elements 3192

( )∗ip . This term is common for all antennas; hence it is computed once then used in channel 3193

estimation formulas of each antenna 3194

• Compute the correlation of the obtained sequence with the received sequence on each antenna 3195

(correlation here can be restated as term by term product followed by the sum over 12 tones) 3196

• Multiply the result by the frequency offset term (see section 7.4): 3197

( ) ( )

−=

FFT

)(ˆ..2exp, NupKjupCFO επ 3198

7.8. CHANNEL ESTIMATION AND ACCUMULATION ON RS 3199

This section is common for ACK/NACK only and ACK/NACK with SRS. 3200

From the frequency compensated samples available along the pilot sequence, a user specific 3201

CAZAC frequency correlation is performed. It is then followed by time compensation with constant 3202

time shift (corresponding to the chosen timing for the implementation ie τA). Then user specific 3203

CAZAC demultiplexing is performed. 3204

( )( ) ( )∑−

=Ρ∆=

1

0,,

*,

0,

0

ˆ..2exp1 M

i

pilotapipiap HpfijT

Mτπα for { }2,1,0∈p and 3205

( )( ) ( )∑−

=+Ρ∆=

1

0,,

*,

0,

0

ˆ..2exp1 M

i

pilotapNipiap HpfijT

Mτπα for { }5,4,3∈p 3206

Here, P(p) is the index of pilot symbols, i.e. P(p=0) = 2 and P(p=5)=11. The notation ( )pip Ρ, 3207

states for the Cazac sequence on tone i on OFDM symbol P(p). 3208

At this stage we have 3 estimations per antenna, user and slot matching the pilot symbols. In order 3209

to descramble users, weighted accumulation with respect to the user’s specific sequence is 3210

performed. 3211

( )aaaRSa www ,2

*2,0,1

*1,0,0

*0,0,0 αααα ++= for the first slot. 3212

( )aaaRSa www ,5

*2,1,4

*1,1,3

*0,1,1 αααα ++= for the second one. 3213

where w is the user specific time scrambling sequence of size 3 for ACK/NACK RS. 3214 3215

Supprimé : 4.0.1

Supprimé : 2







Notice that we do not normalize the result by 3. 3216 We also compute the normalized versions of these quantities which are used for noise and power 3217 computations: 3218 3219

( )aaaa www ,2*

2,0,1*1,0,0

*0,0,0 3

1 αααα ++= for the first slot. 3220

( )aaaa www ,5*2,1,4

*1,1,3

*0,1,1 3

1 αααα ++= for the second one. 3221

3222 3223

DATA INTERFACE 3224 3225


Users uint u8 I 1 Number of PUCCH users

Data_main complex Integer

(16I,16Q) I 12 x 6

Received Main Data for the pilot blocks of the TTI for PUCCH after

CFO compensation with L=1

Data_div complex Integer

(16I,16Q) I 12 x 6

Received Div Data for the pilot blocks of the TTI for PUCCH after


Code_f pointer Int* I 1 Pointer to the frequency codes (p)

Code_t pointer Int* I 1 Pointer to the time codes (w)

Out complex Integer

(16I,16Q) O 2 x 4 x Users 2 channel estimations per

antenna and user

3226 Table 7-1: Interface definition for ACK/NACK channel estimatio n and accumulation on RS 3227

3228 FIX IMPLEMENTATION 3229 Next plot is for CAZAC de-multiplexing and time offset compensation. The CAZAC sequence and 3230 the exponential compensation are stored under Q1.15 format. 3231 1/M0 has Q0.12 (unsigned) format. 3232

Supprimé : 4.0.1

Supprimé : 2







3233 3234

3235 3236 Multiplication by the time spreading sequence is described below: 3237 3238

(16I,16Q)

(16I,16Q)

>>15 Sat16

>>15

Sat16

(16I,16Q)

+ 20

341 (1/12)

>>12 Sat16

*ip

τπ fjie ∆+2

pilotapiH ,,

ˆ

aj ,α

pilot

Supprimé : 4.0.1

Supprimé : 2







3239 3240 3241

7.9. CHANNEL ESTIMATION AND ACCUMULATION ON DATA 3242

7.9.1 DATA ACCUMULATION 3243

Frequency offset compensation is first done, time compensation with constant time shift is also 3244

performed. Then outputs of CAZAC sequence multiplication are accumulated per slot including the 3245

multiplication by the corresponding time spreading values. 3246

The non SRS case is described hereafter: 3247

( )( ) ( )∑ ∑=

−

=∆

∆=

3

0

1

0,

*,

0

*,0,0

0

...2exp1

k

M

i

akikika YpfijT

Mwy τπ for the first slot 3248

and ( )( ) ( )∑ ∑=

−

=++∆

∆=

3

0

1

04,

*,

0

*,1,1

0

..2exp1

k

M

i

akNikika YpfijT

Mwy τπ for the second slot 3249

3250 Notice that we do not normalize the result by the number of data symbols. Here, ∆(k) is the index of 3251 data symbols. 3252 3253

(16I,16Q)

(16I,16Q)

>>15

∗w

aj ,α

aO ),1(α >>15

10922 (1/3 in s15)

Other pilot from same antenna and slot

s16

Sat16

RSaO ),1(α

s17

>>2

Supprimé : 4.0.1

Supprimé : 2







The variables akiY, and a

kNiY 4, ++ are the equivalent of pilotapiH ,,

ˆ and pilotapNiH ,,

ˆ+ for data symbols and 3254

[ ]3,2,1,0, nsnsnsnsns wwwww = is the user specific time domain spreading sequence for 3255

ACK/NACK data on slot 0 or 1 (the code for a given user can be different for slots 0 and 1). 3256 3257

In the case of shortened PUCCH format, a size 3 code is used for the second slot: 3258

( )( ) ( )∑ ∑=

−

=++∆

∆=

2

0

1

04,

*,

0

*,1,1

0

..2exp1

k

M

i

akNikika YpfijT

Mwy τπ and 3259

[ ]2,11,10,11 wwww = correspond to shortened DFT codes. Notice that we do not normalize the 3260

result by 3. 3261 3262

Notice that shortened PUCCH format should be used whenever SRS are configured. SRS are cell-3263 specific, and the SRS configuration are given in 5.5.3.1-1 of specifications document 36.211. The 3264 current working view is that only configuration 0 is used, meaning that the last OFDM symbol of 3265 every TTI is reserved for SRS (which does not mean that SRS are actually transmitted). 3266

This means that the current working view is that only shortened PUCCH format should be used. 3267

The fixed point implementation is the same as for Reference symbols. However, notice that for 3268 slots with four data OFDM symbols (i.e. slot 0, and slot 1 in case of non SRS puncturing), the 3269 multiplication by the time domain code is not explicitly performed since the Hadamard sequences 3270

take their values in { }1,1+− .The rescaling shift (>>15) following these multiplications is then also 3271

removed. 3272

FIX IMPLEMENTATION 3273 3274

The fixed point implementation of accumulation on data for both SRS and non –SRS cases is 3275 described below. The input of this block are the output of the CFO + CTO +CAZAC compensation 3276 which has been described. 3277

3278

No SRS puncturing: 3279

3280

3281

3282

3283

(16I,16Q)

>>1

>>1

>>1

>>1

>>1

>>1

0,0y

Supprimé : 4.0.1

Supprimé : 2







SRS puncturing: 3284

3285

3286

3287

7.9.2 DATA DESCRAMBLING 3288

The data symbols of the slot indexed by sn are multiplied by a complex coefficient ( )snS defined 3289

by: 3290

3291

3292

The parameter )(' Snn is defined in section 5.4.1 of TS 36.211. 3293

Notice that the pilot blocks are not multiplied by this coefficient. Before the processing of data 3294 symbols, we need to perform descrambling. This is done by a multiplication of the data symbols by 3295

( ) 2)( πjs enS −∗ = on the slots verifying 02mod)(' ≠Snn . 3296

Notice that no multiplication is required since this descrambling can be performed by the 3297

transformation ( ) ( )IQQI −→ ,, . 3298

The output of the de-scrambling process is denoted as Dscr,0 ay and Dscr

,1 ay for antenna a on both slots. 3299

7.10. ACK-NACK DETECTION 3300

7.10.1 ALGORITHM DESCRIPTION 3301

The DTX detection is done using both the reference symbols and the data symbols 3302

This can be done by hypothesis testing: 3303

• First, we compute the energies of the possible Ack-Nack configurations 3304

• Keep the configuration that lead to the maximum energy and perform threshold detection 3305

0,0y

16

16

>>15 Sat16

*w

>>2

Supprimé : 4.0.1

Supprimé : 2

=

=otherwise

02mod)('if1)( 2πj

Ss e

nnnS







The symbol detection is obtained for free. 3306

The used strategy is to add coherently RS and data channel estimates slot-wise depending on the 3307

different possible transmitted data. For coherent accumulation, we have to remove the tested Ack-Nack 3308

symbol by multiplying the data channel estimation by its conjugate. Notice that a weighting of the 3309

signals depending on the number of accumulated symbols is automatically done in the formulas below 3310

since we did not normalize by the number of OFDM symbols when we computed the accumulation on 3311

data and RS. This gives for antenna a on the first slot: 3312

( ) ( ) Dscr,0,0,0 . a

RSaa yssZ ∗+= α 3313

( ) ( ) Dscr,1,1,1 . a

RSaa yssZ ∗+= α 3314

In the formula above, the variable s corresponds to the candidate Ack-Nack symbol we want to detect. 3315

This means that the processing described in this section must be done for every possible Ack-Nack 3316

symbol s , i.e. for { }1,1−∈s for single bit Ack-Nack transmission and { }jjs ,,1,1 −−∈ for two bits Ack-3317

Nack transmission. 3318

The corresponding fixed point implementation is described below: 3319

3320

3321

3322

3323

3324

3325

3326

We then compute the following metric: 3327

( ) ( ) ( ) ( ) ( )

+

= ∑∑

−

=

−

=−

1

0

2

,12

0

1

0

2

,02

1

RXRX N

aa

N

aaN sZsZsM σσ 3328

We then end up with 2 metric values for one bit Ack-Nack and 4 metric values for two bits Ack-Nack. 3329

We select the best metric and compare it to a threshold depending on noise: 3330

( ) ( ) ( )21

20

?

opt . max −≥= Ns

sMM σση 3331

If ( ) ( )21

20opt . −< NM σση then a DTX is detected, 3332

If ( ) ( )21

20opt . −≥ NM σση an Ack-Nack transmission is detected, and the estimated Ack-Nack symbol 3333

is equal to: 3334

( )sMssmaxargˆ = 3335

One the optimal Ack-Nack symbol is estimated, and in the case where no DTX is detected, then the 3336 QAM de-mapping is performed. For Double Code word, we have: 3337

3338

1ˆ =s corresponds to b(0)=b(1)=0 (ie NACK/NACK) 3339

RSa,0α

( ) Dscr,0. ays ∗

Sat 15 ( )sZ a,0

Supprimé : 4.0.1

Supprimé : 2







js −=ˆ corresponds to b(0)=0, b(1)=1 (ie NACK/ACK) 3340

js =ˆ corresponds to b(0)=1, b(1)=0 (ie ACK/NACK) 3341

1ˆ −=s corresponds to b(0)=b(1)=1 (ie ACK/ACK) 3342 3343 3344 3345 And for Single Code Word (BPSK): 3346 3347

1ˆ =s corresponds to b(0)= 0 (ie NACK) 3348

1ˆ −=s corresponds to b(0)=1 (ie ACK) 3349 3350

3351

The computation of the metrics for threshold comparison is given below: 3352

3353

3354 The parameter “o” which defines the size of the shift is chosen in order to avoid the loss of LSB as 3355 below. 3356 Let “order” denotes the function computing the number of bits representing the sign before the first 3357 significative bits. Then we compute: 3358

scaling_0 = 31 - order(20σ ); 3359

scaling_1 = 31 - order(2

1−Nσ ); 3360

o=max(scaling_0, scaling_1)-15; 3361

3362 3363

Su

m o

ver a

ll Rx a

nten

na

s

|.|2

|.|2

(16I,16Q)

16

|.|2

|.|2

(16I,16Q) 16

>>o

2nd PRB

1st PRB

0,0Z

∑

∑

=

=−

+Nrx

antant

Nrx

antantN

Z

Z

0

2

,120

0

2

,02

1

σ

σ

NRxZ ,0

u32

30

31

31

Sat 15

>> 15

21−Nσ

… >>log2(NRx)

Supprimé : 4.0.1

Supprimé : 2







The constant threshold is stored under the Q5.10 (unsigned) format. 3364 3365 3366 3367

3368 3369 3370

Thresholds setting 3371 3372 For format 1a, we consider the usual method to derive the threshold. This method consists 3373 in choosing the threshold which satisfies P(DTX -> ACK)=1% (0.9% has been used in the 3374 simulation to have a margin). 3375 3376 For format 1b, we procceed differently to derive the threshold. Actually, P(ACK->NACK) is higher 3377 than for format 1a and is the limiting factor. We will then ensure that this is below 0.1%. 3378 First, we determine the operating point, i.e. SNR value, which guarantees P(ACK->NACK) 3379 =0.1%. Then, we choose the threshold that satisfies P(ACK->DTX) = 1% for the operation 3380 point. The selected threshold should also guarantee close performances for P(ACK->DTX) 3381 and P(DTX-> ACK). 3382 3383 Thresholds for 1Rx case: The threshold η is set to 5.6 dB and 7.66 dB for format 1a and 3384 format 1b respectively for the different bandwidths. 3385 Under Q5.10 unsigned format, this gives for fixed point values: 3386 3718 for format 1a 3387 5974 for format 1b 3388 3389 Thresholds for 2Rx case: Used the appropriate method, the threshold η is set to 7.457dB and 8.9dB 3390 for format 1a and 3391 format 1b respectively for the different bandwidths. 3392 Under Q5.10 unsigned format, this gives for fixed point values: 3393 5702 for format 1a 3394 7949 for format 1b 3395 3396 3397 Thresholds for 4Rx case: The threshold η is set to 9.45dB and 10.5dB for format 1a and 3398 format 1b respectively for the different bandwidths. 3399 Under Q5.10 unsigned format, this gives for fixed point values: 3400 9022 for format 1a 3401 11489 for format 1b 3402 3403 These threshold values assume SRS puncturing. 3404 3405 3406

3407

7.11. SR DETECTION 3408

SR alone 3409

3410

>>o

>>o

>>(16-o) 2

120 . −Nσση

21−Nσ

20σ

u30

u(42-o)

>>8 Sat 31

Supprimé : 4.0.1

Supprimé : 2







It is performed using the same algorithm as for Ack-Nack format 1a but on SR resource. Since the 3411

target performance of SR is the same as for Ack-Nack, we can reuse the same threshold as for 3412

format 1a. The only difference with Ack-Nack detection is that in the case of SR transmission 3413

without Ack-Nack, the detection has to be done using ( )sMM SRSR =∗ for 1=s only 3414

SR multiplexed with Ack-Nack 3415

A special processing has to be done if SR and Ack-Nack are expected on the same TTI. In this 3416

situation, in case of negative SR, the UE transmits its Ack-Nack on its Ack-Nack resource whereas 3417

in the case of positive SR, it transmits its Ack-Nack on its SR resource. Therefore, a double 3418

hypothesis detection has to be performed on both resources. 3419

This consists in computing the Ack-Nack metrics for both resources and performing joint threshold 3420

detection as described below. 3421

Ack-Nack metric on SR resource: compute ( )sMM NackAcks

SR −∗ = maxarg for { }1,1−∈s for format 3422

1a and { }jjs −−∈ ,,1,1 for format 1b. 3423

Ack-Nack metric on Ack-Nack resource: compute ( )sM NackAck− for { }1,1−∈s for format 1a and 3424

{ }jjs −−∈ ,,1,1 for format 1b. Compute then ( )sMM NackAcks

NackAck −∗

− = maxarg . The notation 3425

ANη will denote the threshold corresponding to expected Ack-Nack format, and a1η denotes the 3426

format 1a threshold. 3427

The noise powers are indexed by the AN or SR prefix. 3428

First case: ( )( )21,

20,1 −

∗ < NSRSRaSRM σση and ( )( )21,

20, −

∗− < NANANANNackAckM σση 3429

No Ack-Nack and no SR are detected 3430

Second case: ( )( )21,

20,1 −

∗ < NSRSRaSRM σση and ( )( )21,

20, −

∗− > NANANANNackAckM σση 3431

A negative SR is detected, and Ack-Nack transmission on Ack-Nack resource is detected. 3432

Third case: ( )( )21,

20,1 −

∗ > NSRSRaSRM σση and ( )( )21,

20, −

∗− < NANANANNackAckM σση 3433

A positive SR is detected, and Ack- on SR resource is detected. 3434

Fourth case: ( )( )21,

20,1 −

∗ > NSRSRaSRM σση and ( )( )21,

20, −

∗− > NANANANNackAckM σση 3435

In this case we need to discriminate further. In the case where Ack-Nack and SR are on the same 3436

PRBs, this is done simply by comparing ∗SRM with ∗

−NackAckM 3437

If they are transmitted on different PRBs, then we need to put them at the same scale before 3438 performing a new comparison. The detection rule consists then in comparing 3439

( )( )21,

20, −

∗NANANSRM σσ with ( )( )2

1,2

0, −∗

− NSRSRNackAckM σσ and taking the maximum as the transmitted 3440

signal. Notice that if the SR is selected, then we need to reperform full Ack-Nack detection on SR 3441 resource in order to detect to transmitted Ack-Nack symbol. 3442

Notice that this is not possible to discriminate between the situations where only SR is transmitted 3443 and the case where SR is multipliexed with 1=s . In both cases, this will results in a detection of 3444 NACK for format 1a and (NACK, NACK) for format 1b and trigger a retransmission. 3445

3446

Notice also that to ensure a fair comparison between Ack-Nack and SR resource, the same 3447 threshold has to be used on both. This comment is especially valid in the case where the 3448 performance requirements are not the same for Ack-Nack and SR (see below for example). The 3449

Supprimé : 4.0.1

Supprimé : 2







false SR detection restricted to the cases of Ack-Nack + SR detection will then be equal to the Ack-3450 Nack false detection whatever the nominal SR requirement. 3451

3452

Thresholds setting 3453 If the performance requirement is the same as for Ack-Nack, the same thresholds can be used 3454

since the detection algorithm is the same. We further give below the thresholds in dB 3455

corresponding to the 0.1% false alarm, i.e. P(No SR present -> SR detected ) =0.1%. Notice that 3456

these thresholds assume systematic short PUCCH format. 3457

Floating point value: 3458

Threshold for 0.1% False detection

1Rx 7,0496

2Rx 8,6278

3Rx 9,554

4Rx 10,3127

3459

3460

Fixed point value: 3461

Threshold for 0.1% False detection

1Rx 5191

2Rx 7466

3Rx 9241

4Rx 11004

3462

In order to ease the design and the tests, we can consider a common value across the bands equal 3463

to the max value. 3464

Notice that the reduction of the false alarm rate has the effect of increasing the misdetection 3465

probability P(SR present -> SR not detected). The performance loss is equal to the difference 3466

between the considered thresholds expressed in dB. 3467

3468

Supprimé : 4.0.1

Supprimé : 2







7.12. NOISE ESTIMATION 3469

7.12.1 NOISE ESTIMATION FOR ACK-NACK 3470

It is performed on pilot sequence after CAZAC de-multiplexing with similar approach as for PUSCH 3471

(ie subtraction of the estimated amplitude to the received CAZAC multiplicated signal) but taking 3472

into account the user multiplexing, frequency and timing offset. It is used for MMSE, ACK/NACK 3473

threshold detection and L2 IF (power control). 3474

U denotes the number of multiplexed users on this field. Note the algorithm only works up to 35 (ie 3475

3 10 −M ) users 3476

The noise is estimated by: 3477

for first PRB. 3478

( ) ( )( )( ) ( )( )∑ ∑∑ ∑

−

= =

−

=

−

=

Ρ∆−−−

=1

0

2

0

1

0

2

1

0

,0)(,maxFFT

,0

20

0

,0)()(ˆ..2exp..2exp

3

11 NRx

a p

M

i

U

u

V

au

pPiup

ai

u

pPup

NupKjfijTwX

UMNRx 4444444444444 34444444444444 21

αεπτπσ3479

3480 For the first slot and 3481

( ) ( ) ( )( ) ( )( )( )∑ ∑∑ ∑

−

= =

−

=

−

=+−−

Ρ∆−−−

=1

0

5

3

2

1

0

1

0

,1,maxFFT

,1,10

21

0

)()(ˆ..2exp..2exp3

11 NRx

a p

M

i

U

u

V

au

pPiu

pPp

aiNN

u

upN

upKjfijTwXUMNRx

4444444444444 34444444444444 21

αεπτπσ3482

for For the second slot. 3483

Here, P(p) is the index of pilot symbols, e.g. P(p=0) = 2 and P(p=5)=11. The notation ( )pip Ρ, 3484

states for the Cazac sequence on tone i on OFDM symbol P(p). The CFO complex exponentials 3485 are defined in section 7.4. 3486

3487

The notation paiX , stands for the FFT output and 3488

( )aaaa www ,2*2,1

*1,0

*0,0 3

1 αααα ++= 3489

( )aaaa www ,5*2,4

*1,3

*0,1 3

1 αααα ++= 3490

3491 3492 INTERFACE DEFINITION 3493

3494




(16I,16Q) I 12 x 6

Received Main Data for the pilot blocks of the TTI for PUCCH before CFO compensation

Data_div1 complex Integer

(16I,16Q) I 12 x 6

Received Div Data for the pilot blocks of the TTI for PUCCH before CFO compensation

Data_div2 complex (16I,16Q) I 12 x 6

Received Div Data for the pilot blocks of the TTI for PUCCH

Supprimé : 4.0.1

Supprimé : 2







Integer before CFO compensation


(16I,16Q) I 12 x 6

Received Div Data for the pilot blocks of the TTI for PUCCH before CFO compensation

Alpha complex Integer

(16I,16Q) I 4 x 2 x Users 2 channel estimations per

antenna and user

Phase Integer s8 I Users Normalized frequency offset from

PUSCH

Noise uint u30 O 2 Estimated noise

3495 Table 7-2 : Interface definition for ACK/NACK noise estimation 3496

3497 FIX IMPLEMENTATION 3498

3499

W is the fix representation of UMNrx −03

11 3500

3501 3502

Sat 16 ||.||2 >>6

+ >>15

paiX ,

16

16

uV

16 30 24

25

Other pilot, antenna, tones

32

<<2e Sat 30

W

>>1

Supprimé : 4.0.1

Supprimé : 2







Case number of antenna ports Nrx = 1: 3503 3504 3505

U 1 2 3 4 5 6

W 1872 1928 1986 2048 2114 2185 3506 3507

U 7 8 9 10 11 12

W 2260 2341 2427 2521 2621 2731 3508 3509

U 13 14 15 16 17 18

W 2849 2979 3121 3277 3449 3641 3510 3511

U 19 20 21 22 23 24

W 3855 4096 4369 4681 5041 5461

3512

3513 U 25 26 27 28 29 30

W 5958 6554 7282 8192 9362 10923

3514 3515

U 31 32 33 34 35 36

W 13107 16384 21845 32768 65536 NA

3516 3517 Case number of antenna ports Nrx = 2: 3518

3519 3520

U 1 2 3 4 5 6

W 936 964 993 1024 1057 1092

3521 3522

U 7 8 9 10 11 12

W 1130 1170 1214 1260 1311 1365

3523 3524

U 13 14 15 16 17 18

W 1425 1489 1560 1638 1725 1820

3525 3526

U 19 20 21 22 23 24

W 1928 2048 2185 2341 2521 2731

3527 3528 3529

U 25 26 27 28 29 30

W 2979 3277 3641 4096 4681 5461

3530 3531

U 31 32 33 34 35 36

Supprimé : 4.0.1

Supprimé : 2







W 6554 8192 10923 16384 32768 NA

3532 3533

Case number of antenna ports Nrx = 4: 3534 3535 3536

U 1 2 3 4 5 6

W 468 482 496 512 529 546 3537 3538

U 7 8 9 10 11 12

W 565 585 607 630 655 683 3539 3540

U 13 14 15 16 17 18

W 712 745 780 819 862 910 3541 3542

U 19 20 21 22 23 24

W 964 1024 1092 1170 1260 1365 3543 3544 3545

U 25 26 27 28 29 30

W 1489 1638 1820 2048 2341 2731 3546 3547

U 31 32 33 34 35 36

W 3277 4096 5461 8192 16384 NA

3548 3549

3550 3551 3552 3553

7.12.2 EMPTY PUCCH 3554

In the case where nothing is transmitted in a PUCCH, then we compute the noise estimation on the 3555

empty PRBs by directly summing the squared samples located in this PRB on the PUCCH. This 3556

applies if no Ack-Nack have been detected. 3557

3558

3559

3560

3561

3562

3563

3564 3565

Supprimé : 4.0.1

Supprimé : 2







3566 3567

7.12.3 COMBINING NOISE ESTIMATES FROM BOTH PUCCH 3568

Both noise estimates of the same PRB coming from the two consecutive slots (and thus from two 3569 different PUCCH whatever the PRB configuration is) have to be averaged to provide a single noise 3570

estimation per PRB. They are then exponentially averaged as on PUSCH which yields 21

20 , −Nσσ . 3571

7.12.4 LONG TERM AVERAGING OF NOISE ESTIMATES 3572

After the combination of noise estimates from both slots, we perform a long term averaging of the 3573

noise. The averaging scheme is described below: 3574

3575

3576

3577

3578

3579

3580

3581

3582

3583

3584

3585

3586

3587

3588

3589

3590

>>3 16

1st slot : 12x2x6 values From

other tones

>>3

From 6 other OFDM symbols

>>1

From other antenna

>> log2( Nrx ) 455<<2

>>16

2

2

u30

>>8 >>2

u10

u30

Sat 30

s30 s40 s32

u30

β

Sat 30

z-1

-

2iσ

u30

Supprimé : 4.0.1

Supprimé : 2







7.13. POWER ESTIMATION 3591

It is performed on pilot sequence after UE de-multiplexing, i.e. CAZAC and time domain codes de-3592 multiplexing. This means that we will use the per slot per antenna amplitudes: 3593

( )aaaa www ,2*2,1

*1,0

*0,0 3

1 αααα ++= for the first slot. 3594

( )aaaa www ,5*2,4

*1,3

*0,1 3

1 αααα ++= for the second one. 3595

The noise contribution is subtracted. Averaging over 0M tones and 3 pilot symbols provides a 3596

processing gain of 03M so that the noise power after UE de-multiplexing is equal to 0

20

3M

σand 3597

0

21

3M

σ on slot 0 and 1 respectively. 3598

On each slot, the average power of a,•α is computed, and the noise power is subtracted. The 3599 result is then averaged on the slots and antennas. It is for L2 IF (power control). It applies on all 3600 PUCCH configurations. 3601

For Ack-Nack, the power estimate writes: 3602

( )21

20

00

2

,1

0

2

,06

12

1 σσαα +−

+= ∑∑

== MNP

RXRX N

a

a

N

a

a

RX

3603

3604 3605 FIX IMPLEMENTATION 3606

3607

Supprimé : 4.0.1

Supprimé : 2







3608

3609 3610 3611

7.14. ANTENNA-PATH FAILURE DETECTION AND HANDLING F OR 3612

ACK-NACK ON PUCCH 3613

As stated in section 5.3, the detection of an antenna that has been zeroed out is done using the RSSI 3614 function in the DSP. The modem will however continue processing the incoming signal as for the 3615 nominal configuration (i.e. using the zeroed inputs as if they were valid inputs). 3616 3617 The thresholds need to be updated to ensure that P(DTX -> Ack) is still at the desired level. 3618 All threshold values considered in this section assume SRS puncturing. 3619

3620 3621

The general rule concerning the thresholds update in the case of a RXN receiver with BrokenN 3622

antennas is the following: 3623 3624

• First case : the receiver with BrokenRX NN − is supported (e.g. 2Rx). The modem just 3625

continues processing all received samples from all antennas without antenna selection, but 3626 the values of the different parameters describing the fixed point implementation (shifts, 3627

bitwidth, saturations, etc…) will be the ones of the ( BrokenRX NN − ) Rx mode as well as the 3628

thresholds used for PUCCH / SR detection 3629 3630

• Second case : the receiver with BrokenRX NN − is not supported (e.g. 3Rx). The modem just 3631

continues processing all received samples from all antennas without antenna selection. The 3632 values of the different parameters describing the fixed point implementation (shifts, bitwidth, 3633

||.||2

>>16

a,0α

16 30

1820

>>2+log2(Nrx)

other antennas

>>5

Sat30

||.||2

2nd slot

( )21

20

2

1−+ Nσσ

- +

Sat32

Supprimé : 4.0.1

Supprimé : 2







saturations, etc…) will be the ones of the nominal ( RXN ) Rx mode. However The thresholds 3634

used for PUCCH / SR detection will be multiplied by BrokenRX

RX

NN

N

− 3635

• Special case : We need to consider the special case of 4Rx wit 3 broken antennas. In this 3636

case, the above applies but with asjusted values 2=RXN and 1=BrokenN 3637

3638 3639

8. PUCCH PROCESSING: CQI AND CQI&ACK-NACK 3640


In LA2.0, CQI on PUCCH was not supported, this feature is thus new. Both CQI only and CQI & Ack-3642 Nack multiplexing are supported. 3643

3644

8.2. GENERAL CONSIDERATIONS 3645

In addition to transmitted data, the base station has to demodulate the control symbols of all UEs. 3646

The UL control channel decoding is designed to have a code division multiple (CDM) access 3647

scheme with frequency hopping between two slots in a sub-frame for autonomous feedback. The 3648

slot format of the CQI UL control channel is in the following figures. The green symbols in the 3649

figures are pilot symbols. The UL control channel carries data and non-data associated 3650

information. 3651

A third format exists to carry both ACK/NACK and CQI fields. The format is similar to CQI only 3652

where ACK/NACK is replacing one of the reference symbol. In other words, one of the ACK/NACK 3653

acts as a pilot symbol. The CQI is coded as per CQI only scheme with a known frequency 3654

spreading code. 3655

Following processing applies for each of the configuration: 3656

3657

Figure 8-1CQI and CQI&ACK\NACK Detection processing 3658 3659

3660

ACK/NACK estimation

Channel estimation for CQI

CQI estimation

CQI decoding

ACK/NACK & CQI

Channel estimation for CQI

CQI estimation

CQI decoding

CQI

Supprimé : 4.0.1

Supprimé : 2







3661

Figure 8-2: ACK/NACK&CQI Mux or CQI only transmission 3662 3663

3664

3665

3666

CQI


frequency spreading

CQI


frequency spreading

ACK/NACK

frequency spreading

1 slot

N sub carriers

Supprimé : 4.0.1

Supprimé : 2







8.3. FREQUENCY OFFSET ESTIMATION 3667

None is performed on PUCCH. 3668

8.4. FREQUENCY OFFSET COMPENSATION 3669

Each user is frequency compensated with the latest available estimated offset obtained from 3670

PUSCH with L=1 as per 6.2.1.2. It means that we have as many inputs as users each with their 3671

own frequency compensation. 3672

The values of the complex exponentials are described in section 7.4. 3673

8.5. TIMING OFFSET ESTIMATION 3674

None is carried out on PUCCH. 3675

8.6. TIMING OFFSET COMPENSATION 3676

Please refer to section 7.6 3677

8.7. ORDER OF FRONT END PROCESSING 3678

For sake of optimization, we perform the front-end processing in the same order as for Ack-Nack 3679

processing, see section 7.7. 3680


The user specific CAZAC de-multiplexing is performed. On next equation, the user index is omitted 3682 for sake of simplicity: 3683

( )( )∑−

=

∆=1

0,,

*

0,

0

ˆ..2exp1 M

i

pilotapiiap HpfijT

Mτπα for { }1,0∈p and 3684

( )( )∑−

=+∆=

1

0,,

*

0,

0

ˆ..2exp1 M

i

pilotapNiiap HpfijT

Mτπα for { }3,2∈p 3685

3686 Where: 3687

• ip is the user specific frequency spreading sequence of size 0M (ie 12) including cyclic 3688

shift with Q1.15 representation, 3689 • N is the number of sub carrier in between both PUCCH slots 3690 • ap,α is the estimated amplitude for pilot p and antenna a (we assume that the channel is 3691

sufficiently flat in the frequency domain since timing offset compensation has been applied 3692 which is a bit optimistic) 3693

At this stage we have 2 estimations per antenna, user and slot matching the pilot symbols. In order 3694

to cope with targeted UE speeds, time interpolation is performed so that 3 amplitudes per 3695

slot/antenna/user are provided: one for first data symbol, the next one for the 3 middle symbols and 3696

the other for the last symbol. 3697

+=

−

a

aHjaj I

PMKk

,1

,01

20

20

,ˆαασ

α for the first slot ( { }2,1,0∈j ) 3698

Supprimé : 4.0.1

Supprimé : 2







+=

−−

+a

aNHjaj I

PMKk

,3

,21

20

21

,3ˆαασ

α for the second one ( { }2,1,0∈j ) 3699

With: 3700

• ( ) 12

−+ xIKk Hj being a pre-stored autocorrelation matrix of size 3x2 which is 3701

chosen in a filter bank as per PUSCH. The Doppler frequency used for this 3702 selection is the one derived from PUSCH. If none is available then the highest 3703 Doppler frequency should be considered. The SNR used is derived from PUCCH 3704 noise and power (mean of power and noise on both slots has to be used). 3705

• aj ,α the 6 estimated amplitudes along the sub-frame. 3706

• 21−Nσ , 2

0σ and P are defined in section 8.10 and 8.11 3707

3708 INTERFACE DEFINITION 3709 3710




(16I,16Q) I 12 x 4

Received Main Data for the pilot blocks of the TTI for PUCCH after



(16I,16Q) I 12 x 4




(16I,16Q) I 12 x 4




(16I,16Q) I 12 x 4



df enum {…}

I Users Doppler Frequency of the current PUCCH users for mapping to the

correct auto/cross correlations

Code pointer Int* I 1 Pointer to the frequency codes

Noise uint u30 I 2 One noise per PUCCH PRB

Power uint u30 I Users One power per user

e uint u2 I 1 AGC scaling factor

Out complex Integer

(16I,16Q) O 4 x 6 x Users 6 channel estimations per

antenna and user

3711 Table 8-1: Interface definition for CQI channel estimation 3712

3713 FIX IMPLEMENTATION 3714

3715 Next plot is for CAZAC de-multiplexing and time offset compensation. The CAZAC sequence and 3716 the exponential compensation are stored under Q1.15 format. 3717 1/M0 has Q0.12 (unsigned) format. 3718

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Retrait :Gauche : 3,23 cm, Avec puces+ Niveau : 1 + Alignement : 1,98 cm + Tabulation après : 2,62 cm + Retrait : 2,62 cm,Tabulations :Pas à 2,62 cm







3719

3720 3721 3722 Then we give the implementation for the filtering part where the filter coefficients are stored under 3723 Q2.15 format. 3724 3725

(16I,16Q)

(16I,16Q)

>>15 Sat16

>>15

Sat16

(16I,16Q)

+ 20

341 (1/12)

Sat16

*ip

τπ fjie ∆2

pilotapiH ,,

ˆ

aj ,α

pilot

Supprimé : 4.0.1

Supprimé : 2







3726 3727 3728

Filter selection process is: 3729 3730

3731 3732 3733 3734 3735 3736 3737

3738

E≥(N<<2)

E≥(N<<4)

E≥(N<<3)

7 6 4 5

Y

Y

E≥(N<<5)

E≥N

E≥(N>>1)

3 2 0 1

Y

Y

E≥(N<<1)

Y Y Y

E= P

N= ( )( ) 165461.21,0 >>−Nσ

s16

(16I,16Q)

>>15 Sat16

aj ,α

aj ,α

Supprimé : 4.0.1

Supprimé : 2







8.8.1 CHANNEL ESTIMATION: ACK NACK & CQI MUX 3739

The channel estimate is only for CQI estimation. It is derived from the ACK/NACK detection (see 3740 8.9.2) and is identical to CQI channel estimation procedure (see 8.8) with the difference that the αs 3741 are directly provided by another block (no frequency descrambling is done). 3742 3743 3744 3745 INTERFACE DEFINITION 3746 3747



Speed enum {…}

I Users Speed of the current PUCCH

users for mapping to the correct auto/cross correlations

e uint u2 I 1 AGC scaling factor

Noise uint u30 I 2 One noise per PUCCH PRB

Power uint u30 I Users One power per user

alpha complex Integer

(16I,16Q) I 4x4xUsers CQI/ACK&NACK estimator output

Out complex Integer

(16I,16Q) O 4 x 6 x Users

6 channel estimations per antenna and user

Table 8.1: Interface definition for CQI channel estimation 3748

3749

8.9. DATA ESTIMATION 3750

8.9.1 CQI ONLY 3751

Frequency offset compensation is first done, time compensation with constant time shift is also 3752

performed. Then outputs of CAZAC sequence multiplication are accumulated per symbol. 3753

( )( )∑−

=

∆=1

0,

*

0,

0

..2exp1 M

i

akiiak YpfijT

My τπ for { }4,3,2,1,0∈k and 3754

( )( )∑−

=+∆=

1

0,

*

0,

0

..2exp1 M

i

akNiiak YpfijT

My τπ for { }9,8,7,6,5∈k 3755

This is then proceeded by MMSE where the estimated amplitude corresponding to the data is used 3756

for combination: 3757

2)(

0

1

0

2

),(

1

0,

*),(

1ˆ

ˆ

kp

N

aakm

N

aakakm

k

M

ys

rx

rx

σα

α

+=

∑

∑−

=

−

= for { }9,...,0∈k 3758

Supprimé : 4.0.1

Supprimé : 2







with m(5n) = 3n, m(5n+1)=3n+1, m(5n+2)=3n+1,m(5n+3)=3n+1,m(5n+4)=3n+2 for n={0,1}0 stands 3759 for the first slot, 1 for the second one. 3760 and p(0…4)=0, p(5…9)=N-1 3761 3762 In practice, we can omit the division without performance degradation, so that we can work with: 3763

∑−

=

=1

0,

*),(ˆ

rxN

aakakmk ys α for { }9,...,0∈k 3764

3765

Then QPSK demapping is performed:

==

+ )Im(

)Re(

12

2

kk

kk

st

stfor 90 ≤≤ k 3766

3767 This gives 20 outputs per TTI which are feeding the CQI decoder. 3768

3769





(16I,16Q) I 12 x 10

Received Main Data for the data blocks of the TTI for PUCCH after



(16I,16Q) I 12 x 10

Received Div Data for the data blocks of the TTI for PUCCH after



(16I,16Q) I 12 x 10




(16I,16Q) I 12 x 10




(16I,16Q) I 4 x 6 x Users

6 channel estimations per antenna and user

Code_f pointer Int* I 1 Pointer to the frequency codes (p)

Noise Unsigned

integer u30

I 2 Estimated noise

e Uint u2 I 1 AGC scaling factor

Out Int s16 O 20 x Users CQI soft output

3771 Table 8-2: Interface definition for CQI MMSE 3772

3773 3774 FIX IMPLEMENTATION 3775 3776 Time offset exponential and CAZAC compensation are identical to CQI pilot. 3777 CQI MMSE is given below for the real part. The imaginary part is similar. 3778 3779

Supprimé : 4.0.1

Supprimé : 2







3780

The parameter “ o ” defines the size of the shift chosen in order to get 11 bits at the output. 3781

ks is written as ]...[ 3210 sss over 32 bits, where 0s is the sign bit, 32s the LSB. 3782

For the “ o ” evaluation, let us consider “order” the function computing the number of signed bits before the 3783 first significant bits. 3784

iscaling =31-order( iks ) where { }19,...,0∈i . 3785

[ ] 10max −= ii

scalingo . 3786

3787

8.9.2 ACK NACK & CQI 3788

Two steps are carried out: 3789 1) ACK/NACK estimation, 3790 2) CQI estimation using ACK/NACK outputs. 3791 3792 ACK/NACK estimation 3793

From the frequency compensated samples available along the ACK/NACK sequence, time 3794 compensation with constant time shift is done (corresponding to the chosen timing for the 3795

implementation ie TAττ = in the equations below, cf. section 6.2.2). Then CAZAC de-multiplexing 3796

is performed with candidate sequences. To avoid introducing new notations, we keep pilotapiH ,,

ˆ 3797

although it here addresses the ACK/NACK sequence. 3798

On next equation, the user index is omitted for sake of simplicity: 3799

( )( )∑−

=

∆=1

0,,

*,

0,

0

ˆ...2exp1 M

i

pilotapipiap HpfijT

Mτπα for { }1,0∈p 3800

(16I,16Q)

(16I,16Q)

ajy ,

aj ,α

>>1

31

>>o

11

antennas

SAT32

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Numéros +Niveau : 1 + Style denumérotation : 1, 2, 3, … +Commencer à : 1 + Alignement: Gauche + Alignement : 1,25cm + Tabulation après : 1,89cm + Retrait : 1,89 cm







( )( )∑−

=+∆=

1

0,,

*,

0,

0

ˆ..2exp1 M

i

pilotapNipiap HpfijT

Mτπα for { }3,2∈p 3801

3802

MRC outputs are then computed: ( ) ( )( )

+= ∑

−

=

∗∗1

0,3,2,1,0Reˆ

RXN

aaaaaIs αααα and if QPSK is used 3803

( ) ( )( )

+= ∑

−

=

∗∗1

0,3,2,1,0Imˆ

RXN

aaaaaQs αααα 3804

Then, the demapping is performed as followed for Double Code Word (rotated QPSK): 3805 3806 if ( )QI ss ˆˆ ≥ 3807

if ( )QI ss ˆˆ −≥ 3808

s = 1 (ie NACK/NACK) 3809 else 3810 s = -j (ie NACK/ACK) 3811 else 3812 if ( )QI ss ˆˆ −≥ 3813

s = j (ie ACK/NACK) 3814 else 3815 s = -1 (ie ACK/ACK) 3816 3817 And for Single Code Word (BPSK): 3818 3819 if ( )0ˆ ≥Is 3820

s=1 (ie NACK) 3821 else 3822 s=-1 (ie ACK) 3823 3824 From this ACK/NACK values we compute the following: 3825

=

=

aa

aa

s ,1*

,1

,0,0

αα

ααand

=

=

aa

aa

s ,3*

,3

,2,2

αα

αα 3826

3827 This is provided to the CQI&ACK/NACK channel estimator (see 8.8.1). 3828

3829 INTERFACE DEFINITION 3830 3831




(16I,16Q) I 2 x 12

Received Main Data for the ACK/NACK & pilot blocks of the TTI after CFO compensation with L=1


(16I,16Q) I 2 x 12

Received Div Data for the ACK/NACK & pilot blocks of the TTI after CFO compensation with L=1


(16I,16Q) I 2 x 12



(16I,16Q) I 2 x 12


Supprimé : 4.0.1

Supprimé : 2







sequence Int* 1 O Users Pointer to frequency sequence (p)

Out complex Integer

(I,Q) O 2xUsers ACK NACK hard output


(16I,16Q) O 4xUsersx2 Channel impulse response

estimations

3832 Table 8-3: Interface definition for ACK/NACK data estimation w ith CQI 3833

CQI estimation 3834

Following the channel estimation procedure for CQI&ACK/NACK the CQI data estimation is 3835 performed the same way as CQI only (see 8.9.1). The interface definition is also the same. 3836

3837

8.10. NOISE ESTIMATION 3838

It is performed on pilot sequence after CAZAC de-multiplexing with similar approach as for PUSCH 3839

(ie subtraction of the estimated amplitude to the received CAZAC multiplicated signal) but taking 3840

into account the user multiplexing, frequency and timing offset. It is used for MMSE, and L2 IF 3841 (power control). 3842

U denotes the number of multiplexed users on this field. Notice that LA3.0, we do not allow 3843

simultaneous transmission of CQI and Ack-Nack on the same PUCCH so that these U users 3844 transmit either a CQI or an Ack-Nack. If this assumption was changed in the future, this algorithm 3845

should be revisited. Note the algorithm only works up to 11 (ie 10 −M ) users. 3846

8.10.1 CQI OR CQI&ACK/NACK 3847

For CQI users, we have: 3848

( )( ) ( )( )∑∑∑ ∑= =

−

=

−

=

Ρ∆−−

−=

NuupN

upKjfijTXUMN

rx

u

a p

M

i

U

u

V

apipai

rx 0

1

0

1

0

2

1

0,

FFT,

0

20

0

)()()(ˆ..2exp..2exp1

2

11

44444444444 344444444444 21αεπτπσ3849

for first slot. 3850

( )( ) ( )( )∑∑∑ ∑= =

−

=

−

=+−−

Ρ∆−−

−=

NuupN

upKjfijTXUMN

rx

u

a p

M

i

U

u

V

apip

aiNrx

N0

3

2

2

1

0

1

0,

FFT,1

0

21

0

)()()(ˆ..2exp..2exp1

2

11

44444444444 344444444444 21αεπτπσ for 3851

for the second slot. 3852 The notations are the same as for Ack-Nack, and the CFO complex exponentials are given in 3853 section 7.4. 3854

For CQI&ACK/NACK users, we use the same expression but we have to use the channel estimate 3855

before removal of the Ack-Nack symbol:. 3856

( )( ) ( )( )∑∑∑ ∑= =

−

=

−

=

Ρ∆−−

−=

NuupN

upKjfijTXUMN

rx

u

a p

M

i

U

u

V

apipai

rx 0

1

0

1

0

2

1

0,

FFT,

0

20

0

)()()(ˆ..2exp..2exp1

2

11

44444444444 344444444444 21αεπτπσ3857

3858

Supprimé : 4.0.1

Supprimé : 2






Mis en forme : Anglais(Royaume-Uni)( )( ) ( )( )∑∑∑ ∑

= =

−

=

−

=+−−

Ρ∆−−

−=

NuupN

upKjfijTXUMN

rx

u

a p

M

i

U

u

V

apip

aiNrx

N0

3

2

2

1

0

1

0,

FFT,1

0

21

0

)()()(ˆ..2exp..2exp1

2

11

44444444444 344444444444 21αεπτπσ 3859

3860

In the equation above, we have used the notation paiX , to denote the value of the output of the 3861

sub-carrier de-mapping on pilot number p, tone i and antenna a (i.e. the observations prior to any 3862 CFO compensation, which are thus common to all users). 3863 3864 3865 3866 INTERFACE DEFINITION 3867

3868




(16I,16Q) I 12 x 4

Received Main Data for the data blocks of the TTI for PUCCH before CFO compensation


(16I,16Q) I 12 x 4

Received Div Data for the data blocks of the TTI for PUCCH before CFO compensation


(16I,16Q) I 12 x 4



(16I,16Q) I 12 x 4



(16I,16Q) I 4 x 2 x Users 2 channel estimations per

antenna and user

Phase Integer s8 I Users Normalized frequency offset from PUSCH

Noise Unsigned

integer u30

O 2 Estimated noise

3869 Table 8-4 :Interface definition for CQI noise estimation 3870

3871 FIX IMPLEMENTATION 3872 3873

Supprimé : 4.0.1

Supprimé : 2







3874

W is the fix representation of UMNrx −0

11 3875

1 antenna case: 3876 U 1 2 3 4 5 6 W 2979 3277 3641 4096 4681 5461

3877 U 7 8 9 10 11 W 6554 8192 10923 16384 32768 3878 3879 2 antennas case: 3880 U 1 2 3 4 5 6 W 1489 1638 1820 2048 2341 2731

3881 U 7 8 9 10 11 W 3277 4096 5461 8192 16384 3882

3883 4 antennas case: 3884 U 1 2 3 4 5 6 W 745 819 910 1024 1170 1365

3885 U 7 8 9 10 11 W 1638 2048 2731 4096 8192 3886

3887 As for Ack-Nack, we also perform noise estimation on empty PUCCH PRB, combination of noise 3888 estimates on both slots per TTI, and long term averaging. The implementation is the same and is 3889 described in sections 7.12.2, 7.12.3 and 7.12.4. 3890

||.||2 >>6

+ >>15

paiX ,

16

16

uV

16 30 24

25

Other pilot, antenna, tones

<<2e Sat 30

W

>>1

Supprimé : 4.0.1

Supprimé : 2







8.11. POWER ESTIMATION 3891

It is performed on pilot sequence after CAZAC de-multiplexing on a single amplitude ( ap,α ) per 3892

pilot symbol. On each slot, the average power of ap,α is computed, and the noise power is 3893

subtracted. The result is then averaged on the pilot symbols. It is for L2 IF (power control). It 3894 applies on all PUCCH configurations. 3895

3896 For CQI, we have two pilot symbols per slot, so that the power estimate writes: 3897 3898

( )21

20

00

3

2

2

,0

1

0

2

, 2

1

.4

1−

= == =

+−

+= ∑∑∑∑ N

N

a pap

N

a pap

rx MNP

rxrx

σσαα 3899

3900

FIX IMPLEMENTATION 3901 3902

8.12. DATA DECODING 3903

8.12.1 DESCRAMBLING 3904

Unlike Ack-Nack, CQI bits at the transmitter are bit level scrambled before modulation so that a 3905

descrambling has to be carried out before decoding at the receiver. To do so, the corresponding 3906 scrambling code is first mapped with the transformation { }11 ,10 −→→ , then the resulted 3907 sequence is multiplied term by term by the sequence at the input of the transform Hadamard. 3908

||.||2

>>16

ap,α

16

16

30

5461

>>log2(Nrx)

1st slot

( )21

20

2

1−+ Nσσ

||.||2

||.||2

Other OFDM symbol, antenna port

SAT32

>>1

>>7

2nd slot

Supprimé : 4.0.1

Supprimé : 2







8.12.2 DECODING 3909

The encoding scheme for CQI transmission in PUCCH is done with a uniformly punctured Reed 3910 Muller (RM) code. The generating matrix (11 by 20) of the RM mother code denoted GM is given 3911 in 36.212. 3912

Let us denote K the length of the CQI information bits varying between 1 and 11 , and N=20 as 3913 the considered Reed Muller code length. 3914

Let us notice that the PUCCH CQI coding scheme is a punctured version of the PUSCH CQI 3915 coding scheme, where the last twelve bits are punctured. 3916 3917

Therefore, we can apply the CQI PUSCH decoding algortithm described in section 10.6.1, by 3918 considering that the codeword’s last twelve bits are punctured, 3919

3920

8.12.3 RELIABILITY METRICS FOR CQI 3921

For Reliability metrics definition please refer to 10.6.2. 3922 The thresholds as the performance of the CQI reliability detection algorithm are given below, within 3923 a first transmission BLER target of 1% for CQI and 0.1% for RI. 3924

3925 K False Alarm Probability Missed Detection Probability Threshold 1 3% 3% 21 2 9.7% 9.7% 2.02 4 7.15% 7.15% 1.42 6 6.58% 6.58% 1.3 8 6.07% 6.07% 1.17

3926 Table 8-2 Thresholds and Reliability Metrics for CQI on PUCCH 3927

3928 3929

9. UPLINK CONTROL CHANNELS ON PUSCH 3930


Compared to LA2.0, the maximum supported BLER Target has been increased from 10% to 50%. 3932 As a consequence, the values of the parameters PUSCH

offsetβ will be modified. The exact values are 3933 still TBD, depending on pending simulations. 3934

9.2. NUMBER OF QAM SYMBOLS OCCUPIED BY RI AND ACK-N ACK 3935

The number of QAM symbols occupied by Ack-Nacks and rank indicators is described in TS 3936 36.212 V8.5.0 and is reproduced below. 3937

When the UE transmits Ack-Nack bits or rank indicator bits, it shall determine the number of 3938 QAM symbols Q′ for Ack-Nack or rank indicator as: 3939

⋅

⋅⋅⋅

=′

∑−

=

−−PUSCHscC

rr

PUSCHoffset

initialPUSCHsymb

initialPUSCHsc M

K

NMOQ 4,min

1

0

β 3940

Supprimé : 4.0.1

Supprimé : 2







3941

where O is the number of ACK/NACK bits or rank indicator bits, PUSCHscM is the scheduled 3942

bandwidth for PUSCH transmission in the current sub-frame for the transport block, expressed as 3943

a number of subcarriers and initial-PUSCHsymbN is the number of SC-FDMA symbols per subframe for 3944

initial PUSCH transmission given by ( )( )SRSULsymb

initial-PUSCHsymb 12 NNN −−⋅= , where SRSN is equal 3945

to 1 if UE is configured to send PUSCH and SRS in the same subframe for initial transmission or 3946 if the PUSCH resource allocation for initial transmission even partially overlaps with the cell 3947 specific SRS subframe and bandwidth configuration defined in Section 5.5.3 of TS 36.211 V8.5.0. 3948

Otherwise SRSN is equal to 0. initialPUSCHscM − , C , and rK are obtained from the initial PDCCH for 3949

the same transport block. Notice that for LA3.0, we always have SRSN =1. 3950

For HARQ-ACK information QQQ mACK ′⋅= and [ ACKHARQoffset

PUSCHoffset

−= ββ ], where ACKHARQ

offset−β

is 3951

transmitted by higher layers. 3952

For rank indication QQQ mRI ′⋅= and [ RIoffset

PUSCHoffset ββ = ], where RI

offsetβ is transmitted by higher 3953

layers. 3954

Notice that for LA3.0, since we have only two Rx antennas, the number of Ack-Nack bits can be 3955 equal to 1 or 2, whereas the number of rank Indicator bits is equal to 1. 3956

In LA3.0, the values ACKHARQoffset

−β and RIoffsetβ are cell specific, so that all users uses the same 3957

values. Notice that since we have one value per cell, these values will have to be dimensioned to 3958 the worse case corresponding to the lower authorized MCS configurations. These values have 3959 been determined by simulations to meet the specific requirements below. 3960

• The value of RIoffsetβ have been determined to ensure a RI false detection of 0.1%. This 3961

value is quite low and has been chosen so as to avoid HARQ buffer corruption. Actually, 3962 the Rank indicator value will determine the size of the CQI to be encoded at the UE size, 3963 and will therefore modify the rate matching function. A misdetection of RI will therefore 3964 impact rate matching and will imply HARQ buffer corruption. For LA3.0, the 3965 recommended value is increased compared to LA2.0 due to increase of initial BLER 3966 transmission. The exact value depends on pending simulations and will be given in a 3967 further version of the document 3968

3969

• For Ack-Nack, the value of ACKHARQoffset

−β shall be chosen so as to ensure that both 3970

( )DTXAckP → and ( )AckDTXP → are below 1%. The methodology to set this 3971

parameter is as follows: 3972

o For each candidate value of ACKHARQoffset

−β , compute the threshold ensuring 3973

that ( )AckDTXP → is below 1%. 3974

o For each MCS, compute ( )DTXAckP → with the threshold found above at the 3975

SNR ensuring 10% data BLER 3976

o The selected value of ACKHARQoffset

−β is the smallest one ensuring that for all MCS, 3977

( )DTXAckP → is below 1% at the targeted data BLER SNR 3978

• For LA3.0, the recommended value is increased compared to LA2.0 due to increase of 3979 initial BLER transmission. The exact value depends on pending simulations and will be 3980 given in a further version of the document 3981

Supprimé : 4.0.1

Supprimé : 2







3982

Notice also that whatever the final value of ACKHARQoffset

−β we choose, we cannot guaranty that 3983

both ( )AckDTXP → and ( )DTXAckP → are below 1% at the lowest MCS values. 3984

Therefore, we will ensure that ( )AckDTXP → is below 1% since a detection of Ack in 3985

the DTX case implies a higher layer retransmission that is costly. 3986

3987

9.3. SYMBOL EXTRACTION AND LLR COMPUTATION 3988

The QAM symbols containing Ack-Nack and RI should be first extracted from the PUSCH QAM 3989

symbols. 3990

We promote the same approach as in the previous release, i.e. to perform QPSK-like LLR 3991 computation followed by de-scrambling. 3992

LLR sign inversion 3993

Notice that the LLRs for QPSK as given by section 6.5.3 is ( ) ( )( )

===

0

1log

bP

bPbLLR . 3994

Therefore, in order to make sure that the QPSK LLRs actually corresponds to estimates of the 3995

QAM symbols, we need to invert the sign of the LLRs provided by the FPGA B prior to any 3996 processing. 3997

Multiplexing of data and control information is described in sections 5.2.2.7 and 5.2.2.8 of 36.212, 3998 so that UCI extraction should be compliant with this. 3999

After symbol extraction, we have a vector of estimated QAM symbols for RI and another one for 4000 Ack-Nack. These will be denoted as 4001

( ){ }10 ,ˆAck −′≤≤ ACKQiiS and ( ){ }10 ,ˆ

RI −′≤≤ RIQiiS . 4002

Coding schemes of Ack-Nack and RI symbols are described in section 5.2.2.6 of 36.212. Since 4003 the same coding is used for both, we will do a common description for both in the following and 4004

drop the index, considering a vector of QAM symbols ( ){ }10 ,ˆ −′≤≤ QiiS . However, DTX 4005

detection will be done only for Ack-Nack. 4006

The possible values of ACKQ′ and RIQ′ are defined by 3GPP spec 36.212. 4007

These number depend on the following parameters 4008

4009

9.4. CASE OF ONE BIT TRANSMISSION 4010

For one bit transmission, the coding is done by mapping on constellation edges followed by 4011 repetition,. 4012

For Rank Indication detection, the QAM symbol detection is made by hard decision applied to the 4013 average QAM symbol: 4014

( )∑−′

=′=

1

0RIRI

ˆ1 Q

n

nSQ

m 4015

Notice that we use a BPSK hard detection, i.e. 4016

If ( ) ( ) 0ImRe RIRI >+ mm then jm +=1HardRI else jm −−= 1Hard

RI . 4017

Supprimé : 4.0.1

Supprimé : 2







9.4.1 DTX DETECTION 4018

For Ack-Nack detection, we have to add a DTX detection. To do so, as in the majority method, we 4019 first perform QPSK hard decision (not BPSK as above) and then check if the corresponding 4020 detected symbol is one of the two possible symbols at constellation edge that are used for single 4021 bit Ack-Nack transmission (added when moving from LA0.1 to LA1.0, see [16] and [17] for more 4022 details). 4023

If this is not the case, then a DTX is decided. On the contrary, we compute an estimate of the 4024 mean and variance of the QAM vector and chose the vector that has the lowest variance: 4025

4026

( )∑−′

=′=

1

0AckAck

ˆ1 Q

n

nSQ

m and ( ) ( )∑−′

=

−−′

=1

0

2

AckAck2

Ackˆ

1

1 Q

n

mnSQ

V 4027

4028

Notice that as in LA1.0, the factor ( )1−′Q instead of Q′ is here to ensure a non-biased 4029

estimation of the variance given that the mean of AckS is not known (because of the MMSE + 4030

IDFT factor). 4031

The DTX detection is done by the same threshold detection as in LA1.0, i.e: 4032

If ( )'

'

Ack

Ack thresholdQ

Q

V

m<

then a DTX has been transmitted. 4033

We can rewrite this decision rule as follows for simplification with ( ) ( )( )2'' threshold QQt = : 4034

If ( ) ( ) ( )

−<− ∑

−

=

1

0

2

AckAck'2

Ack''

'

ˆ. .1Q

n

mnSQtmQQ then DTX 4035

One last possibility if we want to remove the division used to compute the average 4036

( )∑−′

=′=

1

0AckAck

ˆ1 Q

n

nSQ

m is to perform the following detection: 4037

If ( ) ( ) ( )

−<− ∑

−

=

1

0

2

Ack'

Ack''2

Ack'''

'

.ˆ.. ..1Q

n

mQnSQQtmQQQ then DTX 4038

HARDWARE IMPLEMENTATION 4039

Notice that for Hardware implementation, the metric above is not suited since it requires to keep 4040 in memory all the QAM symbols to compute their average. 4041

We can compute the variance as below: 4042

( ) ( ) 21

0

2

Ack2

1'

'ˆ1'

1Ack

Q

nAck m

Q

QnS

QV

−−

−= ∑

−′

=

4043

The threshold detection then writes: 4044

If ( ) ( ) ( ) ( )

−<− ∑

−

=

1

0

2

Ack''

2

Ack

2''2

Ack'''

'

..ˆ.. ..1Q

n

mQQnSQQtmQQQ then DTX 4045

4046

Supprimé : 4.0.1

Supprimé : 2








Computation of ( ) ( )∑−

=

1

0

2

Ack

2'

'

ˆ.Q

n

nSQ 4048

4049

Computation of ( ) 2

Ack''' ..1 mQQQ − : 4050

4051

4052

4053

4054

4055

4056

4057

4058

4059

4060

4061

4062

4063

4064

4065

4066

4067

( ) ( )∑−

=

1

0

2

Ack

2'

'

ˆ.Q

n

nSQ

||.||2 ( )0ˆ

AckS

(8I, 8Q)

14+log2(Q’) [Max 22]

||.||2

14

||.||2

14

M M

( )2'Q : u16

( )2'Âck −QS

( )1'Âck −QS

>>(log2(Q’)) >>2.(log2(Q’))

14

15

15

14

( )0ÂckS

(8I, 8Q) ||.||2

M

( )2'Âck −QS

( )1'Âck −QS

>>2.(log2(Q’))

'Q : u8

( 'Q -1): u8

X = ..2

Ack'' mQQ

( ) ..12

Ack''' mQQQ −

X

8+log2(Q’) 14+log2(Q’)

>>(log2(Q’))

15+log2(Q’)

15

15+log2(Q’)

Supprimé : 4.0.1

Supprimé : 2







The threshold is expected to be coded as Q(5,11). This give a maximum threshold value of 16 4068 with a granularity of 0.0156. 4069

Threshold comparison is described below: 4070

4071

4072

4073

4074

4075

4076

4077

4078

4079

4080

• These thresholds will be determined by simulations. For LA3.0, the recommended value 4081 is increased compared to LA2.0 due to increase of initial BLER transmission. The exact 4082 value depends on pending simulations and will be given in a further version of the 4083 document 4084

4085

Notice also that whatever the final value of ACKHARQoffset

−β we choose, we cannot guaranty that 4086

both ( )AckDTXP → and ( )DTXAckP → are below 1% at the lowest MCS values. 4087

Therefore, we will ensure that ( )AckDTXP → is below 1% since a detection of Ack in 4088

the DTX case implies a higher layer retransmission that is costly. 4089

4090

Hereafter we give the thresholds determined by considering P(DTX-->ACK) = 1%. We remind that when 4091 considering AN on PUSCH, a DTX does not mean that nothing was sent but that AN was not multiplexed 4092 inside PUSCH. 4093 4094

I_MCS Thresholds (Floating point)

Thresholds (Q5.11)


Thresholds (Q5.11)

0 4.3765 8963 11 3.8338 7852

1 3.7365 7652 12 3.8809 7948

2 3.3746 6911 13 4.0723 8340

3 3.4484 7062 14 4.1006 8398

4 3.4373 7040 15 4.3139 8835

5 3.5006 7169 16 4.5668 9353

6 3.5948 7362 17 4.7961 9822

7 3.6557 7487 18 4.8180 9867

( ) ( )∑−

=

1

0

2

Ack

2'

'

ˆ.Q

n

nSQ

..2

Ack'' mQQ

( ) ..12

Ack''' mQQQ −

( ) 'Qt

-

+ >>11 Comparison

Supprimé : 4.0.1

Supprimé : 2







8 3.7056 7589 19 4.9729 10184

9 3.7791 7740 20 5.4476 11157

10 3.8142 7812 22 5.4616 11185

Table 9-3: AN 1bit, beta_offset=15.875 4095 4096


Thresholds (Q5.11)


Thresholds (Q5.11)

0 4.3765 8963 11 3.6864 7550

1 3.7908 7764 12 3.7520 7684

2 3.4040 6971 13 3.8573 7900

3 3.3856 6934 14 3.8691 7924

4 3.4077 6979 15 4.0522 8299

5 3.4299 7024 16 4.1412 8481

6 3.4745 7116 17 4.2436 8691

7 3.5457 7262 18 4.2560 8716

8 3.5457 7262 19 4.5924 9405

9 3.5986 7370 20 4.8753 9985

10 3.7365 7652 22 4.9640 10166

Table 9-4: AN 1bit, beta_offset=20 4097 4098 4099 4100


Thresholds (Q5.11)


Thresholds (Q5.11)

0 4.3765 8963 11 3.5044 7177

1 4.0321 8258 12 3.5759 7323

2 3.5608 7292 13 3.5759 7323

3 3.3856 6934 14 3.6557 7487

4 3.3930 6949 15 3.6979 7573

5 3.3636 6889 16 3.7288 7637

6 3.3930 6949 17 3.7830 7748

7 3.4820 7131 18 3.8691 7924

8 3.4225 7009 19 3.9085 8005

9 3.4745 7116 20 4.0401 8274

10 3.5156 7200 22 4.0925 8381

Supprimé : 4.0.1

Supprimé : 2







Table 9-5: AN 1bit, beta_offset=31 4101 4102 4103


Thresholds (Q5.11)


Thresholds (Q5.11)

0 4.3765 8963 11 3.3930 6949

1 4.0321 8258 12 3.4188 7002

2 3.6138 7401 13 3.4857 7139

3 3.8103 7804 14 3.5194 7208

4 3.3930 6949 15 3.5495 7269

5 3.3233 6806 16 3.5231 7215

6 3.3233 6806 17 3.5910 7354

7 3.3379 6836 18 3.6214 7417

8 3.3930 6949 19 3.6481 7471

9 3.3893 6941 20 3.7018 7581

10 3.3856 6934 22 3.6749 7526

Table 9-6: AN 1bit, beta_offset=50 4104 4105

4106

4107

4108

FIXED POINT IMPLEMENTATION : COMMON IMPLEMENTATION WITH 2 BITS CASE 4109

4110

For hardware implementation, it is preferable that the one and two bits Ack-Nack transmission 4111 use the same device. This means that the single bit detection algorithm has to be made as a 4112 particular case of the two bits transmission (even if this makes single bit detection more complex). 4113

4114

This is possible if we change some of the parameters of two bits detection as it is described in 4115 section 9.5. Therefore, we describe below the algorithms parameters that needs to be changed 4116 for 1 bit transmission. 4117

We have only two hypotheses to check which are the following ones: 4118

4119

[ ] [ ]1110 :Η 21000 =⇒= bbboACK 4120

[ ] [ ]1111 :Η 21001 −−−=⇒= bbboACK 4121

4122

The sequence detection is the same as in two bits transmission with these two hypotheses, and 4123 is common to Rank Indicator and Ack-Nack detection. 4124

For DTX detection, we need to skip the step 1 since checking the bits encoding is not necessary 4125 for single bit transmission. 4126

Supprimé : 4.0.1

Supprimé : 2







The threshold detection is the same since the method to compute the variance is generic. 4127

Therefore, we can use exactly the same fixed point implementation as for 2 bits case by changing 4128 the values of the thresholds and the sequence detection. 4129

4130

9.4.2 ACK-NACK AND RI BIT DETECTION 4131

Once the QAM symbol has been estimated, we can detect the corresponding bit as follows: 4132

If ( ) jiS += 1ˆ Hard , then the decided bit is equal to 0 4133

If ( ) jiS −−= 1ˆ Hard , then the decided bit is equal to 1 4134

9.5. CASE OF TWO BITS TRANSMISSION 4135

9.5.1 CODING SCHEME 4136

The common coding scheme for both RI and Ack-Nack is described in the table below: 4137

Qm Encoded HARQ-ACK 2 ] [ 210210

ACKACKACKACKACKACK oooooo

4 x x] x x x x [ 210210ACKACKACKACKACKACK oooooo

4138

where 2mod) ( 102ACKACKACK ooo += . 4139

This means that each couple of bits ), ( 10ACKACK oo is mapped on three QAM symbols (before 4140

repetition). We describe below the corresponding mapping for each possible configuration: 4141

4142

4143

Configuration number

Transmitted Ack-Nack bits Corresponding QAM symbols

0 )0,0(), ( 10 =ACKACK oo

+++

i

i

i

A

1

1

1

.

1 )1,0(), ( 10 =ACKACK oo

−−+−

−

i

i

i

A

1

1

1

.

2 )0,1(), ( 10 =ACKACK oo

−−−+−

i

i

i

A

1

1

1

3 )1,1(), ( 10 =ACKACK oo

+−−−−

i

i

i

A

1

1

1

Supprimé : 4.0.1

Supprimé : 2







4144

Here 2

1=A for QPSK and 10

3=A for 16QAM. For each configuration, the Ack-Nack bits 4145

can obtained from classical QAM de-mapping of the first of the three QAM symbols. 4146

As for the case of single bit transmission, we give below two methods to estimate the transmitted 4147 bits, keeping in mind that, depending on whether decoding is done in DSP or FPGA, we can 4148 chose to embed both or only one of them. 4149

If only one of both methods has to be selected, as for single bit transmission, we recommend to 4150 use the threshold detection. 4151

4152

In order to derive the corresponding algorithms, we first rewrite the observed signal after QPSK 4153 LLR processing. 4154

Let us denote [ ]210 bbb the (-1, +1) values of the bits ),, ( 210ACKACKACK ooo , i.e. 4155

( ) 2 ,1 ,0 ,21 =−= kob ACKkk . 4156

At the receiver, after Frequency Domain Equalizer and IDFT, we perform QPSK LLR computation 4157 to separate the real and imaginary parts of the IDFT output. The obtained vector has a size 2Q’ 4158 and is approximately equal to: 4159

( )( )( )( )

( )( )( )( )

w

b

b

b

b

b

b

A

QS

QS

S

S

Y

k

k

+

≈

−−

=

−1

0

2

1

0

.

1'ˆIm

1'ˆRe

0ˆIm

0ˆRe

M

M 4160

with 2≤k , and ( ){ }1'0 ,ˆ −≤≤ QiiS is the vector of QAM symbols. 4161

The noise w is assumed to be an AWGN vector (this is a good approximation). 4162

For k=0, 1, 2, the bit kb is repeated kM times, where 13

1'2 +

−−= kQM k , and 4163

'2210 QMMM =++ . 4164

The amplitude coefficient A can be explicitely computed as: 4165

( )( )∑

−

= +=

1

0 1

1 M

m mSNR

mSNR

MCA where ( ) ( )

2

2

σmH

mSNR = is the classical signal to noise ratio on the 4166

tone number m, 10/3=C for 16QAM, 2/1=C for QPSK,and M is the total number of tones 4167 allocated to the user (including its data), i.e. M is the IDFT size. 4168

9.5.2 SOFT DETECTION 4169

As in LA1.0, the hard decision may not be sufficient to reach the requirements in all situations, so 4170 that a method more complex and more robust can be needed. 4171

Supprimé : 4.0.1

Supprimé : 2







For i=0, 1, 2, 3, for a hypothesis iΗ corresponding to the transmission of bits ), ( 10ACKACK oo , we 4172

define the following diagonal matrix: 4173

( ) [ ]( )kki bbbbbbDiagB 10210Η −= K 4174

4175

We recall the values of the bits for each hypothesis: 4176

4177

[ ] [ ] [ ] [ ]11100 :Η 210100 =⇒= bbboo ACKACK 4178

[ ] [ ] [ ] [ ]11110 :Η 210101 −=⇒= bbboo ACKACK 4179

[ ] [ ] [ ] [ ]11101 :Η 210102 −−=⇒= bbboo ACKACK 4180

[ ] [ ] [ ] [ ]11111 :Η 210103 −−=⇒= bbboo ACKACK 4181

4182

Under hypothesis iΗ , we have after IDFT: 4183

( ) wBAY Qi += '2.Η. 1 where '2Q1 is a vector of size 2Q’ containing ones. 4184

This can be written equivalently as: 4185

( ) ( )wBAYBY iQii .Η..Η '2 +== 1 4186

The additive noise is still an AWGN. 4187

4188

BITS ESTIMATION 4189

This section is applicable to both RI and Ack-Nack processing. 4190

Under the hypothesis that the transmission has actually occurred (i.e. no DTX case), we know the 4191 transmitted vector corresponds to one of the 4 vectors above and can use this to estimate the 4192 correct configuration. 4193

We first construct the following equivalent statistics 4194

4195

( ) ∑∑−

=

−

=

+==1

0300

1

03

000

.M

ii

M

ii wbMAYY 4196

( ) ∑∑−

=+

−

=+ +==

1

01311

1

013

111

.M

ii

M

ii wbMAYY 4197

( ) ∑∑−

=+

−

=+ +==

1

02322

1

023

222

.M

ii

M

ii wbMAYY 4198

To determine an estimation of the transmitted sequence, we use the maximum likelihood 4199 detection on the vector: 4200

( )

( )

( )

+

=

=

2

1

0

22

11

00

2

1

0

.

εεε

bM

bM

bM

A

Y

Y

Y

Y 4201

Supprimé : 4.0.1

Supprimé : 2







Where ∑−

=+=

1

03

kM

ikik wε , k=0, 1, 2. 4202

4203

Under AWGN assumption, we have ( )( ) ( ) 2IDFTPost

1

0

2

32 . −

−

=+ =Ε=Ε ∑ σε k

M

ikik Mw

k

where 2IDFTPost. −σ 4204

is the post-IDFT noise variance. 4205

Therefore, Maximum Likelihood is equivalent to minimizing the following metric: 4206

( )

∑= −

−=

2

02

IDFTPost

2

H .

..minargH

k k

kkk

M

bMAY

i σ 4207

After straightforward algebraic manipulations, this is equivalent to selecting the hypothesis Η as 4208 follows: 4209

( )

= ∑=

2

0H

.maxargHi

ii Yb

i

4210

The corresponding processing is described below: 4211

4212

The selected configuration among 0Η , 1Η , 2Η , 3Η directly gives the two Ack-Nack bits 4213

)ˆ, ˆ( 10ACKACK oo by the mapping previously defined for each configuration (see above). 4214

4215


0H

3H

2H

1H

( ) ∑−

==

1

03

00M

iiYY

From IDFT + QPSK LLR

Y

( ) ∑−

=+=

1

013

11M

iiYY

( ) ∑−

=+=

1

023

22M

iiYY

0b

MAX

), ( 10ACKACK oo

(8I, 8Q)

1b

2b

Selected configuration

Supprimé : 4.0.1

Supprimé : 2







4217

4218

DTX DETECTION 4219

This section is applicable to Ack-Nack processing only. 4220

Once the configuration with maximum likelihood has been selected, we have to perform DTX 4221 detection. This is done in two steps: 4222

• Step 1 : check bits encoding (done inside the FPGA B) 4223

• Step 2 : threshold detection (done inside the DSP) 4224

The first step is the simple verification that the coding scheme of the selected bits is consistent 4225 with the observations. 4226

To do so, )ˆ, ˆ( 10ACKACK oo denoting the bits obtained from ML scheme of previous section, we 4227

perform the three DTX detections below: 4228

• First bit check: if

≠− ∑

−

=

1

030

0

signˆ21M

ii

ACK Yo then DTX 4229

• Second bit check: if

≠− ∑

−

=+

1

0131

1

signˆ21M

ii

ACK Yo then DTX 4230

• Third bit check: if ( )( )

≠+− ∑

−

=+

1

0232mod10

2

signˆˆ21M

ii

ACKACK Yoo then DTX 4231

Notice that whenever one of the three metrics appearing in the signs above is equal to zero, this 4232 first DTX detection is skipped. 4233

When a DTX is not detected with this first detection test, then the LLR on the Ack-Nack positions 4234 are set to zero in order to avoid HARQ buffer corruption with Ack-Nack. On the other side, when a 4235 DTX is detected, these LLR on Ack-Nack positions are not set to zero since they correspond to 4236 data according to the algorithm. Notice that this is valid only for this first detection and not for the 4237 second detection described below since this is the only one done inside the FPGA B. 4238

0H

3H

2H

1H

( ) ∑−

=

=1

03

00M

iiYY

Y

( ) ∑−

=+=

1

013

11M

iiYY

( ) ∑−

=+=

1

023

22M

iiYY

0b

MAX

), ( 10ACKACK oo

(8I, 8Q)

8+log2(M0) [maxi : 16 bits]

Selected configuration 1b

2b

9+log2(Q’) [maxi : 17 bits]

Supprimé : 4.0.1

Supprimé : 2







4239

This first DTX detection is not foreseen as sufficient to decrease enough the DTX to Ack 4240 probability so that we add a threshold detection on top. 4241

As for LA1.0, the test consists in a threshold comparison between the estimated mean and the 4242

estimated variance of the vector of repetition. This repetition vector is defined as ( )YB .Η . 4243

In case of non-DTX, This vector is equal to ( ) ( )wBAYB Q .Η..Η '2 += 1 , whereas in case of DTX 4244

transmission, the ones are replaced by independent random variables (equal to the +/-QPSK 4245 LLR of data). 4246

In case of non-DTX, the mean of ( )YB .Η is equal to the amplitude A whereas it is equal to 0 in 4247

case of DTX transmission. 4248

The mean of ( )YB .Η has already been computed (up to a scaling coefficient): this is the metric 4249

used for hypothesis selection and it is equal to 4250

( )

= ∑=

2

0

.ˆ'2

1ˆi

ii Yb

QA 4251

The squared variance of ( )YB .Η can be computed as: 4252

( )( )( ) ( )( ) 21'2

0

221'2

0

222 ˆ1'2

'2

1'2

1ˆ1'2

'2.Η

1'2

1.ΗVar A

Q

QY

QA

Q

QYB

QYBV

Q

ii

Q

ii −

−−

=−

−−

== ∑∑−

=

−

=

4253

Notice that in the case where the quantity ( )

( )∑−

= +=

1

0 1

1 M

m mSNR

mSNR

MA is computed after channel 4254

estimation, we can use it instead of A . In this case, since the mean of ( )YB .Η is known a priori 4255

and not estimated on the treated data, the variance is scaled by 2Q’ instead of (2Q’-1). 4256

4257

The threshold detection is based on the comparison of the following random variable with a 4258 threshold that depends on the number of repetitions Q′ and the user’s MCS: 4259

If ( )MCSQ

Q

V

A, threshold

2

ˆ'

'

<

then a DTX has been transmitted. 4260

We can rewrite this decision rule as follows for simplification 4261

with ( ) ( )( )2'' ,threshold , MCSQMCSQK = : 4262

If ( ) ( ) ( )( )( )2''2

''' .212.,' ˆ.2.122 VQQMCSQKAQQQ −<− then DTX 4263

We have ( ) ( ) ( )22

0

''2

''' .ˆ.122ˆ.2.122 ∑=

−=−i

ii YbQQAQQQ and 4264

( ) ( )( ) ( ) ( )22

0

1'2

0

22'2'2' .ˆ'22122 ∑∑=

−

=

−

=−

i

ii

Q

ii YbQYQVQQ 4265

So that finally the threshold comparison can be rewritten: 4266

Supprimé : 4.0.1

Supprimé : 2






Mis en forme : Anglais(Royaume-Uni)If ( ) ( ) ( ) ( ) ( )

−

<− ∑∑∑

=

−

==

22

0

1'2

0

2222

0

'' .ˆ'2'2,'.ˆ.12.2i

ii

Q

ii

i

ii YbQYQMCSQKYbQQ then DTX. 4267

The values of the thresholds ( )MCSQK ,' will be chosen to ensure that 4268

( ) 210−<→ AckDTXP . 4269

4270

4271

4272

4273

4274

FIXED POINT IMPLEMENTATION: 4275

4276

-

+

( ) ∑−

=

=1

03

00M

iiYY

From IDFT + QPSK LLR

Y

( ) ∑−

=+=

1

013

11M

iiYY

( ) ∑−

=+=

1

023

22M

iiYY

0b

∑−

=

1'2

0

2Q

iiY

'22

Q×•

)1'2( −× Q

( )MCSQK ,'× Threshold Comparison

Coding scheme consistency

DTX/NO DTX

( )2'2Q×

1b

2b

Supprimé : 4.0.1

Supprimé : 2







4277

The threshold is expected to be coded as Q(5,11). This give a maximum threshold value of 16 with a 4278

granularity of 0.0156. 4279

Threshold comparison is described below: 4280

4281

4282

4283

4284

4285

4286

4287

4288

4289

( ) MCS ,'QK

>>11 Comparison

-

+

Y : From IDFT + QPSK LLR

∑−

=

1'2

0

2Q

iiY

'2Q : u9

)1'2( −× Q

( )MCSQK ,'× Threshold Comparison

( )2'2Q×

15

( )∑=

2

0

.ˆi

ii Yb

9+log2(Q’)

2•

>>(log2(Q’))

( )22

0

.ˆ'2 ∑=i

ii YbQ

16+log2(Q’)

>>(log2(Q’))

17

>>(log2(Q’))

>>(log2(Q’))

>>(2.log2(Q’))

>>(log2(Q’))

17+log2(Q’)

18+log2(Q’)

17

Supprimé : 4.0.1

Supprimé : 2







These thresholds will be determined by simulations by considering P(DTX-->ACK) = 1%. We 4290 remind that when considering AN on PUSCH, a DTX does not mean that nothing was sent but 4291

that AN was not multiplexed inside PUSCH. 4292

4293

4294


Thresholds (Q5.11)


Thresholds (Q5.11)

0 8.5089 17426 11 7.1717 14688 1 7.1396 14622 12 7.4038 15163 2 7.0543 14447 13 7.4693 15297 3 6.9116 14155 14 7.5295 15420 4 6.9960 14328 15 7.8008 15976 5 6.8278 13983 16 7.8568 16091 6 6.9432 14220 17 7.9693 16321 7 6.9907 14317 18 8.1739 16740 8 7.1824 14710 19 8.4623 17331 9 7.1985 14743 20 8.5966 17606 10 7.1824 14710 22 8.9043 18236

Table 7: AN 2bits, beta_offset = 15.875 4295 4296


Thresholds (Q5.11)


Thresholds (Q5.11)

0 8.5089 17426 11 7.0543 14447 1 7.1396 14622 12 7.0384 14415 2 7.1610 14666 13 7.2469 14842 3 7.1663 14677 14 7.3875 15130 4 6.9960 14328 15 7.5131 15387 5 6.8278 13983 16 7.6231 15612 6 6.8225 13973 17 7.7062 15782 7 6.9380 14209 18 7.7562 15885 8 7.0172 14371 19 7.9637 16310 9 7.0703 14480 20 8.0542 16495 10 7.0543 14447 22 8.2254 16846

Table 9-8: AN 2bits, beta_offset = 20 4297 4298

Supprimé : 4.0.1

Supprimé : 2








Thresholds (Q5.11)


Thresholds (Q5.11)

0 8.5089 17426 11 6.9802 14295 1 7.1396 14622 12 6.9222 14177 2 7.1610 14666 13 7.0066 14350 3 7.7841 15942 14 7.0756 14491 4 6.9960 14328 15 7.1610 14666 5 6.8278 13983 16 7.1717 14688 6 6.7185 13759 17 7.2954 14941 7 6.8487 14026 18 7.2630 14875 8 6.9432 14220 19 7.4420 15241 9 6.8435 14015 20 7.4311 15219 10 6.9432 14220 22 7.5240 15409

Table 9-9 : AN 2bits, beta_offset = 31 4299 4300


Thresholds (Q5.11)


Thresholds (Q5.11)

0 8.5089 17426 11 6.9802 14295 1 7.1396 14622 12 6.9222 14177 2 7.1610 14666 13 6.9222 14177 3 7.7841 15942 14 6.8539 14037 4 6.9960 14328 15 6.9327 14198 5 6.8278 13983 16 6.9116 14155 6 6.7288 13781 17 6.9327 14198 7 6.8487 14026 18 7.0278 14393 8 6.9432 14220 19 7.0013 14339 9 6.8435 14015 20 7.0543 14447 10 6.9432 14220 22 7.1289 14600

Table 9-10 : AN 2bits, beta_offset = 50 4301 4302

4303

Supprimé : 4.0.1

Supprimé : 2







10. CQI ON PUSCH 4304


The symbols extraction below should be applied only in the case where the CQI are multiplexed with the 4306

data. 4307

• Since CQI on PUCCH is supported, we have to support the decoding of block-coded CQI 4308 on PUSCH (the case of 11 bits or less) 4309

• Reliability metrics are computed for CQI block decoding 4310

• In LA2.0, only a small subset of the possible configurations were supported in terms of 4311 (MCS, PRB). This is no more the case in LA3.0. Nevertheless, some restrictions will be 4312 put on these configuration and are currently still under study. The CWV is that the 4313

MCS<6 will be banned for CQI transmission (to be confirmed with UL L2 teams) 4314

• The maximum supported BLER Target has been increased from 10% to 50%. As a 4315

consequence, the values of the parameters PUSCHoffsetβ will be modified. The exact values 4316

are still TBD. 4317

• The DSP – FPGA interface has been enlarged. 4318

4319

As for LA2.0, CQI transmission without data is not supported 4320

4321

10.2. SYMBOLS EXTRACTION 4322

The QAM symbols containing CQI should be first extracted from the PUSCH QAM symbols. 4323

On the contrary of Ack-Nack and RI, when CQI is multiplexed with UL-SCH, the CQI bits are not mapped 4324

on the edge of the modulation. Instead, the same modulation as UL-SCH is used so that LLR computation 4325 is the same for CQI as for UL-SCH symbols. 4326

Multiplexing of UL-SCH and CQI symbols is described in sections 5.2.2.7 of 36.212, so that UCI extraction 4327

should be compliant with this (CQI are placed first, before UL-SCH). 4328

The de-scrambled LLRs are sent to the decoder. Moreover, the CQI bits are rate-matched in a specific way 4329

as described in 5.1.4.2 section of 36.212 and specific rate-de-matching should be performed after symbol 4330 extraction. 4331

10.3. NUMBER OF QAM SYMBOLS OCCUPIED BY CQI 4332

10.3.1 GENERAL CONSIDERATIONS 4333

The number of CQI bits is given in TS 36.212 and is reproduced below: 4334

When the UE transmits CQI bits, it shall determine the number of coded symbols Q′ for CQI as 4335

Supprimé : 4.0.1

Supprimé : 2







−⋅

⋅⋅⋅+=′

∑−

=

−−

m

RIPUSCHsymb

PUSCHscC

rr

PUSCHoffset

initialPUSCHsymb

initialPUSCHsc

Q

QNM

K

NMLOQ ,

)(min

1

0

βwhere O is the number 4336

of CQI bits, L is the number of CRC bits given by ≤

=otherwise8

110 OL , QQQ mCQI ′⋅= and 4337

[ CQIoffset

PUSCHoffset ββ = ], where CQI

offsetβ is transmitted by higher layers. If rank indicator is not transmitted then 4338

0=RIQ . Notice that in LA2.0, we always had O>11 so that L=8. In LA3.0, both L=0 and L=8 are 4339

supported. 4340 initialPUSCH

scM − , C , and rK are obtained from the initial PDCCH for the same transport block and 4341 initial-PUSCH

symbN is the number of SC-FDMA symbols per subframe for initial PUSCH transmission. 4342

For UL-SCH data information RICQIm QQQMNG −−⋅⋅= PUSCHsc

PUSCHsymb , where PUSCH

scM is the scheduled 4343

bandwidth for PUSCH transmission in the current sub-frame for the transport block, and PUSCHsymbN is the 4344

number of SC-FDMA symbols in the current PUSCH transmission sub-frame given by 4345

( )( )SRSNNN −−⋅= 12 ULsymb

PUSCHsymb , where SRSN is equal to 1 if UE is configured to send PUSCH and SRS in the 4346

same subframe for the current subframe or if the PUSCH resource allocation for the current subframe even 4347 partially overlaps with the cell specific SRS subframe and bandwidth configuration defined in Section 5.5.3 4348 of TS36.211. For LA3.0, we always have SRSN =1. 4349

The parameter CQIoffsetβ is chosen so as to reach a CQI BLER of 1% when the UL-SCH BLER is equal to the 4350

target data BLER. In LA2.0, we had 25.2=CQIoffsetβ but this value will be increased in LA3.0 4351

10.3.2 LIMITATIONS FROM THE DSP/FPGA B INTERFACE 4352

We have to take into account the fact that the rate de-matching is done inside the DSP and that the CQI 4353 LLRs computation are done in the FPGA B. Therefore, one limitation comes from the dimensioning of this 4354 interface. 4355

The current working view for the exact size of this interface is that this is equal to 6560LLR per TTI.Notice 4356 that this includes both A-CQI and P-CQI. 4357

10.3.3 NUMBER OF CQI BITS FOR THE DIFFERENT MODES 4358

10.3.3.1 APERIODIC CQI 4359

The number of CQI bits for LA5.0 are given below. 4360

There is no A-CQI for 1.4MHz case. 4361

Mode Corresponding values of RI

CQI bits

1-2 0/1 12

3-0 0/1 12

3-1 0 14

3-1 1 25

Table 10-1 Number of CQI bits for the 3MHz case. 4362 4363

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 3







4364

4365


CQI bits

3-0 0/1 18

3-1 0 20

3-1 1 37

4366



CQI bits

3-0 0/1 22

3-1 0 24

3-1 1 45

4369



CQI bits

3-0 0/1 24

3-1 0 26

3-1 1 49

4372



CQI bits

3-0 0/1 30

3-1 0 32

3-1 1 61

4375


10.3.3.2 PERIODIC CQI 4378

For pediodic CQI, the number of bits is independent of the band and is given in the table below for two 4379 antennas ports. 4380

4381


CQI bits

1-0 0/1 4

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 2

Supprimé : 1

Supprimé : 3

Supprimé : 2

Supprimé : 4

Supprimé : 3

Supprimé : 5

Supprimé : 4







1-1 0 6

1-1 1 8

4382

Table 10-6: Number of CQI bits. 4383 4384

10.4. CQI RATE DE-MATCHING 4385

For both periodic and aperiodic CQI, Rate De-Matching is performed prior the Decoding and it accumulates 4386

the '.QQm CQI LLRs computed by the FPGA-B into CQIBitsN

_ CQI Soft Bits. 4387

CQIBitsN

_ denotes the code length, that is the number of CQI bits just after coding. 4388

For i=0, … 1_

−CQIBits

N , denoting Y the soft bits that feed the decoder, the ith softbit is computed as: 4389

∑−

=⋅+=

1

0_

N

kNkii CQiBits

XY for i = [0, min ( CQiBitsN

_ , '.QQm )] 4390

Where: 4391

• N is the repetition number of the encoded CQI =

CQIBitsN

m QQ

_

'. 4392

• X is the LLR vector from FPGA B + padding with '._

QQNmCQIBits

− zeros. 4393

Fixed Point Implementation 4394

Hereafter, the Scheme shows the fixed point implementation. Notice that this scheme allows to compute 4395 only one CQI soft bits. So it should be repeated CQIBits

N_ times in order to build the output sequence ( at 4396

each iteration ‘i’ is incremented by one) 4397

4398

4399

4400

4401

4402

4403

4404

4405

4406

Possible values for N 4407

Nmax, the maximum value of N, is set from the System Design requirements and from FPGA-B limitation. 4408 Above we state that the minimum number of encoded bits for one CQI message is 30 bits , corresponding 4409

MN LLRs from FPGA B s8

s(8+log2(N)))

>> ( ) N2log

To decoder Figure 10-1 Fixed Point Implementation of rate de-matching

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Retrait :Gauche : 0,63 cm, Suspendu :0,63 cm, Avec puces + Niveau: 1 + Alignement : 0,63 cm +Tabulation après : 1,27 cm +Retrait : 1,27 cm

Supprimé : 5







to a Downlink Transmission Mode 2 in 20MHZ) and the EMIF SIM specification documents restricts the 4410 number of CQI LLRs transferred by the FPGA B to the DSP. Since this has to be revisited for 20MHz, we 4411 will give the new value in a further version of the document. 4412

For the case of aperiodic CQI, the parameter CQIBitsN

_ denotes the number of CQI bits just after 4413

Convolutionnal coding, i.e. prior to repetition. It is given by the table below: 4414

4415

Mode CQI bits CQIBits

N_

1-2 12 60

3-0 12 60

3-1 14 66

3-1 25 99


4418


N_

3-0 18 78

3-1 20 84

3-1 37 135


4421


N_

3-0 22 90

3-1 24 96

3-1 45 159

Table10-9 Number of CQI bits for the 10MHz case. 4422 4423

4424


N_

3-0 24 96

3-1 26 102

3-1 49 171


4427

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Anglais(États-Unis)

Supprimé : 5

Supprimé : 6

Supprimé : 7








N_

3-0 30 114

3-1 32 120

3-1 61 207

Table10-11 Number of CQI bits for the 20MHz case. 4428 4429

4430

For the periodic CQI case, all cases are considered for the number of CQI bits less than 12. 4431

In addition, CQIBitsN

_ is of fixed length namely 32, since CQI is block encoded by a Reed muller code. 4432

4433

10.5. CQI DECODING : CASE OF 12BITS OR MORE 4434

When the number of bits is more than 11bits, the bits are coded with a tail-biting convolutional code. At the 4435 encoder, the initial value of the shift register shall be set to the values corresponding to the last 6 4436

information bits in the input stream so that the initial and final states of the shift register are the same. This 4437

is thus different from a classical convolutional encoder where the initial state of the shift register is equal to 4438 0. 4439

The Tail-Biting LTE Encoder is a binary, rate 1/3 encoder with a constraint length of 7: 4440 4441 4442

4443 Tail baiting LTE convolutional encoder for Aperiodic CQI 4444

4445 Using a classical Viterbi without any modification is thus not possible since classical Viterbi algorithm 4446

requires the knowledge of the initial state of the shift register. 4447

However, from an implementation point of view, we want to use a dedicated coprocessor (currently 4448 unused) that performs classical Viterbi decoding. 4449

The idea is then to add a cyclic prefix and postfix to the data stream at the input of the decoder. Since 4450

original state register is unknown to the decoder, the prefix and postfix allows for initial state data to be 4451 reached. The prefix is taken from the end of the stream, and the postfix from the beginning, as shown 4452 below: 4453

4454

4455

4456

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 8







4457

4458

4459

4460

4461

4462

4463

4464

4465 Prefix and postfix construction prior to decoding 4466

4467

The lengths of the prefix and postfix are both taken equal to 5 times the constraint length minus one: 4468 Postfix, prefix length: = 5·(K-1) = 30 bits. 4469 This extended block is sent to the classical Viterbi decoder. We apply the inverse transformation to the 4470 sequence at the output of the decoder, i.e. we discard the prefix and postfix to retrieve the decoded data. 4471

The CQI transmission includes a CRC. Therefore, CRC check has to be computed, and a CRC flag has to 4472

be given to higher layers. 4473

4474

4475

10.6. CQI DECODING : CASE OF 11BITS OR LESS 4476

10.6.1 DECODING 4477

For 11bits or less, the encoding scheme for CQI transmission on PUSCH is done with a Reed 4478 Muller (RM) code. The generating matrix (11 by 32) of the RM mother code denoted GM is given 4479 in the 36.212 3GPP specifications. 4480

The particular structure of the coding matrix enable to perform decoding using Fast Hadamard 4481 Transforms (FHT). 4482

LLR sign inversion 4483

Notice that the LLRs as given by section 6.5.3 is ( ) ( )( )

===

0

1log

bP

bPbLLR . 4484

Therefore, in order to make sure that the LLRs definition is compliant with the usage of the FHT 4485

decoder as described in this section, we need to invert the sign of the LLRs provided by the 4486 FPGA B prior to FHT. Notice that this is applied for P-CQI only since the decoding of the A-CQI is 4487 done in a Coprocessor that has the same convention as the Xilinx decoder. 4488

4489

Notations: 4490

Let us denote: 4491

Y : The soft received codeword of length 32_ =CQIBitsN . This is the output of the rate dem-4492

matching. 4493

Original block to be decoded

prefix postfix

Expanded block

Supprimé : 4.0.1

Supprimé : 2







{ }0,26,30,31,5,17,22,232,5,8,11,1,28,29,16,1,24,25,2718,19,20,2,12,13,14,4,6,7,9,10,3,1=Ι

K : The number of CQI information bits 4494

( )0,6max −= KsNmaskedbit : The number of masked bits. 4495

X : The CQI information word 4496

If K>6, [ ]110543210 −= sNmaskedbitmmmxxxxxxX K and sNmaskedbitNK += 6 . 4497

If K<7, [ ]KxxxX K10= 4498

In the vectors above, the bits are ordered the same way as in the 36.212 3GPP specifications 4499

sNmaskedbitFHTN 2= : The number of performed FHT transforms. 4500

32H : Hadamard matrix of size 32. 4501

12 −KH : When 6<K , 12 −KH is an Hadamard matrix of size 12 −K . 4502

4503

Two kinds of decoding algorithms are considered depending on wether K is more or less than 6 4504 bits. 4505

4506 K>6: 4507

4508 The decoding is done as explained below: 4509

4510

1. Enumerate exhaustively the sNmaskedbitFHTN 2= mask values. 4511

2. For each mask value ]1:0[ −∈ FHTNm of binary representation ]...[ 10 maskedbitsNmmm 4512

compute the related codeword belonging to the codebook of the shortened mother Reed 4513 Muller code reduced to the masked bits indexes: 4514

4515

×+= ∑−

=

2,)6,(mod)(1

0

0i

sNmaskedbit

iMm

miGY ll for ]31:0[∈l 4516

4517

3. Compute term by term ( )01 21 mm YYY −×= . 4518

4. 1mY is interleaved as in the following 4519

for i=0:31 )())(( 12 iYiIY mm = . 4520

Using the following pattern : 4521 4523

5. Size 32 Hadamard transform of the sequence 2mY is performed to obtain the thm column of 4526

the correlation matrix ∆ of dimension FHTN×32 4527

)()(:, 2mYFHTm =∆ . 4528

6. The maximum element of the matrix ∆ is searched. The couple ),( maxmax ml denotes the 4529

row and column where this maximum is located The row index maxl takes its values in 4530

{0,…,31} and the column index maxm takes its values in {0,…, 12 −sNmaskedbit }. 4531

7. Mask Bits decision: )(]...[ max10 mbinmmmmaskedbitsN = . Where ()bin is the function 4532

converting a decimal number into its binary representation. 4533

8. First Bit decision { }),( maxmax0 msignx l∆= . 4534

Supprimé : 4.0.1

Supprimé : 2







{ }0,26,30,31,5,17,22,232,5,8,11,1,28,29,16,1,24,25,2718,19,20,2,12,13,14,4,6,7,9,10,3,1=Ι

9. Remaining 5 bits decision: The index maxl is decomposed in binary 4535

][ 43210 aaaaa with àa being the MSB. Then the 5 bits decision is taken as 4536

][][ 0123454321 aaaaaxxxxx = 4537

10. The output of the decoding is written as ]:[ 10543210 −sNmaskedbitmmxxxxxx , where 0x is 4538

the MSB. 4539 4540

4541 4542

4543 4544

Block Diagram of Periodic CQI decoding for K>6 4545 4546 4547

K<7 4548

4549 4550 The decoding is done as explained below: 4551

4552 1. Y is interleaved as in the following 4553

for i=0:31 )())((1 iYiIY = . 4554

Using the following pattern : 4555 4557

4559 4561

2. Construction of 2Y an array of dimension )2( 1−K as follows: 4562

∑−

=

−−

×+=12

0

112

6

)2()(K

j

KjiYiY for ]12:0[ 1 −∈ −Ki 4563

Let us call this processing reshaping module. 4564 4565

3. Size 12 −K Hadamard transform of the sequence 2Y is performed to obtain the correlation 4566

array ∆ of dimension 12 −K . 4567

)( 2YFHT=∆ ( Using the Hadamard Matrix 12 −KH ). 4568

Detected CQI

Y

Mask(0)

Mask(m)

Mask( 1−NFHT )

Interleaver

Interleaver

Interleaver

FHT32

FHT32

FHT32

Max

Max

Max

0l

il

1−NFHTl

MAX

maxl

maxm

0∆

m∆

1−∆NFHT

max0∆

maxm∆

max1−∆NFHT Sign( max

maxm∆ )

Decision &

Mapping

2mY

21−NFHTY

20Y 1

0Y

1mY

11−NFHTY

Reliability detection

Reliability Metric

Supprimé : 4.0.1

Supprimé : 2







4. The maximum of ∆ is searched, with index maxl is taking its values in {0,…, 12 −K } 4569

5. First Bit decision { })( max0 l∆= signx where 0x is the MSB. 4570

6. Remaining )1( −K bits decision: The index maxl is decomposed in binary 4571

[ ]20 −Kaa K with àa being the MSB. Then the K-1 bits decision is taken as 4572

[ ] [ ]0211 aaxx KK KK −− = 4573

7. The output of the decoding can be written ]...[ 10 −Kxx where 0x is the MSB. 4574

4575

4576 4577

Block Diagram of Periodic CQI decoding for K<6 4578 4579 4580 4581

10.6.2 RELIABILITY METRICS 4582

A reliability indication is produced by the decoder. This measure allows the scheduler to discard 4583 CQI reports that might have been decoded incorrectly. To do so, the soft information produced 4584 during the decoding stage is used to produce reliability metrics evaluating the CQI reliability 4585 detection algorithm. 4586

4587 Besides, the performance of the reliability is measured through the following statistics: 4588

4589 � False Alarm Probability )( KOOKP : Probability that the decoded CQI is judged to be 4590

reliable given that it is decoded wrongly also denoted FAP . 4591

� Missed Detection Probability )( OKKOP : Probability that the decoded CQI is judged to 4592

be unreliable given that it is decoded correctly also denoted MDP . 4593

4594 We introduce a reliability metric based on the codeword correlation metric generated during 4595

decoding. Let us consider the couple ),( maxmax ml describing the position of the maximum in the 4596

matrix )(Re ∆al . 4597

Let us denote max1θ as the highest magnitude namely )),((Re maxmax mal l∆ . 4598

Let us denote max2θ the second highest magnitude value of the matrix )(Re ∆al . 4599

The reliability metric is defined by the ratio 2max

1max /θθ . The decoding judged more reliable when this 4600

metric is high, and less reliable when it close to 1 This is evaluated through threshold detection. 4601 4602 The decoded CQI is Judged to be reliable if : 4603

Y Interleaver FHT12 −K Max

maxl ∆

Sign max∆ Decision &

Mapping

Reliability Metric

Reliability detection

Reshaping Detected CQI 2Y 1Y

Supprimé : 4.0.1

Supprimé : 2







2max

1max θγθ ×> Th 4604

Where Thγ is a threshold parameter greater than 1. 4605

4606

For the special case where K=1 the metric is evaluated as 4607

21max )))(Re((∑×= iYNθ 4608

and 4609

∑= 22max ))(Re( iYθ . 4610

4611 Where N describes the codelength namely 32 for the PUCSH case , and 20 for the PUCCH case 4612 (described further). 4613 4614 The thresholds as the performance of the CQI reliability detection algorithm are given below, within 4615 a transmission BLER target of 1% for CQI and 0.1% for RI(K=1). 4616

4617 4618

K FAP MDP Threshold Thγ

1 2.4% 2.4% 41 2 10.8% 10.8% 2.08 4 6.96% 6.96% 1.49 6 5.93% 5.93% 1.355 8 6.28% 6.28% 1.224

4619 Table 10-6 Reliability Performance and Thresholds for periodic CQI over PUSCH 4620

4621

Notice that the threshold can be ajusted if one of the probabilitites among FAP or MDP is judged more 4622

important than the other. However, reducing one of these probabilities implying increasing the other one. 4623 Here, we made the implicit hypothesis that both have the same importance. 4624

11. SOUNDING REFERENCE SIGNALS: ECEM 4625

IMPLEMENTATION 4626

4627

11.1. INTRODUCTION 4628

This section corresponds to the legacy SRS processing of previous release (LA3.0 drop 2 version here). 4629 This contains some considerations that are common to eCEM and bCEM implementations, but all fixed 4630

point implementations of this chapter 11 are specific to eCEM. The bCEM implementation is described in 4631

chapter 12. 4632

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 9







11.2. SRS STRUCTURE 4633

11.2.1 GENERAL ASSUMPTIONS 4634

In this section, we describe the main current assumptions for SRS transmission in LA3.0. Notice 4635

that these assumptions constitute the most up to date current working version taken from UL 4636 scheduler design, and have to be updated in case of change of L2 design. 4637

The current working assumptions are: 4638

• Only wideband SRS are used. The SRS bandwidth is equal to 4 PRB for 1.4MHz, 12PRB 4639 for 3MHz, 24PRB for 5MHz, 48 PRB for 10MHz, 72PRB for 15MHz case, and 96PRB for 4640

20MHz case. The same central frequency domain position is used by all users, this 4641

position being configured by the frequency domain position index. For 1.4MHz, the SRS 4642 occupies the band from PRB 1 to 4, the PRBs 0, and 5 being empty For 3MHz, the SRS 4643 occupies the band from PRB 1 to 12, the PRB 0, 13, and 14 being empty. For 5MHz, the 4644

SRS occupies the band from PRB 0 to 23, the PRB 24 being empty. For 15MHz, the SRS 4645 occupies the band from PRB 1 to 72, the PRBs 0, 73, and 74 being empty. 4646

• The UL scheduler shall ensure (if needed) that there is no PUSCH and SRS on the same 4647

OFDM symbol (on different RBs). On the contrary of LA2.0, both combs can be used in a 4648 given TTI. A maximum of 4 UE per comb shall be multiplexed. The combs used depend 4649

on a variable “Transmission comb” taking values in {0, 1}. 4650

• The configuration period (as defined in table 5.5.3.3-1 of [1]) is equal to 1 (i.e. 4651 srsSubframeConfiguration = 0), which means the last symbol of all subframes are 4652

reserved for SRS in the cell. Notice that in debug L1 mode only, the 4653

srsSubframeConfiguration may be equal to 15 (meaning no SRS configured) in order to 4654 permit sensitivity compliance tests of TS 36.104. 4655

• Both combs can be used. A maximum of 4 UE per comb shall be multiplexed. Notice that 4656

LA4.0.1 algorithm for timing offset estimation on SRS requires that consecutive cyclic 4657 shifts are not used. Therefore, even in case on single comb occupation, we still have up 4658

to 4UE multiplexed on this single comb. We can go up to 8UE per comb if we de-activate 4659 timing offset algorithm on SRS. 4660

• Since we can have up to 4UE per comb, the number of users multiplexed inside a SRS 4661

symbol can go up to 8. 4662

• When present, the SRS is located on the last SC-FDMA symbol of the sub-frame 4663

In the frequency domain, SRS is allocated a comb-shaped spectrum, which means that each UE 4664

uses only every other tone for transmission. The different UEs of the same comb use the same 4665

tones. The following figure illustrates the case of Transmission comb=0. 4666

Supprimé : 4.0.1

Supprimé : 2







4667 4668

The following tables lists the different SRS parameters with their default values: 4669

4670

sounding reference signal parameters 36.211 5.5.3.2

UE/cell specific parameter Supported Values comment

nSRS 0,1,2,3,4,5,6,7 Defines SRS cyclic shift

Default shifts for 5 multiplexed SRS: {0,2,4,6,7}

srsBandwidth b=Bsrs

b=Bsrs = 0 defines UE SRS bandwidth with cell- specific srsBbandwidthConfiguration

parameter

transmissionComb kTC

0,1 Default=0

Both transmission comb are supported, but same comb is used for

SRS multiplexed on same RBs

SRSHoppingBandwidth bhop bhop = Bsrs = b = 0 bhop >= Bsrs so that SRS hopping not

enable see SRD-7271-264

UE specific

frequencyDomainPosition nRRC Not needed

Used to offset the frequency domain starting position.

Because only wide band SRS is used, Nb=1, so nb = 0 whatever nRRC

cell specific

srsBbandwidthConfiguration Csrs

3,2,0 Cell specific parameter defining SRS

bandwidth with SRS-bandwidth parameter

4671 4672

4673

Here are the parameters (UE) needed for 213 and the corresponding configurations. 4674

UE-specific parameter SRS Configuration

Index ISRS

SRS Periodicity TSRS (ms)

SRS Subframe Offset Toffset

LA2.0

0 – 1 2 ISRS Not required 2 – 6 5 ISRS – 2 required

7 – 16 10 ISRS – 7 Not required

Tones used by the UE transmitting SRS on this comb

Tones left for other comb

Supprimé : 4.0.1

Supprimé : 2







17 – 36 20 ISRS – 17 Not required 37 – 76 40 ISRS – 37 Not required

77 – 156 80 ISRS – 77 Not required 157 – 316 160 ISRS – 157 Not required 317 – 636 320 ISRS – 317 Not required 637 – 1023 reserved reserved Not required

4675

SRS Configuration Index ISRS

SRS Periodicity TSRS (ms)

SRS Subframe Offset Toffset

SRS transmission instances

(subframes) 2 5 0 0,5 3 5 1 1,6 4 5 2 2,7 5 5 3 3,8 6 5 4 4,9

4676

11.2.2 USER MULTIPLEXING STRATEGY 4677

To multiplex the users on both combs, we need to define which cyclic shifts will be allocated on 4678 each comb and the order in which this will be done. 4679

In order to limit the interference from one comb to the other, we recommend that we first fill the 4 4680 positions of a given comb before starting allocation on the second comb. 4681

Moreover, we recommend to use interleaved cyclic shifts for the same reason. 4682

Therefore, to satisfy the constraints above, we recommend that the cyclic shifts for the first comb 4683 are filled in the order [0, 4, 2, 6] and for the second comb in the order [3, 1, 5, 7]. 4684

4685

11.2.3 SEQUENCE GENERATION 4686

The sounding reference signal sequence ( ) ( ) ( )nrenrnr vunj

vu ,)(

,SRS αα == is defined by Section 4687

5.5.1 of [1]. 4688

The variable { }29,...,1,0∈u is the PUCCH sequence-group number defined in Section 5.5.1.3 of 4689

36.211 and { }1,0∈v is the base sequence number within the group defined in Section 5.5.1.4, and 4690

are given by higher layers. 4691

The cyclic shift SRSα of the sounding reference signal is given as 8

n2 SRSπα = , where

SRSn is 4692

configured for each UE by higher layers and 7,6,5,4,3,2,1,0nSRS = . 4693

The mother CAZAC sequence is the sequence with zero cyclic shift ( ) 1,...,0 ,, −= Mnnr vu 4694

where M denotes the number of SRS tones. This sequence can be constructed as follows: 4695

MnNnXnr qvu <≤= 0),mod ()( RSZC, 4696

4697

where the thq root Zadoff-Chu sequence is defined by 4698

( ) 10,)1(

exp RSZCRS

ZC

−≤≤

+−= NmN

mmqjmX q

π 4699

4700

Supprimé : 4.0.1

Supprimé : 2







and the length RSZCN of the Zadoff-Chu sequence is given by the largest prime number such 4701

that MN ≤RSZC 4702

The values of RSZCN for the different bandwidth are given below: 4703

For 1.4MHz, we use 4PRB, so 24=M and 23RSZC =N 4704

For 3MHz, we use 12PRB, so 72=M and 17RSZC =N 4705





The root sequence q is computed from u and v as below: 4710

31)1(

)1(21RSZC

2

+⋅=

−⋅++=

uNq

vqq q

4711

Notice that the group number and the base sequence number within the group shall not be used in 4712 the rest of document, so that in the following of the document, the variable u shall be used to 4713 numerate user. 4714

On a given cell, the different users have the same CAZAC sequence (which means that they have 4715 the same root q ) and are separated by their different cyclic shifts α . 4716

The SRS signal is multiplied by an amplitude scaling factor SRSβ before transmission, so that the 4717

user u transmits the sequence MnnR <≤0,)(SRS αβ . 4718

SRSβ is set to be compliant to the power control procedure. 4719

At the receiver, we will use a processing based on IDFT/DFT with a size denoted by M≥DFTM . 4720

To chose the DFT size, we take into account three constraints: 4721

First, the DFT size has to be compliant with the Xilinx implementation constraints: the IDFT/DFT 4722 size must be representable by 2^M*3^P*5^Q where M,P,Q are non-negative integers. This is 4723 because the Xilinx IDFT/DFT supports the PUSCH PRB allocations indicated in section 5.3.3 4724 Transform Precoding of 36.211. 4725

Secondly, the DFT size has to be a multiple of the number of cyclic shifts to ease the user 4726 separation in the time domain, i.e 8 4727

The third constraint is that we need to take a size that is bigger than M in order to use zero 4728 padding. 4729

4730

Therefore we chose 192=DFTM for 5MHz, 360=DFTM for 10MHz, 576=DFTM for 15MHz, 4731

and 600=DFTM for 20MHz. Current Working View is that we use 96=DFTM for 3MHz, 4732

48=DFTM for 1.4MHz. 4733

4734

Supprimé : 4.0.1

Supprimé : 2

Supprimé : , i.e

1151RSZC =N .¶

Supprimé : ,







11.2.4 CYCLIC SHIFT BASED USER INDEXING 4735

For SRS processing only, we redefine the user numbering based on the different cyclic shifts. This 4736 numbering is used for time domain filter construction only. 4737

The cyclic shifts at the transmitter are defined as follows: for cyclic shift =k/8, 8

2πα k= , k=0, …, 4738

7 4739

We first transform the cyclic shifts so that they are negative. This is done by subtracting π2 to 4740 them, which does not change the definitions of the reference signals. We obtain: for cyclic shift 4741

=k/8,

−= 18

2kπα , k=0, …, 7 4742

The users are then indexed in the increasing order of the absolute value of their cyclic shifts, i.e. 4743

user u has cyclic shift ( )8

2u

u πα −= 4744

For example, if all cyclic shifts are used, this gives the table below: 4745

4746

User number Cyclic shift values u=0 cyclic shift =0/8, 0=α u=7 cyclic shift =1/8,

8

72πα −=

u=6 cyclic shift =2/8, 8

62πα −=


52πα −=


42πα −=


32πα −=


22πα −=


12πα −=

4747

If less cyclic shifts than 8 are allocated, then the numbering is done only for those users currently 4748 transmitting SRS. 4749

11.2.5 TIME DOMAIN USER DISTRIBUTION 4750

We define a corresponding equivalent time domain delay 4751

( ) ( )88

222

DFTDFTDFT Mu

uMuMud ==−= π

ππα

. This delay corresponds to a time domain cyclic 4752

shift of the time domain CAZAC sequence 0r obtained from 0R by size DFTM IDFT. Actually, if 4753

( ) Mnnru <≤0, is the IDFT of the sequence ( ) )(nR uα padded with ( )M−DFTM zeros at the 4754

end, then we have: ( )

+=DFTM

DFTu

Munrnr

mod0 8

. 4755

Supprimé : 4.0.1

Supprimé : 2







As in LA2.0, let us denote ( )uτ the effective timing position of user u, giving the position of the 4756

center of gravity of the channel of user u. This is the result of the timing offset estimation of this 4757 user. If we are sufficiently confident on the reactivity of the timing advance loop, we can replace 4758

( )uτ by TAτ (which is assumed to be CP/2). 4759

We define the equivalent delay corresponding to the IDFT/DFT window length as: 4760

( ) ( )

≈

=

FFT

DFTTA

FFT

DFTDFT N

M

N

Mu .2.round.u2.round τττ 4761

Where round states for an approximation to the closest integer. The multiplication by two stands for 4762

the fact that we use only half of the tones, and the division by FFTN results from the fact that the 4763

timing offset ( )uτ is relative to the FFT window length whereas ( )uDFTτ is relative to the DFT 4764

window length. 4765

After compensation by mother CAZAC sequence and IDFT, the position of the center of gravity of 4766 the user u is located at the sample: 4767

( ) ( ) ( ) ( )

+=+−=

FFT

DFTDFTDFT

DFT

N

MMuu

uMud .u2.round

82ττ

πα

4768

In practice however, when a UE receives a timing advance command, it takes a while before the 4769 eNB is able to have an accurate estimate of the new time offset, so that during this time period an 4770 incorrect time offset might be used in the SRS algorithm. Therefore, it is better to use the targeted 4771

timing offset TAτ as described in section 6.2.2., leading to 4772

( )

+=

FFT

DFTTA

DFT

N

MMuud .2.round

8τ 4773

The second term in the sum is constant, common for all users and can thus be pre-computed. 4774

11.3. SRS SEPARATION 4775

11.3.1 ALGORITHM DESCRIPTION 4776

The user separation algorithm is similar to the MIMO user separation except that we have an 4777 additional zero padding. It consists in the following steps: FFT + sub-carrier de-mapping + CAZAC 4778 compensation + zero padding + IDFT + time domain filtering + circular shift + DFT + zero removal. 4779

This processing should be done for each comb independently. 4780

In the figure 8.1 below, we illustrate the global SRS processing for a given antenna (processing 4781 related to synchronization will be described in the following section). 4782

4783

4784

4785

4786

4787

4788

4789

4790

4791

Supprimé : 4.0.1

Supprimé : 2







4792

4793

4794

4795

4796 4797 4798 4799

Figure 11-11-1 SRS processing of all users for a given antenna 4800 4801

The CAZAC compensation is done with respect to the zero cyclic shifted sequence 0R . The 4802

content of the different blocks is detailed in the following sections. 4803

SRS

4

~H

SRS

u

~H

SRS

0~H

SRS

4H SRSˆuH SRS

0H

SRSH

mx

SRSX

( )mx0

FFT


CP removal

Conjugate mother CAZAC=( )∗0R

Zeros padding + IDFT

Time domain filtering for

UE 0


UE u + circular shift


UE 4 + circular shift

K K

DFT + zeros removal

Per PRB power

estimation

CFO de-rotation (only if MIMO)

DFT + zeros removal

DFT + zeros removal

Per PRB power

estimation


Per PRB power

estimation


SRSh

SRSh0ˆ SRS

uh SRS

h4ˆ

Time domain noise removal



Post DFT AGC

Post DFT AGC

Post DFT AGC

Supprimé : 4.0.1

Supprimé : 2







11.3.2 CAZAC COMPENSATION 4804

This function is called for each antenna and only for the SRS symbol (i.e. last symbol of the 4805 subframe) after CP removal, FFT and sub-carrier demapping. This consists in a term by term 4806 multiplication of the received signal by the complex conjugate of the mother CAZAC sequence 4807 elements. 4808 4809 The interface signals and their specifications for this block are given in the next table below. 4810


M Integer uX(see below)

I 1 Number of SRS subcarriers

CAZAC Complex Fractional

(12I,12Q) I M SRS Reference Sequence

X_PerAnt Complex Integer

(18I,18Q) I M SRS symbol per antenna


(18I,18Q) O M Compensated SRS symbol per antenna

4811 Table 11-1: Interface definition for SRS CAZAC comp ensation 4812

4813

X=8, 9, 9, 10 for 5, 10, 15 and 20MHz. 4814

The fixed point implementation is described below. 4815

4816

4817

4818

4819

4820

4821

4822

4823

4824

4825

4826

4827

11.3.3 ZERO PADDING 4828

The aim of zero padding is only to enhance the channel estimation at the band edges. 4829

Actually, time domain filtering is equivalent to a frequency domain circular convolution that 4830

degrades the estimation accuracy at the edges because of the discontinuity. Without zero padding, 4831

the Gibbs phenomenon will have greater impact than with zero padding. 4832

(18I,18Q)

(12I,12Q)

>>11 Sat 18 18

29

30

19

-

Supprimé : 4.0.1

Supprimé : 2







This IDFT/DFT size will be denoted as MM DFT > . Zero padding consists in adding 4833

( )MM DFT − zeros at the end of the sequence before IDFT. 4834

Potentially, we may need to modify the zero-padding function by another form of padding in order 4835

to reduce the degradation of the estimations on the edge of the band. This is FFS for the time 4836 being. 4837



M Integer

uX (X=8, 9, 9, 10

for 5/10/15/2

0MHz)


DFTM Integer

uX (X=8, 9, 10, 10

for 5/10/15/2

0MHz)

I 1 DFT size


(18I,18Q) I M Compensated SRS symbol per antenna

HDFT_PerAnt Complex Integer

(18I,18Q) O DFTM IDFT input

4839 Table 11-2: Interface definition for Zero Padding 4840

4841

11.3.4 IDFT 4842

The IDFT is performed in the same way as in section 6.5.2, i.e.: 4843

( ) ( )∑−

=

=

1

0

2expˆˆDFTM

i DFT

SRSSRS

M

ikjiHkh π 4844

Notice that there is no half length cyclic shifting of the resulting sequence as this is the case for the 4845 front-end FFT. Fixed point implementation is the same as in section 14.3.3. Notice that on the 4846

contrary of section 6.5.2 and 14.3.3, there is no left shift (<<p) before the IDFT since the input is 4847

already on 18 bits. 4848


4850


DFTM Integer

uX (X=8, 9, 10, 10

for 5/10/15/2

0MHz)

I 1 DFT size

Supprimé : 4.0.1

Supprimé : 2








(18I,18Q) I/O MDFT samples (time domain)


(18I,18Q) I/O MDFT samples (freq domain)

Table 11-3 : SRS DFT interface definition 4851

4852


11.3.5.1 REDUCED FILTER LENGTH 4854

4855

For LA3.0, time domain filtering construction are updated with respect to LA2.0. Time domain filters 4856 construction is similar to LA2.0, but with a modification of the filters lengths. In LA2.0 construction, 4857 whatever the number of users simultaneously transmitting SRS, the different filters span all the 4858 time domain axis. This resulted in the fact that for a small number of simultaneous SRS 4859 transmissions (e.g. only 2 users), the filters were not limited enough in time so that many noise 4860 samples were collected before DFT. This is avoided here by strictly limiting the filters to the portion 4861 of the time where signal samples are expected. To do so, we simply do as if all the cyclic shifts 4862 were allocated, i.e. we construct 8 time domain filters instead of U filters. These 8 filters are 4863 constructed using LA2.0 time domain filter construction. 4864

To highlight the updated construction, the filters on a given comb will be indexed by w=0… 7, with 4865

filter w corresponding to a cyclic shift ( )8

2w

w πα −= . This means that the indexes w are not 4866

associated to transmitting users but to cyclic shifts (occupied or not). 4867

For example, if we have two users with cyclic shifts 8

22π and

8

52π , this corresponds to cyclic 4868

shift indexes w=6 and w=3 respectively, and we consider the construction of all filters from w=0 to 4869 w=7 before taking the corresponding useful filters for both users. 4870

In this construction, for those cyclic shift not allocated to users (different from 3 and 6 in the 4871

example above), we use the value TAτ instead of the value DFTτ . For the allocated users, we can 4872

use both. 4873

The time domain filter construction is described in the figures below. 4874

Supprimé : 4.0.1

Supprimé : 2







4875

Figure 11-2 Construction of the time domain filter of users (w-1) in red, w in black, and w+1 in blue. 4876

4877

Figure 11-3 Time domain filter for w=0. 4878

In the case where we use the value TAτ instead of the value DFTτ , the construction becomes much simpler 4879

since, as ( )

+=

FFT

DFTTA

DFT

N

MMwwd .2.round

8τ , we get 4880

( )162

)1()( DFTMwdwd =−− so that we finally get the filters of the following figure: 4881

1

0 )2( −wd )(wd )1( +wd )2( +wd )1( −wd

( )2

)1()()(

−−− wdwdwd

( )2

)()1()(

wdwdwd

−++

t

uL

( ) ( ) ( )( )2

1707

−−++ ddMd DFT ( )

2

)0(1)0(

ddd

−+

1

0 1−DFTM ( )1d ( )7d ( )0d

t

Supprimé : 4.0.1

Supprimé : 2







4882

11.3.5.2 MARGIN FOR TIMING OFFSET ERROR 4883

In order to cope with timing offset errors, an extra margin on each side of the filter will be added, 4884 corresponding to one TA command granularity (i.e. 16 samples for 20MHz). 4885

4886 This filter extension in a given direction (left or right) will be added only if the neighbour cyclic shift 4887 is empty, otherwise the filter remains unchanged. Notice that in LA3.0, since we have a maximum 4888 of 4 UE per comb, this extension will always be performed. 4889 4890 4891 4892 4893 4894 4895 4896 4897 4898 4899 4900 4901 4902 4903 4904 4905 4906 4907 4908 4909 4910 4911 4912 4913

11.3.6 TIME DOMAIN NOISE REMOVAL 4914

4915

16)( DFTM

wd −

1

0 )2( −wd )(wd )1( +wd )2( +wd )1( −wd

16)( DFTM

wd +

t

8DFT

u

ML =

1

0 )(wd )1( +wd )1( −wd

t

8DFT

u

ML =

Nominal filter

Extended filter

16 samples 20MHz extension if CS on the right is empty

16 samples 20MHz extension if CS on the left is empty

Supprimé : 4.0.1

Supprimé : 2







The reduction of the time domain filter window length permits to remove noise outside the selected 4916 window. However, there remains some noise inside this user’s window that has to be removed. 4917 This is done by a threshold comparison, meaning that only those samples emerging from noise will 4918 be kept. 4919 4920 The noise power before IDFT in the SRS symbol is assumed to be equal to the noise estimates 4921 coming from both DMRS of the sub-frame containing the SRS. The corresponding post-IDFT noise 4922 power is computed using the IDFT which writes for noise components 4923 4924

( ) ( ) ( )∑∑−

=

−

=

=

=

1

0

1

0

2exp,2exp,,M

i DFT

M

i DFTIDFT M

ikjain

M

ikjainakn

DFT

ππ , k=0, …, 1−DFTM 4925

4926 Where a denotes the antenna index. The last equality comes from zero padding. Since the pre-4927 IDFT noise is assumed to be white, we get: 4928 4929

( ) ( ){ } ( ){ }∑−

=Ε=Ε=

1

0

222 ,,ˆM

iIDFTIDFT ainaknaσ 4930

4931 Since we have 6 samples per PRB, the time domain noise power is equal to 6 times the sum of the 4932 noise estimates over the PRBs containing SRS: 4933 4934

( ) ( ){ } ( )∑−+

=

=Ε=1

62

,

22 ˆ6,ˆ

Mn

nPRBaPRBIDFTIDFT

b

b

akna σσ where ( )2,ˆ aPRBσ is the noise estimate per PRB per 4935

antenna coming from the DMRS. If some scaling is performed on the IDFT, it has to be taken into 4936 account in the noise power (this is not the case with the Xilinx IDFT). 4937

Notice that the noise estimate ( )2,ˆ aPRBσ can be associated to any allocation between SIMO, MIMO 4938

or empty PRB. 4939 From an implementation point of view, see fixed point implementation, the per antenna noise 4940 variance has to be normalized by the agc_shift (per user and unallocated PRBs) associated with 4941 the SRS PRBs before summation. 4942

Denoting thK the selected threshold, we keep before DFT only these samples with power greater 4943

than ( )aK IDFTth2ˆ.σ , i.e for index i varying inside the user’s window, if ( ) ( )aKaih IDFTth

SRS

u2

2

ˆ.,ˆ σ≤ , 4944

then set ( ) 0,ˆ =aihSRS

u . We shall use the value 4=thK as the current working view. 4945

Notice that this noise removal does not remove noise samples that are hidden in the signal 4946 samples so that noise removal in the frequency domain for signal power cancellation remains 4947 necessary. 4948

In the following, we will denote as ( )aLuth the number of samples kept before DFT, i.e. the number 4949

of antenna a samples satisfying ( ) ( )aKaih IDFTth

SRS

u2

2

ˆ.,ˆ σ> . 4950

Notice that since the parameter ( )aLuth relies on the comparison between per user per antenna 4951

values, it is also per user per antenna. 4952 4953 The fixed point implementation of the post-IDFT noise estimation is given below. Notice that the 4954 value of p below is equal to zero for drop 2 since the input of the IDFT is already on 18bits. 4955 4956 In order to find the right scale for the noise to be removed, we recall below the different scalings 4957 during the IDFT/DFT processing: 4958 4959

Supprimé : 4.0.1

Supprimé : 2







4960 4961 Therefore, the noise to be subtracted has to be compliant to the scaling above before subtraction. 4962 4963 Moreover, for the noise estimates coming from DMRS, there is no division by 2 to take into account 4964 the two pilot blocks so that the noise has to first be divided by 2 before time domain noise removal. 4965 Notice that with respect to drop 1, there is no need to compensate by the post-FFT AGC SRS since 4966 it is equal to 0. 4967 4968 We need to: 4969

• Perform >>1 4970 • Invert the per PRB PUSCH AGC scaling 4971 • Apply the SRS AGC scaling (scaling is 0 for drop 2) 4972 • Apply the internal Xilinx IDFT scaling 4973 4974

These operations are described in the figure below (this is only a conceptual description to show 4975 the different scalings, actual fixed point implementation is given after). 4976

4977 Special case : for a given antenna, in the case where all the channel samples are such that 4978

( ) ( )aKaih IDFTth

SRS

u2

2

ˆ.,ˆ σ≤ , then applying the time domain noise removal lead to a channel 4979

estimate equal to zero. Therefore, some defense mechanism shall be applied when performing the 4980 SRS SNR to process correctly the zero values. This can be done e.g. by adding 1 to the values of 4981 the power before SNR computation, or give a special ouput for the 0 input. 4982 4983 Notice that this is a degenerate case that is not expected to happen in practice, and only at very 4984 low SNR. 4985 The actual fixed point implementation is given below for drop 2. 4986 4987 4988

( )2,ˆ aPRBσ ( ) )1 2( −<< PRBe

M

M

All PRBs

( )sfp −<< 2

( )aIDFT2σ

( )SRSe 2>>

left shift

<< 0

iDFT xilinx

DFT xilinx

rescalin

g

1.0 1.0 2-sf M.2-sf-si 1.0 scale :

Supprimé : 4.0.1

Supprimé : 2







4989 Fixed point implementation of post-IDFT noise estim ation: drop 2 4990

4991

The fixed point implementation of the comparison the ( ) ( )aKaih IDFTth

SRS

u2

2

ˆ.,ˆ σ≤ is described 4992

below: 4993 4994

4995 We give below the values of ( ) 6log 2 M for the different bands: 4996

4997

For 1.4MHz, M=24 and ( ) 6log 2 M = 2 4998

For 3MHz, M=72 and ( ) 6log 2 M = 4 4999

For 5MHz, M=144 and ( ) 6log 2 M = 5 5000

For 10MHz, M=288 and ( ) 6log 2 M = 6 5001

For 15MHz, M=432 and ( ) 6log 2 M = 7 5002

For 20MHz, M=576 and ( ) 6log 2 M = 7 5003

5004 5005 5006

( )2

,ˆ aihSRS

u

36

( ) 6log36 2 M−

( ) 6log 2 M>> To comparison

( )2,ˆ aPRBσ ( ) )1 2( −<< PRBe

M

M24

37

( ) 6log37 2 M+

Sat

( ) 6log36 2 M−

( )sfM ×+>> 2 )6/(log2

To comparison

sf×− 242

6× thK

( ) 6log42 2 M+

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Retrait :Gauche : 1,25 cm

Supprimé : ¶







11.3.7 TIME DOMAIN CYCLIC SHIFT 5007

This function is the same as in LA2.0. After time domain filtering, we have to remove the time 5008 domain delay of the different users by shifting them to the left. Notice that we have to remove only 5009 the shift caused by CAZAC sequence cyclic shift and not the shift caused by their timing offset. 5010

Therefore, we have to cyclically shift the user u by ( )

82DFTM

uuM =−

πα

samples on the left. 5011

Notice that this shift is not performed by user 0 since it was compensated with its own sequence. 5012

In the figure below, we describe the cyclic shift of the non-zero coefficients after time domain 5013 filtering for the user u. 5014

5015

Figure 11-4 : Time domain cyclic shift of user u before DFT 5016 5017


5019


DFTM Integer

uX (X=8, 9, 10, 10

for 5/10/15/2

0MHz)

I 1 DFT size

hDFT_PerAnt Complex Integer

(18I,18Q) I MDFT samples per antenna (time

domain)

hDFT_PerUe Complex Integer

(18I,18Q) O MDFT samples per Ue per antenna

(time domain)

Table 11-4 : Time domain cyclic shift interface def inition 5020 5021

1

0 M-1

Cyclic shift by 8DFTM

u samples

( )ud

Position of the samples after time domain cyclic shit and before DFT

t

Supprimé : 4.0.1

Supprimé : 2







11.3.8 IDFT/DFT SCALING 5022

From LA3.0 drops 2 and beyond, the IDFT/DFT scaling differs from the MIMO scaling of section 5023 13. The DFT is the same as in section 14.3.3 using the Xilinx engine. After DFT, we have to rescale the 5024 signal to take into account the IDFT and DFT internal scaling factors, and the required normalization factor 5025

DFTM/1 . 5026

We recall below the different scalings during the IDFT/DFT processing: 5027 5028

5029 5030 The rescaling to be applied on each sample’s real and imaginary parts is therefore [x (2sf+si / MDFT)]. 5031 This can be simplified to a multiplication followed by a right shifit [x q / 2r] with q and r defined as: 5032 5033

DFT

Mfloorq

Mq

DFTs ))((log22 +

= 5034

)( if sshr +−= , ))((log2 DFTs Mfloorqh += , 5035

qs=14 5036

5037

Transmission BW MDFT q r

1.4MHz 48 10923 19– (sf + si)

3MHz 96 10923 20– (sf + si)

5MHz 192 10923 21 – (sf + si) 10MHz 360 11651 22 – (sf + si) 15MHz 576 14564 23 – (sf + si) 20MHz 600 13981 23– (sf + si)

5038 5039

5040 5041

11.3.9 ZERO REMOVAL 5042

After DFT, we remove the ( )MM DFT − samples at the end of the sequence to go back to a 5043

sequence of length M. 5044


iDFT xilinx

DFT xilinx

rescalin

g

1.0 2-sf MDFT.2-sf-si 1.0 scale :

18

>>r

18

q

u14

32

Sat18

Supprimé : 4.0.1

Supprimé : 2







5046


DFTM Integer

uX (X=8, 9, 10, 10

for 5/10/15/2

0MHz)

I 1 DFT size

M Integer

uX (X=8, 9, 9, 10

for 5/10/15/2

0MHz)


HDFT_PerUe Complex Integer

(18I,18Q) I/O MDFT samples per antenna (frequency domain)

Table 11-5 : Zero removal interface definition 5047 5048

11.4. POST-DFT AGC 5049

This feature is added for drop 2 to reduce the number of bits at the DFT output from 18 to 12. One 5050 AGC shift per UE is applied, this shift is common for all antennas. 5051

For a given UE, we determine the upper position of the MSB for the DFT outputs of all Rx antennas 5052

and keep twelve bits starting from this determined MSB. The resulting shift per UE should be sent 5053 to the DSP as an SRS AGC shift value on a per UE basis. 5054

Let “order” denotes the function computing the number of bits representing the sign before the first 5055

significative bits. 5056

Let ( )iH au

SRS

,ˆ denote the value (on 18bits) of the DFT output, for tone i, user u, antenna a. . Then 5057

we compute: 5058

( ) ( )

( ) ( )

−=

−=

iHiauSi

iHiauSr

au

au

SRS

,

SRS

,

ˆImorder18,,

ˆReorder18,, 5059

The AGC shift for user u is then computed as: 5060

( )( ) ( )( ) 11,,max,,,maxmax)(,,

−

= iauSiiauSruAGCshift

iaia 5061

If the result above is negative, then the AGC shift is equal to zero. 5062

Notice that the method above does not describe the way the shift computation should be done, its 5063

intent is only to give a mathematical description. Any equivalent method that is suitable to the 5064 designer can be used to compute the upper MSB position. 5065


As for MIMO, the synchronization is done after separation. Actually, the CFO introduces a time 5067 domain multiplication by a complex exponential and do not have any influence on separation since 5068

Supprimé : 4.0.1

Supprimé : 2







the users are separated in the time domain. Notice also that when only the power of the reference 5069 signal is needed (i.e. when we are not in a MIMO configuration) then the CFO de-rotation which 5070 only consists in this case of a complex multiplication by a complex exponential can be omitted. 5071

Furthermore, even in the case where ICI is needed, we do not have to perform it for SIMO users 5072 since the ICI matrix is orthogonal and do not change the energy of the user. 5073

In a MIMO configuration, we have to provide the scheduler with the value of the channels and not 5074 only its power so that this CFO de-rotation has to be carried out. 5075

Since MIMO users are supposed to be low speed users, we can make the assumption that ICI is 5076 not needed for these users. 5077

Therefore, the CFO de-rotation is done for MIMO users only the same way as for data blocks (see 5078 section 6.2.1.2) on a user by user basis without frequency domain filtering. This consists in a 5079

multiplication of the user’s signal by the complex exponential ( ) ( )

Ι−

FFT

ˆ..2exp Numj επ as 5080

defined in 6.2.1.2. 5081

Notice that from an implementation point of view, it is simpler that the metrics corresponding to 5082 MIMO users are computed for all users, so this is what is done. 5083


5085


M Integer

uX (X=8, 9, 9, 10

for 5/10/15/2

0MHz)


CFOexp Complex Integer

(12I,12Q) I M CFO de-rotation complex

exponential signal (frequency domain)


(12I,12Q) I/O M SRS Ue samples per antenna

(frequency domain)

Table 11-6 : Synchronization interface definition 5086 5087

11.6. SNR ESTIMATION 5088

As for demodulation reference signal processing, we do not compute the actual ratio but provide 5089 instead the scheduler with the instantaneous values of the noise and the useful power per PRB and 5090 per antenna for all users. 5091

We give below a description of the fixed point implementation of the SRS SNR computation. 5092 However, it is not to be performed inside the FPGA, but only in the DSP for UL scheduling purpose. 5093

11.6.1 NOISE POWER ESTIMATION 5094

As for MIMO, cf. section 14.6, the time domain filtering to separate the users leads to a non-white 5095

Gaussian noise so that noise power estimation using the same algorithm as for DMRS leads to an 5096 over-estimated noise power. We could use the same algorithm as for MIMO applied to the 8 5097 different users with the corresponding filter lengths. However, it is simpler to use the noise 5098

estimated from both DMRS present in the same sub-frame as the SRS. Since no Code Domain 5099

Supprimé : 4.0.1

Supprimé : 2







Multiplexing should be involved (except for MIMO) the corresponding noise estimates should be 5100 more reliable. For non-allocated PRBs, we can simply compute the average of the squared 5101

samples. 5102

As described in section 6.3.2, the noise estimation for non-allocated PRBs is performed on all TTIs 5103 by averaging the squared samples on the PRB of interest. 5104

11.6.2 SIGNAL POWER ESTIMATION 5105

The signal power estimation on a given PRB can be computed by subtracting the noise power to 5106 the mean of the squared samples of the SRS channel estimate on that PRB. Notice however that 5107 the time domain filtering and the time domain noise removal operations remove a part of the noise 5108 and that should be taken into account in the noise subtraction. Therefore, we have to weight the 5109 noise power by a scaling factor taking into account the different operations done on noise. 5110

Since before DFT only thL non-zero samples are present, and taking into account the definition of 5111

the used DFT which is the scaled version ( ) ( )∑−

=

−=1

0

2exp,ˆ1,ˆ

DFTM

k

SRSu

DFT

SRSu M

ikjakh

MaiH π , 5112

The mathematical expression of the post-DFT noise power (assuming no time domain noise 5113 removal is applied) is equal to (the sum below is over the tones, and the noise variances on the 5114 right sides are the pre-IDFT noise variance as given by the PUSCH noise estimation):. 5115

( ) ( )( ) ( ) ( ) ( )

( )

( )∑

≠

−

−

+=ij

DFT

DFT

uth

aj

DFT

ai

DFT

uth

DFT

M

ij

M

Lij

MM

Liau 2

2

2

,2

2

,2

2

2

sin

sin

ˆ1

ˆ,,ˆπ

π

σσσ 5116

In practice, the expression above is too complex to be performed and we use instead the 5117 approximation: 5118

( ) ( )2,2 ˆ,,ˆ aPRB

DFT

uth

DFT M

LPRBau σσ ≈ 5119

The corresponding power estimation is slightly biased (noise is slightly underestimated) but 5120 simulations show that the residual bias is negligible. 5121

5122

Notice that post-DFT noise variance is per user per antenna and not just per antenna since uthL is 5123

per user per antenna. 5124

We can then perform noise sub-traction by: 5125

( ) ( )auHaE DFTPRBi

aiu ,ˆ~

6

1 22SRS,,u PRB, σ−

= ∑∈

5126

The sum above contains only 6 terms since only every other tone is allocated. 5127

Moreover, once the signal has been unbiased by noise removal, we have to rescale the signal so 5128 that it is at the same scale as the PUSCH signal. Appropriate scaling should be done at L2 level 5129

since SRS are used only for L2 processing (to take SRSβ into account). 5130

Notice that the output of the SRS sent as input of the scheduler are the values of the channel 5131

powers ( )aE u PRB, of all users over all PRBs and antennas. Notice also that on the contrary of 5132

Supprimé : 4.0.1

Supprimé : 2







DMRS processing, SRS provides one energy estimate per PRB and per user, while DMRS 5133 provides the energy estimate only for the user to which the PRB is allocated to. 5134


5136


NoisePRB Integer u24 I nPBR Noise estimation per PRB per antenna


(12I,12Q) I M SRS samples per antenna (frequency domain)

SRSPwrPRB Integer u24 O nPBR SRS power per PRB and per

antenna

DFTM Integer

uX (X=8, 9, 10, 10

for 5/10/15/2

0MHz)

I 1 DFT size

L Integer u10 I 1 Time domain filter length

Table 11-7 : Signal power interface definition 5137 5138

The fixed point implementation to compute the measure ∑∈PRBi

aiuH2SRS

,,

~ sent to the DSP is described 5139

below. 5140

5141

The fixed point implementation of SRS debiasing is given below: 5142

5143

(12I,12Q) |.|2

|.|2

>>2

12

22

26

24

Sat 24

Supprimé : 4.0.1

Supprimé : 2







5144

The actual implementation of the multiplication by Lu/MDTF is detailed in the scheme below: 5145 5146 5147 5148

5149 5150 5151 The coefficient Inv_DFT corresponds to the inverse of the DFT size coded over 31bits. The values 5152 for the different bandwidths are given below: 5153 5154

<<(2.E(PUSCH))

Lu

ST Noise

Inv_DFT

u16

>>16

>>18

To subtraction

u31

( )∑=

5

0

2

iSRS iH

<<(2.E(SRS))

<<(2.E(PUSCH))

5461

>>15

Lu/MDTF

ST Noise >>3

Unbiased SRS Energy

-

+

Supprimé : 4.0.1

Supprimé : 2







Transmission BW MDFT Inv_DFT

1.4MHz 48 44739243

3MHz 96 22369621

5MHz 192 11184811 10MHz 360 5965232 15MHz 576 3728270 20MHz 600 3579139

5155 5156

The computation of the SRS SINR for UL scheduler is done as follows for eCEM: 5157

5158

5159

Notice that with respect to LA3.0 drop 1 implementation, we removed the shift (>>1) applied to the 5160 long term noise. This is because for drop 2, all long term measures include an initial (>>2) instead 5161 of (>>1) so that the noise is already at the right scale compared to SRS power. 5162

Notice also that this is valid for eCEM only. 5163

11.7. SRS ABSOLUTE POWER 5164

In this section we describe the way to compute the SRS absolute power expressed in dBm based 5165 on the L1 SNR outputs. 5166

The first step is to map the samples received by the L1 to powers in dBm. This is done as per 5167 section 5.2 by using the UL RF antenna gain and the digital gain. 5168 5169 On top of that, we need to include the FFT gain and the channel element gain CE_Gain, i.e. the 5170 scaling introduced by the L1 receiver along the processing. For SRS, after FFT, this includes the 5171 post-FFT shift and the final shift of >>2. Therefore, the conversion formula to obtain the SRS power 5172 in dBm write: 5173 5174

SRS_Power_dBm = 10log10 (SRS_Power) - 10*log10(Imax²) – (UL RF receiver gain) – (UL Digital 5175 gain) - 10*log10(FFT_Gain) - 10*log10(CE_Gain) 5176 5177

LT Noise

SRS SNR to ULS

Unbiased SRS Energy

••

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Gauche

Mis en forme : Gauche







The FFT gain is equal to the FFT size since this is an unscaled FFT. 5178

The channel element gain CE_Gain is equal to 4

2 Shift FFTPost−

with 1 Shift FFTPost =− i.e. 5179

CE_Gain = 0.5 5180

As for section 5.2 we have Imax² = 2^28 5181

5182

The different numerical values that are to be used in this equation are thus given below: 5183

10*log10(Imax²) = 10*log10(2^28) = 84.28 5184

10*log10(CE_Gain) = - 3.01 5185

10*log10(FFT_Gain) = 21.07 / 24.08 / 27.09 / 30.10 / 31.86 / 33.11 respectively for 1.4 / 3 / 5 / 5186 10 / 15 / 20 MHz 5187

5188

Notice that in the formula above, when considering the value given by the DSP, we should not pay 5189 attention to the effective format in QX.Y of the power estimates but only consider the integer value. 5190 Stated differently, in the formula above should be considered as coded in QX.0 format. 5191 5192

11.8. SRS CHANNEL AMPLITUDE ESTIMATION 5193

In LA1.0specifications, the amplitude was to be computed for MIMO users only. However, when 5194 processing the SRS, the FPGA does not necessarily have this information of MIMO or SIMO users. 5195 Therefore, the processing below is expected to be done for every user. 5196

We provide the scheduler with one channel coefficient per antenna, per PRB and per user. To 5197

compute this channel coefficient, we simply average the values of SRS

u

~H over the PRB of interest to 5198

compute

∑∈PRBi

aiuH SRS,,

~

6

1. The users indexes for which these channel coefficients have to be 5199

provided are given by the scheduler. Notice that the 1/6 scaling is done in the DSP so that only the 5200 summation is done inside the FPGA. 5201

As for signal power, we have to rescale the signal so that it is at the same scale as the PUSCH 5202 signal. Appropriate scaling should be done at L2 level since SRS are used only for L2 processing 5203 (to take βsrs into account). Next plot gives the fixed point implementation: 5204

5205

If we want to have SRS,,

~aiuH at the same scale as the SRS SNR given to the Uplink Scheduler, we have to 5206

also remove the SRS AGC shift as done above for the energy. This is done by a shift on the left by the 5207 SRS AGC shift. 5208

SRS,,

~aiuH

(12I, 12Q)

M

M

5461

>>15

∑∈PRBi

aiuH SRS,,

~

6

1

Supprimé : 4.0.1

Supprimé : 2







5209

11.9. SRS FREQUENCY DOMAIN CORRELATION 5210

This feature is new compared to LA2.0. In LA3.0, a frequency domain correlation is performed on a per 5211 user per antenna basis and transmitted to the DSP for timing offset estimation purpose. We describe in this 5212 section only the correlation computation done inside the FPGA C. 5213

The input of this block is the channel estimate post CFO derotation, i.e. SRS

u

~H with the notations above. 5214

5215 Since this is known that the performance of SRS is worse on the edges, a pre-defined (and fixed) number 5216 of tones will be skipped on each side of the band when computing the frequency domain correlations. This 5217

number will be denoted as SRSSkN . The current working view is that 6=SRS

SkN but this value can be 5218

updated in the future. 5219 The min and max indexes used for frequency domain correlations are thus, assuming index starts at 0 and 5220 recalling that M is the total number of SRS tones: 5221 5222

−−−=

=

12max

min

sNMI

NI

SRSSk

SRSSk

5223

The value of s is the same as for PUSCH and is defined in section 6.2.2 5224 5225 The correlation is computed as below for antenna a: 5226 5227

( ) ∑=

∗

+

=

max

min,

2,

,,

~~,,

I

Ii

SRS

as

iu

SRSaiu

SRST HHausC 5228

5229 On the contrary of PUSCH, we propose to avoid computing the correlation on a PRB basis due to the 5230 following reasons: 5231

• The number of tones available for correlation computation is increased if the average is done 5232 across PRBs. This means that the correlation is more precise, and this is highly needed for low 5233 power users for SRS 5234

• The SRS assignement is static, therefore we do not need to parameterize the PRB used for each 5235 user 5236

• This reduces the amount of data transmitted to the DSP. 5237 5238

For the first reason above, we do not average over the number of tones and receive antennas but compute 5239 only the summation. 5240 5241 Notice that since we have a comb allocation, the intertone distance in the correlation above is twice smaller 5242 than for PUSCH. 5243 5244 5245 Fixed point implementation 5246 5247 5248 We give below the fixed point implementation for a given antenna (only real part processing is described, 5249

the same holds for imaginary part). Notice that the correlations ( )ausCSRST ,, are computed with full 5250

∑∈PRBi

aiuH SRS,,

~

6

1<<e(SRS) SRS MIMO channel metric to ULS

Supprimé : 4.0.1

Supprimé : 2







precision, meaning no right shift and authorization of bitwidth expansion. However, we have to ensure that 5251 the correlations do not go beyond 32bits so that a final shift of 2 bits will be performed. 5252 5253 5254

5255 5256

The foreseen value for SRSSkN is 6, so that we have 5257

5258

−−2

2log2

sNM SRS

Sk =4bits for 1.4MHz, 6bits for 3MHz, 8bits for 5 and 10MHz, 9bits for 10MHz and 5259

15MHz, 10bits for 20MHz 5260 5261

12. SOUNDING REFERENCE SIGNALS: BCEM 5262

IMPLEMENTATION 5263

12.1. INTRODUCTION 5264

This section corresponds to the bCEM SRS implementation. All considerations common to eCEM and 5265

bCEM are not re-described here and can be found in chapter 11. 5266

5267

12.2. BCEM VS ECEM IMPLEMENTATIONS 5268

In bCEM implementation, the SRS processing is done in DSP. More precisely, the FPGA BC extracts the 5269 SRS tones after front-end processing (CP removal, front-end FFT, sub-carrier demapping and pre-AGC 5270

processing) and sends the samples to the DSP cores. Before transmission from FPGA to the DSP, a 5271 saturation to (16I, 16Q) shall be done by the FPGA (instead of 18 for FPGA processing). The only effect of 5272

this sat 16 at the input of the DSP is a reduction in the dynamic range we can support for SRS processing 5273

but the resulting maximum supported SNR is highly sufficient for normal operation. 5274

12.3. CAZAC COMPENSATION 5275

The values of the CAZAC sequences can be fully precomputed and stored in tables of constants. 5276 Compared to eCEM implementation, the precision of those values shall be extended to Q(1.15) format. 5277

(12I,12Q)

(12I,12Q)

−−2

2s

NM SRSSk tones

23

24

DSP

1st antenna *

24+

−−2

2log2

sNM SRS

Sk

>>2 32

Supprimé : 4.0.1

Supprimé : 2







The new interface is given in the table below. 5278 5279


M Integer uX(see below)


CAZAC Complex Fractional

(16I,16Q) I M SRS Reference Sequence

X_PerAnt Complex Integer

(16I,16Q) I M SRS symbol per antenna


(16I,16Q) O M Compensated SRS symbol per antenna

5280 Table 12-1: Interface definition for SRS CAZAC comp ensation 5281

5282

X=8, 9, 9, 10 for 5, 10, 15 and 20MHz. 5283

The bCEM fixed point implementation is described below. 5284

5285

5286

5287

5288

5289

5290

5291

5292

5293

5294

5295

12.4. ZERO PADDING 5296

This is exactly the same as for eCEM, see 11.3.3. 5297

12.5. IDFT 5298

The IDFT is performed inside the DFTPE block of the MAPLE. The input and the output of the IDFT 5299

are in the format (16I, 16Q). Notice that as for the Xilinx implementation in eCEM, the MAPLE IDFT 5300 has an internal scaling. For each iDFT, the internal scaling factor (“sf”) applied by the MAPLE can 5301

be fetched. Notice that as for the Xilinx engine, the IDFT is not normalized by the IDFT size. 5302

5303

XSRS (16I,16Q)

Complex conjugate of the mother CAZAC

(16I,16Q) (stored on 2 x 16

bits)

>>15 Sat16 -

s31 (stored on 32 bits)

s32 (stored on 32

bits)

Right bit shift with rounding.

HSRS s16

(stored on 16 bits)

Supprimé : 4.0.1

Supprimé : 2







12.6. TIME DOMAIN FILTERING 5304

Time domain filters definitions are the same as for eCEM implementation. However, there are 5305

some DSP specific constraints to take into account which lead some slight modifications on the 5306 filters edges. 5307

We denote the sample indexes of the beginning and end of the filter by respectively (beginIndex) 5308

and (endIndex) for a given cyclic shift. These values can be fully precomputed (for all cyclic shifts) 5309 and put in tables of constants so that we only have to read the two values beginIndex and 5310

endIndex at that stage for the concerned UE. 5311

There are 32 possible values for d(u) (see chapter 11 for definition). 5312

8 (8 cyclic shifts) 5313

x 4 ( 4 MDFT values: 192 (5MHz), 360 (10MHz), 576 (15MHz) or 600 (20MHz) 5314

+ 4 corresponding NFFT values: 512 (5MHz), 1024 (10MHz), 1536 (15MHz) or 2048 (20MHz) 5315

+ 4 corresponding τTA values in number of samples: 18 samples (5MHz), 36 samples (10MHz), 54 5316

samples (15MHz) or 72 samples (20MHz) ) 5317

= 32 d(u) values . 5318

The τTA values (CP/2) in number of samples (cf. the possible values above) allow to compute τDFT = 5319

round(2.τTA.MDFT/NFFT). 5320

The d(u) values are already known in the DSP today (existing table with all d(u) values to be 5321 reused). 5322

- For cyclic shift u (u = 0..7), beginIndex and endIndex can be precomputed as follows: 5323

Step-1): 5324

- beginIndex is (MDFT/16 + offseterror) indexes before the index d(u) by considering a circular table, 5325

i.e.: 5326

beginIndex = (MDFT + d(u) - MDFT/16 - offseterror) % MDFT, 5327

- endIndex is (MDFT/16 + offseterror) indexes after the index d(u) by considering a circular table, i.e.: 5328

endIndex = (d(u) + MDFT/16 + offseterror) % MDFT 5329

with offseterror = 4 samples (5MHz), 8 samples (10MHz), 12 samples (15MHz) or 16 samples 5330 (20MHz). 5331

Step-2): 5332

To fulfill DSP-intrinsic constraints (8-byte address alignment), to simplify the implementation and to 5333 allow better computing performances, the above computed values of beginIndex and endIndex 5334

shall then be modified as follows: 5335

- if beginIndex is even and endIndex is even: endIndex = (endIndex + 1) % MDFT, 5336

- if beginIndex is odd and endIndex is odd: beginIndex = (MDFT + beginIndex - 1) % MDFT, 5337

- if beginIndex is even and endIndex is odd: no modification needs to be brought, 5338

- if beginIndex is odd and endIndex is even (this case does not happen with current values 5339 – quoted just for completeness reason): beginIndex = (MDFT + beginIndex - 1) % MDFT and 5340

endIndex = (MDFT + endIndex - 1) % MDFT. 5341

Supprimé : 4.0.1

Supprimé : 2







With those modifications, beginIndex will always be even and endIndex will always be odd, so the 5342 number of SRS samples to handle will always be even, fulfilling the above implementation 5343

constraints/goals. 5344

As consecutive cyclic shifts are not allocated, offseterror is always > 0. 5345

Note: in future releases, if consecutive cyclic shifts are allocated, “offseterror_left/right” = 0 will 5346

have to be considered if the left/right CS(s) is/are allocated so that the tables will therefore have to 5347 be duplicated according to neighbour left/right CS allocation or not (4 more tables)) and those 5348 tables will be used in a dynamic way (the table values to be used can change each time a user is 5349

added or discarded from the SRS comb). 5350

- With those formulas, the number of samples in each filter will be: 34 samples for 5MHz, 62 5351 samples for 10MHz, 98 samples for 15MHz and 108 samples for 20MHz (corresponding to MDFT/8 5352

+ 2 x offseterror + x, with x = 1 or 2). 5353

5354

12.7. TIME DOMAIN NOISE REMOVAL 5355

The fixed point implementation is updated compared to eCEM implementation. 5356

The fixed point implementation for comparison to noise threshold is given below (in red the 5357

changes compared to eCEM implementation): 5358

5359

5360

5361

5362

5363

5364

Notice that an alternative approach to do the scaling above before comparison is to do an 5365 equivalent left shift on the noise instead of doing a right shift on the signal. This alternative 5366

approach is implemented in the DSP. 5367

For each couple (user,antenna), Luth(a) shall be computed as an output of this step and used in the 5368

signal power estimation step. 5369

Note: a defense mechanism should be added so that the particular case where all the values are 5370

considered as noise and zeroed out shall be properly supported. 5371

Fixed point implementation for noise threshold computation is given in the figure below (in red the 5372

changes compared to eCEM implementation). Notice that this implementation is specific to the 5373

case where Kth is equal to 4, which is the current working view. However, based on further 5374 considerations (e.g. tests in real conditions, further studies, etc…) we highlight that this value of 4 5375

may be updated to another value, so that the fixed point chain below should then be updated. 5376

5377

5378

5379

5380

( )2

,ˆ aihSRS

u

s32 (or u31) (stored on 32 bits)

>> 3 + log2(M/6) To comparison (comparison of signed 32-bit values)

s(29-log2(M/6)) (stored on 32 bits)

Supprimé : 4.0.1

Supprimé : 2







5381

5382

5383

5384

5385

5386

5387

5388

5389

5390

5391

5392

5393

5394

- The inputs values (σ^PRB,a)² for each PRB are provided by the FPGA to the DSP each subframe 5395

(short-term noise variance, filled each ms by the FPGA at the address specified by the DSP (HIF)). 5396

- “e(PRB)” is the PUSCH AGC shift provided by the FPGA to the DSP each subframe. In LA 4.0.1, 5397

the max value of e(PRB) is 7. 5398

- “sf” is the iDFT scaling factor (scaling factor previously applied by the MAPLE on the output iDFT 5399 samples due to adaptative scaling) to be applied before comparison. 5400

- Kth is hardcoded to 4, M/6 = number of PRBs, on log M/6 bits, [>> log M/6] is there so that the 5401

sample values and the noise threshold have the same unit. 5402

Notes: 5403

- The precision loss versus eCEM implementation (additional [>>3]) is assumed to be acceptable 5404 since it has no impact on the final precision of the data (results used for threshold comparisons 5405 only). 5406

5407

12.8. DFT 5408

The DFT is performed inside the DFTPE block of the MAPLE. The input and the output of the DFT 5409

are in the format (16I, 16Q). Notice that as for the Xilinx implementation in eCEM, the MAPLE DFT 5410 has an internal scaling. For each DFT, the internal scaling factor (“si”) applied by the MAPLE can 5411 be fetched. Notice that as for the Xilinx engine, the DFT is not normalized by the DFT size. 5412

12.9. IDFT/DFT SCALING 5413

As for eCEM implementation, after DFT, we have to rescale the signal to take into account the 5414

IDFT and DFT internal scaling factors, and the required normalization factor DFTM/1 . 5415

( )2,ˆ aPRBσ << (2e(PRB) – 1)

M

u24 (stored on 32 bits)

>> (2 x sf)

s40

s38 (stored on signed 40 bits: Freescale “Word40” type)

Ms(40 – log2(M/6))

s40

(sf is the Freescale iDFT scaling factor to be applied before comparison. 2 x sf

because squared samples are compared to the noise threshold)

Right bit shift without rounding allowing to keep on 40 bits

without saturating in the following steps.

(If possible, to be gathered with the previous PRB-dependent

left shift in the implementation)

>> log2(M/6) - 2 x 3 >>2

Sat32


Acceptable saturation is Sat(29-log2(M/6)) <

Sat32

s38

To comparison (comparison of signed 32-bit values)

Right bit shift without

rounding.

[>> 2 followed by x 3] <=> [>> 5 followed by

x 6Kth i.e. x 24]

s(40-2sf) (stored on signed 40 bits: Freescale “Word40” type)

Supprimé : 4.0.1

Supprimé : 2







The algorithm to apply is quite similar to the Xilinx iDFT/DFT rescaling process (done by the FPGA 5416 for eCEM), but with the new 16-bit Freescale constraint. We recall below the different scalings 5417 during the IDFT/DFT processing: 5418 5419

5420

5421

5422

5423

5424

The rescaling to be applied on each sample’s real and imaginary parts is therefore [x (2sf+si / MDFT)]. 5425 This can be simplified to a multiplication followed by a right shifit [x q / 2r] with q and r defined as: 5426 5427

DFT

Mfloorq

Mq

DFTs ))((log22 +

= 5428

)( if sshr +−= , ))((log2 DFTs Mfloorqh += , 5429

qs=14 5430

5431

5432

Transmission BW MDFT q r

5MHz 192 10923 21 – (sf + si)

10MHz 360 11651 22 – (sf + si)

15MHz 576 14564 23 – (sf + si)

20MHz 600 13981 23 – (sf + si)

5433 5434

Notice that a defense may have to be planned for 10MHz for the (very unrealistic) case “sf=si=11”. 5435 5436 The rescaling process is described below for the real part (this is the same for imaginary part): 5437

5438 5439

5440

5441

5442

5443

For a precision purpose, the rescaling is done by each DSP core and not inside the MAPLE (where 5444 only a bit shift would be possible, not a division by MDFT). 5445


Notice that for bCEM, the CFO processing and SRS AGC are inverted with respect to eCEM 5447

because this eases the design. There is no expected performance impact. 5448

iDFT/DFT rescaling

2p-sf MDFT.2-sf-si 1.0 scale : 1.0

iDFT

Freescale

DFT

Freescale

s16

>> r s16

q

s16

s31

Sat16

Supprimé : 4.0.1

Supprimé : 2







The “CFO de-rotation” will be applied to all UEs, not only to the UE identified as MIMO users. This 5449 will be transparent for the “Signal power estimation” step but will be played anyway to simplify the 5450

design: a small imprecision will thus be introduced in the power estimation due to complex 5451

exponential fixed point finit precision. 5452

The CFO de-rotation is done the same way as for data blocks (see section 6.2.1.2) on a user by user 5453 basis without frequency domain filtering. This consists in a multiplication of the user’s signal by the 5454

complex exponential ( ) ( )

Ι−

FFT

ˆ..2exp Numj επ as defined in 6.2.1.2. 5455

The values of ( )mΙ are listed in the Annex 8, and m=13 shall be used for SRS. Only the Q(1.12) 5456

values for 20 MHz are listed but the same values can be used for any bandwidth since the ratio 5457 ( )

FFTNmΙ is constant for all transmission BWs. 5458

For eCEM, the format of the complex exponentials is Q(1.12). m=13 is also valid for all transmission 5459 BWs. For bCEM, the precision of those values shall be extended to Q(1.15) format 5460

Those complex values are new in the DSP (only present in the FPGA today). 5461 Fixed point implementation is described below for the real part (same for imaginary part with a 5462

subtraction instead of an addition): 5463

5464

5465

5466

5467

5468

5469

5470

5471

5472

5473

12.11. POST-DFT AGC 5474

We use the same approach as for eCEM but with 16bits instead of 18 so that we reduce the 5475

number of bits at the DFT output from 16 to 12. One AGC shift per UE is applied, this shift is 5476

common for all antennas. 5477

For a given UE, we determine the upper position of the MSB for the DFT outputs of all Rx antennas 5478

and keep twelve bits starting from this determined MSB. The resulting shift E(SRS) per UE will be 5479

used in the “Signal power estimation” step (see below), in the “SRS Channel amplitude estimation” 5480 step (see below), and in the “Combination of DMRS and SRS frequency domain correlations” for 5481

timing offset estimation. 5482

Let “order” denotes the function computing the number of bits representing the sign before the first 5483 significative bits. 5484

Let ( )iH au

SRS

,ˆ denote the value (on 16bits) of the DFT output, for tone i, user u, antenna a. . Then 5485

we compute: 5486

(16I,16Q) (stored on 2 x 16 bits)

>>15 Sat16






One complex coefficient (16I,16Q)

(stored on 2 x 16 bits)

Supprimé : 4.0.1

Supprimé : 2

Supprimé : ¶¶







( ) ( )

( ) ( )

−=

−=

iHiauSi

iHiauSr

au

au

SRS

,

SRS

,

ˆImorder16,,

ˆReorder16,, 5487

The AGC shift for user u is then computed as: 5488

( )( ) ( )( ) 11,,max,,,maxmax)(,,

−

= iauSiiauSruAGCshift

iaia 5489

If the result above is negative, then the AGC shift is equal to zero. 5490

Notice that the method above does not describe the way the shift computation should be done, its 5491 intent is only to give a mathematical description. Any equivalent method that is suitable to the 5492

designer can be used to compute the upper MSB position. 5493

The real and imaginary parts are shifted from 16 bits to 12 bits according to E(SRS), if E(SRS) > 0: 5494

5495

5496

5497

5498

The right bit shift on real and imaginary parts is done with rounding to avoid accuracy loss. 5499

Without the “post-DFT AGC” step (i.e. if we keep 16 bits in all the next steps), the approximations 5500 in the “SRS frequency domain correlation” step would not be acceptable (a [>> 8] bit shift would be 5501

required to keep on 40 bits, which is a too big loss of accuracy). 5502

12.12. SIGNAL POWER ESTIMATION 5503

For a given comb, the output of this step is one real value provided per UE and per group of ULSG 5504

PRBs (common for all antennas). For each couple (UE, group of ULSG PRBs), that output to ULS is 5505

the sum of the NbAntennas final SNR values computed for each antenna 5506 5507

The number of PRBs per SRS group is equal to ULSG =1 for 1.4MHz, ULSG =2 for 3 / 5MHz, and 5508

ULSG =4 for 10 / 15 / 20MHz. 5509

5510 The fixed point implementation for Signal power estimation is described below: 5511 5512

5513 5514 5515 5516 5517 5518 5519 5520 5521 5522

>> E(SRS) (16I,16Q)

(12I,12Q)

(12I,12Q)

|.|2 s12

|.|2

s24 (or u23) (stored on 32 bits)

s12

s29 (because a group of max 4 PRBs is considered)

(stored on 32 bits)

M

( )∑−

=

16

0

2ULSG

iSRS iH

One group of PRBs ⇒ 6x4 = 24 subcarrier

values (10, 15, 20MHz case)

Those are the values for one given antenna.

Supprimé : 4.0.1

Supprimé : 2







5523 5524 5525 5526 5527 5528 5529

The fixed point implementation of SRS debiasing is given below. 5530

5531

For short-term noise values, “E(PUSCH)” is equal to “e(PRB)” (see above for its definition – 5532

the same variable is just named differently here). This successive scaling factors are 5533

[<< 2.e(PRB) - 1] (scaling factors previously applied on this PRB value), then another shift 5534 [>> 2] is applied (because of the format expected by the ULS, also applied on the SRS 5535

sample modules in this figure), and a shift of [>> 15] is applied to compensate for the 5536

multiplication with the Q(1.15) number Luth(a)/MDFT. At last, a shift [>> 2] (10MHz, 15MHz 5537 or 20MHz case) or [>>1] (5MHz case) is applied on the sums (because average on a group 5538 of 4 or 2 PRBs is considered, also applied on the SRS sample modules in this figure). 5539

5540

Supprimé : 4.0.1

Supprimé : 2







5541

5542 5543

5544 A possible (future) improvement for better precision, can be the removal of the [>> 2] at the output of 5545 SRS energy computation, provided the ULS code is modified in all its functions using those values 5546 (no reverse [<< 2] needed any longer). If so an adaptation of the way ST Noise and LT Noise are 5547 used should also be done. For the moment, to simplify, we will not do this improvement. 5548 5549 The computation of the SRS SINR for UL scheduler is done as follows for eCEM: 5550

Short-term noise ( )2

,ˆ aPRBσ for one the

PRBs of the group ⇒

ULSG PRB values (10, 15, 20MHz ca

se). Those are the values for one given antenna.

>> 18 - 2.E(PUSCH)

Unbiased SRS energy

1365

s29

-

Right bit shift without rounding.

Extra right bit shift of

( )ULSG2log without

rounding because average on a group of

ULSG PRBs is

considered (*) .

>> (13 + 2 + ( )ULSG2log – 2.E(SRS))


Right bit shift of 2 without rounding to fit the former implementation

( )∑−

=

16

0

2ULSG

iSRS iH


Right bit shift without rounding. Equivalent to [<< 2.e(PRB) - 1] (scaling factors previously applied on this PRB

value) + [>> 15] due to Q(1.15) multiplication format + [>> 2] (because of the [>> 2]

shift applied on the samples above due to format expected by the ULS) + [>> ( )ULSG2log ] (because average on a group of

ULSG PRBs is considered)

u24

Luth(a)/MDFT


s16 (Q(1.15)) (stored on 16 bits)

s36 s38 (or s37 for 5MHz)

(stored on signed 40 bits: Freescale “Word40” type)

MM

>> ( )ULSG2log and Sat32

s32

Sum values of all antennas and Sat32 s32

(stored on 32 bits)

NOTE ( )ULSG2log =0 for

1.4, =1 for 3 and 5MHz, =2 for 10/15/20MHz

Other antenna values for this PRB group

(If ever the difference is negative, 0 is then

considered for this PRB group and this antenna)

Supprimé : 4.0.1

Supprimé : 2







5551

On the contrary of eCEM, for bCEM, the (>>2) at the input of the long term noise averaging is 5552 discarded. This means that the noise is 4 times bigger than for both LA4.0.1 eCEM and LA3.0 5553 drop2 implementation. 5554

Therefore, a (>>2) has to be added to LT noise before SNR computation. 5555

5556

12.13. SRS CHANNEL AMPLITUDE ESTIMATION 5557

The goal of the “SRS Channel amplitude estimation” step is to compute the SRS MIMO channel 5558

metric (list of complex numbers) for the UL scheduler in P4080. This step shall be played for all 5559 UEs (SIMO or MIMO) as needed by the ULS today. 5560

For a given comb, the output of this step is one complex value provided per PRB, per UE and per 5561

antenna. For each couple (UE, PRB), NbAntennas outputs to ULS are therefore generated. An 5562 average computation is done on the 6 subcarrier complex values associated to each PRB. 5563

5564

This SRS Channel amplitude estimation is already implemented in the DSP today for the division 5565 by 6, but the sum of the 6 PRB values will be new in the DSP for bCEM implementation (previously 5566 done by the FPGA). 5567

5568

5569

5570

5571

5572

5573

5574

LT Noise

Unbiased SRS Energy

••

>>2



(Long-term noise, common to all UEs)

s31 (stored on 32 bits) SRS SNR to ULS

(the final scaling factor is unchanged compared to previous releases).

s32 on DSP-ULS interface.

(Note: average on all antennas is done in the existing DSP SW: when there are 4 Rx antennas, this is now doable by using u32 accumulation format)

Supprimé : 4.0.1

Supprimé : 2







5575

5576

5577

5578

5579

5580

5581

5582

5583

5584

5585

5586

5587

12.14. SRS FREQUENCY DOMAIN CORRELATION 5588

The bCEM fixed point implementation of the SRS frequency domain correlation is given below: 5589

5590

5591 5592

5593 5594 5595 5596 5597 5598 5599 5600 5601

Value of tone i (12I,12Q)

−−2

2s

NM SRSSk samples between Imin and Imax

1st antenna *

>>2

s32

s23

s24

s(24+

−−2

2log2

sNM SRS

Sk )

(stored on signed 40 bits: Freescale “Word40” type)

32-bit input value for the combination of DMRS and SRS frequency domain correlations used for Timing offset

estimation .The scaling factor [>>2] will be compensated for in this step.

Right bit shift without rounding

(final shift in order not to exceed 32

bits) Complex conjugate of value of tone

i+s/2 (12I,12Q)

SRS,,

~aiuH

(12I, 12Q)

M

M

5461

>>(15-E(SRS))

∑

∈PRBiaiuH SRS

,,

~6

1

Sum over 6 values

(15I, 15Q) Stored on 2x16bits


Right bit shift on real and imaginary parts without

rounding


Sat16 Sat16 to be done on

both real and imaginary parts (to

keep the existing s16 DSP-ULS interface)

SRS MIMO channel metric to ULS (the final scaling factor is unchanged compared to

previous releases). (s16,s16) on DSP-ULS interface.

Supprimé : 4.0.1

Supprimé : 2







13. RACH PROCESSING 5602

13.1. OVERVIEW 5603

The main purpose of the E-UTRA random access (RA) procedure is to obtain uplink time synchronization 5604 within a fraction of the uplink cyclic prefix. In WCDMA, the random access is non-orthogonal to the uplink 5605 data transmission. This provides the benefit of not having to specifically allocate any resources for random 5606 access, but requires a power ramping procedure to control the inter-UE interference. In LTE, the random 5607 access is made orthogonal to the scheduled data transmissions by reserving a certain bandwidth 5608 specifically for random access for one sub-frame at regular intervals. When a UE needs to access the 5609 network, it transmits a random access preamble. 5610

Upon receiving the random access preamble, the Node B will estimate the timing and frequency offset of 5611 this preamble. The Node B may send a timing advance commands to the UE as part of the RA response 5612 message to allow the UE to adjust its uplink transmission timing accordingly. The frequency offset 5613 information obtained from the random access process can be used as the reference value for the 5614 frequency offset estimation/compensation block in the receiver chain. 5615

5616

13.2. SIGNAL STRUCTURE 5617

The standard has agreed to adopt a Random Access preamble format with cyclic prefix to enable 5618 frequency domain processing. Several burst formats are defined to operate for cells with different sizes. 5619

5620

13.2.1 RANDOM ACCESS PREAMBLE FORMATS 5621

5622 The generic structure of a RACH is shown in Figure 13-1 below. It consists of a cyclic prefix followed by a 5623 sequence that is one or more copies of a preamble followed by a gap that conceptually extends the RACH 5624 to the next subframe boundary. The values Ncp and Nseq are shown in Table 13-1 and are in samples at the 5625 appropriate sample rate. 5626 5627 The RACH occupies a bandwidth of 1.08 MHz (6 RBs) and its length is a multiple of 1ms. The location in 5628 the frequency domain is controlled by the parameter k0, configured by higher layers (see Table 13-1). 5629

5630

5631

5632

Figure 13-1 Generic RACH Structure 5633 5634 There are 4 formats defined for the Type 1 RACH shown in Figure 13-2 below. 5635

5636 5637

CP Sequence Gap

Ncp Ngap Nseq

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Police :NonGras, Anglais (États-Unis)

Mis en forme : Police :NonGras, Anglais (États-Unis),Vérifier l'orthographe et lagrammaire

Mis en forme : Police :NonGras, Anglais (États-Unis)

Mis en forme : Police :NonGras, Anglais (États-Unis),Vérifier l'orthographe et lagrammaire

Supprimé : Figure 13-1

Supprimé : Table 13-1









5638 5639

Figure 13-2 Type 1 RACH Formats 5640 5641 Figure 13-2 above shows the RACH message 1 formats for FDD as defined in 36.211. 5642 5643 Format 0 was designed for cells up to approximately 14 Km. 5644 Format 1 was designed for cells up to approximately 77 Km with no link budget issues. 5645 Format 2 was designed for cells up to approximately 28 Km supporting high data rates. 5646 Format 3 was designed for cells up to approximately 100 Km. 5647 5648 See section 13.6.5 for further discussion of the relation between RACH size and distance. 5649 5650 5651 5652

Table 13-1 – Rach Parameters 36.211 Section 7.5.1 5653 5 MHz 10 MHz 20 MHz Burst

Format Ncp Npre Ncp Npre Ncp Npre 0 792 6144 1584 12288 3168 24576 1 5256 6144 10512 12288 21024 24576 2 1560 12288 3120 24576 6240 49152 3 5256 12288 10512 24576 21024 49152

5654 5655 The random access preambles are generated from Zadoff-Chu sequences with zero correlation zone, (ZC-5656 ZCZ) generated from one or several root Zadoff-Chu sequences. The network configures the set of 5657 preamble sequences the UE is allowed to use. The uth root Zadoff-Chu sequence is defined by 5658

( ) 10

1

−≤≤=+π−

ZCN

)n(uni

u Nn,enx ZC Nzc = 839 5659

From the thu root Zadoff-Chu sequence, random access preambles with zero correlation zone are defined 5660 by cyclic shifts of multiples of vC according to 5661

5662 )Nmod)Cn((x)n(x ZCvuv,u += (13-1) 5663

where vC is configurable by the upper layers: 5664

5665

( )

−+=+−=

=SetsstrictedRennn,,,vNnModvn/vd

SetsedUnrestrictN/N,,,vvNC RA

shiftRAgroup

RAshiftcs

RAshift

RAshiftstart

cszccsv 110

110

K

K 5666

5667

CP

CP

CP

CP

Sequence

Sequence

Sequence 1

Sequence 1

Format 0

Format 2

Format 3

Format 1

1 Subframe 1 Subframe 1 Subframe

Sequence 2

Sequence 2

Supprimé : 4.0.1

Supprimé : 2








See section 5.7.2. of 36.211 for details. The value of Ncs is determined by the Ncs configuration shown in 5668 Table : 5669 5670

5671 Table 13-2 - Ncs Configuration 5672

Ncs value NCS

Configuration Unrestricted Set

Restricted Set

0 0 15

1 13 18

2 15 22

3 18 26

4 22 32

5 26 38

6 32 46

7 38 55

8 46 68

9 59 82

10 76 100

11 93 128

12 119 158

13 167 202

14 279 237

15 419 -

5673 5674

13.2.2 BASEBAND RACH SIGNAL 5675

5676 The time-continuous random access signal )(ts is defined by 5677

5678

( ) ( )( ) ( )∑−

=

−∆++ϕ+πβ=1

0

2 21

0

ZC

CPRA

N

k

TtfkKkjPRACH e)k(xts (13-2) 5679

5680 5681

where ∑−

=

π−=1

0

2zc

zc

N

n

N/nkiv,u e)n(x)k(x is the DFT of the shifted ZC sequence defined by equation (13-2) and 5682

CPPRE0 TTt +<≤ . The parameters in (13-2) are defined in Table below: 5683

5684 5685

Table 13-3 －－－－ RACH Parameters from 36.211 5686

Parameter Value Description

PRACHβ Amplitude scaling factor.

k0 2NNNnk RBsc

ULRB

RBsc

RAPRB0 −= Controls the location of the RACH signal

in the frequency band. RAPRBn 6Nn0 UL

RBRAPRB −≤≤ Expressed as a resource block number,

is configured by higher layers.

RAffK ∆∆= 12 Ratio of uplink subcarrier spacing and RACH subcarrier spacing

Supprimé : 4.0.1

Supprimé : 2

Supprimé : (13-2)

Supprimé : (13-2)







ϕ 7 Fixed offset of RACH carrier within a resource block.

5687 5688

13.2.3 LARGE CELL OPERATION 5689

For large cells with radius more than 15km, either extended mode or repeated mode Rach access should 5690 be used. As we mentioned before, when repeated mode is used, at the receiving end, we may accumulate 5691 repeated preambles or for simplicity in initial development, we may only take one of the repeated copies. 5692 Further study is needed on this subject. 5693 5694 For extended mode and non-accumulating repeated mode, the Rach receive processing is identical to the 5695 normal mode except for the start timing (due to different value of Tcp). 5696 5697 The other receiving option for repeated Rach mode is to accumulate the correlation results at the output of 5698 1024 IFFT across multiple 0.8ms intervals. In this case, the receiver block diagram will remain the same 5699 except for the additional accumulation across multiple preambles. This will improve link level performance. 5700 However, how much of the link level gain translates to system level gain depends on operation scenario 5701 which is still not clear at this point. Therefore whether accumulating repeated Rach is needed should 5702 remain open. 5703 5704 Based on the requirement for LA4.0, large cell option with accumulating Rach is not necessary. Thus, only 5705 normal mode, extended mode and non-accumulating repeat mode are supported. 5706 5707

13.2.4 RANDOM ACCESS BURST CONFIGURATION 5708

Table lists the subframes in which RACH transmission is allowed for a given configuration in frame 5709 structure 1. The start of the RACH is aligned with the corresponding uplink subframes at the Ue assuming 5710 a timing advance of zero. 5711 5712

Table 13-4 –Frame Structure Type 1 Rach Timings for Format 0 5713 5714 5715 5716 5717 5718 5719

5720

5721

5722 5723 5724 5725 5726 5727 5728 5729 5730 5731 5732 5733 5734 5735 5736

PRACH configuration

Index

System Frame Number

Subrame Number

0 Even 1 1 Even 4 2 Even 7 3 Any 1

4 Any 4

5 Any 7 6 Any 1,6

7 Any 2,7 8 Any 3,8

9 Any 1, 4, 7 10 Any 2, 5, 8 11 Any 3, 6, 9

12 Any 0, 2, 4, 6, 8 13 Any 1, 3, 5, 7, 9

14 Any 0, 1, 2, 3, 4, 5, 6, 7, 8, 9 15 Even 9

Supprimé : 4.0.1

Supprimé : 2







Table 13-5 –Frame Structure Type 1 Rach Timings for Format 2 5737 5738 5739 5740 5741 5742 5743

5744

5745

5746 5747 5748 5749 5750 5751 5752 5753 5754 5755 5756 5757 5758 5759 5760 For multiple cell operation, we assume the timing of the RA burst is NOT aligned among the cells. They will 5761 have evenly distributed offsets so that the Rach processing load for the eNB is evenly distributed as well. In 5762 other word, we assume a configuration like the following: 5763 5764

Cell 1

Cell 2

Cell 3

10 ms 5765

Figure 13-3 RA burst configuration 5766 5767

PRACH configuration

Index

System Frame Number

Subrame Number

32 Even 1

33 Even 4

34 Even 7 35 Any 1

36 Any 4 37 Any 7 38 Any 1, 6 39 Any 2 ,7

40 Any 3, 8

41 Any 1, 4, 7 42 Any 2, 5, 8

43 Any 3, 6, 9 44 Any 0, 2, 4, 6, 8

45 Any 1, 3, 5, 7, 9 46 N/A N/A 47 Even 9

Supprimé : 4.0.1

Supprimé : 2







13.2.5 OPERATION IN HIGH MOBILITY ENVIRONMENT 5768

High mobility environment may result in high frequency offset. The maximum frequency offset UL,offsetf 5769

seen at the eNB receiver is obtained as 5770

max_DopplerUEBSUL,offset ffff ×+∆+∆= 2 5771

where BSf∆ , UEf∆ , and max_Dopplerf denote the base station frequency drift, UE frequency error, and the 5772

maximum Doppler frequency, respectively. While it has not been decided in LTE, as a reference, In the 5773 UMTS W-CDMA system, the frequency error requirements at the base station are 0.05 ppm and 0.1 ppm of 5774 the carrier frequency. We suppose that for carrier frequency of 2.6 GHz, the maximum frequency offset is 5775 1000 Hz for a UE moving at the velocity 120 kmph. 5776 5777 Rach preamble detection is done in effect by a CAZAC sequence correlator implemented in frequency 5778 domain. Two issues need to be taken care of when operating in high frequency offset environment, as 5779 explained in the following: 5780 5781 1. frequency offset may result in additional correlation peaks in time domain. 5782 5783

The separation of the dominant additional peak from the normal correlation peak depends on the root 5784 index. The effect of the additional correlation peaks can be reduced by sequence restrictions meaning 5785 that the set of available root sequences and their cyclic shifts are limited in such a way that false 5786 detections due to the additional peaks overlap in time with other shifts’ peaks can be avoided by only 5787 allocate a subset of otherwise Nzc available shifts. From L1 perspective, we will assume the allocation 5788 restriction has been properly taken care of by upper layer scheduling. However, we may utilize the 5789 knowledge of possible additional peaks to aid detection. 5790

5791 2. Detection performance degradation due to frequency offset. 5792 5793

Frequency offsets affects on the detection performance and false alarm rate when the baseline 5794 preamble sequence is CAZAC with circular shifts. If there exists frequency offset at the receiver (Node-5795 B) due to Doppler spread or residual frequency offset, we can see that the frequency sampling position 5796 is not aligned with subcarrier position to result in mixed signal with neighbor subcarriers, as shown in 5797 above figure. 5798

5799 5800

5801 Frequency offset results in additional correlation peaks at multiple of coff offsets, where coff = (Nzcm-1)/u for 5802 the uth Cazac root sequence. Here m is smallest positive integer for which coff is integer . coff 5803 corresponding to the peak produced by Doppler shift fDopp =1/TPRE and it depends on the root index u. 5804 While the values of additional shifts depend on the frequency offset values, the location of the additional 5805 peaks only depend on the root Cazac sequence index u. Further, we notice that for a given frequency 5806 offset values, the large peaks concentrates in a limited number of locations. Thus, for a given the CAZAC 5807 sequence index u, and for a user whose nominal shift is d, we can consider the triple peaks at locations d 5808 and (d+/-coff )mod Nzc, and using proper combining to combat the scattering effect resulting from frequency 5809 offset. 5810 5811 The ZC sequence of odd length is given as 5812

)N

)k(kujexp()k(a

ZCu

1+π−= , 5813

where u is the index of the root sequence, NZC the length of the sequence, and k=0, 1, … NZC-is the index 5814 of the samples. 5815 5816 In the following, let ( ) Nmoddka)k(a ZCud,u −= refer to the d'th cyclic shift of the root sequence u. k is the 5817

index in time that is of interest (i.e., in the timing uncertainty window) to the use with nominal shift d. The 5818 correlation values of the cyclic shift triplet { })k(a),k(a),k(a )Nmodcd(,ud,u)Nmodcd(,u GoffGoff +− is shown in Figure 5819

13-4. 5820 5821

Supprimé : 4.0.1

Supprimé : 2







5822

5823 5824

Figure 13-4 Peak triplet due to frequency offset 5825 5826 5827

5828

13.2.6 HIGH LEVEL REQUIREMENT 5829

The RACH L1 processing is dimensioned to support the following: 5830 1. support 3 Sector 5MHz and 1 Sector 10MHz, 15MHz and 20MHz operation 5831 2. support normal RACH operation and high mobility operation 5832 3. support 15km cell size with RACH format 0, 30km cell size with RACH format 2, and up to 120km/h 5833

mobility 5834 4. support up to 64 CAZAC sequences per cell 5835

5836 An important issue is how many root sequence a cell must support. The number is decided by 3 aspects: 5837 1. cell size 5838 2. mobility scenario 5839 3. root sequence ordering and network planning scheme (how to assign the sequences to each cell in a 5840 large network. 5841

5842 Cell size: 5843 Cell size decides Nzc. For 15km cell size, the time uncertainty window = 5us + 6.67us/km * 15km = 105us. 5844 Thus, we should choose Ncs = 119 from the Ncs table. Thus, for low mobility case, each root can support 7 5845 signatures and it requires a total of 10 roots to support 64 signatures. 5846 As for RACH format 2, for 30km cell size, the time uncertainty window = 5us + 6.67us/km * 30km = 205us. 5847 Thus, we should choose Ncs = 279 from the Ncs table. Thus, for low mobility case, each root can support 3 5848 signatures and it requires a total of 22 roots to support 64 signatures. 5849

5850 Mobility scenario: 5851 As mentioned above, for carrier frequency of 2.6 GHz, the maximum frequency offset is around 1000 Hz for 5852 a UE moving at the velocity 120 kmph. In the case, if not frequency offset mitigation scheme is applied, not 5853 only the probability of miss will increase significantly due to signal attenuation, but the chance that a 5854 frequency offset image being detected erroneously as another signature will increase even more 5855 significantly as well. Therefore, we must consider this as a high mobility scenario and apply frequency 5856 offset mitigation schemes. 5857 5858 Frequency offset mitigation consists of two aspects: on one hand, the Rach-PD needs to take into account 5859 the additional images due to frequency offset; on the other hand, usable shifts must be restricted to make 5860 sure the images due to frequency offset from one signature does fall into the timing uncertainty window of 5861 another signature. 5862 5863 It turns out, for a given Ncs, the number of signature supportable by a root sequence in high mobility case 5864 depends on the root sequence index u. For Ncs = 119, there are 194 roots that can support 2 signatures 5865 per root, 290 roots can support 1 signature and the rest are not usable. 5866

5867 Root sequence ordering and sequence planning in network: 5868

Supprimé : 4.0.1

Supprimé : 2







The number of roots needed to support 64 signatures depends on which root sequences are assigned to 5869 the cell. If we only assign those that can support 2 signatures, we only need 32 roots. But if all of the roots 5870 assigned to the cell can only support 1 signature, we need 64 roots to support 64 signatures. 5871 However, we can’t assign any roots arbitrarily to a cell. To save signaling overhead, roots are pre-ordered 5872 into a table shared by both UE and the eNB and only the index for the first root is communicated from the 5873 cell to the UE. The roots corresponding to the next 64 signatures are automatically chosen to be used by 5874 both the UE and the eNB as defined in Table 5.7.2-4 of 36.211 (v8.7). 5875

13.3. RACH PROCESSING 5876

The RACH processing is logically partitioned into front end processing and back end processing. In the 5877 front end section, the RACH signals from each diversity antenna are processed identically and 5878 independently to extract the RACH signal. In the back end, the two antennas are brought together, 5879 equalized and combined to form a single array of metrics that is searched for correlation peaks. Back end 5880 processing is performed for each ZC root specified in the Signature Table provided by the DSP via the 5881 SRIO-A interface. 5882

13.3.1 RACH FRONT END PROCESSING 5883

Normal mode Rach access should be used for cells with radius less than 15km. The block diagram for 5884 normal operation is given in Figure 13-5. It applies for 5 MHz, 10 MHz, 15 MHz,and 20 MHz 5885

Supprimé : 4.0.1

Supprimé : 2







5886 Figure 13-5 Block Diagram for RACH Front End Proce ssing 5887

5888 5889 5890

After all data is captured, on a per antenna basis, for the latest arriving preamble duration (TPRE), RACH 5891 cell processing first performs RACH front end processing to exact Rach signal. First, scaling is performed 5892 to reduce the bit width. Then the sample rate received signal goes through frequency shift, low pass 5893 filtering, cyclic removal and down sampling to finally become a length 2048 sequence for RACH format 0&1, 5894 or a length 2*2048 sequence for RACH format 2&3. The front end processing portions are different for 5895 5MHz, 10 MHz, 15 MHz, and 20M Hz cases all generate a sequence of length 2048/2*2048 depending on 5896 RACH format. Each antenna-preamble for the current cell is then converted to the frequency domain via 5897 FFT. RACH subcarriers are then extracted and stored in the Pre-IFFT buffer. For every root CAZAC 5898 sequence of interest, each antenna-preamble source is multiplied by the conjugate of the root CAZAC 5899 sequence in the frequency domain and converted back into the time domain correlation via IFFT. The 5900 result is the (circular) correlation between received signal and root CAZAC sequence in time domain. 5901

5902 5903

Bit Selection

Band Shift

Filter/ Decimate

Pre 2048 pt FFT Scaling

2048 pt FFT

Extract ZC Sequence

Scale Extracted

Data

To Backend Processing

16-bit Rx Data

Supprimé : 4.0.1

Supprimé : 2







13.3.2 INPUTS TO RACH FRONT END PROCESSING 5904

The inputs to RACH front end processing are supplied by the DSP through the SRIO-A interface: 5905 5906 5907

Table 13-6 – Inputs to RACH Front End Processing 5908

Signal Name Source Format Size Description

DIV_MODE SRIO-A u2 1

Diversity mode: 0: Antenna A and B 1: Antenna A only 2: Antenna B only 3: all Four antennas

SECTOR SRIO-A u1 1 Sector 0, 1, or 2 3 is reserved

N_SHIFT SRIO-A u8 1 Bandshift factor

PRE_FORMAT SRIO-A u3 1 RACH format 0, 1, 2, or 3 4..7 reserved.

N_SCALE SRIO-A u22 1 Xilinx FFT scaling schedule “hardwired” to 0x155555

rx_sample RX_B_IF (16I,16Q) N x NRX per RA burst

N is Received samples per antenna, where N=7680 for 5MHz N=15360 for 10MHz N=23040 for 15MHz N=30720 for 20MHz NRX is the number of Rx Antennas , where NRX=1 for 1 Rx antenna NRX=2 for 2 Rx antennas NRX=4 for 4 Rx antennas

RX_SHIFT SRIO-A u2 1 Select front end bit selection. It can be 0, 1, 2 and 3.

5909 5910 5911

5912 5913

Figure 13-6 Bit Selection and Max Mag Calculation 5914 5915

Bit Selection

Average

Max Mag

15 12

MaxMag

To Rach Processing Xk

Average NumOvrFlow

Yk

Supprimé : 4.0.1

Supprimé : 2







13.3.3 FRONTEND BIT SELECTION 5916

Interface: 5917 Table 13-7 Inputs and outputs of Frontend Bit Selection 5918


rx_sample Vector of

complex Integer

(16I,16Q)

I N x NRX per RA

burst

Received sample per antenna

N=7680 for 5MHz N=15360 for 10MHz N=23040 for 15MHz N=30720 for 20MHz NRX is the number of Rx Antennas , where NRX=1 for 1 Rx antenna NRX=2 for 2 Rx antennas NRX=4 for 4 Rx antennas

rach_sample Vector of

complex Integer

(12I,12Q)

O N x NRX per RA

burst

Output of front end bit selection

N=7680 for 5MHz N=15360 for 10MHz N=23040 for 15MHz N=30720 for 20MHz NRX is the number of Rx Antennas , where NRX=1 for 1 Rx antenna NRX=2 for 2 Rx antennas NRX=4 for 4 Rx antennas

5919 5920

As defined in the LTE Channel Element eCEM Internal Interface Document sections 3.4.4, 3.4.5, and 3.4.6 5921 for 5MHz, 10MHz, 15MHz and 20MHz respectively, each rx data sample is a 16 bit word of the form: 5922

5923

5924 5925 5926

The data is actually 15 bits with the MSB (denoted by x above) being the extension of the sign bit (denoted 5927 by s above). For current Rach processing, a bit width of 12 bits has been chosen. The following pseudo 5928 code shows the method to select the 12 bits from the 16 shown above, 5929 5930 5931

Y = Saturate12( RoundStrip( X, RX_SELECT ) ) 5932

5933

X represents the signed 15-bit I or Q samples from one of the two antennas. Y is the 12-bit signed output 5934

from bit selection. Overflow is saturated to 12 bits and a counter is updated for each overflow. Samples are 5935

processed from the start of RACH data acquisition and proceeds to the last acquired RACH sample. 5936

With [ ]0, 1Rxn N∈ − and NRX=1,2,4 for the cases with 1, 2 and 4 Rx antennae respectively,NumOvrFlowIn, 5937

NumOveFlowQn, 5938

x s b0

b7

b5

b2

b4

b3

b6

b1

b8

b9

b10

b11

b12

b13

Sign bit

Extension of sign bit

Supprimé : 4.0.1

Supprimé : 2







are the 16-bit unsigned overflow counters corresponding to the nth Rx antenna that will be returned to the 5939

DSP at the end of RACH processing. 5940

5941 5942 The current working view is that no AGC processing is done at this point. Two scaling functions are 5943 inserted into the processing path to maintain dynamic range. 5944 5945 5946 5947

13.3.4 MAX MAGNITUDE CALCULATION 5948

MaxMag calculations are performed on the 12-bit signed integers after bit selection. The purpose of 5949 MaxMag is to detect data underflow and to provide the ability to monitor the long term magnitude of the 5950 received data. Here Y[k] is either the real or imaginary part of the data after bit selection. 5951 5952 The calculation begins at the start of RACH data acquisition and ends after the last RACH sample has 5953 been read. 5954 5955 MaxMag Calculation 5956 5957 MaxMag = 0; 5958 5959 for( k=0; k<N; k++) 5960 MaxMag |= Abs( Y[k] ); 5961 5962

where N refers to the Received samples per antenna given in Table 13-6. 5963

The 12-bit MaxMag value for each I and Q channel for each antenna is returned to the DSP as a diagnostic 5964 at the end of RACH processing. 5965 5966 5967

13.3.5 FREQUENCY SHIFT TO BASEBAND 5968

In order to filter and decimate the received RACH signal it is necessary to shift the RACH into the 5969 passband of the decimation filter. If X[k] are the digitized sampled of the RACH data, 5970

( ) expshift N/nNje)k(XnY π= 2 (13-3) 5971

where, 1 1 ink k ,..., k N 1= + − and inn 0,..., N 1= − are indices for the input rx_sample1 and output 5972

rx_sample2 respectively. k1 indicates the input position where the operation begins. The normalized 5973 frequency shift is: 5974 5975

RA ULshift PRB RB

124 18 3 2n N

Nυ

= − + − where

20MHzfor 45762

15MHzfor 84321

10MHzfor 22881

5MHzfor 1446

3MHzfor 3072

N

= 5976

5977 Substituting the bandwidth dependent parameters we get for the shift defined by equation (13-3): 5978 5979

Supprimé : 4.0.1

Supprimé : 2


Supprimé :

6144 for 5 MHz

12288 for 10 MHzN

18432 for 15 MHz

24576 for 20 MHz

=

Supprimé : (13-3)







( )( )( )

( )( )

20MHzfor n473

15MHzfor n2692

10MHzfor n2232

5MHzfor n21932

3MHzfor n1832

N

RAPRB

RAPRB

RAPRB

RAPRB

RAPRB

shift

−×

−×

−×

−×

−×

= and Nexp = 512 5980

5981 The value of Nshift is computed by the DSP and delivered to the RACH HW through the register 5982 A_LRACH_FSHIFT__NSHIFT_n (n = 1, 2, or 3) during RACH configuration. Note that in the HW, a single 5983 sinusoid table of length 512 can accommodate the shifts for all three bandwidths. 5984 5985 After the frequency shift, the RACH will be centered in the baseband with 420 bins on the positive side and 5986 419 bins on the negative side (see Figure 13-16). 5987 5988 Since this is a simple phase rotation, bit width should not increase with proper scaling. The bit width of the 5989

real and imaginary parts of the phasors used for the frequency shift sequence expshift N/nNie π2 is 10 bits. The 5990 integer part is two bits (one for sign and one bit to represent 1) and an eight bit fraction. 5991 5992 5993

.s

Sign bit

8-bit fraction

2-bit integer part

5994 5995 5996

Figure 13-7 10-bit Format for Frequency Shift Phas ors 5997 5998 1.4MHz system has 6PRBs which corresponds to RACH bandwidth , as a consequence the received signal 5999 is already centred.So, there is no need to bandshifting. 6000 6001 6002

6003

13.3.6 UPSAMPLING, FILTERING AND DECIMATION 6004

After bandshifting the input data to place the RACH signal in the center of the band, we filter and decimate 6005 to reduce the sample rate to 2.56 MHz. The input sample rate of a 20 MHz system is 30.72 M samples/sec, 6006 the input sample rate of a 15 MHz system is 23.04 M samples/sec, the input sample rate of a 10 MHz 6007 system is 15.36 M samples/sec and the input sample rate of a 5 MHz system is 7.68 M samples/sec. Thus 6008 the input samples for the 20 MHz, 15 MHz, 10 MHz, and 5 MHz systems need to be decimated by a factor 6009 of 12, 9, 6, and 3 respectively. We accommodate this by inserting an extra decimation-by-4 filtering block in 6010 the 20 MHz processing chain, an extra decimation-by-3 filtering block in the 15 MHz processing chain and 6011 a decimation-by-2 filtering block in the 10 MHz processing chain: 6012 6013

6014

6015

6016

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Normal,Gauche

Mis en forme : Police :(Pardéfaut) Arial

Supprimé :

((

((473

692

32

1932

Nshift

×

×

×

×

=


Supprimé : UPSAMPLING,







6017

6018

6019

6020

Figure 13-8 20 MHz Processing Chain 6021

6022

6023

6024 6025 6026 6027 6028 6029 6030

6031 Figure 13-9 15 MHz Processing Chain 6032

6033

6034

6035 6036 6037 6038 6039

6040 Figure 13-10 10 MHz Processing Chain 6041

6042 6043 6044 6045 6046 6047 6048 6049 6050

6051 6052 6053 6054

Figure 13-11 5 MHz Processing Chain 6055 6056 6057 6058 6059

6060 6061 6062

Figure 13- 12 3 MHz Processing Chain 6063 6064

6065

3.84 MHz Bandshift to Center

Signal

Filter and Decimate

by 3

Bit Scale 16 bits to 12 Bits

2.56 MHz 3.84 MHz Antenna Samples

2

7.68 MHz

Bandshift to Center

Signal

Antenna

Samples

Filter and Decimate

by 3

Filter and Decimate

by 2


2.56 MHz 15.36 MHz 7.68 MHz 15.36 MHz

7.68 MHz

Bandshift to Center

Signal

Filter and Decimate

by 3


2.56 MHz 7.68 MHz

Antenna

Samples

Bandshift to Center

Signal

Antenna

Samples

Filter and Decimate

by 3

Filter and Decimate

by 4


2.56 MHz 30.72 MHz 7.68 MHz 30.72 MHz

Bandshift to Center

Signal

Antenna

Samples

Filter and Decimate

by 3

Filter and Decimate

by 3


2.56 MHz 23.04 MHz 7.68 MHz 23.04 MHz

Supprimé : 4.0.1

Supprimé : 2

Supprimé : <sp>

Supprimé : 11

Supprimé : 3 MHz Processing Chain







6066 6067 6068

6069 6070

Figure 13- 13 1.4 MHz Processing Chain 6071 6072

13.3.6.1 Filter Design Methodology 6073

6074 Figure 13- 14 Filter Design Methodology 6075

Figure 13-14is a schematic of the filter design process for the FDD LTE frontend. Based on the decimation 6076 rate and filter stage, a set of filter specifications is generated. The filter specifications are input to the 6077 Remez filter design program. 6078 6079 The output of the Remez program is the set of floating point coefficients for the designed FIR filter. The 6080 coefficients are reformatted as a .coe file for input to the Xilinx coregen filter compiler. 6081 6082 The .coe file together with the coregen parameters (Table 13-8 and Table 13-9) generate (besides VHDL) a 6083 .mif file which is a binary representation of the actual filter coefficients used by the Xilinx IP. The binary 6084 values in the .mif file are converted to hex numbers for use by the System Model and a corresponding file 6085 of scaled floating point coefficients for analysis and documentation. 6086 6087 Note that the coefficients in the .coe file will differ slightly from the final coefficients generated from the .mif 6088 file due to quantization of the filter coefficients. 6089 6090 6091 6092 6093 6094

Table 13-8 – Coregen Parameters Common to all Filte rs 6095 Coregen Parameter Value

1.92 MHz Filter and Decimate

by 3


2.56 MHz 1.92 MHz Antenna Samples

4

7.68 MHz

Filter Specs

Floating values in Alg Doc

Hex values to FPGA Filter

Remez Filter

Design

Generate .coe file

Convert to

float

coregen filter file

Convert to hex

.mif file

coregen parameters

Supprimé : 4.0.1

Supprimé : 2

Supprimé : <sp>

Supprimé : 11

Supprimé : 1.4 MHz Processing Chain

Supprimé : 12










coefficient_width 16 data_sign Signed data_width 12 filter_type Decimation output_rounding_mode Symmetric_Rounding_to_Zero output_width 12 quantization Maximize_Dynamic_Range

6096 6097 6098

Table 13-9 – Filter Specific Coregen Parameters 6099 Coregen Parameter DecBy2 DecBy3 DecBy4

coefficient_structure Half_Band Symmetric Symmetric decimation_rate 2 3 4

6100

13.3.6.2 Calculations for Format 0 6101

13.3.6.2.1 Upsampling for 1.4 MHz (Format 0) 6102

For 1.4 MHz, an upsampling by a factor of 4 is applied in order to increase sample rate from 1.92 MHz to 6103 7.68 MHz and apply 5MHz filter. 6104

The upsampling method considered is : 6105

- Zero-order hold interpolation: each sample is replicated three times: 6106

=L

kXkY )( ( )inNLkkk ×+= 11,..., 6107

With: 6108

- X is the output of Frontend Bit Selection block 6109

- 4=L 6110

- 1921 =k 6111

- 1542=inN 6112

6113

13.3.6.2.2 Upsampling for 3 MHz (Format 0) 6114

For 3 MHz, an upsampling by a factor of 2 is applied in order to increase sample rate from 1.92 MHz to 6115 7.68 MHz and apply 5MHz filter. 6116



=L

kXkY )( ( )inNLk ×= ,...,0 6119

With: 6120

- X is the output of Frequency Shift to Baseband block 6121

- 2=L 6122

Supprimé : 4.0.1

Supprimé : 2

Mise en forme : Puces etnuméros











- 3083=inN 6123

For 3MHz case, the value of the parameter 1k , used for bandshifting, is equal to 384. 6124

6125

6126

13.3.6.2.3 Calculations for 5 MHz Filtering (Format 0) 6127

These calculations pertain to RACH format 0 for either 5 MHz filtering or the second stage of 10 MHz, 15 6128 MHz and 20 MHz filtering. Here we want the final number of output samples to be 2048. 6129

6130 Nout = 2048 we want the final number of output points to be 2048. 6131 Ntaps = 25 we have chosen a 25 tap FIR filter for this stage of decimation. 6132 L = 12 the half-width of the filter 6133 D = 3 the decimation factor for this stage of filtering. 6134

From these values we calculate the required number of input samples 6135

6136

Nin = Dx( Nout – 1 ) + Ntaps = 3 x (2048 – 1 ) + 25 = 6166 6137

6138

For the 5 MHz case, the total number of available input samples is: 6139

6140

Nr = Ncp + Npre = 792 + 6144 = 6936 6141

6142

The nominal position of the central filter tap for the first filtered sample is: 6143

6144

kc = ( Nr – 1 – L ) – Dx( Nout – 1 ) = 782 6145

6146

At this point we note that the distance from kc to the start of the preamble is 792 – 782 = 10. Since we are 6147 decimating by a factor of 3, we back up two more samples to kc = 780 so this distance is a multiple of 3: 6148

792 – 780 = 12 = 4·3 6149

6150

In this case the multiple of the decimation factor is 4. We will adjust for the offset of kc from the start of the 6151 preamble before taking the 2048 point FFT by a circular shift of 4 samples (section 13.3.10). 6152

6153

Summary: 6154

kc = 780 the (adjusted) index of the central tap for the first filtering operation. 6155

k1 = kc – L = 780 – 12 = 768 the index of the first sample to be filtered. 6156

6157


For 10 MHz filtering, a factor-of-2 decimation stage precedes the same factor-of-3 decimation stage 6159 described in the previous section 13.3.6.2.3. Since the final decimation by 3 requires an input of 6166 6160 points, this will be the required number of output points from the first stage of decimation: 6161

6162

Supprimé : 4.0.1

Supprimé : 2



Supprimé : 13.3.6.2.1







Nnout = 6166 the number of output points from this first stage of filtering. 6163 Ntaps = 7 we have chosen a 7 tap FIR filter for this stage of decimation. 6164 L = 3 the half-width of the filter 6165 D = 2 the decimation factor for this stage of filtering. 6166

6167


6169


6171


6173

Nr = Ncp + Npre = 1584 + 12288 = 13872 6174

6175


6177

kc = ( Nr – 1 – L ) – Dx( Nout – 1 ) = 1538 6178

6179

At this point we note that the distance from kc to the start of the preamble is 1584– 1538 = 46. Since we 6180 are decimating by a factor of 6, we back up two more samples to kc = 1536 so this distance is a multiple of 6181 6: 6182

6183

1584– 1536= 48= 8·6 6184

6185

In this case the multiple of the decimation factor is 8. We will adjust for the offset of kc from the start of the 6186 preamble before taking the 2048 point FFT by a circular shift of 4 samples. 6187

6188

After filtering and decimation by 2, the location of the start of the preamble, kpre, will be: 6189

6190

242

153615842

kNk ccp

pre =−=−

= 6191

6192

The following decimate by 3 filter will be centered at index k’c = 12 so the distance from the start of the 6193 prefix will be 6194

6195

kpre – k’c = 24 – 12 = 12 6196

6197

After decimating by 3 this will correspond to a time shift of 4 samples (section 13.3.10). 6198

6199

Summary: 6200

6201


6203

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Supprimé : 2

Supprimé : Knout








6205

13.3.6.2.5 Calculations for 15 MHz Filtering (Format 0-3) 6206


6210 Nnout = 6166 the number of output points from this first stage of filtering. 6211 Ntaps = 15 we have chosen a 15 tap FIR filter for this stage of decimation. 6212 L = 7 the half-width of the filter 6213 D = 3 the decimation factor for this stage of filtering. 6214

6215



6218


Format0: Nr = Ncp + Npre = 2376 + 18432 = 20808 6220




6224


Format 0; kc = ( Nr – 1 – L ) – Dx( Nout – 1 ) = 2305 6226




6230

To ensure that the distance kc to the start of the (second, if applicable) preamble is a multiple of 9, the 6231 adjusted index of the central tap of the first stage filter is given as: 6232

Format 0: 2376 – 2304(adjusted kc) = 72 = 8·9 6233

Format 1: 15768 – 15696(adjusted kc) = 72 = 8·9 6234

Format 2: 4680 + 18432(first copy of the preample) – 23040(adjusted kc) = 72 = 8·9 6235

Format 3: 15768 + 18432(first copy of the preample) –34128(adjusted kc) = 72 = 8·9 6236

The first sample output from the decimate-by-3 filter will be at offset 2304

7683

= at the 15 MHz rate. This 6237

corresponds to the offset of the first sample for next bank of the decimate-by-3 filter k1 (see section 6238 1.3.6.2). 6239


cp cpre

N k 2376 2304k 24

3 3

− −= = = 6241

6242

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Supprimé : 2


Supprimé : 13.3.6.2.1

Supprimé : Knout








6245

kpre – k’c = 24 – 12 = 12 6246

6247


6249

Summary: 6250

Format 0: kc = 2304 the (adjusted) index of the central tap for the first filtering operation. 6251




6255

Format 0: k1 = kc – L = 2304 – 7 = 2297 the index of the first sample to be filtered. 6256





For 20 MHz filtering, a factor-of-4 decimation stage precedes the same factor-of-3 decimation stage 6261 described in the previous section. Since the final decimation by 3 requires an input of 6166 points, this will 6262 be the required number of output points from the first stage of decimation: 6263


6269


6271


6273


6275

Nr = Ncp + Npre = 3168 + 24576 = 27744 6276

6277


6279

kc = ( Nr – 1 – L ) – Dx( Nout – 1 ) = 3073 6280

6281

At this point we note that the distance from kc to the start of the preamble is 3168–3073 = 95. Since we are 6282 decimating by a factor of 12, we back up one sample to kc = 3072 so this distance is a multiple of 12: 6283

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6284

3168–3072 = 96= 8·12 6285

6286


3072= at the 5 MHz rate. This 6287

corresponds to the offset of the first sample for the decimate-by-3 filter k1 (see section 13.3.6.2.3). 6288

6289

Summary: 6290

6291


6293


13.3.6.3 Calculations for Format 2 6295

13.3.6.3.1 Upsampling for 1.4 MHz (Format 2) 6296

For 1.4 MHz, an upsampling by a factor of 4 is applied in order to increase sample rate from 1.92 MHz to 6297 7.68 MHz and apply 5MHz filter. 6298



=L

kXkY )( ( )inNLkkk ×+= 11,..., 6301

With: 6302

- X is the output of Frontend Bit Selection block 6303

- 4=L 6304

- 3841 =k 6305

- 3078=inN 6306

6307

13.3.6.3.2 Upsampling for 3 MHz (Format 0) 6308

For 3 MHz, an upsampling by a factor of 2 is applied in order to increase sample rate from 1.92 MHz to 6309 7.68 MHz and apply 5MHz filter. 6310



=L

kXkY )( ( )inNLk ×= ,...,0 6313

With: 6314

- X is the output of Frequency Shift to Baseband block 6315

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Mis en forme : Retrait :Gauche : 0 cm, Première ligne: 0 cm, Hiérarchisation +Niveau : 4 + Style denumérotation : 1, 2, 3, … +Commencer à : 1 + Alignement: Gauche + Alignement : 0 cm+ Tabulation après : 2,54 cm+ Retrait : 1,52 cm,Tabulations :Pas à 2,54 cm






Supprimé : 13.3.6.2.1

Supprimé : ¶







- 2=L 6316

- 6155=inN 6317

For 3MHz case, the value of the parameter 1k , used for bandshifting, is equal to 768. 6318

6319


These calculations pertain to RACH format 2 for either 5 MHz filtering or the second stage of 10 MHz, 15 6321 MHz and 20 MHz filtering. Here we want the final number of output samples to be 4096 due to the 2 6322 repititions inherent in the signal of RACH format 2. 6323

6324 Nout = 4096 we want the final number of output points to be 4096. 6325 Ntaps = 25 we have chosen a 25 tap FIR filter for this stage of decimation. 6326 L = 12 the half-width of the filter 6327 D = 3 the decimation factor for this stage of filtering. 6328


6330


6332


6334

Nr = Ncp + Npre = 1560 + 2*6144 = 13848 6335

6336


6338

kc = ( Nr – 1 – L ) – Dx( Nout – 1 ) = 1550 6339

6340

At this point we note that the distance from kc to the start of the preamble is 1560 – 1550 = 10. Since we 6341 are decimating by a factor of 3, we back up two more samples to kc = 1548 so this distance is a multiple of 6342 3: 6343

1560 – 1548 = 12 = 4·3 6344

6345

In this case the multiple of the decimation factor is 4. We will adjust for the offset of kc from the start of the 6346 preamble before taking the 2048 point FFT twice for the 2 repetitions of RACH preambles by a circular shift 6347 of 4 samples (section 13.3.10). 6348

6349

Summary: 6350



6353

Supprimé : 4.0.1

Supprimé : 2











6363


6365


6367


6369

Nr = Ncp + Npre = 3120 + 24576 = 27696 6370

6371


6373

kc = ( Nr – 1 – L ) – Dx( Nout – 1 ) = 3074 6374

6375

At this point we note that the distance from kc to the start of the preamble is 3120– 3074 = 46. Since we 6376 are decimating by a factor of 6, we back up two more samples to kc = 3072 so this distance is a multiple of 6377 6: 6378

6379

3120– 3072= 48= 8·6 6380

6381

In this case the multiple of the decimation factor is 8. We will adjust for the offset of kc from the start of the 6382 preamble before taking the 2048 point FFT by a circular shift of 4 samples. 6383

6384


6386

cp cpre

N k 3120 3072k 24

2 2

− −= = = 6387

6388


6391

kpre – k’c = 24 – 12 = 12 6392

6393


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Supprimé : 2


Supprimé : 13.3.6.2.1

Supprimé : Knout







6395

Summary: 6396

6397


6399


6401




6411



6414


Nr = Ncp + Npre = 4680 + 36864 = 41544 6416

6417

6418


kc = ( Nr – 1 – L ) – Dx( Nout – 1 ) = 4609 6420

6421

6422

At this point we note that the distance from kc to the start of the preamble is 4680– 4609 = 71. Since we 6423 are decimating by a factor of 9, we back up one more samples to kc = 4608 so this distance is a multiple of 6424 6: 6425

6426

4680– 4608= 72= 8·9 6427


15363


corresponds to the offset of the first sample for next bank of the decimate-by-3 filter k1 (see 6429 section13.3.6.3.3). 6430


cp cpre

N k 4680 4608k 24

3 3

− −= = = 6432

6433

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Supprimé : 2


Supprimé : 13.3.6.2.1

Supprimé : Knout

Supprimé : 13.3.6.3.1








6436

kpre – k’c = 24 – 12 = 12 6437

6438


6440

Summary: 6441




For 20 MHz filtering, a factor-of-4 decimation stage precedes the same factor-of-3 decimation stage 6445 described in the previous section. Since the final decimation by 3 requires an input of 12310 points, this will 6446 be the required number of output points from the first stage of decimation: 6447


6453


6455

Nin = Dx( Nout – 1 ) + Ntaps = 4 x (12310– 1 ) + 21 = 49257 6456

6457


6459

Nr = Ncp + Npre = 6240 + 2*24576 = 55392 6460

6461


6463

kc = ( Nr – 1 – L ) – Dx( Nout – 1 ) = 6145 6464

6465

At this point we note that the distance from kc to the start of the preamble is 6240–6145 = 95. Since we are 6466 decimating by a factor of 12, we back up one sample to kc = 6144 so this distance is a multiple of 12: 6467

6468

6240–6144 = 96= 8·12 6469

6470


15364


corresponds to the offset of the first sample for the decimate-by-3 filter k1 (see section 13.3.6.3.3). 6472

6473

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Supprimé : 2


Supprimé : Knout

Supprimé : 13.3.6.3.1







Summary: 6474

6475


6477


6479

13.3.7 SUMMARY OF FILTERING PARAMETERS 6480

The tables below is the summary of the parameters for RACH filtering and decimation. The 10 MHz column 6481 are the parameters for the first stage of filtering which is decimation by 2. 6482 6483

Table 13-10 – Upsampling Parameters for 1.4MHz 6485 Variable Format 0 Format 2 k1 192 384

6486 Table 13-11 – Upsampling Parameters for 3MHz 6488

Variable Format 0 Format 2 k1 384 768

6489

6490 Table 13-12 – Adjusted Filtering Parameters for 5MHz 6491

Variable Format 0 Format 1 Format 2 Format 3 Nr 6936 11400 13848 17544 kc 780 5244 1548 11388 k1 768 5232 1536 11376 k2 6933 11397 13845 17541

6492

6493

6494 Table 13-13 – Adjusted Filtering Parameters for 10 MHz 6495


6496 6497

6498 Table 13-14 – Adjusted Filtering Parameters for 15MHz 6499


6500 6501

Table 13-15 – Adjusted Filtering Parameters for 20 MHz 6502 Variable Format 0 Format 1 Format 2 Format 3 Nr 27744 45600 55392 70176 kc 3072 20928 6144 45504 k1 3062 20918 6134 45494 k2 27742 45598 55390 70174

Supprimé : 4.0.1

Supprimé : 2


Supprimé : Table 13-10 – ¶Variable

Supprimé : Table 13-10 – Upsampling Parameters for 3MHz¶Variable

Supprimé : ¶

Supprimé : 12

Supprimé : 10

Supprimé : 13

Supprimé : 11

Supprimé : 14

Supprimé : 12

Supprimé : 15

Supprimé : 13

... [4]

... [5]







6503 6504 6505

Table 13-16 – Filtering Parameters Common to Format 0 6508 Variable 1.4 MHz 3 MHz 5 MHz 10 MHz 15 MHz 20 MHz D ---- ---- 3 2 3 4 Ntaps ---- ---- 25 7 15 21 L ---- ---- 12 3 7 10 Nout ---- ---- 2048 6166 6166 6166 Nin 1542 3083 6166 12337 18510 24681

6509 Table 13-17 – Filtering Parameters Common to Format 2 6510

Variable 1.4 MHz 3 MHz 5 MHz 10 MHz 15 MHz 20 MHz D ---- ---- 3 2 3 4 Ntaps ---- ---- 25 7 15 21 L ---- ---- 12 3 7 10 Nout ---- ---- 4096 12310 12310 12310 Nin 3078 6155 12310 24625 36942 49257

6511 6513 Note: k2 is the last sample into the filter = k1 + Nin −1. 6514 6515 6516

13.3.7.1 Specifications for Decimation by 2 filter 6517

For decimating by 2 we would want the passband to end at 420/12288 and the stopband to start at 1/2 – 6518 420/12288 in normalized frequencies. However, in order to give the RACH signal a bit more protection from 6519 out-of-band signals we will increase the passband and decrease the stop band by about 30 bins at the 6520 6144 rate. We choose for the passband and stop band normalized frequencies: 6521

6522

3036458.012288

448Fpass == 64.0

12288448

Fstop 635412

1 =−= 6523

6524 6525 6526 6527

Table 13-18 - Input Specifications for Decimation by 2 Filter 6528 Number of taps 7 Number of bands 2 Filter type 1 Parameters for band 1 Band start 0 Band end 0.036458333 Functional value 1 Weight function 1 Parameters for band 2 Band start 0.463541666 Band end 0.5 Functional value 0 Weight function 1

6529 6530 The final coregen results are: 6531 6532

Supprimé : 4.0.1

Supprimé : 2

Supprimé : ¶

Supprimé : Table 13-14 – Filtering Parameters Common to Format 0¶Variable

Supprimé : Table 13-14 – Filtering Parameters Common to Format 0¶Variable

Supprimé : 15

... [6]

... [7]







H(1) = 0xf7ae = H(7) = -6.500244140625e-02 6533 H(2) = 0x0000 = H(6) = 0.000000000000e+00 6534 H(3) = 0x4850 = H(5) = 5.649414062500e-01 6535 H(4) = 0x7fff = H(4) = 9.999694824219e-01 6536 6537 The corresponding floating point values (scaled by 2-15): 6538 6539 Maximum passband deviation –77.4742 dB 6540 Maximum stopband deviation –78.2095 dB 6541 6542

13.3.7.2 Specifications for Decimation by 3 filter 6543

For decimating by 3 we would want the passband to end at 420/6144 and the stopband to start at 1/3 – 6544 420/6144 in normalized frequencies. However, in order to give the RACH signal a bit more protection from 6545 out-of-band signals we will increase the passband and decrease the stop band by about 30 bins at the 6546 6144 rate. We choose for the passband and stop band normalized frequencies: 6547 6548

6072916.06144448

Fpass == 6260416.06144448

Fstop =−=3

1 6549

6550 6551 When decimating by 3, two sections of the frequency band are aliased into the passband. If we assume 6552 that the levels in these bands are of the same order of magnitude as the RACH signal itself, we will want to 6553 give twice the weight to the stopband attenuation as to the passband deviation. Our target passband 6554 deviation is –77 dB (12 bit input samples) and the target stopband attenuation is –77 –6 = –83 dB (–6 from 6555 the other two bands). 6556

6557 6558

Table 13-19 - Input Specifications for Decimate by 3 Filter 6559 Number of taps 25 Number of bands 2 Filter type 1 Parameters for band 1 Band start 0 Band end 0.0729167 Functional value 1 Weight function 1 Parameters for band 2 Band start 0.260417 Band end 0.5 Functional value 0 Weight function 5.5463

6560 6561

The final coregen results are: 6562 6563 6564 H( 1) = 0xffd5 = H(25) -1.312255859375e-03 6565 H( 2) = 0xff69 = H(24) -4.608154296875e-03 6566 H( 3) = 0xff5f = H(23) -4.913330078125e-03 6567 H( 4) = 0x00f0 = H(22) 7.324218750000e-03 6568 H( 5) = 0x03dd = H(21) 3.018188476562e-02 6569 H( 6) = 0x0455 = H(20) 3.384399414062e-02 6570 H( 7) = 0xfd8e = H(19) -1.910400390625e-02 6571 H( 8) = 0xf1a0 = H(18) -1.123046875000e-01 6572 H( 9) = 0xee3b = H(17) -1.388244628906e-01 6573 H(10) = 0x041f = H(16) 3.219604492188e-02 6574

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 16







H(11) = 0x34cf = H(15) 4.125671386719e-01 6575 H(12) = 0x693c = H(14) 8.221435546875e-01 6576 H(13) = 0x7fff = H(13) 9.999694824219e-01 6577

6578 The corresponding floating point values (scaled by 2-15): 6579 6580 6581

Maximum passband deviation –76.7655 dB 6582 Maximum stopband deviation –84.2884 dB 6583 6584 6585

13.3.7.3 Specifications for 1st bank of Decimation by 3 filter for 15MHz 6586

For decimating by 3 we would want the passband to end at 420/ 18432 and the stopband to start at 1/3 – 6587 420/ 18432 in normalized frequencies. However, in order to give the RACH signal a bit more protection 6588 from out-of-band signals we will increase the passband and decrease the stop band by about 30 bins at the 6589 6144 rate. We choose for the passband and stop band normalized frequencies: 6590 6591

pass

448F 0.024305555555556

18432= = stop

1 448F 0.309027777777778

3 18432= − = 6592

6593 6594 When decimating by 3, two sections of the frequency band are aliased into the passband. If we assume 6595 that the levels in these bands are of the same order of magnitude as the RACH signal itself, we will want to 6596 give twice the weight to the stopband attenuation as to the passband deviation. Our target passband 6597 deviation is –77 dB (12 bit input samples) and the target stopband attenuation is –77 –6 = –83 dB (–6 from 6598 the other two bands). 6599

6600 6601

Table 13-20 - Input Specifications for Decimate by 3 Filter 6602 Number of taps 15 Number of bands 2 Filter type 1 Parameters for band 1 Band start 0 Band end 0.024305555555556 Functional value 1 Weight function 1 Parameters for band 2 Band start 0.309027777777778 Band end 0.5 Functional value 0 Weight function 5

6603 The final MATLAB firpm output results are: 6604

H( 1) = 0xFF8C= H(15) -0.003534988510241 6605 H( 2) = 0xFD10= H(14) -0.022949047828030 6606 H( 3) = 0xF84D = H(13) -0.060148550265929 6607 H( 4) = 0xF849 = H(12) -0.060268203654491 6608 H( 5) = 0x0B23 = H(11) 0.086983035066565 6609 H( 6) = 0x36F2 = H(10) 0.429230316302889 6610 H( 7) = 0x6950 = H(9) 0.822755237222125 6611 H( 8) = 0x7FFF = H(8) 0.9999694824219 6612 6613

The corresponding floating point values (scaled by 2-15): 6614 6615

Maximum passband deviation –72.1422 dB 6616

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 17







Maximum stopband deviation –91.8669 dB 6617 6618 6619

13.3.7.4 Specifications for Decimation by 4 Filter 6620

For decimating by 4 we would want the passband to end at 420/24576 and the stopband to start at 1/2 – 6621 420/24576 in normalized frequencies. However, in order to give the RACH signal a bit more protection from 6622 out-of-band signals we will increase the passband and decrease the stop band by 6 bins at the 6144 rate 6623 which corresponds to half the RACH guard band. We choose for the passband and stop band normalized 6624 frequencies: 6625

6626

0173339843.024576

426Fpass == 6627

6628

[ ]0.267334,326662.024576

426

4

1Fstop1 =±= 482666.0

24576

426

2

1Fstop2 =−= 6629

6630 In addition to these three bands, we also specify a fourth band in the “don’t care” region with a smaller 6631 attenuation to guard against leakage of strong out-of-band signals: 6632 6633

]482.0,268.0[Ftrans = 6634

6635 6636 6637

Table 13-21 - Input Specifications for Decimation by 4 Filter 6638 Number of taps 21 Number of bands 4 Filter type 1 Parameters for band 1 Band start 0 Band end 0.017333984 Functional value 1 Weight function 1 Parameters for band 2 Band start 0.2326660156 Band end 0.2673339844 Functional value 0 Weight function 4 Parameters for band 3 Band start 0.268 Band end 0.482 Functional value 0 Weight function 0.025 Parameters for band 4 Band start 0.482666016 Band end 0.5 Functional value 0 Weight function 4

6639

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 18







6640 The final coregen results are: 6641 6642 H( 1) = 0x009d = H(21) = 4.791259765625e-03 6643 H( 2) = 0x00d6 = H(20) = 6.530761718750e-03 6644 H( 3) = 0xff59 = H(19) = -5.096435546875e-03 6645 H( 4) = 0xfb6e = H(18) = -3.570556640625e-02 6646 H( 5) = 0xf5c2 = H(17) = -8.001708984375e-02 6647 H( 6) = 0xf56b = H(16) = -8.267211914062e-02 6648 H( 7) = 0x0280 = H(15) = 1.953125000000e-02 6649 H( 8) = 0x2125 = H(14) = 2.589416503906e-01 6650 H( 9) = 0x4b79 = H(13) = 5.896301269531e-01 6651 H(10) = 0x70dc = H(12) = 8.817138671875e-01 6652 H(11) = 0x7fff = H(11) = 9.999694824219e-01 6653 6654 The corresponding floating point values (scaled by 2-15): 6655 6656 Maximum passband deviation –77.64 dB 6657 Maximum stopband 1 deviation –89.69 dB 6658 Maximum stopband 2 deviation –89.69 dB 6659 6660

13.3.8 SEQUENCE SEPARATION FOR FORMAT 2 6661

As shown in Figure 13-2, the preamble for Format 2 consists of two identical sequences that are the same 6662

as the single preamble of Format 0. In order to explore the time diversity oder inherent in the two 6663

sequences, after bandshifting and filtering the 2 sequences are separated from each other and fed to the 6664

pre 2048-point FFT scaling block respectively. 6665

6666

6667

6668

6669

6670

6671

Figure 13-15 Sequence Separation for RACH format 2 6672

13.3.9 PRE 2048-POINT FFT SCALING 6673

At this point in the processing we have potentially lost some dynamic range so we rescale the data for each 6674 antenna. The (twos-complement) absolute value of the real and imaginary parts of the data is or’ed into a 6675 single variable. The location of the most significant one bit is located and the shift amount is computed 6676 based on this location. The real and imaginary parts are shifted left to full 12-bit values. 6677 6678 val_mask = 0; 6679 6680 for( k=0; k<2048; k++) 6681 { val_mask |= Abs( real( fftbuf2048[k] ) ); 6682

val_mask |= Abs( imag( fftbuf2048[k] ) ); 6683 } 6684 6685 nshift = 12 - LeftmostBit( val_mask ) - 2; 6686 6687 for( k=0; k<2048; k++) 6688 { real( fftbuf2048[k] ) <<= nshift; 6689

Band Shift

Filter/ Decimate

Demulplex

Sequence 2

Sequence 1 pre

2048-FFT scaling

pre 2048-FFT

scaling

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Supprimé : 2


Supprimé : s

Supprimé : s

Supprimé : 13







imag( fftbuf2048[k] ) <<= nshift; 6690 } 6691 6692 LeftmostBit() returns the location of the most significant bit such that if bit 0 is the leftmost 1 bit then 6693 LeftmostBit() returns zero. In the case where the argument is zero, it also returns zero. Each antenna is 6694 scaled separately and the shift amount for each antenna is stored for post processing. The value mask for 6695 each diversity is saved and output as a diagnostic parameter to the DSP at the end of RACH processing. 6696 6697 Outputs for N=2 case: 6698 6699 Nshift2048A – shift amount for antenna A 6700 Nshift2048B – shift amount for antenna B 6701 ValMask2048A − 12-bit unsigned value mask going into 2048 FFT for antenna A 6702 ValMask2048B − 12-bit unsigned value mask going into 2048 FFT for antenna B 6703 6704 Outputs for N=4 case: 6705 6706 Nshift2048A – shift amount for antenna A 6707 Nshift2048B – shift amount for antenna B 6708 Nshift2048C – shift amount for antenna C 6709 Nshift2048D – shift amount for antenna D 6710 ValMask2048A − 12-bit unsigned value mask going into 2048 FFT for antenna A 6711 ValMask2048B − 12-bit unsigned value mask going into 2048 FFT for antenna B 6712 ValMask2048C − 12-bit unsigned value mask going into 2048 FFT for antenna C 6713 ValMask2048D − 12-bit unsigned value mask going into 2048 FFT for antenna D 6714 6715 Outputs for N=8 case: 6716 6717 Here, N refers to the degree of freedom order of the Chi-square distribution which will be elaborated in 6718 13.4.1. As stated in 13.3.8, the two sequences contained in the signal of RACH format 2 are retrieved and 6719 fed to pre 2048-FFT scaling separately. Thus N = 8 corresponds to 4 Rx diversity with RACH format 2. 6720 6721 Nshift2048A1 – shift amount of the 1st sequence for antenna A 6722 Nshift2048A2 – shift amount of the 2nd sequence for antenna A 6723 Nshift2048B1 – shift amount of the 1st sequence for antenna B 6724 Nshift2048B2 – shift amount of the 2nd sequence for antenna B 6725 Nshift2048C1 – shift amount of the 1st sequence for antenna C 6726 Nshift2048C2 – shift amount of the 2nd sequence for antenna C 6727 Nshift2048D1 – shift amount of the 1st sequence for antenna D 6728 Nshift2048D2 – shift amount of the 2nd sequence for antenna D 6729 ValMask2048A1 − 12-bit unsigned value mask of the 1st sequence going into 2048 FFT for antenna A 6730 ValMask2048A2 − 12-bit unsigned value mask of the 2nd sequence going into 2048 FFT for antenna A 6731 ValMask2048B1 − 12-bit unsigned value mask of the 1st sequence going into 2048 FFT for antenna B 6732 ValMask2048B2 − 12-bit unsigned value mask of the 2nd sequence going into 2048 FFT for antenna B 6733 ValMask2048C1 − 12-bit unsigned value mask of the 1st sequence going into 2048 FFT for antenna C 6734 ValMask2048C2 − 12-bit unsigned value mask of the 2nd sequence going into 2048 FFT for antenna C 6735 ValMask2048D1 − 12-bit unsigned value mask of the 1st sequence going into 2048 FFT for antenna D 6736 ValMask2048D2 − 12-bit unsigned value mask of the 2nd sequence going into 2048 FFT for antenna D 6737 6738 6739

13.3.10 2048-POINT FORWARD FFT 6740

Regardless of the input sample rate, after filtering and decimation the number of RACH samples is 2048. 6741

Although we are still in time domain at this point, the spectrum of the signal is shown below 6742

6743

Supprimé : 4.0.1

Supprimé : 2







6744

6745 6746

6747 6748

6749 Extract 6750 6751 6752 6753

6754 Figure 13- 16 Spectrum of Decimate Rach Signal 6755

Before taking the 2048 point FFT, we adjust for the offset from the preamble due to the filtering operation. 6756 The offset is 4 samples and we perform a circular shift to adjust for this offset. Pseudo-code for this 6757 operation is shown below: 6758 6759

6760 6761 n = 4; 6762 for( k=0; k<2048; k++) 6763 { 6764 fftbuf2048[k] = filter_output[n++]; 6765 if( n >= 2048 )n = 0; 6766 } 6767 6768

6769 Now perform the 2048-point FFT on fftbuf2048[]. After the FFT, each FFT bin contains a complex sample 6770 of the DFT of the shifted ZC sequence. 6771 6772

13.3.11 ZC SEQUENCE EXTRACTION 6773

The 839 samples of the DFT of the ZC sequence are extracted from the 2048 FFT output buffer to the 6774 1024 FFT input buffer. 6775 6776

6777 6778

6779 6780

Figure 13- 17 ZC Sequence Extraction 6781

2047

419 420

0 1629 419

420

0 419

419

1023 605

2048 FFT Buffer

2048 FFT Buffer

First sample 1629→605

Last sample 420→420

negative frequencies

419 420

0 2047

FFT[419] = Xu,v(Nzc-1)

positive frequencies Nyquist

FFT[1629] = Xu,v(0)

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 14

Supprimé : 15







6782 6783

The pseudo-code for this operation: 6784 6785

6786 ksrc = 1629; 6787 kdst = 1024-419; // = 605 6788 6789 for( k=0; k<839; k++) 6790 { fftbuf1024[kdst++] = fftbuf2048[ksrc++]; 6791 if( ksrc >= 2048 )ksrc = 0; 6792 if( kdst >= 1024 )kdst = 0; 6793 } 6794 6795

6796

Note that the index that corresponds to zc_seq[0] is FFT buffer index 605, wrap around to the beginning of 6797 the buffer and end at index 419. Indices 420 through 604 in the 1024 point FFT input buffer are zero 6798 padded. These samples will be multiplied by the conjugate of the DFT of the u’th-root ZC sequence for 6799 each root. 6800 6801

13.3.12 SCALING OF EXTRACTED ZC SEQUENCE 6802

At this point in the processing we have potentially lost some dynamic range so we rescale the data for each 6803 antenna. The (twos-complement) absolute value of the real and imaginary parts of the data is or’ed into a 6804 single variable. The location of the most significant one bit is located and the shift amount is computed 6805 based on this location. The real and imaginary parts are shifted left to full 12-bit values. 6806 6807 val_mask = 0; 6808 6809 for( k=0; k<1024; k++) 6810 { val_mask |= Abs( real( fftbuf1024[k] ) ); 6811

val_mask |= Abs( imag( fftbuf1024[k] ) ); 6812 } 6813 6814 nshift = 12 - LeftmostBit( val_mask ) - 2; 6815 6816 for( k=0; k<1024; k++) 6817 { real( fftbuf1024[k] ) <<= nshift; 6818

imag( fftbuf1024[k] ) <<= nshift; 6819 } 6820 6821 LeftmostBit() returns the location of the most significant bit such that if bit 0 is the leftmost 1 bit then 6822 LeftmostBit() returns zero. Each antenna is scaled separately and the shift amount for each antenna is 6823 stored for post processing. The value mask for each diversity is saved and output as a diagnostic 6824 parameter to the DSP at the end of RACH processing. 6825 6826 6827 6828 Outputs for N=2 case: 6829 6830 Nshift1024A – shift amount for antenna A 6831 Nshift1024B – shift amount for antenna B 6832 ValMask1024A − 12-bit unsigned value mask going into 1024 FFT for antenna A 6833 ValMask1024B − 12-bit unsigned value mask going into 1024 FFT for antenna B 6834 Outputs for N=4 case: 6835 6836 Nshift1024A – shift amount for antenna A 6837 Nshift1024B – shift amount for antenna B 6838

Supprimé : 4.0.1

Supprimé : 2







Nshift1024C – shift amount for antenna C 6839 Nshift1024D – shift amount for antenna D 6840 ValMask1024A − 12-bit unsigned value mask going into 1024 FFT for antenna A 6841 ValMask1024B − 12-bit unsigned value mask going into 1024 FFT for antenna B 6842 ValMask1024C − 12-bit unsigned value mask going into 1024 FFT for antenna C 6843 ValMask1024D − 12-bit unsigned value mask going into 1024 FFT for antenna D 6844 6845 Outputs for N=8 case: 6846 6847 Here, N refers to the degree of freedom order of the Chi-square distribution which will be elaborated in 6848 13.4.1. As stated in 13.3.8, the two sequences contained in the signal of RACH format 2 are retrieved and 6849 processed separately. Thus N = 8 corresponds to 4 Rx diversity with RACH format 2. 6850 6851 Nshift2048A1 – shift amount of the 1st sequence for antenna A 6852 Nshift2048A2 – shift amount of the 2nd sequence for antenna A 6853 Nshift2048B1 – shift amount of the 1st sequence for antenna B 6854 Nshift2048B2 – shift amount of the 2nd sequence for antenna B 6855 Nshift2048C1 – shift amount of the 1st sequence for antenna C 6856 Nshift2048C2 – shift amount of the 2nd sequence for antenna C 6857 Nshift2048D1 – shift amount of the 1st sequence for antenna D 6858 Nshift2048D2 – shift amount of the 2nd sequence for antenna D 6859 ValMask2048A1 − 12-bit unsigned value mask of the 1st sequence going into 2048 FFT for antenna A 6860 ValMask2048A2 − 12-bit unsigned value mask of the 2nd sequence going into 2048 FFT for antenna A 6861 ValMask2048B1 − 12-bit unsigned value mask of the 1st sequence going into 2048 FFT for antenna B 6862 ValMask2048B2 − 12-bit unsigned value mask of the 2nd sequence going into 2048 FFT for antenna B 6863 ValMask2048C1 − 12-bit unsigned value mask of the 1st sequence going into 2048 FFT for antenna C 6864 ValMask2048C2 − 12-bit unsigned value mask of the 2nd sequence going into 2048 FFT for antenna C 6865 ValMask2048D1 − 12-bit unsigned value mask of the 1st sequence going into 2048 FFT for antenna D 6866 ValMask2048D2 − 12-bit unsigned value mask of the 2nd sequence going into 2048 FFT for antenna D 6867 6868

Supprimé : 4.0.1

Supprimé : 2







6869

13.4. RACH BACK END PROCESSING 6870

6871

6872 6873

Figure 13- 18 RACH Back End Processing Flow 6874 6875 Figure 13-18 is a diagram of the RACH backend processing. This flow is executed for each root in the 6876 RACH root table. 6877

1024-pt FFT

Detect Peaks

Estimate Threshold

Analyze CFD

Compute Histogram

Compute Metrics

Rescale/ Equalize

Multiply by ZC

Sequence

Generate ZC

Sequence

Extracted Sequence

These functions are executed once per ZC root as specified in the Signature Table passed though the EmifA interface.

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 16








Table 13-22 – Inputs to RACH Back End Processing 6878 6879

Signal Name Source Format I/O Size Description

UN_WINDOW SRIO-A u10 I 1 Length of a single search window in bins

U_LAST SRIO-A u6 I 1 Last root index => number of roots = U_LAST + 1

DETECT_TH SRIO-A u16 I 1 Detection threshold

Signature Table SRIO-A I Definition for each signature

FFT1024n[],

internal (12I, 12Q) I 1024 complex

Extracted ZC sequence for nth sequence (for RACH format 0, n is just the antenna number; for RACH format 2, n is the sequence number for all the sequences contained in the RACH signals from all Rx antennas).

Nfn internal u4 I 1 Shift amount for 2048-FFT stage for nth sequence. .

Nin internal u4 I 1 Shift amount for 1024-FFT stage for nth sequence.

6880

The Signature Table consists of U_LAST+1 signature blocks, each block consisting of a Root Block Header 6881 and a list of Root Block Shifts.Table shows the contents of the Root Block Header. 6882 6883

Table 13-23 – Contents of Root Block Header 6884


U SRIO-A u10 I 1 ZC root [1..838]

INV_U SRIO-A u10 I 1 Multiplicative inverse of U Mod Nzc

V_LAST SRIO-A u6 I 1 Index of last shift for the\is root. Thus the number of shifts = V_LAST + 1

6885

Following the Root Block Header for each root, there are V_LAST+1 Root Block Shift entries each of which 6886

contains the following parameters: 6887

6888

Table 13-24 – Contents of Root Block Shift Entry 6889


UWS_0 SRIO-A u10 I 1 Start of search window for frequency hypothesis ∆f = 0

UWS_1 SRIO-A u10 I 1 Start of search window for frequency hypothesis +∆f

UWS_2 SRIO-A u10 I 1 Start of search window for

frequency hypothesis －∆f

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 19

Supprimé : a, Nib

Supprimé : 20

Supprimé : 21







6890

6891

6892

13.4.1 OVERVIEW OF BACK END PROCESSING 6893

The output of the 1024 point FFT is the cross-correlation between the reference root Cazac sequence and 6894 received the signal, properly down sampled to 839 point per 0.8ms to match the root Cazac sequence and 6895 then further upsampled to 1024 points. 6896 6897 For each user, the transmitted preamble signature is a circular shift of a root sequence. In TS36.211, for 6898 unrestricted sets, it is specified the shift is a multiple of Ncs, i.e., vNcs. For each signature, the hardware 6899 will be configured with 3 timing uncertainty windows corresponding to the 3 frequency offsets in the form of 6900 {u, s-1, -1}, {u, s0, 0} and {u, s+1, +1}. 6901 6902 There are 3xNum_signature uncertainty windows grouped into Num_signature triplets. Each uncertainty 6903 window contains correlation values for UN_WINDOW offsets. 6904 6905 As stated in chapter 13.3.8, the 2 sequence contained in the preamble for RACH format 2 are separated 6906 after the filtering/decimation block. Then the separated 2 sequences will go through the processing blocks 6907 afterwards repectively like the signal received from 2 different Rx antenna until they arrive at the "compute 6908 metrics" . Incoherent Combining is introduced to combine samples from the 2 sequences together to 6909 construct the final decision metric, which is elaborated in 13.4.6.2. 6910

6911 In the meanwhile, because RACH format 2 is introduced since LA4.0.1, so N in the following content refers 6912 to the degree of freedom order of the Chi-square distribution instead of the exact Rx antenna number. For 6913 instance , as to format 0, N is just the Rx antenna number. But for format 2, N is double of Rx antenna 6914 number. Whereas, for the sake of simplicity, it will not explicitly distguish N with the exact Rx antenna 6915 number in the specification. 6916 6917

13.4.2 GENERATE FREQUENCY DOMAIN ROOT ZC SEQUENCE 6918

For each of the (up to 64) roots, we generate the conjugate DFT of the ZC sequence and multiply it by the 6919 extracted RACH sequence of section 13.3.11. This section describes the method for generating the 6920 sequence. 6921 6922 According to TS 36.211, a random access preambles xu,v(k) is a CAZAC sequence generated by cyclic 6923 shifting the u-th root ZC sequence xu(k), where the root sequence xu(k) is defined by 6924

ZCN/)n(uniu e)n(x 1+π−= , n = 0 .. Nzc -1, and Nzc=839 6925

6926 Thus, the frequency domain root sequence is: 6927 6928

∑−

=

−=1

0

/2)()(ZC

zc

N

k

Nnkiuu enxkX π . k = 0 .. Nzc -1, (13-4) 6929

6930 6931

A straightforward implementation will require performing an 839 point DFT to generate the frequency 6932 domain root sequence Xu(k). However it can be shown that equation (13-4) can be evaluated in closed 6933 form: 6934 6935

2uu)k(k

N2π

i

u

knN2π

i-1N

0nuu

ZCZC

ZC

(0)eX(n)ex(k)X

+−

=

== ∑ (13-5) 6936

6937

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 13-4







6938

where the calculation of u

)uk(k2

+ Mod NZC is performed over GF(NZC). For example, if k = 1, u = 100, then, 6939

u)uk(k

2

+ mod NZC = 592, not 0.505. The factor (0)Xu is a complex constant depending only on u and can 6940

be ignored for correlation processing. The algorithm to compute the DFT of the ZC sequence is thus 6941

reduced to finding an algorithm for generating the indices ZCNMod2u

)uk(k)k(n

+= for a given u and using 6942

these to index into a table of precomputed sinusoids. 6943 6944

We use an iterative method to generate the indices. Given numbers (in this case integers) A, B, and C, the 6945 discrete function 6946 6947

CBkAkx 2k ++= k = 0, 1, ..... 6948

6949 is a solution of the first order system of linear difference equations: 6950 6951

att

txx

k1k

kk1k

+=+=

+

+ with initial conditions Cx0 = , BAt0 += , and 2Aa = 6952

6953 6954

If we expand the index k21

k2u1

2uu)k(k

n(k) 2 +=+= and identify coefficients of the powers of k we have 6955

6956

2u1

A = , 21

B = , and 0C = 6957

6958 The update variable, ZCNModu/2Aa µ=== 1 where µ is the integer that is the multiplicative inverse of 6959

ZCNModu that is, ZCNMod1u =µ . With these results, we can recursively calculate the index sequence 6960

)k(n , with the following algorithm: 6961

6962 1) For a given ZC root, u, the multiplicative inverse, µ, is calculated by the DSP. 6963 2) Initialize the index, n(0), the auxiliary variable, t(0), and the update variable, 6964 6965

n = 0, t = 2

1 µ+ Mod Nzc, a = µ 6966

6967 3) The successive indices, n, are generated recursively as 6968 6969

t = t + a 6970 t = Mod( t, NZC) 6971 6972 n = n + t 6973 n = Mod( n, NZC) 6974

6975

13.4.3 MULTIPLY RACH SIGNAL WITH ZC SEQUENCE 6976

For a given ZC root, u, we have generated the indices into a look-up table of complex phasors that 6977 comprise the DFT of the ZC sequence: 6978 6979

zcN/n(k)-i2u e)k(X π= where n(k) is the sequence of indices generated in the last section. 6980

6981

Supprimé : 4.0.1

Supprimé : 2







zcN/n(k)-i2e π is implemented as a table of sines and cosines in the same 10-bit format as the bandshift 6982 phasors described in section 13.3.5. The 1024-point input buffer is multiplied by this sequence. Note that 6983 the first sample in the 1024 point FFT buffer to be multiplied is fftbuf1024[605] i.e.: 6984 6985 for( k=0; k<839; k++) 6986

fftbuf1024[ (605 + k) Mod 1024 ] *= X u[k]; 6987 6988 6989

13.4.4 1024-POINT FFT 6990

From the last section, the extracted ZC sequence has been recentered into the 1024 point FFT buffer and 6991 multiplied by the conjugate of the root ZC sequence. We now take a forward 1024 point FFT. One would 6992 expect an inverse FFT at this point but since the shifted ZC sequence is defined as: 6993 6994

xu, v (n) = xu((n + Nshift ) ModNzc ) 6995

6996

shift is backwards. Note that we are taking a 1024-point FFT instead of an 839-point FFT. This has the 6997 effect of resampling the correlation function so that a peak associated with a shift of, say, 100 will actually 6998

appear at correlation bin 122839

1024100 ≈× . 6999

7000 The output of the 1024-point FFT is the circular correlation of the received RACH signal with the specified 7001 root u ZC sequence resampled to 1024 points. 7002 7003

13.4.5 RESCALE/EQUALIZE 7004

13.4.5.1 RESCALE FOR N=2 CASE 7005

Inputs : 7006 7007 Nfa 4-bit shift amount for 2048-FFT stage, antenna A. 7008 Nfb 4-bit shift amount for 2048-FFT stage, antenna B. 7009 Nia 4-bit shift amount for 1024-FFT stage, antenna A. 7010 Nib 4-bit shift amount for 1024-FFT stage, antenna B. 7011 Xa 12-bit complex output of 1024 point FFT for antenna A. 7012 Xb 12-bit complex output of 1024 point FFT for antenna B. 7013 7014 7015 Overview: 7016 7017 There were two scaling operations in the RACH processing flow that operated on each antenna separately. 7018 Here we calculate the overall scale and "equalize" the two antennas in the case where they were scaled 7019 differently. 7020 7021 In the case where there is only one antenna, this function is bypassed – no rescaling is done. 7022 7023 7024 ShiftA = Nfa + Nia Combine front-end shifts for each antenna. 7025 ShiftB = Nfb + Nib ShiftA and ShiftB 5 bit unsigned 7026 7027 if( ShiftA > ShiftB ) Find the larger shift. Then compute the relative shift 7028

{ ShiftB = 0; between the two antennas. The antenna with the 7029 ShiftA -= ShiftB; larger shift will be shifted right relative to the antenna 7030 with the smaller shift. 7031

Supprimé : 4.0.1

Supprimé : 2







if( ShiftA > 11 ) Limit the shift to 11 bits. 7032 ShiftA = 11; 7033

7034 } 7035 else 7036 { ShiftA = 0; 7037

ShiftB -= ShiftA; 7038 7039

if( ShiftB > 11 ) 7040 ShiftB = 11; 7041

} 7042 7043 7044 for(k=0; k<1024; k++) Shift the two channels. Then proceed to metric 7045 { Xa[k].real >>= ShiftA calculation. 7046 Xa[k].imag >>= ShiftA 7047 Xb[k].real >>= ShiftB 7048 Xb[k].imag >>= ShiftB 7049 } 7050 7051 7052 Outputs: 7053 7054 Xa[0..1023] 12-bit complex rescaled output of 1024 point FFT for antenna A 7055 Xb[0..1023] 12-bit complex rescaled output of 1024 point FFT for antenna B 7056


Inputs : 7058 7059 Nfn 4-bit shift amount for 2048-FFT stage, antenna n. 7060 Nin 4-bit shift amount for 1024-FFT stage, antenna n. 7061 Xn 12-bit complex output of 1024 point FFT for antenna n. 7062 ,where n is {0,1,2,3} representing the 4 RX antennas {A,B,C,D} respectively 7063 7064 7065 Overview: 7066 7067 There were four scaling operations in the RACH processing flow that operated on each antenna separately. 7068 Here we calculate the overall scale and "equalize" the four antennas in the case where they were scaled 7069 differently. 7070 7071 In the case where there is only one antenna, this function is bypassed – no rescaling is done. 7072 7073 The pseudo-code for this operation is generalized with RX antenna number NRX, with NRX =4 for the case 7074 treated here. 7075 7076 7077 for (n=0;n< N;n++) 7078 { 7079

Shift[n] = Nfn + Nin; Combine front-end shifts for each antenna. 7080 Shift[n] 5 bit unsigned 7081 if n==0 Find the minimum shift values 7082

ShiftMIN = a[n]; 7083 else 7084

if(ShiftMIN >a[n]) 7085 ShiftMIN = a[n]; 7086

} 7087

7088

Supprimé : 4.0.1

Supprimé : 2







for (n=0;n< N-1;n++) Then compute the relative shift to the minimum one for each 7089 { antenna. It is obvious that the one exhibiting the miminum shift 7090

Shift[n] -= ShiftMIN; yields a zero.The antenna with the largershift will be shifted right 7091 relative to the antenna with the minimum shift. 7092

if( Shift[n] > 11 ) Limit the shift to 11 bits. 7093 Shift[n] = 11; 7094

7095 7096

for(k=0; k<1024; k++) Shift the each channel. Then proceed to metric calculation. 7097 { X[n][k].real >>= Shift[n]; 7098 X[n][k].imag >>= Shift[n]; 7099 } 7100 7101 } 7102 7103 7104 Outputs: 7105 7106 Xn[0..1023] 12-bit complex rescaled output of 1024 point FFT for antenna n, 7107 ,where n is {0,1,2,3} representing the 4 RX antennas {A,B,C,D} respectively 7108 7109


With respect to the discussion in 13.3.9 and13.3.12, N = 8 corresponds to 4 Rx diversity with RACH format 7111 2. 7112 7113 Inputs : 7114 7115 Nfn the nth 4-bit shift amount for 2048-FFT stage. 7116 Nin the nth 4-bit shift amount for 1024-FFT stage. 7117 Xn the nth 12-bit complex output of 1024 point FFT. 7118 ,where n is {0,1,..,7} representing the sequences contained in the signal of RACH format 2 from 4 RX 7119 antennas {A,B,C,D} respectively 7120 7121 7122 Overview: 7123 7124 There were eight scaling operations in the RACH processing flow that operated on each antenna 7125 separately. Here we calculate the overall scale and "equalize" the eight antennas in the case where they 7126 were scaled differently. 7127 7128 In the case where there is only one antenna, this function is bypassed – no rescaling is done. 7129 7130 The pseudo-code for this operation is generalized with RX antenna number NRX, with NRX =4 for the case 7131

treated here. Since there are 2 sequences in the RACH signal of RACH format 2, N=2×NRX=8. 7132 7133 7134 for (n=0;n< N;n++) 7135 { 7136

Shift[n] = Nfn + Nin; Combine front-end shifts for each sequence from each antenna. 7137 Shift[n] 5 bit unsigned 7138 if n==0 Find the minimum shift values 7139

ShiftMIN = a[n]; 7140 else 7141

if(ShiftMIN >a[n]) 7142 ShiftMIN = a[n]; 7143

} 7144

Supprimé : 4.0.1

Supprimé : 2







7145 for (n=0;n< NRX-1;n++) Then compute the relative shift to the minimum one for each 7146 { sequence from each antenna. It is obvious that the one exhibiting 7147

Shift[n] -= ShiftMIN; the miminum shift yields a zero.The antenna with the largershift will 7148 be shifted right relative to the antenna with the minimum shift. 7149

if( Shift[n] > 11 ) Limit the shift to 11 bits. 7150 Shift[n] = 11; 7151

7152 7153

for(k=0; k<1024; k++) Shift the each channel. Then proceed to metric calculation. 7154 { X[n][k].real >>= Shift[n]; 7155 X[n][k].imag >>= Shift[n]; 7156 } 7157 7158 } 7159 7160 7161 Outputs: 7162 7163 Xn[0..1023] the nth 12-bit complex rescaled output of 1024 point FFT for antenna n, 7164 , where n is {0,1,..,7} representing the sequences contained in the signal of RACH format 2 from 4 RX 7165 antennas {A,B,C,D} respectively. 7166 7167 7168 7169 7170

13.4.6 COMPUTE METRICS 7171

13.4.6.1 METRICS FOR FORMAT 0,1 7172

13.4.6.1.1 Metrics of N=2 case 7173

Inputs: 7174 7175 Xa 12-bit complex rescaled output of 1024 point FFT for antenna A 7176 Xb 12-bit complex rescaled output of 1024 point FFT for antenna B 7177 7178 Overview : 7179 7180 The rescaled outputs of the 1024-point FFT are the (interpolated) correlation function of the RACH input 7181 data with a ZC sequence for each antenna. Here the two antennas are combined to form a single array of 7182 24-bit unsigned correlation metrics: 7183 7184

+= 22

2

1bak XXM k = 0, 1, …, 1023 7185

7186

In the single diversity case the metric is 2

XMk = k = 0, 1, …, 1023 7187

7188 Processing: 7189 7190 par = Xa[k].real x Xa[k].real Square the real and imaginary parts of the 7191 pai = Xa[k].imag x Xa[k].imag 12-bit complex correlation data to form 7192 pbr = Xb[k].real x Xb[k].real 23-bit unsigned products. 7193 pbi = Xb[k].imag x Xb[k].imag 7194 7195

Supprimé : 4.0.1

Supprimé : 2







7196 sa = par + pai Sum the real and imaginary parts of the 7197 sb = pbr + pbi products for each antenna to form 24-bit 7198 unsigned magnitudes. 7199 7200 7201 M[k] = ( sa + sb )/2 Average the squared magnitudes of each 7202

antenna to form the 24-bit unsigned 7203 correlation metrics. Divide by 2 is implemented with shift 7204 and truncate. 7205

7206 For the single diversity case: 7207 7208 M[k] = sa or Depending on mode A or mode B. 7209 M[k] = sb 7210 7211 7212 Outputs: 7213 7214 M[0..1023] 24-bit unsigned metrics to be searched for correlation peaks. 7215 7216 7217

13.4.6.1.2 Metrics of N=4 case 7218

Inputs: 7219 7220 Xa 12-bit complex rescaled output of 1024 point FFT for antenna A 7221 Xb 12-bit complex rescaled output of 1024 point FFT for antenna B 7222 Xc 12-bit complex rescaled output of 1024 point FFT for antenna C 7223 Xd 12-bit complex rescaled output of 1024 point FFT for antenna D 7224 7225 Overview : 7226 7227 The rescaled outputs of the 1024-point FFT are the (interpolated) correlation function of the RACH input 7228 data with a ZC sequence for each antenna. Here the four antennas are combined to form a single array of 7229 24-bit unsigned correlation metrics: 7230 7231

( )2 2 2 21

4k a b c dM X X X X= + + + k = 0, 1, …, 1023 7232

7233 7234 Processing: 7235 7236 par = Xa[k].real x Xa[k].real Square the real and imaginary parts of the 7237 pai = Xa[k].imag x Xa[k].imag 12-bit complex correlation data to form 7238 pbr = Xb[k].real x Xb[k].real 23-bit unsigned products. 7239 pbi = Xb[k].imag x Xb[k].imag 7240 pcr = Xc[k].real x Xc[k].real 7241 pci = Xc[k].imag x Xc[k].imag 7242 pdr = Xd[k].real x Xd[k].real 7243 pdi = Xd[k].imag x Xd[k].imag 7244 7245 sa = par + pai Sum the real and imaginary parts of the 7246 sb = pbr + pbi products for each antenna to form 24-bit 7247 sc = pcr + pci unsigned magnitudes. 7248 sd = pdr + pdi 7249 7250

Supprimé : 4.0.1

Supprimé : 2








M[k] = ( sa + sb+sc+sd )/4 Average the squared magnitudes of each 7251 antenna to form the 24-bit unsigned 7252 correlation metrics. Divide by 4 is implemented with shift 7253 and truncate. 7254

7255 Outputs: 7256 7257 M[0..1023] 24-bit unsigned metrics to be searched for correlation peaks 7258

13.4.6.2 METRICS FOR FORMAT 2 7259

Inputs: 7260 7261 Xnm 12-bit complex rescaled output of 1024 point FFT for the mth sequence of the preamble of format 2 7262

from the nth Rx antenna A, where m=[1,2], n = [1,…, RxN ], RxN is the Rx antenna number 7263

7264 Overview : 7265 7266 The rescaled outputs of the 1024-point FFT are the (interpolated) correlation function of the RACH input 7267 data with a ZC sequence for each antenna. Here the four antennas and the 2 sequence of the preamble of 7268 format 2 are combined to form a single array of 24-bit unsigned correlation metrics: 7269 7270

22

1 1

1

2

RXN

k nmm nRX

M XN = =

= ∑∑ k = 0, 1, …, 1023 7271

7272 7273 Processing: 7274 7275 M[k] = 0; 7276 7277 7278 for (n=0;n< NRX;n++) 7279 { 7280 s = 0; 7281 for (m=0;n< 2;k++) 7282

{ 7283 7284

pr = X[n][m][k].real × X[n][m][k].real Square the real and imaginary parts 7285

pi = X[n][m][k].imag × X[n][m][k].imag of the 12-bit complex correlation to 7286 form 23-bit unsigned products. 7287 7288

s += (pr + pi) Sum the real and imaginary parts of the 7289 products for each antenna to form 24-bit 7290 unsigned magnitudes. 7291

} 7292 7293

M[k] += s Sum the real and imaginary parts of the 7294 products for all antennae 7295 7296

} 7297 7298 M[k] = M[k]/n/m Average the squared magnitudes of each 7299

antenna to form the 24-bit unsigned 7300

correlation metrics. Divide by n×m is implemented with 7301 shift and truncate. 7302

7303

Supprimé : 4.0.1

Supprimé : 2







Outputs: 7304 7305 M[0..1023] 24-bit unsigned metrics to be searched for correlation peaks 7306 7307

The two sequences contained in the preamble of RACH format 2 are kept separated since the "sequence 7308

separation" and come together in metric combining as well as the other antennas. 7309

In 2Rx diversity, the 4 data paths are equalized and combined into a singlemetric as if this were a 4-7310 diversity Format 0 RACH. After the metric calculation processing is the same as before. This is obviously 7311 done for each root. 7312 7313 Thus from the output of filtering and decimation onwards, this method is the same as if the number of 7314

antennas was doubled. As to the 2Rx diversity, the metrics kM for format 2 have a Chi-squared distribution 7315

with 4 degree of freedom order whereas it is 2 degree of freedom oder for format 0. 7316 7317 It is called incoherent combining because the squared magnitudes of the Seq1 and Seq2are being 7318 summed instead of the bipolar complex samples. Combining the squared magnitudes does not give as 7319 much gain as combining the raw samples in the case where the RACH signal is static but it removes the 7320 phase from the two sequences so that it is more robust when the phases are changing across the two 7321 sequences as when there is a frequency offset and/or fading. 7322 7323 Once decision metrics have been computed, following bank-end processing is the same for both format 0 7324 and format 2, except that detection threshold should be set according to the doubled degree of freedom 7325 order in the format 2 case. 7326 7327

13.4.7 COMPUTE HISTOGRAM 7328

Supprimé : 4.0.1

Supprimé : 2







7329 Inputs: 7330 7331 M[0..1023] 24-bit unsigned metrics to be searched for correlation peaks. 7332 7333 7334 Overview: 7335 7336 In order to estimate the noise floor of the correlation metrics, we generate a histogram of the metrics to find 7337 the location of the first quartile. We compute the histogram using the bit location of the MSB of each metric 7338 so that each bin actually corresponds to a value of 2msb. Since each metric is at most 24 bits and there are 7339 1024 metrics, the histogram will be an array of 24 10-bit unsigned integers. 7340 7341 7342 Processing: 7343 7344 MagHist[0..23] <- 0 Clear the 24 element, 10-bit unsigned histogram 7345 kmax = 0 array and initialize k max. 7346 kmax is 5 bit unsigned 7347 7348 for k in [0..1023] ... do 7349 7350 index = HighOrderBit(M k ) The histogram index is the position of the 1st non- 7351

zero bit in the metric. If Mk is zero, return zero. 7352 index is 5 bits unsigned. 7353

7354 MagHist[index]++ Increment the histogram bin for this index. 7355 7356 if( index > k max ) Keep track of the maximum index. 7357 k max = index 7358 7359 Output: 7360 7361 MagHist[0..23] 24 10-bit unsigned values representing the histogram of metric values. 7362

MagHist[k] is the number of metrics whose values lie between 2k-1 and 2k. 7363 7364

kmax 5 bit unsigned index of the maximum histogram index which is a bound on the 7365 maximum metric: MaxMetric < 2kmax 7366 7367 7368

13.4.8 ANALYZE CDF 7369

7370 Input: 7371 7372 MagHist[0..23] 24 10-bit unsigned values representing the histogram of metric values. 7373 7374 7375 Overview: 7376 7377 Ultimately, we want to find the metric value that corresponds to the first quartile of the correlation metrics. 7378 We accumulate the magnitude histogram to compute the cumulative distribution function (CDF). Since 7379 there are 1024 metrics, the first quartile will be the value where the accumulated histogram equals 256. 7380 We find the values where the CDF brackets this number. Note that we don't need to compute the entire 7381 CDF. 7382 7383 7384 Processing: 7385

Supprimé : 4.0.1

Supprimé : 2







7386 cdf2 = 0; Compute the cumulative distribution function from 7387 cdf1 = 0; the histogram and find the index where the CDF 7388 k1 = 0; is greater than or equal to 256 7389 for( k=0; k<24; k++) 7390 { cdf2 += MagHist[k]; 7391 if( cdf2 >= 256 )break; 7392 7393 if( cdf1 != cdf2 ) Remember the previous index and value provided 7394 { cdf1 = cdf2; that it is smaller than the current value. 7395 k1 = k; 7396 } 7397 } 7398 7399 k2 = k; 7400 7401 7402 7403 Output: 7404 7405 cdf2 10-bit unsigned: the first value of the CDF that is ≥ 256 7406 cdf1 10-bit unsigned: the previous value of CDF that is < cdf2 7407 k2 5-bit unsigned index of cdf2 7408 k1 5-bit unsigned index of cdf1 7409 7410

13.4.9 HANDLE SPECIAL CASES 7411

7412 Inputs: 7413 7414 cdf2 10-bit unsigned: the first value of the CDF that is ≥ 256 7415 k2 5-bit unsigned index of cdf2 7416 7417 if( k2 == 0 ) 7418 { q2 = RoundStrip( cdf2, 4 ) 7419 n2 = 0 7420 Goto compute_Threshold 7421 } 7422 7423 Output: 7424 7425 q2 17-bit unsigned 7426 n2 6-bit signed 7427 7428

13.4.10 COMPUTE FIRST QUARTILE 7429

7430 Inputs: 7431 7432 cdf2 11-bit unsigned (the first value of the CDF that is ≥ 256), 256 ≤ cdf2 ≤ 1024 7433 cdf1 8-bit unsigned (the previous value of CDF that is < cdf2), 0 ≤ cdf1 ≤ 255 7434 k2 5-bit unsigned index of cdf2, 1 ≤ k2 ≤ 24 7435 k1 5-bit unsigned index of cdf1, 0 ≤ k1 ≤ 23 7436 Temif 16-bit unsigned: the ratio of the normalized Pfa threshold and the 1st quartile threshold scaled by 211 7437

This value is written by the DSP to register field DETECT_TH. 7438 7439 7440

Supprimé : 4.0.1

Supprimé : 2







112×=q

pfaemif

X

TT )

)

where pfaT)

is the normalized Pfa threshold and qX)

is the normalize 1st quartile value. 7441

7442 7443 Overview: 7444 7445 Interpolate between cdf1 and cdf2 to find the location of the 1st quartile of the data. From this value an 7446 estimate of the noise power is computed. The normalized Pfa threshold is scaled by this noise power to get 7447 the threshold for the current metrics. The interpolated value xq is calculated thus: 7448 7449

( ) 1256

12

1121 −

−−

−+=cdfcdf

cdfxxxxq where: x1 = 2k1 and x2 = 2k2 7450

7451 The last -1 is to adjust the quartile position relative to the CDF value. For the following calculations we 7452 rewrite the equation above as: 7453 7454 7455

( )( )

−−−

+=∆

12

12561212 1

cdfcdf

cdfx

kk

q where: ∆k = k2 – k1 7456

7457 7458 7459 7460 Processing : 7461 7462 ∆k = k2 – k1 Compute the difference between k2 and k1. 7463 5-bit unsigned: 1 ≤ ∆k ≤ 24 7464 7465 num = 256 − cdf1 Compute numerator and denominator. 7466 num: 9-bit unsigned 1 ≤ num ≤ 256 7467 den = cdf2 − cdf1 den: 11-bit unsigned 1 ≤ den ≤ 1024 7468 7469 en = HighOrderBit(num) Compute exponents of numerator and denominator. 7470 ed = HighOrderBit(den) en: 4-bit unsigned 1 ≤ en ≤ 9 7471 ed: 4-bit unsigned 1 ≤ ed ≤ 11 7472 7473 n1 = 15 + ed – en Scale the numerator by n1 to insure that the quotient will have 16 7474 bits of precision. 7475

1n2nummnu ×=′ n1 : 5-bit unsigned 15 ≤ n1 ≤ 24 7476 num’: 26-bit unsigned 215 ≤ num’ ≤ 225 7477 7478 q = Div30by16( num’, den ) Perform the division 7479 q: 16-bit unsigned 214 ≤ q ≤ 216 7480 7481 7482 At this point 7483 7484

( )[ ] 12q2112x 11 ∆kn∆kkq −×−+= +−− 7485

7486 kqqq ∆−×−= 21 q1: 16-bit unsigned 213 ≤ q1 ≤ 216 7487

7488 At this point 7489 7490

211 2q12x 1k

qkn +−×+−= 7491

For the following calculations: q2 is 17-bits unsigned 7492

Supprimé : 4.0.1

Supprimé : 2









n2 is 5-bits signed 7493 7494 7495 if k 1 = 0 7496 q2 = q1 213 < q2 ≤ 216 7497 n2 = n1 – k2 −9 ≤ n2 ≤ 23 7498 goto xq_done xq_done is a label at the end of this procedure. 7499 7500

m = n1 + k1 – k2 – 15 m is used to balance the relative magnitude of q1 and 12k 7501 m: 6-bit signed −23 ≤ m ≤ 8. 7502 7503

If m > 0 This is the case where 12k is larger 7504 n2 = 15 – k1 −8 ≤ n2 ≤ 14 7505 q2 = 215 + q1×2−m 215 < q2 < 216 7506

if( n2 ≥ 0 )q2 −= 22n 214 < q2 < 216 7507 7508 goto scale_q2 7509 7510 7511 Else if m ≤ 0 7512 7513 n2 = n1 – k2 −9 ≤ n2 ≤ 14 7514 q2 = q1 213 < q2 < 214 7515

if( n2 ≥0 )q2 −= 22n 7516 n = n2 + k1 n: 4-bit signed −8 ≤ n ≤ 15 7517 if( n ≥ 0 )q2 += 2n 7518 Fall through to label scale_q2 7519 At this point 214 < q2 < 217 7520 scale_q2: 7521 7522 if( q2 & 0xF0000 ) If q2 is ≥ 216, divide by 2 (shift right 1) and decrement n2 by 1. 7523

{ q2 >>= 1 7524 n2 −= 1 7525 } 7526 7527 7528

xq_done: 7529 7530 while( (q2 & 0x8000) == 0 ) 7531

{ q2 <<= 1 Shift q2 to 16 bits and adjust the scale factor. 7532 n2++ 7533 } 7534

7535 7536 215 ≤ q2 ≤ 216 q2, which is unsigned 17 bits, has been scaled to 16 bits. 7537 −9 ≤ n2 ≤ 25 7538 7539 7540

At this point 222n

q qx −×= 7541

7542

13.4.11 COMPUTE THRESHOLD 7543

7544 et = HighOrderBit( DETECT_TH) et ≤ 16 7545 7546 t1 = q2×DETECT_TH Multiply 16-bit q2 by the 16-bit detection threshold from Signature table. 7547 7548

Supprimé : 4.0.1

Supprimé : 2







t2 = RoundStrip( t1 , et ) Shift the product et bits to the right and round. 7549 t2: 16 bits unsigned 7550 214 ≤ t2 < 216 7551 7552 n3 = et – n2 Initialize exponent. n3: 6-bit signed 7553 −24 ≤ n3 ≤ 25 7554 7555 7556 if( ( t2 & 0x8000 ) == 0 ) Adjust t2 into 16 bits 7557

{ t2 <<= 1 t2: 16-bits unsigned 215 ≤ t2 ≤ 216 7558 n3 −= 1 n3: 6-bit signed -15 ≤ n3 ≤ 25 7559 } 7560

7561 if( n3 < kmax – 17 ) If the value of the maximum metric is too large relative to the threshold, 7562 n3 = kmax – 17 increase the threshold so the maximum peak is not larger than 16 bits. 7563 7564 7565 if( n3 ≥ 19) The threshold is too large. All peaks will be below threshold. 7566

{ r = 0 7567 nr = 0 7568 goto Metric Scaling 7569 } 7570 7571 if( n3 < −4 ) Threshold is too low. All peaks will be above threshold. 7572 { r = 1 7573 nr = 0 7574 goto Metric Scaling 7575 } 7576 7577

7578 At this point the false alarm threshold is 7579 7580

faefa tT 22 ×= where efa = n3 − 10 7581

7582 t2 and efa are saved and returned in the last entry of the signature block as exp and mantissa (see section 7583 13.4.13). 7584 7585

13.4.12 COMPUTE RECIPROCAL OF THRESHOLD 7586

Inputs: 7587 7588 t2 16-bits unsigned 215 ≤ t2 < 216 7589 n3 6-bit signed −4 ≤ n3 ≤ 18 7590 7591 r = Div30by16( 229, t2 ) Reciprocal of the false alarm threshold scaled by 229 7592

r: 16 bits unsigned 213 ≤ r ≤ 214 7593 7594 nr = n3 + 4 Adjust the exponent to make it positive. 7595 nr: 5-bits unsigned 0 ≤ nr ≤ 22 7596 7597 Output: 7598 7599

r 16 bit unsigned reciprocal of threshold such that: 1521 −−×= rn

fa

rT

7600

Supprimé : 4.0.1

Supprimé : 2








13.4.13 PEAK DETECTION AND SCALING 7601

7602 Inputs: 7603 7604 M[0..1023] 24-bit unsigned metrics to be searched for correlation peaks. 7605 Tfa 24-bit threshold calculated from the last section. 7606 Root_block_shift Table of search windows from SRIO-A. 7607 7608 7609 Overview: 7610 7611 For each signature, the metrics that are in the specified search windows are thresholded and collected. To 7612 insure that we get potential peaks from each window we restrict the search in the following manner: 7613 7614

1. Pick the three highest peaks over threshold in the target window UWS_0. 7615 2. Pick the two highest peaks over threshold in the +∆f window UWS_1. 7616 3. Pick the two highest peaks over threshold in the −∆f window UWS_2. 7617

7618 The 8’th entry in the returned peak table will be set to the calculated Tfa. 7619 7620 7621 Processing: 7622 7623 LRACH_Peak_Structure ← 0 Clear the peak structure array. 7624 7625 For each of the three search windows specified by USW_0, USW_1, and USW_2 search UN_WINDOW 7626 consecutive metrics for peaks over threshold. 7627 7628

7629 Threshold the metric in each of the three search windows, for example for UWS_0: 7630 7631 index = UWS_0 7632 for( k=0; k<UN_WINDOW; k++) 7633 { if( Metric[index] > Tfa ) 7634 Put_index_in_peak_list This is only schematic. For UWS_0 7635 { Tmp[k] = Metric[k]; only 3 peaks are collected. For UWS_1 7636 Offset[k] = index – UWS_0; and UWS_2, 2 peaks are collected. 7637 } Note that the offset is relative to 7638 the window start. 7639 index += 1; 7640 if( index >= 1024 )index = 0; 7641 } 7642 7643 Tmp[] is the 24-bit scratch array to hold pre-scaled peaks. 7644 Make sure to wrap back to 0 if the window straddles the end of the metric array boundary. 7645 7646 Collect the three highest peaks in UWS_0 then 7647 Collect the two highest peaks in UWS_1 then 7648 Collect the two highest peaks in UWS_2. 7649 7650

. .. . ... ..

UWS_0 UWS_1 UWS_2

UN_WINDOW UN_WINDOW UN_WINDOW

Supprimé : 4.0.1

Supprimé : 2







7651 for( k=0; k<7; k++) For each of the detected peaks, divide by the threshold 7652 { x = Tmp[k]; and saturate to 16 bits. 7653 Peak[k] = Saturate( x/Tfa, 16 ) 7654 } 7655 7656 p = M×r Multiply 24-bit metric by 17-bit reciprocal of threshold. 7657 7658 y = RoundLte( p, nr ) Scale the product by nr to check for detection. 7659 y: 24-bits unsigned. 7660 7661 if( (y & 0xFF8000) == 0 ) Threshold test: if the high order 9 bits are zero then this peak is 7662 { ignore this metric } under threshold – ignore it. 7663 7664 7665 peak = RoundLte( y, 15 – nfrac ) If the metric passed the threshold test above perform final scaling 7666 and saturation into 16 bits. nfrac is a constant = 4. Hence 7667 15 – nfrac = 11. 7668 Saturate peak to 0xFFFF 7669 7670 7671 Outputs: 7672

Peak[k] k = 0,…, 6 The 7 16-bit scaled, detected peaks. 7673 Offset[k] k = 0, …, 6 The corresponding offsets from the start of the search window. 7674 mantissa 16 bit threshold mantissa. 7675 exp 8 bit signed threshold exponent. 7676 7677

The output values are written to SRIO-A in the following order: 7678

7679

Table 13-25 – RACH Signature Block 7680 7681 7682 7683

7684

7685

7686

7687

7688

7689

7690

7691

The peaks in each set are ordered by magnitude. I.e. UWS_0 peaks are ordered separately from UWS_1 7692 etc. If no detections are made all the peak entries will be zero but the threshold will still be present ( as a 7693 diagnostic ). 7694

7695

D is a one bit detection flag: 1 => detected, 0 => no detection 7696

H is a two bit frequency hypothesis indicator: 7697

“00” => 0 Delta Freq window 7698

“01” => + Delta Freq window 7699

mantissa

D

D

D

D

D

D

D

H

H

H

H

H

H

H

Offset[0]

Offset[1]

Offset[4]

Offset[3]

Offset[2]

Offset[5]

Offset[6]

Peak[0]

Peak[1]

Peak[4]

Peak[3]

Peak[2]

Peak[5]

Peak[6] UWS_2

UWS_1

UWS_0

10 bits 16 bits

exp MsgId Threshold

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 22







“11” => － Delta Freq window 7700

“10” reserved 7701

7702

The last word in the block contains the 8-bit Message Id from the A_LRACK_CTRL word and the computed 7703 threshold. The threshold is returned as a 16 bit mantissa and an 8-bit exponent: 7704

7705

Threshold = mantissa×2exp (13-6) 7706

7707

For further details of these variables see LTE Channel Element DSP Software Interface Manual ( DSIM). 7708 7709

13.4.14 OTHER RACH OUTPUTS 7710

7711

The following are diagnostic outputs from RACH processing. 7712

7713

Table 13-26 – Front End Overflow Counts 7714 7715

Parameter Type Description NumOvrFlowIn 16-bit unsigned The number of words that were saturated during

bit selection for the I channel of the nthantenna A. NumOvrFlowQn 16-bit unsigned The number of words that were saturated during

bit selection for the Q channel of the nthantenna. 7716 7717

Table 13-27 – Input Magnitude Value 7718 7719

Parameter Type Description MaxMagIn 12-bit unsigned Maximum magnitude of I channel for the nth

antenna. MaxMagQn 12-bit unsigned Maximum magnitude of Q channel for the nth

antenna. 7720 7721 7722

Table 13-28 – Scaling Value Masks 7723 7724 7725

Parameter Type Description ValMask2048n 12-bit unsigned Value mask going into 2048 FFT for the nth antenna ValMask2048n 12-bit unsigned Value mask going into 2048 FFT for the nth antenna 7726 7727 7728 The following table is a list of indicator bits. Each is a single bit returned in a status word and if set, 7729 indicates a processing error. 7730 7731

Table 13-29 – Single Bit Diagnostic Indicators 7732 7733

Indicator Bits Description OverFlowIn Bit selection overflow for I channel of the nth antenna OverFlowQn Bit selection overflow for Q channel of the nth antenna

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 23

Supprimé : 24

Supprimé : 25

Supprimé : 26







UnderFlown The nth antenna is enabled but the data is too low to process.

CWV: ValMask2048n and/or ValMask1024n was zero. 7734 7735 7736

13.5. LOW MOBILITY THRESHOLD CALCULATIONS 7737

For the low mobility case we search for detections in 64 windows of length Ncs corresponding to UWS_0 of 7738 section 13.4.13. The RACH detection threshold, DETECT_TH, which is passed to the HW from the DSP, 7739 is a function of the diversity mode, the probability of false alarm and the number of independent correlation 7740 bins that are being searched over. 7741 7742 The length of each search window is determined by the Ncs configuration (Table 13-2) that has been 7743 specified by higher layers so the total number of bins searched will be csNM ×= 64 . 7744

7745

13.5.1 THEORY OF THRESHOLD CALCULATION 7746

We are going to search over a set of M metrics which are real-valued random variables denoted by Y and 7747 declare a detection when the value of Y is above a given threshold, T. The probability of false alarm, Pfa, is 7748 defined as the probability that one or more of the metrics is above threshold in the absence of a signal. 7749 7750 Pfa = Prob[ "one or more metrics" ≥ T ] 7751 7752 The complement of "one or more metrics ≥ threshold" is that all metrics are below threshold so that: 7753 7754 Pfa = Prob[ "one or more metrics" ≥ T ] = 1 − Prob[ "all metrics” < T ] (13-7) 7755 7756 The probability that one metric is below threshold is: 7757 7758 P[ Y < T ] = F( T ) 7759 7760 where F(T) is the single metric CDF. The probability that all M metrics are below threshold assuming the 7761 metrics are statistically independent is: 7762 7763 P[ "all metrics“ < T ] = [ F ( T ) ]M 7764 7765 Then from equation (13-7): 7766 7767 Pfa = 1 − [ F ( T ) ]M = 1 − ( 1 − Q ( T ) )M 7768 7769 Solving for Q( T ) as a function of Pfa: 7770 7771 Q(T ) = 1 − [ 1 − Pfa ]

1/M (13-8) 7772 7773 The false alarm threshold is obtained by solving equation (13-8) for T: 7774 7775 Tfa = Q−1 ( 1 − [ 1 − Pfa ]

1/M ) 7776 7777 7778 This is usually a transcendental equation and must be solved numerically. 7779

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Police :NonGras

Mis en forme : Police :NonGras, Vérifier l'orthographe etla grammaire

Mis en forme :Police :(asiatique) SimSun, NonGras, (Asiatique) Chinois (RPC)

Mis en forme : Police :NonGras

Mis en forme : Police :NonGras, Vérifier l'orthographe etla grammaire

Supprimé : (13-7)

Supprimé : (13-8)







13.5.2 THE N = 1 CASE 7780

7781 7782 7783

Figure 13- 19 CDF of Y 1 7784 7785 7786 7787 M metrics are being thresholded for a single antenna. In the absence of a signal, the metrics, Y, are Chi-7788 squared random variables with two degrees of freedom: 7789 7790 7791 Y1 = | X |2 7792 7793

where X is the complex, zero mean noise with noise power σ2 i.e. <|X|2> = σ 2. The probability density 7794

function is: 7795 7796

2

211 σ−

σ= /ye)y(f and the tail of the CDF is:

2

1σ−= /ye)y(Q . (13-9) 7797

7798

Figure 13-19 shows a plot of F1(y) = 1 – Q1(y) and the location of the 1st quartile, qX for the normalized (σ2 7799

= 1 ) case. 7800

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 17








7801 If Pfa is the desired probability of false alarm and Tfa is the corresponding threshold we will have 7802 7803 Pfa = Q1( Tfa ) 7804 7805 Take the log of equation (13-9) to get: 7806 7807

−y/σ2 = −ρM where [ ]( )M/paM Plog 111 −−−=ρ 7808

7809 7810 The value of y that solves this equation is the probability of false alarm for the one antenna case: 7811 7812 Tfa = σ2 ρM 7813 7814 7815

Setting σ2 = 1 gives the normalized threshold [ ]( )M/fafa PlogT 111 −−−= 7816

7817 7818

13.5.3 THE N = 2 CASE 7819

. 7820

7821 7822

Figure 13- 20 CDF of Y 2 7823 7824 7825 The metrics being thresholded are the average of the square magnitudes of two antennas for format 0 or 1 7826 antenna for format 2. In the absence of a signal, the metrics, Y, will be distributed as a Chi-squared 7827 distribution with 4 degrees of freedom: 7828 7829

Supprimé : 4.0.1

Supprimé : 2

Supprimé : (13-9)

Supprimé : 18







Y2 = 2

1 ( | Xa |

2 + | Xb |2 ) 7830

7831 The probability density function is: 7832 7833

)y(Uey

)y(f /y 22222

22 σ−

σσ= and the tail of the CDF is 7834

7835

σ+= σ−

22

22

12 y

e)y(Q /y (13-10) 7836

7837

Figure 13-20 shows a plot of F2(y) = 1 – Q2(y) and the location of the 1st quartile, qX for the normalized (σ2 7838

= 1) case. 7839 7840 To compute the Pfa threshold, first take the log of the equation for Q2(y): 7841 7842 7843

My

logy

ρ−=

σ++

σ−

22

21

2 where [ ]( )M/

paM Plog 111 −−−=ρ (13-11) 7844

7845 7846 This is a transcendental equation for y and requires numerical methods for its solution. Let x = 2y/σ2 and 7847 rewrite equation (13-11) as: 7848 7849 f ( x ) = x − log( 1 + x ) − ρM 7850 7851 The solution to this equation is the value of x such that f(x) = 0. We will use Newton's method to 7852 approximate the zero of f(x). In order to find a good initial value, we make the following observation: for 7853 values of M in [13..839] and for Pfa ranging from 0.1 to 0.0001 we find that ρM will be in the range [4.5, 16]. 7854 A numerical study of equation (x) shows that x will be in the range [ 6.5, 19]. 7855 7856 With these limits on the parameters, we can approximate the term log( 1+x ) by the line that passes 7857 through the points ( 6.5, log( 1 + 6.5 ) ) and ( 19, log( 1 + 19 ) ) 7858 7859

( ) ( ) 11011 yxxx

yaxaxlog +−

∆∆

=+≈+ (13-12) 7860

7861

( ) ( ) )5.7log(5.65.619

)5.7log()20log(1log +−

−−=+ xx 7862

7863 Performing the indicated computations gives: 7864 7865 a0 = 1.504872 and a1 = 0.078466 7866 7867 7868 Substitute this approximation into equation (13-12) and solve the equation f(x) = 0 for an initial estimate of x: 7869 7870 f ( x ) ≈ x − ( a0 + a1x ) − ρM = 0 7871 7872 7873

MM BA

a

ax ρρ +=

−+=

1

00 1

7874

7875 7876

Supprimé : 4.0.1

Supprimé : 2


Supprimé : (13-11)

Supprimé : (13-12)







Evaluating these expressions ( to 3 places ) gives: 7877 7878 A = 1.633 and B = 1.085 7879 7880

Given this initial value, we then use Newton’s method: )x(f)x(f

xx′

−← (13-13) 7881

7882 f ( x ) = x − log( 1 + x ) − ρM 7883 7884

xx

)x(f+

=′1

7885

7886 Substituting these into (13-13) and simplifying: 7887 7888

( ) 111 −ρ+++← M)xlog(

xx

x 7889

7890 7891 Numerical studies show that two iterations are sufficient. Recall that x was related to the original Chi-7892

squared variables by 2

2

σ=

yx 7893

7894 Summary of N = 2 Case: 7895 7896 To compute the Pfa threshold for the N = 2 case: 7897 We are given M, the number of metrics searched, and Pfa, the probability of false alarm. 7898 7899 7900

1. Compute ρM = −log( 1 − [ 1 − Pfa]1/M ) 7901

7902 2. Initialize x: x = A + BρM A = 1.633 and B = 1.085 7903

7904 7905

3. Perform two Newton iterations: 7906 7907

( ) 111 −ρ+++← M)xlog(

xx

x 7908

7909

( ) 111 −ρ+++← M)xlog(

xx

x 7910

7911

4. Solve for the normalized false alarm threshold 2

xTfa = where x is the result of step 3 above. 7912

13.5.4 THEN N=4 CASE 7913

The metrics being thresholded are the average of the square magnitudes of four antennas for format 0 or 2 7914 antennas for format 2. In the absence of a signal, the metrics, Y, will be distributed as a Chi-squared 7915 distribution with 8 degrees of freedom if the decision variable is constructed as following: 7916

( )4 4 22 2

4 i I Q1 1

1 1

4 4 k kk k

X X= =

= = +∑ ∑Y X (13-14) 7917

The tail function of CDF and PDF of this r.v. are 7918

Supprimé : 4.0.1

Supprimé : 2



Supprimé : (13-13)







( )

( )

2

2

4

4 3

4 20

34

8

1 4

!

128, 0

3

ky

k

yY

yQ y e

k

yp y e y

σ

σ

σ

σ

−

=

−

= = ≥

∑

(13-15) 7919

Figure 13-21 shows a plot of CDF of Y4, e.g. ( ) ( )4 41YF y Q y= − and the location of the 1st quartile, ˆ

qX 7920

for the normalized (σ2 = 1) case. 7921

0 0.5 1 1.5 2 2.5 3 3.5 40

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1CDF in 4 antenna case

7922

Figure 13-21 CDF of Y4 7923

To compute the faP threshold, first take logarithm of ( )4Q y : 7924

( )2 3

4 2 2 4 6

4 4 8 32ln ln 1

3

y y y yQ y

σ σ σ σ

= − + + + + (13-16) 7925

Suppose )(ln 4 yQM −=ρ , then 7926

++++−=−

6

3

4

2

22 3

32841ln

4

σσσσρ yyyy

M

(13-17) 7927

According to the discussion in §13.5.1, it can be concluded that 7928

( )1

ln 1 1M

M faPρ = − − − (13-18) 7929

Let2

4yx

σ= , then rewrite as(13-17) 7930

ˆ 0.63383005qx =

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Police :Arial,Anglais (Royaume-Uni)

Mis en forme : Police :Arial,Anglais (Royaume-Uni), Vérifierl'orthographe et la grammaire

Mis en forme : Police :Arial,Anglais (Royaume-Uni)

Mis en forme : Police :Arial,Anglais (Royaume-Uni), Vérifierl'orthographe et la grammaire


Supprimé : 19

Supprimé : (13-17)







( )2 3

ln 1 2 6 M

x xf x x x ρ

= − + + + − (13-19) 7931

The solution to this equation is the value of x such that ( ) 0f x = . Since (13-19) is a transcendental 7932

function, we must use Newton’s method to approximate the root of ( ) 0f x = . In order to find a good initial 7933

value, we make the following observation: for values of M in [13..839] and for Pfa ranging from 0.1 to 0.0001 7934

we find that Mρ will be in the range [4.5, 16]. A numerical study of equation (x) shows that x will be in the 7935 range [9.88, 23.88]. 7936

In order to get the initial values for Newton iteration method, it ought to find the linear approximation 7937

of2 3

ln 12 6

x xx

+ + +

. Here we use the linear interpolation to estimate the inner point with the known 7938

ends at 0 9.88x = and 1 23.88x = , e.g. 7939

( )2 3

1 0 1ln 12 6

x x yx a x a x x

x

∆+ + + ≈ + = − ∆ (13-20) 7940

Let2 3

( ) ln(1 )2 6

x xg x x= + + + , then it yields 7941

( ) ( ) ( ) ( )2 3

0 10 0

0 1

ln 12 6

g x g xx xx x x g x

x x

− + + + ≈ − + − (13-21) 7942

Performing the indicated computations gives: 7943

a0 = 3.6596 and a1 = 0.1757 7944

Substitute this approximation into equation (13-20)and solve the equation f(x) = 0 for an initial estimate of x: 7945

0)()( 01 =−+−≈ Maxaxxf ρ

(13-22) 7946

M

M BAa

ax ρρ

+=−+

=1

00 1 (13-23) 7947

Evaluating these expressions (to 3 places) gives: 7948

A = 4.4397 and B = 3.6596 7949

Given this initial value, we then use Newton’s method: 7950

)(

)(

xf

xfxx

′−←

(13-24) 7951

For the equation(13-19), it can be concluded that 7952

3

2 36( )

12 6

x

f xx x

x

′ =+ + +

(13-25) 7953

Substituting these into (13-24) and simplifying: 7954

Supprimé : 4.0.1

Supprimé : 2

Supprimé : (13-19)

Supprimé : (13-20)

Supprimé : (13-19)

Supprimé : (13-24)







++−

+

++++++←

2

32

32

663

621ln)

6631(

xx

xxx

xxxx Mρ

7955 (13-26) 7956

Numerical studies show that five iterations are sufficient. Recall that x was related to the original Chi-7957


4yx

σ= 7958

Summary of N = 4 Case: 7959 7960 To compute the Pfa threshold for the N = 4 case: 7961 We are given M, the number of metrics searched, and Pfa, the probability of false alarm. 7962 7963 7964

1. Compute ρM = −log( 1 − [ 1 − Pfa]1/M ) 7965

7966 2. Initialize x: x = A + BρM A = 4.4397 and B = 3.6596 7967

7968 7969

3. Perform five Newton iterations: 7970 7971

++−

+

++++++←

2

32

32

663

621ln)

6631(

xx

xxx

xxxx Mρ

7972 7973 Solve for the normalized false alarm threshold ˆ

4fa

xT = where x is the result of step 3 above. 7974

13.5.5 THE N=8 CASE 7975

The metrics used of calculating threshold are the average of the square magnitudes of four antennas. In 7976 the absence of a signal, the metrics, Y, will be distributed as a Chi-squared distribution with 16 degrees of 7977 freedom if the decision variable is constructed as following: 7978

( )8 8 22 2

81 1

1 1

8 8k kI kQk k

Y X X X= =

= = +∑ ∑ (13-27) 7979

The tail function of cdf and pdf of this r.v. are 7980

∑

=

−

−

=

≥=

7

02

28

8

28

16

78

8

)8

(!

1)(

0,!7

8)(

k

k

y

y

Y

y

keyQ

yey

yP

σ

σ

σ

σ

(13-28) 7981

Figure 2-23 shows a plot of CDF of Y8, e.g. 8 8( ) 1 ( )YF y Q y= − and the location of the 1st quartile, ˆ

qX for 7982

the normalized (σ2 = 1) case. 7983

Supprimé : 4.0.1

Supprimé : 2









7984 Figure 13- 22 CDF of Y8 7985

7986 7987

To compute the faP threshold, first take logarithm of 8( )Q y given in the equation (2.27): 7988

7989 2 2 3 3 4 4 5 5 6 6 7 7

8 2 2 4 6 8 10 12 14

8 8 8 8 8 8 8 8ln ( ) ln(1 )

2 3! 4! 5! 6! 7!

y y y y y y y yQ y

σ σ σ σ σ σ σ σ= − + + + + + + + + 7990

(13-29) 7991 7992

Let2

8yx

σ= , suppose 8ln ( )M Q yρ = − then: 7993

2 3 4 5 6 7

( ) ln(1 )2 3! 4! 5! 6! 7! M

x x x x x xf x x x ρ= − + + + + + + + − (13-30) 7994

7

'2 3 4 5 6 7

7!( )1

2 3! 4! 5! 6! 7!

x

f xx x x x x x

x

=+ + + + + + +

(13-31) 7995

The solution to this equation is the value of x such that ( ) 0f x = . Since (13-19) is a transcendental 7996

function, we must use Newton’s method to approximate the root of ( ) 0f x = . In order to find a good initial 7997

value, we make the following observation: for values of M in [13..839] and for Pfa ranging from 0.1 to 0.0001 7998

we find that Mρ will be in [4.5, 16]. A numerical study of equation (x) shows that x will be in the range, 7999

0 15.82x = , 1 31.97x = and linear interpolation as follows: 8000

3 4 5 6 7

1 0 1ln(1 ) ( )3! 4! 5! 6! 7!

x x x x x yx a x a x x

x

∆+ + + + + + ≈ + = −∆

(13-32) 8001

1 0.2875a = ; 0 6.7742a = 8002

3 4 5 6 7

( ) ln(1 )3! 4! 5! 6! 7!

x x x x xg x x= + + + + + + , then 8003

3 4 5 6 7

1 00 0

1 0

( ) ( )ln(1 ) ( ) ( )

3! 4! 5! 6! 7!

g x g xx x x x xx x x g x

x x

−+ + + + + + ≈ − +

− (13-33) 8004

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 20

Supprimé : (13-19)







Substitute this approximation into equation (13-20)and solve the equation f(x) = 0 for an initial estimate of x: 8005

0)()( 01 =−+−≈ Maxaxxf ρ

(13-34) 8006

M

M BAa

ax ρρ

+=−+

=1

00 1 (13-35) 8007

Evaluating these expressions (to 3 places) gives: 8008

A = 9.5075 and B = 1.4035. 8009 8010 Given this initial value, we then use Newton’s method: 8011

)(

)(

xf

xfxx

′−←

(13-36) 8012 Take (2-29) and (2-30) into (2-35), 8013 8014

7 7 6

7 6

0 0 0

7! ( )[ ln( )] 7!! ! !

i i i

Mi i i

x x xx x x

i i iρ− −

= = =← + −∑ ∑ ∑ (13-37) 8015

8016

Slove for the normalized false alarm threshold ˆ8fa


Numerical studies show that five iterations are sufficient. Recall that x was related to the original Chi-8018


8yx

σ= 8019

Summary of N = 8 Case: 8020 8021 To compute the Pfa threshold for the N = 8 case: 8022 We are given M, the number of metrics searched, and Pfa, the probability of false alarm. 8023 8024

1) Compute ρM = −log( 1 − [ 1 − Pfa]1/M ) 8025

2) Initialize x: x = A + BρM A =9.5075 and B = 1.4035 8026 3) Perform five Newton iterations: 8027

8028 7 7 6

7 6

0 0 0

7! ( )[ ln( )] 7!! ! !

i i i

Mi i i

x x xx x x

i i iρ− −

= = =

← + −∑ ∑ ∑ 8029

8030

Solve for the normalized false alarm threshold ˆ8fa


8032

13.5.6 CALCULATION OF 1ST QUARTILE VALUE 8033

Given a probability distribution, F(x), the value of x for the first quartile is, xq: 8034 8035 F( xq ) = 0.25 8036 8037 8038 N = 1 Case: 8039 8040

From equation (13-9) (with 2σ set to 1) F1( x ) = 1 − e−x 8041 8042 1 − e−x = 0.25 8043

Supprimé : 4.0.1

Supprimé : 2



Supprimé : (13-20)

Supprimé : (13-9)







8044 Solving for x: 8045 8046

qX = −log( 0.75 ) ≈ 0.28768207 the normalized 1st quartile value. 8047

8048 8049 N = 2 Case: 8050 8051

From equation (13-10), substitude 2

y

σ with x, then it can be revised into F1( x ) = 1 − ( 1 8052

+ 2x ) e−2x 8053 8054 1 − ( 1 + 2x ) e−2x = 0.25 8055 8056 This gives rise to the transcendental equation: 8057 8058 −x + log( 1 + x ) = log( 0.75 ) which can be solved using Newton's method. The result is: 8059 8060 8061

qX = 0.48063938 8062

N = 4 Case: 8063 8064

From equation(13-15), substitude 2

y

σ with x, then it can be revised into 8065

( ) ( )3

41

0

11 4

!kx

k

F x e xk

−

=

= − ∑ 8066

8067

( )3

4

0

11 4 0.25

!kx

k

e xk

−

=

− =∑ 8068

8069 This gives rise to the transcendental equation: 8070 8071

( ) ( )3

0

14 log 4 log 0.75

!k

k

x xk=

− + = ∑ which can be solved using Newton's method. The result is: 8072

8073 8074

qX = 0.63383005 8075

N = 8 Case: 8076 8077

From equation(13-15), substitude 2

y

σ with x, then it can be revised into 8078

( ) ( )7

81

0

11 8

!kx

k

F x e xk

−

=

= − ∑ 8079

8080

( )7

8

0

11 8 0.25

!kx

k

e xk

−

=

− =∑ 8081

8082 This gives rise to the transcendental equation: 8083 8084

Supprimé : 4.0.1

Supprimé : 2

Supprimé : (13-10)

Supprimé : (13-15)

Supprimé : (13-15)







( ) ( )7

0

18 log 8 log 0.75

!k

k

x xk=

− + = ∑ which can be solved using Newton's method. The result is: 8085

8086 8087

qX = 0.74451373 8088

8089 8090

13.5.7 THRESHOLD SCALING 8091

8092

Table 13-30 – Example Normalized Thresholds 8093

Pfa 0.001 0.0001

Number of Antennas 1 2 4 1 2 4

M = 832 13.6311 8.2464 5.2843 15.9341 9.4631 5.9443

M = 26816 17.1040 10.0779 6.2757 19.4070 11.2634 6.9221

8094

For the lowest Ncs index (see Table 13-2) the Ncs value = 13 so the number of search bins is M = 64×13 = 8095 832. For the highest Ncs configuration index the Ncs value is 419 so the number of search bins is M = 8096 64×419 = 26816. Table 13-30 shows the normalized thresholds for these three cases for one antenna, two 8097 antennas and four antennas and for false alarm probabilities of 0.001 and 0.0001. 8098

The threshold written to SRIO-A register DETECT_TH by the DSP is a scaled 16 bit integer. To determine 8099

the proper scaling, we get an upper bound on faT . If we examine the table above we find that for the worst 8100

case of M = 26816 we can accommodate a Pfa of 0.001 for one diversity and a Pfa of 0.0001 for two 8101 diversities with a scale factor of 210 8102

For the one antenna case: 60652259.23120.287689

17.10402

1)(nX

T 101010

q

fa ≈×=×=×=

)

8103

and for the two antenna case: 239992436.3224806.0

2634.112

2)(nX

T 101010

q

fa ≈×=×=×=

)

8104

We define the scaling of the detection threshold as: 8105

DETECT_TH =

× 102

q

fa

X

TRound (13-38) 8106

13.5.8 SUMMARY OF THRESHOLD CALCULATIONS 8107

8108 Table 13-31 – Normalized 1 st Quartile Values 8109

Div Xq

N=1 0.28768207

N=2 0.48063938

N=4 0.63383005 N=8 0.74451373

8110

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 27


Supprimé : 28







8111 The table below gives the normalized thresholds and their corresponding scaled values for a false alarm 8112 probability of 0.001. The normalized thresholds for one antenna are given by the relations derived in 8113 section 13.5.2 and the thresholds for the two antenna case are given by the relations derived in section 8114 13.5.3, and the thresholds for the four antenna case are given by the relations derived in section 13.5.4 and 8115 specified in Table 13-33. The scaled hex values are computed from equation (13-38). 8116 8117 8118

Table 13-32 – Thresholds for Pfa = 0.001 when N=1,2 8119 N=1 N=2 Ncs

Config Index

Ncs Value M Normalized

Threshold Scaled

Hex Threshold

Normalized Threshold

Scaled Hex

Threshold 1 13 832 13.63108811 0xbd88 8.24644109 0x44a1 2 15 960 13.77418888 0xbf85 8.32230989 0x4543 3 18 1152 13.95651035 0xc20e 8.41891597 0x4610 4 22 1408 14.15718096 0xc4d8 8.52517290 0x46f3 5 26 1664 14.32423499 0xc72b 8.61357354 0x47af 6 32 2048 14.53187430 0xca0e 8.72338161 0x4899 7 38 2432 14.70372452 0xcc72 8.81420624 0x495b 8 46 2944 14.89477972 0xcf1a 8.91512201 0x4a32 9 59 3776 15.14367573 0xd290 9.04649868 0x4b4a

10 76 4864 15.39687160 0xd615 9.18004254 0x4c66 11 93 5952 15.59873773 0xd8e3 9.28644140 0x4d49 12 119 7616 15.84526171 0xdc51 9.41629408 0x4e5d 13 167 10688 16.18413201 0xe107 9.59464199 0x4fd9 14 279 17856 16.69734996 0xe82a 9.86443888 0x5218 15 419 26816 17.10400909 0xedd1 10.07796465 0x53df

8120 Table 13-33 – Thresholds for Pfa = 0.001 when N=4,8 8121

N=4 N=8 Ncs Config Index

Ncs Value

M Normalized Threshold

Scaled Hex

Threshold

Normalized Threshold

Scaled Hex

Threshold 1 13 832 5.2843558660 0x2159 3.6152893668 0x136c 2 15 960 5.3256879778 0x219c 3.6385620766 0x138c 3 18 1152 5.3782804122 0x21f1 3.6681520568 0x13b5 4 22 1408 5.4360798162 0x224e 3.7006423427 0x13e2 5 26 1664 5.4841293768 0x229c 3.7276290758 0x1407 6 32 2048 5.5437692742 0x22fc 3.7610969524 0x1435 7 38 2432 5.5930613360 0x234c 3.7887345340 0x145b 8 46 2944 5.6477911144 0x23a4 3.8193965460 0x1485 9 59 3776 5.7189807122 0x2417 3.8592422515 0x14bc

10 76 4864 5.7912766294 0x248c 3.8996642730 0x14f4 11 93 5952 5.8488292844 0x24e9 3.9318127549 0x1520 12 119 7616 5.9190121765 0x255b 3.9709807456 0x1556 13 167 10688 6.0153075820 0x25f6 4.0246593447 0x159f 14 279 17856 6.1607704813 0x26e1 4.1056128560 0x160f 15 419 26816 6.2757233392 0x279b 4.1694774293 0x1667

8122 It should be understood that these are nominal values and may need to be adjusted as a result of 8123 laboratory performance testing.8124

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Police :Arial,Ne pas vérifier l'orthographe oula grammaire

Mis en forme : Police :Arial

Mis en forme : Police :Arial


Supprimé : 13-38

Supprimé : 29

Supprimé : 30







8125

13.6. INTERPRETATION OF RACH TIME DELAY OFFSET 8126

This section describes the background and analysis for interpreting the time delay offsets that the RACH 8127 returns to the DSP in the Peak Signature Block. It also describes the procedure for setting the RACH 8128 search window size UN_WINDOW and the search start locations (UWS_0, UWS_1, and UWS_2) that the 8129 DSP writes to the SRIO-A registers when configuring a RACH. 8130 8131

13.6.1 TIME DELAYED RACH SIGNAL 8132

We consider the RACH signal given by equation (13-2) delayed by td seconds and digitized at a sample rate of Fs 8133 (where Fs = 7.68 MHz for a 5 Mhz system, Fs = 15.36 MHz for a 10 MHz system, Fs = 23.04 MHz for a 15 MHz 8134 system and Fs = 30.72 MHz for a 20MHz system). The sampled, time-delayed version of equation (13-2) (without 8135 βprach ) 8136 8137

( ) ( )( ) ( )∑−

=

−−∆++ϕ+π=1

0

2 21

0

ZC

dCPsRA

N

k

tTF/jfkKkje)k(xjs (13-39) 8138

8139

To simplify the calculations we make the following definitions: 8140

8141

s

cpcp F

TN = The Number of samples in the cyclic prefix at sample rate Fs 8142

( )210 /kK ++φ=κ Rach frequency offset in units of ∆fRA 8143

sdFt=τ The time delay in (generally) fractional samples. 8144

N The number of ∆fRA subcarriers in the band Fs. 8145

We note that 1250=N

Fs Hz regardless of the sample rate. (13-40) 8146

8147

With these definitions in equation (13-39), the received, digitized signal is: 8148

8149

[ ]( )∑−

=

τ−−κ+π=1

0

2zc

cp

N

k

N/Njkie)k(x)j(s (13-41) 8150

8151

We offset into s(j) by the cyclic prefix and bandshift to center the ZC sequence about DC: 8152

8153

jicp e)Nj(s)j(R πν−+= 2 where

N

Nzc

2

12 −+κ=ν is the bandshift “frequency”. If we define 8154

2

1−= zc

dcN

k and Nκτπ−=θ 2 then equation (13-41) can be written as: 8155

8156

Supprimé : 4.0.1

Supprimé : 2

Supprimé : (13-2)

Supprimé : (13-2)

Supprimé : (13-39)

Supprimé : (13-41)







( ) N/jkkiN

k

N/kii dc

zc

ee)k(xe)j(R −π−

=

πτ−θ ∑= 21

0

2 (13-42) 8157

8158

)k(x is the DFT of the root-u ZC sequence shifted by v, which in the frequency domain is equivalent to 8159

multiplying by a phasor: 8160

8161

zcN/kviu e)k(x)k(x π= 2 substituting this into (13-42): 8162

8163

( ) N/jkkiN

k

kNN

vi

ui dc

zczc ee)k(xe)j(R −π

−

=

τ−πθ ∑= 2

1

0

2

(13-43) 8164

8165

At this point it one can see that the time delay, τ, and the cyclic shift, v, work in opposite directions. That is, 8166 if the RACH processing is set up so that v shifts the correlation peaks to the right, then the time delay will 8167 shift the peaks to the left. 8168

We can also see that by setting

τ−NN

v

zc

to zero that a shift of zcNv

is equivalent to a time delay of Nτ

. 8169

So solving for td: 8170

zcszcd N

vv

FNN

t×

==1250

(13-44) 8171

8172

where the last substitution has come from equation (13-40). And conversely a time delay of td seconds will 8173 shift the correlation peak 1250×Nzc×td cyclic shifts to the left. 8174

8175

13.6.2 TIME DELAY IMPLEMENTATION ISSUES 8176

A forward FFT is used to uncover the DFT of the ZC sequence, )n(x that appears in equation (13-43 ). In 8177

order to obtain the correlation of the received Rach with a candidate ZC sequence, the received sequence 8178 is multiplied by the conjugate of the DFT of the candidate ZC sequence. To obtain the cyclic correlation, an 8179 inverse DFT would be performed. 8180 8181 However, this inverse DFT would be an 839 point DFT where 839 is a prime number. There are methods 8182 for computing prime number DFTs but they are computationally more expensive than we want to use at this 8183 time so we approximate the 839 point DFT with a 1024 point FFT. 8184 8185 This has the effect of "stretching" the location of the correlations so that a correlation peak that would have 8186

shown up at bin n will appear at offset 839

1024×n . 8187

8188 Also, instead of implementing an inverse FFT we use a forward FFT to get the final correlations. This is so 8189 that cyclic shifts move to correlation peaks to the right i.e. to higher offsets. However, time delay will shift 8190 the peaks to the left. 8191 8192 8193 Summary of Implementation Issues: 8194 8195

Supprimé : 4.0.1

Supprimé : 2

Supprimé : (13-42)

Supprimé : (13-40)

Supprimé : 13-43







1. The correlation offsets returned by or sent to the Rach HW are "stretched" so that the relation 8196 between a theoretical shift and the actual offset is: 8197

8198

839

1024×= shiftoffset 8199

8200 8201

This will affect UN_WINDOW, UWS_0, UWS_1, and UWS_2 as inputs to the Rach ( see section 8202 13.6.4 ) and the OFFSET field in the peak structure table that is output from the Rach. 8203

8204 2. Cyclic shifts will move peaks to the right (higher offsets) and time delay will move peaks to the left 8205

(lower offsets). This means that the zero time delay position will be at the right edge of the search 8206 window. 8207

8208

13.6.3 EXAMPLE OF RACH CORRELATION 8209

8210

Figure 13- 23 Example of RACH Correlation Metrics 8211 8212

8213

Figure 13-23 above shows an example of the following scenario: 8214

In a 5 MHz system, a RACH is transmitted with a cyclic shift of v = 300. The signal is received by the 8215 RACH processor after incurring a delay of 19 µSec. The transmitted cyclic shift manifests itself as a 8216

positive offset of 15.366839

1024300 ≈× correlation bins. From equation (13-44) a time delay of 19 µSec 8217

corresponds to a left shift of 20191250 ≈µ×× SecNzc . 8218

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 21


Supprimé : (13-44)







8219

But a left shift of 20 becomes a negative offset of 424839

102420 .≈× correlation bins. So the correlation peak 8220

shows up at offset 366.15 – 24.4 = 341.75 which rounds up to 342 relative to the start of the correlation 8221 buffer. 8222

13.6.4 SETTING THE RACH SEARCH WINDOW 8223

8224

Figure 13- 24 Example of Setting the RACH Search Window 8225 8226

We use Figure 13-24 above to illustrate how to set the search window start locations (USW_0, UWS_1, and 8227 UWS_2). In this example: 8228

8229

1. Ncs has been specified to be 50. 8230

2. The Ue will transmit the RACH sequence with a cyclic shift of v = 200. 8231

The theoretical (cyclic shift) domain, we would start searching for the correlation peak at cyclic shift location 8232 200 (which is the zero time delay location) and search to the left Ncs = 50 shifts (Mod 839) in the direction 8233 of increasing time delay. In setting up the RACH parameters there are two considerations that affect the 8234 window size and the search start locations: 8235

8236

1. As described in section 13.6.2, the location of the correlation offsets are “stretched” by 1024/839 so 8237 that, in the example above, we search over 02561839102450 ./ ≈× offsets which rounds to 61. And 8238 likewise, the search window will be stretched from [149, 200] to [183, 244]. 8239

2. The RACH HW starts at UWS_0 and searches UN_WINDOW offsets to the right. This means that 8240 the RACH starting location is nominally 183. However, it is not advisable to have the zero delay 8241 offset at the boundary of the search window. It is recommended that the right boundary of the 8242 window be extended two cyclic shifts to the right of the zero time delay value. This can be 8243

149 200

183 224

Ncs = 50

UN_WINDOW = 61

Max time delay location

Zero time delay location

Theoretical cyclic shift domain

Actual correlation search domain

Correlation bins are stretched by 1024/839

UWS_0

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 22








accomplished by expanding the window by two shifts or by translating the window to the right by 8244 two shifts. The CWV is to shift the window but this may be revised in the future. 8245

8246

8247

13.6.5 CALCULATION OF SEARCH WINDOW SIZE 8248

8249

We are given: 8250

R – the cell radius in Km 8251

sτ – nominal delay spread in seconds (typically 5 µSec). 8252

C – speed of light in Km/sec 5103×≈ Km/Sec. 8253

8254

The total time delay is assumed to be the round trip propagation time plus the delay spread: 8255

8256

sdelay CR τ+×=τ 2 (13-45) 8257

8258

We assume a delay spread of sτ = 5µsec. The time delay (in seconds) associated with one “unstretched” 8259

RACH correlation bin is 8260

8391250

1

×=dT seconds. (13-46) 8261

Therefore the search window size (i.e. number of correlation bins) that spans the time delayτ is 8262

τ=

d

delayw T

N 8263

Substituting equations ( )Nmod)Cn((x)n(x ZCvuv,u += (13-1) and (13-46) into the expression for Nw gives 8264

the window size as a function of the cell radius: 8265

245996105103

28391250 6

5.R.

RNw +×=

×+×

×= − (13-47) 8266

8267

Conversely we may solve for R as a function of Nw: 8268

8269

[ ]( )delaydw TNC

R τ−−−= 12

(13-48) 8270

8271

Table 13-34 – Relation Between Ncs Value and Cell Radius for L ow Mobility Case 8272 Ncs Index Ncs Radius Km

0 0 -

1 13 1.0

2 15 1.3

3 18 1.7

Supprimé : 4.0.1

Supprimé : 2

Supprimé : n((x)n(x uv,u +=(13-1

Supprimé : 13-46

Supprimé : 31







4 22 2.3

5 26 2.8

6 32 3.7

7 38 4.5

8 46 5.7

9 59 7.5

10 76 10.0

11 93 12.5

12 119 16.1

13 167 23.0

14 279 39.0

15 419 59.0

8273

Table 13-34 above shows the low mobility column of Table 13-2 with the cell radius computed from 8274 equation (13-48) with Nw set equal to Ncs. 8275

8276

13.6.6 SUMMARY OF SETTING THE RACH SEARCH WINDOW 8277

Inputs: 8278

Ncs The desired Ncs value as defined in 36.211. 8279 v The nominal (zero time delay) cyclic shift. 8280 ui The multiplicative inverse Mod 839 of the ZC root. 8281

8282

Procedure: 8283 8284 Compute the start (i.e. left side) of the search window: 8285 8286 v0 = ( v – Ncs ) Mod 839 8287 8288 Compute the start locations of the three search windows: 8289 8290 uws0 = ( v0 + 2 ) Mod 839 8291 uws1 = ( v0 + ui + 2 ) Mod 839 8292 uws2 = ( v0 - ui + 2 ) Mod 839 8293 8294 Compute the stretched search window size 8295 8296

UN_WINDOW =

×839

1024NcsRound 8297

8298 Compute the stretched start locations: 8299 8300 8301

USW_0 =

×839

10240uwsRound 8302

8303

USW_1 =

×839

10241uwsRound 8304

8305

Supprimé : 4.0.1

Supprimé : 2


Supprimé : 13-48







USW_2 =

×839

10242uwsRound 8306

8307 8308

8309

13.7. SIGNATURE DETECTION 8310

For LA1.0, LA2.0, LA3.0and LA4.0 we respond to only one RACH signature per RACH instance: namely 8311 the signature with the highest detected peak. Since signatures corresponding to different ZC roots will in 8312 general have different thresholds, the peaks must be reconstructed from the returned data in the signature 8313 block. 8314

If M is a metric to be thresholded, we want to perform the test: is M > T . This is implemented in the 8315 hardware by comparing the product of the metric with the reciprocal of the threshold to 1: 8316

M > T => 11 <×T

M 8317

The value returned in the peak list (Table 13-25) is actually a scaled version of T

M1× . In order to get back 8318

the original metric M, we must multiply by the threshold. From section 13.4.13 we have: 8319

P is the 16-bit peak magnitude returned in the signature list (actually the peak divided by the threshold). 8320

Tm is the 16-bit mantissa of the threshold from the last entry in the signature block, 8321

e is the 8-bit signed threshold exponent from the last entry in the signature block. 8322

The reconstructed 24-bit metric, M, is computed as: 8323

4216

1 −×=×

= emTPTM (13-49) 8324

The reconstructed value of M should be no more than 24 bits unsigned but care must be taken in the 8325 calculation to prevent overflow. These are the values to be compared. 8326

The first peak in the signature block is the largest peak in the main (low mobility) window. 8327

1. Check the detect bit for the first entry in the signature block. 8328

2. If this peak is detected, get the threshold word. 8329

3. Reconstruct the metric using equation ( ( ) expshift N/nNje)k(XnY π= 2 ). 8330

4. Scan through the 64 signatures and select the one with the maximum value of M. 8331

8332

13.8. CALCULATION OF N-AVERAGE DETECTION THRESHOLD 8333

8334 This section describes a method to calculate the RACH detection threshold for multiple antennas and 8335 RACH formats given the detection threshold for a single antenna and RACH format 0. The algorithm is 8336 based on the observation that the relationship between the threshold for one antenna and multiple 8337 antennas is nearly linear in the operational region of interest. 8338 8339 The RACH detection threshold is used to detect peaks in an array of metrics that are formed by 8340 incoherently averaging the squared magnitude of correlations over multiple antennas and multiple 8341 sequences (in the case of RACH formats 2 and 3). The probability distribution of a single metric is 8342 (approximately) chi-squared with 2N degrees of freedom where N is the number of averages. The table 8343 below shows the number of correlations incoherently averaged as a function of antennas and RACH 8344 formats: 8345

Supprimé : 4.0.1

Supprimé : 2


Supprimé : ( ) Nje)k(XnY π= 2







Table 13-35 – Number of Correlations Averaged 8346 Number of Antennas Format

1 2 3 4 0,1 1 2 3 4 2,3 2 4 6 8

8347 8348 For example: with 4 receive antennas and RACH format 2, each metric will be the average of the squared 8349 magnitude of 8 correlations ( 4 antennas x 2 sequences ) and hence the noise statistics will be chi-squared 8350 with 16 degrees of freedom. 8351 8352 The supported configurations have 1, 2, or 4 antennas. However, for completeness, we will also consider 3 8353 antennas in the case of antenna outage. Since for RACH formats 2 and 3 there will always be two 8354 sequences and hence two correlations to be averaged, there should never be a case where 5 or 7 8355 correlations are averaged. As a practical matter, however, we will consider these cases too in case the 8356 internal RACH processing discards one of the two sequences. 8357 8358 Since the threshold depends on the number of correlations averaged this note will refer to the number of 8359 averages rather than the number of diversities. The false alarm threshold, T, for searching over M metrics 8360 with probability Pfa is given by the relation: 8361 8362

[ ] MfaPTQ /111)( −−= (1) 8363

8364 where Q(T) is the tail of the relevant probability distribution for a single metric. The number of correlation 8365 metrics searched depends on the Ncs value and the low/high speed configuration: 8366 8367 M = Ncs×64 for the low speed case and M = 3Ncs×64 for the high speed case. 8368 8369 The Ncs values are specified in table 5.7.2-2 of 36.211. M takes on its smallest value for Ncs configuration 8370 1 (Ncs = 13) for unrestricted sets (i.e. low speed) and takes on its largest value for Ncs configuration 14 8371 (Ncs = 237) for restricted sets (i.e. high speed). The Ncs = 0 case for unrestricted sets will not be 8372 considered here. 8373 8374 In each RACH instance there are 64 signatures to be searched. This gives an overall range of M as: 8375 8376

13×64 = 832 ≤ M ≤ 45504 = 3×64×237 8377 8378 8379

Table 13-36 – Values of Q from Equation (1) 8380 Pfa M = 832 M = 45504

0.1 1.26627 × 10 -4 2.31541 × 10 -6

0.01 1.20797 × 10 -5 2.20867 × 10 -7

0.001 1.20252 × 10 -6 2.19871 × 10 -8

0.0001 1.20198 × 10 -7 2.19772 × 10 -9

8381 8382 The table above shows the values of Q for a single metric for the specified system false alarm rates over 8383 the range of search bins, M. Note that the notation being used here is that the subscript fa (as in Pfa) refers 8384 a system probability i.e. the probability associated with thresholding M bins. Probabilities (or probability tails) 8385 without subscripts or with numerical subscripts refer to single metric probabilities. 8386 8387 The normalized tail function for a chi-squared random variable with 2N degrees of freedom is: 8388 8389

( )∑

−

=

−=1

0!

)(N

k

kNy

n kNy

eyQ (2) 8390

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 32

Supprimé : 33







8391 Note that N is the number of complex magnitudes averaged. The number of degrees of freedom is 2N. 8392 8393 We assume that we have chosen a target system false alarm rate, Pfa, together with a RACH configuration 8394 that determines the number of correlation bins, M, to be searched. Using the equation above we compute 8395 the value for Q. Regardless of the number of diversities or formats being processed, this will be the target 8396 probability. 8397 8398 The detection threshold will be obtained by solving the equation: 8399 8400

Qn ( Tn ) = Q 8401 8402 for Tn where n = 1, 2, ..., 8 and Q is given by equation (1) . Qn(Tn) is the function given by equation (2). If 8403 we were given the 1-average threshold, T1, and wanted the corresponding threshold for the n-average case 8404 we could compute Q from 8405 8406 Q = Q1(T1) 8407 8408 Then, having obtained Q, solve Qn(Tn) = Q for Tn. 8409 8410

However, the RACH hardware is passed nT which is a scaled version of the detection threshold Tn: 8411 8412

×=

nnn TT

ξ

102ˆ 8413

8414 where nξ is the value at the 1st Quartile for n averages: 0.25 = 1 - Qn( ξn ). The factor 210 is used to scale 8415

the thresholds for fixed point implementation. This factor is common for all cases and will temporarily be 8416 dropped from the calculations. Let 8417 8418

n

nn

TT

ξ= so that the FPGA threshold is nn TT 102ˆ = 8419

8420 From the above considerations, we can think of thresholds for 2, ..., 8 averages as a function of the 1-8421 average threshold: 8422 8423

( )1TfT nn = 8424 8425

where the function ( )1Tfn is computed using the following procedure: 8426 8427 8428

Given 1T :

111 ξ×= TT Unscale the 1-average threshold by the 1-average 1st quartile value.

)( 11 TQQ = Compute the tail probability for this threshold from the 1-average distribution.

)(1 QQT nn−= With the previously computed probability, numerically invert the n-average distribution

to get the n-average threshold.

n

nn

TT

ξ=

Scale the n-average threshold with the n-average 1st quartile value.

Supprimé : 4.0.1

Supprimé : 2







8429

Table 13-37 – Range of 1T from Table 13- 38 8430

1T Pfa

M = 832 M = 45504 0.1 31.20 45.11 0.01 39.36 53.27 0.001 47.38 61.29 0.0001 55.39 69.30

8431 8432 The table above shows the values of 8433 8434

[ ]( )Mfan PQT /11

1 11 −−= − 8435 8436

for M and Pfa given in Table 13-39. From this table we infer that a reasonable operational range for 1T is 8437

[31.2, 70.0]. 8438 8439

Plotting the functions ( )1Tfn shows that the scaled thresholds for various averages as a function of the 8440

scaled threshold for one average are nearly linear. This suggests fitting the functions ( )1Tfn to a line with 8441

perhaps a small quadratic correction factor: 8442 8443

( ) ( ) ( ) ( ) 2121101 TnaTnanaTfT nn ++== (3) 8444

8445 A least-squares fit yields the following result: 8446 8447

Table 13-40 – Least Squares Fit for Equation 8448 Function a0 a1 a2

f2 1.73843 0.33293 -1.59103 ×10 -4

f3 1.81567 0.20027 -1.55895 ×10 -4

f4 1.77859 0.14525 -1.44545 ×10 -4

f5 1.73015 0.11524 -1.33996 ×10 -4

f6 1.68531 0.09632 -1.25040 ×10 -4

f7 1.64610 0.08327 -1.17493 ×10 -4

f8 1.61214 0.07371 -1.11077 ×10 -4

8449 8450 Fixed-point Implementation 8451 8452 We start with equation (3) and the values in the table above. 8453 8454

212110 TaTaaTn ++= 8455

8456 where it is understood that the coefficients of the quadratic are functions of the number of averages and 8457 rewrite as: 8458 8459

( ) 11210 TTaaaTn −+= 8460

8461 Since a2 is negative for all cases, we make this coefficient positive and explicitly show the subtraction. The 8462

actual threshold, nT , used by the HW is scaled by 210 so multiply this equation by 210: 8463

8464

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 34

Supprimé : 35

Supprimé : 36

Supprimé : 37







( ) 11210

101010 ˆˆ222ˆ TTaaaTT nn ×−+== − 8465

8466 Fiddle with the scaling so that the coefficients have about 16 bits of precision: 8467 8468

[ ]( ) 1122720

11717

010 ˆˆ22222ˆ TTaaaTn ×−+= −− 8469

8470 Define: 8471 8472 A0 = 210×a0 A1 = 217×a1 A2 = 227×| a2 | 8473 8474 This gives the equation that is the basis for the fixed point algorithm: 8475 8476

( ) 11220

117

0ˆˆ22ˆ TTAAATn ×−×+= −− 8477

8478 where A0, A1, and A2 are computed by scaling and rounding the values in Table 13-41: 8479 8480

Table 13-42 – Scaled Coefficients for Equation 8481 n A0 A1 A2 2 0x06f4 0xaa76 0x536a 3 0x0743 0x668a 0x51bc 4 0x071d 0x4a5e 0x4bc8 5 0x06ec 0x3b01 0x4641 6 0x06be 0x3151 0x418f 7 0x0696 0x2aa3 0x3d9a 8 0x0673 0x25bd 0x3a3d

8482 8483 Fixed Point Algorithm 8484 8485 Inputs: 8486 n 4-bit unsigned index indicating the number of averages [2..8]. 8487

1T 16-bit 1-average threshold = A_LRACH_THRESHOLD.DETECT_TH 8488

8489 8490 Processing: 8491 8492 8493 A0, A1, A2 Get the coefficients from Table-n as specified by n.

A0: 11-bit unsigned A1: 16-bit unsigned A2: 15-bit unsigned

p = A2× 1T Multiply 15-bit A2 by the 16-bit 1-average threshold.

t1 = RoundStrip(p,20) Shift the product right 20 bits and round. t1: 11-bit unsigned.

t2 = A1 – t1 t2: 16-bit unsigned.

1220

12ˆ2 TAAt ××−= −

p = t2× 1T Multiply by the 16-bit 1-average threshold.

t3 = RoundStrip(p,17) Shift the product right 17 bits and round. t3: 15-bit unsigned.

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 38

Supprimé : 39







nT = A0 + t3 nT : 15-bit unsigned. This is the final n-average threshold.

8494 Output: 8495

nT 15-bit unsigned detection threhold for n averages. 8496

8497 A numerical study shows that this fixed-point algorithm has a maximum error of 0.13% compared to a 8498 floating point version of the procedure. 8499 8500 8501

14. PUSCH MIMO 8502

14.1. CHANGES WITH RESPECT TO PREVIOUS RELEASE 8503

In this section, we present for information some foreseen modifications for MIMO receiver but for 8504

LA3.x, we restrict ourselves to 3GPP alignement for MIMO which has already been covered 8505

during the LA1.0 release. 8506

Moreover, the zero-padding processing is currently under further evaluation. Notice that FPGA C 8507

team’s feedback is that if zero padding were required for MIMO separation this could be an issue 8508

since MIMO separation processing is in the critical UL processing path. Therefore, the current 8509

working view is that this feature should be proposed for in a future release only if a significant 8510

gain is expected. 8511

We describe the reduction of the time domain filter size for information, but for LA3.0, we consider 8512

only the use of half length filter, i.e. filters of the same size of LA0.x and LA1.x. Therefore, noise 8513

estimation is the same as in LA0.x and L1.x. 8514

14.2. REFERENCE SIGNALS MULTIPLEXING 8515

As for SRS, the reference signals of both UEs are Code Domain Multiplexed. However, the main 8516

difference is that only two users are multiplexed, and that the cyclic shifts are multiples of 12

2π for 8517

PUSCH instead of 8

2π for SRS, see [1]. We assume no pre-defined values for both cyclic shifts of 8518

the multiplexed users, keeping in mind however that a larger separation will give better 8519 performance. 8520

As for SRS, we will possibly use zero padding: M denotes the number of Resource Elements 8521

occupied by both users, and DFTM denotes the IDFT size after zero padding. We assume the 8522

DFTM is a multiple of 12. 8523

We will denote ( ) ( )12

CS2

uu πα = the cyclic shift of reference sequence of user u=0, 1, with 8524

( ) { }11,,1,0CS K∈u . 8525

The reference signal sequence of user u is defined as ( )( ) ( ) ( )nRenR nuju0

..α= where the sequence 8526

( )nR0 is the reference sequence with zero cyclic shift as defined in Section 5.5.1 of [1]. ( )nR0 will 8527

be called the mother Cazac sequence in the following. 8528

Supprimé : 4.0.1

Supprimé : 2







As for SRS, we define a corresponding equivalent time domain delay 8529

( ) ( ) ( ) ( )12

.12

222

DFTDFTDFT MuCS

uCSMuMud ==−= π

ππα

. This delay corresponds to a time 8530

domain cyclic shift of the time domain CAZAC sequence 0r obtained from 0R by size DFTM IDFT. 8531

Actually, if ( )( ) DFTu Mnnr <≤0, is the IDFT of the sequence ( ) )(nR u padded with ( )M−DFTM 8532

zeros at the end, then we have: ( ) ( ) ( )

−=DFTM

DFTu MuCSnrnr

mod0 12

. 8533

As in LA0.1 and LA1.0, let us denote ( )uτ the effective timing position of user u, giving the position 8534

of the center of gravity of the channel of user u. This is the result of the timing offset estimation of 8535 this user. If we are sufficiently confident on the reactivity of the timing advance loop, we can 8536

replace ( )uτ by TAτ (which is assumed to be CP/4). 8537

We define the equivalent delay corresponding to the IDFT/DFT window length as: 8538

( ) ( )

≈

=

FFT

DFTTA

FFT

DFTDFT N

M

N

Mu .round.uround τττ 8539

Where round states for an approximation to the closest integer. After compensation by mother 8540 CAZAC sequence and IDFT, the position of the center of gravity of the user u is located at the 8541 sample: 8542

( ) ( ) ( ) ( ) ( )DFTDFT MFFT

DFTDFT

MDFT

DFT

N

MMuCSu

uMud

modmod

.uround12

.2

+−=

+−= ττπα

8543

As for SRS, we can use the targeted timing offset TAτ as described in section 6.2.2., leading to 8544

( ) ( )DFTMFFT

DFTTA

DFT

N

MMuCSud

mod

.round12

+−= τ 8545

The second term in the sum is constant, common for all users and can thus be pre-computed. 8546

8547

14.3. REFERENCE SIGNALS DEMULTIPLEXING 8548

14.3.1 ALGORITHM OVERVIEW 8549

The processing below should be done on each Rx antenna, for each pilot block, and for MIMO 8550

users only. After de-multiplexing, we have the pilot channel estimation of both UEs, and the rest of 8551 the pilot processing is done exactly the same way as for the SIMO case (The only difference being 8552

the noise and power estimation). 8553

8554 8555 8556 8557 8558 8559 8560

8561

Figure 14-1 Reference signal demultiplexing. 8562

Reference signal demultiplexing

Pilot channel estimation of UE2

Pilot channel estimation of UE1

Supprimé : 4.0.1

Supprimé : 2







The reference signal separation uses the same processing as for SRS except that this is for U=2 8563

users. 8564

At the receiver, the front-end processing is the same as for SIMO, i.e. FFT + sub-carrier de-8565 mapping + pilot channel estimation performed with the mother CAZAC sequence. The code 8566

compensation is done with respect to the mother CAZAC sequence 0R , i.e. this is a term by term 8567

multiplication by its conjugate sequence ( )*0R . 8568

As for SRS, in order to limit the effect of Gibbs phenomenon on the band edges, we can introduce 8569

some zero padding before IDFT. This is done by adding PadM zeros at the end of the 8570

sequence: PadDFT MMM += . The foreseen value for PadM is 24, but this still needs further 8571

evaluation so that it can be modified in a future version of the document. Notice that 0=PadM is 8572

the case where we do not have padding, and this is equivalent to LA2.0 implementation (and 8573 should also be supported in LA3.0). 8574

8575

14.3.2 ZERO PADDING 8576

This block is additional to LA1.x processing, its aim being to enhance the channel estimation at the 8577

band edges (to be further evaluated). As precised at the beginning of this chapter, the current 8578

working view is that this feature should be proposed in a future release only if a significant gain is 8579

expected. 8580

mx ( )mx0

FFT


CP removal

Conjugate CAZAC=( )∗0R

Zeros padding +IDFT


UE 1+ circular shift


UE 2 + circular shift

DFT+ zeros removal

DFT+ zeros removal

pilot0

~=uH of UE0

pilot1

~=uH of UE1

Supprimé : 4.0.1

Supprimé : 2







Actually, time domain filtering is equivalent to a frequency domain circular convolution that 8581

degrades the estimation accuracy at the edges because of the discontinuity. Without zero padding, 8582

the Gibbs phenomenon will have greater impact than with zero padding. 8583

If M denotes the number of tones occupied by the users before zero padding, then zero padding 8584

consists in adding padM zeros, at the end of the sequence: padDFT MMM += . 8585



M Integer u11 I 1 Number of tones occupied by

the users

DFTM Integer u11 I 1 IDFT/DFT size


(12I,12Q) I M Post CAZAC compensation

frequency domain signal

HDFT_PerAnt Complex Integer

(12I,12Q) O DFTM IDFT input

8587 Table 14-1: Interface definition for Zero Padding 8588

8589

14.3.3 IDFT / DFT 8590

IDFT is defined as ∑−

=

=

1

0

2expˆDFTM

i DFT

pilotik M

ikjHh π 8591

DFT is defined as ∑−

=

−=

1

0

_ 2exp1ˆ

M

k DFTk

DFT

demuxpiloti M

ikjh

MH π where h has been time 8592

windowed. 8593 8594


The interface signals and their specifications for this block are given in the next table below for 8596

IDFT and DFT respectively. 8597


DFTM Integer u11 I 1 IDFT size (after zero padding)

FreqSample Complex

Integer

(12I,12Q) I

DFTM Pre-IDFT frequency domain samples

TimeSample Complex

Integer

(12I,12Q) I

DFTM Time domain samples

Table 14-2 : Interface definition user separation : IDFT 8598 8599

8600


Supprimé : 4.0.1

Supprimé : 2







DFTM Integer u11 I 1 DFT size

TimeSample0 Complex

Integer

(12I,12Q) I

DFTM samples (time domain) for pilot0

TimeSample1 Complex

Integer

(12I,12Q) I

DFTM samples (time domain) for pilot1

FreqSample0 Complex

Integer

(12I,12Q) O

DFTM samples (freq domain) for pilot 0

FreqSample1 Complex

Integer

(12I,12Q) O

DFTM samples (freq domain) for pilot1

Table 14-3 : Interface definition user separation : DFT 8601


8603 Here are the Xilinx’s block parameters : 8604

n2 Power of 2 prime decomposition n3 Power of 3 prime decomposition n5 Power of 5 prime decomposition

DFT

direction 1 (forward) n2 Power of 2 prime decomposition n3 Power of 3 prime decomposition n5 Power of 5 prime decomposition

iDFT

direction 0 (inverse)

8605

The figure below shows the scaling of data during operations : 8606

8607

left shift

<< p

iDFT xilinx

DFT xilinx

rescalin

g

1.0 2p 2p-sf M.2p-sf-si 1.0 scale :

DFT

radix-2 radix-3 radix-5

M inputs

(12I,12Q) (12I,12Q)

Rescaling

Time

domain window

zero

padding

iDFT

radix-2 radix-3 radix-5

(18 I, 18 Q) (18 I, 18 Q) (18 I, 18 Q)

exponent sf exponent

si

Bit width extension 12 to 18 bits << 6

Supprimé : 4.0.1

Supprimé : 2







Figure 14-2 Scaling of data during sequence compen sation separation 8608 8609

The rescaling (last step in figure above) consist in a multiplication by : 8610

)(2.

1if ssp

DFTM +− 8611

, 8612

which is equivalent to ))((log)(

))((log

2

2

2

12DFTsif

DFTs

MfloorqsspDFT

Mfloorq

M +++−

+

⋅ 8613

which is equivalent to r

q2

1⋅ , 8614

where DFT

Mfloorq

Mq

DFTs ))((log22 +

= , ))((log)( 2 DFTsif Mfloorqsspr +++−= 8615

Finally, the rescaling consist in a multiplication by r

q2

1⋅ where 8616

DFT

Mfloorq

Mq

DFTs ))((log22 +

= 8617

)( if sshr +−= , ))((log2 DFTs Mfloorqph ++= , qs=14, p=6 8618

8619

So, rescaling is done for each pilot sample for both real and imaginary parts: 8620

8621

8622 8623

)( if sshr +−= , )( DFTMh is tabulated , )( DFTMq is tabulated 8624

8625

input size prime decomposition n2 n3 n5 h q 12 12=2x2x3 2 1 0 23 10923 24 24=2x2x2x3 3 1 0 24 10923 36 36=2x2x3x3 2 2 0 25 14564 48 48=2x2x2x2x3 4 1 0 25 10923 60 60=2x2x3x5 2 1 1 25 8738 72 72=2x2x2x3x3 3 2 0 26 14564 96 96=2x2x2x2x2x3 5 1 0 26 10923 108 108=2x2x3x3x3 4 3 0 26 9709 120 120=2x2x2x3x5 3 1 1 26 8738 144 144=2x2x2x2x3x3 4 2 0 27 14564 180 180=2x2x3x3x5 2 2 1 27 11651 192 192=2x2x2x2x2x2x3 6 1 0 27 10923 216 216=2x2x2x3x3x3 3 3 0 27 9709 240 240=2x2x2x2x3x5 4 1 1 27 8738

18

>>r

12

q

u14

32

Sat12

Supprimé : 4.0.1

Supprimé : 2







288 288=2x2x2x2x2x3x3 5 2 0 28 14564 300 300=2x2x3x5x5 2 1 2 28 13981 324 324=2x2x3x3x3x3 2 4 0 28 12945 360 360=2x2x2x3x3x5 3 2 1 28 11651 384 384=2x2x2x2x2x2x2x3 7 1 0 28 10923 432 432=2x2x2x2x3x3x3 4 3 0 28 9709 480 480=2x2x2x2x2x3x5 5 1 1 28 8738 540 540=2x2x3x3x3x5 2 3 1 29 15534 576 576=2x2x2x2x2x2x3x3 6 2 0 29 14564 600 600=2x2x2x3x5x5 3 1 2 29 13981 648 648=2x2x2x3x3x3x3 3 4 0 29 12945 720 720=2x2x2x2x3x3x5 4 2 1 29 11651 768 768=2x2x2x2x2x2x2x2x3 8 1 0 30 10923 864 864=2x2x2x2x2x3x3x3 5 3 0 30 9709 900 900=2x2x3x3x5x5 2 2 2 30 9321 960 960=2x2x2x2x2x2x3x5 6 1 1 30 8738 972 972=2x2x3x3x3x3x3 2 5 0 30 8630 1080 1080=2x2x2x3x3x3x5 3 3 1 30 15534 1152 1152=2x2x2x2x2x2x2x3x3 7 2 0 30 14564 1200 1200=2x2x2x2x3x5x5 4 1 2 30 13981

8626


The construction of the time domain filters from the variables ( )ud is done exactly the same way 8628

as for SRS. However, there is a difference with both LA1.0 and SRS. In LA1.0, the filters occupied 8629 half of the IDFT output, whereas here the size of the time domain filters is configurable. In V1.1 of 8630

this document, we limited the filter lengths to an equivalent of FT µs only to remove much of the 8631

noise. This FT µs window should be chosen sufficiently large to capture most of the channel’s 8632

energy. 8633

The current working version is to use the same time domain filter length as for LA1.x in LA3.0, and 8634 to eventually reduce this filter in future releases depending on further analysis. Therefore, the 8635 reduced filter length description below is only given for information and should not be implemented 8636 in LA3.x. 8637

Therefore, the filters cover half of the DFT size, which corresponds to µsTF 34.33= . For post 8638

LA3.0, the foreseen value for FT is 8µs. However, this filter size reduction is possible only if the 8639

multipath profile is resolvable in time, which means that this is not possible for small PRB sizes. 8640 Therefore, we should define a PRB size under which classical full length filters will continue to be 8641 used. 8642

In the following, we describe the general case of variable filter length, even if in LA3.0 we use the 8643 same length as in LA1.0. 8644

On the contrary of the SRS, time domain noise removal is not performed because it would 8645 complicate the noise estimation algorithm. If we have some margin at the end, the processing 8646 could be enhanced by adding time domain noise removal feature. 8647

Time domain filters are thus constructed by cyclically selecting the equivalent of FT µs on each 8648

side of ( )ud . 8649

The half length filter corresponds to a number of samples equal to 8650 ( )

DFTFDFT MML ×=××= 25.07.68/5122/T0 for both 5MHz and 10MHz (same as in 8651

LA0.x). 8652

For drop 2, ( ) DFTFDFT MML ×=××= 06.07.68/5122/T0 8653

8654

Supprimé : 4.0.1

Supprimé : 2







8655

8656 Figure 14-3 Time domain filtering for UE separation 8657

8658 8659

14.3.5 TIME DOMAIN CYCLIC SHIFT 8660

As for SRS, after time domain filtering, we have to remove the time domain delay of the different 8661 users by shifting them. On the contrary of SRS however, the users are shifted to the right because 8662 we did not do any user re-ordering (the user re-ordering was done for SRS to ensure maximum 8663 reuse from LA1.0). 8664

Notice that as for SRS, we have to remove only the shift caused by CAZAC sequence cyclic shift 8665 and not the shift caused by their timing offset. Therefore, we have to cyclically shift the user u by 8666

( )12

DFTMuCS samples on the right. 8667

In the figure below, we describe the cyclic shift of the non-zero coefficients after time domain 8668 filtering for the user u. 8669

1

0 1−DFTM ( )ud

( )0L ( )0L

DFTτ

( )02L

Supprimé : 4.0.1

Supprimé : 2







8670

Figure 14-4 : Time domain cyclic shift of user u before DFT 8671 8672

The lengths of the right and left part of the non-zero signal after shift are equal to: 8673

( ) ( ) ( ) ( )uLuL DFTl τ+= 0 8674

( ) ( ) ( ) ( )uLuL DFTr τ−= 0 8675

The interface signals and their specifications for this block are given in the next table below for a 8676

given user. 8677

8678



CS(u) Integer u3 I 1 Cyclic shift index

H0 Complex Integer

(12I,12Q) I MDFT samples per antenna (time

domain)

H1 Complex Integer

(12I,12Q) O MDFT samples per Ue per antenna

(time domain)

Table 14-4 : Time domain cyclic shift interface def inition 8679 8680 8681

14.3.6 DFT 8682

DFT is applied per user after time domain cyclic shift. 8683

14.3.7 ZERO REMOVAL 8684

After DFT, we remove the ( )MM DFT − samples at the end of the sequence to go back to a 8685

sequence of length M . 8686

1

0 1−DFTM

Cyclic shift by ( )12

DFTMuCS samples

( )ud

Position of the samples after time domain cyclic shit and before DFT

t

( )( )uL l ( )( )uL r

Supprimé : 4.0.1

Supprimé : 2







The interface signals and their specifications for this block are given in the next table below on a 8687

per user basis. 8688

8689



M Integer u11 I 1 Number of allocated tones


(12I,12Q) I MDFT samples per antenna (frequency domain)


(12I,12Q) O M samples per antenna (frequency domain)

Table 14-5 : Zero removal interface definition 8690


14.4.1 CARRIER FREQUENCY OFFSET (CFO) PROCESSING 8692

14.4.1.1 FREQUENCY OFFSET ESTIMATION 8693

The frequency offset estimation is done on a UE by UE basis, the same way as for the SIMO case 8694

after users pilot signals separation. 8695

14.4.1.2 FREQUENCY OFFSET COMPENSATION FOR PILOT 8696

BLOCKS 8697

Pilot block frequency offset compensation is done exactly the same way as for the SIMO case. We 8698

can use a variable length for frequency domain filtering, the single tap case being a particular case 8699

of multi-tap. Notice that since MIMO is only supported for low speeds, only phase rotation is 8700

performed, and the multi-tap filtering to remove inter-carrier interference is not applied. 8701

14.4.2 TIMING OFFSET ESTIMATION 8702

The timing offset estimation is done on a UE by UE basis, the same way as for the SIMO case after 8703

users pilot signals separation. 8704


Except the de-multiplexing part, the time and frequency domain filtering are the same as in the 8706

SIMO case independently for each user. 8707

First, frequency domain filtering is done on both pilot blocks. Then time domain filtering is done by 8708

Time domain MMSE with filters depending on speed and SNR. 8709

In LA0.1 and LA1.0, MIMO was reserved for pedestrian users, so that the Lut corresponding to 8710

3km/h was always used irrespective of the estimated speed of the user. We keep the same 8711

approach for LA3.0. 8712

Supprimé : 4.0.1

Supprimé : 2







14.6. MEASUREMENTS 8713

The different measurements are done on a UE by UE basis after user’s reference signals 8714

separation. 8715

14.6.1 NOISE ESTIMATION FOR FDE 8716

In this section we describe the noise estimation as seen prior to reference signal separation, i.e. as 8717

seen at the input of the Frequency Domain Equalize (FDE) in the data blocks. 8718

Notice that this FDE noise estimate is the noise that is sent to higher layers for long term noise 8719

measurements and SNR computations. 8720

The description of the noise estimation is different from LA0.x since it uses a configurable time 8721

domain filter size. However, in practice, for LA3.0, the implementation is the same. 8722

The general methodology is the same as in LA1.0, i.e. first perform SIMO noise estimation and 8723

then correct the result by a scaling factor. This approach is valid for every time domain filter length. 8724

As for LA1.0, both noise estimates of both users should be averaged after per-user noise 8725

estimation. 8726

As for LA1.0, the SIMO noise estimation algorithm assumes that the noise at the output of the user 8727 separation is white. This hypothesis is not satisfied, and this has non-negligible consequences in 8728 practice since the noise power has been observed to be under-estimated. We will thus have to 8729 further multiply the estimated noise by a factor which depends on the window size and shape 8730 applied in time domain for UE separation.. In practice, this scaling coefficient has been identified as 8731

an increasing function of the number of PRB, that will be denoted as uκ . This is done for each UE. 8732

8733 We define the following quantities: 8734 8735

( )( ) ( )( ) ( )( )DFTDFT

rl

M

uL

M

uLuL 0

0

2=+=ρ which is nothing but the ratio between the time domain filter 8736

length and the DFT size (i.e. the proportion of noise kept by the filter). 8737 8738

( )( ) ( )( )2 ,1 ,

sin.2

12sin12sin

=

++

−

= n

M

nM

LM

nL

M

n

DFTDFT

r

DFT

l

DFTn

π

ππρ 8739

8740 We then define the parameter: 8741

DFTM

M

+−= 210 9

2

9

8

3

2 ρρρθ 8742

Notice that this parameter should depend on the user through the timing offset, but it can be 8743

common to both users if this offset is assumed equal to TAτ . 8744

8745 Since SIMO like noise estimation if the same for both users, we drop the user index and consider 8746

pilot~H the output of user separation for any user. As for SIMO, we consider the frequency domain 8747

filtering of pilot~H with 1Noise =K and 1Noise =G , which result will be denoted as [ ]a

iai

a

i 1,0, ,ζζζ = 8748

as for SIMO. 8749 MIMO noise estimation is based on the fact that: 8750

Supprimé : 4.0.1

Supprimé : 2







8751

22

pilot .~ σθζ =

−Ε H 8752

Where 2σ is equal to the noise power we want to estimate, i.e. the noise power before separation. 8753 8754 To keep the same implementation as in LA1.0, we can first perform SIMO like noise estimation to 8755 obtain: 8756 8757

( )

−Ε

+−

≈−= ∑ ∑=

−

=

2pilot

NoiseNoise

1

0

2

1

2

,pilot

,,2 ~

21

1

1~2

1ˆ

0

ζζσ H

GK

HT p

M

i

apiapiSIMO 8758

8759

( )

−Ε≈

2pilot2 ~

2

3ˆ ζσ HSIMO 8760

8761 8762 We can then compute: 8763 8764

( ) ( )22pilot2 ˆ

3

2~1ˆ SIMOMIMO H σ

θζ

θσ =

−Ε= 8765

8766 This means that finally, we can keep the same implementation as in LA1.0 with 8767 8768 8769

M

M DFTu ×

+−×==

210 9

2

9

8

3

23

2

3

2

ρρρθκ 8770

8771

i.e.: M

M DFTu ×

+−=

210 43

3

ρρρκ 8772

8773

With the variables nρ defined as above. 8774

Notice that we have one such noise estimate per UE. Since these two estimates correspond to the 8775 same noise power, we can average them to obtain a single estimate of the noise power prior to 8776 separation: 8777

8778

( ) ( ) ( ){ } ( ) ( ){ }22uSIMO2

21uSIMO1

22uSIMO2

21uSIMO1

2MIMO ˆˆ

2

1ˆˆ

2

1ˆ =

==

==

==

= +=+= σκσκσκσκσ uuuu 8779

8780

Notice also that the SIMO noise estimation can be per antenna or common to all antennas 8781

depending on the chosen option. 8782

8783 In a future version of the document with reduced filter sizes not equal to half the DFT size, the new 8784

values of the multiplicative parameter 210 43

3

ρρρ +− will be given as a look-up table depending 8785

on the number of PRBs. 8786 8787 For drop 1, we use the same filter size as for LA0.x. This results in the following parameters: 8788

Supprimé : 4.0.1

Supprimé : 2







• 72.1143

3

210

=+− ρρρ

for an allocation of 1 PRB (fix point representation 1500) 8789

• 82.1243

3

210

=+− ρρρ

for an allocation of 2 PRBs (fix point representation 1640) 8790

• 22.1343

3

210

=+− ρρρ

for an allocation of more than 2 PRBs (fix point representation 8791

1692) 8792

If we do not use zero padding before IDFT, this results in the same values of uκ as in LA0.x. 8793

Supprimé : 4.0.1

Supprimé : 2







8794 8795 8796

14.6.2 NOISE ESTIMATION FOR TIME DOMAIN MMSE CHANNEL 8797

ESTIMATION 8798

The noise at the input of the time domain channel estimation has to be derived from the 8799

FDE noise taking account three effects: 8800

UE separation

User 1, Frequency domain filtering with

1NoiseNoise == GK

followed by noise estimation (same as

for SIMO)

1=× uκ 2=× uκ

1/2

From mother CAZAC compensation ( )2

MIMOσ to be

estimated


1NoiseNoise == GK


for SIMO)

One estimation of ( )2MIMOσ

common to both users

To Frequency Domain Equalizer

Supprimé : 4.0.1

Supprimé : 2







• The zero padding from M to DFTM which reduces the per sample noise power at 8801

the IDFT output compared to the non-zero inputs 8802

• Time domain filtering to separate the users reduces the amount of noise 8803

experienced after user separation 8804

• Frequency domain filtering which, as in LA1.0, introduces a processing gain of 8805

(K+2G) 8806

8807

These three effects can be taken into account by multiplying the FDE noise power by the 8808

factor ( )

+

GKM

L

M

M

DFTDFT 2

12 0

, i.e: 8809

( )( )

( )20

2MMSE-TMIMO ˆ

2

12ˆ MIMO

DFTDFT GKM

L

M

M σσ

+

= 8810

For LA2.x, ( )

+=

+

DFTDFTDFT M

M

GKGKM

L

M

M

2

1

2

1

2

12 0

so that the only difference with 8811

LA1.x lies in the factor

DFTM

M coming from zero-padding. Therefore, if zero-padding is not 8812

applied, the processing is exactly the same as for LA1.x. This is the case for LA2.x. 8813

The diagram below describes the global noise processing for MIMO, for both FDE and time domain 8814

MMSE. 8815

8816

8817

8818

8819

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Retrait :Gauche : 3,14 cm, Suspendu :0,63 cm, Avec puces + Niveau: 1 + Alignement : 3,14 cm +Tabulation après : 3,77 cm +Retrait : 3,77 cm







8820 Figure 14-5 Noise estimation in a MIMO configuration 8821

8822 8823


Since for LA3.0 the implementation is the same as in LA1.x, the fixed point implementation 8825

is still valid. 8826

8827

8828

UE separation


1NoiseNoise == GK


for SIMO)

1=× uκ 2=× uκ

1/2

From mother CAZAC compensation ( )2

MIMOσ to be

estimated


1NoiseNoise == GK


for SIMO)

One estimation of ( )2MIMOσ

common to both users To Frequency Domain Equalizer

To higher layers for long term noise averaging

To T-MMSE filter selection

( )

+

+

GKM

L

M

M

DFTDFT 2

112 0

Supprimé : 4.0.1

Supprimé : 2





Code de champ modifié

Code de champ modifié


Supprimé : 14

Supprimé : 5







8829

8830

8831

8832

8833

8834

8835

8836

8837

8838

8839

8840

8841

8842

8843

8844

8845

8846

8847

14.7. DEMODULATION 8848

14.7.1 JOINT FREQUENCY DOMAIN EQUALIZATION AND CFO 8849

COMPENSATION FOR DATA BLOCKS 8850

This function is called for each scheduled UE, once per data OFDM symbol. Its inputs are the two 8851

outputs sequences [ ]uMuu XXX ,1,0 −= K of the sub-carrier demapping for data blocks (one 8852

sequence per user), the CFO estimations and channel estimations for both UEs, and its output is 8853 the DFT-spread symbol estimations, sent to the QAM demapper. 8854 In order to limit the receiver complexity, we will consider single tap filtering for data blocks 8855

frequency offset compensation. This means that we correct only the global rotation of the OFDM 8856

symbol since the beginning of the TTI, but neglect the effect of Inter-Carrier Interference (ICI). 8857

On the contrary of what happens for SIMO, since the signals of both UE are transmitted on the 8858

same physical channel, frequency offset compensation and frequency domain equalization should 8859

be done in a single step in theory. However, we will see that de-rotation can be done on a UE by 8860

UE basis after frequency domain equalization. 8861

Let us denote 1ε and 2ε the relative frequency offsets (i.e. frequency offsets normalized by the 8862

sub-carrier spacing) of UE 1 and 2, with 0.0667<uε , u=1, 2. 8863

Since no ICI is removed, the MMSE is still done on a tone by tone basis as in the SIMO case, 8864 except that the output will be a vector of size 2x1. 8865

>>1

>>11

Ku

1/(K+2G)

DSP

Noise (prb,0)

UE 1

>>2 205

u24

u28

u23

u20 Noise for T-MMSE

Noise (prb,0)

UE 0

u11

>>2

>>5

>>5

u24

>>2

u24

>>2

u24

>>5

>>5

u23

Sat 23

Noise (prb,0)

UE 1

Sat 23

DSP

u23

Noise

(prb,0)

UE 0

Ku

u11

u24

u28

Supprimé : 4.0.1

Supprimé : 2







The reference for rotation is the middle of the OFDM symbol. The exponentials corresponding to 8866 the CFO rotation for the current OFDM symbol (from the beginning of the TTI to the middle of the 8867 current OFDM symbol) are equal to 8868

( )

Ι

FFTN

mj επ .2exp 8869

where ( )mΙ is equal to the number of time domain samples from the beginning of the TTI until the 8870

middle of the m-th OFDM symbol, including all CP and pilot blocks. We recall that FFTN is the size 8871

of the front-end FFT. 8872

In the following, we will ignore the tone index. Let us denote uaH , the frequency domain channel 8873

coefficient from user u to Rx antenna a for the tone of interest. The 2x2 channel matrix used in the 8874 MIMO receiver model is equal to: 8875

( )( ) ( )

( ) ( )

( )

( ) ( )212

1

2221

1211

222

211

122

111

21

,..2

exp0

0.2

exp

.2exp

.2exp

.2exp

.2exp

,

εεεπ

επ

επεπ

επεπ

εε

Φ=

Ι

Ι

=

Ι

Ι

Ι

Ι

=

H

N

mj

N

mj

HH

HH

HN

mjH

N

mj

HN

mjH

N

mj

FFT

FFT

FFTFFT

FFTFFTmH

8876

Where ( )( )

( ) ..2

exp0

0.2

exp

, ,2

1

212221

1211

Ι

Ι

=Φ

=

FFT

FFT

N

mj

N

mj

HH

HHH

επ

επ

εε 8877

Neglecting ICI, the signal at the output can by be written as : 8878

( ) ( ) nSPHnSPX m +Φ=+= 2121 ,., εεεεH 8879

8880

Where : 8881

•

=

2

1

X

XX is the input signal vector from antenna 1 and 2 respectively. 8882

•

=

2

1

s

ss is the vector containing the post-DFT information of user 1 and user 2 respectively. As 8883

for the SIMO case, we assume that these symbols have unit energy. 8884

•

=

2

1

n

nn is the noise vector on antenna 1 and 2 respectively, ),0(~ 2

aa Nn σ , for a = 1 and 2. Let 8885

us denote:

=

22

21

0

0

σσ

D . 8886

8887

Supprimé : 4.0.1

Supprimé : 2







•

=

2

1

0

0

P

PP is the transmit power matrix for user 1 and 2, known by the receiver, and 8888

=

2

1

0

0

P

PP . Notice that the total resulting powers 1P and 2P can be included in the 8889

channels, so that we can use the model with 121 == PP without loss of generality. 8890

Let us denote ( ) ( )22

21 σσγ = . Then we have: 8891

8892

( )( ) ( ){ } ( )( ) XDHIHDHs HHHH 121

1

211

21 ,,,ˆ −−− Φ+ΦΦ= γεεγεεβγεε 8893

8894 8895

Since the matrix ( )21,εεΦ is orthogonal, this estimation is equal to: 8896

8897

( )( ) { } XDHIHDHs HHH 11121 .,ˆ −−− +Φ= γγγεε 8898

8899 8900 This equation means that we can proceed by first computing the MMSE prior to any CFO 8901

correction, which corresponds to { } XDHIHDH HH 111 −−− + γγγ , and then do a CFO de-rotation 8902

on a UE by UE basis, which corresponds to the multiplication by the diagonal de-rotation matrix 8903

( )( )( )

( )

Ι−

Ι−=Φ

FFT

FFTH

N

mj

N

mj

2

1

21.2

exp0

0.2

exp

,επ

επ

εε . 8904

8905

8906 8907

Figure 14-6 MMSE and CFO de-rotation in a MIMO conf iguration 8908 8909

8910 In the following, we describe the details of the MIMO MMSE implementation, corresponding to the 8911

processing{ } XDHIHDH HH 111 −−− + γγγ , i.e. before CFO de-rotation. 8912

8913

The matrix ( )

( )

=−

21

221

0

0

σσγ D involves no division. We can thus also compute the inversion 8914

using a single division as explained below. 8915 8916 8917 We then have: 8918 8919

Frequency domain

equalizer : 2x2 MMSE

CFO de-rotation, UE1

CFO de-rotation, UE2

Demodulation, UE1

Demodulation, UE2

Rx 1

Rx 2

Supprimé : 4.0.1

Supprimé : 2







( ) ( ){ } ( ) ( ) ( ) ( ) ( ) ( ){ }( ) ( ) ( ) ( ){ } ( ) ( ){ } ( ) ( )

++++++

=+

∗∗

∗∗

−

22

21

2

222

1

2

122

222212

112112

2

21222

111122

22

22

1

2

212

1

2

112

2

1

σσσσσσσσσσσσ

γγ

HHHHHH

HHHHHH

IHDH H

8920

The inverse of the matrix above can be compute easily by the general formula: 8921

8922

8923

Which gives in this particular case: 8924

( )( ) ( ){ } ( ) ( ) ( ) ( ) ( ) ( ){ }

( ) ( ) ( ) ( ){ } ( ) ( ){ } ( ) ( )

+++−+−++

=+

∗∗

∗∗

−−

22

21

2

212

1

2

112

222212

112112

2

21222

111122

22

22

1

2

222

1

2

122

2

11

1

σσσσσσσσσσσσ

γγ

HHHHHH

HHHHHH

D

IHDH H

8925

8926

8927

where 8928

( ) ( ){ } ( ) ( ){ } ( ) ( ){ } ( ) ( ){ }( ) ( ) ( ) ( ) 2

22212

112112

2

22

21

2

212

1

2

112

22

22

1

2

222

1

2

122

2

∗∗ +−

++++=

HHHH

HHHHD

σσ

σσσσσσσσ 8929

We also have: 8930

( ) ( ) ( ) ( )( ) ( ) ( ) ( )

++= ∗∗

∗∗−

2222

11122

2

2212

11112

21

XHXH

XHXHXDH H

σσσσγ 8931

8932

Simplified implementation 8933

With the hypothesis of identical noise variance on both antennas, and with 8934

( ) ( )( )22

21

2 ˆˆ2

1ˆ σσσ += : 8935

{ } { }{ } XHIHH

XHIHHXDHIHDH

HH

HHHH

12

2142111

−

−−−−

+=

=+=+

σ

σσσγγγ 8936

So that the final formula simplifies to: 8937

( ){ } ( ) ( ){ }

( ) ( ){ } { }

+++−+−++

=+

∗∗

∗∗

−

22

21

2

1122211211

2122111222

22

2

12

12

1

σσ

σ

HHHHHH

HHHHHH

D

IHH H

8938

And: 8939

Supprimé : 4.0.1

Supprimé : 2

( )

−−

=

−

1121

12221

2221

1211

det

1

MM

MM

MMM

MM







{ }{ } ( ) ( ) 2

2221121122

21

2

1122

22

2

12∗∗ +−++++= HHHHHHHHD σσ8940

( ) ( )( ) ( )

++= ∗∗

∗∗

222112

221111

XHXH

XHXHXH H

8941

8942

Figure 14-7 Frequency domain equalization + derotation for MIMO case. This is done for each data OFDM 8943 symbol where MIMO users are transmitting. 8944




NoiseAv Integer u30 (u24) I 1 Noise power averaged over

Main and Div antennae


H_Main1 Complex Integer

(12I, 12Q) I

(12 / N_mmse_output)

x ( 0M /K) Estimated Channel on the

Main antenna for Ue1


(12I, 12Q) I

(12 / N_mmse_output)

x ( 0M /K) Estimated Channel on the Div

antenna for Ue1

H_Main2 Complex Integer

(12I, 12Q) I

(12 / N_mmse_output) x ( 0M /K)

Estimated Channel on the Main antenna for Ue2


(12I, 12Q) I

(12 / N_mmse_output) x ( 0M /K)

Estimated Channel on the Div antenna for Ue2

K Integer u3 I 1 Number of sub-carriers over

which the channel is assumed to be constant

N_mmse_output Integer u3 I 1 Number of OFDM symbols over which the channel is

assumed constant

Filter 1 Complex Integer

(8I,8Q) O

2x(12 / N_mmse_output) x

( 0M /K) Filter coefficients for user 1


(8I,8Q) O

2x(12 / N_mmse_output) x

( 0M /K) Filter coefficients for user 2

m Integer u5

O 2x(12 /

N_mmse_output) x ( 0M /K)

Scaling factor (order)

8947

Frequency domain equalizer :

2x2 MMSE

Signals from Rx 1 and 2 on

tone of interest

(=output of sub-carrier demapping)

Symbol estimation for UE1 (iDFT)

Symbol estimation for UE2 (iDFT)

To UE by UE CFO de-

rotation + QAM

demapping

Supprimé : 4.0.1

Supprimé : 2







Table 14-6 :Interface definition for MMSE MIMO filt er computation 8948 8949


m Integer u4

I 2x(12 /

N_mmse_output) x ( 0M /K)

Scaling factor (order)

DataMain Complex Integer

(12I, 12Q) I 12 x 0M

Received Data bits for the12 data blocks of the TTI on the

Main antenna.

DataDiv Complex Integer

(12I,12Q) I 12 x 0M

Received Data bits for the12 data blocks of the TTI on the

Div antenna.

K Integer u3 I 1 Number of sub-carriers over

which the channel is assumed to be constant

N_mmse_output Integer u3 I 1 Number of OFDM symbols over which the channel is

assumed constant


(8I,8Q) I

2x(12 / N_mmse_output)

x ( 0M /K) Filter coefficients for user 1


(8I,8Q) I

2x(12 / N_mmse_output)

x ( 0M /K) Filter coefficients for user 2

FilteredData1 Complex Integer

(8I, 8Q) O 12 x 0M Filtered Data through MMSE

for Ue1

FilteredData2 Complex Integer

(8I, 8Q) O 12 x 0M Filtered Data through MMSE

for Ue2

8950 Table 14-7 :Interface definition for MMSE MIMO filt ering 8951

8952

PRACTICAL IMPLEMENTATION 8953

As in the SIMO case, the 2x2 MMSE should be applied in two steps, one step computing the filters(this is 8954

done in the DSP), and the second step applying these filters (in the FPGA). 8955

8956 To compute the filters, we can use the fact that the channel estimation is constant over time-frequency 8957

rectangles in order to limit the number of divisions (see section 14.5). As in the SIMO case, we will have 8958

Filters computations for Frequency domain

Equalizer

FPGA

Channel estimates (after Time domain filtering) Filter applications to get pre-IDFT estimated symbols

8 I, 8Q

12 I, 12Q

Supprimé : 4.0.1

Supprimé : 2







one division per KxL rectangle. Since we have two users, the size of these rectangles over which the 8959

filters are constant is equal to the minimum size of both users. 8960

Correlation Matrix computation 8961

8962 (*) g=3 if short term measurement (recommended behaviour) and g=2*e+1 if long term measurement. 8963

8964 8965

8966 8967

Normalize M11, M22, M21 & M12 8968

12

12

(12I,12Q)

(12I,12Q)

(12I,12Q)

(12I,12Q)

32

23

24

25

M12

|.|2

|.|2

|.|2

|.|2

(12I,12Q)

(12I,12Q)

12

30 (from DSP) or 23 (if ST)

Sat23

32

u22 u23

u24

u23

u25

M11,M22

>> g (*)

Supprimé : 4.0.1

Supprimé : 2







8969 8970

Order is defined in section 6.5.1. 8971

8972 8973

8974 8975

Inverse Determinant 8976

>> (17-o) if o < 17 << (o-17) else

o

u16

u32

Sat 16

M11

M22

u32

u32

|.|

|.|

(32I,32Q) >> (17-o) if o < 17

<< (o-17) else

o

s16

u32

Sat 16

M12

u32

|.|

|.|

(32I,32Q)

order o u32

u32

M11

M22

M12

MAX

Supprimé : 4.0.1

Supprimé : 2







8977

Inverse Matrix computation 8978

8979 Filter computation 8980

8981 8982

8983

Normalize coefficients 8984

(16I,16Q)

(16I,16Q)

(13I,13Q)

(13I,13Q)

28

Filter

29

30

s32

u16

|.|2

|.|2

(16I,16Q) order

>> (16-d) if d <= 16 << (d-16) else

d

0x7fffffff/(.) u16

u30

u32

u16

Sat 16

M11

M22

M12

1/det(M)

d

u8

16

16

>>16

16

32

1/det(M)

M i,j M i,j/det(M)

u16

Supprimé : 4.0.1

Supprimé : 2







8985

8986 8987

8988

8989

s32

order c u32

MAX u8

Sat 8

Filter

>> (25-c) if c < 25 << (c-25) else

s32

c

Filter

s8

Supprimé : 4.0.1

Supprimé : 2







Filtering 8990

8991 8992 Frequency offset is given by: 8993 8994

Performed for the whole sub-frame and for all antennae/user, it is basically the concatenation of 8995 both following blocks: 8996

8997 sincos computation 8998

Use the 20MHz table in Annex 8 to compensate for the global rotation of the OFDM symbol from 8999 the beginning of the TTI. The sine and cosine values directly represent the multiply factor 9000

( )

Ι−

FFT

ˆ..2exp Nmj επ since the beginning of the TTI for the current block. The parameter 9001

“quant” is the current phase (from -8 to +8) and the parameter “m” is the number of blocks before 9002 the current one since the beginning of the TTI. 9003

For information, the value of ( )mΙ is also given in this annex. 9004 9005

Frequency correction 9006

Performs the multiplication of sub carrier signal by a complex exponential coded on 13 bits (Q1.12) 9007

sincos Frequency

compensation

(8Q,8I) (8I,8Q)

>>(19-m)

8

Sat 8

(12I,12Q)

(12I,12Q)

(8I,8Q) (8I,8Q)

Data 12

m = d + o - c

Filtered Data

u4

Supprimé : 4.0.1

Supprimé : 2







9008

9009

15. L1/L2 INTERFACE 9010

The L1/L2 interface defines the protocol primitives for communication between L1 and L2 protocols. The 9011

L1/L2 protocol primitives contains several functions in the following, 9012

• The data flow mapping between the L2 logical channels and L1 transport channels defined in 3GPP 9013

TS36.301 9014

• The required functions and entities for communication between L1 Physical Layer processing and L2 9015

MAC processing functions 9016

• The communication protocol between L1 and L2 for request, response, and abnormal handling of any 9017

control and on-demand information processing. 9018

• The required entities for the cell diagnostic monitor and OA&M. 9019

The emphasis of this specification is the required functions and entities between L1 and L2 processing 9020

functions. The primary function and entities of L1/L2 interface are the L1 measurements to provide L2 9021

MAC for DL and UL scheduling. 9022

The L1/L2 interface for DL and UL scheduling contains functions in the following 9023

• L2 specifies the L1 configuration parameters based on the decision from the UL/DL scheduling and 9024

radio resource management (RRM). The initial radio resource allocation and configuration parameters 9025

in L1 are set up through the call processing control processor (Power PC) at the channel card during 9026

the call setup and handover and not specified in the L1/L2 interface. The L1/L2 interface addresses 9027

the configuration parameters and radio resource dynamically allocated for each user based on the 9028

scheduling or L2 RRM decision. 9029

• L1 sends the L1 processing outputs to the L2 as references for the scheduling and RRM algorithms. 9030

The L1 processing outputs include L1 control messages, L1 measurements, and quality matrix 9031

associated with the L1 measurements. 9032

(8I,8Q)

(13I,13Q)

>>12 Sat8 8

20

21

9

M times

Supprimé : 4.0.1

Supprimé : 2







15.1. L1/L2 INTERFACE FOR DL SCHEDULING 9033

1. L1 message to L2 – 9034

L1 Measurements -> L2 Type Report Type Description

DL CQI Reports Integer Array

Periodic/On-demand Decoded CQI feedback per sub-band per user

Ack/Nack/Erasure Integer Periodic Ack/Nack per HARQ processor per user

CQI CRC CRC flag Periodic/On-demand The CRC of the

decoded CQI feedback per user

Doppler Integer On-Demand

User Doppler Estimation output (one

speed estimate per user)

9035

2. L2 control information to L1 - 9036

• DL-SCH / PDSCH control information 9037

L2 Information -> L1 Type Report Type Description

MAC Id Integer On-demand MAC Id of the transport

block, used for CRC computation

Resource Block Assignment

Bitmap(25 or 50 bits, depending

on bandwidth)

On-demand Scheduled Radio

Resource Assignment per user

MIMO scheme Integer On-demand SISO / TxDiv /

SpatialMux2Layers per user

Cyclic Delay Diversity scheme

Boolean On-demand Either zero, short or long delay CDD, per

user

Codebook element for precoding

Integer On-demand In case of CDD,

precoding matrix to be used, per user

Modulation and coding scheme CW0 (Codeword0)

Integer On-demand

Including transport block size and

modulations used, for codeword 1 for each

user

Redundancy Version Index CW0

Integer On-demand

Index to be used in the rate matching stage by L1, for codeword 1 for

each user

Supprimé : 4.0.1

Supprimé : 2







Modulation and coding scheme CW1 (Codeword1)

Integer On-demand

Including transport block size and

modulations used, for codeword 2 (if any) for

each user

Redundancy Version Index CW1

Integer On-demand

Index to be used in the rate matching stage by L1, for codeword 2 (if

any) for each user

RB assignment Integer Mapping

On-demand PRB allocation for the considered UE

9038

• DL-CCH / PDCCH control information: 9039

For each DCI message in the TTI: 9040


MAC Id Integer On-demand MAC Id of the transport block, used for CRC

computation

CCE Start Integer On-demand First CCE allocated to that UE

CCE Stop Integer On-demand Last CCE allocated to that UE

Numbers of PDCCH CCE Integer On demand For TLA2.0, For subframe0/5, 55

CCE and no UL grant For subframe1/6,4/9, 50

CCEs ,PHICH group number =13, Ng=1

DCI Integer On-demand The message

9041

• Higher layer information 9042

9043


Reference power Integer On-demand

Reference power to refer to for all absolute

power values computation

PBCH power Integer On-demand Power offset to apply on PBCH

PDCCH power Integer On-demand Power offset to apply on PDCCH

PCFICH power Integer On-demand Power offset to apply on

PCFICH

Reference signals power Integer On-demand Power offset to apply on reference signals

Synchronization signals power Integer On-demand

Power offset to apply on synchronization signals

Supprimé : 4.0.1

Supprimé : 2







Pa and Pb power offsets Integer On-demand Power offsets to apply

on PDSCH with respect to reference signals

Permutation offset for PDCCH symbols (Ncss)

Boolean On-demand

Number of symbols to use for PDCCH

Integer On-demand Either 1, 2 or 3.

Number of symbols to use for PDCCH

Integer On-demand Either 1, 2 or 3.

9044

15.2. L1/L2 INTERFACE FOR UL SCHEDULING 9045

1. L1 message to L2 - 9046

L1 Measurements -> L2 Type Report Type Description

UL Scheduling Request Integer On-demand UL Scheduling Request

per user

PUSCH Noise Variance Integer Periodic/On-demand

The estimated noise power per PRB,

antenna and TTI, measured from PUSCH.

PUSCH Signal power Integer Periodic/On-demand

The estimated signal power per PRB antenna and TTI, measured from

PUSCH.

PUCCH Noise Variance Integer Periodic/On-demand The estimated noise per

PRB and per antenna and TTI.

PUCCH Signal power Integer Periodic/On-demand The estimated useful

power per antenna, per user and TTI.

UL Control Channel Resource block Indexes

Integer On-demand

Assigned resource block

（（ (1),0PUCCHn or

(1),1PUCCHn ） for the UL

control channel transmission per user,

which is reported to L2 only when ACK/NACK multiplexing with M=2.

PUCCH ACK/NACK detection results

Enum Periodic/On-demand

Indicate the results of PUCCH ACK/NACK detection per user.

SRS signal power measurement

Integer Periodic/On-demand The estimated signal power per PRB,

Supprimé : 4.0.1

Supprimé : 2







antenna, user and SRS symbol.

Complex channels for MIMO scheduling

Complex

On-demand (the scheduler indicates for

which UE this measure is requested)

For the UEs pointed by the scheduler, L1

provides one complex channel amplitude per PRB and per antenna,

computed from the SRS.

Doppler Integer On-demand

User Doppler information (i.e. one speed estimation per

user).

Timing offset Integer Periodic/On-demand Estimated timing offset of the different users

CRC Integer On-demand The CRC check

associated with each HARQ process per user

9047

2. L2 control information to L1 - 9048


UL Traffic Resource Block Assignment Integer On-demand

Scheduled Radio Resource Assignment

per user UL Traffic Scheduling Tx

Time Integer On-demand Scheduled

Transmission time per user

UL Traffic Modulation Type Integer On-demand Scheduled tx

modulation per user

UL traffic Coding Type Integer On-demand Scheduled coding scheme per user

UL traffic HARQ Type Integer On-demand The HARQ type per

user

UL Traffic Tx Power Integer On-demand Scheduled Tx Power associated with the

selected MCS scheme

UL Traffic Outer loop QoS control

Integer On-demand The mapping of

sounding SNR estimate to CQI index

Sounding Reference Starting Time

Integer On-demand Scheduling starting Tx time of UL sounding

reference

Sounding Reference Integer On-demand Scheduling interval

Supprimé : 4.0.1

Supprimé : 2







Period between the tx sounding reference

Sounding Reference CAZAC Code

Integer On-demand CAZAC sequence per

user

Sounding Reference PRB Integer On-demand Resource block

allocation for each user

Sounding Reference Tx Power

Integer On-demand Tx power level of the

scheduling CQI pilot per user

UL Control Channel Resource block

Integer On-demand Assigned resource

block for the UL control channel transmission.

UL Control Channel Hopping pattern

Integer On-demand Hopping pattern of the

UL control channel

UL Control Channel CAZAC sequence

Integer On-demand Control channel

masking sequence for each user

UL Control Channel Tx Time

Integer On-demand Starting time of the UL

control channel Tx

UL Control Channel Cycle Integer On-demand Scheduled CQI feedback cycle

UL Control Channel Tx power

Integer On-demand Tx power of the UL

control channel per user

UL Control Channel CQI type

Integer On-demand DL CQI feedback type

in the UL control channel per user

Shorten ACK/NACK Boolean On-demand Ture when

simultaneousANandSRS

always is ture in TLA2.0.

TDD ACK/NACK Feedback Mode

Enum On-demand

Indicates one of the two TDD ACK/NACK feedback modes:

Bundling and Multiplexing, see TS

36.213 [23, 7.3].

UL Control Channel Resource block Indexes

Integer On-demand

Assigned resource block （ (1)

,0PUCCHn and (1)

,1PUCCHn ） for the UL control channel

transmission when ACK/NACK multiplexing

with M=2 per user

UL Persistent Scheduling Integer On-demand Starting time of the persistent radio

Supprimé : 4.0.1

Supprimé : 2







Starting time resource allocation

UL Persistent Cycle Integer On-demand Scheduled Tx cycle of

persistent radio resource assignment

UL Persistent Scheduling ending time

Integer On-demand Ending time of the

persistent radio resource allocation

MIMO Traffic Resource Block Assignment Integer On-demand

Scheduled Radio Resource Assignment

per MIMO user MIMO Traffic Scheduling

Tx Time Integer On-demand Scheduled

Transmission time per user for MIMO

MIMO Traffic Modulation Type Integer On-demand

Scheduled tx modulation per user for

MIMO

MIMO traffic Coding Type Integer On-demand Scheduled coding

scheme per user for MIMO

MIMO Decoding Type Integer On-demand The Decoding type per

user for MIMO

9049 9050

16. TRANSMITTER ALGORITHMS 9051

16.1. OVERVIEW 9052

16.1.1 CHAPTER ORGANIZATION 9053

In this section, we give the specification of the DL physical layer parameters. Then we will first 9054

describe all the elementary engines, which are common to all channel processing algorithms (for 9055

avoiding multiple descriptions). Afterwards come a description channel per channel of all involved 9056

processing, using as much as possible elementary engines. 9057

16.1.2 PHYSICAL LAYER PARAMETERS 9058

The main physical layer parameters are summarized in the following table. 9059 9060

Parameter Value Comment Transmission bandwidth

1.4MHz/3MHz/5MHz/10MHz/ 15 MHz/20 MHz

Sampling frequency

1.92MHz/3.84MHz/7.68MHz/15.36MHz/23.04MHz/30.72 MHz

IFFT/FFT size

128/256/512/1024//1536/2048 samples

Number of active

72/180/300/600/900/1200

Centered around DC

Supprimé : 4.0.1

Supprimé : 2


Supprimé : table for LA3.0 case







subcarriers subcarrier DC subcarrier not counted and not allocated

Frame length 10ms Generic frame structure for FDD

Subframe length

1ms

Slot length 0.5ms Cyclic prefix length ( sµ )

5.21 x 1 4.69 x 6

Normal cyclic prefix

Number of OFDM symbols per slot

7

Normal cyclic prefix

Number of consecutive subcarriers per resource unit

12 Normal cyclic prefix

Number of resource blocks per subframe

6/15/25/50/75/100

Number of antenna ports

2 2 antenna ports, except for debug mode

Table 16-1 physical layer parameters for LA5.0 9061 9062

9063 9064

16.1.3 PHYSICAL CHANNELS AND SIGNALS 9065

The supported physical channels and signals together with supported modulation schemes are 9066 summarized in the table below. 9067 9068

Physical Channels Modulation Scheme Comment Physical Downlink Shared Channel PDSCH

QPSK, 16QAM, 64QAM Carries data from higher layers

Physical Broadcast Channel PBCCH

QPSK Broadcast channel, carrying higher layer control information

Physical Downlink Control Channel PDCCH

QPSK L1/L2 control channel

Physical Control Format Indicator Channel PCFICH

QPSK Indicates PDCCH span in OFDM symbols

Physical Hybrid ARQ Indicator Channel PHICH

BPSK UL packets acknowledgement

Physical Signals Modulation Scheme Comment Reference Signal QPSK Required for demodulation and

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 3

Supprimé : .1

Supprimé : The main physical layer parameters are summarized in the following table for LA3.0 case.

Supprimé : in LA3.0







measurements Positioning Reference Signal QPSK Required for UE positioning Synchronization Signal Required to derive frame and

symbol timing 9069

16.2. ELEMENTARY ENGINES 9070

16.2.1 CRC ENCODING 9071

Three CRC encoding engines (CRC24A, CRC24B and CRC16) are defined for transport 9072 channels processing in DL, characterized by their respective generator polynomials. This 9073 section is fully specified in [R3] section 5.1.1. 9074

There is no configuration parameter for this engine. The output, as described in [R3] will be 9075 the concatenation of input sequence and CRC. Here is the interface description for this 9076 block: 9077

9078

Figure 16-1 - CRC encoding interface 9079 9080

In some situations (DCI transport channel encoding for instance), the CRC bits are 9081 scrambled (masked) with the UE identity. This is fully specified in [R3] section 5.3.3.2. This 9082 scrambled CRC block has two inputs, the input stream and the 16 bits UE identity, as 9083 shown in the figure below: 9084

9085

Figure 16-2 - UE masked CRC encoding interface 9086

16.2.2 CODE BLOCK SEGMENTATION 9087

The segmentation is fully described in [R3 section 5.1.2. It consists into splitting the input 9088

sequence into blocks, if the input sequence size is higher than Z=6144 bits. There are at 9089

this engine output C+ sequences of length K+ and C- sequences of length K-, appending F 9090

filler bits to the first sequence and attaching CRC24B to each of the resulting sequences 9091

(note that the size of the sequence is K+ or K- after CRC attachment). For details, please 9092

refer to [R3]. 9093

There is no configuration parameter for this engine. The interface description for this block 9094

is provided in the figure below:. 9095

Supprimé : 4.0.1

Supprimé : 2







9096

Figure 16-3 - Segmentation interface 9097

16.2.3 CHANNEL ENCODING 9098

Two engines are defined for the channel coding: either a turbo encoder or a convolutional 9099

encoder. These are fully defined in [R3] section 5.1.3. The output 3 data streams 9100

correspond to encoded bits with a coding rate of 1/3. There is no configuration parameter 9101

to setup for these engines and the interface is described in the following figure: 9102

9103

Figure 16-4 - Channel encoding interface 9104

16.2.4 RATE MATCHING 9105

Two different engines are defined for rate matching, depending on the channel encoder 9106

adopted: either turbo encoder or convolutional encoder. These are fully defined in section 9107

5.1.4 of [R3]. Again, there is no configuration parameter for this block and the interface is 9108

then simply the following: 9109

9110

Figure 16-5 - Rate matching interface 9111 9112

Note that TC refers here to turbo codes rate matching and CC to convolutional codes rate 9113 matching. 9114

Note that the TC rate matching is dependant on the total number of soft channel bits that 9115 the UE can handle (according to its UE category). 9116

16.2.5 CODE BLOCK CONCATENATION 9117

This is fully specified in section 5.1.5 of [R3]. The output is simply concatenating all input 9118

streams into one output stream. There is no configuration parameter for this block. Here is 9119

the interface definition for this engine. 9120

Supprimé : 4.0.1

Supprimé : 2







9121

Figure 16-6 - Code block concatenation interface 9122

16.2.6 SCRAMBLING 9123

The general principles of the scrambling operation are defined in the sections 6.3.1 and 7.2 9124

of [R2]. It uses Gold sequences resulting from the concatenation of two m-sequences 9125

defined as: 9126

( )( )( ) 2mod)()1()2()3()31(

2mod)()3()31(

2mod)()()(

22222

111

21

nxnxnxnxnx

nxnxnx

NnxNnxnc CC

++++++=+++=+

+++= 9127

where the offset 1600=CN 9128

The upper register is always initialized with the following pattern: 9129

0)1(...)32( 11 === XX and 1)0(1 =X . 9130

The lower register X2 initial value is a configuration parameter for this block. The resulting 9131 interface description is the following: 9132

9133

Figure 16-7 - Scrambling interfaec 9134 Note that the scrambling sequence is initialized every subframe except for PBCH where it 9135 is initialized for every frame where SFN modulo 4 = 0. 9136

16.2.7 MODULATION 9137

Modulation is defined in section 7 of [R2]. Four different modulators are defined: BPSK, 9138

QPSK, 16 QAM and 64 QAM. No configuration parameter is to be specified for this block. 9139

The interfaces of the different modulators are giver below: 9140

Supprimé : 4.0.1

Supprimé : 2







9141

Figure 16-8 - Modulation interface 9142 9143

Note here that the input stream, of size bitM bits is transformed into an output stream of 9144

symbM complex symbols. Each of the complex symbols is coded on modN bits for the real 9145

part and modN for the imaginary part, using 2’s complement coding. 9146

Thanks to implementation choice, the value of modN has been directly set up to 18 bits, 9147

without any need for further optimisation of bit width. 9148

9149

modN = 18, Q(1,17) signed format used

9150

To take advantage of the full input dynamic range, the signal amplitude used in this block 9151 are up scaled versions of the amplitudes specified in [R2] sections 7.1.1 to 7.1.4. The 9152 modulation symbols are scaled to meet the input range of the iFFT described in section 9153 16.2.11 (maximum module of real part and of imaginary part at input is 1). This scaling is 9154 equivalent to applying a power offset to each modulation symbol at the modulation stage. 9155 These equivalent power offsets are reminded in the following table: 9156

9157

Modulation Equivalent power offset (in dB)

Corresponding linear offset

BPSK +3 2

QPSK +3 2

16 QAM +0.4

3

101010

4.0

≈

64 QAM -0.7

7

4210 10

4.0

≈−

9158

These power offsets are to be compensated for in the power block described in section 9159 16.2.10. 9160

The resulting quantized signal amplitudes are given in the following tables, for each 9161 modulation. 9162

Supprimé : 4.0.1

Supprimé : 2







9163

I/Q scaled fix point I/Q (x 217)

2/1 131071

2/1− -131071

Table 16-2- QPSK fix point stored scaled versions 9164 9165


10/1 43402

10/1− -43402

10/3 130206

10/3− -130206

Table 16-3 - 16 QAM fix point stored scaled versions 9166 9167


42/1 18659

42/1− -18659

42/3 55977

42/3− -55977

42/5 93294

42/5− -93294

42/7 130612

42/7− -130612

Table 16-4 - 64 QAM fix point stored scaled versions 9168 9169

16.2.8 LAYER MAPPING 9170

The following layer mapping schemes will be used (corresponding to two transmitter 9171

antennas): 9172

• 1 code word to 1 layer mapping (denoted 11⇒ hereafter), used in spatial 9173

multiplexing scheme (closed-loop case only). 9174

• 1 code word to 2 layers mapping (denoted 21⇒ hereafter) , used in transmit 9175

diversity scheme 9176

• 2 code words to 2 layers mapping (denoted 22⇒ hereafter), used in spatial 9177

multiplexing scheme (SU-MIMO) 9178

Here is a functional description of these layer mapping schemes: 9179

Supprimé : 4.0.1

Supprimé : 2







9180

Figure 16-9 - Layer mapping interface for 2 transmit antennas 9181 9182

In the figure, we have the following: 9183

• (0)d (resp. (1)d ) represents the codeword number 0 (resp. 1). It is a modulated 9184

symbol according to section16.2.7, so it is a complex number using modN bits for 9185

real part and modN bits for imaginary part. 9186

• (0)x (resp. (1)x ) represents the symbol mapped on layer 0 (rep. 1). 9187

• Index i in previous equations represents time index, either at the input or at the 9188 output of the block. 9189

Note that 11⇒ and 22⇒ blocks are basically empty blocks. 9190

16.2.9 PRECODING 9191

The following precoding schemes will be used (corresponding to two transmitter antennas): 9192

• Precoding for transmit diversity. 9193

• Precoding for open-loop spatial multiplexing using large delay CDD . 9194

• Precoding for closed-loop spatial multiplexing on 1 layer. 9195

• Precoding for closed-loop spatial multiplexing on 2 layers. 9196

9197

Supprimé : 4.0.1

Supprimé : 2







TRANSMIT DIVERSITY 9198

The precoding operation for transmitter diversity is shown hereafter: 9199

( )( )( )( )

−

−=

++

)(Im

)(Im

)(Re

)(Re

001

010

010

001

2

1

)12(

)12(

)2(

)2(

)1(

)0(

)1(

)0(

)1(

)0(

)1(

)0(

ix

ix

ix

ix

j

j

j

j

iy

iy

iy

iy

9200

The interface definition for this block is the following one. 9201

9202 Figure 16-X – precoding transmit diversity 9203

9204

OPEN-LOOP SPATIAL MULTIPLEXING WITH LARGE DELAY CDD 9205

The precoding operation for open-loop spatial multiplexing uses large delay CDD (induced 9206

by the matrix D(i) and U) and is shown hereafter: 9207

=

)(

)(

)()(

)(

)(

)1(

)0(

)1(

)0(

ix

ix

UiDiW

iy

iy

MM 9208

With 9209

9210

(18I,18Q)

I

Q

(18I,18Q)

I

(18I,18Q)

I

Q

(0)

Y

(1)

Y

Q (2i)

(18I,18Q)

I

Q

(18I,18Q)

I

(18I,18Q)

I

Q

(1) (0)

Y

(1)

Y

Q (2i+1)

(2i)

(2i+1)

(i) X

(0)

(i) X

Supprimé : 4.0.1

Supprimé : 2







W(i) =

10

01

2

1 D(i) =

− 220

01ije π U =

− 221

11

2

1πje

9211

The precoding procedure can be simplified to: 9212

−=

)(

)(

11

112/1

)(

)(

)1(

)0(

)1(

)0(

ix

ix

iy

iy

MM for even values of i 9213

9214

−=

)(

)(

11

112/1

)(

)(

)1(

)0(

)1(

)0(

ix

ix

iy

iy

MM for odd values of i 9215

9216

The interface definition for these blocks is the following one. 9217

9218 9219

Figure 16-X – precoding SM OL large CDD (even values of i) 9220 9221 9222

9223 9224

Figure 16-X – precoding SM OL large CDD (odd values of i) 9225 9226

(18I,18Q) >>1

>>1

I

Q

(18I,18Q) >>1

>>1

I

Q

(18I,18Q)

I

(18I,18Q)

I

Q

Q

(0)

X (0)

Y

(1)

Y X (1)

(18I,18Q) >>1

>>1

I

Q

(18I,18Q) >>1

>>1

I

Q

(18I,18Q)

I

(18I,18Q)

I

Q

Q

(0)

X (0)

Y

(1)

Y X (1)

Supprimé : 4.0.1

Supprimé : 2







9227

CLOSED-LOOP SPATIAL MULTIPLEXING ON ONE LAYER 9228

The precoding operation for closed-loop spatial multiplexing uses no delay CDD. It is 9229 shown hereafter: 9230

[ ])()(

)(

)()0(

)1(

)0(

ixiW

iy

iy

=

M 9231

The four possible precoding matrix W(i) are enumerated below. For each case, the 9232

interface definition for the equivalent block is described afterwards. 9233

9234

W(i) =

1

1

2

1 9235

9236 Figure 16-X – precoding SM CL 1 layer PMI=0 9237

9238 9239 9240

9241

9242

W(i) =

−1

1

2

1 9243


9246 9247 9248 9249 9250

(18I,18Q)

I

Q

(18I,18Q)

I

(18I,18Q)

I

Q

(0)

X (0)

Y

(1)

Y

Q

(0)

X

(0)

Y

(1)

Y

(18I,18Q)

Supprimé : 4.0.1

Supprimé : 2







W(i) =

j

1

2

1 9251


9254

9255 9256 9257

W(i) =

− j

1

2

1 9258

9259

Figure 16-X – precoding SM CL 1 layer PMI=3 9260 9261

CLOSED-LOOP SPATIAL MULTIPLEXING ON TWO LAYERS 9262

The precoding operation for closed-loop spatial multiplexing uses no delay CDD. It is 9263

shown hereafter: 9264

=

)(

)(

)(

)(

)(

)1(

)0(

)1(

)0(

ix

ix

iW

iy

iy

MM 9265

With the precoding matrix W(i) chosen among: 9266

(18I,18Q)

I

Q

(18I,18Q)

I

(18I,18Q)

I

Q

(0)

X (0)

Y

(1)

Y

Q

(18I,18Q)

I

Q

(18I,18Q)

I

(18I,18Q)

I

Q

(0)

X (0)

Y

(1)

Y

Q

Supprimé : 4.0.1

Supprimé : 2







9267

W(i) =

−11

11

2

1. The interface definition for this block is given in the next figure. 9268

9269 Figure 16-X – precoding SM CL 2 layers PMI 1 9270

9271

9272

or W(i) =

− jj

11

2

1. The interface definition for this block is given in the next figure. 9273

9274

9275 Figure 16-X – precoding SM CL 2 layers PMI 2 9276

Note that: 9277

• the scaling by 2

1 of the precoding stage for transmit diversity and spatial 9278

multiplexing closed loop one layer will be applied at the power offset application 9279 stage by including a -3dB factor to the power offset to apply. 9280

• All input symbols are complex numbers as described in section 16.2.8 9281

• The output symbols of this block are a complex numbers. Each of the real and 9282 imaginary part of these numbers is coded in modN bits, using 2’s complement 9283

encoding. 9284

precoding Power offset to apply

(18I,18Q) >>1

>>1

I

Q

(18I,18Q) >>1

>>1

I

Q

(18I,18Q)

I

(18I,18Q)

I

Q

Q

(0)

X (0)

Y

(1)

Y X (1)

(18I,18Q) >>1

>>1

I

Q

(18I,18Q) >>1

>>1

I

Q

(18I,18Q)

I

(18I,18Q)

I

Q

Q

(0)

X (0)

Y

(1)

Y (1)

X

Supprimé : 4.0.1

Supprimé : 2







Transmit diversity -3 dB

Spatial multiplexing open loop 0 dB

Spatial multiplexing closed loop one layer -3 dB

Spatial multiplexing closed loop two layer 0 dB

Table 16-5 - Compensation power offsets to apply depending on the precoding 9285 9286

16.2.10 POWER SCALING 9287

This block applies the configured power for each transport channel. Here is the interface 9288

for this block: 9289

9290

Figure 16-10 - Power scaling interface 9291 9292

The PDSCH between brackets tells us that the power block for the PDSCH channel is 9293 slightly different as we will see hereafter. Note here that there are some power scaling 9294 factors to take account for. Those depend on the used modulation (see 16.2.7). The 9295 algorithmic description of this block is presented in Figure 16-11 for the non PDSCH case 9296 and in Figure 16-12 in case of PDSCH case. 9297

Note that throughout all this power section, we consider that a signal having 0 dBm is a 9298 complex signal which average power is equal to 1. This is not mandatory, and any 9299 translation of this “0 dBm signal” can be operated at the implementation level, with all the 9300 necessary scaling to meet any implementation constraint. 9301

POWER COMPENSATION STEP 9302

Note here that the power offset compensation step performs here the following operations: 9303

• Depending on the precoding scheme a 0 dB or a -3 dB supplementary power offset 9304

(defined in section 16.2.9) is added to idxP . 9305

• Depending on the modulations, the power offset specified in section 16.2.7 is 9306 applied on top of the input power offset. 9307

• Specifically for phich processing a compensation offset is applied based on the 9308 number of UE multiplexed in a phich group 9309

Supprimé : 4.0.1

Supprimé : 2









• Note that the resulting input power can not be higher than 0 dB power offset, 9310 otherwise we would saturate iFFT input. 9311

Ex1: Input power offset is 0 dB, and 64 QAM used in spatial multiplexing open loop mode. 9312 Then we would have a total power offset to apply of 0 - 0 + 0.7 = 0.7 dB, which is not 9313 possible. This configuration should be forbidden, and if such configuration comes to FPGA 9314 A, clipping to 0 dB can be performed and some warning for bad input released if possible. 9315

Ex2: Input power offset is 1dB, transmit diversity mode configured, and 64 QAM used. 9316 Then total power offset to apply is 1 – 3 + 0.7 = -1.3 dB. This is a valid power offset. 9317

9318

Modulation Power offset compensation to apply (in dB)

QPSK -3

16 QAM -0.4

64 QAM +0.7

BPSK (PHICH) -3

9319 Table 16-6 - Compensation power offsets to apply depending on the modulation 9320

. 9321

POWER LUT 9322

The unsigned linear counterpart of the decibel specified power offset computed by power 9323 compensation step is stored into a LUT. This LUT is encoded in 0.1 dB precision over the 9324 whole input power dynamics. L1 simulations show that with the given implementation 9325 constraints (maximum possible output bit width of 15 bits due to FPGA combiner input bit 9326 width), the maximum applicable power offset at the input of the power stage is -40 dB. 9327 However, thanks to implementation choice, this power offset can go up to 102.3 dB. Please 9328 note that any power below the defined algorithmic range has no sense and should not be 9329

configured. The power offset to apply is encoded on powN bits. The linear to log table of all 9330

power offsets to apply is specified in appendix. 9331

Supprimé : 4.0.1

Supprimé : 2







9332

Figure 16-11 - Power scaling algorithm for non PDSCH channels 9333 9334

PDSCH SPECIFIC PROCESSING 9335

The power block for PDSCH channel is quite different, since we have to apply a different 9336 power on symbols )(iy that are mapped on OFDM symbols containing reference symbols 9337

and on symbols )(iy that are mapped on OFDM symbols which do not contain reference 9338

symbols. 9339

The modified block diagram is: 9340

Supprimé : 4.0.1

Supprimé : 2







9341

Figure 16-12 - Power scaling algorithm for PDSCH channel 9342 9343

The two differences with previous block are the presence of a block determining whether 9344 the symbol )(iy is in an OFDM symbol containing RS or not and the precoding mode used 9345

for PDSCH transmission, and chooses accordingly the correct power index to apply. 9346

SPECIFIC POWER MANAGEMENT FOR PHICH 9347

A power offset depending on the number of actual UE multiplexed in the same pHICH 9348 group is applied on top of all the other power offsets compensation. 9349

Please note that there is one such block processing per phich group and per TTI. 9350

9351

Supprimé : 4.0.1

Supprimé : 2







9352

The UE dependant power offset compensates the Q.17 scaling applied in section 15.4.5 in 9353 order to have a constant UE BPSK symbol power per RE which ever the phich group load. 9354

SUMMARY OF POWER MANAGEMENT AND OUTCOMES 9355

To better emphasize how the power scaling and modulation stage works, we give here a 9356 diagram giving a different insight into all this (described in section 16.2.10). 9357

9358

Figure 16-13 - Power scaling example 9359 9360

The idea of this implementation is to store expanded versions of the modulation symbols, 9361 to gain as much precision as possible. Then, since we have to apply a precise power 9362 command on top of the transmitted modulation, we have to scale back everything. For 9363 instance, to have a 0 dB power command correctly applied on top of the modulated 9364 symbol, we have to scale back the expanded symbol to its original shape as part of the 9365 power scaling management, by applying a negative -3 dB power offset. 9366

Please note that here, the maximum power offsets to be applied on each of the input 9367 modulated symbols are different and depend on the modulation used. This has some 9368 impact on the interfaces, which have to ensure the maximum power offset is not exceeded. 9369

Please note also that throughout this document, the power offsets are decreasing from 9370 0dB. This interface is different from the one assumed by the scheduler for instance, who 9371 has to apply a power offset on top of the reference signals level (which can be positive or 9372 negative). Somehow and somewhere, the conversion has to be done. 9373

PHICH BPSK symbol power (received from DSP)

Number of antenna ports

PHICH power offset

MIN(0,x)

-30 (BPSK power compensation)

Power clipping warning to DSP

X>0

Supprimé : 4.0.1

Supprimé : 2







Please note also that the interface with “external word” (TRM, RRH) has also to be taken 9374 care of. Because we have here a “0 dBm” signal level that is fixed and internal to the 9375 xCEM. This “0 dBm” level has to be converted somehow to actual absolute power value. 9376

BIT WIDTH 9377

Thanks to implementation assumptions, the bit widths to use within this block are the 9378 following: 9379

idxN = 10 (1024 power offsets, in 0.1 dB precision, from 0dB to -102.3dB)

16=powN

Stored values in power LUT are in Q(0,16) unsigned format

Output of power scaling is in Q(1,33) signed format on 34mod =+ powNN bits

16.2.11 IFFT PROCESSING 9380

This is the OFDM signal generation from the time frequency RE description. The OFDM 9381 symbols are generated in a similar way to the section 6.12 of [R2]. The IFFT block to use is 9382 the one by Xilinx, and is similar to the block used in UL. 9383

The algorithmic view of this block is described in the figure below and consists into an input 9384

scaling block, converting the output of the modulator into the iFFT input width. Then iFFT 9385 processing followed by an output scaling block, converting the unscaled output of the iFFT 9386

to the desired output width (dictated by the FPGA combiner input bit width). 9387

9388

9389

Figure 16-14 - iFFT block 9390

IFFT 9391

The parameters to use for the Xilinx IP are the following (extracted from [R15], describing 9392

the bit accurate C model): 9393

9394

Generic parameter Description Value

C_NFFT_MAX Maximum transform length size 11

C_ARCH Architecture 3 (implementation choice)

C_HAS_NFFT Use of run time configurable transform length

1 (run time configuration)

C_INPUT_WIDTH Input data width 15

Supprimé : 4.0.1

Supprimé : 2







C_TWIDDLE_WIDTH Phase factor bit width 16*

C_HAS_SCALING Are internal computations scaled to reduce output width

0 (no scaling)

C_HAS_BFP Scaling option 0 (unused)

C_HAS_ROUNDING Truncation or convergent rounding

0 (truncation used)

*: please note that 15 is here enough, but according to transform synthesis, 16 bits is less 9395 FPGA slices consuming, so recommended value is then obviously 16… 9396

INPUT SCALING 9397

The input bit width has to be converted into the iFFT input bit width. Scaling is performed 9398 accordingly to convert the output of power stage into the input format of the iFFT. 9399

Power stage output is of size 34mod =+ powNN (multiplication of the signed modulation 9400

on modN bits by the unsigned linear power offset on powN bits). Format of power output is 9401

Q1.33. 9402

Recommended IFFT input format for almost no performance loss should be Q1.14, thus 9403

conversion consists in 19 bits right shift (please note that this is a symmetrical shift). 9404

OUTPUT SCALING 9405

Unscaled iFFT output format is Q9.14, Q10.14, Q11.14, Q12.14 and Q13.14 9406 ( 27,26,25,24,23=fftOutN ) for bandwidth 1.4 MHz, 3 MHz, 5 MHz, 10 MHz and 20 MHz 9407 respectively. For 15 MHz, we also make the assumption that the iDFT is unscaled and that 9408

the output format is Q13.14. At the fpga output we want to yield I/Q samples in 15 bits 9409 format. For all bandwidth but 1.4 MHz, conversion is done by right shift with 5 bits and 9410

saturation to 15 bits. 9411

Specifically for 1.4 MHz bandwidth, conversion is done by right shift with 4 bits and 9412 saturation to 15 bits. 9413

See the annex for further details. 9414

16.3. TRANSPORT CHANNELS PROCESSING 9415

We give here the processing chain described for each of the transport channels supported. 9416 The elementary blocks are to be taken from the ones defined in the sections 16.2.1 to 9417

16.2.5. 9418

16.3.1 DL-SCH AND PCH PROCESSING 9419

The complete DL-SCH processing is summarized in the figure below. This processing is 9420

repeated for each DL-SCH packet. This means that if SU-MIMO is configured, and two 9421 code words are to be transmitted onto two layers, this processing is to be applied on each 9422 code word independently. 9423

Supprimé : 4.0.1

Supprimé : 2

Supprimé : . C

Supprimé : This shall be applied for all system bandwidth 5, 10, 15, 20 MHz. Compared to previous version of this document, the right shift 6 has been replaced by a right shift 5 bits. At the input of iFFT, a global power offset reduction of 6 dB is applied in order to set the power to the right scale.¶







9424

Figure 16-15 - DL-SCH and PCH processing 9425

16.3.2 DCI PROCESSING 9426

Four DCI frame formats are defined, depending on the information embedded in the DCI 9427

channel. Here is the summary of the DCI frame size (in bits), as a function of the DCI 9428 format, as described in section 5.3.3.1 of [R3]: 9429

9430

9431 Table 16-7 - Number of bits used for DCI encoding for DCI format 0 9432

9433 9434

Supprimé : 4.0.1

Supprimé : 2

Supprimé :








9437 9438

9439 Table 16-9 - Number of bits used for DCI encoding for DCI format 1A 9440

9441 9442


9445

9446

Supprimé : 4.0.1

Supprimé : 2

Supprimé :

Supprimé :

Supprimé :

Supprimé :







Table 16-11 - Number of bits used for DCI encoding for DCI format 2A 9447 9448 9449

The complete DCI processing is summarized in next figure. This processing is to be 9450

repeated for each DCI packet independently as there might be multiple DCI packets per 9451 sub-frame. The number of bits at the output of the rate matching stage is dictated by the 9452

PDCCH format that will be used to transport the DCI channel. The table below describes 9453

the mapping between PDCCH format and rate matching output number of bits: 9454

Aggregation level Number of bits

1 72

2 144

4 288

8 576

Table 16-12 - Number of bits used for DCI transmission for each PDCCH format 9455 9456

The resulting block diagram of the DCI channel processing is the following: 9457

9458

Figure 16-16 - DCI processing 9459 9460

16.3.3 BCH PROCESSING 9461

This processing is summarized hereafter: 9462

9463

Figure 16-17 - BCH processing 9464 Please note that the CRC is masked with 0xFFFF to signal the dual antenna mode 9465 configurations used for transmission. 9466

16.3.4 CFI PROCESSING 9467

This processing is quite simple, and only consists into channel encoding 2 CFI information 9468 bits with a C(2,32) channel code. This is fully described in the section 5.3.4 of [R3]. 9469

Supprimé : 4.0.1

Supprimé : 2







Additionally for 1.4 MHz, the CFI value to encode is CFI-1, allowing a range of PDCCH 9470 OFDM symbols ={2,3,4}. 9471

16.3.5 HI PROCESSING 9472

This is as simple as the CFI processing: just one channel encoding with a thrice repetition 9473 of the HARQ bit, as described in section 5.3.5 of [R3]. 9474

16.4. PHYSICAL CHANNELS PROCESSING 9475

We give here the description of all physical channels processing. We start the description 9476

by giving the list of transport channels that are mapped on the considered physical 9477 channel, then we describe the specific processing. Please note that some blocks, such as 9478

the RE mapping block are also described as part of these sections (for blocks that are 9479

different from one channel to the other, this description was not included in the elementary 9480 engines description). 9481

9482

16.4.1 PDSCH PROCESSING 9483

The following transport channels are mapped onto the PDSCH: 9484

• DL-SCH 9485

• PCH 9486

The processing chain of the PDSCH channel depends on the multiple antenna scheme 9487

chosen, either TxDiv or SU-MIMO. Please note that: 9488

• PCH can only be mapped on a TxDiv scheme, and never on SU-MIMO 9489

• For SU-MIMO, the zero delay CDD precoding is used. 9490

The different processing are described in the figures below depending on the MIMO 9491 scheme. Please note that modulations used here can be QPSK, 16 QAM or 64 QAM. 9492

Supprimé : 4.0.1

Supprimé : 2







9493

9494

Figure 16-18 - SU-MIMO PDSCH processing 9495

9496

Figure 16-19 - TxDiv PDSCH processing 9497

Supprimé : 4.0.1

Supprimé : 2







9498

Figure 16-20 – SM CL 1 layer PDSCH processing 9499 9500 9501

The RE mapping step consists in mapping the symbols )1(),...,0( )()( −symbkk Myy into the 9502

RE time frequency grid for antenna port k. 9503

The mapping into RE(k,l) is done by: 9504

• Increasing frequency index k then by 9505

• Increasing symbol index l 9506

Over all RE not dedicated to reference and synchronization signals transmissions or 9507 common control channels (PDCCH, PBCH, PCFICH and PHICH) 9508

16.4.2 PDCCH PROCESSING 9509

One DCI channel can be mapped onto one PDCCH channel. So as there can be multiple 9510

DCI channels transported in one sub-frame, there might be multiple PDCCH channels in 9511

one sub-frame. 9512

The PDCCH channels processing is summarized in the diagram shown below. 9513

For further details refer to section 6.8 of [R2]. 9514

Note that PDCCH use the RE groups not assigned to PHICH and PCFICH. As PHICH 9515 group number varies between different downlink subframes, RE groups that are available 9516 to PDCCH varies between different downlink subframes, too. 9517

Supprimé : 4.0.1

Supprimé : 2







9518

Figure 16-21 - PDCCH processing 9519

Supprimé : 4.0.1

Supprimé : 2







16.4.3 PBCH PROCESSING 9520

Only BCH channel is mapped on the PBCH. The whole processing chain is given in the 9521

Figure 16-22. Modulation used here is QPSK. 9522

The RE mapping step in the two antenna case consists, as usual in mapping the symbols 9523

)1(),...,0( )()( −symbkk Myy into the RE time frequency grid for antenna port k. 9524

The mapping into RE(k,l) is done by: 9525

• Increasing frequency index k then by 9526

• Increasing symbol index l then by 9527

• Increasing frame index 9528

Over: 9529

• All RE not dedicated to reference signals transmission 9530

• Index 352

362

RBsc

DLRB

RBsc

DLRB +≤≤−

NNk

NN 9531

• Indexes l=0 to 3 in slot 1 of sub frame 0. 9532

• Each sub frame 0 of a radio frame, for 4 consecutive radio frames. 9533

• Assuming 4 antenna RS transmission whatever the actuel RS transmission mode. 9534

The RE mapping is illustrated for the two antenna case in Figure 16-23 9535

Supprimé : 4.0.1

Supprimé : 2









9536

Figure 16-22 - PBCH processing 9537

Supprimé : 4.0.1

Supprimé : 2







9538

Figure 16-23 - PBCH RE mapping 9539

Supprimé : 4.0.1

Supprimé : 2







The mono antenna case will not be described here for sake of concision, but is quite 9540 straightforward to deduce. 9541

16.4.4 PCFICH PROCESSING 9542

CFI transport channel is mapped on PCFICH. The processing of this block is shown in the 9543 figure below. Note that we have two information bits, and the coding rate is of 1/16, 9544

resulting in 32 bits to send over the air. The modulation is a QPSK, resulting in 16 9545 modulated symbols, which are split thanks to the layer mapping, then duplicated with the 9546

precoding stage. 9547

The PCFICH processing is fully described in §6.7 of [R2]. 9548

9549

9550

Figure 16-24 - PCFICH processing 9551 9552

16.4.5 PHICH PROCESSING 9553

The phich is fully described in §6.9 of [R2]. This section only brings some details on the 9554

bitwidth and power management. 9555

Supprimé : 4.0.1

Supprimé : 2







Bit width 9556

Thanks to implementation choice the sequences can be coded on 18 bits with Q1.17 9557 format. 9558

Note that each of the real part and imaginary part of )()( ky p belongs to the set 9559

{ }8,7,6,5,4,3,2,1,0 ±±±±±±±± . So for the symbol modulation of PHICH we can limit to the 9560

representation of these numbers in fixed point. We however scale these values to meet the 9561 iFFT input width. According to the actual number of phich resources multiplexed in a same 9562 phich group, the mapping to Q.1.17 format is as follows: 9563

9564

I Q amplitude \ nb_ue 1 2 3 4 5 6 7 8

0 0 0 0 0 0 0 0 0 1 131071 65536 43691 32768 26214 21845 18725 16384 -1 -131071 -65536 -43691 -32768 -26214 -21845 -18725 -16384 2 131071 87381 65536 52429 43691 37449 32768 -2 -131071 -87381 -65536 -52429 -43691 -37449 -32768 3 131071 98304 78643 65536 56174 49152 -3 -131071 -98304 -78643 -65536 -56174 -49152 4 131071 104858 87381 74898 65536 -4 -131071 -104858 -87381 -74898 -65536 5 131071 109227 93623 81920 -5 -131071 -109227 -93623 -81920 6 131071 112347 98304 -6 -131071 -112347 -98304 7 131071 114688 -7 -131071 -114688 8 131071 -8 -131071

Please note that a variable compensation factor, depending on the number of multiplexed 9565 UE, must be used in order to have a constant power per user whichever the phich load. 9566

9567

Supprimé : 4.0.1

Supprimé : 2







9568

Figure 16-25 - PHICH processing 9569

Supprimé : 4.0.1

Supprimé : 2







9570

16.5. SIGNALS PROCESSING 9571

Associated to the regular physical channels, some signals are defines, which help the UE 9572 to estimate the propagation channel helping the demodulation, estimation of channel 9573

quality indication (CQI) and to recover the synchronization. We describe in the sequel the 9574

processing associated to reference signals and to synchronization signals. 9575

16.5.1 REFERENCE SIGNALS PROCESSING 9576

Reference signals used are cell specific reference signals fully described in section 9577 6.10.1.1 of [R2]. 9578

Bit width 9579

In LA3.0, reference signals are QPSK modulated symbols. As such, reference signals 9580 sequences can be coded on 18 bits with Q1.17 format. 9581

16.5.2 POSITIONING REFERENCE SIGNALS PROCESSING 9582

Positioning Reference signals used are described in section 6.10.4 of [R2]. 9583 9584

Bit width 9585

Positioning reference signals are QPSK modulated symbols. As such, reference signals 9586 sequences can be coded on 18 bits with Q1.17 format. 9587

9588

PORT 6 MAPPING TO PHYSICAL PORTS 9589

Two modes shall be made available: 9590

- map on physical port 0 only, 9591

- map on all physical ports (identical signal with same power) 9592

Due to the absence of transmit diversity, the second option may be necessary to prevent destructive fading on specific 9593 parts of the spectrum, typically for UE in LOS. 9594

16.5.3 SYNCHRONIZATION SIGNALS PROCESSING 9595

Two different synchronization signals are defined: primary signals used to recover the half-9596

frame synchronization, and the secondary, used to recover the frame synchronization. 9597

PRIMARY SYCHRONIZATION SIGNALS 9598

They are generated by length 62 Zadoff-Chu sequences. There are three such ones, 9599 indexed by the physical layer cell identity group, as described in section 6.11.1 of [R2]. 9600

The primary synchronization symbols )(nd are mapped into the OFDM symbol number 9601

6=l of the slots 0 and 10 in LA3.0 by: 9602

• Increasing frequency index, in the range of 2

31RBsc

DLRB NN

nk +−= (n is symbol 9603

index ranging from 0 to 61) 9604

Supprimé : 4.0.1

Supprimé : 2







• then by increasing slot number (0 then 10) 9605

Note that RE(k,6) for 66,...,62,1,...,5,2

31 −−=+−= nNN

nkRBsc

DLRB are not used for 9606

transmission of synchronization signals nor any channels. This is illustrated in Figure 9607 16-28. 9608

We have to determine the fix point representation of these signals, for the implementation. 9609

Bit width 9610

Thanks to implementation choice the sequences can be coded on 18 bits with Q1.17 9611 format. The fix point representation of the used synchronization sequences is given in 9612 appendix. 9613

9614

SECONDARY SYNCHRONIZATION SIGNALS 9615

There are two different sequences generated to be used in slot 0 and 10 of a radio frame in 9616

LA3.0 based on shifted and interleaved PN sequences, as described in section 10.11.2 of 9617 [R2]. These are indexed by the physical layer identity. 9618

We describe hereafter, in a slightly different way than in [R2] the way the secondary 9619 synchronization is generated. 9620

Three different m-sequences ( )(~ ns , )(~ nc and )(~ nz ) of length 31 each are defined. They 9621

are generated by the following LFSR: 9622

9623

Figure 16-26 - Secondary synchronization signals LFSR 9624 9625

Supprimé : 4.0.1

Supprimé : 2








Two shift parameters are defined for these sequences, called 0m and 1m , depending on 9626

the physical layer cell identity group. Their exact definition is given in section 6.11.2.1 of 9627

[R2]. Let’s recall that the physical layer call identity group is denoted )1(IDN and the physical 9628

layer identity within this group is denoted )2(IDN . With these notations, we give below a 9629

diagram explaining how to generate the synchronization sequences for slot 0. The 9630 processing of slot 10 synchronization sequences is analogous and can be found in section 9631 6.11.2.1 of [R2]. 9632

The mapping of secondary signals into REs is the same as for the primary synchronization, 9633 except that the l=5 OFDM symbols is used in slots 0 and 10 instead of l=6. The mapping is 9634 depicted in Figure 16-28 (note that we assumed here only one antenna port configured for 9635 RS symbol positions). 9636

Bit width 9637

This sequence is binary. Same bit width as for primary sequence is to be used (18 bits, 9638 Q(1,17) format). 9639

9640

)(id I (x 217)

1 131071

-1 -131071

MAPPING TO PHYSICAL PORTS 9641

Two modes shall be made available: 9642

- map on physical port 0 only, 9643

- map on all physical ports (identical signal with same power) 9644

Due to the absence of transmit diversity, the second option may be necessary to prevent destructive fading on specific 9645 parts of the spectrum, typically for UE in LOS. 9646

9647

Supprimé : 4.0.1

Supprimé : 2








9648

Figure 16-27 - Slot 0/10 secondary synchronization signals generation 9649 9650

Supprimé : 4.0.1

Supprimé : 2







9651

Figure 16-28 - RE mapping of synchronization signals in LA3.0 9652 9653

Supprimé : 4.0.1

Supprimé : 2







17. OPEN ISSUES & FURTHER STUDIES 9654

17.1. OPEN ISSUES 9655

• The value of the PUSCH DTX threshold will be provided later in a further release of the 9656

document. 9657

17.2. FURTHER STUDIES (POST DROP 1) 9658

• LLR & post-IDFT SNR weighting 9659

• Improved channel estimation 9660

• Higher speed 9661

• Intra TTI hopping 9662

• 64 QAM 9663

• User separation : implementation in freq. domain with filters can be added. 9664

18. ANNEX 9665

Annex 1. Calculation of coff 9666

Without loss of generality, we shall assume that the transmitted RACH preamble is defined by the zero shift of the ZC 9667 sequence xu(k). Then the samples of the received RACH preamble r(k), k=0,1,…,Nzc-1, after the Doppler frequency 9668 shift of fDopp =1/TPRE=1.25KHz can be represented as 9669

r(k)= kTfj

usymDoppekx

π2)( = k

u Wkx )( , where W= czNje /2π− 9670

9671

where Tsym is the duration of each RACH symbol in RACH preamble, Tsym = TPRE / Nzc = 0.95us. Then, 9672

r(k) = [ ] 2/)/1(2)1( kukkuW ++ 9673

=2/)1(2/)2( 22 +−++++ offoffoffoffoff cuccckckku

WW 9674

=[ ] 2/)1(2/)1()1(( +−+++++ offoffoffoffoff cucckcckku

WW 9675

=2/)1(2/)1)(( +−+++ offoffoffoff cucckcku

WW 9676

= uuoffu Wckx 2/)1()( +−+ , 9677

here, coff =1/u, or equivalently coff =(N·m-1)/u, for the smallest m such that coff is an integer. 9678

The last expression shows that the received RACH preamble after the Doppler frequency shift fDopp =1/TPRE is equal 9679 to the transmitted RACH preamble cyclically shifted by coff where the transmitted RACH preamble is obtained from 9680 u-th root ZC sequence. The complex scaling constant in the received RACH preamble has unit magnitude and thus 9681 does not influence the correlation detector in the receiver. 9682

Supprimé : 4.0.1

Supprimé : 2







9683

Annex 2. Proof for equation (2) 9684

For notation simplicity, let W= Nczje /2π− . Then, the DFT of xu(n), which we denote Xu(k) is 9685

Xu(k) = ∑−

=

1

0

)(Ncz

n

nku Wnx 9686

First, look at a single term inside the summation: 9687

nku Wnx )( =

2/2)1(

++ nu

knnu

W = [ ] 2/2)1( dnnnuW ++ , where Nczu

kd mod= . 9688

which can be further derived as: 9689

nku Wnx )( = [ ]

2/)1(

2/)1(2/2)1(

+

+++

dud

duddnnnu

W

WW = 2/)1( +− dudW 2/)1)(( +++ dndnuW 9690

= )(2

)(

dkxW uu

ukk

++

−. (B) 9691

Thus, substitute (B) back to equation (A), we have: 9692

Xu(k) = ∑−

=

+−

+1

0

2

)(

)(Ncz

nu

u

ukk

dnxW , for u

nd = mod Ncz, 9693

Note that ∑−

=

+=1

0

)()0(Ncz

nuu dnxX , by letting k=0. Then, we have: 9694

9695

Xu(k) = )0(2

)(

uu

ukk

XW+

−= )0(

)(

uuN

ukkj

Xe CZ

+π

. 9696

9697

Annex 3. CAZAC sequences 9698

We give below the values coded as Q12 for the different possible CAZAC sizes of the quantities: 9699 9700

− p

N RSZC

π2cos followed by

− p

N RSZC

π2sin , p=0,.., 12/ +RS

ZCN 9701

9702 =============================================================== 9703

31=RSZCN 9704

9705 9706 2047 2006 1882 1681 1411 1083 711 310 -104 -513 -902 -1254 -1554 -1791 9707 -1954 -2037 9708 9709 0 -412 -808 -1170 -1484 -1738 -1921 -2024 - 2045 -1983 -1839 -1620 -1334 9710 -994 -613 -207 9711 9712 =============================================================== 9713 9714

47=RSZCN 9715

9716 9717 2047 2030 1975 1885 1762 1607 1424 1215 984 736 475 205 -68 -341 -607 9718 -862 -1102 -1322 -1519 -1688 -1828 -1935 -20 07 -2043 9719 9720 9721 0 -273 -541 -800 -1044 -1269 -1472 -1649 -1 796 -1911 -1992 -2038 -2047 9722

Supprimé : 4.0.1

Supprimé : 2







-2019 -1956 -1858 -1726 -1564 -1374 -1159 -9 24 -672 -408 -137 9723 9724 =============================================================== 9725 9726 9727

59=RSZCN 9728

9729 9730 2047 2036 2002 1944 1865 1764 1644 1505 134 9 1177 992 796 591 379 163 9731 -55 -272 -486 -695 -896 -1086 -1265 -1429 - 1577 -1707 -1817 -1907 -1976 9732 -2022 -2045 9733 9734 0 -218 -433 -643 -846 -1040 -1221 -1389 -15 41 -1676 -1792 -1887 -1961 9735 -2013 -2041 -2047 -2030 -1989 -1927 -1842 -1 736 -1611 -1467 -1307 -1132 9736 -944 -746 -539 -326 -109 9737 9738 =============================================================== 9739 9740

71=RSZCN 9741

9742 9743 2047 2040 2016 1976 1921 1851 1766 1667 155 6 1432 1297 1152 998 836 667 494 316 9744 136 -45 -226 -405 -581 -752 -918 -1076 -122 6 -1366 -1495 -1613 9745 -1718 -1810 -1888 -1951 -1998 -2030 -2046 9746 9747 0 -181 -361 -537 -710 -877 -1037 -1189 -133 2 -1464 -1585 -1693 -1789 9748 -1870 -1936 -1988 -2023 -2043 -2047 -2035 -2 008 -1964 -1905 -1831 -1743 9749 -1641 -1526 -1399 -1262 -1114 -958 -794 -624 -449 -271 -91 9750 9751 =============================================================== 9752 9753

89=RSZCN 9754

9755 9756 2047 2043 2028 2002 1967 1922 1867 1803 173 0 1648 1558 1461 1356 1244 1127 1003 9757 875 742 605 466 324 180 36 -108 -252 -395 -536 -674 -809 -939 -1065 -1186 -1301 -9758 1409 -1511 -1604 -1690 -1768 -1836 -1896 -19 46 -1986 9759 -2016 -2037 -2047 9760 9761 0 -144 -288 -431 -571 -708 -842 -971 -1096 -1215 -1329 -1435 -1535 -1627 -1710 -9762 1786 -1852 -1909 -1957 -1994 -2022 -2040 -20 48 -2045 -2032 -2010 9763 -1977 -1934 -1882 -1820 -1749 -1670 -1582 -1 486 -1383 -1273 -1157 -1034 9764 -907 -775 -640 -501 -360 -216 -72 9765 9766 =============================================================== 9767 9768

107=RSZCN 9769

9770 9771 2047 2044 2034 2016 1992 1960 1922 1877 182 6 1769 1705 1635 1560 1480 1394 1304 9772 1209 1110 1007 900 791 678 564 447 329 210 90 -30 -150 -270 -388 -506 -621 -735 9773 -846 -954 -1059 -1160 -1257 -1349 -1437 -152 1 -1599 9774 -1671 -1738 -1798 -1853 -1901 -1942 -1977 -2 005 -2026 -2040 -2047 9775 9776 0 -120 -240 -359 -477 -593 -707 -818 -927 -1033 -1135 -1233 -1327 -1416 -1500 -1580 9777 -1653 -1721 -1784 -1840 -1889 -1932 -1969 -1 999 -2021 -2037 9778 -2046 -2048 -2042 -2030 -2011 -1985 -1951 -1 912 -1865 -1812 -1753 -1688 9779 -1617 -1541 -1459 -1372 -1280 -1184 -1084 -9 80 -873 -763 -650 -535 -418 9780 -300 -180 -60 9781 9782 =============================================================== 9783

113=RSZCN 9784

9785 9786 2047 2045 2035 2020 1998 1969 1935 1895 184 9 1797 1739 1677 1609 1536 1458 1376 9787 1290 1199 1105 1008 907 803 697 589 479 36 8 256 142 28 -85 9788

Supprimé : 4.0.1

Supprimé : 2







-199 -312 -424 -535 -644 -751 -855 -958 -10 57 -1153 -1245 -1333 -1418 9789 -1498 -1573 -1643 -1709 -1769 -1823 -1872 -1 916 -1953 -1984 -2009 -2028 9790 -2041 -2047 9791 9792 0 -114 -227 -340 -452 -562 -671 -777 -881 -983 -1081 -1176 -1267 -1355 9793 -1438 -1517 -1591 -1660 -1724 -1783 -1836 -1 884 -1926 -1961 -1991 -2015 9794 -2032 -2043 -2048 -2046 -2038 -2024 -2004 -1 977 -1944 -1905 -1861 -1810 9795 -1754 -1693 -1626 -1555 -1478 -1397 -1312 -1 222 -1129 -1032 -932 -830 -724 -617 -507 9796 -396 -284 -171 -57 9797 =============================================================== 9798

139=RSZCN 9799

9800 9801 2047 2046 2040 2029 2015 1996 1973 1946 191 6 1881 1842 1800 1754 1704 1651 1595 9802 1535 1472 1407 1338 1267 1193 1116 1037 956 874 789 703 615 526 436 345 254 162 9803 69 -23 -116 -208 -300 -391 -481 -571 -659 -746 -832 -915 -997 -1077 -1155 -1230 -9804 1303 -1373 -1440 -1504 -1566 -1624 -1678 9805 -1730 -1777 -1822 -1862 -1899 -1931 -1960 -1 985 -2006 -2022 -2035 -2043 9806 -2047 9807 9808 0 -93 -185 -277 -368 -459 -549 -637 -725 - 810 -895 -977 -1057 -1135 9809 -1211 -1285 -1355 -1423 -1488 -1551 -1609 - 1665 -1717 -1766 -1811 -1852 9810 -1890 -1924 -1953 -1979 -2001 -2019 -2032 -2 042 -2047 -2048 -2045 -2037 9811 -2026 -2010 -1991 -1967 -1939 -1907 -1872 -1 832 -1789 -1742 -1692 -1638 9812 -1580 -1520 -1456 -1390 -1320 -1248 -1174 -1 097 -1017 -936 -853 -768 -681 -593 -504 9813 -414 -323 -231 -139 -46 9814 =============================================================== 9815

179=RSZCN 9816

9817 9818 2047 2047 2043 2037 2028 2017 2003 1986 196 8 1947 1923 1897 1869 1838 1806 1771 9819 1733 1694 1653 1609 1564 1516 1467 1416 136 3 1309 1253 1195 1136 1075 1014 951 886 9820 821 755 687 619 550 481 411 340 269 197 1 26 54 -18 -90 -162 -233 -304 -375 -446 9821 -516 -585 -653 -721 -788 -854 -919 -982 -10 45 -1106 -1166 -1224 -1281 -1336 -1390 -9822 1442 -1492 -1540 -1587 -1631 9823 -1674 -1714 -1752 -1788 -1822 -1854 -1883 - 1910 -1935 -1958 -1977 -1995 9824 -2010 -2023 -2033 -2040 -2045 -2048 9825 9826 0 -72 -144 -215 -287 -358 -428 -498 -568 - 636 -704 -771 -837 -902 -966 -1029 -1091 9827 -1151 -1210 -1267 -1323 -1377 -1429 -1480 -1 528 -1575 -1620 9828 -1663 -1704 -1743 -1780 -1814 -1846 -1876 -1 904 -1929 -1952 -1973 -1991 9829 -2006 -2020 -2030 -2038 -2044 -2047 -2048 -2 046 -2042 -2035 -2025 -2013 9830 -1999 -1982 -1963 -1941 -1917 -1890 -1862 - 1830 -1797 -1762 -1724 -1684 9831 -1642 -1598 -1552 -1504 -1454 -1403 -1350 - 1295 -1238 -1180 -1121 -1060 9832 -998 -935 -870 -804 -738 -670 -602 -533 -4 63 -393 -322 -251 -179 -108 9833 -36 9834 =============================================================== 9835

191=RSZCN 9836

9837 9838 2047 2047 2044 2038 2030 2020 2008 1994 197 7 1959 1938 1915 1890 1864 1835 1804 9839 1771 1736 1699 1661 1621 1578 1535 1489 144 2 1394 1343 1292 1239 1185 1129 1072 9840 1014 955 895 834 772 709 646 581 516 451 385 319 252 185 118 51 -17 -84 -151 -9841 219 -285 -352 -418 -484 -549 -614 -678 9842 -741 -803 -865 -925 -985 -1043 -1101 -1157 -1212 -1266 -1318 -1369 -1418 -1466 -1512 9843 -1557 -1600 -1641 -1680 -1718 -1754 -1787 -1 819 -1849 -1877 9844 -1903 -1927 -1949 -1968 -1986 -2001 -2015 -2 026 -2034 -2041 -2046 -2048 9845 9846 9847 0 -67 -135 -202 -269 -335 -402 -467 -533 - 598 -662 -725 -788 -849 -910 -970 -1029 9848 -1087 -1143 -1198 -1252 -1305 -1356 -1406 -1 454 -1501 -1546 9849 -1589 -1631 -1671 -1709 -1745 -1779 -1812 -1 842 -1870 -1897 -1921 -1944 9850 -1964 -1982 -1998 -2011 -2023 -2032 -2040 - 2045 -2047 -2048 -2046 -2042 9851

-2036 -2028 -2018 -2005 -1990 -1973 -1954 -1 933 -1909 -1884 -1857 -1827 -1796 -1762 9852 -1727 -1690 -1651 -1610 -1568 -1523 -1478 -1 430 -1381 -1331 9853

-1279 -1225 -1171 -1115 -1058 -1000 -940 -8 80 -819 -756 -693 -630 -565 -500 -435 9854 -369 -302 -235 -168 -101 -34 9855 9856 =============================================================== 9857

211=RSZCN 9858

Supprimé : 4.0.1

Supprimé : 2







9859 9860 2047 2047 2044 2040 2033 2025 2015 2004 199 0 1975 1958 1939 1919 1896 1873 1847 9861 1820 1791 1761 1729 1695 1660 1624 1586 154 7 1506 1464 1421 1376 1331 1284 1236 9862 1186 1136 1085 1033 980 926 871 815 759 70 2 644 586 528 468 409 349 289 228 168 9863 107 46 -15 -76 -137 -198 -258 -319 -379 9864 -439 -498 -557 -615 -673 -731 -787 -843 -89 8 -953 -1006 -1059 -1111 -1161 -1211 -9865 1260 -1307 -1354 -1399 -1443 -1485 -1527 -15 67 -1605 -1642 -1678 9866 -1712 -1745 -1776 -1806 -1834 -1860 -1885 -1 908 -1929 -1949 -1967 -1983 9867 -1997 -2010 -2021 -2030 -2037 -2042 -2046 - 2048 9868 9869 9870 0 -61 -122 -183 -243 -304 -364 -424 -483 - 542 -601 -659 -716 -773 -829 -885 -939 9871 -993 -1046 -1098 -1149 -1199 -1248 -1296 -13 42 -1388 -1432 -1475 -1517 -1557 -1596 -9872 1633 -1669 -1704 -1737 -1769 -1798 -1827 -18 54 -1879 9873 -1902 -1924 -1944 -1962 -1979 -1994 -2007 - 2018 -2028 -2035 -2041 -2045 9874 -2047 -2048 -2047 -2043 -2038 -2032 -2023 - 2013 -2000 -1987 -1971 -1953 9875 -1934 -1913 -1891 -1866 -1840 -1813 -1784 - 1753 -1721 -1687 -1651 -1615 9876 -1576 -1537 -1496 -1454 -1410 -1365 -1319 - 1272 -1223 -1174 -1124 -1072 9877 -1020 -966 -912 -857 -801 -745 -688 -630 -5 72 -513 -454 -394 -334 -274 9878 -213 -152 -91 -30 9879 9880 =============================================================== 9881

239=RSZCN 9882

9883 9884 2047 2047 2045 2042 2037 2030 2023 2013 200 3 1991 1978 1963 1947 1930 1911 1891 9885 1869 1847 1823 1798 1771 1744 1715 1685 165 4 1621 1588 1553 1518 1481 1443 1405 9886 1365 1324 1283 1241 1197 1153 1108 1063 101 6 969 921 873 824 774 724 674 623 571 9887 519 467 414 362 308 255 202 148 94 40 -13 -67 -121 -175 -228 -282 -335 -388 -441 9888 -493 -545 -597 -648 -699 -749 9889 -799 -849 -897 -945 -993 -1040 -1086 -1131 -1175 -1219 -1262 -1304 -1345 -1385 -1424 9890 -1462 -1500 -1536 -1571 -1605 -1638 -1669 -1 700 -1729 -1758 9891 -1785 -1811 -1835 -1858 -1880 -1901 -1920 -1 938 -1955 -1970 -1984 -1997 9892 -2008 -2018 -2027 -2034 -2039 -2044 -2046 -2 048 9893 9894 9895 0 -54 -108 -161 -215 -268 -322 -375 -428 - 480 -532 -584 -635 -686 -737 -787 -836 9896 -885 -933 -981 -1028 -1074 -1120 -1164 -1208 -1251 -1293 -1335 -1375 -1414 -1453 -9897 1490 -1527 -1562 -1596 -1630 -1662 -1692 -17 22 -1751 9898 -1778 -1804 -1829 -1853 -1875 -1896 -1916 -1 934 -1951 -1967 -1981 -1994 9899 -2006 -2016 -2025 -2032 -2038 -2043 -2046 -2 048 -2048 -2047 -2044 -2041 9900 -2035 -2029 -2020 -2011 -2000 -1988 -1974 -1 959 -1943 -1925 -1906 -1886 9901 -1864 -1841 -1817 -1791 -1765 -1737 -1708 -1 677 -1646 -1613 -1579 -1545 9902 -1509 -1472 -1434 -1395 -1355 -1314 -1272 -1 230 -1186 -1142 -1097 -1051 9903 -1005 -957 -909 -861 -812 -762 -712 -661 -6 10 -558 -506 -454 -401 -348 9904 -295 -242 -188 -135 -81 -27 9905 9906 =============================================================== 9907

283=RSZCN 9908

9909 9910 2047 2047 2046 2043 2040 2035 2030 2023 201 6 2007 1998 1987 1976 1963 1950 1935 9911 1920 1904 1887 1868 1849 1829 1809 1787 176 4 1741 1716 1691 1665 1638 1610 1582 9912 1553 1522 1492 1460 1428 1395 1361 1327 129 2 1256 1220 1183 1146 1108 1070 1031 9913 991 951 910 870 828 786 744 702 659 616 5 72 528 484 440 395 351 306 261 216 170 9914 125 80 34 -11 -57 -102 -148 9915 -193 -238 -283 -328 -373 -418 -462 -506 -55 0 -594 -637 -680 -723 -765 9916 -807 -849 -890 -931 -971 -1011 -1050 -1089 -1127 -1165 -1202 -1238 -1274 -1310 -1344 9917 -1378 -1412 -1444 -1476 -1507 -1538 -1567 -1 596 -1624 -1652 9918 -1678 -1704 -1728 -1752 -1776 -1798 -1819 - 1840 -1859 -1878 -1895 -1912 9919 -1928 -1943 -1957 -1970 -1982 -1993 -2003 - 2012 -2020 -2027 -2033 -2038 9920 -2042 -2045 -2047 -2048 9921 9922 9923 0 -45 -91 -136 -182 -227 -272 -317 -362 -4 07 -451 -495 -539 -583 -626 9924 -670 -712 -755 -797 -839 -880 -921 -961 -10 01 -1040 -1079 -1118 -1155 9925 -1193 -1229 -1265 -1301 -1336 -1370 -1403 -1 436 -1468 -1499 -1530 -1560 9926 -1589 -1617 -1645 -1671 -1697 -1722 -1747 - 1770 -1792 -1814 -1834 -1854 9927 -1873 -1891 -1908 -1924 -1939 -1953 -1966 - 1979 -1990 -2000 -2009 -2018 9928 -2025 -2031 -2037 -2041 -2044 -2046 -2048 - 2048 -2047 -2045 -2043 -2039 9929 -2034 -2028 -2022 -2014 -2005 -1995 -1984 - 1973 -1960 -1946 -1932 -1916 9930 -1900 -1882 -1864 -1844 -1824 -1803 -1781 - 1758 -1735 -1710 -1684 -1658 9931

Supprimé : 4.0.1

Supprimé : 2







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293=RSZCN 9938

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431=RSZCN 10070

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479=RSZCN 10108

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523=RSZCN 10152

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10196 =============================================================== 10197

571=RSZCN 10198

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1069=RSZCN 10733

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=============================================================== 10810

1151=RSZCN 10811

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Supprimé : 4.0.1

Supprimé : 2







-1189 -1180 -1171 -1162 -1153 -1143 -1134 -1 125 -1115 -1106 -1097 -1087 -1078 -1068 10884 -1058 -1049 -1039 -1030 -1020 -1010 -1001 -9 91 -981 -971 -961 -951 -941 -932 -922 -10885 912 -902 -891 -881 -871 -861 -851 -841 -831 -820 -810 -800 -790 -779 -769 -758 -10886 748 -738 -727 -717 -706 -696 -685 -675 -664 -654 -643 -632 -622 -611 -600 -590 -10887 579 -568 -557 -547 -536 -525 -514 -503 -493 -482 -471 -460 -449 -438 -427 -416 -10888 405 -394 -383 -372 -361 -350 -339 -328 -317 -306 -295 -284 -273 -262 -251 -240 -10889 229 -218 -206 -195 -184 -173 -162 -151 -140 -128 -117 -106 -95 -84 -73 -61 -50 -10890 39 -28 -17 -6 10891 =============================================================== 10892

10893

Annex 4. Table of complex exponential with the resi dual frequency offset 10894

The estimated frequency offset ε is a s8 with range in [-0.0667 … +0.0667[ . It is represented as [-128 … 10895 +127]. The equivallent frequency range is [-1000 Hz ; +1000 Hz[ and in radian [.;[ Π+Π− 10896

10897

The next table provides the values of ( )

∆

FFT

S

N

nNj

επˆ.

2exp on 16 bits. The cos() value for each quantized 10898

version of ( )nε∆ in the range [0 ; 64] (i.e. angle in the range [ ]2/;0 Π ) is given below. The sin() and 10899

values for other angles are obtained by symetrical properties. 10900 10901

( )

∆

FFT

S

N

nNj

επˆ.

2cos 10902

10903 32767, 32758, 32729, 32679, 32610, 32522, 32413, 32286, 32138, 31972, 31786, 31581, 10904 31357, 31114, 30853, 30572, 30274, 29957, 29622, 29269, 28899, 28511, 28106, 27684, 10905 27246, 26791, 26320, 25833, 25330, 24812, 24279, 23732, 23170, 22595, 22006, 21403, 10906 20788, 20160, 19520, 18868, 18205, 17531, 16846, 16151, 15447, 14733, 14010, 13279, 10907 12540, 11793, 11039, 10279, 9512, 8740, 7962, 7180, 6393, 5602, 4808, 4011, 3212, 10908 2411, 1608, 804, 0 10909 10910 10911

Annex 5. Pre stored filter for T-MMSE PUSCH 10912

10913 # define N_SNR 8 // {-3,0,3,6,9,12,15,18, 21, 24, 27, 30} 10914 10915 // 3kmph 10916 10917

Filter3[N_SNR][1] = 10918

{ 10919

{256},{341},{409},{455},{482},{496},{504},{508},{5 10},{511},{511},{512} 10920

}; 10921

// 3kmph 10922

Filter25[N_SNR][2] = 10923

{ 10924

{257, 254}, {344, 338}, {414, 404},{465, 444}, 10925

{502, 461}, {536, 457}, {577, 431},{636, 380} 10926

{714, 306}, {803, 219}, {883, 140},{942, 81} 10927

Supprimé : 4.0.1

Supprimé : 2







10928

}; 10929

// 25kmph 10930

Filter50[N_SNR][2] = 10931

{ 10932

{260, 249}, {350, 329}, {428, 387},{493, 414}, 10933

{553, 407}, {623, 368}, {707, 299},{798, 216} 10934

{880, 138}, {940, 81}, {977, 45},{999, 24} 10935

}; 10936

// 50kmph 10937

Filter75[N_SNR][4] = 10938

{ 10939

{267, 261, 250, 230}, {368, 352, 332, 297}, 10940

{468, 430, 392, 332}, {571, 492, 423, 320}, 10941

{689, 545, 425, 256}, {827, 595, 406, 147}, 10942

{971, 640, 377, 18}, {1095, 677, 348, -97}, 10943

{1183, 704, 326, -181}, {1237, 720, 312, -233}, 10944

{1267, 728, 305, -262}, {1283, 733, 301, -278}, 10945

}; 10946

// 75kmph 10947

Filter100[N_SNR][4] = 10948

{ 10949

{274, 264, 246, 211}, {386, 359, 325, 265}, 10950

{505, 444, 381, 281}, {634, 515, 405, 242}, 10951

{779, 576, 401, 151}, {929, 629, 378, 31}, 10952

{1060, 672, 352, -85}, {1155, 701, 331, -172}, 10953

{1217, 721, 316, -228}, {1251, 731, 308, -260}, 10954

{1269, 737, 304, -277}, {1278, 740, 302, -286}, 10955

}; 10956

// 100kmph 10957

Filter125[N_SNR][4] = 10958

{ 10959

{283, 267, 241, 188}, {406, 367, 317, 228}, 10960

{543, 459, 369, 225}, {693, 538, 388, 165}, 10961

Supprimé : 4.0.1

Supprimé : 2







{850, 604, 380, 62}, {995, 656, 359, -53}, 10962

{1106, 694, 338, -149}, {1179, 718, 323, -214} 10963

{1224, 733, 314, -254}, {1248, 741, 309, -275}, 10964

{1260, 745, 306, -286}, {1266, 747, 305, -292} 10965

}; 10966

// 125kmph 10967

Filter150[N_SNR][4] = 10968

{ 10969

{292, 272, 235, 163}, {427, 378, 309, 188}, 10970

{580, 477, 358, 167}, {745, 561, 373, 93}, 10971

{904, 630, 365, -12}, {1037, 680, 348, -115}, 10972

{1130, 714, 332, -192}, {1187, 734, 321, -241} 10973

{1219, 744, 315, -269}, {1236, 750, 312, -284}, 10974

{1244, 753, 310, -292}, {1249, 755, 309, -295} 10975

}; 10976

// 150kmph 10977

Filter175[N_SNR][4] = 10978

{ 10979

{301, 277, 228, 134}, {446, 388, 300, 146}, 10980

{612, 493, 346, 111}, {783, 583, 361, 30}, 10981

{938, 652, 355, -71}, {1057, 700, 342, -160}, 10982

{1134, 729, 330, -222}, {1180, 746, 323, -259} 10983

{1205, 755, 318, -280}, {1218, 760, 316, -290}, 10984

{1224, 763, 315, -296}, {1228, 764, 314, -299} 10985

}; 10986

// 175kmph 10987

Filterr200[N_SNR][4] = 10988

{ 10989

{308, 282, 220, 105}, {462, 398, 290, 104}, 10990

{637, 510, 335, 59}, {810, 603, 351, -25}, 10991

{957, 672, 348, -118}, {1062, 717, 339, -193}, 10992

{1128, 744, 331, -243}, {1165, 759, 326, -272} 10993

{1185, 767, 323, -288}, {1195, 771, 322, -296}, 10994

{1200, 773, 321, -300}, {1203, 774, 321, -302} 10995

Supprimé : 4.0.1

Supprimé : 2







}; 10996

// 200kmph 10997

Filterr225[N_SNR][4] = 10998

{ 10999

{315, 286, 212, 75}, {475, 408, 280, 63}, 11000

{654, 525, 326, 10}, {826, 621, 343, -72}, 11001

{963, 690 344, -155}, {1056, 734, 339, -218}, 11002

{1113, 759 334, -259}, {1144, 772, 331, -282} 11003

{1160, 780, 330, -294}, {1169, 783, 329, -300}, 11004

{1173, 785, 328, -303}, {1175, 786, 328 -305} 11005

}; 11006

// 225kmph 11007

Filterr250[N_SNR][4] = 11008

{ 11009

{319, 291, 203, 45}, {484, 417, 271, 23}, 11010

{665, 540, 317, -34}, {832, 639, 338, -112}, 11011

{959, 707, 343, -185}, {1043, 750, 341, -238}, 11012

{1091, 774, 339, -271}, {1118, 786, 338, -289} 11013

{1132, 793, 337, -299}, {1139, 796, 336, -304}, 11014

{1143, 798, 336, -306}, {1144, 799, 336, -308} 11015

}; 11016

// 250kmph 11017

11018

11019

Annex 6. Pre stored filter for T-MMSE PUCCH 11020

#define N_SNR 8 11021

#define LENGTH 3 11022

11023

K_3kmph[N_SNR][LENGTH] = { 11024

{ 5479 , 5479 , 5478}, 11025

{ 8202 , 8202 , 8201}, 11026

{ 10923 , 10923 , 10922}, 11027

{ 13102 , 13101 , 13100}, 11028

{ 14558 , 14556 , 14553}, 11029

Supprimé : 4.0.1

Supprimé : 2







{ 15418 , 15414 , 15409}, 11030

{ 15892 , 15883 , 15873}, 11031

{ 16148 , 16129 , 16110} 11032

}; 11033

11034


{ 5483 , 5478 , 5462}, 11036

{ 8215 , 8202 , 8173}, 11037

{ 10956 , 10924 , 10871}, 11038

{ 13176 , 13104 , 13007}, 11039

{ 14714 , 14561 , 14379}, 11040

{ 15737 , 15420 , 15073}, 11041

{ 16528 , 15889 , 15220}, 11042

{ 17391 , 16136 , 14849} 11043

}; 11044

11045


{ 5497 , 5475 , 5412}, 11047

{ 8254 , 8202 , 8085}, 11048

{ 11055 , 10929 , 10719}, 11049

{ 13397 , 13115 , 12731}, 11050

{ 15174 , 14575 , 13863}, 11051

{ 16651 , 15437 , 14104}, 11052

{ 18264 , 15909 , 13430}, 11053

{ 20485 , 16156 , 11701} 11054

}; 11055

11056


{ 5519 , 5472 , 5329}, 11058

{ 8319 , 8202 , 7942}, 11059

{ 11215 , 10937 , 10468}, 11060

{ 13751 , 13131 , 12283}, 11061

{ 15895 , 14600 , 13051}, 11062

{ 18015 , 15467 , 12649}, 11063

Supprimé : 4.0.1

Supprimé : 2







{ 20629 , 15941 , 10975}, 11064

{ 24102 , 16190 , 7996} 11065

}; 11066

11067


{ 5549 , 5466 , 5214}, 11069

{ 8407 , 8201 , 7743}, 11070

{ 11432 , 10947 , 10126}, 11071

{ 14221 , 13155 , 11683}, 11072

{ 16816 , 14634 , 12000}, 11073

{ 19645 , 15508 , 10892}, 11074

{ 23152 , 15986 , 8327}, 11075

{ 27350 , 16237 , 4623} 11076

}; 11077

11078


{ 5587 , 5459 , 5069}, 11080

{ 8516 , 8201 , 7493}, 11081

{ 11697 , 10961 , 9699}, 11082

{ 14784 , 13185 , 10953}, 11083

{ 17870 , 14678 , 10781}, 11084

{ 21367 , 15560 , 9007}, 11085

{ 25508 , 16044 , 5810}, 11086

{ 29928 , 16298 , 1886} 11087

}; 11088


{5630, 5450, 4895}, 11090

{ 8642, 8201, 7196}, 11091

{12002, 10978, 9200}, 11092

{15412, 13222, 10125}, 11093

{18985, 14732, 9467}, 11094

{23035, 15626, 7144}, 11095

{27522, 16116, 3603}, 11096

{31826, 16373, -203} 11097

Supprimé : 4.0.1

Supprimé : 2







}; 11098


{ 5680, 5439, 4693}, 11100

{ 8784, 8200, 6854}, 11101

{ 12338, 10997, 8635}, 11102

{ 16082, 13265, 9216}, 11103

{ 20110, 14795, 8105}, 11104

{ 24579, 15702, 5367}, 11105

{ 29187, 16200, 1709}, 11106

{ 33214, 16462, -1819} 11107

}; 11108


{ 5733, 5426, 4466}, 11110

{ 8936, 8198, 6472}, 11111

{ 12695, 11018, 8015}, 11112

{ 16770, 13315, 8255}, 11113

{ 21196, 14867, 6747}, 11114

{ 25942, 15790, 3736}, 11115

{ 30505, 16297, 126}, 11116

{ 34192, 16564, -3060} 11117

}; 11118


{ 5789, 5411, 4215}, 11120

{ 9096, 8195, 6054}, 11121

{ 13062, 11042, 7352}, 11122

{ 17452, 13370, 7265}, 11123

{ 22209, 14949, 5429}, 11124

{ 27107, 15890, 2270}, 11125

{ 31520, 16408, -1187}, 11126

{ 34860, 16680, -4023} 11127

}; 11128


{ 5847, 5394, 3943}, 11130

{ 9260, 8191, 5606}, 11131

Supprimé : 4.0.1

Supprimé : 2







{ 13430, 11067, 6656}, 11132

{ 18110, 13431, 6266}, 11133

{ 23125, 15040, 4173}, 11134

{ 28073, 16002, 969}, 11135

{ 32279, 16531, -2276}, 11136

{ 35292, 16810, -4783} 11137

}; 11138

11139

Annex 7. sincos table (cfo) for SIMO, common for al l bandwidths 11140

The values are given only for negative values of quantized CFO. We use symmetry of cos and sin to compute the 11141 other half of the table (same values for cos, opposite values for sin) 11142 11143 20MHz 11144 cos 11145 m=0 m=1 m=2 m=3 m=4 m=5 m=6 m=7 m=8 m=9 m=10 m=11 m=12 m=13 11146 quant = -8 4095 3685 2546 904 -917 -2557 -3691 -40 96 -3685 -2546 -904 917 2557 3691 11147 quant = -7 4095 3780 2890 1561 -5 -1571 -2897 -378 4 -4096 -3781 -2892 -1563 3 1569 11148 quant = -6 4095 3863 3198 2174 907 -461 -1778 -289 6 -3694 -4071 -3992 -3466 -2552 -1351 11149 quant = -5 4095 3934 3465 2726 1774 684 -460 -1568 -2559 -3344 -3868 -4090 -3993 -3584 11150 quant = -4 4095 3992 3688 3200 2552 1776 911 0 -91 7 -1781 -2557 -3204 -3691 -3993 11151 quant = -3 4095 4037 3865 3584 3201 2728 2179 1567 907 226 -461 -1136 -1778 -2371 11152 quant = -2 4095 4070 3993 3866 3690 3468 3202 2896 2552 2177 1775 1351 910 458 11153 quant = -1 4095 4089 4070 4038 3993 3936 3866 3784 3690 3584 3468 3340 3202 3054 11154 quant = 0 4095 4095 4095 4095 4095 4095 4095 4095 4095 4095 4095 4095 4095 4095 11155 11156 sin 11157 m=0 m=1 m=2 m=3 m=4 m=5 m=6 m=7 m=8 m=9 m=10 m=11 m=12 m=13 11158 quant = -8 0 1788 3208 3995 3992 3200 1775 -0 -178 8 -3208 -3995 -3992 -3200 -1775 11159 quant = -7 0 1577 2902 3787 4095 3783 2895 1568 -1 0 -1575 -2901 -3786 -4096 -3784 11160 quant = -6 0 1361 2559 3471 3994 4070 3690 2896 17 69 451 -917 -2183 -3204 -3867 11161 quant = -5 0 1141 2184 3057 3692 4039 4070 3784 31 98 2365 1348 226 -914 -1982 11162 quant = -4 0 917 1781 2557 3204 3691 3994 4095 399 2 3688 3200 2552 1776 911 11163 quant = -3 0 690 1356 1984 2555 3055 3469 3784 399 4 4090 4070 3935 3690 3340 11164 quant = -2 0 461 914 1355 1779 2180 2554 2896 3204 3469 3691 3867 3994 4070 11165 quant = -1 0 231 460 687 912 1134 1353 1567 1778 1 982 2180 2371 2554 2729 11166 quant = 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 11167 11168 Values of I(m) for SIMO 20MHz: 11169 I(m 0->13)= 11170 0 2208 4400 6592 8784 10976 13168 15360 17568 19760 21952 24144 26336 28528 11171 11172 11173

Annex 8. sincos table (cfo) for MIMO, common for al l bandwidths 11174

The values are given only for negative values of quantized CFO. We use symmetry of cos and sin to compute the 11175 other half of the table (same values for cos, opposite values for sin) 11176 11177 20MHz 11178 cos 11179 m=0 m=1 m=2 m=3 m=4 m=5 m=6 m=7 m=8 m=9 m=10 m=11 m=12 m=13 11180 quant = -8 3976 3158 1715 -67 -1836 -3241 -4007 -3 976 -3158 -1715 67 1836 3241 4007 11181 quant = -7 4004 3371 2225 742 -855 -2321 -3435 -40 29 -4005 -3372 -2227 -743 853 2320 11182 quant = -6 4029 3559 2691 1521 181 -1179 -2408 -33 72 -3950 -4086 -3765 -3021 -1940 -641 11183 quant = -5 4049 3720 3102 2241 1205 75 -1060 -2120 -3007 -3659 -4025 -4078 -3812 -3249 11184 quant = -4 4066 3854 3450 2873 2152 1323 428 -495 -1386 -2208 -2920 -3485 -3876 -4074 11185 quant = -3 4079 3959 3728 3392 2960 2444 1860 1218 546 -141 -824 -1484 -2101 -2660 11186 quant = -2 4088 4035 3931 3778 3577 3331 3044 2716 2356 1966 1552 1118 670 214 11187 quant = -1 4094 4081 4055 4016 3964 3900 3824 3735 3635 3524 3401 3268 3124 2971 11188 quant = 0 4095 4095 4095 4095 4095 4095 4095 4095 4095 4095 4095 4095 4095 4095 11189 11190 sin 11191

Supprimé : 4.0.1

Supprimé : 2







m=0 m=1 m=2 m=3 m=4 m=5 m=6 m=7 m=8 m=9 m=10 m=11 m=12 m=13 11192 quant = -8 982 2609 3720 4095 3662 2504 852 -982 - 2609 -3720 -4095 -3662 -2504 -851 11193 quant = -7 861 2327 3439 4028 4006 3375 2231 737 - 860 -2325 -3438 -4028 -4006 -3376 11194 quant = -6 740 2028 3088 3803 4092 3923 3314 2326 1083 -281 -1614 -2766 -3608 -4046 11195 quant = -5 618 1713 2675 3429 3915 4095 3956 3505 2782 1842 758 -385 -1498 -2494 11196 quant = -4 495 1386 2208 2920 3485 3876 4074 4066 3854 3450 2873 2152 1323 428 11197 quant = -3 371 1049 1697 2296 2832 3287 3650 3911 4059 4094 4012 3818 3516 3115 11198 quant = -2 248 704 1150 1583 1996 2383 2741 3066 3 351 3593 3791 3940 4041 4090 11199 quant = -1 124 353 581 807 1031 1251 1468 1681 188 8 2088 2282 2469 2649 2819 11200 quant = 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 11201 11202 11203 Values of I(m) for MIMO 20MHz: 11204 I(m 0->13)= 11205 1184 3376 5568 7760 9952 12144 14336 16544 18736 20 928 23120 25312 27504 29696 11206 11207

Annex 9. c(k) , 5 TAPS 11208

Band 1.4 MHz 11209 epsilon=-0.066667, quant(epsilon)=-8 11210 c(-2).real=-136 11211 c(-1).real=-283 11212 c(0).real=3979 11213 c(1).real=250 11214 c(2).real=130 11215 c(-2).imag=-36 11216 c(-1).imag=-67 11217 c(0).imag=839 11218 c(1).imag=46 11219 c(2).imag=21 11220 epsilon=-0.058333, quant(epsilon)=-7 11221 c(-2).real=-119 11222 c(-1).real=-247 11223 c(0).real=4006 11224 c(1).real=222 11225 c(2).real=114 11226 c(-2).imag=-28 11227 c(-1).imag=-52 11228 c(0).imag=737 11229 c(1).imag=35 11230 c(2).imag=15 11231 epsilon=-0.050000, quant(epsilon)=-6 11232 c(-2).real=-102 11233 c(-1).real=-211 11234 c(0).real=4030 11235 c(1).real=193 11236 c(2).real=99 11237 c(-2).imag=-21 11238 c(-1).imag=-39 11239 c(0).imag=633 11240 c(1).imag=25 11241 c(2).imag=11 11242 epsilon=-0.041667, quant(epsilon)=-5 11243 c(-2).real=-86 11244 c(-1).real=-175 11245 c(0).real=4050 11246 c(1).real=162 11247 c(2).real=83 11248 c(-2).imag=-15 11249 c(-1).imag=-27 11250 c(0).imag=529 11251 c(1).imag=17 11252

Supprimé : 4.0.1

Supprimé : 2

Mis en forme : Police :Gras,Soulignement

Mis en forme :Police :(asiatique) SimSun,Gras, Soulignement

Mis en forme : Normal, Sansnumérotation ni puces

Mis en forme : Police :(Pardéfaut) Times New Roman

Mis en forme : Police :(Pardéfaut) Times New Roman,Anglais (Royaume-Uni)

Mis en forme : Police :(Pardéfaut) Times New Roman,Allemand (Allemagne)
















c(2).imag=7 11253 epsilon=-0.033333, quant(epsilon)=-4 11254 c(-2).real=-69 11255 c(-1).real=-140 11256 c(0).real=4066 11257 c(1).real=131 11258 c(2).real=67 11259 c(-2).imag=-11 11260 c(-1).imag=-18 11261 c(0).imag=424 11262 c(1).imag=10 11263 c(2).imag=4 11264 epsilon=-0.025000, quant(epsilon)=-3 11265 c(-2).real=-51 11266 c(-1).real=-104 11267 c(0).real=4079 11268 c(1).real=100 11269 c(2).real=51 11270 c(-2).imag=-7 11271 c(-1).imag=-11 11272 c(0).imag=319 11273 c(1).imag=5 11274 c(2).imag=1 11275 epsilon=-0.016667, quant(epsilon)=-2 11276 c(-2).real=-34 11277 c(-1).real=-69 11278 c(0).real=4089 11279 c(1).real=67 11280 c(2).real=34 11281 c(-2).imag=-3 11282 c(-1).imag=-5 11283 c(0).imag=213 11284 c(1).imag=2 11285 c(2).imag=0 11286 epsilon=-0.008333, quant(epsilon)=-1 11287 c(-2).real=-17 11288 c(-1).real=-34 11289 c(0).real=4094 11290 c(1).real=34 11291 c(2).real=17 11292 c(-2).imag=-1 11293 c(-1).imag=-2 11294 c(0).imag=106 11295 c(1).imag=0 11296 c(2).imag=0 11297 epsilon=0.000000, quant(epsilon)=0 11298 c(-2).real=0 11299 c(-1).real=0 11300 c(0).real=4095 11301 c(1).real=0 11302 c(2).real=0 11303 c(-2).imag=0 11304 c(-1).imag=0 11305 c(0).imag=0 11306 c(1).imag=0 11307 c(2).imag=0 11308 epsilon=0.008333, quant(epsilon)=1 11309 c(-2).real=17 11310 c(-1).real=34 11311

Supprimé : 4.0.1

Supprimé : 2

























c(0).real=4094 11312 c(1).real=-34 11313 c(2).real=-17 11314 c(-2).imag=0 11315 c(-1).imag=0 11316 c(0).imag=-106 11317 c(1).imag=2 11318 c(2).imag=1 11319 epsilon=0.016667, quant(epsilon)=2 11320 c(-2).real=34 11321 c(-1).real=67 11322 c(0).real=4089 11323 c(1).real=-69 11324 c(2).real=-34 11325 c(-2).imag=0 11326 c(-1).imag=-2 11327 c(0).imag=-213 11328 c(1).imag=5 11329 c(2).imag=3 11330 epsilon=0.025000, quant(epsilon)=3 11331 c(-2).real=51 11332 c(-1).real=100 11333 c(0).real=4079 11334 c(1).real=-104 11335 c(2).real=-51 11336 c(-2).imag=-1 11337 c(-1).imag=-5 11338 c(0).imag=-319 11339 c(1).imag=11 11340 c(2).imag=7 11341 epsilon=0.033333, quant(epsilon)=4 11342 c(-2).real=67 11343 c(-1).real=131 11344 c(0).real=4066 11345 c(1).real=-140 11346 c(2).real=-69 11347 c(-2).imag=-4 11348 c(-1).imag=-10 11349 c(0).imag=-424 11350 c(1).imag=18 11351 c(2).imag=11 11352 epsilon=0.041667, quant(epsilon)=5 11353 c(-2).real=83 11354 c(-1).real=162 11355 c(0).real=4050 11356 c(1).real=-175 11357 c(2).real=-86 11358 c(-2).imag=-7 11359 c(-1).imag=-17 11360 c(0).imag=-529 11361 c(1).imag=27 11362 c(2).imag=15 11363 epsilon=0.050000, quant(epsilon)=6 11364 c(-2).real=99 11365 c(-1).real=193 11366 c(0).real=4030 11367 c(1).real=-211 11368 c(2).real=-102 11369 c(-2).imag=-11 11370

Supprimé : 4.0.1

Supprimé : 2






















c(-1).imag=-25 11371 c(0).imag=-633 11372 c(1).imag=39 11373 c(2).imag=21 11374 epsilon=0.058333, quant(epsilon)=7 11375 c(-2).real=114 11376 c(-1).real=222 11377 c(0).real=4006 11378 c(1).real=-247 11379 c(2).real=-119 11380 c(-2).imag=-15 11381 c(-1).imag=-35 11382 c(0).imag=-737 11383 c(1).imag=52 11384 c(2).imag=28 11385 epsilon=0.066667, quant(epsilon)=8 11386 c(-2).real=130 11387 c(-1).real=250 11388 c(0).real=3979 11389 c(1).real=-283 11390 c(2).real=-136 11391 c(-2).imag=-21 11392 c(-1).imag=-46 11393 c(0).imag=-839 11394 c(1).imag=67 11395 c(2).imag=36 11396 11397 3MHz 11398 epsilon=-0.066667, quant(epsilon)=-8 11399 c(-2).real=-136 11400 c(-1).real=-283 11401 c(0).real=3978 11402 c(1).real=249 11403 c(2).real=129 11404 c(-2).imag=-32 11405 c(-1).imag=-64 11406 c(0).imag=842 11407 c(1).imag=50 11408 c(2).imag=24 11409 epsilon=-0.058333, quant(epsilon)=-7 11410 c(-2).real=-120 11411 c(-1).real=-248 11412 c(0).real=4005 11413 c(1).real=221 11414 c(2).real=114 11415 c(-2).imag=-25 11416 c(-1).imag=-49 11417 c(0).imag=739 11418 c(1).imag=38 11419 c(2).imag=18 11420 epsilon=-0.050000, quant(epsilon)=-6 11421 c(-2).real=-103 11422 c(-1).real=-212 11423 c(0).real=4029 11424 c(1).real=192 11425 c(2).real=99 11426 c(-2).imag=-19 11427 c(-1).imag=-36 11428 c(0).imag=636 11429

Supprimé : 4.0.1

Supprimé : 2









Mis en forme : Police :Gras,Soulignement
















c(1).imag=28 11430 c(2).imag=13 11431 epsilon=-0.041667, quant(epsilon)=-5 11432 c(-2).real=-86 11433 c(-1).real=-176 11434 c(0).real=4050 11435 c(1).real=162 11436 c(2).real=83 11437 c(-2).imag=-13 11438 c(-1).imag=-25 11439 c(0).imag=531 11440 c(1).imag=19 11441 c(2).imag=9 11442 epsilon=-0.033333, quant(epsilon)=-4 11443 c(-2).real=-69 11444 c(-1).real=-140 11445 c(0).real=4066 11446 c(1).real=131 11447 c(2).real=67 11448 c(-2).imag=-9 11449 c(-1).imag=-16 11450 c(0).imag=426 11451 c(1).imag=12 11452 c(2).imag=5 11453 epsilon=-0.025000, quant(epsilon)=-3 11454 c(-2).real=-52 11455 c(-1).real=-104 11456 c(0).real=4079 11457 c(1).real=100 11458 c(2).real=50 11459 c(-2).imag=-5 11460 c(-1).imag=-9 11461 c(0).imag=320 11462 c(1).imag=7 11463 c(2).imag=3 11464 epsilon=-0.016667, quant(epsilon)=-2 11465 c(-2).real=-34 11466 c(-1).real=-69 11467 c(0).real=4089 11468 c(1).real=67 11469 c(2).real=34 11470 c(-2).imag=-3 11471 c(-1).imag=-4 11472 c(0).imag=213 11473 c(1).imag=3 11474 c(2).imag=1 11475 epsilon=-0.008333, quant(epsilon)=-1 11476 c(-2).real=-17 11477 c(-1).real=-34 11478 c(0).real=4094 11479 c(1).real=34 11480 c(2).real=17 11481 c(-2).imag=-1 11482 c(-1).imag=-1 11483 c(0).imag=107 11484 c(1).imag=0 11485 c(2).imag=0 11486 epsilon=0.000000, quant(epsilon)=0 11487 c(-2).real=0 11488

Supprimé : 4.0.1

Supprimé : 2
























c(-1).real=0 11489 c(0).real=4095 11490 c(1).real=0 11491 c(2).real=0 11492 c(-2).imag=0 11493 c(-1).imag=0 11494 c(0).imag=0 11495 c(1).imag=0 11496 c(2).imag=0 11497 epsilon=0.008333, quant(epsilon)=1 11498 c(-2).real=17 11499 c(-1).real=34 11500 c(0).real=4094 11501 c(1).real=-34 11502 c(2).real=-17 11503 c(-2).imag=0 11504 c(-1).imag=0 11505 c(0).imag=-107 11506 c(1).imag=1 11507 c(2).imag=1 11508 epsilon=0.016667, quant(epsilon)=2 11509 c(-2).real=34 11510 c(-1).real=67 11511 c(0).real=4089 11512 c(1).real=-69 11513 c(2).real=-34 11514 c(-2).imag=-1 11515 c(-1).imag=-3 11516 c(0).imag=-213 11517 c(1).imag=4 11518 c(2).imag=3 11519 epsilon=0.025000, quant(epsilon)=3 11520 c(-2).real=50 11521 c(-1).real=100 11522 c(0).real=4079 11523 c(1).real=-104 11524 c(2).real=-52 11525 c(-2).imag=-3 11526 c(-1).imag=-7 11527 c(0).imag=-320 11528 c(1).imag=9 11529 c(2).imag=5 11530 epsilon=0.033333, quant(epsilon)=4 11531 c(-2).real=67 11532 c(-1).real=131 11533 c(0).real=4066 11534 c(1).real=-140 11535 c(2).real=-69 11536 c(-2).imag=-5 11537 c(-1).imag=-12 11538 c(0).imag=-426 11539 c(1).imag=16 11540 c(2).imag=9 11541 epsilon=0.041667, quant(epsilon)=5 11542 c(-2).real=83 11543 c(-1).real=162 11544 c(0).real=4050 11545 c(1).real=-176 11546 c(2).real=-86 11547

Supprimé : 4.0.1

Supprimé : 2























c(-2).imag=-9 11548 c(-1).imag=-19 11549 c(0).imag=-531 11550 c(1).imag=25 11551 c(2).imag=13 11552 epsilon=0.050000, quant(epsilon)=6 11553 c(-2).real=99 11554 c(-1).real=192 11555 c(0).real=4029 11556 c(1).real=-212 11557 c(2).real=-103 11558 c(-2).imag=-13 11559 c(-1).imag=-28 11560 c(0).imag=-636 11561 c(1).imag=36 11562 c(2).imag=19 11563 epsilon=0.058333, quant(epsilon)=7 11564 c(-2).real=114 11565 c(-1).real=221 11566 c(0).real=4005 11567 c(1).real=-248 11568 c(2).real=-120 11569 c(-2).imag=-18 11570 c(-1).imag=-38 11571 c(0).imag=-739 11572 c(1).imag=49 11573 c(2).imag=25 11574 epsilon=0.066667, quant(epsilon)=8 11575 c(-2).real=129 11576 c(-1).real=249 11577 c(0).real=3978 11578 c(1).real=-283 11579 c(2).real=-136 11580 c(-2).imag=-24 11581 c(-1).imag=-50 11582 c(0).imag=-842 11583 c(1).imag=64 11584 c(2).imag=32 11585 11586 5MHz 11587 epsilon=-0.066667, quant(epsilon)=-8 11588 c(-2).real=-137 11589 c(-1).real=-284 11590 c(0).real=3978 11591 c(1).real=249 11592 c(2).real=129 11593 c(-2).imag=-31 11594 c(-1).imag=-62 11595 c(0).imag=844 11596 c(1).imag=51 11597 c(2).imag=26 11598 epsilon=-0.058333, quant(epsilon)=-7 11599 c(-2).real=-120 11600 c(-1).real=-248 11601 c(0).real=4005 11602 c(1).real=221 11603 c(2).real=114 11604 c(-2).imag=-24 11605 c(-1).imag=-47 11606 c(0).imag=741 11607

Supprimé : 4.0.1

Supprimé : 2












Mis en forme :Police :(asiatique) SimSun

Mis en forme : Normal, Sansnumérotation ni puces








c(1).imag=39 11608 c(2).imag=20 11609 epsilon=-0.050000, quant(epsilon)=-6 11610 c(-2).real=-103 11611 c(-1).real=-212 11612 c(0).real=4029 11613 c(1).real=192 11614 c(2).real=98 11615 c(-2).imag=-18 11616 c(-1).imag=-35 11617 c(0).imag=637 11618 c(1).imag=29 11619 c(2).imag=14 11620 epsilon=-0.041667, quant(epsilon)=-5 11621 c(-2).real=-86 11622 c(-1).real=-176 11623 c(0).real=4050 11624 c(1).real=162 11625 c(2).real=83 11626 c(-2).imag=-12 11627 c(-1).imag=-24 11628 c(0).imag=532 11629 c(1).imag=20 11630 c(2).imag=10 11631 epsilon=-0.033333, quant(epsilon)=-4 11632 c(-2).real=-69 11633 c(-1).real=-140 11634 c(0).real=4066 11635 c(1).real=131 11636 c(2).real=67 11637 c(-2).imag=-8 11638 c(-1).imag=-16 11639 c(0).imag=427 11640 c(1).imag=13 11641 c(2).imag=6 11642 epsilon=-0.025000, quant(epsilon)=-3 11643 c(-2).real=-52 11644 c(-1).real=-105 11645 c(0).real=4079 11646 c(1).real=100 11647 c(2).real=50 11648 c(-2).imag=-5 11649 c(-1).imag=-9 11650 c(0).imag=320 11651 c(1).imag=7 11652 c(2).imag=3 11653 epsilon=-0.016667, quant(epsilon)=-2 11654 c(-2).real=-34 11655 c(-1).real=-69 11656 c(0).real=4089 11657 c(1).real=67 11658 c(2).real=34 11659 c(-2).imag=-2 11660 c(-1).imag=-4 11661 c(0).imag=214 11662 c(1).imag=3 11663 c(2).imag=1 11664 epsilon=-0.008333, quant(epsilon)=-1 11665 c(-2).real=-17 11666

Supprimé : 4.0.1

Supprimé : 2







c(-1).real=-34 11667 c(0).real=4094 11668 c(1).real=34 11669 c(2).real=17 11670 c(-2).imag=-1 11671 c(-1).imag=-1 11672 c(0).imag=107 11673 c(1).imag=1 11674 c(2).imag=0 11675 epsilon=0.000000, quant(epsilon)=0 11676 c(-2).real=0 11677 c(-1).real=0 11678 c(0).real=4095 11679 c(1).real=0 11680 c(2).real=0 11681 c(-2).imag=0 11682 c(-1).imag=0 11683 c(0).imag=0 11684 c(1).imag=0 11685 c(2).imag=0 11686 epsilon=0.008333, quant(epsilon)=1 11687 c(-2).real=17 11688 c(-1).real=34 11689 c(0).real=4094 11690 c(1).real=-34 11691 c(2).real=-17 11692 c(-2).imag=0 11693 c(-1).imag=-1 11694 c(0).imag=-107 11695 c(1).imag=1 11696 c(2).imag=1 11697 epsilon=0.016667, quant(epsilon)=2 11698 c(-2).real=34 11699 c(-1).real=67 11700 c(0).real=4089 11701 c(1).real=-69 11702 c(2).real=-34 11703 c(-2).imag=-1 11704 c(-1).imag=-3 11705 c(0).imag=-214 11706 c(1).imag=4 11707 c(2).imag=2 11708 epsilon=0.025000, quant(epsilon)=3 11709 c(-2).real=50 11710 c(-1).real=100 11711 c(0).real=4079 11712 c(1).real=-105 11713 c(2).real=-52 11714 c(-2).imag=-3 11715 c(-1).imag=-7 11716 c(0).imag=-320 11717 c(1).imag=9 11718 c(2).imag=5 11719 epsilon=0.033333, quant(epsilon)=4 11720 c(-2).real=67 11721 c(-1).real=131 11722 c(0).real=4066 11723 c(1).real=-140 11724 c(2).real=-69 11725

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 0







c(-2).imag=-6 11726 c(-1).imag=-13 11727 c(0).imag=-427 11728 c(1).imag=16 11729 c(2).imag=8 11730 epsilon=0.041667, quant(epsilon)=5 11731 c(-2).real=83 11732 c(-1).real=162 11733 c(0).real=4050 11734 c(1).real=-176 11735 c(2).real=-86 11736 c(-2).imag=-10 11737 c(-1).imag=-20 11738 c(0).imag=-532 11739 c(1).imag=24 11740 c(2).imag=12 11741 epsilon=0.050000, quant(epsilon)=6 11742 c(-2).real=98 11743 c(-1).real=192 11744 c(0).real=4029 11745 c(1).real=-212 11746 c(2).real=-103 11747 c(-2).imag=-14 11748 c(-1).imag=-29 11749 c(0).imag=-637 11750 c(1).imag=35 11751 c(2).imag=18 11752 epsilon=0.058333, quant(epsilon)=7 11753 c(-2).real=114 11754 c(-1).real=221 11755 c(0).real=4005 11756 c(1).real=-248 11757 c(2).real=-120 11758 c(-2).imag=-20 11759 c(-1).imag=-39 11760 c(0).imag=-741 11761 c(1).imag=47 11762 c(2).imag=24 11763 epsilon=0.066667, quant(epsilon)=8 11764 c(-2).real=129 11765 c(-1).real=249 11766 c(0).real=3978 11767 c(1).real=-284 11768 c(2).real=-137 11769 c(-2).imag=-26 11770 c(-1).imag=-51 11771 c(0).imag=-844 11772 c(1).imag=62 11773 c(2).imag=31 11774 11775 11776 10MHz 11777 epsilon=-0.066667, quant(epsilon)=-8 11778 c(-2).real=-137 11779 c(-1).real=-284 11780 c(0).real=3977 11781 c(1).real=249 11782 c(2).real=128 11783 c(-2).imag=-30 11784 c(-1).imag=-61 11785

Supprimé : 4.0.1

Supprimé : 2







c(0).imag=845 11786 c(1).imag=52 11787 c(2).imag=26 11788 epsilon=-0.058333, quant(epsilon)=-7 11789 c(-2).real=-120 11790 c(-1).real=-248 11791 c(0).real=4005 11792 c(1).real=221 11793 c(2).real=114 11794 c(-2).imag=-23 11795 c(-1).imag=-47 11796 c(0).imag=742 11797 c(1).imag=40 11798 c(2).imag=20 11799 epsilon=-0.050000, quant(epsilon)=-6 11800 c(-2).real=-103 11801 c(-1).real=-212 11802 c(0).real=4029 11803 c(1).real=192 11804 c(2).real=98 11805 c(-2).imag=-17 11806 c(-1).imag=-34 11807 c(0).imag=638 11808 c(1).imag=30 11809 c(2).imag=15 11810 epsilon=-0.041667, quant(epsilon)=-5 11811 c(-2).real=-86 11812 c(-1).real=-176 11813 c(0).real=4049 11814 c(1).real=162 11815 c(2).real=83 11816 c(-2).imag=-12 11817 c(-1).imag=-24 11818 c(0).imag=533 11819 c(1).imag=21 11820 c(2).imag=10 11821 epsilon=-0.033333, quant(epsilon)=-4 11822 c(-2).real=-69 11823 c(-1).real=-140 11824 c(0).real=4066 11825 c(1).real=131 11826 c(2).real=67 11827 c(-2).imag=-8 11828 c(-1).imag=-15 11829 c(0).imag=427 11830 c(1).imag=13 11831 c(2).imag=7 11832 epsilon=-0.025000, quant(epsilon)=-3 11833 c(-2).real=-52 11834 c(-1).real=-105 11835 c(0).real=4079 11836 c(1).real=100 11837 c(2).real=50 11838 c(-2).imag=-4 11839 c(-1).imag=-9 11840 c(0).imag=321 11841 c(1).imag=8 11842 c(2).imag=4 11843 epsilon=-0.016667, quant(epsilon)=-2 11844

Supprimé : 4.0.1

Supprimé : 2







c(-2).real=-34 11845 c(-1).real=-69 11846 c(0).real=4089 11847 c(1).real=67 11848 c(2).real=34 11849 c(-2).imag=-2 11850 c(-1).imag=-4 11851 c(0).imag=214 11852 c(1).imag=3 11853 c(2).imag=2 11854 epsilon=-0.008333, quant(epsilon)=-1 11855 c(-2).real=-17 11856 c(-1).real=-34 11857 c(0).real=4094 11858 c(1).real=34 11859 c(2).real=17 11860 c(-2).imag=-1 11861 c(-1).imag=-1 11862 c(0).imag=107 11863 c(1).imag=1 11864 c(2).imag=0 11865 epsilon=0.000000, quant(epsilon)=0 11866 c(-2).real=0 11867 c(-1).real=0 11868 c(0).real=4095 11869 c(1).real=0 11870 c(2).real=0 11871 c(-2).imag=0 11872 c(-1).imag=0 11873 c(0).imag=0 11874 c(1).imag=0 11875 c(2).imag=0 11876 epsilon=0.008333, quant(epsilon)=1 11877 c(-2).real=17 11878 c(-1).real=34 11879 c(0).real=4094 11880 c(1).real=-34 11881 c(2).real=-17 11882 c(-2).imag=0 11883 c(-1).imag=-1 11884 c(0).imag=-107 11885 c(1).imag=1 11886 c(2).imag=1 11887 epsilon=0.016667, quant(epsilon)=2 11888 c(-2).real=34 11889 c(-1).real=67 11890 c(0).real=4089 11891 c(1).real=-69 11892 c(2).real=-34 11893 c(-2).imag=-2 11894 c(-1).imag=-3 11895 c(0).imag=-214 11896 c(1).imag=4 11897 c(2).imag=2 11898 epsilon=0.025000, quant(epsilon)=3 11899 c(-2).real=50 11900 c(-1).real=100 11901 c(0).real=4079 11902 c(1).real=-105 11903

Supprimé : 4.0.1

Supprimé : 2

Supprimé : 0







c(2).real=-52 11904 c(-2).imag=-4 11905 c(-1).imag=-8 11906 c(0).imag=-321 11907 c(1).imag=9 11908 c(2).imag=4 11909 epsilon=0.033333, quant(epsilon)=4 11910 c(-2).real=67 11911 c(-1).real=131 11912 c(0).real=4066 11913 c(1).real=-140 11914 c(2).real=-69 11915 c(-2).imag=-7 11916 c(-1).imag=-13 11917 c(0).imag=-427 11918 c(1).imag=15 11919 c(2).imag=8 11920 epsilon=0.041667, quant(epsilon)=5 11921 c(-2).real=83 11922 c(-1).real=162 11923 c(0).real=4049 11924 c(1).real=-176 11925 c(2).real=-86 11926 c(-2).imag=-10 11927 c(-1).imag=-21 11928 c(0).imag=-533 11929 c(1).imag=24 11930 c(2).imag=12 11931 epsilon=0.050000, quant(epsilon)=6 11932 c(-2).real=98 11933 c(-1).real=192 11934 c(0).real=4029 11935 c(1).real=-212 11936 c(2).real=-103 11937 c(-2).imag=-15 11938 c(-1).imag=-30 11939 c(0).imag=-638 11940 c(1).imag=34 11941 c(2).imag=17 11942 epsilon=0.058333, quant(epsilon)=7 11943 c(-2).real=114 11944 c(-1).real=221 11945 c(0).real=4005 11946 c(1).real=-248 11947 c(2).real=-120 11948 c(-2).imag=-20 11949 c(-1).imag=-40 11950 c(0).imag=-742 11951 c(1).imag=47 11952 c(2).imag=23 11953 epsilon=0.066667, quant(epsilon)=8 11954 c(-2).real=128 11955 c(-1).real=249 11956 c(0).real=3977 11957 c(1).real=-284 11958 c(2).real=-137 11959 c(-2).imag=-26 11960 c(-1).imag=-52 11961 c(0).imag=-845 11962

Supprimé : 4.0.1

Supprimé : 2







c(1).imag=61 11963 c(2).imag=30 11964 11965 11966 15MHz : 11967 epsilon=-0.066667 , quant(espilon)=-8 11968 c(-2).real = -137 11969 c(-1).real = -284 11970 c(-0).real = 3977 11971 c(1).real = 249 11972 c(2).real = 128 11973 c(-2).imag = -30 11974 c(-1).imag = -61 11975 c(-0).imag = 845 11976 c(1).imag = 52 11977 c(2).imag = 26 11978 epsilon=-0.058333 , quant(espilon)=-7 11979 c(-2).real = -120 11980 c(-1).real = -248 11981 c(-0).real = 4005 11982 c(1).real = 221 11983 c(2).real = 114 11984 c(-2).imag = -23 11985 c(-1).imag = -46 11986 c(-0).imag = 742 11987 c(1).imag = 40 11988 c(2).imag = 21 11989 epsilon=-0.050000 , quant(espilon)=-6 11990 c(-2).real = -103 11991 c(-1).real = -212 11992 c(-0).real = 4029 11993 c(1).real = 192 11994 c(2).real = 98 11995 c(-2).imag = -17 11996 c(-1).imag = -34 11997 c(-0).imag = 638 11998 c(1).imag = 30 11999 c(2).imag = 15 12000 epsilon=-0.041667 , quant(espilon)=-5 12001 c(-2).real = -86 12002 c(-1).real = -176 12003 c(-0).real = 4049 12004 c(1).real = 162 12005 c(2).real = 83 12006 c(-2).imag = -12 12007 c(-1).imag = -24 12008 c(-0).imag = 533 12009 c(1).imag = 21 12010 c(2).imag = 11 12011 epsilon=-0.033333 , quant(espilon)=-4 12012 c(-2).real = -69 12013 c(-1).real = -140 12014 c(-0).real = 4066 12015 c(1).real = 131 12016 c(2).real = 67 12017 c(-2).imag = -8 12018 c(-1).imag = -15 12019 c(-0).imag = 427 12020 c(1).imag = 14 12021 c(2).imag = 7 12022

Supprimé : 4.0.1

Supprimé : 2







epsilon=-0.025000 , quant(espilon)=-3 12023 c(-2).real = -52 12024 c(-1).real = -105 12025 c(-0).real = 4079 12026 c(1).real = 100 12027 c(2).real = 50 12028 c(-2).imag = -4 12029 c(-1).imag = -8 12030 c(-0).imag = 321 12031 c(1).imag = 8 12032 c(2).imag = 4 12033 epsilon=-0.016667 , quant(espilon)=-2 12034 c(-2).real = -34 12035 c(-1).real = -69 12036 c(-0).real = 4089 12037 c(1).real = 67 12038 c(2).real = 34 12039 c(-2).imag = -2 12040 c(-1).imag = -4 12041 c(-0).imag = 214 12042 c(1).imag = 3 12043 c(2).imag = 2 12044 epsilon=-0.008333 , quant(espilon)=-1 12045 c(-2).real = -17 12046 c(-1).real = -34 12047 c(-0).real = 4094 12048 c(1).real = 34 12049 c(2).real = 17 12050 c(-2).imag = -1 12051 c(-1).imag = -1 12052 c(-0).imag = 107 12053 c(1).imag = 1 12054 c(2).imag = 0 12055 epsilon=0.000000 , quant(espilon)=0 12056 c(-2).real = -0 12057 c(-1).real = -0 12058 c(0).real = 4095 12059 c(1).real = -0 12060 c(2).real = -0 12061 c(-2).imag = -0 12062 c(-1).imag = -0 12063 c(0).imag = 0 12064 c(1).imag = 0 12065 c(2).imag = 0 12066 epsilon=0.008333 , quant(espilon)=1 12067 c(-2).real = 17 12068 c(-1).real = 34 12069 c(-0).real = 4094 12070 c(1).real = -34 12071 c(2).real = -17 12072 c(-2).imag = -0 12073 c(-1).imag = -1 12074 c(-0).imag = -107 12075 c(1).imag = 1 12076 c(2).imag = 1 12077 epsilon=0.016667 , quant(espilon)=2 12078 c(-2).real = 34 12079 c(-1).real = 67 12080 c(-0).real = 4089 12081 c(1).real = -69 12082

Supprimé : 4.0.1

Supprimé : 2







c(2).real = -34 12083 c(-2).imag = -2 12084 c(-1).imag = -3 12085 c(-0).imag = -214 12086 c(1).imag = 4 12087 c(2).imag = 2 12088 epsilon=0.025000 , quant(espilon)=3 12089 c(-2).real = 50 12090 c(-1).real = 100 12091 c(-0).real = 4079 12092 c(1).real = -105 12093 c(2).real = -52 12094 c(-2).imag = -4 12095 c(-1).imag = -8 12096 c(-0).imag = -321 12097 c(1).imag = 8 12098 c(2).imag = 4 12099 epsilon=0.033333 , quant(espilon)=4 12100

c(-2).real = 67 12101 c(-1).real = 131 12102 c(-0).real = 4066 12103 c(1).real = -140 12104 c(2).real = -69 12105 c(-2).imag = -7 12106 c(-1).imag = -14 12107 c(-0).imag = -427 12108 c(1).imag = 15 12109 c(2).imag = 8 12110 epsilon=0.041667 , quant(espilon)=5 12111 c(-2).real = 83 12112 c(-1).real = 162 12113 c(-0).real = 4049 12114 c(1).real = -176 12115 c(2).real = -86 12116 c(-2).imag = -11 12117 c(-1).imag = -21 12118 c(-0).imag = -533 12119 c(1).imag = 24 12120 c(2).imag = 12 12121 epsilon=0.050000 , quant(espilon)=6 12122 c(-2).real = 98 12123 c(-1).real = 192 12124 c(-0).real = 4029 12125 c(1).real = -212 12126 c(2).real = -103 12127 c(-2).imag = -15 12128 c(-1).imag = -30 12129 c(-0).imag = -638 12130 c(1).imag = 34 12131 c(2).imag = 17 12132 epsilon=0.058333 , quant(espilon)=7 12133 c(-2).real = 114 12134 c(-1).real = 221 12135 c(-0).real = 4005 12136 c(1).real = -248 12137 c(2).real = -120 12138 c(-2).imag = -21 12139 c(-1).imag = -40 12140 c(-0).imag = -742 12141 c(1).imag = 46 12142

Supprimé : 4.0.1

Supprimé : 2







c(2).imag = 23 12143 epsilon=0.066667 , quant(espilon)=8 12144 c(-2).real = 128 12145 c(-1).real = 249 12146 c(-0).real = 3977 12147 c(1).real = -284 12148 c(2).real = -137 12149 c(-2).imag = -27 12150 c(-1).imag = -52 12151 c(-0).imag = -845 12152 c(1).imag = 61 12153 c(2).imag = 30 12154 12155 12156 20MHz : 12157 epsilon=-0.066667 , quant(espilon)=-8 12158 c(-2).real = -137 12159 c(-1).real = -284 12160 c(-0).real = 3977 12161 c(1).real = 249 12162 c(2).real = 128 12163 c(-2).imag = -30 12164 c(-1).imag = -61 12165 c(-0).imag = 845 12166 c(1).imag = 52 12167 c(2).imag = 27 12168 epsilon=-0.058333 , quant(espilon)=-7 12169 c(-2).real = -120 12170 c(-1).real = -248 12171 c(-0).real = 4005 12172 c(1).real = 221 12173 c(2).real = 114 12174 c(-2).imag = -23 12175 c(-1).imag = -46 12176 c(-0).imag = 742 12177 c(1).imag = 41 12178 c(2).imag = 21 12179 epsilon=-0.050000 , quant(espilon)=-6 12180 c(-2).real = -103 12181 c(-1).real = -212 12182 c(-0).real = 4029 12183 c(1).real = 192 12184 c(2).real = 98 12185 c(-2).imag = -17 12186 c(-1).imag = -34 12187 c(-0).imag = 638 12188 c(1).imag = 30 12189 c(2).imag = 15 12190 epsilon=-0.041667 , quant(espilon)=-5 12191 c(-2).real = -86 12192 c(-1).real = -176 12193 c(-0).real = 4049 12194 c(1).real = 162 12195 c(2).real = 83 12196 c(-2).imag = -12 12197 c(-1).imag = -23 12198 c(-0).imag = 533 12199 c(1).imag = 21 12200 c(2).imag = 11 12201 epsilon=-0.033333 , quant(espilon)=-4 12202

Supprimé : 4.0.1

Supprimé : 2







c(-2).real = -69 12203 c(-1).real = -140 12204 c(-0).real = 4066 12205 c(1).real = 131 12206 c(2).real = 67 12207 c(-2).imag = -7 12208 c(-1).imag = -15 12209 c(-0).imag = 427 12210 c(1).imag = 14 12211 c(2).imag = 7 12212 epsilon=-0.025000 , quant(espilon)=-3 12213 c(-2).real = -52 12214 c(-1).real = -105 12215 c(-0).real = 4079 12216 c(1).real = 100 12217 c(2).real = 50 12218 c(-2).imag = -4 12219 c(-1).imag = -8 12220 c(-0).imag = 321 12221 c(1).imag = 8 12222 c(2).imag = 4 12223 epsilon=-0.016667 , quant(espilon)=-2 12224 c(-2).real = -34 12225 c(-1).real = -69 12226 c(-0).real = 4089 12227 c(1).real = 67 12228 c(2).real = 34 12229 c(-2).imag = -2 12230 c(-1).imag = -4 12231 c(-0).imag = 214 12232 c(1).imag = 3 12233 c(2).imag = 2 12234 epsilon=-0.008333 , quant(espilon)=-1 12235 c(-2).real = -17 12236 c(-1).real = -34 12237 c(-0).real = 4094 12238 c(1).real = 34 12239 c(2).real = 17 12240 c(-2).imag = -1 12241 c(-1).imag = -1 12242 c(-0).imag = 107 12243 c(1).imag = 1 12244 c(2).imag = 0 12245 epsilon=0.000000 , quant(espilon)=0 12246 c(-2).real = -0 12247 c(-1).real = -0 12248 c(0).real = 4095 12249 c(1).real = -0 12250 c(2).real = -0 12251 c(-2).imag = -0 12252 c(-1).imag = -0 12253 c(0).imag = 0 12254 c(1).imag = 0 12255 c(2).imag = 0 12256 epsilon=0.008333 , quant(espilon)=1 12257 c(-2).real = 17 12258 c(-1).real = 34 12259 c(-0).real = 4094 12260 c(1).real = -34 12261 c(2).real = -17 12262

Supprimé : 4.0.1

Supprimé : 2







c(-2).imag = -0 12263 c(-1).imag = -1 12264 c(-0).imag = -107 12265 c(1).imag = 1 12266 c(2).imag = 1 12267 epsilon=0.016667 , quant(espilon)=2 12268 c(-2).real = 34 12269 c(-1).real = 67 12270 c(-0).real = 4089 12271 c(1).real = -69 12272 c(2).real = -34 12273 c(-2).imag = -2 12274 c(-1).imag = -3 12275 c(-0).imag = -214 12276 c(1).imag = 4 12277 c(2).imag = 2 12278 epsilon=0.025000 , quant(espilon)=3 12279 c(-2).real = 50 12280 c(-1).real = 100 12281 c(-0).real = 4079 12282 c(1).real = -105 12283 c(2).real = -52 12284 c(-2).imag = -4 12285 c(-1).imag = -8 12286 c(-0).imag = -321 12287 c(1).imag = 8 12288 c(2).imag = 4 12289 epsilon=0.033333 , quant(espilon)=4 12290 c(-2).real = 67 12291 c(-1).real = 131 12292 c(-0).real = 4066 12293 c(1).real = -140 12294 c(2).real = -69 12295 c(-2).imag = -7 12296 c(-1).imag = -14 12297 c(-0).imag = -427 12298 c(1).imag = 15 12299 c(2).imag = 7 12300 epsilon=0.041667 , quant(espilon)=5 12301 c(-2).real = 83 12302 c(-1).real = 162 12303 c(-0).real = 4049 12304 c(1).real = -176 12305 c(2).real = -86 12306 c(-2).imag = -11 12307 c(-1).imag = -21 12308 c(-0).imag = -533 12309 c(1).imag = 23 12310 c(2).imag = 12 12311 epsilon=0.050000 , quant(espilon)=6 12312 c(-2).real = 98 12313 c(-1).real = 192 12314 c(-0).real = 4029 12315 c(1).real = -212 12316 c(2).real = -103 12317 c(-2).imag = -15 12318 c(-1).imag = -30 12319 c(-0).imag = -638 12320 c(1).imag = 34 12321 c(2).imag = 17 12322

Supprimé : 4.0.1

Supprimé : 2







epsilon=0.058333 , quant(espilon)=7 12323 c(-2).real = 114 12324 c(-1).real = 221 12325 c(-0).real = 4005 12326 c(1).real = -248 12327 c(2).real = -120 12328 c(-2).imag = -21 12329 c(-1).imag = -41 12330 c(-0).imag = -742 12331 c(1).imag = 46 12332 c(2).imag = 23 12333 epsilon=0.066667 , quant(espilon)=8 12334 c(-2).real = 128 12335 c(-1).real = 249 12336 c(-0).real = 3977 12337 c(1).real = -284 12338 c(2).real = -137 12339 c(-2).imag = -27 12340 c(-1).imag = -52 12341 c(-0).imag = -845 12342 c(1).imag = 61 12343 c(2).imag = 30 12344

12345

Annex 10. Synchro sequences fix point implementatio n 12346

12347 131070 0 // cos(0 Pi/63) -sin(0 Pi/63) 12348 130910 -6534 // cos(1 Pi/63) -sin(1 Pi/63) 12349 130420 -13050 // cos(2 Pi/63) -sin(2 Pi/63) 12350 129608 -19536 // cos(3 Pi/63) -sin(3 Pi/63) 12351 128474 -25972 // cos(4 Pi/63) -sin(4 Pi/63) 12352 127018 -32342 // cos(5 Pi/63) -sin(5 Pi/63) 12353 125248 -38634 // cos(6 Pi/63) -sin(6 Pi/63) 12354 123168 -44830 // cos(7 Pi/63) -sin(7 Pi/63) 12355 120780 -50912 // cos(8 Pi/63) -sin(8 Pi/63) 12356 118092 -56870 // cos(9 Pi/63) -sin(9 Pi/63) 12357 115110 -62686 // cos(10 Pi/63) -sin(10 Pi/63) 12358 111842 -68346 // cos(11 Pi/63) -sin(11 Pi/63) 12359 108296 -73836 // cos(12 Pi/63) -sin(12 Pi/63) 12360 104482 -79142 // cos(13 Pi/63) -sin(13 Pi/63) 12361 100406 -84252 // cos(14 Pi/63) -sin(14 Pi/63) 12362 96082 -89152 // cos(15 Pi/63) -sin(15 Pi/63) 12363 91520 -93830 // cos(16 Pi/63) -sin(16 Pi/63) 12364 86728 -98276 // cos(17 Pi/63) -sin(17 Pi/63) 12365 81722 -102476 // cos(18 Pi/63) -sin(18 Pi/63) 12366 76512 -106422 // cos(19 Pi/63) -sin(19 Pi/63) 12367 71112 -110104 // cos(20 Pi/63) -sin(20 Pi/63) 12368 65536 -113512 // cos(21 Pi/63) -sin(21 Pi/63) 12369 59796 -116638 // cos(22 Pi/63) -sin(22 Pi/63) 12370 53908 -119472 // cos(23 Pi/63) -sin(23 Pi/63) 12371 47886 -122012 // cos(24 Pi/63) -sin(24 Pi/63) 12372 41744 -124246 // cos(25 Pi/63) -sin(25 Pi/63) 12373 35500 -126174 // cos(26 Pi/63) -sin(26 Pi/63) 12374 29166 -127786 // cos(27 Pi/63) -sin(27 Pi/63) 12375 22760 -129080 // cos(28 Pi/63) -sin(28 Pi/63) 12376 16298 -130054 // cos(29 Pi/63) -sin(29 Pi/63) 12377 9796 -130706 // cos(30 Pi/63) -sin(30 Pi/63) 12378 3268 -131032 // cos(31 Pi/63) -sin(31 Pi/63) 12379 -3268 -131032 // cos(32 Pi/63) -sin(32 Pi/63) 12380 -9796 -130706 // cos(33 Pi/63) -sin(33 Pi/63) 12381 -16298 -130054 // cos(34 Pi/63) -sin(34 Pi/63) 12382 -22760 -129080 // cos(35 Pi/63) -sin(35 Pi/63) 12383 -29166 -127786 // cos(36 Pi/63) -sin(36 Pi/63) 12384 -35500 -126174 // cos(37 Pi/63) -sin(37 Pi/63) 12385 -41744 -124246 // cos(38 Pi/63) -sin(38 Pi/63) 12386 -47886 -122012 // cos(39 Pi/63) -sin(39 Pi/63) 12387 -53908 -119472 // cos(40 Pi/63) -sin(40 Pi/63) 12388 -59796 -116638 // cos(41 Pi/63) -sin(41 Pi/63) 12389

Supprimé : 4.0.1

Supprimé : 2







-65536 -113512 // cos(42 Pi/63) -sin(42 Pi/63) 12390 -71112 -110104 // cos(43 Pi/63) -sin(43 Pi/63) 12391 -76512 -106422 // cos(44 Pi/63) -sin(44 Pi/63) 12392 -81722 -102476 // cos(45 Pi/63) -sin(45 Pi/63) 12393 -86728 -98276 // cos(46 Pi/63) -sin(46 Pi/63) 12394 -91520 -93830 // cos(47 Pi/63) -sin(47 Pi/63) 12395 -96082 -89152 // cos(48 Pi/63) -sin(48 Pi/63) 12396 -100406 -84252 // cos(49 Pi/63) -sin(49 Pi/63) 12397 -104482 -79142 // cos(50 Pi/63) -sin(50 Pi/63) 12398 -108296 -73836 // cos(51 Pi/63) -sin(51 Pi/63) 12399 -111842 -68346 // cos(52 Pi/63) -sin(52 Pi/63) 12400 -115110 -62686 // cos(53 Pi/63) -sin(53 Pi/63) 12401 -118092 -56870 // cos(54 Pi/63) -sin(54 Pi/63) 12402 -120780 -50912 // cos(55 Pi/63) -sin(55 Pi/63) 12403 -123168 -44830 // cos(56 Pi/63) -sin(56 Pi/63) 12404 -125248 -38634 // cos(57 Pi/63) -sin(57 Pi/63) 12405 -127018 -32342 // cos(58 Pi/63) -sin(58 Pi/63) 12406 -128474 -25972 // cos(59 Pi/63) -sin(59 Pi/63) 12407 -129608 -19536 // cos(60 Pi/63) -sin(60 Pi/63) 12408 -130420 -13050 // cos(61 Pi/63) -sin(61 Pi/63) 12409 -130910 -6534 // cos(62 Pi/63) -sin(62 Pi/63) 12410 -131071 0 // cos(63 Pi/63) -sin(63 Pi/63) 12411 -130910 6534 // cos(64 Pi/63) -sin(64 Pi/63) 12412 -130420 13050 // cos(65 Pi/63) -sin(65 Pi/63) 12413 -129608 19536 // cos(66 Pi/63) -sin(66 Pi/63) 12414 -128474 25972 // cos(67 Pi/63) -sin(67 Pi/63) 12415 -127018 32342 // cos(68 Pi/63) -sin(68 Pi/63) 12416 -125248 38634 // cos(69 Pi/63) -sin(69 Pi/63) 12417 -123168 44830 // cos(70 Pi/63) -sin(70 Pi/63) 12418 -120780 50912 // cos(71 Pi/63) -sin(71 Pi/63) 12419 -118092 56870 // cos(72 Pi/63) -sin(72 Pi/63) 12420 -115110 62686 // cos(73 Pi/63) -sin(73 Pi/63) 12421 -111842 68346 // cos(74 Pi/63) -sin(74 Pi/63) 12422 -108296 73836 // cos(75 Pi/63) -sin(75 Pi/63) 12423 -104482 79142 // cos(76 Pi/63) -sin(76 Pi/63) 12424 -100406 84252 // cos(77 Pi/63) -sin(77 Pi/63) 12425 -96082 89152 // cos(78 Pi/63) -sin(78 Pi/63) 12426 -91520 93830 // cos(79 Pi/63) -sin(79 Pi/63) 12427 -86728 98276 // cos(80 Pi/63) -sin(80 Pi/63) 12428 -81722 102476 // cos(81 Pi/63) -sin(81 Pi/63) 12429 -76512 106422 // cos(82 Pi/63) -sin(82 Pi/63) 12430 -71112 110104 // cos(83 Pi/63) -sin(83 Pi/63) 12431 -65536 113512 // cos(84 Pi/63) -sin(84 Pi/63) 12432 -59796 116638 // cos(85 Pi/63) -sin(85 Pi/63) 12433 -53908 119472 // cos(86 Pi/63) -sin(86 Pi/63) 12434 -47886 122012 // cos(87 Pi/63) -sin(87 Pi/63) 12435 -41744 124246 // cos(88 Pi/63) -sin(88 Pi/63) 12436 -35500 126174 // cos(89 Pi/63) -sin(89 Pi/63) 12437 -29166 127786 // cos(90 Pi/63) -sin(90 Pi/63) 12438 -22760 129080 // cos(91 Pi/63) -sin(91 Pi/63) 12439 -16298 130054 // cos(92 Pi/63) -sin(92 Pi/63) 12440 -9796 130706 // cos(93 Pi/63) -sin(93 Pi/63) 12441 -3268 131032 // cos(94 Pi/63) -sin(94 Pi/63) 12442 3268 131032 // cos(95 Pi/63) -sin(95 Pi/63) 12443 9796 130706 // cos(96 Pi/63) -sin(96 Pi/63) 12444 16298 130054 // cos(97 Pi/63) -sin(97 Pi/63) 12445 22760 129080 // cos(98 Pi/63) -sin(98 Pi/63) 12446 29166 127786 // cos(99 Pi/63) -sin(99 Pi/63) 12447 35500 126174 // cos(100 Pi/63) -sin(100 Pi/63) 12448 41744 124246 // cos(101 Pi/63) -sin(101 Pi/63) 12449 47886 122012 // cos(102 Pi/63) -sin(102 Pi/63) 12450 53908 119472 // cos(103 Pi/63) -sin(103 Pi/63) 12451 59796 116638 // cos(104 Pi/63) -sin(104 Pi/63) 12452 65536 113512 // cos(105 Pi/63) -sin(105 Pi/63) 12453 71112 110104 // cos(106 Pi/63) -sin(106 Pi/63) 12454 76512 106422 // cos(107 Pi/63) -sin(107 Pi/63) 12455 81722 102476 // cos(108 Pi/63) -sin(108 Pi/63) 12456 86728 98276 // cos(109 Pi/63) -sin(109 Pi/63) 12457 91520 93830 // cos(110 Pi/63) -sin(110 Pi/63) 12458 96082 89152 // cos(111 Pi/63) -sin(111 Pi/63) 12459 100406 84252 // cos(112 Pi/63) -sin(112 Pi/63) 12460 104482 79142 // cos(113 Pi/63) -sin(113 Pi/63) 12461 108296 73836 // cos(114 Pi/63) -sin(114 Pi/63) 12462 111842 68346 // cos(115 Pi/63) -sin(115 Pi/63) 12463 115110 62686 // cos(116 Pi/63) -sin(116 Pi/63) 12464 118092 56870 // cos(117 Pi/63) -sin(117 Pi/63) 12465

Supprimé : 4.0.1

Supprimé : 2







120780 50912 // cos(118 Pi/63) -sin(118 Pi/63) 12466 123168 44830 // cos(119 Pi/63) -sin(119 Pi/63) 12467 125248 38634 // cos(120 Pi/63) -sin(120 Pi/63) 12468 127018 32342 // cos(121 Pi/63) -sin(121 Pi/63) 12469 128474 25972 // cos(122 Pi/63) -sin(122 Pi/63) 12470 129608 19536 // cos(123 Pi/63) -sin(123 Pi/63) 12471 130420 13050 // cos(124 Pi/63) -sin(124 Pi/63) 12472 130910 6534 // cos(125 Pi/63) -sin(125 Pi/63) 12473

Annex 11. Donwlink fix point power offsets 12474 65536 // 0.0 dB 3440 // 25.6 dB 181 // 51 .2 dB 10 // 76.8 dB 12475 64786 // 0.1 dB 3400 // 25.7 dB 179 // 51 .3 dB 9 // 76.9 dB 12476 64044 // 0.2 dB 3361 // 25.8 dB 177 // 51 .4 dB 9 // 77.0 dB 12477 63311 // 0.3 dB 3323 // 25.9 dB 174 // 51 .5 dB 9 // 77.1 dB 12478 62587 // 0.4 dB 3285 // 26.0 dB 173 // 51 .6 dB 9 // 77.2 dB 12479 61870 // 0.5 dB 3247 // 26.1 dB 171 // 51 .7 dB 9 // 77.3 dB 12480 61162 // 0.6 dB 3210 // 26.2 dB 169 // 51 .8 dB 9 // 77.4 dB 12481 60462 // 0.7 dB 3173 // 26.3 dB 167 // 51 .9 dB 9 // 77.5 dB 12482 59770 // 0.8 dB 3137 // 26.4 dB 165 // 52 .0 dB 9 // 77.6 dB 12483 59085 // 0.9 dB 3101 // 26.5 dB 163 // 52 .1 dB 9 // 77.7 dB 12484 58409 // 1.0 dB 3065 // 26.6 dB 161 // 52 .2 dB 9 // 77.8 dB 12485 57741 // 1.1 dB 3030 // 26.7 dB 159 // 52 .3 dB 8 // 77.9 dB 12486 57080 // 1.2 dB 2996 // 26.8 dB 157 // 52 .4 dB 8 // 78.0 dB 12487 56426 // 1.3 dB 2961 // 26.9 dB 156 // 52 .5 dB 8 // 78.1 dB 12488 55780 // 1.4 dB 2928 // 27.0 dB 154 // 52 .6 dB 8 // 78.2 dB 12489 55142 // 1.5 dB 2894 // 27.1 dB 152 // 52 .7 dB 8 // 78.3 dB 12490 54511 // 1.6 dB 2861 // 27.2 dB 150 // 52 .8 dB 8 // 78.4 dB 12491 53887 // 1.7 dB 2828 // 27.3 dB 149 // 52 .9 dB 8 // 78.5 dB 12492 53270 // 1.8 dB 2796 // 27.4 dB 147 // 53 .0 dB 8 // 78.6 dB 12493 52660 // 1.9 dB 2764 // 27.5 dB 145 // 53 .1 dB 8 // 78.7 dB 12494 52057 // 2.0 dB 2732 // 27.6 dB 144 // 53 .2 dB 8 // 78.8 dB 12495 51461 // 2.1 dB 2701 // 27.7 dB 142 // 53 .3 dB 8 // 78.9 dB 12496 50872 // 2.2 dB 2670 // 27.8 dB 140 // 53 .4 dB 7 // 79.0 dB 12497 50290 // 2.3 dB 2639 // 27.9 dB 139 // 53 .5 dB 7 // 79.1 dB 12498 49714 // 2.4 dB 2609 // 28.0 dB 137 // 53 .6 dB 7 // 79.2 dB 12499 49145 // 2.5 dB 2579 // 28.1 dB 135 // 53 .7 dB 7 // 79.3 dB 12500 48583 // 2.6 dB 2550 // 28.2 dB 134 // 53 .8 dB 7 // 79.4 dB 12501 48027 // 2.7 dB 2521 // 28.3 dB 132 // 53 .9 dB 7 // 79.5 dB 12502 47477 // 2.8 dB 2492 // 28.4 dB 131 // 54 .0 dB 7 // 79.6 dB 12503 46933 // 2.9 dB 2463 // 28.5 dB 129 // 54 .1 dB 7 // 79.7 dB 12504 46396 // 3.0 dB 2435 // 28.6 dB 128 // 54 .2 dB 7 // 79.8 dB 12505 45865 // 3.1 dB 2407 // 28.7 dB 126 // 54 .3 dB 7 // 79.9 dB 12506 45340 // 3.2 dB 2380 // 28.8 dB 125 // 54 .4 dB 7 // 80.0 dB 12507 44821 // 3.3 dB 2352 // 28.9 dB 124 // 54 .5 dB 7 // 80.1 dB 12508 44308 // 3.4 dB 2325 // 29.0 dB 122 // 54 .6 dB 7 // 80.2 dB 12509 43801 // 3.5 dB 2299 // 29.1 dB 121 // 54 .7 dB 6 // 80.3 dB 12510 43299 // 3.6 dB 2272 // 29.2 dB 119 // 54 .8 dB 6 // 80.4 dB 12511 42804 // 3.7 dB 2246 // 29.3 dB 118 // 54 .9 dB 6 // 80.5 dB 12512 42314 // 3.8 dB 2221 // 29.4 dB 117 // 55 .0 dB 6 // 80.6 dB 12513 41829 // 3.9 dB 2195 // 29.5 dB 115 // 55 .1 dB 6 // 80.7 dB 12514 41351 // 4.0 dB 2170 // 29.6 dB 114 // 55 .2 dB 6 // 80.8 dB 12515 40877 // 4.1 dB 2145 // 29.7 dB 113 // 55 .3 dB 6 // 80.9 dB 12516 40409 // 4.2 dB 2121 // 29.8 dB 111 // 55 .4 dB 6 // 81.0 dB 12517 39947 // 4.3 dB 2097 // 29.9 dB 110 // 55 .5 dB 6 // 81.1 dB 12518 39489 // 4.4 dB 2073 // 30.0 dB 109 // 55 .6 dB 6 // 81.2 dB 12519 39037 // 4.5 dB 2049 // 30.1 dB 108 // 55 .7 dB 6 // 81.3 dB 12520 38591 // 4.6 dB 2025 // 30.2 dB 106 // 55 .8 dB 6 // 81.4 dB 12521 38149 // 4.7 dB 2002 // 30.3 dB 105 // 55 .9 dB 6 // 81.5 dB 12522 37712 // 4.8 dB 1979 // 30.4 dB 104 // 56 .0 dB 6 // 81.6 dB 12523 37280 // 4.9 dB 1957 // 30.5 dB 103 // 56 .1 dB 6 // 81.7 dB 12524 36854 // 5.0 dB 1934 // 30.6 dB 102 // 56 .2 dB 5 // 81.8 dB 12525 36432 // 5.1 dB 1912 // 30.7 dB 100 // 56 .3 dB 5 // 81.9 dB 12526 36015 // 5.2 dB 1890 // 30.8 dB 99 // 56. 4 dB 5 // 82.0 dB 12527 35603 // 5.3 dB 1869 // 30.9 dB 98 // 56. 5 dB 5 // 82.1 dB 12528 35195 // 5.4 dB 1847 // 31.0 dB 97 // 56. 6 dB 5 // 82.2 dB 12529 34792 // 5.5 dB 1826 // 31.1 dB 96 // 56. 7 dB 5 // 82.3 dB 12530 34394 // 5.6 dB 1805 // 31.2 dB 95 // 56. 8 dB 5 // 82.4 dB 12531 34000 // 5.7 dB 1784 // 31.3 dB 94 // 56. 9 dB 5 // 82.5 dB 12532 33611 // 5.8 dB 1764 // 31.4 dB 93 // 57. 0 dB 5 // 82.6 dB 12533 33226 // 5.9 dB 1744 // 31.5 dB 92 // 57. 1 dB 5 // 82.7 dB 12534 32846 // 6.0 dB 1724 // 31.6 dB 91 // 57. 2 dB 5 // 82.8 dB 12535 32470 // 6.1 dB 1704 // 31.7 dB 90 // 57. 3 dB 5 // 82.9 dB 12536 32098 // 6.2 dB 1685 // 31.8 dB 89 // 57. 4 dB 5 // 83.0 dB 12537 31731 // 6.3 dB 1665 // 31.9 dB 88 // 57. 5 dB 5 // 83.1 dB 12538

Supprimé : 4.0.1

Supprimé : 2







31368 // 6.4 dB 1646 // 32.0 dB 87 // 57. 6 dB 5 // 83.2 dB 12539 31009 // 6.5 dB 1627 // 32.1 dB 86 // 57. 7 dB 5 // 83.3 dB 12540 30654 // 6.6 dB 1609 // 32.2 dB 85 // 57. 8 dB 5 // 83.4 dB 12541 30303 // 6.7 dB 1590 // 32.3 dB 84 // 57. 9 dB 5 // 83.5 dB 12542 29956 // 6.8 dB 1572 // 32.4 dB 83 // 58. 0 dB 4 // 83.6 dB 12543 29613 // 6.9 dB 1554 // 32.5 dB 82 // 58. 1 dB 4 // 83.7 dB 12544 29274 // 7.0 dB 1536 // 32.6 dB 81 // 58. 2 dB 4 // 83.8 dB 12545 28939 // 7.1 dB 1519 // 32.7 dB 80 // 58. 3 dB 4 // 83.9 dB 12546 28608 // 7.2 dB 1501 // 32.8 dB 79 // 58. 4 dB 4 // 84.0 dB 12547 28280 // 7.3 dB 1484 // 32.9 dB 78 // 58. 5 dB 4 // 84.1 dB 12548 27956 // 7.4 dB 1467 // 33.0 dB 77 // 58. 6 dB 4 // 84.2 dB 12549 27636 // 7.5 dB 1450 // 33.1 dB 76 // 58. 7 dB 4 // 84.3 dB 12550 27320 // 7.6 dB 1434 // 33.2 dB 75 // 58. 8 dB 4 // 84.4 dB 12551 27007 // 7.7 dB 1417 // 33.3 dB 75 // 58. 9 dB 4 // 84.5 dB 12552 26698 // 7.8 dB 1401 // 33.4 dB 74 // 59. 0 dB 4 // 84.6 dB 12553 26393 // 7.9 dB 1385 // 33.5 dB 73 // 59. 1 dB 4 // 84.7 dB 12554 26090 // 8.0 dB 1369 // 33.6 dB 72 // 59. 2 dB 4 // 84.8 dB 12555 25792 // 8.1 dB 1354 // 33.7 dB 71 // 59. 3 dB 4 // 84.9 dB 12556 25497 // 8.2 dB 1338 // 33.8 dB 70 // 59. 4 dB 4 // 85.0 dB 12557 25205 // 8.3 dB 1323 // 33.9 dB 70 // 59. 5 dB 4 // 85.1 dB 12558 24916 // 8.4 dB 1308 // 34.0 dB 69 // 59. 6 dB 4 // 85.2 dB 12559 24631 // 8.5 dB 1293 // 34.1 dB 68 // 59. 7 dB 4 // 85.3 dB 12560 24349 // 8.6 dB 1278 // 34.2 dB 67 // 59. 8 dB 4 // 85.4 dB 12561 24070 // 8.7 dB 1263 // 34.3 dB 66 // 59. 9 dB 4 // 85.5 dB 12562 23795 // 8.8 dB 1249 // 34.4 dB 66 // 60. 0 dB 4 // 85.6 dB 12563 23522 // 8.9 dB 1235 // 34.5 dB 65 // 60. 1 dB 4 // 85.7 dB 12564 23253 // 9.0 dB 1220 // 34.6 dB 64 // 60. 2 dB 3 // 85.8 dB 12565 22987 // 9.1 dB 1206 // 34.7 dB 63 // 60. 3 dB 3 // 85.9 dB 12566 22724 // 9.2 dB 1193 // 34.8 dB 63 // 60. 4 dB 3 // 86.0 dB 12567 22464 // 9.3 dB 1179 // 34.9 dB 62 // 60. 5 dB 3 // 86.1 dB 12568 22207 // 9.4 dB 1166 // 35.0 dB 61 // 60. 6 dB 3 // 86.2 dB 12569 21952 // 9.5 dB 1152 // 35.1 dB 61 // 60. 7 dB 3 // 86.3 dB 12570 21701 // 9.6 dB 1139 // 35.2 dB 60 // 60. 8 dB 3 // 86.4 dB 12571 21453 // 9.7 dB 1126 // 35.3 dB 59 // 60. 9 dB 3 // 86.5 dB 12572 21207 // 9.8 dB 1113 // 35.4 dB 59 // 61. 0 dB 3 // 86.6 dB 12573 20964 // 9.9 dB 1100 // 35.5 dB 58 // 61. 1 dB 3 // 86.7 dB 12574 20724 // 10.0 dB 1088 // 35.6 dB 57 // 61 .2 dB 3 // 86.8 dB 12575 20487 // 10.1 dB 1075 // 35.7 dB 57 // 61 .3 dB 3 // 86.9 dB 12576 20253 // 10.2 dB 1063 // 35.8 dB 56 // 61 .4 dB 3 // 87.0 dB 12577 20021 // 10.3 dB 1051 // 35.9 dB 55 // 61 .5 dB 3 // 87.1 dB 12578 19792 // 10.4 dB 1039 // 36.0 dB 55 // 61 .6 dB 3 // 87.2 dB 12579 19565 // 10.5 dB 1027 // 36.1 dB 54 // 61 .7 dB 3 // 87.3 dB 12580 19341 // 10.6 dB 1015 // 36.2 dB 53 // 61 .8 dB 3 // 87.4 dB 12581 19120 // 10.7 dB 1004 // 36.3 dB 53 // 61 .9 dB 3 // 87.5 dB 12582 18901 // 10.8 dB 992 // 36.4 dB 52 // 62. 0 dB 3 // 87.6 dB 12583 18685 // 10.9 dB 981 // 36.5 dB 52 // 62. 1 dB 3 // 87.7 dB 12584 18471 // 11.0 dB 969 // 36.6 dB 51 // 62. 2 dB 3 // 87.8 dB 12585 18259 // 11.1 dB 958 // 36.7 dB 50 // 62. 3 dB 3 // 87.9 dB 12586 18050 // 11.2 dB 947 // 36.8 dB 50 // 62. 4 dB 3 // 88.0 dB 12587 17844 // 11.3 dB 937 // 36.9 dB 49 // 62. 5 dB 3 // 88.1 dB 12588 17639 // 11.4 dB 926 // 37.0 dB 49 // 62. 6 dB 3 // 88.2 dB 12589 17437 // 11.5 dB 915 // 37.1 dB 48 // 62. 7 dB 3 // 88.3 dB 12590 17238 // 11.6 dB 905 // 37.2 dB 48 // 62. 8 dB 3 // 88.4 dB 12591 17041 // 11.7 dB 894 // 37.3 dB 47 // 62. 9 dB 3 // 88.5 dB 12592 16845 // 11.8 dB 884 // 37.4 dB 47 // 63. 0 dB 3 // 88.6 dB 12593 16653 // 11.9 dB 874 // 37.5 dB 46 // 63. 1 dB 3 // 88.7 dB 12594 16462 // 12.0 dB 864 // 37.6 dB 45 // 63. 2 dB 3 // 88.8 dB 12595 16274 // 12.1 dB 854 // 37.7 dB 45 // 63. 3 dB 2 // 88.9 dB 12596 16087 // 12.2 dB 844 // 37.8 dB 44 // 63. 4 dB 2 // 89.0 dB 12597 15903 // 12.3 dB 835 // 37.9 dB 44 // 63. 5 dB 2 // 89.1 dB 12598 15721 // 12.4 dB 825 // 38.0 dB 43 // 63. 6 dB 2 // 89.2 dB 12599 15541 // 12.5 dB 816 // 38.1 dB 43 // 63. 7 dB 2 // 89.3 dB 12600 15363 // 12.6 dB 806 // 38.2 dB 42 // 63. 8 dB 2 // 89.4 dB 12601 15187 // 12.7 dB 797 // 38.3 dB 42 // 63. 9 dB 2 // 89.5 dB 12602 15014 // 12.8 dB 788 // 38.4 dB 41 // 64. 0 dB 2 // 89.6 dB 12603 14842 // 12.9 dB 779 // 38.5 dB 41 // 64. 1 dB 2 // 89.7 dB 12604 14672 // 13.0 dB 770 // 38.6 dB 41 // 64. 2 dB 2 // 89.8 dB 12605 14504 // 13.1 dB 761 // 38.7 dB 40 // 64. 3 dB 2 // 89.9 dB 12606 14338 // 13.2 dB 753 // 38.8 dB 40 // 64. 4 dB 2 // 90.0 dB 12607 14174 // 13.3 dB 744 // 38.9 dB 39 // 64. 5 dB 2 // 90.1 dB 12608 14011 // 13.4 dB 735 // 39.0 dB 39 // 64. 6 dB 2 // 90.2 dB 12609 13851 // 13.5 dB 727 // 39.1 dB 38 // 64. 7 dB 2 // 90.3 dB 12610 13693 // 13.6 dB 719 // 39.2 dB 38 // 64. 8 dB 2 // 90.4 dB 12611 13536 // 13.7 dB 710 // 39.3 dB 37 // 64. 9 dB 2 // 90.5 dB 12612 13381 // 13.8 dB 702 // 39.4 dB 37 // 65. 0 dB 2 // 90.6 dB 12613 13228 // 13.9 dB 694 // 39.5 dB 37 // 65. 1 dB 2 // 90.7 dB 12614

Supprimé : 4.0.1

Supprimé : 2







13076 // 14.0 dB 686 // 39.6 dB 36 // 65. 2 dB 2 // 90.8 dB 12615 12927 // 14.1 dB 679 // 39.7 dB 36 // 65. 3 dB 2 // 90.9 dB 12616 12779 // 14.2 dB 671 // 39.8 dB 35 // 65. 4 dB 2 // 91.0 dB 12617 12632 // 14.3 dB 663 // 39.9 dB 35 // 65. 5 dB 2 // 91.1 dB 12618 12488 // 14.4 dB 655 // 40.0 dB 35 // 65. 6 dB 2 // 91.2 dB 12619 12345 // 14.5 dB 648 // 40.1 dB 34 // 65. 7 dB 2 // 91.3 dB 12620 12203 // 14.6 dB 641 // 40.2 dB 34 // 65. 8 dB 2 // 91.4 dB 12621 12064 // 14.7 dB 633 // 40.3 dB 33 // 65. 9 dB 2 // 91.5 dB 12622 11926 // 14.8 dB 626 // 40.4 dB 33 // 66. 0 dB 2 // 91.6 dB 12623 11789 // 14.9 dB 619 // 40.5 dB 33 // 66. 1 dB 2 // 91.7 dB 12624 11654 // 15.0 dB 612 // 40.6 dB 32 // 66. 2 dB 2 // 91.8 dB 12625 11521 // 15.1 dB 605 // 40.7 dB 32 // 66. 3 dB 2 // 91.9 dB 12626 11389 // 15.2 dB 598 // 40.8 dB 31 // 66. 4 dB 2 // 92.0 dB 12627 11259 // 15.3 dB 591 // 40.9 dB 31 // 66. 5 dB 2 // 92.1 dB 12628 11130 // 15.4 dB 584 // 41.0 dB 31 // 66. 6 dB 2 // 92.2 dB 12629 11002 // 15.5 dB 578 // 41.1 dB 30 // 66. 7 dB 2 // 92.3 dB 12630 10876 // 15.6 dB 571 // 41.2 dB 30 // 66. 8 dB 2 // 92.4 dB 12631 10752 // 15.7 dB 564 // 41.3 dB 30 // 66. 9 dB 2 // 92.5 dB 12632 10629 // 15.8 dB 558 // 41.4 dB 29 // 67. 0 dB 2 // 92.6 dB 12633 10507 // 15.9 dB 552 // 41.5 dB 29 // 67. 1 dB 2 // 92.7 dB 12634 10387 // 16.0 dB 545 // 41.6 dB 29 // 67. 2 dB 2 // 92.8 dB 12635 10268 // 16.1 dB 539 // 41.7 dB 28 // 67. 3 dB 2 // 92.9 dB 12636 10150 // 16.2 dB 533 // 41.8 dB 28 // 67. 4 dB 2 // 93.0 dB 12637 10034 // 16.3 dB 527 // 41.9 dB 28 // 67. 5 dB 2 // 93.1 dB 12638 9919 // 16.4 dB 521 // 42.0 dB 27 // 67.6 dB 2 // 93.2 dB 12639 9806 // 16.5 dB 515 // 42.1 dB 27 // 67.7 dB 2 // 93.3 dB 12640 9694 // 16.6 dB 509 // 42.2 dB 27 // 67.8 dB 2 // 93.4 dB 12641 9583 // 16.7 dB 503 // 42.3 dB 27 // 67.9 dB 2 // 93.5 dB 12642 9473 // 16.8 dB 497 // 42.4 dB 26 // 68.0 dB 1 // 93.6 dB 12643 9365 // 16.9 dB 492 // 42.5 dB 26 // 68.1 dB 1 // 93.7 dB 12644 9257 // 17.0 dB 486 // 42.6 dB 26 // 68.2 dB 1 // 93.8 dB 12645 9151 // 17.1 dB 480 // 42.7 dB 25 // 68.3 dB 1 // 93.9 dB 12646 9047 // 17.2 dB 475 // 42.8 dB 25 // 68.4 dB 1 // 94.0 dB 12647 8943 // 17.3 dB 469 // 42.9 dB 25 // 68.5 dB 1 // 94.1 dB 12648 8841 // 17.4 dB 464 // 43.0 dB 24 // 68.6 dB 1 // 94.2 dB 12649 8739 // 17.5 dB 459 // 43.1 dB 24 // 68.7 dB 1 // 94.3 dB 12650 8639 // 17.6 dB 454 // 43.2 dB 24 // 68.8 dB 1 // 94.4 dB 12651 8541 // 17.7 dB 448 // 43.3 dB 24 // 68.9 dB 1 // 94.5 dB 12652 8443 // 17.8 dB 443 // 43.4 dB 23 // 69.0 dB 1 // 94.6 dB 12653 8346 // 17.9 dB 438 // 43.5 dB 23 // 69.1 dB 1 // 94.7 dB 12654 8251 // 18.0 dB 433 // 43.6 dB 23 // 69.2 dB 1 // 94.8 dB 12655 8156 // 18.1 dB 428 // 43.7 dB 23 // 69.3 dB 1 // 94.9 dB 12656 8063 // 18.2 dB 423 // 43.8 dB 22 // 69.4 dB 1 // 95.0 dB 12657 7971 // 18.3 dB 418 // 43.9 dB 22 // 69.5 dB 1 // 95.1 dB 12658 7879 // 18.4 dB 414 // 44.0 dB 22 // 69.6 dB 1 // 95.2 dB 12659 7789 // 18.5 dB 409 // 44.1 dB 22 // 69.7 dB 1 // 95.3 dB 12660 7700 // 18.6 dB 404 // 44.2 dB 21 // 69.8 dB 1 // 95.4 dB 12661 7612 // 18.7 dB 400 // 44.3 dB 21 // 69.9 dB 1 // 95.5 dB 12662 7525 // 18.8 dB 395 // 44.4 dB 21 // 70.0 dB 1 // 95.6 dB 12663 7439 // 18.9 dB 390 // 44.5 dB 21 // 70.1 dB 1 // 95.7 dB 12664 7353 // 19.0 dB 386 // 44.6 dB 20 // 70.2 dB 1 // 95.8 dB 12665 7269 // 19.1 dB 382 // 44.7 dB 20 // 70.3 dB 1 // 95.9 dB 12666 7186 // 19.2 dB 377 // 44.8 dB 20 // 70.4 dB 1 // 96.0 dB 12667 7104 // 19.3 dB 373 // 44.9 dB 20 // 70.5 dB 1 // 96.1 dB 12668 7022 // 19.4 dB 369 // 45.0 dB 19 // 70.6 dB 1 // 96.2 dB 12669 6942 // 19.5 dB 364 // 45.1 dB 19 // 70.7 dB 1 // 96.3 dB 12670 6863 // 19.6 dB 360 // 45.2 dB 19 // 70.8 dB 1 // 96.4 dB 12671 6784 // 19.7 dB 356 // 45.3 dB 19 // 70.9 dB 1 // 96.5 dB 12672 6706 // 19.8 dB 352 // 45.4 dB 19 // 71.0 dB 1 // 96.6 dB 12673 6630 // 19.9 dB 348 // 45.5 dB 18 // 71.1 dB 1 // 96.7 dB 12674 6554 // 20.0 dB 344 // 45.6 dB 18 // 71.2 dB 1 // 96.8 dB 12675 6479 // 20.1 dB 340 // 45.7 dB 18 // 71.3 dB 1 // 96.9 dB 12676 6405 // 20.2 dB 336 // 45.8 dB 18 // 71.4 dB 1 // 97.0 dB 12677 6331 // 20.3 dB 332 // 45.9 dB 18 // 71.5 dB 1 // 97.1 dB 12678 6259 // 20.4 dB 329 // 46.0 dB 17 // 71.6 dB 1 // 97.2 dB 12679 6187 // 20.5 dB 325 // 46.1 dB 17 // 71.7 dB 1 // 97.3 dB 12680 6116 // 20.6 dB 321 // 46.2 dB 17 // 71.8 dB 1 // 97.4 dB 12681 6046 // 20.7 dB 317 // 46.3 dB 17 // 71.9 dB 1 // 97.5 dB 12682 5977 // 20.8 dB 314 // 46.4 dB 17 // 72.0 dB 1 // 97.6 dB 12683 5909 // 20.9 dB 310 // 46.5 dB 16 // 72.1 dB 1 // 97.7 dB 12684 5841 // 21.0 dB 307 // 46.6 dB 16 // 72.2 dB 1 // 97.8 dB 12685 5774 // 21.1 dB 303 // 46.7 dB 16 // 72.3 dB 1 // 97.9 dB 12686 5708 // 21.2 dB 300 // 46.8 dB 16 // 72.4 dB 1 // 98.0 dB 12687 5643 // 21.3 dB 296 // 46.9 dB 16 // 72.5 dB 1 // 98.1 dB 12688 5578 // 21.4 dB 293 // 47.0 dB 15 // 72.6 dB 1 // 98.2 dB 12689 5514 // 21.5 dB 290 // 47.1 dB 15 // 72.7 dB 1 // 98.3 dB 12690

Supprimé : 4.0.1

Supprimé : 2







5451 // 21.6 dB 286 // 47.2 dB 15 // 72.8 dB 1 // 98.4 dB 12691 5389 // 21.7 dB 283 // 47.3 dB 15 // 72.9 dB 1 // 98.5 dB 12692 5327 // 21.8 dB 280 // 47.4 dB 15 // 73.0 dB 1 // 98.6 dB 12693 5266 // 21.9 dB 276 // 47.5 dB 15 // 73.1 dB 1 // 98.7 dB 12694 5206 // 22.0 dB 273 // 47.6 dB 14 // 73.2 dB 1 // 98.8 dB 12695 5146 // 22.1 dB 270 // 47.7 dB 14 // 73.3 dB 1 // 98.9 dB 12696 5087 // 22.2 dB 267 // 47.8 dB 14 // 73.4 dB 1 // 99.0 dB 12697 5029 // 22.3 dB 264 // 47.9 dB 14 // 73.5 dB 1 // 99.1 dB 12698 4972 // 22.4 dB 261 // 48.0 dB 14 // 73.6 dB 1 // 99.2 dB 12699 4915 // 22.5 dB 258 // 48.1 dB 14 // 73.7 dB 1 // 99.3 dB 12700 4858 // 22.6 dB 255 // 48.2 dB 14 // 73.8 dB 1 // 99.4 dB 12701 4803 // 22.7 dB 252 // 48.3 dB 13 // 73.9 dB 1 // 99.5 dB 12702 4748 // 22.8 dB 249 // 48.4 dB 13 // 74.0 dB 1 // 99.6 dB 12703 4693 // 22.9 dB 246 // 48.5 dB 13 // 74.1 dB 1 // 99.7 dB 12704 4640 // 23.0 dB 244 // 48.6 dB 13 // 74.2 dB 1 // 99.8 dB 12705 4587 // 23.1 dB 241 // 48.7 dB 13 // 74.3 dB 1 // 99.9 dB 12706 4534 // 23.2 dB 238 // 48.8 dB 13 // 74.4 dB 1 // 100.0 dB 12707 4482 // 23.3 dB 235 // 48.9 dB 12 // 74.5 dB 1 // 100.1 dB 12708 4431 // 23.4 dB 233 // 49.0 dB 12 // 74.6 dB 1 // 100.2 dB 12709 4380 // 23.5 dB 230 // 49.1 dB 12 // 74.7 dB 1 // 100.3 dB 12710 4330 // 23.6 dB 227 // 49.2 dB 12 // 74.8 dB 1 // 100.4 dB 12711 4280 // 23.7 dB 225 // 49.3 dB 12 // 74.9 dB 1 // 100.5 dB 12712 4231 // 23.8 dB 222 // 49.4 dB 12 // 75.0 dB 1 // 100.6 dB 12713 4183 // 23.9 dB 220 // 49.5 dB 12 // 75.1 dB 1 // 100.7 dB 12714 4135 // 24.0 dB 217 // 49.6 dB 12 // 75.2 dB 1 // 100.8 dB 12715 4088 // 24.1 dB 215 // 49.7 dB 11 // 75.3 dB 1 // 100.9 dB 12716 4041 // 24.2 dB 212 // 49.8 dB 11 // 75.4 dB 1 // 101.0 dB 12717 3995 // 24.3 dB 210 // 49.9 dB 11 // 75.5 dB 1 // 101.1 dB 12718 3949 // 24.4 dB 207 // 50.0 dB 11 // 75.6 dB 1 // 101.2 dB 12719 3904 // 24.5 dB 205 // 50.1 dB 11 // 75.7 dB 1 // 101.3 dB 12720 3859 // 24.6 dB 203 // 50.2 dB 11 // 75.8 dB 1 // 101.4 dB 12721 3815 // 24.7 dB 200 // 50.3 dB 11 // 75.9 dB 1 // 101.5 dB 12722 3771 // 24.8 dB 198 // 50.4 dB 11 // 76.0 dB 1 // 101.6 dB 12723 3728 // 24.9 dB 196 // 50.5 dB 10 // 76.1 dB 1 // 101.7 dB 12724 3685 // 25.0 dB 194 // 50.6 dB 10 // 76.2 dB 1 // 101.8 dB 12725 3643 // 25.1 dB 191 // 50.7 dB 10 // 76.3 dB 1 // 101.9 dB 12726 3602 // 25.2 dB 189 // 50.8 dB 10 // 76.4 dB 1 // 102.0 dB 12727 3560 // 25.3 dB 187 // 50.9 dB 10 // 76.5 dB 1 // 102.1 dB 12728 3520 // 25.4 dB 185 // 51.0 dB 10 // 76.6 dB 1 // 102.2 dB 12729 3479 // 25.5 dB 183 // 51.1 dB 10 // 76.7 dB 1 // 102.3 dB 12730 12731 12732

Annex 12. LUT for 7.5kHz compensation 12733

20MHz bandwidth (used also for 1.4, 3, 5 & 10MHz) 12734 k=0, cos=16383, sin=0 //k=1, cos=16383, sin=-25 //k=2, cos=16383, sin=-50 //k=3, cos=16383, sin=-75 //k=4, 12735 cos=16383, sin=-101 //k=5, cos=16383, sin=-126 //k=6, cos=16382, sin=-151 //k=7, cos=16382, sin=-176 //k=8, 12736 cos=16382, sin=-201 //k=9, cos=16381, sin=-226 //k=10, cos=16381, sin=-251 //k=11, cos=16381, sin=-276 //k=12, 12737 cos=16380, sin=-302 //k=13, cos=16380, sin=-327 //k=14, cos=16379, sin=-352 //k=15, cos=16379, sin=-377 //k=16, 12738 cos=16378, sin=-402 //k=17, cos=16377, sin=-427 //k=18, cos=16377, sin=-452 //k=19, cos=16376, sin=-477 //k=20, 12739 cos=16375, sin=-503 //k=21, cos=16375, sin=-528 //k=22, cos=16374, sin=-553 //k=23, cos=16373, sin=-578 //k=24, 12740 cos=16372, sin=-603 //k=25, cos=16371, sin=-628 //k=26, cos=16370, sin=-653 //k=27, cos=16369, sin=-678 //k=28, 12741 cos=16368, sin=-704 //k=29, cos=16367, sin=-729 //k=30, cos=16366, sin=-754 //k=31, cos=16364, sin=-779 //k=32, 12742 cos=16363, sin=-804 //k=33, cos=16362, sin=-829 //k=34, cos=16361, sin=-854 //k=35, cos=16359, sin=-879 //k=36, 12743 cos=16358, sin=-904 //k=37, cos=16357, sin=-929 //k=38, cos=16355, sin=-955 //k=39, cos=16354, sin=-980 //k=40, 12744 cos=16352, sin=-1005 //k=41, cos=16351, sin=-1030 //k=42, cos=16349, sin=-1055 //k=43, cos=16347, sin=-1080 12745 //k=44, cos=16346, sin=-1105 //k=45, cos=16344, sin=-1130 //k=46, cos=16342, sin=-1155 //k=47, cos=16340, sin=-12746 1180 //k=48, cos=16339, sin=-1205 //k=49, cos=16337, sin=-1230 //k=50, cos=16335, sin=-1255 //k=51, cos=16333, 12747 sin=-1280 //k=52, cos=16331, sin=-1306 //k=53, cos=16329, sin=-1331 //k=54, cos=16327, sin=-1356 //k=55, 12748 cos=16325, sin=-1381 //k=56, cos=16323, sin=-1406 //k=57, cos=16320, sin=-1431 //k=58, cos=16318, sin=-1456 12749 //k=59, cos=16316, sin=-1481 //k=60, cos=16314, sin=-1506 //k=61, cos=16311, sin=-1531 //k=62, cos=16309, sin=-12750 1556 //k=63, cos=16307, sin=-1581 //k=64, cos=16304, sin=-1606 //k=65, cos=16302, sin=-1631 //k=66, cos=16299, 12751 sin=-1656 //k=67, cos=16297, sin=-1681 //k=68, cos=16294, sin=-1706 //k=69, cos=16291, sin=-1731 //k=70, 12752 cos=16289, sin=-1756 //k=71, cos=16286, sin=-1781 //k=72, cos=16283, sin=-1806 //k=73, cos=16280, sin=-1831 12753 //k=74, cos=16278, sin=-1856 //k=75, cos=16275, sin=-1881 //k=76, cos=16272, sin=-1906 //k=77, cos=16269, sin=-12754 1931 //k=78, cos=16266, sin=-1956 //k=79, cos=16263, sin=-1981 //k=80, cos=16260, sin=-2006 //k=81, cos=16257, 12755 sin=-2031 //k=82, cos=16254, sin=-2055 //k=83, cos=16250, sin=-2080 //k=84, cos=16247, sin=-2105 //k=85, 12756

Supprimé : 4.0.1

Supprimé : 2







cos=16244, sin=-2130 //k=86, cos=16241, sin=-2155 //k=87, cos=16237, sin=-2180 //k=88, cos=16234, sin=-2205 12757 //k=89, cos=16231, sin=-2230 //k=90, cos=16227, sin=-2255 //k=91, cos=16224, sin=-2280 //k=92, cos=16220, sin=-12758 2305 //k=93, cos=16217, sin=-2329 //k=94, cos=16213, sin=-2354 //k=95, cos=16209, sin=-2379 //k=96, cos=16206, 12759 sin=-2404 //k=97, cos=16202, sin=-2429 //k=98, cos=16198, sin=-2454 //k=99, cos=16194, sin=-2479 //k=100, 12760 cos=16191, sin=-2503 //k=101, cos=16187, sin=-2528 //k=102, cos=16183, sin=-2553 //k=103, cos=16179, sin=-2578 12761 //k=104, cos=16175, sin=-2603 //k=105, cos=16171, sin=-2628 //k=106, cos=16167, sin=-2652 //k=107, cos=16163, 12762 sin=-2677 //k=108, cos=16159, sin=-2702 //k=109, cos=16155, sin=-2727 //k=110, cos=16150, sin=-2752 //k=111, 12763 cos=16146, sin=-2776 //k=112, cos=16142, sin=-2801 //k=113, cos=16137, sin=-2826 //k=114, cos=16133, sin=-2851 12764 //k=115, cos=16129, sin=-2875 //k=116, cos=16124, sin=-2900 //k=117, cos=16120, sin=-2925 //k=118, cos=16115, 12765 sin=-2949 //k=119, cos=16111, sin=-2974 //k=120, cos=16106, sin=-2999 //k=121, cos=16102, sin=-3024 //k=122, 12766 cos=16097, sin=-3048 //k=123, cos=16092, sin=-3073 //k=124, cos=16088, sin=-3098 //k=125, cos=16083, sin=-3122 12767 //k=126, cos=16078, sin=-3147 //k=127, cos=16073, sin=-3172 //k=128, cos=16068, sin=-3196 //k=129, cos=16063, 12768 sin=-3221 //k=130, cos=16058, sin=-3246 //k=131, cos=16053, sin=-3270 //k=132, cos=16048, sin=-3295 //k=133, 12769 cos=16043, sin=-3320 //k=134, cos=16038, sin=-3344 //k=135, cos=16033, sin=-3369 //k=136, cos=16028, sin=-3393 12770 //k=137, cos=16023, sin=-3418 //k=138, cos=16017, sin=-3442 //k=139, cos=16012, sin=-3467 //k=140, cos=16007, 12771 sin=-3492 //k=141, cos=16001, sin=-3516 //k=142, cos=15996, sin=-3541 //k=143, cos=15990, sin=-3565 //k=144, 12772 cos=15985, sin=-3590 //k=145, cos=15979, sin=-3614 //k=146, cos=15974, sin=-3639 //k=147, cos=15968, sin=-3663 12773 //k=148, cos=15963, sin=-3688 //k=149, cos=15957, sin=-3712 //k=150, cos=15951, sin=-3737 //k=151, cos=15945, 12774 sin=-3761 //k=152, cos=15940, sin=-3786 //k=153, cos=15934, sin=-3810 //k=154, cos=15928, sin=-3835 //k=155, 12775 cos=15922, sin=-3859 //k=156, cos=15916, sin=-3883 //k=157, cos=15910, sin=-3908 //k=158, cos=15904, sin=-3932 12776 //k=159, cos=15898, sin=-3957 //k=160, cos=15892, sin=-3981 //k=161, cos=15886, sin=-4005 //k=162, cos=15880, 12777 sin=-4030 //k=163, cos=15874, sin=-4054 //k=164, cos=15867, sin=-4078 //k=165, cos=15861, sin=-4103 //k=166, 12778 cos=15855, sin=-4127 //k=167, cos=15848, sin=-4151 //k=168, cos=15842, sin=-4176 //k=169, cos=15836, sin=-4200 12779 //k=170, cos=15829, sin=-4224 //k=171, cos=15823, sin=-4249 //k=172, cos=15816, sin=-4273 //k=173, cos=15809, 12780 sin=-4297 //k=174, cos=15803, sin=-4321 //k=175, cos=15796, sin=-4346 //k=176, cos=15790, sin=-4370 //k=177, 12781 cos=15783, sin=-4394 //k=178, cos=15776, sin=-4418 //k=179, cos=15769, sin=-4442 //k=180, cos=15762, sin=-4467 12782 //k=181, cos=15756, sin=-4491 //k=182, cos=15749, sin=-4515 //k=183, cos=15742, sin=-4539 //k=184, cos=15735, 12783 sin=-4563 //k=185, cos=15728, sin=-4587 //k=186, cos=15721, sin=-4612 //k=187, cos=15714, sin=-4636 //k=188, 12784 cos=15706, sin=-4660 //k=189, cos=15699, sin=-4684 //k=190, cos=15692, sin=-4708 //k=191, cos=15685, sin=-4732 12785 //k=192, cos=15678, sin=-4756 //k=193, cos=15670, sin=-4780 //k=194, cos=15663, sin=-4804 //k=195, cos=15655, 12786 sin=-4828 //k=196, cos=15648, sin=-4852 //k=197, cos=15641, sin=-4876 //k=198, cos=15633, sin=-4900 //k=199, 12787 cos=15626, sin=-4924 //k=200, cos=15618, sin=-4948 //k=201, cos=15610, sin=-4972 //k=202, cos=15603, sin=-4996 12788 //k=203, cos=15595, sin=-5020 //k=204, cos=15587, sin=-5044 //k=205, cos=15580, sin=-5068 //k=206, cos=15572, 12789 sin=-5092 //k=207, cos=15564, sin=-5115 //k=208, cos=15556, sin=-5139 //k=209, cos=15548, sin=-5163 //k=210, 12790 cos=15540, sin=-5187 //k=211, cos=15532, sin=-5211 //k=212, cos=15524, sin=-5235 //k=213, cos=15516, sin=-5259 12791 //k=214, cos=15508, sin=-5282 //k=215, cos=15500, sin=-5306 //k=216, cos=15492, sin=-5330 //k=217, cos=15484, 12792 sin=-5354 //k=218, cos=15475, sin=-5377 //k=219, cos=15467, sin=-5401 //k=220, cos=15459, sin=-5425 //k=221, 12793 cos=15451, sin=-5449 //k=222, cos=15442, sin=-5472 //k=223, cos=15434, sin=-5496 //k=224, cos=15425, sin=-5520 12794 //k=225, cos=15417, sin=-5543 //k=226, cos=15408, sin=-5567 //k=227, cos=15400, sin=-5591 //k=228, cos=15391, 12795 sin=-5614 //k=229, cos=15383, sin=-5638 //k=230, cos=15374, sin=-5661 //k=231, cos=15365, sin=-5685 //k=232, 12796 cos=15356, sin=-5708 //k=233, cos=15348, sin=-5732 //k=234, cos=15339, sin=-5756 //k=235, cos=15330, sin=-5779 12797 //k=236, cos=15321, sin=-5803 //k=237, cos=15312, sin=-5826 //k=238, cos=15303, sin=-5850 //k=239, cos=15294, 12798 sin=-5873 //k=240, cos=15285, sin=-5897 //k=241, cos=15276, sin=-5920 //k=242, cos=15267, sin=-5943 //k=243, 12799 cos=15258, sin=-5967 //k=244, cos=15249, sin=-5990 //k=245, cos=15240, sin=-6014 //k=246, cos=15230, sin=-6037 12800 //k=247, cos=15221, sin=-6060 //k=248, cos=15212, sin=-6084 //k=249, cos=15202, sin=-6107 //k=250, cos=15193, 12801 sin=-6130 //k=251, cos=15184, sin=-6154 //k=252, cos=15174, sin=-6177 //k=253, cos=15165, sin=-6200 //k=254, 12802 cos=15155, sin=-6223 //k=255, cos=15145, sin=-6247 //k=256, cos=15136, sin=-6270 //k=257, cos=15126, sin=-6293 12803 //k=258, cos=15117, sin=-6316 //k=259, cos=15107, sin=-6339 //k=260, cos=15097, sin=-6363 //k=261, cos=15087, 12804 sin=-6386 //k=262, cos=15078, sin=-6409 //k=263, cos=15068, sin=-6432 //k=264, cos=15058, sin=-6455 //k=265, 12805 cos=15048, sin=-6478 //k=266, cos=15038, sin=-6501 //k=267, cos=15028, sin=-6524 //k=268, cos=15018, sin=-6547 12806 //k=269, cos=15008, sin=-6570 //k=270, cos=14998, sin=-6593 //k=271, cos=14988, sin=-6616 //k=272, cos=14977, 12807 sin=-6639 //k=273, cos=14967, sin=-6662 //k=274, cos=14957, sin=-6685 //k=275, cos=14947, sin=-6708 //k=276, 12808 cos=14936, sin=-6731 //k=277, cos=14926, sin=-6754 //k=278, cos=14916, sin=-6777 //k=279, cos=14905, sin=-6800 12809 //k=280, cos=14895, sin=-6823 //k=281, cos=14884, sin=-6846 //k=282, cos=14874, sin=-6868 //k=283, cos=14863, 12810 sin=-6891 //k=284, cos=14853, sin=-6914 //k=285, cos=14842, sin=-6937 //k=286, cos=14831, sin=-6960 //k=287, 12811 cos=14821, sin=-6982 //k=288, cos=14810, sin=-7005 //k=289, cos=14799, sin=-7028 //k=290, cos=14788, sin=-7050 12812 //k=291, cos=14778, sin=-7073 //k=292, cos=14767, sin=-7096 //k=293, cos=14756, sin=-7118 //k=294, cos=14745, 12813 sin=-7141 //k=295, cos=14734, sin=-7164 //k=296, cos=14723, sin=-7186 //k=297, cos=14712, sin=-7209 //k=298, 12814 cos=14701, sin=-7231 //k=299, cos=14690, sin=-7254 //k=300, cos=14679, sin=-7276 //k=301, cos=14667, sin=-7299 12815 //k=302, cos=14656, sin=-7321 //k=303, cos=14645, sin=-7344 //k=304, cos=14634, sin=-7366 //k=305, cos=14622, 12816

Supprimé : 4.0.1

Supprimé : 2







sin=-7389 //k=306, cos=14611, sin=-7411 //k=307, cos=14600, sin=-7434 //k=308, cos=14588, sin=-7456 //k=309, 12817 cos=14577, sin=-7478 //k=310, cos=14565, sin=-7501 //k=311, cos=14554, sin=-7523 //k=312, cos=14542, sin=-7545 12818 //k=313, cos=14531, sin=-7568 //k=314, cos=14519, sin=-7590 //k=315, cos=14507, sin=-7612 //k=316, cos=14496, 12819 sin=-7635 //k=317, cos=14484, sin=-7657 //k=318, cos=14472, sin=-7679 //k=319, cos=14460, sin=-7701 //k=320, 12820 cos=14448, sin=-7723 //k=321, cos=14437, sin=-7746 //k=322, cos=14425, sin=-7768 //k=323, cos=14413, sin=-7790 12821 //k=324, cos=14401, sin=-7812 //k=325, cos=14389, sin=-7834 //k=326, cos=14377, sin=-7856 //k=327, cos=14365, 12822 sin=-7878 //k=328, cos=14353, sin=-7900 //k=329, cos=14340, sin=-7922 //k=330, cos=14328, sin=-7944 //k=331, 12823 cos=14316, sin=-7966 //k=332, cos=14304, sin=-7988 //k=333, cos=14292, sin=-8010 //k=334, cos=14279, sin=-8032 12824 //k=335, cos=14267, sin=-8054 //k=336, cos=14255, sin=-8076 //k=337, cos=14242, sin=-8098 //k=338, cos=14230, 12825 sin=-8119 //k=339, cos=14217, sin=-8141 //k=340, cos=14205, sin=-8163 //k=341, cos=14192, sin=-8185 //k=342, 12826 cos=14180, sin=-8207 //k=343, cos=14167, sin=-8228 //k=344, cos=14154, sin=-8250 //k=345, cos=14142, sin=-8272 12827 //k=346, cos=14129, sin=-8293 //k=347, cos=14116, sin=-8315 //k=348, cos=14104, sin=-8337 //k=349, cos=14091, 12828 sin=-8358 //k=350, cos=14078, sin=-8380 //k=351, cos=14065, sin=-8401 //k=352, cos=14052, sin=-8423 //k=353, 12829 cos=14039, sin=-8445 //k=354, cos=14026, sin=-8466 //k=355, cos=14013, sin=-8488 //k=356, cos=14000, sin=-8509 12830 //k=357, cos=13987, sin=-8531 //k=358, cos=13974, sin=-8552 //k=359, cos=13961, sin=-8573 //k=360, cos=13948, 12831 sin=-8595 //k=361, cos=13934, sin=-8616 //k=362, cos=13921, sin=-8638 //k=363, cos=13908, sin=-8659 //k=364, 12832 cos=13895, sin=-8680 //k=365, cos=13881, sin=-8702 //k=366, cos=13868, sin=-8723 //k=367, cos=13855, sin=-8744 12833 //k=368, cos=13841, sin=-8765 //k=369, cos=13828, sin=-8787 //k=370, cos=13814, sin=-8808 //k=371, cos=13801, 12834 sin=-8829 //k=372, cos=13787, sin=-8850 //k=373, cos=13774, sin=-8871 //k=374, cos=13760, sin=-8892 //k=375, 12835 cos=13746, sin=-8914 //k=376, cos=13733, sin=-8935 //k=377, cos=13719, sin=-8956 //k=378, cos=13705, sin=-8977 12836 //k=379, cos=13691, sin=-8998 //k=380, cos=13677, sin=-9019 //k=381, cos=13664, sin=-9040 //k=382, cos=13650, 12837 sin=-9061 //k=383, cos=13636, sin=-9082 //k=384, cos=13622, sin=-9102 //k=385, cos=13608, sin=-9123 //k=386, 12838 cos=13594, sin=-9144 //k=387, cos=13580, sin=-9165 //k=388, cos=13566, sin=-9186 //k=389, cos=13552, sin=-9207 12839 //k=390, cos=13538, sin=-9227 //k=391, cos=13523, sin=-9248 //k=392, cos=13509, sin=-9269 //k=393, cos=13495, 12840 sin=-9290 //k=394, cos=13481, sin=-9310 //k=395, cos=13466, sin=-9331 //k=396, cos=13452, sin=-9352 //k=397, 12841 cos=13438, sin=-9372 //k=398, cos=13423, sin=-9393 //k=399, cos=13409, sin=-9413 //k=400, cos=13394, sin=-9434 12842 //k=401, cos=13380, sin=-9455 //k=402, cos=13365, sin=-9475 //k=403, cos=13351, sin=-9496 //k=404, cos=13336, 12843 sin=-9516 //k=405, cos=13322, sin=-9537 //k=406, cos=13307, sin=-9557 //k=407, cos=13292, sin=-9577 //k=408, 12844 cos=13278, sin=-9598 //k=409, cos=13263, sin=-9618 //k=410, cos=13248, sin=-9638 //k=411, cos=13233, sin=-9659 12845 //k=412, cos=13218, sin=-9679 //k=413, cos=13204, sin=-9699 //k=414, cos=13189, sin=-9720 //k=415, cos=13174, 12846 sin=-9740 //k=416, cos=13159, sin=-9760 //k=417, cos=13144, sin=-9780 //k=418, cos=13129, sin=-9800 //k=419, 12847 cos=13114, sin=-9820 //k=420, cos=13099, sin=-9841 //k=421, cos=13084, sin=-9861 //k=422, cos=13068, sin=-9881 12848 //k=423, cos=13053, sin=-9901 //k=424, cos=13038, sin=-9921 //k=425, cos=13023, sin=-9941 //k=426, cos=13008, 12849 sin=-9961 //k=427, cos=12992, sin=-9981 //k=428, cos=12977, sin=-10001 //k=429, cos=12962, sin=-10020 //k=430, 12850 cos=12946, sin=-10040 //k=431, cos=12931, sin=-10060 //k=432, cos=12915, sin=-10080 //k=433, cos=12900, sin=-12851 10100 //k=434, cos=12884, sin=-10120 //k=435, cos=12869, sin=-10139 //k=436, cos=12853, sin=-10159 //k=437, 12852 cos=12838, sin=-10179 //k=438, cos=12822, sin=-10198 //k=439, cos=12806, sin=-10218 //k=440, cos=12791, sin=-12853 10238 //k=441, cos=12775, sin=-10257 //k=442, cos=12759, sin=-10277 //k=443, cos=12743, sin=-10296 //k=444, 12854 cos=12728, sin=-10316 //k=445, cos=12712, sin=-10336 //k=446, cos=12696, sin=-10355 //k=447, cos=12680, sin=-12855 10374 //k=448, cos=12664, sin=-10394 //k=449, cos=12648, sin=-10413 //k=450, cos=12632, sin=-10433 //k=451, 12856 cos=12616, sin=-10452 //k=452, cos=12600, sin=-10471 //k=453, cos=12584, sin=-10491 //k=454, cos=12568, sin=-12857 10510 //k=455, cos=12552, sin=-10529 //k=456, cos=12536, sin=-10549 //k=457, cos=12519, sin=-10568 //k=458, 12858 cos=12503, sin=-10587 //k=459, cos=12487, sin=-10606 //k=460, cos=12471, sin=-10625 //k=461, cos=12454, sin=-12859 10644 //k=462, cos=12438, sin=-10663 //k=463, cos=12422, sin=-10683 //k=464, cos=12405, sin=-10702 //k=465, 12860 cos=12389, sin=-10721 //k=466, cos=12372, sin=-10740 //k=467, cos=12356, sin=-10759 //k=468, cos=12339, sin=-12861 10778 //k=469, cos=12323, sin=-10796 //k=470, cos=12306, sin=-10815 //k=471, cos=12290, sin=-10834 //k=472, 12862 cos=12273, sin=-10853 //k=473, cos=12256, sin=-10872 //k=474, cos=12240, sin=-10891 //k=475, cos=12223, sin=-12863 10909 //k=476, cos=12206, sin=-10928 //k=477, cos=12189, sin=-10947 //k=478, cos=12173, sin=-10966 //k=479, 12864 cos=12156, sin=-10984 //k=480, cos=12139, sin=-11003 //k=481, cos=12122, sin=-11021 //k=482, cos=12105, sin=-12865 11040 //k=483, cos=12088, sin=-11059 //k=484, cos=12071, sin=-11077 //k=485, cos=12054, sin=-11096 //k=486, 12866 cos=12037, sin=-11114 //k=487, cos=12020, sin=-11133 //k=488, cos=12003, sin=-11151 //k=489, cos=11986, sin=-12867 11169 //k=490, cos=11969, sin=-11188 //k=491, cos=11951, sin=-11206 //k=492, cos=11934, sin=-11224 //k=493, 12868 cos=11917, sin=-11243 //k=494, cos=11900, sin=-11261 //k=495, cos=11883, sin=-11279 //k=496, cos=11865, sin=-12869 11297 //k=497, cos=11848, sin=-11316 //k=498, cos=11830, sin=-11334 //k=499, cos=11813, sin=-11352 //k=500, 12870 cos=11796, sin=-11370 //k=501, cos=11778, sin=-11388 //k=502, cos=11761, sin=-11406 //k=503, cos=11743, sin=-12871 11424 //k=504, cos=11726, sin=-11442 //k=505, cos=11708, sin=-11460 //k=506, cos=11691, sin=-11478 //k=507, 12872 cos=11673, sin=-11496 //k=508, cos=11655, sin=-11514 //k=509, cos=11638, sin=-11532 //k=510, cos=11620, sin=-12873 11550 //k=511, cos=11602, sin=-11567 //k=512, cos=11584, sin=-11585 //k=513, cos=11567, sin=-11603 //k=514, 12874 cos=11549, sin=-11621 //k=515, cos=11531, sin=-11638 //k=516, cos=11513, sin=-11656 //k=517, cos=11495, sin=-12875 11674 //k=518, cos=11477, sin=-11691 //k=519, cos=11459, sin=-11709 //k=520, cos=11441, sin=-11727 //k=521, 12876

Supprimé : 4.0.1

Supprimé : 2







cos=11423, sin=-11744 //k=522, cos=11405, sin=-11762 //k=523, cos=11387, sin=-11779 //k=524, cos=11369, sin=-12877 11797 //k=525, cos=11351, sin=-11814 //k=526, cos=11333, sin=-11831 //k=527, cos=11315, sin=-11849 //k=528, 12878 cos=11297, sin=-11866 //k=529, cos=11278, sin=-11883 //k=530, cos=11260, sin=-11901 //k=531, cos=11242, sin=-12879 11918 //k=532, cos=11224, sin=-11935 //k=533, cos=11205, sin=-11952 //k=534, cos=11187, sin=-11970 //k=535, 12880 cos=11169, sin=-11987 //k=536, cos=11150, sin=-12004 //k=537, cos=11132, sin=-12021 //k=538, cos=11113, sin=-12881 12038 //k=539, cos=11095, sin=-12055 //k=540, cos=11076, sin=-12072 //k=541, cos=11058, sin=-12089 //k=542, 12882 cos=11039, sin=-12106 //k=543, cos=11021, sin=-12123 //k=544, cos=11002, sin=-12140 //k=545, cos=10983, sin=-12883 12157 //k=546, cos=10965, sin=-12173 //k=547, cos=10946, sin=-12190 //k=548, cos=10927, sin=-12207 //k=549, 12884 cos=10909, sin=-12224 //k=550, cos=10890, sin=-12240 //k=551, cos=10871, sin=-12257 //k=552, cos=10852, sin=-12885 12274 //k=553, cos=10833, sin=-12290 //k=554, cos=10814, sin=-12307 //k=555, cos=10796, sin=-12324 //k=556, 12886 cos=10777, sin=-12340 //k=557, cos=10758, sin=-12357 //k=558, cos=10739, sin=-12373 //k=559, cos=10720, sin=-12887 12390 //k=560, cos=10701, sin=-12406 //k=561, cos=10682, sin=-12423 //k=562, cos=10663, sin=-12439 //k=563, 12888 cos=10644, sin=-12455 //k=564, cos=10624, sin=-12472 //k=565, cos=10605, sin=-12488 //k=566, cos=10586, sin=-12889 12504 //k=567, cos=10567, sin=-12520 //k=568, cos=10548, sin=-12537 //k=569, cos=10528, sin=-12553 //k=570, 12890 cos=10509, sin=-12569 //k=571, cos=10490, sin=-12585 //k=572, cos=10471, sin=-12601 //k=573, cos=10451, sin=-12891 12617 //k=574, cos=10432, sin=-12633 //k=575, cos=10412, sin=-12649 //k=576, cos=10393, sin=-12665 //k=577, 12892 cos=10374, sin=-12681 //k=578, cos=10354, sin=-12697 //k=579, cos=10335, sin=-12713 //k=580, cos=10315, sin=-12893 12729 //k=581, cos=10296, sin=-12744 //k=582, cos=10276, sin=-12760 //k=583, cos=10256, sin=-12776 //k=584, 12894 cos=10237, sin=-12792 //k=585, cos=10217, sin=-12807 //k=586, cos=10198, sin=-12823 //k=587, cos=10178, sin=-12895 12839 //k=588, cos=10158, sin=-12854 //k=589, cos=10138, sin=-12870 //k=590, cos=10119, sin=-12885 //k=591, 12896 cos=10099, sin=-12901 //k=592, cos=10079, sin=-12916 //k=593, cos=10059, sin=-12932 //k=594, cos=10039, sin=-12897 12947 //k=595, cos=10020, sin=-12963 //k=596, cos=10000, sin=-12978 //k=597, cos=9980, sin=-12993 //k=598, 12898 cos=9960, sin=-13008 //k=599, cos=9940, sin=-13024 //k=600, cos=9920, sin=-13039 //k=601, cos=9900, sin=-13054 12899 //k=602, cos=9880, sin=-13069 //k=603, cos=9860, sin=-13085 //k=604, cos=9840, sin=-13100 //k=605, cos=9820, 12900 sin=-13115 //k=606, cos=9799, sin=-13130 //k=607, cos=9779, sin=-13145 //k=608, cos=9759, sin=-13160 //k=609, 12901 cos=9739, sin=-13175 //k=610, cos=9719, sin=-13190 //k=611, cos=9698, sin=-13205 //k=612, cos=9678, sin=-13219 12902 //k=613, cos=9658, sin=-13234 //k=614, cos=9638, sin=-13249 //k=615, cos=9617, sin=-13264 //k=616, cos=9597, 12903 sin=-13279 //k=617, cos=9577, sin=-13293 //k=618, cos=9556, sin=-13308 //k=619, cos=9536, sin=-13323 //k=620, 12904 cos=9515, sin=-13337 //k=621, cos=9495, sin=-13352 //k=622, cos=9474, sin=-13366 //k=623, cos=9454, sin=-13381 12905 //k=624, cos=9433, sin=-13395 //k=625, cos=9413, sin=-13410 //k=626, cos=9392, sin=-13424 //k=627, cos=9372, 12906 sin=-13439 //k=628, cos=9351, sin=-13453 //k=629, cos=9330, sin=-13467 //k=630, cos=9310, sin=-13482 //k=631, 12907 cos=9289, sin=-13496 //k=632, cos=9268, sin=-13510 //k=633, cos=9247, sin=-13524 //k=634, cos=9227, sin=-13538 12908 //k=635, cos=9206, sin=-13553 //k=636, cos=9185, sin=-13567 //k=637, cos=9164, sin=-13581 //k=638, cos=9143, 12909 sin=-13595 //k=639, cos=9123, sin=-13609 //k=640, cos=9102, sin=-13623 //k=641, cos=9081, sin=-13637 //k=642, 12910 cos=9060, sin=-13651 //k=643, cos=9039, sin=-13665 //k=644, cos=9018, sin=-13678 //k=645, cos=8997, sin=-13692 12911 //k=646, cos=8976, sin=-13706 //k=647, cos=8955, sin=-13720 //k=648, cos=8934, sin=-13733 //k=649, cos=8913, 12912 sin=-13747 //k=650, cos=8892, sin=-13761 //k=651, cos=8871, sin=-13774 //k=652, cos=8849, sin=-13788 //k=653, 12913 cos=8828, sin=-13802 //k=654, cos=8807, sin=-13815 //k=655, cos=8786, sin=-13829 //k=656, cos=8765, sin=-13842 12914 //k=657, cos=8743, sin=-13856 //k=658, cos=8722, sin=-13869 //k=659, cos=8701, sin=-13882 //k=660, cos=8680, 12915 sin=-13896 //k=661, cos=8658, sin=-13909 //k=662, cos=8637, sin=-13922 //k=663, cos=8616, sin=-13935 //k=664, 12916 cos=8594, sin=-13949 //k=665, cos=8573, sin=-13962 //k=666, cos=8551, sin=-13975 //k=667, cos=8530, sin=-13988 12917 //k=668, cos=8508, sin=-14001 //k=669, cos=8487, sin=-14014 //k=670, cos=8465, sin=-14027 //k=671, cos=8444, 12918 sin=-14040 //k=672, cos=8422, sin=-14053 //k=673, cos=8401, sin=-14066 //k=674, cos=8379, sin=-14079 //k=675, 12919 cos=8358, sin=-14092 //k=676, cos=8336, sin=-14104 //k=677, cos=8314, sin=-14117 //k=678, cos=8293, sin=-14130 12920 //k=679, cos=8271, sin=-14143 //k=680, cos=8249, sin=-14155 //k=681, cos=8227, sin=-14168 //k=682, cos=8206, 12921 sin=-14181 //k=683, cos=8184, sin=-14193 //k=684, cos=8162, sin=-14206 //k=685, cos=8140, sin=-14218 //k=686, 12922 cos=8119, sin=-14231 //k=687, cos=8097, sin=-14243 //k=688, cos=8075, sin=-14256 //k=689, cos=8053, sin=-14268 12923 //k=690, cos=8031, sin=-14280 //k=691, cos=8009, sin=-14293 //k=692, cos=7987, sin=-14305 //k=693, cos=7965, 12924 sin=-14317 //k=694, cos=7943, sin=-14329 //k=695, cos=7921, sin=-14341 //k=696, cos=7899, sin=-14354 //k=697, 12925 cos=7877, sin=-14366 //k=698, cos=7855, sin=-14378 //k=699, cos=7833, sin=-14390 //k=700, cos=7811, sin=-14402 12926 //k=701, cos=7789, sin=-14414 //k=702, cos=7767, sin=-14426 //k=703, cos=7745, sin=-14438 //k=704, cos=7723, 12927 sin=-14449 //k=705, cos=7700, sin=-14461 //k=706, cos=7678, sin=-14473 //k=707, cos=7656, sin=-14485 //k=708, 12928 cos=7634, sin=-14497 //k=709, cos=7612, sin=-14508 //k=710, cos=7589, sin=-14520 //k=711, cos=7567, sin=-14531 12929 //k=712, cos=7545, sin=-14543 //k=713, cos=7522, sin=-14555 //k=714, cos=7500, sin=-14566 //k=715, cos=7478, 12930 sin=-14578 //k=716, cos=7455, sin=-14589 //k=717, cos=7433, sin=-14601 //k=718, cos=7411, sin=-14612 //k=719, 12931 cos=7388, sin=-14623 //k=720, cos=7366, sin=-14635 //k=721, cos=7343, sin=-14646 //k=722, cos=7321, sin=-14657 12932 //k=723, cos=7298, sin=-14668 //k=724, cos=7276, sin=-14680 //k=725, cos=7253, sin=-14691 //k=726, cos=7231, 12933 sin=-14702 //k=727, cos=7208, sin=-14713 //k=728, cos=7186, sin=-14724 //k=729, cos=7163, sin=-14735 //k=730, 12934 cos=7140, sin=-14746 //k=731, cos=7118, sin=-14757 //k=732, cos=7095, sin=-14768 //k=733, cos=7072, sin=-14779 12935 //k=734, cos=7050, sin=-14789 //k=735, cos=7027, sin=-14800 //k=736, cos=7004, sin=-14811 //k=737, cos=6982, 12936

Supprimé : 4.0.1

Supprimé : 2







sin=-14822 //k=738, cos=6959, sin=-14832 //k=739, cos=6936, sin=-14843 //k=740, cos=6913, sin=-14854 //k=741, 12937 cos=6891, sin=-14864 //k=742, cos=6868, sin=-14875 //k=743, cos=6845, sin=-14885 //k=744, cos=6822, sin=-14896 12938 //k=745, cos=6799, sin=-14906 //k=746, cos=6776, sin=-14917 //k=747, cos=6753, sin=-14927 //k=748, cos=6731, 12939 sin=-14937 //k=749, cos=6708, sin=-14948 //k=750, cos=6685, sin=-14958 //k=751, cos=6662, sin=-14968 //k=752, 12940 cos=6639, sin=-14978 //k=753, cos=6616, sin=-14989 //k=754, cos=6593, sin=-14999 //k=755, cos=6570, sin=-15009 12941 //k=756, cos=6547, sin=-15019 //k=757, cos=6524, sin=-15029 //k=758, cos=6501, sin=-15039 //k=759, cos=6478, 12942 sin=-15049 //k=760, cos=6454, sin=-15059 //k=761, cos=6431, sin=-15069 //k=762, cos=6408, sin=-15078 //k=763, 12943 cos=6385, sin=-15088 //k=764, cos=6362, sin=-15098 //k=765, cos=6339, sin=-15108 //k=766, cos=6316, sin=-15118 12944 //k=767, cos=6292, sin=-15127 //k=768, cos=6269, sin=-15137 //k=769, cos=6246, sin=-15146 //k=770, cos=6223, 12945 sin=-15156 //k=771, cos=6199, sin=-15166 //k=772, cos=6176, sin=-15175 //k=773, cos=6153, sin=-15184 //k=774, 12946 cos=6130, sin=-15194 //k=775, cos=6106, sin=-15203 //k=776, cos=6083, sin=-15213 //k=777, cos=6060, sin=-15222 12947 //k=778, cos=6036, sin=-15231 //k=779, cos=6013, sin=-15240 //k=780, cos=5990, sin=-15250 //k=781, cos=5966, 12948 sin=-15259 //k=782, cos=5943, sin=-15268 //k=783, cos=5919, sin=-15277 //k=784, cos=5896, sin=-15286 //k=785, 12949 cos=5872, sin=-15295 //k=786, cos=5849, sin=-15304 //k=787, cos=5825, sin=-15313 //k=788, cos=5802, sin=-15322 12950 //k=789, cos=5778, sin=-15331 //k=790, cos=5755, sin=-15340 //k=791, cos=5731, sin=-15349 //k=792, cos=5708, 12951 sin=-15357 //k=793, cos=5684, sin=-15366 //k=794, cos=5661, sin=-15375 //k=795, cos=5637, sin=-15383 //k=796, 12952 cos=5613, sin=-15392 //k=797, cos=5590, sin=-15401 //k=798, cos=5566, sin=-15409 //k=799, cos=5543, sin=-15418 12953 //k=800, cos=5519, sin=-15426 //k=801, cos=5495, sin=-15435 //k=802, cos=5472, sin=-15443 //k=803, cos=5448, 12954 sin=-15451 //k=804, cos=5424, sin=-15460 //k=805, cos=5400, sin=-15468 //k=806, cos=5377, sin=-15476 //k=807, 12955 cos=5353, sin=-15485 //k=808, cos=5329, sin=-15493 //k=809, cos=5305, sin=-15501 //k=810, cos=5282, sin=-15509 12956 //k=811, cos=5258, sin=-15517 //k=812, cos=5234, sin=-15525 //k=813, cos=5210, sin=-15533 //k=814, cos=5186, 12957 sin=-15541 //k=815, cos=5163, sin=-15549 //k=816, cos=5139, sin=-15557 //k=817, cos=5115, sin=-15565 //k=818, 12958 cos=5091, sin=-15573 //k=819, cos=5067, sin=-15581 //k=820, cos=5043, sin=-15588 //k=821, cos=5019, sin=-15596 12959 //k=822, cos=4995, sin=-15604 //k=823, cos=4971, sin=-15611 //k=824, cos=4947, sin=-15619 //k=825, cos=4923, 12960 sin=-15627 //k=826, cos=4899, sin=-15634 //k=827, cos=4875, sin=-15642 //k=828, cos=4851, sin=-15649 //k=829, 12961 cos=4827, sin=-15656 //k=830, cos=4803, sin=-15664 //k=831, cos=4779, sin=-15671 //k=832, cos=4755, sin=-15679 12962 //k=833, cos=4731, sin=-15686 //k=834, cos=4707, sin=-15693 //k=835, cos=4683, sin=-15700 //k=836, cos=4659, 12963 sin=-15707 //k=837, cos=4635, sin=-15715 //k=838, cos=4611, sin=-15722 //k=839, cos=4587, sin=-15729 //k=840, 12964 cos=4563, sin=-15736 //k=841, cos=4538, sin=-15743 //k=842, cos=4514, sin=-15750 //k=843, cos=4490, sin=-15757 12965 //k=844, cos=4466, sin=-15763 //k=845, cos=4442, sin=-15770 //k=846, cos=4418, sin=-15777 //k=847, cos=4393, 12966 sin=-15784 //k=848, cos=4369, sin=-15791 //k=849, cos=4345, sin=-15797 //k=850, cos=4321, sin=-15804 //k=851, 12967 cos=4296, sin=-15810 //k=852, cos=4272, sin=-15817 //k=853, cos=4248, sin=-15824 //k=854, cos=4224, sin=-15830 12968 //k=855, cos=4199, sin=-15837 //k=856, cos=4175, sin=-15843 //k=857, cos=4151, sin=-15849 //k=858, cos=4126, 12969 sin=-15856 //k=859, cos=4102, sin=-15862 //k=860, cos=4078, sin=-15868 //k=861, cos=4053, sin=-15875 //k=862, 12970 cos=4029, sin=-15881 //k=863, cos=4005, sin=-15887 //k=864, cos=3980, sin=-15893 //k=865, cos=3956, sin=-15899 12971 //k=866, cos=3932, sin=-15905 //k=867, cos=3907, sin=-15911 //k=868, cos=3883, sin=-15917 //k=869, cos=3858, 12972 sin=-15923 //k=870, cos=3834, sin=-15929 //k=871, cos=3809, sin=-15935 //k=872, cos=3785, sin=-15941 //k=873, 12973 cos=3761, sin=-15946 //k=874, cos=3736, sin=-15952 //k=875, cos=3712, sin=-15958 //k=876, cos=3687, sin=-15964 12974 //k=877, cos=3663, sin=-15969 //k=878, cos=3638, sin=-15975 //k=879, cos=3614, sin=-15980 //k=880, cos=3589, 12975 sin=-15986 //k=881, cos=3565, sin=-15991 //k=882, cos=3540, sin=-15997 //k=883, cos=3516, sin=-16002 //k=884, 12976 cos=3491, sin=-16008 //k=885, cos=3466, sin=-16013 //k=886, cos=3442, sin=-16018 //k=887, cos=3417, sin=-16024 12977 //k=888, cos=3393, sin=-16029 //k=889, cos=3368, sin=-16034 //k=890, cos=3344, sin=-16039 //k=891, cos=3319, 12978 sin=-16044 //k=892, cos=3294, sin=-16049 //k=893, cos=3270, sin=-16054 //k=894, cos=3245, sin=-16059 //k=895, 12979 cos=3220, sin=-16064 //k=896, cos=3196, sin=-16069 //k=897, cos=3171, sin=-16074 //k=898, cos=3146, sin=-16079 12980 //k=899, cos=3122, sin=-16084 //k=900, cos=3097, sin=-16088 //k=901, cos=3072, sin=-16093 //k=902, cos=3048, 12981 sin=-16098 //k=903, cos=3023, sin=-16103 //k=904, cos=2998, sin=-16107 //k=905, cos=2974, sin=-16112 //k=906, 12982 cos=2949, sin=-16116 //k=907, cos=2924, sin=-16121 //k=908, cos=2899, sin=-16125 //k=909, cos=2875, sin=-16130 12983 //k=910, cos=2850, sin=-16134 //k=911, cos=2825, sin=-16138 //k=912, cos=2800, sin=-16143 //k=913, cos=2776, 12984 sin=-16147 //k=914, cos=2751, sin=-16151 //k=915, cos=2726, sin=-16156 //k=916, cos=2701, sin=-16160 //k=917, 12985 cos=2677, sin=-16164 //k=918, cos=2652, sin=-16168 //k=919, cos=2627, sin=-16172 //k=920, cos=2602, sin=-16176 12986 //k=921, cos=2577, sin=-16180 //k=922, cos=2553, sin=-16184 //k=923, cos=2528, sin=-16188 //k=924, cos=2503, 12987 sin=-16192 //k=925, cos=2478, sin=-16195 //k=926, cos=2453, sin=-16199 //k=927, cos=2428, sin=-16203 //k=928, 12988 cos=2403, sin=-16207 //k=929, cos=2379, sin=-16210 //k=930, cos=2354, sin=-16214 //k=931, cos=2329, sin=-16218 12989 //k=932, cos=2304, sin=-16221 //k=933, cos=2279, sin=-16225 //k=934, cos=2254, sin=-16228 //k=935, cos=2229, 12990 sin=-16232 //k=936, cos=2204, sin=-16235 //k=937, cos=2179, sin=-16238 //k=938, cos=2155, sin=-16242 //k=939, 12991 cos=2130, sin=-16245 //k=940, cos=2105, sin=-16248 //k=941, cos=2080, sin=-16251 //k=942, cos=2055, sin=-16255 12992 //k=943, cos=2030, sin=-16258 //k=944, cos=2005, sin=-16261 //k=945, cos=1980, sin=-16264 //k=946, cos=1955, 12993 sin=-16267 //k=947, cos=1930, sin=-16270 //k=948, cos=1905, sin=-16273 //k=949, cos=1880, sin=-16276 //k=950, 12994 cos=1855, sin=-16279 //k=951, cos=1830, sin=-16281 //k=952, cos=1805, sin=-16284 //k=953, cos=1780, sin=-16287 12995 //k=954, cos=1755, sin=-16290 //k=955, cos=1730, sin=-16292 //k=956, cos=1705, sin=-16295 //k=957, cos=1680, 12996

Supprimé : 4.0.1

Supprimé : 2







sin=-16298 //k=958, cos=1655, sin=-16300 //k=959, cos=1630, sin=-16303 //k=960, cos=1605, sin=-16305 //k=961, 12997 cos=1580, sin=-16308 //k=962, cos=1555, sin=-16310 //k=963, cos=1530, sin=-16312 //k=964, cos=1505, sin=-16315 12998 //k=965, cos=1480, sin=-16317 //k=966, cos=1455, sin=-16319 //k=967, cos=1430, sin=-16321 //k=968, cos=1405, 12999 sin=-16324 //k=969, cos=1380, sin=-16326 //k=970, cos=1355, sin=-16328 //k=971, cos=1330, sin=-16330 //k=972, 13000 cos=1305, sin=-16332 //k=973, cos=1280, sin=-16334 //k=974, cos=1255, sin=-16336 //k=975, cos=1230, sin=-16338 13001 //k=976, cos=1205, sin=-16340 //k=977, cos=1180, sin=-16341 //k=978, cos=1155, sin=-16343 //k=979, cos=1130, 13002 sin=-16345 //k=980, cos=1104, sin=-16347 //k=981, cos=1079, sin=-16348 //k=982, cos=1054, sin=-16350 //k=983, 13003 cos=1029, sin=-16352 //k=984, cos=1004, sin=-16353 //k=985, cos=979, sin=-16355 //k=986, cos=954, sin=-16356 13004 //k=987, cos=929, sin=-16358 //k=988, cos=904, sin=-16359 //k=989, cos=879, sin=-16360 //k=990, cos=854, sin=-13005 16362 //k=991, cos=829, sin=-16363 //k=992, cos=803, sin=-16364 //k=993, cos=778, sin=-16365 //k=994, cos=753, 13006 sin=-16367 //k=995, cos=728, sin=-16368 //k=996, cos=703, sin=-16369 //k=997, cos=678, sin=-16370 //k=998, 13007 cos=653, sin=-16371 //k=999, cos=628, sin=-16372 //k=1000, cos=603, sin=-16373 //k=1001, cos=577, sin=-16374 13008 //k=1002, cos=552, sin=-16375 //k=1003, cos=527, sin=-16375 //k=1004, cos=502, sin=-16376 //k=1005, cos=477, 13009 sin=-16377 //k=1006, cos=452, sin=-16378 //k=1007, cos=427, sin=-16378 //k=1008, cos=402, sin=-16379 //k=1009, 13010 cos=376, sin=-16380 //k=1010, cos=351, sin=-16380 //k=1011, cos=326, sin=-16381 //k=1012, cos=301, sin=-16381 13011 //k=1013, cos=276, sin=-16382 //k=1014, cos=251, sin=-16382 //k=1015, cos=226, sin=-16382 //k=1016, cos=201, 13012 sin=-16383 //k=1017, cos=175, sin=-16383 //k=1018, cos=150, sin=-16383 //k=1019, cos=125, sin=-16384 //k=1020, 13013 cos=100, sin=-16384 //k=1021, cos=75, sin=-16384 //k=1022, cos=50, sin=-16384 //k=1023, cos=25, sin=-16384 //>> 13014 13015 15MHz bandwidth 13016 k=0, cos=16383, sin=0 //k=1, cos=16383, sin=-34 //k=2, cos=16383, sin=-67 //k=3, cos=16383, sin=-101 //k=4, 13017 cos=16382, sin=-134 //k=5, cos=16382, sin=-168 //k=6, cos=16382, sin=-201 //k=7, cos=16381, sin=-235 //k=8, 13018 cos=16381, sin=-268 //k=9, cos=16380, sin=-302 //k=10, cos=16380, sin=-335 //k=11, cos=16379, sin=-369 //k=12, 13019 cos=16378, sin=-402 //k=13, cos=16377, sin=-436 //k=14, cos=16376, sin=-469 //k=15, cos=16375, sin=-503 //k=16, 13020 cos=16374, sin=-536 //k=17, cos=16373, sin=-570 //k=18, cos=16372, sin=-603 //k=19, cos=16371, sin=-637 //k=20, 13021 cos=16369, sin=-670 //k=21, cos=16368, sin=-704 //k=22, cos=16366, sin=-737 //k=23, cos=16365, sin=-770 //k=24, 13022 cos=16363, sin=-804 //k=25, cos=16362, sin=-837 //k=26, cos=16360, sin=-871 //k=27, cos=16358, sin=-904 //k=28, 13023 cos=16356, sin=-938 //k=29, cos=16354, sin=-971 //k=30, cos=16352, sin=-1005 //k=31, cos=16350, sin=-1038 13024 //k=32, cos=16348, sin=-1072 //k=33, cos=16346, sin=-1105 //k=34, cos=16343, sin=-1138 //k=35, cos=16341, sin=-13025 1172 //k=36, cos=16339, sin=-1205 //k=37, cos=16336, sin=-1239 //k=38, cos=16334, sin=-1272 //k=39, cos=16331, 13026 sin=-1306 //k=40, cos=16328, sin=-1339 //k=41, cos=16325, sin=-1372 //k=42, cos=16323, sin=-1406 //k=43, 13027 cos=16320, sin=-1439 //k=44, cos=16317, sin=-1472 //k=45, cos=16314, sin=-1506 //k=46, cos=16311, sin=-1539 13028 //k=47, cos=16307, sin=-1573 //k=48, cos=16304, sin=-1606 //k=49, cos=16301, sin=-1639 //k=50, cos=16297, sin=-13029 1673 //k=51, cos=16294, sin=-1706 //k=52, cos=16290, sin=-1739 //k=53, cos=16287, sin=-1773 //k=54, cos=16283, 13030 sin=-1806 //k=55, cos=16279, sin=-1839 //k=56, cos=16276, sin=-1872 //k=57, cos=16272, sin=-1906 //k=58, 13031 cos=16268, sin=-1939 //k=59, cos=16264, sin=-1972 //k=60, cos=16260, sin=-2006 //k=61, cos=16256, sin=-2039 13032 //k=62, cos=16251, sin=-2072 //k=63, cos=16247, sin=-2105 //k=64, cos=16243, sin=-2139 //k=65, cos=16238, sin=-13033 2172 //k=66, cos=16234, sin=-2205 //k=67, cos=16229, sin=-2238 //k=68, cos=16225, sin=-2271 //k=69, cos=16220, 13034 sin=-2305 //k=70, cos=16215, sin=-2338 //k=71, cos=16211, sin=-2371 //k=72, cos=16206, sin=-2404 //k=73, 13035 cos=16201, sin=-2437 //k=74, cos=16196, sin=-2470 //k=75, cos=16191, sin=-2503 //k=76, cos=16185, sin=-2537 13036 //k=77, cos=16180, sin=-2570 //k=78, cos=16175, sin=-2603 //k=79, cos=16170, sin=-2636 //k=80, cos=16164, sin=-13037 2669 //k=81, cos=16159, sin=-2702 //k=82, cos=16153, sin=-2735 //k=83, cos=16147, sin=-2768 //k=84, cos=16142, 13038 sin=-2801 //k=85, cos=16136, sin=-2834 //k=86, cos=16130, sin=-2867 //k=87, cos=16124, sin=-2900 //k=88, 13039 cos=16118, sin=-2933 //k=89, cos=16112, sin=-2966 //k=90, cos=16106, sin=-2999 //k=91, cos=16100, sin=-3032 13040 //k=92, cos=16094, sin=-3065 //k=93, cos=16088, sin=-3098 //k=94, cos=16081, sin=-3131 //k=95, cos=16075, sin=-13041 3163 //k=96, cos=16068, sin=-3196 //k=97, cos=16062, sin=-3229 //k=98, cos=16055, sin=-3262 //k=99, cos=16048, 13042 sin=-3295 //k=100, cos=16042, sin=-3328 //k=101, cos=16035, sin=-3361 //k=102, cos=16028, sin=-3393 //k=103, 13043 cos=16021, sin=-3426 //k=104, cos=16014, sin=-3459 //k=105, cos=16007, sin=-3492 //k=106, cos=15999, sin=-3524 13044 //k=107, cos=15992, sin=-3557 //k=108, cos=15985, sin=-3590 //k=109, cos=15978, sin=-3622 //k=110, cos=15970, 13045 sin=-3655 //k=111, cos=15963, sin=-3688 //k=112, cos=15955, sin=-3720 //k=113, cos=15947, sin=-3753 //k=114, 13046 cos=15940, sin=-3786 //k=115, cos=15932, sin=-3818 //k=116, cos=15924, sin=-3851 //k=117, cos=15916, sin=-3883 13047 //k=118, cos=15908, sin=-3916 //k=119, cos=15900, sin=-3948 //k=120, cos=15892, sin=-3981 //k=121, cos=15884, 13048 sin=-4013 //k=122, cos=15876, sin=-4046 //k=123, cos=15867, sin=-4078 //k=124, cos=15859, sin=-4111 //k=125, 13049 cos=15850, sin=-4143 //k=126, cos=15842, sin=-4176 //k=127, cos=15833, sin=-4208 //k=128, cos=15825, sin=-4240 13050 //k=129, cos=15816, sin=-4273 //k=130, cos=15807, sin=-4305 //k=131, cos=15798, sin=-4338 //k=132, cos=15790, 13051 sin=-4370 //k=133, cos=15781, sin=-4402 //k=134, cos=15772, sin=-4434 //k=135, cos=15762, sin=-4467 //k=136, 13052 cos=15753, sin=-4499 //k=137, cos=15744, sin=-4531 //k=138, cos=15735, sin=-4563 //k=139, cos=15725, sin=-4595 13053 //k=140, cos=15716, sin=-4628 //k=141, cos=15706, sin=-4660 //k=142, cos=15697, sin=-4692 //k=143, cos=15687, 13054 sin=-4724 //k=144, cos=15678, sin=-4756 //k=145, cos=15668, sin=-4788 //k=146, cos=15658, sin=-4820 //k=147, 13055

Supprimé : 4.0.1

Supprimé : 2







cos=15648, sin=-4852 //k=148, cos=15638, sin=-4884 //k=149, cos=15628, sin=-4916 //k=150, cos=15618, sin=-4948 13056 //k=151, cos=15608, sin=-4980 //k=152, cos=15598, sin=-5012 //k=153, cos=15587, sin=-5044 //k=154, cos=15577, 13057 sin=-5076 //k=155, cos=15567, sin=-5108 //k=156, cos=15556, sin=-5139 //k=157, cos=15546, sin=-5171 //k=158, 13058 cos=15535, sin=-5203 //k=159, cos=15524, sin=-5235 //k=160, cos=15514, sin=-5266 //k=161, cos=15503, sin=-5298 13059 //k=162, cos=15492, sin=-5330 //k=163, cos=15481, sin=-5362 //k=164, cos=15470, sin=-5393 //k=165, cos=15459, 13060 sin=-5425 //k=166, cos=15448, sin=-5456 //k=167, cos=15437, sin=-5488 //k=168, cos=15425, sin=-5520 //k=169, 13061 cos=15414, sin=-5551 //k=170, cos=15403, sin=-5583 //k=171, cos=15391, sin=-5614 //k=172, cos=15380, sin=-5646 13062 //k=173, cos=15368, sin=-5677 //k=174, cos=15356, sin=-5708 //k=175, cos=15345, sin=-5740 //k=176, cos=15333, 13063 sin=-5771 //k=177, cos=15321, sin=-5803 //k=178, cos=15309, sin=-5834 //k=179, cos=15297, sin=-5865 //k=180, 13064 cos=15285, sin=-5897 //k=181, cos=15273, sin=-5928 //k=182, cos=15261, sin=-5959 //k=183, cos=15249, sin=-5990 13065 //k=184, cos=15236, sin=-6021 //k=185, cos=15224, sin=-6053 //k=186, cos=15212, sin=-6084 //k=187, cos=15199, 13066 sin=-6115 //k=188, cos=15187, sin=-6146 //k=189, cos=15174, sin=-6177 //k=190, cos=15161, sin=-6208 //k=191, 13067 cos=15149, sin=-6239 //k=192, cos=15136, sin=-6270 //k=193, cos=15123, sin=-6301 //k=194, cos=15110, sin=-6332 13068 //k=195, cos=15097, sin=-6363 //k=196, cos=15084, sin=-6394 //k=197, cos=15071, sin=-6424 //k=198, cos=15058, 13069 sin=-6455 //k=199, cos=15045, sin=-6486 //k=200, cos=15031, sin=-6517 //k=201, cos=15018, sin=-6547 //k=202, 13070 cos=15004, sin=-6578 //k=203, cos=14991, sin=-6609 //k=204, cos=14977, sin=-6639 //k=205, cos=14964, sin=-6670 13071 //k=206, cos=14950, sin=-6701 //k=207, cos=14936, sin=-6731 //k=208, cos=14923, sin=-6762 //k=209, cos=14909, 13072 sin=-6792 //k=210, cos=14895, sin=-6823 //k=211, cos=14881, sin=-6853 //k=212, cos=14867, sin=-6884 //k=213, 13073 cos=14853, sin=-6914 //k=214, cos=14839, sin=-6944 //k=215, cos=14824, sin=-6975 //k=216, cos=14810, sin=-7005 13074 //k=217, cos=14796, sin=-7035 //k=218, cos=14781, sin=-7066 //k=219, cos=14767, sin=-7096 //k=220, cos=14752, 13075 sin=-7126 //k=221, cos=14738, sin=-7156 //k=222, cos=14723, sin=-7186 //k=223, cos=14708, sin=-7216 //k=224, 13076 cos=14693, sin=-7246 //k=225, cos=14679, sin=-7276 //k=226, cos=14664, sin=-7307 //k=227, cos=14649, sin=-7336 13077 //k=228, cos=14634, sin=-7366 //k=229, cos=14619, sin=-7396 //k=230, cos=14603, sin=-7426 //k=231, cos=14588, 13078 sin=-7456 //k=232, cos=14573, sin=-7486 //k=233, cos=14558, sin=-7516 //k=234, cos=14542, sin=-7545 //k=235, 13079 cos=14527, sin=-7575 //k=236, cos=14511, sin=-7605 //k=237, cos=14496, sin=-7635 //k=238, cos=14480, sin=-7664 13080 //k=239, cos=14464, sin=-7694 //k=240, cos=14448, sin=-7723 //k=241, cos=14433, sin=-7753 //k=242, cos=14417, 13081 sin=-7782 //k=243, cos=14401, sin=-7812 //k=244, cos=14385, sin=-7841 //k=245, cos=14369, sin=-7871 //k=246, 13082 cos=14353, sin=-7900 //k=247, cos=14336, sin=-7929 //k=248, cos=14320, sin=-7959 //k=249, cos=14304, sin=-7988 13083 //k=250, cos=14287, sin=-8017 //k=251, cos=14271, sin=-8046 //k=252, cos=14255, sin=-8076 //k=253, cos=14238, 13084 sin=-8105 //k=254, cos=14221, sin=-8134 //k=255, cos=14205, sin=-8163 //k=256, cos=14188, sin=-8192 //k=257, 13085 cos=14171, sin=-8221 //k=258, cos=14154, sin=-8250 //k=259, cos=14137, sin=-8279 //k=260, cos=14121, sin=-8308 13086 //k=261, cos=14104, sin=-8337 //k=262, cos=14086, sin=-8366 //k=263, cos=14069, sin=-8394 //k=264, cos=14052, 13087 sin=-8423 //k=265, cos=14035, sin=-8452 //k=266, cos=14018, sin=-8480 //k=267, cos=14000, sin=-8509 //k=268, 13088 cos=13983, sin=-8538 //k=269, cos=13965, sin=-8566 //k=270, cos=13948, sin=-8595 //k=271, cos=13930, sin=-8623 13089 //k=272, cos=13912, sin=-8652 //k=273, cos=13895, sin=-8680 //k=274, cos=13877, sin=-8709 //k=275, cos=13859, 13090 sin=-8737 //k=276, cos=13841, sin=-8765 //k=277, cos=13823, sin=-8794 //k=278, cos=13805, sin=-8822 //k=279, 13091 cos=13787, sin=-8850 //k=280, cos=13769, sin=-8878 //k=281, cos=13751, sin=-8906 //k=282, cos=13733, sin=-8935 13092 //k=283, cos=13714, sin=-8963 //k=284, cos=13696, sin=-8991 //k=285, cos=13677, sin=-9019 //k=286, cos=13659, 13093 sin=-9047 //k=287, cos=13640, sin=-9075 //k=288, cos=13622, sin=-9102 //k=289, cos=13603, sin=-9130 //k=290, 13094 cos=13585, sin=-9158 //k=291, cos=13566, sin=-9186 //k=292, cos=13547, sin=-9214 //k=293, cos=13528, sin=-9241 13095 //k=294, cos=13509, sin=-9269 //k=295, cos=13490, sin=-9297 //k=296, cos=13471, sin=-9324 //k=297, cos=13452, 13096 sin=-9352 //k=298, cos=13433, sin=-9379 //k=299, cos=13414, sin=-9407 //k=300, cos=13394, sin=-9434 //k=301, 13097 cos=13375, sin=-9461 //k=302, cos=13356, sin=-9489 //k=303, cos=13336, sin=-9516 //k=304, cos=13317, sin=-9543 13098 //k=305, cos=13297, sin=-9571 //k=306, cos=13278, sin=-9598 //k=307, cos=13258, sin=-9625 //k=308, cos=13238, 13099 sin=-9652 //k=309, cos=13218, sin=-9679 //k=310, cos=13199, sin=-9706 //k=311, cos=13179, sin=-9733 //k=312, 13100 cos=13159, sin=-9760 //k=313, cos=13139, sin=-9787 //k=314, cos=13119, sin=-9814 //k=315, cos=13099, sin=-9841 13101 //k=316, cos=13079, sin=-9867 //k=317, cos=13058, sin=-9894 //k=318, cos=13038, sin=-9921 //k=319, cos=13018, 13102 sin=-9947 //k=320, cos=12997, sin=-9974 //k=321, cos=12977, sin=-10001 //k=322, cos=12956, sin=-10027 //k=323, 13103 cos=12936, sin=-10054 //k=324, cos=12915, sin=-10080 //k=325, cos=12895, sin=-10106 //k=326, cos=12874, sin=-13104 10133 //k=327, cos=12853, sin=-10159 //k=328, cos=12832, sin=-10185 //k=329, cos=12812, sin=-10212 //k=330, 13105 cos=12791, sin=-10238 //k=331, cos=12770, sin=-10264 //k=332, cos=12749, sin=-10290 //k=333, cos=12728, sin=-13106 10316 //k=334, cos=12707, sin=-10342 //k=335, cos=12685, sin=-10368 //k=336, cos=12664, sin=-10394 //k=337, 13107 cos=12643, sin=-10420 //k=338, cos=12621, sin=-10446 //k=339, cos=12600, sin=-10471 //k=340, cos=12579, sin=-13108 10497 //k=341, cos=12557, sin=-10523 //k=342, cos=12536, sin=-10549 //k=343, cos=12514, sin=-10574 //k=344, 13109 cos=12492, sin=-10600 //k=345, cos=12471, sin=-10625 //k=346, cos=12449, sin=-10651 //k=347, cos=12427, sin=-13110 10676 //k=348, cos=12405, sin=-10702 //k=349, cos=12383, sin=-10727 //k=350, cos=12361, sin=-10752 //k=351, 13111 cos=12339, sin=-10778 //k=352, cos=12317, sin=-10803 //k=353, cos=12295, sin=-10828 //k=354, cos=12273, sin=-13112 10853 //k=355, cos=12251, sin=-10878 //k=356, cos=12228, sin=-10903 //k=357, cos=12206, sin=-10928 //k=358, 13113 cos=12184, sin=-10953 //k=359, cos=12161, sin=-10978 //k=360, cos=12139, sin=-11003 //k=361, cos=12116, sin=-13114 11028 //k=362, cos=12094, sin=-11052 //k=363, cos=12071, sin=-11077 //k=364, cos=12048, sin=-11102 //k=365, 13115

Supprimé : 4.0.1

Supprimé : 2







cos=12026, sin=-11126 //k=366, cos=12003, sin=-11151 //k=367, cos=11980, sin=-11175 //k=368, cos=11957, sin=-13116 11200 //k=369, cos=11934, sin=-11224 //k=370, cos=11911, sin=-11249 //k=371, cos=11888, sin=-11273 //k=372, 13117 cos=11865, sin=-11297 //k=373, cos=11842, sin=-11322 //k=374, cos=11819, sin=-11346 //k=375, cos=11796, sin=-13118 11370 //k=376, cos=11772, sin=-11394 //k=377, cos=11749, sin=-11418 //k=378, cos=11726, sin=-11442 //k=379, 13119 cos=11702, sin=-11466 //k=380, cos=11679, sin=-11490 //k=381, cos=11655, sin=-11514 //k=382, cos=11632, sin=-13120 11538 //k=383, cos=11608, sin=-11562 //k=384, cos=11584, sin=-11585 //k=385, cos=11561, sin=-11609 //k=386, 13121 cos=11537, sin=-11633 //k=387, cos=11513, sin=-11656 //k=388, cos=11489, sin=-11680 //k=389, cos=11465, sin=-13122 11703 //k=390, cos=11441, sin=-11727 //k=391, cos=11417, sin=-11750 //k=392, cos=11393, sin=-11773 //k=393, 13123 cos=11369, sin=-11797 //k=394, cos=11345, sin=-11820 //k=395, cos=11321, sin=-11843 //k=396, cos=11297, sin=-13124 11866 //k=397, cos=11272, sin=-11889 //k=398, cos=11248, sin=-11912 //k=399, cos=11224, sin=-11935 //k=400, 13125 cos=11199, sin=-11958 //k=401, cos=11175, sin=-11981 //k=402, cos=11150, sin=-12004 //k=403, cos=11126, sin=-13126 12027 //k=404, cos=11101, sin=-12049 //k=405, cos=11076, sin=-12072 //k=406, cos=11052, sin=-12095 //k=407, 13127 cos=11027, sin=-12117 //k=408, cos=11002, sin=-12140 //k=409, cos=10977, sin=-12162 //k=410, cos=10952, sin=-13128 12185 //k=411, cos=10927, sin=-12207 //k=412, cos=10902, sin=-12229 //k=413, cos=10877, sin=-12252 //k=414, 13129 cos=10852, sin=-12274 //k=415, cos=10827, sin=-12296 //k=416, cos=10802, sin=-12318 //k=417, cos=10777, sin=-13130 12340 //k=418, cos=10751, sin=-12362 //k=419, cos=10726, sin=-12384 //k=420, cos=10701, sin=-12406 //k=421, 13131 cos=10675, sin=-12428 //k=422, cos=10650, sin=-12450 //k=423, cos=10624, sin=-12472 //k=424, cos=10599, sin=-13132 12493 //k=425, cos=10573, sin=-12515 //k=426, cos=10548, sin=-12537 //k=427, cos=10522, sin=-12558 //k=428, 13133 cos=10496, sin=-12580 //k=429, cos=10471, sin=-12601 //k=430, cos=10445, sin=-12622 //k=431, cos=10419, sin=-13134 12644 //k=432, cos=10393, sin=-12665 //k=433, cos=10367, sin=-12686 //k=434, cos=10341, sin=-12707 //k=435, 13135 cos=10315, sin=-12729 //k=436, cos=10289, sin=-12750 //k=437, cos=10263, sin=-12771 //k=438, cos=10237, sin=-13136 12792 //k=439, cos=10211, sin=-12813 //k=440, cos=10184, sin=-12833 //k=441, cos=10158, sin=-12854 //k=442, 13137 cos=10132, sin=-12875 //k=443, cos=10106, sin=-12896 //k=444, cos=10079, sin=-12916 //k=445, cos=10053, sin=-13138 12937 //k=446, cos=10026, sin=-12957 //k=447, cos=10000, sin=-12978 //k=448, cos=9973, sin=-12998 //k=449, 13139 cos=9947, sin=-13019 //k=450, cos=9920, sin=-13039 //k=451, cos=9893, sin=-13059 //k=452, cos=9866, sin=-13079 13140 //k=453, cos=9840, sin=-13100 //k=454, cos=9813, sin=-13120 //k=455, cos=9786, sin=-13140 //k=456, cos=9759, 13141 sin=-13160 //k=457, cos=9732, sin=-13180 //k=458, cos=9705, sin=-13200 //k=459, cos=9678, sin=-13219 //k=460, 13142 cos=9651, sin=-13239 //k=461, cos=9624, sin=-13259 //k=462, cos=9597, sin=-13279 //k=463, cos=9570, sin=-13298 13143 //k=464, cos=9543, sin=-13318 //k=465, cos=9515, sin=-13337 //k=466, cos=9488, sin=-13357 //k=467, cos=9461, 13144 sin=-13376 //k=468, cos=9433, sin=-13395 //k=469, cos=9406, sin=-13415 //k=470, cos=9378, sin=-13434 //k=471, 13145 cos=9351, sin=-13453 //k=472, cos=9323, sin=-13472 //k=473, cos=9296, sin=-13491 //k=474, cos=9268, sin=-13510 13146 //k=475, cos=9241, sin=-13529 //k=476, cos=9213, sin=-13548 //k=477, cos=9185, sin=-13567 //k=478, cos=9157, 13147 sin=-13585 //k=479, cos=9130, sin=-13604 //k=480, cos=9102, sin=-13623 //k=481, cos=9074, sin=-13641 //k=482, 13148 cos=9046, sin=-13660 //k=483, cos=9018, sin=-13678 //k=484, cos=8990, sin=-13697 //k=485, cos=8962, sin=-13715 13149 //k=486, cos=8934, sin=-13733 //k=487, cos=8906, sin=-13752 //k=488, cos=8878, sin=-13770 //k=489, cos=8849, 13150 sin=-13788 //k=490, cos=8821, sin=-13806 //k=491, cos=8793, sin=-13824 //k=492, cos=8765, sin=-13842 //k=493, 13151 cos=8736, sin=-13860 //k=494, cos=8708, sin=-13878 //k=495, cos=8680, sin=-13896 //k=496, cos=8651, sin=-13913 13152 //k=497, cos=8623, sin=-13931 //k=498, cos=8594, sin=-13949 //k=499, cos=8566, sin=-13966 //k=500, cos=8537, 13153 sin=-13984 //k=501, cos=8508, sin=-14001 //k=502, cos=8480, sin=-14018 //k=503, cos=8451, sin=-14036 //k=504, 13154 cos=8422, sin=-14053 //k=505, cos=8394, sin=-14070 //k=506, cos=8365, sin=-14087 //k=507, cos=8336, sin=-14104 13155 //k=508, cos=8307, sin=-14121 //k=509, cos=8278, sin=-14138 //k=510, cos=8249, sin=-14155 //k=511, cos=8220, 13156 sin=-14172 //k=512, cos=8191, sin=-14189 //k=513, cos=8162, sin=-14206 //k=514, cos=8133, sin=-14222 //k=515, 13157 cos=8104, sin=-14239 //k=516, cos=8075, sin=-14256 //k=517, cos=8046, sin=-14272 //k=518, cos=8017, sin=-14288 13158 //k=519, cos=7987, sin=-14305 //k=520, cos=7958, sin=-14321 //k=521, cos=7929, sin=-14337 //k=522, cos=7899, 13159 sin=-14354 //k=523, cos=7870, sin=-14370 //k=524, cos=7841, sin=-14386 //k=525, cos=7811, sin=-14402 //k=526, 13160 cos=7782, sin=-14418 //k=527, cos=7752, sin=-14434 //k=528, cos=7723, sin=-14449 //k=529, cos=7693, sin=-14465 13161 //k=530, cos=7663, sin=-14481 //k=531, cos=7634, sin=-14497 //k=532, cos=7604, sin=-14512 //k=533, cos=7574, 13162 sin=-14528 //k=534, cos=7545, sin=-14543 //k=535, cos=7515, sin=-14558 //k=536, cos=7485, sin=-14574 //k=537, 13163 cos=7455, sin=-14589 //k=538, cos=7426, sin=-14604 //k=539, cos=7396, sin=-14619 //k=540, cos=7366, sin=-14635 13164 //k=541, cos=7336, sin=-14650 //k=542, cos=7306, sin=-14665 //k=543, cos=7276, sin=-14680 //k=544, cos=7246, 13165 sin=-14694 //k=545, cos=7216, sin=-14709 //k=546, cos=7186, sin=-14724 //k=547, cos=7155, sin=-14739 //k=548, 13166 cos=7125, sin=-14753 //k=549, cos=7095, sin=-14768 //k=550, cos=7065, sin=-14782 //k=551, cos=7035, sin=-14797 13167 //k=552, cos=7004, sin=-14811 //k=553, cos=6974, sin=-14825 //k=554, cos=6944, sin=-14839 //k=555, cos=6913, 13168 sin=-14854 //k=556, cos=6883, sin=-14868 //k=557, cos=6853, sin=-14882 //k=558, cos=6822, sin=-14896 //k=559, 13169 cos=6792, sin=-14910 //k=560, cos=6761, sin=-14924 //k=561, cos=6731, sin=-14937 //k=562, cos=6700, sin=-14951 13170 //k=563, cos=6669, sin=-14965 //k=564, cos=6639, sin=-14978 //k=565, cos=6608, sin=-14992 //k=566, cos=6577, 13171 sin=-15005 //k=567, cos=6547, sin=-15019 //k=568, cos=6516, sin=-15032 //k=569, cos=6485, sin=-15046 //k=570, 13172 cos=6454, sin=-15059 //k=571, cos=6424, sin=-15072 //k=572, cos=6393, sin=-15085 //k=573, cos=6362, sin=-15098 13173 //k=574, cos=6331, sin=-15111 //k=575, cos=6300, sin=-15124 //k=576, cos=6269, sin=-15137 //k=577, cos=6238, 13174 sin=-15150 //k=578, cos=6207, sin=-15162 //k=579, cos=6176, sin=-15175 //k=580, cos=6145, sin=-15188 //k=581, 13175

Supprimé : 4.0.1

Supprimé : 2







cos=6114, sin=-15200 //k=582, cos=6083, sin=-15213 //k=583, cos=6052, sin=-15225 //k=584, cos=6021, sin=-15237 13176 //k=585, cos=5990, sin=-15250 //k=586, cos=5958, sin=-15262 //k=587, cos=5927, sin=-15274 //k=588, cos=5896, 13177 sin=-15286 //k=589, cos=5865, sin=-15298 //k=590, cos=5833, sin=-15310 //k=591, cos=5802, sin=-15322 //k=592, 13178 cos=5771, sin=-15334 //k=593, cos=5739, sin=-15346 //k=594, cos=5708, sin=-15357 //k=595, cos=5676, sin=-15369 13179 //k=596, cos=5645, sin=-15381 //k=597, cos=5613, sin=-15392 //k=598, cos=5582, sin=-15404 //k=599, cos=5550, 13180 sin=-15415 //k=600, cos=5519, sin=-15426 //k=601, cos=5487, sin=-15438 //k=602, cos=5456, sin=-15449 //k=603, 13181 cos=5424, sin=-15460 //k=604, cos=5393, sin=-15471 //k=605, cos=5361, sin=-15482 //k=606, cos=5329, sin=-15493 13182 //k=607, cos=5298, sin=-15504 //k=608, cos=5266, sin=-15515 //k=609, cos=5234, sin=-15525 //k=610, cos=5202, 13183 sin=-15536 //k=611, cos=5171, sin=-15547 //k=612, cos=5139, sin=-15557 //k=613, cos=5107, sin=-15568 //k=614, 13184 cos=5075, sin=-15578 //k=615, cos=5043, sin=-15588 //k=616, cos=5011, sin=-15599 //k=617, cos=4979, sin=-15609 13185 //k=618, cos=4947, sin=-15619 //k=619, cos=4915, sin=-15629 //k=620, cos=4883, sin=-15639 //k=621, cos=4851, 13186 sin=-15649 //k=622, cos=4819, sin=-15659 //k=623, cos=4787, sin=-15669 //k=624, cos=4755, sin=-15679 //k=625, 13187 cos=4723, sin=-15688 //k=626, cos=4691, sin=-15698 //k=627, cos=4659, sin=-15707 //k=628, cos=4627, sin=-15717 13188 //k=629, cos=4595, sin=-15726 //k=630, cos=4563, sin=-15736 //k=631, cos=4530, sin=-15745 //k=632, cos=4498, 13189 sin=-15754 //k=633, cos=4466, sin=-15763 //k=634, cos=4434, sin=-15772 //k=635, cos=4401, sin=-15782 //k=636, 13190 cos=4369, sin=-15791 //k=637, cos=4337, sin=-15799 //k=638, cos=4305, sin=-15808 //k=639, cos=4272, sin=-15817 13191 //k=640, cos=4240, sin=-15826 //k=641, cos=4207, sin=-15834 //k=642, cos=4175, sin=-15843 //k=643, cos=4143, 13192 sin=-15851 //k=644, cos=4110, sin=-15860 //k=645, cos=4078, sin=-15868 //k=646, cos=4045, sin=-15877 //k=647, 13193 cos=4013, sin=-15885 //k=648, cos=3980, sin=-15893 //k=649, cos=3948, sin=-15901 //k=650, cos=3915, sin=-15909 13194 //k=651, cos=3883, sin=-15917 //k=652, cos=3850, sin=-15925 //k=653, cos=3818, sin=-15933 //k=654, cos=3785, 13195 sin=-15941 //k=655, cos=3752, sin=-15948 //k=656, cos=3720, sin=-15956 //k=657, cos=3687, sin=-15964 //k=658, 13196 cos=3655, sin=-15971 //k=659, cos=3622, sin=-15979 //k=660, cos=3589, sin=-15986 //k=661, cos=3556, sin=-15993 13197 //k=662, cos=3524, sin=-16000 //k=663, cos=3491, sin=-16008 //k=664, cos=3458, sin=-16015 //k=665, cos=3425, 13198 sin=-16022 //k=666, cos=3393, sin=-16029 //k=667, cos=3360, sin=-16036 //k=668, cos=3327, sin=-16042 //k=669, 13199 cos=3294, sin=-16049 //k=670, cos=3261, sin=-16056 //k=671, cos=3229, sin=-16063 //k=672, cos=3196, sin=-16069 13200 //k=673, cos=3163, sin=-16076 //k=674, cos=3130, sin=-16082 //k=675, cos=3097, sin=-16088 //k=676, cos=3064, 13201 sin=-16095 //k=677, cos=3031, sin=-16101 //k=678, cos=2998, sin=-16107 //k=679, cos=2965, sin=-16113 //k=680, 13202 cos=2932, sin=-16119 //k=681, cos=2899, sin=-16125 //k=682, cos=2866, sin=-16131 //k=683, cos=2833, sin=-16137 13203 //k=684, cos=2800, sin=-16143 //k=685, cos=2767, sin=-16148 //k=686, cos=2734, sin=-16154 //k=687, cos=2701, 13204 sin=-16160 //k=688, cos=2668, sin=-16165 //k=689, cos=2635, sin=-16171 //k=690, cos=2602, sin=-16176 //k=691, 13205 cos=2569, sin=-16181 //k=692, cos=2536, sin=-16186 //k=693, cos=2503, sin=-16192 //k=694, cos=2470, sin=-16197 13206 //k=695, cos=2437, sin=-16202 //k=696, cos=2403, sin=-16207 //k=697, cos=2370, sin=-16212 //k=698, cos=2337, 13207 sin=-16216 //k=699, cos=2304, sin=-16221 //k=700, cos=2271, sin=-16226 //k=701, cos=2238, sin=-16230 //k=702, 13208 cos=2204, sin=-16235 //k=703, cos=2171, sin=-16239 //k=704, cos=2138, sin=-16244 //k=705, cos=2105, sin=-16248 13209 //k=706, cos=2072, sin=-16252 //k=707, cos=2038, sin=-16257 //k=708, cos=2005, sin=-16261 //k=709, cos=1972, 13210 sin=-16265 //k=710, cos=1938, sin=-16269 //k=711, cos=1905, sin=-16273 //k=712, cos=1872, sin=-16277 //k=713, 13211 cos=1839, sin=-16280 //k=714, cos=1805, sin=-16284 //k=715, cos=1772, sin=-16288 //k=716, cos=1739, sin=-16291 13212 //k=717, cos=1705, sin=-16295 //k=718, cos=1672, sin=-16298 //k=719, cos=1639, sin=-16302 //k=720, cos=1605, 13213 sin=-16305 //k=721, cos=1572, sin=-16308 //k=722, cos=1539, sin=-16312 //k=723, cos=1505, sin=-16315 //k=724, 13214 cos=1472, sin=-16318 //k=725, cos=1439, sin=-16321 //k=726, cos=1405, sin=-16324 //k=727, cos=1372, sin=-16326 13215 //k=728, cos=1338, sin=-16329 //k=729, cos=1305, sin=-16332 //k=730, cos=1272, sin=-16335 //k=731, cos=1238, 13216 sin=-16337 //k=732, cos=1205, sin=-16340 //k=733, cos=1171, sin=-16342 //k=734, cos=1138, sin=-16344 //k=735, 13217 cos=1104, sin=-16347 //k=736, cos=1071, sin=-16349 //k=737, cos=1038, sin=-16351 //k=738, cos=1004, sin=-16353 13218 //k=739, cos=971, sin=-16355 //k=740, cos=937, sin=-16357 //k=741, cos=904, sin=-16359 //k=742, cos=870, sin=-13219 16361 //k=743, cos=837, sin=-16363 //k=744, cos=803, sin=-16364 //k=745, cos=770, sin=-16366 //k=746, cos=736, 13220 sin=-16367 //k=747, cos=703, sin=-16369 //k=748, cos=669, sin=-16370 //k=749, cos=636, sin=-16372 //k=750, 13221 cos=603, sin=-16373 //k=751, cos=569, sin=-16374 //k=752, cos=536, sin=-16375 //k=753, cos=502, sin=-16376 13222 //k=754, cos=469, sin=-16377 //k=755, cos=435, sin=-16378 //k=756, cos=402, sin=-16379 //k=757, cos=368, sin=-13223 16380 //k=758, cos=335, sin=-16381 //k=759, cos=301, sin=-16381 //k=760, cos=268, sin=-16382 //k=761, cos=234, 13224 sin=-16382 //k=762, cos=201, sin=-16383 //k=763, cos=167, sin=-16383 //k=764, cos=134, sin=-16383 //k=765, 13225 cos=100, sin=-16384 //k=766, cos=67, sin=-16384 //k=767, cos=33, sin=-16384 // 13226 13227

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Mis en forme : Anglais(Royaume-Uni)Annex 13. Power commands mediation in iBTS 13228

13229 Introduction 13230 This appendix contains power computations to adjust power level of signals output by the CE. The 13231 objective is to determine the maximum input power level and the power offsets to apply at the input of the 13232 CE on top of the power commands to scale correctly the signals that are going to the RRH through the 13233 FPGA combiner. 13234 13235 RRH interpretation of CE signals 13236 The power relative to full scale is computed. It corresponds to the average power per received sample, 13237 normalized to the full scale power. Here is an example, of how this computation is performed: 13238 On CPRI link for LTE the I & Q samples are 15 bits encoded from [-16384 to 16383] so 2^14 (16383) is our 13239 full scale. 13240 Lets take an example of 3 samples: 13241 Signal = [(103 - 450*j) (-178 + 239*j) (-4320 +423*j)] 13242 The instantaneous digital power is (I^2 + Q^2) is: 13243 Power = [ (103 ^2 + 450^2) (178^2 + 239^2) (4320^2 + 423^2)] 13244 Power = 213109 88805 18841329 13245 The power relative to Fullscale is: 13246 Power = [213109 88805 18841329] / ( 16384)^2 13247 Power = [213109 88805 18841329] / 2^28 13248 The power in dBFS is : 13249 Power_dBFS = 10 * log10( mean (Power) ) = -16.24 dBFs 13250 A margin is taken at RRH side as follows: 13251 A parameter called HSSL_dBFS (digital backoff) is defined, such as if the computed Power_dBFS is equal 13252

to HSSL_dBFS, the RRH output power will be the maximum power called hereafter maxP . This power level 13253 is denoted HSSL

dBFSP in the sequel. 13254

In the example, if the backoff is -12 dB, then the output power would be Power_dBFS+12 dB below maxP . In 13255 the previous example, we would have maxP +12-16.24= maxP -4.24. 13256

In the example, if the backoff is -12 dB, then the output power would be Power_dBFS+12 dB below maxP. In 13257

the previous example, we would have maxP+12-16.24= maxP

-4.24. 13258 The signal output power level has to be adjusted in the CE such as to meet this behaviour. 13259 13260 CE Power computations: 13261

Case 3 MHz / 5 MHz / 10 MHz / 15MHz / 20MHz bandwidth 13262

Let’s consider the signal levels inside the CE. The CE model is reminded in the following figure: 13263 3 MHz: 13264

13265

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5 MHz 13266

13267

10 MHz 13268

13269

13270

15 MHz 13271

13272

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Supprimé :







20 MHz 13273

13274 13275 For an input power offset to CE of 0dB, the power observed at the output of the power scaling stage will be 13276 0 dBm (as per our design choice). 13277

Let’s consider only one RE containing some energy, the power observed after power scaling is denoted '

aP 13278

(this is the linear power). 13279 The 19 bits shift does not change the power level (only precision is lost going from Q(1,33) representation 13280 to Q(1,14) representation). 13281

So the input power level at iFFT input is the same '

aa PP =. 13282

The chosen iFFT architecture is an unscaled one, so the output power will be: 13283

afftn

b PP _2=. 13284

Again the shift does not change the average power level. We suppose that the saturation does not occur 13285 often enough to modify the average power level. So we can consider that saturation does not change 13286 power level too. 13287 The average power level at the FFT output will then be: 13288

afftna

fftn

b PP

P == _

_

2

2

13289 The output of the saturation step is represented in Q(6,9) format. This means that the full scale signal level 13290

is 52 , and the full scale per sample power is 102 . Thus the average power to full scale level observed at 13291 the output of the FFT is: 13292

)2

log(.1010a

dBFS

PP = 13293

13294 This is the contribution of one RE to the output signal, if more are allocated and contain energy, then the 13295 power to full scale would be: 13296

)2

log(.1010Tot

dBFS

PP = , where

∑=RE

aTot REPP )(. 13297

13298 Power settings: 13299 We can derive from the previous two sections, the power settings that can be used within the CE. 13300

To achieve a power level of maxP at the output of the RRH, the power at the output of the CE should exhibit 13301 an average power to full scale level of HSSL

dBFSP ( HSSLdBFSP is a negative quantity). 13302

Thus, for a power level of aP at the input of the CE, the output equivalent power at RRH level would be: 13303

( ) HSSLdBFS

adBFS

HSSLdBFS

Txa P

PPPPPP −

⋅+=−−=

10maxmax2

210log10 13304

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aP depends on the power command received by CE. We know that for 0dB power command, 0 dBm is 13305

observed at the input of the CE (as per design choice), so ( ) dBadBma PPP ∆=⋅= 10log10 where dBP∆ is 13306

the (negative) power command in dB received by the CE for the considered RE. 13307

We then have: 13308

( ) HSSLdBFSdB

Txa PPPP −⋅−∆+= 10

max2 210log10 . 13309

2TxaP represents the total power per RE, observed at the RRH output. Please note that this considers the 13310

total power radiated by both transmit antennas. So to derive the radiated power per antenna for that RE, 3 13311 dB must be subtracted. 13312

( ) 3210log10 10max −−⋅−∆+= HSSL

dBFSdBTx

a PPPP 13313

As an example let’s consider a power command of 0dB, a backoff of -12 dB and 30W output power 13314 (44.77dBm), we have then an output power contribution of this RE of: 13315

67.261212.3077.442 =+−=TxaP dBm. 13316

Having the maximum power offset commands that are available for each modulation, we can deduce the 13317 maximum input power per RE, in a 30W RRH case: 13318

( ) 67.32667.262max, =+=QPSKPTx

a dBm (per RE) 13319

( ) 07.304.367.26162max, =+=QAMPTx


( ) 97.283.267.26642max, =+=QAMPTx


( ) =+= 667.262max, HICHPTx

a 32.67dBm (per RE) 13322

( ) 67.29367.262max, =+=RSPTx


( ) 67.26,2max, =SSSPSSPTx

a dBm (per RE), please note that PSS and SSS are transmitted on one antenna. 13324

13325

Considering no fast power control is allowed on PDSCH, this means that the output power per RE on 13326 PDSCH channel should be restricted to 22.95 dBm (in our example). 13327

Suppose as an example that we setup maximum power for all the RE allocated to a PDSCH channel (600 13328 RE in 10MHz), we would have an output power per antenna of: 13329

( ) 74.50397.2830010log.10 =−+=TxaP dBm, which is above the 30W. So with this restriction on 13330

maximum power per RE, the 30W max power per RRH transmit antenna is achievable. 13331

This assesses that with current settings, maximum power on PDSCH can be obtained with current 13332 restrictions on maximum input power level at the CE. 13333

13334

Summary: 13335

The relationship between power commands sent to CE and actual applied power per RE at RRH level 13336 (counting for both transmit antennas) is clarified in previous section, so all commands coming to CE should 13337 take into account for the following relationship: 13338

( ) HSSLdBFSdB

Txa PPPP −⋅−∆+= 10

max2 210log10 13339

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Here 2TxaP shall be the transmit power per resource element setup at OAM level for the given channel, and 13340

dBP∆ is the power command from the FPGA A point of view. This mediation shall be operated at the DSP 13341

level. 13342

Following restrictions apply on power offsets to send to CE, per channel: 13343

• PDSCH: dBPdB 3.2max +=∆ , dBPdB 45min −=∆ (CWV) 13344

• PHICH: dBPdB 0max =∆ , dBPdB 45min −=∆ (CWV) 13345

• RS: dBPdB 3max =∆ , dBPdB 45min −=∆ (CWV) 13346

• PSS/SSS: dBPdB 0max =∆ , dBPdB 45min −=∆ (CWV) 13347

Please note that the restriction to a minimum of dBP∆min is a layer one performance limitation (is out 13348

passed, performance loss might be observed, but as a matter of fact, outpassing might be authorized). 13349

13350

Case 1.4 MHz bandwidth 13351

Let’s consider the signal levels inside the CE. The CE model is reminded in the following figure: 13352

13353

The transmission power relationship becomes: 13354

( ) HSSLdBFSdB

Txa PPPP −⋅−∆+= 8

max2 210log10 13355

13356

The downlink scheduler should check the following constraints for power offsets. 13357

To remain in a generic case for the PA max power, the power offsets are expressed relative to the 13358 cellDlTotalPower. In other words the constraint PDCCH_max_rel_power = -12.12 dB in the table below for 13359 10 MHze case matches 44.77-12.12 = 32.65 dBm. 13360

13361

Pay attention to the fact that the PDSCH power offsets given in this table are per antenna while the 13362 pCFICH, pBCH, pHICH and PDCCH power offsets are for both antennas 13363

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Supprimé :







13364

19. ABREVIATIONS AND DEFINITIONS 13365

19.1. ABREVIATIONS 13366

CFO Carrier Frequency Offset 13367

CP Cyclic Prefix 13368

CQI Channel Quality Indication 13369

CPICH Common Pilot Channel 13370

CRC Cyclic Redundancy Check 13371

DCH Dedicated channel 13372

DFT Discrete Fourier Transform 13373

DMRS Demodulation Reference Signal 13374

FEC Forward Error Correction 13375

FFT Fast Fourier Transform 13376

FDD Frequency Division Duplex 13377

FDM Frequency Division Multiplexing 13378

HSOPA High Speed OFDM Packet Access 13379

HARQ Hybrid Automatic Repeat and Request 13380

ICI Inter Carrier Interference 13381

IDFT Inverse Discrete Fourier Transform 13382

IFFT Inverse Fast Fourier Transform 13383

IP Internet Protocol 13384

LLR Log Likelihood Ratio 13385

MAC Medium Access Control 13386

MIMO Multiple Input Multiple Output 13387

MMSE Minimum Mean Squared Error 13388

MRC Maximum Ratio Combiner 13389

OFDM Orthogonal Frequency Division Multiplexing 13390

PRB Physical resource block (= 12 consecutive sub-carriers for UL) 13391

QPP Quadratic Polynomial Permutation 13392

RV Redundancy Version 13393

SCH Shared Channel 13394

SC-FDMA Single Carrier – Frequency Division Multiple Access 13395

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SRS Sounding Reference Signal 13396

TrCh Transport Channel 13397

UE User Equipment 13398

ZF Zero Forcing 13399

13400 13401

�� END OF DOCUMENT 13402

13403 13404

13405

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1. INTRODUCTION..........................................................................................................................11

1.1. SCOPE OF THIS DOCUMENT .....................................................................................................11

1.2. AUDIENCE FOR THIS DOCUMENT ..............................................................................................11

1.3. AUTHOR CONTACTS................................................................................................................12

2. RELATED DOCUMENTS ............................................................................................................12

2.1. REFERENCE DOCUMENTS ........................................................................................................12

3. SYSTEM PARAMETER ...............................................................................................................13

3.1. FRAME/SLOT STRUCTURE .....................................................................................................13

3.2. SYSTEM PARAMETERS .............................................................................................................13

4. OVERVIEW ..................................................................................................................................13

4.1. RECEIVER STRUCTURE.........................................................................................................13

4.1.1 Pilot blocks ....................................................................................................................14 4.1.2 data blocks ....................................................................................................................14

4.2. TRANSMITTER STRUCTURE ......................................................................................................15

4.3. METHODOLOGY TO DETERMINE BITWIDTHS FOR BLOCK INTERFACES ..........................................15

5. FRONT END ................................................................................................................................16

5.1. INPUT DATA ............................................................................................................................16

5.2. RSSI COMPUTATION ...............................................................................................................17

5.3. ANTENNA-PATH FAILURE DETECTION AND HANDLING..................................................................20

5.4. CP REMOVAL..........................................................................................................................21

5.5. 7.5KHZ FREQUENCY OFFSET COMPENSATION ..........................................................................22

5.6. FFT AND SUB-CARRIER DEMAPPING .........................................................................................23

5.7. POST-FFT AGC.....................................................................................................................26

6. PUSCH SIMO...............................................................................................................................30

6.1. PILOT CHANNEL ESTIMATION....................................................................................................30

6.1.1 Reference signal compensation....................................................................................31 6.1.2 Reference sequence generation ...................................................................................32 6.1.2.1 generation of the cyclic shift exponentials.................................................................33 6.1.2.2 Generation of MnNnxnr qvu <≤= 0),mod()( RS

ZC, ..........................................34

Computer generated sequences ..............................................................................................34 6.2. SYNCHRONIZATION .................................................................................................................35

6.2.1 Carrier Frequency Offset (CFO) processing .................................................................35 6.2.1.1 Frequency offset estimation ......................................................................................35 6.2.1.1.1 algorithm description .................................................................................................35 6.2.1.1.2 case of ONE TTI latency (NOT implementED)..........................................................38 6.2.1.1.3 case of two TTI latency ( CURRENT implementation) ..............................................40 6.2.1.1.4 fixed point implementation.........................................................................................43 6.2.1.2 Frequency offset compensation ................................................................................53 6.2.2 Timing offset estimation ................................................................................................58

6.3. MEASUREMENTS.....................................................................................................................75

6.3.1 Long term averaging .....................................................................................................75 6.3.2 Noise and power estimation..........................................................................................78 6.3.2.1 introduction ................................................................................................................78 6.3.2.2 noise variance estimation for scheduled PRB...........................................................78 6.3.2.3 noise variance estimation for un-scheduled PRB......................................................83 6.3.2.4 instantaneous power estimation ................................................................................84 6.3.2.5 Noise and power absolute values (ecem only)..........................................................85 6.3.3 metrics exchange between FPGA and DSP .................................................................86 6.3.4 metrics used in L1 processing for demodulation...........................................................92 6.3.5 Speed estimation...........................................................................................................93 6.3.5.1 Theoretical background .............................................................................................93 6.3.5.2 practical implementation............................................................................................94 6.3.6 DTX detection................................................................................................................98 6.3.7 Post iDFT SNR estimation ............................................................................................99 6.3.7.1 Theoretical computation ..........................................................................................100 6.3.7.2 algorithm simplifications ..........................................................................................102 6.3.7.3 fallback solution .......................................................................................................103

6.4. CHANNEL ESTIMATION...........................................................................................................103

6.4.1 Frequency domain filtering..........................................................................................103 6.4.2 Time domain filtering...................................................................................................107 6.4.2.1 Algorithm overview ..................................................................................................107 6.4.2.2 Filters coefficients computation ...............................................................................109 6.4.2.3 Filter application ......................................................................................................113

6.5. DEMODULATION ....................................................................................................................115

6.5.1 Frequency domain equalizer.......................................................................................115 6.5.2 iDFT.............................................................................................................................124 6.5.3 QAM demapping and LLR computation......................................................................128 6.5.4 De-interleaving, de-rate matching and H-ARQ recombination ...................................134 6.5.5 Decoder.......................................................................................................................134

7. PUCCH PROCESSING: ACK-NACK AND SR .........................................................................135

7.1. CHANGES WITH RESPECT TO LA2.0........................................................................................135

7.2. GENERAL CONSIDERATIONS ...................................................................................................135

7.3. FREQUENCY OFFSET ESTIMATION ..........................................................................................136

7.4. FREQUENCY OFFSET COMPENSATION.....................................................................................136

7.5. TIMING OFFSET ESTIMATION ..................................................................................................137

7.6. TIMING OFFSET COMPENSATION.............................................................................................137

7.7. ORDER OF FRONT END PROCESSING .....................................................................................138

7.8. CHANNEL ESTIMATION AND ACCUMULATION ON RS.................................................................138

7.9. CHANNEL ESTIMATION AND ACCUMULATION ON DATA ..............................................................141

7.9.1 data accumulation .......................................................................................................141 7.9.2 data Descrambling ......................................................................................................143

7.10. ACK-NACK DETECTION ..........................................................................................................143

7.10.1 algorithm description ...................................................................................................143 7.11. SR DETECTION .....................................................................................................................146

7.12. NOISE ESTIMATION................................................................................................................149

7.12.1 Noise estimation for Ack-Nack ....................................................................................149 7.12.2 empty pucch ................................................................................................................152 7.12.3 combining noise estimates from both pucch...............................................................153 7.12.4 long term averaging of noise estimates ......................................................................153

7.13. POWER ESTIMATION..............................................................................................................154

7.14. ANTENNA-PATH FAILURE DETECTION AND HANDLING FOR ACK-NACK ON PUCCH.....................155

8. PUCCH PROCESSING: CQI AND CQI&ACK-NACK ..............................................................156

8.1. CHANGES WITH RESPECT TO LA2.0........................................................................................156

8.2. GENERAL CONSIDERATIONS ...................................................................................................156

8.3. FREQUENCY OFFSET ESTIMATION ..........................................................................................158

8.4. FREQUENCY OFFSET COMPENSATION.....................................................................................158

8.5. TIMING OFFSET ESTIMATION ..................................................................................................158

8.6. TIMING OFFSET COMPENSATION.............................................................................................158

8.7. ORDER OF FRONT END PROCESSING .....................................................................................158


8.8.1 Channel Estimation: ACK NACK & CQI Mux..............................................................162 8.9. DATA ESTIMATION .................................................................................................................162

8.9.1 CQI only ......................................................................................................................162 The parameter “ o ” defines the size of the shift chosen in order to get 11 bits at the output. ...164 8.9.2 ACK NACK & CQI .......................................................................................................164

8.10. NOISE ESTIMATION................................................................................................................166

8.10.1 CQI or CQI&ACK/NACK .............................................................................................166 8.11. POWER ESTIMATION..............................................................................................................169

8.12. DATA DECODING ...................................................................................................................169

8.12.1 descrambling ...............................................................................................................169 8.12.2 decoding......................................................................................................................170 8.12.3 Reliability Metrics for CQI............................................................................................170

9. UPLINK CONTROL CHANNELS ON PUSCH ..........................................................................170

9.1. CHANGES WITH RESPECT TO LA2.0 .......................................................................................170

9.2. NUMBER OF QAM SYMBOLS OCCUPIED BY RI AND ACK-NACK.................................................170

9.3. SYMBOL EXTRACTION AND LLR COMPUTATION........................................................................172

9.4. CASE OF ONE BIT TRANSMISSION ...........................................................................................172

9.4.1 DTX Detection .............................................................................................................173 9.4.2 Ack-nack and RI bit detection .....................................................................................178

9.5. CASE OF TWO BITS TRANSMISSION.........................................................................................178

9.5.1 coding scheme ............................................................................................................178 9.5.2 soft detection ...............................................................................................................179

10. CQI ON PUSCH .........................................................................................................................188

10.1. CHANGES WITH RESPECT TO LA2.0 .......................................................................................188

10.2. SYMBOLS EXTRACTION ..........................................................................................................188

10.3. NUMBER OF QAM SYMBOLS OCCUPIED BY CQI ......................................................................188

10.3.1 General considerations ...............................................................................................188 10.3.2 limitations from the DSP/FPGA B interface ................................................................189 10.3.3 Number of CQI bits for the different modes ................................................................189 10.3.3.1 aperiodic Cqi............................................................................................................189 10.3.3.2 periodic Cqi..............................................................................................................190

10.4. CQI RATE DE-MATCHING .......................................................................................................190

10.5. CQI DECODING : CASE OF 12BITS OR MORE............................................................................192

10.6. CQI DECODING : CASE OF 11BITS OR LESS .............................................................................194

10.6.1 decoding......................................................................................................................194 10.6.2 Reliability Metrics ........................................................................................................197

11. SOUNDING REFERENCE SIGNALS: ECEM IMPLEMENTATION .........................................198

11.1. INTRODUCTION......................................................................................................................198

11.2. SRS STRUCTURE..................................................................................................................198

11.2.1 general assumptions ...................................................................................................198 11.2.2 User multiplexing strategy...........................................................................................200 11.2.3 sequence generation...................................................................................................201 11.2.4 cyclic shift based user indexing...................................................................................202 11.2.5 time domain user distribution ......................................................................................203

11.3. SRS SEPARATION.................................................................................................................204

11.3.1 algorithm description ...................................................................................................204 11.3.2 CAZAC Compensation................................................................................................205 11.3.3 zero padding................................................................................................................206 11.3.4 IDFT ............................................................................................................................206 11.3.5 time domain filtering ....................................................................................................207 11.3.5.1 reduced filter length .................................................................................................207 11.3.5.2 margin for timing offset error ...................................................................................209 11.3.6 time domain noise removal .........................................................................................209 11.3.7 time domain cyclic shift ...............................................................................................212 11.3.8 IDFT/DFT scaling ........................................................................................................213 11.3.9 zero removal................................................................................................................214

11.4. POST-DFT AGC ..................................................................................................................215

11.5. SYNCHRONIZATION................................................................................................................215

11.6. SNR ESTIMATION..................................................................................................................216

11.6.1 noise power estimation ...............................................................................................216 11.6.2 Signal power estimation ..............................................................................................216

11.7. SRS ABSOLUTE POWER ........................................................................................................220

11.8. SRS CHANNEL AMPLITUDE ESTIMATION..................................................................................221

11.9. SRS FREQUENCY DOMAIN CORRELATION ...............................................................................222

12. SOUNDING REFERENCE SIGNALS: BCEM IMPLEMENTATION .........................................223

12.1. INTRODUCTION......................................................................................................................223

12.2. BCEM VS ECEM IMPLEMENTATIONS .......................................................................................223

12.3. CAZAC COMPENSATION .................................................................................................223

12.4. ZERO PADDING......................................................................................................................224

12.5. IDFT .....................................................................................................................................224

12.6. TIME DOMAIN FILTERING ........................................................................................................225

12.7. TIME DOMAIN NOISE REMOVAL ...............................................................................................226

12.8. DFT ......................................................................................................................................227

12.9. IDFT/DFT SCALING...........................................................................................................227

12.10. SYNCHRONIZATION............................................................................................................228

12.11. POST-DFT AGC...............................................................................................................229

12.12. SIGNAL POWER ESTIMATION...............................................................................................230

12.13. SRS CHANNEL AMPLITUDE ESTIMATION ..............................................................................233

12.14. SRS FREQUENCY DOMAIN CORRELATION............................................................................234

13. RACH PROCESSING ................................................................................................................235

13.1. OVERVIEW............................................................................................................................235

13.2. SIGNAL STRUCTURE .............................................................................................................235

13.2.1 Random Access Preamble Formats ...........................................................................235 13.2.2 Baseband RACH Signal ..............................................................................................237 13.2.3 Large Cell Operation ...................................................................................................238 13.2.4 Random Access Burst Configuration ..........................................................................238 13.2.5 Operation in High Mobility Environment......................................................................240 13.2.6 High Level Requirement..............................................................................................241

13.3. RACH PROCESSING.........................................................................................................242

13.3.1 RACH Front End Processing ......................................................................................242 13.3.2 Inputs to RACH Front End Processing........................................................................244 13.3.3 Frontend Bit Selection.................................................................................................245 13.3.4 Max Magnitude CALCULation.....................................................................................246 13.3.5 Frequency Shift to Baseband......................................................................................246 13.3.6 Filtering and Decimation .............................................................................................247 13.3.7 Summary of Filtering Parameters ...............................................................................258 13.3.8 Sequence Separation for Format 2 .............................................................................263 13.3.9 Pre 2048-point FFT Scaling ........................................................................................263 13.3.10 2048-point Forward FFT..........................................................................................264 13.3.11 ZC Sequence Extraction..........................................................................................265 13.3.12 Scaling of Extracted ZC Sequence..........................................................................266

13.4. RACH BACK END PROCESSING ............................................................................................268

13.4.1 Overview of Back End Processing ..............................................................................270 13.4.2 Generate Frequency Domain Root ZC Sequence ......................................................270 13.4.3 Multiply RACH Signal with ZC Sequence ...................................................................271 13.4.4 1024-Point FFT ...........................................................................................................272 13.4.5 Rescale/Equalize.........................................................................................................272 13.4.5.1 Rescale for N=2 case ..............................................................................................272 13.4.5.2 Rescale for N=4 case ..............................................................................................273 13.4.5.3 Rescale for N=8 case ..............................................................................................274 13.4.6 Compute Metrics .........................................................................................................275 13.4.6.1 Metrics for Format 0,1 .............................................................................................275 13.4.6.2 Metrics for Format 2 ................................................................................................277 13.4.7 Compute Histogram ....................................................................................................278 13.4.8 Analyze CDF ...............................................................................................................279 13.4.9 Handle Special cases..................................................................................................280 13.4.10 Compute FIRst Quartile...........................................................................................280 13.4.11 Compute Threshold .................................................................................................282 13.4.12 Compute Reciprocal of Threshold...........................................................................283 13.4.13 Peak Detection and Scaling ....................................................................................284 13.4.14 Other Rach Outputs.................................................................................................286

13.5. LOW MOBILITY THRESHOLD CALCULATIONS ..........................................................................287

13.5.1 Theory of Threshold Calculation .................................................................................287 13.5.2 The N = 1 Case ...........................................................................................................288 13.5.3 The N = 2 Case ...........................................................................................................289 13.5.4 Then N=4 Case ..........................................................................................................291 13.5.5 The N=8 Case ............................................................................................................294 13.5.6 Calculation of 1st Quartile Value..................................................................................296 13.5.7 Threshold Scaling........................................................................................................298 13.5.8 Summary of Threshold Calculations ...........................................................................298

13.6. INTERPRETATION OF RACH TIME DELAY OFFSET...................................................................300

13.6.1 Time Delayed RACH Signal........................................................................................300 13.6.2 Time Delay Implementation Issues .............................................................................301 13.6.3 Example of RACH Correlation ....................................................................................302 13.6.4 Setting the RACH Search Window..............................................................................303 13.6.5 Calculation of Search Window Size ............................................................................304 13.6.6 Summary of Setting the RACH Search Window.........................................................305

13.7. SIGNATURE DETECTION ........................................................................................................306

13.8. CALCULATION OF N-AVERAGE DETECTION THRESHOLD..........................................................306

14. PUSCH MIMO............................................................................................................................311

14.1. CHANGES WITH RESPECT TO PREVIOUS RELEASE ...................................................................311

14.2. REFERENCE SIGNALS MULTIPLEXING ......................................................................................311

14.3. REFERENCE SIGNALS DEMULTIPLEXING ..................................................................................312

14.3.1 algorithm Overview......................................................................................................312 14.3.2 zero padding................................................................................................................313 14.3.3 idft / DFT......................................................................................................................314 14.3.4 time domain filtering ....................................................................................................317 14.3.5 time domain cyclic shift ...............................................................................................318 14.3.6 DFT .............................................................................................................................319 14.3.7 zero removal................................................................................................................319

14.4. SYNCHRONIZATION...............................................................................................................320

14.4.1 Carrier Frequency Offset (CFO) processing ...............................................................320 14.4.1.1 Frequency offset estimation ....................................................................................320 14.4.1.2 Frequency offset compensation for pilot blocks ......................................................320 14.4.2 Timing offset estimation ..............................................................................................320


14.6. MEASUREMENTS ...................................................................................................................321

14.6.1 noise estimation for FDE.............................................................................................321 14.6.2 noise estimation for time domain MMSE channel estimation .....................................324

14.7. DEMODULATION....................................................................................................................327

14.7.1 Joint frequency domain equalization and CFO compensation for data blocks...........327

15. L1/L2 INTERFACE ....................................................................................................................338

15.1. L1/L2 INTERFACE FOR DL SCHEDULING .................................................................................339

15.2. L1/L2 INTERFACE FOR UL SCHEDULING .................................................................................341

16. TRANSMITTER ALGORITHMS ................................................................................................344

16.1. OVERVIEW............................................................................................................................344

16.1.1 Chapter organization...................................................................................................344 16.1.2 Physical Layer Parameters .........................................................................................344 16.1.3 Physical Channels And Signals ..................................................................................345

16.2. ELEMENTARY ENGINES..........................................................................................................345

16.2.1 CRC Encoding.............................................................................................................345 16.2.2 Code block Segmentation ...........................................................................................346 16.2.3 Channel encoding .......................................................................................................346 16.2.4 Rate matching .............................................................................................................347 16.2.5 Code block concatenation...........................................................................................347 16.2.6 Scrambling ..................................................................................................................347 16.2.7 Modulation...................................................................................................................348 16.2.8 Layer mapping.............................................................................................................350 16.2.9 Precoding ....................................................................................................................351 16.2.10 Power scaling ..........................................................................................................357 16.2.11 IFFT processing.......................................................................................................361

16.3. TRANSPORT CHANNELS PROCESSING.....................................................................................362

16.3.1 DL-SCH and PCH processing.....................................................................................362 16.3.2 DCI processing............................................................................................................363 16.3.3 BCH processing ..........................................................................................................365 16.3.4 CFI processing ............................................................................................................366 16.3.5 HI processing ..............................................................................................................366

16.4. PHYSICAL CHANNELS PROCESSING ........................................................................................366

16.4.1 PDSCH processing .....................................................................................................366 16.4.2 PDCCH processing .....................................................................................................368 16.4.3 PBCH processing........................................................................................................370 16.4.4 PCFICH processing.....................................................................................................373 16.4.5 PHICH processing.......................................................................................................373

16.5. SIGNALS PROCESSING...........................................................................................................376

16.5.1 Reference signals processing .....................................................................................376 16.5.2 Positioning Reference signals processing ..................................................................376 16.5.3 Synchronization signals processing............................................................................376

17. OPEN ISSUES & FURTHER STUDIES ....................................................................................381

17.1. OPEN ISSUES ...................................................................................................................381

17.2. FURTHER STUDIES (POST DROP 1)...................................................................................381

18. ANNEX .......................................................................................................................................381

ANNEX 1. CALCULATION OF COFF................................................................................................381

ANNEX 2. PROOF FOR EQUATION (2) .........................................................................................382

ANNEX 3. CAZAC SEQUENCES ....................................................................................................382

ANNEX 4. TABLE OF COMPLEX EXPONENTIAL WITH THE RESIDUAL FREQ UENCY OFFSET399

ANNEX 5. PRE STORED FILTER FOR T-MMSE PUSCH .............................................................399

ANNEX 6. PRE STORED FILTER FOR T-MMSE PUCCH .............................................................402

ANNEX 7. SINCOS TABLE (CFO) FOR SIMO, COMMON FOR ALL BANDWID THS..................406

ANNEX 8. SINCOS TABLE (CFO) FOR MIMO, COMMON FOR ALL BANDWID THS .................406

ANNEX 9. C(K) , 5 TAPS .................................................................................................................407

ANNEX 10. SYNCHRO SEQUENCES FIX POINT IMPLEMENTATION ......................................420

ANNEX 11. DONWLINK FIX POINT POWER OFFSETS .............................................................421

ANNEX 12. LUT FOR 7.5KHZ COMPENSATION .........................................................................425

ANNEX 13. POWER COMMANDS MEDIATION IN IBTS .............................................................434

CASE 5 MHZ / 10 MHZ / 15MHZ / 20MHZ BANDWIDTH ...................................................................434

SUMMARY: ..........................................................................................................................................437

19. ABREVIATIONS AND DEFINITIONS .......................................................................................439

19.1. ABREVIATIONS ......................................................................................................................439

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Retrait : Suspendu : 0,13 cm, Numéros + Niveau : 1 + Style de numérotation : 1, 2, 3, … + Commencer à : 1 + Alignement : Gauche + Alignement : 2,5 cm + Tabulation après : 3,13 cm + Retrait : 3,13 cm

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Couleur de police : Noir

Page 260: [4] Supprimé hbilel 21/12/2010 11:16:00

Table 13-10 – Variable Format 0 Format 2 k1 192 384


Table 13-10 – Upsampling Parameters for 3MHz Variable Format 0 Format 2 k1 384 768


Table 13-14 – Filtering Parameters Common to Format 0 Variable 1.4 MHz 3 MHz 5 MHz 10 MHz 15 MHz 20 MHz D ---- ---- 3 2 3 4 Ntaps ---- ---- 25 7 15 21 L ---- ---- 12 3 7 10 Nout ---- ---- 2048 6166 6166 6166 Nin 1542 3083 6166 12337 18510 24681

Table 13-14 – Filtering Parameters Common to Format 2

Variable 1.4 MHz 3 MHz 5 MHz 10 MHz 15 MHz 20 MHz D ---- ---- 3 2 3 4 Ntaps ---- ---- 25 7 15 21 L ---- ---- 12 3 7 10 Nout ---- ---- 4096 12310 12310 12310 Nin 3078 6155 12310 24625 36942 49257


Table 13-14 – Filtering Parameters Common to Format 0 Variable 5 MHz 10 MHz 15 MHz 20 MHz D 3 2 3 4 Ntaps 25 7 15 21 L 12 3 7 10 Nout 2048 6166 6166 6166 Nin 6166 12337 18510 24681

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