ieee/asme transactions on mechatronics 1 analysis and...

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IEEE/ASME TRANSACTIONS ON MECHATRONICS 1

Analysis and Development of Novel Three-PhaseHybrid Magnetic Paths Switched Reluctance

Motors Using Modular and SegmentalStructures for EV Applications

Wen Ding, Member, IEEE, Yanfang Hu, and Luming Wu

Abstract—The classical switched reluctance motors (SRMs) of-ten suffer from drawbacks such as low power and torque densi-ties, high torque ripple, mutual coupling, etc., which limit theirindustrial applications. This paper presents the analysis and de-velopment of two novel three-phase SRMs with hybrid magneticpaths comprising six E-shaped modular stators and three segmen-tal common rotors, termed as the modular SRMs (MSRMs), forelectric vehicle applications. The machine topologies with differ-ent winding arrangements are described. The voltage and outputpower equations are analytically derived, and some design par-ticularities and parameters are discussed. The field distributionsand static magnetic characteristics of an MSRM with double coilare analyzed by using 3-D finite-element method. After that, twoMSRMs with different winding arrangements, namely a double-coil MSRM and single-coil MSRM, are analyzed and compared toevaluate the distinct features of this novel MSRM, accompaniedwith a classical three-phase 6/4 SRM. The comparison includesstatic magnetic characteristics, mass of iron core, normal dynamic,and fault-tolerant performances. It is shown that the double-coilMSRM appears to have better characteristics such as higher torqueproduction capability, lower torque ripple and cost, higher torqueand output power densities, and higher reliability and fault toler-ance. For experimental verification, laboratory testing of a double-coil MSRM is developed, and the simulated and measured staticinductance characteristics and dynamic performances correlatewell.

Index Terms—Double coil, hybrid magnetic paths, single coil,switched reluctance motor (SRM).

I. INTRODUCTION

W ITH the development of electrical and mechanical en-gineering, more and more novel mechatronics systems,

such as carbon nanotube shuttles, brushless dc (BLDC) mo-tors, piezoelectric rotary motor, actuator system, and multidiskspherical electromechanical brake [1]–[5], have been applied for

Manuscript received June 25, 2014; revised September 29, 2014; acceptedNovember 17, 2014. Recommended by Technical Editor S. K. Dwivedi. Thiswork was supported in part by the National Natural Science Foundation ofChina under Grant 51477130, Natural Science Basic Research Plan in ShaanxiProvince under Grant 2014JM7247, and State Key Laboratory of ElectricalInsulation and Power Equipment under Grant EIPE14315.

The authors are with the State Key Laboratory of Electrical Insulation andPower Equipment, School of Electrical Engineering, Xi’an Jiaotong Univer-sity, Xi’an 710049, China (e-mail: [email protected]; [email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TMECH.2014.2383615

industrial purposes. Switched reluctance motor (SRM) is a vari-able reluctance stepping motor that is developed to efficientlyconvert energy. Although the structure of the SRM is similar tothat of the step motor [6], [7], it has the following features thatare different from the step motor: fewer poles; larger steppingangle; usually one tooth per pole, higher power output capability,etc. The SRM is gaining widespread interest as good candidatesfor many applications such as electrical aircraft, electric vehicles(EV), wind turbines, etc., due to its simple structure, rugged-ness, high starting torque, ability of fault tolerance, high-speedoperation, and low manufacturing cost [8]–[10]. However, itsdisadvantages are also obvious: such as low power and torquedensities, high torque ripple, mutual coupling, etc., which limittheir industrial applications.

There are several efficacious methods have been investigatedto improve the SRM drive system and make it more meritoriousto industrial application, which can be sorted into three clas-sifications. The first way is an easy way, which is to increasethe stator/rotor pole or phase numbers in SRMs. In [11]–[13],design considerations and comprehensive evaluations for SRMswith a higher number of rotor poles for the drivetrain have beenpresented. Owing to the higher number of poles, the conduc-tion time in each phase current period is much smaller than theclassical SRMs. The static torque production capability is muchbetter and the torque ripple is lower than the classical SRMs.The second method is to optimize the control strategies for theSRM drive. The torque ripple can be minimized by finding anoptimum current profile, and then, following that through a high-bandwidth current regulator. Various current profiling and phasetorque distribution methods to minimize the torque ripple appearin the literature [14] and [15]. Another method to minimize thetorque ripple is called as the direct instantaneous torque control.It uses novel converter and does not require any current profilesfor shaping the phase currents or torque profiles and commutat-ing waveforms. It only uses simple control schemes of torquehysteresis control to reduce the torque ripple [16], [17]. Thethird method is to employ the redundant technique in the SRMtopology. In [18], a novel dual-channel SRM (DCSRM) wasproposed for high reliability operation by dividing the windingsinto two symmetrical three-phase winding units in an classi-cal 12/8 SRM topology. In this connection, a winding or phasefault in one channel will not influence the operation of the samephase in another channel or of other phases. This DCSRM is aredundant machine, which is suitable for developing integrated

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2 IEEE/ASME TRANSACTIONS ON MECHATRONICS

starter/generator in some safety critical applications such as EV,more electric aircraft and power systems [19]–[21]. The lastway of increasing the applied performance is by means of spe-cial stator or rotor configuration design such as segmental andmultilayer structure. In [22], Mecrow proposed a novel SRMthat combines a segmental rotor with short-pitched windings.The short-pitched windings are placed around a single tooth.The flux linkage of the machine is largely increased. Thus, thetorque density and average torque are also more than a classicalSRM. In [23], a novel segmented SRM with U-shaped coresstator is proposed for safety-critical application. Owing to themodular construction of the U-core stators, the fault tolerance ofthe machine is increased. The independence of the phases is alsoincreased and the shorter magnetic flux paths are formed insidethe machine. It also has less iron losses as compared to the clas-sical SRM because of shorter magnetic flux paths. In [24] and[25], a new topology for the two-phase SRM with segmentedE-core stators was proposed for improved performance, simplestructure, and cost reduction. It has 20% steel savings and higherperformances such as higher power density and efficiency, andlower core losses. In [26], a novel SRM with stepped-skewingrotor is proposed to improve efficiency. The designed SRM isdivided into three stacks, and only rotor is skewed. The skew an-gle has been designed to reduce both of torque ripple and radialforce. In [27] and [28], novel SRMs with different C-core statorsand hybrid magnetic paths were developed for low-cost produc-tion and wind power generation. In comparison with classicalSRMs, they have better characteristics such as higher averagetorque and lower weight of rotor. The phase windings are com-pletely separated and there is no mutual coupling between eachother. More novel designs were focused on the axial-flux SRMs(AFSRMs). In [29], a novel five-phase pancake-shaped AFSRMis designed and prototyped for EV application. The stator of thisAFSRM is comprised of 15 C-cores and the rotor has 12 cubes.It is a good fit in the EV to improve its fault tolerance. In [30], anovel three-phase, 12/16 pole, dual-rotor, and single-stator axialflux segmented rotor SRM is proposed for EV applications. Thenumber of rotor segments is higher than the stator slots, thusthe overall volume of the motor is reduced. The torque den-sity is increased and the winding overhang length is reduced.Some other novel types of multilayer SRM configurations withspecific features that consist of magnetically independent mod-ules or layers for phases were also proposed for higher torqueand efficiency, and lower torque ripple and acoustic noise [31],[32]. In addition, some hybrid reluctance motors such as doublysalient permanent-magnet (PM) motor [33] and flux switchingPM motor [34], [35] that combine the virtues of SRM (simplestructure of the rotor) with the virtues of BLDC motors and syn-chronous PM motors (high power density) are proposed for anattractive alternative in the industry and new EV applications.

This paper presents the analysis and development of two newSRMs with hybrid magnetic paths using modular and segmentalstructures, termed as modular SRMs (MSRMs), for EV appli-cations. Section II gives the MSRM topologies with differentwinding arrangements and some theoretical equations and de-sign particularities. Section III simulated the field distributionand flux linkages of a double-coil MSRM using finite-element

Fig. 1. Machine topologies of the proposed MSRM with different windingarrangements. (a) Single-coil MSRM. (b) Double-coil MSRM.

method (FEM) analysis. The characteristics of two differentMSRMs with single-coil and double-coil arrangements are com-pared with a classical 6/4 SRM to evaluate the distinct featuresof this novel MSRM in Section IV. The experimental valida-tion obtained from a prototype double-coil MSRM are shownin Section V, followed by conclusions in Section VI.

II. MACHINE DESCRIPTION AND DESIGN ANALYSIS

A. Machine Description

This so-called MSRM is a kind of special machine. Fig. 1shows the machine structure and topologies for two three-phaseMSRMs with modular and segmental structures. The stator inthese two MSRMs consists of six separate modular E-shapedcores. The rotor is composed by three separate and magneti-cally isolated segmental common rotors. These three segmentalrotors are placed in one shaft and surrounded by the six annularmodular E-shaped stators. It can be found from Fig. 1 that thelayout and connection of windings in the E-shaped modular sta-tor are different in these two MSRMs. In Fig. 1(a), there is onlyone coil that is twisted in the middle pole of one E-shaped mod-ular stator. This configuration is called as “single-coil” MSRM.The coils are series connection on diametric modular stators toform one-phase windings. In another configuration, there aretwo coils that are twisted in the upper yoke and lower yoke ofeach E-shaped stator, respectively. This configuration is calledas “double-coil” MSRM. The four coils on the E-shaped mod-ular stators are series or parallel connection to form one-phasewindings. Figs. 2 and 3 show the different flux paths of thesetwo MSRMs with single coil and double coil.

In conclusion, the structures of the stator and rotor in thesetwo MSRMs are the same excluding the layout and connectionof the coils. Because of this difference, it shows that the parts ofthe coils in the double-coil MSRM are outside the stators. Onthe other hand, the coils in the single-coil MSRM are twisted inthe middle tooth of the E-shaped modular stator, thus the volumeof the single-coil MSRM is smaller than that of the double-coilMSRM. It may be a good choice for the special applicationswith the strict requirement of space limitations. However, theflexibility of coils arrangement and connection will be increasedand the risk of winding failure may be reduced in the double-coilMSRM thanks to the more coils.


DING et al.: ANALYSIS AND DEVELOPMENT OF NOVEL THREE-PHASE HYBRID MAGNETIC PATHS SWITCHED RELUCTANCE MOTORS 3

Fig. 2. (a) Cut view and (b) front view of the single-coil MSRM with fluxpaths when one phase is excited.

Fig. 3. (a) Cut view and (b) front view of the double-coil MSRM with fluxpaths when one phase is excited.

B. Machine Design Analysis

In general, the voltage and output power equations are firstderived for an SRM design [36]. The area enclosed by the fluxlinkage/current (λ/i) trajectory of an SRM, determines the outputenergy conversion of the motor for a given machine geometryand current control strategy in one period. If the phase resistivevoltage drop in the MSRM winding is neglected, the appliedvoltage in one phase can be expressed as the rate of change offlux linkages, as

V ∼= dλ

dt=

λa − λu

Δt=

(La − Lu )iΔt

(1)

where λa and λu , La and Lu are the flux linkages and in-ductances at the aligned and unaligned positions, respectively.Δt = βs/ωm is the time taken for the rotor to move from theunaligned to aligned position, βs is the stator pole arc in radian,and ωm is the rotor speed in rad/s.

The relationship of the aligned and unaligned inductances inan SRM can be expressed as

δ = La/Lu . (2)

Substituting Δt and (2) into (1), the applied voltage becomes

V =(La − Lu )i

Δt=

Lai(1 − 1/δ)ωm

βs. (3)

Generally, the flux linkage λa at the aligned position is afunction of inductance and current, as

λa = Lai = φN = BAspN (4)

τ

ββ

τ

Fig. 4. Geometry parameters of the proposed MSRM. (a) One stator. (b) Onerotor segment.

where φ, Asp , N, and B are the aligned flux, the area of the statorpole, the number of turns per phase, and the average flux densityat the stator pole face in the MSRM, respectively.

In order to derive the voltage and output power equations,the physical characteristics and key geometric parameters aredefined as shown in Fig. 4. As can be seen that the cross-sectional area for the stator pole Asp in one E-core module isexpressed as

Asp = D(Ls1 + Ls2 + Ls3)βs/2 = DLeqβs/2 (5)

where D is the bore diameter, i.e., the inner diameter of stator,which is equal to 2rsi in Fig. 4 (rsi is the inner radius of stator).Leq is the effective axial stack length of stator poles, which isalso equal to the active stack length of three rotors. It can seenform Figs. 2 and 3 that the magnetic fluxes in the middle toothof one pair of modular stators are always in the same direction,thus the total flux in the middle tooth is also larger than thatof the two end teeth. So in order to keep the flux densities inthe bodies of three rotors/teeth the same, the width of middlerotor/tooth is also designed equal to the sum of those of theupper and lower ones.

Then, the voltage equation for the proposed MSRM can beexpressed as follows by substituting (4) and (5) into (3):

V =Lai(1 − 1/δ)ωm

βs= BDLeqN(1 − 1/δ)ωm /2. (6)

Normally, the stator phase current is relative to the specificelectric loading As , which is expressed as

As =2mNi

πD(7)

where m is the number of conducting phases at the same time.In the case of a 6/4 SRM, it often operates with single-pulse

control. The output power equation at this time is given as

Pd = kekdV · i · m (8)

where ke is the motor efficiency factor, usually in the range of0.8–0.93, and kd is the duty cycle which is given as

kd =θiNphNr

2π(9)

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TABLE IGEOMETRY PARAMETERS OF THE PROTOTYPE MSRM (UNIT: MM)

Stator outer radius, rs o 45 Stator total length, Ls 80Stator yoke width, ys 9 Upper stator pole length, Ls 1 12Rotor yoke width, yr 9 Middle stator pole length, Ls 2 24Stator pole arc, βs 32° Lower stator pole length, Ls 3 12Rotor pole arc, βr 32° Stator pole width, τ s 10.73Airgap length, mm 0.25 Rotor pole width, τ r 10.60

Fig. 5. Magnetization B–H curve used in the 3-D FEA program.

Fig. 6. (a) 3-D FEA meshing model and (b) flux distributions at the alignedposition when one phase is excited.

where θi is the mechanical angle during which the current isflowing through one-phase winding. Nph and Nr are the numberof stator phases rotor poles, respectively.

Finally, the output power equation can be obtained by com-bining the voltage equation (6) and the current equation in (7)into (8) as follows:

Pd = kekdV · i · m = kekd(1 − 1/δ)BAsπD2Leqωm /4(10)

In the aforementioned equation, the active length Leq is oftendefined as a multiple of stator inner diameter D as follows:

Leq = k · D (11)

Substituting (11) into (10), the output power equation is ob-tained as

Pd ∝ k2 · D3 (12)

In the aforementioned equation, D is evaluated if the ratedpower Pd , rated speed ωm , B, As , ke , kd , δ, and k are known.

From the aforementioned equations, the proposed MSRMdesign can be started with iterative process. The geometric pa-rameters of the MSRM prototype are summarized as in Table I.

III. FINITE-ELEMENT MODELING AND SIMULATION OF THE

MSRM WITH DOUBLE COIL

A. Finite-Element Modeling and Magnetic Characteristics

From aforementioned descriptions, there are hybrid magneticpaths with both radial and axial magnetic paths in these twoMSRMs, which result in 3-D field distribution. In order to solvethis problem, it requires an effective way of 3-D FEM analysis tocompute the field distribution and magnetic characteristics in theMSRM design. The FEM analysis package Maxwell 3-D is usedin the magnetic field analysis. It is a well-known software thatcan be used to compute the magnetic vector potential (MVP) onstructures with a complex geometry and with nonlinear magneticmaterial characteristics. In addition, in order to reduce the lengthof this paper, only the 3-D FEM analysis of the double-coilMSRM is studied in this section. The single-coil MSRM andclassical SRM will be investigated in Section IV.

It can be seen from Fig. 3 that there are four coils on the op-posite modular E-shaped stators of one phase; the arrangementof the windings is flexible. In this study, the upper and lower twocoils on the opposing stators are connected to form one winding,respectively. These two windings are placed in parallel to buildup one phase, as shown in Fig. 3(a). When one phase is excited,it means that the four coils are excited simultaneously. Owing toits independent magnetic structure of each phase in the double-coil MSRM, the magnetic structure of one phase is selected asthe solved domain, i.e., only two E-shaped stators with threerotors are used to calculate the MSRM’s entire characteristicsin one half period, where the rotor moves from the unaligned tothe fully aligned positions.

The magnetic material used in the 3-D FEA program of thedouble-coil MSRM is the nonoriented silicon steel DW540-50.The nonlinear magnetization B–H curve of this silicon–iron isshown in Fig. 5 and the lamination thickness of the stator androtor is 0.5 mm. The relative magnetic permeability (RMP) ofthe air region is set to 1. And the RMP of the copper windingand aluminum shaft in this double-coil MSRM are also set to1. With the given geometric parameters, the 3-D FEM meshingmodel of the double-coil MSRM and its flux distributions atthe aligned position are shown in Fig. 7, where only one phaseis excited at iA1 = iA2 = 2.5 A. In this 3-D FEM analysis,the elements are meshed as tetrahedron defined by four nodes.Maxwell software modeled the potential in each element as annth polynomial in a given space coordinate. The MVP of eachelement is obtained in the 3-D FEA of the MSRM. In general, theaccuracy and the computation times of this solution are relativeto the field attribute, the quantity of the meshed elements and theorder of the polynomial. Maxwell provides a smart size optionto discretize the node elements. In this smart operation, one canidentify and refine the areas of the mesh with a higher degreeof changing the magnetic quantities through an adaptive andautomatically process. Owing to the adaptive mesh refinement,the quantity and the number of node elements generated for eachexcitation current or for the same excitation current at differentrotor positions will be also different. For example, the meshconsists of 398250 tetrahedrons in the case of 2.5-A excitationcurrent at the aligned position. However, in the same case of

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Fig. 7. Flux distributions contours at the (a) aligned position and (b) unalignedpositions when one phase is excited.

2.5-A current at the unaligned position, the mesh consists ofonly 345725 tetrahedrons. Obviously, the hybrid magnetic pathswith axial and radial fluxes in the stator and rotor appear in the3-D flux distribution, as shown in Fig. 6.

Additionally, the magnetic flux distribution contours withiA1 = iA2 = 2.5 A at the unaligned and aligned positions areshown in Fig. 7. It can be seen that the double-coil MSRM is ableto produce large flux density in the stator and rotor at this currentexcitation. For example, the maximum flux density in the statoryoke is about 2.0 T at the aligned position, and the magnitude ofthe flux density in the same area is about 0.8 T at the unalignedposition. On the another hand, the maximum flux densities instator/rotor poles when the rotor moves from unaligned positionto fully aligned position are increased from about 0.4 to 1.4 T atthis current. It means that the power of generation force betweenthe stator and rotor poles could develop high torque value.

Generally, the phase flux linkage and inductance of an SRMare a function of excitation current and rotor position owingto the doubly salient structure and magnetic saturation. At thealigned position or at higher current levels, the magnetic mate-rial in the rotor and stator poles is easy to become to be saturated.The self flux linkage curves of winding-A1 obtained from the3-D simulations with respect to the current levels at differentrotor positions when only winding-A1 is excited are shown inFig. 8(a). It consists of 16 curves, corresponding to the vari-ous rotor positions between 0° and 45° with step of 3°. Thismagnetostatic problem is solved for 16 rotor positions and for15 current levels from 0 to 3.5 A in step of 0.25 A for eachposition, i.e., 240 simulations are carried out. Accordingly, thewinding-A1 flux linkage curves when winding-A1 and A2 areboth excited simultaneously are shown in Fig. 8(b). The exci-tation current of x-coordinate in Fig. 8(b) means each winding

Fig. 8. Comparisons of winding-A1 flux linkage characteristics under differ-ent excitations. (a) Single-winding excitation. (b) Two-winding excitation.

TABLE IICOMPARISON OF WINDING FLUX LINKAGE AT DIFFERENT CURRENTS

(UNIT: WB)

Currents 1 A 1.5 A 2 A 2.5 A 3 A

Winding fluxlinkage undersingle-windingexcitation at 0°

0.024 0.036 0.048 0.060 0.073

Winding fluxlinkage undertwo-windingexcitation at 0°

0.017 0.025 0.033 0.042 0.050

Winding fluxlinkage undersingle-windingexcitation at22.5°

0.090 0.103 0.109 0.113 0.117

Winding fluxlinkage undertwo-windingexcitation at22.5°

0.071 0.097 0.103 0.108 0.113

current (A1 or A2), that is half of the phase current. Table II givesthe winding-A1 flux linkages with different excitation modes atdifferent current levels. As can be seen that the flux linkage ofwinding-A1 under simultaneous two-winding (A1 and A2) ex-citation is lower than that under single-winding (A1) excitation.Because the magnetic fluxes in the upper and lower yokes suchas regions 1, 3 and 2, 4 produced by the currents of windingsA1 and A2 are in the opposite directions under two-windingexcitation, so their fluxes in these two windings are partiallyoffset. For example, the self-flux linkage λA1 at iA1 = 1 Aand iA1 = 2 A under single-winding excitation at the alignedposition are 0.090 and 0.109 Wb, respectively, Whereas, the



Fig. 9. Comparison of static torque characteristics under different excitations.(a) Single-winding excitation. (b) Two-winding excitation.

TABLE IIICOMPARISON OF STATIC TORQUE FOR DIFFERENT CURRENTS (UNIT: N·M)

Currents 1 A 1.5 A 2 A 2.5 A 3 A

Average torqueundersingle-windingexcitation

0.045 0.085 0.124 0.160 0.188

Average torqueundertwo-windingexcitation

0.063 0.139 0.230 0.318 0.384

Maximumtorque undersingle-phaseexcitation at 15°

0.076 0.172 0.304 0.446 0.507

Maximumtorque undertwo-phaseexcitation at 15°

0.100 0.226 0.401 0.627 0.858

flux linkage λA1 at iA1 = iA2 = 1 A and iA1 = iA2 = 2 A un-der simultaneous two-winding excitation at the aligned positionare 0.071 and 0.103 Wb, respectively. λA1,max at iA1 = 3.5A under single-winding excitation at the unaligned position is0.085 Wb, which is 146.5% of λA1,max at iA1 = iA2 = 3.5 Aunder two-winding excitation.

Similarly, the static torque is also in terms of rotor posi-tion and phase current owing to the doubly salient structureand magnetic saturation. The static torque characteristics of thedouble-coil MSRM under single- and two-winding excitationsare calculated by the 3-D model, as shown in Fig. 9. This pro-file consists of 15 curves, corresponding to the various currentsbetween 0.2 and 3 A with step of 0.2 A. And Table III shows

Fig. 10. (a) 3-D-FEA model of one phase. (b) Drive circuit in analysis.

the average and maximum torques during one excitation cycleunder single-winding and two-winding excitation modes. It canbe found that the maximum torque appears at around 15° rotorposition. The total and maximum torque produced by wind-ings A1 and A2 with two-winding excitation is more than twotimes of the winding A1 torque at the same currents. For exam-ple, the average torque Tav ,A1 produced by one winding withiA1 = 2A is 0.124 N·m, which is 53.9% of Tav ,A1A2 producedby two windings with iA1 = iA2 = 2 A. The maximum torquesat 3 A under single- and two-winding excitations are 0.501 and0.858 N·m, respectively.

B. Dynamic Behavior Performed by 3-D FEA

Owing to the highly nonlinear nature and hybrid magneticfluxes in the MSRM, 3-D FEA is found to be an optimum so-lution. However, 3-D FEA dynamic simulation process usuallyconsumes significant computational time. In order to reduce the3-D FEA model size and save the computational times, onlyone-phase computation of the double-coil MSRM is performedin this simulation. This processing is reasonable because anyphase E-core magnetic circuit is completely separated from theother phase E-cores. Due to the sufficient spacing between anytwo phases, the mutual coupling between adjacent phases can,therefore, be neglected. A classical single-phase asymmetricbridge converter is linked to the 3-D FEA model, which is usedto provide the electric excitation to the phase winding on theE-core stators. This operation is based on the single-pulse con-trol that usually applies to an SRM. It should be noted that inthis simulation the upper and lower four coils on the opposingstators are divided in two windings and connected in parallelto form one phase, as shown in Fig. 3(a). Accordingly, the 3-DFEA model of one phase and the analysis circuit of single-phaseasymmetric bridge converter are shown in Fig. 10. The dc-busvoltage is applied across the phase windings, and the voltageequation of the single phase can be given by

V = rph · i +dλ(i, θ)

dt(13)

where rph denotes the phase resistance and λ denotes theflux linkage. The flux linkage is a nonlinear function of ro-tor position and phase current, as shown in Fig. 8. It can also



Fig. 11. Comparison of 3-D FEA results obtained by varying the firing angle positions in accordance with different operation speeds. (a) Phase current. (b) Phasetorque. (c) Energy conversion loop.

TABLE IVTURN-ON AND TURN-OFF ANGLES FOR VARIOUS OPERATION SPEEDS

Speed (r/min) 800 1200 1600 2000

Turn-on angle (Deg) 2 0 −2 −4Turn-off angle (Deg) 25 30 32 34

be expressed as the product of phase inductance and current,i.e., λ = L · i. Thus, the voltage equation (13) can also be de-scribed by the phase inductance L and the angular velocity ωas

V = rph · i + L(i, θ) · di

dt+ i · ω · dL(i, θ)

dθ. (14)

In this equation, the phase voltage consists of a resis-tive drop, an inductive drop, and a back electromotive force(EMF).

Furthermore, the electromagnetic torque produced by onephase can be derived from coenergy of that phase as

Tph =∂W (i, θ)

∂θ|i=const =

∂∫ i

0 λ(i, θ)di

∂θ(15)

where W is the coenergy.The simulated phase current and phase torque at various op-

eration speeds are shown in Fig. 11(a) and (b). The operationspeeds and turn-on and turn-off angles were given in Table IV.These analyses are carried out about 60 mechanical degrees andit takes several days to finish these 3-D FEA computations. Ac-cordingly, the different energy conversion loops in one phasecycle are shown in Fig. 11(c). It can be seen that at low speedssuch as 800 r/min, the phase current rises up rapidly to reach ahigh level. This is because that the back EMF is very small andmost of the dc-bus voltage is applied to the resistance and in-ductance in this case. A relatively larger coenergy area enclosedby the current trajectory will also be resulted in the magnetiza-tion characteristics. Accordingly, the torque will also reach thehigher possible level. In this case, the current has to be regu-lated or limited in order to get the desired torque at low speed.At high speeds such as 2000 r/min, the back EMF is increased,thus there is no need to limit the current. On the contrary, the

TABLE VCOMPARISON OF DIMENSIONS OF THREE TYPES OF SRMS

DimensionClassical

SRM

Double-coil

MSRM

Single-coil

MSRM

Stator outer diameter(mm)

90 90 90

Stator inner diameter(mm)

40.5 40.5 40.5

Stator yoke width (mm) 9 9 9Rotor yoke width (mm) 9 9 9Stator and rotor pole arcs 32°/32° 32°/32° 32°/32°Stator pole length (mm) 48 12,24,12 12,24,12Rotor length (mm) 48 12,24,12 12,24,12Volume (mm × mm) φ80 ×

62φ96 ×

80φ80 ×

80Minimum airgap (mm) 0.25 0.25 0.25Coil turns per pole 100 300 300Stator yoke mass (kg) 0.857 0.381 0.381Stator iron mass (kg) 1.25 0.776 0.776Rotor iron mass (kg) 0.304 0.304 0.304Winding mass (kg) 0.213 0.246 0.209Total machine mass (kg) 1.767 1.326 1.289

turn-on angle should be advanced in order to allow the currentto reach its desired value before being limited by the back EMF.

IV. DISTINCT FEATURES OF MSRM AND COMPARISON WITH

CLASSICAL SRM

In the previous section, the static and dynamic character-istics of the double-coil MSRM are analyzed by using the3-D FEM. These design and analysis techniques can be em-ployed within other modular E-shaped SRMs. In this section,a classical 6/4 SRM and two MSRMs with double coil andsingle coil are analyzed and compared using the FEM. Themain dimensions and some mass characteristics are presented inTable V.

A. Magnetic Circuit and Other Characterizations

First, the flux distributions of the classical 6/4 SRM and thesingle-coil MSRM when one phase is excited at the alignedposition are simulated in Fig. 12 to compare with that of thedouble-coil MSRM. It is evident that the flux path in the classical

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Fig. 12. Flux distribution. (a) Classical 6/4 SRM, (b) single-coil MSRM.

TABLE VICOMPARISON OF STATIC AVERAGE AND MAXIMUM TORQUES FOR THREE

SRMS AT DIFFERENT CURRENT LEVELS (UNIT: N·M)

Currents Classical SRM Double-coil MSRM Single-coil MSRM

Ave. Max. Ave. Max. Ave. Max.

i = 1A 0.027 0.045 0.068 0.099 0.064 0.101i = 1.5A 0.061 0.102 0.139 0.222 0.142 0.223i = 2A 0.109 0.182 0.230 0.394 0.230 0.397i = 2.5A 0.170 0.282 0.321 0.603 0.324 0.611i = 3A 0.245 0.404 0.407 0.835 0.421 0.837

SRM is a long flux path pattern, which is different with theMSRMs. From Table VI, it can be found that the outer diameterof the double-coil MSRM is larger than those of other twoSRMs because the coils of the double-coil MSRM are twistedin the yokes of E-shaped stators axially and parts of the coilsare outside the stators. The axial lengths of the two MSRMsare also longer than that of the classical SRM because of thedistance between two neighboring rotor segments. Thus, thevolume of the classical SRM looks like smaller than those oftwo MSRMs, but the real material and mass of the stator in theclassical SRM are more than those of two MSRMs. Because afully enclosed stator back iron in these MSRMs is eliminated, itleads to significant cost reduction. For example, the stator yokemass in two MSRMs is only 0.381 kg, which is only 44.5% ofthat in the classical SRM.

The reduction of the iron-core material in the stators of theMSRM can also be explained by the occupied volume of themean flux paths in the iron-core of these SRMs. The meanflux paths of the classical SRM and MSRM when one phase isexcited are shown in Fig. 13. It should be noted that the statorsand rotors in Fig. 13(b) are taken from the side and front viewsfor the 2-D presentation, respectively.

The lengths l1 and l2 are equal in these two motors, respec-tively. The volumes (Vcy and Vmy ) of mean flux-path in thestator yoke of classical SRM and MSRM, as shown in Fig. 13,are

{Vcy = 2π(l1 + l2)yscLsc

Vmy = 4l3τsm ysm

(16)

Fig. 13. Mean flux paths for the (a) classical 6/4 SRM and (b) MSRM.

Fig. 14. Static winding flux linkage characteristics in the (a) classical 6/4SRM and (b) single-coil MSRM.

where Lsc and ysc are the stack length and stator yoke width ofthe classical SRM, respectively, and τsm and ysm are the stackpole width and yoke width of the MSRM.

Additionally, in these two SRMs, l1 = 10.5 mm, l2 = 30 mm,τsm = 11 mm, ysc = ysm = 9 mm, Lsc = 48 mm, l3 = 40 mm,the ratio of volumes for the mean flux paths in stator yoke oftwo SRMs is

Vmy

Vcy=

4l3τsm ysm

2π(l1 + l2)yscLsc≈ 0.145. (17)

Considering the stator yokes of other two phases in theMSRM, it can be obtained that the stator yoke in the MSRMonly utilizes 43.5% of the core material compared to the classi-cal 6/4 SRM, which is coincided with the aforementioned resultin Table V.

Second, the static winding flux linkage characteristics of theclassical SRM and the single-coil MSRM from the unalignedto aligned position are calculated, as shown in Fig. 14. The



Fig. 15. Static torque characteristics in the (a) classical 6/4 SRM, (b) double-coil MSRM, and (c) single-coil MSRM.

Fig. 16. Comparison of static average torque versus current characteristics ofthe single-coil and double-coil MSRMs and classical SRM.

excitation current of x-coordinate represents each winding cur-rent for these two SRMs. It can be seen from Figs. 8 and 14that when one winding carries current over 3 A, the windingflux linkage of the classical SRM becomes saturated around thealigned position, while when the windings carry current over1.5 A, the winding flux linkages of the MSRMs becomes satu-rated around the same position.

Additionally, the static torque characteristics of the classical6/4 SRM and the two MSRMs at five different current levels arecompared in Fig. 15. The average torque–current characteristicsin these three SRMs for one half of the excitation cycle are shownin Fig. 16. It can be seen that the shape of static torque in theMSRM is different with that of the classical one at higher currentlevels. The torque curves of these two MSRMs are almost thesame in most regions, and the maximum and average torquesproduced by them is higher than that of the classical SRM atthe same current. For example, the maximum torque of the twoMSRMs at i = 3 A is near 0.837 N·m, which is more than twotimes of that in the classical SRM at the same excitation. One ofthe primary reasons is that the number of turns-per-pole in theMSRM is three time of that in the classical SRM, which woulddevelop more phase magnetomotive force and higher torque,and also cause the MSRM is more easily saturated at the samelarge current levels. The average and maximum torques of thesethree SRMs at different current levels are compared, as shownin Table VI.

Fig. 17. Comparison of simulated steady-state phase current and torque wave-forms in dynamic operation with single-pulse control at 1500 r/min. (a) Single-coil MSRM. (b) Double-coil MSRM. (c) Classical 6/4 SRM.

B. Normal Dynamic Performance

The normal dynamic performances of the classical SRM andtwo MSRMs are compared at a constant speed of 1500 r/min,which are conducted with single-pulse control. In this case, thedc-link voltage is 35 V, and the turn-on and turn-off anglesare 0° and 33°, respectively. The simulated phase currents andtotal torque waveforms are shown in Fig. 17. It should be noted



TABLE VIISUMMARY OF THE PREDICTED DYNAMIC PERFORMANCES FOR THREE SRMS

Single-coil MSRM Double-coil MSRM Classical SRM

Speed, r/min 1500 1500 1500Dc-bus voltage, V 35 35 35RMS phase current, A 0.520 1.986 1.233Maximum phase current, A 1.375 6.853 4.027Average torque, N·m 0.118 0.435 0.266Maximum torque, N·m 0.165 0.537 0.463Torque ripple,% 78.1% 62.3% 108.3%Torque density, N·m/kg 0.092 0.328 0.151Output power density, W/kg 14.45 51.67 23.71Copper loss, W 3.17 13.61 18.24Copper loss per torque, W/N·m 26.86 31.28 68.57

Fig. 18. Comparison of output average torque-versus-speed characteristicsfor three SRMs at dynamic operation.

again that in the simulation of the double-coil MSRM, the upperand lower four coils on the opposing stators are divided in twowindings and connected in parallel to form one phase, as shownin Fig. 3(a), whereas in the classical SRM and the single-coilMSRM, the two coils on the opposing stators are connected inseries to form one phase.

Table VII presents the summary of the predicted dynamicperformances in the classical SRM and MSRMs at 1500 r/min.It could be found that the output torque developed by the double-coil MSRM is the largest in these three SRMs, whereas itstorque ripple is the lowest at the same condition. For example,the output average torque and torque ripple developed by thedouble-coil MSRM at 1500 r/min are 0.435 N·m and 62.3%,whereas they are 0.266 N·m and 108.3% in the classical SRMas well as 0.118 N·m and 78.1% in single-coil MSRM with atthe same condition. Accordingly, the output power and torquedensities of the double-coil MSRM are the largest in these threeSRMs. Moreover, it could be found that the copper loss of thedouble-coil MSRM is higher than that of the single-coil MSRMand lower than that of the classical SRM even though the turns-per-pole of the two MSRMs is more than that of the classicalSRM. The main reason for this is that the copper loss is the sumof the product of square of phase current and resistance. Andthe sum in the two MSRMs is smaller than that in the classicalSRM so that the copper loss is also diminished. It can also beseen that the copper loss per torque in the two MSRMs is lowerthan that of the classical SRM, i.e., for the same output torque,

Fig. 19. Simulated faulty performances of the double-coil MSRM under lackof windings with single-pulse control at 1200 r/min. (a) Lack of winding A1.(b) Lack of windings A1 and B1. (c) Lack of windings A1 and A2.

Fig. 20. Simulated faulty performances of the (a) single-coil MSRM and (b)classical SRM under lack of one phase (A) at 1200 r/min.



Fig. 21. Prototype of the double-coil MSRM: (a) Six E-shaped modular stators, (c) three segmental rotors, (b) and (d) assembled stators and rotors.

TABLE VIIISIMULATED AVERAGE TORQUES OF THE THREE SRMS OPERATION UNDER DIFFERENT CONDITIONS

Condition Double-coil MSRM Single-coil MSRM Classical SRM

Averagetorque,

N·m

Percentageof normal

torque

Averagetorque,

N·m

Percentageof normal

torque

Averagetorque,

N·m

Percentageof normal

torque

Normal 0.432 100% 0.123 100% 0.245 100%One windingopened (A1)

0.357 82.6% 0.079 64.3% 0.157 64.1%

Two windingsopened (A1and B1)

0.306 70.8% 0.041 33.2% 0.080 32.7%

Two windingsopened (A1and A2)

0.289 66.9% / / / /

the amount of copper loss of the two MSRMs is lower than thatof the classical SRM.

Finally, Fig. 18 shows the comparison of simulated averagetorque–speed characteristics of these three SRMs at the sameconditions. These simulations were executed for a dc voltageof 25 V and the speed was changed from 800 to 1600 r/minwith single-pulse operation. It can be seen that the double-coilMSRM has the largest torque and the single-coil MSRM has thelowest torque at the same condition, respectively.

C. Faulty Performances Under Lack of Diverse Windings

This novel MSRM inherently offers high fault-tolerant fea-ture, it means that the motor will continue to run in a satisfactoryoperation under fault conditions. Fig. 19 shows the simulatedfaulty performances for the double-coil MSRM drive operationunder lacks of one winding (A1), two-winding (A1 and B1) andone phase (i.e., both A1 and A2). In these simulations, the speedis not close-loop control. The dc-bus voltage is fixed at 30 V andthe speed of the machine is fixed at 1200 r/min, and the turn-onand turn-off angles are also fixed.

In order to investigate the faulty performances of these twoMSRMs with the classical SRM at the same condition undersingle-pulse control, Fig. 20 shows the simulated faulty phasecurrents and torque waveforms of the single-coil MSRM andthe classical SRM operation under lack of one phase whenthe dc-bus voltage and rotor speed are fixed at 30 V and1200 r/min. Table VIII shows a comparison of the averagetorque of these three SRMs at normal and open-circuit fault op-erations. It can be found that the average torque produced by the

double-coil MSRM can reach as high as 0.357 and 0.306 N·m,82.6% and 70.8% of the normal condition torque, under lacksof one-winding (A1) and one-phase (A1 and A2) operation. Thesingle-coil MSRM and classical SRM can only develop the av-erage torques of 0.079 and 0.157 N·m, which are 64.3% and64.1% of the normal condition torques, under lack of one phase(A). It means that the average torque of the double-coil MSRMis greater than those of the classical SRM and single-coil MSRMat the same condition. Furthermore, the double-coil MSRM hassuperior fault-tolerant performance over other two SRMs.

V. EXPERIMENTAL VALIDATION OF DOUBLE-COIL MSRM

In this section, a prototype of double-coil MSRM is developedfor experimental verification by using the design and analysisobtained from the aforementioned simulations. Its main specifi-cations are shown in Table I. Fig. 21 shows the prototype of thedouble-coil MSRM including the six E-shaped modular statorsand three segmental rotors. And the assembled stators with therotor are inserted into a prefabricated sleeve-type fixture.

A. Experimental Determination of Inductance

Generally, the calculation or measurement of the inductanceand flux linkage characteristics is an essential step in the SRMdesign and dynamic simulation. An indirect method, as in [19]and [20], is used to measure the flux linkage and inductancecharacteristics of this MSRM. Fig. 22 shows the experimentalsetup and circuit for measurement. In this static measurement,a dc voltage is applied to one-phase winding, while the rotor



Fig. 22. Block diagram of the experimental system for measuring the magneticcharacteristics.

Fig. 23. (a) Measured winding voltage and current for the simultaneous two-winding (A1 and A2) excitation. (b) Measured winding voltage, mutually in-duced voltage and current when only one winding (A1) is excited.

is locked at a certain position by an indexing head. Thus, theinstantaneous phase voltage uj and current ij in the phase canbe measured and recorded. The flux linkage λj and inductanceLj in the MSRM phase can be estimated as follows:

λj =∫ t

0{uj (t) − r · ij (t)} dt (18)

Lj = λj /ij for ij �= 0 (19)

where r is the winding resistance.After that, the rotor position is partially changed with a step

of 3° and the same processes are repeated until the rotor polemoves from the unaligned (θ = 0°) to aligned (θ = 45°) po-sition. Fig. 23 shows two typical phase voltage and currentwaveforms that are recorded at the aligned position. During thismeasurement process, the inductance data with single-winding(A1) excitation and simultaneous two-winding (A1 and A2)excitation are both tested and measured. Fig. 24 shows the com-parison of the measured data with 3-D FEA model results atthe unaligned and aligned positions. Furthermore, a comparisonof the inductances with single-winding (A1) excitation obtainedby different methods at the aligned position is given in Table IX.An excellent agreement can be seen in Fig. 24 and Table IX be-tween the 3-D FEA results and measured data. The maximumrelative errors between the 3-D FEA and measured inductance

Fig. 24. Comparison of inductances in winding-A1 (a) when only one windingis excited, (b) for simultaneous two-winding (A1 and A2) excitation.

TABLE IXCOMPARISON OF THE INDUCTANCES IN WINDING-A1 WITH SINGLE-WINDING

(A1) EXCITATION AT THE ALIGNED POSITION

Method 1 A 2 A 3 A 4 A

3-D-FEA [H] 0.0704 0.0513 0.0376 0.0299Measured [H] 0.0628 0.0493 0.0378 0.0308Relative error [%] 10.79% 3.81% 0.71% 3.09%

Fig. 25. Experimental setup for dynamic operation. (a) Double-coil MSRM,(b) Power converter and driver, (c) dSPACE, (d) Transducer, (e) Hysteresisbrake.

are lower than 11% and 3.9% at the aligned and unaligned rotorpositions, respectively.

B. Normal and Fault-Tolerant Performances

The three-phase double-coil MSRM drive is set up on an in-strumented test rig as shown in Fig. 25. The double-coil MSRMis loaded using a hysteresis brake coupled through a torquetransducer. A dSPACE 1103 is used as the digital controller.The controller and drive circuits are used for developing controlprogram and providing proper excitation signals to control the



Fig. 26. Simulated and measured waveforms with single-pulse operationwhen ω = 1220 r/min and Vdc = 30 V. (a) Simulated. (b) Measured.

Fig. 27. Simulated and measured phase-current waveforms with single-pulseoperation at when ω = 1000 r/min and Vdc = 28 V.

MSRM through a three-phase asymmetric half-bridge converter.The terminal variables such as phase voltage, phase current, ro-tor position, shaft torque, and dc-bus voltage and current arerecorded by sensors.

The normal steady-state performances with single-pulse op-eration are first tested and performed. Fig. 26 shows the sim-ulated and measured steady-state waveforms of phase voltageand phase currents with single-pulse operation. The dc-bus volt-age was set as 30 V and the speed was 1220 r/min. During thisoperation, the experiment was performed using the same settingas that of simulation. It can be found that the simulated wave-forms agree well with the experimental results, which validatesthe modeling and analysis of the MSRM. Another validationwas done for the MSRM running with a dc-link voltage of 28 Vand a speed of 1000 r/min under single-pulse operation. Thesimulated and measured steady-state phase current waveformsare shown in Fig. 27. A good agreement can also be found inFig. 27 between the simulation and experimental results. Ad-

TABLE XCOMPARISON OF THE SIMULATED AND MEASURED PHASE CURRENTS

Parameter Measured Simulated Measured Simulated

Speed, r/min 1220 1220 1000 1000Peak phase current, A 4.76 4.89 6.61 6.82Peak phase current error, A — +0.13 — +0.21RMS phase current, A 1.39 1.48 1.67 1.78RMS phase current error, A — +0.09 — +0.11

Fig. 28. Speed step response with closed-loop control for 2000 r/min. (a)Transient performance (i = 10 A/div). (b) Steady-state phase currents (i = 2.5A/div).

ditionally, the values of phase current at these two conditionsas obtained from simulation and measurement are given in Ta-ble X for comparison. The values include peak phase current,RMS phase current, peak phase current error, and RMS phasecurrent error. An inspection of the values in Table IX revealsgood agreement between the simulation and measurement, andthe current errors are very small. For example, the errors ofpeak phase current at speeds of 1220 and 1000 r/min are about2.75% and 3.18%, respectively. And the errors of RMS phasecurrent at speeds of 1220 and 1000 r/min are about 6.47% and6.58%, respectively. These comparisons verified the aforemen-tioned analysis and simulation again.

Second, the dynamic response with speed closed-loop con-trol under normal condition is measured. Fig. 28 shows thetransient speed step response and steady-state current wave-forms of the MSRM drive for a speed step of 2000 r/min at alight load. As expected, the fast speed response is shown andthe motor accelerates rapidly from standstill to command speedof 2000 r/min. Another speed step response for 1000 r/min un-der closed-loop control is performed and shown in Fig. 29(a).Furthermore, Fig. 29(b) shows the transient rotor speed andcorresponding phase currents at step speed command changes(1000→800→1200 r/min). The test result shows that the rotorspeed can quickly track the command speeds under the speedclosed-loop control and the speed ripple is also very small.

Finally, fault-tolerant operations of the double-coil MSRMdrive with open-circuit faults under speed closed-loopcontrol were tested. Fig. 30(a) shows the transient responseof the MSRM drive when one winding (A1) is open circuited at1500 r/min. Fig. 30(b) shows the steady-state phase currents af-ter open-circuit fault. Additionally, Fig. 31 shows the transientresponse and steady-state phase currents when two windings(A1 and B1) are simultaneously open circuited at 1500 r/min. It



Fig. 29. (a) Speed step response for 1000 r/min. (i = 10 A/div). (b) Transientresponse at step speed command changes 1000→800→1200 r/min. (i = 5 A/divand T = 0.25 N·m /div).

Fig. 30. Measured waveforms before and after one winding (A1) opencircuit. (a) Transient response. (b) Steady-state phase currents after fault.(i = 2.5 A/div).

Fig. 31. Measured waveforms before and after two windings (A1 and B1)open-circuit. (a) Transient response. (b) Steady-state phase currents after fault.(i = 2.5 A/div).

can be seen from these figures that after the open-circuit faults,the transient changes in speed are quite small, while the speedregulation is very good.

VI. CONCLUSION

With the development of EV applications, there are strongdemands for developing electric motors with low cost and highperformances. The SRM is gaining widespread interest as agood candidate for EV applications. In order to overcome thedrawbacks of the SRM such as low torque/power density andhigh torque ripple, this paper proposed a novel MSRM withhybrid magnetic paths for simple structure, cost reduction, andimproved dynamic performance. This novel MSRM using mod-ular and segmental structures has been presented by analyticaldesign, numerical simulation, and laboratory test. The voltage

and output power equations are analytically derived, and somedesign particularities and parameters are discussed. The fielddistribution, magnetic characteristic, and dynamic behavior aresimulated by using FEM analysis. In order to evaluate the dis-tinct features of this novel SRM, two MSRMs called as single-coil MSRM and double-coil MSRM are compared with a classi-cal 6/4 SRM. The comparison shows that the double-coil MSRMappears to have better characteristics such as higher torque pro-duction capability, lower torque ripple and stator iron-core mass,and higher torque and output power densities. Furthermore, thesteady-state faulty performances of these two MSRMs and theclassical SRM under lack of diverse windings/phases conditionshave been simulated and compared. The double-coil MSRM hasalso superior fault-tolerant performance over other two SRMs.Finally, a prototype MSRM with double-coil has been builtand tested in the lab. The experimental results of inductancecharacteristics and dynamic performances under normal andopen-circuit fault operations are presented to validate the anal-ysis and simulation. Both simulation and experimental resultsconfirm that the prospect for this novel MSRM will be bright ina wide several of industrial applications with the requirementsof high torque and power densities, high reliability, and faulttolerance..

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Wen Ding (M’11) was born in Changde, China. Hereceived the B.S. degree from the Xi’an Universityof Technology, Xi’an, China, in 2003, and the M.S.and Ph.D. degrees from Xi’an Jiaotong University,Xi’an, in 2006 and 2009, respectively, all in electricalengineering.

Since then, he has been with the Department ofElectrical Machinery and State Key Laboratory ofElectrical Insulation and Power Equipment, Schoolof Electrical Engineering, Xi’an Jiaotong University,where he is currently an Associate Professor. His

research interests include switched reluctance machines, electrical drives, andpower electronics.

Yanfang Hu was born in Shangqiu, China. She re-ceived the B.S. degree from the Shenyang Universityof Chemical Technology, Shenyang, China, in 2011.

In 2011, she joined the Hangzhou Hangfa PowerEquipment Co., Ltd, Hangzhou, China, where shewas involved in research and development of induc-tion and synchronous motors and turbine generatorfor small and medium-sized power equipment. Sheis currently involved in a switched reluctance motorproject as a Graduate Student at the School of Elec-trical Engineering, Xi’an Jiaotong University, Xi’an,

China.

Luming Wu was born in Ganzhou, China. He re-ceived the B.S. degree from Xi’an Jiaotong Univer-sity, Xi’an, China, in 2010.

In 2010, he joined the CPI Jiangxi Nuclear PowerCo. Ltd., Jiujiang, China, where he was involved in re-search and development of large nuclear power plant.He is currently involved in a switched reluctance mo-tor project as a Graduate Student at the School ofElectrical Engineering, Xi’an Jiaotong University.

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