Experimental High Performance Control of

Two Permanent Magnet Synchronous Machines in an

Integrated Drive for Automotive Applications Lixin Tang

Oak Ridge Associated Universities

Knoxville, TN 37932, U.S.A.

tangl@ornl.gov

Gui-jia Su

Oak Ridge National Laboratory

Knoxville, TN 37932, U.S.A.

sugj@ornl.gov

Xianghui Huang

GE Global Research

Niskayuna, NY 12309, U.S.A

huang@research.ge.com

Abstract—Closed-loop control of an integrated dual-inverter which is able to drive two permanent magnet motors independently is presented and evaluated experimentally. By utilizing the neutral point of the main traction motor, only two inverter legs are needed for the two-phase auxiliary motor. A modified field oriented control scheme for this integrated inverter was introduced and employed in real-time control. Experimental results show that the inverter is able to control two drives independently. An integrated, component-count-reduced drive is achieved.

Keywords-Permanent magnet motors; closed-loop systems

I. INTRODUCTION

Due to the increasingly limited amount of petroleum in the earth and the developing needs for it around the world, the energy efficiency of automobiles is becoming more and more important. The general public is becoming concerned about increased gas mileage and reduced air pollution. Hybrid electric vehicles (HEVs) and electric vehicles (EVs) are becoming more and more attractive as they are able to achieve much higher fuel efficiency and lower levels of emissions than their counterparts powered only by internal combustion engines (ICEs). However, high prices still prevent these vehicles from being widely accepted in the marketplace. In the EVs, HEVs, and fuel-cell powered vehicles, two distinct electric drives are needed. One is the traction drive, and the other is the compressor drive for the air conditioning (AC) or climate-control system. Normally, the first one has a power rating about 30-100 kW and requires a fast dynamic response [1]. The second drive uses only a few kW and does not require fast dynamic responses. Power electronics inverters and control circuits are needed for these two drives. Typically, two three-leg inverters are used for the drives to achieve independent control.

For automotive applications, reliability and cost are major concerns of both manufacturers and consumers. Generally, component-reduced topologies can reduce the size and cost of the circuit. Furthermore, reliability can also be improved by reducing the component count [2]. Component-reduced circuits with the same or similar functionalities are desired in automotive applications.

In order to minimize the size and cost of inverters for AC drive applications, different topologies can be used. Much work has been done on a low-cost induction motor drive based on a four-switch three-phase inverter (FSTPI) by using a split-capacitor bank, as indicated in Fig. 1. A space-vector modulated four-switch inverter (SVMFSI) without zero-voltage vector was proposed and verified for this topology [3]; the middle point voltage variation can also be compensated by an improved SVM scheme [4]. By using this topology, two of the six power switches in a three-phase inverter can be eliminated. A disadvantage to this reduction is a larger current distortion which results in 14% torque ripple at a 4 kHz switching frequency, doubled-direct current (dc) bus voltage, and increased capacitor number count. For automotive applications, a larger current distortion means lower efficiency and a larger capacitor count means reduced system reliability, all of which are undesirable. Therefore, the split-capacitor bank is not appropriate for automobile applications.

Figure 1. FSTPI based on a split-capacitor bank.

Another method for reducing the inverter switch number count is to use apportioning technologies [5]. This scheme does not require extra hardware; it can drive one three-phase induction motor and one two-phase induction motor with five inverter legs. However, it should be noted that the apportioning technology introduces extra distortion, which may cause degraded performance in both drives. For automotive applications, the main traction motor consumes much more power and requires better performance and higher efficiency. Therefore, this scheme is not very desirable.

For this purpose, an integrated-dual inverter without a split-dc bus capacitor was proposed [6] and verified with induction motors whose system diagram is shown in Fig. 2. In this novel

* Prepared by Oak Ridge National Laboratory, managed by UT-Battelle, LLC, for the U.S. Dept. of Energy under contract DE-AC05-00OR22725.

The submitted manuscript has been authored by a contractor of the U.S. Government under contract DE-AC05-00OR22725. Accordingly, the U.S. Government retains a nonexclusive, royalty-free license to publish or reproduce the published form of this contribution, or allow others to do so, for U.S. Government purposes.

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2006 IEEE COMPEL Workshop, Rensselaer Polytechnic Institute, Troy, NY, USA, July 16-19, 2006

0-7803-9725-8/06/$20.00 ©2006 IEEE.

integrated inverter, the same reduction of the auxiliary inverter switches can be achieved without introducing more capacitors. The neutral point for the main traction motor is utilized as a return path of the two-phase compressor drive. Experimental results [6–7] showed that the zero-sequence current from the auxiliary drive did not affect the operation of the main traction motor and both motors can be controlled independently at different speeds. The limitation of this scheme is that it is only applicable to Y-connected motors.

Figure 2. System diagram of the dual inverter for automotive applications.

It should be noted that all the control methods in the previous works were implemented in an open-loop voltage controlled manner. In this paper, current-control-based rotor flux oriented control (RFOC) is employed for the dual integrated drives for the first time. Furthermore, considering the desirable features of permanent magnet synchronous machines (PMSM) such as high efficiency, high torque/volume ratio, and fast dynamic response, the idea is verified with PMSMs in this paper instead of with induction motors that have typically been used with this kind of inverter. To date, most of these reduced component topologies/algorithms have not yet been tested with PMSMs.

This paper is arranged as follows: Section II explains the control algorithms for the two PMSM drives and the starting control scheme, Section III shows the hardware setup used in this project, Section IV shows modeling and experimental results, Section V explains a possible extension of the scheme to two three-phase PM motor, and Section VI provides the conclusions.

II. RFOC OF PM MOTORS WITH THE DUAL INVERTER

A. RFOC of Two PM Motors

The main traction motor will be controlled by the RFOC algorithm as indicated in Fig. 3. The control algorithm is executed in the rotor-reference frame which can be calculated from the position sensor. Two current controllers are employed, namely the d- and q-axis current controller, both of which are proportional-integral types. The normal SVM scheme is adopted to generate pulse-width modulation (PWM) pulses for current control. Because of the current controllers, it can be assumed that the motor is connected to three current sources. For a standalone three-phase balanced motor, there will be no zero-sequence current flowing in the main motor itself and the sum of the three-phase currents, iu, iv, and iw, will be zero, i.e.

0iii wvu =++ . (1)

,iα β

di

qirθ

*

di

*

qi

Figure 3. Block diagram of RFOC of the main traction motor based on current control in rotating frame.

In the integrated inverter, however, because the two motors are connected together via the neutral point of the main motor, as indicated in Fig. 4, a zero-sequence current flows in the main traction motor,

wvuba0 iii)ii(i ++=+−= . (2)

Figure 4. Currents flowing in the two PM machines.

Hence, there are two components in the currents of the main traction motor. One is from the RFOC itself, which is a balanced three-phase current without zero-sequence components; the other is from the auxiliary motor, which is a zero-sequence current to the main traction motor. Therefore, in each phase of the main PM motor there will be an equivalent zero-sequence current. The RFOC current controllers take this into account and generate reference currents which are able to drive the main motor and supply a path for the auxiliary-motor current. It should be noted that the measured currents in phases U and W (phase V current can be calculated from the other two currents) are not used directly for motor control feedback. iu

and iw must be mathematically derived to account for the zero-sequence current, as indicated in Eqs. (3) and (4). By doing so, the control independence of the main drive can be assured. The main motor offers paths for the auxiliary drive currents without sacrificing its own performance.

3

))mea(i)mea(i()mea(i)fbk(i ba

uu

++= . (3)

3

))mea(i)mea(i()mea(i)fbk(i ba

ww

++= . (4)

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The integrated RFOC of the dual inverter is shown in Fig. 5. For the main PM motor, it is seen that the modified RFOC scheme, as indicated in Fig. 5, is almost the same as the RFOC for a three-phase PM motor in Fig. 3. The extra time spent on calculation of the feedback currents in the main inverter real-time control algorithm is negligible, especially with high speed digital signal processor (DSP) chips.

*

dMi

*

qMi

*

dAi

*

qAi

dMi

qMi

dAi

qAi

Mθ

Aθ

Miαβ

Aiαβ

Figure 5. System diagram of RFOC for the dual inverter.

The auxiliary-motor drive is a two-phase PM motor with a smaller power rating. The measured phase currents are used for feedback of the RFOC. Phase A and B windings are orthogonal and therefore can be used for Park transformation directly. It should be noted that current-control schemes with and without a position sensor were developed in this project. However, due to the limit of the length of the paper, the sensor-less control part will be explained in a future publication. In this paper, position signals from two encoders were used for both PM motors.

B. Starting the Motors

In order to control the PM motors smoothly, the rotor position has to be known continuously so that motor currents can be calculated and controlled correctly. Absolute encoders can provide continuous rotor position information. However, the cost of an absolute encoder is much higher than that of an incremental encoder. An incremental encoder performs similarly to an absolute encoder, except that it does not supply an absolute reference angle. Its output signals are incremental. In each cycle it generates a pulse (index pulse) at a fixed position, if the fixed position is known, the rotor’s absolute position can be inferred from an incremental encoder with the help of index signal. In order to use an incremental encoder for a PM motor control, initial position detection or starting schemes are needed.

Two starting methods were used in this project. For the main traction motor, the index signal was used to calculate the absolute rotor position. During motor starting, the controller generates an open-loop rotating current vector to turn the rotor until the index signal of the encoder is detected. Both the amplitude of the current and the frequency are much smaller than the rated values. The angle between the index and rotor d-axis must be measured in advance so that the controller will continuously know the absolute position of the rotor after the index signal is received. It should be noted that the main drive is under open-loop control until the index pulse is received.

There might be some small oscillation or disturbance during this process, which are not desirable. The objective of using this scheme is to achieve a lower cost for our Lab test-setup. For HEV applications, resolvers are often used to provide rotor position information.

For the auxiliary drive, the current controller generates a specific current vector so that the rotor will be aligned and locked at a certain position (d- or q-axis). The controller will then jump to the normal operational program to start the motor at the specific position. The index pulse is not needed under this scheme. There might be some disturbance or small oscillation during this transition (i.e. the rotor moves from arbitrary position to d- or q-axis); however, the duration is quite short.

III. EXPERIMENTAL SETUP

An integrated (dual) inverter was constructed for experimental verification; it had two kinds of insulated gate bipolar transistor (IGBT) modules—one with a large power rating for the main drive and the other with a smaller rating for the auxiliary drive. The power modules were mounted on a liquid-cooled cold plate. The dc bus voltage was 325 V, and four Hall-effect current sensors were employed to detect the currents. Film capacitors rather than electrolytic capacitors were used for dc bus filtering, which was desirable because of their higher reliability. The switching frequency was 15 kHz for both inverters.

The system diagram of the experimental setup is shown in Fig. 6. The main traction motor was an 8.2 kW PMSM, and a 100 HP induction motor was utilized as a generator to load the main motor at various speeds. Because it is very difficult to find a two-phase PM motor on the market, a three-phase 2 kW PMSM, manufactured by Allen-Bradley, was modified into a two-phase PM motor for this project. The phase B’ and C’ windings are simply connected in series and disconnected from the phase A’ winding to construct a two-phase motor, as indicated in Fig. 7. An eddy-current dynamometer was used to load the two-phase PM motor. The parameters of the two PM motors are listed in Table 1. It should be noted that the auxiliary motor was modified to a two-phase PM motor, while the parameters were measured before the modification.

Figure 6. Experimental setup for the dual inverter project.

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Figure 7. The structure of a two-phase PM motor.

Table 1. Parameters of the two PM motor used in this project

Name of the Parameters Main Motor Auxiliary

Motor

Rated output power (Watt) 8200 5500

Rated speed (rev/min) 2000 4000

Rated voltage (Volt, rms) 200 230

Rated current (A, rms) 52.7 18.9

Back EMF constant (V/1000rpm, L-L voltage, 0 to peak)

88.1 71.25

Poles 8 8

Rated torque (Nm) 39.4 13.1

Stator resistance ( ) 0.03 0.2

Stator inductance (mH) 0. 99 1.5

Inertia (Kg.m²) 0.019 0.00147

Texas Instrument’s TMS320F2812 DSP was used in this project as the real-time controller. A TMS320F2812 board designed at the Oak Ridge National Laboratory (ORNL) was used as the controller, whose diagram is indicated in Fig. 8. There were two event managers that are especially suitable for dual PWM generation for the main and auxiliary inverter. The analog-to-digital converter (ADC) circuit was used to read the voltage and currents of the inverter; the four-channel digital-to-analog (DA) converter was used to display the real-time waveforms. The encoder interface enabled easy and reliable connections between the encoders and the controller. The joint test action group (JTAG) interface was used to upload programs and perform real-time debugging.

The software was designed according to the diagram shown earlier in this paper. It was written in C language using a multi-phase incremental system build method. A DSP emulator was used to test and debug the real-time code. In order to run the DSP at the maximum speed, the real-time code must be loaded into the random assess memory (RAM) of the DSP.

It should also be noted that the amplitude of phase A and B currents are not equal because the numbers of turns in the modified motor’s phases are different. Phase B winding only required 0.577 of the current in the phase A winding to generate the same magnetic-motive force.

Figure 8. Diagram of the TMS320F2812 controller board.

IV. MODELING AND EXPERIMENTAL RESULTS

An eight-channel oscilloscope was used to record the current waveforms of the motors. The CH1–CH3 were defined as U, V, and W phase currents of the main motor; and CH5–CH7 were A and B phase currents of the auxiliary motor and current to the neutral point, respectively, as indicated in Table 2. .

Table 2. Definitions of the currents in the waveforms.

Channels Definition and scale Details

Ch1 Phase U current, 50A/div

Ch2 Phase V current, 50A/div

Ch3 Phase W current, 50A/div

Main motor phase currents.

Ch5 Phase A current, 50A/div

Ch6 Phase B current, 50A/div

Auxiliary motor phase currents.

Ch7 Current to the neutral point, 50A/div

Between two motors.

Simulation results are compared with the experimental results. The simulation results obtained by using the method described in Ref. [6]. Psim® software was employed to carry out the simulation.

Figure 9 shows motor currents when the main motor was running at 2000 rev/min with 40 Nm load and the auxiliary motor stopped. There is no current in the auxiliary motor, while the three-phase current in the main motor were sinusoidal and symmetrical. Similarly, in Fig. 10 the main motor was stopped and the auxiliary motor was running at 750 rev/min with 10 Nm load torque. It should be noted that the amplitude of phase A and B currents are not equal because of the number of turns of them were different, as explained in the previous part. Although the main motor was stopped, however, there were currents flowing in it. The three-phase windings had the same instantaneous value and the net output torque was zero. These currents are zero sequence currents.

Figures 11–14 show current waveforms of the two motors at different speeds with the experimental results in Figs. 11 and 13 compared with modeling results in Figs. 12 and 14. In Fig. 11, the main motor ran at 750 rpm with a 30 Nm load, while the auxiliary motor was running at 505 rpm with a

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10 Nm load. The currents in the auxiliary motor are sinusoidal; however, the currents in the main motor were imbalanced due to zero-sequence currents from the auxiliary motor. The simulation results in Fig. 12 show a close match with the experimental waveforms in Fig. 11.

Figure 9. Main motor running at 2000 rev/min with 40 Nm load, auxiliary motor stopped.

Figure 10. Auxiliary motor running at 750 rev/min with 10 Nm load, main motor stopped.

Figure 11. Main motor running at 750 rpm with auxiliary motor

running at 505 rpm, TLm = 30.0 Nm, Pm = 2.4kW, TLaux =

10.0 Nm, and Paux = 529 W-experimetnal results.

-50

0

50

-50

0

50

0.01 0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.01

-50

0

50 Current to the neutral point of the main motor

Currents in the auxiliary motor

Currents in the main motor

Time (sec.)

Figure 12. Main motor running at 750 rpm with auxiliary motor running at 505 rpm, TLm = 30.0 Nm, TLaux = 10.0 Nm-modeling results.

In Fig. 13, the main motor ran at 1004 rpm with a 40 Nm load, while the auxiliary motor was running at 750 rpm with a 12 Nm load. Figure 14 is the modeling results under the same speed and load torque of Fig. 13. Again, the waveforms match well, except some ripples in phase A current of the auxiliary motor. It may be noted that the speed of the two PM motors can be independently controlled to their reference values at different load conditions through the experiments. All these close matches between the modeling and experimental results

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shown in Fig 11–14 confirms the effectiveness of the proposed RFOC algorithm for the integrated dual inverter.

Figure 13. Main motor running at 1004 rpm with auxiliary motor

running at 800 rpm, TLm= 40.0 Nm, Pm = 4.2 kW, TLaux =

12.0 Nm, and Paux = 1003 W-experimental results.

0 5 10 15 20 25 30 35 40 45 50

-50

0

50

0 5 10 15 20 25 30 35 40 45 50

-50

0

50

0 5 10 15 20 25 30 35 40 45 50

-50

0

50

Time (ms)

Currents in the main motor

Currents in the auxiliary motor

Current to the neutral point of the main motor

Figure 14. Main motor running at 1004 rpm with auxiliary motor running at 800 rpm, TLm= 40.0 Nm, TLaux = 12.0 Nm-modeling results.

Obviously, these zero-sequence currents observed in Fig. 11–14 would cause extra copper loss in the main motor. In reality, the main traction motor normally has a larger power rating (more than 30kW), which makes the zero-sequence portions of currents a relatively smaller percentage of the total currents. The extra copper loss is not significant [6]. The experimental results showed that the two drives could be controlled independently at various speeds.

V. POSSIBLE EXTENDSION

It may be noted that the integrated (dual) inverter is used to drive a three-phase main motor and a two-phase auxiliary motor in this paper. However, it is possible to use it drive two three-phase motors, as indicated in Fig. 15. The neutral point of the main motor can be used to connect one phase of auxiliary motor. Techniques discussed in [3–4] may be used.

Figure 15. Using the proposed inverter to control two three-phase motors.

VI. CONCLUSIONS

In this paper, the DSP control algorithm for the integrated (dual) inverter PMSM drive is presented along with experimental and simulation results. It is an extension of a closed-loop control version of the scheme proposed for induction machines in Ref. [6]. Experimental results at various loads and speeds of two motors are shown to confirm that both motors can be controlled independently.

A low-cost, reduced component count integrated drive for simultaneously controlling two motors for automotive applications has been achieved.

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