APPLICATIONS OF SURFACE-ACOUSTIC-WAVE DEVICES IN

SATELLITE COMMUNICATION SYSTEMS

Jeannine HENAFF

ABSTRACT: TypicaZ performances of surface-acoustic-wave (SAW) devices offerseveral advantages in the construction of digital communication networks.This paper describes the subunits of terrestrial equipments involved in asatellite communication Zink where some SAW devices improve significantZytheir design. ExperimentaZ examples of deZay-Zines, fiZters, osciZZators,...used for the modulation, the frequency conversion and the demodulation ofn-phase-shift-keyed (PSK) digital signals are described and present resuZtsare reported. These SAW devices operate in the range 70 MHz to 1 GHz wherethe surface-acoustic-wave technology aZlows reduction in size and weightcombined with ruggedness and reZiability.

INTRODUCT IONDue to their extremely Zow veZocity about 105times slower than

electromagnetic waves, surface acoustic waves (SAW) present a great interestfor signal processing applications. Because of the low velocity v, SAW alsopossess extremeZy smaZZ wavelengths X, when compared with electromagneticwaves of the same frequency F (A = v/F). The reduction in size is again ofthe order of 10-5: SAW devices, therefore offer drastic reductions in sizeand weight. In addition, SAW devices are fabricated on the surface of a crys-tal, so that they are also generally rugged and reZiable. Furthermore, SAWdevices are compatible with integrated circuit technology and their fabri-cation is obtained by photolithographic techniques. Devices using these wa-ves can therefore be mass produced at relatively low cost with precise andreproducible characteristics.

SAW devices can be designed with a center frequency of operationwhich may lie from MHz to GHz that is, in the VHF or UHF range. Since thesize of a circuit element is proportional to the wavelength, the lower fre-quency limit is governed by the size of available substrates, and the upperlimit occurs because of fabrication difficulties.

In this paper, we will restrict our study to a few examples ofapplication in satellite communication systems.v F7io71I- SHORT DESCRIPTION OF A SATELLITE DIGITAL

COMMUNICATION LINKA block diagram identifying the

subunits of terrestrial equipments involvedin a conventional satellite digital communi-cation link is described in Fig.1. At thetransmitting end, we first meet an interface(a) between the line carrying the digital in-formation and the actual modulator. It produ-ces the conversion of the HDB 3-bipolar si-gnal on-line to a (binary + clock) signal,the scrambling and the differential encoding.Then the signal is processed by a modulator(b) where it is 2 or 4-phase-shift-keyed atan intermediate frequency (i.f) to generatethe modulated signal. The modulator is follo-wed by an i.f.filter (c) limiting the spec-tral occupancy and "whitening" the spectrum.The 70 MHz modulated signal is then up con-verted (d) to the microwave carrier in the 6CNET PAB departement DTS/MAE92131 Issy-les-Moulineaux, FRANCE

70 MHz whiteningand bandwidthlifniting fitter

Fig. Block-diagram of a satellitedigital communication link

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GHz range, and a power amplifier (e) provides the antenna with the requiredmicrowave power.

At the receiving end, the signal coming from the antenna is firstamplified in a 4 GHz parametrizc amplifier (f), then down converted (g) fromthe microwave range to the 70 MHz i.f., and it is next amplified at the le-vel required at the input of the demodulator in a AGC 70 MHz amplifier (h).To limit the noise bandwidth, and in this way the noise power collected bythe demodulator, an i.f. filter (i) is inserted in front of the digitaZ de-modulator (j). The demodulation associated with bit timing recovery circuits,may be of the differential or coherent type for 2-PSK signals since the biterror rate versus the (energy per bit/noise spectral density) is about thesame for the two implementations, but for 4-PSK signals the coherent demodu-lation is obviously preferable and requires a carrier recovery circuit. Fi-nally another interface (k) between the digital line equipments and the de-modulator is needed. It performs the differential decoding in the case ofcoherent demodulation, and the final conversion of (binary signal + clock)to HDB3 bipolar signal.

SAW devices_delay-lines, filters, oscillators,....can improve si-gnificantly the design of the following subsystems: (b) modulator, (c) i.f. transmit filter, (d) and (g) up and down frequency converter (i) i.f. re-ceive filter and (j) demodulator. Some examples will now be described in amore detailed form.II. TRANSMITTING END

II.1. Digital modulator

The ability of SAW oscillators to give a stable frequency direc-tly in the VHF-UHF frequency ranges permits the realization of simple andreliable all digital 4-PSK modulators [1] (cf.Fig.2). Starting from a 280MHz-SAW oscillator (point A,Fig.2) a digital divider by 2 gives square wave-forms at 140 MHz. Then, two other divisions by 2 of the square waveform(point B) and of its complement (that is to say with a ir phaseshift, point C)give the 4 possible phases of the i.f. signal on 4 outputs (points D,E,F andG). A suitable switching of these 4 outputs provides the desired phase-sta-te according to the binary streams coming from an odd/even splitting andencoding device. The whole device is implemented using only "off-the-shelf"logical circuits and without any analog circuit liable to drift. The fourphase-states are keyed at + 10 and the levels are equal within + 0.5 dB. The-se results are about the same as for analog modulators but the temperaturebehaviour and the aging of this digital modulator are distinctly better.

An all digital 2-PSK modulator is obviously easier to implementsince only one divider-by-two (to bring down the frequency from 140 MHz to70 MHz) and half the switches are required. A dBt1~~~~~~~~~~~~~~~~~/ / I?-\

Fe-2/T| F.-3/2T illi || Fo-l/2T Pu P.+l/ZT || I,|K Poo3/2T P.eu2/Te~~~~~~~~~~~61,53 63,64 65,6? 67,86 70D 72,12 74,24 76,36 76,47

D 7OMHz~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~~~~~~~~H

cJJg,- Example. 8.448BMbin/s -4CfPSK Fo =70Mfz MHz.

E~~~~~~~ PT Fig. 3. (o)_ _{aSpectral density of the 4 pfhcse-P'SK modulated signetof the output of the modulator.F ~ piu mltd sfeunyrsos hTe

G 3T/P2 transmit filter.-- ______ ()c Actuolamplitude Vufrequency response of the SAWtransmit filter.

Fig. 2. Block diagram of an all digita 4 phase PSK modulaitor _ (d) Spectra density of the trensmitted 4 phOse - P SKstarting from a-80MHz SAWoscillator signar.

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v5 \ \ At JF ~~~~~~~~~~~~~~~~~~~~~~mulclkrbyS 6 .(2611= 0336\. control volage

\olta9e controlle-dSAW oscillatlor

io

Fig 4 Bit rn ntei (BER)wesusenergy pr bit (E/N.) . 5. Up- converter.: /oco/ osci/lo/ornoise spectral densitY4 phoee -PSK ddiffentol encoding and coherent demodution.

II.2. I.f. transmitting filter [2]

In digital transmission, the Nyquist criteria are generally expres-sed in terms of signaling with impulses. This is particularly convenient be-cause the spectrum of an impulse is a constant at all frequency (white spec-trum). In fact, we use rectangular pulses and therefore the density of a n-PSK signal is (sin iTFT/ 1TFT)2distributed, where F is the frequency and T isequal to the symbol length ; that is, 1 bit for 2-PSK signals and 2 bits for4-PSK signals.

As the first Nyquist criterion restricts the required bandwidth toF9 + 1/2T, the spectral occupancy can be limited by filtering the i.f. PSKsignal. Moreover, it is possible to correct the transmitting filter, givingit a ( wfFT/sinfr FT) amplitude response (cf.Fig.3) to obtain a white spectrumat the transmitting end. This "double-humped" filter is sometimes termed awhitening filter [3] .

SAW bandpass filters can very easily perform this function [4].Moreover, they can be designed with a linear phase response (symmetric fini-te impulse response) and so have constant group delay. This point is parti-cularly important since group delay equalizers will not be needed. Sincesuch equalizers are expensive to manufacture and to adjust, and are often asource of disturbance in the case of low digital rates (narrow bandwidtharound 70 MHz), it is important to avoid their use. The experimental andtheoretical (DIRAC pulses) bit error rate (BER) versus the energy per bit/noi-se spectral density (E/No) are shown on Fig.4 for a complete transmit-recei-ve unit using such a ( orFT/sin mfFT)2 filter at the transmitter end. Noticethat the discrepancy between the theoretical and experimental results is on-ly equal to IdB which is considered as exceptionally good.II.3. Up-converter

A frequency shift, using a transmitter local oscillator and a mi-xer, is necessary to convert the 70 MHz modulated signal to the microwavesignal sent to the antenna. In the fixed-frequency radio-links, this localoscillator is generally obtained starting from a bulk quartz oscillator fol-lowed by successive amplifications and multiplications. But, the satellitesystem operators have to change their transmission frequency according tothe satellite in operation, to the digital rate, to the allocated repeater,to the location of their carrier within the frequency plan, etc...

To avoid ceaseless changing of the quartz oscillator and readjus-tement of the multipliers, the operators prefer the use of a frequency syn-thesizer as primary pilot, which gives so-called "agile" frequency-shifts.For this purpose, SAW devices provide a very simple and efficient solutionsince voltage controlled stable oscillators are obtained which can be tunedover a sufficient frequency range. Moreover, they oscillate directly on thefundamental in the UHF range. Thus the multiplication factor is low and thefiltering of spurious spectral lines becomes easier.

As an example, one can describe the up-converter for a 6-4 GHzsatellite system: in the earth station, the transmit frequency is in the

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,0-2

range 6.2 + 0.2 GHz and therefore the pilot frequency has to be tuned overa + 0.2/6.2 = + 3.2 % range.

This can be done using a 1.033 GHz SAW oscillator voltage control-led to give the + 33.3 MHz tuning range and followed by a multiplier by 6.Fig.5 shows the transmitter local oscillator arrangement.III. RECEIVING END

III. 1. Down-converterThe advantages and the block-diagram of the reception down-conver-

ter are obviously about the same as those of the transmit up-converter. Des-cribing the same example as in § II, the down link frequency is in the ran-ge of 4 GHz, the multiplier by 6 has simply to be replaced by a multiplierby 4 if the tuning range of the pilot is increased to + 0.2 = + 5 %, whichis a little difficult to realize. Another possible solutiin is-to keep themultiplier of 6 and choose for the voltage controlled SAW oscillator a fre-quency around 0.67 GHz which is easier to implement but leads to a light ex-tra-degradation of the phase noise ( 3dB).

III2. I.f. receiving filterEven though, at the reception end, "double-humped" bandpass filters

are not required since the need is only to limit the noise bandwidth to theNyquist bandwidth, the usual advantages of the SAW filters remain still im-portant : easy reproducibility, long term stability (no adjustement), narrowbandwidth design for low digital rates and especially linearity of the phaseversus frequency which excludes the expensive and difficult to adjust groupdelay equalizer.

III3. Digital demodulatorBesides the previous examples of SAW applications several very in-

teresting solutions of digital demodulation by means of SAW devices will nowbe discussed.

Demodulation of n-PSK signals is achieved either by comparison withthe preceding bit, which is called differential demodulation, or, in a moreoptimal way, through coherent demodulation which requires generating a localreference wave in phase with the carrier (or with the signal shifted to theintermediate frequency) without modulation.

The voltage controlled SAW oscillators (VCO) are particularly ad-vantageous for producing such a reference wave thanks to their high oscilla-tion frequency (SAW oscillators offer good performances at frequencies ran-ging from about 100 MHz to 1 or 2 GHz) and to the broad tuning frequency ran-ge covered.

III.3.1.IDifferential demodulation [5,6]Let us first consider the case of 2-PSK signal, generally use for

low digital rates (2.048 Mbit/s) because its spectral occupancy is twice asmuch as 4-PSK signals. Differential demodulation is commonly used for 2-PSKsignals because of its very simple, and hence reliable, implementation andalso because of the low loss of performance between differential and coherentdemodulation in this case (0.5 db).

Fig.6 a shows the block-diagram of a 2-PSK differential demodula-~~~~~~~~~~phase discriminatort;!,~~binwy l ; At'*xduroion of a syrmbo (2bils)r

signal I;r.an biytnncrt sMoon <nil1t)T220

I E tI;- I> j Got1;| ~1;2) :1rZ2dlphase lockedp ~~~ ~~~~~~~~~~~~~~~~~~~~~~~~I loop

I ~ ~~ ~ ~ ~ ~ ~ ~ ~ I r / ittitinginaAnryIItrsatn L. SA VCO Z8OOMHzJ

loJ _____Jk1]I~ -T~ I4PKLr a c 0ea-7=TMMHz = a 70 MHa .70SHz

Hz~~~~~~~~~~~~~~~~~~~~~70M 70MM.~L, 2t Lr-eltTl nwlT- -"

b) or,,~~~~~~~~~~~~~~~~~~~~~bttmng bnr propsrlyuilli: ~~~~~~~~~~~~~~~~~~~~~~~~1.0-7y biay. socalled2 photeosPSK demodulolor gnd SAW rlemeneiion of tthe delaryrs A ..irg sro rn denoduoloor

eA clock

Fig 6 n phase-PPSK differenticl demodulctors - L.J

Lr~2oTr.a4 Fig, 7 i. f. carrier recovery circuits using a phase - locked -loop.b 4 ph-PSK dtlOdoictor cnd SAW iptplrmentrtion of t. da stoy% with 2MiMSAW VCOQ

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tor: a phase descriminator receives on one input the modulated signal and onthe second one the modulated signal delayed approximately by one bit duration(T= 1/2.048 106) by a delay line T, therefore its output will provide theoriginal bit stream after regeneration. The delayc has to be also an integralnumber of half periods of the 70 MHz i.f. Thus we chose -r=68. r7o%T(1-5.1C0=486 nS. For high rate communication systems, the demodulator delay-l'ines areeasily realized with coaxial cables ; but here for lower bit rates, this ap-proach is no longer valid due to the length (= 1OOm) and the attenuation ofthe required cables. An attractive solution to this problem is provided bySAW delay-lines.Fig.6a shows also the implementation of the delay: the pie-zoelectric substrate is a 9x9 mm YZ lithium niobate, 1mm thick, leading toa low-cost, small size and easy-to-adjust device. Differential demodulationof 4-PSK signals can be achieved in the same way. The block-diagram of 4-PSKdifferential demodulator and SAW implementation of the delays is shown onFig.6b. A very simple, cheap and reliable solution may still be obtained. Butin order to get the same bit error rate BER, a 2 dB increase of the E/Nora-tio (energy per bit/noise spectral density) must be available with differen-tial demdodulation with regard to coherent demodulation.III3.2. Coherent demodulation[7,8]

For coherent demodulation, mainly used for 4-PSK signals, a carrierrecovery circuit is necessary. The 70 MHz i.f. is first multiplied by 4 toeliminate any modulation by changing the phase back to 2kir whatever the ori-ginal phase-shift. After filtering, the 70 MHz recovered carrier is obtainedthrough division by 4 by means of low-cost ECL logic circuits which requiresno adjustement. As a matter of fact, frequency shift in the satellite may re-sult in an important variation of the i.f. in the repeater, and so a phase-locked-loop is generally preferred for the filtering of the 280 MHz spectralline. This gives narrower filtering and wider tracking range of the interme-diate carrier frequency at the price of a larger response time.

Again SAW devices bring an important advantage. Analog multipliersare devices calling for filtering of the required harmonic frequency, whereasfrequency dividers can be fully digitalized and are therefore inexpensive andmore reliable. Since it is very easy to realize voltage-controlled-SAW-oscil-lators working directly on the fundamental at 280 MHz, the block-diagram ofthe i.f. carrier recovery circuit becomes the one that is shown on Fig.7 u-sing a divider by 4 instead of a multiplier by 4 in the conventional solution

Such a phase-locked-loop provides very interesting results:- + 60 kHz tracking range for a OdB E/No ratio, the noise power being appliedbefore the signal.

- < + 20 residual phase error of the recovered carrier in the 70 MHz + 60kHzrange.

- 2.5° mean phase deviation for a OdB E/Noratio.- less than 1 cycle slipping of the recovered carrier a day, always for aOdB E/No ratio.

Such characteristics are obviously very important for a satelli-te digital communication link protected by a convolutional code and using aViterbi algorithm decoder (Remember that a OdB e/No ratio will give only atheoretical BER equal to 0.15 when using coherent demodulation of 4-PSK dif-ferential encoded s ignal).

IV. CONCLUSIONAlready,some of these devices have been successfully used for pic-

turephone applications and 30 voice channels PCM multiplex via the Symphoniesatellite. Furthermore, a radio-link between Bercenay-en-Othe (home-France)and La Riviere des Pluies (Island of la Reunion, France in the Indian Ocean)via Symphonie and/or Intelsat will soon take advantage of most ot these SAWdevices to transmit 240 voice channels.

In conclusion, SAW techniques have proved to be a fruitful solutionfor extending the use of differential demodulation at low digital rates, and

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for improving the design of coherent demodulation. They provide whiteningor noise-bandwidth-limiting linear phase i.f. filters but also very narrow-bandwidth minimum phase VHF filters for carrier recovery circuits. In addi-tional, SAW techniques can supply 2 or 4 PSK all digital modulators withfixed frequency oscillators, and up and down frequency converters with vol-tage-controlled oscillators. This results from their adaptiveness to therequisite frequency ranges and from their favourable stability as well aselectronic tuning properties.References

1V1 Jeannine HENAFF, Proc.79 IEEE Ultrasonics Symp. cat.79 CH1482-9SU.pp. 855-860.

2. Jeannine HENAFF and M. FELDMANN, Electron. Lett., 1980, vol.16, n04,pp. 124-125.

3. D.R. DUPONTEIL and D.A. LOMBARD, Proc. ICC74, ppv36 D1-5.4. J. HENAFF and M. FELDMANN, Proc. 79 ISCAS, cat.IEEE 79 CH 1421-7CAS,

pp. 617-620.5. J. HENAFF, M. CAREL, G.LAINEY and M. LABASSE , Electron. Lett.,

15 thsept.1977, vol.13, N09, pp.586-588.6. P.BROSSARD, J.HENAFF and D.LOMBARD, Proc.77 IEEE Ultrasonics Symp.,

cat.77 CH1264-ISU, pp.532-536.7. P. BROSSARD et J. HENAFF, French patent n°76 30684.8. Jeannine HENAFF, l'Onde Electrique, vol.59, n°8-9 aouft-sept.79,

pp.95-101.

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