high efficiency metal halide discharge lamp ballast with an internal igniter and low ripple lamp...

High efficiency metal halide discharge lamp ballastwith an internal igniter and low ripple lamp current

I.K. Lee and B.H. Cho

Abstract: This is a proposal for a half-bridge type metal halide discharge (MHD) lamp ballast witha coupled inductor and a frequency-controlled synchronous rectifier. To avoid using an externaligniter, the internal LC resonance of a buck converter is used to generate a high-voltage pulsefor the ignition. A coupled inductor filter is used for steady state ripple cancellation. This filterallows the MHD lamp to avoid the acoustic resonance phenomenon. To improve the efficiencyof the ballast, a synchronous buck converter is used for the DC/DC converter stage and a frequencycontrol method is proposed. This scheme reduces the circulation current and turn-off loss of themetal-oxide semiconductor field effect transistor (MOSFET) switch in the constant poweroperation, which results in an increase of the efficiency of the ballast system compared to fixedfrequency control. A front-end power factor correction circuit is included in the ballast. Thisproposal zis verified with hardware experiments.

1 Introduction

Metal halide discharge (MHD) lamps are receiving con-siderable attention as multi-purpose light sources becauseof their high efficiency, long lifespan and natural colour.Like many other discharge lamps, the MHD lamp hascomplex operating characteristics both when it is initiallyturned on and also in its steady state. MHD lamps initiallyneed a high-voltage pulse for ignition (1–3 kV), a propercurrent for the glow-to-arc transition and a warm-upcurrent to reach their steady state. In the steady state, thevoltage of each MHD lamp is different (normally 65–110 V). This voltage increases with operating time. For along lifetime and constant light output, the power to thelamp should be controlled at the rated value in spite ofthe lamp’s voltage variation. In addition, the ripple com-ponent of the drive current should be small in order toavoid a fluctuation of the light output and the acoustic res-onance phenomenon. Also, an alternating current is neededin order to prevent the lamp from consuming each of itselectrodes unequally in the steady state [1–3].In order to simplify the ballast system, there have been

many efforts that involve the combining of power stages.One approach is to unify the power factor correction(PFC) circuit and the DC/DC converter. Another approachis to combine the DC/DC converter and the full-bridgeinverter [4–6]. In a commercial AC line application, abuck-type converter is usually used for the DC/DC conver-sion stage of the MHD lamp ballast, because the lampvoltage is lower than the rectified DC voltage from a com-mercial AC line. Therefore the second scheme is a goodchoice, because the buck converter is easily implementedwith a full-bridge or a half-bridge inverter without anyadditional components.

# The Institution of Engineering and Technology 2007

doi:10.1049/iet-epa:20060315

Paper first received 29th May 2006 and in revised from 11th January 2007

The authors are with the School of Electrical Engineering, Seoul NationalUniversity, San 56-1, Shinlim-Dong, Kwanak-Gu, Seoul, 151-742, Korea

E-mail: [email protected]

IET Electr. Power Appl., 2007, 1, (3), pp. 291–298

In addition, to improve the efficiency of the buck conver-ter, a soft switching method, such as zero voltage switching(ZVS) or zero current switching of its active or passiveswitch, is usually used. A synchronous rectifier buck con-verter or a discontinuous conduction mode buck converteris a good choice for the soft switching operation. Theyhave another advantage in that they are easily combinedto a full-bridge or a half-bridge inverter for the convertingand inverting functions. However, these types of convertershave a larger peak inductor current than the continuous con-duction mode buck converter for the same output averagecurrent. This large peak inductor current can be filtered bya large output capacitor. This large capacitor makes thesystem bulky. Also the converter poses difficulty in operat-ing as a DC current source. If a half-bridge inverter isadopted for simplicity, difficulty arises in igniting thelamp because the LC resonance causes a large switchcurrent for the internal ignition. Also, the open circuitvoltage cannot be easily obtained with the half-bridge inver-ter for the external ignition.

To obtain a low ripple output current, internal LCresonant ignition, and high efficiency of the ballast, a newhalf-bridge ballast is proposed. It operates as a frequency-controlled synchronous rectifier and low frequency inverterwith a coupled inductor and an auxiliary inductor. The oper-ation mode is analysed, and the ripple-cancellation methodis proposed. Also, the frequency control of the synchronousrectifier with a constant power load is presented.

2 Conventional two-stage ballasts and theproposed ballast

For the conventional unified half-bridge and full-bridgetype ballasts, it is difficult to obtain a high-voltage ignitionpulse with the internal output LC filter. To limit the switchcurrent when the LC resonance circuit generates a high-voltage pulse, the characteristic impedance of the LCfilter should be high. Because the LC circuit acts as anoutput filter that reduces the output current ripple to avoidthe acoustic resonance of the lamp in the steady state, thecapacitance should be sufficiently large. Therefore a high

291

characteristic impedance could not be easily obtained [4].This can be improved by using a buck-type ballast with afourth-order LC filter [5]. It has the advantage of eliminat-ing the external igniter and being able to filter the outputcurrent by means of the use of a second-order LCnetwork. However, two separate inductors are necessaryfor sufficient filtering of the lamp current. One of themhas a large inductance to avoid a large resonant currentduring the ignition phase. This results in an increase insize and weight. As a result, conventional half-bridge andfull-bridge type ballasts have difficulties to be simul-taneously implemented both an internal igniter and outputripple current filter in one topology.

Fig. 1 shows the proposed half-bridge type ballast. It con-tains a coupled inductor, an output filter capacitor, C2, anauxiliary inductor, Laux, and a triac switch, St. L11 and C1

are used to generate a high-voltage pulse for ignition (Stswitch is open). In the steady state, the output currentripple is cancelled through the coupled inductor, the auxili-ary inductor and C2 (St switch is closed).

3 Inverting and converting control method

As shown in Fig. 1b, one of the MOSFETs acts as a mainswitch and the other acts as a diode for the buck converteroperation. After half of an inverting cycle, the roles of thetwo MOSFET switches are exchanged, which commutateslamp current. As a result, in order to achieve the ZVSturn-on operation of the main switch, the circuit simul-taneously operates as a synchronous rectifier, as well as a

Fig. 1 Proposed ballast and ripple-cancellation operation

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low-frequency inverter. To achieve this operation, a newinverting method is proposed as follows.

3.1 Principle of the proposed converting andinverting control method – an open loopdescription

During the low-frequency inverting operation, the commu-tated current has a large overshoot on the rising part or aringing on around zero crossing point of the low-frequencysquare wave current, because the output filter capacitorcannot change its polarity abruptly. The overshoot resultsin the switching operation of the converter going out ofthe soft switching condition. The ringing results in increas-ing the restrike voltage of the lamp. And these reduce theefficiency and the lamp’s lifespan [7, 8].In Fig. 2, an open loop description of a control method is

proposed to prevent the reverse biasing of C2 during theinverting operation. At the comparator, the ramp is com-pared with a current shaping waveform, which has aproper slope during the inversion of polarity of thecurrent. This results in the changing of the duty cycle pro-portional to the current shaping slope (D , 0.5 ! D ¼0.5 ! D . 0.5). The output capacitor voltage is decreasedgradually to zero and then increased to its steady state value.Therefore a monotonically increasing lamp current can beachieved without overshoot or ringing. Fig. 2 shows simu-lation results with a resistive load. Figs 2c and d illustratewhere the current is inverted instantaneously and where itis inverted by the proposed current shaping wave. Fig. 2cshows ringing at the current’s peak or at the zero crossingpoint of the inductor current during inverting. Fig. 2dshows that the lamp current commutates monotonicallyand the turn-on loss of the MOSFET is minimised.

3.2 Feedback control-inverting control with acurrent reference

An MHD lamp ballast should control the lamp power, at itsrated value, to provide a long lifespan for the lamp. Tocontrol the power and accomplish proper inverting, feed-back control is proposed, as shown in Fig. 3. A currentshaping signal is applied to the error amplifier as a currentreference that controls the rated lamp power with thelamp voltage and the commutation of the lamp current.And the controller is simplified because that controls boththe converting and the inverting operations simultaneously.

4 Analysis of the operation mode

4.1 Ignition phase

An internal LC resonance is used for the ignition, where as St,in Fig. 1, is open. A square wave voltage is applied to the LCresonant network by the half-bridge switching action. Fig. 4shows the equivalent circuit at the ignition mode. Theignition voltage and the input peak current are given by (1)and (2). A sufficiently high-ignition voltage (Vign) pulse canbe obtained by the switching at fign in (1) and (3), the switch-ing frequency in the ignition mode, which is near the resonantfrequency. As shown in (2), the resonant capacitance C1

could be designed to be small for a high characteristic impe-dance, Z in (2), to reduce the peak current in MOSFETs (S1,S2 in Fig. 1). Thus, the value of C1 should be determined withthe value of Iin,pk and the rated current of the MOSFETs (S1,S2) taken into consideration.Once the ignition takes place and the lamp turns on, the

switching frequency of the MOSFETs is adjusted to the

IET Electr. Power Appl., Vol. 1, No. 3, May 2007

Fig. 2 Open loop description of the proposed control scheme and simulation results

a PWM with a current shaping waveb Control signal and gate pulse during invertingc Instantaneous invertingd Inverting by a modulation with a current shaping wave

(1)

(2)

value at the steady state

Vign ¼4

p

Vin

2

� �

�1=sC1 þ Rc

sL11 þ Rl þ 1=sC1 þ Rc

��S¼j2pfign

;

4

p

Vin

2

� �1

Rl þ RC

ffiffiffiffiffiffiffiL11C1

s

Iin,pk ¼(Vin=2)

Zwhere Z ¼

ffiffiffiffiffiffiffiL11C1

s

Fig. 3 Proposed inverting control method


fign ¼ fO

ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi1þ

4=p (Vin=2)

Vign

swhere fO ¼

1

2pffiffiffiffiffiffiffiffiffiffiffiffiL11C1

p (3)

4.2 Steady state ripple cancellation – a coupledinductor

After the ignition, St is turned on to operate the coupled induc-tor filter. Using only the coupled inductor, the ripple current ofthe lamp can be cancelled by adjusting the coupling coeffi-cient, k, and the effective turns ratio, n, in (4) [9]

k ¼LMffiffiffiffiffiffiffiffiffiffiffiffiffiL11L22

p n ¼ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiL11=L22

pwhere LM ¼ mutual inductance

(4)

Fig. 5 shows the equivalent circuit after the ignitionwithout the auxiliary inductance in Fig. 1, along with itsT-equivalent circuit, where L11 and L22 are the primaryand secondary side self-inductances, respectively. Inaddition, Lefe, Lefp and Lefs are defined as the effectiveentire inductance, the effective primary inductance andthe effective secondary inductance of the coupled inductor,respectively. This circuit is the conventional buck converterwith a coupled inductor filter [10–12]. In this circuit, thelamp ripple current flowing through L11 should be mini-mised to avoid the acoustic resonance of the MHD lamp.

For Fig. 5a, if the value of the output filter capacitor, C2,is infinity and the turns ratio of the coupled inductor, n, isone-to-one, all AC components of the input voltage are

293

applied to the secondary side of the coupled inductor. Thisvoltage is applied to the primary side, which cancels theripple current of the primary side. As a result, the outputcurrent ideally has a zero ripple component.

Equations (5)–(9) show the principles of operation. Theload ripple current can be expressed as (9). Therefore nmust be nearly equal to 1/k to make Lefp approach a verylarge value

LS ¼ L22 � LM (5)

LP ¼ L11 � LM (6)

Lefp ¼ L11 þLPLS

LM ¼ L111� k2

1� kn(7)

Lefs ¼ L22 þLSLP

LM ¼ L221� k2

1� k=n(8)

Di1 ¼(Vin=2)� VO

LefpDTS (9)

4.3 Effect of the finite filter capacitance

The input voltage (Vs in Fig. 5) to output ripple voltagetransfer function of the circuit, with the finite output capaci-tor in Fig. 5b, is given by

voVs

¼1

(LMLPC2=RLAMP)s3 þ C2LMs

2 þ (L11=RLAMP)sþ 1

(10)

Fig. 4 Equivalent circuit of the ignition mode

294

A capacitor connected in parallel with the load is necessaryfor the unified structure ballast that has an internal igniter asshown in Fig. 1. Fig. 5c shows the equivalent circuit withthis capacitor, and (11) is the transfer function of the circuit

voVs

¼1

LPLMC1C2s4þ (LMLPC2=RLAMP)s

3

þ{(C1 þ C2)LM þ LPC1}s2þ (L11=RLAMP)sþ 1

(11)

As shown in (11), because the capacitance of C1 should besmall for the small ignition current, its contribution to theattenuation of the ripple current at the switching frequencyis negligibly small.If n ’ 1, then k ’ 1 and LP ¼ (L11 – LM) ¼ 0 because

L11 ’ LM. Therefore if the turns ratio, n, is unity, thethird-order term in the denominator of the transfer functionin (10) becomes zero when C2 has finite value. The rippleattenuation cannot become 260 dB/decade. In this case,(10) becomes the transfer function of a conventional buckconverter [9, 10]. As a result, the ripple-cancellation oper-ation is not sufficiently effective. Therefore the effectiveturns ratio of the coupled inductor should not be unity forthe ripple-cancellation operation that attenuates the ripplecurrent at 260 dB/decade.If the turns ratio of the coupled inductor is designed as

n , 1, then LP is not zero value, and the ripple attenuationsignificantly improves in comparison with that in the caseillustrated in Fig. 5, (10) and (11). A leakage inductanceadjustment method is also one of the efforts made to solvethis problem. In this case, k , 1 and n . 1 and the ripple-cancellation operation can be achieved. However, theadjustment of n and k may be too sensitive. Whenprimary and secondary coils are wound on the samebobbin with a ferromagnetic core, if k is nearly equal to1, then n is also nearly equal to 1. As an example,k ¼ 0.99 and then n ¼ 1.01. Thus, it is too difficult toadjust n to 1/k accurately.Another problem is the voltage stress of the triac switch,

St, in Fig. 1 in the ignition mode. Because the voltage of theopen switch is equal to that of the lamp, in the ignition modewhen n ’ 1, the ignition voltage (1–3 kV) is applied to thetriac switch, which destroys the triac.

Fig. 5 Equivalent circuit in the steady state and its T-equivalent circuits

a Equivalent circuit in the steady stateb T-equivalent circuits with C2

c T-equivalent circuits with C1 and C2


Also, in order to limit the resonant current at a low valuein the ignition mode, the characteristic impedance should behigh. Therefore L11 is large and C1 is small. Since thevoltage of St should be low, n cannot be large. If L11 islarge and n ’ 1, then the entire effective inductance, Lefe,in Fig. 5a, is large. As a result, the switching frequencyshould be low for the ZVS condition. This causes thelarge current ripple of the lamp.To resolve the problems described above, an auxiliary

inductance, Laux, can be added on the secondary side ofthe coupled inductor as shown in Fig. 6. In this figure, Vais defined as Vo – Vs (during S1 off and S2 on) and Vo(during S1 off and S2 on), respectively. The effect of theimprovement in the ripple cancellation will be explainedas follows.

4.4 Adding an auxiliary inductor on the secondaryside

A T-equivalent circuit model with Laux, shown in Fig. 6b, isused to obtain the zero ripple condition for an infinite outputfilter capacitor. This condition can be explained by (5)–(9),and (12). The leakage inductance can be included in Laux forthis analysis. The coupling coefficient, k, is assumed to bealmost unity. This assumption is valid when the primaryand secondary coils are wound on the same bobbin with aferromagnetic core

Lefs ¼ L22 þ Laux þ LS=LP

where LS ¼ L22 þ Laux � LM(12)

If L22þ Laux is designed to be equal to LM, LS, in (12),becomes zero and Lefp, the effective inductance seen byC1, approaches infinity. This causes the lamp ripplecurrent to be nearly equal to zero. In this case, as Laux isadded to the circuit, the circuit has several merits asfollows. If Laux is added to the secondary side of thecoupled inductor, the zero ripple condition on the primaryside can be controlled not by k and n, but by Laux. Inaddition, for the case of the finite output filter capacitor in(11), the turns ratio, n, is easily controlled with L22 andLaux satisfying the zero ripple condition.For the synchronous rectifier buck converter with the

half-bridge topology, shown in Fig. 1, L22 can be designedas a reduced value, which also results in the reduction ofLM. The effective inductance of the buck converter, Lefe,is reduced by the same amount as LM in Fig. 6. Becauseof the small inductance, the switching frequency of the con-verter can be increased by satisfying the ZVS condition ofthe synchronous rectifier. This makes the switching currentripple low and the efficiency high because the negative peakcurrent of the inductor can be decreased as a result of thesmaller inductance. This results in the reduction of the cir-culation energy. And also, this causes the AC component ofthe inductor current to be small. Thus, the core size of thecoupled and the auxiliary inductors is not increased in com-parison with the case where only the coupled inductor isused. Furthermore, in the ignition mode, as the ignitionvoltage is divided into St, L22 and Laux, the voltage stressof the triac switch is greatly reduced. Therefore the failureproblem of St can be avoided.

4.5 Sensitivity for the coupled inductor mismatch

In Fig. 6, Laux can be much larger than the leakage induc-tance of the coupled inductor. Here, k is nearly equal to1. Therefore the adjustment of the condition L22 þ Laux ¼LM is accomplished by the design of Laux. In addition, in


(12), Lp of the numerator is much larger than 12 k2. Forthese reasons, the sensitivity problem can be improved bythe addition of Laux

voVs

¼b2s

2þ 1

a4s4 þ a3s

3 þ a2s2 þ a1sþ 1

(13)

Along with Table 1, (13) shows the transfer function whenLaux = LM – L22 and LS = 0. Because of this mismatch,the attenuation operation somewhat falls off. For thedesign example, as shown in Fig. 7, at 140 kHz, theseattenuations are 251, 243 and 239 dB/decade for aperfect match, 5% mismatch and 10% mismatch, respect-ively. These are still useful for the ballast output filter.The ripple component to the total output current is lessthan 5% [1–3].

5 Frequency control of a synchronous rectifierbuck converter

The steady state terminal voltage of theMHD lamp varies in awide range even from the same lamp manufacturer (normally65–110 V). The lamp impedance and current are different foreach lamp in the constant power operation. It changes theoperating point of the synchronous rectifier buck converter.The inductor current contains a much greater circulating com-ponentwhen the lamp voltage is high as shown in Fig. 8c. Thiscirculating current reduces the efficiency of the ballast. Also,the higher peak-to-peak value of the inductor current increasesthe switch turn-off loss [11, 12].

The relation of the buck converter inductance, switchingfrequency, peak-to-peak inductor current, ILpkpk, and lamp

Fig. 6 Output filter of a buck converter with a coupled inductor

a Additional capacitor and an auxiliary inductorb T-equivalent circuit with an infinite output capacitorc T-equivalent circuit with a finite output capacitor

295

impedance can be understood through Fig. 8. To obtain theZVS operation, the inductor current, ILmin in (14), satisfiesthe condition in (15).

I0 ¼1

2ILpkpk � ILmin (14)

1

2LI2Lmin � CdsV

2IN (15)

In the fixed frequency operation, the peak-to-peak inductorcurrent of a high-voltage lamp (e.g. VLAMP ¼ 100 V) islarger than that of the low voltage one. This is because ofthe constant power control of the lamp. The switching fre-quency of the converter can be changed according to thelamp voltage for the fixed inductance in order to minimiseILmin. The peak-to-peak inductor current, ILpkpk, and thusthe circulating energy and the switching loss can bereduced.

Fig. 8d shows the change of the inductor current becauseof the increasing switching frequency. To operate with the

Fig. 7 Plots of the transfer function in (13)

Table 1: Parameters of the transfer function (13)

a4 f(LP þ LS)LM þ LPLSgC1C2

a3 f(LP þ LS)LM þ LPLSg/RLAMPC2)

a2 f(C1þ C2)LM þ LPC1þ LSC2g

a1 L11

b2 LSC

296

ZVS condition for the various lamp voltages, the inductanceof the buck converter should be designed for the case of alow-voltage lamp. If a high-voltage lamp is used for sucha ballast, the controller detects the lamp voltage andincreases the switching frequency of the converter to mini-mise ILmin in (15).

6 Design example and experimental results

In these experiments, 70 W MHD lamps manufactured byOsram are used. The lamp voltages are 70 V and 100 V.The first step of designing the MHD lamp ballast in Fig. 1is to decide L11 and C1 for the proper switch current inthe ignition mode. For a smaller switch current during theignition mode, L11 should be larger. For a fixed value ofLM, which is the effective inductance of the half-bridgeinverter in the steady state operation, increase of L11results in a smaller value of L22. If the inductance of L22is too small compared to that of L11, the problem of sensi-tivity could arise as mentioned previously in Section 4.5.Also, when the value of L11 is too large, the core size ofthe coupled inductor gets larger. C1 is determined alongfor the peak current rating of the MOSFETs. In this exper-iment, the values of L11, C2 and C1 are designed as 1.2 mH,0.22 mF and 2.2 nF, respectively. The peak switch current iscalculated as 2.7 A in the ignition mode with these par-ameters when the input voltage of the half-bridge inverterin Fig. 1 is 400 V DC.Considering the margin of ZVS operation of MOSFET

switch, L22 is designed for the switching frequency of100 kHz for the low-voltage lamp (VLAMP ¼ 65 V)and 160 kHz for the high-voltage lamp (VLAMP ¼ 110 V),which results in the peak-to-peak inductor current of6.0 A and 2.8 A, respectively. The designed values areL22 ¼ 20 mH, Laux ¼ 135 mH and n ¼ 0.129. The 70 Vand 100 V lamps are used for the actual experiment.Fig. 9 shows that the voltage of St is 14% of the lamp

voltage when Laux is not applied. The reduction of thevoltage can protect the triac switch from its over-voltagefailure. In addition, the peak inductor current is limited to1.7 A, which is a little smaller value than the one calculated.This is due to the damping by the parasitic resistance.

Fig. 8 Synchronous rectifier buck converter and its inductor current

a Synchronous rectifier buck converterb Inductor currentc Inductor current for various lamp voltagesd Reduction of the circulation current


Fig. 9 Operating waveforms in the ignition phase

Fig. 10 Operating waveforms in the steady state

a Converting waveformsb Waveforms during invertingc Inverting waveforms


Therefore the MOSFET switches, S1 and S2, can avoid theover-current failure in the ignition mode.

Fig. 10a shows the output current of the buck converter. Itshows that the ripple current of the lamp is 20 mApk–pk,which is about 3% of the lamp current. It can be seenfrom the inductor current that the circuit operates in theZVS condition of the synchronous rectifier. Fig. 10bshows the inductor current and the lamp current controlledby the proposed inverting method. The inductor currentshows that the circuit operates in the ZVS condition, justafter the inverting transition. Also, the inverting operationin the steady state is shown in Fig. 10c. The lamp currentincreases monotonically and the restrike voltage of thelamp is minimised. The current crest factor is measured as1.07 and 1.10 for 100 and 70 V lamps, respectively.

Fig. 11 shows the efficiency of various lamp voltages andswitching frequencies. The efficiency is measured for thehalf-bridge circuit with internal igniter excepting afront-end PFC circuit. Considering the lower bound of thelamp voltage (i.e. 65 V), the fixed switching frequency isdesigned as 90 kHz for the ZVS operation for the entirelamp voltage range. Two types of lamps (70 and 100 Vlamp) are used for this experiment. For the higher lampvoltage, the efficiency increases as the switching frequencyincreases because of the reduced circulating current andturn-off loss. For example, for the 100 V lamp, the effi-ciency increases about 2.5% for the variable frequencycontrol compared to the 100 kHz fixed frequency control.

7 Conclusion

A new high-efficiency ballast with an internal ignitercapable of lowering cost and size is proposed. Differentfrom the conventional half-bridge type ballasts, the pro-posed ballast contains an internal igniter and an outputripple current filter in one topology. The internal LC reson-ance of the buck converter and the coupled inductor filterare used for steady state ripple cancellation. To improvethe current commutation characteristics of the inverter, thePWM controller modulates the output current with asquare wave and the inverting current commutates monoto-nically. Therefore the proposed method prevents the

Fig. 11 Efficiency for various lamp voltages and switchingfrequency

297

inverter from having an overshoot and a ringing problem onthe inverted lamp current. As a result, the ZVS condition ismaintained and the restrike voltage of the lamp is mini-mised. A synchronous buck converter is applied for theDC/DC conversion stage. In this research, the operationmodes of the ballast are analysed. To improve the efficiencyof the ballast, we proposed a frequency control method.This method reduces the peak-to-peak inductor currentand the circulating current of the converter. This increasesthe efficiency of the ballast system by about 2.5% in com-parison with the fixed frequency control. The performanceof the proposed ballast has been verified by hardware exper-iments using 70 W MHD lamps.

8 References

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3 Denneman, J.W.: ‘Acoustic resonance in high frequency operated lowwattage metal halide lamps’, Philips J. Res., 1983, 38, pp. 263–272

4 Nishimura, H., Nagase, H., Uchihashi, K., Shiomi, T., and Fukuhara,M.: ‘A new electronic ballast for HID lamps’, J. Illum. Eng. Soc.,1988, pp. 70–76

5 Sun, Y., and Goriki, T.: ‘Driving circuit for high intensity dischargelamp electronic ballast’. US Patent 6,020,691

6 Shen, M., Qian, Z., and Peng, F.Z.: ‘Design of a two-stagelow-frequency square-wave electronic ballast for HID lamps’, IEEETrans. Indust. Appl., 2003, 39, (2), pp. 424–430

7 Partlow, W.D.‘The interaction of metal halide lamps and ballastduring reignition’, J. Illum. Eng. Soc., 1981, pp. 114–119

8 Franke, A., Gungle, W.C., Ring, J.F., andWaymouth, J.F.: ‘Reignitioncharacteristics of metal halide lamps and their effect on ballastdesign’, J. Illum. Eng. Soc., 1967, pp. 204–213

9 Cuk, S., and Middlebrook, R.D.: ‘Advances in switched-mode powerconversion’. (Tesla Co., Pasadena, CA, 1983), Vol. I–II)

10 Schutten, M.J., Steigerwald, R.L., and Sabate, J.A.: ‘Ripple currentcancellation circuit’. Proc. IEEE APEC, 2003, pp. 464–470

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high efficiency metal halide discharge lamp ballast with an internal igniter and low ripple lamp...

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