high-efficiency dcdc converter with high voltage

High-efficiency DC/DC converter with high voltagegain

R.J. Wai and R.Y. Duan

Abstract: A high-efficiency converter with high voltage gain applied to a step-up power conversionis presented. In the proposed strategy, a high magnetising current charges the primary winding ofthe coupled inductor, and the clamped capacitor is discharged to the auxiliary capacitor when theswitch is turned on. In contrast, the magnetising current flows continuously to boost the voltage inthe secondary winding of the coupled inductor, and the voltages across the secondary winding ofthe coupled inductor, the clamped capacitor and the auxiliary capacitor are connected in series tocharge the output circuit. Thus, the related voltage gain is higher than in conventional convertercircuits. Moreover, this scheme has soft-switching and voltage-clamped properties, i.e. the switch isturned on under zero-current switching and its sustainable voltage is comparatively lower than theoutput voltage, so that it can select low-voltage low-conduction-loss devices and there are noreverse-recovery currents within the diodes in this circuit. In addition, closed-loop controlmethodology is utilised in the proposed scheme to overcome the voltage drift problem of the powersource under the load variations. As a result, the proposed converter topology can promote thevoltage gain for a conventional boost converter with a single inductor, and deal with the problemof the leakage inductor and demagnetisation of the transformer for a coupled-inductor-basedconverter. Some experimental results via examples of a proton exchange membrane fuel cell powersource and a traditional battery are given to demonstrate the effectiveness of the proposed powerconversion strategy.

1 Introduction

DC/DC converters with steep voltage ratio are required inmany industrial applications. Example include: the front-end stage for clean-energy sources, the DC back-up energysystem for an uninterruptible power supply (UPS), high-intensity discharge lamps for automobile headlamps, andthe telecommunication industry [1�4]. The conventionalboost converter cannot provide such a high DC voltagegain, even for an extreme duty cycle. It also may result in aserious reverse-recovery problem and increase the rating ofall devices. As a result, the conversion efficiency is degradedand the electromagnetic interference (EMI) problem issevere in this situation [5]. To increase the conversionefficiency and voltage gain, many modified step-upconverter topologies have been investigated [6�12].

Although voltage-clamped techniques are manipulated inthe converter design to overcome the severe reverse-recoveryproblem of the output diode in high-level voltage applica-tions, overlarge switch voltage stresses remain and thevoltage gain is limited by the turn-on time of the auxiliaryswitch [6, 7]. Da Silva et al. [8] have presented a boost soft-single-switch converter, which has only a single activeswitch. It is able to operate with soft switching in a pulse-

width-modulation (PWM) way without high voltage andcurrent stresses. Unfortunately, the voltage gain is limitedbelow four in order to achieve the soft switching. Hirachiet al. [9] and Roh et al. [10] have employed coupledinductors to provide a high step-up ratio and to reduce theswitch voltage stress substantially, and they have alsoalleviated efficiently reverse-recovery problem of the outputdiode. In those cases, the leakage energy of the coupledinductor was another problem when the switch was turnedoff. It will result in high-voltage ripple across the switch dueto the resonant phenomenon caused by the leakage current.To protect the power devices, either a high-voltage-rateddevice with higher RDS(on) or a snubber circuit is usuallyadopted to deplete the leakage energy. These methods, willdegrade the power conversion efficiency. Zhao and Lee [11]have introduced a family of high-efficiency, high step-upDC/DC converters by adding only one additional diodeand a small capacitor. It can recycle the leakage energy andalleviate the reverse-recovery problem. In this scheme, themagnetic core can be regarded as a flyback transformer andmost of the energy is stored in the magnetic inductor.

Fuel cells appear to be one of the most efficient andeffective solutions to the environmental pollution problem[13�19]. A fuel cell is an energy conversion device thatproduces electricity by electrochemically combining fuel(hydrogen) and oxidant (oxygen from the air) gases throughelectrodes, across an ion conduction electrolyte. Thisprocess produces much higher conversion efficiency thanany conventional thermal–mechanical system because thesystem operates without combustion and extracts moreelectricity from the same amount of fuel. It has the meritsof: high efficiency, energy security, reliability, being pollu-tion-free and operating quietly. Fuel cells have been knownto science for more than 160 years and have recently

R.J. Wai is with the Department of Electrical Engineering, Yuan Ze University,Chung Li, Taiwan, R.O.C.

R.Y. Duan is with the Department of Industrial Safety and Health, HungKuang University, Tai Chung, Taiwan, R.O.C.

E-mail: [email protected]

r IEE, 2005

IEE Proceedings online no. 20045067

doi:10.1049/ip-epa:20045067

Paper received 22nd June 2004. Originally published online: 20th April 2005

IEE Proc.-Electr. Power Appl., Vol. 152, No. 4, July 2005 793

become the subject of intense research and development.Up to now, many demonstration projects have shown fuelcell systems to be feasible for portable power, transporta-tion, utility power and onsite power generation in a varietyof building applications.

For portable power, a fuel cell with a fuel container canoffer a higher energy density and more convenience thanconventional battery systems. Moreover, portable powerpacks using fuel cells can be lighter and smaller in volumefor an equivalent amount of energy. In transportationapplications, fuel cells offer higher efficiency and better part-load performance than conventional engines. In stationarypower applications, low emissions permit fuel cells to belocated in high-power requirement areas where they cansupplement the existing utility grid. Using fuel cells andhydrogen technology, electrical power can be deliveredwhen and where it is required, cleanly, efficiently andsustainably. Most research interest has focused on theproton exchange membrane (PEM) and solid oxide cellstacks. In particular, the proton exchange membrane fuelcell (PEMFC) has promising characteristics the following:first, the by-product waste is water; secondly, it has low-temperature operation; and thirdly, it uses a solid polymeras the electrolyte that reduces concerns related to construc-tion, transportation and safety issues [18]. Thus, it seems tobe a good alternative source for distributed generationsystems.

The aim of this study is to design a high-efficiency high-voltage-gain converter with a coupled inductor to regulate astable constant DC voltage. To achieve this goal, themanipulation of a coupled inductor, a lower-voltage-ratedswitch, a clamped circuit and an auxiliary circuit is adoptedto promote the voltage gain. Moreover, the problems of thestray inductance energy and the diode reverse-recoverycurrent in conventional converter strategies can also besolved so that it can achieve high-efficiency powerconversion. In addition, the feedback control methodologyis utilised in the proposed converter to overcome the voltagedrift problem of the power source under the load variations.

2 Converter design and analyses

The system configuration of the proposed convertertopology is shown in Fig. 1. It contains eight parts including

a DC input circuit, a primary-side circuit, a secondary-sidecircuit, a clamped circuit, an auxiliary circuit, a filter circuit,a DC output circuit and a feedback control mechanism.

The major symbol representations are summarised asfollows: VIN and II denote DC input voltage and current,respectively, and CIN is an input filter capacitor in the DCinput circuit. L1 and L2 represent individual inductors in theprimary and secondary sides of the coupled inductor (Tr),respectively. Q is a switch in the primary-side circuit and TQ

is a switch trigger signal produced by the feedback controlmechanism. C1 and D1 are clamped capacitor and diode,respectivelly, in the clamped circuit; C2 and D2 are high-voltage capacitor and diode in the secondary-side circuit; C3

and L3 are auxiliary capacitor and inductor in the auxiliarycircuit; DO and CO are output diode and filter capacitor,respectively, in the filter circuit. RO, VO and IO denoteoutput load, voltage and current in the DC output circuit,respectively.

The characteristic waveforms of the proposed high-efficiency converter are shown in Fig. 2. Figure 3 illustratesthe topological modes in one switching cycle and thedetailed operation stages will be introduced later. In Fig. 3,the coupled inductor in Fig. 1 is modelled as an ideal

L1

TQ

Tr

Q D1

II

VIN CIN

C1

L3

TQ

C3 L2 D2 Do

C2 Co Ro Vo

Io

−

+

−

+

driving circuit andtrigger signal

proportional-integral control

and PWM

voltage command

voltage feedback

feedback control mechanisamauxiliarycircuit

DC inputcircuit

DC outputcircuit

filtercircuit

clampedcircuit

primary-sidecircuit

secondary-sidecircuit

Fig. 1 System configuration of high-efficiency converter

t

t

t

t

t

t

t

t

t

mode 1

mode 2

mode 3

mode 4

mode 5

mode 6

t 0t 1t 0

iC2

iC1

iC3

iL1

iL2

iL3

iL1

iL3

iL2

iDO

iDS

iDS

iLm

iLm

�C1−3

�DS

�DS

�GS

�C1

�C2

�DO

�DO

�C3

iC3

iC1 iC 3

VO

iDO

iC1

iC2

t 2 t 3 t 4 t 5

Fig. 2 Characteristic waveforms

794 IEE Proc.-Electr. Power Appl., Vol. 152, No. 4, July 2005

transformer, a magnetising inductor (Lm), and a leakageinductor (Lk). The turns ratio (n) and coupling coefficient(k) of this ideal transformer are defined as

n ¼ N2=N1 ð1Þ

k ¼ Lm=ðLk þ LmÞ ð2Þwhere N1 and N2 are the winding turns in the primary andsecondary sides, respectively. For simplicity, the DC inputcircuit in Fig. 1 is denoted as a voltage source, VS. Thevoltages across the switch, the primary and secondarywinding of the ideal transformer, and the leakage inductorare denoted as vDS, vLm, vL2 and vLk, respectively. Moreover,the primary current of the coupled inductor is composed ofthe magnetising current (iLm) and the primary leakageinductor current (iL1). The secondary current (iL2) is formedby the magnetising current (iLm) through the idealtransformer, and its value is related to the turns ratio (n).In addition, the conductive voltage drops of the switch (Q)

and all diodes (DO, D1 and D2) are neglected to simplifycircuit analyses.

2.1 Operation stages

2.1.1 Mode 1 (t0–t1) (Fig. 3a): In this mode, theswitch (Q) was turned on for a span. Because themagnetising inductor (Lm) is charged by the input voltagesource (VS), the magnetising current (iLm) increasesgradually in an approximately linear way like the leakageinductor current (iL1). The clamped capacitor (C1) isdischarged to the auxiliary inductor (L3) and auxiliarycapacitor (C3) through the switch (Q). Thus, the auxiliaryinductor current (iL3) increases linearly at the change rate ofthe clamped capacitor voltage (vC1), and the switch current(iDS) is the summation of the currents iLm and iL3.According to the reflected positive voltage across thesecondary winding of the coupled inductor, the high-voltage diode (D2) is reverse-biased and the secondary

a

+

Q

+

−

−−

+

− +

+

+−

−CO VO

νLm

iDS D1

C3C2

iLm

Lk

iL1N1

N2 DO

D2

L3

VS

C1

L2

b

Q

+

−

+

− +

−

CO VOVS

iLm

LkLm

iL1 N1

DO

L3

C1

D2L2

C3C2

D1

N2

e

Q

+− +−

CO

+

−VO

+

−

VS

Lk Lm

iLmiL1

N1

L3

C1

C2

L2C3

D1

DOD2N2

f

Q

CO

+

−VO

+

−

VS

+−Lk Lm

iLmiL1

N1

L3

C1

C2

L2C3

D1

DO+− D2N2

c

Q

+

−CO

+

−VO

VS

Lk LmiLm

iL1N1

L3

C1

C2

D2L2

C3

D1

DON2

d

Q

−

CO

+

−

VO

+

−VS

Lk Lm

iLmiL1

N1

DO

L3

C1

D2L2

C3C2

D1

N2

Fig. 3 Topological modesa mode 1 [t0–t1];b mode 2 [t1–t2];c mode 3 [t2–t3];d mode 4 [t3–t4];e mode 5 [t4–t5];f mode 6 [t5–t0]


current (iL2) is equal to zero. During this period, the voltageof the clamped diode (D1), that is equal to the clampedcapacitor voltage (vC1) under reverse-biased, belongs to thelow-voltage level. Therefore, the Schottky diode can beadopted to further alleviate the conduction loss and reverse-recovery current.

2.1.2 Mode 2 (t1–t2) (Fig. 3b): At time t¼ t1, theswitch (Q) is turned off. At this time, the leakage inductorcurrent (iL1) and auxiliary inductor current (iL3) start tocharge the parasitic capacitor of the switch. This intervalends when the switch voltage (vDS) is charged up and equalto the clamped capacitor voltage (vC1).

2.1.3 Mode 3 (t2–t3) (Fig. 3c): When the switchvoltage (vDS) is higher than the clamped capacitor voltage(vC1), the clamped diode (D1) conducts to transmit theenergy of the leakage inductor (Lk) into the clampedcapacitor (C1). In the meantime, the auxiliary circuitcurrents (iC3 and iL3) start to discharge into the outputterminal through the high-voltage capacitor (C2) and outputdiode (DO). In performing the analysis, the clampedcapacitor (C1) and auxiliary capacitor (C3) are assumed tobe large enough so that the related across voltages can beconsidered to be ripple-free. During this mode, the leakageinductor current (iL1) is shared out between the auxiliarycapacitor current (iC3) and clamped diode current (iD1). Toalleviate the spike voltage of the switch (Q) caused by thestray inductance, the switch (Q), clamped capacitor (C1)and clamped diode (D1) must be closely connected. Itshould be noted that the clamped diode (D1) should be afast conductive device for clamping the switch voltage, andthe voltage ratings of vD1 and vDS are the same as theclamped capacitor voltage (vC1). Thus, the Schottky diodewith low consumptive power and conductive voltage maybe a better choice.

2.1.4 Mode 4 (t3–t4) (Fig. 3d): To release theenergy of the leakage inductor (Lk) persistently, the currentiL1 has to decrease continuously. At the start of this mode,the high-voltage diode (D2) is forward-biased due to thereflected negative voltage across the secondary winding ofthe coupled inductor, and the secondary current (iL2)increases in a linearly to charge the output filter capacitor(CO) through the diodes DO and D2. Thus, the summationof the currents iC2 and iL2 is passed through the outputdiode (DO). When the leakage inductor current (iL1) decaysto a negative value and the secondary current (iL2) increasesto its maximal magnitude gradually, the current status ofthe high-voltage capacitor (C2) changes from the dischargestate to the charge state at the end of this mode. This modeends when the current magnitude of iC2 is equal to thesecondary current (iL2) and the output diode (DO) is cut offsimultaneously.

2.1.5 Mode 5 (t4–t5) (Fig. 3e): The auxiliary in-ductor (L3) starts to discharge into the auxiliary capacitor(C3) through the primary side of the coupled inductor andthe input voltage source (VS) at the beginning of this mode.The primary current, which comprises the magnetisingcurrent (iLm) and the auxiliary inductor current (iL3), isinducted to the secondary side of the coupled inductorthrough the ideal transformer, and the current of iL2 chargesthe high-voltage capacitor (C2). It should be noted that thecomponent of iL3 which passed through the input voltagesource (VS) belongs to the circulating current. Fortunately,this circulating phenomenon will not result in seriousconversion loss because the current of iL3 is much smaller

than the switch current (iDS), the voltage source is a lowvoltage type, and most of the auxiliary inductor energy istransmitted effectively to the output circuit.

2.1.6 Mode 6 (t5–t6) (Fig. 3f): At time t¼ t5, theswitch (Q) is turned on. Since the clamped diode (D1) is alow-voltage-rated Schottky diode, it will be cut off promptlywithout reverse-recovery current when the switch (Q) isturned on. Moreover, the reversal primary leakage current(iL1) is derived from the auxiliary circuit (iC3 or iL3) at timet¼ t5, and the currents in the series path cannot changeimmediately due to the continuous flow property of theprimary-side leakage inductor (Lk). Therefore, one cannotderive any current from these three paths including theprimary-side circuit, auxiliary circuit and clamped circuit.Consequently, the switch (Q) is turned on under zero-current switching (ZCS) and this soft switching property ishelpful for alleviating the switching loss. Due to thesummation of the voltages vDS, vC3, vC2 and vDO are equalto the output voltage (VO), and the voltages of vC2 and vC3

are ripple-free, the voltage rating of the output diode (DO) isthe same as that of the switch (Q). Thus, a low-voltage-rated Schottky diode also can be adopted. After time t¼ t5,the leakage inductor current (iL1) raises quickly and thesecondary current (iL2) decays synchronously since theleakage inductor (Lk) is charged again by the input voltagesource (VS). When the current of iL1 is equal to themagnetising current (iLm), the high-voltage diode (D2) isreverse-biased. It begins the next switching cycle and repeatsthe operation in mode 1.

Remark 1: Because the clamped capacitor (C1) isdischarged into the auxiliary capacitor (C3) in series duringthe modes in Fig. 3a, b, e and f, the charge current of C1 isequal to the discharge current of C3 in modes 3 and 4 (i.e.iC1¼�iC3) according to the capacitor charge balance.Moreover, the current iC2 is approximately equal to iC3 inmode 3 since the relations of iC2¼ iC3+iL3 and iL3{iC3hold. Therefore, the initial current of the high-voltagecapacitor (C2) at t¼ t2 is iC2D0.5iLm via the relation ofiLm¼�iC1+iL3+iC3D�iC1+iC3¼ 2iC3D2iC2 in mode 3.

2.2 Formula derivationWhen the switch (Q) is turned on, the voltages across themagnetising inductor (Lm) can be denoted via (2) as

vLm ¼ kVS ð3ÞBy applying the volt–second balance of the magnetisinginductor, when the switch (Q) is turned off, its voltage isgiven by

vLm ¼ DkVS=ð1� DÞ ð4Þwhere D is the duty cycle of the switch (Q). Due to theconcept of zero average voltage across the leakage inductor

iLm

iC2

A1 = A2

A2

iLm / 2

iLm / n

A1

iLK

tL tC

TS

D . TS

∆t∆t

Fig. 4 Relationship between charge and discharge currents of high-voltage capacitor


(Lk) over one period, the reset time of the leakage inductor(Lk) is related to its voltage (vLk) during mode 4. Figure 4shows the relationship between charge and dischargecurrents of the high-voltage capacitor (C2) by assumingthat the discharge current at the beginning is equal to half ofthe magnetising current (iLm). Because the high-voltagecapacitor current (iC2) needs to maintain the conservation ofenergy, the following relation can be found by making thecharge area equal to the discharge area:

DL ¼ tL=TS ¼ 4½Dt þ ð1� DÞTS �=½ðnþ 4ÞTS � ð5Þwhere DL is the reset duty cycle, TS is the switching period,tL is the time of mode 4, and Dt is the transition timebetween constant charge current and zero current. Since thetransition time (Dt) is much smaller than the switch dutycycle (D), for simplicity, the reset duty cycle in (5) can berewritten as

DL ¼ tL=TS ¼ 4ð1� DÞ=ðnþ 4Þ ð6ÞDuring mode 4, the voltage of vLk can be calculatedaccording to the volt–second balance of the leakageinductor (Lk) as

vLk ¼Dðnþ 4Þð1� kÞ

4ð1� DÞ VS ð7Þ

Thus, the clamped capacitor voltage (vC1) can be repre-sented via (4) and (7) as

vC1 ¼ vDS ¼ VS þ vLk þ vLm

¼ VS

1� D1þ nD

4ð1� kÞ

� � ð8Þ

Assume that the capacitors of C1 and C3 are sufficientlylarge, then their voltages (vC1 and vC3) can be considered tobe ripple-free and described as

vC3 ¼ DvC1 ð9ÞNote that, the voltage of vC1 is equal to the switch voltage(vDS), and the leakage inductor voltage (vLk) will increasethe voltage stress of the switch (Q), especially under heavyload. Moreover, the energy of the clamped capacitor (C1) isdischarged to the auxiliary capacitor (C3) when the switch isturned on so that the voltages of the auxiliary and outputcircuits can be increased relatively.

When the switch (Q) is turned off, the voltage across thesecondary winding of the ideal transformer can berepresented via (4) as

vL2 ¼ vC2 ¼ nvLm ¼ DknVS=ð1� DÞ ð10ÞBecause the voltages of vDS, vC3, and vL2 charge the outputcapacitor (CO) and load (RO) during modes 3 and 4, theoutput voltage (VO) can be calculated as

VO ¼ vC1 þ vC3 þ vL2

¼ 1þ Dþ Dnk1� D

VS þnDð1� kÞð1þ DÞ

1� DVS ð11Þ

Therefore, the voltage gain of the propose high-efficiencyconverter can be expressed as

GV ¼VO

VS¼ 1þ Dþ Dnk

1� Dþ nDð1� kÞð1þ DÞ

1� Dð12Þ

Substituting k¼ 0.98 and n¼ 3, 5, 7 into (12), the curve ofthe voltage gain (GV) with respect to the duty cycle (D) isdepicted in Fig. 5a, where the lines labelled with starsdenote the voltage gain curve of the proposed converter andthe solid lines represent the one in [11]. As can be seen fromthis Figure, the voltage gain of the high-efficiency converteris higher than a coupled-inductor-based converter in [11],especially in the region within larger duty cycle. Moreover,

the voltage gain curve by substituting k¼ 0.9–1 and n¼ 5into (12) is depicted in Fig. 5b. This Figure shows that thevoltage gain (GV) is less sensitive to the coupling coefficient,k. For simplicity, the coupling coefficient (k) is set at one,then (8) and (12) can be rewritten as

vDS ¼ VO=ð1þ Dþ DnÞ ð13Þ

GV ¼VO

VS¼ 1þ Dþ Dn

1� Dð14Þ

Moreover, the switch duty cycle (D) can be calculated via(14) as

D ¼ GV � 1

GV þ 1þ nð15Þ

3 Fuel cell operating principle

The basic fuel cell concept involves converting chemicalenergy directly into electrical energy. It produces electricityby electrochemically combining fuel (hydrogen) and oxidant(oxygen from the air) gases through electrodes and acrossan ion-conducting electrolyte. The fuel cell is composed oftwo electrodes, an anode, cathode, the catalyst and anelectrolyte, as illustrated in Fig. 6 [15]. The main function ofthe electrode is to bring about a reaction between thereactant and the electrolyte. The anode, used as the negative

- coupledinductor- based converter in [11]

* proposed high step-up converter

coupling coefficient k = 0.98

n = 7

n = 5

n=3

40

35

30

25

20

15

10

5

00.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9

volta

ge g

ain,

G v

volta

ge g

ain,

Gv

duty cycle, D

a

turns ratio n = 5k = 0.98

k = 1

k = 0.9

k = 0.95

0.36 0.38 0.4 0.42 0.44 0.46 0.48 0.5 0.52 0.54 0.56

duty cycle, D

b

5

5.5

6.5

7.5

8.5

9.5

9

10

7

8

6

Fig. 5 Voltage gain curvea Coupling coefficient k¼ 0.98b Turns ratio n¼ 5


post in the fuel cell, disperses the hydrogen gas equally overthe entire catalyst surface and conducts the electrons for useas power in an external circuit. The cathode, used as thepositive post in the fuel cell, distributes the oxygen fed to itonto the catalyst surface and conducts the electrons backfrom the external circuit. The catalyst is a special materialused to facilitate the oxygen and hydrogen reaction.

According to the chemical characteristics of the electro-lyte used as the ion conductor in the cells, the mostpromising types are classified as: first, PEMFC and directmethanol fuel cell (DMFC), which use a polymermembrane as the electrolyte. Secondly, a phosphoric acidfuel cell (PAFC), which uses pure phosphoric acid as theelectrolyte. Thirdly, a molten carbonate fuel cell (MCFC),which uses a molten mixture, sodium, and potassium car-bonates as the electrolyte. Finally, a solid oxide fuel cell(SOFC), which uses a ceramic material as the electrolyte [17].

The PEMFC is one of the most promising fuel cell types,and is often considered a potential replacement for theinternal combustion engine in transportation applications[17]. The PEMFC consists of porous carbon electrodesbound to a thin sulphonated polymer membrane. Theanode, cathode, and net cell reactions of the PEMFC can berepresented as

anode reaction H2 ! 2Hþ þ 2e� ð16Þ

cathode reaction1

2O2 þ 2Hþ þ 2e� ! H2O ð17Þ

net cell reaction H2 þ1

2O2 ! H2O ð18Þ

where the mobile ion is H+. The membrane electrodeassembly (MEA) is sandwiched between two collectorplates that provide an electrical path from the electrodes tothe external circuit. Flow channels cut into the collectorplates distribute reactant gases over the surface of theelectrodes. Individual cells consisting of collector plates andMEAs are assembled in series to form a fuel cell stack.

Fuel cell generation systems have been receiving moreattention in recent years due to the advantages of their highefficiency, low aggression to the environment, no movingparts and superior reliability and durability. Due to theelectrochemical reaction, fuel cell has the power quality oflow voltage and high current. However, the fuel cell stackwith high output voltage is difficult to fabricate and it mayfail when any single cell is inactive. In addition, the outputvoltage is easily varied with respect to the load variations.To satisfy the requirement of high-voltage demand, a stableboost converter with high voltage gain and superiorconversion efficiency is necessary to utilise the fuel cellenergy more efficiently and satisfy the requirement of high-voltage demand in specific applications. The validity of theproposed converter in Section 2 is verified by the followingexperimental results.

4 Experimental results

To verify the effectiveness of the designed topology, aPEMFC system and a traditional 24 V-battery are utilisedfor low-voltage power sources in the proposed high-efficiency converter. The PEMFC system used in this studyis the Nexa power module manufactured by the BallardCompany. The system operates on ambient air and cleanpressurised hydrogen fuel. The fuel cell system consists of a48-cell stack of the PEM type, mechanical auxiliaries, andelectronic control module.

In experimentation, the high-efficiency converter isdesigned initially to operate from the fuel cell variabilityDC input, VIND26�38 V, to deliver a constant DC output,VO¼ 200 V. The key design step is to determine the turnsratio of the coupled inductor, the duty cycle and maximalvoltage of the switch for the PEMFC source with variabilityDC voltage. By considering VO¼ 200 V, k¼ 1, and (13)–(15), Figure 7 depicts the relationship of the duty cycle (D)and switch voltage (vDS) with respect to the input voltage(VS) under different turns ratios. Because the maximal fuelcell voltage is near to 40 V, it is assumed that the maximalvalue of the switch voltage can be clamped at 63 Vaccording to Fig. 7b. Although a higher turns ratio has alower switch voltage, it will result in a smaller duty cycle anda higher peak switch current. For a compromise, the turnsratio (n) and the duty cycle (D) are chosen as n¼ 5 andD¼ 0.5 in this study. To solve the problem of the fuel cell

A

e− e−

fuel inlet(hydrogen)

depletedfuel out

gaschannel

gaschannel

useful power

or+ ions

− ions

electrolyte

anode cathode

oxidant inlet(from humid air)

unused air andproduct water

Fig. 6 Fuel cell basic configuration

15 20 25 30 35 40 45 50

switc

hvol

tage

, VD

S

input voltage, Vsb

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.115 20 25 30 35 40 45 50

duty

cyc

le, D

input voltage, Vs

a

D

Vo = 200 k = 1

design point

Vo= 200

DL

k = 1

n = 34

4 5 6

56

n = 3

n = 7

n = 7

n = 3

design point

100

90

80

70

60

50

40

30

20

10

0

546 n = 7

Fig. 7 Relationship of duty cycle and switch voltage with respect toinput voltage under different turns ratios


output voltage varied with the load variations, the proposedconverter with DC voltage feedback control is utilised toensure system stability, and a PWM control IC TL494 is

adopted to achieve this goal of feedback control. Aprototype with the following specifications has been de-signed to illustrate the design procedure given in Section 2.

iL1

�DS

(25 A/div)

(20 V/div)

(2 µs/div)

0 V

0 A

iD2

�D2

(5 A/div)

(200 V/div)

(2 µs/div)0 V

0 A

iL1

iL2

�DS

(25 A/div)

(10 A/div)

(50 V/div)

(2 µs/div)0 V

0 A

0 A

iL3

�D1

�DS (50 V/div)

(50 V/div)

(2 µs/div)

(2 A/div)

0 V

0 A

0 V

iL1

�D1

�DO

(50 V/div)

(50 V/div)

(2 µs/div)

(25 A/div)

0 V

0 A

0 V

iL1

VIN

II (10 A/div)(10 V/div)(2 µs/div)

(25 A/div)

0 V A

0 A

0

a b

c d

e f

iL3

�C3

�C1

(2 A/div)

(20 V/div)

(2 µs/div)0 V

0 A

VO

�C2

(50 V/div)

(5 µs/div)0 V

IO

VO

(200 µs/div)

(1 A/div)

(50 V/div)

0 V

0 A

70 W ↔ 780 W

g h

i

Fig. 8 Experimental voltage and current responses of high-efficiency converter for PEMFCPO¼ 700 W, VO¼ 200 V


switchingfrequency fS¼ 100 kHz

coupledinductor L1¼ 11.6 mH, L2¼ 290 mH, N1:N2¼ 3:15,

k¼ 0.98, EE-55 core, L3¼ 25 mHcapacitor CIN¼ 3300 mF/50 V*2, C1¼ 10 mF/100 V,

C2¼ 20 mF/250 V, C3¼ 10 mF/100 V,CO¼ 47 mF/250 V

switch Q: IRFP2907 (75 V/209 A, RDS(on)¼ 4.5mO), TO-247

diode D1, DO: Schottky diode STPS20H100CT,TO-220AB (100 V/2*10 A) D2: SFA1606G,TO-220AB (400 V/16 A)

The experimental voltage and current responses of theproposed high-efficiency converter operating at 700 W-output power is depicted in Fig. 8. From Fig. 8a, it can beseen that the switch voltage (vDS) which is clamped below 65V is much smaller than the output voltage, VO¼ 200 V.Figure 8c, the fuel cell output current (II) keeps a low-ripplewaveform, and the ZCS property of the switch (Q) iseffective to retard the spike current. According to Fig. 8d�f,the reverse-recovery current in the high-voltage diode (D2)can be alleviated effectively, and the voltages of the diodes(DO and D1) are clamped near to the switch voltage (vDS).The ringing phenomenon in Fig. 8f is caused by theresonance between the leakage inductor (Lk) of the coupledinductor and the parasitic capacitor of the output diode(DO) when the switch (Q) is turned on; therefore, a suitablesnubber circuit is necessary to alleviate the peak voltage ofthe high-voltage diode (D2). From Figs. 8g and 8h, thevoltages of the clamped capacitor (C1), the high-voltagecapacitor (C2) and the auxiliary capacitor (C3) are close toconstant values. To examine the robust performance of theproposed converter scheme, the experimental result of the

output voltage (VO) and output current (IO) under the stepload variation between light load (70 W) and heavy load(780 W) is depicted in Fig. 8i. As can be seen from thisFigure, the converter output voltage, VO¼ 200 V, is lesssensitive to the load variations due to the utilisation a smallcoupled-inductor and a closed-loop control, and the outputvoltage ripple is also extremely slight as a result of the highswitching frequency.

For the sake of verifying the effectiveness of the proposedconverter for different output powers, the experimentalswitch voltage and current responses at 70 W, 328 W, 502W and 710 W-output powers are given in Fig. 9. As can beseen from these results, the duty cycle (D) needs to be raisedto keep a constant output voltage (VO¼ 200 V) since thefuel cell voltage drops when the output power increases.

Po = 710 w

(2 µs/div)

(50 V/div)

(25 A/div)

0A

0V

VDS

iL1

d

Po = 328 w (2 µs /div)

(20 V/div)

(10 A/div)

0A

0V

VDS

iL1

b

Po = 70 w

(2 µs /div)

(20 V/div)

(10 A/div)

0A

0V

VDS

iL1

a

Po = 502 w

(2 µs /div)

(50 V/div)

(25 A/div)0A

0V

VDS

iL1

c

Fig. 9 Experimental switch voltage and current curves of high voltage gain converter for PEMFCVO¼ 200 Va PO¼ 70 Wb PO¼ 328 Wc PO¼ 502 Wd PO¼ 710 W

98

96

94

92

90

88

86

840 100 200 300 400 500 600 700 800

conv

ersi

on e

ffici

ency

, %

40

38

36

34

32

30

28

26

inpu

t vol

tage

, V

output power, W

* conversion efficiency-output powero input voltage-output power

Fig. 10 Conversion efficiency and fuel cell voltage for PEMFCVO¼ 200 V under different output powers


Moreover, the maximal switch voltage is also clampedbelow 65 V. It should be noted that the oscillated switchvoltage in Fig. 9a is caused by the resonance between theleakage inductance (Lk) and the switch parasitic capacitor atlow power output [20]. It is helpful to alleviate the switchingloss in mode 6. Figure 10 summarises the experimentalconversion efficiency of the proposed converter and fuel cellvoltage under different output powers. From the experi-mental results, the output voltage of the fuel cell decreasesas the output power increases, and is easily varied withrespect to the load variations. To solve this phenomenon,the proposed high-efficiency converter with DC voltagefeedback control is utilised in this study to ensure the systemstability. In addition, the conversion efficiency at 40 Woutput power is over 92.5% and the maximal efficiency isabout 97% at 410 W output power, which is comparativelyhigher than conventional converters.

In the following experimentation, a traditional battery isadopted for another low-voltage power source to furtherexamine the characteristics of high voltage gain andsuperior conversion efficiency of the proposed converterstrategy. The high-efficiency converter is designed to operatefrom the battery DC input, VIN¼ 24 V, to deliver aconstant DC output, VO¼ 200 V. The experimental switchvoltage and current responses at 70 W, 200 W, 400 W and800 W-output powers are given in Fig. 11. These experi-mental results agree well with those obtained from thePEMFC system given in Fig. 9. Comparing these Figures, itis obvious that the switch clamped voltages in Fig. 11 arelower than those in Fig. 9 since the switch voltage (vDS) isproportional to the input voltage (VS) according to Fig. 7b.Figure 12 summarises the experimental conversion effi-ciency of the proposed converter under different outputpowers. At the examined condition, the maximal efficiencyof over 96.5% is slightly smaller than the one in Fig. 10 as

the voltage gain in Fig. 12 is larger than 8%. Consequently,the designed object of high-efficiency, high step-up andvoltage-clamped characteristics can be achieved accordingto the aforementioned experimental results.

Zhao and Lee [11] investigated a family of high-efficiency,high step-up DC/DC converters with simple topologies,where four paralleled 250 V voltage-rating switches(RDS(on)¼ 0.28O) were adopted for high-end server systems.Though the switch voltage rate is lower than the outputvoltage (380 V), it is greater than the maximum inputvoltage (75 V). In general, the selection of high-voltage-rateMOSFET switches will result in higher conduction lossunder the same current amount. Although the proposedscheme has additional clamped and auxiliary circuits incomparison with [11], the clamped circuit is helpful to curb

po = 800 W

(2 µs/div)

po = 200 W

(2 µs/div)

(20 V/div)

(10 A /div)

0A

0V

<< main:4k >>

(50 V/div)

(25 A /div)

0A

0V

<< main:4k >>

po = 70 W

(2 µs /div)

(20 V/div)

(10 A /div)

0A

0V

<< main:4k >>

po = 400 W

VDSVDS

VDSVDS

iL1iL1

iL1iL1

(2 µs /div)

(50 V/div)

(25 A /div)

0A

0V

<< main:4k >>

a

dc

b

Fig. 11 Experimental voltage and current responses of high-efficiency converter for 24 V batteryVO¼ 200 Va PO¼ 70 Wb PO¼ 200 Wc PO¼ 400 Wd PO¼ 800 W

98

96

94

92

90

88

86

840 100 200 300 400 500 600

conv

ersi

on e

ffici

ency

, %

output power, W

VIN = 24 V Vo = 200 V

Fig. 12 Conversion efficiency for 24 V batteryVO¼ 200 V under different output powers


the switch voltage stress, and the voltage of the auxiliarycircuit is effective to further decrease the switch voltagecapacity. Moreover, the proposed circuit has much lessRDS(on) than does the power switch in [11], and the switchconduction loss caused by the smaller inductor current (iL3)can nearly be ignored. Thus, the proposed scheme possesseshigh conversion efficiency over a wide load range.

5 Conclusions

This study has successfully developed a high-efficiency DC/DC converter with high voltage gain which has beensuccessfully applied to a PEMFC system and a traditionalbattery. According to the experimental results, the maximalefficiency was measured to be over 96.5%, which iscomparatively higher than conventional converters withthe same voltage gain. The efficiency of this circuit can befurther optimised by building the circuit more compactly viaa PCB layout. The newly designed converter circuit has thefollowing improvement compared to the previous works:first, the stray energy can be recycled by a clamped circuit tocontribute toward the step-up and voltage-clamped objectso that the circuit layout is easy in practical applications.Secondly, because the switch voltage stress is proportionalto the DC input voltage, it is suitable for a DC powerconversion mechanism with high voltage gain. Thirdly, thereverse-recovery problems of diodes can be solved due tothe utilisation of Schottky diodes and the limitation of thecurrent changing rate by the leakage inductor. Finally, thevoltage drift problem of the power source under the loadvariations can be coped with by utilising a small coupled-inductor and a closed-loop control. This high-efficiencyconverter topology provides designers with an alternativechoice to convert renewable energy efficiently, and it alsocan be extended easily to other power conversion systems tosatisfy high-voltage demands.

6 Acknowledgments

The authors acknowledge the financial support of theNational Science Council of Taiwan, Republic of Chinathrough grant NSC 92-2623-7-155-014 and the Ministry ofEconomic Affairs of Taiwan, Republic of China throughgrant 92-EC-17-A-05-S1-0012. The authors would also liketo express their gratitude to the referees and the associateeditor for their useful comments and suggestions.

7 References

1 Barbi, I., and Gules, R.: ‘Isolated DC-DC converters with high-outputvoltage for TWTA telecommunication satellite applications’, IEEETrans. Power Electron., 2003, 18, pp. 975–984

2 Abutbul, O., Gherlitz, A., Berkovich, Y., and Ioinovici, A.: ‘Step-upswitching-mode converter with high voltage gain using a switched-capacitor circuit’, IEEE Trans. Circuits Syst. I, 2003, 50, pp. 1098–1102

3 Tseng, K.C., and Liang, T.J.: ‘Novel high-efficiency step-upconverter’, IEE Proc., Electr. Power Appl., 2004, 151, pp. 182–190

4 Harry, J.E., and Hoare, D.W.: ‘Electronic power supplies for high-density discharge (HID) lamps’, IEE Eng. Sci. Educ. J., 2000,pp. 203– 206

5 Mohan, N., Undeland, T.M., and Robbins, W.P.: ‘Power electronics:converters, applications, and design’ (Wiley, New York, 1995)

6 Jovanovic, M.M., and Jang, Y.: ‘A new soft-switched boost converterwith isolated active snubber’, IEEE Trans. Ind. Appl., 1999, 35,pp. 496–502

7 Duarte, C.M.C., and Barbi, I.: ‘An improved family of ZVS-PWMactive-clamping DC-to-DC converters’, IEEE Trans. Power Electron.,2002, 17, pp. 1 7

8 Da Silva, E.S., Dos Reis Barbosa, L., Vieira, J.B., De Freitas, L.C.,and Farias, V.J.: ‘An improved boost PWM soft-single-switchedconverter with low voltage and current stresses’, IEEE Trans. Ind.Electron., 2001, 48, (6), pp. 1174–1179

9 Hirachi, K., Yamanaka, M., Kajiyama, K., and Isokane, S.: ‘Circuitconfiguration of bidirectional DC/DC converter specific for smallscale load leveling system’. Proc. IEE Power Conversion Conf., 2002,pp. 609

10 Roh, C.W., Han, S.H., and Youn, M.J.: ‘Dual coupled inductor fedisolated boost converter for low input voltage applications’, Electron.Lett., 1999, 35, pp. 1791–1792

11 Zhao, Q., and Lee, F.C.: ‘High-efficiency, high step-up DC-DCconverters’, IEEE Trans. Power Electron., 2003, 18, (1), pp. 65–73

12 Luo, F.L.: ‘Re-lift circuit: a new DC-DC step-up converter’, Electron.Lett., 1997, 33, pp. 5–7

13 Kyoungsoo, R., and Rahman, S.: ‘Two-loop controller for maximiz-ing performance of a grid-connected photovoltaic-fuel cell hybridpower plant’, IEEE Trans. Energy Convers., 1998, 13, (3), pp. 276–281

14 Lukas, M.D., Lee, K.Y., and Ghezel-Ayagh, H.: ‘Development of astack simulation model for control study on direct reforming moltencarbonate fuel cell power plant’, IEEE Trans. Energy Convers., 1999,14, (4), pp. 1651–1657

15 Lukas, M.D., Lee, K.Y., and Ghezel-Ayagh, H.: ‘An explicit dynamicmodel for direct reforming carbonate fuel cell stack’, IEEE Trans.Energy Convers., 2001, 16, (3), pp. 289–295

16 Boudghene Stambouli, A., and Traversa, E.: ‘Fuel cells, an alternativeto standard sources of energy’, Renew. Sustain. Energy Rev., 2002, 6,pp. 297–306

17 Ellis, M.W., Von Spakovsky, M.R., and Nelson, D.J.: ‘Fuel cellsystems: efficient, flexible energy conversion for the 21st century’, Proc.IEEE, 2001, 89, (12), pp. 1808–1818

18 Correa, J.M., Farret, F.A., Gomes, J.R., and Simoes, M.G.:‘Simulation of fuel-cell stacks using a computer-controlled powerrectifier with the purposes of actual high-power injection applications’,IEEE Trans. Ind. Applicat., 2003, 39, (4), pp. 1136–1142

19 Green, K., and Wilson, J.C.: ‘Future power sources for mobilecommunications’, IEE Electron. Commun. Eng. J., 2001, pp. 43–47

20 Lu, D.C., Cheng, D.K.W., and Lee, Y.S.: ‘A single-switch continuous-conduction-mode boost converter with reduced reverse-recovery andswitching losses’, IEEE Trans. Ind. Electron., 2003, 50, pp. 767–776


high-efficiency dcdc converter with high voltage

Documents