Analog Front End Design For ADSLR. K. Hester
Texas Instruments IncorporatedDallas, Texas, USA
Outline
• Canonical ADSL System Diagram
• Signal Characteristics
• Impairments
• Line Coupling Circuits
• Transmitter Design
• Receiver Design
ADSL System
Transmitter
Receiver
Transmitter
Receiver
DOWNSTREAM <= 8 Mbps
UPSTREAM <= 800 kbps
POTSSplitter
POTSSplitter
up to 19 kft
Ethernet,PCI, USB, ...
BackboneDigital
Network
Central Office Remote Terminal
POTS
Subscriber Loop
Unlike V.xx dial-up modems, the signal is not terminated by a voice-band line card
Signal Characteristics
• DMT modulation
• Frequency plan
• Capacity
• Peak-to-Average Ratio and BER
• Continuous-time/Discrete-time PSD
Discrete Multi-Tone Signaling(ODFM)
• Employs many narrow-band (4.3125 kHz) sub-carriers
• Low baud frequency (4 kHz)
• Data dynamically assigned to sub-carrier according to SNR
• IFFT/FFT used to modulate/demodulate in blocks
time
Frame k+1
cyclic prefix
Frame k
40 (D)5 (U)
samples
512 (D)64 (U)
samples
40 (D)5 (U)
samples
512 (D)64 (U)
samples
cyclic prefix
IFFTprepend
prefix FFTstrip
prefixTEQ
Frequency Plan
Upstream DownstreamPOTS
1.1 MHzf
Power Spectrum
Frequency Division Duplexed
26 KHz
Bi-directional DownstreamPOTS
1.1 MHzf
Power Spectrum
Echo-Cancelled
26 KHz
ITU G.hs training protocol guarantees FDD-EC modem interoperability
Annex A Frequency Plan Example
6 31 38 255
• Sub-carrier frequencies: n*4.3125 kHz
• Upstream n= 6, …, 31
• Echo-canceling downstream n= 6, …, 255
• Frequency division downstream n= 33, …, 255
• Modulation: 4QAM … 32768QAM
• Subcarrier data rate capacity: 2-15 bits per baud period
Subcarrier Capacity versus SNR
10 20 30 40 50 60 70
2
4
6
8
10
12
14
SNR (dB)
Cap
acity
(bits
/bau
d)
- 140 - 130 - 120 - 110
500000
1 ×106
1.5×106
2 ×106
2.5×106
3 ×106
Effective Noise Floor at Tip/Ring (dBm/Hz)
Cap
acity
(b/s
ec)
upstream
downstream
Maximum 15kft 26awg Data Rate Capacity
More on Frequency Plan
Annex Upstream Downstream
• A (POTS) 6 – 31 6 (33) – 255
• A+ 6 – 31 6 (33) – 511
• B (ISDN) 29 – 63 29 (60) – 255
• B+ 29 – 63 29 (60) – 511
• C (TCM-ISDN) 33 – 63 33 – 255
• H (TCM-ISDN) 6 – 255 6 – 255
• I 1 – 31 1 – 255
• J 1 – 31 (63) 1 – 255
Peak-to-RMS Ratio • Subcarriers are statistically independent, so sum of N has Gaussian probability
distribution in time-domain, where variance is 0.43152* N volt2.
• When N=250, the RMS signal is 3.28V, and a 16dB peak-to-RMS ratio corresponds to 41.4 VPP.
0.0001 0.0002 0.0003 0.0004 0.0005 0.0006 0.0007
- 15
- 10
- 5
5
10
15
11.3dB
Sign
al (V
olts
)
Time
Gaussian Signal Statistics• Sub-carriers are statistically independent, so sum of N sub-carriers has
Gaussian probability (central limits theorem)
• Signal clipping, either analog or digital, transmitter or receiver, wrecks SNR and creates transmission errors
• Must support signal swing that corresponds to desired bit error rate
•16dB peak-to-RMS ratio ( peaks to 6.3 VRMS) corresponds to a 10-7 BER.
- 7.5 - 5 - 2.5 2.5 5 7.5
0.1
0.2
0.3
0.4
Sub-Carrier Power Spectral Density (dBm/Hz)
Sub-carrier Power =
= -3.65dBm (432µW) downstream and -1.65dBm (544µW) upstream.
- 40000 - 20000 20000 40000
- 100
- 80
- 70
- 60
- 50
- 40
�∞∞- df sdp
�∞
∞−df psd
ITU/ANSI PSD Masks
Downstream PSD With Continuous Time Sub-Carrier GenerationPS
D (d
Bm
/Hz)
1 ×106 2×106 3×106 4 ×106 5×106
- 110
- 100
- 90
- 80
- 70
- 60
- 50
- 40
Frequency
20000 40000 60000 80000 100000 120000 140000
- 70
- 60
- 50
- 40
- 30
DSP Transmitter Signal (Minimum Sample Rate )
PSD
(dB
m/H
z)Frequency
• Minimum Annex A downstream 256-point IFFT with 4312.5Hz resolution (one bin per subcarrier) produces 2208kS/sec sample rate.
• Minimum Annex A upstream 32-point IFFT with 4312.5Hz resolution (one bin per subcarrier) produces 276kS/sec sample rate.
200000 400000 600000 800000 1´ 10 6
- 70
- 60
- 50
- 40
- 30
Downstream PSD With Minimum Sample Rate
PSD
dB
m/H
z
Frequency2×106 4 ×106 6×106 8×106
- 100
- 80
- 60
- 40
• Conversion of downstream signal to continuous-time at 2208kS/sec produces in-band droop and significant standard’s non-compliance.
Upstream PSD With Minimum Sample Rate (276kS/sec)
PSD
dB
m/H
z
Frequency1×106 2×106 3×106 4×106
- 100
- 80
- 60
- 40
• Conversion of upstream signal to continuous-time at 276kS/sec produces in-band droop and significant standard’s non-compliance.
Impairments
• Loop attenuation
• Loop variability
• Crosstalk
26awg Loop Attenuation
200000 400000 600000 800000 1×106
- 100
- 80
- 60
- 40
- 201kft
5kft
10kft
15kft
Frequency
Att
enua
tion
(dB
)
Downstream PSD on 26awg Loops
200000 400000 600000 800000 1×106 1.2×106
- 160
- 140
- 120
- 100
- 80
- 60
- 40
Frequency
PSD
(dB
m/H
z)
1kft
5kft
10kft
15kft
Loop Variability
500000 1 ´ 10 6 1.5 ´ 10 6 2´ 10 6
10
20
30
40
ETSI Loop Loss at 1 kilometer Versus Frequency
0.32mm
0.40mm
0.50mm
0.63mm
0.90mm
Inse
rtio
n L
oss
(dB
)
Frequency
Loop Impedance
500000 1´ 10 6 1.5 ´ 10 6 2´ 10 6
100
120
140
160
180
ETSI Loop Impedance Versus Frequency
Impe
danc
e M
agni
tude
(ohm
)
Frequency
0.32mm
0.40mm
0.50mm0.63mm
0.90mm
4.5 5 5.5 6 6.5 7
100
150
200
250
Bridged Taps
ClientC OA
bs[Z
in]
Log10[frequency]
CSA4 Zin from client26awg characteristic impedance
CSA4 Zin from central office
Far End Cross Talk (FEXT)
ClientC O
ClientC O
Near End Cross Talk (NEXT)
ClientC O
ClientC O
Far End Crosstalk Transfer Function (FEXT)
250000 500000 750000 1´ 10 6 1.25 ´ 10 6 1.5 ´ 10 6
- 120
- 110
- 100
- 80
- 70
- 60
1 km
2 km
3 km
24
10
24
10
24
10
( )( )2
26.049n20
f,dossinsertionL
fd10*8 ∗−
H (d
B)
Frequency (Hz)
250000 500000 750000 1´ 10 6 1.25 ´ 10 6 1.5 ´ 10 6
- 160
- 140
- 120
- 100
- 80
- 60
- 40
24-Self FEXT Example
1 km2 km3 km
1 km
2 km
3 km
~50dB
Downstream Signal
24 self-FEXT
~60dB
~70dB
H (d
Bm
/Hz)
Frequency (Hz)
250000 500000 750000 1´ 10 6 1.25 ´ 10 6 1.5 ´ 10 6
- 70
- 60
- 50
- 40
Near End Crosstalk Transfer Function (NEXT)
24
10
1
( ) 5.1fnX ∗
H (d
B)
Frequency (Hz)
5.963*10-14243.510*10-14102.020*10-1448.818*10-151
X(n)n
20000 40000 60000 80000 100000 120000 140000
- 180
- 160
- 140
- 120
- 100
1-, 10- and 24-Self NEXT ExampleH
(dB
m/H
z)
Frequency (Hz)
Line Coupling Circuit
• Splitter
• Active termination
• Hybrid
ADSLModem
POTS/ISDN
Splitter
Bi-directionalLow-Pass
Filter
Bi-directionalHigh-Pass
Filter
• In the POT/ISDN frequency band, the high-pass, generally built into modem transformer circuit, presents high impedance to loop and reduces ADSL modem-generated noise.
• In the ADSL frequency band, the low-pass presents high impedance to loop and reduces POTS/ISDN-generated noise.
Annex B Splitter Low-Pass Example
ToLoop
88.7u94.1u86.1u64.7u
2.7p4.7p8.2n
8.2n 18n 22n 22n
• Very high quality (low distortion) components are required
• DC current feed (POTS) may not saturate transformers
• Cost is high, both the bill of materials and the PCB area
Active Termination
ZL
VTX
VRX
R
I
gg*I*R
(((( ))))Rg1ZVZI
VZ LgRZR
VRX
L0V
RXin
L
RXTX
++++====−−−−−−−−
−−−−====−−−−−−−−====++++++++====
• Reduces driver power supply voltage requirement• Downside: Reduces receiver gain by 20Log10[(1+g)]
Transmitter Echo
TX
RX
2-wireto
4-wire(hybrid)
transmit
receive
echoOn long loops, the echo power is greater than the receive power
A B
C D
Z4
Z2
Vrxm
Vtxp
Vtxm
RT
RT
Z1
Z3
Z1
Z2
Z3
Vrxp
1 : N
Z4
Eg
ZL
2nd Order High-Pass with Passive Termination and Hybrid
( )
���
����
����
����
�−=−
++=
1
3grxmrxp
L31T2
L324
ZZ
N4
EVV yielding
ZZZRN2
ZZZ Z:Balance Hybrid
IL
( )( )txmtxpgL
LT
VVNEZI
N
ZR
−−��
���
�=
=
21
yielding
2 :Match Impedance
L
2
3rd Order High-Pass With Passive Termination and Hybrid
Vtxp
Vtxm
ZL
1 : N
Za
Za Zb
Zb
hybrid
RXamp
RT
RT
LT2
L
a
b
ZRNZ
ZZ
++++====
Hybrid Balance Condition
LT2 ZRN ====
Impedance Match Condition
3rd Order High-Pass With Active Termination and Hybrid
Subscriber line
Vtxp
Vtxm
Eg
ZL
1 : N
H(f)
Active Hybrid Transfer Function
200000 400000 600000 800000 1 ´ 10 6
- 24
- 22
- 20
Magnitude
Ideal transfer function from driver output to termination resistor voltage when received signal is not present
Best fit to a 3-zero, 4-pole active hybrid
Hybrid Rejection Definitions
(((( ))))
(((( )))) ����
��������
====
����������������
����������������
����
====
present is hybrid when tip/ringat signalr transmitteremoved is hybrid when tip/ringat signalr transmitteLog20thR
outputamplifier receiver to tip/ring from gain signal receivedoutputamplifer receiver at signalr transmitte
tip/ringat signalr transmitteLog20expR
10H
10H
Echo Response
����
��������
====amplifierreceiver at signalr transmitteoutputdriver at signalr transmitteLog20H 10Echo
Hybrid Rejection Example
4.5 5.5 6
30
40
50
60
Hybrid Rejection , Theorist and Experimentalist Definitions
Log10[frequency]
Hecho
RH(exp)
RH(th)
Transmitter Signal Path
• Over-sampling
• Managing PAR
• MTPR requirement
• Line Drivers
Downstream PSD With Minimum Sample Rate
PSD
dB
m/H
z
Frequency2×106 4 ×106 6×106 8×106
- 100
- 80
- 60
- 40
• Conversion of downstream signal to continuous-time at 2208kS/sec produces in-band droop and significant standard’s non-compliance.
Upstream PSD With Minimum Sample Rate (276kS/sec)
PSD
dB
m/H
z
Frequency1×106 2×106 3×106 4×106
- 100
- 80
- 60
- 40
• Conversion of upstream signal to continuous-time at 276kS/sec produces in-band droop and significant standard’s non-compliance.
Low-Pass Reconstruction Filter Architectures
IFFT 2.2MSpsDAC
ALPF
IFFT N*2.2MSpsDAC ALPFDLPF
• High order
• Trimmed to provide precision roll-off at 1.1MHz as well as image rejection at 2.2MHz.
•Low order
•Trimming not required for image rejection at N*2.2MHz.
Provides precision roll-off at 1.1MHz.
(A) Nyquist rate DAC (Analog filter)
(B) Over-sampled DAC (Digital/Analog filter combination)
Helping Out the Central Office Hybrid
IFFT N*2.2MSpsDAC AHPFDLPF
Provides precision rejection of downstream subcarrier side lobes in upstream band
DHPF ALPF
Provides rejection of DAC distortion components in upstream band
Digital High-Pass Output Spectrum(downstream bin 32-255 employed)
50000 100000 150000 200000 250000
- 180
- 160
- 140
- 120
- 100
- 80
- 60
- 40
Frequency
PS
D (d
Bm
/Hz)
Digital Low-Pass Filter Output Spectrum
2 ×106 4 ×106 6 ×106 8×106
- 100
- 80
- 60
- 40
Frequency
PS
D (d
Bm
/Hz)
Analog Low-Pass Filter Output SpectrumPS
D (d
Bm
/Hz)
2×106 4×106 6×106 8×106
- 100
- 80
- 60
- 40
Frequency
Transmitter Template
• DHPF • High order DHPF is critical to downstream transmitter signal suppression in FDD modems• Reduced-NEXT compliance in echo-canceling modems employing non-overlapping spectra
• DLPF• High order is critical to upstream transmitter signal suppression in FDD modems• Interpolating stages increase sample rate, reducing signal droop and relaxing analog low-pass
• DAC• Precision (<12 effective bits) depends upon analog filtering, hybrid rejection and receiver sensitivity
•AHPF• Suppresses DAC noise/distortion in upstream band in FDD central office modems.
•ALPF• Eliminates signal images centered at multiples of DAC sample rate.• Suppresses DAC noise/distortion in downstream band in FDD client modems.
• Line Driver• High voltage technology. • Very low distortion required.
Corner frequency precisionZero additive noise
Suppresses DAC noise/distortion and images
LineDriver
DACFromDSP
DigitalHigh-Pass
Filter
DigitalLow-Pass
Filter
AnalogLow-Pass
Filter
AnalogHigh-Pass
Filter
ToHybrid
Managing Peak Signal to Average Signal Ratio
• Bit Error Rate is proportional to the frequency of signal clipping
• Clipping rate of gaussian signal is determined by PAR
• PAR is determined by circuit NOT a property of signal
• Managing PAR in digital or analog domains – node by node• Usually desirable to limit the number of clipping nodes to one (driver)• For given signal swing (volts or bits), PAR is increased (decreased) by decreasing (increasing) signal power• For given signal power, PAR is increased (decreased) by increasing (decreasing) swing (volts or bits)
LineDriver
DACFromDSP
DigitalHigh-Pass
Filter
DigitalLow-Pass
Filter
AnalogLow-Pass
Filter
AnalogHigh-Pass
Filter
ToHybrid
PARDHPF PARDAC PARAHPF PARAHPFPARDLPF PARLD
Missing Tone Power Ratio Test
• Equivalent of spurious free dynamic range in narrow band systems
DownstreamUpstream
MTPRrx MTPRtx
FDD (EC) Central Office Example (frequency-independent echo response)MTPRtx=65 (75)MTPRrx=75 (75)
Virtue of High Voltage IC ProcessesClient Reconstruction Low-Pass Filter Example(138kHz corner frequency, -140dBm/Hz noise in downstream band)
g LPF 1
15V
LPF g
15V3.3V
020406080
100
0 0.2 0.4 0.6 0.8 1
Vo pk-pk / (Vcc-Vee)
Effi
cien
cy (%
)
Class AB Output Stage
VCC
VEE
-1
-0.5
0
0.5
1
0 10 20 30 40 50
Vol
tage
(V
)
<20% efficiency
Time
Class G Topology
• Multiple supply output stage
• For VOUT low, current comes from +5 V
• For VOUT high, current comes from +15 V
• Current is mostly drawn from low supply
• Challenge in DSL is low distortion at 1 MHz
Ip15
Vout
Iout
Ip5
Rs
Rl
Vout
+5V+15V
Ip5Ip15
Iout
-5V-15V
Continuous Class G Circuit
Vout
Q1
I2
D4
D3
Q5D6
Q3
Q2
Q0
D2
D1
Q4D5
I1
+15V
-15V
Vin
+15V
-5V
-15V
+5V
“Zero Overhead” Class G Concept –Block Diagram
VPSWLogic
Interface
LineDriver
TXsignal
+8V
+4V
-4V
-8V
TVEE
TVCC
“Zero Overhead” Class G Concept -Waveforms
TVCC
DMT
TVEE
VPSW
+4V
-4V
+8V
-8V
ZOCG Design Challenges
• Peak prediction– Analog signal blocks change peak positions and magnitudes
• Low voltage drop switches– Mus swing as close as possible to the rail
• Shoot through currents– Must minimize the time that switches to both supplies are on
• Switching edge coupling– Changing supply AC can couple into signal output
Receiver Signal Path
• Signals
• Minimizing ADC requirements
• ADC precision
• PAR management
Anti-Alias
FilterADCGain
Receiver Noise/Distortion Floor
• Inherent loop noise
•Loop impedance is ~100 ohms, corresponding to <-173 dBm/Hz
• Measured loop noise is usually in the neighborhood of –140 dBm/Hz
• Central off ice noise is much higher• ISDN, T1, SHDSL, ….
• Typical noise floor targets for ADSL front ends
• Central office: <-120 dBm/Hz at loop
• Client: <-145 dBm/Hz at loop
ADC Resolution Versus Conversion Rate
2.5 ´ 10 6 5´ 10 6 7.5 ´ 10 6 1´ 10 7 1.25 ´ 10 7 1.5 ´ 10 7 1.75 ´ 10 7
0.0005
0.001
0.0015
0.002
0.0025
0.003
LSB
effre
ferr
ed to
mod
em in
put (
volt)
ADC Conversion Rate (Sps)
-120 dBm/Hz (CO)
-140 dBm/Hz (Client)
1000 2000 3000 4000 5000 6000
- 80
- 60
- 40
- 20
Signal Power Reaching ReceiverPo
wer
(dB
m)
Loop Length (meters)
Upstream
Downstream
A
BA
B
Largest voltage on null loop: 11.2Vpp at client and 19.95Vpp at central office
Minimum ADC Neff versus Conversion Rate
2.5 ´ 10 6 5´ 10 6 7.5 ´ 10 6 1´ 10 7 1.25 ´ 10 7 1.5 ´ 10 7 1.75 ´ 10 7
13
14
15
16
17
Eff
ectiv
e nu
mbe
r of
bits
, Nef
f
ADC Conversion Rate (Sps)
Client Vin=11.22V
Central office Vin=19.95V
• Input voltages are too high to integrate – must reduce gain from loop to ADC• Reducing channel gain drives ADC lsb too small
Received Input Peak-to-Peak Voltage versus Loop Length
Loop Length (ft)
5000 10000 15000 20000
5
10
15
20
25
30
35
Peak
-to-
Peak
Vol
tage
(vol
ts)
Need signal attenuation on short loops and gain on long ones
downstream
upstream
Programmable Gain to the Rescue
Anti-Alias
FilterADCGain
• The noise floor targets were set using loop noise measurements,and represent reasonable goals for long loops where signals are small
• On short loops we only need noise floors sufficient to handle 15 bits per subcarrier, 12.5Neff
• Use programmable gain to increase the channel gain for small signals from long loop
• gives low channel gain when signals are large• gives high channel gain when signals are small
• Remaining problem: the echo
1000 2000 3000 4000 5000 6000
- 125
- 100
- 75
- 50
- 25
Total Signal Power at ReceiverPo
wer
(dB
m)
Loop Length (meters)
Upstream
Downstream
A
BA
B
Largest voltage on null loop: 11.2Vpp at client and 19.95Vpp at central office
Killing the Echo
• Better
• Best
Anti-Alias
FilterADCGain
Anti-Echo
Filter
Anti-Alias
FilterADCGain
Anti-Echo
FilterGain
Receiver Architectures
+ PreampAnalog
High-PassFilter
Analog Low-Pass
FilterFrom
Hybrid
FromSense
Resistor
Client
PGA ADC
+ PreampAnalog
Low-PassFilter
FromHybrid
FromSense
Resistor
Central Office
PGA ADC
Receiver Programmable Gain
• Receiver gain improves SNR only when the ADC quantization noise is dominant. • Typically 20 dB gain will amplify the receiver analog noise above the ADC noise.
Additional gain will not improve the SNR, so the typical programmable gain range is ~ -20dB to +20dB at the remote terminal and ~ -12dB to +20dB at the central office.
5000 10000 15000 20000
20
40
60
Gai
n (d
B)
Loop Length (ft)
downstream
upstream
Equalization
200000 400000 600000 800000 1×106
- 120
- 110
- 100
- 90
- 80
- 70
- 60
- 50
Frequency
PSD
(dB
m/H
z)
AmplifiedDownstream signal at 9kft
ADC noise floor
SNR sufficient for 15b/bit/frame
• Equalization is frequency dependent gain• Improves data rate when
• SNR is limited by noise source subsequent to equalizer• Low-Frequency subcarrier SNR exceeds that required to support 15b/frame
• Programmable gain should follow equalizer
Generic Template
• AHPF • In client FDD modems, it may reject virtually all of the upstream echo power and enable higher PGA gain.
• Frequently employed in Annex B central office modem to reject ISDN power in front of PGA.• Equalizer
• Provides frequency dependent gain to amplify weak high-frequency subcarriers above ADC noise.• Not useful in central office receiver
•ALPF• Provides anti-alias filtering for ADC.• In central office FDD modems, it also rejects upstream echo and enables higher PGA gain.
•PGAs• Placed both before and after filters when echo is rejected by integrated filter.
• ADC• Resolution depends upon duplexing, EC being more demanding than FDD because echo power
is dominant on long loops. •DLPF
• Decimates sample rate to that of FFT.
PGAADCDigital
Low-PassFilter
AnalogLow-Pass
Filter
AnalogHigh-Pass
FilterEqualizer PGA
Design Example:Echo-Canceling Central Office Modem
• Deviations from generic topologies• Analog transmitter high-pass deleted to support echo cancellation• Programmable attenuator added to support power cutback on short loops• Programmable gain amplifiers precede and follow receiver analog low-pass
to adjust signal to 3Vpp before non-overlapping portion of echo is removed• Equalizer is not necessary on central office receivers
• DSP sample rates• 2208kS/sec transmitter in full rate mode and 1104kS/sec in “lite” mode• 276kS/sec receiver
LineDriver
DAC
DSP
DigitalHigh-Pass
Filter
DigitalLow-Pass
Filter
ProgrammableAttenuator
AnalogLow-Pass
Filter
Hybrid
ADCDigital
Low-PassFilter
AnalogLow-Pass
Filter
AnalogHigh-Pass
Filter
PGA(coarse)
PGA(fine)
AD16
AD16
PGA0-6dBgain
in 1dBsteps
ALPF
138kH zto
172kH zcorner,
(tr imm ed)
PGA2.5-11.5dB
gainin 1dBsteps
PGA-20dB
to +16dBgain
in 6dBsteps
DAC
14 bits
8832kS/secsample rate
DLPF1(f33)
1104kHzor
552kHzcorner
DLPF2(f9)
1104kHzor
552kHzcorner
INTRP22X
zeroinsertion
ALPF
1104 -1725kHz
corner
(1380kHz+/-25%)
PA
0dB to-24dB in
1dB steps
TRANSMITTER PATH
RECEIVER PATH
E-C
lite
dmt
INTRP32X
zeroinsertion
8832kS/secsample rate
lite
INTRP12X
zeroinsertion
4416kS/ssample rate
lite
dmtdmt
4416kS/sec2208kS/sec
lite dmt
8832kS/sec
lite dmt
INTRP02X
zeroinsertion
2208kS/secsample rate
dmt
litelite
DHPF
138kHzcorner
2208kS/secsample rate
P-FDM P-FDM
DLPF3(f7)
552kHzcorner
8832kS/secsample rate
ADC
2208kH z14 bits
DLPF
138kHzcorner,
0dB gain
SampleRate
Select
D/A Converter
VDD
CALx
VA
VB
VMVDD
CALx
VA
VB VBM31
VM
1 1VB
1/2VB
1/4VB
1/4
DD C/HDD C/H
M1
DD
M32
DD C/H
M33 M34
HOLD CAL
CASCODE ARRAYS(5+2)b LSBs
CURRENT SOURCEARRAY 7b MSBs
C/H
Iout-Iout Ihold Ical
DD CAL
Signal DAC Current Source Bias and Calibration
Calibrating CellReplicaCell (4X)
ReplicaCell (8X)
ReplicaCell (4X)
Vcm 16-1MUX
4b
1.5Vbandgapderived
VA
VB
ReferenceCurrent
Transmitter Static Integral Non-Linearity(digital high-pass bypassed)
-0.6
-0.4
-0.2
0
0.2
0.4
0.6
-8191 0 8191
Code
INL
(14b
LS
Bs)
Receiver PGA/Low-Pass Combination
t0
t1
t2
t0
t1
t2
t0
t1
t2
f6
f5
f4
x CMOS transmission gate conducting when logic bit x = 1
cx bits set gain -20dB to +16dB in 6dB steps,fx bits set gain 0dB to +6dB in 1dB steps, andtx bits trim corner frequency.
f3f1
f2
c1
c7
Ext.
PSD at Input to Analog Low-Pass Filter(15kft, 26awg, RH=18dB)
500000 1 ×106 1.5×106 2×106
- 140
- 120
- 100
- 80
- 60
- 40
PSD
(dB
m/H
z)
Frequency
Signal
Echo
PSD at Input to ADC(15kft, 26awg, RH=18dB)
200000 400000 600000 800000 1×106 1.2×106
- 140
- 120
- 100
- 80
- 60
- 40
PSD
(dB
m/H
z)
Frequency
Signal
Echo
Pipelined Analog-to-Digital Converter
Stage 11.5b
8b trim
TrimSignMUX
d2
1
1
1
1
2
2Vin+
Vref8
Vref8
Vin-
d0
d1
d0
d2
Vref+
Vref-
2pF
2pF
2pF
2pF
Stage 21.5b
8b trim
Stage 31.5b
7b trim
Stage 41.5b
7b trim
Stage 51.5b
7b trim
Stage 61.5b
no trim2b
flash
Stage 121.5b
no trim
X represents CMOS switchconducting during clock phase X
d0
d1
d2Vcm +
Vcm -
7-Bit Capacitor Trimming Array
t4t4 t6t6t5t5t1t1 t2t2t0t0 t3t3
AVSS
ts ts
Trim sign MUX
Cu
Cu
2Cu 4Cu Cu 2Cu 4Cu 8Cu
Cu
Cu = 104fF
Ceff = 98fF in 0.77fF steps +-
Receiver INL
-2
-1.5
-1
-0.5
0
0.5
1
1.5
2
0 4096 8192 12288 16384
CODE
INL
PSD at Output of Digital Low-Pass Filter(15kft, 26awg, RH=18dB)
50000 100000 150000 200000 250000 300000
- 140
- 120
- 100
- 80
- 60
- 40
PSD
(dB
m/H
z)
Frequency
Signal
Echo
Summary
• The ADSL AFE requirements depend upon multiple system parameters such as • Duplexing method• Upstream and downstream frequency allocations• DSP transmit and receive sample rates• Receiver sensitivity • Hybrid rejection
• An AFE design example provides the optimum integrated solution for modems employing echo cancellation.