doctoral thesis - tesis doctoral contribution to...

UNIVERSIDAD POLITÉCNICA DE MADRID

ESCUELA TÉCNICA SUPERIORDE INGENIEROS DE TELECOMUNICACIÓN

DOCTORAL THESIS - TESIS DOCTORAL

CONTRIBUTION TO ACTIVE ARRAY

ANTENNAS AT MICROWAVE BANDS

CONTRIBUCIÓN A LOS ARRAYS DEANTENAS ACTIVOS EN MICROONDAS

Gonzalo Expósito DomínguezIngeniero de Telecomunicación

2013

UNIVERSIDAD POLITÉCNICA DE MADRID

ESCUELA TÉCNICA SUPERIOR DE INGENIEROS DE TELECOMUNICACIÓN

DEPARTAMENTO DE SEÑALES, SISTEMAS Y RADIOCOMUNICACIONES

GRUPO DE RADIACIÓN

DOCTORAL THESIS - TESIS DOCTORAL

CONTRIBUTION TO ACTIVE ARRAY

ANTENNAS AT MICROWAVE BANDS

CONTRIBUCIÓN A LOS ARRAYS DEANTENAS ACTIVOS EN MICROONDAS

Autor :

Gonzalo Expósito Domínguez

Ingeniero de Telecomunicación

Directores:

Manuel Sierra-Castañer

Doctor Ingeniero de Telecomunicación

Profesor Titular de Universidad

José-Manuel Fernández-González


Contratado doctor

Madrid, 2013

TESIS DOCTORAL: Contribution to active array antennas at microwave bands.

Contribución a los arrays de antenas activos en microondas.

AUTOR: Gonzalo Expósito Domínguez

Ingeniero de Telecomunicación

DIRECTORES: Manuel Sierra-Castañer


Profesor Titular de Universidad

José-Manuel Fernández-González


Contratado doctor

DEPARTAMENTO: Señales, Sistemas y Radiocomunicaciones

Univerisdad Politécnica de Madrid

El Tribunal de Calicación, compuesto por:

PRESIDENTE:

Prof. Dr.

VOCALES:

Prof. Dr.

Prof. Dr.

Prof. Dr.

VOCAL SECRETARIO:

Prof. Dr.

VOCALES SUPLENTES:

Prof. Dr.

Prof. Dr.

Realizado el acto de defensa y lectura de la Tesis, acuerda otorgarle la CALIFI-

CACIÓN de:

Madrid, a de de 2013.

A mi sobrino Martín

"Serenidad para aceptar las cosas que no puedo cambiar,

valor para cambiar las cosas que puedo

y sabiduría para poder diferenciarlas"

REINHOLD NIEBUHR

Resumen

Esta tesis que tiene por título Contribución a los arrays de antenas activos en

banda X, ha sido desarrollada por el estudiante de doctorado Gonzalo Expósito

Domínguez, ingeniero de telecomunicación en el Grupo de Radiación del Departa-

mento de Señales, Sistemas y Radiocomunicaciones de la ETSI de Telecomunicación

de la Universidad Politécnica de Madrid bajo la dirección de los doctores Manuel

Sierra Castañer y José Manuel Fernández González.

Esta tesis contiene un profundo estudio del arte en materia de antenas activas en

el campo de apuntamiento electrónico. Este estudio comprende desde los fundamen-

tos de este tipo de antenas, problemas de operación y limitaciones hasta los sistemas

actuales más avanzados. En ella se identican las partes críticas en el diseño y pos-

teriormente se llevan a la práctica con el diseño, simulación y construcción de un

subarray de una antena integrada en el fuselaje de un avión para comunicaciones

multimedia por satélite que funciona en banda X. El prototipo consta de una red

de distribución multihaz de banda ancha y una antena planar.

El objetivo de esta tesis es el de aplicar nuevas técnicas al diseño de antenas

de apuntamiento electrónico. Es por eso que las contribuciones originales son la

aplicación de barreras electromagnéticas entre elementos radiantes para reducir los

acoplamientos mutuos en arrays de exploración electrónica y el diseño de redes des-

fasadoras sencillas en las que no son necesarios complejos desfasadores para antenas

multihaz.

Hasta la fecha, las barreras electromagnéticas, Electronic Band Gap (EBG),

se construyen en sustratos de permitividad alta con el n de aumentar el espacio

disponible entre elementos radiantes y reducir el tamaño de estas estructuras. Sin

embargo, la utilización de sustratos de alta permitividad aumenta la propagación

por ondas de supercie y con ellas el acoplo mutuo. Utilizando sustratos multicapa

y colocando la via de las estructuras en su borde, en vez de en su centro, se consigue

reducir el tamaño sin necesidad de usar sustratos de alta permitividad, reducir

i

ii

la eciencia de radiación de la antena o aumentar la propagación por ondas de

supercie.

La última parte de la tesis se dedica a las redes conmutadoras y desfasadoras

para antenas multihaz. El diseño de las redes de distribución para antenas son una

parte crítica ya que se comportan como un atenuador a la entrada de la cadena

receptora, modicando en gran medida la gura de ruido del sistema. Las pérdidas

de un desfasador digital varían con el desfase introducido, por ese motivo es necesario

caracterizar y calibrar los dispositivos correctamente. Los trabajos presentados en

este manuscrito constan de un desfasador reectivo con un conmutador doble serie

paralelo para igualar las pérdidas de inserción en los dos estados y también un

conmutador de una entrada y dos salidas cuyos puertos están adaptados en todo

momento independientemente del camino del conmutador para evitar las reexiones

y fugas entre redes o elementos radiantes.

El tomo naliza con un resumen de las publicaciones en revistas cientícas y

ponencias en congresos, nacionales e internacionales, el marco de trabajo en el que se

ha desarrollado, las colaboraciones que se han realizado y las líneas de investigación

futuras.

Abstract

This thesis was carried out in the Radiation Group of the Signals, Systems and

Radiocomunications department of ETSI de Telecomunicación from Technical Uni-

versity of Madrid. Its title is Contribution to active array antennas at X band and

it is developed by Gonzalo Expósito Domínguez, Electrical Engineer MsC. under the

supervision of Prof. Dr. Manuel Sierra Castañer and Dr. José Manuel Fernández

González.

This thesis is focused on active antennas, specically multibeam and electronic

steering antenas. In the rst part of the thesis a thorough description of the state

of the art is presented. This study compiles the fundamentals of this antennas,

operation problems and limits, up to the breakthrough applications. The critical

design problems are described to use them eventually in the design, simulation and

prototyping of an airborne steering array antenna for satellite communication at X

band.

The main objective of this thesis is to apply new techniques to the design of

electronically steering antennas. Therefore the new original contributions are the

application of Electromagnetic Band Gap materials (EBG) between radiating ele-

ments to reduce the mutual coupling when phase shift between elements exist and

phase shifting networks where special characteristics are required.

So far, the EBG structures have been constructed with high permitivity substra-

tes in order to increase the available space between radiating elements and reduce

the size of the structures. However, the surface wave propagation modes are enhan-

ced and therefore the mutual coupling increases when high permitivity substrates

are used. By using multilayered substrates and edge location via, the size is redu-

ced meanwhile low permitivity substrates are used without reducing the radiation

eciency or enhancing the surface propagation modes.

The last part of the thesis is focused on the phase shifting distribution networks

for multibeam antennas. This is a critical part in the antenna design because the

iii

iv

insertion loss in the distribution network behaves as an attenuator located in the rst

place in a receiver chain. The insertion loss will aect directly to the receiver noise

gure and the insertion loss in a phase shifter vary with the phase shift. Therefore

the devices must be well characterized and calibrated in order to obtain a properly

operation. The work developed in this thesis are a reective phase shifter with a

series-shunt switch in order to make symmetrical the insertion loss for the two states

and a complex Single Pole Double Through (SPDT) with matched ports in order to

reduce the reections and leakage between feeding networks and radiating elements.

The end of this Ph D. dissertation concludes with a summary of the publications

in national and international conferences and scientic journals, the collaborations

carried out along the thesis and the future research lines.

Contents

Resumen i

Abstract iii

List of Figures x

List of Tables xi

1. Introduction 1

1.1. Motivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

1.2. Objectives . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

1.3. Outline of the thesis . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

2. State of the art on multibeam and recongurable steering array

antennas 23

2.1. Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

2.2. Multibeam passive antennas . . . . . . . . . . . . . . . . . . . . . . . 33

2.2.1. Wide band Butler matrix network at X band . . . . . . . . . . 35

2.3. Scanning active antennas . . . . . . . . . . . . . . . . . . . . . . . . . 38

2.3.1. Airborne steering antenna . . . . . . . . . . . . . . . . . . . . 39

2.4. Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48

References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55

3. Mutual coupling reduction using EBG in steering antennas 57

3.1. Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57

3.2. EBG theory fundamentals . . . . . . . . . . . . . . . . . . . . . . . . 60

3.3. Surface wave supression . . . . . . . . . . . . . . . . . . . . . . . . . 63

3.4. Mutual coupling reduction . . . . . . . . . . . . . . . . . . . . . . . . 66

3.5. Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75

v

vi CONTENTS

3.6. Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80

References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84

4. Phase shifting distribution networks for switchable beam antennas 85

4.1. Fundamentals of phase shifters . . . . . . . . . . . . . . . . . . . . . . 85

4.2. Topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88

4.3. Switchable beam antenna . . . . . . . . . . . . . . . . . . . . . . . . 92

4.3.1. Symmetric reective phase shifter . . . . . . . . . . . . . . . . 95

4.3.2. Matched SPDT switch . . . . . . . . . . . . . . . . . . . . . . 98

4.4. Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106

References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111

5. Conclusions 113

5.1. Original contributions . . . . . . . . . . . . . . . . . . . . . . . . . . 114

5.2. Future research lines . . . . . . . . . . . . . . . . . . . . . . . . . . . 115

5.3. Publications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115

5.3.1. Journal publications . . . . . . . . . . . . . . . . . . . . . . . 115

5.3.2. Conference contributions . . . . . . . . . . . . . . . . . . . . . 116

List of Figures

1.1. Astronomic interferometry clusters [1.6]. . . . . . . . . . . . . . . . . 3

1.2. Classication of metamaterials [1.17]. . . . . . . . . . . . . . . . . . . 6

1.3. Schematic view of a Field Eect Transtistor. . . . . . . . . . . . . . . 9

1.4. Micro ElectroMechanical Systems schematic view. . . . . . . . . . . . 10

1.5. PIN diode and equivalent circuit. . . . . . . . . . . . . . . . . . . . . 11

1.6. Ferrite core phase shifter. . . . . . . . . . . . . . . . . . . . . . . . . . 13

1.7. Varactor diode outline. . . . . . . . . . . . . . . . . . . . . . . . . . . 13

2.1. Array splitting problem. . . . . . . . . . . . . . . . . . . . . . . . . . 24

2.2. Wavefront combination to get electronic steering. . . . . . . . . . . . 25

2.3. Visible margin of an array in ψ and θ. . . . . . . . . . . . . . . . . . 26

2.4. HPBW vs scanning angle. . . . . . . . . . . . . . . . . . . . . . . . . 30

2.5. Directivity reduction vs scanning angle. . . . . . . . . . . . . . . . . . 31

2.6. Crossing point losses vs number of beams. . . . . . . . . . . . . . . . 31

2.7. Quantization noise oor vs number of bits. . . . . . . . . . . . . . . . 32

2.8. Appearance of grating lobes due to failures in the phase feeding law. . 33

2.9. Wide band Butler Matrix network . . . . . . . . . . . . . . . . . . . . 37

2.10. Phase of the scattering parameters of a wideband Butler matrix net-

work. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

2.11. Airborne steering antenna for satellite communications [2.40]. . . . . 39

2.12. Double stacked patch with 3dB/90o hybrid coupler. . . . . . . . . . . 41

2.13. Block diagram system. . . . . . . . . . . . . . . . . . . . . . . . . . . 41

2.14. Radiation pattern of 16x1 array. . . . . . . . . . . . . . . . . . . . . . 42

2.15. Amplitude and phase of the active network elements. . . . . . . . . . 43

2.16. 1 to 4 unbalanced network divider. . . . . . . . . . . . . . . . . . . . 43

2.17. S parameters of 1 to 4 unbalance network divider. . . . . . . . . . . . 44

2.18. 3 dB/90o Hybrid coupler miniaturization. . . . . . . . . . . . . . . . . 45

2.19. Hybrid coupler S parameters. . . . . . . . . . . . . . . . . . . . . . . 46

vii

viii LIST OF FIGURES

2.20. Radiation pattern of azimuth plane. . . . . . . . . . . . . . . . . . . . 46

2.21. 4x4 array array prototype [2.40]. . . . . . . . . . . . . . . . . . . . . . 47

2.22. S parameters measurements of 4x4 array prototype (8 ports). . . . . . 48

2.23. Steering radiation pattern of 4x4 subarray. . . . . . . . . . . . . . . . 48

3.1. High impedance surface and its model with parallel resonant LC cir-

cuit. The substrate is transparent in order to get better visualization

of metallic vias. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61

3.2. LC model for the mushroom like EBG structure. . . . . . . . . . . . . 61

3.3. Layer and top views of traditional EBG structures (left hand side)

and multilayered F structure (right hand side). . . . . . . . . . . . . . 63

3.4. Unit cell scheme for eigenmode solutions. Dimensions in mm. . . . 64

3.5. Brillouin diagrams of the EBG unit cell. . . . . . . . . . . . . . . . . 65

3.6. Simulation scheme for transmission parameters S21 analysis. . . . . . 66

3.7. Parametric study of isolation characteristics (S21 for original shape

mushrooms) when size and number of elements are swept. . . . . . . 67

3.8. Parametric study of isolation characteristics (S21 for original shape

mushrooms) when substrate thickness, via diameter and gap size are

sweept. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68

3.9. Parametric study of width W and length L for F shape mushrooms.

Transmission S parameters (S21). . . . . . . . . . . . . . . . . . . . . 69

3.10. Samples of single and multilayered EBG mushrooms with dierent

shapes and number of elements. . . . . . . . . . . . . . . . . . . . . . 69

3.11. Comparison between measurements and simulations of transmission

parameters S21 for dierent types of EBG mushrooms. . . . . . . . . 70

3.12. Multilayered mushroom with rectangular shape, 4 elements and edge-

located via (F-shape). . . . . . . . . . . . . . . . . . . . . . . . . . . 71

3.13. |E| eld simulation of two round patches with dual circular polarization. 72

3.14. 2x1 test array of circular patch antennas with and without EBG F

shape mushrooms. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72

3.15. Specic dimensions of 2x1 test array. . . . . . . . . . . . . . . . . . . 73

LIST OF FIGURES ix

3.16. Comparison of measurements and simulated S parameters for 2x1 array. 73

3.17. Radiation patterns of 2x1 test patch antenna array with and without

EBG structures. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74

3.18. Layer view of the 4x4 array with EBG structures construction. . . . . 76

3.19. S parameters measurements of 4x4 array with EBG structures. . . . . 77

3.20. 4x4 array and Butler matrix network connection for LHCP congu-

ration. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77

3.21. 4x4 steering array radiation pattern for RHCP, at the center frequency

(7.825GHz). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78

3.22. 4x4 steering array axial ratio for frequencies 7.25 GHz and 8.4 GHz,

RH and LH circular polarizations over the scanning angles. . . . . . . 79

3.23. 4x4 steering array axial ratio for RH and LH circular polarizations

and dierent pointing directions over the working frequency. . . . . . 79

4.1. Single switches. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89

4.2. Compound switches. . . . . . . . . . . . . . . . . . . . . . . . . . . . 89

4.3. Tuned λ/4 switches. . . . . . . . . . . . . . . . . . . . . . . . . . . . 90

4.4. Quadrature matched hybrid switches. . . . . . . . . . . . . . . . . . . 91

4.5. Switched line phase shifter. . . . . . . . . . . . . . . . . . . . . . . . . 92

4.6. Uniform radiation pattern beam shape. . . . . . . . . . . . . . . . . . 93

4.7. Multibeam radiation pattern unbalance description. . . . . . . . . . . 94

4.8. Multibeam radiation pattern isolation description. . . . . . . . . . . . 94

4.9. Reective phase shifter construction. . . . . . . . . . . . . . . . . . . 95

4.10. Reective phase shifter measurements. . . . . . . . . . . . . . . . . . 97

4.11. Insertion loss variation due to the PIN diode parameters model. . . . 98

4.12. Reective congurations. . . . . . . . . . . . . . . . . . . . . . . . . . 98

4.13. Series-shunt reective phase shifter results. . . . . . . . . . . . . . . . 99

4.14. Single and Double PIN diode based SPDT switch. . . . . . . . . . . . 99

4.15. Graphic equivalent of the double SPDT operation. . . . . . . . . . . . 100

4.16. Comparison of the features of the single and double SPDT. . . . . . . 101

4.17. Double PIN diode SPDT switch Photograph. . . . . . . . . . . . . . . 101

x LIST OF FIGURES

4.18. Double PIN diode SPDT switch measurements. . . . . . . . . . . . . 102

4.19. SPDT full measurements at 24 GHz. . . . . . . . . . . . . . . . . . . 102

4.20. SPDT switch measurements over the temperature. . . . . . . . . . . . 104

4.21. Coupling between networks due to the open circuit of the isolated

path of the conventional SPDT switch. . . . . . . . . . . . . . . . . . 104

4.22. Complex SPDT switch with matched output ports. . . . . . . . . . . 105

4.23. Insertion loss, isolation and return loss of the complex SPDT switch

with matched output ports. . . . . . . . . . . . . . . . . . . . . . . . 107

List of Tables

1.1. Characteristics summary of the digital switches. . . . . . . . . . . . . 12

1.2. Characteristics summary of the analog phase shifters [1.31]. . . . . . . 14

2.1. Gain reduction and angle widening for scanning angles. . . . . . . . . 30

2.2. Characteristics summary of the passive electronic steerable techniques. 36

2.3. Amplitude coecients of the passive network. . . . . . . . . . . . . . 44

3.1. Comparison of PEC, PMC, and EBG ground planes for low prole

antenna designs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59

3.2. Antenna specications. . . . . . . . . . . . . . . . . . . . . . . . . . 75

4.1. Summary of formulas for SPST Switches. . . . . . . . . . . . . . . . . 90

4.2. Reective phase shifter [S] matrix. . . . . . . . . . . . . . . . . . . . . 96

4.3. Complex SPDT [S] matrix. . . . . . . . . . . . . . . . . . . . . . . . . 106

xi

Chapter 1

Introduction

Since the discovery of electromagnetic wave propagation in 1888 by Hertz, an-

tennas have been used to transmit information from one point to another through

the free space. The IEEE Standards Denitions of Terms for Antennas [1.1] de-

nes the antenna or aerial as a means for radiating or receiving radio waves. In

other words an antenna is a transducer which adapts the impedance of a guiding

device to the free space impedance and allows the transmission or reception of elec-

tromagnetic energy. During the last decades, there has been a growing interest in

developing more complex antenna systems. Nowadays, these antenna systems can

be used as sensors such as vigilance, detectors or RADAR, in personal communica-

tions (xed or mobile), instrumentation equipment, measurement purposes, satellite

communications or space exploration.

The work carried out since September 2009 in the Radiation Group, of the Tech-

nical University of Madrid (UPM) has been focused on the research of the application

of new technologies to antenna arrays and radio frequency circuits. This doctoral

thesis parts fundamentally from the necessity to nd more exible solutions that

could be apply in the analysis, design and construction of passive and active anten-

na systems.

1.1. Motivation

Since the early beginnings of telecommunications, some of the research eort in

the antenna eld has focused on getting higher directivities in order to reach longer

distances or transmit higher amount of information. The old days where big anten-

nas which work at low frequencies using earth surface wave or ionosferic propagation

1

2 INTRODUCTION

have nished due to the large dimension and low exibility of the antennas. Mo-

reover, by using these frequencies the operation bandwidth is very narrow and high

data rate systems can not be used. If higher frequency bands are used, the physical

dimensions of the antenna decrease and the available bandwidth increase. However,

the free space losses increase too, and the distances that they can reach are shorter.

Therefore the quest to achieve higher directivities keeps on. The directivity of an

antenna is its ability to gather the energy in a specic region of the space, to say

so, if the energy is focused in a smaller beam, this electromagnetic waves will reach

longer distances. Nevertheless, not only for reaching higher distances the directi-

vity can be used, but also for avoid external interference sources. In example in

radioastronomy, the antennas have very large directivities in order to obtain narrow

beams for deep space exploration where to avoid another heavenly bodies is very

important. As long as de directivity is directly proportional to the efective area of

the antenna (eq. 1.1, directivity for parabolic antennas) [1.2], the antennas tended

to follow the rule of thumb "the bigger, the better".

D =

(πD

λ

)2

εap (1.1)

Where D is the antenna diameter, λ is the wavelength and εap is the aperture

eciency.

For 29 years the Eelsberg Radio Telescope [1.3] was the largest fully steerable

radio telescope on Earth. In 2000 it was surpassed by the Green Bank Telescope

[1.4], placed in West Virginia with an elliptical 100 by 110-metre aperture. The

biggest antenna in the world is placed in Arecibo, Puerto Rico with 305 m [1.5],

nevertheless, this antenna does not have movement capabilities.

The construction, maintenance and operation aspects of these huge antennas

make them very complex and expensive, consequently other solutions were explo-

red. An astronomical interferometer achieves high-resolution observations using the

technique of aperture synthesis, mixing signals from a cluster of comparatively small

telescopes rather than a single very expensive monolithic telescope.

Examples of Astronomic interferometry clusters are the Very Large Array (VLA),

Motivation 3

(a) Very Large Array (VLA) New Mexico. (b) Atacama Large Millimeter Array (AL-

MA).

Figure 1.1: Astronomic interferometry clusters [1.6].

in New Mexico [1.7] or Atacama Large Millimeter Array (ALMA) [1.8]. The VLA

consists of twenty-seven 25-m-diameter antennas arranged in a Y-shaped array. Each

arm of the Y is approximately 21 km long and the antennas can be moved to various

positions on the arms by a rail-mounted transporter and it gets the smallest angular

resolution of 0.05 arcseconds at a wavelength of 7 mm. On the other hand ALMA is

compound by 66 12-meter and 7-metter diameter radio telescopes observing at milli-

meter and sub-millimeter wavelengths placed at 5000 meters altitude over distance

of 16 km. With a spatial resolution of 10 milliarcseconds is 10 times better than the

Very Large Array (VLA) and 5 times better than the Hubble Space Telescope. The

Square Kilometer Array (SKA) [1.9] is the biggest project of interferometry teles-

cope under developement, it will be deployed in Australia and South Africa and it

will have a total collecting area of approximately one square kilometer. The SKA

will combine the signals received from thousands of small antennas (15 m diameter

dish) spread over a distance of more than 3000 km. These antennas cover from 70

MHz to 30 GHz and simulate a single giant radio telescope capable of extremely

high sensitivity and angular resolution.

The main advantage of the antenna arrays over the parabolic dishes is the mul-

tibeam or beam steering capability. If the signals are properly combined with a

determined amplitude distribution or phase dierence, the group of antennas can

transmit or receive dierent levels of power from dierent points of the space wit-

4 INTRODUCTION

hout moving physically the antenna. This is very useful for high interference envi-

ronments, adaptive systems, etc. Nowadays this type of antennas, which work at

higher frequencies to achieve size reduction, are integrated in mobile devices. Howe-

ver the design and the integration of this type of systems is really complex because

neither the technology nor the theoretical foundations can handle the problems that

appear in the radiation pattern composition of the antenna elements.

The purpose of the work carried out in the rst part of the thesis is triple: in

the rst place, to identify and describe the problems related to active antennas

focusing on steering systems. Secondly, to study the solutions already proposed

exploring several dierent paths and techniques and nally to contribute with new

solutions. Nowadays broadband communication systems are required everywhere,

therefore there is a growing interes on COTM (Communications On The Move)

systems. They are installed in cars, ships and more recently in aircrafts. To achieve

large data rates in such a high frequency, it is necesary high signal to noise ratio.

In order to get more radiated or received power, the gain of the antenna is increase

but at the same time the beam is reduced, therefore it is necessary to accurately

steer the antenna. The antennas can be mechanically or electronically oriented.

By means of a thorough study of the art in steering systems, simulations and the

identication of the main problems and constraints, an airborne sub array antenna

for satellite communications is built. This prototype will be used to illustrate the

capabilities of a broad band, dual circular polarized 4 x 4 array. This antenna

includes miniaturized hybrid couplers and it is electronically steered by means of a

wideband butler network.

The second part of the thesis is related to the use of metamaterials in acti-

ve antennas. Since the discovery of metamaterials, the interest on it has grown

explosively. More than 10 years have took place since the emergence of the rst

metamaterials and since then they have proved to be exceptionally promising for

both research and applications. The investigation of metamaterials is currently one

of the most active topics in engineering and physics. A number of detailed review

articles and books have been published recently [1.101.15].

These complex materials allow us to achieve extraordinary electromagnetic pro-

Motivation 5

perties which are sometimes even not available in natural materials. The name

for articial complex materials that agree with this characteristic of unconventio-

nal properties can be called metamaterials. Metamaterials are dicult to dene

and to classify. A unique denition for metamaterial does not exist within the re-

search community and in the literature. One formal denition proposed in [1.16]

is macroscopic composites having man-made, three-dimensional, periodic cellular

architecture designed to produce an optimized combination, not available in nature,

of two or more responses to specic excitation. Another denition in [1.13]: a

metamaterial is an articial structure of material that, in a certain frequency ran-

ge presents unusual electromagnetic properties (as propagation of backward waves,

negative refraction, presence of forbidden zones, etc.), which gains its properties

from its structure rather than directly form its composition. Therefore, in order to

clarify the situation, a global denition for metamaterials that might satisfy most

researchers is artitial engineered complex electromagnetic functional structured

materials, by placing them in a periodic manner, have a superior electromagnetic

properties that can not be observed in the constituent materials used to manufacture

them.

The classication of these materials (Fig. 1.2) is also a hard task since nowadays

combination of fabrication techniques, properties and applications are combined in

order to nd new materials. However the concept of metamaterials is treated quite

general among the researchers with topics such as frequency selective surfaces (FSS),

electromagnetic/photonic bandgap (EBG/PBG) structures, left-handed (LH) ma-

terials, articial magnetic conductors (AMC) or high-impedance surfaces (HIS) or

hard/soft surfaces, articial dielectrics (AD), and plasmonic medias. A raw descrip-

tion of these materials is the following:

Frequency selective surfaces (FSS) are dielectric layers of very large ex-

tent, which contain planar conductive elements on its true side and which

backside is free. If the true side of the selective surface is illuminated by

harmonic waves of various frequencies, some waves are transmitted with a mi-

nimum attenuation, some waves are totally reected back to the half-space of

6 INTRODUCTION

Figure 1.2: Classication of metamaterials [1.17].

the source, and some waves are partially transmitted and partially reected.

So, a FSS can be viewed as a lter for plane waves at any angle of inciden-

ce [1.18].

Articial magnetic conductors (AMC) (or high impedance surfaces (HIS))

have new emerged properties as a magnetic response, even if the component

materials are non-magnetic. We can associate the hard and soft surfaces in

the class of the AMC because their behaviour can be used as AMC and perfect

electric conductors (PEC).

Left-handed (LH) materials have been called by many names as negative

refractive index (NRI) materia or backward wave media. These properties have

simultaneously negative permittivity and permeability. Composite right/left-

handed (CRLH) concept is an articial transmission line (TL) approach that

describe the behavior (RH or LH) of these medium depending on the frequency

range of working.

Articial Dielectrics (AD) consist of a large number of subwavelength con-

ducting obstacles embedded in a homogeneous host medium [1.19]. Calling

Motivation 7

such dielectrics as metamaterials make sense because the conducting proper-

ties of metals are being changed to a dielectric-type behavior in the macros-

copic properties. One of the kind of AD can be the magnetic materials (mu

negative (MNG) media).

Plasmonic medias (or epsilon negative (ENG) media) is the name given to

a discipline seeking to benet from the resonant interaction obtained under

certain conditions between an electromagnetic radiation and the free electrons

with the interface between a metal and a dielectric material. This interaction

generates density waves of electrons, behaving like waves and called plasmons

or surface plasmons at optical frequencies [1.20].

Electromagnetic/photonic bandgap (EBG/PBG) materials are also

known as photonic crystals. They are periodic structures that can be made

by metallic, dielectric or metallodielectric elements. These structures are used

to control and manipulate the propagation of electromagnetic waves. The

EBG structures have two important attributes that are to create a bandgap

operation and to localize frequency windows in the bandgap by breaking the

periodicity of the structure. The rst property is useful in using EBG as a

spatial and frequency lter, while the second property is useful in propagating

the EM wave in a desired frequency and direction. There exist 1D, 2D or

3D periodic structures in which the propagation of electromagnetic waves is

inhibited in some frequency bands (called bandgaps or stopbands) or directions

that are determined by the periodicity of the materials and their dielectric

constants.

Therefore, metamaterials can contribute to enhance the performance of active

antennas. The surface wave supression properties of the last type of metamaterials

described, EBG materials are used in this work to reduce the mutual coupling bet-

ween radiating elements. The reason for the integration of EBG between radiating

elements as electromagnetic barriers is to replace the cavities which are heavier and

more expensive.

8 INTRODUCTION

The last part of this thesis is dedicated to the distribution networks that feed the

radiating elements in active antennas. The behaviour of these distribution networks

can be modied by placing components which change their capacitance, resistance,

permeability, etc. by means of a control signal. They can split, divert, block or

allow the current and therefore they change the array antenna features. This work is

oriented to explore the technologies, topologies and concepts regarding phase shifting

in the distribution networks. There are several ways to classify this components,

but from the point of view of steering antennas there are two main groups: digital,

which actuates as switches between dierent paths, and analog, which can modify

the characteristic impedance of the transmission line, thus the propagation speed is

change and therefore a phase shift is obtained.

Digital:There are two main switching techniques based on semiconductor bia-

sed: CMOS and PIN diodes. These devices change their conductor or isola-

tion behavior as a function of its chemistry composition. On the other hand,

MEMS recently attracts more interest, due to its mechanical operation and its

excellent features.

-MOSFET transistors use an electric eld to control the shape and hence

the conductivity of a channel of one type of charge carrier in a semiconductor

material. These devices has three terminals, gate, drain and source, in such

a way that a negative gate-to-source voltage causes a depletion region on the

channel for a n type. If the depletion region expands to completely close the

channel, the resistance of the channel from source to drain becomes large and

the FET is eectively turned o like a switch. Likewise a positive gate-to-

source voltage increases the channel size and allows electrons to ow easily

(Fig. 1.3).

CMOS switches Complementary metal-oxide-semiconductor is a technology

for constructing integrated circuits. The most important characteristics of

CMOS devices are high noise immunity and low static power consumption.

Since one transistor of the pair is always o, the series combination draws

signicant power only momentarily during switching between on and o states.

Motivation 9

Figure 1.3: Schematic view of a Field Eect Transtistor.

Consequently, CMOS devices do not produce waste heat and therefore the

power consumption is lower. However, the main drawback is that nowadays

the higher frequencies where they can work do not reach eectively 20 GHz.

There are several materials which the transistors are made from, the switches

are commonly designed using III-V semiconductor based such as AlP or more

recently GaAs, which are used in high power and high throughput applications

[1.21]. New dierent materials are currently tested, such as Silicon-on-Sapphire

(SOS) [1.22], which obtains higher frequency operation ranges but can not

handle high power.

The main advantage of the FET is its high input resistance. Thus, it is a

voltage-controlled device, and shows a high degree of isolation between input

and output. It has low noise level, it exhibits no oset voltage at zero drain

current and it has good thermal stability. Its main disadvantage is the inte-

gration of device in the systems, in order to reduce the insertion losses and

the noise gure, the introduction of these devices has to be made during the

photolitography processes, the only way to make protable the systems is for

mass production. However this problems are being currently addressed [1.23].

-The Micro ElectroMechanical Systems (MEMS) are micro mechanized

surfaces which use a mechanical movement to obtain a short or open circuit

in a radio frequency transmission line, with very low losses. The movement is

caused by an electro static force between to electrodes which are fed with a

large diference voltage. The MEMS can be used from µ-wave frequencies up

10 INTRODUCTION

to mm-wave frequencies (0.1-100 GHz) and they have a great performance in

high frequencies up to 30 - 40 GHz, however, in higher frequencies due to the

fabrication process the package capacitances yields higher insertion losses.

RF MEMS switches oer higher performance than pin diodes or FET diode

switches and those can work up to 120 GHz. However, important issues such as

long reliability, packaging techniques and production costs, are currently being

addressed. The switches are either fabricated using an attached membrane

xed in two points or a oating cantilever, and are modeled as mechanical

springs with an equivalent spring constant k [N/m]. The actuation mechanism

is achieved using an electrostatic force between the top and bottom electrodes.

This electrostatic force depends on the geometry of the switch, but mainly on

the actuation voltage [1.24]. Depending on the topology there are two main

types of MEMS: cantilever (Fig. 1.4(a)), where the application requires a

series switch and membrane or bridge (Fig. 1.4(b)) for shunt schemes.

(a) Cantilever [1.25]. (b) Capacitor bridge [1.24].

Figure 1.4: Micro ElectroMechanical Systems schematic view.

Due to the physical constrains the advantages of this systems are the near-zero

power consumption (only 10 - 100 nJ per switching cycle), very high isolation

thanks to its low o-state capacitances (2-4 fF) and very low insertion loss due

to its low series resistance up to 30 GHz. However, MEMS have low switching

speed, the power that the element can handle is not too large, the actuation

voltage is very high and the long term reliability is only up to 10 billion cycles.

Therefore this components can full the requirements in special applications

systems such as defense and ground station satellite communications, but are

expensive for terminal users.

The main advantages of this devices are the low insertion losses, linearity,

Motivation 11

null power compsumtion, and high isolation. On the other hand there are

some drawbacks such as low operation speed, high voltage operation, power

handling, limited life operation time or packaging.

-The P-I-N diode is a current controlled resistor at radio and microwave

frequencies. A unique feature of these diodes is their ability to control large

amounts of RF power with much lower levels of DC. It is a silicon semiconduc-

tor diode in which a high resistivity intrinsic I-region is between a P-type and

a N-type region. When the PIN diode is forward biased holes and electrons are

injected into the I-region. These charges do not immediately annihilate each

other. These charges stay alive (τ , carrier lifetime) and lowers the eective

resistance of the I-region to a value RS. When the PIN diode is at zero or

reverse bias there is no stored charge in the I-region and the diode appears as

a capacitor, CT , shunted by a parallel resistance RP [1.26].

By varying the I-region (Fig. 1.5(a)) width or area it is possible to construct

PIN diodes of dierent shapes but similar RS and CT characteristics. However,

the thicker I-region diode would have a higher bulk or RF breakdown voltage

and better distortion properties. On the other hand the thinner device would

have faster switching speed [1.27] - [1.28].

(a) PIN diode outline. (b) Forward and reverse bias.

Figure 1.5: PIN diode and equivalent circuit.

Another important issue for these components is the biasing. The higher IF ,

the lower RS and therefore the lower the insertion losses, when the series

topology is used. For this same series topology, when the reverse biased is

applied, the higher -VR, the lower the CT , and the higher isolation is obtained

12 INTRODUCTION

Table 1.1: Characteristics summary of the digital switches.

Parameter GaAs pHEMT GaAs PIN Si PIN RF-MEMS

Insertion loss [dB] 0.35-0.65 0.3-0.6 0.4-1.2 0.1-0.5

Isolation [dB] >30 12-25 34-55 20-30

Actuation Voltage [V] 5 5 2-5 30-80

Switching speed [µs] 0.02-0.06 0.02 0.02 2-10

Power consumption [µW] <100 <100 <1 0

Power handling [dBm] 36 33 30 23-27

Linearity IP3 [dBm] 55-72 65 70 65

[1.29]. However due to system restrictions, not always high IF and -VR are

possible.

Finally, in order to have an overview, in Table 1.1 a comparison between the three

technologies is shown. Due to the requirements of our design in terms of operation

frequency, power handle, insertion losses and isolation, the technology chose is PIN

diodes.

Analog:The phase shifters based on analog devices such as varactor diodes

or ferrite core have the main advantage of not to have quantization errors and

very low insertion loss. These devices follow a continuous signal (voltage or

current) which yields a behavior change in the conducting material properties

without steps. However the control signals are less robust against noise.

-The operation of FERRITE cores phase shifters is relatively easy. By

changing the magnetic eld inside a waveguide, the propagation constant can

be modied, and a phase dierence obtained. The current that ows along

a coil with a ferrite core is electronically adjusted (Fig. 1.6), obtaining with

that 360o phase variation [1.30].

-VARACTOR diodes are P-N diodes that changes its capacitance and the

Motivation 13

Figure 1.6: Ferrite core phase shifter.

series resistance as the reverse bias applied to the diode is varied. No current

ows but since the thickness of the depletion zone varies with the applied bias

voltage, the capacitance of the diode can be made to vary. Generally, the

depletion region thickness is proportional to the square root of the applied

voltage, capacitance is inversely proportional to the depletion region thickness

(Fig. 1.7). Thus, the capacitance is inversely proportional to the square root

of the applied voltage. This property of capacitance variation is utilized to

achieve a change of phase response or line impedance. Therefore, they can be

used as a phase propagation in phased array antenna applications.

Figure 1.7: Varactor diode outline.

Table 1.2 shows a summary of the analog phase shifters devices. In orther to

establish a comparison with digital devices the same parameters have been shown.

The quantitative gures in these type of devices are determined by the construction

process, therefore only qualitative values are presented.

By presenting this historical evolution of the active antennas focusing on the

14 INTRODUCTION

Table 1.2: Characteristics summary of the analog phase shifters [1.31].

Parameter Varactor Ferrite

Loss High Medium

Actuation Voltage [V] 0-30 100

Switching speed Nano seconds Milli seconds

Power consumption Negligible Negligible

Mounting complexity Low Medium/high

Cost Medium High

steering systems and the problems so far that are currently being addressed the

following objectives are pursued in this thesis.

1.2. Objectives

The eld of active antennas is very wide in surveillance radar technology, ATC

(Air Trac Control), SAR (Synthetic Aperture Radar), etc. Applications of active

antennas are already well identied what is still under investigation are the techno-

logies, materials, switching networks, etc. Considering that a lot of experiences in

the area of steering antennas are achieved, the objective of this doctoral thesis is

trying to contribute to an important series of aspects in the scope of the potential

application of these concepts in the design, analysis and prototyping in the eld of

active antennas, where the Radiation Group has wide experience. The electronic

steerable antennas can be divided in three dierent parts.

First, the selection of group topology, secondly the radiating elements itself, and

nally, the phase shifters placed in the distribution networks.

This thesis is divided in three main parts as well, where the following objetives

are pursued:

Objetive 1: this thesis analyzes the array antennas and their problems re-

Objectives 15

garding size, materials, construction or topology for steerable antennas. A

thorough study is carried out and trade o between angular resolution, direc-

tivity reduction in squinted angles, cross beam losses and number of bits of the

phase shifters are decided for a real satellite communications steering antenna

in X band (7.25-8.4 GHz). After a large number of simulations, the best op-

tion is identied and a 4x4 sub array prototype is constructed and measured

to demonstrate its operation.

Objetive 2: this thesis explores the use of metamaterials for mutual coupling

reduction purposes. By using double layer and via edge location, mushroom

EBG materials are reduced by 30% and placed between radiating elements

in low permittivity substrates. In this way the surface waves that appears in

planar antennas are suppressed in the same way that cavities work but this

solution is lighter and it can be integrated in a photolitography process. These

electromagnetic barriers are used in the previous sub array in order to enhance

its radiation properties.

Objetive 3: this thesis proposes the combination of the well known topolo-

gies and technologies regarding switches and phase shifters to get important

features for phased arrays antennas such as match or unmatch ports, maxi-

mum isolation between ports, balanced or unbalanced power distribution and

of course phase dierence accuracy.

With these objetives, the general purpose of this work is to be a practical appli-

cation of the proposed methods to real antenna systems. Therefore, this doctoral

thesis allows to extend the knowledge of the analysis, design and operation of active

antennas focusing on steerable ones, and proposes possible solutions that help to

improve the steerable antenna performances.

As observed from the description of the main objetives of the thesis, it can

be noticed that a similar methodology is followed in the three areas in order to

accomplish each of the objetives. First of all, a thorough study of the state of art in

the topic of interest is given, in order to know the situation of research in the area

of active antennas. Afterwards, theoretical studies are done in order to propose a

16 INTRODUCTION

solution, to construct and measure the prototype and identify the advantages and

drawbacks that the proposed solution may have. These last two parts, prototyping

and measuring, are the key aspects in this thesis which is not focused in theoretical

formulations but practical solutions. Finally, the analysis of the results is done, and

the main conclusions of each of the three areas of study are given.

1.3. Outline of the thesis

The main document is organized in ve chapters. Chapter 1, already exposed,

presents the motivation which leads this work. Furthermore, the main objetives are

highlighted and an outline of the thesis is detailed.

In Chapter 2, a complete revision of the array antenna basics is illustrated.

Issues such as the amplitude and phase feeding laws, and antenna feeding networks

are shown along this chapter. Through the design of a real airborne steerable anten-

na for satellite communications in X band, the most common problems regarding the

number of beams, bits selected, widening of the main beam and consequently direc-

tivity losses are addressed. At the end of this chapter measurements of a subarray

prototype are shown.

Chapter 3 presents the fundamental theory of Electromagnetic Band Gap (EBG)

Materials and their application for mutual coupling reduction. In this chapter, an

analytical design method is extended to double layer structures for size reduction.

These new smaller structures are used in steering antennas to cancel surface wave

propagation modes and reduce mutual coupling in planar antennas.

Chapter 4 illustrates a thorough revision of the topologies that are used in the

design of switches and phase shifters so far. In order to overcome the demanding

requirements of switchable beam antennas used for automovile RADAR a combina-

tion of switch topologies to obtain equal insertion losses in a reective phase shifter

is presented. Besides, a matched ports SPDT based on the previous reective phase

shifter is proposed.

Finally, Chapter 5 gives the conclusions in the form of the original contributions

of this Ph.D. dissertation and the future research lines that it has given rise to.

Outline of the thesis 17

Furthermore, the framework in which this work has been carried out (research pro-

jects, nancial support, international collaborations...) are reported together with

the publications this Ph.D. dissertation has generated.

References

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01, 1983.

[1.2] A. W. Love. Reector Antennas. IEEE Press, 1978.

[1.3] O. Hachenberg, B.H. Grahl, and R. Wielebinski. The 100-meter radio telescope

at Eelsberg. Proceedings of the IEEE, 61(9):1288 1295, Sept. 1973.

[1.4] S. Srikanth, R. Norrod, L. King, and D. Parker. An overview of the green

bank telescope. In Antennas and Propagation Society International Symposium,

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[1.5] P.A. Castleberg and K.M. Xilouris. The Arecibo observatory. Potentials, IEEE,

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and performance of a modern synthesis radio telescope. Proceedings of the

IEEE, 71(11):1295 1320, Nov. 1983.

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red, Millimeter and Terahertz Waves (IRMMW-THz), 2011 36th International

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pagation, IEEE Transactions on, 53(1):27, Jan. 2005.

19

20 REFERENCES

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and Engineering Exploration. Wiley and IEEE Press, 2006.

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comunicación, Technical University of Madrid (UPM), 2008.

[1.18] B.A. Munk. Frequency selective surfaces: theory and design. John Wiley and

sons IEEE press, 2000.

[1.19] R.E. Collin. Field theory of guided waves. New York: IEEE Press, 1991.

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REFERENCES 21

[1.22] D. Kelly, C. Brindle, C. Kemerling, and M. Stuber. The state-of-the-art of

silicon-on-sapphire cmos RF switches. In Compound Semiconductor Integrated

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Chapter 2

State of the art on multibeam and

recongurable steering array

antennas

Nowadays the communication systems are more and more demanding, the re-

quirements of these systems in terms of the data throughput, number and location

of users force the designers to explore new techniques to develop the systems. An-

tennas have a very important role in telecommunication systems because they can

focus the transmited power or radiate it in all the directions, to avoid another user

signals or to explore dierent angles. To say so, the antennas can vary their beha-

viour in order to adapt themselves to the environment and to oer the best possible

conguration. Those antennas are known as well as ADAPTIVE ANTENNAS.

2.1. Introduction

By applying the superposition principle due to the linearity of the Maxwell's

equations and considering that the current in each element is the same as that of

the isolated element (neglecting coupling), the total eld of the array is determined

by the vector addition of the elds radiated by the individual elements. This is

usually not the case and depends on the separation between the elements. To

provide very directive patterns, it is necessary that the elds from the elements of

the array interference constructively (add) in the desired regions and interference

destructively (cancel each other) in the other regions. If the radiating elements of

23

24 STATE OF THE ART ON MULTIBEAM AND . . .

the array are identical, there are ve parameters that can be used to shape the

overall radiation pattern of the antenna:

1. The geometrical conguration of the overall array (linear, circular, rectangular,

spherical, etc).

2. The relative displacement between the elements.

3. The excitation amplitude of the individual elements.

4. The excitation phase of the individual elements.

5. The relative pattern of the individual elements.

Usually, the elements are equal and they have the same spatial orientation, the-

refore they are completely equivalent by a simple translation (Fig. 2.1). In this

case, the array can be described by the relative position of the radiating elements,

the feeding currents and the radiation pattern of each element. The eld radiated

of each element can be described by means of the radiation pattern of the radiating

element placed in the origin axis fed by a reference current I0 and a phase term,

which takes into account the displacement respect to the origin axis (eq. 2.1).

En(r, θ, φ) = Ee(r, θ, φ)InI0ejk0rrn (2.1)

Figure 2.1: Array splitting problem.

Introduction 25

(a) In transmission. (b) In reception.

Figure 2.2: Wavefront combination to get electronic steering.

The introduction of a phase shift α between two consecutive elements in an

antenna array allows to adjust the steering direction of the main beam without

moving the antenna physically. Through the reciprocity theorem, this eect can be

explain in two dierent ways:

In transmission: The phase shifters placed in the distribution network create

a delay between the wavefront of each element. The combination of the wave-

fronts of all the radiating elements along the time shows the desired steering

direction (Fig. 2.2(a)).

In reception: The wavefront of the desired direction travels with a determined

velocity, however, this does not reach all the radiating elements at the same

time. Those radiating elements, which receive earlier the wavefront, must

delay their contribution to the summation in order to sum all the contributions

appropiately (Fig. 2.2(b)).

From the eq. 2.2, a phase shift α can be introduced, as a step forward respect

to the previous radiating element: An = anejnα

FA(θ, φ) =∑n

anejnαejnk0dcosθ =

∑n

anejn(k0dcosθ+α) =

∑n

anejnψ (2.2)

Applying the coordinate transformation shown in eq. 2.3, it is observed that the

array factor as function of ψ is the inverse DFT of the magnitude feeding law.

ψ = k0dcosθ + α (2.3)


Similarly to eq. 2.3, where the periodicity of the function is 2π, in eq. 2.2 the

variable ψ has a period 2π. However, the radiation pattern has specic limits due

to the real values of the variable θ. The visible margin is the collection of the array

factor values which correspond directly with the radiation pattern of the antenna.

Its lenght is 2k0d and its center is ψ = α.

0 ≤ θ ≤ π → −k0d+ α < ψ < k0d+ α (2.4)

Figure 2.3: Visible margin of an array in ψ and θ.

In Fig. 2.3 the visible margin of a generic steering array is shown. From this

gure, three types of antennas can be named as function of the phase shift α:

Broadside array: the maximum of the radiation pattern is perpendicular to

the plane where the radiating elements are placed (θmax = π/2), in this case

Introduction 27

α = 0 and the visible margin is −k0d < ψ < k0d.

Exploration array: the maximum of the radiation pattern points to θmax

which is controlled by the progressive phase shift α (eq. 2.5).

θmax→ψ=0 = arccos

(−αk0d

)(2.5)

End re array: the maximum of the radiation pattern is located in the

surface where the radiating elements are placed (θmax = 0 or π). In this

case (θmax = 0) the value of α is equal to −k0d and the visible margin is

−2k0d < ψ < 0.

There are some important aspects which must be taken into account when an

exploration array is design. If the the distance between the elements d and/or the

phase shift α are higher than specic values, a replica of the main lobe could appear

in the radiation pattern, this phenomena is called grating lobes. This eect reduce

the separation between elements to 0.6 - 0.8λ for broadside arrays and 0.4 - 0.5 λ

for end-re arrays.

The recongurable antennas are dened as antennas that can radiate on demand

at several predetermined frequencies, or create rejection notches at various frequen-

cies, they can change their polarization or their radiation patterns on demand. The

motivation of this type of systems is to congure or to adjust the operation under

variable conditions, recover their functionality under any adverse or anormal condi-

tions by self-conguration (self-healing) such as degraded components that can be

recovered or damaged components that can be bypassed.

This is a very wide topic because it has a lot of application areas from personal

devices to high complexity facilities. It can be divided in its kind of application,

instant of change, used devices, parameters of change and exibility limits. These

systems are very wide spread because they are used in military phased arrays, mobile

base stations, broadcast facilities and industrial or domestic sensors.

Regarding instant of change the recongurable antennas can be divided in two

groups.


The ones which change the property or propperties of the antenna dinamically,

that is to say, during their operation

and the ones which can be adjust before the operation.

An example for the rst ones are the phased arrays which recongures beam

pointing direction during operation to look for targets over wide area of sky. A

mobile base station antenna is mechanically tilted when they are set in order to

adjust for correct coverage before turn it on. The main advantages of the dynamic

systems are the speed and exibility, however they have some drawbacks such as

high cost, additional DC power and additive RF losses that not all the systems can

handle.

There are several ways to control the operation of a recongurable system. On

the one hand, the ones controlled mechanically (hand, motor, hydraulically) which

are slow but with a low cost and they do not add additional RF losses, and on

the other hand the ones controlled electronically such as switches based on MEMS,

PIN diodes or CMOS and analog or digital phase shifters which are faster but their

drawbacks are the high cost and the additional RF losses.

The exibility limit is an important parameter which directly aects on the

complexity of the system. For example, in a phased array, the fewer the pointing

beam steps, the simpler the system is. This issue limits the improvement of the

system regarding steering accuracy or gain in the desired direction.

Finally, there are many groups in which the recongurable antennas can be di-

vided in terms of the parameter change. They can change their radiation pattern,

beam direction (steering antennas) [2.1], beamwith or coverage angle (reector an-

tennas which are fed by several horn antennas in order to obtain dierent beam

shapes to earth coverage) [2.2], sidelobe level (in systems where the noise level can

change) [2.3], null steering (to avoid interferences) or single/multiple or split beam

(reector antennas) [2.4]. Another parameter of change is the polarization, it can

change from vertical to horizontal [2.52.7] or change the type of circular Left/Right

Handed Polarization (LHCP/RHCP) [2.8]. By feeding a radiating element with two

orthogonal modes which have a phase dierence of 90o the circular polarization is

Introduction 29

obtained, i.e. a patch antenna fed by a 90o 3dB branch line coupler ([2.9]). And

the last one is the operation frequency band [2.10], there are Ultra Wide Band

(UWB) antennas [2.11], there are systems which can broadering the band (large

number of military radios from HF to UHF) or systems that can use multiple bands

(the mobile phones use nowadays 5 or more frequency bands) [2.12]. The methods

in order to achieve these recongurable characteristics are not formal established,

they can change the antenna geometry, feeding points, add additional reactance,

recongurable matching circuits [2.13], etc.

There are no real design guidelines or theories for the operation of recongurable

antennas and some clear goals to advance the theoretical foundation of these anten-

nas are needed. In this way, many more recongurable antennas could be included

in complex and multifunctional systems.

The main target of this work is to study the steerable systems and to propose new

techniques to enhance their performance. In the next section, the main problems

and constructing strategies when using scanning arrays are described.

As it was previously mentioned by controlling the phase dierence between the

elements, the maximum radiation can be squinted in any desired direction to form

a scanning array. Since in the phased array technology the scanning must be conti-

nuous, the system should be capable of continuously varying the progressive phase

between the elements. In practice, this is accomplished electronically by the use of

analog or digital phase shifters. However, analog systems are less exible and more

complex than digital ones. Therefore, systems which are based on beam switching

are preferred. These systems are simpler but several drawbacks have to be taken

into account in the design process.

The Half Power BeamWidth (HPBW) of an array is dened as the angle where

the directivity in the main lobe is below the maximum 3 dB. In an array the direc-

tivity is directly related to the radiating element directivity and the array factor.

When electronic steering is applied, the visible margin (Fig. 2.3) distort the ψ to θ

transformation and the mean beam gets wider (2.6).

BW ∝ 1

cosθ(2.6)


Table 2.1: Gain reduction and angle widening for scanning angles.

Angle 90o 70o 50o 30o 10o

Gain (dB) 30 29.72 28.84 26.98 22.39

Beamwidth (o) 14.32 15.24 18.69 28.64 82.48

Figure 2.4: HPBW vs scanning angle.

The eect of reduction in squinted angles is directly related to the previous one.

There is a directivity reduction for the farther steering angles which is a consequence

of the wider beamwidth. In Fig. 2.5 it is shown a directivity of 2 dB when the

steering angle is ±50o. Therefore, the farther the scanning angle, the higher are the

directivity losses. In table 2.1 the numerical values of gain and beamwidth for a 16

rows of elements array with scanning elevation are shown.

In a discrete scanning system, it is very important the number of beams that

this is going to use. As it can be seen in Fig. 2.6 there are two diferent systems with

±20o scanning angles. However, one system has four beams meanwhile the other

one has eight beams. For the four beams system the crossing point between beams

is under 15 dB, which means that the link budget could not be satised for certain

Introduction 31

Figure 2.5: Directivity reduction vs scanning angle.

scanning angles. Therefore, the system with eigth beams has better behaviour, since

the crossing point between beams is only 4 dB below the maximum.

Figure 2.6: Crossing point losses vs number of beams.


In this case the dierent number of bits in the phase shifters is analyzed. When

discrete number of shift angles are used, each radiating element is not fed with

the necessary phase, but with one which is aproximated. In Fig 2.7, the radiation

patterns of three systems with ±30o but dierent number of bits are shown. It can

be noticed that the lower number of bits, the higher quantization noise. Eventhough

the same scanning angles are obtained, in systems where the SNR is important for

data rate transmission, high accuracy phase shifters are needed.

Figure 2.7: Quantization noise oor vs number of bits.

The antenna arrays can be divided in two groups depending on the way in which

the radiating elements are fed. On the one hand the passive arrays where the

radiating elements are fed by only one source, and the amplitude and phase feeding

are function of a propagation medium (i.e. the slot arrays antennas which are fed by

a waveguide, used for high power transmission). And on the other hand the active

arrays where each radiating element is fed by an independent source. This last group

is more exible because the feeding law can be controled in a easier way. But this is

not the only one advantage, these systems are more robust against failures since the

wrong operation of a radiating element do not aect signicantly the total radiation

pattern.

Multibeam passive antennas 33

(a) Steering angle of 70o (b) Steering angle of 50o

Figure 2.8: Appearance of grating lobes due to failures in the phase feeding law.

The last eect considered try to illustrate the problem when some phase shifters

do not work properly. There is a situation where the failure of certain parts could

be critical, in the case where a periodicity exists in the phase feeding law, grating

lobes can appear. In Fig. 2.8(a) and Fig. 2.8(b) the radiation patterns of three

systems with dierent failures are shown. It can be noticed that when the radiating

elements are fed in groups of two with the same phase, it is similar to place the

elements with double separation between them and therefore, when steer capability

is used, grating lobes appear. The same situation takes place when the elements are

fed in groups of three with the same phase.

In the next sections the dierent techniques to obtain the phase shift between

the radiating elements are described.

2.2. Multibeam passive antennas

There are several passive methods and devices in order to obtain the steering

angles of an array. After a thorough literature revision here are presented most of

the well known passive techniques.


Butler Matrix Network

The Butler matrix is a passive network with 2n inputs, 2n outputs, 2n−1log22n

hybrid couplers, crossovers and phase shifters [2.14] [2.15]. The function of a Butler

matrix is to combine signals in phase going to or coming from an antenna array. It

produces 2n beams with constant angular separation. Each output signal at n port

(Sn) can be expressed as follows:

Sn =n∑

m=1

Amejαmn (2.7)

where Am is the input signal at port m and αmn is the phase dierence bet-

ween input ports. There are several dierent congurations in order to obtain the

broadside beam [2.16, 2.17], group a large number of radiating elements [2.18, 2.19]

or to get multiple beams in a hexagonal planar array with triangular distribution

[2.20]. In [2.21] a reective Butler network is proposed, its peculiarity resides on

the aditional phase shifts in order to get a symmetric structure, consequently, this

device can be used in transmission and reception at the same time.

Blass Matrix

The Blass matrix [2.22] consists in a determined number of interconnected feeding

lines which feed an antenna array. Two groups of interconnected lines by directional

couplers form the network. The applied signal in the input terminals travels along

the dierent paths which yields a phase shift. The amplitude can be modied by

the directional couplers design [2.23]. In [2.24, 2.25] a synthesis method for the

directional couplers is shown. In [2.26, 2.27] dierent congurations are presented

when applying to antenna arrays.

Nolen Matrix

The Nolen matrix combines the properties from both the Butler and the Blass

one and it was introduced in [2.28]. The Nolen matrix is actually lossless like the

Butler one and serial feeding network like the Blass one. This latter characteristic

is of great interest to avoid the crossovers. However being lossless, the Nolen matrix


is limited like the Butler one to orthogonal set of output excitations while the Blass

matrix has no constrains on the output excitations. Therefore, a Nolen matrix can be

seen as a lossless Blass matrix [2.29]. As a counterpart, this lossless characteristic

imposes orthogonality on the output excitations that are not found on the Blass

matrices. In fact, the less orthogonal the output excitations of a Blass matrix the

higher the losses.

Rotman lens

The Rotman lens [2.30] are feeding networks which obtain in a easy way a

discrete main beam steering. The fundamental of these lens resides in the time

delay due to the path dierence, consequently the scanning angles are frequency

independent. In [2.31] an eight element array are fed by a Rotman lens.

The weight of dielectric lens does not allow the use of these lens at µwave fre-

quencies, however these are nowadays important for EHF and mmwave frequencies.

The parallel metallic lens solve the weight problem [2.32], where the spacing between

parallel plates is substituted by a guided medium and the medium permittivity is

directly related to the distance between parallel plates.

Summary of passive electronic steerable techniques

In table 2.2 the main characteristics of the passive techniques in order to get

scanning angles are shown. In this way, it is easy to compare the advantages and

the disadvantages of these systems. It can be concluded for all the systems that the

higher number of beams, the bigger the lattice. Therefore the space constraints in

the system will limit its exibility.

2.2.1. Wide band Butler matrix network at X band

In this work, a wideband Butler matrix [2.33, 2.34] with four inputs and four

outputs has been designed. The output phase dierences are ±45 and ±135, and

crossovers are implemented with two hybrid couplers in cascade. The hybrid coupler

is the key design part since the rest of the elements are based on it. By applying


Table 2.2: Characteristics summary of the passive electronic steerable techniques.

Type Scanning

range

Aperture

size

SLL Bandwidth Eciency

Rotman ±45 10λ -20dB 4:1 >63%

Hybrid lens ±10 130λ -16dB - >60%

Gradual index lens ±360 20λ -13dB 9:1 -

Hybrid reector ±30 230λ -22dB - >76%

Blass matrix ±60 15λ -13dB <1% 75%

Butler matrix ±60 16λ -13dB >2:1 40%

Barlett's [2.35, 2.36] theorem to two stages hybrid coupler resolution the following

equations are drawn:

(Z02

Z0

)2

=2Z01Z03

Z20 + Z2

01(Z01

Z0

c0 − 1

)= 1− c20 (2.8)

where c20 is the power ratio between ports 2 and 3. In this case, the design

condition is c20 = 12. There are three missing values (Z01, Z02 and Z03) and only two

equations. Therefore a degree of freedom exists. In this case, the periodic solution

(Z02 = Z0) is selected.

The prototype of the Butler network can be seen in Fig. 2.9(a) [2.37]. As it

can be seen in Fig. 2.9(b), a plain response over the frequency is obtained for

transmission parameters (Sji=-6.5±2). Besides, the reection coecient and the

isolation are adequate (Sii and Sjj < −14dB). The phase dierence between the

outputs is ±45 for port 1 (Fig.2.10(a)) and ±135 for port 2 (Fig.2.10(b)) along

the whole bandwidth.


(a) Construction (b) Amplitude Scattering parameters

Figure 2.9: Wide band Butler Matrix network

(a) Port 1 (b) Port2

Figure 2.10: Phase of the scattering parameters of a wideband Butler matrix net-

work.


2.3. Scanning active antennas

The complexity of the scanning active antennas which obtain a continuous beam

steering (instead the discrete group of beams obtained with a multibeam passive

antenna) resides on its feeding distribution network. In it every element (for 2D

scanning arrays), row (for elevation scanning) or column (for azimuth scanning)

is feed with a dierent phase. This phase is provided by an element called phase

shifter.

A phase shifter is a device used in automation, conversion technology, and mea-

suring technology to change the phase of electromagnetic oscillations. The design

of a phase shifter depends on the range of operating frequencies, the limits of the

phase change, and the accuracy of the equipment. They can be analog or digital.

Electrically controlled analog phase shifters can be realized with varactor diodes

that change capacitance with voltage or ferrite phase shifters where the current

that ows along a coil with a ferrite core changes the speed propagation inside a

waveguide.

Most phase shifters are of the digital variety, as they are more immune to noise on

their voltage control lines. Digital phase shifters provide a discrete set of phase states

that are controlled by two-state "phase bits."The highest order bit is 180 degrees,

the next highest is 90 degrees, then 45 degrees, etc., as 360 degrees is divided into

smaller and smaller binary steps. A three bit phase shifter would have a 45 degree

least signicant bit (LSB), while a six bit phase shifter would have a 5.625 least

signicant bit. These set of phases are produced in two ways, one posibility is to

obtained dierent values of L and C with discrete components such as capacitors

and inductors or by switching lines of dierent lengths which are commuted with

CMOS switches, MEMS or PIN diodes, this last group is know as well as true time

delays because they do not change the phase dierence between the outputs with

the frequency variance.

Scanning active antennas 39

2.3.1. Airborne steering antenna

The purpose of this work is to build a broadband airborne satellite communi-

cation system for low observability airplanes. The system will be able to transmit

and receive at X band MPEG-2 video format with IP-v.6 encapsulation under the

DVB-S2 standar.

The antenna developed for this system must full the recommendations ITU-R

S580-6 [2.38] and ITU-R S.465-5 [2.39]. The antenna consists of double stacked

printed elements grouped in an array, this terminal works in a frequency band from

7.25 up to 8.4 GHz (15% of bandwidth), where both bands transmission (7.9-8.4

GHz, RHCP) and reception (7.25-7.75 GHz, LHCP) are included simultaneously

(full duplex system). The antenna has a gain of 31 dBi to ensure 99.9% system

operation (the gain limit of the visible region is 28 dBi for θ=40o), and it has a

radiation pattern with a beamwidth smaler than 10o. The antenna operates with

dual circular polarization and it has the capability to steer in elevation from 90o to

40o electronically and 360o in azimuth with a motorized junction. A diagram of the

antenna can be seen in Fig. 2.11.

Figure 2.11: Airborne steering antenna for satellite communications [2.40].

A rectangular structure is selected formed by an array of 16x24 elements. These

elements are treated in two dierent ways:

Rows: First grouped in rows, separated a distance between them of 0.85λ|fminin order to increase the eective area and consequently the gain. In case of


using an uniform excitation the side lobe levels exceed the limit of the recom-

mendations mentioned before. Due to that restriction, a progresive reduction

of the feeding amplitude from the centre to the edges of each row wil be im-

plemented with a passive unbalanced feeding network.

Columns: Thus, 16 rows of 24 elements are grouped vertically and separated

0.5λ|fmin. The steering direction of the main beam is achieved due to the

phase shifting feeding between each row. The terminal will be integrated in

the fuselage and because of that, it will be necessary to steer close to endre

locations. Therefore to avoid the closer angles to endre, where the directivity

is reduced, the planar antenna is raised. With the installation of a wedge of

30o the nal steering directions will be from 90o to 40o. The antenna nal

dimensions are 33 x 85 x 20 cm.

Radiating element

Each radiating element is composed of two stacked patches (as shown in Fig.

2.3.1): the upper one is fed by electromagnetic coupling (thickness substrate 1

=0.254 mm), meanwhile the bottom one (thickness substrate 2=1.143 mm) is fed

by two via holes (diameter=1.1 mm) to get the circular polarization. These vias are

connected with a two-stages miniaturized hybrid coupler [2.41] which enhances the

bandwidth of the circuit, permits two circular polarizations (RHCP and LHCP) at

the same time, and increases the isolation between both channels (TX and RX). The

two patches are separated by one foam layer (thickness=4 mm) in order to get higher

bandwidth (Fig. 2.3.1). In the bottom part (thickness substrate 3=0.254 mm), the

feeding distribution network and connectors are placed. It is necessary to remark

that specications and space constraints make the network design a challenging one,

since the hybrid couplers are especially shaped according to the available space. In

order to introduce the distribution lines, and 1 to 4 dividers, the double stage hy-

brid couplers were miniaturized. This miniaturization reduces the dimensions from

10.90 mm x 14.41 mm to 8.07 mm x 10.67 mm (26 %). Also the connection with the

radiating array has to be carefully dened (in this case by means of via holes). The


(a) Radiating element 3D view. (b) Radiating element layer view.

Figure 2.12: Double stacked patch with 3dB/90o hybrid coupler.

(a) Transmiter. (b) Receiver.

Figure 2.13: Block diagram system.

substrate permittivity is 2.17 in order to get good radiation of the antenna. Despite

the reduction of the Q factor, the substrates and foam thickness are high in order

to enhance the bandwidth (∼ 15 %).

Active feeding network

The active feeding network is presented in Fig. 2.13(a) and Fig. 2.13(b). Each

subsystem will be the responsible for adapting the signal in amplitude, noise or

phase. In both cases, transmitter and receiver, the subsystem in charge for the

beam steering will be the phase shifters, these, can be digital or analogical, but in

both cases TTD (true-time-delay) in order to cover the whole frequency operation

band.

In the receiver system, another lter is added after the antenna in order to

minimize the power transmitted, coupled in the receiver system. The necessary

steering range in elevation goes from 10o to 60o above the horizon. However, the

plane in which the antenna is supported is tilted, thus, the nal steering range is from

40o to 90o. Electromagnetic simulations (in dashed line) and the analytic simulations

(continuous line) for the dierent steering directions are presented in Fig. 2.14. It


can be seen that there is a directivity reduction between the perpendicular steering

direction (90o, or broadside) and the worst case (40o).

Figure 2.14: Radiation pattern of 16x1 array.

The feeding amplitude and phase coecients were calculated in Fig. 2.15 for

reducing the side lobe levels, and obtain the desired steering angles.

The most important decision in the active chain is the election of the phase

shifter. For this high performance communication system the noise gure it is very

important because it is desirable to have high data rates. In order to reduce the

noise oor it is necessary to have a phase shifter with the higher number of bits as

possible because those are directly related to the noise oor. Therefore a MAPS-

010166 of MA-COM with 6 bits and least signicant bit =5.625o, able to operate

from 8 to 12 GHz is choose. With this devide the continuous mean beam exploration

is ensured (there are no cross beam losses) and periodic phase feeding failures are

avoid since each row of the antenna is fed by a dierent phase shifter.

Passive feeding network

The antenna has two dierent distribution networks, the passive one which feed

each all the elements (or columns) inside a row and an active one which feeds every

row. The mask that the radiation pattern must full states that the side lobe level

must be 42 dB below the main beam directivity at ±48o for the azimuth plane.


Figure 2.15: Amplitude and phase of the active network elements.

Therefore, enourmous tapering is design for the feeding distribution network. In

order to make feasible the construction of the transmision lines, the characteristic

impedance of the radiating elements is reduce to Z0 =25Ω.

In Fig. 2.16 the 1 to 4 element distribution network is presented. The unbalanced

power distribution between the output ports in order to obtain the rigorous azimuth

radiation pattern can be seen in the width of lines.

Figure 2.16: 1 to 4 unbalanced network divider.

The results of this circuit are presented in Fig. 2.3.1. The dierent output power

for each of the output ports is shown in Fig. 2.17(a). However, all the dividers

and transmission lines that reach each radiating element must have the exact same

lenght, so all the radiating elements are feed with the same phase Fig. 2.17(b).

In the same way the unbalanced 1 to 3 power divider was design to gather three

of the unbalanced 1 to 4 power divider. This group of power dividers will feed 12

elements, and by means of a balanced 1 to 2 power divider the rest of the elements


are feed with the exact same network but with a transversal symmetry. Finally the

following coecients are obtained 2.3.

(a) Module. (b) Phase.

Figure 2.17: S parameters of 1 to 4 unbalance network divider.

Table 2.3: Amplitude coecients of the passive network.

Element Amplitude Element Amplitude

1 &24 0.299 7 & 18 0.682

2 &23 0.363 8 & 17 0.745

3 &22 0.427 9 & 16 0.809

4 &21 0.491 10 & 15 0.873

5 &20 0.555 11 & 14 0.936

6 &19 0.618 12 & 13 1


(a) Quarter-wavelength transmis-

sion line T-model equivalent.

(b) Construction of a miniaturized

hybrid coupler.

Figure 2.18: 3 dB/90o Hybrid coupler miniaturization.

The available space in the passive feeding network is one problem for this kind of

antenna, because the vertical separation between the rows 0.5λ|7,25GHz is 20.68 mm

and to t the distribution network and the hybrid coupler is a hard issue. For this

reason, a miniaturized hybrid coupler was desing. Basically, the reduction consists

in the substitution of each line in the model of a branch line for its equivalent

quarter-wavelength transmission line of T-model [2.41].

A hybrid coupler was designed, simulated and constructed following these gui-

delines. Final dimensions are smaller than 8 mm x 10 mm (Fig.2.18(b))

Electromagnetic optimizations were carried out to improve the hybrid operation.

These simulations and measurements can be seen in Fig.2.19(a) (Module of S pa-

rameters) and Fig.2.19(b) (phase of S parameters). In Fig. 2.19(b) it can be seen

that the dierence between the phases S21 and S31 is ≈90o±5o.

The nal simulation results of the distribution network for the 1 x 24 elements

array is presented in Fig 2.20. The side lobe level slightly surpasses the mask,

however the average energy surpassed is low enough following the ITU-R S.732

[2.42].



Figure 2.19: Hybrid coupler S parameters.

Figure 2.20: Radiation pattern of azimuth plane.

4x4 sub array construction

In this section a 4x4 element array for satellite communications in X band is

designed, constructed and measured to prove the feasibility of the system above

describe. This antenna is a subarray of a larger system which is compact, narrow

beamwidth and reaches a gain of 16 dBi. It has the capability to steer in elevation

to 45o, 75o, 105o and 135o electronically by means of a butler matrix (section 2.2).


Fig.2.21(b) shows the bottom part of the construction of the 16 elements array. In

this gure, it can be seen the uniform feeding distribution network, the miniaturized

hybrid couplers and the connectors, which will be connected to the steering network.

If the Butler network is connected to left side ports, the antenna radiation will be

LHCP and vice versa. The rest of the ports will be loaded with 50Ω loads.

(a) Top view (b) Bottom view

Figure 2.21: 4x4 array array prototype [2.40].

Fig. 2.22 presents eight input ports of the sixteen elements array. The continuous

line show the left side ports, the dashed line represents the right side ports and the

marked line corresponds to the isolation between two ports together. The isolation

is proper enough with a value under 15 dB for the whole band (SLR < −15 dB) and

the reection coecient for all the ports is under -10 dB (Sii < −10 dB).

In Fig. 2.23(a) and Fig. 2.23(b) the radiation patterns at the center frequency of

each band (7.5 GHz and 8.15 GHz) for every steering angle are shown. The steering

angles −45, −15, +15 and +45 correspond to phase dierence feed α = 135,

α = 45, α = −45 and α = −135 respectively. The continuous line presents

the measured data, while the dashed one shows simulated data, which are in good

agreement.

It can be seen in both gures (Fig. 2.23(a) and Fig. 2.23(b)) a gain reduction

in the main beam far from the broadside direction as it was previously shown in

section. This is reasonable due to the main beam widening. The level dierences

between measurements (continuous line) and simulations (dashed line) are because

of the fact that loss tangent in the dielectric substrate, cables and connectors were


Figure 2.22: S parameters measurements of 4x4 array prototype (8 ports).

(a) LHCP 7.5 GHz. (b) RHCP 8.15 GHz.

Figure 2.23: Steering radiation pattern of 4x4 subarray.

not accurately taken into account.

The integration of all the necessary elements, passive networks and active devices

to full the system requirements in such a small space is a hard task and trade o

must be done. After a large amount of simulations and analytical calculations, this

proposed design full all the requirements and it yields good performance.

2.4. Conclusions

This chapter gives a theoretical and practical overview of the factors that have to

be taken into account in the array exploration design process. This in an important

Conclusions 49

starting point to understand thoroughly the basics of this antennas, to look for the

principal contributions that have been already made, and what it is more important

to identify the elds in which some new contributions can be made. A design of an

steering antenna for a satellite communications in X band is carried out. In this

system the link budget requirements and antenna specications are very demanding

and state of the art technologies and techniques are requided in order to full the

high performance in terms of radiation eciency and reduced available space.

One of the detected problems in printed antennas is the mutual coupling bet-

ween the radiating elements when a phase shift (α=0o) is applied to get the steering

direction. It has been observed that the surface wave modes along the substrate are

enhanced when the radiating elements are not fed in phase. To avoid this problem

the use of cavity patch antennas is commonly utilized, however, this technique in-

creases the weight of the system, the mounting process becomes more dicult and

therefore the cost rises. To solve this problem without increasing the complexity of

the antenna fabrication the use of Electromagnitic Band Gap (EBG) metamaterials

is explored. By placing these structures in between the radiating elements the sur-

face wave modes along the substrate are suppressed, obtaining therefore a similar

eect to cavity back radiating elements.

Another problem of the phased array antennas resides in their distribution fee-

ding networks. There are several kind of phase shifters, and the election of the

devices and the circuit topologies is directly related to the return and insertion

losses and the isolation between radiating elements. Therefore, several designs com-

bining topologies and technologies are proposed and developed in order to obtain

dierent properties in the feeding distribution network. Features such as match or

unmatch ports, maximum isolation between ports, balanced or unbalanced power

distribution and of course phase dierence.

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.

Chapter 3

Mutual coupling reduction using

EBG in steering antennas

The behaviour of radiating elements varies when they are placed surrounded

by other elements. This eect is due to some amount of energy transfered between

them whether one and/or several are transmiting or receiving. The amount depends

basically on the radiation characteristics of each, relative separation between them

and the relative orientation of each. There are many dierent mechanisms that can

cause this interchange of energy. For example, even if both antennas are transmit-

ting, some of the energy radiated from each will be received by the other because of

the non-ideal directional characteristcs of practical antennas. Part of the incident

energy on one or both antennas may be re-scatered in dierent directions allowing

them to behave as secondary transmitters. This interchange of energy is known as

MUTUAL COUPLING, and in many cases it complicates the analysis and design

of an antenna.

3.1. Introduction

In microstrip antennas, not only the radiating elements contribute to the mutual

coupling, but also the feeding networks and feeding probes (if exist). These antennas

are printed in a thin layer of metal (18-35µm) on a substrate by chemical process or

laser etching. The relative permittivity εr and the thickness of the substrate are very

important because they change the size and the properties of the radiating elements.

It is desirable to use thick substrates to increase the operation bandwidth and high

εr to reduce the size of the printed antennas. However, by using this strategy, the

57

58 MUTUAL COUPLING REDUCTION USING EBG IN . . .

surface wave propagation modes are enhanced because of the thick substrate, and

the electromagnetic elds instead of being radiated, are conned in the substrate

due to the high εr which lead to a stronger interaction and therefore, higher mutual

coupling between radiating elements.

This undesired energy transfer characteristic causes several damages in the ra-

diation pattern of the antenna such as directivity drop, power losses and higher

side lobe level (SLL) and crosspolar (XP) component.The most common mutual

coupling reduction techniques are cavity structures [3.1], non uniform feeding distri-

bution [3.2], unequally space distribution [3.3] and Defected Ground Plane (DGP)

[3.4], but lately Electromagnetic Band Gap (EBG) structures are being used. The

introduction of several rows of EBG structures between two printed antennas has

been proved to increase the isolation [3.5].

EBG structures are usually realized by periodic arrangement of dielectric mate-

rials and metallic conductors. In general, they can be categorized into three groups

according to their geometric conguration: three-dimensional transmission lines,

two-dimensional planar surfaces and one-dimensional transmission lines. This work

is focused on the 2-D EBG surfaces, which have the advantages of low prole, light

weight, low fabrication cost, and are widely considered in antenna engineering.

Among others, EBG structures are used as ground planes emulating Articial

Magnetic Conductors (AMC) in a small frequency range and they are also used to

reduce the mutual coupling between elements.

EBG as Articial Magnetic Conductor (AMC) structure has one important

feature: the in-phase reection coecient for plane waves [3.6]. This property

can be used to design low-prole wire antennas [3.7]. The low-prole design

usually refers to the antenna structures whose height is less than one-tenth.

In chapter 12 of [3.8], a comparison between PEC, PMC, and EBG ground

planes is carried out. The conclusions are drawn in table 3.1.

In [3.9], EBG ground plane is used to reduce the eect of multipath by blocking

the propagation of surface waves in a low prole GALILEO antenna. Its

performance is similar to a classical choke ring antenna.

Introduction 59

Table 3.1: Comparison of PEC, PMC, and EBG ground planes for low prole an-

tenna designs.

Ground plane Reection phase Comments

PEC 180 Reverse image.

PMC 0 In-phase reection.

EBG from 180 to −180 with frequency Suitable frequency band.

as Filters, thanks to its frequency selective feature, this EBG structure can

be used as a lter. By introducing three elements as a ground plane for a

microstrip line, in [3.10] isolation higher than 55 dB in the rst frequency

band (2.1 GHz) and 40 dB in the second one (2.45 GHz) are achieved.

as Mutual Coupling barriers: Patch antennas are found to have very strong

mutual coupling due to the severe surface waves on thick and high permitti-

vity substrates. In the literature, it can be found a variety of works which

apply metamaterials for the reduction of this eect. In [3.5], four rows of EBG

mushrooms are inserted between the patch antennas in a εr=10.2 and thick-

ness (h=2 mm) substrate. With this conguration, 8 dB of mutual coupling

reduction is obtained. In [3.11], dierent substrates are combined: radia-

ting elements are suspended over a thick foam layer in order to increase the

bandwith, meanwhile EBG structures are printed in a thin high permitti-

vity substrate for size reduction and surface wave suppression. In [3.12], by

using edge-located vias, the size of mushroom-type EBG is reduced by 20 %.

Among other strategies, in [3.13], a fork shape is used. The area occupied by

the fork-like structure is less than 25% of the mushroom-like structure. Be-

sides, MicroElectroMechanical Systems (MEMS) are used, and recongurable

stop band is obtained. Another studied technique is the use of metal strips.

Basically, the idea is to combine the EBG concept with soft surfaces. A com-

parison of mushroom-type EBG surfaces and corrugated and strip-type soft


surfaces is shown in [3.14]. This stripe-type is used in [3.15] and in [3.16] to

reduce mutual coupling. Finally, dual band planar soft surfaces are developed

in [3.17]: two sizes of strips are mixed in order to get dual forbidden band.

In this chapter, EBG structures as electromagnetic barriers for mutual coupling

reduction are studied. In order to solve the problem for printed radiating elements

which are very close, a new type of EBG is developed. By combining several tech-

niques the size of these structures is reduced meanwhile the barrier characteristics

are preserved.

3.2. EBG theory fundamentals

EBG based on Frequency Selective Surfaces (FSS) [3.18] are one type of meta-

materials with particular electrical properties [3.19]. EBG technique appears as an

application of truncated frequency selective surface (FSS) [3.20]. These structures

consist of an array of metal protrusions on a at metal sheet and can be visualized

as mushrooms protruding from the surface as shown in Fig.3.1. When the period p

is small compared to the wavelength of interest (p << λ), it is possible to analyze

the material as an eective medium, with a surface impedance. These mushrooms

present very high impedance for vertical an horizontal modes at certain frequencies.

The behavior of this structure is similar to a LC circuit, eq. 3.1. Below the reso-

nance frequency, the surface is inductive and supports TM modes meanwhile above

resonance frequency, the surface is capacitive and TE surface waves are supported.

ω0 =1√LC

(3.1)

Nearby ω0, the surface impedance (Zs) is much higher than the impedance of free

space (Z0=120π), as eq. 3.2 depicts. Therefore, no vertical or horizontal propagation

modes are allowed. The high impedance surface also ensures that a plane wave will

be reected without the phase reversal that occurs on a perfect electric conductor

(PEC).

EBG theory fundamentals 61

Figure 3.1: High impedance surface and its model with parallel resonant LC circuit.

The substrate is transparent in order to get better visualization of metallic vias.

(a) EBG parameters. (b) LC model.

Figure 3.2: LC model for the mushroom like EBG structure.

Zs =jωL

1− ω2LC(3.2)

The value of the capacitor is given by the fringing capacitance between neigh-

boring coplanar metal plates. This can be derived using conformal mapping, this

technique establish two-dimensional electrostatic eld distributions. The derivation

starts with a pair of semi-innite plates separated by a gap and then truncates them

with a nite patch size. Finally, the edge capacitance for the the proximity of the

metal plates C, according to [3.20] is given in eq. 3.3:

C =

[wε0(εeff )

π

]cosh−1

(2w

g0

)(3.3)

The value of the inductor is derived from the current loop in Fig. 3.2, consisting


of the vias and metal sheets. For a solenoid current, the magnetic eld can be

calculated using Ampere's law. The equivalent inductor is then computed from the

stored magnetic eld energy and the excitation current. After a simple derivation,

the inductance L is expressed as eq. 3.4 which depends only on the thickness t of

the structure and the permeability of the free space µ0:

L = µ0t (3.4)

In [3.20], Yang et al. propose the analitical solution when the radius of the via

is not r << t eq. 3.5.

L = 2 ∗ 10−7t

[ln

(2t

r

)+ 0,5

(2r

t

)− 0,75

](3.5)

Substituting (3.3) and (3.4) or (3.5), into (3.1) and (3.2), the surface impedance

and resonant frequency can be computed. Other characteristic parameters, such

as the reection phase, can also be derived accordingly. This LC model is easy to

understand, but the static eld approximations limit its accuracy.

Analytically, it can be noticed that the band gap operation is proportional to√L/C. Thus, for a xed separation of the metal plates the value of the capacitance

is set, and the working bandwidth can be improved by increasing the length of the

via and therefore the thickness of the substrate, which is adequate for wideband

antennas. On the other hand, when the thickness of the substrate is xed and the

gap size can not be reduced, C can be increased by using multilayer circuit boards

with overlapping plates, in order to decrease the frequency. In Fig. 3.3, on the left

hand side, traditional mushrooms are shown, meanwhile on the right hand side a

multilayered mushroom structure is presented.

Finally, EGB structures can be seen as bend corrugations, where the length of

the corrugation is the sum of the via and the patch two times, as it is depicted

in Fig. 3.3. The band gap operation of the corrugations starts and nishes when

the length is similar to λ/4 and λ/2 respectively, which yields an octave regarding

operation frequency band [3.21].

Surface wave supression 63

Figure 3.3: Layer and top views of traditional EBG structures (left hand side) and

multilayered F structure (right hand side).

3.3. Surface wave supression

Thanks to the high impedance surface, horizontal or vertical modes are not

allowed in the mushroom structures at certain frequencies. In order to nd the

allowed frequencies for each wave vector, a single unit cell with periodic conditions

is simulated with CST and the Eigenmode solver. The simulation is carried out

by means of a variable sweep which goes across the unit cell (from the x point to

m and p) in the two dimensions x and y. The x-axis of the graphs represents the

dimensions inside the unit cell (x-m-p) in which the boundary conditions are fullled

at certain frequencies for the transmission modes of the structure. In Fig. 3.4(a)

and Fig. 3.4(b) the unit cell is symmetric respect to the via, thus, only three traces

have to be computed. However, for the F and H shape cases in Fig. 3.4(c) and Fig.

3.4(d) respectively, the EBG structure does not have the same dimensions in x and

y. Therefore, ve traces have to be analyzed to describe all the possible propagation

modes for these boundary conditions.

All the studied structures are developed in a low permittivity substrate εr=2.17

and thickness of 1.143 mm. For the traditional mushroom, the size of the original

patch in X band (7.25 - 8.4 GHz) is 3 mm x 3 mm, on the other hand for multilayer

mushrooms 2.8 mm x 2.8 mm size is used. For this last conguration and the rest of

multilayer congurations, two stacked substrates of 0.762 mm and 0.381 mm thick


(a) Original mushroom. (b) Double layer mushroom.

(c) Double layer and edge-location

via mushroom (F shape).

(d) Double layer interdigitated

mushroom (H shape).

Figure 3.4: Unit cell scheme for eigenmode solutions. Dimensions in mm.

are used.

The electric eld is described in terms of an eigenvalue equation, which is solved

numerically. In Fig. 3.5(a), Fig. 3.5(b), Fig. 3.5(c) and Fig. 3.5(d) the mode solu-

tions that satisfy the boundary conditions are shown. The abscissa value represents

the wave number which fulls the requirements at a certain frequency. The lowest

line is the TM mode, the second and third lines are TE modes. A frequency band

gap (dashed lines), in which the surface does not support surface wave propagation

of either polarization, horizontal nor vertical, extends from the top of the TM band

to the point where TE band crosses the light line. The comparison between original

shape and double layer mushrooms yields the result of 2 GHz bandwidth for the

original shape and 1.2 GHz for the multilayer solution.

Surface wave supression 65

(a) Original mushroom. (b) Double layer mushroom.

(c) Double layer and edge-location via mush-

room (F shape).

(d) Double layer interdigitated mushroom (H

shape).

Figure 3.5: Brillouin diagrams of the EBG unit cell.

In this work, the combination of multilayer structure and edge-location via [3.22]

for mushroom size reduction is discussed. In order to keep the frequency working

band (15%), and radiation eciency, the same substrate (εr=2.17 and thickness of

1.143 mm) is used. As long as the inductance depends on the substrate thickness,

which is xed already, the only available parameter is the capacitance C. In order

to increase this parameter, a multilayered structure with edge-location [3.23] via is

presented. The dimensions of the patches are 2.1 mm x 3.6 mm (this value means

30% size reduction). The Brillouin diagram for this conguration is presented in Fig.

3.5(c). In this case, the bandwidth remains its value but signicant size reduction is

noticed. Another dierent attempt to reduce the size of the mushroom is to increase

the capacitance by constructing H shape mushrooms. The main idea is to avoid the


multilayered congurations by using interdigitated elements. However, due to the

construction limits and frequency of operation, this solution is not feasible (Fig.

3.5(d)).

3.4. Mutual coupling reduction

The size of the mushroom patches and the necessary number of periods for mu-

tual coupling reduction is higher than the available space between radiating elements

for low permittivity substrates. To study this space constraint, a simulation scheme

for S21 analysis is proposed in Fig. 3.6. In this scheme, a 50 Ω transmission line is

placed over an EBG ground plane. In this simulation, dierent topologies (original,

double or F shape), size of the mushroom w, gap size g and number of elements n

are tested in order to obtain the surface wave suppression behavior.

Figure 3.6: Simulation scheme for transmission parameters S21 analysis.

In order to keep the study in the same frequency operation band, the thickness

of the substrate is the same that in the previous section (t=1.143 mm) and the

gap size is x to (g=0.2 mm) to reduce the space. In Fig. 3.7(a), a comparison

between the mushroom size w and the working operation band is shown. It can be

noticed that an increase of operation bandwidth is obtained for higher frequencies

Mutual coupling reduction 67

when L is xed and C is reduced because of w reduction, as it was forementioned

in section 3.2. In the same way, a study of the required number of elements n is

carried out and the results are drawn in Fig. 3.7(b). Now, with the mushroom size

x to w=3 mm, dierent number of periods are placed. In the original study seven

elements are used, when this number is reduced, it can be observed how the isolation

characteristic decrease.

(a) Isolation S21 for dierent EBG sizes (w). (b) Isolation S21 for dierent number of EBG

periods (n).

Figure 3.7: Parametric study of isolation characteristics (S21 for original shape mush-

rooms) when size and number of elements are swept.

Other simulations are done maintaining the xed values of n=4 and w=3 mm.

In Fig. 3.8(a) a swept of the substrate thickness is shown. In this case of study

the values for g and dvia are 0.2 mm and 0.4 mm respectively. It can be noticed

that the behavior of resonance frequency tends to move to a higher values when the

substrate is thinner. The reason for that behavior is the dependance of L with the

thickness of the substrate (as it is explained in eq. 3.4). With values of t=1.143

mm and g=0.2 mm a variation of the via diameter from 0.2 mm to 1.2 mm of the

mushroom structure is carried out in Fig. 3.8(b). With this parameter variation,

the eq. 3.5 and the dependence of the resonance frequency with the via diameter

of the structure can be proved. By examining this gure, it can be noticed that for

dvia=0.2 mm the bandwidth operation is roughly 1.5 GHz meanwhile for dvia=1.2

mm, the bandwidth operation is increased to ≈2 GHz. At the end of section 3.2 it

is stated that the bandwidth operation is proportional to√L/C. However, here,


through these simulations L is changed and this property regarding the bandwidth

is proved. Eventually, in Fig. 3.8(c), the variation response of S21 transmission

parameter is shown. Just like in the other gures, a frequency operation change can

be observed. This change is due to the value of C in the mushroom structure, which

is proportional to the gap between the parallel plates. Therefore, when the gap size

increases, C decreases, and the band gap appears at higher frequencies.

(a) Isolation S21 for dierent substrate thick-

ness (t).

(b) Isolation S21 for dierent via diameter

(dvia).

(c) Isolation S21 for dierent gap size (g).

Figure 3.8: Parametric study of isolation characteristics (S21 for original shape mush-

rooms) when substrate thickness, via diameter and gap size are sweept.

All the substrates have permittivity of εr=2.17. However four dierent thickness

values are used. The TLs impedance is 50 Ω and those TLs are printed in a 0.254

mm thick substrate. Mushrooms in single layer case are printed in a 1.143 mm thick

substrate, meanwhile double layer case are printed in 0.762 mm (bottom layer) and


0.381 mm thick (upper layer) substrates. Therefore, total thickness value maintains

its value as it can be seen in Fig. 3.12(a).

F shape solution is deeply developed and a parametric study regarding the sizes

of the mushrooms is carried out. In Fig. 3.9 the deviation of the isolation response

is shown. When the size is very small, the band gap tends to appear in higher

frequencies, and it can be seen how the sum of both dimensions, W and L must

remain under certain values.

Figure 3.9: Parametric study of width W and length L for F shape mushrooms.

Transmission S parameters (S21).

Figure 3.10: Samples of single and multilayered EBG mushrooms with dierent

shapes and number of elements.

In order to validate the whole process, prototypes of four and seven rows are


built. Six prototype Transmission Lines (TL) with EBG ground plane are shown in

Fig. 3.10. On the left hand side, the two circuits are single layered, the two circuits

in the middle are double layered, and the last two circuits on the right hand side

combine double layer with edge-location via (F-shape).

(a) Original shape mushroom. (b) Double layer mushroom.

(c) F shape mushroom.

Figure 3.11: Comparison between measurements and simulations of transmission

parameters S21 for dierent types of EBG mushrooms.

In Fig. 3.11(a), Fig. 3.11(b) and Fig. 3.11(c), comparisons between the simula-

ted structures and the measured prototypes are shown. Ensembling diculties due

to the small size of the circuits (size circuit approximately 1 cm) add some dieren-

ces between simulations and measurements. The fabrication process in PTFE, such

as metal vias and stack up process is a hard task. However, the overall behavior

of a LC lter in a certain frequency band is observed. It can be noticed, compa-

ring the gures, that traditional mushrooms have larger frequency operation band


than F shape mushrooms. However, F shape fulls the bandwidth and isolation

requirements.

Finally, the trade o solution between isolation and available space is carried

out, being the necessary number of periods for surface wave suppression: n=4. In

Fig. 3.12(a) and Fig. 3.12(b), the chosen topology is shown. With nal dimensions

of 2.1 x 3.6 mm, four elements, the double layered structure and edge-location via

fulls the requirements of available space (10.2 mm) between two printed antennas

separated 0.6λ0 in a εr=2.17 and 1.143 mm thick substrate. A bandwidth operation

in X band, from 7.1 GHz to 8.2 GHz (15%) and 10 dB of isolation are obtained.

(a) Schematic (b) Prototype

Figure 3.12: Multilayered mushroom with rectangular shape, 4 elements and edge-

located via (F-shape).

In order to prove the eectiveness of these electromagnetic barriers, four rows of

double layered edge-located via EBG mushrooms are introduced between two round

patches with double circular polarization. The radiating elements are integrated

in the same substrate and they are circular polarized. The elements are fed by a

90/3dB branch line coupler, in order to get the double circular polarization.

In Fig. 3.13(a), |E| eld simulation for the two patches is shown for Left Handed

Circular Polarization (LHCP). In Fig. 3.13(b), it can be seen graphically, how |E|

eld value decays quicker when using double layer edge-location via EBG structures.


(a) Without EBG structures (b) With EBG structures

Figure 3.13: |E| eld simulation of two round patches with dual circular polarization.

Thus, two circular patches fed by the 90/3dB branch line coupler with and

without four rows of EBG F-shape mushroom are built. These printed antennas, se-

parated 0.6λ0, are mounted with double stacked patch with permitivity of εr=2.17

and a foam layer between them of 4 mm in order to cover the whole bandwidth

(20%). The thicknesses of the substrates are 1.143 mm for the bottom patch and

0.254 mm for the upper patch and the feeding network. Fig. 3.14 shows the cons-

truction process of both test antennas.

Figure 3.14: 2x1 test array of circular patch antennas with and without EBG F

shape mushrooms.

The radius for the via fed and the coupled fed patches are 7.31 mm and 7.17 mm


Figure 3.15: Specic dimensions of 2x1 test array.

respectively as it can be seen in Fig. 3.15. With these prototypes, a comparison

between simulations and measurements of S21 parameters for a 2x1 array antenna is

presented in Fig. 3.16(b). LHCP ports are excited and RHCP ports are loaded with

50 Ω loads. Similar behavior between simulations and measurements is achieved.

There is a mutual coupling reduction between the two patch antennas of approxi-

mately 5 dB and an improvement of reection coecient Sii of ∼3 dB in most of the

operation band. This coupling reduction remains for the dierent phase feedings of

the elements.

(a) Matching S11 parameters (b) Isolation S21 parameters

Figure 3.16: Comparison of measurements and simulated S parameters for 2x1 array.

In Fig. 3.17, the radiation patterns for φ cuts are presented. It can be noticed

that for the 90 pattern, the beamwidth is reduced for the EBG case. EBG structures

surrounding microstrip antennas tend to make narrower the radiation pattern of


the radiating element. This beamwidth reduction increases the directivity of the

antenna, since the eect of the EGB barriers is similar to the eect of cavities.

On the other hand, due to the structures placed on the edge of the antenna, a ∼5

dB of back lobe reduction is obtained. The radiation pattern for φ=90 presents

asymmetry due to the measurement setup in the anechoic chamber.

Figure 3.17: Radiation patterns of 2x1 test patch antenna array with and without

EBG structures.

The maximum separation between elements for avoiding grating lobes in steering

antennas is a double problem. First, by placing the radiating elements very close,

|E| eld interaction between radiating elements is stronger. In the second place, the

eective area of the antenna is reduced. For multimedia applications, broadband

capability is required, and thick substrate is used. Thus, the surface wave propaga-

tion modes are enhanced and the mutual coupling between elements grows. For that

reason, the steerable antenna prototype from [3.24] has been built to measure the

radiation patterns and to verify the features and performances when mushroom F

shape structures are introduced between elements in order to suppress surface wave

propagation and, consequently, to reduce mutual coupling.

Applications 75

3.5. Applications

In this particular electronic steering antenna, the separation between radiating

elements must not be higher than 0.6λ0 in order to avoid grating lobes in the visible

region when the beam is not pointed to broadside, as it was explained in Chapter

2. In this section, a module of a wideband planar array antenna with dual circular

polarization (LHCP and RHCP) and electrical elevation steering for satellite com-

munication systems is provided (Table 3.2). In order to reduce the mutual coupling

between radiating elements, EBG structures are introduced.

Table 3.2: Antenna specications.

Parameter Value Units

Frequency RX 7.25 − 7.75 GHz.

Frequency TX 7.9 − 8.4 GHz.

Polarization RX LHCP* * Interchangeable.

Polarization TX RHCP*

Gain ∼16 dBi.

Elevation steering ±10 and ±40 Degrees

Dimensions <0.2 m.

Antenna Eciency >60 %.

Axial Ratio <3 dB.

CP/XP >25 dB.

Matching Sii >13 dB.

Isolation Sij >15 dB.

For the antenna, it is important to have good radiation eciency, therefore, a

εr=2.17 substrate is used. The available space between radiating elements under

this circumstance is not enough to place original mushroom shape EBGs. Therefore,


multilayer and edge location via techniques are combined to reduce the size of the

mushrooms by 30% in the vertical direction where the elements get the steering

elevation.

In Fig. 3.18, the printed array layers are presented. In the bottom layer (0.254

mm thick) the feeding distribution network and 90/3dB hybrid couplers are printed.

In the second layer (0.762 mm thick) the rst mushrooms and the vias for the patches

are placed. On top of it, the third layer (0.381 mm thick) contains the via fed circular

patches and the second mushrooms, which are stacked with the rst mushrooms in

the previous layer. On the top layer (0.254mm thick) parasitic fed patches are

printed. Finally, between via feed patches and parasitic fed patches a 4 mm foam

layer is placed in order to enhance the bandwidth [3.25]. The distance between the

radiating elements is 0.6λ0 for vertical axis in order to avoid grating lobes when

electronic steering is used. However for horizontal axis 0.85λ0 is used in order to

reduce the number of elements and to get the required directivity.

Figure 3.18: Layer view of the 4x4 array with EBG structures construction.

In Fig. 3.19(a) and Fig. 3.19(b) the measurements for eight input ports of

the sixteen elements array are presented. The continuous lines show the left side

ports, meanwhile the dashed lines represent the right side ports. The dotted lines

Applications 77

correspond to the isolation between two closer ports. The measured isolation is

adequate enough with a value below 15 dB for the whole band and the measured

reection coecient for all the ports is below 13 dB, thanks to the EBG structures,

otherwise these values do not reach 10 dB. The rest of the ports (ports 3 and 4) are

not shown but they have similar results.

(a) Port 1: right and left inputs (b) Port 2: right and left inputs

Figure 3.19: S parameters measurements of 4x4 array with EBG structures.

Figure 3.20: 4x4 array and Butler matrix network connection for LHCP congura-

tion.

In order to get the beam steering in elevation, the passive Butler matrix network,

(Chapter 2) is connected to the 4x4 subarray with EBG barriers as it can be seen

in Fig. 3.20. If the Butler network is connected to left side ports the antenna

radiation will be LHCP and vice versa, therefore, this system obtains double circular


polarization. The rest of the ports are loaded with 50Ω loads. Butler Matrix network

yields -45, + 135, -135 and +45 phase shift between output ports when exciting

ports 1 to 4 respectively.

These phase shifts between elements (α= ±45 and ±135) are used to calculate

the pointing directions of the antenna. Those directions are calculated by applying

exploration beam equation. The steering directions for the antenna: +10, -40,

+40 and -10 are obtained.

Figure 3.21: 4x4 steering array radiation pattern for RHCP, at the center frequency

(7.825GHz).

In Fig. 3.21, the steering vertical plane radiation patterns for RHCP at the

center frequency (7.825 GHz) and dierent pointing directions are presented. Good

agreement between simulations (dashed lines) and measurements (continuous lines)

is obtained. The gain dierences between simulations and measurements are due

to the connectors and cables used to connect the Butler matrix network with the

antenna, which are not introduced in the simulations. CP/XP ratio is better than

15 dB. Due to the uniform distribution, Side Lobe Level (SLL) is under 12 dB for

±10 and under 7 dB for ±40 beams.

Finally, in Fig. 3.22(a) and Fig. 3.22(b), the axial ratio for the dierent beams

and dierent steering angles at the end and the beginning of the frequency band are

shown. In this case, dierent frequencies, polarizations and steering directions are

Applications 79

shown in order to prove the good features of the antenna.

(a) +10 LHCP and -40 RHCP beams (b) -10 LHCP and +40 RHCP beams

Figure 3.22: 4x4 steering array axial ratio for frequencies 7.25 GHz and 8.4 GHz,

RH and LH circular polarizations over the scanning angles.

As it was expected, the introduction of EBG structures between the radiating

elements has not signicant inuence in the circular polarization of the antenna,

and only slightly dierences are appreciated. Finally, in Fig. 3.23, the axial ratio

of the antenna is shown, related to the frequency. The purity of the polarization is

better than 3 dB over the whole frequency band [3.26].

Figure 3.23: 4x4 steering array axial ratio for RH and LH circular polarizations and

dierent pointing directions over the working frequency.


3.6. Conclusions

In the beginning of this chapter, the basic metamaterial properties are overvie-

wed. Nevertheless, the main objective is to summarize the EBG structures which

are one type of metamaterials based on FSS. By understanding its electromagne-

tic behavior, a design method is proposed for calculating the size of the structures

analytically. These structures can block the surface waves when they are introduced

in between of two printed antennas and therefore its properties as electromagnetic

barriers are thoroughly described. In the rst place, the allowed propagating modes

are presented by means of Brillouin diagrams. Secondly, a electromagnetic simula-

tion scheme is built and parameterized in CST. With this scenario, the inuence of

the dierent parameters such as, gap g, patch size w, via diameter dvia or substrate

thickness t is analyzed.

So far, the solution in order to introduce the EBG structures was to increase

the substrate permittivity which leads to a size reduction of the printed radiating

elements, however, this strategy reduces the radiation eciency and enhance the

surface wave propagation modes.

In order to reduce the mutual coupling between radiating elements of a steering

array, using low permittivity substrates and therefore maintaining good radiation

eciency a new smaller mushroom shape is proposed. This structure combines

multilayer substrate and via edge location reducing the eective size by 30%.

Finally the new EBG structures are inserted in a 4x4 steering array showing how

these electronic barriers reduce the mutual coupling between elements of dierent

rows and increase the performance of the antenna in terms of directivity and back

lobe reduction.

References

[3.1] R. Mailloux. On the use of metallized cavities in printed slot arrays with dielec-

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[3.2] C. A. Balanis. Antenna Theory, analysis and design. Wiley india, third edition

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[3.3] B.P. Kumar and G.R. Branner. Design of unequally spaced arrays for per-

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[3.4] S. Mohsen, M. Alireza, T. Ahad, and H. Teimur. Mutual coupling reduction

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[3.6] J.M. Fernández-González. Application of metamaterial structures in the design,

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[3.7] F. Yang and Y. Rahmat-Samii. Reection phase characterizations of the EBG

ground plane for low prole wire antenna applications. Antennas and Propaga-

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[3.8] N. Engheta and R. Ziolkowski. Metamaterials, physics and engineering explo-

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[3.11] E. Rajo-Iglesias, O. Quevedo-Teruel, and L. Inclán-Sánchez. Mutual coupling

reduction in patch antenna arrays by using a planar EBG structure and a

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[3.12] E. Rajo-Iglesias, L. Inclán-Sanchez, J.-L. Vázquez-Roy, and E. García-Muñoz.

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[3.14] E. Rajo-Iglesias, M. Caiazzo, L. Inclán-Sanchez, and P.-S. Kildal. Comparison

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IET, 3(5):742748, Aug. 2009.

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Chapter 4

Phase shifting distribution networks

for switchable beam antennas

In electrical engineering, a switch is an electrical component that can break

an electrical circuit, interrupting the current or diverting it from one conductor to

another. A Single Pole Single Throw (SPST) can be used to enable or disable specic

parts of the circuit meanwhile a Single Pole Double Throw (SPDT) is used to select

one of the two posible paths. In active steering antennas, they can be used to switch

between to dierent distribution networks, feeding points of the radiating elements

to change their polarization or phase shifters.

4.1. Fundamentals of phase shifters

The phase shifter is a two-port device used to produce either a xed or con-

trollable phase shift to the input signal. Phase shifters are used in a variety of

communication and radar systems. They are essential components in antenna sys-

tems and phased arrays, where the radiating elements must be fed with appropiate

phases so as to create the required radiated beam. In general, xed phase shifters

can potentially become variable phase shifters by replacing some xed reactances

with electronically tunable ones.

In the ideal case, a phase shifter can be dened as a two-port device perfectly

matched (S11=S22=0) and without loss (thus |S21|=|S12|), where the output signal

undergoes a prescribed phase shift ∆φ with respect to a given reference. The phase

shift of the output signal is usually evaluated with reference to the phase shift of

85

86 PHASE SHIFTING DISTRIBUTION NETWORKS FOR . . .

a line section of equal length as the device. Being the lenght of such a line section

and β its phase constant; the phase shift is thus evaluated as:

∆φ = θ − βl (4.1)

Depending on the frequency behavior of ∆φ(f), there are two fundamental types

of phase shifters:

The true− phase phase shifter, whose ∆φ(f) is constant with the frequency.

The true − time − delay (TTD) phase shifter, whose ∆φ(f) varies linearly

with frequency, so that the device behaves as a delay line.

In phased array antennas, it is very important to use TTD, because the elements

are placed physically at a certain distance, but the eective distance changes with

the frequency. The delay line behavior of TTD adjust the phase dierence in each

frequency which is very appropiate. The work presented in this thesis is focused on

TTD phase shifters.

Phase shifters can be broadly classied as mechanical or electronic devices, de-

pending whether the phase control is achieved through mechanical or electronic

tuning. Depending on the type of operation, they can be categorized as analog or

digital, having reciprocal or nonreciprocal characteristics. Other classications can

be in terms of the type of employed transmission structure to realize the phase shif-

ter (waveguide, planar transmission line, dielectric guide, etc.) and the technology

adopted for fabrication (semiconductors, monolithic or ferrites).

An important sector where this technology can be used is automotive radars.

These systems integrated in cars can be widely used with a relative low cost design.

There are two frequency bands reserved for automotive applications, at 24 GHz

a Butler matrix network is modied with CMOS chips in order to get continuous

scanning ±90o [4.1]. In [4.2] by changing the phase of 18 power ampliers the

steering beam is obtained with a directivity of 12 dB and 5o beamwidth. 60 GHz

is another working frequency band for automotive applications in [4.3] a 4 elements

array with ±25o steering angles is shown.

Fundamentals of phase shifters 87

The use of PIN diodes in steering antennas is widely known. Examples of the

use of these devices can be found in [4.4] where PIN diodes are use to change

the radiation pattern of a Yagui-Uda and a planar inverted F antenna. In [4.5]

the diodes are applied to a grid antenna which obtains a ±10o and ±20o steering

angles in elevation and azimuth respectively. This antenna has 2 dB insertion loss

at 3 GHz. Finally, in [4.6] the purpose of the work is focused on the design low

series resistivity PIN diodes, the reason for that is to reduce the insertion loss in a

reectarray compound of dipoles circularly placed which get circular polarization at

18.2 GHz.

The good linearity of the MEMS devices allows to build switches with an ope-

ration band from 0.1 to 120 GHz. In [4.7] two phase shifters, the rst one with

two bits and 0.6 dB of insertion loss and the second one with four bits and 1.2 dB

insertion loss at 10 GHz. Another example in [4.8] shows a transmission line with

the ability to change the propagation speed, by changing the line impedance with

the capacitance variation. These concepts have been applied to electronic steering

antennas. In [4.9] a Ka band antenna constructed with 6 layers LTCC technology is

shown. This antenna has a 8 dBi gain and ±15o steerable beam. Meanwhile in [4.10]

the antenna is able to steer the beam in both directions, (azimuth and elevation)

and it works in V band (40-75GHz). These devices can be used also in reectarrays

such as [4.11] where the device acts as a periodic structure, it obtains a phase shift

of 150o, and it has between 0.4-1.5 dB insertion losses at 2 GHz. In [4.12] a patch

antenna is coupled fed by an slot, by changing the slot dimensions with MEMS the

phase shift of the radiating elements can be changed. High performance antennas

for radar applications are developed with MEMS devices too [4.13]. This antenna

has a scan range of ±60o, eective area of 0.4 m2 and it works at X band.

In [4.14] a beam steering traveling wave antenna is designed. It is compound

by interconnected patch antennas with transmission lines where varactor diodes are

placed. By changing the capacitance of the diodes, the eective distance between

the radiating elements change and consequently the phase feeding law and the stee-

ring direction. Similarly, in [4.15] the eective lenght of a meander line antenna at

X band is electronically controlled with varactor diodes. The 60 GHz band is used


for wireless communication, many private networks and indoor multi-channel trans-

mission are being developed. For this purpose, a ±7o steering antenna composed of

24 radiating elements is proposed in [4.16]. This antenna is fed by phase shifters

based on transmission lines nished with varactor diodes.

The problem of ferrite based phase shifters is their poor calibration accuracy.

Hysteresis can cause changes of 30o in the phase shift prevailing at a given applied

magnetic eld. In some cases, thermal drifts may also alter the obtained phase shift.

This is an important issue that can be handle with a servo system [4.17]. Another

important issue is to build these devices at very high frequencies due to the space

constraints. However, in [4.18], ferrite based phase shifters are designed at 35 and

94 GHz. In order to avoid the space and weight problems, a low cost microstrip

line based ferrite phase shifter is proposed in [4.19]. The design is based on three

microstrip lines arranged and fed with phase dierences so as to produce circular

polarization in the ferrite region. This obtains a phase shift of 360o in less than an

eective wavelength yielding an important size and weight reduction.

4.2. Topologies

Depending on the circuit topology, the device can be considered as transmission

or reection phase shifters, depending on where the phase shift is produced. In

transmission, the signal phase shift is through the device, while the reection one

exploits the phase shift produced by the reection from a reactive load. Below, the

well known topologies with its advantages and drawbacks are thoroughly discussed.

Series-shunt

There are two fundamental methods of connecting switches to a transmission

line to provide a switching function: in series with the transmission line so that RF

power is conducted when the switch is forward biased (Fig. 4.1(a)) and reected

when reverse biased; or in shunt with the transmission line so that the RF power

is conducted when the switch is reverse biased and reected when forward biased

Topologies 89

(Fig. 4.1(b)). A simple reective SPST switch can be designed utilizing one or more

switches in either conguration as shown in Fig. 4.1. The combination of several

devices will increase the isolation regarding the diversion of the current, however it

will introduce higher insertion losses.

(a) Series switch. (b) Shunt switch.

Figure 4.1: Single switches.

Compound

A multi-throw microwave switch essentially consists of combination of SPST

switches connected to a common junction and biased so that each switch port can

be enabled individually. The common junction of the switch must be designed to

minimize the resistive and reactive loading presented by the OFF ports in order to

obtain low insertion loss and VSWR for the ON port. By using series mounted PIN

diodes connected to the common junction, a path is selected by forward biasing its

series diode and simultaneously reverse biasing all the other diodes. This provides

the desired low-loss path for the ON port with a minimum of loading from the OFF

ports. The two most common compound switch conguration are Series-Shunt or

TEE designs. The schematic diagrams for both switches are shown in Fig. 4.2.

(a) Series-Shunt SPST switch. (b) TEE SPST switch.

Figure 4.2: Compound switches.


Table 4.1: Summary of formulas for SPST Switches.

Type Isolation Insertion loss

Series 1 +(XC

2Z0

)2 (1 + RS

2Z0

)2Shunt

(1 + Z0

2RS

)21 +

(Z0

2XC

)2Series-Shunt

(1 + Z0

2RS

)2+(XC

2Z0

)2 (1 + Z0

RS

)2 (1 + RS

2Z0

)2+(Z0+RS

2XC

)2TEE

[1 +

(XC

Z0

)2] [(1 + Z0

2RS

)2+(XC

2RS

)2] [(1 + RS

Z0

)2] [1 +

(Z0+RS

XC

)2]

A summary of formulas used for calculating insertion loss and isolation for com-

pound and simple switches is given in table 4.1.

Tuned

This method utilizes shunt mounted PIN diodes located a quarter wavelength

from the junction. The diode(s) of the selected ON port is reverse biased while the

OFF ports are forward biased to create a short circuit across the transmission line.

As a result of the quarter wavelength spacing, the short circuits are transformed

to open circuits at the junction (Fig. 4.3). By proper choice of transmission line

impedances and minimization of stray reactance it is possible to construct a switch

of this type with low insertion loss and VSWR over a three to one bandwidth.

(a) Tuned series switch. (b) Tuned shunt switch.

Figure 4.3: Tuned λ/4 switches.

Topologies 91

Reective

It is often desirable to have a switch that presents a low VSWR in its OFF

position as well as in its ON state in order to maintain desired system performance.

Reective or also known as absorptive switch incorporate 50Ω terminations in each

of the output ports. Fig. 4.4 shows the two possible congurations. The operation of

this switch is as follows: The power incident on port A divides equally between ports

B and C, port D is isolated. The mismatch produced by the switch in series (Fig.

4.4(a)) or parallel (Fig. 4.4(b)) with the load resistance at ports B and C reects

the power. The reected power exits through port D isolating port A. Therefore, A

appears matched to the input signal.

Both of the above types of hybrid switches oer good features. The series con-

nected diode conguration is, however, recommended for attenuators used primarily

at high attenuation levels while the shunt mounted diode conguration is better for

low attenuation ranges.

(a) series connection. (b) shunt connection.

Figure 4.4: Quadrature matched hybrid switches.

Switched transmission lines

A basic example of a switched line phase shifter circuit is shown in Fig. 4.5.

In this design, two SPDT switches employing PIN diodes are used to change the

electrical length of transmission line by some length ∆φ. The phase shift obtained

from this circuit varies with frequency and is a direct function of this dierential

line length as ∆φ = 2π∆l/λ. The switched line phase shifter is inherently a broad


band circuit producing true time delay, with the actual phase shift dependent only

on ∆l.

Figure 4.5: Switched line phase shifter.

There are some other techniques based on the reactive part variation. By chan-

ging the biasing of the switch device (i.e PIN diode) the XC changes and therefore

there is phase shift produced along the transmision line depending on the biasing.

This topology is called loaded line phase shifter and it main advantages are high

power handling and continuous phase shift. However the biasing is very sensitive

and the insertion loss are higher. This topology is not further developed because its

phase shift is dependant with the frequency. It is not a TTD and therefore is not

adequate for steering array antennas.

4.3. Switchable beam antenna

So far the topic of the thesis has been focussed on steerable systems, the descrip-

tion of the system requirements and antenna features were shown and the problems

about the radiating elements and their mutual coupling has been addressed, howe-

ver, none of the phase distribution networks have been presented. In this section

the implementation of several feeding networks, based on phase shifters or switches,

able to achieve the adequate phase distribution are presented.

Standard users can not aord high accuracy scanning antennas in daily used

systems such as satellite television, WIFI connection or automotive RADAR and

therefore another dierent techniques not based in active phased arrays antennas

must be explored. By means of digital signal processing the SNR can be enhanced

Switchable beam antenna 93

when using several radiation patterns such as multibeam systems. These systems do

not require narrow step scanning angles, and consequently the distribution networks

are cheaper due to the lack of phase shifters with large number of bits. However,

the radiation pattern must be well dened, the insertion loss must be equal for all

the beams, and the isolation between dierent networks or signal paths must be as

large as possible to avoid cross talk interferences.

Switchable beam antennas can be used in RADAR systems to avoid ambiguity

when two objets are at the same distance but in two dierent azimuth or elevation

positions in the same temporal window [4.20, 4.21]. However, the following issues

must be taken into account carefully:

Crossing beam losses. When using an specic beam it is desirable to transmit

or receive the same amount power regardless θ or φ angles (Fig. 4.6(b)). The

cross beam losses could yield a position or velocity miscalculation (Fig. 4.6(a)).

(a) Received power as a function of the

incoming signal direction.

(b) Equal received power for two dierent

incoming signals.

Figure 4.6: Uniform radiation pattern beam shape.

Dierent insertion loss for each feeding network. The construction of the

distribution networks can contain dierent components in each of the signal

paths yielding dierent insertion loss for each beam and therefore a reduction

in its gain (Fig. 4.7).


(a) Desired Scenario. (b) Dierent beam insertion loss scenario.

Figure 4.7: Multibeam radiation pattern unbalance description.

Isolation between networks. Eventhought a beam is selected, the rest of the

beams are presented at the same time. The gain level of the undesired beams is

dened by the isolation between the networks, therefore the lower the leakage

is in the distribution network switches, the better performance the system has

(Fig. 4.8(a)).

(a) Gain level of the isolated beams. (b) Worst case scenario.

Figure 4.8: Multibeam radiation pattern isolation description.

In Fig. 4.8(b) the worst case is presented were undesired clutter appears in

the isolated beams and the desired signal is weakly received because of higher

insertion losses. To overcome this problem, the insertion losses in the distri-

bution network must be equal for every path and clutter reduction techniques


based on digital correlation signal must be applied.

4.3.1. Symmetric reective phase shifter

As a key component for a distribution network, a 120o reective phase shifter at

Ka band (24 GHz) is designed and constructed (Fig. 4.9). This device is construc-

ted with microstrip technology on a ROGERS 4350B substrate,ε=3.66, 0.25 mm

thickness and tested with two dierent PIN diodes MA912 and MA200, to nd the

best response under the same biasing conditions (IF=100 mA and VR=0V). Deco-

upling capacitors and lter structures are place to isolate RF and DC biasing and

the connectors used are rosenberger mini SMP able to operate up to 40 GHz.

Figure 4.9: Reective phase shifter construction.

The operation of this circuit is similar to a reective switch (section 4.2) but with

a transmission line which introduce a phase shift. The phase dierence is obtained

when the PIN diode is reverse or forward biased. The following equations show the

behaviour when the PIN diodes are reverse bias.


Table 4.2: Reective phase shifter [S] matrix.

ON OFF 0 ejπ/2

ejπ/2 0

0 ej(π/2+2φ)

ej(π/2+2φ) 0

[S]Refl =−1√

2

0 j 1 0

j 0 0 1

1 0 0 j

0 1 j 0

· ejφ · ejφ · −1√

2

0 j 1 0

j 0 0 1

1 0 0 j

0 1 j 0

(4.2)

S21 · ejφ · ejφ · S42 + S31 · ejφ · ejφ · S43 =

−1√2j · ejφ · ejφ−1√

21 +−1√

21 · ejφ · ejφ−1√

2j =

j

2· ej2φ +

j

2· ej2φ = j · ej2φ = ej(π/2+2φ) (4.3)

And the case for the PIN diodes forward biasing is:

[S]Refl =−1√

2

0 j 1 0

j 0 0 1

1 0 0 j

0 1 j 0

·

0 j 1 0

j 0 0 1

1 0 0 j

0 1 j 0

(4.4)

S21 · S42 + S31 · S43 =

−1√2j−1√

21 +−1√

21−1√

2j =

j

2+j

2= j = ejπ/2 (4.5)

Therefore, the [S] matrix of the reective phase shifter for the OFF and ON case

are:

The amplitude measurements (Fig. 4.10(a)) show a frequency shift (22.5 GHz

instead 24 GHz) in the crossing point between the OFF and the ON state. The


insertion loss are 1.5 ±0.5dB for the whole operation band (BW=1 GHz, 4%) and

the phase shift between the two states (Fig. 4.10(b)) is 120±10.


Figure 4.10: Reective phase shifter measurements.

This frequency shift is due to the inaccurate PIN diode model, which series

resistance and total capacitance for the ON and OFF states, depend on the bias

current (IF ) and voltage (VR), and physical dimensions. The non-linear model obey

the following equations:

Q = IF τ (4.6)

RS =W 2

(µp + µn)Q(4.7)

Where τ is the carrier lifetime,W is the Intrinsic region width, and µn and µp are

the electron and hole mobility respectively. In Fig. 4.11, the insertion loss variation

due to the PIN diode parameters model is presented. After a thorough simulation,

by comparing the simulation expected from Fig. 4.10 with the measurements, the

diode parameters for MA200 are found and accurate model is parametrized for the

next designs. CT=40pF, L=0.1nH, Rp=1000Ω and Rs=5Ω.

To overcome the problem that the dierent insertion loss yield on each beam

gain of the radiation pattern a series-shunt switch is introduced in the outputs (B

and C) of the reective phase shifter (Fig. 4.12(b)).


(a) ON state series resistance varia-

tion.

(b) OFF state total capacitance va-

riation.

Figure 4.11: Insertion loss variation due to the PIN diode parameters model.

(a) Reective switch. (b) Reective series-shunt switch.

Figure 4.12: Reective congurations.

With this circuitry topology, the overall insertion loss are slightly increased but

the dierence between ON and OFF states are signicantly reduced while maintai-

ning the desired frequency shift. To symmetrize both paths regarding insertion loss

is very important because this phase shifter is placed in the distribution network

several times and the current path to each radiating element must remain constant

at all times [4.22].

4.3.2. Matched SPDT switch

To feed an array antenna with several distribution networks is a well known

technique in order to obtain multiple radiation patterns. However, it is very impor-

tant to ensure an adequate isolation between the dierent distribution networks and


(a) Amplitude. (b) Phase.

Figure 4.13: Series-shunt reective phase shifter results.

reduce the leakage as much as possible. When using a Single Pole Double Through

switch for diverting the current from one input to one of the two possible outputs

it is very important to take good care of the technology utilized and the chosen

circuitry topology.

For a two beam array antenna, which is fed by means of two feeding networks a

SPDT based on PIN diodes is designed and built. The diodes utilized are MA 200,

which are well characterized for the biasing conditions at the operation frequency

(24 GHz). To choose the better topology for the system, the isolation, leakage and

insertion loss where study for the cases of single and double SPDT switch.

(a) Single. (b) Double.

Figure 4.14: Single and Double PIN diode based SPDT switch.

In Fig. 4.14, two schematics for single and double PIN diode based SPDT switch

are shown. The operation of both circuits is controlled biasing one of the output with

forward current (IF ) meanwhile the other output is reversed biased (VR). The rst

diode is placed λ/2 away from the junction, in this way the diode high impedance

produced in the reversed bias output is translated to the junction as a open circuit.

The transmission lines where the diodes are placed are Z0 = 50Ω. In contrast to a


wilkinson divider, at any time one of the branches is an open circuit and therefore

the output impedance do not change. This technique enhance the isolation between

the outputs and solve the possible leakage of placing diodes too close.

Due to the limitations of the biasing the equivalent series resistance (RS) and the

total capacitance (CT ) of the PIN diodes do not yield perfect open or short circuits

and the possibility of introducing other PIN diode in each output is considered. This

second diode must be placed λ/4 away from the rst diode, in this way the isolation

performance is increased by introducing a second pole in the transfer funtion (Fig.

4.15). In these set of gures the operation of the switch is shown. The PIN diodes

are shunt placed to reduce the insertion loss during the operation.

(a) Schematic equivalent. (b) Diodes in output 1 are reversed bia-

sed and forward biased in output 2.

(c) The second diode is an open circuit in

series with the output.

(d) The rst diode is shunt short circuit

with the output.

Figure 4.15: Graphic equivalent of the double SPDT operation.


Figure 4.16: Comparison of the features of the single and double SPDT.

In Fig. 4.16 the comparison of the single and double SPDT switches is shown.

At the operation frequency, the insertion loss for both cases are -1 dB, however it

can be see how the isolation and the leakage are ≈-12 dB for the single diode SPDT

meanwhile these values for the double diode SPDT are ≈-22 dB. The increase of

complexity in the design and cost is worth in comparison with the features obtained.

Figure 4.17: Double PIN diode SPDT switch Photograph.

The construction of one SPDT switch with two PIN diodes in each output is

shown in Fig. 4.17. The PIN diodes are shunt placed and are fed by high impedance

λ/2 lines. The forward current that fed the diodes is limited by polarization resistors,

in this way the series resistance of the diode is characterized accurately at all the


times. Several circuits were constructed with dierent λ/2 lengths in order to obtain

a precise short circuit at the operation frequency (Fig. 4.18).

(a) Isolation. (b) Transmission.

Figure 4.18: Double PIN diode SPDT switch measurements.

In Fig. 4.19 the matching, isolation and transmission parameters for the circuit

adjusted to 24 GHz are shown. The insertion loss are smaller than -2 dB when the

eect of the connectors is taken into account and the matching of the ports is better

than -10 dB over the whole operation band. The isolation obtained (<-40 dB) shows

that double SPDT circuit model presented in Fig. 4.15 works properly.

Figure 4.19: SPDT full measurements at 24 GHz.

As long as these devices must operate under a wide range of circumstances, a


temperature study is carried out. In order to study the behaviour of the device with

the temperature is necessary to remember that the thicker intrinsic region of the

PIN diode W , the lower the switching time, ts, however the higher power handling

capacity. The operation or ambient temperature, TA, is directly related with the

maximum allowable power dissipation, PD in Watts, which is determined by:

PD =Tj − TA

θ(4.8)

where Tj is the maximum allowable junction temperature (usually 175). Power

dissipation may be computed as the product of the RF current squared multiplied by

the diode resistance, RS. For CW applications the value of the thermal resistance,

θ, used is the average thermal resistance, θAV . However in most pulsed RF and

microwave applications where the duty factor, DF, is less than 10 % and the pulse

width, tp, is less than the thermal time constant of the diode, good approximation

of the eective value of θ in C/W is obtained (eq. 4.9).

θ = DF · θAV + θtp (4.9)

Where θtp is the thermal impedance of the diode for the time interval correspon-

ding to tp.

The double PIN SPDT switch is tested over temperature (Fig. 4.20), under the

same biasing conditions but with a temperature variation from -20C to 80C. An

insertion loss rise can be noticed when the temperature is raised (Fig. 4.20(a)).

This makes sense because the power that the diode can handle is reduced when the

ambient temperature raises and therefore its behaviour as a short circuit is worse.

As long as the diodes are placed as a shunt, more current goes through the isolated

path, hence the overall insertion loss to the desired output are increased.

When the outputs of a SPDT switch are connected to an antenna array the iso-

lated port is an open circuit for the distribution network (network 2), therefore all

the power is reected and re-radiated or coupled to the other network (network 1)

through the radiating element (Fig.4.21). This situation could yield strong interfe-

rences and modify the desired radiation pattern.


(a) Isolation. (b) Transmission.

Figure 4.20: SPDT switch measurements over the temperature.

Figure 4.21: Coupling between networks due to the open circuit of the isolated path

of the conventional SPDT switch.

To overcome this problem, a combination of several circuits that select one from

two possible outputs matching all the ports at any time is proposed [4.23]. A

Wilkinson divider splits the input power equally in two outputs paths. The rst

ouptut is connected to a 270 delay line meanwhile the other output of the Wilkinson

divider is connnected to a 180 reective phase shifter. Both paths are connected

to the input (A) and isolated (C) ports of a branch line 3dB/90 . The distribution

networks are connected to the outputs of the branch line (B, D). Fig. 4.22 shows a

circuit schematic of the complex SPDT with matched output ports.

Thus, the operation of the complex SPDT can be expressed in form of [S] para-

meters in the following form for the OFF state and the electrical length of the delay

lines is φ = 90o:


Figure 4.22: Complex SPDT switch with matched output ports.

[S]SPDT =−j√

2

0 1 1

1 0 0

1 0 0

·e

j3π/2 · −1√2

0 j 1 0

j 0 0 1

1 0 0 j

0 1 j 0

+ ej2π · −1√2

0 j 1 0

j 0 0 1

1 0 0 j

0 1 j 0

(4.10)

Output 1:

S21,Wil · e−j3π/2 · S21,Refl + S31,Wil · e−j2π · S24,Refl =

−j√2

1 · e−j3π/2 · −1√2j +−j√

21 · e−j2π−1√

21 =

j

2· e−j3π/2 − 1

2e−j2π = 0 (4.11)

Output 2:

S21,Wil · ej3π/2 · S31,Refl + S31,Wil · ej2π · S34,Refl =

−j√2

1 · e−j3π/2 · −1√2

1− −j√2

1 · e−j2π−1√2j =

1

2· e−jπ − 1

2e−j2π = 1 (4.12)

Therefore, the output power goes to the output port 2, the output port 1 is

isolated and the input and the outputs are matched at anytime. Similar development

can be done for the ON state:


Table 4.3: Complex SPDT [S] matrix.

ON OFF0 1 0

1 0 0

0 0 0

0 0 1

0 0 0

1 0 0

Output1:

S21,Wil · e−jπ/2 · S21,Refl + S31,Wil · e−j2π · S24,Refl =

−j√2

1 · e−jπ/2 · −1√2j +−j√

21 · e−j2π−1√

21 =

−1

2· e−jπ/2 +

1

2e−j3π/2 = 1 (4.13)

Output 2:

S21,Wil · ejπ/2 · S31,Refl + S31,Wil · ej2π · S34,Refl =

−j√2

1 · e−jπ/2 · −1√2

1− −j√2

1 · e−j2π−1√2j =

1

2· e−j0 − 1

2e−j2π = 0 (4.14)

Finally the [S] of the complex SPDT is shown in Table 4.3:

In Fig. 4.23, the simulation results for the complex SPDT switch with matched

output ports are shown. The insertion loss are 1.1±0.15 dB and the return loss of

all the ports, isolation and leakage between the outputs have values below -23 dB

for the whole operation frequency band.

4.4. Conclusions

In this chapter, the phase shifting distribution networks are thoroughly studied

in order to apply them to switchable beam antennas. A briey fundamentals des-

cription, which is essential to understand the operation of this networks is presented.

The technology overview (MOSFET, MEMS, PIN and Varactor diodes and ferri-

te) employed in switches and phase shifters is presented in the introduction. This

Conclusions 107

Figure 4.23: Insertion loss, isolation and return loss of the complex SPDT switch

with matched output ports.

chapter focuses on the development of distribution networks with the capability of

phase shifting between radiating elements to obtain a steering direction, therefore a

thorough description of switch and phase shifter topologies is shown.

The application of this circuits to automotive RADARs at the frequency band

of 24 GHz requires high performance in terms of isolation, insertion loss, leakage,

temperature behaviour, etc. In this work, a symmetric reective phase shifter with

series-shunt switch at the outputs gets a small dierence of 0.2 dB for the ON and

OFF states while maintaining the desired 120 degrees. The design task carried out

here can be exported to any frequency because the circuit topology is not frequency

dependant.

Diverting the input to one of the two dierent output networks with a single

SPDT can involve low features in terms of matching, isolation and leakage. In order

to solve this problem, a complex SPDT with matched ports regardless operation

constrains is proposed. This circuit combines a 180 reective phase shifter, a

Wilkinson divider, a 270 delay line and a 3 dB 90. The results show a 1.1 dB

insertion loss and return loss, isolation and leakage below -23 dB.

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Chapter 5

Conclusions

This thesis has been carried out in the Radiation Group of the Signal, Systems

and Radiocommunication Department, in the Escuela Técnica Superior de Ingenie-

ros de Telecomunicación of the Technical University of Madrid from October 2009

to September 2013. The work presented in this thesis has been supervised by Pro-

fessor Dr. Manuel Sierra Castañer and co-directed by Dr. José Manuel Fernández

González.

Part of the work reported in this Ph. D. dissertation (specically, chapter 4) was

carried out within a research period (January-June 2012) at the following foreign

institution: Institute of Mobile and Satellite Communications Technology (IMST),

Kamp-Lintfort, Nordrhein-Westfalen, Germany.

A three month stay during 2013 was accomplished in Colorado University at

Boulder, United States of America, under the supervision of professor Dr. Zoya

Popovic. In this stay waveguide to microstrip transtitions at G band (140-325 GHz)

for power ampliers based on Double Heterojunction Bipolar Transistors (DHBT)

were desgined.

The research projects for which the work reported in this Ph. D. dissertation

has generated results are:

Project title: Communication systems for emergency environments (SICOMO-

RO) TEC2011-28789-C02-01. Financial institution: Ministerio de Educación

y Ciencia. Research institutions: Universidad Politécnica de Madrid.Duration:

2011 - 2014. Research in chief: Belén Galocha Iragüen.

Project title: Caracterización de Canales Radio, Optimización y Calibrado de

la Antena GEODA para Comunicaciones Espaciales (CROCANTE) TEC2008-

113

114 CONCLUSIONS

06736-C03-01. Financial institution: Ministerio de Educación y Ciencia. Re-

search institutions: Universidad Politécnica de Madrid Universidad de Vi-

go.Duration: 2009 - 2011. Research in chief: Leandro de Haro y Ariet.

Project title: STEALTH, plan Avanza I+D TSI-020100-2009-76. Financial

institution: Ministerio de Educación y Ciencia. Research institutions: Univer-

sidad Politécnica de Madrid TTI Santander.Duration: 2009 - 2010. Research

in chief at UPM: Manuel Sierra Castañer.

The main nancial support for the developement of this Ph. D. has come from

a grant of Technical University of Madrid CH/003/2011.

5.1. Original contributions

This thesis has yielded the following original contributions:

The design of a wideband steering antenna for satellite communication at X

band. In order to full the coverages and data rate throughput under the space

constraints and demanding antenna features several miniaturization and band-

width enhancement techniques are utilized. By combining these techniques,

new printed circuits and radiating elements are generated.

The size reduction of the mushrooms EBG structures to use them as mutual

coupling barriers. So far, the utilization of these structures is made in high

permitivity substrates. The use of high permitivity substrates enhances the

surface wave modes and lower the radiation eciency of the antenna, hence,

it is not very useful. With the EBG miniaturization originally presented in

this thesis the structures can be placed between the radiating elements in low

permitivity substrates.

The design of a symmetric 120 reective phase shifter, which includes a series-

shunt switch to obtain equal insertion loss for the both states (ON and OFF).

With this circuit the amplitude dierence between the switchable lobes is

minimized.

Future research lines 115

The design of a complex SPDT. The purpose of this circuit is to reduce the

leakage, increase the isolation and to match the three ports (input and outputs)

at any time regardles operation constrains.

5.2. Future research lines

Integration and interoperability of steerable antennas in X, Ku and Ka band

for airplanes. This task will be carried out in Airbus Military. The designs

must full the requirements in terms of EMC and EMI in order to operate

safely under any operation scheme.

Scale and development of the reective phase shifter and complex SPDT pre-

sented in Chapter 4 from 24 GHz to X band. In this way, future multibeam

antennas could be developed.

Design of an hybrid 3 dB/90 at 220 GHz in aWR4.3 based on the waveguide to

microstrip transition developed during the short stay in University of Colorado.

5.3. Publications

This Ph. D. dissertation has given rise to the following publications:

5.3.1. Journal publications

P. Padilla, J.M. Fernández-González, J.L. Padilla, G. Expósito-Domínguez, M.

Sierra-Castañer and B. Galocha-Iraguen. 'Comparison of dierent methods for

the experimental antenna phase center determination using a planar acquisi-

tion system ,' in Progress in Electromagnetics Research, Vol. 135. pp.331-346,

2013.

G. Expósito-Domínguez, J.M. Fernández-González, P. Padilla and M. Sierra-

Castañer. 'EBG size reduction for low permittivity substrates,' in Internatio-

nal Journal of Antennas and Propagation, Metamaterials special issue, 2012.

116 CONCLUSIONS


Castañer. 'Mutual coupling reduction using EBG in steering antennas,' in

IEEE Antennas and Wireless propagation Letters, Vol.11. pp. 1265-1268,

2012.

J.M. Fernández-González, P. Padilla, G. Expósito-Domínguez and M. Sierra-

Castañer. 'Lightweight portable planar slot array antenna for satellite com-

munications in X band,' in Antennas and Wireless Propagation letters, Vol.10,

pp. 1409-1412, 2011.


Castañer. 'Dual circular polarized steering antenna for satellite communica-

tions in X band,' in Progress In Electromagnetics Research, Vol. 122. pp.

61-76, 2012.

5.3.2. Conference contributions

5.3.2.1. International

G. Expósito-Domínguez, J.M. Fernández-González, M. Sierra-Castañer and

M. Martínez-Vázquez. 'Switches and phase shifters characterization for au-

tomotive switchable antennas,' in COST VISTA MC Meeting and Workshop.

Greece, Thessaloniki, May. 2013.


Castañer. 'Revision of EBG metamaterials and active antennas,' in 6th Euro-

pean Conference on antennas and propagation - EuCAP 2012. Prague, Czech

Republic, Mar. 2012.


Castañer. 'New EBG solutions for mutual coupling reduction,' in 6th European

Conference on antennas and propagation - EuCAP 2012. Prague, Czech Re-

public, Mar. 2012.

Publications 117


Castañer. 'Mutual coupling reduction techniques in electronic steering an-

tennas in X band,' in 33rd ESA Antenna Workshop on Challenges for Space

Antenna Systems, Noordwijk, Oct. 2011.


Castañer. 'Electronic steering antenna onboard for satellite communications

in X band,' in 5th European Conference on antennas and propagation - EuCAP

2011, Rome, Italy, pp. 2245 2248, Apr. 2011.

5.3.2.2. National

G. Expósito-Domínguez, B. Sanadgol, B. Zhou. 'Phase shifters for switchable

antenna arrays,' in XXVII Simposium Nacional de la Unión Cientíca Inter-

nacional de Radio - URSI 2012, Eche, Spain, Sep. 2012.


Castañer. 'Matriz de Butler de banda ancha en banda X para antenas de haz

recongurable,' in XXVI Simposium Nacional de la Unión Cientíca Interna-

cional de Radio - URSI 2011, Leganes, Spain, Sep. 2011.

G. Expósito-Domínguez, A. Sánchez, N. Ortiz and M. Sierra-Castañer. 'An-

tena impresa con escaneo electrónico en sistemas embarcados para comuni-

caciones por satélite en banda X,' in XXV Simposium Nacional de la Unión

Cientíca Internacional de Radio - URSI 2010, Bilbao, Spain, Sep. 2010.

doctoral thesis - tesis doctoral contribution to...

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