Common Mode Filter Inductor Analysis

Document 200-1

Specifications subject to change without notice. Document 200-1 Revised 11/19/02

AbstractNoise limits set by regulatory agencies make solutionsto common mode EMI a necessary consideration in themanufacture and use of electronic equipment. Commonmode filters are generally relied upon to suppress lineconducted common mode interference. When properlydesigned, these filters successfully and reliably reducecommon mode noise. However, successful design ofcommon mode filters requires foresight into the nonidealcharacter of filter components the inductor in particu-lar. It is the aim of this paper to provide filter designersthe knowledge required to identify those characteristicscritical to desired filter performance.

IntroductionThe filtering of common mode noise is typically not aswell understood as its differential counterpart and thispaper deals with the practical aspects of common modefilters as related specifically to the common mode in-ductor. Common mode noise occurs simultaneously onboth lines of a conductor pair with respect to a commonground, whereas differential noise occurs between con-ductor paths. The frequency response characteristicsof filters incorporating different common mode chokeconstructions is examined. The filter designer shouldthen have a better understanding of common modeinductors and be able to choose the common modeinductor construction which will yield the required at-tenuation characteristic without the additional cost ofover-designing or the failure of under-designing the com-ponent.

I. Types of Noise and Noise SourcesPower converters are often major sources of noise inany equipment. Power converters normally producecommon mode and differential mode noise at harmon-ics of the switching frequency while some wide banddifferential mode noise is usually also produced.1

Conducted emissions from power converters are attrib-utable to a number of causes. The nature of converteroperation (the rectification of the line frequency, andswitching waveforms, for example) and circuit magnet-ics contribute several unique types of noise; also, thecapacitive effects of components and overall mechani-cal structures, such as cases, and the semiconductorcomponents themselves add their own disruptive volt-ages. An input L-C smoothing filter is generally required

in off-line switching regulators, but these inductors andcapacitors may themselves be sources of EMI. If theinductor is constructed with a relatively high Q material,it will display substantial ringing and produce spectralnoise energy. Also, switching noise of the converter maybe coupled back into the line through the distributedcapacitance of the inductor. The power transformer mayalso ring and couple in ways similar to the filter inductorand produce its own EMI.There are semiconductor noise sources associated withtemperature (thermal noise) within the junction of differ-ing materials (contact noise) and electron-hole move-ment in junction devices (shot noise). There exists lowfrequency noise attributable to dc current carrying elec-tronic devices (modulation, flicker, or 1/f noise), due tothe non-ohmic behavior of semiconductors at high fields(hot carrier noise), the generation and recombination ofcharge carriers (generation-recombination noise), andinducted noise at the gate of an FET due to the alter-ation of the source to drain currents by the inducedcharge at the gate.2

II. Real vs Observed Filter Frequency ResponseReal filters do not follow the theoretical expectationsprovided by standard filter alignments (Bessel,Chebyschev and Butterworth See Appendix A). Forexample the expected frequency response of a typicalsecond order low pass filter is shown in Figure 1. How-ever, due to the self-resonant behavior of common mode

Figure 1. Ideal frequency response predicted for secondorder filter with Coilcraft E3493 common mode

inductor and 0.005 F capacitors.

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Document 200-2

Specifications subject to change without notice. Document 200-2 Revised 11/19/02

inductors (and any inductor), the observed response isactually similar to that shown in Figure 2.At frequencies below resonance, where the filter com-ponents are nearly ideal, standard filter configurationsyield very nearly theoretical results. Inductors, ideally,provide an impedance which is directly proportional tofrequency, but this is only true for a long, single layer air-core cylindrically wound coil with large gauge wire (i.e.,the perfect coil).At frequencies above self-resonance, an inductor be-gins to display the full effects of its parasitic compo-nents, particularly the distributed capacitance (Cd).The Cd describes the effective capacity across an in-ductor and is caused by individual turns of wire in closeproximity (Figure 3). It is the distributed capacitancethat gives the inductor its characteristic self-resonantfrequency ( 1 LC2 ).

Other non-ideal aspects of inductors include leakageinductance, which acts as inductance in series witheach winding. All multiple winding chokes display leak-age inductance. The leakage inductance of a winding isthe amount of inductance which is not coupled to any

other windings through a shared core and is undesir-able in transformers because it stores energy withouttransforming it to other windings in the structure. In alow pass filter, however, leakage inductance adds to theattenuation of the filter. In line frequency common modechokes (i.e. where the differential signal passesunattenuated due to coupling of the winds) the un-coupled leakage inductance will aid in the suppressionof high frequency differential noise.Resistive losses such as copper (I2R) and core lossalso affect attenuation. The diameter of wire used in achoke is determined by the amount of current which itwill be required to handle. The larger the current, thelarger the wire. For example, at a line frequency currentof 1 Ampere, 26 AWG wire is required to provide 250circular mils to support the current. As frequency in-creases, the amount of cross sectional area (for a singlestrand of wire) used by the current decreases (skineffect). For frequencies above about 100 kHz, multi-stranded wire (litz wire, with each strand insulated)should be used if the high frequency current is to besupported. For a low pass inductive filter that needs topass only the line frequencies, further attenuation dueto skin effect is actually desirable.Capacitors exhibit parasitics of their own. For filter appli-cations, mylar, mica, and ceramic capacitors are themost useful because they exhibit high self-resonantfrequencies due to minimization of their parasitics (se-ries inductance and resistance, and parallel resistance).

III. Winding ConfigurationsThree winding configurations for inductors are shown inFigure 4. The most simple and least prone to distributedcapacitance of all the standard configurations is thesingle layer wind. The starts and finishes of a singlelayer wind are as far from one another as possible, thusreducing capacitive coupling. A multilayer wind (two ormore single layers) provides capacitance between lay-ers as well as from the start lead to the finish lead (thatare generally close to one another with the finish leadending where the start lead began). The multilayer con-

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Figure 2. Measured frequency response of second orderfilter with Coilcraft E3493 common mode inductor

and 0.005 F capacitors.

Figure 3. Model of an inductor (one common modeinductor winding).

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Figure 4. The tree winding configurations examined

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figuration displays the greatest capacitance of the wind-ing configurations (thus the lowest attenuation at higherfrequencies for an inductive low pass filter).A compromise between the single and multi-layer wind-ing configurations is the third configuration: the pro-gressive (banked) wind. A progressive wind isaccomplished by winding a few turns, then backing up acouple of turns and repeating the process (three stepsforward and two steps back). The start and finish leadsare thus as far apart as possible, with the same numberof turns as a multilayer wind, but without the interleav-ing. The distributed capacitance of a progressive wind-ing configuration is between that of a single layer windand a multilayer wind.

IV. Real Data and DescriptionIn general, above the self-resonant frequency, chokesbecome largely dissipative, yielding roughly constantand substantial resistance for a very limited frequencyband (a few octaves at most); above this limited band,the chokes become predominately capacitive (Figure 5).

The self-resonant frequency of a filters inductor deter-mines the resonant frequency of the filter itself; maxi-mum attenuation is normally achieved near thisfrequency.The capacitor (in a second order filter configuration) de-termines the attenuating behavior of the filter at frequen-cies above resonance (Figure 6), and with very littlecapacitance, the filter response exhibits the rapid de-crease of attenuation due to the inductor response. Alarger value capacitor will increase the slope of attenu-ation after resonance for a limited band of frequencies.A very large filter capacitor, used to maintain stability

out to high frequencies, will cause the post resonantattenuation to increase over that achieved at resonance.Table 1 shows the electrical characteristics of one windof each of the common mode chokes.

Part L@10kHz (mH) Freq (MHz)E3499 20.70 0.2 to 0.3G6252 17.70 0.2 to 0.3E3490 10.90 0.3 to 0.4E5705 6.44 0.5 to 1.0P104 4.00 0.5 to 1.0E3493 3.30 0.5 to 1.0A-S 2.60 1.0 to 2.0A-P 2.50 1.0 to 2.0F5806 1.70 1.0 to 2.0F5593 1.50 0.5 to 1.0B-S 1.20 1.0 to 2.0B-P 1.20 1.0 to 2.0B-D 1.20 1.0 to 2.0F3495 1.08 1.0 to 2.0C-D 0.85 0.5 to 1.0C-P 0.84 1.0 to 2.0C-S 0.80 1.0 to 2.0E3506 0.71 1.0 to 2.0D-D 0.27 2.5D-S 0.27 4.0E-S 0.26 5.0 to 6.0E-D 0.23 3.0 to 4.0E-P 0.21 5.0 to 6.0Table 1. Self-resonant frequency ranges and initialinductance values for the common mode chokes

examined Note the change in resonant frequency ofthe double layer (-D) compared to the equivalently

constructed single layer (-S) and progressivelywound (-P) chokes.

Figure 5. Typical inductor impedance characteristic(Coilcraft E3490).

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Figure 6. Frequency response of a 2nd order filter withCoilcraft E3495 for various capacitor values.

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Document 200-3

Specifications subject to change without notice. Document 200-3 Revised 11/19/02

For similar constructions, self-resonance (the useful andtheoretically predictable frequency limit of inductance)generally decreases as initial inductance (measured be-tween 10 kHz and 20 kHz) increases. Moreover, the self-resonant frequency of a layer wound inductor decreasesas the number of layers are increased (while maintainingthe same turn count).The progressive or banked windings (-P) of the tabledisplay the same self-resonance as the single layer (-S)versions. Progressive winding allows the increased turnsof a multilayer wind while maintaining the optimum char-acteristics of a single layer wind.Because our examination was to determine the attenu-ation effected by a typical EMI filter configuration due todiffering chokes and choke constructions, we used thecircuit shown in Figure 7, maintaining all circuit compo-nents constant except for the inductive element.

We initially felt it necessary to determine whether differ-ential power (60 Hz) applied to the common mode cir-cuit would affect the high frequency common modeattenuation of the filter. Presumably, the differential op-eration of the common mode circuit, with inductorscoupled to each other and their relative polarity suchthat their equal and opposite (differentially produced)flux lines cancel, no inductive reactance should be en-countered by the differential signal and saturation of thecore should therefore not occur from the differentialsignal. To prove this we used the circuit shown in Figure8 with a load, RL, to provide the rated current throughthe choke.

The LISN (see Appendix B) of the power circuit wassplit at 2 MHz between the standard 50 Henry/5 Ohmarrangement (below 2 MHz) and the 50 Farad/50 Ohmarrangement (providing 50 Ohms above 2 MHz to thenoise source). Splitting the LISN (and later splicing theattenuation curves at 2 MHz) provided a more accuratecomposite LISN than either arrangement alone wouldhave with the required power components.Neglecting measurement error (approximately 4 deci-bels), the differential input did not appear to affect thecommon mode attenuation of the circuit even at highfrequencies (through 10 MHz).Figures 9 and 10 show the common mode attenuationby second order filters with the standard chokes and0.005 F capacitors and using the LISN load.

Figure 7. Test circuit used to measure common modesignal attenuation.

LISN0.005 F

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Figure 8. Test circuit used to apply common mode signaland differential (60 Hz) power simultaneously.

Figure 9. Common mode attenuation of second orderfilters with 0.005 F capacitors and various

toroid inductors.

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Specifications subject to change without notice. Document 200-4 Revised 11/19/02

The attenuation plots vary greatly below 10 MHz andabove 70 MHz, but provide very similar attenuation be-tween these frequencies. Above 10 MHz, the capacitorand its anomalies dominate the filters attenuation, andabove 70 MHz the distributed capacitance of the chokeshunts increasingly more noise signal across the chokeas frequency increases. If the distributed capacitanceof a choke is increased (e.g., by multi-layer) the filterattenuation further decreases and displays this decreaseat lower frequencies.Figures 11 and 12 display the effect of increased distrib-uted capacity (by double layering) of various chokes andconstructions.

When comparing the attenuation data of the three con-figurations (S=single layer; P=progressive wind; D=doublelayer), the difference between the single and progressivewound chokes are not very significant; the double layerversions do, however, display less attenuation at highfrequencies than either the progressive or double layerchokes.Figure 13 shows the effect of leakage inductance onattenuation. It was expected that increased leakage in-ductance may slightly increase the high frequency com-mon mode attenuation. From the data taken, however, nosuch trend was apparent, and it appears that typicalchanges in the leakage inductance do not significantlyaffect the performance of common mode inductors.

Figure 10. Common mode attenuation of second orderfilters with 0.005 F capacitors and various

E core inductors.

E3495A(1.2 mH)

P104 (4.0 mH)

E3400(11 mH)

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Figure 11. The effect on attenuation due to the differingdistributed capacitance of single and double

layered winds.

D (16.7 pF)

P (4.4 pF)S (4.8 pF)

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Figure 12. The effect on attenuation due to the differingdistributed capacitance of single and double

layered winds.

D - Single Layer (5.8 pF)

D - Double Layer (7.6 pF)

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Figure 13. The effect on attenuation due to leakageinductance.

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Document 200-5

Specifications subject to change without notice. Document 200-5 Revised 11/19/02

V. DiscussionThe data taken can be used in the following ways:1. The data show that common mode filter response is

the same as that expected for the more familiardifferential mode L-C filter; common mode filter re-sponse can be fairly accurately predicted using stan-dard L-C calculations except where the componentsused exhibit non-ideal characteristics. For example,Figure 2 shows the theoretically calculated fre-quency response for an L-C filter using a 3.3 mHcommon mode choke, a 0.005 F capacitor, andthe LISN load. With these component values onewould expect a high frequency rate of attenuation of20 dB per decade. It can be seen from the actualresponse of the same filter (shown in Figure 2) thatthe measured rate of attenuation for frequenciesbelow resonance (within an octave of resonance)agrees quite well with the expected slope and ap-proximate value. This is useful in that for frequen-cies below inductor self-resonance, the componentvalues necessary to achieve a desired level of at-tenuation may simply be calculated.

2. The data also show that common mode filtersachieve a maximum value of attenuation at the self-resonant frequency of the common mode inductor.The self-resonant frequency of the inductor thusbecomes an easily used indicator of whether oneshould adjust the capacitor value or the inductorvalue to achieve greater attenuation at a specificfrequency or frequency band. For example, Figure6 shows the differences in attenuation caused bychanging the filter capacitor value. If one were inter-ested in attenuation at 4 MHz, it is seen that theattenuation can be increased from 55 dB to 85 dBby increasing the capacitor value from 50 nF to 100nF; whereas even a relatively large change in in-ductance would have negligible affect.

3. In general common mode filter response can bebroken down into three frequency regions of inter-est: (A) the region below inductor self-resonance inwhich calculations based on component values holdtrue; (B) the region near inductor self-resonance inwhich the filter achieves the maximum attenuation;and (C) the region above inductor self-resonance inwhich the response is dominated by the filter ca-pacitor.

ConclusionsCommon mode filter response differs substantially fromtheoretically predicted performance. Filter performancecan be explained and successfully manipulated if

nonideal component response is taken into account. Thecommon mode inductor is a primary component in deter-mining the response of a typical filter circuit. The com-mon mode inductor affects the magnitude (maximumattainable attenuation) and the shape (resonant fre-quency) of the frequency response of the filter.A filter designer should carefully consider filter responsefrom approximately 1 MHz to 30 MHz to determinewhether the slightly diminished attenuation of a multiplelayer inductor is acceptable. If one were to specify theinductor to be single layer it may result in unnecessarycost and size penalties. The winding pattern of the indi-vidual windings (Cd) is far more significant than therelationship of the two windings (leakage inductance).The distributed capacitance of a choke decreases at-tenuation at high frequencies and multilayering increasesthe distributed capacitance of an inductor. Progressivewinding allows the equivalent number of turns of wire asa multilayer and usually far more turns than a singlelayer could accommodate. A progressively wound in-ductor will display a distributed capacitance similar to asingle layer. To attenuate noise voltages which occurabove the limits prescribed by the FCC and VDE, thefilter designer may opt for the progressively wound in-ductor.

Appendix APassive filters serve as a very good means of eliminat-ing the majority of conducted noise into a device (or outof a device and back into the line) when relatively highcurrent is flowing. Many filter configurations exist andeach has its own advantages. Commonly consideredfilter alignments are the Chebyshev, the Butterworth,and the Bessel.The ideal Chebyshev low pass filter alignments allow acompromise between the amount of ripple in the pass-band (and damping) and the slope of attenuation at thecutoff frequency. The stable behavior of the time re-sponse of Chebyschev filters is related to the dampingfactor, the allowed ripple, and the slope of attenuation atthe cutoff frequency; as the slope of the attenuation atthe cutoff is increased, the transient response becomesless stable and prone to ringing, and the phase re-sponse becomes much less linear.Butterworth low pass filter alignments are Chebyshevfilters designed for minimum ripple and ideally provide aflat response, no attenuation prior to the cutoff frequency,and a damping factor of approximately 0.7. After thecutoff frequency, the attenuation begins and continuesat 20 times the order in decibels per decade. The timeresponse for the Butterworth filter exhibits some ringing

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Specifications subject to change without notice. Document 200-6 Revised 11/19/02

and the phase response is not ideal, but predictable. Theoverall response of Butterworth filters is well suited toquick and easy methods of approximation. Ideally, Besselfilters maintain a very stable, linear phase response butat the expense of frequency response; the time responseof Bessel filters is well behaved without substantial over-shoot or ringing.In their use as noise suppressors, filters must be able toeliminate as much noise as possible over a predeter-mined frequency band. As long as the inherent ringingof the filter is low (does not itself become a noise source)it is the frequency rather than the time response of afilter that is an important concern. In a common modefilter, the differential signal does not encounter the filter,thus any problems associated with phase or time re-sponse affect only the common mode noise. When thefilter is used to keep noise from entering the powermains from the device, phase and time responses be-come trivial to even the differential voltage.Butterworth alignments for the design of EMI filters arean appropriate starting point; they are easily modeled,readily approximated and constructed, and provide goodfrequency response with relatively little ringing.

Appendix BA major object of testing a component part or device isthe ability to repeat the testing with the same resultsregardless of the time or place of the test. When testingrequires the power line to be stable, or at least standardfor a test, adjustments must usually be made or spuri-ous results may occur.Power lines may vary in output impedance by as muchas 40 Ohms from location to location, making the re-peatability of line based evaluations very unreliable be-tween differing locations. A Line Impedance StabilizingNetwork (LISN, or Artificial Mains Network: AMN) al-lows uniform line based testing regardless of locality. Aline impedance standard has been devised and is speci-fied by several licensing and independent safety agen-cies (e.g., the FCC and VDE; see Figure 14).3

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Figure 14. LISN characteristic specified by FCC Docket20780 and VDE 0876 Part 1.

(1) Schneider, Len M. Noise Sources Equivalent CircuitModel for Off-Line Converters and Its use in Input FilterDesign. Proceedings of Powercon 10, March 1983, P. C-1.(2) Scidmore, A.K. Noise in Amplifiers. Lecture Notesfor ECE-341Linear Active Circuits, University of Wis-consin: 1982.(3) Nave, Mark Line Impedance Stabilizing Networks:Theory and Use. RF Design,April 1985, P. 54.Kendall, C.M. and Schmid, A.A. Characteristics and Con-trol of EMI in a Switching Regulator Power Converter.Proceedings of Powercon 10, March 1983, P. C-4.Head, Mike Know RF-Emission Regulations Pertainingto Your Design. EDN, August 1981.Kociecki, John Predicting the Performance of Common-Mode Inductors. Powerconversion International, March1984, P. 56.Carsten, Bruce W. Design Techniques for the InherentReduction of Power Converter EMI. Proceedings ofPowercon 11, March 1984, P. D-2.Nye, J.F. Physical Properties of Crystals. Oxford Univer-sity Press, 1972.Williams, Arthur B. Electronic Filter Design Handbook.McGraw-Hill, 1981.

Document 200-7

Specifications subject to change without notice. Document 200-7 Revised 11/19/02