development of 50-kv 100-kw three-phase resonant converter...

6674 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 63, NO. 11, NOVEMBER 2016

Development of 50-kV 100-kW Three-PhaseResonant Converter for 95-GHz Gyrotron

Sung-Roc Jang, Jung-Ho Seo, and Hong-Je Ryoo

Abstract—This paper describes the development of a50-kV 100-kW cathode power supply (CPS) for the operationof a 30-kW 95-GHz gyrotron. For stable operation of the gy-rotron, the requirements of CPS include low output voltageripple and low arc energy less than 1% and 10 J, respec-tively. Depending on required specifications, a three-phaseseries-parallel resonant converter (SPRC) is proposed fordesigning CPS. In addition to high-efficiency performanceof SPRC, three-phase operation provides the reduction ofthe output voltage ripple through a minimized output filtercomponent that is closely related to the arc energy. For al-lowing symmetrical resonant current from three-phase res-onant inverter, the high-voltage transformers are configuredas star connection with floated neutral node. This facilitatesbalanced voltage on each secondary winding. In addition,distinctive design of the high-voltage rectifier is introduced,taking into consideration the effective series stacking ofdiodes by means of the parallel resonant capacitor. In partic-ular, the implementation of the high-voltage part includingtransformer and rectifier is presented in detail. For provid-ing high power density and high reliability, effective meth-ods for winding the high-voltage transformer and stackingrectifier diodes are discussed. Finally, the developed CPSachieves 95.5% of maximum efficiency, 0.92 of maximumpower factor, 500 W/liter of power density, 0.6% of outputvoltage ripple, with 8.3-J arc energy.

Index Terms—DC–DC power converters, gyrotrons,pulsed power supplies.

I. INTRODUCTION

R ESEARCH on high-power vacuum devices, such as mag-netron, klystron, and gyrotron, has grown in recent times

owing to the development of various industrial applications in-cluding medical, military, environmental, and aerospace. Therecent proliferation of applications requiring high voltage andpower has led to a greater focus on the development of the high-voltage power supplies [1]–[22]. In addition to the general re-quirements of high-voltage power supplies, efficient operation,

Manuscript received January 12, 2016; revised March 14, 2016; ac-cepted May 23, 2016. Date of publication June 29, 2016; date ofcurrent version October 7, 2016. This work was supported by theKorea Electrotechnology Research Institute Primary Research Programof MSIP/NST (16-12-N0101-49).

S. R. Jang is with the Electric Propulsion Research Center, KoreaElectrotechnology Research Institute, Changwon 641-120, South Korea,and also with the University of Science and Technology, Daejeon 13557,South Korea (e-mail: [email protected]).

J.-H. Seo is with the Department of Energy and Power ConversionEngineering, University of Science and Technology, Daejeon 13557,South Korea (e-mail: [email protected]).

H.-J. Ryoo is with the School of Energy Systems Engineering, Chung-Ang University, Seoul 06974, South Korea (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are availableonline at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2016.2586021

and high power density, the operation of high-power vacuumdevices requires low output voltage ripple with low arc energy.This is because the output voltage ripple and the arc energyare closely related to the stability of the output power of theelectron beam and the safety of the device, respectively. It isclear that a higher value of the output filter capacitor allowsa lower value of the output voltage ripple. On the other hand,the energy stored in the power supply output, which may in-stantaneously be discharged to vacuum devices because of thearc, is proportional to the value of the output filter capacitor. Inorder to achieve low output voltage ripple with low arc energy,a pulse step modulator (PSM) has been proposed to operate agyrotron [1], [2]. Compared with other proposed designs, basedlargely on alternating current (ac) voltage regulation or a star-point controller, the PSM-based high-voltage power supply isexpected to be highly reliable owing to low stored energy. Otherapproaches that use additional solid-state switches to protect theload against the arc have been proposed [7]. A crowbar circuitconnected in parallel with the load helps us to limit the energyfrom the power supply to the load, and a crowbar switch usinga solid-state device exhibits short response time such that theload can be effectively protected. Another method used to limitthe arc energy by means of a solid-state switch is to insert a fastopening switch between the power supply output and the load.Further research on the reduction of the ripple as well as the arcenergy has been presented for electrostatic precipitator appli-cation [21]. Presented inductive adder topology which consistsof high-voltage converter modules in series, and uses the phaseshifting control technique between each module shows manyadvantages including decrease of the output ripple as well as thestored energy.

Based on the basic concept of the inductive adder [21] andthe operating principle of the series-parallel resonant converter(SPRC) [22], the design and the implementation of a 50-kV100-kW CPS for a 30-kW 95-GHz gyrotron are described in thispaper. A three-phase resonant converter based on half-bridgeSPRC module is proposed for achieving desired specifications.For balanced three-phase operation, the star configuration withfloated neutral node is suggested for three high-voltage trans-formers that are connected to each of three half-bridge resonantinverters. In addition, a distinctive design of a high-voltagerectifier is introduced for minimizing component count.

Generally, the parallel resonant capacitor is connected inparallel with the transformer primary or secondary winding[22]. On the other hand, the proposed rectifier circuit forCPS uses the capacitors which are connected in parallel withrectifier diodes for balancing voltage of each diode as wellas for implementing the parallel resonant capacitor. With

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JANG et al.: DEVELOPMENT OF 50-KV 100-KW THREE-PHASE RESONANT CONVERTER FOR 95-GHZ GYROTRON 6675

TABLE ISPECIFICATIONS OF CPS FOR A 30-KW95-GHZ GYROTRON

Input Voltage 380 Vac ± 10%

Maximum output voltage, Vo , m a x −50 kVMaximum output current, Io , m a x −2 AMaximum output power, Po , m a x 100 kWMaximum pulse width, PWm a x 3 sMaximum duty cycle, Dm a x 50%Output voltage ripple at rated voltage 0.6%Arc energy, EA rc 8.3 JMaximum Efficiency, ηm a x 95.5%Maximum Power Factor, PFm a x 0.92Protections Arc (overcurrent)

OvertemperatureOvervoltage

considerations such as high-voltage insulation and coolingof components, the detailed implementation of the proposedcircuit is discussed to improve power density with reliableoperation. Especially for high-voltage part which is immersedin insulation oil, compact design and arrangement are presentedincluding the high-voltage transformer and rectifier.

Section II presents the design of CPS based on three-phaseSPRC with respect to the required specifications summarizedin Table I. Feasibility and performance of the proposedcircuit is verified by PSpice simulation. In Section III, thedetailed implementation including a special winding methodfor high-voltage transformer and a compact design of rectifiercircuit is described. Finally, the performance of the developedCPS is experimentally proven from the point of view of itsefficiency, power factor, output voltage ripple, and arc energy.

II. DETAILED CIRCUIT DESIGN OF CPS

SPRC which operates in continuous-conduction mode andhas above-resonance switching frequency range has the advan-tages of zero voltage (ZV) turn-on and relatively low conductionloss compared with other resonant converter mode of opera-tions. Moreover, the lossless snubber capacitor which can beconnected in parallel with the semiconductor switches reducesturn off switching loss by decreasing the slope of the voltagerise across the switches. However, it should be noted that there isadditional consideration in choosing the value of snubber capac-itance for achieving ZVS. This is because the snubber capacitorshould be discharged before turning on the switch by means ofstored energy in resonant inductor.

For designing high-voltage and high-power converters, theSPRC has many advantages compared to the other resonant con-verter topologies that use different resonant tank structure suchas series resonant converter (SRC), parallel resonant converter(PRC), and LLC resonant converter. First of all, the SPRC hasthe current source characteristic of SRC and the intrinsic voltageboost-up characteristic of the PRC. Also, the parasitic capaci-tance of the high-voltage transformer can play a role of a parallelresonant capacitor instead of causing adverse effect. In addition,the SPRC provides relatively wide controllable output voltage

Fig. 1. Proposed circuit of SPRC module for CPS.

range so that CPS can control the required output voltage from−5 to −50 kV.

The operating principle of SPRC have been already intro-duced and well known [13], [15], [19], [20], [22]. Therefore,this paper intends to omit the description about the operatingprinciple of SPRC and deal with a detailed design of CPS.

A. Design of SPRC Module for CPS

The proposed circuit of SPRC module is shown in Fig. 1.Compared with the conventional circuit of SPRC, it is worthnoting that the parallel resonant capacitor (Cp , dashed line) is ac-tually not connected in parallel with the transformer primary orsecondary winding. For implementing the parallel resonant ca-pacitor, the proposed circuit uses the capacitors (CD1 1−CD1 N )that are originally installed for balancing the voltage betweenseries stacked diodes. Thus, the capacitors which are connectedin parallel with the diodes play the role of the parallel reso-nant capacitor. Because it should be discharged and chargedfor forward and reverse biasing of diode, respectively. Fromthe transformer secondary side, equivalent parallel capacitorcan be regarded as the parallel of two capacitors (CD1 , CD2)where CD1 is equivalent capacitor for series connection ofCD1 1−CD1 N (CD1 1/N). Accordingly, the value of Cp is cal-culated as n2×(CD1 + CD2) where n is the transformer turnsratio (n2/n1). By choosing suitable value of CD1 and CD2 , ef-fective balancing of voltage between the series stacked diodesas well as inserting the parallel resonant capacitor can be simul-taneously achieved with minimized component count. Anotherpoint worth mentioning is that the proposed circuit does not onlycontrol the switching frequency, but also adjusts the dead timewhich represents the time duration between the turning off of oneMOSFET and the turning ON of another MOSFET. According togeneral control characteristic of SPRC, it is clear that the outputvoltage can be controlled by switching frequency modulation.Besides controlling the switching frequency, the proposed cir-cuit changes the dead time for allowing ZV turn on of MOSFET

independent of the load condition [13]–[15], [19], [20].As shown in Fig. 1, the gate–source signal of MOSFET (VGS )

has less pulse width compared with the frequency modulatedsignal (Vsw ) from the controller.


Fig. 2. Designed circuit of CPS based on the three-phase SPRC.

Fig. 3. Analysis of the proposed three-phase SPRC for CPS. (a) Operating waveforms depending on operational mode. (b) Operational modeanalysis.

Thus, the gate drive (GD) circuit is designed for detectingZV condition of MOSFET and providing the switching signalso that ZV turn ON of MOSFET can be always achievedirrespective of the load condition. When SPRC use rela-tively high value of lossless snubber capacitors (Csn1 , Csn2)for reducing turn off loss, this additional function of theproposed GD circuit provides high efficiency and reliableoperation.

B. Analysis and Design of CPS Based on Three-PhaseSPRC

Based on the proposed SPRC module, CPS is designedas shown in Fig. 2. In order to minimize both arc energyand output voltage ripple, CPS is designed based on a three-phase resonant inverter. Three switching legs consisting ofMOSFET (S1/S1′ − S3/S3′) operate with a 120° phase delayas shown in Fig. 3(a). High-frequency ac power is transferred


from each switching leg to the high-voltage transformer (TR1–TR3) through a series resonant inductor (Ls1−Ls3) and ca-pacitor (Cs1−Cs3). The voltage on the transformer’s sec-ondary winding is rectified by a voltage-doubled rectifierconsisting of the diodes (Dr1 − Dr6) and the filter capaci-tors (Cf1-Cf6). The three rectifiers are connected in seriesto generate −50 kV of output voltage, which is six timesthe value of the voltage on secondary winding. The recti-fying diodes (Dr1 − Dr6) are practically implemented by aseries stacking of high-frequency diodes, and the capacitors(Cp1−Cp6) represent the equivalent capacitance of all capaci-tors connected in parallel with each high-frequency diode [19].In addition, Cp1−Cp6 act as parallel resonant capacitors con-nected in parallel with the transformer’s secondary winding.Thus, the proposed converter can take advantage of an intrin-sic voltage boost-up function owing to the parallel resonantcapacitor.

As depicted in Fig. 2, the star configuration of three high-voltage transformers provides neutral node (TR_N). The neutralnode of transformer (TR_N) is typically connected to the neutralnode of dc-link capacitor (DC_N) in order to operate three-phaselegs as individual three half-bridge inverter. In other words, in-dependent operation of three-phase half-bridge SPRC can beachieved when two neutral nodes (TR_N, DC_N) are connectedtogether. Although the aforementioned configuration is a generalapproach for three-phase system [20], the proposed converterdifferentiates between the two neutral nodes in order to over-come the disadvantages of the star with neutral configuration.One disadvantage of star configuration with two neutrals con-nected together is that the rms value of high-frequency currentflowing between these two neutral nodes is relatively high. Thisis due to the fact that SPRC does not generate pure sinusoidalwaveform of resonant current, but trapezoidal waveform [15],[19], [20]. Therefore, the sum of three-phase resonant current isnot zero. This neutral current poses a challenge in choosing suit-able dc-link capacitor (Cdc1 , Cdc2) due to high-frequency rmscurrent. Another demerit is the unbalanced operation betweeneach phase. It was experimentally found that the output voltageof each converter module could be different due to the smalldifference in resonant tank parameters. In addition to the differ-ences in the three rectified voltages that are connected in series,the unbalanced shape of three resonant currents causes signifi-cant adverse effect on the control as well as the losses. In orderto solve these problems, the star configuration of transformerwith floating neutral node is suggested. When two neutral nodesare floated, it is not necessary to consider high-frequency neu-tral current. Moreover, the proposed three-phase SPRC is nolonger the individual three half-bridge converters because fulldc-link voltage (Vd ) is applied to two resonant tanks and twotransformers. The detailed operating principle of the proposedcircuit can be analyzed from Fig. 3. It includes the operatingwaveforms of each operational mode depending on the currentflowing.

The meaning of solid and dashed lines in Fig. 3(b) that rep-resent current flowing path are defined as follows:

1) Solid line at inverter side: current flowing path from dcinput (Vd ) to the transformer.

2) Dashed line at inverter side: freewheeling current flowingpath for allowing ZVS during switching transition.

3) Solid line at rectifier side: current flowing path for charg-ing the output filter capacitor and supplying the loadcurrent.

4) Dashed line at rectifier side: current flowing path forcharging parallel resonant capacitor.

When S1′ is turned OFF, Mode1 starts with conducting an-tiparallel diode (D1) and turning ON of S1 with ZV. Actually,there is an elliptical mode that charge and discharge the snubbercapacitors (C1, C1′) between the turning off of S1′ and the turn-ing ON of S1. During Mode1 operation, freewheeling current ofswitching leg1 flows through S3, TR1, and TR3 as shown inFig. 3(b) (dashed line at Mode1).

Owing to continuous current flowing through TR1, the recti-fier diode Dr1 maintains forward biased and charges the outputfilter (Cf1). The solid line in Fig. 3(b) represents the currentflowing path for powering that apply input voltage (Vd ) to se-ries connected two resonant tanks and two transformers. Theseoperations imply that the three-phase resonant inverter operatesnot independently as half bridge. The operation is similar tofull-bridge circuit that supplies full input voltage to two reso-nant tanks and transformers simultaneously. In other words, theresonant current always flows through the combined resonanttank.

The resonant current, ILs3 , is rectified, and it charges thefilter capacitor (Cf6). On the other hand, additional operatingmode of rectifier is observed for the secondary side of TR2during Mode1 operation. When the resonant current changesits polarity, it is also necessary to change the conducting diodesdepending on the operating principle of the voltage doubled rec-tifier. As shown in Fig. 3(a), the resonant current, ILs2 , changesits polarity from positive to negative at the starting of Mode1.Thus, the secondary-side current flows to discharge Cp3 andcharge Cp4, respectively. During this time duration, the res-onant current has different frequency because two capacitors(Cp3, Cp4) play the role of the parallel resonant capacitor.

Mode2 starts with S1 conducting, and the subsequent flowof positive current from the dc input to TR1. This transition inthe current polarity also requires the charging and dischargingof Cp1 and Cp2, respectively. During Mode2 operation, S3 isturned off and the freewheeling current flows through the an-tiparallel diode D3′. At Mode3, the power from the switchingleg1 is finally transferred to the load by means of diode Dr2.Although the explanation of the operational mode is made fo-cusing on the switching leg1, all the operations of the proposedcircuit can be analyzed based on the conducting devices andwaveforms depicted in Fig. 3.

In addition to the analysis and the design of SPRC, the de-tailed design procedure and equations of three-phase SPRC havebeen presented in [23]–[27]. Depending on the operation ofthe proposed three-phase SPRC, the brief design guideline andconsideration for CPS are as follows.

For designing the resonant tank parameters of CPS, the min-imum switching frequency should be determined dependingon the requirements of the output voltage ripple and arc en-ergy. From the determined minimum switching frequency, the


TABLE IISUMMARY OF DESIGN PARAMETERS FOR CPS

Resonant inductor (Ls1−Ls3) 5 μHSeries resonant capacitor (Cs1−Cs3) 30 μFParallel resonant capacitor (Cp1−Cp6) 0.38 nFSnubber capacitor (C 1−C3) 2 nFTransformer turns ratio (N1:N2) 12 : 400Output filter capacitor (Cf1−Cf6) 40 nF

Fig. 4. Simulation results of designed CPS at rated operation.

Fig. 5. Comparison of operating waveforms between star configura-tions with floated neutral and connected neutral.

required characteristic impedance as well as resonant frequencycan be calculated for achieving maximum output power. Itshould be noted that the proposed SPRC mainly uses the res-onance between the series resonant inductor and the parallelresonant capacitor. Accordingly, the characteristic impedancethat is calculated from the series resonant inductor and the par-allel resonant capacitor determines the peak value of resonant

current. After charging the parallel resonant capacitor, the reso-nant current depends on the series resonant tank that can providerelatively low frequency compared to the resonant frequency byparallel resonant capacitor. The value of the parallel resonantcapacitor is closely related to not only the voltage gain, but alsothe controllable load range. For choosing the value of the snub-ber capacitor, the minimum stored energy in resonant inductorshould be considered. When the resonant tank parameters aredetermined, the required transformer ratio can be calculateddepending on the maximum output voltage.

For designing high-voltage transformer, relatively high valueof leakage inductance due to the insulation distance between pri-mary and secondary winding is inevitable. Thus, it is preferableto implement the resonant inductor by means of transformerleakage inductor. Especially for high-power resonant converter,loss and size of resonant inductor cannot not be ignored due tothe relatively high value of ac resonant current. Therefore, highpower density and high efficiency can be achieved by usingleakage inductance instead of additional inductor. For design-ing transformer, the value of desired leakage inductance can beconsidered [17].

Based on aforementioned design guideline and considerationof the proposed circuit, all the designed parameters for a 50-kV100-kW CPS are summarized in Table II. The value of parallelresonant capacitor (Cp) is calculated from Cp1 which is actuallyimplemented using 18 pieces of 6.8-nF capacitor in series. Tocontrol the output voltage and the current with desired loadrange, the operating switching frequency is determined from arange of 45 to 200 kHz. The value of the equivalent output filtercapacitance is 6.67 nF, and it provides 8.3 J of maximum arcenergy at the rated output voltage.

C. PSpice Simulation of Designed CPS

The detailed design of CPS for the 30-kW 95-GHz gyrotrondescribed in the foregone sections was verified using PSpicesimulation. Based on the designed circuit and parameters inFig. 2 and Table II, the CPS was simulated with a resistor loadand the results are shown in Fig. 4. With 25-kΩ load condition,simulated waveforms including the resonant current, the MOS-FET switching voltage, the output voltage, and power confirmthe fact that the rated operation of designed CPS is achievedwith 500 V of input voltage, 45 kHz of switching frequency,and 300 A of resonant current peak value.

From the switching voltage (VDS ) and the current through theanti-parallel diode, ZV turning on of S1 is confirmed becausethe gate–source voltage increases after conducting antiparalleldiode. In addition, the output voltage ripple measurement wave-form shown in Fig. 4 shows a 0.3% ripple rate with designedfilter. It is clear that the frequency of output voltage ripple issix times that of the resonant current due to the three-phasehigh-frequency operation.

For comparing the operational waveforms between star con-figurations with floated neutral and connected neutral, PSpicesimulation is performed with the same operating parameters in-cluding the resonant tank, the switching frequency, and the loadcondition. As depicted in Fig. 5, the proposed circuit generates


three-phase sinusoidal waveform which provides three-phasebalanced resonant current. On the other hand, star configurationwith neutral connection exhibits relatively high value of neutralcurrent because the three-phase resonant current has trapezoidalwaveform. Compared with the waveform of the proposed cir-cuit, trapezoidal waveform of the resonant current provides anadvantage related to the conduction loss because rated outputpower can be achieved with less rms value of resonant current.However, high-frequency neutral current causes an additionalproblem for designing input filter. As shown in Fig. 5, 100 A ofrms current should be considered when choosing high-frequencyfilter capacitor (Cdc1 , Cdc2).

III. IMPLEMENTATION OF CPS

From the designed parameters summarized in Table II, the de-tailed implementation of high-voltage components is describedfor achieving high power density as well as high reliability.

A. Implementation of High-Voltage Transformer

Depending on the design performed in Section II, the trans-formers with 12:400 of turns ratio are implemented by meansof UU shape ferrite core (cross-sectional area: 12 cm2) for pre-venting saturation and generating 50 kV of maximum outputvoltage. To implement the high-voltage transformer, provid-ing insulation distance between the primary and the secondarywinding is essential. In addition, securing sufficient insulationbetween each wire of secondary winding is also important whena number of winding layers are necessary for satisfying designedturns ratio. Generally, the secondary winding of the high-voltagetransformer uses multiple layer structure to wind a number ofwires within restricted winding area.

Fig. 6(a) shows an example of a conventional winding methodfor high-voltage transformer secondary winding. In order towind 60 turns of secondary wire within restricted length ofbobbin (Dbobbin), five layers are used for providing insulationdistance between layers. By assuming 250 V per each turn, themaximum voltage difference between layer 1 and layer 2 is 9 kV,as shown in Fig. 6(a). Therefore, it is necessary to provide insu-lation distance with respect to 9 kV, and insulation material mayneed to be inserted between layers. Accordingly, it is inevitablethat the height of the secondary winding will increase due tothe reduction of effective winding area. On the other hand, theproposed winding method shown in Fig. 6(b) provides more ef-fective winding area by minimizing voltage difference betweenadjacent wires. Compared with the conventional method, theproposed winding method limits the maximum voltage differ-ence within 2 kV. Therefore, it is not necessary to provide theinsulation distance between adjacent layers and to insert insu-lation material when the wire has 1 kV of intrinsic insulationstrength. Also, the insulation between each winding section isguaranteed by partitioning flanges which is made by insulationmaterial such as Teflon. As compared in Fig. 6, the proposedwinding method facilitates 80 turns of secondary wire within thesame length of bobbin (Dbobbin ). It should be noted that the sim-ple drawing depicted in Fig. 6 is just to explain the advantages ofthe proposed winding method by roughly comparing it with the

Fig. 6. Comparison of winding method between conventional and pro-posed transformer. (a) Conventional method for secondary winding.(b) Proposed method for secondary winding.

Fig. 7. Structure of designed bobbin for a CPS transformer.

conventional winding method without necessarily consideringthe exact insulation distance with respect to the voltage.

A detailed design structure of bobbin for CPS transformershown in Fig. 7 includes seven partitioning flanges for pro-viding eight winding area on which each secondary windingis wound. The primary winding is wound on the core and thebobbin envelops the primary winding in order to provide insu-lation as well as to minimize leakage inductance. Two smallsections on the top and bottom of bobbin are specially designedfor providing insulation distance against surface discharging be-tween the secondary winding and the primary winding. Basedon the aforementioned winding method, a transformer for CPSis implemented as shown in Fig. 8 and measured magnetizingand leakage inductances are 1.2 mH and 5 μH, respectively.By creating space between primary and secondary winding, the


Fig. 8. Picture of implemented transformer for CPS.

Fig. 9. Circuit of high-voltage rectifier PCB module.

desired value of leakage inductance that is exactly matched withthe designed value of resonant inductor is implemented.

Another point worth mentioning is that we use only one sideof core for the primary and secondary winding. That is es-pecially for effective arrangement of three transformers whileproviding sufficient insulation between them. Consequently, theproposed winding method allows compact design as well as highreliability of high-voltage transformer.

B. Implementation of High-Voltage Rectifier

A high-voltage rectifier is basically designed based on thevoltage doubled circuit as shown in Fig. 9. From the secondarywinding which is connected to P3 and P4, the filter capacitors(C37, C38) are charged through series stacked diodes (D1–D18,D19–D36). For rectifying one third of the maximum outputvoltage, 18 pieces of diodes are stacked in series. The capaci-tors (C1–C36) which are connected in parallel with each diodeprovide voltage balancing between series stacked diodes. Theother role of these capacitors is to serve as the parallel reso-nant capacitor that is connected in parallel with load [19]. Bydesigning suitable value of these capacitors, it is possible toimplement series-parallel resonant tank structure without usingadditional capacitor on the transformer primary or secondaryside. Therefore, the balance of voltage as well as the implemen-tation of parallel resonant capacitor can be effectively achievedwith reduced component count. The sensing circuit by means

Fig. 10. Picture of implemented high-voltage rectifier for CPS.

Fig. 11. Picture of implemented CPS.

of resistor and capacitor divider is also designed in the rectifierPCB as depicted in Fig. 9 (R1– R2, C39–C58).

As shown in Fig. 2, one node (P5) of a high-voltage sensingcircuit is connected to the high-voltage output which has samepotential with connector P1 on Fig. 9. That is, there are only oneconnection between a high-voltage sensing circuit and a rectifieramong three PCBs. Although, the other node of a high-voltagesensing circuit is not connected with a rectifier circuit, a sensingcircuit can be implemented very closely with a rectifier circuit byminimizing the voltage difference between neighboring nodesas depicted in Fig. 10. In addition, there are slits between theseries stacked diode groups for preventing surface discharging.These implementation methods for a rectifier PCB allows com-pact design and provides high reliability against high-voltagebreakdown.

C. Compact Assemble of CPS

In order to achieve high power density and high reliabilityof CPS, the arrangement and the structural design are shownin Fig. 11. They include the resonant capacitor (Cr1−Cr3),three-phase inverter with MOSFETs, the GD circuit, oil im-mersed transformers, and rectifiers. Three MOSFET modules areattached to the heat sink located at the back side of CPS. TheGD circuit which drives six MOSFETs is located on top of the


Fig. 12. Experimental waveforms of the unbalanced operation whentwo neutral nodes (TR_N and DC_N) are connected (switching voltageof S1: 50 V/div., resonant current (ILs1 , ILs2 , ILs3 ): 20 A/div., 5 μs/div.).

MOSFETs. Three-phase resonant capacitors are arranged as “U”shape for reducing the space and allowing effective cooling.There is no additional resonant inductor because it is imple-mented by means of leakage inductance of transformer. TwoFANs at the rear of CPS generate forced air which flows to thetop front side for the cooling all components except the high-voltage part. Three high-voltage transformers and rectifiers areimmersed in insulation oil. By using only one side of the corefor the primary and secondary winding, three transformers canbe arranged effectively while considering insulation distancebetween each secondary winding. In addition, the insulationdistance from the secondary winding to the grounded core isalso guaranteed with reduced volume. The secondary windingof TR1 is located near a Rec1 (in Figs. 2 and 11) for providingsimilar potential between neighboring components.

Likewise, the components including Rec3 and TR3 whichhave less voltage potential are situated near the grounded oiltank case. For the cooling of high-voltage components, the oilfilled tank made of duralumin material functions as a heat sink.For more effective cooling, additional heat sink is attached to thetop cover of oil tank with FAN. From these compact design andeffective cooling structure of CPS, a 50-kV 100-kW power sup-ply is developed within 60 L (width: 440 mm, depth: 490 mm,height: 270 mm). Finally, 500W/L of power density is achievedwith an additional input filter for reducing low-frequency ripple.

IV. EXPERIMENTAL RESULTS OF DEVELOPED CPS

A prototype of CPS for a 30-kW 95-GHz gyrotron was testedwith a dummy resistor load, and the feasibility and performanceof design was verified. Fig. 12 shows the experimental wave-forms of the unbalanced operation when two neutral nodes(TR_N and DC_N) are connected. Three different resonantcurrent waveforms with same operating switching frequencyrepresent the difference in voltage between three rectifies. Fur-thermore, the triangular shape of the resonant current (ILs1)signifies that the rectified voltage is almost zero. On the otherhand, the proposed configurations that share the resonant tankbetween each phase solve the problem depicted in Fig. 12. Toverify the effective voltage balancing between the series stacked

Fig. 13. Experimental waveforms of measured voltage balancing be-tween series stacked diodes (1 kV/div., 1 μs/div., inverted polarity).

Fig. 14. Experimental waveforms at CPS-rated operation (output volt-age: 10 kV/div., resonant current: 200 A/div., switching voltage (VDS ):500 V/div., 10 μs/div.).

diodes, the diodes reverse biased voltage for low-voltage-siderectifier (rectifier connected with TR3 in Fig. 2, Rec3 inFig. 11) was measured during −15-kV operation. As shownin Fig. 13, three voltage waveforms that has the value of 1.73,3.39, 5.09 kV are measured from ground to anode of diode D31,D25, D19 (see Fig. 9), respectively. This experimental waveformverifies that voltage balancing between series stacked diodes isachieved without equalizing resistors. The value of each capac-itor (C1–C36 in Fig. 9.) is 6800 pF with ±10% tolerance. Therising and falling of the voltage waveform represent chargingof capacitors for reverse biasing and discharging of capacitorsfor forward biasing, respectively. Therefore, C1–C36 can playa role of a parallel resonant capacitor.

Rated operation of CPS is shown in Fig. 14 which was testedwith 25 kΩ of resistor load. The experimental waveforms that areexactly matched with the simulation depicted in Fig. 4 validatethe design described in this paper. The result of the measuredoutput voltage ripple shown in Fig. 15 verifies the superiority ofthe proposed three-phase operation.

The efficiency and the power factor were measured by poweranalyzer and Fig. 16 shows 95.5% of maximum efficiency and


Fig. 15. Experimental waveforms of output voltage ripple at CPS-ratedoperation (output voltage ripple: 200 V/div., resonant current: 200 A/div.,switching voltage (VDS ): 500 V/div., 10 μs/div.).

Fig. 16. Measured efficiency and power factor versus output power.

Fig. 17. Measured output voltage and power depending on the switch-ing frequency.

92% of maximum power factor at rated operation. In addition,the control characteristic of output voltage depending on theswitching frequency was measured as shown in Fig. 17. It wasverified that the SPRC provides wide controllable output voltageand load range.

Finally, a preliminary test of the developed CPS for operatingthe 30-kW 95-GHz gyrotron was completed, and the design was

verified with a resistor load. Subsequently, the developed CPSwill need to be tested with the gyrotron to further improve theprototype.

V. CONCLUSION

In this paper, the design and implementation of CPS tooperate a 30-kW 95-GHz gyrotron were described. Depend-ing on given design considerations and specifications, a three-phase SPRC was proposed, and the detailed design and im-plementation were presented to lower output voltage ripplewith low arc energy. Finally, the developed CPS was testedwith a resistor load, and the proof of its superiority included95.5% of maximum efficiency, 92% of maximum power factor,500W/L of power density, and 0.6% of output voltage ripplewith 8.3 J of arc energy. For future research, we intend to im-prove our prototype by testing it with an actual 30-kW 95-GHzgyrotron.

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Sung-Roc Jang was born in Daegu, SouthKorea, in 1983. He received the B.S. degreefrom Kyungpook National University, Daegu, in2008, and the M.S. and Ph.D. degrees in elec-tronic engineering from the University of Scienceand Technology (UST), Daejeon, South Korea,in 2011.

Since 2011, he has been with the KoreaElectrotechnology Research Institute, Chang-won, South Korea, as a Senior Researcher of theElectric Propulsion Research Center. In 2015,

he became an Assistant Professor in the Department of Energy Conver-sion Technology, UST. His current research interests include high-voltageresonant converters and solid-state pulsed power modulators and theirindustrial applications.

Dr. Jang received the Young Scientist Award at the third Euro-AsianPulsed Power Conference in 2010, and the IEEE Nuclear Plasma Sci-ence Society Best Student Paper Award at the IEEE International PulsedPower Conference in 2011.

Jung-Ho Seo was born in Seoul, South Korea, in1986. He received the B.S. degree from Kyung-Hee University, Suwon, South Korea, in 2011.He is currently working toward the M.S. degreein electronic engineering at the University ofScience and Technology, Daejeon, South Korea.

His current research interests include soft-switched resonant converter applications andhigh-voltage pulsed power supply systems andtheir industrial applications.

Hong-Je Ryoo received the B.S., M.S., andPh.D. degrees in electrical engineering fromSungKyunkwan University, Seoul, South Korea,in 1991, 1995, and 2001, respectively.

From 2004 to 2005, he was with WEM-PEC at the University of Wisconsin–Madisonas a Visiting Scholar for his postdoctoral study.Since 1996, he joined the Korea Electrotech-nology Research Institute, Changwon, Korea,where in 2008, he became a Principal Re-search Engineer at the Electric Propulsion

Research Division and a Leader of the Pulsed Power World Class Labo-ratory. He was also a Professor in the Department of Energy ConversionTechnology, University of Science and Technology, Daejeon, South Ko-rea. Since 2015, he has been with the School of Energy Systems Engi-neering, Chung-Ang University, Seoul, where he is currently a Professor.His current research interests include pulsed power systems and theirapplications, as well as high-power and high-voltage conversions.

Prof. Ryoo is a Member of the Korean Institute of Power Electronicsand the Korean Institute of Electrical Engineers.

development of 50-kv 100-kw three-phase resonant converter...

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