design optimization of high frequency transformer for dual
TRANSCRIPT
This is a repository copy of Design Optimization of High Frequency Transformer for Dual Active Bridge DC-DC Converter.
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Version: Accepted Version
Proceedings Paper:Hoang, K.D. orcid.org/0000-0001-7463-9681 and Wang, J. orcid.org/0000-0003-4870-3744 (2012) Design Optimization of High Frequency Transformer for Dual Active Bridge DC-DC Converter. In: XXth International Conference onElectrical Machines (ICEM). XXth International Conference on Electrical Machines, 02 - 05September, 2012, Marseille, France. IEEE , pp. 2311-2317. ISBN 978-1-4673-0142-8
https://doi.org/10.1109/ICElMach.2012.6350205
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Design Optimization of High Frequency Transformer for Dual Active Bridge DC-DC Converter
K. D. Hoang and J. Wang
Abstract -- This paper presents a design optimization
procedure for high frequency transformer (HFT) employed in
bidirectional dual active bridge (DAB) isolated DC-DC
converter. It is shown that leakage inductance, phase-shifted
angle, skin and proximity effects have to be taken into account
together with the HFT voltage-ampere rating to minimize
total losses. It is also demonstrated that the leakage inductance
required for zero voltage switching operation can be realized
under the proposed design procedure without employing extra
inductor. The proposed design methodology is experimentally
validated by measurements on a prototype HFT.
Index Terms--AC losses, converter losses, design
optimization, dual active bridge DC-DC converter, eddy
current effects, high frequency transformer, Litz-wire,
MOSFET, zero voltage switching operation.
I. INTRODUCTION
CENTLY, investigations in high-power DC-DC
conversion have been significantly increased due to its
essential role in electric vehicle applications and
battery based energy storage systems [1], [2]. Among DC-
DC converter topologies, bidirectional dual active bridge
(DAB) isolated DC-DC converter [3], [4] shown in Fig.
1(a) is often considered as a preferred candidate for such
applications because of its advantages on galvanic
isolation, switching loss reduction, electromagnetic
interference (EMI) improvement, and efficiency gain.
Generally, this topology consists of two voltage source
active bridges linked by a galvanically isolated high-
frequency transformer (HFT). Each bridge is controlled to
produce a high-frequency square wave voltage at its
transformer terminals and the power flow from one DC
source to the other is regulated via appropriately controlling
the phase-shifted angle between these two square wave
voltages [4].
It is desirable for a DAB isolated DC-DC converter to
operate at high switching frequency to achieve high-power
density and light-weight. However, it is essential that soft-
switching techniques with zero or low switching losses are
employed. In [4], it was demonstrated that leakage
inductance value of the isolated HFT needs to be carefully
selected for zero voltage switching (ZVS). As leakage
inductance under the HFT design is normally minimized
for conducting loss reduction, additional inductors were
utilized [4] which results in increase in HFT size and
weight.
On the other hand, based on the primary referred
equivalent circuit of the DAB isolated DC-DC converter
shown in Fig. 1(b) [3], it is evident that the power transfer
via the HFT in the DC-DC converter relies on both its
This work was supported in part by the European Commission under
Grant No. 260087. K. D. Hoang and J. Wang are with the Department of Electronic and
Electrical Engineering, The University of Sheffield, Mappin Street, Sheffield, United Kingdom (e-mail: [email protected]).
leakage inductance and the phase-shifted angle between its
terminal square wave voltages. As a result, there is always
a trade-off between the leakage inductance and the phase-
shifted angle when maximizing the power conversion.
Additionally, at high switching frequency, high AC
losses in the HFT windings are inevitable due to the skin-
and proximity-effects [5], [6], and [7]. The leakage
inductance and the magnitude of the AC loss are influenced
by the transformer geometry and winding layout.
Therefore, the leakage inductance, phase-shifted angle, and
the skin and proximity effects need to be taken into
consideration in the HFT design procedure for minimizing
electrical losses.
In [8], [9], [10], [11], and [12], design procedures for
high frequency transformer (HFT) were presented.
However, all these proposed methods start with the core
dimensions being determined based on voltage-ampere
(VA) rating of the transformer. The winding arrangement
based on selected core dimensions is subsequently
suggested. As a result, the influence of leakage inductance
and phase-shifted angle on the design are neglected under
these conventional design methodologies.
R
1V
11S
12S
13S
14S
21 : nn21S
22S
23S
24S
2V1Ti 2Ti
1Bi 2Bi
1Tv 2Tv
XXS
XXD XXCXXM
HFT
(a)
1V
11S
12S
13S
14S
21S
22S
23S
24S
'
2V1Ti
1Bi'
2Bi
1Tv '
2Tv
lkL
HFT
( b)
Fig. 1. DAB DC-DC converter [3]; each switch SXX is implemented by a power MOSFET MXX, a diode DXX, and a snubber capacitor CXX; iB1-V1 and iB2-V2 are, respectively, the primary and secondary DAB current-voltage; iT1-vT1 and iT2-vT2 are, respectively, the primary and secondary HFT current-voltage. (a) Topology. (b) Primary referred equivalent circuit.
This paper presents a design optimization procedure for
HFT which takes into consideration the leakage inductance,
phase-shifted angle, skin and proximity effects for
minimizing total losses of DAB isolated DC-DC converter.
The proposed design method is experimentally validated by
measurements on a prototype HFT.
II. PRINCIPLES OF DAB DC-DC CONVERTER OPERATION
A. Operation of DAB DC-DC Converter
.
1V
1Tv
1V
1V
'
2Tv
'
2V
'
2V'
2V
1Ti
2
t
'
2V
1Bi
'
2Bi
3
1V
'
2V
Fig. 2. Idealized operating waveforms for DAB DC-DC converter [3]; の
is the angular frequency; h is the current zero-crossing angle; and 轄 is the
phase-shifted angle.
TABLE I CONDUCTION MODE OF DAB DC-DC CONVERTER
0
Bridge 1 Bridge 2 Bridge 1 Bridge 2 Bridge 1 Bridge 2
1411D 2322D 1411M 2322M
1411M 2421D
Assuming that the power is transferred from bridge 1 to
bridge 2 as denoted in Fig. 1(b), idealized operating
waveforms and conduction mode of a DAB isolated DC-
DC converter [3] can be illustrated in Fig. 2 and Table I,
respectively. Based on these operating waveforms,
instantaneous value of HFT current, iT1=i(し), can be
computed from the current slope associated with the
conducting modes lkL
VVA
'
21 ;lkL
VVB
'
21 as follows.
For 0 (mode 1)
0)( IAi (1)
2
0
2
0
32
2
132
1IAI
AI rms (2)
0
2
122
1I
AIav (3)
For (mode 2)
)()( Ai (4)
)()(
32
1 2332
2
2
AA
I rms (5)
2
2 )(22
1
A
Iav (6)
For (mode 3)
IBi )()( (7)
))()(()(
32
1 332
2
3 BIBIB
I rms (8)
))(()(
22
1 22
3 BI
BIav (9)
)(2 2
3
2
2
2
11 rmsrmsrmsrmsT IIII (10)
Id
L
VI
lk
)]2([2
10
(11)
1
2
1 dL
VP
lk
O (12)
1
2
1
'
2
V
VN
V
Vd ;
2
1
n
nN ;
A
I0 (13)
where d is the conversion ratio; i(し) is the instantaneous
HFT current at the angular し; I0 and Iヾ is, respectively, the
HFT current at し = 0 and し = ヾ; Iavi and Irmsi is,
respectively, the average and rms values of HFT current
associated with the ith conduction mode; IT1rms is the total
rms value of HFT current; Llk is the leakage inductance; N
is the turn ratio; n1 and n2 is, respectively, the primary turn
number and secondary turn number; PO is the transferred
power; V1 and V2� is, respectively, the primary voltage
value and secondary voltage referred primary.
Zero voltage switching occurs when an active switch in
the DAB is turned on while its anti-parallel diode is free-
wheeling. Thus the condition for ZVS operation is given
by:
00 I ; (14) 0I
B. Conduction and Switching Losses of DAB DC-DC Converter
Due to its fast switching characteristics, MOSFET is
often selected for DAB isolated DC-DC converters.
According to [9] and [13], the conduction and switching
losses of one MOSFET device of the converter can be
expressed as:
For 0 (mode 1)
(15) DoffsFrmsDFavDDpri EfIRIVP 2
1.
(16) DoffsFrmsDFavDD EfINRNIVP 22
1.sec
For (mode 2)
(17) MonsMrmsMMpri EfIRP 2
2.
(18) )(22
2.sec MoffMonsMrmsMM EEfINRP
For (mode 3)
(19) MoffsMrmsMMpri EfIRP 2
3.
(20) 22
3.sec FrmsDFavDD INRNIVP
rrDSonMonDSon
fvri
Mon QVIVtt
E
2
)( (21)
MoffDSoff
firv
Moff IVtt
E2
)( (22)
rrDSonDoff QVE (23)
)(4 3.sec2.sec1.sec3.2.1.. DMDMpriMpriDpriLossS PPPPPPP (24)
where EDoff is the reverse recovery charge energy of free-
wheeling diode; EMon and EMoff is, respectively, the switch-
on and switch-off energy of the MOSFET; fs is the
switching frequency; IFav and IFrms is, respectively, the
average and rms forward current of the free-wheeling
diode; IMon and IMoff is, respectively, the MOSFET current at
on-state and off-state; IMrms is the rms value of the
MOSFET current; PD and PM is, respectively, the switching
losses of the free-wheeling diode and MOSFET; PS.Lossぇ is
the total conduction and switching losses; Qrr is the reverse
recovery charge; RD and RM is, respectively, the free-
wheeling diode and the MOSFET on-state resistance; tri, tfi,
and trv, tfv is, respectively, the MOSFET current and voltage
rise-time and fall-time; VD is the freewheeling diode
forward voltage drop; VDSon and VDSoff is, respectively, the
MOSFET Drain-Source voltage at on-state and off-state.
For completeness, aforementioned definitions for
switching loss calculation are diagrammatically represented
in Fig. 3 [13]. Appropriate currents expressed in (1) to (11)
are employed to compute the converter conduction and
switching losses. It is worth noting that for the ZVS
operation, EDoff and EMon become zero.
)(thVGS
onSwitch offSwitch
fvt
2fvt1fvt
rit
1rvt2rvt
rvt fit
onP offP
Goni
Goffi
GSv GSv
DSvDSv
Doni Doffi
t
rit fvtrrt
rrQ
rrQ
DSv
Fi Doni EffectRecovery Reverse
Fig. 3. Switching losses of power MOSFET [13].
In the next section, the conduction and switching loss
model is utilized for optimizing HFT design with particular
emphasis on minimizing the total losses in the HFT and
DAB.
III. OPTIMIZED HFT DESIGN
The objective of the HFT design is to find the optimal
values of the leakage inductance, phase-shifted angle, core
dimensions, turn ratio, and winding information in order to
minimize the total converter losses, PC.Lossぇ, (conduction
and switching losses, winding losses, and core losses) of
the DAB isolated DC-DC converter. Normally, for high
switching frequency applications, ferrite core should be
selected to eliminate eddy current loss in the core [14]. The
design specification of the HFT is shown in Table II. The
output voltage range from 90V to 140V corresponds to the
variation of battery voltage when the converter is used as a
bidirectional battery charger. The chosen switching device
is a power MOSFET (STW55NM60ND) which is
recommended for bridge topologies, in particular, ZVS
phase-shifted converters [15]. The design process can be
categorized into 4 steps as follows.
TABLE II DESIGN SPECIFICATION OF HFT
Output power 2.2kW
Input voltage 380V
Output voltage 90-140V
Nominal output voltage 120V
Switching frequency 40kHz
Flux density in the core, Bmax 0.25T
Switching device STW55NM60ND [15]
A. Step 1: Optimal Phase-shifted Angle
The phase-shifted angle of the DAB isolated DC-DC
converter varies from 0 to 90 degrees [4], Fig. 2. Therefore,
for the given design specification PO = 2.2kW, V1 = 380V,
and fs = 40kHz, the phase-shifted angle 轄 is varied from 0
to 90 degrees to minimize HFT current associated with a
given output voltage. Since the output voltage varies from
90V to 140V, which corresponds to the conversion ratio d
varies from 0.8 to 1.5. For each value of 轄, the resultant
leakage inductance for the power transfer and the HFT
current are computed, respectively, using (12) and (10).
Fig. 4(a) shows that a phase-shifted angle around 30
degrees can result in minimum HFT current for a full-range
conversion ratio from 0.8 to 1.5. Fig. 4(b) illustrates the
variation of the leakage inductance from 90ȝH to 170ȝH
when d varies from 0.8 to 1.5 for the phase-shifted angle of
30 degrees.
0.8 0.9 1 1.1 1.2 1.3 1.4 1.54
6
8
10
12
Conversion Ratio - d
Cu
rren
t (A
)
100
300
500
700
900
(a) HFT current
0.8 0.9 1 1.1 1.2 1.3 1.4 1.50
100
200
300
400
Conversion Ratio - d
Lea
kag
e In
du
ctan
ce (
H)
100
300
500
700
900
(b) Leakage inductance
Fig. 4. Variation HFT current and leakage inductance with phase-shifted
angle and conversion ratio.
B. Step 2: Optimal Leakage Inductance and Conversion Ratio
In step 2, for a given leakage inductance value between
90ȝH and 170ȝH, PO = 2.2kW, V1 = 380V, and fs = 40kHz,
the variations of phase-shifted angle, HFT current, and
ZVS operation range with the conversion ratio d (0.8~1.5)
are investigated according to (12), (10), and (14).
Fig. 5(a) shows that when the leakage inductance is
around 90ȝH, the HFT current becomes minimum for a
conversion ratio of 1.07. Fig. 5(b) shows that the phase-
shifted angle associated with Llk = 90H and d = 1.07 is
around 22 degrees, whilst Fig. 5(c) illustrates the ZVS
operation range, where �1� indicates ZVS, and �0�
otherwise. It can be seen that d = 1.07 satisfies the ZVS
requirement for the design specification given in Table II.
0.8 0.9 1 1.1 1.2 1.3 1.4 1.56
7
8
9
10
11
Conversion Ratio - d
Cu
rren
t (A
)
90H 110H 130H 150H 170HLlk
(a) HFT current
0.8 0.9 1 1.1 1.2 1.3 1.4 1.50
20
40
60
80
100
Conversion Ratio - d
(d
egre
es)
90H 110H 130H 150H 170HLlk
(b) Phase-shifted angle
0.8 0.9 1 1.1 1.2 1.3 1.4 1.50
1
Conversion Ratio - d
ZV
S C
on
dti
on
90H 110H 130H 150H 170H
ZVS
Llk
(c) ZVS operation range
Fig. 5. Variation HFT current and phase-shifted angle and ZVS range with
leakage inductance and conversion ratio.
C. Step 3: Optimal Core Dimensions
In this step, core dimensions are optimized for minimum
HFT weight and electrical losses with PO = 2.2kW; Llk =
90ȝH; d = 1.07; and 轄 = 22 degrees.
Assuming the chosen core is an EE ferrite core with
dimensions shown in Fig. 6, the leakage inductance and
maximum flux density, Bmax, can be computed by [9]:
b
hMLTnLlk
2
103
1 (25)
1
1max
4 nfA
VBB
score
(26)
where Acore is the core area; MLT is the mean-length per
turn which is estimated by assuming that bobbin occupies
20% of window area.
;adA Tcore )2(8.0)(2 bdaMLT T (27)
It is worth noting that the top-bottom winding
arrangement is chosen for the designed HFT in order to
maximize the attainable leakage inductance [16].
a2
a
2
a
h
a bb2
a
2
a
MLT
Bobbin
b8.0
Winding WindingCoreCore Core
Td
Td
Fig. 6. Core dimensions used in HFT design investigation.
By defining two factors K1 and K2 as
a
dK T1 ;
b
hK 2 (28)
Substituting (28) into (25) and (26) gives
(29) 1max
2
11 4 VfBaKn s
(30) lkLMLTKn 320
2
1 In addition, the relation between the core window area,
Aw, and the primary conductor area [9] can be expressed as
max
11
2
2
2
J
I
knbKA rmsT
f
w (31)
where kf is the filling factor chosen as 0.3 for Litz-wire and
Jmax is the current density given as 5.5 A.m-2 according to
the thermal dissipation [9].
Equations (29), (30), and (31) can be used to study the
optimized core dimensions for minimizing both core
weight, CW, and winding weight, WW.
(32) ]2))((2[ 2
221 bKbKabaaKCC FeVFeW MLTAkW WfCuW (33)
where とFe and とCu is, respectively, the ferrite mass density
and copper mass density; CV is the core volume.
5 10 15 20 25 30 35 400
100
200
300
400
500
a (mm)
Wei
gh
t (g
)
CW
-K1:0.5
WW
-K1:0.5
TW
-K1:0.5
CW
-K1:3
WW
-K1:3
TW
-K1:3
Fig. 7. Variations of HFT weight with a and K1.
Fig. 7 shows the variations of the core weight, the
winding weight, and the total HFT weight (TW = CW + WW)
with a for K1 = 0.5 and 3. As can be seen, minimum total
HFT weight can be achieved when a is between 10mm and
23mm. Fig. 8 shows variations of the other core
dimensions, b and K2, the mean length per turn, MLT, and
the number of turns of primary winding n1 with a and K1.
When an optimal a for a given K1 which minimize the total
HFT losses can be determined, then the other dimensions,
MLT and n1 are found from Fig. 8.
8 14 20 26 320
10
20
30
40
50
a (mm)
b (
mm
)
0.5
1
1.5
2
2.5
3
K1
(a)
8 14 20 26 320
2
4
6
8
10
a (mm)
K2
0.5
1
1.5
2
2.5
3
K1
(b)
8 14 20 26 320
50
100
150
200
a (mm)
ML
T (
mm
)
0.5 1 1.5 2 2.5 3K1
(c)
8 14 20 26 320
50
100
150
a (mm)
n1 (
turn
s)
0.5 1 1.5 2 2.5 3K1
(d)
Fig. 8. Variations of core dimensions, MLT and n1 with a.
Further, optimal round Litz-wire design needs to be
determined to minimize the total winding losses. According
to [17] and [18], optimal number of Litz-wire strands
should be chosen to have its AC resistance, RAC, being
around 2 times of its DC resistance. This optimal AC
resistance in per unit length can be computed as follows:
(34) 3535.05113.14128.2 2 cc dd
s
sf
075.0 ;
2
12 42
11
0
sbbd
(35)
2
2
222
1 )24
116(24
cdmb
(36)
4
2
20
2
)24
116(24
1
34
m
n (37)
5
14
3
812
1
3
1
2
1
3
1122)(
(38)
7
12
3
522
1
2
1
2
1)( (39)
)]()24
116(24
)([2
22
20
2
1
00
1
m
n
dn
nR
s
CuAC (40)
where dc is the conductor diameter [9] computed from core
window dimensions, HFT current, and current density; d0 is
the optimized strand diameter; m is the number of winding
layers; n0 is the number of strands per Litz-wire conductor;
く is the Litz-wire packing factor; hs is the skin depth.
8 14 20 26 320
10
20
30
a (mm)
HF
T L
oss
es (
W)
0.5 1 1.5 2 2.5 3K1
(a) Total HFT losses
8 14 20 26 3240
80
120
160
200
a (mm)
Tem
per
atu
re (
0C
)
1 1.5 2 2.5 3K1
(b) Temperature
Fig. 9. Variation of HFT losses and temperature with a and K1.
Based on the AC loss model of (40), the variations of
total HFT losses, including DC and AC losses and core loss
with the core dimension a and K1 is shown in Fig. 9(a). The
resultant temperature rise [9] can be determined as follows:
])15.273()15.273[(1.5
1044
8
,
asCR
radTTA
TR (41)
25.0
25.0
,)(34.1
)(
TA
dR
CR
vertconv
;convrad
convrad
saRR
RRR
,,
,,
,
(42)
CAC
sa
HFT PPR
TP
,
; ;2
1rmsTACAC IRP VVC CPP (43)
(44)
abbbKabKba
abKaKbaaKACR
422))((4
)(2)(4
22
22
211
where ACR is the crossectional area; dvert is the HFT vertical
height; PAC is the winding loss; PC and PV is, respectively,
the core loss and the relative core loss; PHFTぇ is the total
HFT losses; Rし,conv and Rし,rad are the thermal resistances due
to convective and radiative heat transfer, respectively; Rし,sa
is the total thermal resistance; Ta and Ts is the ambient and
surface temperature in 0C, respectively; ∆T = Ts - Ta.
Fig. 9(b) shows the resultant temperature variations with
a and K1 assuming Ta = 250C. From Fig. 9(b), it can be seen that the surface temperature is minimized when a is between 14mm to 17mm. This results together with the relevant K1 presented in Fig. 9(a) and the minimum MLT value shown in Fig. 8(c) are utilized to determine the suitable ferrite core (E55/28/21). Table III compares the dimensions of optimized design and that of the selected core. It is evident that the proposed design optimization procedure leads to appropriate selection of available cores.
TABLE III COMPARISON BETWEEN OPTIMIZED DESIGN AND CHOSEN CORE
Optimized design E55/28/21 (N87)
a = 17 (mm), K1 = 1.25 a = 17.2 (mm), K1 = 1.22
MLT = 112 (mm) MLT = 124 (mm)
K2 = 2.78 K2 = 3.645
b = 8.63 (mm) b = 10.1 (mm)
n1 = 27 (turns) n1 = 27 (turns)
D. Step 4: Optimal Number of Turns of the Secondary Winding
Fig. 10 shows the numerical results of the total converter
losses (PC.Lossぇ = PS.Lossぇ + PHFTぇ) with the output voltage
and transformer turn ratio when d = 1.07, Llk = 90ȝH
realized with E55/28/21 core; 轄 = 22 degrees, and PO =
2200W. For the purpose of illustration, ZVS range at 90V
(dashed blue line with crosses) and 140V (dashed grey line
with circles) are also shown. As will be seen, the total
losses decrease with the increases in turn ratio, but ZVS
operation at 140V is lost when the turn ratio is greater than
3.4. Thus the optimal turn ratio for minimum total
converter losses and full ZVS operation over the 90V-140V
output voltage range is 3.4. The resultant number of turns
of the secondary winding is n2 = 8. Fig. 11 shows the
variations of minimum leakage inductance for ZVS
operation according to (14) with the output voltage for PO =
2200W and 500W with n2 = 8. It is obvious that ZVS
operation is achieved for full range output voltage at PO =
2200W with Llk = 90ȝH. At lower output power, modified
modulation techniques should be considered [2] to facilitate
ZVS operation.
2.8 3 3.2 3.4 3.6 3.8 4 4.260
80
100
120
140
160
180
To
tal
Lo
sses
(W
)
2.8 3 3.2 3.4 3.6 3.8 4 4.20
1
Turn Ratio
ZV
S
2.8 3 3.2 3.4 3.6 3.8 4 4.20
1
2.8 3 3.2 3.4 3.6 3.8 4 4.20
1
140V
115V
90V
ZVS
Fig. 10. Variation of total converter losses and ZVS range with turn ratio
and output voltage.
90 100 110 120 130 1400
100
200
300
400
Output Voltage (V)
Lea
kag
e In
du
ctan
ce (
H)
PO
=2200W
PO
=500W
Non-ZVS region
ZVS region
Non-ZVS regionZVS region
Fig. 11. Variation of minimum leakage inductance for ZVS with output
voltage for PO = 2200W and 500W with n2 = 8.
IV. PROTOTYPE AND MEASUREMENTS
The outcomes of the proposed HFT design optimization,
which is realized with core E55/28/21, are summarized in
Table IV. It is worth noting that leakage inductance
computed from (25) should be rescaled by a factor 0.7 due
to three-dimensional and high frequency effects [19]. The
winding arrangement for the HFT design is shown in Fig.
12(a). A prototype transformer has been constructed as per
the optimal design, Fig. 12(b). The magnetizing and
leakage inductances seen on both the primary and
secondary sides are measured using an impedance analyzer.
Fig. 13 shows the variations of resultant total converter
losses with output voltage and power. As can be seen, the
maximum total losses occur at 90V and PO = 2200W is
about 6.59% of the rated output power. At the nominal
output voltage, the total losses are reduced to less than 5%.
The measurement results are shown in Fig. 14 and validate
the predicted magnetizing and leakage inductance, Table
IV. This implies that ZVS operation can be ensured with
the proposed design method without employing extra
inductor.
TABLE IV OPTIMIZED DESIGN OF HFT
Bmax 0.24848 (T)
Primary winding 27 turns, 162 strads-AWG38
Secondary winding 8 turns, 300 strands-AWG35
Magnetizing inductance Pri./Sec. 5.231/0.46 (mH)
Leakage inductance Pri./Sec. 90/10.5 (ȝH)
(a) (b)
Fig. 12. HFT design. (a) Winding arrangement. (b) Prototype.
90 100 110 120 130 1400
40
80
120
160
Output Voltage (V)
To
tal
Lo
sses
(W
)
2200W 1500W 1000W 500WPO
Fig. 13. Variation of total converter losses with output voltage and power.
V. CONCLUSION
In this paper, a design optimization procedure for the
HFT employed in DAB isolated DC-DC converter has been
described and experimentally validated. It has been shown
that by considering leakage inductance, phase-shifted angle
together with HFT VA rating, essential design equations
for minimizing total losses of the DAB isolated DC-DC
converter can be derived. It also has been demonstrated that
leakage inductance requirement for ZVS operation can be
achieved under the proposed design method without
employing an extra inductor. Comparative study of DAB
HFT design with conventional and the proposed design
methods will be performed in the near future.
(a) Primary magnetizing inductance
(b) Primary leakage inductance
(c) Secondary magnetizing inductance
(d) Secondary leakage inductance
Fig. 14. Measured inductances of prototype HFT.
VI. ACKNOWLEDGMENT
The authors gratefully acknowledge the European
Commission for financial support and the P-MOB project
partners for permission of the publication of this paper.
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VIII. BIOGRAPHIES
Khoa Dang Hoang (S�10) received the B.Eng. and M.Eng. degrees from Ho Chi Minh City University of Technology, Ho Chi Minh City, Vietnam, in 2002 and 2005, respectively, and the Ph.D. degree from the University of Sheffield, Sheffield, U.K., in 2011, all in electrical and electronics engineering. He is currently working as a post-doctoral Research Associate at the University of Sheffield. His key research interests include power conversion and advanced control techniques for electrical drives.
Jiabin Wang (SM�03) received the B.Eng. and M.Eng. degrees from Jiangsu University of Science and Technology, Zhengjiang, China, in 1982 and 1986, respectively, and the Ph.D. degree from the University of East London, London, U.K., in 1996, all in electrical and electronic engineering. Currently, he is a Professor in Electrical Engineering at the University of Sheffield, Sheffield, U.K. His research interests range from motion control to electromagnetic devices and their associated drives in applications ranging from automotive, household appliances to aerospace sectors.