design of a multi-mhz resonant driver chip for high-voltage d ......wouter faelens d-mode gan hemts...

Wouter Faelens

D-mode GaN HEMTsDesign of a multi-MHz resonant driver chip for high-voltage

Academic year 2017-2018Faculty of Engineering and ArchitectureChair: Prof. dr. ir. Koen De BosschereDepartment of Electronics and Information Systems

Master of Science in Electrical EngineeringMaster's dissertation submitted in order to obtain the academic degree of

Supervisors: Prof. dr. ir. Jan Doutreloigne, Dr. ir. Pieter Bauwens

Permission of use on loan

”The author gives permission to make this master dissertation available for consultation and to

copy parts of this master dissertation for personal use.

In the case of any other use, the copyright terms have to be respected, in particular with regard to

the obligation to state expressly the source when quoting results from this master dissertation.”

January 2018

Preface

First of all I would like to thank my supervisor Jan Doutreloigne for advising me throughout

this year. He challenged me to really dig into the matter and understand it at a fundamental

level. He also helped me along when I was stuck on certain problems, and provided helpful

suggestions where needed. Lastly, I enjoyed his work ethic greatly: doing what needs to be

done, with simple down to earth methods and explanations, without the need for rigid official

systems.

Secondly, I would like to thank all my friends for giving me joy and energy. First off my

flatmates Jasper and Lukas, for giving spice to my life and putting up with my lesser sides.

Secondly, thank you Wouter and Rori for inspiring me to work hard both on my thesis as in

general, as well as thinking critically about life and society. Lastly, I would like to thank ’de

duivenmelkers’ for giving me so many good times in life, both as gaming buddies as well as in

real life.

Finally, I would like to thank my family in supporting me for all this time. Thank you

mama for showing me the joys in life, like travelling, and supporting me no matter what. Thank

you papa for driving me to not settle for good enough but strive for perfection. Challenging,

encouraging and helping me to give the best of myself, while still keeping in mind what is really

important in life. Thank you Femke for giving me joy and being awesome, you are the one I

would turn to in hard times. Thank you Ruben for being my ’big brother’: a living example of

things to achieve in life. Lastly, thank you Linde for reminding me why we strive everyday to

make this world a better place. Special thanks to my dad and Ruben for giving suggestions on

this thesis, improving it so many times over, and elevating it to something I am proud of!

Wouter Faelens, janauri 2018

Design of a multi-MHz resonant driver chip

for high-voltage D-mode GaN HEMTsWouter Faelens

Master’s dissertation submitted in order to obtain the academic degree of

Master of Science in Electrical Engineering

Academic year 2017-2018

Supervisors: Prof. dr. ir. Jan Doutreloigne, Dr. ir. Pieter Bauwens

Faculty of Engineering and Architecture

Ghent University

Department of Electronics and Information Systems

Chair: Prof. dr. ir. Koen De Bosschere

Abstract

Depletion mode (D-mode) gallium nitride (GaN) High Electron Mobility Transistors (HEMT)

can be used as the switching element in boost converters, a type of switched-mode power supply

(SMPS). Charging the gate capacitance of the switching element at high frequency provides

an important contribution to the power losses in this application. By using an inductor and

recovery path to recycle this energy, the power efficiency can be improved.

To evaluate the power efficiency, a resonant and a direct driver are designed that drive the

HEMT directly at its gate. We also incorporate an internal negative voltage generator so the

driver can be controlled via positive signals and a single positive voltage supply.

Through simulation, the power loss and efficiency of the direct and resonant driver are compared

under different parameter values: duty cycle (0.2 - 0.9), switching period (200 - 1000 ns), resonant

inductance (1 - 10 µH ), diode types (Schottky vs normal diodes), booster supply voltage (1 -50 V) and load impedance (5 - 500 Ω).

By driving resonantly, driver power losses can be lowered significantly, especially at higher

frequencies. At 5MHZ switching frequency, the resonant driver consumes 89.6mW while the

direct driver consumes 197.4mW. The boost converter with the highest efficiency has an efficiency

of 84%, 2% more than in direct drive with identical parameters. Typical efficiency increases for

most parameters are around 1% - 2%. The largest efficiency increase comparing resonant and

direct drive is an improvement of 17% (59.5% versus 42.5%).

The simulation results show that a resonant driver is more efficient than a direct driver. This

result is robust under varying parameter values. However, a resonant driver requires extra off-

chip components, making it more costly to produce. It will depend on the application if the

benefits outweigh the costs.

Design of a multi-MHz resonant driver chip forhigh-voltage D-mode GaN HEMTs

Wouter Faelens

Supervisor(s): Prof. dr. ir. Jan Doutreloigne, Dr. ir. Pieter Bauwens

Abstract—This research tests the efficiency improvement of driving a D-mode GaN HEMT in a resonant way. It does this by designing a resonantdriver and direct driver and comparing their efficiency under different pa-rameters via simulations. The proposed resonant driver is predicted to havea higher power efficiency than a direct driver for any switching frequency,booster voltage, load, and for duty cycles ¡ 0.5. Efficiency increases of upto 17% were shown (42.5% to 59.5%), although typical efficiency increaseswere around 1% - 2%. Driver power losses can be reduced up to 50%, espe-cially at higher frequencies: 89.6mW resonant driver loss versus 197.4mWin the direct driver, this at 5MHz switching frequency. However, a resonantdriver requires extra off-chip components, making it more costly to pro-duce. It will depend on the application if the benefits outweigh the costs.

Keywords—Depletion mode, GaN HEMT, Switched-mode power supply(SMPS), resonant driver

I. INTRODUCTION

EVERY electronic circuit needs power. This power can bedirectly drawn from the source like batteries or a poweroutlet, but more often than not a special power circuit is usedto convert the incoming power to a more suitable power for therest of the circuit. In an electric car for example, this could betransforming a low battery voltage towards a high voltage usableby the motor.

One of the most common power converters are switched modepower supplies (SMPS). These power supplies utilize a switch-ing element to convert power efficiently. Instead of using a lin-ear amplifier, the pass transistor of a switching- mode supplycontinually switches between low-dissipation full-on and full-off states, and spends very little time in the high dissipationtransitions, which minimizes wasted energy. Voltage regulationis achieved by varying the ratio of on-to-off time, called dutycycle. In contrast, a linear power supply regulates the outputvoltage by continually dissipating power in the pass transistor.

In this abstract, we will use a GaN HEMT as the switchingelement. This switching element needs a driver to turn it on andoff. This driver consumes some power, lowering the efficiencyof the total system. In this paper, we aim to lower the power con-sumption of this driver by recycling some energy via a resonantcircuit.

II. CONCEPTS

A. Boost converter

One of the main power converters in use is a boost converter.This is a DC/DC power converter that boosts a low voltage to-wards a higher voltage. A circuit of a typical boost convertercan be seen in figure 1. When the transistor is on, the inductorexperiences a positive voltage and current is built up. After ashort while we turn the transistor off, and the inductor current isforced through the diode towards the load. This current charges

the output capacitor to a high voltage. Because of this high volt-age, the inductor experiences a negative voltage and the currentgoes down again. Charges are pushed towards the high-voltageload until either the transistor turns on again (continuous mode)or the inductor current falls to zero (discontinuous mode). Thecharging cycle can repeat again.

While the transistor is off and the inductor current builds up,the output load is powered by the output capacitor. The outputvoltage goes down during this time. This means the output volt-age goes down slightly when the transistor is on, and the voltagegoes up again when it is off. The larger the output capacitance,the less this voltage will bounce up and down.

The faster the transistor switches, the smaller the inductor andoutput capacitor can be. A faster switching time means the in-ductor has less time to build up its current, with a lower averagecurrent as a result. This means the booster circuit will lose lesspower due to conduction losses. On the other hand, if we doublethe switching frequency, we can halve the inductor value with-out increasing our average current. The output capacitance canbe lower as well, because it needs to power the load for a shorteramount of time. Lower inductance and capacitor values have theadvantage of being cheaper to produce on an integrated circuit(IC), and having lower parasitics.

In order to switch this transistor on and off, we need a driver.This driver consumes some power that is wasted, reducing theefficiency of our system. In this article, we try to minimizethis driver power loss. We do this in two ways. First, we usea GaN HEMT (Gallium-Nitride high-electron-mobility transis-tor). This is a type of transistor with very low parasitic capac-itance. This means we do not need a lot of charge to turn thetransistor on or off, and hence can switch it faster with an equalamount of current. This also means we can improve the switch-ing frequency. Secondly, we use a resonant circuit inside thedriver. This is a type of circuit using an inductor to recycle someof the energy used to charge and discharge the gate voltage ofthe transistor.

Fig. 1. Circuit boost converter

B. GaN HEMT

GaN is a wide band gap semiconductor (band gap energyEg=3.44eV). This wide band gap translates into an ability tohandle high internal electric fields before electronic breakdownoccurs.

GaN is particularly interesting because of its ability to formheterojunctions with wider band gap semiconductors such asaluminium gallium nitride(up to 6.2 eV). These heterojunctionsresult in the forming of a 2-dimensional-electron-gas (2DEG) atthe interface due to large polarisation in the material which pro-vides a highly dense, majority carrier channel with a large elec-tron [1]. Due to the 2DEG, GaN offers a much larger currentdensity than other materials. This makes it so the gate can besmaller, resulting in lower gate capacitance while still maintain-ing a great on-resistance. This also means there is less chargeneeded for switching, resulting in lower switching losses in thedriver and faster switching.

Due to the presence of this 2DEG, we have a normally-ondevice, or a depletion mode FET (D-mode FET). This meanswe have to apply a negative gate-source voltage in order toturn the HEMT off. Enhancement-mode GaN HEMTs havebeen demonstrated, but their performance is worse than D-modeHEMTS.

There are two ways to generate a negative gate-source voltagein order to turn the HEMT off. First, we can tie the gate toground and raise the voltage of the source. We do this by placinga MOSFET at the source of the HEMT. When we now switchthis MOSFET, we can raise or lower the voltage at the source ofthe HEMT, effectively turning it off and on. The MOSFET doesnot need to be able to handle high voltages, since it only raisesits drain (the HEMT source) high enough until the HEMT stopsconducting and is off. The advantage of driving it this way is thatwe can use conventional gate drivers to drive this MOSFET. Thedisadvantage is that we need a large MOSFET with low enoughon-resistance in order to have good efficiency. This also meanslarger parasitic capacitances and bigger switching losses.

The second way to drive the HEMT is to tie the source of theHEMT to ground and put a negative voltage on its gate. This ismuch more efficient, but the driver will need to be able to useand generate negative voltages to be able to do this. Because itis more efficient, we choose to use this driving scheme.

In this thesis, we use a MGG1T0617T. This D-mode GaNHEMT has a threshold voltage of -3V and low parasitic capac-itances. It can operate at drain-source voltages of up to 600V,meaning we can use this HEMT in high-voltage applications.

C. Resonant driver

The second way we try to improve efficiency is by using aresonant driver [2]. The circuit of the used resonant driver canbe seen in figure 2. When we charge the gate voltage by turningMDR1 on, we also send a current through the inductor. This cur-rent keeps building up, until the gate voltage reaches the uppervoltage. The voltage will not go higher because it will drainaway via the upper diode. The current will now free-wheelthrough MDR1-L-DDR1. When we turn MDR1 off, the currentin the inductor wants to keep flowing. The only way this currentcan flow, is via MDR2-L-DDR1. This is effectively pumping

current from a low voltage towards a high voltage, recoveringthe energy stored in the inductor and destroying its built-up mag-netic fields, lowering the current through it. After some time, thecurrent will have died out and the driving circuit is in rest.

Once we want to shut the transistor off again, we set MDR2on and a current starts flowing from the gate, through L towardsthe low voltage. This charges L again, transferring the energystored on the gate capacitance to the inductor. Once the gate ca-pacitance has discharged completely, the inductor current free-wheels again via DDR2-L-MDR2. When we now shut MDR2off, all the energy stored in the inductor will be used to pushcharge from a low voltage to a high (recuperating energy) viaDDR2-L-MDR1.

This way, we can recover a lot of charges otherwise dissipatedin the circuit, and improve the driver efficiency.

Fig. 2. Circuit resonant driver

III. MAKING THE CIRCUIT

In order to drive the GaN HEMT directly at its gate in a res-onant way, we need a few things. First, we need a negativevoltage to put on the gate. Secondly, we need to translate the ex-ternal positive control signals into negative control signals thatwe can use to drive the HEMT. Thirdly, we need to design theresonant circuit itself. Lastly, we build a boost converter thatuses the HEMT driver in order to test its working and efficiency.

We start with generating negative voltages. We need two neg-ative voltages: a small one to drive the first level shifter of -3V,and a second one to drive a second level shifter and provide thenegative voltage for the gate itself at -5V. This second generatorneeds to be much stronger. We construct two Dickson chargepumps [3], which generate output voltages of -3V and -5V.

The negative charge pumps need a clock signal in order towork. We construct an astable multivibrator [4] to generatethis clock signal on-circuit. Because the dickson charge pump’svoltage output depends on the load, we design some additionalfunctionality. We put OR gates at the input of the charge pumpclocks with as input the original clock signal and an OK sig-nal. This way, when the OK signal is low the clock just passesthrough. When the OK signal is high however, the output of theOR gate will remain high and no pumping action will occur.

The OK signals are generated by two simple opamp compara-tors. When the -3V voltage is low enough, we output a highV3 OK signal, and when the -5V is low enough, we output a

high V5 OK signal. This way, the generators are shut downwhen their output is low enough, and they will stop consumingpower.

Next come the level shifters. These shift the positive controlsignals coming into the driver towards negative voltage signalsusable to drive the DMOS (double-diffused MOS) in the reso-nant circuit. DMOS are a type of MOSFET that has an increasedtolerance for high voltages on the drain. The gate-source volt-age on the other hand has to remain at lower voltage in order forthe DMOS to not break. This is why we use two level shifters:one to drive the upper pDMOS going from -3V to ground and asecond one driving the lower nDMOS from -5V to -2V.

These level shifters power the DMOS in the resonant cir-cuit. The size of these DMOS should be chosen large enoughso that they can provide enough current to charge the HEMTgate quickly. Sizing them too large however enlarges their par-asitics, making it slower and less power efficient to drive theseDMOS.

Then comes the energy recovering element itself: the induc-tor. This also needs to be sized according to the desired switch-ing frequency. The larger the inductor, the more energy it canrecover but the slower the HEMT can switch. This not onlylowers the possible switching frequencies, but can also lowerthe efficiency of the total system by increasing the power lossover the HEMT. When the HEMT is switched slowly, its hardswitching losses increase. This means it dissipates a lot of powerin the time where the drain-source voltage is increasing but the(large) current still flows through the HEMT. This happens whenthe gate voltage is slightly above the switching voltage of theHEMT, or slightly above -3V for the used MGG1T0617T. Wecan conclude that a balance needs to be found in the sizing ofthis inductor: big enough to have enough energy recovery, butnot too large so we maintain sufficient switching speed.

As recovering elements we use schottky diodes. These have alow voltage drop and have very low recovery charges, meaningthey can react very quickly to changing currents.

IV. SIMULATION RESULTS

The efficiency improvement of driving the HEMT in a reso-nant way will now be tested. We do this by comparing the powerconsumption and efficiency of the resonant driver with a conven-tional direct driver. This direct driver is built by removing theresonant circuit in the previously built driver.

We will vary different design parameters and test their influ-ence on the power consumption and efficiency. The parameterswe will vary are the duty cycle D (0.2 - 0.9), switching periodTin (200 - 1000 ns), resonant inductance L (1 - 10 µH ), diodetypes (Schottky vs normal diodes), booster supply voltage Vb(1 - 50 V) and load impedance RL(5 - 500 Ω). Unless other-wise mentioned, the base parameters used in the simulations aregiven in table I

A. Base case

We start by simulating the driver with default parameters toestablish a base performance for our system. A waveform us-ing the default parameters can be seen in figure 3. We noticethat the real duty cycle is higher in both drivers: the HEMTstays on 50ns longer than we expect from the input duty cycle.

TABLE IDEFAULT VALUES FOR THE MAIN DESIGN PARAMETERS

Width clock buffer 200 µmCpump -3V generator 100 pFCpump -5V generator 400 pFSwitching period 500 ns

Pulse width to turn on 80 nsPulse width to turn off 85 ns

Duty Cycle 0.4nDMOS transistor width 200 µmpDMOS transistor width 400 µm

Resonant inductor 2 µ HInput voltage boost converter 3.3 V

output Rload 100 Ω

This is due to the non-linear parasitic capacitance of the usedHEMT. Once the HEMT is on, the parasitic capacitances aremuch larger. This results in most of the switching time beingspent above threshold voltage, for both turn-on and turn-off.

We see that output voltage, booster input power and boosteroutput power are about the same for both drivers : 5.9V, 400mWand 350mW respectively. The driver power consumption how-ever is lower in the resonant case: 68mW versus 78mW in thedirect driver. This results in an efficiency increase of 0.8%, from73.5% in direct drive to 74.3% with resonant drive.

Fig. 3. Waveform of a typical simulation with default parameter values.

B. Duty cyckle

The first parameter we change is the duty cycle, since thishas a big influence on the boost converter performance. We no-tice a few effects for both drivers. First, as expected, the outputvoltage goes up with higher duty cycles. Secondly, the effectiveduty cycle is larger than the theoretical, due to the HEMT stay-ing longer above threshold voltage when switching due to itsnon-linear parasitic capacitance. Lastly, varying the duty cycledoes not change the driver power consumption.

For input duty cycles below 0.5, the resonant driver is moreefficient: 70.5% versus 73% at a duty cycle of 0.3. At low dutycycles below 0.5 the output power is relatively low meaning thepower saving in the resonant driver account for a substantial ef-ficiency increase. For duty cycle above 0.5, the direct driver is

more efficient. This is because the direct driver switches faster,which results in less hard switching losses in the HEMT.

C. Input period

The second thing we vary is the input period. The result ofvarying this from 200ns - 1µs can be seen in figure 4. Twoclear trends are found. First off, the driver power consumptiongoes up with faster switching periods. This is normal, since theHEMT gate will have to be charged and discharged more of-ten, generating higher losses in the circuit. At an input periodof 200ns, a possible reduction of more than 50% (89.6mW ver-sus 197.4mW) was found. Secondly, we confirm that a higherresonance inductor leads to more energy recovery, especially atfaster switching speeds. At an input period of 600ns, an effi-ciency increase up to 5.5% was found: 74.8% efficiency in di-rect drive and 80.3% in resonant drive.

Fig. 4. Driver power versus switching period for different inductance values.L=0 is the direct drive scheme. duty cycle=0,5.

D. Resonant inductance

Third, we vary the value of the resonance inductor. Very lowvalues (≈ 1 µH) resemble the direct drive, both in behaviourand in power consumption. Higher inductance (≈ 5 µH) val-ues work as intended, and too high inductor values (≈ 10 µH)pose problems. These problems include the hard switching ef-fect discussed earlier. The higher the inductor, the better thedriver power consumption. This does not mean total efficiencyhowever, since very high inductor values lead to slower switch-ing speeds, and thus the GaN HEMT consuming more powerdue to this hard switching.

E. diode types

In the design we used schottky diodes, which are not availablein every technology. We now test the influence if we switchout the schottky diodes for standard pn-junctions, which have ahigher voltage drop of 0.6V.

The first schottky diode is found in the resonant circuit. Ifwe switch out this diode to a normal diode with a higher volt-age drop, the driver power consumption only increases by 1mW (56.36mW versus 57.34). This does not seem like a bigdeal. This is however simulated with a very simple diode model.When we use a more complicated model which also modelstransient diode behaviours like reverse recovery charge and -time, we see that a normal diode is not fast enough to work as a

recovery diode. In conclusion, the voltage drop of the diode isnot that important, but its transient behaviour is.

Secondly, we change out the flywheel schottky diode in theboost converter. Doing this leads to an efficiency decrease of5%: from 74.4% down to 69.2%. Since all the output current hasto go through this flywheel diode, the voltage drop has a largeinfluence on the total power loss and thus efficiency. In realitythe driver efficiency will be even worse since the reverse recov-ery time of normal diodes is much higher than those of schottkydiodes, resulting in even bigger losses. Some boost convertershave been shown that do not use flywheel diodes anymore toeliminate these losses.

F. boost converter supply voltage

The driver was designed for and needs a supply voltage of3.3V to work. The boost converter on the other hand can useany voltage it wants. We vary the voltage from 1 to 50V and testthe working of the driver.

We notice that the higher the input voltage, the higher theefficiency. This is because higher input voltages lead to higheroutput voltages, and thus more output power. This also meansthat the driver losses are a bigger portion of the total loss whenthe boost converter input voltage is low. This is why the resonantdriver is more efficient than the direct driver for input voltagesbelow 5V: the power saving in the resonant driver account fora substantial efficiency increase at low voltage. At 2V input,the efficiencies for direct and resonant drive are 53.3% and 50%respectively. At higher voltages, the booster power consumptioninfluences the efficiency the most. Due to the slower switchingspeed of the resonant driver, it has more hard switching lossesin the boost converter and its efficiency is lower. At 10V, theresonant driver has an efficiency of 86.29% and the direct driverone of 88.3%.

The second thing we note is that the driver works up to verylow voltages, but breaks down at very high voltages. At an in-put voltage of 50V for example, the boost converter only pumpsthe output to 60V. Some modifications to the circuit can be doneto let the driver work at these high voltages, albeit at some effi-ciency costs.

G. load impedance

Lastly, the default driver can effectively power loads from20Ω to 500Ω. Loads above 500Ω provide problems becausethey require very large output voltages. The same type of driveradjustments as for high input voltages can be made, since theorigin of the problem is the same: a large output voltage.

When we compare a resonant and direct driver for differentoutput loads, we see that the resonant driver is more efficient athigher output loads, while the direct driver is more efficient atlower load resistances. This once again boils down to the res-onant driver consuming less power but being more susceptibleto losses in the HEMT due to hard switching behaviour. Whenthe booster current is high for low output loads, the direct driveris more efficient due to the HEMT dissipating less power. Oncethe output load is 100Ω or higher, the output power becomeslower and the power driver consumption becomes more impor-tant,resulting in the resonant driver being more efficient, up to11% at an output load of 500Ω.

H. Best achieved results

H.1 Best absolute efficiency

The boost converter with the highest efficiency at an inputvoltage of 3.3V has an efficiency of 84% with a resonant driver,versus 82% in direct drive. This was in a boost converter switch-ing at 1MHz and a duty cycle of 0.5. The resonant inductor usedwas 5µH and the output load 100Ω. At this efficiency, we havean output voltage of 7V, more than two times the input voltage.

When this driver would be designed into real commercialproducts, the efficiency could be much higher. The goal of thisthesis was to test the efficiency increase of resonantly driving aGaN HEMT. We did not focus our efforts on creating a driverwith the best possible efficiency, since this was not the goal ofthis research.

H.2 Best efficiency increase

The strength of resonant drivers lies in their low driver powerconsumption. We tested the best efficiency increase of the res-onant driver in ideal circumstances. We found this to be at aninput voltage of 3.3V, switching period of 400ns, duty cycle of0.25, output load of 500Ω and inductance value of 5µH.

These parameters result in a direct driver with an efficiency of42.5% and a resonant driver with an efficiency of 59.5%: an ef-ficiency increase of 17%! The output voltage with the direct andresonant driver is respectively 6.25V and 7.85V. This proves thatthe resonant driver can deliver a significant efficiency improve-ment in certain applications.

V. CONCLUSION

The goal of this research was to test the possibility and/or ef-ficiency increase of driving a GaN D-mode HEMT in a resonantway. The simulation results show that a resonant driver is moreefficient than a direct driver. This result is robust under varyingparameter values. Resonant driving scheme is possible and im-proves efficiency by typically 1%, although improvements of upto 17% were demonstrated. However, a resonant driver sacri-fices driver simplicity and flexibility, and requires extra off-chipcomponents, making it more costly to produce. It will dependon the application if the benefits outweigh the costs.

The designed driver can be used for a very wide variety ofboost converter input voltages, but the driver itself can only bepowered by 3,3V. The load resistance can vary from 10Ω to500Ω, depending on the application. The designed driver canswitch frequencies up to 5 MHz. Bigger resonance inductor val-ues lead to more energy recovery, and thus lower driver powerconsumption.

ACKNOWLEDGMENTS

I would like to acknowledge the help of my supervisors, giv-ing great suggestions and coaching. I would also like to thankmy family for the never ending support, and my friends for giv-ing me energy and happiness in my life. Special thanks goesout to my dad and brother for the continued interest in this workthroughout the year, as well as reviewing it and giving sugges-tions.

REFERENCES[1] Gan O. Ambacher, J. Smart, J. R. Shealy, N. G. Weimann, K. Chu, M.

Murphy, W. J. Schaff, L. F. Eastman, R. Dimitrov, L. Wittmer, and etal., Two-dimensional electron gases induced by spontaneous and piezo-electric polarization charges in n- and ga-face algan/gan heterostructures,Journal of Applied Physics , vol. 85, no. 6, pp. 3222-3233, 1999. doi:10.1063/1.369664.

[2] Y. Chen, F. C. Lee, L. Amoroso, and H.-P. Wu, A resonant mosfet gatedriver with efficient energy recovery, IEEE Transactions on Power Elec-tronics, vol. 19, no. 2, Mar. 2004

[3] D. Matousek and L. Beran, Comparison of positive and negative dicksoncharge pump and fibonacci charge pump, 2017 International Conference onApplied Electronics (AE), 2017. doi: 10.23919/ae.2017.8053595.

[4] Astable multivibrator.[Online] http://www.electronics-tutorials.ws/ wave-forms/astable.html,

CONTENTS i

Contents

1 Introduction 1

2 Literature study 3

2.1 Power converters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

2.1.1 Types of power converters . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

2.1.2 Example power converter . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

2.2 HEMTs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

2.2.1 GaN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

2.2.2 AlGaN/GaN HEMTs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

2.2.3 HEMT usages . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

2.2.4 HEMT simulation model . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

2.3 HEMT drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

2.3.1 Cascode driven . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

2.3.2 Direct drive . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

2.4 Resonant drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

2.4.1 Circuit Explanation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

2.4.2 Circuit discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

2.5 Negative voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

2.5.1 Negative voltage generator . . . . . . . . . . . . . . . . . . . . . . . . . . 19

3 Driver components 22

3.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

3.2 Clock generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

3.2.1 Ring Oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

3.2.2 Schmit trigger oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

CONTENTS ii

3.2.3 Astable multivibrator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

3.2.4 Efficiency measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

3.3 Negative voltage generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

3.3.1 -3V generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

3.3.2 -5V generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

3.4 Negative level detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32

3.5 Level shifters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

3.6 Resonance circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

3.7 HEMT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

3.8 Boost converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

3.9 Complete system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42

4 Simulation results 46

4.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46

4.2 Base case . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46

4.3 Duty Cycle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48

4.4 Switching period . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51

4.5 Resonant inductance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54

4.6 Diode types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55

4.7 Boost converter input voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56

4.8 Load impedance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57

5 Discussion 61

5.1 Best possible boost converter efficiency . . . . . . . . . . . . . . . . . . . . . . . . 61

5.2 Best possible efficiency increase . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

5.3 Possible improvements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63

5.4 Comparison direct drive versus resonant drive . . . . . . . . . . . . . . . . . . . . 64

5.5 Possible changes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65

6 Conclusion 67

6.1 Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67

6.2 Suggestions for future work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68

Appendix A Datasheet HEMT 72

CONTENTS iii

Appendix B Verilog-A code 78

B.1 Diode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78

B.2 Shottky diode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78

B.3 HEMT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82

B.3.1 Drain-source current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82

B.3.2 Capacitances . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83

LIST OF FIGURES iv

List of Figures

2.1 DC/DC boost converter circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

2.2 The waveforms for continuous and discontinuous driving modes. The current in

continuous mode never drops to zero, while in discontinuous mode it does. . . . . 7

2.3 Typical AlGaN/GaN HEMT structure (not drawn to scale). . . . . . . . . . . . . 10

2.4 Typical band diagram for an AlGaN/GaN HEMT (not drawn to scale). . . . . . 11

2.5 Ideal characteristics Ids-Vds for an AlGaN/GaN at different gate voltages [6]. . . 11

2.6 Two different driving schemes for a D-mode HEMT device. . . . . . . . . . . . . 14

2.7 Resonant circuit for designed for high frequency PWM applications [18] . . . . . 16

2.8 Example implementation of 7-stage negative dickson charge pump. . . . . . . . . 20

3.1 Overview of all the building blocks of the driver. The blue arrows are external

control signals. Almost all blocks are connected to ground and the positive voltage

rail, but these connections aren’t shown. . . . . . . . . . . . . . . . . . . . . . . . 23

3.2 Circuit ring oscillator for 20 MHz clock. It contains 27 stages and delay-time

capacitors of 10pF between every stage. . . . . . . . . . . . . . . . . . . . . . . . 25

3.3 Circuits Schmitt trigger oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

3.4 Astable multivibrator circuit for 20 MHz clock. It also buffers both (complemen-

tary) clock outputs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

3.5 -3V negative voltage generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

3.6 -5V negative voltage generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

3.7 Negative level detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

3.8 Circuit differential opamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

3.9 DC response opamp. Vdd is 3,3V. Vss and Vin+ are tied to ground and Vneg is

varied. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

3.10 Level shifter shifting +3,3V to -5V and 0V to -2V.Vdd=3,3V;Vss=-5V . . . . . . 35

LIST OF FIGURES v

3.11 Level shifter shifting +3,3V to 0V and 0V to -3V. Vdd=3,3V;Vss=-3V . . . . . . 36

3.12 The used resonant circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

3.13 The average and standard deviation of critical parameters . . . . . . . . . . . . 40

3.14 The circuit of the boost converter . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

3.15 The complete circuit of the boost converter, including the driver. . . . . . . . . . 43

3.16 Waveform of the driver and boost converter with default parameters. . . . . . . . 45

4.1 Waveform of the default simulation. . . . . . . . . . . . . . . . . . . . . . . . . . 47

4.2 Driver power and efficiency versus input duty cycle. L=2µH. . . . . . . . . . . . 49

4.3 Driver power versus switching period for different inductance values. L=0 is the

direct drive scheme. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53

4.4 Efficiency versus switching period for different inductance values. L=0 is the

direct drive scheme. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53

4.5 Efficiency at different booster input voltages. . . . . . . . . . . . . . . . . . . . . 57

4.6 Segment of the driver waveform when Rload=1kΩ. . . . . . . . . . . . . . . . . . 59

LIST OF TABLES vi

List of Tables

3.1 Power consumption of the three different clock generators at 20 MHz clock frequency. 28

3.2 OR gate thruth table . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

3.3 Default values for all the design parameters . . . . . . . . . . . . . . . . . . . . . 44

4.1 Power consumptions for resonant and direct driver at default configuration. . . . 47

4.2 Power consumptions for varying duty cycle of the resonant driver. Tin=500ns . . 48

4.3 Power consumptions for varying duty cycle of the direct driver. Tin=500ns . . . 48

4.4 Power consumption for varying duty cycle of the Resonant driver. Tin=1µs . . . 49

4.5 Power consumptions for varying duty cycle of the direct driver. Tin=1 µs . . . . 49

4.6 Power consumptions for varying switching periods of the resonant driver. L=2

µH, duty cycle=0.5 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51


µH, duty cycle=0.5 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51

4.8 Power consumptions for varying switching periods of the direct driver. duty

cycle=0.5 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51


µH, duty cycle=0.3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52


µH, duty cycle=0.3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52

4.11 Power consumptions for varying switching periods of the direct driver. duty

cycle=0.3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52

4.12 Power consumption for varying inductance values in the resonance driver. . . . . 54

4.13 Power consumption of the resonance driver with different diode configurations.

L=5µH . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55

4.14 Power consumptions of the resonant driver for varying booster input voltages. . . 56

LIST OF TABLES vii

4.15 Power consumptions of the direct driver for varying booster input voltages. . . . 56

4.16 Power consumptions of the resonant driver for varying load impedance. . . . . . 58

4.17 Power consumptions of the direct driver for varying load impedance. . . . . . . . 58

5.1 Parameter values to achieve the highest possible efficiency. The parameters not

mentioned are default. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

5.2 Parameter values to achieve the highest possible efficiency increase. The param-

eters not mentioned are default. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63

GLOSSARY viii

Glossary

AC/DC Alternating current and direct current.

MOSFET Metal-oxide-semiconductor field-effect transistor

GaN Gallium nitride

D-mode Depletion mode, or normally-on

HEMT High electron mobility transistor

SMPS Switched mode power supply

CMOS Complementary metal-oxide-semiconductor

2DEG Two-dimensional-electron-gas

HV/LV High voltage and low Voltage

DMOS Double-diffused metal-oxide-semiconductor

nDMOS N-type DMOS

pDMOS P-type DMOS

PWM Pulse-width modulation

Vcc Supply rail voltage

Vg (HEMT) gate voltage

IC Integrated Circuit

EMC Electromagnetic compatibility

INTRODUCTION 1

Chapter 1

Introduction

In modern day life, electronics is something you see everywhere. You wake up with an electrical

alarm clock, work on a computer, watch TV in the evening... All these electronic devices need

power, and this is what the little black boxes are for that we plug into the wall socket. Welcome

to the field of power electronics. It controls power coming from the input (e.g. power socket

or batteries) and converts it into something the electronic circuit can use. One of the most

important metrics in power electronics is the efficiency: the ratio of input power versus useful

output power. A device with bad efficiency uses extra energy which can lead to bad battery life

and excess heat generation.

In the field of power electronics, a big topic is power converters. These convert an input

voltage into a different output voltage. Some example of different outputs are a DC voltage, a

higher or lower voltage or a voltage that can draw a lot of current. Most modern day power

converters use some type of switching element. These types are called switched-mode power

supplies. Most often, the switching element is a power MOSFET, an insulated-gate bipolar

transistor (IGBT) or a thyristor.

In recent years, GaN D-mode HEMTs (Gallium Nitride depletion-mode High Electron Mo-

bility Transistors) have started receiving attention in research and products as the switching

element in power converters. They have high voltage capabilities, high switching speeds and

high electron carrier mobility, resulting in a low ON resistance. The high voltage capability lets

them be used in high-voltage environments and converters with high voltage outputs. The high

switching speeds and high electron carrier mobility on the other hand improve the efficiency

when it is used in a power converter.

The major disadvantage of these GaN HEMTs is that they are depletion mode devices. This

INTRODUCTION 2

means that they are normally-on at zero gate–source voltage. To turn this type of transistor off,

we have to apply a negative gate–source voltage. We can do this by either putting a negative

voltage on the gate of the HEMT, called direct drive, or we can raise the voltage at the source

and hence create a negative source-gate voltage, called cascode drive.

A different method used to improve the efficiency of power converters is driving the switching

element in a resonant way. This means using some sort of component (most often an inductor)

to recover some of the energy used for switching the switch element on and off. This method

has already been proven to be effective in silicium technology, but has not yet been tested for

GaN HEMTs.

This thesis aims to test the efficiency improvement of power converters when we combine

both technologies: driving a GaN HEMT with a resonant driver. We will test the efficiency of

a power converter by first driving it in the conventional way, and then we will test the resonant

driver. The efficiency will be tested for different switching frequencies (1-5 MHz), different load

resistances and different input voltages. A GaN HEMT can be driven in two ways: via a cascode

circuit or directly at its gate. Here, we will drive the HEMT directly at its gate.

This thesis is divided into six chapters. The first chapter is this introduction, stating the

goal and used technologies. Next, we explore what research has already been done, and what we

can use to build upon in chapter 2. We design and build the circuits in simulations in chapter

3. In chapter 4, the results of our simulations are given and explained. In chapter 5, we will

discuss the results in a broader scope. We will finish the thesis by summarizing our results and

giving a final conclusion as well as some directions for further research in chapter 6.

LITERATURE STUDY 3

Chapter 2

Literature study

The first step in designing and testing a new system is understanding what already exists and

what is already known. We do this by making a literature study. First we will discuss in

what type of applications this research can be used: switched-mode power supplies. Next, we

will discuss the used switching element, a GaN HEMT (Gallium nitride high electron mobility

transistor) and its main advantages and disadvantages. We also discuss how we will model

this component. Following, we take a look at the possible driving schemes, and its driving

circuits. Lastly, we take a look at possible negative voltage generators. This is because we want

to generate the required negative voltages ourself, instead of relying on an external negative

voltage power rail.

2.1 Power converters

A big topic in power electronics are power converters. These transform an initial AC or DC

voltage into a different AC or DC voltage, with the difference being in voltage level, frequency

or other properties. A brief discussion of some power converters is given below.

2.1.1 Types of power converters

DC to DC converters are primarily used in portable electronic devices such as cellular phones

and laptop computers, which are supplied with power from batteries primarily. Such electronic

devices often contain several sub-circuits, each with its own voltage level requirement different

from that supplied by the battery or an external supply (sometimes higher or lower than the

supply voltage). Additionally, the battery voltage declines as its stored energy is drained.

2.1 Power converters 4

Switched DC to DC converters offer a method to increase voltage from a partially lowered

battery voltage thereby saving space instead of using multiple batteries to accomplish the same

thing.

AC to AC transformers used for voltage conversion at mains frequencies of 50–60 Hz must

be large and heavy for powers exceeding a few watts. This makes them expensive, and they are

subject to energy losses in their windings and due to eddy currents in their cores. DC-to-DC

techniques that use transformers or inductors work at much higher frequencies, requiring only

much smaller, lighter, and cheaper wound components. Consequently these techniques are used

even where a mains transformer could be used. For example, for domestic electronic appliances

it is preferable to rectify mains voltage to DC, use switch-mode techniques to convert it to high-

frequency AC at the desired voltage, then, usually, rectify to DC. The entire complex circuit is

cheaper and more efficient than a simple mains transformer circuit of the same output.

A switched-mode power supply (SMPS) is an electronic power supply that incorporates a

switching regulator to convert electrical power efficiently. Like other power supplies, a SMPS

transfers power from a DC or AC source (often mains power) to DC loads while converting volt-

age and current characteristics. Unlike a linear power supply, the pass transistor of a switching-

mode supply continually switches between low-dissipation full-on and full-off states, and spends

very little time in the high-dissipation transitions, which minimizes wasted energy. Ideally, a

switched-mode power supply dissipates no power. Voltage regulation is achieved by varying the

ratio of on-to-off time, called duty cycle. In contrast, a linear power supply regulates the output

voltage by continually dissipating power in the pass transistor. The higher power conversion

efficiency is an important advantage of a switched-mode power supply. Switched-mode power

supplies may also be substantially smaller and lighter than a linear supply due to the smaller

transformer size and weight.

Fast rise and fall times of the semiconductor devices are required for efficiency. The shorter

the switch stays in its power-inefficient linear region, the higher the efficiency of the driver will

be. However, a downside of fast switching is that these fast transitions combine with layout

parasitic effects to make circuit design challenging. They can also give rise to EMC problems.

Faster devices also permit higher switching frequencies. This means the requirements for the

low-pass filter at the output of the power converter are less strict and hence can be made more

power-efficient and smaller/cheaper. The filter can use smaller components, which have less

parasitics and can be cheaper. If the components are small enough, we can even incorporate


them on-chip. The low-pass filter at the output is necessary to transform the high-frequency

output to a DC voltage. A (small) voltage ripple will always be present on this ’DC’ output.

The higher efficiency of a switched-mode converter reduces the heatsinking needed, and in-

creases battery endurance of portable equipment. Efficiency has improved since the late 1980s

due to the use of power FETs, which are able to switch more efficiently with lower switching

losses at higher frequencies than power bipolar transistors, and use less complex drive cir-

cuitry. Another important improvement in DC-DC converters is replacing the flywheel diode

by synchronous rectification using a power FET, whose ”on resistance” is much lower, reducing

conduction losses. This improvement however is not used in this project, because it increases

the complexity of the circuit while not providing more information on the topic of a direct driver

versus resonant driver.

We can summarize that power converters are used in almost every piece of modern electronic

equipment. They use switching power semiconductor devices for both DC as well as AC con-

versions. These devices need to be able to handle large voltages and/or currents. Because of

efficiency reasons, they need to be able to switch fast between their on and off state. A last re-

quirement is that their on-resistance should be as low as possible to minimize power dissipation

in the device itself.

2.1.2 Example power converter

One of the simplest power converters is a dc-dc boost converter. Battery power systems often

stack cells in series to achieve higher voltage. However, sufficient stacking of cells is not possible

in many high voltage applications due to lack of space. Boost converters can increase the voltage

and reduce the number of cells. Two battery-powered applications that use boost converters are

electric vehicles and lighting systems.

In this thesis, we test the efficiency increase of driving a power converter with a GaN HEMT

in a resonant way, by incorporating it in a dc-dc boost converter. The circuit of this boost

converter can be seen in figure 2.1. Its advantages are that an (arbitrary) higher output voltage

than the input voltage can be obtained. Secondly, the switch can be driven with respect to

ground as compared to high side or isolated drive required for buck or buck-boost converters.

Lastly, the input current is continuous (no discontinuities) which means it is easy to filter and

meet EMC requirements.

Of course, the circuit has some disadvantages. A large output capacitor is required to reduce


the voltage ripple as the feeding current is pulsating. There is also a slower transient response

and lastly the need for feedback loop compensation in order to achieve a steady voltage.

Figure 2.1: DC/DC boost converter circuit

The key principle that drives the boost converter is the tendency of an inductor to resist

changes in current by creating and destroying a magnetic field. We make use of this tendency

by constantly switching it ON and OFF. When the switch is closed, current flows through the

inductor L in clockwise direction and the inductor stores some energy by generating a magnetic

field. The voltage over the inductor is positive and hence the amount of current flowing through

it will increase.

When the switch is opened, the current running through the inductor cannot go via the

HEMT anymore. This current will now be forced to go through the diode into the output

capacitor and load. The magnetic field previously created will be destroyed to maintain the

current towards the load. Because the output voltage is higher than the input voltage, the

voltage over the inductor will be negative and the current will decrease. Since there is now a

current flowing into the output capacitor Cout, the output voltage will increase.

When we now close the switch again, the voltage over the diode will become negative again

and it will stop conducting. The process of charging the inductor starts anew.

Meanwhile, the output capacitor powers the load, but loses charge while doing so. In order

for the output voltage Vo to remain steady over this period, we need the output capacitor to be

rather large. The load impedance experiences a DC voltage over the complete cycle, with some

ripple present due to the constant charging and discharging of the output capacitor.


(a) Continuous mode (b) Discontinuous mode

Figure 2.2: The waveforms for continuous and discontinuous driving modes. The current in

continuous mode never drops to zero, while in discontinuous mode it does.

There are two modes the boost converter can operate in: continuous and discontinuous

mode. Their respective waveforms can be seen in figure 2.2. When a boost converter operates in

continuous mode, the current through the inductor never falls to zero. On the other hand, if the

ripple amplitude of the current is too high, the inductor may be completely discharged before

the end of a whole cycle. This commonly occurs under light loads. In this case, the current

through the inductor falls to zero during part of the period. Although the difference is slight, it

has a strong effect on the output voltage equation.

In continuous mode, the output voltage is affected by the duty cycle. The output voltage

can be calculated via equation (2.1). In this equation, D is the duty cycle and ranges from

0 (S is never on) to 1 (S is always on). In our example waveforms the duty cycle is equal to

0,5. This equation supposes that all component are ideal. The output voltage equation can

theoretically go to infinite. This is ofcourse not possible in practice, since our components are

not ideal (power loss in switch, power loss over diode, limited current in the diode/FET...).

Discontinuous mode on the other hand has a more complicated output voltage equation

(2.2). The output voltage gain not only depends on the duty cycle (D), but also on the inductor

value (L), the input voltage (Vi), the commutation period (T) and the output current (Io). Once

again this equation supposes ideal components.

Which of the two modes we have depends on a few factors, like output current (in other words

the load impedance), duty cycle and maximum currents through the elements. Continuous mode

2.2 HEMTs 8

has a steadier output voltage and input current, but discontinuous mode is more power efficient,

since less current flows through the system and as a result less power is dissipated in the different

components.

VoVi

=1

1 −D (2.1)

VoVi

= 1 +ViD

2T

2LI0(2.2)

2.2 HEMTs

A HEMT is a high-electron-mobility transistor [1] , also known as heterostructure FET (HFET)

or modulation-doped FET (MODFET). It is a field-effect transistor incorporating a junction

between two materials with different band gaps (i.e. a heterojunction) as the channel instead of

a doped region (as is generally the case for MOSFET).

2.2.1 GaN

Since the invention of the metal-oxide-semiconductor field-effect-transistor (MOS-FET) in 1959,

the semiconductor industry for electronics has been dominated by silicon (Si). This is because

of the ease and cost of growing Silicium-oxide which enables ’easy’ production of complementary

metal-oxide-semiconductor (CMOS) process which has driven electronics forward to where we are

today. Si is however a low band gap material (1.1 eV) and hence not very suitable (although used)

in power electronics. Because of this, new materials are investigated as potential replacements,

such as gallium nitride (GaN), silicon carbide (SiC) and diamond.

GaN based transistors were developed in the early to mid 1990’s and have been extensively

researched and used for high-power high-frequency applications. In more recent years, it has

been rediscovered for use in power-electronics as high-voltage power switches.

GaN is a wide band gap semiconductor (band gap energy Eg=3.44eV) similar to diamond

and SiC. This wide band gap generally translates into an ability to handle high internal electric

fields before electronic breakdown occurs [2]. GaN is particularly interesting because of its

ability to form heterojunctions with wider band gap semiconductors such as aluminium gallium

nitride or aluminium nitride (up to 6.2 eV). This heterojunctions results in the forming of a 2-

dimensional-electron-gas (2DEG) at the interface due to large polarisation in the material which

provides a highly dense, majority carrier channel with a large electron mobility. This results in

2.2 HEMTs 9

a device which is capable of operating as a high-voltage (HV) power switch or high-frequency

power amplifier.

Due to the 2DEG, GaN offers a much larger current density than other materials. This

makes it so the gate can be smaller, resulting in lower gate capacitance. This also means there

is less charge needed for switching, resulting in lower switching losses. Of course, since the GaN

material is more expensive than Si, and the chip production process is much more optimized for

Si wafers, the price of GaN HEMTs is higher than that of Si power FETs.

Because the conduction happens via the 2DEG located in the undoped GaN layer, there is

a very low source-drain resistance. There are few to no lattice deformations and no impurities

to hinder the flow of electrons, creating a very low ON-resistance.

The advantages of using GaN as material can be summarized as follows [2]:

1. GaN has a higher breakdown voltage. The critical electric field is around 300V/µm (vs

Si: 3V/µm [3]), meaning that for electrodes on GaN with a spacing of 1 µm, a theoretical

bias voltage of around 300V could be applied without material breakdown.

2. They have a lower on-state resistance. AlGaN/GaN HEMTs display on resistances of

1mΩcm2 compared to an on-resistance 100mΩcm2 for Si. This leads to lower conduction

losses in the power transistor, improving the efficiency of power converters and reducing

the heat generation.

3. A higher current density due to the 2DEG lets the gate be smaller, resulting in lower gate

capacitance. This means the HEMT can be switched with less gate charges and hence

faster and more efficiently.

4. GaN can withstand higher temperatures. Devices have been shown to work beyond 300°C,

which leads to a lower need for cooling systems and large heat sinks.

2.2.2 AlGaN/GaN HEMTs

Many high power applications use gallium nitride based electronic devices in the form of high

electron mobility transistors (HEMTs). HEMTs will be the type of device used in this thesis

and this section will provide some details on their structure and properties.

A typical GaN based HEMT structure is shown in figure 2.3. This structure is typically grown

by metal organic chemical vapour deposition (MOCVD) or molecular beam epitaxy (MBE).

2.2 HEMTs 10

If a wider band gap material is grown on top of this layer, a heterojunction forms where

electrons can be confined into a quantum well forming a two dimensional electron gas (2DEG).

In this quantum well, electrons are able to move very easily and their mobility can be up to

2000cm2/V s [4]. The wider band gap materials commonly used with the GaN are the semi-

conductor alloy aluminium gallium nitride (AlxGa1−xN), aluminium nitride (AlN) and indium

aluminium nitride (InxAl1−xN) [5]. The most popular of these options is the AlxGa1−xN , and

uses an Al content (x value) of around 20-30%.

The structure shown in figure 2.3 includes some extra layers, such as a thick foreign substrate,

a bonding layer and a cap layer. The substrate layer is used due to the difficulty and cost in

growing native GaN. The typical materials are SiC, sapphire or Si. To reduce thermal stress

and latice mismatch with the GaN layer, a thin nucleation layer of AlN is grown on top of the

substrate. Then the GaN layer comes, on which a wide band-gap layer is put (AlGaN). On

top of the AlGaN barrier a GaN cap layer is grown which helps to reduce gate leakage currents

compared to devices without this cap layer. This layer works by increasing the effective Schottky

barrier height. This structure, in contrast to other HEMTs, does not require impurity doping.

Figure 2.3: Typical AlGaN/GaN HEMT structure (not drawn to scale).

In figure 2.4, the band diagram of the AlGaN/GaN heterostructure is shown. It shows the

wider band gap AlGaN to the left hand side and the narrower band gap GaN on the right. The

difference in conduction band energies at the interface of the materials results in a conduction

band offset ∆EC and a quantum well is formed at the material interface where the electrons

which make up the 2DEG will occupy the conduction band due to preferable (lower) energy.

2.2 HEMTs 11

The 2DEG is typically very narrow, and this is the reason why it is called two dimensional.

Figure 2.4: Typical band diagram for an AlGaN/GaN HEMT (not drawn to scale).

Figure 2.5: Ideal characteristics Ids-Vds for an AlGaN/GaN at different gate voltages [6].

Due to the presence of this 2DEG, we have a normally-on device, or a depletion mode FET

(D-mode FET). This means we have to apply a negative gate-source voltage in order to turn

the HEMT off. An example of the I-V characteristic of a GaN HEMT is given in figure 2.5.

This D-mode device is more difficult to work with than a standard enhancment mode device

(e-mode), such as conventional CMOS technology [7]. Because of this, there has been a push to

devellop e-mode GaN HEMTs. While high-voltage (HV) enhancement mode (normally OFF)

GaN devices have been demonstrated, depletion-mode GaN HEMTs are typically superior in

intrinsic performance. Additional process steps such as the recessed-gate technique and fluorine-

2.2 HEMTs 12

based plasma treatment are required to obtain enhancement-mode GaN HEMTs, which increase

the cost of fabrication and impact the device performance.

2.2.3 HEMT usages

HEMTs have long been used in radio-frequency application because they have high gain, which

makes them useful as amplifiers at high switching speeds. This is achieved because the main

charge carriers in HEMTs are majority carriers, and minority carriers are not significantly in-

volved. Extremely low noise values are achieved because the current variation in these devices

is low compared to other FETs, as a result of the current travelling through undoped material.

However, in recent years the HEMT has been rediscovered in the field of power electronics.

Here, the switched-mode power supplies utilize a fast switching HV FET. Thus a need arises

for more power efficient switching drivers. In DC-DC converters, this switching FET is followed

by a lowpass filter in order to transform the output to DC. The higher the switching frequency,

the more leniency you have on this filter and the smaller you can size your components. Smaller

components are often more efficient as well, with less losses and heat generation. This results

in smaller, cheaper and more powerful electronics.

Because a HEMT has such exceptional carrier mobility and switching speed, we can minimize

the gate size for the same current density compared to a normal FET, and hence minimize the

gate capacitance. This means we have less switching losses, since this loss is equal to equation

(2.3). This also means we can switch faster with the same amount of gate current.

Pgate = QgVGfS (2.3)

2.2.4 HEMT simulation model

This thesis uses simulations for the development of the driver, as well as the efficiency measure-

ments of the different drivers. There is a lot of research being done to accurately simulate GaN

HEMTs, modelling all the charges moving and accumulating inside the material [8], [9], [10],

[11], [12]. However, this research is still ongoing, and an accurate model needs both an accurate

measurements of the device itself, as well as significant computation time.

Because of these two downsides, we prefer to make a simulation model specific for one

HEMT. We choose the GaN HEMT MGG1T0617T, because it suits the needs for this project.

We model most of the properties of this GaN HEMT given in its datasheet, see appendix A

2.3 HEMT drivers 13

[13]. Specifically, we model the drain-source current which is dependant on the gate-source and

drain-source voltage. We also model the voltage-dependant Cds, Cgd and Cgs, since these have

a very big influence on the switching performance of our HEMT. In this model, we neglect the

gate current leakage, which is very small anyway. We also neglect the gate resistance. This

could have an effect, but since the current needed to charge/discharge the gate is quite small,

this effect will be minimal.

Guidelines how to model a voltage-dependant capacitor is given in [14]. Extra care needs to

be taken that you conserve the charge on the component. In other words, the charge function

is continuous, and the current through the component is dqdt .

To model this component, we use the Verilog-A language. Verilog-A is an industry standard

modeling language for analog circuits. It is the continuous-time subset of Verilog-AMS.

2.3 HEMT drivers

In this thesis, we want to be able to drive a depletion-mode HEMT. This means the device is

normally on, and needs a negative gate-source voltage to turn off. As a result, we cannot use the

conventional drivers we use for silicium. There are currently two major ways to drive a HEMT:

either directly via its gate or via a cascode configuration.

2.3.1 Cascode driven

The most popular way of driving a normally ON device is via the cascode-drive structure, shown

in figure 2.6a. A LV (Low Voltage), high current Si MOSFET is connected in series with the HV

(High Voltage) GaN HEMT, and the gate of the HEMT is tied to the source of the MOSFET.

This way, the high voltage is positioned over the HEMT instead of the Si MOSFET, and a

MOSFET with low on-resistance can be used instead of the HV MOSFETs, which have a higher

on-resistance.

The cascode works by elevating the source voltage of the HEMT, so that the gate-source

voltage becomes negative, and the HEMT turns off. The cascode device now has a normally-off

characteristic. A conventional MOSFET driver can be used to drive the LV MOSFET at high

speeds.

The cascade mode has a major disadvantage however. It has higher switching losses due to

the need to charge both the Cgs of the HEMT as well as the Coss of the LV MOSFET, and the

Coss of the LV MOSFET is quite large, because its needs a large gate. Since it is made in Si

2.3 HEMT drivers 14

technology, the gate area has to be big because the large current flows through this MOSFET,

so to achieve a low RON and have efficient operation we need a big gate. Since the goal of this

thesis is to minimize driving power consumption, we will not use this driving scheme.

(a) Cascode HEMT driver (b) Direct HEMT driver

Figure 2.6: Two different driving schemes for a D-mode HEMT device.

2.3.2 Direct drive

The other commonly used way to drive a HEMT is by driving its gate directly with a negative

voltage, shown in figure 2.6b. The major disadvantage of this way is that we need to apply

a negative voltage to the gate. This voltage can be supplied via an external negative supply

rail or generated on-chip. The proposed driver has negative voltage generators, implemented as

dickson charge pumps.

Because the direct-drive scheme is always on, we also add an extra LV MOSFET in cascode

with the HEMT. This MOSFET is there to be able to shut the HEMT off for longer times

without consuming any power, and let the driver do a startup sequence before the HEMT starts

conducting. An UVLO (under voltage lockout) can also be incorporated here. This MOSFET

does not need to switch quickly or often, since its only function is to turn the HEMT switching

operation on or off. This means there is no power loss due to switching. Some small conduction

losses due to the (small) Ron introduced by this MOSFET still remain present.

The cascode and direct drive schemes were previously demonstrated for SiC junction field-

effect transistors (JFETs), which are also normally ON devices. It was concluded that the

cascode driving scheme has larger switching losses, due to the driving of the cascode LV MOSFET

[15], [16]. The cascode driving is also an indirect driving scheme, which offers less controllability

2.4 Resonant drivers 15

over the device switching behaviour.

2.4 Resonant drivers

High power density applications require converters operating at high frequencies. Therefore, the

size of the energy storage elements can be reduced and the losses associated with the equivalent

series resistance of the capacitor and inductor decrease. At higher frequencies however, gate

driver power consumption also increases considerably. The power consumption for a conventional

gate driver is dissipated almost completely via the gate resistance.

With the increase in operating frequency, the gate drive power loss becomes significant

enough to affect the efficiency of the converter. This loss can be reduced by introducing an

inductor which helps in either recovery or recycling of energy. Many authors have proposed

different techniques of energy recovery and energy recycling using the principle of resonance. A

detailed study of different resonant drivers for MOSFETS is given in [17].

Since our goal is low power consumption, we want a circuit that minimizes this as much as

possible. The second thing to take into account is the circuit complexity. At the input of our

circuit we only have positive voltages, but in order to turn the HEMT off we need (switching)

negative voltages. Because of this, we need a level shifter for each switching voltage. This means

we need to add a lot of extra complexity for each extra MOSFET in the original circuit.

With this caveat in mind, we chose the circuit depicted in figure 2.7a. This circuit only uses

two MOSFETS, two diodes and one inductor. It has the advantage of recovering the energy and

the removal of cross conduction losses. It has the disadvantage of only clamping the gate during

energy recovery. In some circuits, it can also cause unreliable ringing. To accomodate for this

ringing, we will generate a negative voltage sufficiently below the cutoff point of our HEMT.

The working of this circuit will now be discussed in more detail.


2.4.1 Circuit Explanation

(a) Circuit(b) Ideal waveforms when diode forward drop is ne-

glected and M1’s gate capacitance is approximated

as linear

Figure 2.7: Resonant circuit for designed for high frequency PWM applications [18]

In this circuit, a complementary driving pair (MDR1 and MDR2) is inherited from the conven-

tional driver, an inductor (LR) is inserted as the basic resonant element, and two more diodes

(DDR1 and DDR2 ) are designed to clamp and recover the driving energy. The timing of the

circuit is set to cycle the inductor current within each driving transition so that the circuit per-

formance will have no dependence on the operational duty cycle. In addition, when the diodes

recover the driving energy an equivalent low impedance path is provided.

The working of the circuit can be best explained with the help of the ideal waveforms given

in figure 2.7b. We begin with VGS M1 = 0 (t < t1) where both mosfets are off and the inductor

current is zero. At time t1, MDR1 is turned on and a voltage step appears at the source of

MDR1. Responding to this voltage, the inductor current ILR and the capacitor voltage VGS M1

both start to rise, until a time t2 when VGS M1 = VDD and ILR = IPeak. If the quality factor

Q of the resonant circuit is high enough, IPeak and the rise time tr (= t2 − t1) can be easilyestimated.

IPeak =VDDZ0

= VDD ·√CG M1LR

(2.4)


tr =π

2ω0=π

2·√LR · CG M1 (2.5)

where CG M1 is the equivalent gate capacitance of M1, Z0 the characteristic impedance of

the resonant circuit and ω0 the resonant frequency.

During the period between t2 and t3, VGS M1 is clamped at VDD by diode DDR1 and iLR flows

freewheeling along MDR1, LR and DDR1. This freewheeling phase should be as short as possible,

since conduction losses exist over all these elements and thus enlarges the power consumption

of the driver.

Once we arrive at time t3, MDR1 is turned off and the energy recovery process starts: the

inductor current turns on the body diode of MDR2 and flows through the path of MDR2 - LR -

DDR1 - VDD. With a constant voltage VDD across LR, the inductor current diminishes linearly

and the recovery time trec (=t4 - t3) = is simply

trec =LR · IPeakVDD

=√LR · CG M1 (2.6)

Because the gate voltage VGS M1 is at the right value, the driving circuit does nothing from

t4 to t5. When we now want to turn the switch off again, we need to discharge this gate voltage.

We do this by turning on MDR2. This imposes a ’negative’ voltage over LR and a current starts

flowing from CG M1 through LR and MDR2 to ground. Once the voltage is back at ground level

(at time t5, the current freewheels along DDR2, LRandMDR2. In reality, the inductor current iLR

will diminish a bit because of voltage drops over DDR2 and MDR2. Lastly, the recovery process

starts again when we shut off MDR2. From t7 - t8, charge is pumped from the low-voltage ground

to the high-voltage source VDD. Now the cycle can start anew

This description explains the operation of the circuit in terms of voltages and currents. We

can also explain the operation in terms of energy processing. From time t1 to t2, energy is

transferred from the power source VDD to the resonant inductor (0.5LRI2Peak) and the gate

capacitor (0.5CG M1V2DD). The subsequent stage (t2 to t3) freewheels the inductor energy, or in

other words takes power from VDD and moves it back to VDD. Lastly, from t3 to t4 the energy

stored in the inductor is returned to the power source.

In the discharging state, the potential energy present on the gate capacitor (0.5CG M1V2DD)

is now transferred to the inductor (discharging the gate capacitor and shutting off the switch)

at time t5 to t6. When we now start the recovery process this energy stored in the inductor is

finally returned to the power source.

2.5 Negative voltage 18

2.4.2 Circuit discussion

Theoretically, this circuit has zero power usage compared to the original power usage shown in

equation (2.3). In practice however, dissipation exists in every component and the power usage

is greater than zero. The dissipation is calculated theoretically in [18]. In this thesis, we will

adapt the circuit so it can drive a D-mode HEMT and simulate this so we can measure the

power usage/efficiency increase quantitatively.

A second advantage of this circuit is that the voltages are clamped via the diodes. This

means we have immunity to false triggering and we do not need extra ’safety circuitry’ to

protect against voltage spikes. The gate voltage will always reside between ground and source

(plus diode voltage).

A third advantage is that this circuit can handle (almost) any duty cycle, and a wide range

of frequencies, depending on the values of LR and CG M1 (because of tr and trec). A higher

inductance value has lower conduction losses (lower IPeak => lower I·V power consumption)but a lower possible operating frequency as well.

Of course the circuit has disadvantages too. The first being that an (ideally large) inductance

is needed. Inductors take up a lot of space (and are hence costly) to incorporate in an IC. An

external inductor will probably be needed. However, this external inductor takes up a lot of

space (landing pads + external connection) and brings a lot of extra parasitics with it, and

hence adds to the gate resistance. Secondly, the forward voltage drop of the clamping diodes

dissipate a lot of power in the freewheeling and recovery stage. We can lower these losses with

the use of Schottky diodes, if their leakage current does not cause a substantial Vgs drift at the

operating frequency. Lastly, the gate voltage Vgs is floating from t4 to t5 and t8 to t1. It is

clamped between VDD and ground but not connected to a specific voltage source. When there

is some leakage current, the exact value can change during this period of time. If this period is

too long, or in other words the switching frequency is too low, the switch could shut off earlier

than expected.

2.5 Negative voltage

As discussed earlier in section 2.3, we need a negative voltage to shut the HEMT off. We can

either bring this negative voltage in externally via a negative voltage supply rail, or generate

this voltage ourself. In this thesis, we choose to generate this voltage on-chip.


A big problem in working with negative voltages on a chip is the possibility of charge leaking

from ground to the negative voltage via the body of our MOSFETs in the driving circuit. If

the source (or drain) has a low enough voltage, the parasitic source-bulk (or drain-bulk) p-n

junction can be forward biased and conduct a lot of charge. In order to prevent these losses,

we have to put the bulk of our chip at the lowest potential, so the lowest voltage. This also

means special care has to be taken in order to not destroy the MOSFETs due to large voltage

differences between its terminals.

In this thesis we work with the Intelligent Interface Technology I3T80 process technology

[19]. This technology has a maximum voltage of 3.6V over the terminals of the LV MOSFETs.

However, MOSFETs with floating body are also present, so we can still provide positive logic if

we float the body of the respective MOSFETs at e.g. ground. In short, care has to be taken that

a technology is chosen in which the MOSFETs can either handle big enough voltages between

their terminals without breaking, or have the option of floating the body of the MOSFETs and

taking special care to never exceed the maximum allowed voltage between terminals.

The I3T80 technology also has DMOS (double-diffused metal–oxide–semiconductor) present,

which has a higher allowed voltage between drain-gate, drain-source and drain-body. They can

be used when we want to sweep the voltage (for example at the HEMT gate) over more than

3,6V. The downside of these DMOS is a larger Ron. This results in either more conduction losses

or a larger gate, and thus more switching losses and more chip area needed.

2.5.1 Negative voltage generator

The classical DC/DC converters are based on inductors or transformers. These converters are

suitable for a high output power, but are too large for our application. Charge pumps are

a sufficient alternative to classical DC/DC converters for a lower output power and a smaller

dimension. A well-known variant of a charge pump is the Dickson charge pump. The output

to input voltage ratio is directly proportional to the number of Dickson charge pump stages.

The Fibonacci charge pump is a suitable alternative to the Dickson charge pump especially for

a higher output to input voltage ratio [20].


Figure 2.8: Example implementation of 7-stage negative dickson charge pump.

An example negative dickson charge pump is given in figure 2.8. The circuit works as

follows: it first inverses and buffers the clock, creating signals clka and clkb. Now we will look

at the voltage over C1 during the clock transition. Suppose the voltage is initially 0V above

C1, and the clock clka transitions from 3V to 0V. Since the voltage over the capacitor cannot

change drastically, the voltage above C1 lowers as well by 3V, so to -3V. Diode D2 will start

conducting, until the voltage is distributed equally between C1 and C2, or a voltage of -1.5V is

at both terminals of D2 (assuming C1=C2 and ideal components).

In the next step, clka goes up again from 0V to 3V, and the voltage ’above’ C1 goes to 1.5V.

This activates D1, discharging the voltage until it is back to 0V, the same as before. However,

in this step clkb goes from 3V to 0V, lowering the voltage above C2 from -1.5V to -4.5V. This

activates D3, and again charge flows until both C2 and C3 are at the same voltage.

This process keeps repeating until eventually the node above C1 switches between 0 and -3V

and cannot pump charge anymore. The final stage is diode D8 with a load capacitor, changing

the switching negative voltage into a continuous negative DC voltage.

By constantly switching the clocks on and off, charge gets pumped towards the ground, and

a negative voltage is generated. Some voltage drops over the diodes is unavoidable, but we can

minimize this by using shottky diodes with a low voltage drop.

The biggest design parameter these voltage generators have are the pumping capacitor size,

the amount of stages and the clock frequency at which they pump their charge. The larger the

capacitors are, the more current the output can draw, or the more current loss is possible in the

following circuit, being the HEMT and the resonance circuit. Adding more stages to the pump

results in a lower voltage being achievable, and at higher voltages there is more current possible

at the output. Adding more stages has the downside of more steps where power can be lost, so a


lower efficiency as a result. Lastly, increasing the clock frequency lets more charges be pumped

towards the load, so a higher output current is possible.

A second design factor is the output storage capacitor. Increasing this makes it so the output

voltage is steadier, but the start-up time will be larger as well. A larger output capacitor has

the downside that it takes up more space on a chip. It can also lead to a larger leakage current.

DRIVER COMPONENTS 22

Chapter 3

Driver components

We test the impact of the resonance driving on the performance by making use of simulations.

The used simulation program is Cadence. The used circuit simulator is Spectre: a SPICE-class

circuit simulator. It provides the basic SPICE analyses and component models. It also s

design of a multi-mhz resonant driver chip for high-voltage d ......wouter faelens d-mode gan hemts...

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