design of a 60 ghz hybrid mode substrate integrated...

Thomas Deckmyn

Waveguide Cavity-Backed Antenna ArrayDesign of a 60 GHz Hybrid Mode Substrate Integrated

Academic year 2014-2015Faculty of Engineering and ArchitectureChairman: Prof. dr. ir. Daniël De ZutterDepartment of Information Technology

Master of Science in Electrical EngineeringMaster's dissertation submitted in order to obtain the academic degree of

Counsellors: Ir. Martijn Huynen, Ir. Gert-Jan Stockman, Ir. Sam AgneessensSupervisors: Prof. dr. ir. Dries Vande Ginste, Prof. dr. ir. Johan Bauwelinck

Preface

First and foremost, I would like to express my gratitude towards prof. dr. ir. Dries Vande

Ginste and prof. dr. ir. Johan Bauwelinck for providing me with the opportunity to carry out

this research at the Electromagnetics Group and the INTEC design group of the Department

of Information Technology. I thank Prof. Vande Ginste for sharing his expertise and giving

splendid advice, and for his never-ending positivism and support throughout the entire year. I

am very grateful as well for the insights and encouragements Prof. Bauwelinck offered.

Two people I would like to thank very profoundly are ir. Martijn Huynen and

ir. Gert-Jan Stockman. Their excellent guidance and exhaustive feedback were indispensable up

until the very last minute. Thanks to both of you, you helped pave the way towards a gratifying

conclusion of five years of studies.

Special thanks goes out to ir. Sam Agneessens for the assistance with the design of the hybrid

mode SIW antenna array. His expertise and proficient counseling enabled me to successfully

complete the design.

Ir. Niels Lambrecht deserves special thanks as well, for offering theoretical and practical

advice on more than one front. Furthermore, I appreciate the helping hand that was extended

to me by everyone at the research group, whenever I needed it.

My fellow thesis students Bob Mertens, Alexander Vindelinckx, Kristof Baes, Jorn Marievoet,

Piet Merckx and Enrico Massoni also deserve an acknowledgement. Thank you for the positive

vibe in the thesis room and the pleasant days (and evenings) I have spent there with you all.

I would like to thank my parents for their support throughout the entire period of my studies.

They have always believed in me and supported every choice I made. Without them, I wouldn’t

be where I am today. Lastly, but definitely not least, a special thank you is in order for Bieke

Keysabyl, for providing moral support when it was needed most.

Thomas Deckmyn, May 2015

Admission to Loan

The author gives permission to make this master’s dissertation available for consultation and

to copy parts of this master’s dissertation for personal use. In the case of any other use, the

limitations of the copyright have to be respected, in particular with regard to the obligation to

state expressly the source when quoting results from this master dissertation.

Thomas Deckmyn, May 2015

Design of a 60 GHz Hybrid Mode SubstrateIntegrated Waveguide Cavity-Backed

Antenna Arrayby

Thomas DECKMYN

Master’s Dissertation submitted to obtain the academic degree of

Master of Science in Electrical Engineering

Academic 2014–2015

Promoters: Prof. dr. ir. Dries VANDE GINSTE, Prof. dr. ir. Johan BAUWELINCK

Supervisors: Ir. Martijn HUYNEN, Ir. Gert-Jan STOCKMAN, Ir. Sam AGNEESSENS

Faculty of Engineering and Architecture

Ghent University

Department of Information Technology

Chairman: Prof. dr. ir. Daniel DE ZUTTER

Summary

The goal of this master’s dissertation is to develop a highly compact and integratable antenna

array that operates in the 60 GHz band, whilst maintaining compatibility with standard printed

circuit board manufacturing processes. Two different antenna technologies, i.e., microstrip patch

and Substrate Integrated Waveguide (SIW), are thoroughly analyzed through extensive simu-

lation procedures and measurements. This leads to the formulation of a founded opinion that

SIW is the best suited technology to leverage in the array configuration. Moreover, to en-

hance the impedance bandwidth of the inherently band limited cavity-backed SIW antennas,

a technique based on the excitation of hybrid modes is exploited. The integration aspect of

this dissertation is fortified by selecting a 50 µm flexible substrate material for the design of the

hybrid mode SIW antenna and the array. Although the substrate material is extremely thin, a

fractional impedance bandwidth of 3.6% is achieved. An antenna gain and directivity of 7.2 dBi

and 12.0 dBi, respectively, are obtained by constructing a four-element Uniform Linear Array

(ULA).

Keywords

Hybrid mode substrate integrated waveguide; bandwidth enhancement; antenna array; integra-

tion; flexible substrate

Design of a 60 GHz Hybrid Mode Cavity-BackedSubstrate Integrated Waveguide Antenna Array

Thomas Deckmyn

Supervisors: prof. dr. ir. D. Vande Ginste, prof. dr. ir. J. Bauwelinck, ir. G.-J. Stockman, ir. M. Huynenand dr. ir. S. Agneessens

Abstract— The goal of this master’s dissertation is to develop a highlycompact and integratable antenna array that operates in the 60 GHz band,whilst maintaining compatibility with standard printed circuit board man-ufacturing processes. Two different antenna technologies, i.e., microstrippatch and Substrate Integrated Waveguide (SIW), are thoroughly analyzedthrough extensive simulation procedures and measurements. This leads tothe formulation of a founded opinion that SIW is the best suited technologyto leverage in the array configuration. Moreover, to enhance the impedancebandwidth of the inherently band limited cavity-backed SIW antennas, atechnique based on the excitation of hybrid modes is exploited. The inte-gration aspect of this dissertation is fortified by selecting a 50 µm flexiblesubstrate material for the design of the hybrid mode SIW antenna and thearray. A simulated fractional impedance bandwidth of 3.6% is achieved. Asimulated antenna gain and directivity of 7.2 dBi and 12.0 dBi, respectively,are obtained by constructing a four-element Uniform Linear Array (ULA).

Keywords— Hybrid mode substrate integrated waveguide; bandwidthenhancement; antenna array; integration; flexible substrate

I. INTRODUCTION

NOWADAYS, the omnipresent use and rapid evolution ofelectronic devices puts ever rising demands on the hard-

ware engineer of today. The swift development towards higherbitrates, to satisfy the presently unquenchable mobile user, posesnovel challenges in terms of bandwidth. Moreover, higher oper-ating frequencies are explored, which brings about the addedpredicament of high frequency effects. Paired with the vastminiaturization of high speed electronics, the design task at handbecomes ever more complicated.

This master’s dissertation focuses on the design of a band-width enhanced Substrate Integrated Waveguide (SIW) antennaarray, operating in the 60 GHz band. Hybrid modes are excitedinside the cavity and merged in the desired frequency range, no-tably increasing the impedance bandwidth. The need for minia-turization is tackled by designing the antenna array on extremelythin, i.e., 50 µm and 100 µm, flexible substrate materials.

In this abstract, first, the bandwidth enhancement techniquebased on hybrid modes is discussed (Section II). The designof the hybrid mode cavity-backed SIW antenna is considered inSection III and in Section III-B a Uniform Linear Array (ULA)is constructed utilizing the previously designed antenna ele-ments. Measurement results are presented in Section IV. Con-clusions and future research are discussed in Section V.

II. HYBRID MODE EXCITATION

The limited bandwidth of an SIW antenna can be amelioratedby simultaneously exciting two distinct resonances, i.e., hybridmodes, inside the cavity. By merging these hybrid modes withinthe desired frequency range, the impedance bandwidth is signif-icantly enhanced. The two resonances are, in essence, two dif-

(a)

(b)

Fig. 1. Field distribution in the SIW cavity: (a) Dominant E-field distribution oflower hybrid mode in largest half cavity; (b) Dominant E-field distributionof higher hybrid mode in smallest half cavity [1].

ferent combinations of a TE110 and TE120 mode. By offsettingthe slot from the center of the cavity, two half parts with differ-ent dimensions are created. The lower frequency hybrid modeis dominant in the largest half cavity and is a combination ofa strong TE110 and a weak TE120, as depicted by the E-fielddistributions in Figure 1(a). The total E-field is in phase in bothhalf cavities, but radiation can still be evoked due to the high dif-ference in magnitude. The higher hybrid mode is dominant inthe smallest half part and consists of a strong TE120 and a weakTE110, as illustrated in Figure 1(b). Here, the field is out ofphase in both half parts. A large electric field is present accrossthe slot, hence radiation is generated.

III. DESIGN OF THE HYBRID MODE SIW ANTENNA ARRAY

A. Hybrid Mode SIW Antenna

A general configuration for a hybrid mode SIW antenna isdepicted in Figure 2. The cavity dimensions Lc and Wc aredetermined as such that the resonant frequency of the TE110

is 60 GHz. The length of the slot Ls is much larger than λ2 ,

hence it is non-resonant, and the width Ws tunes the impedancematching to some extent. Here, it is opted for dc,u > dc,l; thehigher frequency hybrid mode is dominant in the lower half partof the cavity.

If dc,u increases, the resonating area of the dominant field ofthe lower hybrid mode is enlarged, hence its resonant frequencydecreases. The area of the weak field of the higher frequency hy-brid mode increases as well, but only causes a minor drop in fre-quency. Hence, the lower hybrid mode decreases in frequency,while the higher hybrid mode remains practically unchanged.This causes an enhancement of the bandwidth.

Ls Ws

dc,l

dc,u

Lc

Wc

Fig. 2. General configuration of a hybrid mode SIW antenna.

After several design iterations and ample optimization, thesimulated reflection coefficient for the design on the 50 µm sub-strate is as presented in Figure 3. Two distinct resonances areperceived at 59.16 GHz and 60.38 GHz. The total impedancebandwidth amounts to 2.2 GHz, which corresponds to a frac-tional bandwidth of 3.6% at 60 GHz.

58 59 60 61 62

0

-5

-10

-15

-20

-25

-30

-35

59.16 GHz 60.38 GHz

2.2 GHz

Frequency [GHz]

|S11|[

dB]

Fig. 3. Simulated reflection coefficient of the designed hybrid mode SIW an-tenna.

B. Uniform Linear Array Configuration

An additional benefit of adopting the SIW technology for thedesign of the antenna array, is the high isolation due to the metal-lic via rows [2]. The array elements can be spaced very closelytogether, without inducing high mutual coupling. The UniformLinear Array (ULA) is constructed using four hybrid mode SIWantennas, while reusing the via wall for adjacent elements. Thesimulated reflection coefficient of the SIW antennas in the ULAconfiguration very closely resembles that of a separate element,as in Figure 3. A simulated antenna gain and directivity of7.2 dBi and 12.0 dBi, respectively, is obtained.

IV. MEASUREMENTS

The fabricated hybrid mode SIW antenna array is depictedin Figure 4. As Uniform Thickness Copper Plating (UTCP) ofthe vias is not feasible for such a small quantity of prototypes,grounding of the top layer is achieved by injecting conductivepaste into the holes. A detailed inspection of the array revealsthat residue of the conductive paste has completely filled the in-sets and part of the slot. Attempts have been made to clear thespillage from both. Still, some remains, as illustrated in Fig-ure 5.

Fig. 4. Fabricated hybrid mode SIW antenna ULA on 100 µm flexible substrate.

Measurements are performed using an N5242A Pro-grammable Network Analyzer (PNA-X) from Keysight Tech-nologies and indicate that a standing wave is present on thefeed network of the array. This implies that the matching atboth ends of the feed structure is poor. The insets procure a50 Ω Grounded Coplanar Waveguide (GCPW) structure that isperfectly matched to the feed line. Taking into account the re-maining spillage of the paste in the slot, it is clear that the di-mensions of the insets have changed, effectively altering theimpedance matching. The electric performance of the paste bywhich the vias are grounded is not precisely known at 60 GHzeither, hence it is plausible that the impedance of the SIW an-tenna elements is changed significantly.

Fig. 5. Detail of fabricated hybrid mode SIW antenna array.

Measurements indicate that the dimensions of the antenna el-ements and 50 Ω microstrip lines are accurate, hence a fabrica-tion inaccuracy is not to blame for the poor matching. The elec-trical parameters of the substrate material are characterized at1 MHz in [3], thus the loss tangent and permittivity could havenotably changed at 60 GHz. This discrepancy is likely to con-

tribute to the deteriorating of the impedance matching. More-over, due to the extremely thin substrate material, the perfor-mance of the press fit of the connector is likely to have dimin-ished; a good connection is not unquestionably ensured.

To gain more insight into the effects at hand, the characteris-tics of a 50 Ω reference microstrip line are measured. To quan-tify the amount of additional losses introduced by the combina-tion of connector and substrate, the power balance of the lineis calculated, i.e., |S11|2 + |S21|2. For a lossless transmissionline structure, this should equal unity, indicating that all incidentpower is either reflected or transmitted. Calculations reveal thatapproximately 70% of the incident power is dissipated. This isdue to losses in the copper and substrate, as well as to radiation.This confirms that it is indeed the combination of connector is-sues and an insufficiently characterized substrate material thatare to blame for the faulty operation of the antenna system.

V. CONCLUSIONS AND FUTURE RESEARCH

A highly compact and integratable hybrid mode cavity-backed SIW antenna array for the 60 GHz band was succesfullydesigned. Bandwidth enhancement was achieved by exploitingthe excitation of hybrid modes inside the SIW cavity. Elevatedgain and directivity was procured by utilizing a four-elementULA configuration. The integration aspect was pushed to theutmost extent by selecting extremely thin substrates for the de-sign.

The insufficient characterization of the flexible substrate at60 GHz and the non-ideal processing of the minute vias wasdetrimental for the operation of the fabricated prototypes. Fu-ture research on this topic could certainly encompass investigat-ing the characteristics of flexible substrates for antenna design at60 GHz. Even research towards the development of novel flexi-ble substrate materials for use in the Extremely High Frequency(EHF) band could be performed.

In future work, the combination of the hybrid mode SIW arraywith very high speed phase shifters can yield the development ofadaptive antenna systems at 60 GHz. This can have applicationsin, e.g., millimeter wave radar detection systems.

REFERENCES

[1] G. Q. Luo and Z. F. Hu and W. J. Li and X. H. Zhang and L. L. Sun and J.F. Zheng, “Bandwidth-Enhanced Low-Profile Cavity-Backed Slot Antennaby Using Hybrid SIW Cavity Modes”, IEEE Transactions on Antennas andPropagation, vol. 60, no. 4, pp. 1698 - 1704, April 2012.

[2] M. Bozzi and A. Georgiadis and K. Wu, “Review of Substrate IntegratedWaveguide Circuits and Antennas”, IET Microwaves, Antennas and Prop-agation, vol. 5, no. 8, pp. 909 - 920, September 2011.

[3] DuPontTM, “DuPontTMPyralux R© AP flexible cir-cuit materials, Technical Datasheet [Online]”, Avail-able: http://www.dupont.com/content/dam/assets/products-and-services/electronic-electrical-materials/assets/PyraluxAPclad DataSheet.pdf.

List of Abbreviations i

List of Abbreviations

ADS Advanced Design System

CST Computer Simulation Technology

DC Direct Current

DSP Digital Signal Processing

EHF Extremely High Frequency

FEM Finite Element Method

GCPW Grounded Coplanar Waveguide

MWS Microwave Studio

PNA-X Programmable Network Analyzer

SHF Super High Frequency

SIW Substrate Integrated Waveguide

SMA Sub-Micron version A

ULA Uniform Linear Array

UTCP Uniform Thickness Copper Plating

UWB Ultra Wideband

WLAN Wireless Local Access Network

WPD Wilkinson Power Divider

CONTENTS ii

Contents

List of Abbreviations i

1 Introduction 1

1.1 High Frequency Design and Challenges . . . . . . . . . . . . . . . . . . . . . . . . 1

1.2 Goal and Outline . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

2 Antenna Design Aspects 3

2.1 Substrate Material . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

2.2 Microstrip Patch Antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

2.2.1 Operating Mechanism . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

2.2.2 Impedance Matching . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

2.3 Substrate Integrated Waveguide Technology . . . . . . . . . . . . . . . . . . . . . 7


2.3.2 Cavity-Backed SIW Antenna . . . . . . . . . . . . . . . . . . . . . . . . . 8

2.4 Simulation Software . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

3 Deembedding of Connector and Feed Line Structure 10

3.1 Scattering Transfer Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

3.2 Deembedding Algorithm Based on Matrix-Pencil Method . . . . . . . . . . . . . 11

3.2.1 Illustrative Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

3.3 Characteristics of a Stand Alone Antenna . . . . . . . . . . . . . . . . . . . . . . 14

4 Design and Measurement of Antenna Test Structures 16

4.1 Substrate and Connector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

4.2 Microstrip Patch Antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

4.2.1 Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

4.2.2 Corner Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

4.3 Cavity-Backed SIW Antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21


4.3.2 Corner Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

4.4 Transmission Line Test Structures for Deembedding . . . . . . . . . . . . . . . . 25

4.5 Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

4.5.1 Microstrip Patch Antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

4.5.2 Cavity-Backed SIW Antenna . . . . . . . . . . . . . . . . . . . . . . . . . 29

CONTENTS iii

4.6 Topology for Antenna Array . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

5 Hybrid Mode SIW Antenna Array 32

5.1 Hybrid Mode Cavity-Backed SIW Antenna . . . . . . . . . . . . . . . . . . . . . 32


5.1.2 Substrate Material . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

5.1.3 Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34


5.2 Uniform Linear Antenna Array . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

5.3 Array Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

5.3.1 Mutual Coupling Between Array Elements . . . . . . . . . . . . . . . . . . 39


5.3.3 Wilkinson Power Divider for Feed Network . . . . . . . . . . . . . . . . . 42

5.3.4 Influence of Feed Network . . . . . . . . . . . . . . . . . . . . . . . . . . . 44

5.4 Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

6 Conclusion and Future Research 49

Bibliography 51

List of Figures 53

List of Tables 56

INTRODUCTION 1

Chapter 1

Introduction

1.1 High Frequency Design and Challenges

The omnipresent use and rapid evolution of electronic devices places ever rising demands on the

hardware engineer of today. Higher bit rates are needed to offer high speed wireless communica-

tion to the presently unquenchable mobile user, which poses many novel challenges in terms of

bandwidth. Moreover, higher operating frequencies make that high frequency effects come into

play and the design task at hand becomes severely more complicated. The hardware engineer

needs to be fully aware of these effects, which are among others more pronounced parasitic

effects, rising losses with frequency and wave phenomena in interconnection structures.

A second trend is the vast miniaturization of high speed electronics. Hence, compact and easily

integrated antenna systems are of prime interest. Antennas rapidly scale down with rising

frequency, but need to be manufactured with high accuracy to ensure proper millimeter wave

operation. They need to remain low-cost as well, if they are to be used in high volume consumer

applications. This evidently puts a strain on the available manufacturing processes and a proper

design needs to be robust against possible inaccuracies.

If the dimensions of planar electronic structures become electrically large, i.e., of the order of

a wavelength of operation, distributed circuit analysis needs to be performed. Transmission

line effects set in and the characteristic impedance of lines becomes an important parameter.

An discontinuity in the impedance of the signal line will introduce reflections and will result in

additional attenuation. As losses are already strongly present at high frequencies, impedance

matching requires additional attention.

In this master’s dissertation, antenna designs in the Super High Frequency (SHF) and Extremely

High Frequency (EHF) range are performed, hence the high frequency effects mentioned above

are taken into close consideration.

INTRODUCTION 2

1.2 Goal and Outline

The goal of this master’s dissertation is to design a highly compact and easily integrated antenna

array that operates in the unlicensed 60 GHz band [1]. A hybrid mode technique will be used

to enhance the bandwidth, while the array configuration will enable beam forming and yield

elevated gain.

To arrive at a 60 GHz antenna array with optimal performance in the final stage of the design,

a first step is to develop test structures to assess the inherent advantages and disadvantages

of different antenna technologies at various frequencies. By comparing the simulations and

measurements at distinct frequencies, it can be deduced which effects are more severe at higher

frequencies. Concretely, both the microstrip patch and Substrate Integrated Waveguide (SIW)

antenna technology will be exploited to design radiating elements with resonant frequencies

at 15 GHz, 30 GHz and 60 GHz. After manufacturing, intensive measurements will enable a

comparison of both topologies and will provide sufficient arguments to make an appropriate

choice of topology. Moreover, any critical steps in the manufacturing process can be exposed as

the dimensions of the devices miniaturize with rising frequency.

In Chapter 2 the microstrip patch antenna and SIW technologies will be addressed in a theoret-

ical manner. In Chapter 3 a method to eliminate the effects of a connector from S-parameter

measurements will be introduced. The design and measurement of the antenna test structures

is described in Chapter 4, and a reasoned choice of the most suitable technology for this appli-

cation will be made as well. In Chapter 5, the concept of hybrid modes will be introduced and

the array will be designed. Lastly, in Chapter 6 conclusions will be provided and future research

will be considered as well.

ANTENNA DESIGN ASPECTS 3

Chapter 2

Antenna Design Aspects

2.1 Substrate Material

Selecting the best suitable substrate for an antenna is of the utmost importance. By opting for

a proper material, the inherent restrictions of the antenna, such as insertion loss, narrowband

behavior and low efficiency can be ameliorated. The parameters of the substrate, i.e., the relative

dielectric permittivity εr and loss tangent tan δ, have notable effects on the performance of the

device. A high loss tangent implies strong losses in the substrate material, which will result in a

low radiation efficiency. The permittivity determines the speed of wave propagation inside the

material. A high permittivity results in slower propagation and thus a decreased wavelength.

This can be used in antenna miniaturization; if an antenna is placed in a medium with higher

permittivity, the required size for half wavelength resonance decreases and the antenna will be

smaller. Unfortunately, a high permittivity diminishes the available bandwidth.

Another important parameter is the thickness of the substrate material. The ohmic losses in

the copper of a microstrip line, due to the finite conductivity of the conductor, are proportional

to the height of the substrate and inversely proportional to the width of the line [2]. A thin

substrate hence decreases ohmic losses, but it also degrades the radiation efficiency and band-

width of a patch antenna for example. If the substrate material is thick, a larger bandwidth

and more efficiency can be achieved, but the propagation of surface waves will be enhanced.

The fundamental order surface wave has a cut off frequency at Direct Current (DC), but higher

order surface modes will be excited if the substrate thickness is too large [3] [4]. These waves

travel inside the substrate and are scattered at surface discontinuities, degrading the radiation

pattern and polarization characteristics of the antenna. Moreover, the power that is used to

excite these waves, can no longer be radiated, hence the radiation efficiency is degraded as well.

2.2 Microstrip Patch Antenna

Microstrip patch antennas have the advantage of being low-profile and light weight, and can

easily be integrated together with other planar circuits. A patch antenna consists of a metallic


film bonded to a grounded dielectric substrate. The thickness of the substrate layer and the

relative permittivity εr are usually small, hence the patch behaves more like a parallel plate

transmission line. If the patch is fed using a microstrip feed line, as depicted in Figure 2.1,

waves travel from the feed point to the edges of the patch. There, a discontinuity is present

and the fields sense an open circuit, hence a considerable amount of reflections occur. This

suggests that the portion of incident energy that is radiated, is reduced; this in turn leads to

the assumption that a patch antenna can be seen as a resonant (leaky) cavity. The fact that a

patch antenna behaves more like a cavity supports the known limitations of this topology, i.e.,

low efficiency and narrowband behavior.

x

y

z

εr

h

L

W

Figure 2.1: Simple microstrip patch antenna configuration.

2.2.1 Operating Mechanism

A simple microstrip patch configuration is presented in Figure 2.1, with W and L the width

and length of the patch respectively. The substrate material is characterized by its relative

permittivity εr and height h. The radiation mechanism of the microstrip patch antenna is

discussed using a simple aperture model. Typically, the height of the substrate is a small

fraction of the wavelength (order of λ20) and the length L is of the order of λ

2 . As the height h of

the cavity is very small (h λ, with λ the wavelength inside the dielectric material), the fields

are assumed constant with respect to the z-axis. The radiation is caused by the fringing E-fields

emerging from the exposed substrate at the edge of the patch, as depicted in Figure 2.2.


x

z

yL

Fringing fields

εr

Figure 2.2: Fringing E-fields as radiation mechanism for a patch antenna.

Applying the cavity model to the patch antenna topology in Figure 2.1, yields that the resonant

frequency (fr)mnp of the transverse magnetic TMmnp mode [5] is given by

(fr)mnp =c

2π√εr

√(mπh

)2+(nπL

)2+(pπW

)2, (2.1)

with εr the relative permittivity of the substrate material and c the speed of light in vacuum.

The fundamentel order mode, i.e., the lowest in frequency, now depends on the dimensions of

the patch. If L > W h holds, the fundamental mode is the TM010. It follows from (2.1) that

its resonant frequency is given by

(fr)010 =c

2L√εr. (2.2)

A practical width of the patch that yields good radiation efficiencies is [5]

W =c

2fr

√2

εr + 1. (2.3)

When alternatively W > L h, the fundamental mode is the TM001 with resonant frequency

given by

(fr)001 =c

2W√εr. (2.4)

Equivalently, a practical length that yields good radiation efficiency is

L =c

2fr

√2

εr + 1. (2.5)

As stated above, fringing fields are the sources of radiation for the microstrip patch antenna.

From Figure 2.2 one can observe that most field lines of the electric field reside inside the

substrate, but some of the field lines exist in the air. Because some of the lines travel through

the air surrounding the substrate, the fields feel an effective dielectric constant εr,eff that differs


from the εr of the substrate itself. The low frequency value for the effective relative permittivity

εr,eff is given by [6]

εr,eff =

εr+1

2 + εr−12

[1 + 12 h

W

]− 12 , L > W h

εr+12 + εr−1

2

[1 + 12 hL

]− 12 , W > L h

. (2.6)

For higher frequencies the effective dielectric constant starts to increase monotonically with fre-

quency, approaching the dielectric constant of the substrate.

Due to the fringing of the fields the patch is electrically wider and longer as compared to its

physical dimensions, hence this needs to be accounted for in the calculation of the resonant

frequency. A commonly used approximate expression for the increase in length is [7]

∆L = 0.412h(εr,eff + 0.3)

(Wh + 0.264

)(εr,eff − 0.258)

(Wh + 0.8

) . (2.7)

This yields an effective electrical length Leff = L + 2∆L and a resonant frequency for the

dominant TM010 mode given by

(fr)010 =c

2(L+ 2∆L)√εr

=c

2Leff√εr. (2.8)

Equivalently, an effective width Weff can be calculated and used to obtain the resonant frequency

of the TM001 mode.

2.2.2 Impedance Matching

The designed patch antenna will be excited using a 50 Ω microstrip feed line, as already depicted

in Figure 2.1. To avoid high reflections at the junction between patch and feed line, the edge

impedance of the antenna has to be matched to the 50 Ω of the microstrip line. The impedance

at the edge of the patch [8] is, in good approximation, given by

Zin = 90ε2r

ε2r − 1

(L

W

)2

Ω. (2.9)

As is clear, the edge impedance of the antenna can be set to 50 Ω by calculating the appropriate

value for the width W .

A different technique, and of most practical importance, makes use of insets to perform the

matching, as presented in Figure 2.3. The input impedance seen at the edge depends on the

inset depth R as defined in

Z ′in(R) = cos2

(πR

L

)Zin, (2.10)


where Zin can be calculated using (2.9). It can be understood from (2.10) that the input

impedance exhibits a maximum value for R = 0 and decreases monotonically with increasing

depth R, until the minimum value is reached for R = L2 . In the center of the patch the voltage

is zero and the current is maximum, hence the impedance is zero. The width of the inset d can

be used to further tune the resonant input impedance.

L

W

R

d

Figure 2.3: Microstrip patch antenna with insets for impedance matching.

2.3 Substrate Integrated Waveguide Technology

SIWs are planarly integrated structures with a close resemblance to conventional rectangular

waveguides. SIWs are fabricated by connecting two parallel copper planes with electrically con-

ductive vias embedded in the substrate, as illustrated in Figure 2.4. Hence, this technology

allows one to integrate a usually non-planar rectangular waveguide, while maintaining compat-

ibility with conventional planar printed circuit board processing steps. Because of the high

resemblance to rectangular waveguides, SIW structures also exhibit the main advantages of

having a high Q-factor and high electrical shielding. This technology creates the possibility to

integrate passive and active components, and even antennas, onto the same substrate, which is

of prime practical interest.

sd

w

Figure 2.4: SIW structure: conductive vias connecting parallel copper planes.



As stated above, SIWs closely resemble conventional rectangular waveguides, be it in planar

form, but SIW structures also display similar propagation characteristics. The guided modes

are practically equal to the TEn0-modes that propagate in a rectangular waveguide with similar

dimensions, where the dominant mode is TE10. Due to this high similarity, empirical formulas

have been presented that allow one to obtain the effective width weff of a rectangular waveguide

with the same propagation characteristics as the SIW structure. It holds that [9]

weff = w − d2

0.95s, (2.11)

where d and s are the diameter of the vias and the spacing between the vias respectively. The

frequency of the first order guided mode in an SIW structure can be obtained as the frequency

of the TE10 guided mode in a rectangular waveguide of width weff, and hence equals [3]

f10 =c

2√εrweff

(2.12)

=c

2√εr

(w − d2

0.95s

)−1

. (2.13)

The similarity only holds provided that the radiation leakage through the gaps between the

vias is neglegible, i.e., the row of vias needs to approximate a contiguous metallic wall at the

operating frequency. If this is the case, the SIW structure can be treated as being a rectangular

waveguide with width weff as defined in (2.11). Empirical constraints on the dimensions of and

the spacing between the vias such that radiation leakage is neglegible, have been formulated

in [10] as

s ≤ 2d

d ≤ λ10 ,

(2.14)

with λ the wavelength in the substrate material.

2.3.2 Cavity-Backed SIW Antenna

A cavity-backed SIW antenna consists of a cavity resonator constructed in SIW technology. If

the cavity is made leaky by introducing one or more slots, radiation occurs. The cavity is built

up as depicted in Figure 2.5 and is excited using a microstrip feed line. The resonant frequency

of the TEmn0-mode can be found as [11]

fmn0 =c√

2πεr

√(mπ

Weff

)2

+

(nπ

Leff

)2

, (2.15)

where Leff and Weff can be obtained using (2.11), c is the speed of light in vacuum and εr is the

relative permittivity of the substrate material. As this is an empirical expression, tuning will


be necessary to obtain the desired operating frequency. Note that the restrictions on the via

spacing and diameter (2.14) need to be fulfilled to ensure proper operation.

LW

Ls WsRgcpw

dgcpw

Figure 2.5: SIW resonant cavity with microstrip line feed.

Once the cavity is constructed at the right resonant frequency, a slot is added to evoke radiation.

The slot can be etched in the ground plane, which will isolate the radiation caused by the feeding

network from the desired radiation coming from the slot, or it can be etched in the top copper

plane, which will drastically lower backward radiation.

The slot is an important tuning element of the cavity-backed SIW antenna. As described in [12],

the width Ws can be tuned to obtain optimal impedance bandwidth, but the effect on impedance

bandwidth is much less than that of the substrate thickness. The length of the slot Ls can be

used to adjust the resonant frequency by some degree, but the resonance is primarily determined

by the cavity size. The length of the slot also has major influence on the radiation efficiency,

which is at its maximum when the slot length is half a wavelength.

To match the edge impedance of the cavity to the 50 Ω microstrip feed line, the parameters

Rgcpw and dgcpw need to be chosen such that a 50 Ω grounded coplanar waveguide (GCPW)

structure is obtained.

2.4 Simulation Software

When designing millimeter wave antenna systems, one wants to predict the real behaviour of the

device as accurately as possible. A good simulation tool that takes into account all high frequency

effects is thus indispensable. For all designs that follow, the Keysight Technologies Advanced

Design System (ADS) simulation software is used. It incorporates a planar 3D electromagnetic

solver Momentum, which uses a frequency-domain Method of Moments (MoM) to simulate

complete board structures whilst taking parasitic effects, losses and coupling into account.

DEEMBEDDING OF CONNECTOR AND FEED LINE STRUCTURE 10

Chapter 3

Deembedding of Connector and Feed

Line Structure

To connect the antenna test structures that will be designed in the next chapter to measurement

equipment or an exterior circuit, a suitable connector will be needed, i.e. one with excellent high

frequency performance. Still, this connector will introduce a certain amount of reflections, due

to impedance mismatch and insertion loss. In this chapter a method is discussed that enables

extracting the S-parameters of the connector from a series of reference measurements. These

extracted characteristics can then be deembedded from the measured characteristics of the

cascade of connector and antenna, eliminating the added losses and reflections.

3.1 Scattering Transfer Parameters

A different approach to represent the characteristics of a two-port network is by means of T-

parameters [13] (Scattering Transfer Parameters). These are an alternative to the well-known

S-parameters with their prime advantage being cascadeability. The T-parameters relate the

amplitudes of the waves a1 and b1 at the input of the two-port to the amplitudes of the waves

a2 and b2 at the output (see Figure 3.1), while the S-parameters relate the amplitudes of the

reflected waves at both input and output, i.e. b1 and b2, to the incident waves a1 and a2. The

T-parameters are hence defined by the relation

(a1

b1

)=

(T11 T12

T21 T22

)·(b2

a2

). (3.1)

The corresponding T-parameters can be calculated from the two-port’s S-parameters using the


following conversion formulas [14].

T11 = 1

S21

T12 = −S22S21

T21 = S11S21

T22 = −S11S22−S12S21S21

.

(3.2)

As stated above, the main advantage of T-parameters over S-parameters is cascadeability. To

illustrate this, consider a two-port network that consists of 2 two-port networks in a cascade

configuration as depicted in Figure 3.1.

a1

b1

a2

b2

a3

b3

a4

b4

T 1 T 2

Figure 3.1: Cascade of two arbitrary two-port networks.

Taking into account (3.1), it holds for the network of Figure 3.1 that(a1

b1

)=

(T 1

11 T 112

T 121 T 1

22

)·(b2

a2

)(3.3)

and (a3

b3

)=

(T 2

11 T 212

T 221 T 2

22

)·(b4

a4

). (3.4)

It now follows that the scattering transfer matrix for the overall network is defined as the matrix

product of the T-matrices of the cascaded two-port networks [14], i.e.(a1

b1

)=

(T 1

11 T 112

T 121 T 1

22

)·(T 2

11 T 212

T 221 T 2

22

)·(b4

a4

). (3.5)

The previous reasoning can be extended to an arbitrary cascade of N two-port networks, yielding

an expression for the overall scattering transfer matrix T given by

T = T 1 · T 2 · ... · TN . (3.6)

3.2 Deembedding Algorithm Based on Matrix-Pencil Method

As mentioned above, a deembedding algorithm will be used to extract the characteristics of

the connector with simple reference measurements as a starting point. A novel matrix-pencil

two-line method that achieves this was developed in [15], where it was exerted to eliminate the

impact of the connector in performing substrate characterization for wearable antenna systems.


To achieve deembedding, the S-parameters of two microstrip lines of length l1 and l2, with

l2 > l1, as depicted in Figure 3.2, need to be measured. The characteristics of the connectors,

in cascade with a coax-to-microstrip transition, are defined by SA and SB. The piece of ideal

lossy transmission line, with length ∆L = l2 − l1, is defined by SL. The two lines are fully

characterized by their propagation factor γ and characteristic impedance Z0. Prior knowledge

of the impedance Z0 is obsolete as the effects of impedance mismatch are included in the model.

l1

SA SB

l2

SA SBSL

Figure 3.2: Microstrip lines with coax-to-microstrip transitions SA and SB and ideal lossy

transmission line section SL.

Defining the scattering transfer matrix of the short line, with length l1, as T short and the matrix

of the long line, with length l2, as T long one can express both matrices as [15]

T short = TA · TBT long = TA · TL · TB.

(3.7)

Here TA, TB and TL are the T-parameter equivalents of SA, SB and SL respectively, which can

be obtained using the conversion formulas (3.2). If (3.7) is solved, one obtains

T · TA = TA · TL, (3.8)

where T = T long · T−1

short · T long. The S-parameter matrices Slong and Sshort can be obtained

through measurements of both microstrip lines and converted to their T-parameter counterparts

using conversion formulas [13].

The method described in [15] calculates the deembedded complex propagation factor γ of the

piece of transmission line defined by TL based on an eigenvalue equation obtained from (3.8).

Perturbations in the complex propagation constant can yield unphysical results, hence they are

minimized by using the matrix-pencil method as an averaging method. Once the propagation

constant γ is obtained, the scattering transfer parameters TA and TB (corresponding to the coax-

to-microstrip transitions SA and SB respectively) are constructed, hence the characteristics of

the connector are extracted.

It can be seen from Figure 3.2 that the extracted characteristics TA in reality comprise the

characteristics of the connector in cascade with a coax-to-microstrip transition and a piece of


transmission line of length lA = 0.5l1. If a suitable choice is made for the lengths of the reference

microstrip lines, not only the influence of the connector, but of the entire feed line exciting

the microstrip patch antenna in Figure 2.1 can be eliminated as well. This implies that after

measurements of the return loss of the fabricated antennas, the additional losses due to connector

as well as feed line can be removed from the measurements, yielding the characteristics of the

stand-alone antenna. How the characteristics of connector and feedline will be deembedded from

the antenna measurements in a later stage of the design is discussed in Section 3.3.

3.2.1 Illustrative Example

To prove the correctness of the deembedding algorithm presented in Section 3.2, an illustrative

example based on simulations will be given. All simulations are performed using the Keysight

Technologies ADS circuit solver. The algorithm will be used to extract the characteristics of

a connector with T-matrix T conn, which is modeled as a 35 Ω discontinuity in a piece of 50 Ωtransmission line. For this example, a 50 Ω transmission line with a 35 Ω discontinuity at each

side is considered (see Figure 3.3), which corresponds to a transmission line with connectors at

both ends. As discussed in Section 3.2 the S-parameters of two microstrip lines with different

lengths need to be simulated and fed to the deembedding algorithm.

35 Ω TL 50 Ω TL 35 Ω TL

50 ΩVs,1

Port plane

50 Ω

Port plane

Vs,2

Figure 3.3: Piece of 50 Ω transmission line with connector modeled as 35 Ω discontinuity.

To assess the correctness of the algorithm, a second S-parameter simulation is performed solely

on the connector as pictured in Figure 3.4. Now the characteristics of the connector simulated

by the circuit solver can be compared to the result of the deembedding algorithm, as presented

in Figure 3.5. It can be seen that the algorithm yields results that correspond very well to what

is predicted by the simulation.

35 Ω TL

50 ΩVs,1

Port plane

50 Ω

Port plane

Vs,2

Figure 3.4: S-parameter simulation performed on modeled connector.


0 10 20 30 40 50 60

−50

−40

−30

−20

−10

Frequency [GHz]

S11

[dB

]

Figure 3.5: Characteristic of model connector: extracted with deembedding algorithm (solid

line) and simulated with circuit solver (dashed line).

3.3 Characteristics of a Stand Alone Antenna

As the S-parameters of the connector and the piece of feed line are extracted as described in

the previous section, these can now be used to calculate the characteristics of the stand-alone

antenna from the S-parameter measurement of the cascade of connector, feed line and antenna.

As stated, the S-parameter measurements are performed on the system as depicted in Figure

3.6, where a1,2 and b1,2 are incident and reflected power waves respectively.

50 ΩVs

Port plane

a1

b1

a2

b2

Connector

Figure 3.6: Block diagram of measurement to be performed.

For this system it is known that

(b1

a1

)= T conn

(a2

b2

), (3.9)

where T conn is the T-matrix (containing the two-port Scattering Transfer Parameters) of the

connector, which is fully known as a result of the deembedding algorithm. The system in


Figure 3.6 is a one-port, hence the conducted measurement will yield a reflection coefficient Sm,

with

Sm =b1a1. (3.10)

If one wants to retrieve the return loss of the stand-alone antenna, the S-parameter Sa needs to

be extracted, which is defined as

Sa =a2

b2. (3.11)

Combining (3.9) and (3.11) one obtains

(b1

a1

)= T conn

(Sab2

b2

), (3.12)

and evidently

(Sa

1

)b2 = T

−1

conn

(b1

a1

). (3.13)

If one defines the matrix X as the inverse matrix of T conn, the system of equations

Sab2 = X11b1 +X12a1

b2 = X21b1 +X22a1

(3.14)

becomes apparent and provides an expression for the return loss of the antenna, given by

Sa =X11

b1a1

+X12

X21b1a1

+X22

. (3.15)

Taking into account (3.10) the return loss of the antenna can be expressed as a function of the

measured S-parameter Sm of the system defined in Figure 3.6 and the characteristics of the

connector obtained with the deembedding algorithm. One obtains

Sa =X11Sm +X12

X21Sm +X22, (3.16)

with X = T−1

conn.

It is clear that this method enables one to calculate the characteristics of the stand-alone antenna,

eliminating all added losses and reflections due to the presence of the feed line and connector.

DESIGN AND MEASUREMENT OF ANTENNA TEST STRUCTURES 16

Chapter 4

Design and Measurement of Antenna

Test Structures

4.1 Substrate and Connector

The substrate selected for the design and fabrication of the antenna test structures is the

RO4350B® Rogers Corporation high speed laminate [16]. It offers superior high frequency

performance and is compatible with standard low-cost circuit fabrication techniques. The main

benefits are the low dielectric loss and the stability of the dielectric constant over a broad fre-

quency range. The latter makes this laminate an ideal material for broadband applications. The

main characteristics are summarized in Table 4.1.

εr 3.48 (10 GHz)

tan δ 0.0037 (10 GHz)

thickness 168 µm

Cu thickness 35 µm

Table 4.1: Characteristics of Rogers RO4350B High Speed Laminate.

The connectors that will be used to measure the antenna test structures are 1.85 mm Sub-Micron

version A (SMA) end launch connector assemblies fabricated by Southwest Microwave [17] (Fig-

ure 4.1), which are suited for applications up to 67 GHz. A press fit is used to ensure good

connection between connector pin and circuit board, so no soldering is needed. To obtain con-

sistent measurements, the amount of force by which the connector is pressed onto the circuit

board, is controlled with a torque wrench. The end launch connectors are rather bulky and

made out of solid metal, so as a precaution not to interfere with the proper operation of the

antennas, the connector needs to be at a reasonable distance. This implies a long microstrip

feed line towards the patches. An optimized connector footprint will be used as suggested in [18]

and depicted in Figure 4.2. This footprint uses a 50 Ω Grounded Coplanar Waveguide (GCPW)

launch structure to improve the performance of the microstrip and the line is slightly tapered

to match the connector pin.


Figure 4.1: Southwest Microwave 1.85 mm end launch connector.

2.82 mm

280 µm

210 µm

Figure 4.2: Connector footprint with GCPW launch structure.

4.2 Microstrip Patch Antenna

The first step in designing the patch antenna test structures at 15 GHz, 30 GHz and 60 GHz is

using (2.5) to calculate a length L of the patch that yields good radiation efficiencies, with the

desired resonant frequency as a starting point. From this the effective relative permittivity εr,eff

can be determined with (2.6). Once the length L and εr,eff are known, the width extension ∆W

can be obtained using (2.7), where in this case W must be replaced with L. Taking into account

all of the above, the physical width W that will yield the desired resonant frequency for the

TM001-mode is obtained by solving

W =1

2fr√εr,eff√ε0µ0

− 2∆W. (4.1)

The last step is adding insets to obtain good impedance matching to the 50 Ω feed line. The

depth R of the inset, as defined in Figure 2.3, is calculated using (2.10).

After several design iterations, including extensive optimization steps, the final designs for the

patch antenna test structures have dimensions as listed in Table 4.2.


15 GHz 30 GHz 60 GHz

W [mm] 5.95 2.70 1.50

L [mm] 5.21 2.60 1.28

R [mm] 1.83 0.94 0.43

d [mm] 0.20 0.20 0.16

Lfeed [mm] 10.75 10.75 10.75

Table 4.2: Dimensions of designed microstrip patch antenna test structures.

4.2.1 Simulation Results

All simulation results presented below include substrate and copper losses. The simulated re-

flection coefficientes for the designed antennas are depicted in Figure 4.3. As can be seen, all

designs exhibit a well defined resonance peak at the desired frequency. However, there are dis-

tinct differences between the characteristics of the three antennas. The patch antenna at 60 GHz

(Figure 4.3(c)) exhibits much more insertion loss than the one at 15 GHz (Figure 4.3(c)). The

higher insertion loss is to be expected as the attenuation due to both conductor and substrate

losses increases with frequency [3]. The limited bandwidth, which is an inherent disadvantage

of microstrip patch technology, is clearly visible as well. The antennas at 15 GHz, 30 GHz and

60 GHz have a relative impedance bandwidth of 1.2%, 1.8% and 3.5%, respectively.

The simulated directivity and gain in the E-plane are presented in Figure 4.4. The gain is lower

than the directivity in all three cases, because of the substantial losses at these high frequencies.

Directivity is related to gain as [19]

G(θ, φ) =D(θ, φ)

KL, (4.2)

where KL is a real factor, greater than unity and independent of direction, that represents the

power losses in the materials forming the antenna. The radiation efficiency is defined as the

ratio of the gain to the directivity, hence it is equal to 1KL

. As is clear from Figure 4.4, the

efficiency of the three patches is moderate; there is a notable difference between directivity and

gain. The radiation efficiency for all three designs is approximately 60%.

4.2.2 Corner Analysis

As discussed in Chapter 1, the ongoing desire for higher operating frequencies causes a vast

miniaturization of antenna structures and in turn puts a strain on the fabrication processes at

hand. The etching of the antennas will not be infinitely precise, hence an insight into the effects

of possible fabrication flaws is imperative. Consulting Table 4.2, it is clear that the performance

of the patch antenna at 60 GHz is the most sensitive to fabrication inaccuracies because of its

very small dimensions (order of 1 mm). The minimum feature size of the fabrication process is

200 µm and a worst case scenario of a 10% fabrication error is assumed, i.e., a fault of 20 µm.

The dominant mode of the patch antenna at 60 GHz is the TM001 and is determined by the


14 14.5 15 15.5 16

0

-5

-10

-15

-20

-25

170 MHz

Frequency [GHz]

|S11|[

dB

]

(a)

29 29.5 30 30.5 31

0

-5

-10

-15

-20

-25

-30

540 MHz

Frequency [GHz]

|S11|[

dB

]

(b)

58 59 60 61 62

0

-10

-20

-30

2.1 GHz

Frequency [GHz]

|S11|[

dB

]

(c)

Figure 4.3: Simulated reflection coefficient for patch antenna test structures: (a) at 15 GHz; (b)

at 30 GHz and (c) at 60 GHz.

width W (see Table 4.2). Figure 4.5(a) illustrates the effects of the fabrication error on the

resonant frequency of the antenna. Another important feature is the amount of matching, as

it determines the quantity of power that is reflected and thus not radiated. This is mainly

determined by the inset depth R, for which the effects of the fabrication error are depicted

in Figure 4.5(b). It is clear that the most critical parameter is the width of the patch, as it

determines the resonant frequency. An error of 20 µm already causes a notable shift of the

resonance.


−15

−10

−5

0

5

10 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(a)

−15

−10

−5

0

5

10 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(b)

−15

−10

−5

0

5

10 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(c)

Figure 4.4: Simulated directivity (solid) and gain (dotted) in the E-plane of the antenna: (a) at

15 GHz, (b) at 30 GHz and (c) at 60 GHz.

58 59 60 61 62

0

-10

-20

-30

-40

Frequency [GHz]

|S11|[

dB

]

W − 20 µmW + 20 µm

Final Design

(a)

58 59 60 61 62

0

-10

-20

-30

-40

Frequency [GHz]

|S11|[

dB

]

R− 20 µmR+ 20 µm

Final Design

(b)

Figure 4.5: Effect of a 20 µm fabrication error on the 60 GHz patch: (a) error on W affects the

resonant frequency; (b) error on R affects the matching.


4.3 Cavity-Backed SIW Antenna

As stated in Section 2.3, the equivalency between rectangular waveguides and SIW structures

only holds when the restrictions on the via diameter and spacing (2.14) are met. Sizing the vias

is an obligatory first step to ensure proper operation of the SIW antenna. These restrictions

are empirical, hence it is good practice to stay well below these limits. This will make sure the

via wall approximates a continuous conductive wall as good as possible. Table 4.3 summarizes

the via diameter and spacing maxima for each frequency, as well as the values that will be used

during the design. The dimensions at 30 GHz and 60 GHz are the minimal dimensions that still

comply with the fabrication design rules.

f [GHz] dmax [mm] smax [mm] d [mm] s [mm]

15 1.05 2.1 0.5 1

30 0.52 1.04 0.2 0.4

60 0.26 0.52 0.2 0.4

Table 4.3: Via diameter and spacing maxima for SIW operation.

Now, one can start dimensioning the actual cavity-backed SIW antenna as depicted in Figure 2.5.

Firstly, the dimensions of the cavity that correspond to the desired resonant frequency need to

be determined. The slot divides the SIW cavity into two half cavities with equal dimensions

and the radiation is caused by a strong TE120 resonance [20]. The SIW cavities of the antenna

test structures are hence designed to support a TE120 resonance at their desired frequencies of

operation.

From (2.15) the effective widths Weff and lengths Leff of the rectangular waveguides that are

equivalent to the SIW cavities can be determined. Together with the dimensions of the via

diameter and spacing from Table 4.3, the physical dimensions W and L of the SIW cavities can

be calculated using (2.11).

Next, the slot is added in the top copper plane to procure radiation. The slot is resonant to

ameliorate the radiation efficiency to the utmost extent [20], hence it has a length Ls approxi-

mately equal to half a wavelength at the desired operating frequency. The width of the slot Ws is

optimized to slightly improve the impedance bandwidth. Ideal impedance matching is acquired

through dimensioning Rgcpw and dgcpw for a 50 Ω Grounded Coplanar Waveguide (GCPW) feed.

After a considerable number of design iterations and ample optimization, the final dimensions

for the cavity-backed SIW antennas at 15 GHz, 30 GHz and 60 GHz are obtained as given in

Table 4.4.


The simulated reflection coefficients for the cavity-backed SIW antennas are presented in Fig-

ure 4.6. It is clear that all three designs exhibit a well-defined resonance at the desired frequency.


15 GHz 30 GHz 60 GHz

L [mm] 8.00 4.00 2.00

W [mm] 10.00 4.40 2.40

Ls [mm] 5.82 3.05 1.59

Ws [mm] 0.76 0.30 0.21

Rgcpw [mm] 1.98 1.07 0.62

dgcpw [mm] 0.70 0.32 0.21

Table 4.4: Dimensions of designed cavity-backed SIW antenna test structures.

14 14.5 15 15.5 16

0

-5

-10

-15

-20

-25

-30

-35

-40

130 MHz

Frequency [GHz]

|S11|[

dB

]

(a)

29 29.5 30 30.5 31

0

-5

-10

-15

-20

-25

-30

-35

-40

370 MHz

Frequency [GHz]

|S11|[

dB

]

(b)

58 59 60 61 62

0

-5

-10

-15

-20

-25

-30

-35

-40

1.04 GHz

Frequency [GHz]

|S11|[

dB

]

(c)

Figure 4.6: Simulated reflection coefficient for cavity-backed SIW antenna test structures: (a)

at 15 GHz; (b) at 30 GHz and (c) at 60 GHz.

The impedance bandwidths are 1%, 1.2% and 1.7% for the antennas at 15 GHz, 30 GHz and

60 GHz, respectively. This demonstrates the inherent narrowband behavior of the SIW antenna.

Concerning the insertion loss, it is apparent that the difference between the SIW antennas at

15 GHz and 60 GHz is substantially smaller than was the case for the microstrip patch antenna

treated in Section 4.2. This indicates that high frequency losses are less severe when opting for


the SIW technology. Comparing Figure 4.3(c) with Figure 4.6(c), one perceives the significant

amount of approximately 7 dB less insertion loss for the SIW antenna. The metallic via wall

prevents radiation from spreading through the lossy substrate material, hence reducing substrate

losses as compared to a microstrip patch antenna.

The E-field inside the cavity of the SIW antenna at 60 GHz is presented in Figure 4.7. It is clear

from the configuration of the E-field lines that the designed cavity is in TE120 resonance, which

complies with the design strategy outlined in Section 4.3. The electric field at the two sides of

the slot has opposite phase and a large magnitude. Because of this, a transverse electric field

exists across the slot and radiation arises.

The simulated directivity and gain are presented in Figure 4.8. Again, there is a difference

between the gain and directivity, indicating losses in the materials forming the antenna. The

simulated radiation efficiencies are obtained from calculation of the factor KL as defined in (4.2)

and amount to approximately 35% for all three designs. The simulated gain appears to be

substantially less than for the microstrip patch antennas treated above. Additional simulations

of the SIW antenna with Microwave Studio (MWS) from Computer Simulation Technology

(CST), a full 3D solver based on the Finite Element Method (FEM), yield different results. The

simulated directivity corresponds to the results obtained with Advanced Design System (ADS),

but the simulated gain is notably higher. The simulated radiation efficiency is 75%. As stated

above, there is a large electric field across the slot and the substrate losses are considerably less

than for the patch antenna topology. Hence, a higher radiation efficiency is to be expected. In

literature, similar designs, using the cavity-backed SIW antenna topology, show a high radiation

efficiency as well [12].

0 1 2 3 4 5 6 7 8 9

[kVm ]

Figure 4.7: E-field inside the cavity of the SIW antenna at 60 GHz: TE120 resonance.


−15

−10

−5

0

5

10 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(a)

−15

−10

−5

0

5

10 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(b)

−15

−10

−5

0

5

10 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(c)

Figure 4.8: Simulated directivity (solid) and gain (dotted) in the E-plane of the SIW antenna:

(a) at 15 GHz; (b) at 30 GHz and (c) at 60 GHz.

4.3.2 Corner Analysis

In a similar fashion as Section 4.2.2, the sensitivity of the parameters of the SIW antenna at

60 GHz are investigated. Once more, the worst case fabrication error is assumed to be 10% of

the minimum feature size allowed by the manufacturing process, i.e., 20 µm. An error on the

dimensions of the cavity, affects the resonant frequency, as depicted in Figure 4.9(a). An error

on the length of the slot Ls causes a slight shift in resonant frequency and an error on the

width Ws affects the impedance matching. This supports the theoretical analysis performed in

Section 2.3.2. It is clear from Figure 4.9 that the most critical parameters are the dimensions

of the cavity. An inaccuracy of 20 µm on the length of the cavity imposes a resonance shift of

approximately 300 MHz. To ensure that the cavity size is exact, the placement of the via wall

needs to be precise. If the row of vias is skewed, the dimensions of the cavity are altered and

the resonance is shifted. Moreover, if the spacing between the vias no longer complies with the

restrictions (2.14) due to inaccuracies, the equivalence with rectangular waveguides is lost and

the theoretical analysis of Section 2.3.2 no longer holds.


58 59 60 61 62

0

-5

-10

-15

-20

-25

-30

-35

-40

Frequency [GHz]

|S11|[

dB

]

L− 20 µmL+ 20 µm

Final Design

58 59 60 61 62

0

-5

-10

-15

-20

-25

-30

-35

-40

Frequency [GHz]

|S11|[

dB

]

W − 20 µmW + 20 µm

Final Design

(a)

58 59 60 61 62

0

-5

-10

-15

-20

-25

-30

-35

-40

Frequency [GHz]

|S11|[

dB

]

Ls − 20 µmLs + 20 µm

Final Design

58 59 60 61 62

0

-5

-10

-15

-20

-25

-30

-35

-40

Frequency [GHz]

|S11|[

dB

]

Ws − 20 µmWs + 20 µmFinal Design

(b)

Figure 4.9: Effect of a 20 µm fabrication error on 60 GHz SIW antenna: (a) error on dimensions

of the cavity affects the resonant frequency; (b) error on slot length Ls slightly shifts resonant

frequency and error on Ws affects impedance matching.

4.4 Transmission Line Test Structures for Deembedding

As discussed in Chapter 3, the effects of the feed line structure that excites the antenna, and of

the connector used for measurements, can be eliminated by exploiting a deembedding technique.

For this, reference measurements need to be performed on microstrip transmission lines, using

the same measurement connectors (Figure 3.2). It is stated in Section 3.2 that if a suitable

choice is made for the lengths of the lines, the effect of the connector and the entire feed line

can be eliminated; the shortest line needs to have a length equal to twice the length of the

feed line. The length of the feed is equal for all antenna test structures, as is apparent from

Table 4.4. Taking this into account, the reference transmission lines are designed as depicted

in Figure 4.10. The optimized footprint for the connector, as illustrated in Figure 4.2, is used

as well.


l2 = 51.5 mm

l1 = 2Lfeed = 21.5 mm

Figure 4.10: Designed reference transmission lines for deembedding of connector and feed line

structure.

4.5 Measurements

All S-parameter measurements presented in this section are performed with an N5247A Pro-

grammable Network Analyzer (PNA-X) from Keysight Technologies [21]. All reflection coeffi-

cients depicted below are the characteristics of the stand-alone antennas as discussed in Sec-

tion 3.3, i.e., deembedding of the connector and feed line is already performed. The deembedding

algorithm yields the characteristics of the end launch connector, as presented in Figure 4.11. It

is clear from Figure 4.11(b) that the connector introduces a maximum attenuation of 2 dB at

67 GHz. Looking at the overall trend of the reflection coefficient in Figure 4.11(a), it is perceived

that the matching deteriorates for rising frequencies. The fluctuations of the characteristics of

the connector are a result of the deembedding procedure. At each frequency where one of the

reference microstrip lines corresponds to a multiple of half a wavelength, a singularity occurs.

This also causes an additional ripple in the characteristics of the stand-alone antennas (see

further).


10 20 30 40 50 60

0

-10

-20

-30

Frequency [GHz]

|S11|[

dB

]

(a)

10 20 30 40 50 60

0

-0.5

-1

-1.5

-2

-2.5

Frequency [GHz]

|S21|[

dB

]

(b)

Figure 4.11: Characteristics of the end launch connector as obtained with the deembedding

algorithm: (a) reflection coefficient; (b) insertion loss.

4.5.1 Microstrip Patch Antenna

The measured reflection coefficients for the three microstrip patch antenna test structures are

presented in Figure 4.12. For the designs at 15 GHz and 30 GHz a distinct resonant peak is

visible; however, not exactly at the desired frequency. For both patch antennas, the resonance

frequency is too low, which indicates the substrate is not sufficiently characterized at these

high frequencies. Taking into account (2.4), if the resonant frequency is lower than the simu-

lated characteristics, this implies that the specified value for the relative permittivity εr of the

substrate in Table 4.1 is too low. By comparing the simulated reflection coefficients with the

measurements, the correct value of εr for both operating frequencies can be extracted.

As opposed to the designs at 15 GHz and 30 GHz, the resonance frequency of the patch at

60 GHz is shifted upwards. The fabricated prototype at 60 GHz is presented in Figure 4.13.

The manufacturing process involves a copper etching step, which introduces a certain degree of

overetching. Due to the small dimensions of the patch, the overetching can slightly reduce the

width. The latter determines the resonant frequency, hence the inaccuracy shifts the resonance,

as discussed in Section 4.2.2. The width is reduced due to the fabrication process and the

resonant frequency is displaced to higher frequencies, corresponding to (2.4). The patch at

60 GHz is by far the most sensitive to overetching, hence it is plausible that the fabrication

process is a more dominant factor than the inaccurately characterized εr.

The matching has deteriorated as compared to the simulation, as is apparent from Figure 4.12.

The width of the feed line structure exciting the antenna at 60 GHz needs to be 350 µm to obtain

an impedance of 50 Ω. From Figure 4.13 it is clear that the width of the feed line is substantially

smaller than the desired value of 350 µm, which is again caused by the inaccuracy of the etching

process. Moreover, the insets of the fabricated patch have a circular shape, instead of the desired

sharply defined corners. Hence, the isotropic character of the etching affects the matching as

well.


It is apparent from Figure 4.12(c) that the reflection coefficient remains rather low in the re-

gion between 58 GHz and 61 GHz. As discussed above, the matching of the patch to the feed

line structure is expected to be poor and the connector at the other end of the line introduces

mismatch as well. It is plausible that this evokes standing waves in the feed line; hence, the

microstrip is in resonance, which explains the low reflection coefficient. Moreover, the appar-

ent resonance at approximately 62 GHz can very well be caused by the same standing wave

phenomenon instead of a resonance of the patch.

14 14.5 15 15.5 16

0

-5

-10

-15

-20

Frequency [GHz]

|S11|[

dB

]

MeasuredSimulated

(a)

28 29 30 31 32

0

-5

-10

-15

-20

Frequency [GHz]

|S11|[

dB

]MeasuredSimulated

(b)

58 59 60 61 62 63 64

0

-5

-10

-15

-20

-25

-30

Frequency [GHz]

|S11|[

dB

]

MeasuredSimulated

(c)

Figure 4.12: Measured reflection coefficient for microstrip patch test structures: (a) at 15 GHz;

(b) at 30 GHz and (c) at 60 GHz.


300 µm

Overetching

Figure 4.13: Fabricated microstrip patch antenna at 60 GHz.

4.5.2 Cavity-Backed SIW Antenna

The measured reflection coefficients for the cavity-backed SIW antenna test structures are pre-

sented in Figure 4.14. The designs at 15 GHz and 30 GHz both exhibit a distinct resonance,

albeit at a frequency lower than the desired resonance. As already discussed in Section 4.5.1,

the shift of the resonance to lower frequencies is explained by the discrepancy in the relative

permittivity εr of the substrate material. Yet again, the antenna at 60 GHz exhibits the opposite

behavior; the resonance has shifted to a higher frequency. Figure 4.15 depicts the fabricated

SIW antenna at 60 GHz. As the frequency of the resonant TE120 mode (see Section 4.3) is

determined by the size of the cavity, the via placement needs to be very precise to achieve the

desired behavior. It is clear from Figure 4.15 that the via rows are skewed. This means the

cavity no longer has the desired dimensions, hence the resonance shifts.

As opposed to the microstrip patch antenna discussed in Section 4.5.1, the resonant frequency

is no longer subject to the effects of overetching, as it is determined by the cavity size. However,

overetching still affects the matching to the feed line structure; the dimensions of the insets need

to ensure a 50 Ω GCPW structure. It is clear that the isotropic nature of the etching process

influences the shape of the insets. From Figure 4.15 it can also be perceived that the microstrip

part of the feed line structure is too narrow to correspond with a 50 Ω impedance level. At

60 GHz the desired width of a 50 Ω microstrip line is 350 µm for this substrate material, while

the width of the fabricated specimen is approximately 290 µm. The matching of the fabricated

antenna has obviously worsened as compared to the simulations.


14 14.5 15 15.5 16

0

-5

-10

-15

-20

Frequency [GHz]

|S11|[

dB

]

MeasuredSimulated

(a)

29 29.5 30 30.5 31

0

-5

-10

-15

-20

-25

-30

Frequency [GHz]

|S11|[

dB

]

MeasuredSimulated

(b)

59 60 61 62 63 64

0

-5

-10

-15

-20

-25

Frequency [GHz]

|S11|[

dB

]

MeasuredSimulated

(c)

Figure 4.14: Measured reflection coefficient for cavity-backed SIW antenna test structures: (a)

at 15 GHz; (b) at 30 GHz and (c) at 60 GHz.

290 µm

Figure 4.15: Fabricated cavity-backed SIW antenna at 60 GHz.


4.6 Topology for Antenna Array

As discussed in Chapter 1, one of the main goals of this master’s dissertation is to develop a

highly compact and integratable antenna array. If the antenna system is to be integrated together

with other, likely active, electronic components, high isolation is a necessity. This ensures the

integrity of all signals and prevents crosstalk. As mentioned before, the SIW antenna provides

high isolation as a consequence of the metallized via wall forming the cavity. An additional

benefit of the high isolation is that the antenna elements in the array can be spaced closer

together without invoking high mutual coupling, as is clarified in Chapter 5.

A second goal is to ameliorate the bandwidth of the inherently bandlimited antennas. Different

techniques can be applied to enhance the bandwidth of both technologies. E.g., for the SIW

antenna, a technique based on hybrid modes can be exploited, whilst maintaining high isolation

and easy integration.

Even though losses are substantial at 60 GHz, the attenuation of the signal needs to be kept to a

minimum. It is proven through simulation and measurement that the SIW antenna has superior

performance over the patch when it comes to insertion loss. Considering the additional simu-

lation results obtained with MWS from CST, as discussed in Section 4.3.1, and the precedents

found in literature, the radiation efficiency of the SIW antenna is expected to be better as well.

Taking all of the above into account, the SIW technology is elected for the design of the 60 GHz

antenna array in the next chapter.

HYBRID MODE SIW ANTENNA ARRAY 32

Chapter 5

Hybrid Mode SIW Antenna Array

5.1 Hybrid Mode Cavity-Backed SIW Antenna

As proven in Chapter 4, cavity-backed SIW antennas are narrowband devices. The impedance

bandwidth depends on the matching condition, which is proportional to the antenna profile.

Due to the low-profile and high Q-factor of the resonant cavity, it is hard to achieve wideband

impedance matching when the cavity is only resonating at a single frequency. To overcome this

inherent disadvantage, a bandwidth enhancement technique based on hybrid modes is exploited.


The bandwidth of a cavity-backed SIW antenna can be enhanced by simultaneously exciting two

hybrid modes inside the cavity and merging them within the desired frequency range as proposed

in [20]. The dominant fields of the hybrid modes are located in different half parts of the SIW

cavity, which are defined by the location of the slot. If the slot is off-center, the half parts will

have different dimensions and consequently support hybrid modes at different frequencies. By

tuning the size of both half cavities, the hybrid modes can be shifted in frequency and brought

close together. This will notably increase the impedance bandwidth of the antenna.

In essence, the hybrid modes are two different combinations of the TE110 and TE120 cavity

resonances. The hybrid mode with the lowest frequency, which is dominant in the largest

half cavity, is a combination of a strong TE110 and a weak TE120 resonance, as illustrated in

Figure 5.1(a). Accordingly, the hybrid mode with the highest frequency is dominant in the

smallest half cavity and consists of a strong TE120 and a weak TE110 resonance, as presented

in Figure 5.1(b). The field distribution of the higher hybrid mode (Figure 5.1(b)) displays a

large difference in phase and magnitude between the fields in both half cavities, hence effective

radiation can be generated from the slot. Although the fields are in phase for the lower hybrid

mode, radiation can still be generated due to the high difference in magnitude.


(a)

(b)

Figure 5.1: Field distribution in the SIW cavity: (a) Dominant E-field distribution of lower

hybrid mode in largest half cavity; (b) Dominant E-field distribution of higher hybrid mode in

smallest half cavity [20].

5.1.2 Substrate Material

One of the main goals of this master’s dissertation is to develop a compact and highly integratable

antenna system. To push the integration aspect even further, a different substrate material than

for the antenna test structures is selected. For the design of the hybrid mode SIW array, a very

thin and flexible material is opted for. Concretely, DuPont™Pyralux® AP [22] flexible circuit

material is used. The design is made in duplicate; once on Pyralux AP9141 (50 µm substrate

thickness) and once on Pyralux AP8525 (100 µm substrate thickness). This way, the effects of

scaling down the substrate height can be assessed. The main characteristics of both materials

are summarized in Table 5.1.

AP9141 AP8525

εr 3.4 (1 MHz) 3.4 (1 MHz)

tan δ 0.002 (1 MHz) 0.002 (1 MHz)

thickness 100 µm 50 µm

Cu thickness 35 µm 18 µm

Table 5.1: Characteristics of Pyralux AP9141 and AP8525 flexible substrate material.

In [22], the dielectric constant and loss tangent are characterized up until 20 GHz. As the

array will operate at 60 GHz, additional characterization of the substrate material at this high

frequency is needed. To this end, a simple microstrip patch antenna at 60 GHz, with a well

defined resonant peak, will be fabricated on both substrates as well. By comparing the measured

characteristic of the patch with the simulations, the loss tangent and dielectric constant at


60 GHz can be extracted.

One must take into account that the substrate materials used here are much thinner than the

ones used for the antenna test structures in Chapter 4. As discussed in Section 2.1, a thinner

substrate will decrease the maximum achieveable bandwidth.

5.1.3 Design

A general configuration for a hybrid mode cavity-backed SIW antenna is depicted in Figure 5.2.

The design procedure is very similar to that of a common cavity-backed SIW antenna. Firstly,

the dimensions of the cavity are obtained using (2.15), where the resonant frequency of the

TE110 mode needs to be 60 GHz. This yields an effective width Weff and length Leff, from which

the physical length Lc and width Wc of the cavity can be calculated using (2.11). Yet again, the

restrictions on the via spacing and diameter, as given in (2.14), need to be fulfilled to conserve

the equivalence with rectangular waveguides.

Once the cavity is designed, the slot can be added. A non-resonant slot is used, hence the slot

length Ls is larger than half a wavelength. In Chapter 4, the SIW antennas are designed with a

resonant slot, which implies that the resonant frequency can not only be affected by fabrication

errors on the cavity size, but on the slot as well. By designing the hybrid mode SIW antenna

with a slot length that is substantially larger than half a wavelength, an error on Ls will not

induce a shift of the resonance. The position of the slot determines the dimensions dc,u and dc,l

of the upper and lower half cavity, respectively. As stated before, the frequency of both hybrid

modes depends on the size of both half parts. If the slot is placed in the middle of the cavity, i.e.,

dc,u = dc,l, the hybrid modes coincide and a single resonance is perceived. The hybrid modes

can be pulled further apart by offsetting the slot from the center, i.e., by adjusting dc,u and dc,l.

For this design, it is assumed that dc,u > dc,l, hence the hybrid mode in the upper cavity is a

combination of a strong TE110 and a weak TE120. Equivalently, the mode in the lower cavity

consists of a strong TE120 and a weak TE110.

When dc,l increases, the area of the dominant resonant field of the higher hybrid mode increases,

which implies a decrease in resonant frequency. The area of the weak resonant field of the lower

hybrid mode is enlarged as well, but this only causes a slight decrease in resonant frequency. The

higher hybrid mode decreases in frequency, while the lower hybrid mode’s resonance remains

nearly unchanged, hence the net effect is a decrease of the impedance bandwidth. Alternatively,

when dc,u increases, the resonating area of the dominant field of the lower hybrid mode is

enlarged, hence its resonant frequency decreases. The resonating area of the weak field of the

higher frequency hybrid mode increases as well, but only causes a minor drop in frequency. In

this case, the lower hybrid mode decreases in frequency, while the higher hybrid mode remains

practicaly unchanged. This causes an enlargement of the impedance bandwidth.


Ls Ws

dc,l

dc,u

Lc

Wc

Figure 5.2: General configuration of hybrid mode cavity-backed SIW antenna.


The simulated reflection coefficients, depicted in Figure 5.3, clearly show two distinct resonant

peaks, i.e., two hybrid modes. The hybrid modes have resonant frequencies of 59.16 GHz and

60.38 GHz for the design on 50 µm substrate, and 59.18 GHz and 60.75 GHz for the design on

100 µm. The two modes are merged in the vicinity of 60 GHz, and as such, notably increase

the impedance bandwidth of the antennas. The fractional bandwidths are 3.6% and 4.5% for

the 50 µm and 100 µm design respectively. Taking into account that the bandwidth increases

monotonically with the substrate thickness [23], this implies an increase in impedance bandwidth

of 3% and 3.4%, respectively, as compared to the cavity-backed SIW antenna discussed in

Section 4.3.

58 59 60 61 62

0

-5

-10

-15

-20

-25

-30

-35

59.16 GHz 60.38 GHz

2.2 GHz

Frequency [GHz]

|S11|[

dB

]

(a)

58 59 60 61 62

0

-5

-10

-15

-20

-25

-30

-35

59.18 GHz 60.75 GHz

2.8 GHz

Frequency [GHz]

|S11|[

dB

]

(b)

Figure 5.3: Simulated reflection coefficient for hybrid mode SIW antenna: (a) substrate thickness

50 µm; (b) substrate thickness 100 µm.

Figure 5.4 depicts the simulated directivity and gain patterns in the E-plane for each hybrid


mode of both designs. For both substrate thicknesses, the gain and directivity are lower for the

lower hybrid mode. The higher hybrid mode consists of a strong TE120 and weak TE110, and it

has been proven in [12] that elevated gains are achieved when a TE120 resonance is excited in

the cavity.

−15

−10

−5

0

5

10 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(a)

−15

−10

−5

0

5

10 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(b)

−15

−10

−5

0

5

10 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(c)

−15

−10

−5

0

5

10 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(d)

Figure 5.4: Simulated directivity (solid) and gain (dotted) in the E-plane: (a) at 59.16 GHz for

50 µm substrate; (b) at 60.38 GHz for 50 µm substrate; (c) at 59.18 GHz for 100 µm substrate;

(d) at 60.75 GHz for 100 µm substrate.

The E-field and H-field distributions inside the cavity for the 50 µm design are depicted in

Figure 5.5. From the E-field distribution at 59.16 GHz, presented in Figure 5.5(a), it is clear

that the field is dominant in the upper half cavity. As dc,u > dc,l was assumed in Section 5.1.3,

the upper cavity has the largest dimensions; this supports the fact that the lower hybrid mode

is dominant in the upper half cavity. From the H-field distribution for the lower hybrid mode in

Figure 5.5(b), one can ascertain that the fields in the half parts are in phase. Both the E-field

and the H-field distributions corroborate that the mode is indeed a combination of a strong

TE110 and a weak TE120, with a field distribution as already presented in Figure 5.1(a). An

analogous reasoning can be performed for the higher hybrid mode. The E-field distribution at

60.38 GHz in Figure 5.5(c) reveals that the higher hybrid mode is dominant in the smallest, lower

half cavity. The H-field in both half parts, depicted in Figure 5.5(d), is clearly in antiphase. Yet

again, both the electric and magnetic field confirm that the high frequency hybrid mode is a

combination of a strong TE120 and a weak TE110, which coincides with the theoretical analysis

of Section 5.1.1 and Figure 5.1(b).


(a) (b)

(c) (d)

Figure 5.5: Simulated field distributions in the hybrid mode SIW cavity for 50 µm substrate: (a)

E-field at 59.16 GHz; (b) H-field at 59.16 GHz; (c) E-field at 60.38 GHz; (d) H-field at 60.38 GHz.

5.2 Uniform Linear Antenna Array

As demonstrated by the simulations performed in Chapter 4, the gain of a single antenna element

is limited and the radiation pattern is far from directive. High gain can be achieved by increasing

the electrical size of the antenna. By exploiting an array configuration, the gain can be increased

without enlarging the size of the individual elements. An array is, in essence, a collection of

radiating elements, placed in an electric and geometric configuration. Evidently, a new antenna

with a larger aperture size is formed.

The fields of the array elements will influence each other and the characteristics of an antenna

in an array configuration will deviate from the stand-alone characteristics. This is called mutual

coupling, but is often neglected. Vector addition of the fields of the seperate antenna elements

yields the total field of the array. Depending on the direction, the fields will interfere con-

structively or destructively. By electronically controlling the way the distinct elements are fed,

constructive interference can be obtained in any desired direction; this yields a highly directive

radiation pattern.

There is an abundance of possible array configurations. Here, the Uniform Linear Array (ULA)

configuration is adopted. It consists of N identical antenna elements placed along a line with

constant spacing a. If the total far-field radiation vector of a single antenna element is defined as


F (θ, φ) and mutual coupling is indeed neglected, the radiation pattern of the array only differs

from that of a single element by the array factor T (θ, φ). The latter is only valid in the far-field,

i.e., the fields are observed at distances much larger than the dimension of the aperture. The

array factor for a ULA with evenly spaced antenna elements along the y-axis is defined as [24]

T (θ, φ) =N−1∑n=0

Anejβnejnk0a sinφ sin θ, (5.1)

where An and βn are, respectively, the amplitude and phase by which the nth element is excited,

θ the elevation angle and φ the azimuth angle. All array elements will be excited with equal

phase βn = 0 and magnitude An = A, as to construct a broadside array. Moreover, if one limits

the analysis to the azimuth plane (θ = π2 ), the array factor becomes [24]

T (φ) = A

N−1∑n=0

ejnk0a sinφ

= NAejN−1

2(k0a sinφ) sin

(N2 k0a sinφ

)N sin

(12k0a sinφ

) . (5.2)

Focussing on the amplitude of the array factor, the direction of maximum radiation φ0 is given

by

k0a sinφ = 0

=⇒ φ0 = 0°, −π2≤ φ ≤ π

2, (5.3)

which confirms that the array indeed radiates in the broadside direction. The orientation of the

maximum radiation is independent of the antenna element spacing a. Equivalently, the nulls of

the radiation pattern are given by

φ = arcsin

[± λ

2πa

2m

Nπ

], −π

2≤ φ ≤ π

2, (5.4)

with N being the number of antenna elements in the array and m = 1, 2, 3, .... The argument

of the arcsine cannot exceed unity, hence there is a limited number of zeroes. The amount of

zeroes is a function of the element spacing a and the number of antennas in the array N . If a

or N increases, the number of nulls in the radiation pattern will increase as well and the width

of the main beam will evidently decrease.

5.3 Array Design

Four hybrid mode cavity-backed SIW antennas, as developed in Section 5.1, are now exploited

to construct a broadside ULA. The array configuration yields elevated gain and the enlarged

aperture provides a narrower mainbeam, i.e., the directivity is raised as well. The elements are


spaced by a distance a and the ports are numbered in ascending order from left to right, as

depicted in Figure 5.6. All elements are excited with identical phase and amplitude to point the

main beam in the broadside direction.

a aa

1 2 3 4

Figure 5.6: Hybrid mode SIW antennas in a four-element broadside ULA with spacing a.

5.3.1 Mutual Coupling Between Array Elements

As discussed in Section 5.2, an antenna array is designed under the assumption that mutual cou-

pling between the seperate elements can be neglected. For this to be true, the array elements

need to be spaced by a certain distance, hence maintaining low coupling. The spacing of the ele-

ments affects the width of the main beam, i.e., the directivity, as proven in Section 5.2. The core

of the antenna elements is the SIW cavity, which provides high isolation of the fields. Because

of this, it is of practical interest to investigate the minimal spacing of the elements for which

the mutual coupling is still low enough to be neglected. It is most likely that, due to the high

isolation, the elements can be spaced more closely, which is beneficial for the integration aspect

of the antenna system. Figure 5.8 depicts the mutual coupling between the array elements for

various spacings a, for the design on 50 µm substrate. Obviously, the mutual coupling increases

as the element spacing a decreases. Comparing Figure 5.8(a) with Figure 5.8(b), an increase

of approximately 4 dB is visible in the S21 characteristic when a decreases from λ2 to λ

4 . The

S21 signifies the mutual coupling between adjacent elements in the array; as the configuration

is symmetrical, this also equals S32 and S43. Even with a spacing of a quarter wavelength the

mutual coupling remains below −25 dB. It stands to reason that the elements can be brought

even closer together. To exploit the integration aspect to the utmost extent, the mutual coupling

is investigated for the case where the via row is reused for adjacent array elements, as illustrated

in Figure 5.7. The simulated mutual coupling is presented in Figure 5.8(c). The mutual coupling

for the design on the 100 µm substrate is equivalent and an analogous reasoning is performed.

Figure 5.7: Via wall is reused for adjacent array elements.


58 59 60 61 62

-10

-20

-30

-40

-50

Frequency [GHz]

Mutu

alco

upling

[dB

]

|S21||S31||S41|

(a)

58 59 60 61 62

-10

-20

-30

-40

-50

Frequency [GHz]

Mutu

alco

upling

[dB

]

|S21||S31||S41|

(b)

58 59 60 61 62

-10

-20

-30

-40

-50

Frequency [GHz]

Mutu

alco

upling

[dB

]

|S21||S31||S41|

(c)

Figure 5.8: Simulated mutual coupling between array elements for 50 µm substrate: (a) element

spacing a = λ2 ; (b) element spacing a = λ

4 ; (c) reusing via wall for adjacent elements.


It is clear from Figure 5.8(c) that the mutual coupling remains below −20 dB, even if the via

wall is reused for adjoining antennas. This particular configuration is only feasible if the amount

of coupling does not profoundly change the frequency characteristic of the array as compared

to the characteristic of the seperate elements. This comparison is depicted in Figure 5.9. It is

clear that for both designs the elevated mutual coupling, induced by reusing the via wall, does

not detrementally impact the frequency characteristic of the array. Moreover, this configuration

yields a more compact and integrateable antenna system.

Figure 5.10 depicts the simulated directivity and gain of the array configuration. The direction

of the main beam is indeed broadside (φ = 0). The nulls in the radiation pattern due to the array

configuration are clearly visible as well; a more directive, narrower beam is achieved as compared

to the stand-alone hybrid mode SIW antenna (Figure 5.4). Furthermore, a notable increase

in gain and directivity can be perceived. For the 50 µm substrate, the gain and directivity

in the broadside direction are respectively 4.6 dBi and 8.6 dBi for the lower hybrid mode at


59.16 GHz, and 7.2 dBi and 12.0 dBi for the hybrid mode at 60.38 GHz. The higher hybrid

mode achieves more gain due to the strong TE120 resonance, in the array configuration as

well. The array designed for the 100 µm substrate exhibits similar performance, as illustrated

in Figures 5.10(c) and 5.10(d).

58 59 60 61 62

0

-10

-20

-30

Frequency [GHz]

|S11|[

dB

]

Reuse via wallSeperate element

(a)

58 59 60 61 62

0

-10

-20

-30

Frequency [GHz]

|S11|[

dB

]

Reuse via wallSeperate element

(b)

Figure 5.9: Simulated reflection coefficients for the hybrid mode SIW broadside ULA as com-

pared to the reflection coefficient of a seperate element: (a) 50 µm substrate (b) 100 µm substrate.

−10

−5

0

5

10

15 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(a)

−10

−5

0

5

10

15 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(b)

−10

−5

0

5

10

15 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(c)

−10

−5

0

5

10

15 dBi

90o

60o

30o

0o

−30o

−60o

−90o

(d)

Figure 5.10: Simulated directivity (solid) and gain (dotted) in the E-plane of the hybrid mode

SIW array: (a) at 59.16 GHz for 50 µm substrate; (b) at 60.38 GHz for 50 µm substrate; (c) at

59.18 GHz for 100 µm substrate; (d) at 60.75 GHz for 100 µm substrate.


5.3.3 Wilkinson Power Divider for Feed Network

To arrive at a broadside ULA, all four antenna elements need to be fed with equal amplitude

and phase. This implies that the incoming power will need to be equally four-way divided. To

achieve this, the Wilkinson Power Divider (WPD) is exploited in a corporate feed configuration.

Corporate feeding is essentialy parallel feeding, i.e., all excitations have the same amplitude and

phase. Three WPDs are used in the configuration as depicted in Figure 5.11. As the power is

divided into two equal parts by every WPD, each divider stage introduces an additional 3 dB

signal loss.

50 ΩVs

Port plane

WPD

WPD

WPD

Figure 5.11: WPDs in corporate feed network for ULA.

The WPD makes use of a quarter-wave transformer to attain impedance matching at its input

port. Figure 5.12 depicts a Wilkinson Power Divider in microstrip technology. The input port

is matched to the two output ports using a quarter wavelength transmission line section with an

impedance of√

2Z0. Isolation between the two output ports is procured by the added lumped

resistor with impedance 2Z0. If power reflects at port 2, it will be absorbed by the resistor

instead of coupling to port 3 and vice versa.

The designed corporate feed network for the 50 µm substrate is depicted in Figure 5.14. The

characteristics are presented in Figure 5.13. The reflection coefficient in the desired frequency

range at the input port (Figure 5.13(a)) is below −40 dB and at the output ports (Figure 5.13(b))

it remains below −25 dB. Hence, excellent matching is achieved. The coupling between two

adjacent output ports (Figure 5.13(c)) is below −20 dB due to the added lumped resistor. The

additional signal loss from input to output induced by the two stage divider is of the order of

6 dB as expected (Figure 5.13(d)).


√2Z0,

λ4

1

2

3

Z0

Z0

Z0

2Z0

Figure 5.12: Microstrip Wilkinson Power Divider: power is split equally from 1 to 2 and 3.

50 52 54 56 58 60 62 64 66 68 70

-10

-20

-30

-40

-50

Frequency [GHz]

|S11|[

dB

]

(a)

50 52 54 56 58 60 62 64 66 68 70

-10

-15

-20

-25

-30

Frequency [GHz]

S22

[dB

]

(b)

50 52 54 56 58 60 62 64 66 68 70

-10

-15

-20

-25

-30

Frequency [GHz]

S32

[dB

]

(c)

50 52 54 56 58 60 62 64 66 68 70

-6.2

-6.4

-6.6

-6.8

Frequency [GHz]

|S21|[

dB

]

(d)

Figure 5.13: Simulated characteristics of designed corporate feed network with cascaded Wilkin-

son power dividers: (a) reflection coefficient at input port; (b) reflection coefficient at an output

port; (c) coupling between two adjacent output ports; (d) insertion loss from input to output

port.


5.3.4 Influence of Feed Network

The designed hybrid mode SIW array with added corporate feed network is depicted in Fig-

ure 5.14. The feed network obviously introduces additional losses, as proven by simulation in

Section 5.3.3 and presented in Figure 5.15. As the signal first needs to travel through the corpo-

rate feed network, a substantial amount of extra attenuation occurs due to copper, substrate and

radiation losses before it reaches the actual antenna. Moreover, the feed network will introduce

additional radiation, which will interfere with the desired radiation coming from the array.

Figure 5.14: Designed hybrid mode SIW array with added corporate feed network.

58 59 60 61 62

0

-5

-10

-15

-20

-25

-30

-35

Frequency [GHz]

|S11|[

dB

]

Figure 5.15: Simulated reflection coefficient for the hybrid mode SIW array with added corporate

feed network.


5.4 Measurements

The fabricated hybrid mode SIW antenna array on 100 µm flexible substrate is depicted in

Figure 5.16. Measurements indicate that the degree of overetching is notably lower than for the

designs in Section 4.5. The width of the 50 Ω microstrip traces is approximately equal to the

desired 210 µm. The holes for the conductive vias are created with a laser ablation process. For

the limited diameter of 200 µm this yields the highest possible accuracy. Compared to the drilled

via rows of the SIW antenna in Figure 4.15, it is clear that the placement with laser ablation is

notably more accurate. As Uniform Thickness Copper Plating (UTCP) processing of the vias

is not feasible for such a small quantity of prototypes, grounding of the top layer is provided by

injecting electrically conductive paste into the holes and pressing it down with a stencil. The

performance of this conducting paste is not well known at frequencies as high as 60 GHz, hence

some undesired effects are highly plausible to occur.

3 mm

Figure 5.16: Fabricated hybrid mode SIW antenna on 100 µm substrate.

The fabricated antenna array is presented in more detail in Figure 5.17. As explained above, the

conductive paste is pushed into the holes to ensure a good connection. Doing so, some spillage of

the paste ends up in the slot and the insets. After fabrication, the insets are completely sealed,

i.e., reducing the inset depth to zero, which is detrimental for the impedance matching. The

slots are partially occupied by the conductive material as well, plausibly affecting the proper

operation of the antenna array. As explained in Section 2.3.2, the slot is an important tuning

element for the impedance matching. Attempts have been made to clear both the insets and

the slot of the paste spillage. Still, some residue remains, as depicted in Figure 5.17.


paste in inset

paste in slot

850 µm

Figure 5.17: Detail of fabricated hybrid mode SIW antenna array.

Measurements are performed using the SouthWest Microwave end launch connectors, discussed

in Section 4.1, and an N5247A PNA-X from Keysight Technologies [21]. These revealed that

standing waves occur in the corporate feed network of the array, as depicted in Figure 5.18. A

resonance occurs, approximately, every 5 GHz, which corresponds to a wavelength of circa 3.2 cm

inside the substrate material. The combined length of the microstrip feed line and the corporate

feed network corresponds to approximately half a wavelength at 5 GHz. This implies that a

resonance occurs whenever the length of the feed structure is a multiple of half a wavelength. A

standing wave is formed when two waves of the same frequency propagate in opposite directions,

i.e., when reflections occur at both ends of the feed network. Hence, the impedance of the SIW

antenna has changed significantly and is no longer properly matched to the 50 Ω feed line. This

can be attributed to the discrepancy in the size of the insets and the slot. Another hypothesis is

that the impedance of the SIW antenna has drastically changed due to the non-ideal electrical

properties of the paste at these high frequencies. The standing wave signifies that the impedance

matching at the connector end of the line is not sufficient either. As the dimensions of the 50 Ωmicrostrip are accurate, this indicates that there is an amount of mismatch at the junction

between the connector and the circuit. The thickness of the flexible substrate is substantially

less than in Chapter 4. Although proper operation for any board thickness is assured in [17], it

stands to reason that the press fit of the connector no longer ensures a good connection. Also,

the characteristics of the substrate are not precisely known at 60 GHz; if these have gravely

changed as compared to the values in Table 5.1, that could explain the faulty matching as well.


40 45 50 55 60 65

0

-5

-10

-15

-20

-25

-30

-35

Frequency [GHz]

|S11|[

dB

]

Figure 5.18: Measured reflection coefficient for the fabricated hybrid mode SIW array on 50 µm

substrate.

To gain insight into the effects at hand, one of the reference microstrip lines for deembedding the

connector and feed line structure is measured. The characteristics are depicted in Figure 5.19.

The reflection coefficient in Figure 5.19(a) demonstrates that the matching of the line to the con-

nector is poor, i.e., above −10 dB on average. From the measured insertion loss in Figure 5.19(b)

it is clear that a considerable amount of attenuation is introduced. As stated above, the width of

the trace is accurate, hence the faulty matching is not caused by a fabrication inaccuracy. This

implies that the additional losses and inferior matching are due to the combination of unknown

substrate and connector characteristics.

To quantify the amount of additional losses that is introduced, the power balance of the line is

calculated. For an ideal, lossless transmission line it holds that [25]

|S11|2 + |S21|2 = 1, (5.5)

which denotes that all incident power at port 1 is either reflected back to the source or trans-

mitted to port 2. For a lossy transmission line this power balance becomes

|S11|2 + |S21|2 + Ploss + Prad = 1, (5.6)

where Ploss is the normalized power that is dissipated due to substrate and copper losses, and

Prad is the normalized power radiated into free space. Figure 5.20 depicts the power balance for

the measured reference microstrip line. It is clear that a substantial amount of power is lost,

70% on average. This indicates that the substrate material is notably more lossy than suggested

by the characteristics in Table 5.1, hence it is insufficiently characterized at 60 GHz.


50 52 54 56 58 60 62 64 66 68

0

-10

-20

-30

-40

Frequency [GHz]

|S11|[

dB

]

(a)

50 52 54 56 58 60 62 64 66 68

-5

-6

-7

-8

-9

Frequency [GHz]

|S21|[

dB

]

(b)

Figure 5.19: Measured characteristics of the reference microstrip line for deembedding: (a)

reflection coefficient; (b) insertion loss.

50 52 54 56 58 60 62 64 66 680

0.2

0.4

0.6

0.8

1

Frequency [GHz]

|S11|2

+|S

21|2

Figure 5.20: Power balance for the measured reference microstrip line.

CONCLUSION AND FUTURE RESEARCH 49

Chapter 6

Conclusion and Future Research

The goal of this master’s dissertation was to design a highly compact and integratable antenna

array that operates in the 60 GHz band, whilst maintaining compatibility with standard printed

circuit board processing steps. Firstly, antenna test structures, in both microstrip patch and

Substrate Integrated Waveguide (SIW) technology, at different frequencies were developed to

make an assessment of the impact of high frequency effects. This brought forth a comparative

study of both technologies, based on both simulation and measurement results, and enabled the

formulation of a founded opinion that SIW is the most advantageous technology for the design

of the array.

The analysis of the SIW antenna test structures has confirmed the inherent band limited be-

havior of this technology, which is due to the high Q-factor of the resonance and the low profile.

A bandwidth enhancement technique based on hybrid mode excitation was proposed and suc-

cessfully executed. Simulated fractional impedance bandwidths of 3.6% and 4.5% were achieved

at 60 GHz on extremely thin substrates. To push the integration aspect of the dissertation to

the utmost extent, it was opted to design and fabricate the hybrid mode SIW antenna array

on 50 µm and 100 µm flexible substrate material. Moreover, exploiting a Uniform Linear Array

(ULA) configuration with four antenna elements, high gain and directivity were achieved, i.e.,

7.2 dBi and 12.0 dBi, respectively.

The insufficient characterization of the flexible substrate at 60 GHz and the non-ideal processing

of the minute vias were detrimental for the operation of the fabricated prototypes. Future

research on this topic could certainly encompass investigating the characteristics of flexible

substrates for antenna design at 60 GHz, or even research towards the development of novel

flexible substrate materials for use in the Extremely High Frequency (EHF) band.

The fabricaton process aside, future development can certainly cover a beam forming system,

which could be utilized in, for example, systems that provide a Wireless Local Access Network

(WLAN) in the 60 GHz band in office environments. Due to the beam forming, spacial selectivity

is achieved; this enables multiple systems to work side-by-side without causing interference.

Higher bandwidths, and correspondingly higher bit rates, can be offered to the mobile user.

Further research could be performed towards phase shifter circuits operating at 60 GHz as well.

CONCLUSION AND FUTURE RESEARCH 50

Combined with the hybrid mode SIW array and a Digital Signal Processing (DSP) unit, an

adaptive beam steering system could be developed. Adjusting the respective phases of the

excitations of the antennas in the array allows for scanning. This means the direction of maxi-

mum radiation is moved electronically, which could find its application in millimeter wave radar

detection systems.

If the bandwidth of the hybrid mode SIW antenna can be enlarged even further or research is

performed towards Ultra Wideband (UWB) SIW antennas, radar operation based on very short

pulses is possible. Applications could be, for example, an indoor positioning system. Due to

the UWB operation of the antenna, the round trip time of a very short pulse can be measured,

which allows triangulation.

BIBLIOGRAPHY 51

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[3] D. M. Pozar, Microwave Engineering. John Wiley and Sons Inc., 2012.

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[5] C. A. Balanis, Antenna Theory: Analysis and Design. John Wiley and Sons Inc., 2005.

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BIBLIOGRAPHY 52

[14] G. P. Srivastava and V. L. Gupta, Microwave Devices and Circuit Design. Prentice-Hall

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[15] F. Declercq, H. Rogier, and C. Herteleer, “Permittivity and Loss Tangent Characterization

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[17] Southwest Microwave. (2007) Southwest Microwave End Launch Connector [Online]. http:

//mpd.southwestmicrowave.com/products/endLaunch.php.

[18] ——. (2007) Optimizing Test Boards for 50 GHz End Launch Connectors, Datasheet [On-

line]. http://mpd.southwestmicrowave.com/showImage.php?image=439.

[19] R. S. Elliott, Antenna Theory and Design. John Wiley and Sons Inc., 2003.

[20] G. Q. Luo, Z. F. Hu, W. J. Li, X. H. Zhang, L. L. Sun, and J. F. Zheng, “Bandwidth-

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[22] DuPont™. DuPont™Pyralux® AP flexible circuit materials, Technical Datasheet

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[23] D. M. Pozar and D. H. Schaubert, Microstrip Antennas: The Analysis and Design of

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http://mpd.southwestmicrowave.com/products/endLaunch.php

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http://www.dupont.com/content/dam/assets/products-and-services/electronic-electrical-materials/assets/PyraluxAPclad_DataSheet.pdf

LIST OF FIGURES 53

List of Figures

2.1 Simple microstrip patch antenna configuration. . . . . . . . . . . . . . . . . . . . 4

2.2 Fringing E-fields as radiation mechanism for a patch antenna. . . . . . . . . . . . 5

2.3 Microstrip patch antenna with insets for impedance matching. . . . . . . . . . . . 7

2.4 SIW structure: conductive vias connecting parallel copper planes. . . . . . . . . . 7

2.5 SIW resonant cavity with microstrip line feed. . . . . . . . . . . . . . . . . . . . . 9

3.1 Cascade of two arbitrary two-port networks. . . . . . . . . . . . . . . . . . . . . . 11

3.2 Microstrip lines with coax-to-microstrip transitions SA and SB and ideal lossy

transmission line section SL. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

3.3 Piece of 50 Ω transmission line with connector modeled as 35 Ω discontinuity. . . 13

3.4 S-parameter simulation performed on modeled connector. . . . . . . . . . . . . . 13

3.5 Characteristic of model connector: extracted with deembedding algorithm (solid

line) and simulated with circuit solver (dashed line). . . . . . . . . . . . . . . . . 14

3.6 Block diagram of measurement to be performed. . . . . . . . . . . . . . . . . . . 14

4.1 Southwest Microwave 1.85 mm end launch connector. . . . . . . . . . . . . . . . . 17

4.2 Connector footprint with GCPW launch structure. . . . . . . . . . . . . . . . . . 17

4.3 Simulated reflection coefficient for patch antenna test structures: (a) at 15 GHz;

(b) at 30 GHz and (c) at 60 GHz. . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

4.4 Simulated directivity (solid) and gain (dotted) in the E-plane of the antenna: (a)

at 15 GHz, (b) at 30 GHz and (c) at 60 GHz. . . . . . . . . . . . . . . . . . . . . . 20

4.5 Effect of a 20 µm fabrication error on the 60 GHz patch: (a) error on W affects

the resonant frequency; (b) error on R affects the matching. . . . . . . . . . . . . 20

4.6 Simulated reflection coefficient for cavity-backed SIW antenna test structures: (a)

at 15 GHz; (b) at 30 GHz and (c) at 60 GHz. . . . . . . . . . . . . . . . . . . . . . 22

4.7 E-field inside the cavity of the SIW antenna at 60 GHz: TE120 resonance. . . . . 23

4.8 Simulated directivity (solid) and gain (dotted) in the E-plane of the SIW antenna:

(a) at 15 GHz; (b) at 30 GHz and (c) at 60 GHz. . . . . . . . . . . . . . . . . . . 24

LIST OF FIGURES 54

4.9 Effect of a 20 µm fabrication error on 60 GHz SIW antenna: (a) error on dimen-

sions of the cavity affects the resonant frequency; (b) error on slot length Ls

slightly shifts resonant frequency and error on Ws affects impedance matching. . 25

4.10 Designed reference transmission lines for deembedding of connector and feed line

structure. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

4.11 Characteristics of the end launch connector as obtained with the deembedding

algorithm: (a) reflection coefficient; (b) insertion loss. . . . . . . . . . . . . . . . 27

4.12 Measured reflection coefficient for microstrip patch test structures: (a) at 15 GHz;

(b) at 30 GHz and (c) at 60 GHz. . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

4.13 Fabricated microstrip patch antenna at 60 GHz. . . . . . . . . . . . . . . . . . . . 29

4.14 Measured reflection coefficient for cavity-backed SIW antenna test structures: (a)

at 15 GHz; (b) at 30 GHz and (c) at 60 GHz. . . . . . . . . . . . . . . . . . . . . . 30

4.15 Fabricated cavity-backed SIW antenna at 60 GHz. . . . . . . . . . . . . . . . . . 30

5.1 Field distribution in the SIW cavity: (a) Dominant E-field distribution of lower

hybrid mode in largest half cavity; (b) Dominant E-field distribution of higher

hybrid mode in smallest half cavity [20]. . . . . . . . . . . . . . . . . . . . . . . . 33

5.2 General configuration of hybrid mode cavity-backed SIW antenna. . . . . . . . . 35

5.3 Simulated reflection coefficient for hybrid mode SIW antenna: (a) substrate thick-

ness 50 µm; (b) substrate thickness 100 µm. . . . . . . . . . . . . . . . . . . . . . 35

5.4 Simulated directivity (solid) and gain (dotted) in the E-plane: (a) at 59.16 GHz

for 50 µm substrate; (b) at 60.38 GHz for 50 µm substrate; (c) at 59.18 GHz for

100 µm substrate; (d) at 60.75 GHz for 100 µm substrate. . . . . . . . . . . . . . . 36

5.5 Simulated field distributions in the hybrid mode SIW cavity for 50 µm substrate:

(a) E-field at 59.16 GHz; (b) H-field at 59.16 GHz; (c) E-field at 60.38 GHz; (d)

H-field at 60.38 GHz. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

5.6 Hybrid mode SIW antennas in a four-element broadside ULA with spacing a. . . 39

5.7 Via wall is reused for adjacent array elements. . . . . . . . . . . . . . . . . . . . . 39

5.8 Simulated mutual coupling between array elements for 50 µm substrate: (a) ele-

ment spacing a = λ2 ; (b) element spacing a = λ

4 ; (c) reusing via wall for adjacent

elements. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40

5.9 Simulated reflection coefficients for the hybrid mode SIW broadside ULA as com-

pared to the reflection coefficient of a seperate element: (a) 50 µm substrate (b)

100 µm substrate. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

5.10 Simulated directivity (solid) and gain (dotted) in the E-plane of the hybrid mode

SIW array: (a) at 59.16 GHz for 50 µm substrate; (b) at 60.38 GHz for 50 µm

substrate; (c) at 59.18 GHz for 100 µm substrate; (d) at 60.75 GHz for 100 µm

substrate. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

LIST OF FIGURES 55

5.11 WPDs in corporate feed network for ULA. . . . . . . . . . . . . . . . . . . . . . . 42

5.12 Microstrip Wilkinson Power Divider . . . . . . . . . . . . . . . . . . . . . . . . . 43

5.13 Simulated characteristics of designed corporate feed network with cascaded Wilkin-

son power dividers: (a) reflection coefficient at input port; (b) reflection coefficient

at an output port; (c) coupling between two adjacent output ports; (d) insertion

loss from input to output port. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43

5.14 Designed hybrid mode SIW array with added corporate feed network. . . . . . . 44

5.15 Simulated reflection coefficient for the hybrid mode SIW array with added cor-

porate feed network. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44

5.16 Fabricated hybrid mode SIW antenna on 100 µm substrate. . . . . . . . . . . . . 45

5.17 Detail of fabricated hybrid mode SIW antenna array. . . . . . . . . . . . . . . . . 46

5.18 Measured reflection coefficient for the fabricated hybrid mode SIW array on 50 µm

substrate. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47

5.19 Measured characteristics of the reference microstrip line for deembedding: (a)

reflection coefficient; (b) insertion loss. . . . . . . . . . . . . . . . . . . . . . . . . 48

5.20 Power balance for the measured reference microstrip line. . . . . . . . . . . . . . 48

LIST OF TABLES 56

List of Tables

4.1 Characteristics of Rogers RO4350B High Speed Laminate. . . . . . . . . . . . . . 16

4.2 Dimensions of designed microstrip patch antenna test structures. . . . . . . . . . 18

4.3 Via diameter and spacing maxima for SIW operation. . . . . . . . . . . . . . . . 21

4.4 Dimensions of designed cavity-backed SIW antenna test structures. . . . . . . . . 22

5.1 Characteristics of Pyralux AP9141 and AP8525 flexible substrate material. . . . 33

design of a 60 ghz hybrid mode substrate integrated...

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