design feature mmds amplifier s-band amplifier … · microwaves & rf november 2000 55 design...
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MICROWAVES & RF ■ NOVEMBER 2000
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The development of this amplifierincludes the modeling and the charac-terization of the “twin” gallium-arsenide metal-semiconductor-field-effect-transistor (GaAs MESFET)device, NES2427P-60 used in thisdesign.1 This article also presents thedesign methodology of the externalinput- and output-matching circuitsof the GaAs device and a two-armbranch 90-deg. coupler. The perfor-mance of this compact amplifier ispresented and discussed.
The goal was to design a compactlinear PA using the 60-W Class A-BNES2427P-60 from NEC/CEL thatdelivers a power of +40 dBm for eachtone with an IMD3 of less than �40
dBc over the 2.5-to-2.7- GHz, S-bandinstantaneous bandwidth. Therequirement for the typical gain at 1-dB compression (P1dB) was 11 dBwith a gain flatness of better than�0.5 dB. The target for the typicaloutput power at P1dB was +48 dBm.
The amplifier-circuit optimizationfor linearity and the accurate predic-tion of the amplifier’s performancerequire an available device model andcharacterization for a two-tone sig-nal. The device was modeled andcharacterized with a two-tone signalat an output-power level of +40 dBm
Shansong Song andRaymond BassetCalifornia Eastern Laboratories, Inc.,4590 Patrick Henry Dr., Santa Clara,CA 95054-1817.
S-Band AmplifierModeled ForWireless Data A 70-W, 2.5-to-2.7-GHz (S-band)
balanced amplifier offers the linearperformance for next-generationwireless applications.
DESIGN FEATURE
MMDS Amplifier
MULTICHANNEL-MULTIPOINT-DISTRIBUTION-SYSTEM (MMDS)and wireless-data/Internet technologies are the latest transportmedia for television broadcasting and the Internet. Since thesetechnologies require the use of reliable linear power amplifiers
(PAs), a compact amplifier was designed based on device modeling andcharacterization with a two-tone signal for optimum third-order intermod-ulation distortion (IMD3) at a defined output power. The amplifier config-uration was also an important factor in obtaining good linearity, stability,and external matching, along with low circuit loss.
1. A complete balanced, linear-amplifier circuit based around theNES2427P-60 “twin” GaAs MESFET,the device incorporates a splitter andcombiner in the package shown.
FrequencyGHz
2.4
2.5
2.6
2.7
RIN(ohms)
4.9
4.0
8.0
6.7
XIN(ohms)
�9.1
�4.2
1.7
�2.4
ROUT(ohms)
17.2
21.9
18.4
16.8
XOUT(ohms)
16.4
12.3
10.1
11.1
Table 1: Half-device impedancesversus frequency
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DESIGN FEATURE
each tone in the 2.4-to-2.7-GHz band. The design challenge was to match
the device’s optimum output im-pedance for IMD3 to a 50-� load overthe 2.5-to-2.7-GHz bandwidth with anexcellent return loss (better than 18dB) and to minimize the loss of theoutput matching and combining cir-cuit. The input circuit is not as criticalas the output since it does not signifi-cantly affect the output power of theamplifier. Its matching and loss mustbe excellent only at the maximum fre-quency for gain consideration. Stabil-ity is always an issue with high-poweramplifiers. The splitter and combinerhave to be selected for high isolation(20 dB), low loss (0.25 dB), good bal-ance, and ease of integration to theamplifier layout. The choice of theamplifier configuration is important ifa good external match is needed toisolate the driver from the powerstage.
A MESFET MODELThe device used
in this amplifier (aClass A-B MES-FET “twin” de-vice) consists oftwo separatedpairs of chips withtheir internalmatching circuitsmounted in thesame package.The package hastwo gate connec-tions and twodrain connectionswithout any inter-nal connection
between the twosides of the device.These devices ares o m e t i m e simproperly called“ p u s h - p u l l ”devices. This con-cept comes fromlower-frequencya p p l i c a t i o n s —high frequency(HF), very-highfrequency (VHF),and ultra-high fre-quency (UHF)—using silicon (Si)devices with in-
ternal connections between the twosides of the device to take advantageof the virtual ground created by thepush-pull configuration.2,3 The high-power GaAs devices at L-band andhigher frequencies do not have theseinternal connections because theirtransverse dimensions are too large.4
Therefore, these devices are not“push-pull.” They are devices withtwo identical sides called (by CEL)“twin” devices. The configuration ofthe amplifier defines only how thedevices operate. They can be com-bined the same way as single-endedparts in balanced configuration (90-deg. hybrid), in push-pull configura-tion (baluns4-6), and in phase combin-ing (Wilkinson), etc.
To model these devices, only oneside is considered. Simple narrow-band external input- and output-matching circuits with biasing cir-cuits were designed. At eachfrequency, the input circuit wastuned for maximum return loss and
MMDS Amplifier
Vds = +10 VDC, Idsq = 12 A, Circuit optimizedfor IMD from 2.5-to-2.7-GHz instantaneous bandwidth
Test condition:
5045
40
35
30
2520
Frequency—GHz2.4 2.5 2.6 2.7 2.8
2019
18
17
16
15
14
PAEIds
Ids
PAE
PAE—
perc
enta
ge
Ids —A
3. Another performance curve of the NES2427P-60, thisone illustrates the amplifier’s PAE and drain-to-sourcecurrent (Ids) at 1-dB gain compression.
Vds = +10 VDC, Idsq = 12 A, Circuit optimizedfor IMD from 2.5-to-2.7-GHz instantaneous bandwidth
Test condition:
50
48
46
44
42
P1dB
—dB
m
Frequency—GHz2.4 2.5 2.6 2.7 2.8
16
13
10
7
4
G1dB—dB
P1dBG1dB
P1dB
G1dB
2. These curves show the P1dB and G1dB performanceover the 2.4-to-2.8-GHz range of the NES2427P-60 amplifierat a Vds of +10 VDC and an Idsq of 12 A.
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DESIGN FEATURE
MMDS Amplifier
the output circuit was optimized forIMD3. The performance, the source,and the load impedances were mea-sured and recorded. These im-pedances are relatively high and donot present any difficulty in measur-ing to good accuracy because thedevice is internally prematched.
Table 1 shows the half-deviceimpedances versus frequency. Underthese conditions and for half thedevice, the following performancewas obtained in the 2.5-to-2.7-GHzbandwidth—P1dB = +45 dBm, IMD3= �41 dBc to �43 dBc at +37 dBmeach tone, and G1dB = 13.3 dB.
4. The overall IMD performance of the NES2427P-60 amplifier is provided inthese sets of curves for IMD3, IMD5, and IMD7 over the 2.5-to-2.7-GHz band.IMD3 remains below -40 dBc for output-power tones up to +41 dBm.
Vds = +10 VDC, Idsq = 12 A, Circuit optimized for IMD from 2.5 to 2.7 GHz
Test condition:
50
40
48
46
44
42
28.0
22.0
20.0
18.0
16.0
14.0
12.029.0 30.0 31.0 32.0 32.9 34.0 35.0 36.0 37.0 38.0 39.0 40.0
P out
—dB
m
Pin—dBm
Ids
Pout
Ids —A
Pout at 2.5 GHzPout at 2.6 GHzPout at 2.7 GHzIds at 2.5 GHzIds at 2.6 GHzIds at 2.7 GHz
5. This set of curves shows the relationship between Pout, Pin, Ids, and Pin forthe amplifier. The measurements were made at three different frequencies foreach set (2.5, 2.6, and 2.7 GHz).
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DESIGN FEATURE
In addition to the device character-ization with a two-tone signal, thedevice was characterized with one-tone signal at P1dB and its S-param-eters were measured under small-sig-nal conditions in the 1.5-to-3.5-GHzband. The S-parameters were usedfor gain and stability analysis undersmall-signal conditions.
Contrary to popular belief, for abandwidth of less than one octave
there are no advantages in using apush-pull amplifier in terms of band-width and linearity. The disadvan-tages are:
1. Low isolation between the twosides of the device if a classical balunis used (only 6 dB).
2. Poor external input and outputmatch.
On the other hand, the balancedconfiguration has the same perfor-
MMDS Amplifier
6. IMD3 as a function of Pout is shown here for various values of Idsq over the2.5-to-2.7-GHz range. The best IMD3 performance is achieved at the highestlevel of Idsq, 12 A.
7. These curves are similar to those of Fig. 6, except that IMD5 is plotted againstpower. And like Fig. 6, IMD performance improves for higher values of Idsq.
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DESIGN FEATURE
MMDS Amplifier
mance as the push-pull concerningbandwidth and linearity (bandwidthless than one octave) and has the fol-lowing advantages:
• High isolation between the twosides of the device.
• Good external match. • Easy-to-design printed 90-deg.
splitters/combiners for narrowbandapplications that can be easily inte-grated with the layout of the amplifier.
• Good reliability—the failure of onedevice side does not result in total fail-ure. Only the output power drops by 6 dB.
For balanced amplifiers, manykinds of couplers that can be used—Lange couplers, Wilkenson with 90-deg. phase jog on one port, two-armbranch 90-deg. hybrid, etc. In this pro-
ject, a two-arm branch 90-deg. hybridwas selected due to its simple layoutand ease in integrating with the ampli-fier layout. Since the input- and out-put-matching circuits realize thematching of the device to 50-�impedance, the hybrid does not per-form any impedance transformationand all its port impedances are 50 �.
The design of the hybrid was per-formed with Hewlett-Packard (nowAgilent) HP multipoint-distributionsoftware (MDS). Also, HP MOMEN-TUM software was used to performthe electromagnetic (EM) simulation.The comparison of the simulatedresults between MDS and MOMEN-TUM software showed a slight differ-ence in the coupler layout. The finalEM simulation indicates that this 90-
8. The curves of IMD7 with respect to Pout follow the same pattern as those ofFigs. 6 and 7, showing how IMD performance improves for higher values ofdrain current.
FrequencyGHz
2.4
2.5
2.6
2.7
Simulated
6.2 + j4.8
7.1 + j4.9
7.4 + j3.6
4.9 + j2.3
Measured
5.0 + j7.4
7.5 + j7.1
8.0 + j4.0
4.6 + j2.8
Simulated
20.2 � j12.3
19.5 � j11.9
18.8 � j11.5
18.2 � j11.2
Measured
16.4 � j8.7
18.4 � j8.9
19.0 � j10.6
17.9 � j12.4
Table 2: Comparison of simulated andmeasured impedances
Source impedance of halfNES2427P-60
Load impedance of halfNES2427P-60
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deg. hybrid has more than 20 dB of iso-lation between its coupled ports and 20dB of return loss for each port. Its bal-ance is better than 0.1 dB in magnitudeand 90 �1 deg. in phase.
The input-matching circuit wasoptimized for flat gain over the 2.4-to-2.7-GHz bandwidth and to providethe best match to a 50-� impedance atthe highest frequency. It consists oftwo sections using only transmissionlines with all stubs being open. Inorder to cover the 2.4-to-2.7-GHzbandwidth, sections with low qualityfactor (Q) were selected for thedesign, especially the first one. A gainslope of �6.0 dB per octave for thedevice was assumed and the input-matching circuit was designed tocompensate for this slope and toobtain an excellent match at 2.7 GHz.The result is a flat gain and a maxi-mum gain over the desired band-width. From the device modeling andcharacterization data the predictedassociated gain was G1dB = 13.0 dBand the gain flatness was 0.5 dB. Thesimulation also showed an input-return loss of more than 12 dB acrossthe bandwidth for half the device.However, the full device return losswill be higher since a balanced config-uration was selected and the twodevice sides are symmetric.
The design goal for the outputmatching network was to present theoptimal load impedance for a two-tone signal at a defined output powerwith minimum loss. Since theNES2427P-60’s device-output opti-mal impedance for a two-tone signalis not far from 50-� impedance, twosections of one-sixteenth wavelengthChebyshev impedance transformercircuit was selected. This circuit min-imizes the dimensions and loss of thematching network. The simulationshowed that the loss of this outputcircuit was less than 0.2 dB and thereturn loss was better than 19 dBacross the bandwidth.
Figure 1 shows the complete bal-anced-amplifier layout. The two-armbranch 90-deg. hybrid, which uses thesame substrate as the matching cir-cuit, is integrated with the amplifierlayout and does not require any addi-tional connection. All the circuits aredirectly printed on the same Er = 2.2,
MMDS Amplifier
DESIGN FEATURE
(continued on p. 160)
31-mil-thick substrate. Standard 50-�loads are connected to the coupler-iso-lated ports. The amplifier is compactand easy to assemble. The totaldimensions of the amplifier are 13.2 �6.0 cm2.
TEST RESULTSThe source and load impedances
presented by the circuit to the devicedirectly from simulation were mea-sured versus frequency with a vectornetwork analyzer (VNA). The exper-imental results showed good agree-ment between the measured and simulated impedances (Table 2).However, a slight tuning was per-formed with the use of a VNA on theinput- and output-matching circuitsto obtain the exact simulated im-pedance values.
The amplifier was tested in a 50-�system with fixed broadband tuningcorresponding to an optimum IMD3performance in CEL’s high-powerautomated setup.7 The device wasbiased at Vds = +10 VDC and Idsq = 12 A. Figure 2 shows the P1dB andG1dB performance versus frequency.Figure 3 provides power-added effi-ciency (PAE) and Ids versus frequen-cy at 1-dB gain compression. It showsthat the amplifier exhibits a typicalPAE of 40 percent from 2.4 to 2.7GHz.
Figure 4 shows the IMD perfor-mance of the amplifier. These curvesshow that the amplifier has good IMDperformance with an IMD3 lowerthan �42 dBc at +40-dBm outputpower, each tone across the 2.5-to-2.7-GHz MMDS bandwidth. Pout andIds versus Pin and frequency areshown in Fig. 5.
The amplifier-input return lossover the 2.5-to-2.7-GHz bandwidthwas better than 15 dB, which is betterthan the expected return loss for apush-pull configuration. The amplifi-er’s IMD performance was measuredwith fixed tuning at a constant, Vds =+10 VDC, and at constant frequencyof 2.7 GHz, versus Pin and Idsq. Fig-ures 6, 7, and 8 show respectivelyIMD3, IMD5, and IMD7. The curvesshow, as expected, that this amplifierexhibits the best IMD3 performanceat any Pout level when biased at thehighest Idsq, 12 A. The maximum Idsq
is limited only at Vds = +10 VDC bythe maximum recommended channeltemperature which is 150°C.
The mean time to failure (MTTF) ofGaAs power devices is limited bytheir channel temperature. It isimportant for amplifier designers tobe able to calculate this temperaturefor their applications from the devicepower dissipated, the case tempera-ture, and its thermal resistance (Rth).This calculation is not as simple as itmay appear since the Rth of GaAsdevices is a strong function of thedevice-flange temperature and thechannel temperature, or the powerdissipated. The NES2427P-60 datasheet and CEL’s application note8
AN1032 give the information neces-sary to calculate Rth versus the flangeand channel temperatures, or powerdissipated.
The NES2427P-60 data sheet indi-cates Rth = 0.76°K/W maximum for Tf= 25°C, Vds = +10 VDC, Ids = 12 A andrecommends a maximum channeltemperature of 150°C. This tempera-ture corresponds to a MTTF of 2.4 �106 h. From these data and with thehelp of the application note, the max-imum Idsq can be calculated versusthe device-flange temperature.
As an example, for a maximumflange temperature of 52°C, the max-imum power dissipated correspond-ing to a channel temperature of 150°Cis Pdiss. = 120 W. It means the maxi-mum quiescent Idsq for Vds = +10VDC and a flange temperature of52°C is 12 A.
This current is relatively high, butit is lower than half of the device-sat-urated drain current (Idss), which is 36A typical. If the standard definition ofclass A for power devices is used, Idsq= 12 A does not correspond to thisdevice for a Class A operation but toa Class A-B one.
References1. NES2427P-60 data sheet, http://www.cel.com 2. L. Max, “Balanced Transistors: A New Option For RF
Design,” MicroWaves, June 1977.3. L. Max, “Apply Wideband Techniques To Balanced
Amplifiers,” MicroWaves, July 1980.4. G. Sarkissian, “An S-Band Push-Pull 60-Watt GaAs
MESFET For MMDS Applications,” 1997 IEEE MTT-SDigest.
5. R. Basset, “Three Balun Designs For Push-Pull Ampli-fiers,” MicroWaves, July 1980.
6. D. Raicu, “Design of Planar, Single-Layer MicrowaveBaluns,” 1998 IEEE MTT-S International Microwave Sym-posium Digest.
7. R. Basset, “Automated Test Equipment For HighPower Solid State Device Development And Characteriza-tion,” 47th ARFTG Conference Digest, June 1996.
8. Application note AN1032 “Microwave Power GaAsDevice Thermal Resistance Basics,” http://www.cel.com.
MMDS Amplifier
DESIGN FEATURE
(continued from p. 61)
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