design and implementation of a high-performance bidirectional dc/ac converter for advanced evs/hevs

Design and implementation of a high-performancebidirectional DC/AC converter for advanced EVs/HEVs

C.H. Chen and M.Y. Cheng

Abstract: A high-performance bidirectional DC/AC power converter is required in low emissionand high-efficiency propulsion systems such as electric vehicles and hybrid electric vehicles. Ageneral six-switch full-bridge inverter topology with a high efficient interior permanentmagnet alternator is adopted to explore the constant-voltage, constant-current and pulsating-current charging processes. In addition, useful information such as maximum conversion ratio,average armature current, armature current ripple and output voltage ripple are derived based onthe equivalent circuit model where the armature resistance of the alternator and the conductionresistance of the power switch are considered. A 32 bit digital signal processor, TI 2812, is used toimplement the switching strategies and the control algorithms. Experimental results indicate thatthe performance of the proposed control strategy is satisfactory for all of the popular chargingstrategies employed. Compared with the traditional claw pole alternator, the proposed strategyexhibits a significant improvement in output efficiency.

List of symbols

C capacitance of the converter capacitor, FD duty ratio of the converterD0 1-Deu,v,w instantaneous back EMF voltage of phase u, v,

W, VVemf back EMF voltage between lines of each

commutation state, Vfs switching frequency of the converter, Hziu,v,w armature current of phase u, v, w, AIa average capacitor current, AIb average battery current, Aic instantaneous capacitor current, Adia amount variation of the armature current

during the charging process, Adic amount of the variation of the capacitor current

during the charging process, ALs self inductance of each phase, HLm mutual inductance between phases, HLa armature inductance of each commutation

state, HNp equivalent wire turns at the primary side, turnsNs equivalent wire turns at the secondary side,

turnsp differential operatordQc the amount of the variation of the capacitor

charge during the charging process, CRa armature resistance between lines of each

commutation state, ORc_ESR ESR of the converter capacitor, O

Rs armature resistance of each commutationstate, O

Te electromagnetic torque, NmTs switching period, st time, sVu,v,w input phase voltage of phase u, v, w, VVL average inductor voltage drop, VVc average capacitor voltage, VvL instantaneous inductor voltage drop, Vdvc the amount of variation of the capacitor voltage

during the charging process, Vdvr_c the amount of variation of the capacitor voltage

caused by the limited size of the capacitor, Vdvr_ESR the amount of variation of the capacitor voltage

caused by the ESR of the capacitor, Vor shaft angular speed, rad/s

1 Introduction

As the issues of environmental protection and energyconservation receive growing concern around the world,low emission and high efficiency propulsion systems such aselectric vehicles (EVs) and hybrid electric vehicles (HEVs)have emerged into the market recently, e.g. Toyota Prius,Honda Insight etc. In general, there are three majortechniques in developing advanced HEV propulsionsystems [1]:

Recover the deceleration energy (regenerative braking)when the brake pedal is pushed: The integrated starteralternator (ISA) adopted in EVs/HEVs is operated in thealternator mode so that the dynamic energy of the vehicle isrecovered into the battery.

Idle stop: When the vehicle is stationary, the advancedpropulsion system of HEVs will shut down the enginetemporarily so that there is no fuel consumed during this

The authors are with the Department of Electrical Engineering, National ChengKung University, No. 1, Ta-Hsueh Road, Tainan 701, Taiwan, Republic ofChina

r IEE, 2006

IEE Proceedings online no. 20040954

doi:10.1049/ip-epa:20040954

Paper first received 31st October 2003 and in revised form 9th July 2004

140 IEE Proc.-Electr. Power Appl., Vol. 153, No. 1, January 2006

period. Moreover, electric motors can generate maximumtorque at zero speed: it is ideal for accelerating the vehicle inthe lower speed range. Thereby, fuel consumption duringthe starting stage is improved.

Reduction in engine size: General combustion engines aredesigned for matching both the acceleration and cruisingrequirements. However these requirements will make theengine larger than needed in general cruising operations.Larger engines mean more fuel consumption and moreenvironmental pollutions. Nevertheless, thanks to theadvanced HEV power train system, auxiliary power canbe provided by the electric motor during accelerationperiods, hence the engine size can be reduced substantiallyand the fuel consumption efficiency can be much improved.

To implement the aforementioned functions required inthe advanced power train systems of EVs/HEVs, abidirectional DC/AC power converter is needed. Duringthe motor mode (starting, acceleration or power assisting),the torque command is generated from the vehicle controlunit (VCU) when the control system senses the pedal input.The required output magnetic torque is controlled based onthe direction and the amplitude of the motor current, andthe power converter may deliver power from the batterypack to various types of motors (DC motor, AC motor orreluctance motor). During regenerative mode (decelerationor continuous charging), whenever the brake pedal ispushed or the state of charge (SOC) of the battery is low,the VCU will issue a charge command. The dynamic energyfrom the vehicle or the combustion engine is transferred tothe battery via the alternator and the power converter.

From the point of view of power density and outputefficiency, most EV/HEV designers prefer permanentmagnet synchronous machines in their applications [2–4].Permanent magnet synchronous machines with trapezoidalinduced EMF are known as permanent magnet brushlessDC motors (PMBLDCMs). In contrast, other types ofpermanent magnet synchronous machines that havesinusoidal induced EMF are referred to as permanentmagnet synchronous motors (PMSMs) [5]. Compared withPMSMs, PMBLDCMs have the following attractiveadvantages: 1 for the same copper losses, PMBLDCMshave 15% more power density than PMSMs. 2 Six-steppedcommutation is much easier to implement than thecontinuous commutation required in PMSMs. With thesesuperior properties, it is no wonder that the PMBLDCMshave been widely used in EVs. HEVs, exercise equipment,washing machines, air conditioners, pumps, fans etc. [6].

In this study, a 2.0kWPMBLDCM with six-switch full-bridge inverter topology was constructed to evaluateperformance in both motoring and regenerating operation.In addition, useful information such as maximum conver-sion ratio, average armature current, armature currentripple and output voltage ripple were derived based on theequivalent circuit model that includes the armatureresistance of the alternator and the conduction resistanceof the power switch. Considering the charge time and thelifetime of the battery, various charging strategies, e.g.constants-current, constant-voltage and pulsating-currentwere explored with the proposed compact design. Com-pared with two-stage converter topologies and otherpulsating charging circuits, additional components such aspower switches and passive components were eliminated inthis study. In addition, the concept of module-based designcan be adopted in practical implementations. This kind ofmodule-based configurations will reduce not only the costand the complexity of design and implementation, but also

the need for future maintenance. Experimental resultsindicate that a high-efficiency bidirectional DC/AC powerconverter can be realised based on the general six-switchfull-bridge inverter topology without additional compo-nents. It can be operated in the three most popular chargingprocesses: constant-voltage, constant-current and pulsating-charging process. Compared with the traditional claw polealternator, the proposed strategy exhibits a significantimprovement in output efficiency [7].

2 Dynamic model of PMBLDCM

The two axes theory (dq0 transformation) has been adoptedextensively in analysing the PMSMs and the inductionmotors. However, it is not suitable for non-sinusoidalelectrical machines, such as PMBLDCMs, which havetrapezoidal back EMF waveforms. Therefore, per phasevariable [6] was adopted in this study. The voltage equationof PMBLDCMs in matrix form is expressed as

Vuvw ¼ RIuvw þ LpIuvw þ Euvw ð1Þwhere p stands for the differential operator. Vuvw,Iuvw andEuvw denote the phase voltage, the phase currents and thephase back EMF voltage, respectively, where,

Vuvw ¼vu

vv

vw

24

35; Iuvw ¼

iu

iv

iw

24

35;Euvw ¼

eu

ev

ew

24

35 ð2Þ

R and L are 3 3 symmetrical matrices consisting ofarmature resistance Rs, self inductance Ls and mutualinductance between phases Lm, and are defined as

R ¼Rs 0 00 Rs 00 0 Rs

24

35; L ¼ Ls Lm Lm

Lm Ls Lm

Lm Lm Ls

24

35 ð3Þ

According to the principle of energy conservation, theelectromagnetic torque can be written as

Teor ¼ ðeuiu þ eviv þ ewiwÞ ð4Þ

3 Switching strategies

3.1 Motor modeFigure 1 shows the back EMF waveforms of the idealPMBLDCM. If the conducting current is carefully deliveredto each phase, as shown in Fig. 1, theoretically the outputtorque will be constant, according to (4). Observing boththe waveforms of the back EMF and the excited currentshown in Fig. 1, it is found that the characteristics ofPMBLDCMs are fundamentally identical to the DC brushmotors during each 60 commutation electrical degrees. Thisis why it has the name ‘brushless DC motor’.

Figure 2 illustrates the topology of a general six-switchfull-bridge inverter. It has been widely employed in high-performance power converters, general V/F control inver-ters and high-precision servo drives provided with dedicatedswitching strategies [8–10]. Since there are only two legsworking in each of the 601 commutation period, the openleg is not depicted in the following analysis. Figures 3a andb illustrate the details of one of the six motoringcommutation states: state I, defined in Fig. 1. During state1, MOS1 is always closed in either the build-up process orthe chopping process, whereas MOS4 is chopped accordingto the torque command that is generated from the controlunit. The upper anti-parallel power diodes are used toprovide the path for current freewheeling in the choppingprocess of the motor mode, whereas the lower one is used to

IEE Proc.-Electr. Power Appl., Vol. 153, No. 1, January 2006 141

accumulate and release the inductor energy in the alternatormode, shown in Figs. 3c and d, respectively.

Figure 4 shows the steady-state equivalent circuit for themotor mode of each commutation state. In this case, thefull-bridge converter can be considered as an ideal DCtransformer, in which the primary side of the transformer isfed by a battery, and the output voltage at the secondaryside is a function of duty ratio Vs¼DVBatery. Since the dutyratio D is smaller than 1, the turns ratio Np/Ns¼ 1/D of thetransformer will be greater than 1. Note that Np is theequivalent wire turns at the primary side, while Ns

represents the equivalent wire turns at the secondary side.

3.2 Alternator modeWhen the PMBLDCM is operated as an alternator(regenerative braking or continuous charging driven by acombustion engine), as shown in Fig. 5, the absolute valueof the induced line–line voltage of each commutation stateis kept as constant when the speed is fixed.

In general cases, the amplitude of this induced voltage issmaller than that of the battery voltage, i.e. one has topump up the induced voltage high enough to charge thebattery. In [11], a two-stage power converter was introducedto achieve such a goal: however, additional power switchesand passive components are needed. As a result, the costand the volume of the system will be substantially increased.

MOS1 MOS3 MOS5D1 D3 D5

MOS2

C

battery− + MOS4 MOS6D2 D4 D6

Fig. 2 Topology of general six-switch full bridge inverter

phase u

phase v

phase w

I II III IV V VI

Fig. 1 Characteristics of ideal trapezoidal back EMF and excitedcurrent waveforms in motor mode––––back EMF waveformFFexcited current waveform

MOS1 D1 MOS3 D3

MOS2battery D2Ra La

Ia

VemfMOS4 D4

− +

MOS1 D1 MOS3 D3

MOS2battery D2Ra La

Ia

Vemf

MOS4 D4− +

MOS1 D1 MOS3 D3

MOS2battery D2Ra La

Ia

Ic

Vc

Vemf

MOS4 D4− +

MOS1 D1 MOS3 D3

MOS2battery D2Ra La

IaIb

Ic

VemfMOS4 D4

− +

d

c

b

a

Fig. 3 Switching strategies of motor mode and alternator mode forstate 1a Current building process in motor modeb Current freewheeling during chopping process in motor modec Current accumulating process in alternator moded Current releasing path, charging process in alternator mode

battery

1 D

switchingDC/DC transformer

Ra La

Vemf− +−+

Fig. 4 Steady-state equivalent circuit for motor mode of eachcommutation state


Theoretically, the induced voltage can be pumped up to anarbitrary level based on the general full-bridge inverterwithout any additional component if proper switchingstrategies are adopted [12]. The operational mechanism isquite similar to the so-called non-isolated boost converter.

4 Constant-voltage and constant-current charge

Since magnetic saturation and armature reaction of theBLDCMs/PMSMs may result in side effects, such asreduced efficiency and induced voltage, the peak currentshould be limited according to the rated specifications. Toreduce the amplitude of the average armature currentefficiently, the continuous conduction mode (CCM) wasimplemented in this study. Moreover, considering issuessuch as EMI and power factor, the current ripple iscontrolled to be within 10% of the average current value inall operation regions, regardless of the variations in severalparameters such as the speed of the alternator, the inputvoltage of the converter and the state of charge (SOC) of thebattery (i.e. the equivalent load resistance of the batteryduring the charging process).

4.1 Maximum conversion ratioIn the following analysis, state I is used to explain thedetails. As shown in Fig. 3c, when MOS2 is turned on. i.e.0otoDTs (TS is the switching period), according toKirchhoff voltage law, the voltage drops around the closedpath include the induced back EMF Vemf, the inductor drop

vL ¼ Ladiadt and the internal resistor drop va¼ iaRa, which

comprises the armature resistance and the conductionresistance of the power switch. If the armature currentripple and the voltage ripple over the capacitance arerelatively small (which is controlled to be within 10% of theaverage value in this study), the inductor voltage and thecapacitor current for the case that MOS2 is closed duringstate I can be described as

vL ¼ Vemtf iaRa ð5Þ

Ic ¼Vc

Rð6Þ

where R is the equivalent load resistance of the battery, andits detailed derivations will be discussed later in this Section.

Figure 3d shows the equivalent circuit of state I whenMOS2 is opened for DTsotoTs. The armature voltagedrop and the capacitor current are described by

vL ¼ Vemf iaRa Vc ð7Þ

ic ¼ ia ib ð8ÞIn periodic steady state, by the principle of volt–secondbalance, one can conclude that the net change in inductorvoltage is zero, namelyZ tþTs

tvLðtÞdt ¼ DðVemf IaRaÞ

þ D0ðVemf IaRa VcÞ ¼ 0

ð9Þ

where D+D0 ¼ 1. Similarly, in periodic steady state, by theprinciple of capacitor charge balance, one will haveZ tþTs

ticðtÞdt ¼ D Vc

R

þ D0 Ia

Vc

R

¼ 0 ð10Þ

Substituting (10) into (9), the charging voltage Vc can bedescribed in terms of D0, the converter internal resistance Ra

and the load resistance R as follows:

Vc

Vemf¼ 1

D0 þ KD0

9T ðD0Þ ð11Þ

where

K9Ra

Rð12Þ

In Fig. 6, the primary side of the transformer is fed by theline–line induced voltage Vemf of each commutation state,and the equivalent turns ratio is a function of duty ratiodescribed by

Np

Ns¼ 1

T ðD0Þ ð13Þ

To evaluate the maximum conversion ratio of the converter,one can differentiate (11) with respect to D0 and, putting theobtained expression to zero, obtain:

dT ðD0ÞdD0

¼ K D02

ðD0 þ KÞ2¼ 0 ð14Þ

According to (14), the maximum conversion ratio occurs if

D0 ¼ffiffiffiffiKp

ð15Þwhere 0rD0r1 in practical implementations. Substituting(15) into (11) yields the maximum conversion ratio of theconverter:

TmaxðD0Þ ¼ T ðD0ÞjD0¼ ffiffikp ¼ 1

2ffiffiffiffiKp ð16Þ

It should be noted that the maximum conversion ratio issmaller than 1 for K40.25, as shown in Fig. 7. This meansthat the output voltage of the boost converter is smaller

phase u

phase v

phase wI II III IV V VI

Fig. 5 Ideal back EMF and recovery current waveforms inalternator mode––––back EMF waveformFFregenerative current waveform

battery

1

1

Ra La D' D'K

Vemf−+

+

switchingDC/DC transformer

−+

Fig. 6 Steady state equivalent circuit for the alternator mode ofeach commutation state


than the input voltage for K40.25. In that case, the boostconverter will saturate and change into a buck converter.

4.2 Current rippleAs mentioned at the beginning of Section 4, the converterhas several advantages if the armature current ripple issmall. In this study, the switching frequency was controlledas the speed of the alternator and the SOC of the batterywas varied. In the following analysis, both the current rippleratio and the voltage ripple ratio are assumed to be lessthan 10%.

Figure 8 shows the equivalent circuit of Figs. 3c and d.According to Fig. 8a, the net change of the armaturecurrent dia during the accumulating process can beexpressed as

dia ¼ DTsðVemf IaRaÞ

Lað17Þ

where the average armature current Ia can be derived fromthe principle of energy balance:

Ia ¼Vemf

D02Rþ Rað18Þ

The current ripple can be determined by (17) and (18) as:

dia

Ia¼ DD02RTs

Lað19Þ

Equation (19) reveals that the variations in load resistanceand switching period will affect the amplitude of thearmature current ripple significantly. In this study, theswitching frequency fs was controlled by (20) to keep the

current ripple ratio under 10%:

fs 10DD02R

Lað20Þ

4.3 Voltage rippleThe maximum voltage ripple dvc across the capacitor can beevaluated by considering dvr_c (the ripple due to the limitedcapacitance) and dvr_ESR (the ripple caused by the ESR)simultaneously, in which

dvr c ¼dQc

C¼ VcDTs

RCð21Þ

dvr ESR ¼ dicRc ESR ð22ÞThe capacitance and the ESR of the capacitor are chosenbased on the constraint:

dvr c þ dr ESR dvc ð23Þ

4.4 Equivalent load resistance of batteryAccording to (12), (19) and (20), obviously, the performanceof the AC/DC converter is strongly related to the equivalentload resistance R, and the derivation of equivalent loadresistance of the battery under the charging process isintroduced in this Section. The most commonly used modelof the lead–acid battery consists of the open terminalvoltage Vb, and the internal series resistance Rb, as shown inFigs. 9a and b; the terminal voltage equation is given by

Vc ¼ Vb þ IbRb ð24ÞWhen the lead–acid battery is charging, the open terminalvoltage Vb can be modelled as an equivalent resistor, asshown in Fig. 9c. Hence the power dissipation of the lead–acid battery during the charging process can be expressedas:

I2b R ¼ VbIb þ I2b Rb ð25ÞSince Vb¼ IbRc, we can obtain:

R ¼ Rc þ Rb ð26ÞEquations (26) illustrates that the equivalent load resistanceof the battery can be expressed as the sum of the chemicaltransfer resistance Rc and the internal resistance Rb. Thepower absorbed by the chemical resistance will betransformed into chemical energy, which will be stored inthe battery for further utilisation. However, the powerconsumed by the internal resistance will be dissipated asheat and hence increases the temperature of the battery. Theload resistance of the battery can be treated as constantunder each short switching period. However, it is varied asthe charging current or the SOC of the battery is changed.

5

4.1

3.2

2.3

1.41

0.5

0 0.2 0.4 0.6 0.8 1K

Tmax(D' ) = T(D' )D'= k

=1

2 K

Fig. 7 Maximum conversion ratio varied from 5 to 0.5 asK¼ 0.01–1

Ra

a b

La

Vemf

Ra

Rb

La

VbVemf

C

+−

+−

+−

Fig. 8 Equivalent circuit for the constant-voltage charging processa Armature inductor accumulates magnetic energyb Armature inductor releases magnetic energy

Rb Rb

RcVb

Ib Ib

a b c

Ib

battery

Vc Vc

VbVc

−+

−

+

−+

Fig. 9 Equivalent circuit of lead–acid batterya Lead–acid battery during charging processb Equivalent circuit consists of OC voltage in series with resistancec The equivalent circuit consists of load resistance (Rc+Rb), valid forcharging process; load resistance is function of SOC and chargingcurrent of battery


5 Pulsating current charge

To shorten the charging duration, pulsating currentcharging is preferred rather than continuous charging.The main idea behind pulsating current charging is toprovide relaxation time during the charging process. Such adiscontinuous charging process not only shortens thecharging duration but also accelerates the neutralisationof the internal electrolyte of the battery, which will behelpful in prolonging the life cycle of the battery [13]. It wasfound in this study that, even though the armature currentis nearly constant, the pulsating charging current can beimplemented successfully based on the general six-switchfull-bridge inverter topology without any additionalcomponent.

Figure 10 shows the equivalent circuits of Figs. 3c and d,in which the capacitor is eliminated. If the current ripple issmall, the average current of the armature can be derived as:

Vemf IaTs ¼ I2a RaDTs þ I2a ðRa þ RbÞD0Ts þ VbIaD0Ts

) Ia ¼Vemf D0Vb

Ra þ D0Rb¼ Veq

Req

ð27Þ

According to Fig. 10a, when the armature inductoraccumulates magnetic energy, the current ripple can beexpressed as:

dia ¼ DTsVemf IaRa

La

ð28Þ

diaIa¼ DTsðReqVemf VeqRaÞ

LaVeqð29Þ

Figure 11 shows the simulation results of the currentwaveforms of the armature current and the pulsatingcharging current of the battery. The width and theamplitude of the current waveform are controlled by theduty ratio. Theoretically, the output waveform will be adelta function as the duty ratio approaches 100%.

6 DSP based fully digital controller

Figure 12 shows the architecture of a fully digital, DSP-based bidirectional EV/HEV drive. The concept of module-based design is adopted in practical implementations, inwhich the drive is mainly composed of a power switchmodule and a single digital-signal-processor (DSP) con-troller. Such a compact configuration will reduce both thecost and the complexity of design and implementation, andalso the need for future maintenance [14].

A 32-bit, highly integrated DSP, TMS320F2812, whichintegrates the high-performance DSP core with powerfulon-chip peripherals supplied from Texas Instruments,capable of 150MIPs, was used as the system control kernel[15]. The input interface of the DSP contains various types

of feedback signals, including the operational mode (motormode or regenerative braking) from the control unit of theEVs/HEVs via the CAN protocol, the phase currents of thePMBLDCM, the charging/discharging current of thebattery, the voltage of the battery and the shaft positionfrom the Hall-effect sensors. According to the requiredcommands and the feedback signals, the DSP core willexecute the well tuned PI type control algorithm andgenerate proper gating signals for each power switch of thebidirectional converter.

7 Experimental evaluation

Table 1 shows the parameters of the bidirectional DC/ACconverter used in this study. Figure 13 shows theconfiguration of the prototype bidirectional DC/AC con-verter used for experimental verification. A 5kW ACservomotor is used to drive the PMBLDCM at differentspeeds. The battery pack consists of four 12V–17Ah lead–acid batteries. Moreover, a load bank, which consists of10O power resistors, is used to simulate the cases ofdifferent SOC of the battery. Figure 14 shows the line–neutral back-EMF waveforms of each phase; thePMBLDCM is operated as an alternator at a constantspeed, 3000 rpm. Note that the waveform is not perfectlytrapezoidal due to the effect of open slotting of the stator,which is imposed by manufacturing constraints.

Ra

a b

La

Vemf

Ra

Rb

La

Vb

Vemf

+−

+−

+−

Fig. 10 Equivalent circuit for pulsating-current charging processa Armature inductor accumulates magnetic energyb Armature inductor releases magnetic energy

70

60

50

40

30

20

10

09.5m 9.6m 9.7m

a

b

9.8m 9.9m 10m

70

60

50

40

30

20

10

09.5m 9.6m 9.7m 9.8m 9.9m 10m

Fig. 11 Waveforms of pulsating-current charging processa Armature current of alternatorb Charging current of the batteryVemt¼ 30V, Vb¼ 50V, Ra¼ 0.3O, Rb¼ 0.01O, La¼ 100mH, D¼ 0.8


Figures 15a–c show the waveforms of the armaturecurrent and the output charging voltage as the loadresistance is decreased from 60 to 2O. Figure 15d showsthe case for charging the lead–acid battery whereIb¼ 0.3C¼ 5.1A. Experimental results verify that thelead–acid battery can be modelled as an equivalent loadresistance as introduced in Section 4.4. According to (19),the current ripple will not be negligible if the load resistanceis increased. Because of the closed loop control and themain capacitance, no matter whether the armature currentis operated in CCM, BCM, or DCM, the output voltage

ripple in all cases is relatively small as the load resistancevaries. Figures 15e and f show the measured waveformduring the pulsating current charging process. It should benoted that even when the output current is pulsated, thecurrent of the armature is continuous. As mentionedSection 5, the amplitude and the width of the current pulseare controlled by the duty ratio. Figure 16 shows theefficiency of the constant-voltage charging process withdifferent charging rates, where C is 17A in this study.Because the losses of the converter are caused mainly by thearmature resistance and the power switches, the efficiencydeteriorates as the duty ratio is increased. Nevertheless, theaverage output efficiency is higher than 85% over the entireoperating region, which shows significant improvementswhen compared with the traditional claw pole alternator [7].In addition, the prototype converter has been adopted bythe local hybrid electric and pure electric scooter manu-facturers, in which the on-road testing shows promisingresults.

8 Conclusions

A compact and high-efficiency bidirectional DC/AC powerconverter for BLDCMs under a general six-switch full-

current andvoltage feedback

gate signal

charging command

torgue command

shaft encoder

+

+

+

−

−−

M3−

battery

A

A

V

Fig. 12 Architecture of module-based fully digital EV/HEV drive

Table 1: ISA parameters

Name of parameters Value

Maximum speed 9000rpm at 50V

Rated torque 6Nm

Rated power 3Hp

Number of poles 4

Phase inductance 150mH

Phase resistance 20mO

Equivalent conduction resistance ofpower switches

40mO

Main capacitance 20mF

Fig. 13 Configuration of prototype converter developed in thisstudy

Tek stop

Ch3 10.0 V10.0 V 10.0 V 1.00msM A Ch1 −14.4 VCh2Ch1

Fig. 14 Phase – neutral back EMF waveform at 3000 rpm,10V/div.


Tek stop T

T

1

10.0 V 20.0 V

a b

c d

e f

40.0µsM A Ch1 9.20mVCh2Ch1

Tek stop T

T

1

10.0 mV 20.0 V 40.0µsM A Ch1 13.6mVCh2Ch1

Tek stop T

T

1

10.0 mV 20.0 V 40.0µsM A Ch1 7.20mVCh2Ch1

Tek stop T

T

1

10.0 mVΩ 20.0 V 40.0µsM A Ch1 7.40mVCh2Ch1

Tek stop T

T

10.0 mV 40.0 µsM A Ch1 16.0mVCh1

Tek stop T

T

1

10.0 mV 40.0 µsM A Ch1 16.0mVCh1

1

Fig. 15 Measured waveforms for various charging processesa Constant-voltage charging, DCM, R¼ 60O, 5A/div./ch1, 20V/div./ch2b Constant-voltage charging, BCM, R¼ 20O, 5A/div./ch1, 20V/div./ch2c Constant-voltage charging, CCM, R¼ 2O, 50A/div./ch1, 20V/div./ch2d Constant-voltage charging for lead–acid battery, Ib¼ 0.3C¼ 5.1A, 10A/div./ch1, 20V/div./ch2e Pulsating current charging, R¼ 10O, D¼ 0.5, 10A/div.f Pulsating current charging, R¼ 10O, D¼ 0.9, 10A/div.


bridge inverter topology, has been studied and implemen-ted. Compared with two-stage power converters, additionalpower switches and inductors are eliminated in theproposed approach. To improve the performance of thecharging process, i.e. shorten the charging period andprolong the lifetime of the battery, the ability of thepulsating current charging of the general six-switch full-bridge inverter topology has also been explored. Experi-mental results indicate that the proposed control strategyexhibits a satisfactory performance when operated in severalpopular charging processes. Compared with the traditionalclaw pole alternator, the proposed strategy exhibits asignificant improvement in output efficiency. The prototypeconverter has been adopted by the local hybrid electric andpure electric scooter manufacturers, and on-road testingshows promising results.

9 Acknowledgments

The authors would like to acknowledge the support of theE-Ton Dynamics Tech. Industry Co., Ltd., and the Electric

Motor Technology Research Center, National Cheng KungUniversity. We would also like to thank Prof. M.-C. Tsaiand Prof. R.-L. Lin for their valuable suggestions. Specialthanks are dedicated to J. Wu for his assistance with thiswork.

10 References

1 Chan, C.C.: ‘The state of the art of electric and hybrid vehicles’, Proc.IEEE, 2002, 90, (2), pp. 247–275

2 Chu, C.L., Tsai, M.C., and Chen, H.Y.: ‘Torque control of brushlessDC motors applied to electric vehicles,’. Proc. 2001 IEEE Int. Conf.on Electric Machines and Drives, pp. 82–87

3 Tsai, M.C., Chou, M.C., and Chu, C.L.: ‘Control of a variable-winding brushless motor with the application in electric scooters.’.Proc. 2001 IEEE Int. Conf. on Electric Machines and Drives,Cambridge, MA, USA, pp. 922–925

4 Bianchi, N., Bolognani, S., and Zigliotto, M.: ‘High-performance PMsynchronous motor drive for an electrical scooter’, IEEE Trans. Ind.Appl., 2001, 37, (5), pp. 1348–1355

5 Hendershot, J.R., and Miller, THE.: ‘Design of brushless permanent-magnet motors,’, (Oxford University Press, Oxford, UK, 1994)

6 Krishnan, R.: ‘Electric motor drives, modeling, analysis, and control’(Prentice Hall, NJ, USA, 2001)

7 Adler, U., ‘Automotive electric/electronic systems’ (Bosch, Auto-motive Equipment Business Sector) 1994, Germany

8 Erickson, R.W., and Maksimovic, D.: ‘Fundamental of powerelectronics’, 2nd Edn. (Kluwer Academic, MA, USA, 2001)

9 Mohan, N., Undeland, T.M., and Robbins, W.P.: ‘Power electronics,converters, applications and design’ (John Wiley and Son, New York,USA, 1995)

10 Jahns, T.M.: ‘Motion control with permanent magnet ac machine’,Proc. IEEE, 1994, 82, (8), pp. 1241–1252

11 Caricchi, F., Crescimbini, F., Noia G., and Pirolo, D.: ‘Experimentalstudy of a bidirectional dc-dc converter for the dc link voltage controland the regenerative braking in pm motor drives devoted to electricalvehicles’. Proc. 1994 Applied Power Electronics Conf. and Exp.(APEC), Orlando, Florida, USA, Vol. 1, pp. 381–386

12 Becerra, R.C., Ehsani, M., and Jahns, T.M.: ‘Four-quadrant brushlessECM drive with integrated current regulation’, IEEE Trans. Ind.Appl., 1992, 28, (4), pp. 833–841

13 Hua, C.C., and Lin, M.Y.: ‘A study of charging control of lead-acidbatten for electric vehicle’. Proc. 2000 IEEE Int. Symp. on IndustrialElectronics Cholula, Puebla, Mexico, 2000, pp. 135–140

14 Tsai, M.C., and Chen, J.H.: ‘A practical implementation of a linearinduction motor drive using new generation DSP controller’. Proc.1999 IEEE Int. Conf. on Control applications., Hawaii, USA, Vol. 2,pp. 945–949

15 ‘TMS320F2812 DSP controller reference set’ (Texas Instrument Inc.,2002)

1

0.98 Ib = 0.3C

Ib = 1C

Ib = 1.5C

0.96

0.94

0.92

0.9

0.88

0.86

0.84

0.82

0.84000

effic

ienc

y

4500 5000 5500 6000rpm

6500 7000 7500 8000

Fig. 16 System efficiency for different charging rate


design and implementation of a high-performance bidirectional dc/ac converter for advanced evs/hevs

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