comparison of 2.3-kv medium-voltage multilevel converters for industrial medium-voltage drives

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 54, NO. 6, DECEMBER 2007 2979

Comparison of 2.3-kV Medium-Voltage MultilevelConverters for Industrial Medium-Voltage Drives

Dietmar Krug, Steffen Bernet, Member, IEEE, Seyed Saeed Fazel,Kamran Jalili, and Mariusz Malinowski, Member, IEEE

Abstract—This paper compares the expense of powersemiconductors and passive components of a (2.3 kV, 2.4 MVA)two-level, three-level neutral-point-clamped, three-level flying-capacitor, four-level flying-capacitor, and five-level series-connected H-bridge voltage source converter on the basis of thestate-of-the-art 6.5-, 3.3-, 2.5-, and 1.7-kV insulated gate bipolartransistors for industrial medium-voltage drives. The powersemiconductor losses, the loss distribution, the installed switchpower, the design of flying capacitors, and the components of anLC sine filter for retrofit applications are considered.

Index Terms—Filter design, medium voltage, multilevel con-verters, power electronics.

I. INTRODUCTION

TODAY, there is a large variety of converter topologies formedium-voltage drives (MVDs) [1], [31]. Cycloconvert-

ers and load commutated converters (LCI) applying thyristorsare used, particularly in applications with very high powerdemands (e.g., S ≥ 30 MVA).

For low- and medium-power industrial applications (e.g.,S = 300−30 MVA), the majority of the drive manufacturersoffer different topologies of voltage source converters: two-level voltage source converters (2L-VSCs; e.g., Converteam),three-level neutral-point-clamped voltage source converters(3L-NPC VSCs; e.g., ABB, Converteam, Siemens, TMEIC),four-level flying-capacitor voltage source converters (4L-FLCVSCs; e.g., Converteam), and series-connected H-bridge volt-age source converters (SCHB VSCs; Siemens). One manufac-turer (Allen Bradley) offers self-commutated current sourceinverters (CSI).

Whereas 4.5-, 6-, and 6.5-kV integrated gate commutatedthyristors are mainly used in 3L-NPC VSCs and CSIs, respec-tively, 2.5-, 3.3-, 4.5-, and 6.5-kV medium-voltage insulatedgate bipolar transistors (MV-IGBTs) are applied in 2L-VSCs,3L-NPC VSCs, and 4L-FLC VSCs. In contrast, 1.2- and 1.7-kVlow-voltage IGBTs (LV-IGBTs) are usually applied in SCHB

Manuscript received February 26, 2007; revised August 9, 2007. The work ofM. Malinowski was supported by a Foundation for Polish Science Scholarship.

D. Krug is with Siemens AG Automation & Drives (Large Drives), 90441Nuremberg, Germany (e-mail: [email protected]).

S. Bernet is with Dresden University of Technology, 01187 Dresden,Germany.

S. S. Fazel and K. Jalili are with the Berlin University of Technology, 10623Berlin, Germany.

M. Malinowski is with the Institute of Control and Industrial Electronics,Warsaw University of Technology, 00-662 Warsaw, Poland.

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2007.906997

VSCs. To derive specific converter characteristics, the mod-ulation schemes, losses, and harmonic spectra of a 2L-VSC,a 3L-NPC VSC, a 3L-FLC VSC, and a 4L-FLC VSC werecompared [2], [3].

This paper also includes the five-level SCHB VSCs(5L-SCHB VSCs) in the comparison. Thus, all available volt-age source converter topologies for 2.3-kV industrial MVDsare compared regarding the expense of semiconductors andpassive components, converter losses, modulation schemes,and harmonic spectra. It should be noted that the 2L-VSC(e.g., [1] and [4]–[6]), the 3L-NPC VSC (e.g., [1], [7], [8],[17], [24], and [25]), and the 3L/4L-FLC VSC (e.g., [9]–[13]and [33]) can be fed by identical grid side converters since theyoperate at one dc voltage link [Figs. 1(a) and 2]. In contrast,the 5L-SCHB VSC (e.g., [1], [8], [11], and [13]) requires sixinsulated dc voltage links (one per power cell), which are fed bya special multiwinding transformer and corresponding rectifiers[Figs. 1(b) and 2].

A retrofit application demanding an output voltage total har-monic distortion (THD) of less than or equal to 5% accordingto the standard IEEE 519-1999 is chosen to evaluate the qualityof the output spectrum and the size of the passive compo-nents of an LC sine filter. State-of-the-art 6.5-, 3.3-, 2.5-, and1.7-kV IGBTs are assumed. The design of semiconductors,flying capacitors, and passive components of an LC sine outputfilter is described. The calculation of losses and the expense ofpower semiconductors, gate units, capacitors, and inductors atmedium and high switching frequencies are the basis for theconverter comparison.

II. CONVERTER SPECIFICATION

Table I depicts the basic converter data and the conditions ofa medium-voltage converter for the comparison. The converterratings and conditions are closed to that of the commerciallyavailable medium-voltage converters.

The minimum dc-link voltage to achieve an output line-to-line voltage of 2.3 kV using space vector modulation or anatural sampled sine-triangle modulation with one-sixth addedthird harmonics can be calculated by

Vdc,min =√

2 × Vll,1,rms =√

2 × 2.3 kV = 3252.7 V. (1)

To determine the nominal dc-link voltage of the converter, adc-link voltage reserve of 4% is required to cover the voltagedrop across the filter inductor, i.e.,

Vdc,n =1.04 × Vdc,min=1.04 × 3252.7 V=3382.8 V. (2)

0278-0046/$25.00 © 2007 IEEE

2980 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 54, NO. 6, DECEMBER 2007

Fig. 1. Block diagram of medium-voltage drives applying (a) 2L-VSC, 3L-NPC VSC, 3L-FLC VSC, and 4L-FLC VSC and (b) 5L-SCHB VSC.

Fig. 2. Circuit configurations of (a) 2L-VSC, (b) 3L-NPC VSC; (c) 3L-FLC VSC; (d) 4L-FLC VSC; and (e) 5L-SCHB VSC.

III. POWER SEMICONDUCTOR UTILIZATION FOR

CONSTANT CONVERTER POWER AND

SWITCHING FREQUENCY

Considering the nominal device voltage Vcom at 100 FIT ofIGBTs and diodes, where a cosmic ray withstand capability of

100 FIT (1 FIT is equivalent to 1 failure in 109 operation hours)is guaranteed, 6.5-, 3.3-, 2.5-, and 1.7-kV IGBTs/diodes have tobe applied in the 2L-VSC, the 3L-NPC/FLC VSC, the 4L-FLCVSC, and the 5L-SCHB VSC, respectively.

Table II summarizes the design of the power semiconductorsfor the converter specification of Table I, assuming a carrier

KRUG et al.: COMPARISON OF MULTILEVEL CONVERTERS FOR INDUSTRIAL MEDIUM-VOLTAGE DRIVES 2981

TABLE IBASIC CONVERTER DATA

frequency of fC = 750 Hz and a sine-triangle modulation withone-sixth added third harmonics [3], [32].

The constant ratio of commutation voltage Vcom and nominalIGBT/diode voltage Vcom at 100 FIT (Vcom/Vcom at 100 FIT =0.94) shows that the different converters feature the samesemiconductor voltage utilization.

To determine the semiconductor current rating, the IGBT anddiode ON-state voltages VCE/F,x and the switching losses Esw,x

of a device x given in the data sheets are approximated by

VCE/F,x =Vo,x + Acond,x · i(t)Bcond,x (3)

Esw,x =Asw,x · i(t)Bsw,x (4)

where i is the instantaneous value of the device current, Vo,x

and Acond,x are the ON-state voltage parameters (thresholdvoltage, ON-state resistance), and Bcond,x, Asw,x, and Bsw,x

denote the fitting constants of the ON-state voltage and switch-ing losses, respectively [14], [15].

The fitting parameters and thermal resistance of the semicon-ductors being considered can be taken from Table III, wherethe abbreviations T (IGBT) and D (diode) are used for adevice x.

An accurate loss simulation model, which is described indetail in [14], enables the determination of the semiconductorlosses and junction temperatures. The accuracy of the loss andjunction temperature calculation and the thermal model beingapplied is evaluated in [24]. To calculate the ideal current ratingIC,n(IF,n), an ideal parallel connection of commercially avail-able IGBT or diode modules is assumed. It is obvious that ON-state and switching losses are adapted to the ideal rated currentand the corresponding silicon area. Furthermore, also the ther-mal resistance Rth,jc (thermal resistance of IGBT/diode fromjunction to case) and Rth,ch (thermal resistance of IGBT/diodefrom case to heat sink) are adjusted to the rational number ofideally parallel connected modules according to the silicon areaand the module size. The calculated ideal rated IGBT/diodecurrents IC,n/IF,n (Table II) guarantee that the junction tem-perature of the mostly stressed IGBT or diode reaches a value

of Tj = 125 ◦C in one worst case operating point of thefour-quadrant operation. The temperature of the heat sink issupposed to be constant (Th = 80 ◦C).

It is interesting that the required ideal current ratingIC,n/IF,n to enable a converter output current of Iph,1,rms =600 A is very different for the considered topologies (Table II).Reasons for the different current ratings and the correspondingdifferent installed switch powers (SS = VCE,n · IC,n · n, wheren is the number of switch positions containing one IGBTwith one inverse diode) are the different commutation volt-ages, device blocking voltages, effective switching frequencies,and semiconductor loss distributions, which are caused by thedifferent circuit structures and modulations. Compared to the5L-SCHB VSC, the installed switch powers of the 2L-VSC,4L-FLC VSC, and both 3L VSCs are increased by about 55%,28%, and 4%, respectively.

Although the semiconductor utilization is a very importantvalue to evaluate medium-voltage topologies due to the highshare of semiconductor costs in medium-voltage converters,it must be considered that the output voltage spectrum of thetopologies is very different at constant carrier frequency [3]. Toeliminate the influence of the different output voltage spectra,Section V contains a comparison of active (semiconductors,gate units) and passive (inductors, capacitors) power part com-ponents if all converters realize an output voltage THD of lessthan or equal to 5%.

IV. DESIGN OF PASSIVE COMPONENTS

A. Design of Flying Capacitors

Assuming a constant dc output current and reference signalof the modulation, the equation

C =Idc

p · ∆VC · fC(5)

where Idc is the dc output current, P is the number of series-connected flying capacitor cells, ∆VC is the maximum voltageripple across the flying capacitors, and fC is the carrier fre-quency, enables the design of the flying capacitors for a givenconverter structure and a specified capacitor voltage ripple [16].For sinusoidal converter output voltages (sinusoidal referencesignals) and currents, (5) can be used as an approximationif the amplitude of the maximum phase current Iph,rms isapplied, i.e.,

C =Iph,rms

p · ∆VC · fC. (6)

To determine the design of the flying capacitors for sinu-soidal output voltages and currents, the flying capacitor volt-ages and currents were simulated during one period of thereference signal for the (2.3 kV, 2.39 MVA) 3L-FLC VSC and4L-FLC VSC (Figs. 7 and 8). The maximum capacitor voltageripple ∆VC,max was specified to 7.5% of the dc-link voltageVdc. A sine triangle modulation with added third harmonics[3] was chosen to ensure a natural balancing of the flyingcapacitor voltages within one period of the reference signal


TABLE IICONVERTER VOLTAGE AND SEMICONDUCTOR SPECIFICATIONS FOR CONSTANT CONVERTER POWER AND CARRIER FREQUENCY

(Vll,1,rms = 2.3 kV; Iph,1,rms = 600 A; SC = 2.39 MVA; fC = 750 Hz; Tj,max = 125 ◦C)

TABLE IIIFITTING PARAMETERS AND THERMAL RESISTANCE OF SEMICONDUCTORS

Fig. 3. Flying capacitor voltage ripple of a 3L-FLC VSC as a function ofmodulation index and phase shift (Iph,1,rms = 600 A, fC,3L-FLC VSC =1200 Hz, C = 1393 µF).

[34], [35]. A carrier frequency of fC = 1200 Hz was assumedsince commercially available flying capacitor medium-voltageconverters are operated at similar carrier frequencies.

Figs. 3–6 depict the voltage ripple ∆VC and the current ofthe flying capacitors as a function of the modulation index

m =Vll,1,rms√3 · Vdc/2

(7)

Fig. 4. Flying capacitor current of a 3L-FLC VSC as a function ofmodulation index and phase shift (Iph,1,rms = 600 A, fC,3L-FLC VSC =1200 Hz, C = 1393 µF).

and the phase shift ϕ between output voltage and current forthe 3L-FLC VSC and the 4L-FLC VSC, respectively. Themaximum voltage ripple ∆VC,max and the current stress of the3L-FLC VSC occur at a modulation index of m = 0 (Figs. 3and 4). The maximum voltage ripple is identical to the cor-responding value based on (6). The reason for the matchof simulated and calculated capacitor voltage ripple is thatthe flying capacitor of the 3L-FLC VSC is stressed with180◦ rectangular capacitor current parts of the load current


Fig. 5. Flying capacitor voltage ripple of a 4L-FLC VSC as a function ofmodulation index and phase shift (Iph,1,rms = 600 A, fC,4L-FLC VSC =1200 Hz, C1,2 = 928 µF).

Fig. 6. Flying capacitor current of a 4L-FLC VSC as a function ofmodulation index and phase shift (Iph,1,rms = 600 A, fC,4L-FLC VSC =1200 Hz, C1,2 = 928 µF).

(Fig. 7). Thus, the rms capacitor current is identical to therms load current. In contrast, the maximum capacitor voltageripple ∆VC,max of the 4L-FLC VSC occurs at m = 0.808 ata phase shift between output voltage and current of ϕ = 90◦

(Figs. 5 and 6).The maximum capacitor current of the 4L-FLC VSC is

about 18% lower than that of the 3L-FLC VSC due to theoccurring zero capacitor current states, which are causedby the three phase-shifted carrier signals of the modulation.The voltage ripple is about 30% higher compared to thecalculated value according to (6) since maximum positiveand negative capacitor voltages during one-half period of thereference modulation signal are not identical (Fig. 8). Ob-viously, the average capacitor voltage of the 4L-FLC VSCfluctuates during half cycles of the period of the referencesignal.

This deviation between simulated and calculated capacitorvoltage ripple is independent of the carrier frequency fC ifthe frequency ratio of carrier frequency and frequency ofthe reference signal fo is sufficiently large (e.g., fC/fo >4, . . . , 5). Thus, the value of the flying capacitors of the4L-FLC VSC calculated on the basis of (6) has to be in-creased by about 30% to fulfill the specified maximum ripplevoltage.

Table IV summarizes the design of the flying capacitorsaccording to (6) and the simulation results of Figs. 3–8.

Fig. 7. Flying capacitor voltage and current of a 3L-FLC VSC (Iph,1,rms =600 A, cos ϕ = 0, m = 0, fC,3L-FLC VSC = 1200 Hz, C = 1393 µF).

Fig. 8. Flying capacitor voltage and current of a 4L-FLC VSC(Iph,1,rms = 600 A, cos ϕ = 0, m = 0.808, fC,4L-FLC VSC = 1200 Hz,C1,2 = 928 µF).

B. Design of an LC Sine Filter

Disadvantages of directly inverter-fed variable speed drivesare the additional harmonic losses, a high insulation stress ofthe machine windings due to steep dv/dt’s at the inverter out-put, and increased overvoltages at the machine windings if longcables are applied [17], [18]. These drawbacks can be avoidedif the harmonics around the switching frequency and multiplesof the switching frequency are reduced. In particular, for retrofitapplications, MVDs usually apply a low-pass LC sine filter atthe output (Fig. 9), which basically enables sinusoidal machinevoltages and currents.

Minimum costs, losses, size, and weight are typical optimiza-tion criteria of the LC sine filters. For the converter comparisonof Section V, the design and evaluation of the filters have beenperformed for the following two cases:

• converter operation at constant converter efficiency assum-ing a THD of the phase voltage vM ≤ 5% according to thestandard IEEE 519-1999;

• converter operation at maximum switching frequency at aTHD of the phase voltage vM ≤ 5%.


TABLE IVFLYING CAPACITOR DESIGN (Vll,1,rms = 2.3 kV, Iph,1,rms = 600 A, fC = 1200 Hz)

The filter design was realized by MATLAB/Simulink sim-ulations assuming an active damping control [19]. A machineleakage inductance of LM = 20% (1.4 mH), which is typicalfor industrial induction machines, was assumed. The transferfunctions of the harmonics are described by (8)–(10) [2],[20], [21].

The inductance of the LC filter should be chosen accord-ing to (11) to limit the converter current ripple [21]. For alltopologies, the maximum current ripple iripple,peak is specifiedto be equal to or less than 40% of the rated phase currentamplitude, i.e.,

H1(s) =iI(s)vI(s)

=LMCfs2 + 1

LfLMCfs3 + (Lf + LM )s(8)

H2(s) =iM (s)vI(s)

=1


H3(s) =vM (s)vI(s)

=LMs


Lf =Vll/

√3

2 ·√

6 · f1cb · iripple, peak. (11)

To set the resonance frequency to about 0.5 of the firstcarrier band frequency f1cb, the capacitor of the LC filter(Y-connected) is described by the following [22]:

Cf =Lf + LM

4 · π2 · f2res · Lf · LM. (12)

V. CONVERTER COMPARISON

A. Comparison at Constant Efficiency

1) Losses and Semiconductor Loss Distribution: To enablean evaluation of the considered topologies for different applica-tions, it was assumed for a first comparison that converter power(SC = 2.39 MVA) and efficiency are constant (η ≈ 99%). Theefficiency being selected is typical for state-of-the-art medium-voltage converters.

The carrier frequencies, the ideal rated IGBT/diode currents,and, thus, the installed switch powers were determined inan iterative simulation procedure to meet both the efficiencyrequirement at the specified converter power and a junctiontemperature of Tj = 125 ◦C in one worst case operating point.Table V depicts the resulting ideal rated currents, installed

Fig. 9. Block diagram of three-phase LC sine filter connected between IM(induction motor) and pulsewidth modulation VSC.

switch powers, carrier frequencies, and capacitance of theflying capacitors.

The core of the iterative simulation procedure is the losssimulation model described in Section III. In a first step, thecarrier frequencies for a converter efficiency of η ≈ 99% (m =1.11; Iph,1,rms = 600 A; cos ϕ = 0.9) were determined assum-ing an installed switch power of SS = 38.61 MVA (Table II)in all topologies. In a second step, the rated semiconductorcurrents were calculated for the new carrier frequencies toachieve a maximum junction temperature of Tj = 125 ◦C inone worst case operating point of the converter. Since thechange of the semiconductor current rating influences theconverter efficiency, a new carrier frequency and new ratedsemiconductor currents are calculated during a second iteration.The iterations are repeated until a converter efficiency of aboutη ≈ 99% and a maximum junction temperature of Tj = 125 ◦Care achieved. The semiconductor current ratings, carrierfrequencies, efficiencies, and installed switch powers are shownin Table V.

The relative installed switch power SSR is calculated bynormalizing the installed switch power of a certain con-verter topology to the installed switch power of the 3L-NPCVSC, i.e.,

SSR =SS

SS,3L-NPC VSC× 100. (13)

Fig. 10 depicts the corresponding loss distribution of the con-verters being considered. Although the installed switch powersof the 2L-VSC and the 3L-NPC VSC are comparable, the


TABLE VCARRIER FREQUENCY AND CAPACITY OF FLYING CAPACITORS FOR A CONVERTER EFFICIENCY OF 99%

(Vll,1,rms = 2.3 kV, Iph,1,rms = 600 A, SC = 2.39 MVA)

Fig. 10. Loss distribution and converter efficiency (Iph,1,rms =600 A, cos ϕ = 0.9; fC,2L-VSC = 450 Hz, fC,3L-NPC VSC = 1000 Hz,fC,3L-FLC VSC = 500 Hz,fC,4L-FLC VSC = 275 Hz,fC,5L-SCHB VSC =985 Hz, fo = 50 Hz, m = 1.11).

installed switch powers of the 3L-FLC VSC (fC = 500 Hz),the 4L-FLC VSC, and the 5L-SCHB VSC are reduced by 21%,6%, and 16%, respectively, in comparison to the 3L-NPC VSC.Although the 3L-FLC VSC is operated at half of the carrierfrequency of the 3L-NPC VSC, the losses of both converters arenot identical. The reason for this is the lower ideal current ratingof the semiconductors in the 3L-FLC VSC, which is caused bya symmetrical loss distribution in the 3L-FLC VSC in contrastto the unsymmetrical loss distribution in the 3L-NPC VSC[3], [23], [25]. The conduction losses (PconT/D: conductionlosses of IGBTs/diodes) of the 4L-FLC VSC are drasticallyhigher compared to those of the other topologies since thereare always three conducting medium-voltage devices in thecurrent path due to the three series-connected commutationcells. On the other hand, the switching losses (PonT: turn-onlosses of IGBTs; PoffT/D: turnoff losses of IGBTs/diodes) ofthe 2.5-kV IGBTs/diodes of the 4L-FLC VSC are very small.Also, in the case of the 5L-SCHB VSC, the ON-state lossesclearly dominate due to the very low switching losses of the1.7-kV IGBTs/diodes. The 6.5-kV IGBTs/diodes of the2L-VSC realize low conduction but very high switching losseseven at the low carrier frequency of fC = 450 Hz.2) LC Sine Filter: The values of the LC filter components

of the five topologies for a converter efficiency of about η =99% and a THD of the machine phase voltages of THD−VM ≤5% are given in Table VI.

The 2L-VSC realizes a very low frequency f1cb of thefirst harmonic band. To eliminate the low-frequency current

harmonics, an extremely large inductor in the LC filter must beused, which results in a high voltage drop across the inductorand finally a substantially increased dc-link voltage to deliverthe nominal power to the motor. However, an increase in thedc-link voltage is not possible to ensure a high reliabilityof the semiconductors (Vdc,max ≤ Vcom at 100 FIT). Obviously,the 2L-VSC applying 6.5-kV IGBTs/diodes is not a usefulmedium-voltage converter topology for industrial medium-voltage applications if a high efficiency (η = 99%) and a lowTHD of the machine voltage (THD−VM ≤ 5%) are requiredsince a reasonable LC sine filter design cannot be achieved.The 3L-NPC VSC and the 3L-FLC VSC realize an identicaloutput voltage spectrum, which leads to an identical filterdesign of both converters, if the condition fC,3L-FLC VSC =0.5fC,3L-NPC VSC is fulfilled, and a comparable modulation isapplied.

Both first carrier band frequencies f1cb and stored energy inthe LC filter are in the same range for the 3L-NPC VSC, the3L-FLC VSC, and the 4L-FLC VSC. However, the very highfrequency of the first carrier band f1cb of the 5L-SCHB VSCcompared to the other topologies causes a drastic reductionof the filter values. Despite comparable carrier frequencies,the 5L-SCHB VSC enables a reduction of the stored energyof the LC sine filter components by about 84% comparedto both 3L-VSCs and the 4L-FLC VSC. Furthermore, the5L-SCHB VSC causes the minimum THD of the convertercurrent.

Figs. 11–13 present the simulation results of the inverterphase voltage vI , the machine phase voltage vM , the invertercurrent iI , and the machine current iM .

B. Comparison at Maximum Switching Frequency

1) Losses and Semiconductor Loss Distribution: To extendthe converter evaluation to high switching frequency applica-tions, a second converter comparison is realized for the maxi-mum switching frequency of each converter. Additional to theconverter data of Table I, a constant installed switch power ofSS = 38.61 MVA is assumed.

The maximum carrier/switching frequency of a converter isachieved if one of the semiconductors reaches the maximumjunction temperature Tj,max = 125 ◦C in one worst case operat-ing point of the four-quadrant operation. Table VII summarizesthe corresponding carrier frequencies, the frequencies of thefirst harmonic band f1cb, and the converter efficiencies. The


TABLE VICOMPONENT VALUES AND STORED ENERGY OF 2.3-kV 2.39-MVA VSCS AT A CONVERTER EFFICIENCY OF 99%

Fig. 11. Simulation results of 3L-NPC/FLC VSC (fC,3L-NPC VSC =1000 Hz, fC,3L-FLC VSC = 500 Hz). (a) Phase voltage vI (inverter side).(b) Inverter current iI . (c) Phase voltage vM (machine side). (d) Machinecurrent iM .

Fig. 12. Simulation results of 4L-FLC VSC (fC,4L-FLC VSC = 275 Hz).(a) Phase voltage vI (inverter side). (b) Inverter current iI . (c) Phase voltagevM (machine side). (d) Machine current iM .


Fig. 13. Simulation results of 5L-SCHB VSC (fC,5L-SCHB VSC =985 Hz). (a) Phase voltage vI (inverter side). (b) Inverter current iI . (c) Phasevoltage vM (machine side). (d) Machine current iM .

semiconductor loss distribution can be taken from Fig. 14. Itis remarkable that the 3L-FLC VSC enables almost the samecarrier frequency like the 3L-NPC VSC, which means that theresulting switching frequency at the converter output is almostdoubled. This is because of a substantially more equal lossdistribution of the semiconductors of one phase in the 3L-FLCVSC [23].

Whereas also the carrier frequencies of the 4L-FLC VSC(fC = 1500 Hz) and the 5L-SCHB VSC (fC = 4580 Hz)are remarkably high, the maximum carrier frequency of the2L-VSC is very low (fC = 750 Hz) due to the high switching

losses of the 6.5-kV IGBTs/diodes. The maximum frequency ofthe first carrier band of the 4L-FLC VSC (f1cb = 4500 Hz) isabout 25% and 137% higher than that of the 3L-FLC VSC andthe 3L-NPC VSC, respectively. The phase-shifted modulationof the series-connected H-bridges and the low switching lossesof the 1.7-kV IGBTs/diodes cause an extremely high maximumfirst carrier band frequency of the 5L-SCHB VSC (f1cb =18.3 kHz). This frequency is increased by factors of about 9.6and 4 compared to the 3L-NPC VSC and the 4L-FLC VSC,respectively.2) LC Sine Filter: Table VIII shows the values of the filter

components for the maximum switching (carrier) frequencies.The first harmonic carrier band of the 3L-NPC VSC occurs

at the maximum carrier frequency of 1900 Hz. In contrast, thefirst harmonic carrier band of the 3L-FLC VSC is centeredat around twice of its maximum carrier frequency (f1cb =3600 Hz). Obviously, the components of the output filter of the3L-FLC VSC are essentially smaller than those of the 3L-NPCVSC since the inductor values, as well as the capacitor values,decrease with increasing frequency of the first harmonic carrierband according to (11) and (12).

The four-level output voltage and the high frequency of thefirst harmonic band (f1cb = 4500 Hz) are the reasons why theinductance value of the 4L-FLC VSC is about 35% and 67%lower than that of the 3L-FLC VSC and the 3L-NPC VSC,respectively. The capacitance value of the 4L-FLC VSC isabout 34% and 70% lower than that of the 3L-FLC VSC andthe 3L-NPC VSC, respectively.

The very high first carrier band frequency f1cb of the5L-SCHB VSC causes very small filter components and storedenergies. It becomes clear from Table VIII that the increase inthe number of voltage levels enables smaller LC filter valuesand a decrease in the THDs of the machine phase voltages andthe inverter currents.

Simulated waveforms of the inverter phase voltage vI , themachine phase voltage vM , the inverter current iI , and themachine current iM are depicted in Figs. 15–17.

C. Practical Considerations

1) Modularity and Maintenance of the Converters: A mod-ular design of medium-voltage converters is an important re-quirement, which enables a platform based on development,manufacturing, and service of converter systems. Usually,one inverter phase leg depicts one power electronic build-ing block in 3L-NPC VSCs and 4L-FLC VSCs, respectively[8], [27], [28].

In the case of a 2.3-kV IGBT 3L-NPC VSC, a modular phaseleg design can be achieved by three power cards containingtwo 3.3-kV IGBT/diode modules including gate units on oneheat sink (Fig. 18) [27]. The standardization of the power cardsenables a simple assembly of the inverters. Furthermore, thepower cards can be exchanged within a few minutes withoutspecial tools in case of a failure [27].

One phase leg of a 4L-FLC VSC is realized by three IGBTcell modules and three floating capacitor modules [Fig. 19(a)][28]. The IGBT cell modules consist of two IGBT moduleswith separate heat sinks, gate drivers, and bus bars. To enable


TABLE VIICARRIER FREQUENCY AND CAPACITY OF FLYING CAPACITORS FOR MAXIMUM SWITCHING FREQUENCY

(Vll,1,rms = 2.3 kV; Iph,1,rms = 600 A; SC = 2.39 MVA, Tj,max = 125 ◦C)

Fig. 14. Loss distribution and converter efficiency (Iph,1,rms = 600 A,fC,2L-VSC = 450 Hz, fC,3L-NPC VSC = 1900 Hz, fC,3L-FLC VSC =1800 Hz, fC,4L-FLC VSC = 1500 Hz, fC,5L-SCHB VSC = 4580 Hz, fo =50 Hz, m = 1.11, cos ϕ = 0.9).

low inductive commutation circuits, special floating capacitormodules have been designed that offer two pairs of low in-ductance terminals on the top and bottom sides [28]. Also,in this case, the modular design of the IGBT cell and ca-pacitor modules allows a simple replacement of failed powercomponents.

In an SCHB VSC, each converter phase consists of a seriesconnection of isolated low-voltage power cells consisting ofone standardized H-bridge of 1700-V IGBT modules, one dc-link capacitor, and a six-pulse diode rectifier [Fig. 19(b)] [30].Obviously, the voltage rating of the converter can be adjusted bythe number of series-connected power cells. The expense for thereplacement of a failed power cell is comparable to the 3L-NPCVSC and 4L-FLC VSC, respectively. However, the availabilityof the SCHB VSC can be substantially increased if a redundantconverter design is chosen.1 In this case, one additional powercell per phase and a mechanical bypass per cell, which short-circuits a failed power cell during 250 ms, are added to theconverter structure [30].2) Installation: Usually, the 3L-NPC VSC and the 4L-FLC

VSC can be separately ordered from the transformer. Thus,the converter installation includes the operation and test of thetransformer, the converter, and the electric machine.

The requirement of a special transformer in the SCHB VSCis the reason why the transformer is usually integrated inthe converter, which slightly simplifies the installation of theconverter at the customer site.

1The SCHB VSC is the only commercially available converter that offersredundancy as an option.

VI. CONCLUSION

Table IX summarizes the component count and the expenseof active and passive components of the different voltage sourceconverter topologies for an (2.3 kV, 2.39 MVA) industrialMVD. To also include the NPC diodes of the 3L-NPC VSCin the comparison, the total installed switch power SStot of acertain converter topology is defined to be

SStot = VCE,n × IC,n × n + 0.5 × VRRM × IF,n × k (14)

where VCE,n is the rated collector-emitter voltage of IGBTs,IC,n is the ideal rated collector current, n is the number ofIGBTs in the converter, VRRM is the rated repetitive peakreverse voltage of diodes, IF,n is the ideal rated diode forwardcurrent, and k is the number of diodes in the converter.

Considering that the diode silicon area is typically about 50%of that of the IGBTs in IGBT modules, the diodes are weightedwith 50% compared to the IGBTs.

The relative total installed switch power is calculated by nor-malizing the total installed switch power of a certain convertertopology to the total installed switch power of the 3L-NPCVSC, i.e.,

SStot,R =SStot

SStot,3L-NPC VSC× 100. (15)

The 2L-VSC on the basis of the 6.5-kV IGBT modules can-not be applied in applications where a high converter efficiency(e.g., η = 99%) and a low THD of the output voltage (e.g.,THD ≤ 5%) are required since an LC sine filter cannot berealized at low carrier frequencies (e.g., fC = 450 Hz). Further-more, the 2L-VSC is not attractive for high switching frequencymedium-voltage applications since the high switching losses ofthe 6.5-kV IGBT modules strongly limit the switch utilizationand the maximum switching frequency.

The unsymmetrical loss distribution within the 3L-NPC VSCand the additional neutral-point-clamp diodes are the reasonswhy the 3L-NPC VSC (η ≈ 99%, f1cb = 1000 Hz) requiresthe highest installed switch power. On the other hand, theexpense of the LC sine filter is moderate. Assuming an installedswitch power of SS = 38.6 MVA, a maximum first carrier bandfrequency of f1cb = 1900 Hz can be achieved. The high shareof switching losses causes a reduction of the installed switchpower at low switching frequency. A simple grid transformer, asmall dc link capacitor, and the possible modular realization ofcommon dc bus configurations are attractive additional features


TABLE VIIICOMPONENT VALUES AND STORED ENERGY OF 2.3-kV 2.39-MVA VSCS AT MAXIMUM SWITCHING FREQUENCY

Fig. 15. Simulation results of 3L-NPC VSC (fC,3L-NPC VSC = 1900 Hz).(a) Phase voltage vI (inverter side). (b) Inverter current iI . (c) Phase voltagevM (machine side). (d) Machine current iM .

Fig. 16. Simulation results of 4L-FLC VSC (fC,4L-FLC VSC = 1500 Hz).(a) Phase voltage vI (inverter side). (b) Inverter current iI . (c) Phase voltagevM (machine side). (d) Machine current iM .


Fig. 17. Simulation results of 5L-SCHB VSC (fC,5L-SCHB VSC =4580 Hz). (a) Phase voltage vI (inverter side). (b) Inverter current iI . (c) Phasevoltage vM (machine side). (d) Machine current iM .

of this topology [1]. Therefore, the 3L-NPC VSC is a compet-itive topology for a large variety of low and medium switchingfrequency applications (e.g., fC ≤ 1000 Hz).

The high capacitance values and stored energies of the fly-ing capacitors limit the use of the 3L/4L-FLC VSC to highswitching frequency applications (e.g., fC ≥ 1200 Hz) likehigh-speed drives and test benches. In these applications, the3L-FLC VSC is an interesting alternative to the 4L-FLC VSCdue to the reduced expense of flying capacitors. At lower car-rier frequencies (e.g., fC = 250−1000 Hz) and high converter

Fig. 18. One phase leg of 3L-NPC VSC and power card [29].

efficiency (e.g., η = 99%), both topologies are not competitivecompared to the 3L-NPC VSC and the 5L-SCHB VSC.

The 5L-SCHB VSC requires the lowest installed switchpower and stored energy of the LC sine filter. Compared to the3L-NPC VSC, the installed switch power and the stored energyof the 5L-SCHB VSC (η ≈ 99%) are reduced by 28% and 84%,respectively.

Furthermore, an extraordinary high maximum first carrierband frequency of the converter voltage can be achieved ata given installed switch power (e.g., SS = 38.6 MVA, f1cb =18 320 Hz). However, a complicated grid transformer, increaseddc link capacitance values compared to the 3L-NPC VSC, andthe absence of a common dc voltage bus are the disadvantagesof this topology [1]. Nevertheless, the 5L-SCHB VSC is anattractive topology for manifold MVDs including high-speeddrives.

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Fig. 19. Mechanical design structure. (a) One phase leg of a 4L-FLC converter. (b) 5L-SCHB converter.

TABLE IXCOMPARISON OF POWER PART COMPONENTS OF 2.3-kV 2.39-MVA VSCS

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Dietmar Krug received the Diploma degree inelectrical engineering from the Berlin University ofTechnology, Berlin, Germany, in 2001. He is cur-rently working toward the Ph.D. degree.

He joined the Power Electronics Group, BerlinUniversity of Technology in 2002, as a ResearchAssistant. Since July 2007, he has been a Devel-opment Engineer with Siemens AG Automation &Drives (Large Drives), Nuremberg, Germany. Hismajor research interest is the investigation of mul-tilevel voltage source converters for medium voltageapplications.

Steffen Bernet (M’97) received the M.S. degreefrom Dresden University of Technology, Dresden,Germany, in 1990 and the Ph.D. degree fromIlmenau University of Technology, Ilmenau,Germany, in 1995, both in electrical engineering.

During 1995 and 1996, he was with the ECEDepartment, University of Wisconsin, Madison, as aPostdoc. In 1996, he joined ASEA Brown Boverie(ABB) Corporate Research, Heidelberg, Germany,where he led the Electrical Drive Systems Group.From 1999 to 2000, he was responsible for the ABB

research worldwide in the areas of power electronics systems, drives, andelectric machines. From 2001 to 2007, he was a Professor of Power Electronicswith Berlin University of Technology, Berlin, Germany. Since June 2007, he hasbeen a Professor of Power Electronics with Dresden University of Technology.

Seyed Saeed Fazel was born in Iran in 1966. Hereceived the B.Sc. and M.Sc. degrees in electricalengineering from Iran University of Science andTechnology, Tehran, Iran, in 1990 and 1993, respec-tively, and is currently working toward the Ph.D.degree at Berlin University of Technology, Berlin,Germany.

He spent five years as an Engineer with JahadDaneshgahi Elm Va Sanat and was involved in high-voltage transformer-rectifier in electrostatic precip-itator applications. From 1998 to 2003, he was an

Assistant Professor with Iran University of Science and Technology. Hisresearch interests include power electronics, medium-voltage converter topolo-gies, and electrical machines.

Kamran Jalili was born in Iran in 1974. He receivedthe B.Sc. degree in power engineering from Powerand Water Institute of Technology, Tehran, Iran, in1997 and the M.Sc. degree in control engineeringfrom Tehran University, Tehran, in 2000. He is cur-rently working toward the Ph.D. degree at BerlinUniversity of Technology, Berlin, Germany.

His research interests include electrical drives andcontrol of converters with LCL filters.

Mariusz Malinowski (S’99–M’03) received theM.Sc.E.E. and Ph.D. degrees in electrical engineer-ing (with awards) from the Institute of Control andIndustrial Electronics, Warsaw University of Tech-nology (WUT), Warsaw, Poland, in 1997 and 2001,respectively.

He was a Visiting Scholar at Ålborg University,Ålborg, Denmark, at the University of Nevada, Reno,and at the Technical University of Berlin, Berlin,Germany. He is currently with the Institute of Con-trol and Industrial Electronics, WUT. He is the holder

of two patents, an author of 80 technical papers, and a coauthor of two bookchapters in Control in Power Electronics (New York: Academic, 2002). Hiscurrent research activities include control of PWM rectifiers and active filters,modulation techniques, and DSP applications.

Dr. Malinowski is the recipient of the Siemens Prize for his Ph.D. dis-sertation, the WUT President Prize, the Paper Award at the IEEE IndustrialElectronics, Control, and Instrumentation Conference 2000, and the PolishMinister of Education Award. He is also a Scholar of the Foundation for PolishScience and an Associate Editor of the IEEE TRANSACTIONS ON INDUSTRIAL

ELECTRONICS.

comparison of 2.3-kv medium-voltage multilevel converters for industrial medium-voltage drives

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medium voltage

mediumvoltage drives

minimum dclink voltage

nominal dclink voltage

dclink voltage reserve

insulated dc voltage

npc vscs

flc vscs