CERN – EUROPEAN ORGANIZATION FOR NUCLEAR RESEARCH

DESIGN OF COMPACT HIGH POWER RF COMPONENTS AT X-BAND

Alexej GRUDIEV

European Organization for Nuclear Research, Geneva, Switzerland

Abstract In this note, designs of compact high power X-band RF components are presented. Their scaling to C-and S-band is straightforward and in many case could solve a problem of size encountered in previous designs.

Geneva, Switzerland 17th May 2016

CLIC – Note – 1067

Design of compact high power RF components at X-band Alexej Grudiev, CERN

Abstract In this note, designs of compact high power X-band RF components are presented. Their scaling to C-and S-band is straightforward and in many case could solve a problem of size encountered in previous designs.

Introduction CLIC stands for Compact Linear Collider. Its compactness is reflected in many subsystem and components. For example, in the main linac practically everything must be compact and there the proposed components have already been used or could be used in the future modifications. Moreover, with the arrival of Toshiba 6MW X-band klystrons on the market an RF unit becomes more compact. This raises the need for more compact RF components. Furthermore, combining these lower power RF units requires more RF components. This makes more compact RF components even more desirable. In addition, scaling solutions proposed below to C- and S-band is straightforward and results in smaller size components which might be a great advantage at lower frequencies. In this note, the RF designs of several compact RF components are given including hybrids, rotating circular TE11 mode launcher, variable phase shifters and attenuators as well as a circular TE11 mode launcher with variable polarization.

Compact E-hybrids The compact hybrid presented in Fig 1 has been designed in 2010 for feeding X-band power from one PETS to two accelerating structures (AS) forming a SuperAS for CLIC CDR. It has actually been build and it is in high power operation in CTF3 CLEX. It is called an E-hybrid since the RF power splitting is done in the plane which is tangential to electric field. All four ports are WR90. The S-parameters are presented in Fig 2.

Fig 1. E-hybrid for CLIC SAS: geometry and E-field distribution at 1 W input power.

Fig 2. S-parameters of E-hybrid for CLIC SAS.

An alternative new design is presented in Fig 3. It is different from the CLIC one in the way it splits the power. In the new design the two outgoing power branches exit in the same direction. As can be seen from Fig 3 it is even more compact and from Fig. 4 it has larger bandwidth. Table 1 summarizes the main RF parameters of both designs.

Fig 3. New E-hybrid: geometry and E-field distribution at 1 W input power

Fig 4. S-parameters of the new E-hybrid.

Table 1. RF parameters of CLIC E-hybrid and the new E-hybrid.

CLIC E-hybrid New E-hybrid

Bandwidth at -30 dB [MHz] [MHz]

S11 187 312

S12 187 311

1-2Conversion - 220

Maximum Surface Fields at 100MW input power

E [MV/m] 34 43

H [kA/m] 91 76

S [MW/mm2] 1.1 1

In the design of the new E-hybrid a single goal function called Conversion has been introduced as defined below:

Conversion = |Re{S13}Im{S14} + Im{S13}Re{S14}| (1)

This goal function is equal 0.5 in the case of ideal hybrid for which three conditions are met:

1. |S13|2=0.5 2. |S14|2=0.5 3. |arg{S14}-arg{S13}| = π/2+nπ

The new goal function is more convenient and more efficient in the design especially using semi-automatic optimization routings available in commercial RF codes, like HFSS. In Fig. 4, (1-2Conversion) is plotted in green. This curve show smaller bandwidth in comparison to the bandwidth of S11 and S12 since it also includes the bandwidth of the 90 degree phase difference (condition 3 in the list above). This is actually the true bandwidth of a hybrid.

Compact E-plane circular TE11 rotating mode launcher (E-Rotator) Using the design of the new E-hybrid as a starting point, the design of a compact circular waveguide TE11 rotating mode launcher has been made by short circuiting the forward branches and attaching the circular waveguide in the middle as it is shown in Fig 5. Injecting power from port 1 results in circular polarized TE11 mode at port 3 rotating in negative direction (clockwise in Fig 5) and injecting power from port 2 results in the same mode rotating in positive direction (anti-clockwise in Fig 5). Since the rotating mode in the port 3 is a superposition of two degenerate TE11 circular waveguide modes with vertical and horizontal polarization and 90 degree phase shift the E-rotator is a 3 port hybrid with 4x4 S-matrix, where S13X and S13Y are the two degenerate TE11 circular waveguide modes. In the design of the E-rotator it has been essential to use the new hybrid conversion goal function defined similar to Eq. (1) as it is shown below:

Conversion = |Re{S13X}Im{S13Y} + Im{S13X}Re{S13Y}| (2)

Fig 6 shows the S-parameters frequency sweep and Table 2 summarizes the RF parameters of the E-rotator.

Fig 5. E-rotator: geometry and E-field distribution at 1 W input power injected from port 1 (left) or from port 2 (right).

Fig 6. S-parameters of the E-rotator.

Table 2. RF parameters of the E-rotator.

Bandwidth at -30 dB [MHz] Max Surface Fields at 100MW input power

S11 78 E [MV/m] 51

S12 65 H [kA/m] 180

1-2Conversion 50 S [MW/mm^2] 2

Compact variable phase shifter Taking the E-rotator as a starting building block and adding a movable short circuit to the circular waveguide port makes a variable phase shifter. To do this, a new compact movable short circuit has been designed as shown Fig 7. It is based on the transition from the diameter of the E-rotator circular port of 16 mm to a circular pipe of 22 mm diameter where a movable piston is placed. In order to isolate the coaxial port formed by the movable piston and the pipe a compact choke is introduced inside of the piston. Fig 8 shows the RF phase variation (upper plot ) and the isolation of the coaxial port (bottom plot) versus the piston displacement. The RF phase variation is ~20 degree/mm of piston displacement. When piston comes closer to the circular waveguide transition the isolation deteriorates, so the minimum distance between the transition and the piston is limited to ~20 mm. If it is respected the movable short shows good isolation with large bandwidth presented in Fig 9.

Fig 7. Movable short for circular waveguide: geometry of ¼ and the E-field distribution for 1W input power.

Fig 8. RF phase variation (top) and coaxial port isolation (bottom) versus the piston position.

Fig 9. Coaxial port isolation bandwidth.

This concept works as well in the rectangular waveguide, for example, in WR90 as shown in Fig 10.

Fig 10. Movable short for WR90 waveguide: geometry and E-field distribution at 1W input power.

Fig 11. Variable phase shifter: geometry and E-field distribution at 1 W input power.

Coming back to the phase shifter design we can complete it by connecting the movable short to the E-rotator as it is shown in Fig 11. In Fig 12 (top), the variation of RF phase of S12 is plotted with obviously the same sensitivity as for movable short: ~20 degree/mm. In Fig 12 (bottom), reflection (red) and isolations of the coaxial port are plotted versus the piston displacement demonstrating excellent performance.

Fig 12. Variation of RF phase advance from port 1 to port 2 (top) and the reflection and isolation (bottom) versus position of the piston in the movable short.

Variable power splitter Combining the variable RF phase shifter with two E-hybrid an ultra-compact variable power splitter is designed as shown in Fig 13. Power from port 1 is split between port 2 and 3 depending on the position of the piston in the variable phase shifter. Port 4 is always isolated (if no reflections come back from ports 2 and 3) and can be terminated by an RF load and/or used for vacuum pumping. Fig 14 shown bandwidth of the S11 and S14. In Fig 15, all four S-parameters are plotted versus the position of the piston.

Fig 13. Ultra-compact variable power splitter: geometry and E-field distribution at 1W input power.

Fig 14. Reflection and transmission to port 4.

Fig 15. S-parameters of the variable power splitter versus piston position.

Variable polarization circular waveguide TE11 mode launcher

Fig 16. E-rotator as a linear polarized mode launcher.

Another use of the E-rotator is proposed in the following section. As demonstrated in Fig 5, if power is injected from port 1, a circular polarized mode is launched in port 3. If it is injected from port 2, a circular polarized mode rotating in opposite direction is launched in port 3. Injecting the same

amount of power to both port 1 and port 2, linearly polarized mode is launched in the port 3, as is shown in Fig 16. The orientation of the linearly polarized mode depends on the RF phase between the waves injected in port 1 and port 2, which can be controlled by means of the variable phase shifter described above. Changing the phase difference between port 1 and port 2 by 180 degrees results in rotation of the polarization by 90 degree from vertical to horizontal. The overall layout is presented in Fig 17 illustrating how this device can be used as variable polarization mode launcher into a transverse deflecting structure with variable orientation of deflecting kick.

Fig 17. Overall layout of the variable polarization circular TE11 mode launcher into transverse deflecting structure.

Compact X-band all-metal RF load

Fig 18. Geometry of the compact X-band RF load

A new design of a compact all-metal high-power X-band RF load is proposed. The overall shape and conceptual design of the cooling and vacuum pumping are shown in Fig 18. The input power is split in two equal parts and then combined with the 90 degree RF phase difference in a dual feed coupler to launcher it into a corrugated waveguide which has very high attenuation. The power absorbed in the corrugations heats the outer wall which is cooled by a water cooling circuit placed very close by. A rather larger (compared to the wavelength) diameter of the corrugated waveguide helps both distribute heat load over larger surface of the outer wall and give efficient vacuum pumping via a standard CF flange at the far end of the load. The distributions of the electric and magnetic fields as

well as the Poynting vector are shown in Fig 19. Table 3 summarizes their values at 50 MW input power level.

Table 3 Maximum surface field values at 50 MW input power

Pin Esurf_max Hsurf_max Sc_max

50MW 55 MV/m 120 kA/m 0.6 MW/mm2

In Fig 20, reflection from the load are shown as a function of frequency for different values of surface roughness Ra in the range from 0 to 50 μm assuming RF load material to be Ti6V4Al titanium alloy with electrical conductivity of 600000 S/m. The roughness model built into ANSYS HFSS has been used to calculate these curves in order to have an idea about influence of the surface roughness on the load performance. As shown in Fig 20, roughness gives a non-negligible positive impact. The roughness has an effect not only on the reflection but also on the field distribution in the load as it is illustrated in Fig 21. Indeed, in the presented design increased roughness helps to redistribution the surface fields from a hot spot thus reducing the maximum surface field compared to zero-roughness design. The expected roughness in the additive manufacturing which is a potential way for load fabrication is about few tens of microns which would certainly improve the RF load performance compared to the case of no roughness.

Fig 19 Distributions of electric (top) and magnetic (middle) fields and Poynting vector (bottom) in the mid plane of the RF load at 2W input power.

Fig 20 Reflection from the load for different values of surface roughness: Ra = 0 – 50 μm.

Fig 21. Magnetic field distribution in the RF load for different values of the surface roughness.

Conclusions In conclusion, a number of innovative compact solutions have been found for high power RF waveguide components. They will help in building chipper and more compact RF units and combining them together if necessary. Furthermore, due to their compactness, all this solution have a great potential in scaling down to lower frequency at C- and S-bands.

Acknowledgements The author is thankful to Igor Syratchev for stimulating discussions to Walter Wuensch for his careful reading the manuscript useful comments.