ca3160-a - ca3160, ca3160a - 4mhz, bimos operational amplifier

FN976Rev 5.00

June 2004

CA31604MHz, BiMOS Operational Amplifier with MOSFET Input/CMOS Output

DATASHEET

The CA3160 is an operational amplifier that combines the advantages of both CMOS and bipolar transistors on a monolithic chip. The CA3160 series are frequency compensated versions of the popular CA3130 series.

Gate protected P-Channel MOSFET (PMOS) transistors are used in the input circuit to provide very high input impedance, very low input current, and exceptional speed performance. The use of PMOS field effect transistors in the input stage results in common-mode input voltage capability down to 0.5V below the negative supply terminal, an important attribute in single supply applications.

A complementary symmetry MOS (CMOS) transistor-pair, capable of swinging the output voltage to within 10mV of either supply voltage terminal (at very high values of load impedance), is employed as the output circuit.

The CA3160 Series circuits operate at supply voltages ranging from 5V to 16V, or 2.5V to 8V when using split supplies, and have terminals for adjustment of offset voltage for applications requiring offset null capability. Terminal provisions are also made to permit strobing of the output stage.

Features

• MOSFET Input Stage Provides:

- Very High ZI . . . . . . . . . . . . . 1.5T (1.5 x 1012) (Typ)- Very Low II . . . . . . . . . . . . . 5pA (Typ) at 15V Operation

. . . . . . . . . . . . . . . . . . . . . . . 2pA (Typ) at 5V Operation

• Common-Mode Input Voltage Range Includes Negative Supply Rail; Input Terminals Can Be Swung 0.5V Below Negative Supply Rail

• CMOS Output Stage Permits Signal Swing to Either (or Both) Supply Rails

Applications

• Ground Referenced Single Supply Amplifiers

• Fast Sample Hold Amplifiers

• Long Duration Timers/Monostables

• High Input Impedance Wideband Amplifiers

• Voltage Followers (e.g., Follower for Single SupplyD/A Converter)

• Wien-Bridge Oscillators

• Voltage Controlled Oscillators

• Photo Diode Sensor Amplifiers

PinoutCA3160 (PDIP)

TOP VIEW

NOTE: CA3160 Series devices have an on-chip frequency compensation network. Supplementary phase compensation or frequency roll-off (if desired) can be connected externally between Terminals 1 and 8.

Ordering Information

PART NUMBERTEMP.

RANGE (oC) PACKAGEPKG. NO.

CA3160E -55 to 125 8 Ld PDIP E8.3

OFFSET NULL

INV.

NON-INV.

V-

1

2

3

4

8

7

6

5

STROBE

V+

OUTPUT

OFFSET NULL

-+

INPUT

INPUT

FN976 Rev 5.00 of 18June 2004

CA3160

Absolute Maximum Ratings Thermal Information

Supply Voltage (Between V+ and V- Terminals) . . . . . . . . . . . .+16VDifferential Mode Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . .8VInput Voltage . . . . . . . . . . . . . . . . . . . . . . . . . (V+ +8V) to (V- -0.5V)Input Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1mAOutput Short Circuit Duration (Note 2). . . . . . . . . . . . . . . . Indefinite

Operating Conditions

Temperature Range. . . . . . . . . . . . . . . . . . . . . . . . . -55oC to 125oC

Thermal Resistance (Typical, Note 1) JA (oC/W) JC (oC/W)PDIP Package . . . . . . . . . . . . . . . . . . . 115 N/A

Maximum Junction Temperature (Plastic Package) . . . . . . . .150oCMaximum Storage Temperature Range . . . . . . . . . -65oC to 150oCMaximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC

CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of thedevice at these or any other conditions above those indicated in the operational sections of this specification is not implied.

NOTES:

1. JA is measured with the component mounted on an evaluation PC board in free air.

2. Short Circuit may be applied to ground or to either supply.

Electrical Specifications TA = 25oC, V+ = 15V, V- = 0V, Unless Otherwise Specified

PARAMETER SYMBOL TEST CONDITIONS

CA3160

UNITSMIN TYP MAX

Input Offset Voltage |VIO| VS = 7.5V - 6 15 mV

Input Offset Current |IIO| VS = 7.5V - 0.5 30 pA

Input Current II VS = 7.5V - 5 50 pA

Large-Signal Voltage Gain AOL VO = 10VP-P, RL = 2k 50 320 - kV/V

94 110 - dB

Common-Mode Rejection Ratio CMRR 70 90 - dB

Common-Mode Input-Voltage Range VlCR 0 -0.5 to 12 10 V

Power-Supply Rejection Ratio PSRR VIO/VS, VS = 7.5V - 32 320 V/V

Maximum Output Voltage VOM+ RL = 2k 12 13.3 - V

VOM- - 0.002 0.01 V

VOM+ RL = 14.99 15 - V

VOM- - 0 0.01 V

Maximum Output Current IOM+ VO = 0V (Source) 12 22 45 mA

IOM- VO = 15V (Sink) 12 20 45 mA

Supply Current (Note 3) I+ VO = 7.5V, R L = - 10 15 mA

VO = 0V, R L = - 2 3 mA

Input Offset Voltage Temperature Drift VIO/T - 8 - V/oC

Electrical Specifications For Design Guidance, VSUPPLY = 7.5V, TA = 25oC, Unless Otherwise Specified

PARAMETER SYMBOL TEST CONDITIONS TYP UNITS

Input Offset Voltage Adjustment Range 10k Across Terminals 4 and 5 or Terminals 4 and 1 22 mV

Input Resistance RI 1.5 T

Input Capacitance CI f = 1MHz 4.3 pF

Equivalent Input Noise Voltage eN BW = 0.2MHz RS = 1M 40 V

RS = 10M 50 V

Equivalent Input Noise Voltage eN RS = 100 1kHz 72 nV/Hz

10kHz 30 nV/Hz

Unity Gain Crossover Frequency fT 4 MHz

Slew Rate SR 10 V/s

FN976 Rev 5.00 of 18June 2004

CA3160

Block Diagram

Transient Response Rise and Fall Time tr CL = 25pF, RL = 2k, (Voltage Follower) 0.09 s

Overshoot OS 10 %

Settling Time tS CL = 25pF, RL = 2k, (Voltage Follower) To <0.1%, VIN = 4VP-P

1.8 s

Electrical Specifications For Design Guidance, VSUPPLY = 7.5V, TA = 25oC, Unless Otherwise Specified (Continued)


Electrical Specifications For Design Guidance, V+ = +5V, V- = 0V, TA = 25oC, Unless Otherwise Specified


Input Offset Voltage VIO 6 mV

Input Offset Current IIO 0.1 pA

Input Current Il 2 pA

Common-Mode Rejection Ratio CMRR 80 dB

Large Signal Voltage Gain AOL VO = 4VP-P, RL = 5k 100 kV/V

100 dB

Common-Mode Input Voltage Range VlCR 0 to 2.8 V

Supply Current I+ VO = 5V, RL = 300 A

VO = 2.5V, RL = 500 A

Power Supply Rejection Ratio PSRR VIO/V+ 200 V/V

NOTE:

3. ICC typically increases by 1.5mA/MHz during operation.

6

BIAS CKT.

200A 1.35mA 200A

78mA(NOTE 4)

OUTPUT

AV 30XAV 6000X

STROBE

V-

V+

OFFSETNULL

COMPENSATION(WHEN DESIRED)

+

-

INPUT AV 5X

CC

NOTES:

4. Total supply voltage (for indicated voltage gains) = 15V with input terminals biased so that Terminal 6 potential is +7.5V above Terminal 4.

5. Total supply voltage (for indicated voltage gains) = 15V with output terminal driven to either supply rail.

0mA(NOTE 5)

4

815

2

3

FN976 Rev 5.00 of 18June 2004

CA3160

Schematic Diagram

Application Information

Circuit Description

Refer to the Block Diagram of the CA3160 series CMOS Operational Amplifiers. The input terminals may be operated down to 0.5V below the negative supply rail, and the output can be swung very close to either supply rail in many applications. Consequently, the CA3160 series circuits are ideal for single supply operation. Three class A amplifier stages, having the individual gain capability and current consumption shown in the Block Diagram provide the total gain of the CA3160. A biasing circuit provides two potentials for common use in the first and second stages. Terminals 8 and 1 can be used to supplement the internal phase compensation network if additional phase compensation or frequency roll-off is desired. Terminals 8 and 4 can also be used to strobe the output stage into a low quiescent current state. When Terminal 8 is tied to the negative supply rail (Terminal 4) by mechanical or electrical means, the output potential at Terminal 6 essentially rises to the positive supply-rail potential at Terminal 7. This condition of essentially zero current drain in the output stage under the strobed “OFF” condition can only be achieved when the ohmic load resistance presented to

the amplifier is very high (e.g., when the amplifier output is used to drive MOS digital circuits in comparator applications).

Input Stage - The circuit of the CA3160 is shown in the Schematic Diagram. It consists of a differential-input stage using PMOS field-effect transistors (Q6, Q7) working into a mirror-pair of bipolar transistors (Q9, Q10) functioning as load resistors together with resistors R3 through R6. The mirror-pair transistors also function as a differential-to-single-ended converter to provide base drive to the second-stage bipolar transistor (Q11). Offset nulling, when desired, can be effected by connecting a 100,000 potentiometer across Terminals 1 and 5 and the potentiometer slider arm to Terminal 4. Cascode-connected PMOS transistors Q2, Q4, are the constant-current source for the input stage. The biasing circuit for the constant-current source is subsequently described. The small diodes D5 through D7 provide gate-oxide protection against high-voltage transients, including static electricity during handling for Q6 and Q7.

Second-Stage - Most of the voltage gain in the CA3160 is provided by the second amplifier stage, consisting of bipolar transistor Q11 and its cascode-connected load resistance provided by PMOS transistors Q3 and Q5. The source of bias

7

4815

2

3

BIAS CURRENT “CURRENT SOURCELOAD” FOR Q11

Q2

D1

D2

D3

D4

Z18.3V

Q1

R140k

Q4

R25k

INPUT STAGE

D5

NON-INV.INPUT

INV. INPUT

+

-

Q6

R31k

Q9 Q10

R51k

R61k

R41k

Q7

D6D7

Q3

OFFSET NULL

Q11

SUPPLEMENTARYCOMP IF DESIRED STROBING

SECOND

OUTPUTQ8

Q12

STAGE

STAGE

Q5

V+

2k

30pF

6

OUTPUT

NOTE: Diodes D5 Through D7 Provide Gate Oxide Protection For MOSFET Input Stage.

CURRENT SOURCEFOR Q6 AND Q7

FN976 Rev 5.00 of 18June 2004

CA3160

potentials for these PMOS transistors is described later. Miller Effect compensation (roll off) is accomplished by means of the 30pF capacitor and 2k resistor connected between the base and collector of transistor Q11. These internal components provide sufficient compensation for unity gain operation in most applications. However, additional compensation, if desired, may be used between Terminals 1 and 8.

Bias-Source Circuit - At total supply voltages, somewhat above 8.3V, resistor R2 and zener diode Z1 serve to establish a voltage of 8.3V across the series-connected circuit, consisting of resistor R1, diodes D1 through D4, and PMOS transistor Q1. A tap at the junction of resistor R1 and diode D4 provides a gate-bias potential of about 4.5V for PMOS transistors Q4 and Q5 with respect to Terminal 7. A potential of about 2.2V is developed across diode-connected PMOS transistor Q1 with respect to Terminal 7 to provide gate bias for PMOS transistors Q2 and Q3. It should be noted that Q1 is “mirror-connected” to both Q2 and Q3. Since transistors Q1, Q2, Q3 are designed to be identical, the approximately 200A current in Q1 establishes a similar current in Q2 and Q3 as constant-current sources for both the first and second amplifier stages, respectively.

At total supply voltages somewhat less than 8.3V, zener diode Z1 becomes nonconductive and the potential, developed across series-connected R1, D1 - D4, and Q1, varies directly with variations in supply voltage. Consequently, the gate bias for Q4, Q5 and Q2, Q3 varies in accordance with supply-voltage variations. This variation results in deterioration of the power-supply-rejection ratio (PSRR) at total supply voltages below 8.3V. Operation at total supply voltages below about 4.5V results in seriously degraded performance.

Output Stage - The output stage consists of a drain-loaded inverting amplifier using CMOS transistors operating in the Class A mode. When operating into very high resistance loads, the output can be swung within millivolts of either supply rail. Because the output stage is a drain-loaded amplifier, its gain is dependent upon the load impedance. The transfer characteristics of the output stage for a load returned to the negative supply rail are shown in Figure 17. Typical op amp loads are readily driven by the output stage. Because large-signal excursions are non-linear, requiring feedback for good waveform reproduction, transient delays may be encountered. As a voltage follower, the amplifier can achieve 0.01% accuracy levels, including the negative supply rail.

Offset Nulling

Offset-voltage nulling is usually accomplished with a 100,000 potentiometer connected across Terminals 1 and 5 and with the potentiometer slider arm connected to Terminal 4. A fine offset-null adjustment usually can be effected with the slider arm positioned in the mid-point of the potentiometer's total range.

Input Current Variation with Common Mode Input Voltage

As shown in the Electrical Specifications, the input current for the CA3160 Series Op Amps is typically 5pA at TA = 25oC when Terminals 2 and 3 are at a common-mode potential of +7.5V with respect to negative supply Terminal 4. Figure 23 contains data showing the variation of input current as a function of common-mode input voltage at TA = 25oC. These data show that circuit designers can advantageously exploit these characteristics to design circuits which typically require an input current of less than 1pA, provided the common-mode input voltage does not exceed 2V. As previously noted, the input current is essentially the result of the leakage current through the gate-protection diodes in the input circuit and, therefore, a function of the applied voltage. Although the finite resistance of the glass terminal-to-case insulator of the metal can package also contributes an increment of leakage current, there are useful compensating factors. Because the gate-protection network functions as if it is connected to Terminal 4 potential, and the metal can case of the CA3160 is also internally tied to Terminal 4, input Terminal 3 is essentially “guarded” from spurious leakage currents.

Input-Current Variation with Temperature

The input current of the CA3160 Series circuits is typically 5pA at 25oC. The major portion of this input current is due to leakage current through the gate-protective diodes in the input circuit. As with any semiconductor junction device, including op amps with a junction-FET input stage, the leakage current approximately doubles for every 10oC increase in temperature. Figure 24 provides data on the typical variation of input bias current as a function of temperature in the CA3160.

In applications requiring the lowest practical input current and incremental increases in current because of “warm-up” effects, it is suggested that an appropriate heat sink be used with the CA3160. In addition, when “sinking” or “sourcing” significant output current the chip temperature increases, causing an increase in the input current. In such cases, heat-sinking can also very markedly reduce and stabilize input current variations.

Input Offset Voltage (VIO) Variation with DC Bias vs Device Operating Life

It is well known that the characteristics of a MOSFET device can change slightly when a DC gate-source bias potential is applied to the device for extended time periods. The magnitude of the change is increased at high temperatures. Users of the CA3160 should be alert to the possible impacts of this effect if the application of the device involves extended operation at high temperatures with a significant differential DC bias voltage applied across Terminals 2 and 3. Figure 25 shows typical data pertinent to shifts in offset voltage encountered with CA3160 devices in metal can packages during life testing. At lower temperatures (metal can and plastic) for example at 85oC, this change in voltage is considerably less. In typical linear applications where the differential voltage is small and symmetrical, these incremental changes are of about the same magnitude as those encountered in an operational amplifier employing a bipolar transistor input stage. The 2V differential voltage example

FN976 Rev 5.00 of 18June 2004

CA3160

represents conditions when the amplifier output state is “toggled”, e.g., as in comparator applications.

Power Supply Considerations

Because the CA3160 is very useful in single supply applications, it is pertinent to review some considerations relating to power supply current consumption under both single and dual supply service. Figures 1A and 1B show the CA3160 connected for both dual and single supply operation.

Dual-supply operation: When the output voltage at Terminal 6 is 0V, the currents supplied by the two power supplies are equal. When the gate terminals of Q8 and Q12 are driven increasingly positive with respect to ground, current flow through Q12 (from the negative supply) to the load is increased and current flow through Q8 (from the positive supply) decreases correspondingly. When the gate terminals of Q8 and Q12 are driven increasingly negative with respect to ground, current flow through Q8 is increased and current flow through Q12 is decreased accordingly.

Single supply operation: Initially, let it be assumed that the value of RL is very high (or disconnected), and that the input-terminal bias (Terminals 2 and 3) is such that the output terminal (No. 6) voltage is at V+/2, i.e., the voltage-drops across Q8 and Q12 are of equal magnitude. Figure 18 shows typical quiescent supply-current vs supply voltage for the CA3160 operated under these conditions.

Since the output stage is operating as a Class A amplifier, the supply current will remain constant under dynamic operating conditions as long as the transistors are operated in the linear portion of their voltage-transfer characteristics (see Figure 17). If either Q8 or Q12 are swung out of their linear regions toward cutoff (a non-linear region), there will be a corresponding reduction in supply-current. In the extreme case, e.g., with Terminal 8 swung down to ground potential (or tied to ground), NMOS transistor Q12 is completely cut off and the supply current to series connected transistors Q8, Q12 goes essentially to zero. The two preceding stages in the CA3160, however, continue to draw modest supply-current (see the lower curve in Figure 18) even though the output stage is strobed off. Figure 1A shows a dual-supply arrangement for the output stage that can also be strobed off, assuming RL = , by pulling the potential of Terminal 8 down to that of Terminal 4.

Let it now-be assumed that a load resistance of nominal value (e.g., 2k) is connected between Terminal 6 and ground in the circuit of Figure 1B. Let it further be assumed again that the input-terminal bias (Terminals 2 and 3) is such that the output terminal (No. 6) voltage is at V+/2. Since PMOS transistor Q8 must now supply quiescent current to both RL and transistor Q12, it should be apparent that under these conditions the supply current must increase as an inverse function of the RL magnitude. Figure 20 shows the voltage-drop across PMOS transistor Q8 as a function of load current at several supply voltages. Figure 17 shows the voltage transfer characteristics of the output stage for several values of load resistance.

Wideband Noise

From the standpoint of low-noise performance considerations, the use of the CA3160 is most advantageous in applications where in the source resistance of the input signal is on the order of 1M or more. In this case, the total input-referred noise voltage is typically only 40V when the test circuit amplifier of Figure 2 is operated at a total supply voltage of 15V. This value of total input-referred noise remains essentially constant, even though the value of source resistance is raised by an order of magnitude. This characteristic is due to the fact that reactance of the input capacitance becomes a significant factor in shunting the source resistance. It should be noted, however, that for values of source resistance very much greater than 1M, the total noise voltage generated can be dominated by the thermal noise contributions of both the feedback and source resistors.

73

2

8

4

6

V+

-

+

CA3160

NEGATIVESUPPLY

OUTPUT

RL

STAGEQ12

Q8

V-

FIGURE 1A. DUAL POWER SUPPLY OPERATION

73

2

8

4

6

-

+

CA3160OUTPUT

RL

STAGEQ12

Q8

V+

FIGURE 1B. SINGLE POWER SUPPLY OPERATION

FIGURE 1. CA3160 OUTPUT STAGE IN DUAL AND SINGLE POWER SUPPLY OPERATION

FN976 Rev 5.00 of 18June 2004

CA3160

CA3160 6

73

42

+7.5V

0.01F

NOISEVOLTAGEOUTPUT

30.1k0.01F

1k

-7.5V

RS

1M+

-

BW (3dB) = 200kHzTOTAL NOISE VOLTAGE(INPUT REFERRED = 40V (TYP)

FIGURE 2. TEST CIRCUIT AMPLIFIER (30dB GAIN) USED FOR WIDEBAND NOISE MEASUREMENTS

Typical Performance Curves

FIGURE 3A.

FIGURE 3B. SMALL SIGNAL RESPONSE FIGURE 3C. INPUT-OUTPUT DIFFERENCE SIGNAL SHOWINGSETTLING TIME

FIGURE 3. DUAL SUPPLY VOLTAGE FOLLOWER WITH ASSOCIATED WAVEFORMS

CA3160 6

73

42

+7.5V

0.01F

0.01F

-7.5V

10k+

-2k

25pFSIMULATEDLOADCAPACITANCE

0.1FBW (-3dB) = 4MHzSR = 10V/s

2k

Top Trace: OutputBottom Trace: Input

Top Trace: Output SignalCenter Trace: Difference Signal 5mV/Div.

Bottom Trace: Input Signal

FN976 Rev 5.00 of 18June 2004

CA3160

Typical Applications

Voltage Followers

Operational amplifiers with very high input resistances, like the CA3160, are particularly suited to service as voltage followers. Figure 3 shows the circuit of a classical voltage follower, together with pertinent waveforms using the CA3160 in a split-supply configuration.

A voltage follower, operated from a single supply, is shown in Figure 4 together with related waveforms. This follower circuit is linear over a wide dynamic range, as illustrated by the reproduction of the output waveform in Figure 4B with input-signal ramping. The waveforms in Figure 4C show that the follower does not lose its input-to-output phase-sense, even though the input is being swung 7.5V below ground potential. This unique characteristic is an important attribute in both operational amplifier and comparator applications. Figure 4C also shows the manner in which the COS/MOS output stage permits the output signal to swing down to the negative supply-rail potential (i.e., ground in the case shown). The digital-to-analog converter (DAC) circuit, described in the following section, illustrates the practical use of the CA3160 in a single supply voltage follower application.

9-Bit CMOS DAC

A typical circuit of a 9-bit Digital-to-Analog Converter (DAC) (see Note 6) is shown in Figure 5. This system combines the concepts of multiple-switch CMOS lCs, a low-cost ladder network of discrete metal-oxide-film resistors, a CA3160 op amp connected as a follower, and an inexpensive monolithic regulator in a simple single power-supply arrangement. An additional feature of the DAC is that it is readily interfaced with CMOS input logic, e.g., 10V logic levels are used in the circuit of Figure 5.

The circuit uses an R/2R voltage-ladder network, with the output-potential obtained directly by terminating the ladder arms at either the positive or the negative power supply terminal. Each CD4007A contains three inverters, each inverter functioning as a single-pole double-throw switch to terminate an arm of the R/2R network at either the positive or negative power-supply terminal. The resistor ladder is an assembly of 1% tolerance metal-oxide film resistors. The five arms requiring the highest accuracy are assembled with series and parallel combinations of 806,000 resistors from the same manufacturing lot.

A single 15V supply provides a positive bus for the CA3160 follower amplifier and feeds the CA3085 voltage regulator. A “scale-adjust” function is provided by the regulator output control, set to a nominal 10V level in this system. The line-voltage regulation (approximately 0.2%) permits a 9-bit accuracy to be maintained with variations of several volts in the supply. The flexibility afforded by the CMOS building blocks simplifies the design of DAC systems tailored to particular needs.

NOTE:6. “Digital-to-Analog Conversion Using the Intersil CD4007A

COS/MOS lC”, Application Note AN6080.

FIGURE 4A.

FIGURE 4B. OUTPUT WAVEFORM WITH GROUND REFERENCE SINE WAVE INPUT

FIGURE 4C. OUTPUT SIGNAL WITH INPUT SIGNAL RAMPING

FIGURE 4. SINGLE SUPPLY VOLTAGE FOLLOWER WITH ASSOCIATED WAVEFORMS. (E.G., FOR USE IN SINGLE SUPPLY D/A CONVERTER; SEE FIGURE 9 IN AN6080)

CA3160 6

73

42

+15V

0.01F

10k+

-

0.1FBW (-3dB) = 4MHzSR = 10V/s

15

2k

OFFSETADJUST

Top Trace: OutputBottom Trace: Input

FN976 Rev 5.00 of 18June 2004

CA3160

Error-Amplifier in Regulated Power Supplies

The CA3160 is an ideal choice for error-amplifier service in regulated power supplies since it can function as an error-amplifier when the regulated output voltage is required to approach zero.

The circuit shown in Figure 6 uses a CA3160 as an error amplifier in a continuously adjustable 1A power supply. One of the key features of this circuit is its ability to regulate down to the vicinity of 0V with only one DC power supply input.

An RC network, connected between the base of the output drive transistor and the input voltage, prevents “turn-on overshoot”, a condition typical of many operational amplifier regulator circuits. As the amplifier becomes operational, this RC network ceases to have any influence on the regulator performance.

Precision Voltage-Controlled Oscillator

The circuit diagram of a precision voltage-controlled oscillator is shown in Figure 7. The oscillator operates with a tracking error in the order of 0.02% and a temperature coefficient of 0.01%/oC. A multivibrator (A1) generates pulses of constant amplitude (V) and width (T2). Since the output (Terminal 6) of A1 (a CA3130) can swing within about 10mV of either supply-rail, the output pulse amplitude (V) is essentially equal to V+. The average output voltage (EAVG = V T2/T1) is applied to the non-inverting Input terminal of comparator A2 via an integrating network R3, C2. Comparator A2 operates to establish circuit conditions such that

EAVG = V1. This circuit condition is accomplished by feeding an output signal from Terminal 6 of A2 through R4, D4 to the inverting terminal (Terminal 2) of A1, thereby adjusting the multivibrator interval, T3.

Voltmeter With High Input Resistance

The voltmeter circuit shown in Figure 8 illustrates an application in which a number of the CA3160 characteristics are exploited. Range-switch SW1 is ganged between input and output circuitry to permit selection of the proper output voltage for feedback to Terminal 2 via 10k current-limiting resistor. The circuit is powered by a single 8.4V mercury battery. With zero input signal, the circuit consumes somewhat less than 500A plus the meter current required to indicate a given voltage. Thus, at full scale input, the total supply current rises to slightly more than 1500A.

Function Generator

A function generator having a wide tuning range is shown in Figure 9. The adjustment range, in excess of 1,000,000/1, is accomplished by a single potentiometer. Three operational amplifiers are utilized: a CA3160 as a voltage follower, a CA3080 as a high speed comparator, and a second CA3080A as a programmable current source. Three variable capacitors C1, C2, and C3 shape the triangular signal between 500kHz and 1MHz. Capacitors C4, C5, and the trimmer potentiometer in series with C5 maintain essentially constant (+10%) amplitude up to 1MHz.

CA3085

12

63

0.001F

47

6

13

8

1

5

12

CD4007A“SWITCHES”

103

14

11

2

9

4

7

806K1%

402K1%

200K1%

806K1%

6

13

8

1 12


103

100K1%

(2)806K

6

13

8

12


103

806K1%

806K1%

1%

(4)806K1%

(8)806K1%

1

5

806K1%

750K1%

10V LOGIC INPUTS

PARALLELEDRESISTORS

+10.010V

LSB MSB

9 8 7 6 5 4 3 2 1

8+10.010V

22.1K1%

1K

3.83K1%

2F25V

+

-

+15V 62

VOLTAGEREGULATOR

REGULATEDVOLTAGEADJUST

CA3160

3

4 2

0.1F

+

-

15

6

2K

+15V

100K

VOLTAGEFOLLOWER

7

OFFSETNULL

OUTPUT

LOAD

10K

BITREQUIRED

RATIO-MATCH

1 Standard

2 0.1%

3 0.2%

4 0.4%

5 0.8%

6 - 9 1% ABS.

FIGURE 5. 9-BIT DAC USING CMOS DIGITAL SWITCHES AND CA3160

806K

1%5

FN976 Rev 5.00 of 18June 2004

CA3160

5

41

6

7

8

+

-

CA3160

2

3

1

2

3

40V INPUT+

2.4k1W

100F25V

CA3086

+

-+

-

2.2k

5F

10 11 2 1

9 357

6 4

12 14

13

0.2F

TURNON

DELAY

100k 1.5k1W

1k

62k-

2N6385POWER DARLINGTON SHORT-CIRCUIT CURRENT

LIMIT ADJUSTMENT1

10k

1k2N2102

1N914 56pF 43k

10k

-

+

OUTPUT0V TO 35VAT 1A

10k

4.7k

0.01F

8.2k

100F

100k50k

2k

-

Hum and Noise Output <250VRMS; Regulation (No Load to Full Load) <0.005%; Input Regulation <0.01%/V

FIGURE 6. VOLTAGE REGULATOR CIRCUIT (0.1V TO 35V AT 1A)

1k

2N2102

8

4

7+

-

MULTI-

3

R5

VIBRATORCA3130

2

6

+

-

COMPARATORCA3160

3 4

1

2 7

6

+15V

100K

R6100K

C1500pF

D1

D20.1F

R1182K

100K

10K

fO

EAVG = V T2/T1R31M

0.01F1M

R210K

VCO CONTROL VOLTAGE (VI)(0V - 10V)

(SENSITIVITY = 1kHz/V)

C20.01F

D1 - D5 = 1N914

R4

R7100K

0.01F

D5

T2 T3

V

T1

+15V+15V

D3

D4

FIGURE 7. VOLTAGE CONTROLLED OSCILLATOR

3K

5

FN976 Rev 5.00 of 18June 2004

CA3160

300V

100V

30V

10V

1V

300V

100V

30V

10V

3V

1V

300V

100V

30V

10V

3V

300V

100V

30V

10V

100M

1.02

SW1A

INPUT

M

CA3160

73

42

15

6

+

-

30V

10V

3V

1V

300mV

100V

300V

30mV

10mV

100mV

SW1B

SW1C

SW1D

M

30V

10V

1V

100V

300V

3V

300mV

30mV

10mV

100mV9k

900

100

BATTERYTEST

OFF ON

3 POSITIONSLIDE SWITCH

BATTERY

100k

ZEROADJUST

10k

9.1k

1V CAL.

820 200

2.7k3V CAL.

500

+9VBATTERY

500F

0-1mA

+

-

0.001F22M

9.9k

FIGURE 8. HIGH INPUT RESISTANCE DC VOLTMETER

1k

3

7

5

CA3080A

+

-42

6 3

CA3160

+

-2

7

4

6 2

CA3080

+

-

3

7

6

5

20pF

8.2k

+7.5VVOLTAGE-CONTROLLED

CURRENT SOURCE

1k

2M-7.5V

4.7k

-7.5V

SYMMETRY

+7.5V

10k

MAX FREQSET

500FREQ

ADJUST

MIN FREQ.SET

-7.5V

EXTERNALSWEEPING INPUT

10-80pF

C2

100k

+7.5V

0.9 - 7pFC1

6.2k

HIGHFREQ.SHAPE

4 - 60pFC3

BUFFERVOLTAGE FOLLOWER

+7.5V

0.1F

CENTERING100k

430pF

10k

2k

C44 - 60pF

HIGH FREQLEVEL

ADJUST

6.8M

-7.5V +7.5V

30k

10k

50k

C515 - 115pF

0.1F

-7.5V

THRESHOLDDETECTOR

2-1N914

+7.5V

-7.5V

6.2k 500

FIGURE 9A. 1,000,000/1 SINGLE CONTROL FUNCTION GENERATOR: 1Hz to 1MHz

FN976 Rev 5.00 of 18June 2004

CA3160

NOTE: A square wave signal modulates the external sweeping input to produce 1Hz and 1MHz, showing the 1,000,000/1 frequency range of the Function Generator.

FIGURE 9B. TWO-TONE OUTPUT SIGNAL FROM THE FUNCTION GENERATOR

NOTE: The bottom trace is the sweeping signal and the top trace is the actual generator output. The center trace displays the 1MHz signal via delayed oscilloscope triggering of the upper swept output signal.

FIGURE 9C. TRIPLE-TRACE OF THE FUNCTION GENERATOR SWEEPING TO 1MHz

FIGURE 9. 1,000,000/1 SINGLE CONTROL FUNCTION GENERATOR: 1Hz to 1MHz

8

3

2

4

7

CA3130

+

-

6 2

3

4

7

CA3160

+

-6 3

2

4

7

CA3130

+

- 8

6

+15V

100k 1M

+15V

15 - 115pFFREQ

ADJUST

MULTIVIBRATOR RETRACE INHIBIT

100k

100k

MULTIVIBRATOR

STEP HEIGHTADJUST4 - 60pF

8.2k

CHARGECOMMUTATING

NETWORK

470pF

+15V

INTEGRATOR

STAIRCASEOUTPUT

10k

2k

1.5

HYSTERESIS SWITCH

+15V

+15V

+15mV TO +10V51k

100k

5.1k 1N914

M

1N914

FIGURE 10A. STAIRCASE GENERATOR CIRCUIT

FN976 Rev 5.00 of 18June 2004

CA3160

Staircase Generator

Figure 10 shows a staircase generator circuit utilizing three CMOS operational amplifiers. Two CA3130s are used; one as a multivibrator, the other as a hysteresis switch. The third amplifier, a CA3160, is used as a linear staircase generator.

Picoammeter Circuit

Figure 11 is a current-to-voltage converter configuration utilizing a CA3160 and CA3140 to provide a picoampere meter for 13pA full scale meter deflection. By placing Terminals 2 and

4 of the CA3160 at ground potential, the CA3160 input is operated in the “guarded mode”. Under this operating condition, even slight leakage resistance present between Terminals 3 and 2 or between Terminals 3 and 4 would result in zero voltage across this leakage resistance, thus substantially reducing the leakage current.

If the CA3160 is operated with the same voltage on input Terminals 3 and 2 as on Terminal 4, a further reduction in the input current to the less than one picoampere level can be achieved as shown in Figure 23.

To further enhance the stability of this circuit, the CA3160 can be operated with its output (Terminal 6) near ground, thus markedly reducing the dissipation by reducing the supply current to the device.

The CA3140 stage serves as a X100 gain stage to provide the required plus and minus output swing for the meter and feedback network. A 100-to-1 voltage divider network consisting of a 9.9k resistor in series with a 100 resistor sets the voltage at the 10G resistor (in series with Terminal 3) to 30mV full-scale deflection. This 30mV signal results from 3V appearing at the top of the voltage divider network which also drives the meter circuitry.

By utilizing a switching technique in the meter circuit and in the 9.9k and 100 network similar to that used in voltmeter circuit shown in Figure 8, a current range of 3pA to 1nA full scale can be handled with the single 10G resistor.

STAIRCASEOUTPUT

2V STEPS

COMPARATOR

OSCILLATOR

Top Trace: Staircase Output 2V StepsCenter Trace: ComparatorBottom Trace: Oscillator

FIGURE 10B. STAIRCASE GENERATOR WAVEFORM

FIGURE 10. STAIRCASE GENERATOR CIRCUIT

4

1

7

CA3160

+

-

6

10M

+15V

100k

10k

5

2

4

6

7

CA3140

+

-

3

1M

M

9.9k 5.6k

500-0-500A

-15V

560k

9.1k

500

100

+15V

10G

10pF

0.1F

2

0.1F

3

-15V

FIGURE 11. CURRENT-TO-VOLTAGE CONVERTER TO PROVIDE A PICOAMMETER WITH 3pA FULL SCALE DEFLECTION

FN976 Rev 5.00 of 18June 2004

CA3160

Single Supply Sample-and-Hold SystemFigure 12 shows a single supply sample-and-hold system using a CA3160 to provide a high input impedance and an input voltage range of 0V to 10V. The output from the input buffer integrator network is coupled to a CA3080A. The CA3080A functions as a strobeable current source for the CA3140 output integrator and storage capacitor. The CA3140 was chosen because of its low output impedance and constant gain-bandwidth product. Pulse “droop” during the hold interval can be reduced to zero by adjusting the 100k bias-voltage potentiometer on the positive input of the CA3140. This zero adjustment sets the CA3080A output voltage at its zero current position. In this sample-and-hold circuit it is essential that the amplifier bias current be reduced to zero to minimize output signal current during the hold mode. Even with 320mV at the

amplifier bias circuit terminal (5) at least 1100pA of output current will be available.

Wien Bridge Oscillator

A simple, single supply Wien Bridge oscillator using a CA3160 is shown in Figure 13. A pair of parallel-connected 1N914 diodes comprise the gain-setting network which standardizes the output voltage at approximately 1.1V. The 500 potentiometer is adjusted so that the oscillator will always start and the oscillation will be maintained. Increasing the amplitude of the voltage may lower the threshold level for starting and for sustaining the oscillation, but will introduce more distortion.

Top Trace: Sampled OutputCenter Trace: Input Signal

Bottom Trace: Sampling Pulses

FIGURE 12B. SAMPLE AND HOLD WAVEFORM

Top Trace: Sampled OutputCenter Trace: Input Signal

Bottom Trace: Sampling Pulses

FIGURE 12C. SAMPLE AND HOLD WAVEFORM

FIGURE 12. SINGLE SUPPLY SAMPLE AND HOLD SYSTEM, INPUT 0V TO 10V

4

1

7

CA3160

+

-

1M

+15V

100k

5

9.1k

2

0.1F

3

6

0.1F

2

6

34

CA3080A

+

- 7

5

2

34

CA3140

+

-7

6

+15V

0.1F39k

2200pF

8.2

1M100k

27k

STROBE INPUT

500A

DROOPZERO

ADJUST

SAMPLE =HOLD =

15V0V

100k

30pF

39k

0.1F

1N914

OFFSETVOLTAGE

ADJUST

8.2k

2k

0.1F

+15V

FIGURE 12A. SINGLE SUPPLY SAMPLE AND HOLD SYSTEM, INPUT 0V TO 10V

8

SAMPLEDOUTPUT

INPUTSIGNAL

SAMPLINGPULSES

SAMPLEDOUTPUT

INPUT 0V-

SAMPLINGPULSES

0V-

FN976 Rev 5.00 of 18June 2004

CA3160

Operation with Output Stage Power Booster

The current sourcing and sinking capability of the CA3160 output stage is easily supplemented to provide power-boost capability. In the circuit of Figure 14, three CMOS transistor-pairs in a single CA3600 lC array are shown parallel-connected with the output stage in the CA3160. In the Class A mode of CA3600E shown, a typical device consumes 20mA of supply current at 15V operation. This arrangement boosts the current-handling capability of the CA3160 output stage by about 2.5X.

The amplifier circuit in Figure 14 employs feedback to establish a closed-loop gain of 20dB. The typical large-signal-bandwidth (-3dB) is 190kHz.

4

CA3160

+

-

7

6

0.01F

2k

f =1

2 (R1 || R2) C1 R3 C2

R1100k

R351k

+15V

R2100k

C110-80pF

680

500

0.1F

C251pF

OUTPUTf = 100kHz2% THD AT 1.1VP-P

2-1N914

3

2

FIGURE 13. SINGLE SUPPLY WEIN BRIDGE OSCILLATOR

8

CA3160

+

-6

1M

2

6

3

7

13

3 10

7 4 9

8 5

1

214 11

12

0.01F

1F

680k

2k

INPUT

1F

20k

CA3600

A = 20dBLARGE SIGNALBW (-3dB) = 190kHz

QN1 QN2 QN3

QP1 QP2 QP3

4

+15V

500F

50100mWAT 10%THD

- +

FIGURE 14. CMOS TRANSISTOR ARRAY (CA3600E) CONNECTED AS POWER BOOSTER IN THE OUTPUT STAGE OF THE CA3160

FN976 Rev 5.00 of 18June 2004

CA3160

Typical Performance Curves

FIGURE 15. OPEN LOOP VOLTAGE GAIN AND PHASE SHIFT vs FREQUENCY

FIGURE 16. OPEN LOOP GAIN vs TEMPERATURE

FIGURE 17. VOLTAGE TRANSFER CHARACTERISTICS OF CMOS OUTPUT STAGE

FIGURE 18. QUIESCENT SUPPLY CURRENT vs SUPPLY VOLTAGE

FIGURE 19. QUIESCENT SUPPLY CURRENT vs SUPPLYVOLTAGE

FIGURE 20. VOLTAGE ACROSS PMOS OUTPUT TRANSISTOR (Q8) vs LOAD CURRENT

120

100

80

60

40

20

0

OP

EN

LO

OP

VO

LTA

GE

GA

IN (

dB

)

0

50

100

150

200

OP

EN

LO

OP

PH

AS

E (

DE

GR

EE

S)

101 102 103 104 105 106 107 108

FREQUENCY (Hz)

VS = ±7.5V

TA = 25oC

CL = 30pF

RL = 2k

OL

150

140

130

120

110

100

90

80-100 -50 0 50 100

OP

EN

LO

OP

VO

LTA

GE

GA

IN (

dB

)

TEMPERATURE (oC)

RL = 2k

17.5

15

12.5

10

7.5

5

2.5

OU

TP

UT

VO

LTA

GE

[T

ER

MS

. 4

AN

D 6

] (V

)

0 2.5 5 7.5 10 12.5 15 17.5 20 22.5

GATE VOLTAGE [TERMINALS 4 AND 8] (V)

V+ = 15V, V- = 0VTA = 25oC

RL = 5k

2k1k500

0

15.0

12.5

10.0

7.5

5.0

2.5

0

QU

IES

CE

NT

SU

PP

LY C

UR

RE

NT

(m

A)

6 8 10 12 14 16 18

POSITIVE SUPPLY VOLTAGE (V)

TA = 25oCRL = V- = 0

HIGH VO = V+OR LOW VO = V-

BALANCEDVO = V+/2

14

12

10

8

6

4

2

0

QU

IES

CE

NT

SU

PP

LY C

UR

RE

NT

(m

A)

0 2 4 6 8 10 12 14 16 18POSITIVE SUPPLY VOLTAGE (V)

VO = V+ / 2V- = 0

TA = -55oC

25oC

125oC

50

10

1

0.1

0.01

0.001

VO

LTA

GE

DR

OP

AC

RO

SS

PM

OS

OU

TP

UT

STA

GE

TR

AN

SIS

TO

R (

Q8)

(V

)

0.001 0.01 0.1 1 10 100MAGNITUDE OF LOAD CURRENT (mA)

V- = 0VTA = 25oC

V+ = 15V10V

5V

FN976 Rev 5.00 of 18June 2004

CA3160

FIGURE 21. VOLTAGE ACROSS NMOS OUTPUT TRANSISTOR (Q12) vs LOAD CURRENT

FIGURE 22. EQUIVALENT NOISE VOLTAGE vs FREQUENCY

FIGURE 23. INPUT CURRENT vs COMMON MODE VOLTAGE FIGURE 24. INPUT CURRENT vs TEMPERATURE

FIGURE 25. TYPICAL INCREMENTAL OFFSET VOLTAGE SHIFT vs OPERATING LIFE

Typical Performance Curves (Continued)

V- = 0VTA = 25oC V+ = 15V

10V5V

50

10

1

0.1

0.01

0.001

VO

LTA

GE

DR

OP

AC

RO

SS

NM

OS

OU

TP

UT

STA

GE

TR

AN

SIS

TO

R (

Q1

2) (

V)

0.001 0.01 0.1 1 10 100

MAGNITUDE OF LOAD CURRENT (mA)

1000

100

10

EN

(n

V

Hz)

1 101 102 103 104 105

FREQUENCY (Hz)

TA = 25oCVS = 7.5V

1

10

7.5

5

2.5

0-1 0 1 2 3 4 5 6 7

INPUT CURRENT (pA)

INP

UT

VO

LTA

GE

(V

)

TA = 25oC

15VTO5V

V+

V-

0VTO

-10V

CA3160

2

3

6

7

84

PA

VIN

4000

1000

100

10

1-80 -60 -40 -20 0 20 40 60 80 100 120 140

VS = 7.5V

TEMPERATURE (oC)

INP

UT

CU

RR

EN

T (

pA

)

7

6

5

4

3

2

1

0

OF

FS

ET

VO

LTA

GE

SH

IFT

(m

V)

0 500 1000 1500 2000 2500 3000 3500 4000

TIME (HOURS)

TA = 125oC FORMETAL CAN PACKAGES

DIFFERENTIAL DC VOLTAGE(ACROSS TERMINALS 2 AND 3) = 0VOUTPUT VOLTAGE = V+ / 2

DIFFERENTIAL DC VOLTAGE(ACROSS TERMINALS 2 AND 3) = 2VOUTPUT STAGE TOGGLED

FN976 Rev 5.00 of 18June 2004

FN976 Rev 5.00 of 18June 2004

CA3160

Intersil products are manufactured, assembled and tested utilizing ISO9001 quality systems as notedin the quality certifications found at www.intersil.com/en/support/qualandreliability.html

Intersil products are sold by description only. Intersil may modify the circuit design and/or specifications of products at any time without notice, provided that such modification does not, in Intersil's sole judgment, affect the form, fit or function of the product. Accordingly, the reader is cautioned to verify that datasheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.

For information regarding Intersil Corporation and its products, see www.intersil.com

For additional products, see www.intersil.com/en/products.html

© Copyright Intersil Americas LLC 2002-2004. All Rights Reserved.All trademarks and registered trademarks are the property of their respective owners.

Dual-In-Line Plastic Packages (PDIP)

CL

E

eA

C

eB

eC

-B-

E1INDEX

1 2 3 N/2

N

AREA

SEATING

BASEPLANE

PLANE

-C-

D1

B1B

e

D

D1

AA2

L

A1

-A-

0.010 (0.25) C AM B S

NOTES:

1. Controlling Dimensions: INCH. In case of conflict between English and Metric dimensions, the inch dimensions control.

2. Dimensioning and tolerancing per ANSI Y14.5M-1982.

3. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication No. 95.

4. Dimensions A, A1 and L are measured with the package seated in JEDEC seating plane gauge GS-3.

5. D, D1, and E1 dimensions do not include mold flash or protru-sions. Mold flash or protrusions shall not exceed 0.010 inch(0.25mm).

6. E and are measured with the leads constrained to be per-pendicular to datum .

7. eB and eC are measured at the lead tips with the leads uncon-strained. eC must be zero or greater.

8. B1 maximum dimensions do not include dambar protrusions. Dambar protrusions shall not exceed 0.010 inch (0.25mm).

9. N is the maximum number of terminal positions.

10. Corner leads (1, N, N/2 and N/2 + 1) for E8.3, E16.3, E18.3, E28.3, E42.6 will have a B1 dimension of 0.030 - 0.045 inch(0.76 - 1.14mm).

eA-C-

E8.3 (JEDEC MS-001-BA ISSUE D)8 LEAD DUAL-IN-LINE PLASTIC PACKAGE

SYMBOL

INCHES MILLIMETERS

NOTESMIN MAX MIN MAX

A - 0.210 - 5.33 4

A1 0.015 - 0.39 - 4

A2 0.115 0.195 2.93 4.95 -

B 0.014 0.022 0.356 0.558 -

B1 0.045 0.070 1.15 1.77 8, 10

C 0.008 0.014 0.204 0.355 -

D 0.355 0.400 9.01 10.16 5

D1 0.005 - 0.13 - 5

E 0.300 0.325 7.62 8.25 6

E1 0.240 0.280 6.10 7.11 5

e 0.100 BSC 2.54 BSC -

eA 0.300 BSC 7.62 BSC 6

eB - 0.430 - 10.92 7

L 0.115 0.150 2.93 3.81 4

N 8 8 9

Rev. 0 12/93

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ca3160-a - ca3160, ca3160a - 4mhz, bimos operational amplifier

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