barton dissertation pdfthe integration of mach-zehnder modulators with sampled grating dbr lasers

7/30/2019 Barton Dissertation PDFThe Integration of Mach-Zehnder Modulators with Sampled Grating DBR Lasers

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3.6 TEMPERATURE INDUCED BANDGAP SHRINKAGE

The bandgap of the material will shrink with increases in temperature. This has

been expressed with the Varshni equations for unstrained materials:

( )T T

K E T E g g +

-=

b a

)0()( [3.21]

where Alpha is 4.9E-4 eV/K Beta = 327K.

The change in bandgap energy with temperature has also been extrapolated

for binary data at 300K for lattice matched quaternary material and expressed

as[350]:

)61.041.018.3(101 24 y y xdT

dE g +--= - [3.22]

The change in bandgap for the tensile strained modulator structure was

measured using a micro-photoluminescence setup as a function of temperature

as shown in fig 3.9.

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=0.432meV/K

Fig 3-10 Temperature dependence of the waveguide composition emission wavelength

The slope of the waveguide composition vs temperature is shown in fig. 3-10.

This is a little higher than the slope shown in the literature[351] at 1.3Q of

0.333meV/K, probably due to the strain in the material. As can be seen in fig

3-10, the material is highly temperature sensitive and linear with respect to

temperature. As the modulator heats up with high optical powers, the bandgap

shrinks and the effective waveguide Q can change from 1.4Q to 1.435Q from

20-70C. Heat crosstalk is an important issue in integrated devices as the laser

benefits from low temperatures with higher gain and lower optical loss, and the

modulator benefits from the higher efficiencies at higher temperatures.

The rise in temperature with bias can be evaluated with the following model:

t d Z P T =D [3.23]

where P d is the power dissipated, and the thermal impedance (Z t) is given by:

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x p Lwh

Z t )/4ln(

= [3.24]

where h is the height of the substrate, w is the width of the device and L is the length. x is thethermal conductivity[350].

The thermal resistivity of InGaAsP has been given by [350] as:

242.3978.5947.11

y yT -+== x r K cm / W [3.25]

The refractive index of InGaAsP as a function of composition, and temperature

has been extended from a model given by Adachi, and fitted to experimental

data and given by:

-

++

D+

+= )300(1)()()(

)(

21

)()(

2/3

T T

y B z f T E

T E z f y An

oo

o g

g r

e

e [3.26]

where A(y) = 8.616 -3.886y [3.27]

B(y) = 6.621 + 3.461y [3.28]

2112)(

z z z z f

--+-= [3.29]

)(T E E

z g

= [3.30]

o g o T E

E z

D+=

)([3.31]

Although InGaAsP data is difficult to come by, for InP near 300K

o xT e

e 41016.5 - =

1/K [3.32]

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As one can see, the temperature dependence stems from the bandgap and the

high frequency dielectric constant terms.

3.7 ACCUMULATION OF EFFECTS

The accumulation of the aforementioned field effects and carrier effects are

plotted for a 300mm long device with three different input powers (4.9mW,

11mW, and 15.6mW) under reverse bias.

Fig. 3.11a Change in refractive index as a function of voltage with input optical power 4.9mW T= 16C l = 1555nm

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Fig. 3.11b Change in refractive index as a function of voltage with input optical power 11mWT = 16C, l = 1555nm

Fig. 3.11c Change in refractive index as a function of voltage with input optical power 15.6mWT = 16C l 1555nm

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The refractive index change was extracted from the absorption curves (fig. 3.6)

and the output power dc extinction curves (fig. 3.3) using equation 3.2 and

shown as DATA on each graph in fig. 3-11a-c. The total change is also plotted

for each case accounting for all of the effects which fit very well the observed

change in index.

A number of conclusions can be made from these plots. First of all, the

dominant effect is clearly the bandfilling effect or in this case bandemptyingdue to the n-doped waveguide. The plasma, linear and quadratic effects all

have fairly similar contributions given the doping profile that was used. The rf

change in index is the total change minus the heating portion as under RF

modulation, the device will not heat up much. This RF line appears to line up

well with the RF Vpi data observed in the next chapter in this case ~ 4V.

From the change in phase due to a change in index, one can determine the

modulator arm length required to achieve a pi phase shift.

n L D

=D

l p

f 2 [3-33]

Modulator Length to achieve pi shift.

||2 n L

D=

l p

[3-34]

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For a 300 mm modulator, the index change to achieve a pi shift for 1550 nm is

approximately is 0.26%.

As can be seen from the previous plots, the index and absorption are strongly

dependent on the optical power at DC as the modulator is heated and

experiences bandgap shrinkage. Under RF modulation, this efficiency is not

very power dependent. As the heating due to the photocurrent absorption

changes the refractive in the same direction as the other effects, at DC this

gives the appearance that the efficiency is better than it is at RF. Obviously, DC

extinction is not a very good indicator of RF performance.

3.8 HIGH SPEED DESIGN

Capacitance and carrier lifetime govern the maximum bandwidth possible for a

modulator. For a lumped modulator with an open termination port, the small-

signal modulation response is given by:

2

21 12

jwRC S

+= [3.35]

assuming that the microwave attenuation is low[5]. Typically, there are three

approaches to achieving high speed operation: low impedance matching 14,

reducing the capacitance and distributing the modulation region 15. The

14 See Chapter 4 Termination Section

15 with T sections as shown in Chapter 4

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capacitance can be decreased by either increasing the intrinsic region in the

waveguide16, lowering the pad capacitance with low k dielectrics, or decreasing

the waveguide area[314,320]17

. An accurate account of the capacitance in thestructure needs to take into account the junction capacitance [C j], parallel plate

capacitance [C pp] of the interconnect region and the fringing capacitance[C f ] for

the geometry as shown in the side-view of the modulator ridge in Fig 3-12.

Fig 3-12 Modulator end-view with different contributions of capacitance

16 Reducing the modulator efficiency and reducing the optical loss

17 Potentially increasing optical loss and or drive voltage

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Next we will look at the minimization of these capacitances separately. In

both lumped and traveling wave devices, one would like to reduce the

capacitance per unit length.

3.5 JUNCTION CAPACITANCE MINIMIZATION

The PN junction capacitance is minimized by using a short device with a

narrow ridge. As shown in Fig. 3-13, the junction capacitance improves for

wider intrinsic region widths and lower doping levels. The material exhibits less

free carrier absorption with low doping particularly Zn. Structures with large

intrinsic regions do not provide high electric fields so

Fig 3-13 Capacitance per unit length [pF/m] for various doping structures - 2 mm ridge

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clearly there is a tradeoff between capacitance as improved with a PIN

structure and efficiency with a PN junction. Since the devices here use a low-

doped PN junction18

, the bandwidth varies considerably with bias as seen inthe variance in fig 3-13 of the capacitance with bias. As an illustration of this,

the small-signal modulation response is shown for a 200 m long lumped MZ at

various biases in fig. 3-14.

Fig. 3-14 Small-Signal normalized modulation response at 1555nm for a 200um long electrode

device.

18 3e17 Si as in Fig 3-9b

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The waveguide is depleted out as the bias induced electric field increases in

the waveguide changing the capacitance and bandwidth as shown in fig 3-

14.

In this work, the ridges were tapered down to 2.5 m (effectively 2.2 m) in the

modulator regions. This did not seem to adversely affect the insertion loss of

the modulators much as was shown in Chap. 1. Below 2m wide, one would

expect propagation losses to increase markedly due to light scattering.

3.6 PARASITIC CAPACITANCE MINIMIZATION

There are a number of different innovative materials that can be used for

providing a low-dielectric constant dielectric layer in the modulator section as

shown in table 3-1.

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Table 3-1 Dielectric MaterialsMaterial Dielectric ConstantNanoporous silica

1.3 2.8Fluorinated organic polymers 1.8 3.0Fluorinated amorphous carbon 2.1 2.3Non-fluorinated organic polymers 2.5 3.5Cyclotene Benzocyclobutene (BCB) 2.65SILK (Dow) 2.65Non-fluorinated polymers 2.7 3.5Inorganic polymers 2.7 3.5Phase separated hybrids 2.8 3.0

Poly-imides 3.2 3.4Fluorinated HDPCVD SiO2

Fluorinated PECVD SiO2 3.5

Thermal SiO2 3.9Plasma deposited SiO 2 4.2Thermal silicon nitride Si3N4 7.9Plasma silicon nitride Si-N-H 7.0 9.0

Application techniques, vary from LPCVD, PECVD, sputtering, to deposition of

low-K liquids by simple spin coating and multiple baking techniques, similar to

photoresist processing. These materials are helpful for a number of reasons.

First of all, the parasitic capacitance in the modulator is reduced due to the low

dielectric constant which is important for high speed. Also, the dielectrics are

useful for planarization over rough topographies on InP wafers particularly

with n-topside contacts.

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Although there are a number of low-k electronic material candidates for

electronic device designs such as oxide-based materials that can handle

temperatures as high as 600C, Cyclotene BCB was chosen for fabrication asthe dielectric material has not only a low dielectric constant (2.65) but is

easily cleaved and easily applied.

Fig 3-15a PhotoBCB planarized ridges Fig. 3-15b Dry-etchable BCB

Although dry-etchable BCB tends to have superior planarity over

photodefinable varieties (see fig 3-15ab) the latter choice avoids excessive

overetches of the BCB that are necessary in order to remove BCB residuals

fully from the surface as shown in Fig. 3-16. The shelf life of Photo-BCB is not

very long however at room temperature 19, so freezing it is a necessity.

19 approximately 1 week

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BCB scum

voids

Fig 3-16. BCB residuals after etch

The reactive ion etcher (RIE) tends to leave a BCB residue scum on the

surface with the etch conditions that were used consisting of 20% CF 4/ 80% O2

with either 250V (W) or 350V (W) conditions as recommended by Dow.

Going to a lower CF4 percentage gives better selectivity between the BCB and

Silicon oxy-nitride layers however is more susceptible to oxide scum and the

etch rate decreases dramatically.

BCB

Fig 3-17 Cyclotene 4024 PhotoBCB defined in only the modulator regions.

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Due to adhesion problems and device heat dissipation issues BCB was

defined to be only under the modulator section pads. This was defined using a

photolithographic stepper tool and the rest developed off using a puddleemersion developer DS-2100 avoiding a 5 m BCB etch. It was found that

adhesion of the pads during wirebonding was not acceptable on the first device

run due to the BCB being etched under the pads which had excessive

roughness and resulted in delamination during wedgebonding. Using a

different approach only etching a via to the ridge and leaving the area under

the pad unetched with a sandwiching layer of SiN yOx on top of the BCB proved

superior not only as a thicker dielectric leaving lower parasitic capacitance

but very good adhesion for wirebonding. See process Appendix C. Photo-BCB

does not have very good definition resolution as can be seen in fig. 17-a with

very sloped sidewalls, however it is sufficient for this application.

3.7 FRINGING CAPACITANCE

Using the basic geometry given in Fig. 3-12, one can calculate the parallel-

plate capacitance C pp of the interconnect segment. However, in interconnect

lines where the wire thickness (t) is comparable in magnitude to the ground-

plane distance (h), fringing electric fields significantly increase the total parasitic

capacitance (fig. 3-1). It has been shown [315] that the influence of fringing

fields increases with the decreasing (w/h) ratio, and that the fringing-field

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capacitance can be as much as 10-20 times larger than the parallel-plate

capacitance. It was mentioned earlier that the sub-micron fabrication

technologies allow the width of the metal lines to be decreased somewhat, yetthe thickness of the line must be preserved in order to ensure structural

integrity. This situation, which involves narrow metal lines with a considerable

vertical thickness, is especially vulnerable to fringing field effects.

A set of simple formulas [315] can be used to estimate the capacitance of the

interconnect structures in which fringing fields complicate the parasitic

capacitance calculation. The following two cases are considered for two

different ranges of line width (w).

+++

+

-

=

2222

1ln

22

t h

t h

t hh

t w

C p

e for 2t

w [3.36]

+

+++

-+= 47.1

2222

1ln

20543.0

1(

t h

t h

t h

ht

hw

C p

e for 2t

w < [3.37]

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where t, h and w are the dimensions as shown in Fig 3-12. These formulas

permit the accurate approximation of the parasitic capacitance values to within

10% error, even for very small values of (t/h).

The other contribution of capacitance is attributed to the parasitic capacitance

of the contact pad. This contribution was measured in Fig 4-3b to be

approximately 0.2pF. Figure 3-18 shows the parasitic capacitance as a

function of dielectric thickness for different dielectrics and modulator lengths.

2BCB 100um deviceBCB 200um deviceBCB 300um deviceSiNx 100um deviceSiNx 200um deviceSiNx 300um device

P ar a

s i t i c P

a d C a p a c i t an c e ( pF

)

1.8

1.6

1.4

1.2

1

0.8

0.6

0.4

0.2

01 2 3 40.5 1.5 2.5 3.5

Dielectric Thickness (m)

Fig. 3-18. Pad Capacitance for different dielectrics and pad sizes w/fringing fields

3.8 MULTI-MODE INTERFERENCE DESIGN

Another very important element to the Mach-Zehnder design is that of the

MMI splitters and combiners[326-331]. General MMI theory states that the

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shortest 1x2 splitter requires a length of 3/8L pi where the beat length of the two

lowest order modes is given by[328]:

o

eff eff wn Ll

p

34 2

= [3-38]

where neff is the effective index of the mode, l is the wavelength, and w eff is the equivalent widthof the MMI

Fig. 3-19a Electric Field profile for the optimized MMI design showing imaging into the two MZbranches (Waveguide 1.4Q @ 1550nm)Fig. 3-19b MMI with curved waveguides (Height = 9um Length = 85um, taper = 20um)

Using Beamprop, an MMI design was optimized with a center wavelength of

1550nm as shown in fig 3-19ab. MMIs have broad optical bandwidth20[328]

much wider than the tuning range of the SGDBRs here (38nm). The length of

the MMI becomes very long for wide widths due to the quadratic dependence

so it is imperative to minimize the width as much as possible. A 9 m wide

MMI was chosen to that the gap between the waveguides could be resolved

with the stepper as shown in Fig. 3-20. Note also the high angle sidewall in

110

20 close to 100nm for 1dB bandwidth


127/237

this gap due to the crystal orientation during the ridge wet etch. This sidewall is

not likely to adversely affect reflections in the device in fact it gives a more

gradual index discontinuity.

Fig 3-20 Gap between waveguides approximately 1m

Curved waveguides were used to extend the separation distance to 16m asshown in Fig 3-21 to minimize the propagation distance. The ridge was defined

using a dry etch/wet etch process where approximately 1m of material is RIE

etched with Methane/Hydrogen/Oxygen with a subsequent 3:1 H 3PO4:HCl wet-

etch to remove the rest of the InP on top of the waveguide. As the radius of

curvature is low, the sidewall roughness appears to be low as shown in fig. 3-

16.

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Fig. 3-21 Sidewall roughness on curved waveguides and MMI taper

3.9 PHASE SHIFTER

A phase shifter electrode was integrated in one branch of the MZ in order to

facilitate changing the phase for different wavelengths. It is best to design the

waveguide structure to achieve a pi-phase shift without bias. Pi-shifted

modulators have been fabricated with one length a multiple of 0.241m 21

longer than the other. The devices in this dissertation utilize a pi-shiftedconfiguration however this is accomplished using one ridge slightly wider

(0.2 m) in the curved waveguide regions than the other to achieve the pi

shift22[300]. Unlike the RF sections, the phase section can be forward biased,

which gives close to 5x the index shift as reverse bias as illustrated in fig. 3-

22.

112

21 for 1550nm

22 This is easier due to fabrication tolerances


129/237

Fig. 3-22 Fiber-coupled power for device #1 as a function of bias on phase section in bothforward and reverse bias

As this device needs to operate over the full C-Band in which the pi-shift will

change with wavelength, designs allowed for the use of a forward biased

electrode to achieve the pi-phase shift. This requires fairly good control over

waveguide widths/thicknesses/compositions in order to achieve from run-to-

run. By biasing this electrode however, it induces a significant amount of loss

in that waveguide as shown in fig. 3-17. Ideally the device is forward biased

slightly as very little current is required ~2mA to achieve the desired phase.

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Fig. 3-23 Normalized Optical Loss vs. wavelength and bias for 100 m long phase electrode

The loss was measured with Device #1 23 where the laser sections are forward

biased, and the SOA is reverse biased to measure the optical power that

makes it through the phase section as a function of bias on the phase

electrode.

3.10 1ST GENERATION DESIGN

The initial design involved the integration of a SGDBR with a passive Mach-

Zehnder modulator as demonstrated in fig 3-24.

23 see Generation 2 designs 3.10

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Fig 3-24 Integrated SGDBR- Mach Zehnder modulator

One branch of the MZ modulator was meant for DC biasing to change the

phase for each wavelength, and the second for RF modulation (Pad #2).

Devices were fabricated with parameters as shown in Table 3-2. These

modulators uses two identical 3dB MMI splitters/combiners that are 98m long

as described in section 3.8.

Table 3-2 1st Generation DevicesMach Zehnder Lengths 550,750,950Waveguide offset 40umWidth1 2um

Width2 2.1 to 2.2 to achieve pi shiftCurve length 185Curve width 20umTrench 15umMMI length 98umMMI width 9um

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Although these devices were fairly long and suffered from high capacitance

due to the problem outlined in fig. 2-7b, the DC extinction and chirp24

characteristics looked promising as shown in fig. 3-25.

-35

-30

-25

-20

-15

-10

-5

-5 -4 -3 -2 -1 0

+0.8Vbias+0.6V+0.3V0V Bias-1V Bias-2V Bias-3V

O u t p u

t P o w e r

( d B m

)

Arm #1 DC Bias Voltage

Fig 3-25 550m long electrode at = 1535nm

3.10 2 ND GENERATION DESIGNS

A number of different improvements were made to the 2 nd generation devices

to improve performance. First, SOAs were integrated before the MZ and inside

the MZ modulator to mitigate the 4-5dB insertion losses. Additionally, the gap

between the two waveguides was reduced from 37um to 16um which allowed

for shorter curved waveguide sections. A 2x2 MMI was placed at the output to

24 As will be shown in Chap 5

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guide away off-state light in a controllable way as shown in fig. 3-20. The

output was curved and flared as well as a front passive detector electrode

placed on the output waveguide to reduce reflections. Also, two RF electrodeswere placed on each device so that push-pull modulation could be possible. In

addition, considerably shorter electrodes were employed to improve the high

speed performance.

Output

Laser Input

1x2 splitter 2x2 combiner

Fig. 3-26 Ridge waveguide structure illustrating the 1x2 and 2x2 MMIs with curved waveguidesand output flares

The first three devices have Dual SOAs as mentioned in Chapter 1. Device 7

and 8 have electrodes at the rear of the modulator for rear resistive termination

as will be elaborated in chapter 4 and 5.

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The first 8 designs use lumped electrodes and are shown for reference:

Table 3-3 Lumped electrode MZ devicesTotal device length = 3200 m

#SOAConfig MZ Electrode Length SOA Length

1 Dual 300 350

2 Dual 250 350

3 Dual 200 350

4 Single 300 400

5 Single 250 400

6 Single 200 400

7 Single 300 400

8 Single 200 400

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[323] Soda H, Furutsu M, Sato K, Okazaki N, Yamazaki S, Nishimoto H,Ishikawa H. High-power and high-speed semi-insulating BH structuremonolithic electroabsorption modulator/DFB laser light source.Electronics Letters, vol.26, no.1, 4 Jan. 1990, pp.9-10.

[324] Jackel JL, Perlmutter P, Johnson J. High-speed low-voltage modulationwith a nonsymmetric Mach-Zehnder interferometer. Journal of LightwaveTechnology, vol.7, no.6, June 1989, pp.937-40.

[325] Rolland C, Mak G, Prosyk KL, Maritan CM, Puetz N. High speed and lowloss, bulk electroabsorption waveguide modulators at 1.3 mu m. IEEE

Photonics Technology Letters, vol.3, no.10, Oct. 1991, pp.894-6.[326] Pennings ECM, van Roijen R, van Stralen MJN, de Waard PJ, Koumans

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[327] Soldano LB, Pennings ECM. Optical multi-mode interference devicesbased on self-imaging: principles and applications. Journal of LightwaveTechnology, vol.13, no.4, April 1995, pp.615-27.

[328] Besse PA, Bachmann M, Melchior H, Soldano LB, Smit MK. Opticalbandwidth and fabrication tolerances of multimode interference couplers.Journal of Lightwave Technology, vol.12, no.6, June 1994, pp.1004-9.

[329] Erasme D, Spiekman LH, Herben CGP, Smit MK, Groen FH.Experimental assessment of the reflection of passive multimodeinterference couplers. IEEE Photonics Technology Letters, vol.9, no.12,Dec. 1997, pp.1604-6.

[330] Leuthold J, Joyner CW. Multimode interference couplers with tunablepower splitting ratios. Journal of Lightwave Technology, vol.19, no.5,

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strongly guiding semiconductor rib waveguide Y-junctions. IEEEPhotonics Technology Letters, vol.2, no.6, June 1990, pp.404-6.

[332] Cartledge JC, Rolland C, Lemerle S, Solheim A. Theoreticalperformance of 10 Gb/s lightwave systems using a III-V semiconductor Mach-Zehnder modulator. IEEE Photonics Technology Letters, vol.6,no.2, Feb. 1994, pp.282-4.

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[345] Adachi S, Oe K. Linear electro-optic effects in zincblende-typesemiconductors: key properties of InGaAsP relevant to device design.Journal of Applied Physics, vol.56, no.1, 1 July 1984, pp.74-80.

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C h a p t e r 4

Series Push-Pull Modulator Designs As first demonstrated by Walker[7,455] and later Spickermann[456] in the

GaAs/AlGaAs material system, if a RF signal is applied across the two MZ

electrodes thereby connecting the diodes in series one achieves superior

performance in terms of optical bandwidth and zero-chirp performance with a

single RF input. This chapter will first look at the design concept and resultsfrom lumped device designs with input side termination. Next, end-terminated

CPS transmission line electrode designs that aim to match the characteristic

impedance are explored with respect to transmission line design and device

characteristics.

MZ#1 p-contact MZ#2 p-contactSiNxOy

BCBn-contact

Vdc2 RVdc1 MZ #1

InGaAs contact layer RF SIGNALMZ #2

SI InP substrate

Figure 4-1 Series push-pull bias configuration

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4.1 LUMPED SERIES PUSH-PULL BANDWIDTH

The series push-pull biasing scheme and cross-section of the modulator are

shown in fig. 4-1. The use of a SI substrate lowers the parasitic capacitance

under the electrode pads and enables series push-pull (SPP) operation.

Ideally, there would be full isolation between the n-contact region in the

modulator and that of the rest of the chip. In the devices presented here, the n-

InP region was etched down to the SI-substrate close to the ridge and the p-

InP region was proton implanted. This leaves a narrow region below theshallow ridge that is not effectively isolated. The resistance between the n-

contact to the modulator and that of the rest of the transmitter typically

measured approximately 150 ohms.

Lumped series push-pull devices were fabricated and tested using special

75mm pitch CPS picoprobes with integrated 50 ohm parallel resistors. In this

case, the 50 ohm termination is on the front end of the device. Later

transmission line based devices use a rear termination. The 3dB optical

bandwidth of three different devices with 200mm, 250mm and 300 mm electrode

lengths are shown in fig. 4-2 and compared to the one-sided modulation. The

probe configuration is shown in fig. 4-3 for the single side and SPPconfigurations.

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GND

GND

SIGNAL

SIGNAL

Fig. 4-2 Biasing for single side and series push pull

Note that using the series push-pull electrode structure almost doubles the

optical bandwidth the loss due to parasitics to ground on one branch of the

MZ which equates to approximately the capacitance from the n-contact to the

backside of the substrate (100 mm thick). The substrate is metalized on the

backside to facilitate soldering to a carrier for good heat conduction.

Alternatively the device could be either flip-chipped without backside

metalization or epoxied to the carrier thereby removing the metallization and

reducing the parasitic capacitance at the expense of reduced thermal

conductivity. Clearly a smaller n-contact region would be beneficial.

One can see in fig 4-3b the capacitance per unit length (1225pF/m and

690pF/m for single side and SPP respectively) as the slope and pad

capacitance as the y-intercept for both single side and SPP modulation.

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RC f dB

p

13

=

Fig. 4-3a Comparison of single side and series push-pull 3dB optical small signal bandwidthFig 4-3b Capacitance vs electrode length for single side and SPP configuration DC -3V Bias

Due to the photocurrent generated in the devices the impedance is reduced

considerably. This means that with a 50ohm parallel resistor, the single side

configuration has approximately 45.5 ohms and the SPP configuration yields

47.6 ohms. The effect of added bandwidth is also evident in the back-to-back

eye diagrams for the comparison of single side to SPP modulation as shown

for a 250mm long electrode device at 10 Gbit/s with a 2 7-1 PRBS in fig. 4-4.

250mm single side lumped 250 mm SPP

Fig. 4-4 Back to back Eyes comparing single side and series push-pull operation with 10dBextinction. Both at using -2V DC bias with 1.5V Vpp. OC-192 with 27-1 PRBS

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4.2 DUAL RF SERIES PUSH-PULL DEVICES

As mentioned before, the Dual RF series push-pull devices take advantage of

the improved bandwidth of the SPP electrode structure and reduced voltage by

having two of them. Figure 4-5 shows the device layout and the parasitic

conduction path between the two n-contact regions. Ideally this path would be

cut or reduced by He implanting in between the sets of electrodes. In the

current layer structure this would be difficult as the n-InP and n-InGaAs layers

are approximately 2.3 m thick as well as the ridge on top (2 m) which isdifficult to achieve without very high implant energies. To do this a quaternary

contact would need to be placed much closer to the waveguide ~ 0.5 m.

GND DATA

GND

N-contact DATAN-contact

Fig 4-5 Dual RF Series push-pull 4 electrode structure

Due to the finite conductivity of the n-layer, the conduction path prefers the

closest GND and even without He implantation, the device operates well at

10Gbit/s.

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Fig 4-6 device layout for Dual RF SPP electrode devices

As can be seen in fig. 4-7, the swing improves considerably with Dual RF

sources. The bandwidth is compromised a bit due to the lack of isolation

between the two n-contacts however not excessively as illustrated in the

back-to-back eyes for single SPP and Dual SPP as shown in fig. 4-7.

Both SPPRisetime:72psFalltime:56ps

One SPPRisetime:65psFalltime:52ps

Fig. 4-7 Optical signal levels for single and dual SPP operation and back-to-back eyediagrams for each with Vpp = 2V with 10Gbit/s PRBS 27-1 signal

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4.3 TRAVELING WAVE MODULATORS

Numerous groups have demonstrated discrete high-speed modulators and

DFB integrated devices utilizing traveling-wave electrode structures [4,7].

Although lumped electrodes can provide fairly good performance with respect

to bandwidth, careful design of the transmission line will provide superior return

loss (S 11) if the loaded transmission line is designed to match the driver [25 or

50 ohms] and/or superior bandwidth if the microwave index is matched. An

assortment of different transmission line structures have been pursued for traveling wave electrodes. Microstrip, Coplanar waveguide (CPW), and

Coplanar strip (CPS) transmission lines are most often employed. Microstrip,

although simple is sometimes regarded as disadvantageous due to

inaccessible ground planes, difficulties in shunt connections between the strip

and ground, limitations on the substrate thickness and exhibit more radiation

with thick substrates. In the case of CPW lines the impedance is mostly

defined by the lateral dimensions and the substrate thickness is not as

important. CPW localizes the electric field reducing spurious coupling,

radiation and dispersion[402].

Unfortunately, both the even and odd modes can exist in CPW which this odd

mode can be suppressed with air bridges.[408] Additionally, parallel plate

modes are supported (microstrip modes) between the CPW and the ground

plane on the bottom which is a cause of energy leakage from the CPW

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[408]. As a general rule the thickness of the substrate must be > 2(2G + W) in

order to suppress the microstrip modes. In order to match the characteristic

impedance of the source, the on-chip loaded transmission lines require fairlylarge unloaded characteristic impedances. Although CPW can easily be made

to match 50ohms, capacitively loaded lines require much larger unloaded

characteristic impedances to yield 50 ohms loaded and these high values

cannot be realized in CPW easily with the current doping restraints of the

integration platform. CPW designed for index matching yields poor

characteristic impedance matching. However, Coplanar Stripline (CPS)

which has a range of possible Z o values twice that of CPW works well for the

matching region. CPS was chosen for this reason, and the compactness of the

transmission lines suitable for further integration such as in a photocurrent-

driven wavelength converter.

However, it is more difficult to make a 50 ohm unloaded section (for the

feedthroughs) without very narrow gaps and wide pads leading to higher

microwave attenuation. The feedthroughs were designed at a linear taper as

this was found to be the best approach in [462]. Also, the phase difference

between the two lines will affect the matching ability at higher frequencies.

Ideally the lines should be excited with equal length feedthrough lines. The

design was chosen to have an input line at 30 degrees to allow probing away

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from the optical waveguide but minimize the phase difference between the

two electrodes.

4.4 TRAVELING WAVE MATCHING

The design of Traveling-Wave (TW) modulators is based on the matching of

the optical and electrical wave velocities. As has been pointed out[454], it is

the group index that should be matched not the phase velocity. In the case

of LiNbO3 modulators, the electrical wave (n eff

4.225) propagates slower thanthe optical wave(neff 2.138)[303,421,424]. To perform matching, one can use

a buffer layer, phase reversal, or a shielding plane to decrease the microwave

effective index of the line[422,423]. Alternatively, one can increase the

electrode thickness, decreasing the effective index further [421]. GaAs and

InP, modulators can have electrical waves that propagate faster than the

optical wave. In order to match the index, often either capacitive coupling or

inductive coupling approaches are employed. According to work done by

Spickermann et al.[461], the inductively coupled slow wave structures have

higher attenuation loss for a given gap width and are harder to model than

capacitively-coupled devices. LiNbO3 devices do not have a PN structure and

do not load the line substantially similarly to devices such as demonstrated by

Spickermann that rely on the electric field between the electrodes to change

the index which usually is far less efficient than the use of a PN structure.

The devices in this dissertation use PN junctions to increase the electric field

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overlap with the optical mode which leads to a very large capacitance per unit

length resulting in a similar situation as the LiNbO3 where the microwave

index is much higher than the optical index.

For a capacitively loaded transmission line, the optimum loading capacitance is

given by Walker:

optoo

cpwoptoloading ncZ

nnC

22-

=

[4.1]

where c is the speed of light, n opto is the optical group index, n cpw is the electrical index and Z o isthe characteristic impedance

However, in order to fabricate high performance SGDBRs, the doping required

typically results in capacitance per unit lengths in the range of 2000pF/m to

2500pF/m for a 3mm wide ridge. The junction capacitance/length of the device

due to the PN or PIN region is considerably larger (x10) than the

capacitance/length of the coplanar line. The result of this is the line is highly

capacitively coupled which both slows the electrical wave and reduces the

characteristic impedance considerably.

First, the optical group index of the modulator section needs to be assessed.

The effective index and group index are shown in fig. 4.8 for various waveguide

compositions.

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3.3.

3E f f e

c t i v eI n d ex

.35

325

1.3Q1.35Q1.4Q1.45Q

3.2 1520 1530 1540 1550 1560 1570Wavelength (nm)

1580

3.43.3.44.4. G

r o u pI n

d ex

24

68

1.3Q1.35Q1.4Q1.45Q

1520 1530 1540 1550 1560 1570 1580

Wavelength (nm)

Fig. 4-8 Effective Index and Group index for different waveguide compositions. Assuming astructure where the waveguide has been etched off halfway.

One can see that not only is the group index significantly higher for waveguide

compositions at approx. 1.45 but the dispersion increases as the operating

wavelength approaches that of the band-edge. One will obtain a superior

velocity match at the lower wavelengths and higher waveguide Q as loaded

transmission lines tend to slow the electrical wave excessively. Matching over

a wide wavelength range becomes more difficult as the waveguide composition

Q increases.

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Next we should consider the group index of the microwave signal. The

electrical signal does not have as much dispersion as the optical signal and is

often approximated as just the phase velocity. The dispersion has been curvefitted from spectral domain data and is given by [415].

)1()()(

1

b

qr qeff eff aF

f f n -+

-

+==

e e

e e

[4.2]

14 11 -=

r

TE h

c f e

[4.3]

where: f = frequency; F = f/f TE

the cutoff frequency for the lowest-order TE mode

)log(

10v

W S

u

a+

= [4.4]

u0.54 0.64q + 0.015q 2 v0.43 0.86q + 0.54q 2 q = log(S/h1)h1 thickness of substrateb = 1.8e q = effective permittivity at the quasi-static limit.

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4.5 TRANSMISSION LINE MODEL

The CPS transmission line in these series push-pull devices can be modeled

as a distributed circuit model along the device as shown in fig. 4-9. Often, the

characteristic impedance of a transmission line is approximated for low

microwave loss in equation 4.5.

cps

cpslosslesso

C

L

Y

Z Z = [4.5]

cpscpslossless C Lc ZY cn =m [4.6]

However, the devices here experience microwave losses due to a number of

sources as outlined in section 4.7 and the transmission line model fits the data

best if the capacitive and inductive loading are accounted for in the model. The

transmission lines in the device are loaded by the depletion capacitance from

each ridge C PN1 and CPN2 in the ridge as is shown in fig. 4-9 as well as a small

amount of inductance due to the T structures. The capacitance shown in fig. 4-

9 is composed of the PN junction capacitance (significant), the CPS

metallization capacitance and the parasitic capacitance.

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Ccps

CPN G CPN

Fig. 4-9 Device cross-section equivalent circuit for smooth CPS

This can be expressed as a distributed circuit model as shown in fig. 4-10.

RZLcps LT

CPN GPN

Y Gn CcpsCPara GPN2 CPN2

LT

Transmission line Equivalent circuit for T-electrode CPS line

Fig. 4-10 Transmission line distributed equivalent circuit. Gn is conductance in n-claddingregion, Cpara is parasitic capacitance to ground, L T is the T-inductance, G PN is the conductancedue to the photocurrent in the ridge C pn is the depletion capacitance

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The equivalent circuit model for the smooth CPS line devices is the same as

given in fig 4-10 except without the inductance contribution of the Ts (LT = 0).

Given the equivalent circuit model in fig. 4-10, the characteristic impedance of the smooth CPS transmission line can be expressed as:

cps L j R Z w += [4.7]

+++

+=

p PN PN

n

ncps smooth

C C C jG

GC jY

21

111 w

w [4.8]

+

+++

+==

p PN PN

n

ncps

cps

smoothosmooth

C C C jG

GC j

L j R

Y Z

Z

21

111

)(

w

w

w

[4.9]

The T structures have some additional inductance as shown in the equivalent

circuit in fig. 4.10

( ) p PN PN PN PN T ncpsT T

C C jGC jG L j

G

C jY

+++

+++

+=

21

112

11

w w w

w [4.10]

( ) p PN PN PN PN T ncpsT

cpsT

T oT

C C jGC jG L j

G

C j

L j R

Y

Z Z

+++

+++

+

+==

21

112

11

)(

w w w

w

w [4.11]

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4.6 CHARACTERISTIC IMPEDANCE COMPARISON

The CPS lines used in this dissertation were modeled using ADS software. As

the lines are considerably capacitively loaded, this means we need to design a

transmission line that has a much larger impedance unloaded in order to

obtain 50ohms loaded. Figure 4.11 shows the unloaded characteristic

impedance for two CPS structures, one with 50 mm Ts and one with a smooth

CPS line 16mm apart with 8mm wide strips.

4.11 Unloaded Characteristic Impedance for smooth CPS and 50 mm T structures from devicesas shown in table 4.2. 1.5 mm thick Au

As can be seen in fig. 4-12a, narrow lines increase the characteristic

impedance by increasing the inductance at the expense of microwave loss.

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A much higher characteristic impedance is possible with the Ts as shown in fig

4-12b for a given electrode width. The width of the T electrodes was chosen at

8 m as a compromise with microwave loss shown in fig 4-12b as design G.

25

30

35

40

45

50

0 5 10 15 20 25 30 35 40

2468101215

C h a r a c t e r

i s t i c I m p e

d a n c e

Frequency (GHz)

16um spacing 1000pF/m loading

38

40

42

44

46

48

50

52

0 5 10 15 20 25 30 35 40

Design G 5umDesignG 8umDesignG 15um

L o a d e d

C h a r a c

t e r i s

t i c I m p e

d a n c e

Frequency (GHz)

Fig 4-12a Characteristic Impedance for different CPS line widths given a 16um spacingFig 4-12b Characteristic impedance for T-section electrodes vs electrode width

From S parameters and the resulting [ABCD] matrix, the characteristic

impedance was extracted for different biases for device #9. After testing the

characteristic impedance of the different devices it was clear that they dont fit

the characteristics shown at low frequencies in fig 4-12b. After analyzing the

expected conductance in the structure, it was obvious that the n-epilayer

conductance for this structure is considerably higher than that of previousstructures done on lower doped or dielectric substrates.

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Based on Hall measurements:The conductivity of the n-layer between the ridges is given by

nq nn InP m s =)( = (1.6E-19 C)(1800cm2/Vs)(1E18 1/cm3) = 288 S/cm (0.032S/cm

in Spickermann)

where the Conductance is

)(2)( Length Area

nG InP InP s = = 1.469 S (compare with 0.01S in Spickermann)

Where the Length is 16 m; Area = (2.3m*314m) for Device #9

Data was taken comparing the characteristic impedance of T structures,

smooth CPS lines and lumped rear terminated electrode devices. The fit fromthe model shown in equations 4.9 and 4.11 are also shown assuming for the Ts

the capacitance per unit length of the transmission line is C T = 2.737e-11F/m,

Inductance per unit length is LT = 1.736e-6 H/m and for the smooth CPS C s =

4.602e-11F/m and L s = 7.3068e-7H/m with Rpn = 500ohms, R = 0.2 ohms, C pn

= 0.5pF, C para = 0.5pF, LT = 5e-12.

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Fig 4.13 Lower 4 lines extracted from Device #7 (single side 300mm long electrode)

Middle 4 lines extracted from Device #18 (Smooth CPS 500mm long electrode)Top 4 lines extracted from Device #7 (CPS Ts 250 mm long electrode)

As can be seen in fig 4-13, the characteristic impedance improves for higher

reverse biases on the electrodes where the depletion region is increased and

the capacitance/unit length decreases. Also, clearly one can see a large

benefit of using a T electrode over the smooth CPS lines in terms of better

characteristic impedance matching as it is much closer to 50 ohms.

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4.7 RF LOSS MECHANISMS

High frequency losses stem from three different mechanisms:

1. Dielectric losses2. Ohmic/conductor losses3. Radiation loss

These losses can be minimized using a number of approaches such as the use

of deep trenches between electrodes or thick dielectric layers below the

electrodes separating them from the substrate. By careful design of the

electrodes, minimization of longitudinal substrate currents may also reduce the

overall microwave attenuation[435]. Most work is done on Semi-insulating InP

and GaAs where the bulk of the electrical attenuation comes from the

conductor and radiation losses at least below 20GHz [450]. However, typical

SGDBR design is performed on n-InP conducting substrates with lossy InGaAs

contact layers. In this case, the dielectric losses are very high and the lines

become highly dispersive. Also, the capacitance between the two lines

increases dramatically - effectively loading the line and dropping the

characteristic impedance significantly. Dielectric loss is given by the following

relationship:

g eff

r D

ql e

d e a tan=

(Np/m) *27.3 for dB/lamda [4.12]

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For a doped semiconductor the loss tangent can be expressed as[450]:

)(")('2)(')("2

)(tan f f f f f f

f s e p

s e p d

+

+= [4.13]

where e and e are the real and imaginary parts of the complex dielectric

permittivity and s and s are the respective parts of the complex conductivity.

Taking the conductivity from the Drude model, we have s = s s/(1-j2pf t m) where

the conductivity can be extracted from Hall measurements.

The attenuation drops linearly with increasing metal thickness up to the point

where the metal depth is 3x skin depth. As the dimensions of the transmission

line increase, the attenuation decreases. There seems to be an optimum w/d

point of approximately 0.40 for InP with 0.25um of gold. If w = 80um that

corresponds to d = 177.8 [405]

In order to match the velocities of the electrical and optical waves, one can

manipulate a few parameters electrode thickness, coplanar gap width, anddistributed capacitance along the line. The electrode thickness highly affects

the microwave loss in the structure as shown in fig. 4-14.

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6

8

10

12

14

16

18

20

2

4

6

8

10

12

14

16

0 2 4 6 8 10 12 14 16

nload (5um)

nload (10um)

nload (20um)

Loss (dB/cm) 5um

Loss (dB/cm) 10um

Loss (dB/cm) 20um

M i c r o w a v e

I n d e x

L o s s ( d B

/ c m

)

Electrode Thickness (um)

Fig. 4-14 Microwave index and loss for loaded CPS line [2000pF/m loading] with differentelectrode widths varying from 5-20 microns

Clearly an electrode thickness exceeding 2m is preferable to reduce both the

microwave index and loss. For the work shown here, the p-metal thickness isapproximately 1.5mm. The loaded-microwave index drops significantly with

electrode thickness although as we have shown before, we would like 3.7-

4.2. This does not take into account the change in effective index when the

area between the center conductor and ground are removed or BCB is

placed below the contacts. Although thickening the electrode improves the

index match, the characteristic impedance is reduced. In order to match the

characteristic impedance and index simultaneously, the capacitance per unit

length of the line must be reduced.

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Table 4-1Material Relative

PermittivityLoss Tangent (tan d ) Typical

ResistivityInP n 12.4,12.6 5x10-5 20E-4 ohm-cmInP SI 12.4 1.5E7 ohm-cmGaAs 12.85 5x10-4High resistivity Si 11.9 1x10-4 at 30GHz 4000ohm-cmStandard Si 11.9 4x10-3 at 30GHz 1000ohm-cmBCB 2.65

Aluminum Nitride 8.9

Conductor loss can be estimated from the unloaded Q factor

f Z w

Q ou

=

2 [4.14]

Good up to 2 GHz

above 2 GHz one must keep in mind the dielectric attenuation is mostly

dependent upon the substrate thickness. The conductor attenuation coefficientis minimized at a particular w/s. This conductor attenuation decreases with

dielectric constant. The dispersion is lower for smaller waveguide dimensions.

The coplanar waveguide design takes into account the dielectric that the lines

reside, the thickness of the metal layer to achieve a 50 ohm line. As the lines

are deposited on a multilayer dielectric not just on the InP surface, one needs

to take into consideration the effective dielectric constant that insues. This can

be calculated analytically using the conformal mapping technique.[407]

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As discussed earlier, conductor losses are reduced by using wider and thicker

electrodes. However, the characteristic impedance is improved by going to

thinner and narrower electrodes. A compromise was made using 8 mm wide

electrodes. The gap between the two ridges was designed to be fairly close

(16mm) to shorten the curved waveguides and reduce propagation losses.

LiNbO3 traveling wave modulators usually use CPW transmission lines as

without careful attention to the electrode gap widths have experienced large RF

losses in CPS structures [453]. It has been found [453] that leakage of the CPS

modes into substrate modes may occur at fairly low frequencies (11 and

22GHz). This leakage is due to the geometry of the device (gaps 0.5mm to

1mm on substrates 0.25-0.5mm thick)

The electrical attenuation was measured for different biases without bias on

the laser or SOA. The loss was extracted from the ABCD parameters [see

appendix] after measuring the S parameters of the device. These values

compare closely with other similar EAM devices that report losses in the range

15-20dB/mm at 40GHz. The microwave loss results with bias are shown in fig.

4-15. As can be seen, the microwave loss decreases considerably with

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reverse bias. This has been attributed to loss due to undepleted material in the

PN junctions25.

Fig 4-15 Microwave loss as a function of frequency and bias from Device #9

As the PN junction depletes out, the loss becomes dominated by ohmic losses

due to the skin depth in the electrodes.

25 Spickermann Dissertation pp 133

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4.8 CPS T-ELECTRODE DEVICES

Much work has been done to velocity match traveling wave Mach Zehnder

structures using T-sections that both increase the characteristic Impedance

and the length thereby reducing the capacitance per unit length and

providing better matching that results in higher bandwidth.

Modulator n-contact

GeAuNiAun-contact

MZ Phase electrode

Semiconductor Optical Amplifier

SGDBR Laser

Fig. 4-16. T-Electrode SPP-MZ-SOA-SGDBR Transmitter Layout

The device layout of these devices is shown in fig. 4-16 above.

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By distributing the capacitance using fins, one can lower the capacitance per

unit length at will. However, as the InP/InGaAsP material has considerable

optical loss, and the mismatch becomes greater between the optical andelectrical waves at longer lengths, the Ts designed in this work did not lengthen

the device much. The periodicity of the tabs is related to the cutoff frequency

for a given phase velocity and width[304].

d

v f phasecutoff 2

=

[4.15]

where d is the spacing of the fins (period) and v phase is the phase velocity.

These Ts are 50 m long with 10m spacing between as shown in fig. 4-17.

Fig 4-17 TW electrode structure with 50 m Ts with 10 m gaps

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This approach is only practical when the capacitance per unit length is already

small. For highly capacitively loaded lines the device length that is required in

order to improve the bandwidth is very large leading to excessive microwaveand optical insertion losses.

In this work a few different CPS transmission line electrode designs were

explored as shown in table 4-2.

Table 4-2 Transmission Line based electrode MZ devices using Dual RF Series Push-pulldrive

#SOAConfig

SOALength

TotalElectrodeLength(um)

ElectrodeWidth

T length(number)

9 Single 400 313 8 50(5)

Activeelectrodelength

Dual 575 400 490.5 8 50(8)

11 Dual 490 500 610 8 50(10)

12 Dual 380 600 734.5 8 50(12)

16 Single 600 400 490.5 8 50(8)

17 Single 500 500 500 15 N/A

18 Single 500 500 500 5 N/A

19 Single 500 500 560 8 100(5)

20 Single 500 500 610 8 50(10)

21 Single 400 600 730.75 8 50(12)

250

10

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4.9 TRAVELING WAVE BANDWIDTH

The bandwidth of a traveling wave modulator is governed by the difference in

the optical and electrical waves and the overlap factor of these two modes, the

frequency dependent attenuation along the device, the termination impedance,

and the length of the device. Accounting for both the attenuation and the

optical-electrical matching and assuming that the device is terminated with thecharacteristic impedance the bandwidth can be approximated as[4]:

22

22

2/

22

2sin

2sinh

)(

+

+

= -l l

l l

e f B l x a

x a

a wherec f

nn op

x [4.16]m 2

][ -=

It is clear from the previous equation that both the attenuation and indexmatching are very important to achieve high bandwidth. Although simple, the

above equation does not take into account mismatches in the characteristic

impedance which is important as it is difficult to reach 50 ohms with such high

loading capacitance.

The small-signal modulation response S 21 can be modeled accounting for theopto-electrical velocity mismatch, microwave attenuation, and impedance

mismatch[5].

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[ ] 221 )(

1)(exp()2exp(

)(

1)(exp

)2exp(1

+

-+-G+

---

-GG-=

L j

j L L

j L

j L

LT

S LS L m

m

m

m

m

m g b

g b g

g b

g b

g [4.17]

where the amplitude transmission into the modulator is : T , The

reflection coefficients at the source and the load are given by:

S G-= 1

)()(

m s

m sS Z Z

Z Z +

-=G [4.18]

)(

)(

m L

m L L

Z Z

Z Z

+

-=G [4.19]

Zm is the characteristic impedance of the modulator, and Z s and ZL are the

impedances of the source and load respec

barton dissertation pdfthe integration of mach-zehnder modulators with sampled grating dbr lasers

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