adf7021-n high performance narrow-band …...high performance narrow-band transceiver ic adf7021-n...

High PerformanceNarrow-Band Transceiver IC

ADF7021-N

Rev. 0 Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2008 Analog Devices, Inc. All rights reserved.

FEATURES Low power, narrow-band transceiver Frequency bands using dual VCO

80 MHz to 650 MHz 842 MHz to 916 MHz

Programmable IF filter bandwidths of 9 kHz, 13.5 kHz, and 18.5 kHz

Modulation schemes: 2FSK, 3FSK, 4FSK, MSK Spectral shaping: Gaussian and raised cosine filtering Data rates supported: 0.05 kbps to 24 kbps 2.3 V to 3.6 V power supply Programmable output power

−16 dBm to +13 dBm in 63 steps Automatic power amplifier (PA) ramp control Receiver sensitivity

−130 dBm at 100 bps, 2FSK −122 dBm at 1 kbps, 2FSK

Patent pending, on-chip image rejection calibration

On-chip VCO and fractional-N PLL On-chip, 7-bit ADC and temperature sensor Fully automatic frequency control loop (AFC) Digital received signal strength indication (RSSI) Integrated Tx/Rx switch 0.1 μA leakage current in power-down mode

APPLICATIONS Narrow-band, short range device (SRD) standards

ARIB STD-T67, ETSI EN 300 220, Korean SRD standard, FCC Part 15, FCC Part 90, FCC Part 95

Low cost, wireless data transfer Remote control/security systems Wireless metering Wireless medical telemetry service (WMTS) Home automation Process and building control Pagers

FUNCTIONAL BLOCK DIAGRAM

Tx/RxCONTROL

AFCCONTROL

2FSK3FSK4FSK

DEMODULATOR

CLOCKAND DATARECOVERY

RSSI/

7-BIT ADC

GAIN

DIV R

RFOUT

LNA

PFDCP

OSC1 OSC2

N/N + 1DIV P

TEMPSENSOR

OSC CLKDIV

CLKOUT

TEST MUX

VCOIN CPOUT

LDO(1:4)

MUXOUTRSET CREG(1:4)

RLNA

RFIN

RFINB

CE

TxRxCLK

SWD

TxRxDATA

SERIALPORT

SLE

SDATA

SREAD

SCLK

IF FILTER

Σ-ΔMODULATOR

PA RAMP

L1 L2

LOG AMP

MUX

2FSK3FSK4FSK

MOD CONTROL

GAUSSIAN/RAISED COSINE

FILTER

3FSKENCODING

AGCCONTROL

MUX

÷1/÷2

VCO1

VCO2

÷2

0724

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1

Figure 1.

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TABLE OF CONTENTS Features .............................................................................................. 1 Applications....................................................................................... 1 Functional Block Diagram .............................................................. 1 Revision History ............................................................................... 2 General Description ......................................................................... 3 Specifications..................................................................................... 4

RF and PLL Specifications........................................................... 4 Transmission Specifications........................................................ 5 Receiver Specifications ................................................................ 6 Digital Specifications ................................................................... 9 General Specifications ............................................................... 10 Timing Characteristics .............................................................. 11 Timing Diagrams........................................................................ 12

Absolute Maximum Ratings.......................................................... 15 ESD Caution................................................................................ 15

Pin Configuration and Function Descriptions........................... 16 Typical Performance Characteristics ........................................... 18 Frequency Synthesizer ................................................................... 22

Reference Input........................................................................... 22 MUXOUT.................................................................................... 23 Voltage Controlled Oscillator (VCO) ...................................... 24 Choosing Channels for Best System Performance................. 25

Transmitter ...................................................................................... 26 RF Output Stage.......................................................................... 26 Modulation Schemes.................................................................. 26 Spectral Shaping ......................................................................... 28 Modulation and Filtering Options ........................................... 29 Transmit Latency ........................................................................ 29 Test Pattern Generator ............................................................... 29

Receiver Section.............................................................................. 30 RF Front End............................................................................... 30 IF Filter......................................................................................... 30 RSSI/AGC.................................................................................... 30

Demodulation, Detection, and CDR....................................... 32 Receiver Setup............................................................................. 34 Demodulator Considerations ................................................... 36 AFC Operation ........................................................................... 36 Automatic Sync Word Detection (SWD)................................ 37

Applications Information .............................................................. 38 IF Filter Bandwidth Calibration ............................................... 38 LNA/PA Matching...................................................................... 39 Image Rejection Calibration ..................................................... 40 Packet Structure and Coding.................................................... 42 Programming After Initial Power-Up ..................................... 42 Applications Circuit ................................................................... 45

Serial Interface ................................................................................ 46 Readback Format........................................................................ 46 Interfacing to a Microcontroller/DSP ..................................... 48 Register 0—N Register............................................................... 49 Register 1—VCO/Oscillator Register ...................................... 50 Register 2—Transmit Modulation Register ............................ 51 Register 3—Transmit/Receive Clock Register........................ 52 Register 4—Demodulator Setup Register ............................... 53 Register 5—IF Filter Setup Register......................................... 54 Register 6—IF Fine Cal Setup Register ................................... 55 Register 7—Readback Setup Register...................................... 56 Register 8—Power-Down Test Register .................................. 57 Register 9—AGC Register......................................................... 58 Register 10—AFC Register ....................................................... 59 Register 11—Sync Word Detect Register ................................ 60 Register 12—SWD/Threshold Setup Register........................ 60 Register 13—3FSK/4FSK Demod Register ............................. 61 Register 14—Test DAC Register............................................... 62 Register 15—Test Mode Register ............................................. 63

Outline Dimensions ....................................................................... 64 Ordering Guide .......................................................................... 64

REVISION HISTORY 2/08—Revision 0: Initial Version

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GENERAL DESCRIPTION The ADF7021-N is a high performance, low power, narrow-band transceiver based on the ADF7021. The ADF7021-N has IF filter bandwidths of 9 kHz, 13.5 kHz, and 18.5 kHz, making it ideally suited to worldwide narrowband standards and particularly those that stipulate 12.5 kHz channel separation.

It is designed to operate in the narrow-band, license-free ISM bands and in the licensed bands with frequency ranges of 80 MHz to 650 MHz and 842 MHz to 916 MHz. The part has both Gaussian and raised cosine transmit data filtering options to improve spectral efficiency for narrow-band applications. It is suitable for circuit applications targeted at the Japanese ARIB STD-T67, the European ETSI EN 300 220, the Korean short range device regulations, the Chinese short range device regulations, and the North American FCC Part 15, Part 90, and Part 95 regulatory standards. A complete transceiver can be built using a small number of external discrete components, making the ADF7021-N very suitable for price-sensitive and area-sensitive applications.

The range of on-chip FSK modulation and data filtering options allows users greater flexibility in their choice of modulation schemes while meeting the tight spectral efficiency requirements. The ADF7021-N also supports protocols that dynamically switch among 2FSK, 3FSK, and 4FSK to maximize communica-tion range and data throughput.

The transmit section contains two voltage controlled oscillators (VCOs) and a low noise fractional-N PLL with an output resolution of <1 ppm. The ADF7021-N has a VCO using an internal LC tank (421 MHz to 458 MHz, 842 MHz to 916 MHz) and a VCO using an external inductor as part of its tank circuit (80 MHz to 650 MHz). The dual VCO design allows dual-band operation where the user can transmit and/or receive at any frequency supported by the internal inductor VCO and can also transmit and/or receive at a particular frequency band supported by the external inductor VCO.

The frequency-agile PLL allows the ADF7021-N to be used in frequency-hopping, spread spectrum (FHSS) systems. Both VCOs operate at twice the fundamental frequency to reduce spurious emissions and frequency pulling problems.

The transmitter output power is programmable in 63 steps from −16 dBm to +13 dBm and has an automatic power ramp control to prevent spectral splatter and help meet regulatory standards. The transceiver RF frequency, channel spacing, and modulation are programmable using a simple 3-wire interface. The device operates with a power supply range of 2.3 V to 3.6 V and can be powered down when not in use.

A low IF architecture is used in the receiver (100 kHz), which minimizes power consumption and the external component count yet avoids dc offset and flicker noise at low frequencies. The IF filter has programmable bandwidths of 9 kHz, 13.5 kHz, and 18.5 kHz. The ADF7021-N supports a wide variety of pro-grammable features including Rx linearity, sensitivity, and IF bandwidth, allowing the user to trade off receiver sensitivity and selectivity against current consumption, depending on the application. The receiver also features a patent-pending automatic frequency control (AFC) loop with programmable pull-in range that allows the PLL to track out the frequency error in the incoming signal.

The receiver achieves an image rejection performance of 56 dB using a patent-pending IR calibration scheme that does not require the use of an external RF source.

An on-chip ADC provides readback of the integrated tempera-ture sensor, external analog input, battery voltage, and RSSI signal, which provides savings on an ADC in some applications. The temperature sensor is accurate to ±10°C over the full oper-ating temperature range of −40°C to +85°C. This accuracy can be improved by performing a 1-point calibration at room temperature and storing the result in memory.

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SPECIFICATIONS VDD = 2.3 V to 3.6 V, GND = 0 V, TA = TMIN to TMAX, unless otherwise noted. Typical specifications are at VDD = 3 V, TA = 25°C. All measurements are performed with the EVAL-ADF7021-NDBxx using the PN9 data sequence, unless otherwise noted.

RF AND PLL SPECIFICATIONS

Table 1. Parameter Min Typ Max Unit Test Conditions/Comments RF CHARACTERISTICS See Table 9 for required VCO_BIAS and

VCO_ADJUST settings Frequency Ranges (Direct Output) 160 650 MHz External inductor VCO 842 916 MHz Internal inductor VCO Frequency Ranges (RF Divide-by-2 Mode) 80 325 MHz External inductor VCO, RF divide-by-2 enabled 421 458 MHz Internal inductor VCO, RF divide-by-2 enabled Phase Frequency Detector (PFD) Frequency1 RF/256 24 MHz

PHASE-LOCKED LOOP (PLL) VCO Gain2

868 MHz, Internal Inductor VCO 67 MHz/V VCO_ADJUST = 0, VCO_BIAS = 8 426 MHz, Internal Inductor VCO 45 MHz/V VCO_ADJUST = 0, VCO_BIAS = 8 426 MHz, External Inductor VCO 27 MHz/V VCO_ADJUST = 0, VCO_BIAS = 3 160 MHz, External Inductor VCO 6 MHz/V VCO_ADJUST = 0, VCO_BIAS = 2

Phase Noise (In-Band) 868 MHz, Internal Inductor VCO −97 dBc/Hz 10 kHz offset, PA = 10 dBm, VDD = 3.0 V,

PFD = 19.68 MHz, VCO_BIAS = 8 433 MHz, Internal Inductor VCO −103 dBc/Hz 10 kHz offset, PA = 10 dBm, VDD = 3.0 V,

PFD = 19.68 MHz, VCO_BIAS = 8 426 MHz, External Inductor VCO −95 dBc/Hz 10 kHz offset, PA = 10 dBm, VDD = 3.0 V,

PFD = 9.84 MHz, VCO_BIAS = 3 Phase Noise (Out-of-Band) −124 dBc/Hz 1 MHz offset, fRF = 433 MHz, PA = 10 dBm,

VDD = 3.0 V, PFD = 19.68 MHz, VCO_BIAS = 8 Normalized In-Band Phase Noise Floor3 −203 dBc/Hz PLL Settling 40 μs Measured for a 10 MHz frequency step to within

5 ppm accuracy, PFD = 19.68 MHz, loop bandwidth (LBW) = 100 kHz

REFERENCE INPUT Crystal Reference4 3.625 24 MHz External Oscillator4, 5 3.625 24 MHz Crystal Start-Up Time6

XTAL Bias = 20 μA 0.930 ms 10 MHz XTAL, 33 pF load capacitors, VDD = 3.0 V XTAL Bias = 35 μA 0.438 ms 10 MHz XTAL, 33 pF load capacitors, VDD = 3.0 V

Input Level for External Oscillator7 OSC1 0.8 V p-p Clipped sine wave OSC2 CMOS levels V

ADC PARAMETERS INL ±0.4 LSB VDD = 2.3 V to 3.6 V, TA = 25°C DNL ±0.4 LSB VDD = 2.3 V to 3.6 V, TA = 25°C

1 The maximum usable PFD at a particular RF frequency is limited by the minimum N divide value. 2 VCO gain measured at a VCO tuning voltage of 0.7 V. The VCO gain varies across the tuning range of the VCO. The software package ADIsimPLL™ can be used to model this

variation. 3 This value can be used to calculate the in-band phase noise for any operating frequency. Use the following equation to calculate the in-band phase noise performance

as seen at the power amplifier (PA) output: −203 + 10 log(fPFD) + 20 logN. 4 Guaranteed by design. Sample tested to ensure compliance. 5 A TCXO, VCXO, or OCXO can be used as an external oscillator. 6 Crystal start-up time is the time from chip enable (CE) being asserted to correct clock frequency on the CLKOUT pin. 7 Refer to the Reference Input section for details on using an external oscillator.

http://www.analog.com/ADIsimPLL

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TRANSMISSION SPECIFICATIONS

Table 2. Parameter Min Typ Max Unit Test Conditions/Comments DATA RATE

2FSK, 3FSK 0.05 18.51 kbps IF_FILTER_BW = 18.5 kHz 4FSK 0.05 24 kbps IF_FILTER_BW = 18.5 kHz

MODULATION Frequency Deviation (fDEV)2 0.056 28.26 kHz PFD = 3.625 MHz

0.306 156 kHz PFD = 20 MHz Deviation Frequency Resolution 56 Hz PFD = 3.625 MHz Gaussian Filter BT 0.5 Raised Cosine Filter Alpha 0.5/0.7 Programmable

TRANSMIT POWER Maximum Transmit Power3 +13 dBm VDD = 3.0 V, TA = 25°C Transmit Power Variation vs.

Temperature ±1 dB −40°C to +85°C

Transmit Power Variation vs. VDD ±1 dB 2.3 V to 3.6 V at 915 MHz, TA = 25°C Transmit Power Flatness ±1 dB 902 MHz to 928 MHz, 3 V, TA = 25°C Programmable Step Size 0.3125 dB −16 dBm to +13 dBm

ADJACENT CHANNEL POWER (ACP) 426 MHz, External Inductor VCO PFD = 9.84 MHz

12.5 kHz Channel Spacing −50 dBc Gaussian 2FSK modulation, measured in a ±4.25 kHz bandwidth at ±12.5 kHz offset, 2.4 kbps PN9 data, 1.2 kHz frequency deviation, compliant with ARIB STD-T67

25 kHz Channel Spacing −50 dBc Gaussian 2FSK modulation, measured in a ±8 kHz bandwidth at ±25 kHz offset, 9.6 kbps PN9 data, 2.4 kHz frequency deviation, compliant with ARIB STD-T67

868 MHz, Internal Inductor VCO PFD = 19.68 MHz 12.5 kHz Channel Spacing −46 dBm Gaussian 2FSK modulation, 10 dBm output power, measured in

a ±6.25 kHz bandwidth at ±12.5 kHz offset, 2.4 kbps PN9 data, 1.2 kHz frequency deviation, compliant with ETSI EN 300 220

25 kHz Channel Spacing −43 dBm Gaussian 2FSK modulation, 10 dBm output power, measured in a ±12.5 kHz bandwidth at ±25 kHz offset, 9.6 kbps PN9 data, 2.4 kHz frequency deviation, compliant with ETSI EN 300 220

433 MHz, Internal Inductor VCO PFD = 19.68 MHz 12.5 kHz Channel Spacing −50 dBm Gaussian 2FSK modulation, 10 dBm output power, measured in

a ±6.25 kHz bandwidth at ±12.5 kHz offset, 2.4 kbps PN9 data, 1.2 kHz frequency deviation, compliant with ETSI EN 300 220

25 kHz Channel Spacing −47 dBm Gaussian 2FSK modulation, 10 dBm output power, measured in a ±12.5 kHz bandwidth at ±25 kHz offset, 9.6 kbps PN9 data, 2.4 kHz frequency deviation, compliant with ETSI EN 300 220

OCCUPIED BANDWIDTH 99.0% of total mean power; 12.5 kHz channel spacing (2.4 kbps PN9 data, 1.2 kHz frequency deviation); 25 kHz channel spacing (9.6 kbps PN9 data, 2.4 kHz frequency deviation)

2FSK Gaussian Data Filtering 12.5 kHz Channel Spacing 3.9 kHz 25 kHz Channel Spacing 9.9 kHz

2FSK Raised Cosine Data Filtering 12.5 kHz Channel Spacing 4.4 kHz 25 kHz Channel Spacing 10.2 kHz

3FSK Raised Cosine Filtering 12.5 kHz Channel Spacing 3.9 kHz 25 kHz Channel Spacing 9.5 kHz

4FSK Raised Cosine Filtering 19.2 kbps PN9 data, 1.2 kHz frequency deviation 25 kHz Channel Spacing 13.2 kHz

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Parameter Min Typ Max Unit Test Conditions/Comments SPURIOUS EMISSIONS

Reference Spurs −65 dBc 100 kHz loop bandwidth HARMONICS4 13 dBm output power, unfiltered conductive/filtered conductive

Second Harmonic −35/−52 dBc Third Harmonic −43/−60 dBc All Other Harmonics −36/−65 dBc

OPTIMUM PA LOAD IMPEDANCE5 fRF = 915 MHz 39 + j61 Ω fRF = 868 MHz 48 + j54 Ω fRF = 450 MHz 98 + j65 Ω fRF = 426 MHz 100 + j65 Ω fRF = 315 MHz 129 + j63 Ω fRF = 175 MHz 173 + j49 Ω

1 Using Gaussian or raised cosine filtering. The frequency deviation should be chosen to ensure that the transmit-occupied signal bandwidth is within the receiver

IF filter bandwidth. 2 For the definition of frequency deviation, refer to the Register 2—Transmit Modulation Register section. 3 Measured as maximum unmodulated power. 4 Conductive filtered harmonic emissions measured on the EVAL-ADF7021-NDBxx, which includes a T-stage harmonic filter (two inductors and one capacitor). 5 For matching details, refer to the LNA/PA Matching section.

RECEIVER SPECIFICATIONS

Table 3. Parameter Min Typ Max Unit Test Conditions/Comments SENSITIVITY Bit error rate (BER) = 10−3, low noise amplifier (LNA)

and power amplifier (PA) matched separately 2FSK

Sensitivity at 0.1 kbps −130 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW = 13.5 kHz


Sensitivity at 1 kbps −122 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW = 13.5 kHz


Gaussian 2FSK Sensitivity at 0.1 kbps −129 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =

13.5 kHz Sensitivity at 0.25 kbps −127 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =

13.5 kHz Sensitivity at 1 kbps −121 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =


18.5 kHz GMSK

Sensitivity at 9.6 kbps −113 dBm fDEV = 2.4 kHz, high sensitivity mode, IF_FILTER_BW = 18.5 kHz

Raised Cosine 2FSK Sensitivity at 0.25 kbps −127 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =

13.5 kHz Sensitivity at 1 kbps −121 dBm fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =


18.5 kHz

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Parameter Min Typ Max Unit Test Conditions/Comments 3FSK

Sensitivity at 9.6 kbps −110 dBm fDEV = 2.4 kHz, high sensitivity mode, IF_FILTER_BW = 18.5 kHz, Viterbi detection on

Raised Cosine 3FSK Sensitivity at 9.6 kbps −110 dBm fDEV = 2.4 kHz, high sensitivity mode, IF_FILTER_BW =

13.5 kHz, alpha = 0.5, Viterbi detection on 4FSK

Sensitivity at 9.6 kbps −112 dBm fDEV (inner) = 1.2 kHz, high sensitivity mode, IF_FILTER_BW = 13.5 kHz

Raised Cosine 4FSK Sensitivity at 9.6 kbps −109 dBm fDEV (inner) = 1.2 kHz, high sensitivity mode,

IF_FILTER_BW = 13.5 kHz, alpha = 0.5 INPUT IP3 Two-tone test, fLO = 860 MHz, F1 = fLO + 100 kHz,

F2 = fLO − 800 kHz Low Gain Enhanced Linearity

Mode −3 dBm LNA_GAIN = 3, MIXER_LINEARITY = 1

Medium Gain Mode −13.5 dBm LNA_GAIN = 10, MIXER_LINEARITY = 0 High Sensitivity Mode −24 dBm LNA_GAIN = 30, MIXER_LINEARITY = 0

ADJACENT CHANNEL REJECTION 868 MHz Wanted signal is 3 dB above the sensitivity point

(BER = 10−3); unmodulated interferer is at the center of the adjacent channel; rejection measured as the difference between the interferer level and the wanted signal level in dB

12.5 kHz Channel Spacing 40 dB 9 kHz IF_FILTER_BW 25 kHz Channel Spacing 39 dB 18.5 kHz IF_FILTER_BW

426 MHz Wanted signal is 3 dB above the reference sensitivity point (BER = 10−2); modulated interferer (same modulation as wanted signal) at the center of the adjacent channel; rejection measured as the difference between the interferer level and reference sensitivity level in dB

12.5 kHz Channel Spacing 40 dB 9 kHz IF_FILTER_BW, compliant with ARIB STD-T67 25 kHz Channel Spacing 39 dB 18.5 kHz IF_FILTER_BW, compliant with ARIB STD-T67

CO-CHANNEL REJECTION Wanted signal (2FSK, 9.6 kbps, ±4 kHz deviation) is 3 dB above the sensitivity point (BER = 10−3), modu-lated interferer

868 MHz −5 dB IMAGE CHANNEL REJECTION Wanted signal (2FSK, 9.6 kbps, ±4 kHz deviation) is

10 dB above the sensitivity point (BER = 10−3); modu-lated interferer (2FSK, 9.6 kbps, ±4 kHz deviation) is placed at the image frequency of fRF − 200 kHz; the interferer level is increased until BER = 10−3

868 MHz 26/39 dB Uncalibrated/calibrated1, VDD = 3.0 V, TA = 25°C 450 MHz, Internal Inductor

VCO 29/50 dB Uncalibrated/calibrated1, VDD = 3.0 V, TA = 25°C

BLOCKING Wanted signal is 10 dB above the input sensitivity level; CW interferer level is increased until BER = 10−3

±1 MHz 69 dB ±2 MHz 75 dB ±5 MHz 78 dB ±10 MHz 78.5 dB

SATURATION (MAXIMUM INPUT LEVEL)

12 dBm 2FSK mode, BER = 10−3

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Parameter Min Typ Max Unit Test Conditions/Comments RSSI

Range at Input2 −120 to −47 dBm Linearity ±2 dB Input power range = −100 dBm to −47 dBm Absolute Accuracy ±3 dB Input power range = −100 dBm to −47 dBm Response Time 390 μs See the RSSI/AGC section

AFC Pull-In Range 0.5 1.5 × IF_

FILTER_BW kHz The range is programmable in Register 10

(R10_DB[24:31]) Response Time 64 Bits Accuracy 0.5 kHz Input power range = −100 dBm to +12 dBm

Rx SPURIOUS EMISSIONS3 Internal Inductor VCO −91/−91 dBm <1 GHz at antenna input, unfiltered conductive/filtered

conductive −52/−70 dBm >1 GHz at antenna input, unfiltered conductive/filtered

conductive External Inductor VCO −62/−72 dBm <1 GHz at antenna input, unfiltered conductive/filtered

conductive −64/−85 dBm >1 GHz at antenna input, unfiltered conductive/filtered

conductive LNA INPUT IMPEDANCE RFIN to RFGND

fRF = 915 MHz 24 − j60 Ω fRF = 868 MHz 26 − j63 Ω fRF = 450 MHz 63 − j129 Ω fRF = 426 MHz 68 − j134 Ω fRF = 315 MHz 96 − j160 Ω fRF = 175 MHz 178 − j190 Ω

1 Calibration of the image rejection used an external RF source. 2 For received signal levels < −100 dBm, it is recommended to average the RSSI readback value over a number of samples to improve the RSSI accuracy at low input powers. 3 Filtered conductive receive spurious emissions are measured on the EVAL-ADF7021-NDBxx, which includes a T-stage harmonic filter (two inductors and one

capacitor).

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DIGITAL SPECIFICATIONS

Table 4. Parameter Min Typ Max Unit Test Conditions/Comments TIMING INFORMATION

Chip Enabled to Regulator Ready 10 μs CREG (1:4) = 100 nF Chip Enabled to Tx Mode 32-bit register write time = 50 μs

TCXO Reference 1 ms XTAL 2 ms

Chip Enabled to Rx Mode 32-bit register write time = 50 μs, IF filter coarse calibration only

TCXO Reference 1.2 ms XTAL 2.2 ms

Tx-to-Rx Turnaround Time 390 μs + (5 × tBIT) Time to synchronized data out, includes AGC settling (three AGC levels)and CDR synchronization; see the AGC Information and Timing section for more details; tBIT = data bit period

LOGIC INPUTS Input High Voltage, VINH 0.7 × VDD V Input Low Voltage, VINL 0.2 × VDD V Input Current, IINH/IINL ±1 μA Input Capacitance, CIN 10 pF Control Clock Input 50 MHz

LOGIC OUTPUTS Output High Voltage, VOH DVDD − 0.4 V IOH = 500 μA Output Low Voltage, VOL 0.4 V IOL = 500 μA CLKOUT Rise/Fall 5 ns CLKOUT Load 10 pF

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GENERAL SPECIFICATIONS

Table 5. Parameter Min Typ Max Unit Test Conditions/Comments TEMPERATURE RANGE (TA) −40 +85 °C POWER SUPPLIES

Voltage Supply, VDD 2.3 3.6 V All VDD pins must be tied together TRANSMIT CURRENT CONSUMPTION1 VDD = 3.0 V, PA is matched into 50 Ω

868 MHz VCO_BIAS = 8 0 dBm 20.2 mA 5 dBm 24.7 mA 10 dBm 32.3 mA

450 MHz, Internal Inductor VCO VCO_BIAS = 8 0 dBm 19.9 mA 5 dBm 23.2 mA 10 dBm 29.2 mA

426 MHz, External Inductor VCO VCO_BIAS = 2 0 dBm 13.5 mA 5 dBm 17 mA 10 dBm 23.3 mA

RECEIVE CURRENT CONSUMPTION VDD = 3.0 V 868 MHz VCO_BIAS = 8

Low Current Mode 22.7 mA High Sensitivity Mode 24.6 mA

433MHz, Internal Inductor VCO VCO_BIAS = 8 Low Current Mode 24.5 mA High Sensitivity Mode 26.4 mA

426 MHz, External Inductor VCO VCO_BIAS = 2 Low Current Mode 17.5 mA High Sensitivity Mode 19.5 mA

POWER-DOWN CURRENT CONSUMPTION Low Power Sleep Mode 0.1 1 μA CE low

1 The transmit current consumption tests used the same combined PA and LNA matching network as that used on the EVAL-ADF7021-NDBxx evaluation boards.

Improved PA efficiency is achieved by using a separate PA matching network.

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TIMING CHARACTERISTICS VDD = 3 V ± 10%, DGND = AGND = 0 V, TA = 25°C, unless otherwise noted. Guaranteed by design but not production tested.

Table 6. Parameter Limit at TMIN to TMAX Unit Test Conditions/Comments t1 >10 ns SDATA to SCLK setup time t2 >10 ns SDATA to SCLK hold time t3 >25 ns SCLK high duration t4 >25 ns SCLK low duration t5 >10 ns SCLK to SLE setup time t6 >20 ns SLE pulse width t8 <25 ns SCLK to SREAD data valid, readback t9 <25 ns SREAD hold time after SCLK, readback t10 >10 ns SCLK to SLE disable time, readback t11 5 < t11 < (¼ × tBIT) ns TxRxCLK negative edge to SLE t12 >5 ns TxRxDATA to TxRxCLK setup time (Tx mode) t13 >5 ns TxRxCLK to TxRxDATA hold time (Tx mode) t14 >¼ × tBIT μs TxRxCLK negative edge to SLE t15 >¼ × tBIT μs SLE positive edge to positive edge of TxRxCLK

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TIMING DIAGRAMS Serial Interface

SCLK

SLE

DB31 (MSB) DB30 DB2 DB1(CONTROL BIT C2)SDATA DB0 (LSB)

(CONTROL BIT C1)

t6

t1 t2

t3 t4

t5 0724

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2

Figure 2. Serial Interface Timing Diagram

t8

t3

t1 t2

t10

t9

X RV16 RV15 RV2

SCLK

SDATA

SLE

SREAD

REG7 DB0(CONTROL BIT C1)

RV1 X

0724

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3

Figure 3. Serial Interface Readback Timing Diagram

2FSK/3FSK Timing

TxRxCLK

DATATxRxDATA

±1 × DATA RATE/32 1/DATA RATE

0724

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4

Figure 4. TxRxDATA/TxRxCLK Timing Diagram in Receive Mode

TxRxCLK

DATATxRxDATA

SAMPLEFETCH

1/DATA RATE

0724

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5

Figure 5. TxRxDATA/TxRxCLK Timing Diagram in Transmit Mode

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4FSK Timing

In 4FSK receive mode, MSB/LSB synchronization should be guaranteed by SWD in the receive bit stream.

Rx SYMBOLMSB

Rx SYMBOLLSB

Rx SYMBOLMSB

Rx SYMBOLLSB

Tx SYMBOLMSB

Tx SYMBOLLSBTxRxDATA

TxRxCLK

SLE

Rx MODE Tx MODE

REGISTER 0 WRITESWITCH FROM Rx TO Tx

Tx/Rx MODE

Tx SYMBOLMSB

t11 t12

t13

tBIT

tSYMBOL

0724

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4

Figure 6. Receive-to-Transmit Timing Diagram in 4FSK Mode

Tx SYMBOLMSB

Tx SYMBOLLSB

Tx SYMBOLMSB

Tx SYMBOLLSB

Rx SYMBOLLSB

Rx SYMBOLMSBTxRxDATA

TxRxCLK

SLE

Tx MODE Rx MODE

REGISTER 0 WRITESWITCH FROM Tx TO Rx

Tx/Rx MODE

t15

t14tBIT

tSYMBOL

0724

6-07

5

Figure 7. Transmit-to-Receive Timing Diagram in 4FSK Mode

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UART/SPI Mode

UART mode is enabled by setting R0_DB28 to 1. SPI mode is enabled by setting R0_DB28 to 1 and setting R15_DB[17:19] to 0x7. The transmit/receive data clock is available on the CLKOUT pin.

Tx BIT Tx BIT Tx BIT Tx BITTxRxCLK

(TRANSMIT DATA INPUTIN UART/SPI MODE.)

CLKOUT(TRANSMIT/RECEIVE DATA

CLOCK IN SPI MODE.NOT USED IN UART MODE.)

Tx MODETx/Rx MODE

TxRxDATA(RECEIVE DATA OUTPUT

IN UART/SPI MODE.) HIGH-Z

Tx BIT

tBIT

FETCH SAMPLE

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2

Figure 8. Transmit Timing Diagram in UART/SPI Mode

Rx BIT Rx BIT Rx BIT Rx BIT

TxRxCLK(TRANSMIT DATA INPUT

IN UART/SPI MODE.)

CLKOUT(TRANSMIT/RECEIVE DATA

CLOCK IN SPI MODE.NOT USED IN UART MODE.)

Rx MODETx/Rx MODE

TxRxDATA(RECEIVE DATA OUTPUT

IN UART/SPI MODE.)

HIGH-Z

Rx BIT

tBIT

FETCH SAMPLE

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8

Figure 9. Receive Timing Diagram in UART/SPI Mode

ADF7021-N

Rev. 0 | of 64

ABSOLUTE MAXIMUM RATINGS TA = 25°C, unless otherwise noted.

Table 7. Parameter Rating VDD to GND1 −0.3 V to +5 V Analog I/O Voltage to GND −0.3 V to AVDD + 0.3 V Digital I/O Voltage to GND −0.3 V to DVDD + 0.3 V Operating Temperature Range

Industrial (B Version) −40°C to +85°C Storage Temperature Range −65°C to +125°C Maximum Junction Temperature 150°C MLF θJA Thermal Impedance 26°C/W Reflow Soldering

Peak Temperature 260°C Time at Peak Temperature 40 sec

1 GND = CPGND = RFGND = DGND = AGND = 0.

Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

This device is a high performance RF integrated circuit with an ESD rating of <2 kV and it is ESD sensitive. Proper precautions should be taken for handling and assembly.

ESD CAUTION

ADF7021-N

Rev. 0 | of 64

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

0724

6-00

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36

35

34

33

32

31

30

29

28

27

26

25

48 47 46 45 44 43 42 41 40 39 38 37

13 14 15 16 17 18 19 20 21 22 23 24

VCOINCREG1

VDD1RFOUTRFGND

RFINRFINB

RLNAVDD4RSET

CREG4GND4

MIX

_IM

IX_I

MIX

_QM

IX_Q

FILT

_IFI

LT_I

GN

D4

FILT

_QFI

LT_Q

GN

D4

TEST

_A CE

CLKOUT

TxRxDATATxRxCLK

SWDVDD2CREG2ADCINGND2SCLKSREADSDATASLE

CVC

OG

ND

1L1 G

ND

L2 VDD

CPO

UT

CR

EG3

VDD

3O

SC1

OSC

2M

UXO

UT

1

2

3

4

5

6

7

8

9

10

11

12

PIN 1INDICATOR

ADF7021-NTOP VIEW

(Not to Scale)

Figure 10. Pin Configuration

Table 8. Pin Function Descriptions Pin No. Mnemonic Description 1 VCOIN The tuning voltage on this pin determines the output frequency of the voltage controlled oscillator (VCO).

The higher the tuning voltage, the higher the output frequency. 2 CREG1 Regulator Voltage for PA Block. Place a series 3.9 Ω resistor and a 100 nF capacitor between this pin and

ground for regulator stability and noise rejection. 3 VDD1 Voltage Supply for PA Block. Place decoupling capacitors of 0.1 μF and 100 pF as close as possible to this pin.

Tie all VDD pins together. 4 RFOUT The modulated signal is available at this pin. Output power levels are from −16 dBm to +13 dBm. The output

should be impedance matched to the desired load using suitable components (see the Transmitter section). 5 RFGND Ground for Output Stage of Transmitter. All GND pins should be tied together. 6 RFIN LNA Input for Receiver Section. Input matching is required between the antenna and the differential LNA

input to ensure maximum power transfer (see the LNA/PA Matching section). 7 RFINB Complementary LNA Input. (See the LNA/PA Matching section.) 8 RLNA External Bias Resistor for LNA. Optimum resistor is 1.1 kΩ with 5% tolerance. 9 VDD4 Voltage Supply for LNA/MIXER Block. This pin should be decoupled to ground with a 10 nF capacitor. 10 RSET External Resistor. Sets charge pump current and some internal bias currents. Use a 3.6 kΩ resistor with 5% tolerance. 11 CREG4 Regulator Voltage for LNA/MIXER Block. Place a 100 nF capacitor between this pin and GND for regulator

stability and noise rejection. 12, 19, 22 GND4 Ground for LNA/MIXER Block. 13 to 18 MIX_I, MIX_I,

MIX_Q, MIX_Q, FILT_I, FILT_I

Signal Chain Test Pins. These pins are high impedance under normal conditions and should be left unconnected.

20, 21, 23 FILT_Q, FILT_Q, TEST_A

Signal Chain Test Pins. These pins are high impedance under normal conditions and should be left unconnected.

24 CE Chip Enable. Bringing CE low puts the ADF7021-N into complete power-down. Register values are lost when CE is low, and the part must be reprogrammed after CE is brought high.

25 SLE Load Enable, CMOS Input. When SLE goes high, the data stored in the shift registers is loaded into one of the four latches. A latch is selected using the control bits.

26 SDATA Serial Data Input. The serial data is loaded MSB first with the four LSBs as the control bits. This pin is a high impedance CMOS input.

27 SREAD Serial Data Output. This pin is used to feed readback data from the ADF7021-N to the microcontroller. The SCLK input is used to clock each readback bit (for example, AFC or ADC) from the SREAD pin.

28 SCLK Serial Clock Input. This serial clock is used to clock in the serial data to the registers. The data is latched into the 32-bit shift register on the CLK rising edge. This pin is a digital CMOS input.

ADF7021-N

Rev. 0 | of 64

Pin No. Mnemonic Description 29 GND2 Ground for Digital Section. 30 ADCIN Analog-to-Digital Converter Input. The internal 7-bit ADC can be accessed through this pin. Full scale is 0 V to

1.9 V. Readback is made using the SREAD pin. 31 CREG2 Regulator Voltage for Digital Block. Place a 100 nF capacitor between this pin and ground for regulator

stability and noise rejection. 32 VDD2 Voltage Supply for Digital Block. Place a decoupling capacitor of 10 nF as close as possible to this pin. 33 SWD Sync Word Detect. The ADF7021-N asserts this pin when it has found a match for the sync word sequence

(see the Register 11—Sync Word Detect Register section). This provides an interrupt for an external microcontroller indicating that valid data is being received.

34 TxRxDATA Transmit Data Input/Received Data Output. This is a digital pin, and normal CMOS levels apply. In UART/SPI mode, this pin provides an output for the received data in receive mode. In transmit UART/SPI mode, this pin is high impedance (see the Interfacing to a Microcontroller/DSP section).

35 TxRxCLK Outputs the data clock in both receive and transmit modes. This is a digital pin, and normal CMOS levels apply. The positive clock edge is matched to the center of the received data. In transmit mode, this pin outputs an accurate clock to latch the data from the microcontroller into the transmit section at the exact required data rate. In UART/SPI mode, this pin is used to input the transmit data in transmit mode. In receive UART/SPI mode, this pin is high impedance (see the Interfacing to a Microcontroller/DSP section).

36 CLKOUT A divided-down version of the crystal reference with output driver. The digital clock output can be used to drive several other CMOS inputs such as a microcontroller clock. The output has a 50:50 mark-space ratio and is inverted with respect to the reference. Place a series 1 kΩ resistor as close as possible to the pin in applications where the CLKOUT feature is being used.

37 MUXOUT Provides the DIGITAL_LOCK_DETECT signal. This signal is used to determine if the PLL is locked to the correct frequency. It also provides other signals such as REGULATOR_READY, which is an indicator of the status of the serial interface regulator (see the MUXOUT section for more information).

38 OSC2 Connect the reference crystal between this pin and OSC1. A TCXO reference can be used by driving this pin with CMOS levels and disabling the internal crystal oscillator.

39 OSC1 Connect the reference crystal between this pin and OSC2. A TCXO reference can be used by driving this pin with ac-coupled 0.8 V p-p levels and by enabling the internal crystal oscillator.

40 VDD3 Voltage Supply for the Charge Pump and PLL Dividers. Decouple this pin to ground with a 10 nF capacitor. 41 CREG3 Regulator Voltage for Charge Pump and PLL Dividers. Place a 100 nF capacitor between this pin and ground

for regulator stability and noise rejection. 42 CPOUT Charge Pump Output. This output generates current pulses that are integrated in the loop filter. The

integrated current changes the control voltage on the input to the VCO. 43 VDD Voltage Supply for VCO Tank Circuit. Decouple this pin to ground with a 10 nF capacitor. 44, 46 L2, L1 External VCO Inductor Pins. If using an external VCO inductor, connect a chip inductor across these pins to set

the VCO operating frequency. If using the internal VCO inductor, these pins can be left floating. See the Voltage Controlled Oscillator (VCO) section for more information.

45, 47 GND, GND1 Grounds for VCO Block. 48 CVCO Place a 22 nF capacitor between this pin and CREG1 to reduce VCO noise.

ADF7021-N

Rev. 0 | of 64

TYPICAL PERFORMANCE CHARACTERISTICS

FREQUENCY OFFSET (kHz)

PHA

SE N

OIS

E (d

Bc/

Hz)

–150

–140

–130

–120

–110

–100

–90

–80

–70

1 10 100 1000 10000

RF FREQ = 900MHzVDD = 2.3VTEMPERATURE = 25°CVCO_BIAS = 8VCO_ADJUST = 3ICP = 0.8mA

ICP = 1.4mA

ICP = 2.2mA

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Figure 11. Phase Noise Response at 900 MHz, VDD = 2.3 V

–40–36–32–28–24–20–16–12–8–4

048

1216

0 4 8 12 16 20 24 28 32 36 40 44 48 52 56 60

PA SETTING

RF

OU

TPU

T PO

WER

(dB

m)

PA_BIAS = 5µA

PA_BIAS = 11µA

PA_BIAS = 9µA

PA_BIAS = 7µA

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Figure 12. RF Output Power vs. PA Setting

VBW 100HzSTART 300MHzRES BW 100Hz SWEEP 385.8ms (601pts)

STOP 3.5GHz

RF FREQ = 440MHzOUTPUT POWER = 10dBmFILTER = T-STAGE LC FILTERMARKER Δ = 52.2dB

1R

1

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Figure 13. PA Output Harmonic Response with T-Stage LC Filter

CENTER 869.5 25MHzRES BW 300Hz SWEEP 2.118s (601pts)

SPAN 50kHz

DR = 9.6kbpsDATA = PRBS9fDEV = 2.4kHzRF FREQ = 869.5MHz

VBW 300Hz

GFSK

2FSK

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7

Figure 14. Output Spectrum in 2FSK and GFSK Modes

CENTER 869.5 25MHzRES BW 300Hz SWEEP 2.118s (601pts)VBW 300Hz

SPAN 50kHz

DR = 9.6kbpsDATA = PRBS9fDEV = 2.4kHzRF FREQ = 869.5MHz

RC2FSK

2FSK

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Figure 15. Output Spectrum in 2FSK and Raised Cosine 2FSK Modes

VBW 300HzCENTER 869.493 8MHzRES BW 300Hz SWEEP 4.237s (601pts)

SPAN 100kHz

SR = 4.8ksym/sDATA = PRBS9fDEV = 2.4kHzRF FREQ = 869.5MHz

RC4FSK

4FSK

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ADF7021-N

Rev. 0 | of 64

RES BW 300HzCENTER 869.5MHz VBW 300Hz SPAN 50kHz

SWEEP2.226s (401pts)

REF 15dBmSAMP LOG 10dB/ ATTEN 25dB

VAVG 100V1 V2S3 FC

DR = 9.6kbpsDATA = PRS9fDEV = 2.4kHzRF FREQ = 869.5MHz

3FSK

RC3FSK

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FREQUENCY OFFSET (kHz)

OU

TPU

T PO

WER

(dB

m)

0

–10

10

–20

–30

–40

–100 –50 500 100

–50

–60

RAMP RATE:CW ONLY256 CODES/BIT128 CODES/BIT64 CODES/BIT32 CODES/BIT

TRACE = MAX HOLDPA ON/OFF RATE = 3HzPA ON/OFF CYCLES = 10,000VDD = 3.0V

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8

Figure 18. Output Spectrum in Maximum Hold

for Various PA Ramp Rate Options

–8

–7

–6

–5

–4

–3

–2

–1

0

–122 –120 –118 –116 –114 –112 –110 –108 –106 –104

3.0V, +25°C

3.6V, –40°C2.3V, +85°C

RF INPUT POWER (dBm)

LOG

BER

DATA RATE = 9.6kbpsfDEV = 4kHzRF FREQ = 868MHzIF BW = 25kHz

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2

Figure 19. 2FSK Sensitivity vs. VDD and Temperature, fRF = 868 MHz

–8

–7

–6

–5

–4

–3

–2

–1

0

–130 –128 –126 –124 –122 –120 –118 –116 –114 –112 –110 –108

3.6V, –40°C

3.0V, +25°C2.3V, +85°C


LOG

BER

DATA RATE = 1kbpsfDEV = 1kHzRF FREQ = 135MHzIF BW = 12.5kHz

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3


–8

–7

–6

–5

–4

–3

–2

–1

0

–120 –115 –110 –105 –100 –95


LOG

BER

2.3V +25°C3.0V +25°C3.6V +25°C2.3V –40°C3.0V –40°C3.6V –40°C2.3V +85°C3.0V +85°C3.6V +85°C

3FSK MODULATIONDATA RATE = 9.6kbpsfDEV = 2.4kHzMOD INDEX = 0.5RF FREQ = 440 MHz

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5


–8

–7

–6

–5

–4

–3

–2

–1

0

–120 –115 –110 –105 –100 –95


LOG

BER

2.3V +25°C3.0V +25°C3.6V +25°C2.3V –40°C3.0V –40°C3.6V –40°C2.3V +85°C3.0V +85°C3.6V +85°C

DATA RATE = 19.6kbpsSYMBOL RATE = 9.8ksym/sfDEV (inner) = 2.4kHzMOD INDEX = 0.5RF FREQ = 420MHzIF BW = 12.5kHz

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ADF7021-N

Rev. 0 | of 64

FREQUENCY OFFSET (MHz)

BLO

CK

ING

(dB

)

–10

0

10

20

30

40

50

60

70

80

90

–22 –18 –14 –10 –6 –2 0 2 6 10 14 18 22

RF FREQ = 868MHz WANTED SIGNAL (10dB ABOVE SENSITIVITY POINT) = 2FSK, fDEV = 4kHz, DATA RATE = 9.8kbpsBLOCKER = 2FSK,

fDEV = 4kHz, DATA RATE = 9.8kbpsVDD = 3.0VTEMPERATURE = 25°C

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Figure 23. Wideband Interference Rejection

–140

–120

–100

–80

–60

–40

–20

–122.5 –112.5 –102.5 –92.5 –82.5 –72.5 –62.5 –52.5 –42.5

ACTUAL RF INPUT LEVEL

RF INPUT (dBm)

RSS

I LEV

EL (d

Bm

)

RSSIREADBACK LEVEL

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Figure 24. Digital RSSI Readback Linearity

–10

0

10

20

30

40

50

60

70

429.80 429.85 429.90 429.95 430.00 430.05 430.10 430.15 430.20

RF FREQ = 430MHzEXTERNAL VCO INDUCTORDATA RATE = 9.6kbpsTEMPERATURE = 25°C, VDD = 3.0V

RF FREQUENCY (MHz)

BLO

CK

ING

(dB

)

CALIBRATED

UNCALIBRATED

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Figure 25. Image Rejection, Uncalibrated vs. Calibrated

0072

46-0

91

2.50

–2.5–5.0–7.5

–10.0–12.5–15.0–17.5–20.0–22.5–25.0–27.5–30.0–32.5–35.0–37.5

90 92 94 96 98 100 102 104 106 108 110

ATT

ENU

ATI

ON

(dB

)

IF FREQUENCY (kHz)

–40°C

+90°C

Figure 26. Variation of IF Filter Response with Temperature

(IF_FILTER_BW = 9 kHz, Temperature Range is −40°C to +90°C in 10° Steps)

MODULATION INDEX

SEN

SITI

VITY

PO

INT

(dB

m)

–118

–116

–114

–112

–110

–108

–106

–104

–102

–100

0 0.2 0.4 0.6 0.8 1.0 1.2

RF FREQ = 860MHz2FSK MODULATIONDATA RATE = 9.6kbpsIF BW = 25kHzVDD = 3.0VTEMPERATURE = 25°C

DISCRIMINATOR BANDWIDTH =1× FSK FREQUENCY DEVIATION

DISCRIMINATOR BANDWIDTH =2× FSK FREQUENCY DEVIATION

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Figure 27. 2FSK Sensitivity vs. Modulation Index vs. Correlator Discriminator Bandwidth

–120 –118 –116 –114 –112 –110 –108 –106 –104 –102 –100

3FSK MODULATIONVDD = 3.0V, TEMP = 25°CDATA RATE = 9.6kbpsfDEV = 2.4kHzRF FREQ = 868MHzIF BW = 18.75kHz

INPUT POWER (dBm)

LOG

BER

–7

–6

–5

–4

–3

–2

–1

0

VITERBI DETECTION

THRESHOLD DETECTION

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Figure 28. 3FSK Receiver Sensitivity Using Viterbi Detection and

Threshold Detection

ADF7021-N

Rev. 0 | of 64

REC

EIVE

R S

YMB

OL

LEVE

L

+1

+3

–1

–3

0

RF I/P LEVEL = –70dBmDATA RATE = 9.7kbpsfDEV (inner) = 1.2kHz

22452 ACQS M 50µs

IF BW = 25kHzPOST DEMOD BW = 12.4kHz

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Figure 29. 4FSK Receiver Eye Diagram Measured Using the Test DAC Output

+1

–1

0

4

20834 ACQS M 20µs C13 1.7V

RF I/P LEVEL = –70dBmDATA RATE = 10kbpsfDEV = 2.5kHz

IF BW = 12.5kHzPOST DEMOD BW = 12.4kHz

REC

EIVE

R S

YMB

OL

LEVE

L

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Figure 30. 3FSK Receiver Eye Diagram Measured Using the Test DAC Output

LNA GAIN, FILTER GAIN

SEN

SITI

VITY

(dB

m)

–130

–120

–110

–100

–90

–80

–70

3, 72(LOW GAIN MODE)

10, 72(MEDIUM GAIN MODE)

30, 72(HIGH GAIN MODE)

HIGH MIXERLINEARITY

DEFAULTMIXER

LINEARITY

MODULATION = 2FSKDATA RATE = 9.6kbpsfDEV = 4kHzIF BW = 12.5kHzDEMOD = CORRELATORSENSITIVITY @ 1E-3 BER

IP3 = –3dBm

IP3= –5dBm

IP3 = –9dBm

IP3 = –13.5dBm

IP3 = –24dBm

IP3 = –20dBm

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Figure 31. Receive Sensitivity vs. LNA/IF Filter Gain and Mixer Linearity Settings (The input IP3 at each setting is also shown)

ADF7021-N

Rev. 0 | of 64

FREQUENCY SYNTHESIZER REFERENCE INPUT The on-board crystal oscillator circuitry (see Figure 32) can use a quartz crystal as the PLL reference. Using a quartz crystal with a frequency tolerance of ≤10 ppm for narrow-band applications is recommended. It is possible to use a quartz crystal with >10 ppm tolerance, but to comply with the absolute frequency error specifications of narrow-band regulations (for example, ARIB STD-T67 and ETSI EN 300 220), compensation for the frequency error of the crystal is necessary.

The oscillator circuit is enabled by setting R1_DB12 high. It is enabled by default on power-up and is disabled by bringing CE low. Errors in the crystal can be corrected by using the automatic frequency control feature or by adjusting the fractional-N value (see the N Counter section).

OSC1

CP1CP2

OSC2

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Figure 32. Oscillator Circuit on the ADF7021-N

Two parallel resonant capacitors are required for oscillation at the correct frequency. Their values are dependent on the crystal specification. They should be chosen to make sure that the series value of capacitance added to the PCB track capacitance adds up to the specified load capacitance of the crystal, usually 12 pF to 20 pF. Track capacitance values vary from 2 pF to 5 pF, depending on board layout. When possible, choose capacitors that have a very low temperature coefficient to ensure stable frequency operation over all conditions.

Using a TCXO Reference

A single-ended reference (TCXO, VCXO, or OCXO) can also be used with the ADF7021-N. This is recommended for applications having absolute frequency accuracy requirements of <10 ppm, such as applications requiring compliance with ARIB STD-T67 or ETSI EN 300 220. The following are two options for interfacing the ADF7021-N to an external reference oscillator.

• An oscillator with CMOS output levels can be applied to OSC2. The internal oscillator circuit should be disabled by setting R1_DB12 low.

• An oscillator with 0.8 V p-p levels can be ac-coupled through a 22 pF capacitor into OSC1. The internal oscillator circuit should be enabled by setting R1_DB12 high.

Programmable Crystal Bias Current

Bias current in the oscillator circuit can be configured between 20 μA and 35 μA by writing to the XTAL_BIAS bits (R1_DB [13:14]). Increasing the bias current allows the crystal oscillator to power up faster.

CLKOUT Divider and Buffer

The CLKOUT circuit takes the reference clock signal from the oscillator section, shown in Figure 32, and supplies a divided-down, 50:50 mark-space signal to the CLKOUT pin. The CLKOUT signal is inverted with respect to the reference clock. An even divide from 2 to 30 is available. This divide number is set in R1_DB[7:10]. On power-up, the CLKOUT defaults to divide-by-8.

DVDD

CLKOUTENABLE BIT

CLKOUTOSC1 DIVIDER1 TO 15

÷2

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Figure 33. CLKOUT Stage

To disable CLKOUT, set the divide number to 0. The output buffer can drive up to a 20 pF load with a 10% rise time at 4.8 MHz. Faster edges can result in some spurious feedthrough to the output. A series resistor (1 kΩ) can be used to slow the clock edges to reduce these spurs at the CLKOUT frequency.

R Counter

The 3-bit R counter divides the reference input frequency by an integer between 1 and 7. The divided-down signal is presented as the reference clock to the phase frequency detector (PFD). The divide ratio is set in R1_DB[4:6]. Maximizing the PFD frequency reduces the N value. This reduces the noise multiplied at a rate of 20 log(N) to the output and reduces occurrences of spurious components.

Register 1 defaults to R = 1 on power-up.

PFD [Hz] = XTAL/R

Loop Filter

The loop filter integrates the current pulses from the charge pump to form a voltage that tunes the output of the VCO to the desired frequency. It also attenuates spurious levels generated by the PLL. A typical loop filter design is shown in Figure 34.

CHARGEPUMP OUT

VCO

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Figure 34. Typical Loop Filter Configuration

The loop should be designed so that the loop bandwidth (LBW) is approximately 100 kHz. This provides a good compromise between in-band phase noise and out-of-band spurious rejection. Widening the LBW excessively reduces the time spent jumping between frequencies, but it can cause insufficient spurious attenua-tion. Narrow-loop bandwidths can result in the loop taking long periods to attain lock and can also result in a higher level of power falling into the adjacent channel. The loop filter design on the

ADF7021-N

Rev. 0 | of 64

EVAL-ADF7021-NDBxx should be used for optimum performance.

The free design tool ADI SRD Design Studio™ can also be used to design loop filters for the ADF7021-N (see the ADI SRD Design Studio web site for details).

N Counter

The feedback divider in the ADF7021-N PLL consists of an 8-bit integer counter (R0_DB[19:26]) and a 15-bit, sigma-delta (Σ-Δ) fractional_N divider (R0_DB[4:18]). The integer counter is the standard pulse-swallow type that is common in PLLs. This sets the minimum integer divide value to 23. The fractional divide value provides very fine resolution at the output, where the output frequency of the PLL is calculated as

⎟⎟⎠

⎞⎜⎜⎝

⎛+×= 152

__

NFractionalNInteger

RXTALfOUT

When RF_DIVIDE_BY_2 (see the Voltage Controlled Oscillator (VCO) section) is selected, this formula becomes

⎟⎠⎞

⎜⎝⎛ +××= 152

_0.5 NFractionalInteger_NR

XTALfOUT

The combination of Integer_N (maximum = 255) and Fractional_N (maximum = 32,768/32,768) gives a maximum N divider of 255 + 1. Therefore, the minimum usable PFD is

[ ]( )1255

Hz+

=FrequencyOutputRequiredMaximum

PFD MIN

For example, when operating in the European 868 MHz to 870 MHz band, PFDMIN = 3.4 MHz.

VCO

4\N

THIRD-ORDERΣ-Δ MODULATOR

PFD/CHARGE

PUMP

4\R

INTEGER_NFRACTIONAL_N

REFERENCE IN

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Figure 35. Fractional_N PLL

Voltage Regulators

The ADF7021-N contains four regulators to supply stable voltages to the part. The nominal regulator voltage is 2.3 V. Regulator 1 requires a 3.9 Ω resistor and a 100 nF capacitor in series between CREG1 and GND, whereas the other regulators require a 100 nF capacitor connected between CREGx and GND. When CE is high, the regulators and other associated circuitry are powered on, drawing a total supply current of 2 mA. Bringing the CE pin low disables the regulators, reduces the supply current to less than 1 μA, and erases all values held in the registers.

The serial interface operates from a regulator supply. Therefore, to write to the part, the user must have CE high and the regulator

voltage must be stabilized. Regulator status (CREG4) can be monitored using the REGULATOR_READY signal from the MUXOUT pin.

MUXOUT The MUXOUT pin allows access to various digital points in the ADF7021-N. The state of MUXOUT is controlled in Register 0 (R0_DB[29:31]).

REGULATOR_READY

REGULATOR_READY is the default setting on MUXOUT after the transceiver is powered up. The power-up time of the regulator is typically 50 μs. Because the serial interface is powered from the regulator, the regulator must be at its nominal voltage before the ADF7021-N can be programmed. The status of the regulator can be monitored at MUXOUT. When the regulator ready signal on MUXOUT is high, programming of the ADF7021-N can begin.

REGULATOR_READY (DEFAULT)

DIGITAL_LOCK_DETECT

RSSI_READY

Tx_Rx

LOGIC_ZERO

TRISTATE

MUX CONTROL

DGND

DVDD

MUXOUT

FILTER_CAL_COMPLETE

LOGIC_ONE

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Figure 36. MUXOUT Circuit

FILTER_CAL_COMPLETE

MUXOUT can be set to FILTER_CAL_COMPLETE. This signal goes low for the duration of both a coarse IF filter calibration and a fine IF filter calibration. It can be used as an interrupt to a microcontroller to signal the end of the IF filter calibration.

DIGITAL_LOCK_DETECT

DIGITAL_LOCK_DETECT indicates when the PLL has locked. The lock detect circuit is located at the PFD. When the phase error on five consecutive cycles is less than 15 ns, lock detect is set high. Lock detect remains high until a 25 ns phase error is detected at the PFD.

RSSI_READY

MUXOUT can be set to RSSI_READY. This indicates that the internal analog RSSI has settled and a digital RSSI readback can be performed.

Tx_Rx

Tx_Rx signifies whether the ADF7021-N is in transmit or receive mode. When in transmit mode, this signal is low. When in receive mode, this signal is high. It can be used to control an external Tx/Rx switch.

http://www.analog.com/ADIsimPLL

http://www.analog.com/srd_design_studio

ADF7021-N

Rev. 0 | of 64

A plot of the VCO operating frequency vs. total external inductance (chip inductor + PCB track) is shown in Figure 38.

VOLTAGE CONTROLLED OSCILLATOR (VCO) The ADF7021-N contains two VCO cores. The first VCO, the internal inductor VCO, uses an internal LC tank and supports 842 MHz to 916 MHz and 421 MHz to 458 MHz operating bands. The second VCO, the external inductor VCO, uses an external inductor as part of its LC tank and supports the RF operating band of 80 MHz to 650 MHz.

0 5 10 15 20 25 30200

250

300

350

400

450

500

550

600

650

700

TOTAL EXTERNAL INDUCTANCE (nH)

FREQ

UEN

CY

(MH

z)

750

fMIN (MHz)

fMAX (MHz)

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To minimize spurious emissions, both VCOs operate at twice the RF frequency. The VCO signal is then divided by 2 inside the synthesizer loop, giving the required frequency for the transmitter and the required local oscillator (LO) frequency for the receiver. A further divide-by-2 (RF_DIVIDE_BY_2) is performed outside the synthesizer loop to allow operation in the 421 MHz to 458 MHz band (internal inductor VCO) and the 80 MHz to 325 MHz band (external inductor VCO).

The VCO needs an external 22 nF capacitor between the CVCO pin and the regulator (CREG1 pin) to reduce internal noise.

Figure 38. Direct RF Output vs. Total External Inductance

The inductance for a PCB track using FR4 material is approxi-mately 0.57 nH/mm. This should be subtracted from the total value to determine the correct chip inductor value.

VCOLOOP FILTER

VCO_BIASR1_DB(19:22)

220µF

CVCO PIN

RF_DIVIDE_BY_2R1_DB18

÷2÷2

MUX

TON DIVIDER

TO PA

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Typically, a particular inductor value allows the ADF7021-N to function over a range of ±6% of the RF operating frequency. When the RF_DIVIDE_BY_2 bit (R1_DB18) is selected, this range becomes ±3%. At 400 MHz, for example, an operating range of ±24 MHz (that is, 376 MHz to 424 MHz) with a single inductor (VCO range centered at 400 MHz) can be expected. Figure 37. Voltage Controlled Oscillator (VCO)

The VCO tuning voltage can be checked for a particular RF output frequency by measuring the voltage on the VCOIN pin when the part is fully powered up in transmit or receive mode.

Internal Inductor VCO

To select the internal inductor VCO, set R1_DB25 to Logic 0, which is the default setting.

The VCO tuning range is 0.2 V to 2 V. The external inductor value should be chosen to ensure that the VCO is operating as close as possible to the center of this tuning range. This is particularly important for RF frequencies <200 MHz, where the VCO gain is reduced and a tuning range of <±6 MHz exists.

VCO bias current can be adjusted using R1_DB[19:22]. To ensure VCO oscillation, the minimum bias current setting under all conditions when using the internal inductor VCO is 0x8.

The VCO should be recentered, depending on the required frequency of operation, by programming the VCO_ADJUST bits (R1_DB[23:24]). This is detailed in Table 9. The VCO operating frequency range can be adjusted by

programming the VCO_ADJUST bits (R1_DB[23:24]). This typically allows the VCO operating range to be shifted up or down by a maximum of 1% of the RF frequency.

External Inductor VCO

When using the external inductor VCO, the center frequency of the VCO is set by the internal varactor capacitance and the combined inductance of the external chip inductor, bond wire, and PCB track. The external inductor is connected between the L2 and L1 pins.

To select the external inductor VCO, set R1_DB25 to Logic 1. The VCO_BIAS should be set depending on the frequency of operation (as indicated in Table 9).

ADF7021-N

Rev. 0 | of 64

Table 9. RF Output Frequency Ranges for Internal and External Inductor VCOs and Required Register Settings Register Settings

RF Frequency Output (MHz)

VCO to Be Used

RF Divide by 2

VCO_INDUCTOR R1_DB25

RF_DIVIDE_BY_2 R1_DB18

VCO_ADJUST R1_DB[23:24]

VCO_BIAS R1_DB[19:22]

870 to 916 Internal L No 0 0 11 8 842 to 870 Internal L No 0 0 00 8 440 to 458 Internal L Yes 0 1 11 8 421 to 440 Internal L Yes 0 1 00 8 450 to 650 External L No 1 0 XX 4 200 to 450 External L No 1 0 XX 3 80 to 200 External L Yes 1 1 XX 2

CHOOSING CHANNELS FOR BEST SYSTEM PERFORMANCE An interaction between the RF VCO frequency and the reference frequency can lead to fractional spur creation. When the synthesizer is in fractional mode (that is, the RF VCO and reference frequencies are not integer related), spurs can appear on the VCO output spectrum at an offset frequency that corresponds to the difference frequency between an integer multiple of the reference and the VCO frequency.

These spurs are attenuated by the loop filter. They are more noticeable on channels close to integer multiples of the reference where the difference frequency may be inside the loop bandwidth; thus, the name integer boundary spurs. The occurrence of these spurs is rare because the integer frequencies are around multiples of the reference, which is typically >10 MHz. To avoid having very small or very large values in the fractional register, choose a suitable reference frequency.

ADF7021-N

Rev. 0 | of 64

TRANSMITTER RF OUTPUT STAGE The power amplifier (PA) of the ADF7021-N is based on a single-ended, controlled current, open-drain amplifier that has been designed to deliver up to 13 dBm into a 50 Ω load at a maximum frequency of 950 MHz.

The PA output current and consequently, the output power, are programmable over a wide range. The PA configuration is shown in Figure 39. The output power is set using R2_DB[13:18].

IDAC

2

6R2_DB(13:18)

R2_DB7

R2_DB(11:12)

+

RFGND

RFOUT

FROM VCO

R0_DB27

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Figure 39. PA Configuration

The PA is equipped with overvoltage protection, which makes it robust in severe mismatch conditions. Depending on the appli-cation, users can design a matching network for the PA to exhibit optimum efficiency at the desired radiated output power level for a wide range of antennas, such as loop or monopole antennas. See the LNA/PA Matching section for more information.

PA Ramping

When the PA is switched on or off quickly, its changing input impedance momentarily disturbs the VCO output frequency. This process is called VCO pulling, and it manifests as spectral splatter or spurs in the output spectrum around the desired carrier frequency. Some radio emissions regulations place limits on these PA transient-induced spurs (for example, the ETSI EN 300 220 regulations). By gradually ramping the PA on and off, PA transient spurs are minimized.

The ADF7021-N has built-in PA ramping configurability. As Figure 40 illustrates, there are eight ramp rate settings, defined as a certain number of PA setting codes per one data bit period. The PA steps through each of its 64 code levels but at different speeds for each setting. The ramp rate is set by configuring R2_DB[8:10].

If the PA is enabled/disabled by the PA_ENABLE bit (R2_DB7), it ramps up and down. If it is enabled/disabled by the Tx/Rx bit (R0_DB27), it ramps up and turns hard off.

DATA BITS

PA RAMP 0(NO RAMP)

PA RAMP 1(256 CODES PER BIT)







1 2 3 4 ... 8 ... 16

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Figure 40. PA Ramping Settings

PA Bias Currents

The PA_BIAS bits (R2_DB[11:12]) facilitate an adjustment of the PA bias current to further extend the output power control range, if necessary. If this feature is not required, the default value of 9 μA is recommended. If output power of greater than 10 dBm is required, a PA bias setting of 11 μA is recommended. The output stage is powered down by resetting R2_DB7.

MODULATION SCHEMES The ADF7021-N supports 2FSK, 3FSK, and 4FSK modulation. The implementation of these modulation schemes is shown in Figure 41.

VCO

÷N

THIRD-ORDERΣ-Δ MODULATOR

PFD/CHARGE

PUMP

REF

INTEGER_NTx_FREQUENCY_DEVIATION

TOPA STAGE

1 – D2 PRSHAPING

4FSKBIT SYMBOL

MAPPER

MUX

TxDATA

2FSK

4FSK

GAUSSIANOR

RAISED COSINEFILTERING

PRE-CODER

3FSK

÷2LOOP FILTER

FRACTIONAL_N

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Figure 41. Transmit Modulation Implementation

ADF7021-N

Rev. 0 | of 64

Setting the Transmit Data Rate

In all modulation modes except oversampled 2FSK mode, an accurate clock is provided on the TxRxCLK pin to latch the data from the microcontroller into the transmit section at the required data rate. The exact frequency of this clock is defined by

32____ ××=

DIVIDECLKCDRDIVIDECLKDEMODXTALCLKDATA

where: XTAL is the crystal or TCXO frequency. DEMOD_CLK_DIVIDE is the divider that sets the demodulator clock rate (R3_DB[6:9]). CDR_CLK_DIVIDE is the divider that sets the CDR clock rate (R3_DB[10:17]).

Refer to the Register 3—Transmit/Receive Clock Register section for more programming information.

Setting the FSK Transmit Deviation Frequency

In all modulation modes, the deviation from the center frequency is set using the Tx_FREQUENCY_DEVIATION bits (R2_DB[19:27]).

The deviation from the center frequency in Hz is as follows:

For direct RF output,

162__]Hz[ DEVIATIONFREQUENCYTxPFDfDEV

×=

For RF_DIVIDE_BY_2 enabled,

162__5.0]Hz[ DEVIATIONFREQUENCYTxPFDfDEV

××=

where Tx_FREQUENCY_DEVIATION is a number from 1 to 511 (R2_DB[19:27]).

In 4FSK modulation, the four symbols (00, 01, 11, 10) are transmitted as ±3 × fDEV and ±1 × fDEV.

Binary Frequency Shift Keying (2FSK)

Two-level frequency shift keying is implemented by setting the N value for the center frequency and then toggling it with the TxDATA line. The deviation from the center frequency is set using the Tx_FREQUENCY_DEVIATION bits, R2_DB[19:27].

2FSK is selected by setting the MODULATION_SCHEME bits (R2_DB[4:6]) to 000.

Minimum shift keying (MSK) or Gaussian minimum shift keying (GMSK) is supported by selecting 2FSK modulation and using a modulation index of 0.5. A modulation index of 0.5 is set up by configuring R2_DB[19:27] for an fDEV = 0.25 × transmit data rate.

3-Level Frequency Shift Keying (3FSK)

In 3-level FSK modulation (also known as modified duobinary FSK), the binary data (Logic 0 and Logic 1) is mapped onto three distinct frequencies: the carrier frequency (fC), the carrier frequency minus a deviation frequency (fC − fDEV), and the carrier frequency plus the deviation frequency (fC + fDEV).

A Logic 0 is mapped to the carrier frequency while a Logic 1 is either mapped onto the fC − fDEV frequency or the fC + fDEV frequency.

fCfC – fDEV fC + fDEV

RF FREQUENCY

0+1–1

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Figure 42. 3FSK Symbol-to-Frequency Mapping

Compared to 2FSK, this bits-to-frequency mapping results in a reduced transmission bandwidth because some energy is removed from the RF sidebands and transferred to the carrier frequency. At low modulation index, 3FSK improves the transmit spectral efficiency by up to 25% when compared to 2FSK.

Bit-to-symbol mapping for 3FSK is implemented using a linear convolutional encoder that also permits Viterbi detection to be used in the receiver. A block diagram of the transmit hardware used to realize this system is shown in Figure 43. The convolu-tional encoder polynomial used to implement the transmit spectral shaping is

P(D) = 1 − D2

where: P is the convolutional encoder polynomial. D is the unit delay operator.

A digital precoder with transfer function 1/P(D) implements an inverse modulo-2 operation of the 1 − D2 shaping filter in the transmitter.

PRECODER1/P(D)

CONVOLUTIONALENCODER

P(D)

FSK MODCONTROL

ANDDATA FILTERING

Tx DATA0, 1

0, +1, –1

0, 1

TON DIVIDER

fCfC + fDEVfC – fDEV

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Figure 43. 3FSK Encoding

ADF7021-N

Rev. 0 | of 64

The signal mapping of the input binary transmit data to the 3-level convolutional output is shown in Table 10. The convolutional encoder restricts the maximum number of sequential +1s or −1s to two and delivers an equal number of +1s and −1s to the FSK modulator, thus ensuring equal spectral energy in both RF sidebands.

Table 10. 3-Level Signal Mapping of the Convolutional Encoder TxDATA 1 0 1 1 0 0 1 0 0 1

Precoder Output 1 0 0 1 0 1 1 1 1 0

Encoder Output +1 0 −1 +1 0 0 +1 0 0 −1

Another property of this encoding scheme is that the transmitted symbol sequence is dc-free, which facilitates symbol detection and frequency measurement in the receiver. In addition, there is no code rate loss associated with this 3-level convolutional encoder; that is, the transmitted symbol rate is equal to the data rate presented at the transmit data input.

3FSK is selected by setting the MODULATION_SCHEME bits (R2_DB[4:6]) to 010. It can also be used with raised cosine filtering to further increase the spectral efficiency of the transmit signal.

4-Level Frequency Shift Keying (4FSK)

In 4FSK modulation, two bits per symbol spectral efficiency is realized by mapping consecutive input bit-pairs in the Tx data bit stream to one of four possible symbols (−3, −1, +1, +3). Thus, the transmitted symbol rate is half of the input bit rate.

By minimizing the separation between symbol frequencies, 4FSK can have high spectral efficiency. The bit-to-symbol mapping for 4FSK is gray coded and is shown in Figure 44.

Tx DATA

SYMBOLFREQUENCIES

f

t

+3fDEV

+fDEV

–fDEV

–3fDEV

0 0 0 1 1 0 1 1

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Figure 44. 4FSK Bit-to-Symbol Mapping

The inner deviation frequencies (+fDEV and −fDEV) are set using the Tx_FREQUENCY_DEVIATION bits, R2_DB[19:27]. The outer deviation frequencies are automatically set to three times the inner deviation frequency.

The transmit clock from Pin TxRxCLK is available after writing to Register 3 in the power-up sequence for receive mode. The MSB of the first symbol should be clocked into the ADF7021-N on the first transmit clock pulse from the ADF7021-N after writing to Register 3. Refer to Figure 6 for more timing information.

Oversampled 2FSK

In oversampled 2FSK, there is no data clock from the TxRxCLK pin. Instead, the transmit data at the TxRxDATA pin is sampled at 32 times the programmed rate.

This is the only modulation mode that can be used with the UART mode interface for data transmission (refer to the Interfacing to a Microcontroller/DSP section for more information).

SPECTRAL SHAPING Gaussian or raised cosine filtering can be used to improve transmit spectral efficiency. The ADF7021-N supports Gaussian filtering (bandwidth time [BT] = 0.5) on 2FSK modulation. Raised cosine filtering can be used with 2FSK, 3FSK, or 4FSK modulation. The roll-off factor (alpha) of the raised cosine filter has programmable options of 0.5 and 0.7. Both the Gaussian and raised cosine filters are implemented using linear phase digital filter architectures that deliver precise control over the BT and alpha filter parameters, and guarantee a transmit spectrum that is very stable over temperature and supply variation.

Gaussian Frequency Shift Keying (GFSK)

Gaussian frequency shift keying reduces the bandwidth occupied by the transmitted spectrum by digitally prefiltering the transmit data. The BT product of the Gaussian filter used is 0.5.

Gaussian filtering can only be used with 2FSK modulation. This is selected by setting R2_DB[4:6] to 001.

Raised Cosine Filtering

Raised cosine filtering provides digital prefiltering of the transmit data by using a raised cosine filter with a roll-off factor (alpha) of either 0.5 or 0.7. The alpha is set to 0.5 by default, but the raised cosine filter bandwidth can be increased to provide less aggressive data filtering by using an alpha of 0.7 (set R2_DB30 to Logic 1). Raised cosine filtering can be used with 2FSK, 3FSK, and 4FSK.

Raised cosine filtering is enabled by setting R2_DB[4:6] as outlined in Table 11.

ADF7021-N

Rev. 0 | of 64

MODULATION AND FILTERING OPTIONS The various modulation and data filtering options are described in Table 11.

Table 11. Modulation and Filtering Options Modulation Data Filtering R2_DB[4:6] BINARY FSK

2FSK None 000 MSK1 None 000 OQPSK with Half Sine Baseband Shaping2

None 000

GFSK Gaussian 001 GMSK3 Gaussian 001 RC2FSK Raised cosine 101 Oversampled 2FSK None 100

3-LEVEL FSK 3FSK None 010 RC3FSK Raised cosine 110

4-LEVEL FSK 4FSK None 011 RC4FSK Raised cosine 111

1 MSK is 2FSK modulation with a modulation index = 0.5. 2 Offset quadrature phase shift keying (OQPSK) with half sine baseband shaping

is spectrally equivalent to MSK. 3 GMSK is GFSK with a modulation index = 0.5.

TRANSMIT LATENCY Transmit latency is the delay time from the sampling of a bit/symbol by the TxRxCLK signal to when that bit/symbol appears at the RF output. The latency without any data filtering is one bit. The addition of data filtering adds a further latency as outlined in Table 12.

It is important that the ADF7021-N be left in transmit mode after the last data bit is sampled by the data clock to account for this latency. The ADF7021-N should stay in transmit mode for a time equal to the number of latency bit periods for the applied modulation scheme. This ensures that all of the data sampled by the TxRxCLK signal appears at RF.

The figures for latency in Table 12 assume that the positive TxRxCLK edge is used to sample data (default). If the TxRxCLK is inverted by setting R2_DB[28:29], an additional 0.5 bit latency can be added to all values in Table 12.

Table 12. Bit/Symbol Latency in Transmit Mode for Various Modulation Schemes Modulation Latency 2FSK 1 bit

GFSK 4 bits RC2FSK, Alpha = 0.5 5 bits RC2FSK, Alpha = 0.7 4 bits

3FSK 1 bit RC3FSK, Alpha = 0.5 5 bits RC3FSK, Alpha = 0.7 4 bits

4FSK 1 symbol RC4FSK, Alpha = 0.5 5 symbols RC4FSK, Alpha = 0.7 4 symbols

TEST PATTERN GENERATOR The ADF7021-N has a number of built-in test pattern generators that can be used to facilitate radio link setup or RF measurement.

A full list of the supported patterns is shown in Table 13. The data rate for these test patterns is the programmed data rate set in Register 3.

The PN9 sequence is suitable for test modulation when carrying out adjacent channel power (ACP) or occupied bandwidth measurements.

Table 13. Transmit Test Pattern Generator Options Test Pattern R15_DB[8:10] Normal 000 Transmit Carrier 001 Transmit + fDEV Tone 010 Transmit − fDEV Tone 011 Transmit 1010 Pattern 100 Transmit PN9 Sequence 101 Transmit SWD Pattern Repeatedly 110

ADF7021-N Error! Unknown document

property name.

Rev. 0 | of 64

RECEIVER SECTION RF FRONT END The ADF7021-N is based on a fully integrated, low IF receiver architecture. The low IF architecture facilitates a very low external component count and does not suffer from powerline-induced interference problems.

Figure 45 shows the structure of the receiver front end. The many programming options allow users to trade off sensitivity, linearity, and current consumption to best suit their application. To achieve a high level of resilience against spurious reception, the low noise amplifier (LNA) features a differential input. Switch SW2 shorts the LNA input when transmit mode is selected (R0_DB27 = 0). This feature facilitates the design of a combined LNA/PA matching network, avoiding the need for an external Rx/Tx switch. See the LNA/PA Matching section for details on the design of the matching network.

SW2 LNA

RFIN

RFINB

Tx/Rx SELECT(R0_DB27)

LNA_MODE(R9_DB25)

LNA_BIAS(R9_DB[26:27])

MIXER LINEARITY(R9_DB28)

LO

I (TO FILTER)

Q (TO FILTER)

LNA_GAIN(R9_DB[20:21])

LNA/MIXER_ENABLE(R8_DB6) 07

246-

017

Figure 45. RF Front End

The LNA is followed by a quadrature downconversion mixer, which converts the RF signal to the IF frequency of 100 kHz. An important consideration is that the output frequency of the synthesizer must be programmed to a value 100 kHz below the center frequency of the received channel. The LNA has two basic operating modes: high gain/low noise mode and low gain/low power mode. To switch between these two modes, use the LNA_MODE bit (R9_DB25). The mixer is also configurable between a low current and an enhanced linearity mode using the MIXER_LINEARITY bit (R9_DB28).

Based on the specific sensitivity and linearity requirements of the application, it is recommended to adjust the LNA_MODE bit and MIXER_LINEARITY bit as outlined in Table 15.

The gain of the LNA is configured by the LNA_GAIN bits (R9_DB[20:21]) and can be set by either the user or the automatic gain control (AGC) logic.

IF FILTER IF Filter Settings

Out-of-band interference is rejected by means of a fifth-order Butterworth polyphase IF filter centered on a frequency of 100 kHz. The bandwidth of the IF filter can be programmed to 9 kHz, 13.5 kHz, or 18.5 kHz by R4_DB[30:31] and should be

chosen as a compromise between interference rejection and attenuation of the desired signal.

If the AGC loop is disabled, the gain of the IF filter can be set to one of three levels by using the FILTER_GAIN bits (R9_DB[22:23]). The filter gain is adjusted automatically if the AGC loop is enabled.

IF Filter Bandwidth and Center Frequency Calibration

To compensate for manufacturing tolerances, the IF filter should be calibrated after power-up to ensure that the bandwidth and center frequency are correct. Coarse and fine calibration schemes are provided to offer a choice between fast calibration (coarse calibration) and high filter centering accuracy (fine calibration). Coarse calibration is enabled by setting R5_DB4 high. Fine calibration is enabled by setting R6_DB4 high.

For details on when it is necessary to perform a filter calibration, and in what applications to use either a coarse calibration or fine calibration, refer to the IF Filter Bandwidth Calibration section.

RSSI/AGC The RSSI is implemented as a successive compression log amp following the baseband (BB) channel filtering. The log amp achieves ±3 dB log linearity. It also doubles as a limiter to convert the signal-to-digital levels for the FSK demodulator. The offset correction circuit uses the BBOS_CLK_DIVIDE bits (R3_DB[4:5]), which should be set between 1 MHz and 2 MHz. The RSSI level is converted for user readback and for digitally controlled AGC by an 80-level (7-bit) flash ADC. This level can be converted to input power in dBm. By default, the AGC is on when powered up in receive mode.

1

IFWR IFWR IFWR IFWR

LATCHA A A

R

CLK

ADC

OFFSETCORRECTION

RSSI

FSKDEMOD

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Figure 46. RSSI Block Diagram

RSSI Thresholds

When the RSSI is above AGC_HIGH_THRESHOLD (R9_DB[11:17]), the gain is reduced. When the RSSI is below AGC_LOW_THRESHOLD (R9_DB[4:10]), the gain is increased. The thresholds default to 30 and 70 on power-up in receive mode. A delay (set by AGC_CLK_DIVIDE, R3_DB[26:31]) is programmed to allow for settling of the loop. A value of 13 is recommended to give an AGC update rate of 7.7 kHz.

ADF7021-N

Rev. 0 | of 64

The user has the option of changing the two threshold values from the defaults of 30 and 70 (Register 9). The default AGC setup values should be adequate for most applications. The threshold values must be more than 30 apart for the AGC to operate correctly.

Offset Correction Clock

In Register 3, the user should set the BBOS_CLK_DIVIDE bits (R3_DB[4:5]) to give a baseband offset clock (BBOS CLK) frequency between 1 MHz and 2 MHz.

BBOS CLK [Hz] = XTAL/(BBOS_CLK_DIVIDE)

where BBOS_CLK_DIVIDE can be set to 4, 8, 16, or 32.

AGC Information and Timing

AGC is selected by default and operates by setting the appropriate LNA and filter gain settings for the measured RSSI level. It is possible to disable AGC by writing to Register 9 if the user wants to enter one of the modes listed in Table 15. The time for the AGC circuit to settle and, therefore, the time it takes to measure the RSSI accurately, is typically 390 μs. However, this depends on how many gain settings the AGC circuit has to cycle through. After each gain change, the AGC loop waits for a programmed time to allow transients to settle. This AGC update rate is set according to

AGC Update Rate [Hz] = DIVIDECLKAGC

DIVIDECLKSEQ__

[Hz]__

where: AGC_CLK_DIVIDE is set by R3_DB[26:31]. A value of 13 is recommended. SEQ_CLK_DIVIDE = 100 kHz (R3_DB[18:25]).

By using the recommended setting for AGC_CLK_DIVIDE, the total AGC settling time is

[Hz][sec]

RateUpdateAGCChangesGainAGCofNumber

TimeSettlingAGC =

The worst case for AGC settling occurs when the AGC control loop has to cycle through all five gain settings, which gives a maximum AGC settling time of 650 μs.

RSSI Formula (Converting to dBm)

The RSSI formula is

Input Power [dBm] = −130 dBm + (Readback Code + Gain Mode Correction) × 0.5

where: Readback Code is given by Bit RV7 to Bit RV1 in the Register 7 readback register (see Figure 58 and the Readback Format section). Gain Mode Correction is given by the values in Table 14.

The LNA gain (LG2, LG1) and filter gain (FG2, FG1) values are also obtained from the readback register, as part of an RSSI readback.

Table 14. Gain Mode Correction LNA Gain (LG2, LG1)

Filter Gain (FG2, FG1)

Gain Mode Correction

H (1, 0) H (1, 0) 0 M (0, 1) H (1, 0) 24 M (0, 1) M (0, 1) 38 M (0, 1) L (0, 0) 58 L (0, 0) L (0, 0) 86 An additional factor should be introduced to account for losses in the front-end-matching network/antenna.

Table 15. LNA/Mixer Modes

Receiver Mode LNA_MODE (R9_DB25)

LNA_GAIN (R9_DB[20:21])

MIXER_LINEARITY(R9_DB28)

Sensitivity (2FSK, DR = 4.8 kbps, fDEV = 4 kHz)

Rx Current Consumption (mA)

Input IP3 (dBm)

High Sensitivity Mode (Default)

0 30 0 −118 24.6 −24

Enhanced Linearity High Gain

0 30 1 −114.5 24.6 −20

Medium Gain 1 10 0 −112 22.1 −13.5

Enhanced Linearity Medium Gain

1 10 1 −105.5 22.1 −9

Low Gain 1 3 0 −100 22.1 −5 Enhanced Linearity Low Gain

1 3 1 −92.3 22.1 −3

ADF7021-N

Rev. 0 | of 64

DEMODULATION, DETECTION, AND CDR System Overview

An overview of the demodulation, detection, and clock and data recovery (CDR) of the received signal on the ADF7021-N is shown in Figure 47.

The quadrature outputs of the IF filter are first limited and then fed to either the correlator FSK demodulator or to the linear FSK demodulator. The correlator demodulator is used to demodulate 2FSK, 3FSK, and 4FSK. The linear demodulator is used for frequency measurement and is enabled when the AFC loop is active. The linear demodulator can also be used to demodulate 2FSK.

Following the demodulator, a digital post demodulator filter removes excess noise from the demodulator signal output. Threshold/slicer detection is used for data recovery of 2FSK and 4FSK. Data recovery of 3FSK can be implemented using either threshold detection or Viterbi detection.

An on-chip CDR PLL is used to resynchronize the received bit stream to a local clock. It outputs the retimed data and clock on the TxRxDATA and TxRxCLK pins, respectively.

POST

DEM

OD

FIL

TER

I

Q

LIMITERS

VITERBIDETECTION

MUX

CLOCKAND

DATARECOVERY

TxRxDATA

TxRxCLK

FREQUENCY CORRELATOR

LINEARDEMODULATOR

MUX

3FSK

THRESHOLDDETECTION

2/3/4FSK

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Figure 47. Overview of Demodulation, Detection, and CDR Process

Correlator Demodulator

The correlator demodulator can be used for 2FSK, 3FSK, and 4FSK demodulation. Figure 48 shows the operation of the correlator demodulator for 2FSK.

IF – fDEV IF + fDEV

I

IFQ

LIMITERS

R4_DB(10:19)

R4_DB7DOT_PRODUCT

FREQUENCY CORRELATOR

DISCRIMINATOR_BWR4_DB9

Rx_INVERT

DISCRIM BW

2FSK = +1, –13FSK = +1, 0, –14FSK = +3, +1, –1, –3

OUTPUT LEVELS:

0724

6-07

9

Figure 48. 2FSK Correlator Demodulator Operation

The quadrature outputs of the IF filter are first limited and then fed to a digital frequency correlator that performs filtering and frequency discrimination of the 2FSK/3FSK/4FSK spectrum.

For 2FSK modulation, data is recovered by comparing the output levels from two correlators. The performance of this frequency discriminator approximates that of a matched filter detector, which is known to provide optimum detection in the presence of additive white Gaussian noise (AWGN). This method of FSK demodulation provides approximately 3 dB to 4 dB better sensitivity than a linear demodulator.

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Linear Demodulator

Figure 49 shows a block diagram of the linear demodulator.

POST

_DEM

OD

_FI

LTER

ENVE

LOPE

DET

ECTO

R SLICER2FSK

FREQUENCY

IFLEVELI

Q

LIMITER

LINEARDISCRIMINATOR

R4_DB(20:29)FREQUENCYREADBACKAND AFC LOOP

+ 2FSK RxDATA

RxCLK

0724

6-07

3

Figure 49. Block Diagram of Linear FSK Demodulator

A digital frequency discriminator provides an output signal that is linearly proportional to the frequency of the limiter outputs. The discriminator output is filtered and averaged using a combined averaging filter and envelope detector. The demodulated 2FSK data from the post demodulator filter is recovered by slicing against the output of the envelope detector, as shown in Figure 49. This method of demodulation corrects for frequency errors between transmitter and receiver when the received spectrum is close to or within the IF bandwidth. This envelope detector output is also used for AFC readback and provides the frequency estimate for the AFC control loop.

Post Demodulator Filter

A second-order, digital low-pass filter removes excess noise from the demodulated bit stream at the output of the discriminator. The bandwidth of this post demodulator filter is programmable and must be optimized for the user’s data rate and received modulation type. If the bandwidth is set too narrow, performance degrades due to intersymbol interference (ISI). If the bandwidth is set too wide, excess noise degrades the performance of the receiver. The POST_DEMOD_BW bits (R4_DB[20:29]) set the bandwidth of this filter.

2FSK Bit Slicer/Threshold Detection

2FSK demodulation can be implemented using the correlator FSK demodulator or the linear FSK demodulator. In both cases, threshold detection is used for data recovery at the output of the post demodulation filter.

The output signal levels of the correlator demodulator are always centered about zero. Therefore, the slicer threshold level can be fixed at zero, and the demodulator performance is independent of the run-length constraints of the transmit data bit stream. This results in robust data recovery that does not suffer from the classic baseline wander problems that exist in the more traditional FSK demodulators.

When the linear demodulator is used for 2FSK demodulation, the output of the envelope detector is used as the slicer threshold, and this output tracks frequency errors that are within the IF filter bandwidth.

3FSK and 4FSK Threshold Detection

4FSK demodulation is implemented using the correlator demodulator followed by the post demodulator filter and threshold detection. The output of the post demodulation filter is a 4-level signal that represents the transmitted symbols (−3, −1, +1, +3). Threshold detection of 4FSK requires three threshold settings, one that is always fixed at 0 and two that are programmable and are symmetrically placed above and below zero using the 3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]).

3FSK demodulation is implemented using the correlator demodu-lator, followed by a post demodulator filter. The output of the post demodulator filter is a 3-level signal that represents the transmitted symbols (−1, 0, +1). Data recovery of 3FSK can be implemented using threshold detection or Viterbi detection. Threshold detection is implemented using two thresholds that are programmable and are symmetrically placed above and below zero using the 3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]).

3FSK Viterbi Detection

Viterbi detection of 3FSK operates on a four-state trellis and is implemented using two interleaved Viterbi detectors operating at half the symbol rate. The Viterbi detector is enabled by R13_DB11.

To facilitate different run length constraints in the transmitted bit stream, the Viterbi path memory length is programmable in steps of 4 bits, 6 bits, 8 bits, or 32 bits by setting the VITERBI_PATH_MEMORY bits (R13_DB[13:14]). This should be set equal to or longer than the maximum number of consecutive 0s in the interleaved transmit bit stream.

When used with Viterbi detection, the receiver sensitivity for 3FSK is typically 3 dB greater than that obtained using threshold detection. When the Viterbi detector is enabled, however, the receiver bit latency is increased by twice the Viterbi path memory length.

Clock Recovery

An oversampled digital clock and data recovery (CDR) PLL is used to resynchronize the received bit stream to a local clock in all modulation modes. The oversampled clock rate of the PLL (CDR CLK) must be set at 32 times the symbol rate (see the Register 3—Transmit/Receive Clock Register section). The maximum data/symbol rate tolerance of the CDR PLL is determined by the number of zero-crossing symbol transitions in the transmitted packet. For example, if using 2FSK with a 101010 preamble, a maximum tolerance of ±3.0% of the data rate is achieved. However, this tolerance is reduced during recovery of the remainder of the packet where symbol transi-tions may not be guaranteed to occur at regular intervals. To maximize the data rate tolerance of the CDR, some form of encoding and/or data scrambling is recommended that guarantees a number of transitions at regular intervals.

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For example, using 2FSK with Manchester-encoded data achieves a data rate tolerance of ±2.0%.

The CDR PLL is designed for fast acquisition of the recovered symbols during preamble and typically achieves bit synchro-nization within 5-symbol transitions of preamble.

In 4FSK modulation, the tolerance using the +3, −3, +3, −3 preamble is ±3% of the symbol rate (or ±1.5% of the data rate). However, this tolerance is reduced during recovery of the remainder of the packet where symbol transitions may not be guaranteed to occur at regular intervals. To maximize the symbol/data rate tolerance, the remainder of the 4FSK packet should be constructed so that the transmitted symbols retain close to dc-free properties by using data scrambling and/or by inserting specific dc balancing symbols that are inserted in the transmitted bit stream at regular intervals such as after every 8 or 16 symbols.

In 3FSK modulation, the linear convolutional encoder scheme guarantees that the transmitted symbol sequence is dc-free, facilitating symbol detection. However, Tx data scrambling is recommended to limit the run length of zero symbols in the transmit bit stream. Using 3FSK, the CDR data rate tolerance is typically ±0.5%.

RECEIVER SETUP Correlator Demodulator Setup

To enable the correlator for various modulation modes, refer to Table 16.

Table 16. Enabling the Correlator Demodulator Received Modulation DEMOD_SCHEME (R4_DB[4:6]) 2FSK 001 3FSK 010 4FSK 011

To optimize receiver sensitivity, the correlator bandwidth must be optimized for the specific deviation frequency and modulation used by the transmitter. The discriminator bandwidth is controlled by R4_DB[10:19] and is defined as

( )310400

_×

×=

KCLKDEMODBWTORDISCRIMINA

where: DEMOD CLK is as defined in the Register 3—Transmit/Receive Clock Register section. K is set for each modulation mode according to the following:

For 2FSK,

⎟⎟⎠

⎞⎜⎜⎝

⎛ ×=

DEVfRoundK

310100

For 3FSK,

⎟⎟⎠

⎞⎜⎜⎝

⎛

××

=DEVf

RoundK2

10100 3

For 4FSK,

⎟⎟⎠

⎞⎜⎜⎝

⎛××

=DEV

FSK fRoundK

410100 3

4

where: Round is rounded to the nearest integer. Round4FSK is rounded to the nearest of the following integers: 32, 31, 28, 27, 24, 23, 20, 19, 16, 15, 12, 11, 8, 7, 4, 3. fDEV is the transmit frequency deviation in Hz. For 4FSK, fDEV is the frequency deviation used for the ±1 symbols (that is, the inner frequency deviations).

To optimize the coefficients of the correlator, R4_DB7 and R4_DB[8:9] must also be assigned. The value of these bits depends on whether K is odd or even. These bits are assigned according to Table 17 and Table 18.

Table 17. Assignment of Correlator K Value for 2FSK and 3FSK K K/2 (K + 1)/2 R4_DB7 R4_DB[8:9] Even Even N/A 0 00 Even Odd N/A 0 10 Odd N/A Even 1 00 Odd N/A Odd 1 10

Table 18. Assignment of Correlator K Value for 4FSK K R4_DB7 R4_DB[8:9] Even 0 00 Odd 1 00

Linear Demodulator Setup

The linear demodulator can be used for 2FSK demodulation. To enable the linear demodulator, set the DEMOD_SCHEME bits (R4_DB[4:6]) to 000.

Post Demodulator Filter Setup

The 3 dB bandwidth of the post demodulator filter should be set according to the received modulation type and data rate. The bandwidth is controlled by R4_DB[20:29] and is given by

CLKDEMODf

BWDEMODPOST CUTOFF××=

π2__

11

where fCUTOFF is the target 3 dB bandwidth in Hz of the post demodulator filter.

Table 19. Post Demodulator Filter Bandwidth Settings for 2FSK/3FSK/4FSK Modulation Schemes Received Modulation

Post Demodulator Filter Bandwidth, fCUTOFF (Hz)

2FSK 0.75 × data rate 3FSK 1 × data rate 4FSK 1.6 × symbol rate (= 0.8 × data rate)

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3FSK Viterbi Detector Setup

The Viterbi detector can be used for 3FSK data detection. This is activated by setting R13_DB11 to Logic 1.

The Viterbi path memory length is programmable in steps of 4, 6, 8, or 32 bits (VITERBI_PATH_MEMORY, R13_DB[13:14]).

The path memory length should be set equal to or greater than the maximum number of consecutive 0s in the interleaved transmit bit stream.

The Viterbi detector also uses threshold levels to implement the maximum likelihood detection algorithm. These thresholds are programmable via the 3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]).

These bits are assigned as follows:

3FSK/4FSK_SLICER_THRESHOLD =

⎟⎟⎠

⎞⎜⎜⎝

⎛×

×× 310100

75KDeviationrequencyTransmit F

where K is the value calculated for correlator discriminator bandwidth.

3FSK Threshold Detector Setup

To activate threshold detection of 3FSK, R13_DB11 should be set to Logic 0. The 3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]) should be set as outlined in the 3FSK Viterbi Detector Setup section.

3FSK CDR Setup

In 3FSK, a transmit preamble of at least 40 bits of continuous 1s is recommended to ensure a maximum number of symbol transitions for the CDR to acquire lock.

The clock and data recovery for 3FSK requires a number of parameters in Register 13 to be set (see Table 20).

4FSK Threshold Detector Setup

The threshold for the 4FSK detector is set using the 3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]). The threshold should be set according to

3FSK/4FSK_SLICER_THRESHOLD =

⎟⎟⎠

⎞⎜⎜⎝

⎛×

×× 310100

87 KDeviationTxOuter4FSK


Table 20. 3FSK CDR Settings Parameter Recommended Setting Purpose PHASE_CORRECTION (R13_DB12) 1 Phase correction is on 3FSK_CDR_THRESHOLD (R13_DB[15:21])

⎟⎠⎞

⎜⎝⎛

××

× 31010062 KDeviationrequencyTransmit F


Sets CDR decision threshold levels

3FSK_PREAMBLE_TIME_VALIDATE (R13_DB [22:25]) 15 Preamble detector time qualifier

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DEMODULATOR CONSIDERATIONS 2FSK Preamble

The recommended preamble bit pattern for 2FSK is a dc-free pattern (such as a 10101010… pattern). Preamble patterns with longer run-length constraints (such as 11001100…) can also be used but result in a longer synchronization time of the received bit stream in the receiver. The preamble needs to allow enough bits for AGC settling of the receiver and CDR acquisition. A minimum of 16 preamble bits is recommended when using the correlator demodulator and 48 bits when using the linear demod-ulator. When the receiver uses the internal AFC, the minimum recommended number of preamble bits is 64.

The remaining fields that follow the preamble header do not have to use dc-free coding. For these fields, the ADF7021-N can accommodate coding schemes with a run length of greater than eight bits without any performance degradation. Refer to Application Note AN-915 for more information.

4FSK Preamble and Data Coding

The recommended preamble bit pattern for 4FSK is a repeating 00100010… bit sequence. This 2-level sequence of repeating −3, +3, −3, +3 symbols is dc-free and maximizes the symbol timing performance and data recovery of the 4FSK preamble in the receiver. The minimum recommended length of the preamble is 32 bits (16 symbols).

The remainder of the 4FSK packet should be constructed so that the transmitted symbols retain close to a dc-free balance by using data scrambling and/or by inserting specific dc balancing symbols in the transmitted bit stream at regular intervals, such as after every 8 or 16 symbols.

Demodulator Tolerance to Frequency Errors Without AFC

The ADF7021-N has a number of options to combat frequency errors that exist due to mismatches between the transmit and receive crystals/TCXOs.

With AFC disabled, the correlator demodulator is tolerant to frequency errors over a ±0.3 × fDEV range, where fDEV is the FSK frequency deviation. For larger frequency errors, the frequency tolerance can be increased by adjusting the value of K and thus doubling the correlator bandwidth.

K should then be calculated as

⎟⎟⎠

⎞⎜⎜⎝

⎛

××

=DEVf

RoundK2

10100 3

The DISCRIMINATOR_BW setting in Register 4 should also be recalculated using the new K value. Doubling the correlator bandwidth to improve frequency error tolerance in this manner typically results in a 1 dB to 2 dB loss in receiver sensitivity.

The linear demodulator (AFC disabled) tracks frequency errors in the receive signal when the receive signal is within the IF filter bandwidth. For example, for a receive signal with an occupied bandwith = 9 kHz, using the 18.5 kHz IF filter bandwidth allows the linear demodulator to track the signal at an error of ±4.75 kHz with no increase in bit errors or loss in sensitivity.

Correlator Demodulator and Low Modulation Indices

The modulation index in 2FSK is defined as

RateDataf

IndexModulation DEV×=

2

The receiver sensitivity performance and receiver frequency tolerance can be maximized at low modulation index by increasing the discriminator bandwidth of the correlator demodulator. For modulation indices of less than 0.4, it is recommended to double the correlator bandwidth by calculating K as follows:

⎟⎟⎠

⎞⎜⎜⎝

⎛×

=DEVf

RoundK2

1003

The DISCRIMINATOR_BW in Register 4 should be recalculated using the new K value. Figure 27 highlights the improved sensitivity that can be achieved for 2FSK modulation, at low modulation indices, by doubling the correlator bandwidth.

AFC OPERATION The ADF7021-N also supports a real-time AFC loop that is used to remove frequency errors due to mismatches between the transmit and receive crystals/TCXOs. The AFC loop uses the linear frequency discriminator block to estimate frequency errors. The linear FSK discriminator output is filtered and averaged to remove the FSK frequency modulation using a combined averaging filter and envelope detector. In receive mode, the output of the envelope detector provides an estimate of the average IF frequency.

Two methods of AFC supported on the ADF7021-N are external AFC and internal AFC.

External AFC

Here, the user reads back the frequency information through the ADF7021-N serial port and applies a frequency correction value to the fractional-N synthesizer-N divider.

The frequency information is obtained by reading the 16-bit signed AFC readback, as described in the Readback Format section, and by applying the following formula:

Frequency Readback [Hz] = (AFC READBACK × DEMOD CLK)/218

Although the AFC READBACK value is a signed number, under normal operating conditions, it is positive. In the absence of frequency errors, the frequency readback value is equal to the IF frequency of 100 kHz.

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Internal AFC When AFC errors are removed using either the internal or external AFC, further improvement in receiver sensitivity can be obtained by reducing the IF filter bandwidth using the IF_FILTER_BW bits (R4_DB[30:31]).

The ADF7021-N supports a real-time, internal, automatic frequency control loop. In this mode, an internal control loop automatically monitors the frequency error and adjusts the synthesizer-N divider using an internal proportional integral (PI) control loop.

AUTOMATIC SYNC WORD DETECTION (SWD) The ADF7021-N also supports automatic detection of the sync or ID fields. To activate this mode, the sync (or ID) word must be preprogrammed into the ADF7021-N. In receive mode, this preprogrammed word is compared to the received bit stream. When a valid match is identified, the external SWD pin is asserted by the ADF7021-N on the next Rx clock pulse.

The internal AFC control loop parameters are controlled in Register 10. The internal AFC loop is activated by setting R10_DB4 to 1. A scaling coefficient must also be entered, based on the crystal frequency in use. This is set up in R10_DB[5:16] and should be calculated using

This feature can be used to alert the microprocessor that a valid channel has been detected. It relaxes the computational requirements of the microprocessor and reduces the overall power consumption.

⎟⎟⎠

⎞⎜⎜⎝

⎛ ×=

XTALRoundFACTORSCALINGAFC

5002__

24

Maximum AFC Range

The maximum frequency correction range of the AFC loop is programmable on the ADF7021-N. This is set by R10_DB[24:31]. The maximum AFC correction range is the difference in frequency between the upper and lower limits of the AFC tuning range. For example, if the maximum AFC correction range is set to 10 kHz, the AFC can adjust the receiver LO within the fLO ± 5 kHz range.

The SWD signal can also be used to frame the received packet by staying high for a preprogrammed number of bytes. The data packet length can be set in R12_DB[8:15].

The SWD pin status can be configured by setting R12_DB[6:7]. R11_DB[4:5] are used to set the length of the sync/ID word, which can be 12, 16, 20, or 24 bits long. A value of 24 bits is recommended to minimize false sync word detection in the receiver that can occur during recovery of the remainder of the packet or when a noise/no signal is present at the receiver input. The transmitter must transmit the sync byte MSB first and the LSB last to ensure proper alignment in the receiver sync-byte-detection hardware.

However, when RF_DIVIDE_BY_2 (R1_DB18) is enabled, the programmed range is halved. The user should account for this halving by doubling the programmed maximum AFC range.

The recommended maximum AFC correction range should be ≤1.5 × IF filter bandwidth. If the maximum frequency correction range is set to be >1.5 × IF filter bandwidth, the attenuation of the IF filter can degrade the AFC loop sensitivity.

An error tolerance parameter can also be programmed that accepts a valid match when up to three bits of the word are incorrect. The error tolerance value is assigned in R11_DB[6:7].

The adjacent channel rejection (ACR) performance of the receivers can be degraded when AFC is enabled and the AFC correction range is close to the IF filter bandwidth. However, because the AFC correction range is programmable, the user can trade off correction range and ACR performance.

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APPLICATIONS INFORMATION IF FILTER BANDWIDTH CALIBRATION The IF filter should be calibrated on every power-up in receive mode to correct for errors in the bandwidth and filter center frequency due to process variations. The automatic calibration requires no external intervention once it is initiated by a write to Register 5. Depending on numerous factors, such as IF filter bandwidth, received signal bandwidth, and temperature variation, the user must determine whether to carry out a coarse calibration or a fine calibration.

The performance of both calibration methods is outlined in Table 21.

Table 21. IF Filter Calibration Specifications Filter Calibration Method

Center Frequency Accuracy1

Calibration Time (Typ)

Coarse Calibration 100 kHz ± 2.5 kHz 200 μs Fine Calibration 100 kHz ± 0.6 kHz 8.2 ms

1 After calibration.

Calibration Setup

IF Filter calibration is initiated by writing to Register 5 and setting the IF_CAL_COARSE bit (R5_DB4). This initiates a coarse filter calibration. If the IF_FINE_CAL bit (R6_DB4) has already been configured high, the coarse calibration is followed by a fine calibration, otherwise the calibration ends.

Once initiated by writing to the part, the calibration is performed automatically without any user intervention. Calibration time is 200 μs for coarse calibration and a few milliseconds for fine calibration, during which time the ADF7021-N should not be accessed. The IF filter calibration logic requires that the IF_FILTER_DIVIDER bits (R5_DB[5:13]) be set such that

kHz50__[Hz]

=DIVIDERFILTERIF

XTAL

The fine calibration uses two internally generated tones at certain offsets around the IF filter. The two tones are attenuated by the IF filter, and the level of this attenuation is measured using the RSSI. The filter center frequency is adjusted to allow equal attenuation of both tones. The attenuation of the two test tones is then remeasured. This continues for a maximum of 10 RSSI measurements, at which stage the calibration algorithm sets the IF filter center frequency to within 0.6 kHz of 100 kHz.

The frequency of these tones is set by the IF_CAL_LOWER_ TONE_DIVIDE (R6_DB[5:12]) and IF_CAL_UPPER_TONE_ DIVIDE (R6_DB[13:20]) bits, outlined in the following equations:

Lower Tone Frequency (kHz)

2VIDEER_TONE_DIIF_CAL_LOW

XTAL

×

Upper Tone Frequency (kHz)

2VIDEER_TONE_DIIF_CAL_UPP

XTAL

×

It is recommended to place the lower tone and upper tone as outlined in Table 22.

Table 22. IF Filter Fine Calibration Tone Frequencies IF Filter Bandwidth

Lower Tone Frequency

Upper Tone Frequency

9 kHz 78.1 kHz 116.3 kHz 13.5 kHz 79.4 kHz 116.3 kHz 18.5 kHz 78.1 kHz 119 kHz

Because the filter attenuation is slightly asymmetrical, it is necessary to have a small offset in the filter center frequency to give near equal rejection at the upper and lower adjacent channels. The calibration tones given in Table 22 give this small positive offset in the IF filter center frequency.

In some applications, an offset may not be required, and the user may wish to center the IF filter exactly at 100 kHz. In this case, the user can alter the tone frequencies from those given in Table 22 to adjust the fine calibration result.

The calibration algorithm adjusts the filter center frequency and measures the RSSI 10 times during the calibration. The time for an adjustment plus RSSI measurement is given by

CLKSEQLL_TIMEIF_CAL_DWETimenCalibratioToneIF =

It is recommended that the IF tone calibration time be at least 800 μs. The total time for the IF filter fine calibration is given by

IF Filter Fine Calibration Time = IF Tone Calibration Time × 10

When to Use Coarse Calibration

It is recommended to perform a coarse calibration on every receive mode power-up. This calibration typically takes 200 μs. The FILTER_CAL_COMPLETE signal from MUXOUT can be used to monitor the filter calibration duration or to signal the end of calibration. The ADF7021-N should not be accessed during calibration.

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When to Use a Fine Calibration

In cases where the receive signal bandwidth is very close to the bandwidth of the IF filter, it is recommended to perform a fine filter calibration every time the unit powers up in receive mode.

A fine calibration should be performed if

OBW + Coarse Calibration Variation > IF_FILTER_BW

where: OBW is the 99% occupied bandwidth of the transmit signal. Coarse Calibration Variation is 2.5 kHz. IF_FILTER_BW is set by R4_DB[30:31].

The FILTER_CAL_COMPLETE signal from MUXOUT (set by R0_DB[29:31]) can be used to monitor the filter calibration duration or to signal the end of calibration. A coarse filter calibration is automatically performed prior to a fine filter calibration.

When to Use Single Fine Calibration

In applications where the receiver powers up numerous times in a short period, it is only necessary to perform a one-time fine calibration on the initial receiver power-up.

After the initial coarse calibration and fine calibration, the result of the fine calibration can be read back through the serial interface using the FILTER_CAL_READBACK result (refer to the Filter Bandwidth Calibration Readback section). On subsequent power-ups in receive mode, the filter is manually adjusted using the previous fine filter calibration result. This manual adjust is performed using the IF_FILTER_ADJUST bits (R5_DB[14:19]).

This method should only be used if the successive power-ups in receive mode are over a short duration, during which time there is little variation in temperature (<15°C).

IF Filter Variation with Temperature

When calibrated, the filter center frequency can vary with changes in temperature. If the ADF7021-N is used in an application where it remains in receive mode for a considerable length of time, the user must consider this variation of filter center frequency with temperature. This variation is typically 1 kHz per 20°C, which means that if a coarse filter calibration and fine filter calibration are performed at 25°C, the initial maximum error is ±0.5 kHz, and the maximum possible change in the filter center frequency over temperature (−40°C to +85°C) is ±3.25 kHz. This gives a total error of ±3.75 kHz.

If the receive signal occupied bandwidth is considerably less than the IF filter bandwidth, the variation of filter center frequency over the operating temperature range may not be an issue. Alternatively, if the IF filter bandwidth is not wide enough to tolerate the variation with temperature, a periodic filter calibration can be performed or, alternatively, the on-chip temperature sensor can be used to determine when a filter cali-bration is necessary by monitoring for changes in temperature.

LNA/PA MATCHING The ADF7021-N exhibits optimum performance in terms of sensitivity, transmit power, and current consumption, only if its RF input and output ports are properly matched to the antenna impedance. For cost-sensitive applications, the ADF7021-N is equipped with an internal Rx/Tx switch that facilitates the use of a simple, combined passive PA/LNA matching network. Alternatively, an external Rx/Tx switch such as the ADG919 can be used, which yields a slightly improved receiver sensitivity and lower transmitter power consumption.

Internal Rx/Tx Switch

Figure 50 shows the ADF7021-N in a configuration where the internal Rx/Tx switch is used with a combined LNA/PA matching network. This is the configuration used on the EVAL-ADF7021-NDBxx evaluation board. For most applications, the slight performance degradation of 1 dB to 2 dB caused by the internal Rx/Tx switch is acceptable, allowing the user to take advantage of the cost saving potential of this solution. The design of the combined matching network must compensate for the reactance presented by the networks in the Tx and the Rx paths, taking the state of the Rx/Tx switch into consideration.

PA

LNA

PA_OUT

RFIN

RFINB

VBAT

L1

ADF7021-N

OPTIONALBPF OR LPF

LA

CA

C1

CB

ZIN_RFIN

ZOPT_PA

ZIN_RFIN

ANTENNA

0724

6-02

2

Figure 50. ADF7021-N with Internal Rx/Tx Switch

The procedure typically requires several iterations until an acceptable compromise has been reached. The successful imple-mentation of a combined LNA/PA matching network for the ADF7021-N is critically dependent on the availability of an accurate electrical model for the PCB. In this context, the use of a suitable CAD package is strongly recommended. To avoid this effort, a small form-factor reference design for the ADF7021-N is provided, including matching and harmonic filter components. The design is on a 2-layer PCB to minimize cost. Gerber files are available at www.analog.com.

http://www.analog.com/ADG919

http://www.analog.com/

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External Rx/Tx Switch Depending on the antenna configuration, the user may need a harmonic filter at the PA output to satisfy the spurious emission requirement of the applicable government regulations. The harmonic filter can be implemented in various ways, for example, a discrete LC pi or T-stage filter. The immunity of the ADF7021-N to strong out-of-band interference can be improved by adding a band-pass filter in the Rx path. Alternatively, the ADF7021-N blocking performance can be improved by selecting one of the enhanced linearity modes, as described in Table 15.

Figure 51 shows a configuration using an external Rx/Tx switch. This configuration allows an independent optimization of the matching and filter network in the transmit and receive path. Therefore, it is more flexible and less difficult to design than the configuration using the internal Rx/Tx switch. The PA is biased through Inductor L1, while C1 blocks dc current. Together, L1 and C1 form the matching network that transforms the source impedance into the optimum PA load impedance, ZOPT_PA.

IMAGE REJECTION CALIBRATION

PA

LNA

PA_OUT

RFIN

RFINB

VBAT

L1

ADF7021-N

ADG919

OPTIONALBPF

(SAW)

OPTIONALLPF

LA

CA

C1

CB

ZIN_RFIN

ZOPT_PA

ZIN_RFIN

ANTENNA

Rx/Tx – SELECT

0724

6-02

1

The image channel in the ADF7021-N is 200 kHz below the desired signal. The polyphase filter rejects this image with an asymmetric frequency response. The image rejection performance of the receiver is dependent on how well matched the I and Q signals are in amplitude and how well matched the quadrature is between them (that is, how close to 90° apart they are). The uncalibrated image rejection performance is approximately 29 dB (at 450 MHz). However, it is possible to improve on this performance by as much as 20 dB by finding the optimum I/Q gain and phase adjust settings.

Calibration Using Internal RF Source Figure 51. ADF7021-N with External Rx/Tx Switch

With the LNA powered off, an on-chip generated, low level RF tone is applied to the mixer inputs. The LO is adjusted to make the tone fall at the image frequency where it is attenuated by the image rejection of the IF filter. The power level of this tone is then measured using the RSSI readback. The I/Q gain and phase adjust DACs (R5_DB[20:31]) are adjusted and the RSSI is remeasured. This process is repeated until the optimum values for the gain and phase adjust are found that provide the lowest RSSI readback level, thereby maximizing the image rejection performance of the receiver.

ZOPT_PA depends on various factors, such as the required output power, the frequency range, the supply voltage range, and the temperature range. Selecting an appropriate ZOPT_PA helps to minimize the Tx current consumption in the application. Application Note AN-764 and Application Note AN-859 contain a number of ZOPT_PA values for representative conditions. Under certain conditions, however, it is recommended to obtain a suitable ZOPT_PA value by means of a load-pull measurement.

Due to the differential LNA input, the LNA matching network must be designed to provide both a single-ended-to-differential conversion and a complex, conjugate impedance match. The network with the lowest component count that can satisfy these requirements is the configuration shown in Figure 51, consisting of two capacitors and one inductor.

ADF7021-N

Rev. 0 | of 64

INTERNALSIGNALSOURCE

MUX

RFIN

RFINBLNA

ADF7021-N

POLYPHASEIF FILTER

PHASE ADJUST

GA

IN A

DJU

ST

I Q

FROM LO

GAIN ADJUSTREGISTER 5

PHASE ADJUSTREGISTER 5

SERIALINTERFACE

4

MICROCONTROLLER

4

RSSI/LOG AMP

7-BIT ADC

RSSI READBACK

I/Q GAIN/PHASE ADJUST ANDRSSI MEASUREMENT

ALGORITHM

0724

6-07

2

Figure 52. Image Rejection Calibration Using the Internal Calibration Source and a Microcontroller

Using the internal RF source, the RF frequencies that can be used for image calibration are programmable and are odd multiples of the reference frequency.

Calibration Using External RF Source

IR calibration can also be implemented using an external RF source. The IR calibration procedure is the same as that used for the internal RF source, except that an RF tone is applied to the LNA input.

Calibration Procedure and Setup

The IR calibration algorithm available from Analog Devices, Inc., is based on a low complexity, 2D optimization algorithm that can be implemented in an external microprocessor or microcontroller.

To enable the internal RF source, the IR_CAL_SOURCE_ DRIVE_LEVEL bits (R6_DB[28:29]) should be set to the maximum level. The LNA should be set to its minimum gain setting, and the AGC should be disabled if the internal source is being used. Alternatively, an external RF source can be used.

The magnitude of the phase adjust is set by using the IR_PHASE_ ADJUST_MAG bits (R5_DB[20:23]). This correction can be applied to either the I channel or Q channel, depending on the value of the IR_PHASE_ADJUST_DIRECTION bit (R5_DB24).

The magnitude of the I/Q gain is adjusted by the IR_GAIN_ ADJUST_MAG bits (R5_DB[25:29]). This correction can be applied to either the I or Q channel, depending on the value of

IR_GAIN_ADJUST_I/Q bit (R5_DB30), whereas the IR_GAIN_ADJUST_UP/DN bit (R5_DB31) sets whether the gain adjustment defines a gain or an attenuation adjust.

The calibration results are valid over changes in the ADF7021-N supply voltage. However, there is some variation with temperature. A typical plot of variation in image rejection over temperature after initial calibrations at −40°C, +25°C, and +85°C is shown in Figure 53. The internal temperature sensor on the ADF7021-N can be used to determine if a new IR calibration is required.

0

10

20

30

40

50

60

–60 –40 –20 0 20 40 60 80 100

VDD = 3.0VIF BW = 25kHz

WANTED SIGNAL:RF FREQ = 430MHzMODULATION = 2FSKDATA RATE = 9.6kbps, PRBS9fDEV = 4kHzLEVEL= –100dBm

INTERFERER SIGNAL:RF FREQ = 429.8MHzMODULATION = 2FSKDATA RATE = 9.6kbps, PRBS11fDEV = 4kHz

TEMPERATURE (°C)

IMA

GE

REJ

ECTI

ON

(dB

)

CAL AT +25°C

CAL AT +85°CCAL AT –40°C

0724

6-06

7

Figure 53. Image Rejection Variation with Temperature After Initial

Calibrations at −40°C, +25°C, and +85°C

ADF7021-N

Rev. 0 | of 64

PACKET STRUCTURE AND CODING The suggested packet structure to use with the ADF7021-N is shown in Figure 54.

PREAMBLESYNCWORD

IDFIELD DATA FIELD CRC

0724

6-02

3

Figure 54. Typical Format of a Transmit Protocol

Refer to the Receiver Setup section for information on the required preamble structure and length for the various modulation schemes.

PROGRAMMING AFTER INITIAL POWER-UP Table 23 lists the minimum number of writes needed to set up the ADF7021-N in either Tx or Rx mode after CE is brought high. Additional registers can also be written to tailor the part

to a particular application, such as setting up sync byte detection or enabling AFC. When going from Tx to Rx or vice versa, the user needs to toggle the Tx/Rx bit and write only to Register 0 to alter the LO by 100 kHz.

Table 23. Minimum Register Writes Required for Tx/Rx Setup Mode Registers Tx Reg 1 Reg 3 Reg 0 Reg 2 Rx Reg 1 Reg 3 Reg 0 Reg 5 Reg 4 Tx to Rx and Rx to Tx Reg 0

The recommended programming sequences for transmit and receive are shown in Figure 55 and Figure 56, respectively. The difference in the power-up routine for a TCXO and XTAL reference is shown in these figures.

ADF7021-N

Rev. 0 | of 64

POWER-DOWNCE LOW

XTALREFERENCE

TCXOREFERENCE

CE HIGHWAIT 10µs (REGULATOR POWER-UP)

WRITE TO REGISTER 1 (TURNS ON VCO)WAIT 0.7ms (TYPICAL VCO SETTLING)

WRITE TO REGISTER 0 (TURNS ON PLL)WAIT 40µs (TYPICAL PLL SETTLING)

WRITE TO REGISTER 2 (TURNS ON PA)WAIT FOR PA TO RAMP UP (ONLY IF PA RAMP ENABLED)

WAIT FOR Tx LATENCY NUMBER OF BITS(REFER TO TABLE 12)

WRITE TO REGISTER 2 (TURNS OFF PA)WAIT FOR PA TO RAMP DOWN

WRITE TO REGISTER 3 (TURNS ON Tx/Rx CLOCKS)

CE HIGHWAIT 10µs + 1ms

(REGULATOR POWER-UP + TYPICAL XTAL SETTLING)

CE LOWPOWER-DOWN

Tx MODE

OPTIONAL. ONLY NECESSARY IF PARAMP DOWN IS REQUIRED. 07

246-

086

Figure 55. Power-Up Sequence for Transmit Mode

ADF7021-N

Rev. 0 | of 64

POWER-DOWNCE LOW

WRITE TO REGISTER 5 (STARTS IF FILTER CALIBRATION)WAIT 0.2ms (COARSE CAL) OR WAIT 8.2ms

(COARSE CALIBRATION + FINE CALIBRATION)

WRITE TO REGISTER 11 (SET UP SWD)WRITE TO REGISTER 12 (ENABLE SWD)

WRITE TO REGISTER 6 (SETS UP IF FILTER CALIBRATION)

CE LOWPOWER-DOWN

Rx MODE

WRITE TO REGISTER 3 (TURNS ON Tx/Rx CLOCKS)

WRITE TO REGISTER 4 (TURNS ON DEMOD)

WRITE TO REGISTER 10 (TURNS ON AFC)

OPTIONAL.

WRITE TO REGISTER 0 (TURNS ON PLL)WAIT 40µs (TYPICAL PLL SETTLING)

CE HIGHWAIT 10µs (REGULATOR POWER-UP)

WRITE TO REGISTER 1 (TURNS ON VCO)WAIT 0.7ms (TYPICAL VCO SETTLING)

CE HIGHWAIT 10µs + 1ms

(REGULATOR POWER-UP + TYPICAL XTAL SETTLING)

XTALREFERENCE

TCXOREFERENCE

OPTIONAL:ONLY NECESSARY IFAFC IS REQUIRED.

OPTIONAL:ONLY NECESSARY IFSWD IS REQUIRED.

OPTIONAL:ONLY NECESSARY IFIF FILTER FINE CAL IS REQUIRED.

0724

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7

Figure 56. Power-Up Sequence for Receive Mode

ADF7021-N

Rev. 0 | of 64

APPLICATIONS CIRCUIT The ADF7021-N requires very few external components for operation. Figure 57 shows the recommended application circuit. Note that the power supply decoupling and regulator capacitors are omitted for clarity.

For recommended component values, refer to the ADF7021-N evaluation board data sheet and AN-859 application note accessible from the ADF7021-N product page. Follow the reference design schematic closely to ensure optimum performance in narrow-band applications.

48 47 46 45 44 43 42 41 40 39 38 37ADF7021-N

VCOINCREG1VDD1RFOUTRFGNDRFINRFINBRLNAVDD4

RSETCREG4GND4

CVC

OG

ND

1 L1G

ND L2

VDD

CPO

UT

CR

EG3

VDD

3O

SC1

OSC

2M

UXO

UT

1

2

3

4

5

6

7

8

9

10

11

12

13 14 15 16 17 18 19 20 21 22 23 24

MIX

_IM

IX_I

MIX

_QM

IX_Q

FILT

_IFI

LT_I

GN

D4

FILT

_QFI

LT_Q

GN

D4

TEST

_AC

E

CLKOUT

TxRxDATA TxRxCLK

SWDVDD2

CREG2ADCINGND2SCLK

SREADSDATA

SLE

36

35

34

33

32

31

30

29

28

27

26

25

VDD

VDD

VDD

TCXO

VDD

VDD

VDDANTENNA

CONNECTION

TOMICROCONTROLLERTx/Rx SIGNALINTERFACE

TOMICROCONTROLLERCONFIGURATIONINTERFACE

T-STAGE LCFILTER

MATCHING

LOOP FILTER

CVCOCAP

EXT VCO L*

REFERENCE

RSETRESISTOR

RLNARESISTOR

CHIP ENABLETO MICROCONTROLLER

*PIN 44 AND PIN 46 CAN BE LEFT FLOATING IF EXTERNAL INDUCTOR VCO IS NOT USED.

NOTES1. PINS [13:18], PINS [20:21], AND PIN 23 ARE TEST PINS AND ARE NOT USED IN NORMAL OPERATION. 07

246-

084

Figure 57. Typical Application Circuit (Regulator Capacitors and Power Supply Decoupling Not Shown)

ADF7021-N

Rev. 0 | of 64

SERIAL INTERFACE The serial interface allows the user to program the 16-/32-bit registers using a 3-wire interface (SCLK, SDATA, and SLE). It consists of a level shifter, 32-bit shift register, and 16 latches. Signals should be CMOS compatible. The serial interface is powered by the regulator, and, therefore, is inactive when CE is low.

Data is clocked into the register, MSB first, on the rising edge of each clock (SCLK). Data is transferred to one of 16 latches on the rising edge of SLE. The destination latch is determined by the value of the four control bits (C4 to C1); these are the bottom 4 LSBs, DB3 to DB0, as shown in Figure 2. Data can also be read back on the SREAD pin.

READBACK FORMAT The readback operation is initiated by writing a valid control word to the readback register and enabling the READBACK bit (R7_DB8 = 1). The readback can begin after the control word has been latched with the SLE signal. SLE must be kept high while the data is being read out. Each active edge at the SCLK pin successively clocks the readback word out at the SREAD pin, as shown in Figure 58, starting with the MSB first. The data appearing at the first clock cycle following the latch operation must be ignored. An extra clock cycle is needed after the 16th readback bit to return the SREAD pin to tristate. Therefore, 18 total clock cycles are needed for each read back. After the 18th clock cycle, SLE should be brought low.

AFC Readback

The AFC readback is valid only during the reception of FSK signals with either the linear or correlator demodulator active. The AFC readback value is formatted as a signed 16-bit integer comprising Bit RV1 to Bit RV16 and is scaled according to the following formula:

FREQ RB [Hz] = (AFC_READBACK × DEMOD CLK)/218

In the absence of frequency errors, FREQ RB is equal to the IF frequency of 100 kHz. Note that, for the AFC readback to yield a valid result, the downconverted input signal must not fall outside the bandwidth of the analog IF filter. At low input signal levels, the variation in the readback value can be improved by averaging.

RSSI Readback

The format of the readback word is shown in Figure 58. It comprises the RSSI-level information (Bit RV1 to Bit RV7), the current filter gain (FG1, FG2), and the current LNA gain (LG1, LG2) setting. The filter and LNA gain are coded in accordance with the definitions in the Register 9—AGC Register section. For signal levels below −100 dBm, averaging the measured RSSI values improves accuracy. The input power can be calculated from the RSSI readback value as outlined in the RSSI/AGC section.

READBACK MODE

AFC READBACK

DB15

RV16

X

X

RV16

0

RSSI READBACK

BATTERY VOLTAGE/ADCIN/TEMP. SENSOR READBACK

SILICON REVISION

FILTER CAL READBACK

READBACK VALUE

DB14

RV15

X

X

RV15

0

DB13

RV14

X

X

RV14

0

DB12

RV13

X

X

RV13

0

DB11

RV12

X

X

RV12

0

DB10

RV11

LG2

X

RV11

0

DB9

RV10

LG1

X

RV10

0

DB8

RV9

FG2

X

RV9

0

DB7

RV8

FG1

X

RV8

RV8

DB6

RV7

RV7

RV7

RV7

RV7

DB5

RV6

RV6

RV6

RV6

RV6

DB4

RV5

RV5

RV5

RV5

RV5

DB3

RV4

RV4

RV4

RV4

RV4

DB2

RV3

RV3

RV3

RV3

RV3

DB1

RV2

RV2

RV2

RV2

RV2

DB0

RV1

RV1

RV1

RV1

RV1

0724

6-02

9

Figure 58. Readback Value Table

ADF7021-N

Rev. 0 | of 64

Battery Voltage/ADCIN/Temperature Sensor Readback

The battery voltage is measured at Pin VDD4. The readback information is contained in Bit RV1 to Bit RV7. This also applies to the readback of the voltage at the ADCIN pin and the temperature sensor. From the readback information, the battery or ADCIN voltage can be determined using

VBATTERY = (BATTERY VOLTAGE READBACK)/21.1

VADCIN = (ADCIN VOLTAGE READBACK)/42.1

The temperature can be calculated using

Temp [°C] = −40 + (68.4 − TEMP READBACK) × 9.32

Silicon Revision Readback

The silicon revision readback word is valid without setting any other registers. The silicon revision word is coded with four quartets in BCD format. The product code (PC) is coded with three quartets extending from Bit RV5 to Bit RV16. The revision code (RC) is coded with one quartet extending from Bit RV1 to Bit RV4. The product code for the ADF7021-N should read back as PC = 0x211. The current revision code should read as RC = 0x1.

Filter Bandwidth Calibration Readback

The filter calibration readback word is contained in Bit RV1 to Bit RV8 (see Figure 58). This readback can be used for manual filter adjust, thereby avoiding the need to do an IF filter calibration in some instances. The manual adjust value is programmed by R5_DB[14:19]. To calculate the manual adjust based on a filter calibration readback, use the following formula:

IF_FILTER_ADJUST = FILTER_CAL_READBACK − 128

The result should be programmed into R5_DB[14:19] as outlined in the Register 5—IF Filter Setup Register section.

ADF7021-N

Rev. 0 | of 64

INTERFACING TO A MICROCONTROLLER/DSP Standard Transmit/Receive Data Interface

The standard transmit/receive signal and configuration interface to a microcontroller is shown in Figure 59. In transmit mode, the ADF7021-N provides the data clock on the TxRxCLK pin, and the TxRxDATA pin is used as the data input. The transmit data is clocked into the ADF7021-N on the rising edge of TxRxCLK.

MISO

ADuC84x ADF7021-N

MOSISCLOCK

SSP3.7

P3.2/INT0P2.4P2.5

TxRxDATA

TxRxCLK

CESWDSREADSLE

P2.6P2.7

SDATASCLK

GPIO07

246-

026

Figure 59. ADuC84x to ADF7021-N Connection Diagram

In receive mode, the ADF7021-N provides the synchronized data clock on the TxRxCLK pin. The receive data is available on the TxRxDATA pin. The rising edge of TxRxCLK should be used to clock the receive data into the microcontroller. Refer to Figure 4 and Figure 5 for the relevant timing diagrams.

In 4FSK transmit mode, the MSB of the transmit symbol is clocked into the ADF7021-N on the first rising edge of the data clock from the TxRxCLK pin. In 4FSK receive mode, the MSB of the first payload symbol is clocked out on the first negative edge of the data clock after the SWD and should be clocked into the microcontroller on the following rising edge. Refer to Figure 6 and Figure 7 for the relevant timing diagrams.

UART Mode

In UART mode, the TxRxCLK pin is configured to input transmit data in transmit mode. In receive mode, the receive data is available on the TxRxDATA pin, thus providing an asynchronous data interface. The UART mode can only be used with oversampled 2FSK. Figure 60 shows a possible interface to a microcontroller using the UART mode of the ADF7021-N. To enable this UART interface mode, set R0_DB28 high. Figure 8 and Figure 9 show the relevant timing diagrams for UART mode.

UART

ADF7021-NTxRxCLKTxRxDATA

TxDATARxDATA

CESWDSREADSLESDATASCLK

GPIO

MICROCONTROLLER

0724

6-08

5

Figure 60. ADF7021-N (UART Mode) to

Asynchronous Microcontroller Interface

SPI Mode

In SPI mode, the TxRxCLK pin is configured to input transmit data in transmit mode. In receive mode, the receive data is available on the TxRxDATA pin. The data clock in both transmit and receive modes is available on the CLKOUT pin. In transmit mode, data is clocked into the ADF7021-N on the positive edge of CLKOUT. In receive mode, the TxRxDATA data pin should be sampled by the microcontroller on the positive edge of the CLKOUT.

SPI

ADF7021-NTxRxCLK

TxRxDATAMISO

MOSI

CESWDSREADSLESDATASCLK

GPIO

MICROCONTROLLER

SCLK CLKOUT

0724

6-07

6

Figure 61. ADF7021-N (SPI Mode) to Microcontroller Interface

To enable SPI interface mode, set R0_DB28 high and set R15_DB[17:19] to 0x7. Figure 8 and Figure 9 show the relevant timing diagrams for SPI mode, while Figure 61 shows the recommended interface to a microcontroller using the SPI mode of the ADF7021-N.

ADSP-BF533 interface

The suggested method of interfacing to the Blackfin® ADSP-BF533 is given in Figure 62.

MOSI

ADSP-BF533 ADF7021-N

MISOPF5

RSCLK1DT1PRIDR1PRI

RFS1PF6

SDATA

SLE

TxRxDATA

SWDCE

SCK SCLK

SREAD

TxRxCLK

0724

6-02

7

Figure 62. ADSP-BF533 to ADF7021-N Connection Diagram

ADF7021-N

Rev. 0 | of 64

REGISTER 0—N REGISTER

TR1 Tx/Rx0 TRANSMIT

RECEIVE1

M3 M2 M1 MUXOUT0 REGULATOR_READY (DEFAULT)

FILTER_CAL_COMPLETE00 DIGITAL_LOCK_DETECT0 RSSI_READY1 Tx_Rx1 LOGIC_ZERO1 TRISTATE1

00110011

01010101 LOGIC_ONE

U1 UART_MODE0 DISABLED1 ENABLED

N8 N7 N6 N5 N4 N3 N2 N10 230 24...1 253

1 254

1

00...1

1

1

00...

.

.

.1

1

1

11

1

1

1

.

.

.

01

1

1

1

.

.

.

10

1

1

1

.

.

.

10

0

1

1

.

.

.

.

.

.

10

1

0

1 255

FRACTIONAL_NINTEGER_NTx/R

x

UA

RT_

MO

DE

MUXOUTADDRESS

BITS

N5

N4

N8

M5

M6

M7

M8

M12

M13

M15N1

N2

N3

M14 M9

M10

M11 M4

M3

TR1

U1

M1

M3

M2

C2

(0)

C1

(0)

C3

(0)

C4

(0)

M1

M2

N7

N6

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

DB

24

DB

26

DB

27

DB

28

DB

25

DB

1

DB

0

DB

2

DB

3

DB

29

DB

30

DB

31

FRACTIONAL_NDIVIDE RATIO012...32764327653276632767

M15000...1111

M14000...1111

M13000...1111

...

...

...

...

...

...

...

...

...

...

...

M3000...1111

M2001...0011

M1010...0101

0724

6-03

0

INTEGER_NDIVIDE RATIO

Figure 63. Register 0—N Register Map

• The RF output frequency is calculated by the following:

For the direct output

⎟⎠⎞

⎜⎝⎛ +×= 152

__ NFractionalNIntegerPFDRFOUT

For the RF_DIVIDE_BY_2 (R1_DB18) selected

⎟⎠⎞

⎜⎝⎛ +××= 152

__5.0 NFractionalNIntegerPFDRFOUT

• In UART/SPI mode, the TxRxCLK pin is used to input the Tx data. The Rx Data is available on the TxRxDATA pin.

• FILTER_CAL_ COMPLETE in the MUXOUT map in Figure 63 indicates when a coarse or coarse plus fine IF filter calibration has finished. DIGITAL_ LOCK_DETECT indicates when the PLL has locked. RSSI_READY indicates that the RSSI signal has settled and an RSSI readback can be performed.

• Tx_Rx gives the status of DB27 in this register, which can be used to control an external Tx/Rx switch.

ADF7021-N

Rev. 0 | of 64

REGISTER 1—VCO/OSCILLATOR REGISTER

R3 R2 R100...1

12...7

10...1

01...1

X1 XOSC_ENABLE0 OFF1 ON

VA2 VA1VCO CENTERFREQ ADJUST

0 NOMINAL0 VCO ADJUST UP 11 VCO ADJUST UP 21

0101 VCO ADJUST UP 3

D1XTAL_DOUBLER

0 DISABLEENABLED1

CP2CP1RSET

ICP (mA)3.6kΩ

0 0 0.30 1 0.91 0 1.51 1 2.1

VB4 VB3 VB2 VB1VCO_BIASCURRENT

0 0.25mA0 0.5mA.1

10.1

01.1

00.1 3.75mA

CL4 CL3 CL2 CL1CLKOUT_DIVIDE RATIO

0 OFF00...1

010...1

24...

001...1

000...1 30

VCO_BIAS CP_

CU

RR

ENT

RF_

DIV

IDE_

B

Y_2

XOSC

_EN

AB

LE

VCO

_EN

AB

LE ADDRESSBITSXT

AL_

DO

UB

LER

XTAL_BIASVC

O_

AD

JUST

VA1

VB4

CL1

CL2

CL3

CL4

CP1

CP2

RFD

1

VB1

VB2

VB3

VE1

X1XB1

XB2

D1

R3

C2

(0)

C1

(1)

C3

(0)

C4

(0)

R1

R2

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

VA2

DB

24

DB

1

DB

0

DB

2

DB

3

XB2 XB1XTAL_BIAS

0 20µA0 25µA1 30µA1

0101 35µA

RFD1 RF_DIVIDE_BY_20 OFF

ON1

LOOPCONDITIONVCO OFFVCO ON

VE101

DB

25VC

L1 V

CO

_IN

DU

CTO

R

VCO_INDUCTORINTERNAL L VCOEXTERNAL L VCO

VCL101

R_COUNTERCLKOUT_DIVIDE

0724

6-03

1

RF R_COUNTERDIVIDE RATIO

Figure 64. Register 1—VCO/Oscillator Register Map

• The R_COUNTER and XTAL_DOUBLER relationship is as follows:

If XTAL_DOUBLER = 0, COUNTERRXTALPFD

_=

If XTAL_DOUBLER =1, COUNTERRXTALPFD

_2×

=

• CLOCKOUT_DIVIDE is a divided-down and inverted version of the XTAL and is available on Pin 36 (CLKOUT).

• Set XOSC_ENABLE high when using an external crystal. If using an external oscillator (such as TCXO) with CMOS-level outputs into Pin OSC2, set XOSC_ENABLE low. If using an external oscillator with a 0.8 V p-p clipped sine wave output into Pin OSC1, set XOSC_ENABLE high.

• The VCO_BIAS bits should be set according to Table 9.

• The VCO_ADJUST bits adjust the center of the VCO operating band. Each bit typically adjusts the VCO band up by 1% of the RF operating frequency (0.5% if RF_DIVIDE_BY_2 is enabled).

• Setting VCO_INDUCTOR to external allows the use of the external inductor VCO, which gives RF operating frequencies of 80 MHz to 650 MHz. If the internal inductor VCO is being used for operation, set this bit low.

ADF7021-N

Rev. 0 | of 64

REGISTER 2—TRANSMIT MODULATION REGISTER

P60000..1

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.1

P20011..1

P10101..1

0 (PA OFF)1 (–16.0 dBm)23.. 63 (13 dBm)

TFD90000.1

TFD30000.1

...

...

...

...

...

...

...

TFD20011.1

TFD10101.1

0123.511

Tx_FREQUENCY_DEVIATION POWER_AMPLIFIERTxDATA_INVERT PA_BIAS PA_RAMP

MODULATION_SCHEME

ADDRESSBITSPA

_EN

AB

LE

PE101

PA_ENABLEDOFFON

PA20011

PA10101

PA_BIAS5µA7µA9µA11µA

DI20011

DI10101

TxDATA_INVERTNORMALINVERT CLKINVERT DATAINV CLK AND DATA

S300001111

S200110011

MODULATION_SCHEME2FSKGAUSSIAN 2FSK3FSK4FSKOVERSAMPLED 2FSKRAISED COSINE 2FSKRAISED COSINE 3FSKRAISED COSINE 4FSK

S101010101

PR300001111

PR200110011

NO RAMP256 CODES/BIT128 CODES/BIT64 CODES/BIT32 CODES/BIT16 CODES/BIT8 CODES/BIT4 CODES/BIT

PR1 PA_RAMP RATE01010101

TFD

5

TFD

4

TFD

8

PR1

PR2

PR3

PA1

P3P4P6TFD

1

TFD

2

TFD

3

P5 PA2

P1P2 PE1

S3TFD

9

DI1

DI2

C2

(1)

C1

(0)

C3

(0)

C4

(0)

S1S2TFD

7

TFD

6

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

DB

24

DB

26

DB

27

DB

28

DB

25

DB

1

DB

0

DB

2

DB

3

DB

29

R-C

OSI

NE_

ALP

HA

DB

30N

RC

1

NRC1

01

R-COSINE_ALPHA

0.70.5 (Default)

fDEV

POWER_AMPLIFIER

0724

6-03

2

Figure 65. Register 2—Transmit Modulation Register Map

• The 2FSK/3FSK/4FSK frequency deviation is expressed by the following:

Direct output

Frequency Deviation [Hz] =

162PFDONCY_DEVIATITx_FREQUEN ×

With RF_DIVIDE_BY_2 (R1_DB18) enabled

Frequency Deviation [Hz] =

1625.0 PFDONCY_DEVIATITx_FREQUEN ××

where Tx_FREQUENCY_DEVIATION is set by R2_DB[19:27] and PFD is the PFD frequency.

• In the case of 4FSK, there are tones at ±3 × the frequency deviation and at ±1 × the deviation.

• The power amplifier (PA) ramps at the programmed rate (R2_DB[8:10]) until it reaches its programmed level (R2_DB[13:18]). If the PA is enabled/disabled by the PA_ENABLE bit (R2_DB7), it ramps up and down. If it is enabled/disabled by the Tx/Rx bit (R0_DB27), it ramps up and turns hard off.

• R-COSINE_ALPHA sets the roll-off factor (alpha) of the raised cosine data filter to either 0.5 or 0.7. The alpha is set to 0.5 by default, but the raised cosine filter bandwidth can be increased to provide less aggressive data filtering by using an alpha of 0.7.

ADF7021-N

Rev. 0 | of 64

REGISTER 3—TRANSMIT/RECEIVE CLOCK REGISTER

FS800.11

FS700.11

FS300.11

...

...

...

...

...

...

FS201.11

FS110.01

CDR_CLK_ DIVIDE12.254255

BK20011

BK10101

BBOS_CLK_DIVIDE481632

SK800.11

SK700.11

SK300.11

...

...

...

...

...

...

SK201.11

SK110.01

SEQ_CLK_DIVIDE12.254255

OK200...1

OK101...1

DEMOD_CLK_DIVIDEINVALID1...15

SEQ_CLK_DIVIDEAGC_CLK_DIVIDE CDR_CLK_DIVIDE

BB

OS_

CLK

_D

IVID

E

DEMOD_CLK_DIVIDE

ADDRESSBITS

GD600...1

GD500...1

GD300...1

GD400...1

GD200...1

GD101...1

AGC_CLK_DIVIDEINVALID1...63

SK8

SK7

FS1

FS2

FS3

FS4

FS8

SK1

SK3

SK4

SK5

SK6

SK2

FS5

FS6

FS7

OK

2

OK

1

OK

4

OK

3

C2

(1)

C1

(1)

C3

(0)

C4

(0)

BK

1

BK

2

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

GD

6

GD

5

GD

1

GD

2

GD

3

GD

4

DB

24

DB

25

DB

28

DB

27

DB

26

DB

29

DB

30

DB

31

DB

1

DB

0

DB

2

DB

3

OK300...1

00...1

OK4

0724

6-03

3

Figure 66. Register 3—Transmit/Receive Clock Register Map

• Baseband offset clock frequency (BBOS CLK) must be greater than 1 MHz and less than 2 MHz, where

DIVIDECLKBBOSXTALCLKBBOS

__=

• Set the demodulator clock (DEMOD CLK) such that 2 MHz ≤ DEMOD CLK ≤ 15 MHz, where

DIVIDECLKDEMODXTALCLKDEMOD

__=

• For 2FSK/3FSK, the data/clock recovery frequency (CDR

CLK) needs to be within 2% of (32 × data rate). For 4FSK, the CDR CLK needs to be within 2% of (32 × symbol rate).

DIVIDECLKCDRCLKDEMOD

CLKCDR__

=

• The sequencer clock (SEQ CLK) supplies the clock to the digital receive block. It should be as close to 100 kHz as possible.

DIVIDECLKSEQXTALCLKSEQ

__=

• The time allowed for each AGC step to settle is determined by the AGC update rate. It should be set close to 8 kHz.

DIVIDECLKAGCCLKSEQ

RateUpdateAGC__

[Hz]=

ADF7021-N

Rev. 0 | of 64

REGISTER 4—DEMODULATOR SETUP REGISTER

DISCRIMINATOR_BW

DO

T_PR

OD

UC

T

POST_DEMOD_BWRx_

INVERT

IF_F

ILTE

R_B

W

ADDRESSBITS

TD4

TD3

RI1

RI2

DW

1

DW

2

DW

6

DW

7

DW

9

DW

10

TD1

TD2

DW

8

DW

3

DW

4

DW

5

DP1

DS3

C2(

0)

C1(

0)

C3(

1)

C4(

0)

DS1

DS2

TD6

TD5

TD10

TD9

TD7

TD8

IFB

2

IFB

1

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

DB

24

DB

26

DB

27

DB

28

DB

25

DB

1

DB

0

DB

2

DB

3

DB

29

DB

30

DB

31

DS101010101

DEMOD_SCHEME

2FSK LINEAR DEMODULATOR2FSK CORRELATOR DEMODULATOR3FSK DEMOD4FSK DEMODRESERVEDRESERVEDRESERVEDRESERVED

DP101

DOT_PRODUCTCROSS_PRODUCTDOT_PRODUCTD

RI10101

Rx_INVERTNORMALINVERT CLKINVERT DATAINVERT CLK/DATA

IFB10101

IF_FILTER _BW

DS2DS300110011

00001111

0011

RI2

IFB20011

DW300....1

DW110....1

POST_DEMOD_BW12....1023

DW201....1

DW1000....1

DW600....1

.

.

.

.

.

.

.

.

DW500....1

DW400....1

TD300....1

TD110....0

DISCRIMINATOR_BW12....660

TD201....0

TD1000....1

TD600....0

.

.

.

.

.

.

.

.

TD500....1

TD400....0

DEMOD_SCHEME

0724

6-03

4

9 kHz13.5 kHz18.5 kHzINVALID

Figure 67. Register 4—Demodulator Setup Register Map

• To solve for DISCRIMINATOR_BW, use the following equation:

DISCRIMINATOR_BW = 310400××KCLKDEMOD

where the maximum value = 660.

• For 2FSK,

⎟⎟⎠

⎞⎜⎜⎝

⎛ ×=

DEVfRoundK

310100

• For 3FSK,

⎟⎟⎠

⎞⎜⎜⎝

⎛

××

=DEVf

RoundK2

10100 3

• For 4FSK,

⎟⎟⎠

⎞⎜⎜⎝

⎛

××

=DEV

FSK fRoundK

410100 3

4

where: Round is rounded to the nearest integer. Round4FSK is rounded to the nearest of the following integers: 32, 31, 28, 27, 24, 23, 20, 19, 16, 15, 12, 11, 8, 7, 4, 3. fDEV is the transmit frequency deviation in Hz. For 4FSK, fDEV is the frequency deviation used for the ±1 symbols (that is, the inner frequency deviations).

• Rx_INVERT (R4_DB[8:9]) and DOT_PRODUCT (R4_DB7) need to be set as outlined in Table 17 and Table 18.

CLKDEMODf

_BWPOST_DEMOD CUTOFF××=

π211

where the cutoff frequency (fCUTOFF) of the post demod-ulator filter should typically be 0.75 × the data rate in 2FSK. In 3FSK, it should be set equal to the data rate, while in 4FSK, it should be set equal to 1.6 × symbol rate.

ADF7021-N

Rev. 0 | of 64

REGISTER 5—IF FILTER SETUP REGISTER

IR_PHASE_ADJUST_MAGIR

_PH

ASE

_A

DJU

ST_D

IREC

TIO

N

IR G

AIN

_A

DJU

ST_I

/Q

IR_G

AIN

_A

DJU

ST_U

P/D

N

IR_GAIN_ADJUST_MAG IF_FILTER_DIVIDER

IF_C

AL_

CO

AR

SE

IF_FILTER_ADJUSTADDRESS

BITS

IFA

1

IFD

9

IFD

5

IFD

6

PM2

PM3

GM

1

GM

2

GM

4

GM

5

IFD

7

IFD

8

GM

3

PM4

PD1

IFD

4

IFD

3

C2

(0)

C1

(1)

C3

(1)

C4

(0)

IFD

1

CC

1

IFD

2

IFA

3

IFA

2

PM1

IFA

6

IFA

4

IFA

5

GA

1

GQ

1

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

DB

24

DB

26

DB

27

DB

28

DB

25

DB

1

DB

0

DB

2

DB

3

DB

29

DB

30

DB

31

CC1 IF_CAL_COARSE01

NO CALDO CAL

PD101

IR_PHASE_ADJUST_DIRECTIONADJUST I CHADJUST Q CH

GA101

IR_GAIN_ADJUST_UP/DNGAINATTENUATE

GQ101

IR_GAIN_ADJUST_I/QADJUST I CHADJUST Q CH

IFD300....1

IFD1IF_FILTER_DIVIDER

10....1

12....511

IFD201....1

IFA2

001..1001.1

IFA6

000..011111

GM3IR_GAIN_

ADJUST_MAGGM5

000.1

IFD900....1

IFD600....1

.

.

.

.

.

.

.

.

IFD500....1

IFD400....1

010..1010.1

IFA1 IF_FILTER_ADJUST..............................

...0+1+2 ...+310–1–2...–31

GM4 GM2 GM1

000.1

000.1

001.1

010.1

012...31

PM2IR PHASEADJUSTPM3 PM1 PM1

000.1

000.1

001.1

010.1

012...15

IFA5

000..1000.1

0724

6-03

5

Figure 68. Register 5—IF Filter Setup Register Map

• A coarse IF filter calibration is performed when the IF_CAL_COARSE bit (R5_DB4) is set. If the IF_FINE_ CAL bit (R6_DB4) has been previously set, a fine IF filter calibration is automatically performed after the coarse calibration.

• Set IF_FILTER_DIVIDER such that

kHz50__

=DIVIDERFILTERIF

XTAL

• IF_FILTER_ADJUST allows the IF fine filter calibration result to be programmed directly on subsequent receiver power-ups, thereby saving on the need to redo a fine filter calibration in some instances. Refer to the Filter Bandwidth Calibration Readback section for information about using the IF_FILTER_ ADJUST bits.

• R5_DB[20:31] are used for image rejection calibration. Refer to the Image Rejection Calibration section for details on how to program these parameters.

ADF7021-N

Rev. 0 | of 64

REGISTER 6—IF FINE CAL SETUP REGISTER

IF_CAL_LOWER_TONE_DIVIDEIF_CAL_UPPER_TONE_DIVIDEIF_CAL_DWELL_TIME

IF_F

INE_

CA

L ADDRESSBITS

CD

3

CD

2

CD

6

LT4

LT5

LT6

LT7

UT3

UT4

UT6

UT7

UT8

CD

1

UT5 LT8

UT1

UT2 LT3

LT2

CD

7

C2

(1)

C1

(0)

C3

(1)

C4

(0)

FC1

LT1

CD

5

CD

4

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

DB

24

DB

26

DB

27

DB

25

DB

1

DB

0

DB

2

DB

3

FC101

IF_FINE_CALDISABLEDENABLED

UT3000..1

UT1101..1

IF_CAL_UPPER_TONE_DIVIDE123..

UT2011..1

UT8000..0

...

...

...

...

...

...

... 127

LT3000..1

LT1101..1

123..

LT2011..1

LT8000..1

...

...

...

...

...

...

... 255

IF_CAL_LOWER_TONE_DIVIDE

CD3000..1

CD1101..1

IF_CAL_DWELL_TIME123..

CD2011..1

CD7000..1

...

...

...

...

...

...

... 127

DB

28IR

C1

DB

29

DB

30

IRC

2

IRD

1

IR_C

AL_

SOU

RC

E_D

RIV

E_LE

VEL

IR_C

AL_

SOU

RC

E ÷

2

IRC10101

IR_CAL_SOURCE_DRIVE_LEVELIRC2

0011

OFFLOWMEDHIGH

IRD101

IR_CAL_SOURCE ÷2SOURCE ÷2 OFFSOURCE ÷2 ON

000..1

LT7

000..1

UT7

0724

6-03

6

Figure 69. Register 6—IF Fine Cal Setup Register Map

• A fine IF filter calibration is set by enabling the IF_FINE_CAL Bit (R6_DB4). A fine calibration is then carried out only when Register 5 is written to and R5_DB4 is set.

Lower Tone Frequency (kHz) =

2VIDEER_TONE_DIIF_CAL_LOW

XTAL

×

Upper Tone Frequency (kHz) =

2VIDEER_TONE_DIIF_CAL_UPP

XTAL

×

It is recommended to place the lower tone and upper tone as outlined in Table 24.

Table 24. IF Filter Fine Calibration Tone Frequencies IF Filter Bandwidth

Lower Tone Frequency

Upper Tone Frequency

9 kHz 78.1 kHz 116.3 kHz 13.5 kHz 79.4 kHz 116.3 kHz 18.5 kHz 78.1 kHz 119 kHz

• The IF tone calibration time is the amount of time that is spent at an IF calibration tone. It is dependent on the sequencer clock. For best practice, is recommended to have the IF tone calibration time be at least 500 μs.

IF Tone Calibration Time = CLKSEQ

TIMEDWELLCALIF ___

The total time for a fine IF filter calibration is

IF Tone Calibration Time × 10

• R6_DB[28:30] control the internal source for the image rejection (IR) calibration. The IR_CAL_SOURCE_ DRIVE_LEVEL bits (R6_DB[28:29]) set the drive strength of the source, whereas the IR_CAL_SOURCE_÷2 bit (R6_DB30) allows the frequency of the internal signal source to be divided by 2.

ADF7021-N

Rev. 0 | of 64

REGISTER 7—READBACK SETUP REGISTER

AD1AD2RB1RB2RB3

DB8 DB7 DB6 DB5 DB4 DB3 DB2

C2 (1) C1 (1)

CONTROLBITS

DB1 DB0

C3 (1)C4 (0)

READBACK_SELECT

ADC_MODE

AD20011

AD10101

ADC_MODEMEASURE RSSIBATTERY VOLTAGETEMP SENSORTO EXTERNAL PIN

RB20011

RB10101

READBACK MODEAFC WORDADC OUTPUTFILTER CALSILICON REV

RB301

READBACK_SELECTDISABLEDENABLED

0724

6-03

7

Figure 70. Register 7—Readback Setup Register Map

• Readback of the measured RSSI value is valid only in Rx mode. Readback of the battery voltage, temperature sensor, or voltage at the external pin is not valid in Rx mode.

• To read back the battery voltage, the temperature sensor, or the voltage at the external pin in Tx mode, users should first power up the ADC using R8_DB8 because it is turned off by default in Tx mode to save power.

• For AFC readback, use the following equations (see the Readback Format section):

FREQ RB [Hz] = (AFC READBACK × DEMOD CLK)/218

VBATTERY = BATTERY VOLTAGE READBACK/21.1

VADCIN = ADCIN VOLTAGE READBACK/42.1

Temperature [°C] = −40 + (68.4 − TEMP READBACK) × 9.32

ADF7021-N

Rev. 0 | of 64

REGISTER 8—POWER-DOWN TEST REGISTER

PD1PD3PD4PD5

DB8 DB7 DB6 DB5 DB4 DB3 DB2

C2 (0) C1 (0)

CONTROLBITS

DB1 DB0

C3 (0)C4 (1)

LOG

_AM

P_EN

AB

LE

SYN

TH_

ENA

BLE

RES

ERVE

D

LNA

/MIX

ER_

ENA

BLE

FILT

ER_

ENA

BLE

AD

C_

ENA

BLE

DEM

OD

_EN

AB

LE

Tx/R

x_SW

ITC

H_

ENA

BLE

PA_E

NA

BLE

_R

x_M

OD

E

CO

UN

TER

_R

ESET

Rx_RESET

CR1

DB15 DB14 DB13 DB12 DB11

LE1 PD6

DB10 DB9

SW1PD7

PD701

PA (Rx MODE)PA OFFPA ON

CR101

COUNTER_RESETNORMALRESET

DEMODRESET

CDRRESET

SW101

Tx/Rx SWITCHDEFAULT (ON)OFF

PD601

DEMOD_ENABLEDEMOD OFFDEMOD ON

PD501

ADC_ENABLEADC OFFADC ON

LE101

LOG_AMP_ENABLELOG AMP OFFLOG AMP ON

PD401

FILTER_ENABLEFILTER OFFFILTER ON

PD301

LNA/MIXER_ENABLELNA/MIXER OFFLNA/MIXER ON

PD101

SYNTH_ENABLESYNTH OFFSYNTH ON

0724

6-03

8

Figure 71. Register 8—Power-Down Test Register Map

• It is not necessary to write to this register under normal operating conditions.

• For a combined LNA/PA matching network, R8_DB11 should always be set to 0, which enables the internal Tx/Rx switch. This is the power-up default condition.

ADF7021-N

Rev. 0 | of 64

REGISTER 9—AGC REGISTER

AGC_HIGH_THRESHOLDLNA_GAIN

AGC_MODE

FILTER_GAINLN

A_

BIA

S

FILT

ER_

CU

RR

ENT

MIX

ER_

LIN

EAR

ITY

LNA_

MO

DE

AGC_LOW_THRESHOLDADDRESS

BITS

FG2

FG1

GL5

GL6

GL7

GH

1

GH

5

GH

6

GM

1

GM

2

LG1

LG2

GH

7

GH

2

GH

3

GH

4

GL4

GL3

C2

(0)

C1

(1)

C3

(0)

C4

(1)

GL1

GL2FI1

LG1

ML1 LI1

LI2

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

DB

24

DB

26

DB

27

DB

28

DB

25

DB

1

DB

0

DB

2

DB

3

FI101

FILTER_CURRENTLOWHIGH

0123

AGC_MODEAUTO AGCMANUAL AGCFREEZE AGCRESERVED

FG20011

FG1 FILTER_GAIN0101

82472INVALID

LG20011

LG10101

LNA_GAIN31030INVALID

GL30001...111

GL11010...101

AGC_LOW_THRESHOLD1234...616263

GL20110...011

GL70000...111

GL60000...111

GL50000...111

GL40000...111

GH30001...110

GH11010...010

AGC_HIGH_THRESHOLD1234...787980

GH20110...110

GH70000...111

GH60000...000

GH50000...001

GH40000...110

LI20

LI10

LNA_BIAS800µA (DEFAULT)

LG101

LNA_MODEDEFAULTREDUCED GAIN

ML101

MIXER_LINEARITYDEFAULTHIGH

0724

6-03

9

Figure 72. Register 9—AGC Register Map

• It is necessary to program this register only if AGC settings, other than the defaults, are required.

• In receive mode, AGC is set to automatic AGC by default on power-up. The default thresholds are AGC_ LOW_ THRESHOLD = 30 and AGC_HIGH_ THRESHOLD = 70. See the RSSI/AGC section for details.

• AGC high and low settings must be more than 30 apart to ensure correct operation.

• An LNA gain of 30 is available only if LNA_MODE (R9_DB25) is set to 0.

ADF7021-N

Rev. 0 | of 64

REGISTER 10—AFC REGISTER

KIKP AFC_SCALING_FACTORMAX_AFC_RANGE

AFC

_EN

ADDRESSBITS

KP3

KP2

MA

3

M4

M5

M6

M7

M11

M12KI2

KI3

KI4

KP1 KI1 M8

M9

M10 M3

M2

MA

4

MA

5

C2

(1)

C1

(0)

C3

(0)

C4

(1)

AE1M1

MA

2

MA

6

MA

7

MA

8

MA

1

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

DB

24

DB

26

DB

27

DB

28

DB

25

DB

1

DB

0

DB

2

DB

3

DB

29

DB

30

DB

31

KP101.1

2^02^1...2^7

AE101

AFC_ENOFFAFC ON

MA30001...111

MA11010...101

MAX_AFC_RANGE1234...253254255

MA20110...011

MA80000...111

...

...

...

...

...

...

...

...

...

...

...

00.1

00.1

KP2KP3 KP KI101.1

2^02^1...2^15

00.1

00.1

KI2KI3 KIKI400.1

M30001...111

M11010...101

AFC_SCALING_FACTOR1234...409340944095

M20110...011

M120000...111

...

...

...

...

...

...

...

...

...

...

...

0724

6-04

0

Figure 73. Register 10—AFC Register Map

• The AFC_SCALING_FACTOR can be expressed as

⎟⎟⎠

⎞⎜⎜⎝

⎛ ×=

XTALRoundFACTORSCALINGAFC 5002__

24

• The settings for KI and KP affect the AFC settling time and

AFC accuracy. The allowable range of each parameter is KI > 6 and KP < 7.

• The recommended settings to give optimal AFC performance are KI = 11 and KP = 4. To trade off between AFC settling time and AFC accuracy, the KI and KP parameters can be adjusted from the recommended settings (staying within the allowable range) such that

AFC Correction Range = MAX_AFC_RANGE × 500 Hz

• When the RF_DIVIDE_BY_2 (R1_DB18) is enabled, the programmed AFC correction range is halved. The user accounts for this halving by doubling the programmed MAX_AFC_RANGE value.

• Signals that are within the AFC pull-in range but outside the IF filter bandwidth are attenuated by the IF filter. As a result, the signal can be below the sensitivity point of the receiver and, therefore, not detectable by the AFC.

ADF7021-N

Rev. 0 | of 64

REGISTER 11—SYNC WORD DETECT REGISTER

PL20011

PL10101

SYNC_BYTE_LENGTH12 BITS16 BITS20 BITS24 BITS

MT20011

MT10101

MATCHING_TOLERANCEACCEPT 0 ERRORSACCEPT 1 ERRORACCEPT 2 ERRORSACCEPT 3 ERRORS

SYNC_BYTE_SEQUENCECONTROL

BITS

SYN

C_B

YTE_

LEN

GTH

MA

TCH

ING

_TO

LER

AN

CE

MT2

SB1

SB2

SB3

SB4

SB5

SB6

SB7

SB8

SB9

SB10

SB11

SB12

SB13

SB14

SB15

SB16

SB17

SB18

SB19

SB20

SB21

SB22

SB23

SB24

MT1

C2

(1)

C1

(1)

C3

(0)

C4

(1)

PL1

PL2

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

DB

24

DB

26

DB

27

DB

28

DB

25

DB

1

DB

0

DB

2

DB

3

DB

29

DB

30

DB

31

0724

6-04

1

Figure 74. Register 11—Sync Word Detect Register Map

REGISTER 12—SWD/THRESHOLD SETUP REGISTER

DATA_PACKET_LENGTHCONTROL

BITSLOC

K_

THR

ESH

OLD

_M

OD

E

SWD

_MO

DE

IL2

IL1

C2

(0)

C1

(0)

C3

(1)

C4

(1)

LM1

LM2

DB

15

DB

14

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DP8

DP7

DP6

DP5

DP4

DP3

DP2

DP1

DB

7

DB

6

DB

5

DB

4

DB

1

DB

0

DB

2

DB

3

LOCK_THRESHOLD_MODE0 THRESHOLD FREE RUNNING1 LOCK THRESHOLD AFTER NEXT SYNCWORD2 LOCK THRESHOLD AFTER NEXT SYNCWORD FOR DATA PACKET LENGTH NUMBER OF BYTES3 LOCK THRESHOLD

DATA_PACKET_LENGTH0 INVALID1 1 BYTE... ...255 255 BYTES

SWD_MODE0 SWD PIN LOW1 SWD PIN HIGH AFTER NEXT SYNCWORD2 SWD PIN HIGH AFTER NEXT SYNCWORD FOR DATA PACKET LENGTH NUMBER OF BYTES3 INTERRUPT PIN HIGH

0724

6-04

2

Figure 75. Register 12—SWD/Threshold Setup Register Map

Lock threshold locks the threshold of the envelope detector. This has the effect of locking the slicer in linear demodulation and locking the AFC and AGC loops when using linear or correlator demodulation.

ADF7021-N

Rev. 0 | of 64

REGISTER 13—3FSK/4FSK DEMOD REGISTER Refer to the Receiver Setup section for information about programming these settings.

3FSK_CDR_THRESHOLD VITE

RB

I_PA

TH_

MEM

OR

Y

3FSK/4FSK_SLICER_THRESHOLD

CONTROLBITS

3FSK

_VIT

ERB

I_D

ETEC

TOR

PHA

SE_

CO

RR

ECTI

ON

ST4

ST5

ST6

ST7

VD1

PC1

VM1

VM2

VT1

VT2

VT3

VT4

VT5

VT6

VT7

ST3

C2

(0)

C1

(1)

C3

(1)

C4

(1)

ST1

ST2

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

1

DB

0

DB

2

DB

3

3FSK_PREAMBLE_TIME_VALIDATE

PTV1

PTV2

PTV3

PTV4

DB

24

DB

23

DB

22

DB

25

4 BITS0011

VM2VITERBI_PATH _MEMORYVM1

0101

6 BITS8 BITS32 BITS

PHASE_CORRECTION

0 DISABLED1 ENABLED

PC1

3FSK_VITERBI_DETECTOR

0 DISABLED1 ENABLED

VD1

ST3

000..1

ST1

101..1

SLICERTHRESHOLD

123..

ST2

011..1

ST7

000..1

...

...

...

...

...

...

... 127

0 00 ... 0 OFF

VT3

000..1

VT1

101..1

3FSK_CDR_THRESHOLD

123..

VT2

011..1

VT7

000..1

...

...

...

...

...

...

... 127

0 00 ... 0 OFF

PTV3

000..1

PTV1

101..1

3FSK_PREMABLE_TIME_VALIDATE

123..

PTV2

011..1

PTV4

000..1 15

0 00 0 0

0724

6-04

3

Figure 76. Register 13—3FSK/4FSK Demod Register Map

ADF7021-N

Rev. 0 | of 64

REGISTER 14—TEST DAC REGISTER

TEST_DAC_GAIN TEST_DAC_OFFSET TEST

_TD

AC

_EN

ED_P

EAK

_R

ESPO

NSE

ED_L

EAK

_FA

CTO

R

PULS

E_EX

TEN

SIO

N

ADDRESSBITS

TG3

TG2

ER2

TO4

TO5

TO6

TO7

TO11

TO12

TO14

TO15

TO16

TG1

TO13

TO8

TO9

TO10

TO3

TO2

EF1

EF2

C2

(1)

C1

(0)

C3

(1)

C4

(1)

TE1

TO1

ER1

EF3

PE1

PE2

TG4

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

DB

24

DB

26

DB

27

DB

28

DB

25

DB

1

DB

0

DB

2

DB

3

DB

29

DB

30

DB

31

ED_PEAK_RESPONSE0123

FULL RESPONSE TO PEAK0.5 RESPONSE TO PEAK0.25 RESPONSE TO PEAK0.125 RESPONSE TO PEAK

TEST_DAC_GAIN01...15

NO GAIN× 2^1...× 2^15

ED_LEAK_FACTOR

01234567

LEAKAGE =2^–82^–92^–102^–112^–122^–132^–142^–15

0724

6-04

4

PULSE_EXTENSION0123

NO PULSE EXTENSIONEXTENDED BY 1EXTENDED BY 2EXTENDED BY 3

Figure 77. Register 14—Test DAC Register Map

The demodulator tuning parameters, PULSE_EXTENSION, ED_LEAK_FACTOR, and ED_PEAK_RESPONSE, can be enabled only by setting R15_DB[4:7] to 0x9.

Using the Test DAC to Implement Analog FM DEMOD and Measuring SNR

For detailed information about using the test DAC, see Application Note AN-852.

The test DAC allows the post demodulator filter out for both linear and correlator demodulators to be viewed externally. The test DAC also takes the 16-bit filter output and converts it to a high frequency, single-bit output using a second-order, error feedback Σ-Δ converter. The output can be viewed on the SWD pin. This signal, when filtered appropriately, can then be used to do the following:

• Monitor the signals at the FSK post demodulator filter output. This allows the demodulator output SNR to be measured. Eye diagrams of the received bit stream can also be constructed to measure the received signal quality.

• Provide analog FM demodulation.

While the correlators and filters are clocked by DEMOD CLK, CDR CLK clocks the test DAC. Note that although the test DAC functions in regular user mode, the best performance is achieved when the CDR CLK is increased to or above the frequency of DEMOD CLK. The CDR block does not function when this condition exists.

Programming Register 14 enables the test DAC. Both the linear and correlator/demodulator outputs can be multiplexed into the DAC.

Register 14 allows a fixed offset term to be removed from the signal (to remove the IF component in the ddt case). It also has a signal gain term to allow the usage of the maximum dynamic range of the DAC.

ADF7021-N

Rev. 0 | of 64

REGISTER 15—TEST MODE REGISTER

Rx_TEST_MODES

Tx_TEST_MODES

Σ-Δ_TEST_MODES

PFD

/CP_

TEST

_M

OD

ES

PLL_TEST_MODES

ANALOG_TEST_MODES CLK_MUX

FOR

CE_

LDH

IGH

REG

1_P

D

CA

L_O

VER

RID

E

ADDRESSBITS

PM4

PM3

AM

3

TM1

TM2

TM3

SD1

PC2

PC3

CM

2

CM

3

PM1

PM2

CM

1

SD2

SD3

PC1

RT4

RT3

AM

4

FH1

RD

1

CO

2

CO

1

C2

(1)

C1

(1)

C3

(1)

C4

(1)

RT1

RT2

AM

2

AM

1

DB

16

DB

15

DB

14

DB

17

DB

20

DB

19

DB

18

DB

21

DB

13

DB

12

DB

11

DB

10

DB

9

DB

8

DB

7

DB

6

DB

5

DB

4

DB

22

DB

23

DB

24

DB

26

DB

27

DB

28

DB

25

DB

1

DB

0

DB

2

DB

3

DB

29

DB

30

DB

31

ANALOG_TEST_MODES0 BAND GAP VOLTGE1 40µA CURRENT FROM REG42 FILTER I CHANNEL: STAGE 13 FILTER I CHANNEL: STAGE 24 FILTER I CHANNEL: STAGE 15 FILTER Q CHANNEL: STAGE 16 FILTER Q CHANNEL: STAGE 27 FILTER Q CHANNEL: STAGE 18 ADC REFERENCE VOLTAGE9 BIAS CURRENT FROM RSSI 5µA10 FILTER COARSE CAL OSCILLATOR O/P11 ANALOG RSSI I CHANNEL12 OSET LOOP +VE FBACK V (I CH)13 SUMMED O/P OF RSSI RECTIFIER+14 SUMMED O/P OF RSSI RECTIFIER–15 BIAS CURRENT FROM BB FILTER

Rx_TEST_MODES0 NORMAL1 SCLK, SDATA -> I, Q2 REVERSE I,Q3

LINEAR SLICER ON RXDATA

4CORRELATOR SLICER ON TxRxDATA

ADDITIONAL FILTERING ON I, Q

ENVELOPE DETECTOR WATCHDOG DISABLEDRESERVED

ENABLE REG 14 DEMOD PARAMETERS

PROHIBIT CALACTIVE

ENABLE DEMOD DURING CALFORCE CALACTIVE

56

I,Q TO TxRxCLK, TxRxDATA

789101112

POWER DOWN DDT AND ED IN T/4 MODE

131415

Tx_TEST_MODES0

Tx CARRIER ONLY1Tx +VE TONE ONLY2Tx –VE TONE ONLY3Tx "1010" PATTERN4Tx PN9 DATA, AT PROGRAMED RATE5Tx SYNC BYTE REPEATEDLY6

Σ-Δ_TEST_MODES0 DEFAULT, 3RD ORDER SD, NO DITHER1 1ST ORDER SD2 2ND ORDER SD3 DITHER TO FIRST STAGE4 DITHER TO SECOND STAGE5 DITHER TO THIRD STAGE6 DITHER × 87 DITHER × 32

0 NORMAPLL_TEST_MODES

L OPERATION1 R DIV2 N DIV3 RCNTR/2 ON MUXOUT4 NCNTR/2 ON MUXOUT5 ACNTR TO MUXOUT6 PFD PUMP UP TO MUXOUT7 PFD PUMP DN TO MUXOUT8 SDATA TO MUXOUT (OR SREAD?)9 ANALOG LOCK DETECT ON MUXOUT10 END OF COARSE CAL ON MUXOUT11 END OF FINE CAL ON MUXOUT12

13 TEST MUX SELECTS DATA14 LOCK DETECT PERCISION15 RESERVED

PFD/CP_TEST_MODES0 DEFAULT, NO BLEED1 (+VE) CONSTANT BLEED2 (–VE) CONSTANT BLEED3 (–VE) PULSED BLEED4 (–VE) PULSE BLD, DELAY UP?5 CP PUMP UP6 CP TRI-STATE7 CP PUMP DN

CLK MUXES ON CLKOUT PIN01234567

CAL_OVERRIDE0 AUTO CAL1 OVERRIDE GAIN2 OVERRIDE BW3 OVERRIDE BW AND GAIN

FORCE_LD_HIGH0 NORMAL1 FORCE

REG1_PD0 NORMAL1 PWR DWN

SDATA TO CDR

FORCE NEW PRESCALER CONFIG.FOR ALL N

NORMAL OPERATION

NORMAL, NO OUTPUTDEMOD CLKCDR CLKSEQ CLKBB OFFSET CLKSIGMA DELTA CLKADC CLKTxRxCLK

3FSK SLICER ON TxRxDATA

0724

6-04

5

Figure 78. Register 15—Test Mode Register Map

• Analog RSSI can be viewed on the Test_A pin by setting ANALOG_TEST_MODES to 11.

• Tx_TEST_MODES can be used to enable test modulation.

• The CDR block can be bypassed by setting Rx_TEST_ MODES to 4, 5, or 6, depending on the demodulator used.

ADF7021-N

Rev. 0 | of 64

OUTLINE DIMENSIONS

PIN 1INDICATOR

TOPVIEW

6.75BSC SQ

7.00BSC SQ

148

1213

3736

2425

4.254.10 SQ3.95

0.500.400.30

0.300.230.18

0.50 BSC

12° MAX

0.20 REF

0.80 MAX0.65 TYP

1.000.850.80

5.50REF

0.05 MAX0.02 NOM

0.60 MAX0.60 MAX PIN 1

INDICATOR

COPLANARITY0.08

SEATINGPLANE

0.25 MIN

EXPOSEDPAD

(BOTTOM VIEW)

COMPLIANT TO JEDEC STANDARDS MO-220-VKKD-2 Figure 79. 48-Lead Lead Frame Chip Scale Package [LFCSP_VQ]

7 mm × 7 mm Body, Very Thin Quad (CP-48-3)

Dimensions shown in millimeters

ORDERING GUIDE Model Temperature Range Package Description Package Option ADF7021-NBCPZ1 −40°C to +85°C 48-Lead Lead Frame Chip Scale Package [LFCSP_VQ] CP-48-3 ADF7021-NBCPZ-RL1 −40°C to +85°C 48-Lead Lead Frame Chip Scale Package [LFCSP_VQ] CP-48-3 ADF7021-NBCPZ-RL71 −40°C to +85°C 48-Lead Lead Frame Chip Scale Package [LFCSP_VQ] CP-48-3 ADF7021-NDF −40°C to +85°C Die on Film EVAL-ADF70XXMBZ21 Evaluation Platform Mother Board EVAL-ADF7021-NDBIZ1 426 MHz to 429 MHz Daughter Board EVAL-ADF7021-NDBEZ1 426 MHz to 429 MHz Daughter Board EVAL-ADF7021-NDBZ21 860 MHz to 870 MHz Daughter Board EVAL-ADF7021-NDBZ51 Matching Unpopulated Daughter Board 1 Z = RoHS Compliant Part.

©2008 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D07246-0-2/08(0)

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