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CATR. The design procedure, based on the one-dimensional quasi-op-tical approach, despite of its simplicity and approximations that aremade regarding the internal lens reflections, is quite efficient for com-puting the sectorial lens profile. The axisymmetry of the lens makesit easy, in principle, to fabricate. Nevertheless the choice of the di-electric material is important since it has to be compatible with themilling technique for ensuring a good surface quality of finish. There-fore, polyurethane material is preferred to PVC. Very good agreementhas been obtained between measurements and simulations carried outwith SRSRD. This is a key point because the lens design procedure ne-glects some important lens features such as the internal reflections orstep shadowing. From this knowledge, we can perform quite accuratesimulations using SRSRD. The above-mentioned effects were quanti-fied and their influences on the final CATR setup were studied.

REFERENCES

[1] S. Qi and K. Wu, “Leakage and resonance characteristics of radiatingcylindrical dielectric structure suitable for use as a feeder for high-ef-ficient omnidirectional/sectorial antenna,” IEEE Trans. AntennasPropag., vol. 46, no. 11, pp. 1767–1773, Sep. 1998.

[2] T. Hirvonen, J. Tuovinen, and A. V. Räisänen, “Lens-type compactantenna test range at mm-waves,” in Proc. 21st Eur. Microw. Conf.,Stuttgart, Germany, Oct. 1991, vol. 2, pp. 1079–1083.

[3] B. D. Nguyen, C. Migliaccio, Ch. Pichot, K. Yalmamoto, and N.Yonemoto, “W-band fresnel zone plate reflector for helicopter colli-sion avoidance radar,” IEEE Trans. Antennas Propag., vol. 55, no. 5,pp. 1452–1456, May 2007.

[4] W. Mayer, M. Meilchen, W. Grabherr, P. Nüchter, and R. Gühl, “Eightchannel 77 GHz front-end module with high-performance synthesizedsignal generator for FM-CW sensor applications,” IEEE Microw.Theory Tech., vol. 52, no. 3, pp. 993–1000, Mar. 2004.

[5] E. K. Walton and J. D. Young, “The Ohio state university compactrange cross-section measurement range,” IEEE Trans. AntennasPropag., vol. 32, no. 11, pp. 1218–1233, Nov. 1984.

[6] C. W. I. Pistorius, G. C. Clerici, and W. D. Burnside, “A dual chamberGregorian subreflector system for compact range applications,” IEEETrans. Antennas Propag., vol. 37, no. 3, pp. 305–313, Mar. 1989.

[7] G. Forma, D. Dubruel, J. Marti-Canales, M. Paquay, G. Crone, J.Tauber, M. Sandri, F. Villa, and I. Ristorcelli, “30-70-100-320 GHzradiation measurements for the radio frequency qualification modelof the Planck satellite,” presented at the 1st Eur. Conf. on AntennasPropag., (EuCAP2006), Nice, France, Nov. 6–10, 2006, paper 349443.

[8] J. Meltaus, J. Salo, E. Noponen, M. M. Salomaa, V. Viikari, A. Lön-nqvist, T. Koskinen, J. Sáily, J. Hakli, J. Ala-Laurinaho, J. Mallat, andA. V. Räisänen, “Millimeter-wave beam shaping using holograms,”IEEE Microw. Theory Tech., vol. 51, no. 4, pp. 1274–1279, Apr. 2003.

[9] J. Häkli, T. Koskinen, A. Lönnqvist, J. Säily, V. Viikari, J. Mallat, J.Ala-Laurinaho, J. Tuovinen, and A. V. Räisänen, “Testing of a 1.5-mreflector antenna at 322 GHz in a CATR based on a hologram,” IEEETrans. Antennas Propag., vol. 53, no. 10, pp. 3142–3150, Oct. 2005.

[10] A. Lönnqvist, T. Koskinen, J. Hakli, J. Sáily, J. Ala-Laurinaho, J.Mallat, V. Viikari, J. Tuovinen, and A. V. Räisänen, “Hologram-basedcompact range for submillimeter-wave antenna testing,” IEEE Trans.Antennas Propag., vol. 53, no. 10, pp. 3151–3159, Oct. 2005.

[11] Menzel and B. Huder, “Compact range for millimeter-wave frequen-cies using a dielectric lens,” Electron. Lett., vol. 20, pp. 768–769, Sep.1984.

[12] G. Bruhat, Optique. France: Masson, 1959, 5ème edition,RO30013068.

[13] A. Berthon and R. Bills, “Integral equation analysis for radiating struc-tures of revolution,” IEEE Trans. Antennas Propag., vol. 49, no. 4, pp.159–170, Feb. 1989.

[14] C. Migliaccio, J.-Y. Dauvignac, L. Brochier, J.-L. Le Sonn, and Ch.Pichot, “W-band high gain lens antenna for metrology and radar appli-cations,” Electron. Lett., vol. 40, no. 22, pp. 1394–1396, Oct. 28, 2004.

[15] A. P. Pavacic, D. L. Del Rio, J. R. Mosig, and G. V. Eleftheriades,“Three dimensional ray tracing theory to model internal reflections inoff-axis lens antennas,” IEEE Trans. Antennas Propag., vol. 54, no. 2,pp. 604–612, Feb. 2006.

[16] R. Leberer and W. Menzel, “A dual planar reflectarray with synthesizedphase and amplitude distribution,” IEEE Trans. Antennas Propag., vol.53, no. 11, pp. 3534–3539, Nov. 2005.

[17] [Online]. Available: www.scilab.org[18] J. Lanteri, C. Migliaccio, J.-Y. Dauvignac, and Ch. Pichot, “Reflec-

tarray using a prolate feed at 94 GHz,” in Proc. IEEE AP-S, San Diego,Jul. 5–11, 2008, pp. 1–4.

[19] M. Multari, C. Migliaccio, J.-Y. Dauvignac, L. Brochier, J.-L. Le Sonn,Ch. Pichot, W. Menzel, and J.-L. Desvilles, “Investigation of low-costcompact range W-band,” in Proc. EuCAP, Nice, Nov. 6–10, 2006, pp.1–6.

[20] R. C. Rudduck and C. L. J. Chen, “New plane wave spectrum formu-lation for the near-field of circular apertures,” IEEE Trans. AntennasPropag., vol. 24, no. 4, pp. 438–449, Jul. 1976.

A Single-Layer Ultrawideband Microstrip Antenna

Qi Wu, Ronghong Jin, and Junping Geng

Abstract—A single-layer microstrip antenna for ultrawideband (UWB)applications, in which an array of rectangular microstrip patches was ar-ranged in the log-periodic way and proximity-coupled to the microstripfeeding line, is presented. In order to reduce the number of microstrippatches in the UWB log-periodic arrays, a large scale factor � � � � wasfirstly reported and proved to be highly effective. Furthermore, instead ofusing an absorbing terminal loading, a novel loss-free compensating stubwas proposed. Detailed parameters study was also presented for better un-derstanding of the proposed antennas. The impedance bandwidth (mea-sured ���� � ) of an example antenna with only 11 elements isfrom 2.26–6.85 GHz with a ratio of about 3.03:1. Both numerical and ex-perimental results show that the proposed antenna has stable directionalradiation patterns, very low-profile and low fabrication cost, which are suit-able for various broadband applications.

Index Terms—Directional antennas, log-periodic antennas, microstripantennas, ultrawideband (UWB) antennas.

I. INTRODUCTION

Currently, there are increasing demands for novel ultrawideband(UWB) antennas with low-profile structures and constant directionalradiation patterns for both commercial and military applications [1],[2]. Unfortunately, most of the mature UWB antennas like equiangularand Archimedean spirals [3], planar monopoles [4], [5] and wide slotantennas [6], [7] have inherently bi-directional or omnidirectionalradiation patterns, which were unsuitable for conformal placement oncertain platforms. Cavity-backed log-periodic slot antennas [8] couldbe integrated compactly into various aircrafts, but they could onlyprovide end-fire radiation patterns and have somewhat high profile.

On the other hand, microstrip antennas have some attractive meritslike very low-profile and broadside radiation patterns with mediumgains, which have been considered as excellent conformal radiators[9] for a long time. However, a traditional single-element microstripantenna has inherently narrow impedance bandwidth. In the 1980s, the

Manuscript received January 06, 2009; revised May 10, 2009. First publishedJuly 14, 2009; current version published January 04, 2010.

The authors are with the Department of Electronic Engineering, ShanghaiJiao Tong University, Shanghai 200240, China (e-mail: wuqi_2004@sjtu.edu.cn).

Color versions of one or more of the figures in this communication are avail-able online at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TAP.2009.2027728

0018-926X/$26.00 © 2009 IEEE

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212 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 1, JANUARY 2010

series-fed log-periodic microstrip antennas (LPMAs) were firstly intro-duced for bandwidth enhancement [10], [11]. After that, other feedingstructures like ���-microstrip feeding [12], [13], direct feeding withvias [14], slot-coupled feeding [15], inset feeding [16] for LPMAs werealso reported. However, all of the LPMAs mentioned above were basedon the combination of two substrate layers, in which very accurate col-location between the two layers were required.

A UWB traveling-wave LPMA was usually terminated by a matchedload to absorb the terminal energy and prevent reflections from the endof feeding line, which could improve its impedance matching and radi-ation patterns considerably at the band edges. However, the absorbingloads could remarkably degrade the efficiency of LPMAs over their op-erating bandwidths, especially at the band edges by the order of 10 %or more [10].

It is also well known that the operating bandwidth of a LPMA couldbe determined by the scale factor k [10] and the number of microstripelements n, in a simple expression as ����. Therefore, for the LMPAwith a small scale factor, a large number of microstrip patches wererequired to achieve ultrawide operating bandwidth and make the an-tenna unacceptably large. Obviously, the number of elements in a UWBLMPA could be significantly reduced with a larger scale factor. For ex-ample, with a scale factor k as 1.05 or below like the reported LPMAsin [10]–[16], about 21 elements were needed to achieve a 2.6:1 band-width; however, for the LMPA with a scale factor k as 1.1, 11 microstripelements could be enough for the same bandwidth with a size reductionof over 40%.

This communication presents a single-layer proximity-coupled log-periodic microstrip antenna for UWB applications. A large scale factoras 1.1 for UWB LPMAs is firstly reported and discussed. A novelloss-free compensating stub is also proposed for the termination andbandwidth enhancement of the proposed antennas. The configurationof the proposed LPMA is described in Section II. Numerical results arepresented in Section III for better understanding of this antenna, whilethe measured results are presented and discussed in Section IV. Thiscommunication is concluded in Section V.

II. ANTENNA CONFIGURATION

The proposed single-layer LPMA, as illustrated in Fig. 1, wascomposed of a 50 �–100 � impedance transition, a 100-� microstripfeeding line and an array of proximity-coupled rectangular microstrippatches, which were all etched on a Teflon-based substrate with arelative permittivity of 2.65 and a thickness of 3 mm. The microstripelements were arranged in a transposed log-periodic way with thesame coupling gap G. The log-periodic scale factor k was defined as

� ���

����

���

����

�������

������

� (1)

In (1), ������ was the center distance between patch ���� and ����

and �� were respectively the width and length of the patch ��. Theproposed LPMA was terminated by a novel compensating stub withlength T, instead of a matched load or open-circuit. The LPMAs weredesigned by a semi-empirical way like the design procedure of LPDAs[3] except some parameters of the LPMAs should be optimized withthe assistance of full-wave simulator HFSS. Following discussions ofthe proposed LPMAs were all based on the optimized parameters inTable I.

III. NUMERICAL RESULTS

Coupling gap G between the microstrip patches and the microstripfeeding line could determine the power transmission efficiency of thefeeding line thus should be carefully optimized. Fig. 2 shows the simu-lated results of the proposed LPMA at the reference plane ��� (see

Fig. 1. Structural configuration of the proposed LPMA and the coordinatesystem.

TABLE IOPTIMIZED PARAMETERS OF THE PROPOSED LPMA WITH COMPENSATING

STUB

Fig. 2. Simulated reflection coefficient and antenna efficiency of the proposedantennas with compensating stubs at the reference plane �� �� � ����.

Fig. 1), and the influence of the coupling gap G on the impedancematching and antenna efficiency could be easily observed. The antennawith� � � mm was no longer a travelling-wave array and exhibited amultiresonant behavior. The antennas with � � ��� mm and � � ���mm, which could be respectively referred as “critical coupled” and“under coupled” [17], have very similar reflection coefficient and ef-ficiency above 2.55 GHz because the last several microstrip patchescould “absorb” the residual energy which was not properly coupledto the corresponding patches due to the “under coupled” effects. So,it is not surprising to find that the “critical coupled” one was betterimpedance matched in the band from 2.25–2.55 GHz than the “undercoupled” one because this frequency band corresponds to the last mi-crostrip elements �� .

The characteristics of compensating stubs with positive or negativeT value were different: when the stub went beyond the last patch ��(known as positive T), the stub was actually an open-circuit microstripstub and behaved like a loss-free “series capacitive load,” the stub withnegative T could be considered as a loss-free “series inductive load.”The compensating effects of the stub could be clearly observed by ex-amining the simulated input impedance of the proposed LPMAs at thereference plane ��� as shown in Fig. 3. The input impedance of theproposed antenna without compensating (� � � mm) was heavily

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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 1, JANUARY 2010 213

Fig. 3. Simulated input impedance of the proposed antennas with compen-sating stubs at the reference plane �� �� � ����.

capacitive just after its first resonant frequency at 2.36 GHz. The ca-pacitive components were well compensated by the “series inductiveload” (� � �� mm), and thus the impedance matching was signif-icantly improved; but the situation got worse if the “series capacitiveload” (� � � mm) was applied. After the first resonant frequency, theinfluence of different stub T was very slight, thus it could be observedthat the antennas were all well impedance matched.

The performance of two identical LPMAs with the compensatingand absorbing loads could be also evaluated and compared. The ab-sorbing load was assumed to be positioned beyond the last patch ��with � � �� mm and set as a “Lumped RLC boundary” with 100-�resistance and a dimension of 2.4� 3.0 mm� in HFSS. Generally bothof the antennas were well impedance matched in the frequency bandbetween 2.4 and 6.8 GHz as shown in Fig. 4. In the frequency bandbelow 2.3 GHz, the one with an absorbing load still has good VSWRfor its terminal load could absorb all of the residual energy and preventpossible terminal reflections, which also significantly degrades its effi-ciency as shown in Table II. The radiation patterns of the LPMAs withcompensating and absorbing loads were very similar in the frequencyband above 3.3 GHz as shown in Fig. 5. But the patterns have somedifferences in the x-z plane at 2.3 GHz, in which the pattern of the onewith absorbing load has narrower beamwidth and higher directivity.That difference was mainly caused by the different dealing methodsof the residual energy: for the absorbing load, all of the residual en-ergy was absorbed at the terminal, thus good radiation patterns couldbe observed; for the compensating case, the residual energy was totallyreflected at the termination, thus some disordered modes of patch ��appeared and the patterns were also influenced. Furthermore, althoughthe LPMA with absorbing load has higher directivity below 3.3 GHz,its gain is obviously lower than the one with compensating stub for itsextremely low efficiency.

In addition, the performance of the LPMAs with � � ��� shouldalso be examined. The simulated impedance bandwidth of the LPMAwith � � ���, defined by �� � ��, was from 3.35–6.95 GHz.The radiation patterns and gain of the proposed LPMA with � � ���

were respectively illustrated in Fig. 5 and Table II. Generally the twoLPMAs with � � ��� and � � ��� have very similar radiation pat-terns, and the one with � � ��� has higher directivity, 0.65 dB onaverage, than the one with � � ���, which shows the similar trend asthe log-periodic dipole array (LPDA) [18] and could be considered asthe main cost of large scale factor. Besides, the trend of patch gap����

was also found to comply with the best design curve for LPDAs, and

Fig. 4. Simulated and measured VSWR and gain of the proposed antennas withcompensating and absorbing terminations �� � ����.

Fig. 5. Simulated radiation patterns of the proposed LPMAs with compen-sating and absorbing stubs: (a) x-z plane, compensating stub, � � ���; (b) x-zplane, absorbing stub, � � ���; (c) y-z plane, compensating stub, � � ���; (d)y-z plane, absorbing stub, � � ���; (e) x-z plane, compensating stub, � � ����;(f) y-z plane, compensating stub, � � ����.

thus the well-established theory of LPMAs could be an important guid-ance in the design of LPMAs.

IV. EXPERIMENTAL RESULTS AND DISCUSSION

An example antenna with � � ��� was fabricated based on the opti-mized parameters in Table I. The impedance bandwidth was measuredby using an Agilent 8722ES Vector Network Analyzer (VNA) and the

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214 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 1, JANUARY 2010

TABLE IISIMULATED DIRECTIVITY, GAIN AND EFFICIENCY OF THE PROPOSED LPMAS

Fig. 6. Measured radiation patterns of the proposed antenna with compensatingstub (k=1.1): (a) x-z plane; (b) y-z plane.

results were shown in Fig. 4. It shows good agreement between thesimulated and measured results, and the little difference between themmay be caused by the soldering effect of the SMA connector and itsmechanical tolerance. Its measured impedance bandwidth defined by���� � ��� is from 2.26–6.85 GHz with a ratio of about 3.03:1.The radiation patterns of the example antenna were illustrated in Fig. 6.Generally, the measured patterns agree well with the simulated one,and the patterns were reasonably stable at the broadside. The gain wasalso measured by using the comparison method with a corrugated horn

as the standard antenna, and the results could be found in Fig. 4. Thesimulated gains agree reasonably with the measured one, and the dif-ferences between them may be caused by the numerical errors and thecalibration errors of the UWB standard horn. The gain bandwidth, de-fined by the simulated and measured gain better than 6.5 dB, was from2.4–6.6 GHz with a radio bandwidth 2.75:1. In its gain bandwidth, thefluctuations of simulated and measured gain were also small and eval-uated to be 2.4–2.0 dB, respectively.

V. CONCLUSION

A single-layer log-periodic microstrip antenna for UWB applica-tions was presented. A large scale factor � � � was firstly reportedand proved to be highly effective for the purpose of size reduction.Furthermore, instead of using an absorbing terminal loading, a novelloss-free compensating stub was also proposed. The impedance band-width (with measured���� � ���) of the example antenna with only11 elements is from 2.26–6.85 GHz with a ratio of about 3.03:1. Bothnumerical and experimental results show that the proposed antenna hasstable directional radiation patterns, very low-profile and low fabrica-tion cost, which are suitable for various broadband applications.

REFERENCES

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