9.a nine-level grid-connected converter topology for single-phase transformerless pv systems.pdf

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 8, AUGUST 2014 3951

A Nine-Level Grid-Connected Converter Topologyfor Single-Phase Transformerless PV Systems

Giampaolo Buticchi, Member, IEEE, Davide Barater, Student Member, IEEE, Emilio Lorenzani, Member, IEEE,Carlo Concari, Member, IEEE, and Giovanni Franceschini

Abstract—This paper presents a single-phase transformerlessgrid-connected photovoltaic converter based on two cascaded fullbridges with different dc-link voltages. The converter can syn-thesize up to nine voltage levels with a single dc bus, since oneof the full bridges is supplied by a flying capacitor. The mul-tilevel output reduces harmonic distortion and electromagneticinterference. A suitable switching strategy is employed to regulatethe flying-capacitor voltage, improve the efficiency (most devicesswitch at the grid frequency), and minimize the common-modeleakage current with the help of a novel dedicated circuit (tran-sient circuit). Simulations and experiments confirm the feasibilityand good performance of the proposed converter.

Index Terms—Leakage current, multilevel systems, photo-voltaic (PV) systems, pulsewidth modulation (PWM) inverters.

I. INTRODUCTION

GRID-CONNECTED photovoltaic (PV) converters repre-sent the most widespread solution for residential renew-

able energy generation. While classical designs of PV convertersfeature a grid frequency transformer, which is a typically heavyand costly component, at the interface between the converterand the electrical grid, researchers are now considering trans-formerless architectures in order to reduce costs and weight andimprove efficiency. Removing the grid frequency transformerentails all the benefits above but worsens the output power qual-ity, allowing the injection of dc current into the grid [1], [2] andgiving rise to the problem of ground leakage current [3], [4].

Although the active parts of PV modules might be electri-cally insulated from the ground-connected mounting frame, apath for ac ground leakage currents generally exists due to aparasitic capacitance between the modules and the frame andto the connection between the neutral wire and the ground,usually realized at the low-voltage/medium-voltage (LV/MV)transformer [3]. In addition to deteriorating power quality, theground leakage current increases the generation of electromag-netic interference and can represent a safety hazard, so that

Manuscript received March 4, 2013; revised June 13, 2013 and August 1,2013; accepted September 16, 2013. Date of publication October 21, 2013;date of current version February 7, 2014.

G. Buticchi, D. Barater, C. Concari, and G. Franceschini are with the De-partment of Information Engineering, University of Parma, 43124 Parma, Italy(e-mail: [email protected]; [email protected]; [email protected]; [email protected]).

E. Lorenzani is with the Department of Science and Methods for Engineer-ing, University of Modena and Reggio Emilia, Modena, Italy (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2013.2286562

international regulations pose strict limits to its magnitude. Thisissue must be confronted in all transformerless PV converters,regardless of architecture. In particular, in full-bridge-basedtopologies, the ground leakage current is mainly due to high-frequency variations of the common-mode voltage at the outputof the power converter [4]. Several solutions can be found inliterature aiming at the reduction of the common-mode voltageharmonic content [5]–[7]. Once the grid frequency transformeris removed from a PV converter, the bulkiest wound and re-active components that remain are those that form the outputfilter used to clean the output voltage and current from high-frequency switching components. Further reduction in costand weight and improvement in efficiency can be achievedby reducing the filter size, and this is the goal of multilevelconverters.

Multilevel converters have been investigated for years [8],but only recently have the results of such researches found theirway to commercial PV converters. Since they can synthesizethe output voltages using more levels, multilevel converters out-perform conventional two- and three-level converters in termsof harmonic distortion. Moreover, multilevel converters subdi-vide the input voltage among several power devices, allowingfor the use of more efficient devices. Multilevel converters wereinitially employed in high-voltage industrial and powertrainapplications. They were first introduced in renewable energyconverters inside utility-scale plants, in which they are stilllargely employed [9]–[13]. Recently, they have found theirway to residential-scale single-phase PV converters, where theycurrently represent a hot research topic [14]–[29]. Single-phasemultilevel converters can be roughly divided into three cate-gories based on design: neutral point clamped (NPC), cascadedfull bridge (CFB), and custom.

In NPC topologies, the electrical potential between the PVcells and the ground is fixed by connecting the neutral wire ofthe grid to a constant potential, resulting from a dc-link capac-itive divider [15]. A huge advantage is that single-phase NPCconverters are virtually immune from ground leakage currents,although the same is not true for three-phase NPC converters[12], [30]. A recent paper has proposed an interesting NPCdesign for exploiting next-generation devices such as superjunction or SiC MOSFETs [16]. The main drawback of NPCdesigns, with respect to full bridge, is that they need twice thedc-link voltage.

CFBs make for highly modular designs. Usually, each fullbridge inside a CFB converter needs an insulated power supply,matching well with multistring PV fields [17]. In this case,sequential permutation of the full bridges can be used to evenly

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3952 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 8, AUGUST 2014

share power among the parts and to mitigate the effects ofpartial shading [17]–[20]. As an alternative, only one powersupply can be used if the output voltage is obtained through atransformer [21], [31]. CFB converters have also been proposedfor stand-alone applications [17], [22]. CFBs give developersmany degrees of freedom for the control strategy. Togetherwith the aforementioned sequential permutation and with phaseshifting [19], artificial neural networks [23] and predictive con-trol [24] have been proposed to minimize harmonic distortionand achieve maximum power point tracking (MPPT). A CFBmade up of n full bridges (and at least 4n power switches)can synthesize 2n+ 1 voltage levels when the supply voltageis the same for each full bridge. Custom architectures cangenerally provide more output levels with a given number ofactive devices, and custom converters generally need custompulsewidth modulation (PWM) and control schemes [25]–[27],although unified control schemes for different types of multi-level converters have been proposed [28]. In addition to usingless switches, custom architectures can be devised so that someof the switches commutate at the grid frequency, thus improv-ing the efficiency [29]. Reduction in the switches-per-output-voltage-level ratio can be achieved in CFB structures if differentsupply voltages are chosen for each full bridge (asymmetricalCFBs) [32], [33]. The topology proposed in this paper consistsof two asymmetrical CFBs, generating nine output voltagelevels. In the proposed converter, the dc voltage source suppliesone of the full bridges, whereas a flying capacitor supplies theother one. By suitably controlling the ratio between the twovoltages, different sets of output levels can be obtained.

Moreover, the flying capacitor used as a secondary energysource allows for limited voltage boosting, as it will resultclear in the following section. The number of output levels perswitch (eight switches, nine levels) is comparable to what canbe achieved using custom architectures. In fairness, it shouldbe noted that two additional very low power switches and aline frequency switching device [transient circuit (TC)] wereincluded in the final topology in order to reduce the groundleakage current. The custom converter proposed in [29] gen-erates five levels with six switches but has no intrinsic boostingcapability. In [25], Rahim et al. used three dc-bus capacitors inseries together with two bidirectional switches (diode bridge +unidirectional switch) and an H-bridge to generate seven outputlevels; however, they give no explanations on how they keepthe capacitor voltages balanced. In [27], five switches, fourdiodes, and two dc-bus capacitors in series are used to generatefive levels with boosting capability. Again, no mention is madeabout how the capacitors are kept balanced.

In PV applications, the PV field dc voltage is constantlychanging due to variations of solar radiation and to the MPPTalgorithm, but the output voltage has to be controlled regardlessof the voltage ratio. This problem was studied in [34]–[36],measuring the separate full-bridge voltages and computing on-line the duty cycles needed to balance the different voltages, andanalyzing also the power balance between the separate cells.A similar approach is followed in this paper. Moreover, thedeveloped PWM strategy, in addition to controlling the flying-capacitor voltage, with the help of the specific TC illustratedin Section IV, minimizes the ground leakage current. Finally, it

Fig. 1. CFB with a flying capacitor.

is important to put in evidence that the proposed converter canwork at any power factor as reported in Section III, while notall the alternative proposals can continuously supply reactivepower [37], [38]

The proposed topology was presented by the authors in aprevious paper [39]. With respect to the previous work, thispaper was rewritten and presents a better organization and anew set of simulation and experimental results with differentsetups. This paper is organized as follows: Section II presentsthe power converter topology and the PWM control strategychosen in order to maximize the performance with the use ofa low-cost digital signal processor (DSP). Section III describesthe regulation of the flying capacitor used to supply the secondfull bridge of the CFB topology. Section IV describes theprinciple of operation of the additional components able toreduce the ground leakage current. Sections V and VI showthe simulation and experimental results, whereas Section VIIreports the concluding remarks.

II. NINE-LEVEL CONVERTER AND

PWM CONTROL STRATEGY

The proposed converter is composed of two CFBs, one ofwhich is supplied by a flying capacitor (see Fig. 1). Thisbasic topology was already presented in [34]. In this paper, adifferent PWM strategy was developed in order to allow grid-connected operation with no galvanic isolation (transformerlesssolution) for this basic topology. Since the PWM strategy aloneis not sufficient to maintain a low ground leakage current, othercomponents were added as will be shown in Section IV. As itwill be described in the following, the proposed PWM strategystretches the efficiency by using, for the two legs where PWM-frequency switching does not occur, devices with extremelylow voltage drop, such as MOSFETs lacking a fast recoverydiode. In fact, the low commutation frequency of those two legsallows, even in a reverse conduction state, the conduction inthe channel instead of the body diode (i.e., active rectification).Insulated-gate bipolar transistors (IGBTs) with fast antiparalleldiodes are required in the legs where high-frequency hard-switching commutations occur. In grid-connected operation,one full-bridge leg is directly connected to the grid neutral wire,whereas the phase wire is connected to the converter through anLC filter.

BUTICCHI et al.: GRID-CONNECTED CONVERTER TOPOLOGY FOR TRANSFORMERLESS PV SYSTEMS 3953

TABLE IDESCRIPTION OF THE CONVERTER OPERATING ZONES

As it will be described and justified in the following section,flying-capacitor voltage Vfc is kept lower, at steady state, thandc-link voltage VDC. Accordingly, the full bridge supplied bythe dc link is called the high-voltage full bridge (HVFB),whereas the one with the flying capacitor is the low-voltagefull-bridge (LVFB).

The CFB topology allows certain degrees of freedom in thecontrol, so that different PWM schemes can be considered;however, the chosen solution needs to satisfy the followingrequirements.

1) Most commutations must take place in the LVFB to limitthe switching losses.

2) The neutral-connected leg of the HVFB needs to switchat grid frequency to reduce the ground leakage current.

3) The redundant states of the converter must be properlyused to control the flying-capacitor voltage.

4) The driving signals must be obtained from a single carrierfor a low-cost DSP to be used as a controller.

The switching pattern described in Table I was developedstarting from the above requirements. Requirement 2), in partic-ular, is due to the aforementioned parasitic capacitive couplingbetween the PV panels and their frames, usually connectedto the earth. Capacitive coupling renders the common-modecurrent inversely proportional to the switching frequency of theneutral-connected leg.

The converter can operate in different output voltage zones,where the output voltage switches between two specific levels.The operating zone boundaries vary according to the dc-linkand flying-capacitor voltages, and adjacent zones can overlap(see Fig. 2).

In zones labeled A, the contribution of the flying-capacitorvoltage to the converter output voltage is positive, whereasit is negative in B zones. Constructive cascading of the twofull bridges can, therefore, result in limited output voltageboosting. Depending on the Vfc/VDC ratio, one of the (a) or(b) situations in Fig. 2 can ensue; nevertheless, the operationof the converter does not differ much in the two cases. Iftwo overlapping operating zones can supply the same outputvoltage, the operating zone to be used is determined takinginto account the regulation of Vfc, as will be described inSection III.

As mentioned in the introduction, the duty cycles are calcu-lated on-line by a simple equation, similarly to the approachpresented in [34]. The switching pattern depends on the instan-taneous fundamental component of output voltage V ∗

out and onthe measured values of Vfc and VDC.

Fig. 2. Operating zones under different Vfc ranges. (a) Vfc < 0.5VDC.(b) Vfc > 0.5VDC.

If Vfc = VDC/3, the converter can synthesize nine equallyspaced output voltage levels. Fig. 3 refers to this case and showsthe theoretical waveforms, where one leg of the HVFB operatesat grid frequency and one leg of the LVFB at five times the gridfrequency.

Moreover, apart from zone 2, no high-frequency commuta-tions occur in the whole HVFB (see Fig. 2). Since the voltageregulation of the flying capacitor takes place in zone 2, thezone-2 behavior is more articulated and will be described indetail in the following section.

III. FLYING-CAPACITOR VOLTAGE REGULATION

Since the main task facing a grid-connected PV converter isthe transfer of active power to the electrical grid, controlling thevoltage of the flying capacitor is critical.

Flying-capacitor voltage Vfc is regulated by suitably choos-ing the operating zone of the converter depending on the in-stantaneous output voltage request. Depending on the operatingzone of the converter (see Fig. 2), Vfc can be added to


Fig. 3. Theoretical waveforms of the proposed converter. (a) PWM switchingpatterns. (b) Output voltages.

(A zones) or subtracted from (B zones) the HVFB output volt-age, charging or discharging the flying capacitor. In particular,considering a positive value of the current injected into the grid,the flying capacitor is discharged in A zones and charged inB zones. Since a number of redundant switch configurationscan be used to synthesize the same output voltage waveform,it is possible to control the voltage of the flying capacitor,forcing the converter to operate more in A zones when theflying-capacitor voltage is higher than a reference value or morein B zones when it is lower than a reference value. Similarconsiderations hold in case of a negative injected grid current.In each case, some commutations between nonadjacent outputlevels must inevitably occur (level skipping), with the drawbackof a certain increase in the output current ripple.

The voltage control of the flying capacitor (which determinesthe zone-A or zone-B operation) is realized by a simple hystere-sis control.

Fig. 4. Converter configurations for the regulation of the flying capacitor.(a) Flying-capacitor charge. (b) Flying-capacitor discharge.

Fig. 4 illustrates the regulation of Vfc supposing a positivegrid current with Vout > 0 and Vfc < 0.5VDC. If Vfc is too low,output level Vfc can be replaced by VDC − Vfc, thus switchingbetween the 0 and VDC − Vfc output levels [zone 2B, Fig. 4(a)].Similarly, if Vfc is too high, VDC − Vfc can be replaced with Vfc,causing the converter to switch between the Vfc and VDC outputlevels [zone 2A, Fig. 4(b)]. In Fig. 4, the devices switchingat low frequency are short circuited when on and not shownwhen off.

Similar Vfc regulation strategies can be likewise developedfor the case when Vfc > 0.5VDC.

If Vfc < 0.5VDC, in order to minimize the current ripple,zone 2 is chosen only when Vfc < V ∗

out < VDC − Vfc (zones 3are otherwise chosen), limiting level skipping. Level skippingalways occurs if Vfc > 0.5VDC; hence, any A or B zone can bechosen according to the voltage regulation algorithm.

Since the dc-link voltage can go through sudden variationsdue to the MPPT strategy, it is important that the converter isable to work in any [VDC, Vfc] condition. While the distortionof the output voltage is minimized through the on-line duty-cycle computation, it is important to assess the capability ofthe converter to regulate the flying-capacitor voltage underdifferent operating conditions.

The ability to control the flying-capacitor voltage throughthe proposed PWM strategy has been studied in simulation bydetermining the average flying-capacitor current under a largespan of VDC and Vfc values. In the simulations, grid voltagevgrid is sinusoidal with an amplitude of 230

√2 V; however,

the same results hold even for different voltages if the ratioVgrid/VDC remains constant.

The results in the case of unity power factor are summarizedin Fig. 5. The white area covers the range over which Vfc

is fully controllable, whereas it cannot be controlled in thegray and black regions. In particular, in the black region, Vfc

cannot be decreased, whereas in the gray region, it cannot beincreased.


Fig. 5. Flying-capacitor voltage regulation regions.

Therefore, the white region located between the gray andblack ones is a stable and safe operating area for the converter.Even if Vfc was not actively controlled, it would be constrainedinside the white region, ensuring that the flying capacitor cannotbe over charged nor completely discharged.

The results are not affected by the amplitude of the gridcurrent. Nevertheless, the power factor affects the results: alower power factor determines widening of the controllablearea. When the converter supplies only reactive power, Vfc iscontrollable in the entire [VDC, Vfc] domain.

IV. APPLICATION TO TRANSFORMERLESS

PV CONVERTERS—TC

A particular feature of the commutation pattern of Table Iis that T3 and T4 switch at grid frequency, commutating atevery zero crossing of vgrid. If the zero crossing with a negativederivative is considered, T4 opens and T3 closes, changing theneutral wire voltage (and thus the voltage across the parasiticcapacitance of the PV field) from zero to VDC. For this reason,the commutation can cause a large surge of leakage current thatcan decrease the power quality and damage the PV modules. Aproper TC was designed to decrease these surge currents.

Fig. 6(a) shows the proposed converter topology; it is con-stituted of the two-cell CFB described in Fig. 1 with theaddition of the TC components. In order to better understandthe behavior of the TC, the distributed parasitic capacitance ofthe PV source was modeled with a simple equivalent parasiticcapacitance, i.e., Cp, connected between the negative pole ofthe dc link and the ground.

The TC consists of two low-power MOSFETs M1 and M2,bidirectional switch T9, and resistor RT . When the converterenters operating zone 1, the HVFB output voltage must bezero, obtained by switching T1 and T3 or T2 and T4 on.Nevertheless, to operate the TC, when entering zone 1, T1, T2,T3, and T4 are all kept off, while T9 is on. This keeps theneutral potential floating, so that the voltage on the parasitic

Fig. 6. Ground leakage current limitation circuit topology and behavior.(a) TC topology. (b) TC operation. (c) TC waveforms.

capacitor vground stays constant [see Fig. 6(b)]. At this point,one of M1 and M2 is turned on (M1 if the slope of the zerocrossing is negative and M2 if positive). So doing, Cp is chargedthrough RT with a first-order transient [see Fig. 6(c)], limitingthe current surge.

Whereas the TC introduces additional components, they canbe selected with current ratings much lower than the devices ofthe CFB. Moreover, the power loss due to the added resistor isnegligible. Estimating the energy lost charging and discharginga capacitor Cp to VDC averaged over a line period T by Ptc =CpV

2DC/T , with Cp = 200 nF and VDC = 300 V, a dissipation

of about 1 W is obtained. The operation of the TC is not affectedby the power factor because in grid-connected operation, theoutput voltage is always very close to the grid voltage. The cor-rect operation of the TC requires the grid voltage instantaneousangle that can be obtained with a phase-locked loop (PLL) fedby the grid voltage [40].


Fig. 7. Simulation results with VDC = 300 V.

V. SIMULATION RESULTS

The proposed converter and PWM were extensively sim-ulated under MATLAB/Simulink, using the PLECS toolbox.The simulations cover a large range of active and reactivepower injected into the grid, dc-link voltage, and equivalent PVparasitic capacitance.

A dc-link voltage VDC = 300 V was used in the simulations,unless otherwise specified. The grid was represented by a sinu-soidal voltage source at 50 Hz of amplitude vgrid = 230 V. Theoutput filter was composed of a capacitor Cf = 1 μF and aninductor Lf = 1.5 mH. An additional inductor Lgrid = 40 μHrepresented the total distributed grid inductance. The PWMfrequency was fs = 20 kHz, and the flying capacitor had acapacitance of Cfc = 500 μF. The surge limiting resistanceRT was selected as 1.5 kΩ. The current injected into the gridwas regulated through a proportional-integral regulator plusfeedforward at igrid = 8.5 A rms.

As stated above, the injection of both active and reactivepower was simulated; however, the switches being ideal andthe commutations instantaneous, performance did not dependon the power factor. For this reason, only the unity power factorsimulation results are reported. In the simulations, the gridvoltage angle information is available; hence, a PLL was notemployed.

Fig. 7 shows the output voltage and current under differentconditions of the dc voltage ratio. As expected, the THD of thegrid current increases with the dc voltage ratio, being 2.7%, 3%,and 3.3%, respectively, for Vfc/VDC = 0.33, Vfc/VDC = 0.5,and Vfc/VDC = 0.66.

Fig. 8 shows the performance of the TC with a parasiticcapacitance of the PV field of Cp = 200 nF. The groundleakage current results iground = 30 mA rms. Please note that

Fig. 8. TC behavior with a 200 nF parasitic capacitor.

only a common-mode inductor of 1 mH was employed in thissetup. The ground leakage current could be further reduced bya more accurate design of the common-mode filter.

In order to obtain further indications about the regulation ofthe flying-capacitor voltage, Fig. 9 reports the result of a stepvariation of Vfc from 150 to 190 V (inside the controllableregion in Fig. 5) occurring at time 0.1 s. As it can be seen,the average value of Vfc rapidly (in about 25 ms) rises to thereference value without any overshoot.

VI. EXPERIMENTAL RESULTS

A prototype nine-level converter was designed and built inorder to test the performance of the proposed solution. Theprototype is based on the Texas Instruments TMS320F28335microcontroller and features the two full bridges, the TC, thesensors, and the output filter. Infineon IKW30N60H3 IGBTs


Fig. 9. Vfc step variation response.

Fig. 10. Block scheme of the delay-based PLL.

Fig. 11. Schematic of the test bed employed for the experiments.

and ST Microelectronics STW55NM60ND MOSFETs wereemployed as active devices. The microcontroller implementsthe PWM signal generation, the current controller, and theproposed modulation. Synchronization with the grid is realizedwith a transport delay-based PLL structure (see Fig. 10). Theschematic is the same as that of the d–q PLL in a synchronousreference frame, except that the quadrature input signal andthe cosine of the estimated angle are realized with a constantdelay equal to 1/4 of the nominal grid voltage period T . Asdemonstrated in [41], this modification allows to obtain a zerosteady-state error for small-frequency variations of the inputsignal with respect to the nominal grid voltage one.

Fig. 11 shows a schematic of the test bed. The converterwas powered by an Agilent Technologies dc power supply,and its output was directly connected to the grid. In order tosimulate the parasitic capacitance of the PV field, a capacitorwas connected between the negative dc-link terminal and theneutral wire. Although in actual PV fields the parasitic capaci-

Fig. 12. Experimental results with unity power factor and Vfc = 0.33VDC.Vout (300 V/div), igrid (10 A/div), and vgrid (200 V/div). Time base 5 ms/div.



tance is distributed between both the positive and negative sidesof the dc link, for analysis purposes, it can be considered asconcentrated only on the negative side.

For consistency, the same circuit parameters were chosenas in the simulations, and the output current and voltage arereported for different dc voltage ratios. Figs. 12–14 refer to Vfc/VDC=0.33, Vfc/VDC=0.5, and Vfc/VDC=0.66, respectively.

For completeness, the output voltage of the HVFB, i.e., V HVout ,

and of the LVFB, i.e., V LVout is presented in Fig. 15. It can be

noted that the high-frequency voltage switching of the HVFBhappens only in a limited time interval during a period of thegrid voltage.

The experimental results show a low-frequency crossoverdistortion, absent in the simulations, that is more evident forVfc/VDC = 0.33. This behavior can be explained with the pres-ence of dead times between the commutations of the devices.As was shown in [32], extreme values of duty cycles lead to


Fig. 15. Experimental results with unity power factor and Vfc = 0.5VDC.Vout (300 V/div), V HV

out (300 V/div), and V LVout (200 V/div). Time base 5 ms/div.

Fig. 16. Experimental results with power factor 0.85 and Vfc = 0.5VDC.Vout (300 V/div), igrid (10 A/div), and vgrid (200 V/div). Time base 5 ms/div.

higher current distortion. When the dc voltage ratio is low,the duty cycle is forced to go all the way from 0% to 100%,leading to higher THD. The best THD performance is obtainedat Vfc/VDC = 0.5.

The other component of the low-frequency distortion canbe explained with the current controller chosen. In fact, itis clear from the experimental results that the grid voltagepresents a marked third-harmonic distortion, typical of LVdistribution grids. The finite rejection to disturbances of thecurrent controller is the cause of the low-frequency distortionnear the peaks of the grid voltage.

Reactive power operation is shown in Fig. 16 for Vfc/VDC =0.5. With respect to the simulations, the THD deteriorates dueto the low-frequency distortion. As there is no difference in thebehavior with inductive and capacitive power factor, only themeasurement with capacitive power factor is shown.

The TC waveforms are shown in Fig. 17, with a Cp = 200 nFcapacitor accounting for the parasitic capacitance of the PVfield. The obtained rms current of about 30 mA is in line withthe simulations.

The dynamic behavior of the proposed solution was evalu-ated imposing step variations of the desired injected grid currentand of the floating-capacitor voltage. To be consistent with thesimulation results, the test was run in the same conditions asin Fig. 9. Fig. 18 shows the dynamic system response in thepresence of a step variation of the injected grid current set pointfrom 1.2Arms to 7.4Arms. It is important to note that the currentstep variation, which can happen over a grid voltage period dueto a change in solar radiation, is typically lower. The higher stepvariation puts in evidence the effectiveness of the solution.

Fig. 17. TC behavior. vground (200 V/div) and iground (500 mA/div). Timebase 5 ms/div.

Fig. 18. Experimental results with step variation of the injected grid currentfrom 1.2Arms to 7.4Arms. igrid (10 A/div) and vgrid (100 V/div). Time base50 ms/div.

Fig. 19. Experimental results with step variation of Vfc from 150 to 190 V.igrid (20 A/div), vgrid (200 V/div), and Vfc (100 V/div). Time base 20 ms/div.

Fig. 19 shows the evolution of the flying-capacitor voltagewhen a step variation of its desired value is performed. Thedesired variation of Vfc is satisfied by the floating-capacitorvoltage regulation realized inside the PWM modulator, asdescribed in Section III. This experiment was carried out inorder to evaluate the dynamic behavior of the floating-capacitorvoltage regulation system. It can be seen that the desired Vfc

value (190 V) is reached after only one period of the gridvoltage. This result is in agreement with the simulations. Asimilar dynamic behavior can be obtained in the case of stepvariation of the input dc voltage VDC. The step variation of VDC

was not conducted because of test bed limitations. The ripple ofthe flying-capacitor voltage in Fig. 19 is due to the small valueof capacitance chosen (Cfc = 500 μF). Inverter manufacturersusually choose higher values of electrolytic capacitance in orderto increase the capacitor lifetime. However, the performance ofthe converter is good even with this small capacitor.


TABLE IIEXPERIMENTAL MEASUREMENTS OF EFFICIENCY η AND THD

Fig. 20. Spectrum of vout (20 dB/div) in the case of the proposed nine-levelconverter. Frequency base 10 kHz/div.

The efficiency and THD measurements were performedusing an N4L PPA5530 power analyzer without taking intoaccount the power consumed by the control logic or the gatedriver circuits. The results are summarized in Table II. The mea-surements were performed without any form of compensationfor dead times or voltage drops on the power switches.

Finally, the output voltage spectrum of the proposed solu-tion is shown in Fig. 20. This spectrum shows that the mainswitching harmonics are located at multiples of the switchingfrequency. A feature that can be seen from the FFT analysisis that several spectral lines are located near the multiples ofthe switching frequency, resulting in some degree of spread-spectrum output. This behavior is due to the switching patternvariations deriving from the regulation of the flying capacitor.

VII. CONCLUSION

This paper has proposed a novel nine-level grid-connectedtransformerless PV converter based on a CFB topology with twofull bridges, one of which is supplied by a floating capacitor.

A suitable PWM strategy was developed in order to improveefficiency (most power devices commutate at low frequency)and, with the help of a specific TC, minimize the groundleakage current.

The proposed PWM strategy can regulate the voltage acrossthe flying capacitor. Simulations were performed to assess theability to regulate the flying-capacitor voltage in a wide rangeof operating conditions.

Extensive simulations and experiments confirm the resultsof the theoretical analysis and show the good performance ofthe converter as far as ground leakage current and harmonicdistortion are concerned. Despite the use of traditional powerdevices for the laboratory prototype, the experimentally mea-sured efficiency was fairly good.

The proposed converter can continuously operate at arbitrarypower factors, has limited boosting capability, and can producenine output voltage levels with 11 power switches, of whichthree are low power switches for the TC and only four need tobe controlled by PWM.

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Giampaolo Buticchi (S’10–M’13) was born inParma, Italy, in 1985. He received the Master’s de-gree in electronics engineering and the Ph.D. degreein information technology from the University ofParma, Parma, Italy, in 2009 and 2013, respectively.

He is currently a Postdoctoral Research Associatewith the University of Parma. His research area isfocused on power electronics for renewable energysystems, static energy conversion, and motor drives.

Davide Barater (S’11) was born in Pontremoli,Italy, in 1983. He received the M.S. degree in elec-tronics engineering from the University of Parma,Parma, Italy, in 2009, where he is currently workingtoward the Ph.D. degree in information technology.

His research area is focused on power electronicsand static energy conversion.

Emilio Lorenzani (S’03–M’07) was born in Parma,Italy, in 1976. He received the Master’s degree inelectronics engineering and the Ph.D. degree in in-formation technology from the University of Parma,Parma, in 2002 and 2006, respectively.

Since 2011, he has been with the Departmentof Science and Engineering Methods, University ofModena and Reggio Emilia, Modena, Italy. He isthe author or coauthor of more than 50 technicalpapers and holds four industrial patents. His researchactivity is mainly focused on power electronics for

renewable energy resources, electric drives, and electric motor diagnostics.

Carlo Concari (S’98–M’06) was born in SanSecondo Parmense, Italy, in 1976. He received theM.S. degree in electronics engineering and the Ph.D.degree in information technology from the Uni-versity of Parma, Parma, Italy, in 2002 and 2006,respectively.

Since 2006, he has been an Assistant Professorwith the Department of Information Engineering,University of Parma. He is the author or coauthor ofmore than 40 technical papers. His research activityis mainly focused on power electronics, digital drive

control, static power converters, and electric machine diagnostics.

Giovanni Franceschini was born in Reggio Emilia,Italy, in 1960. He received the Master’s degreein electronics engineering from the University ofBologna, Bologna, Italy.

Since 1990, he has been with the Departmentof Information Engineering, University of Parma,Parma, Italy, where he was first an Assistant Pro-fessor and is currently a Full Professor of electricmachines and drives. His research interests in-clude high-performance electric drives and diagnos-tic techniques for industrial electric systems.

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