8vsbbook

8-VSB For DTV

By Doc Daugherty

Senior Broadcast Technology Instructor

Harris Corporation, Broadcast Division

11/19/03

Table of Contents11/19/03

Table of Contents1 8-VSB Overview. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

1.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .1

1.1.1 One-way Modem. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .1

1.1.2 Cliff Effect . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .1

1.2 MPEG Transport Layer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2

1.3 Transmission Stream . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2

1.4 Transport Stream and Transmission Stream Terminology . . . . . . . . . . . . . . . . . .3

1.5 Efficient Use of Frequency Spectrum . . . . . . . . . . . . . . . . . . . . . . . . . .3

2 8-VSB Transmitter Block Diagram Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

2.1 The Digital Exciter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5

2.2 The RF Up Converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5

2.2.1 +/- 54 Hz Transport Stream Bit Rate Variation . . . . . . . . . . . . . . . . . . . . . . . . .5

2.3 Digital Exciter Adaptive Equalization and the Down Converter . . . . . . . . . . . . . . . .6

2.4 The RF Power Amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6

2.5 The DTV Mask Filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6

3 The 8-VSB Exciter Block Diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

3.1 8-VSB Clock Recovery Block . . . . . . . . . . . . . . . . . . . . . . . . . . . . .7

3.1.1 Related 8-VSB Numbers. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .7

3.2 Data Randomizer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .8

3.3 Reed-Solomon Encoder. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .8

3.4 Data Interleaver . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .9

3.5 Variations of the DTV System . . . . . . . . . . . . . . . . . . . . . . . . . . . . .9

3.6 Trellis Encoder . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

3.7 Multiplexer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

3.8 Vestigial Sideband Modulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

3.8.1 Nyquist Filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

3.9 8-VSB Similar to Single Sideband Transmission . . . . . . . . . . . . . . . . . . . . . 12

4 Modulation Waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .13

4.1 Fitting Symbols Into a 6-MHz Channel . . . . . . . . . . . . . . . . . . . . . . . . 13

4.2 Developing the 8-VSB Eye pattern . . . . . . . . . . . . . . . . . . . . . . . . . . 15

4.2.1 Error Vector Magnitude . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

4.3 Constellation Pattern . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

4.3.1 Drawing An Eye Pattern . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

4.4 Noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

4.5 Linear Distortions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

Page: i


4.5.1 Amplitude Verses Frequency Response . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

4.5.2 Group Delay . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

4.6 Nonlinear Distortions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

4.6.1 Linearity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

4.6.2 ICPM (Incidental Carrier Phase Modulation) . . . . . . . . . . . . . . . . . . . . . . . . . 24

5 Achieving FCC Mask Specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .27

5.1 The New (Current) FCC Mask . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

5.2 Effects of Mask Filter on Adjacent Channel Intermods . . . . . . . . . . . . . . . . . . 29

5.3 Peak to Average Power Rating of DTV Signals and Amplifiers . . . . . . . . . . . . . . . 30

5.3.1 DTV Signal Peak to Average Power Rating. . . . . . . . . . . . . . . . . . . . . . . . . . 30

5.3.2 Amplifier Peak to Average Power Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

5.4 Amplifier Nonlinearity, Chief Cause of Intermod products . . . . . . . . . . . . . . . . . 30

5.5 Peak To Average Power Ratio vs Shoulder Intermod Levels . . . . . . . . . . . . . . . . 31

5.6 Resolution Bandwidth and Offsets for FCC Mask Measurements . . . . . . . . . . . . . . 34

5.6.1 Calculating the Offset . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

5.6.1.1 Shoulders Passed, What About the Rest of the Response . . . . . . . . . . . . . 35

5.7 Standard D Mask Filter Responses . . . . . . . . . . . . . . . . . . . . . . . . . . 36

5.7.1 Group Delay and Amplitude Response Correction . . . . . . . . . . . . . . . . . . . . . . 36

5.8 Sharp Tuned Filter Responses . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

5.8.1 Sharp Tuned Filter Group Delay. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

5.9 EVM, Eye Patterns, Amplitude Response and Group Delay Interactions . . . . . . . . . . . 38

5.9.1 PA Nonlinearity vs EVM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

5.9.2 Amplitude Response and Group Delay vs EVM . . . . . . . . . . . . . . . . . . . . . . . 39

5.10 Creating The FCC Mask Performance Exhibit . . . . . . . . . . . . . . . . . . . . . . 40

6 Dual Sideband Suppressed Carrier and Quadrature Modulation . . . . . . . . . . . . . . . . . . . . . . . . .45

6.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

6.2 Amplitude Modulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

6.2.1 Amplitude Modulation Sideband Frequencies. . . . . . . . . . . . . . . . . . . . . . . . . 46

6.2.2 Power Level of Sidebands and Carrier . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49

6.3 Generation Of Dual Sideband Suppressed Carrier Signals . . . . . . . . . . . . . . . . . 50

6.4 Operation Of A Diode Ring Modulator. . . . . . . . . . . . . . . . . . . . . . . . . 50

6.4.1 PIN Diode Mode of Operation of the Diode Ring Modulator . . . . . . . . . . . . . . . . . 50

6.5 Doubly Balanced Modulator Waveforms . . . . . . . . . . . . . . . . . . . . . . . . 54

6.6 Demodulating The Dual Sideband Suppressed Carrier Signal . . . . . . . . . . . . . . . . 55

6.7 Summary Of Dual Sideband Suppressed Carrier Modulation . . . . . . . . . . . . . . . . 55

6.8 Amplitude Modulation Produced From A Doubly Balanced Modulator . . . . . . . . . . . . 56

Page: ii


6.9 Phase Modulation Produced From A Doubly Balanced Modulator . . . . . . . . . . . . . . 58

6.10 Frequency Modulation Produced From A Phase Modulator. . . . . . . . . . . . . . . . . 60

6.10.1 FM and NTSC TV Aural Contain Both FM and PM . . . . . . . . . . . . . . . . . . . . . 60

6.11 Single Sideband And Quadrature Distortion. . . . . . . . . . . . . . . . . . . . . . . 61

6.12 QAM (Quadrature Amplitude Modulation) . . . . . . . . . . . . . . . . . . . . . . . 62

6.12.1 16 QAM Modulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63

6.13 THE SYNCHRONOUS DETECTOR . . . . . . . . . . . . . . . . . . . . . . . . . 64

6.14 The Quadrature Demodulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66

6.14.1 The I and Q Synchronous Detectors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66

7 Switch Mode of Operation of the Diode Ring Modulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . .67

7.1 Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68

7.2 Conclusion. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68

7.3 Conclusion. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71

7.3.1 Performance Test Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71

Appendix A Additional 8-VSB Topics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .73

A.1 Details of the Transport Stream and Symbol Rates . . . . . . . . . . . . . . . . . . . . 73

A.2 More About Segments and Fields . . . . . . . . . . . . . . . . . . . . . . . . . . . 73

A.2.1 Data Segment Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74

A.2.2 Field Sync Segment Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75

A.3 More About The Spectrum . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

A.3.1 Efficient Use of Power. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

A.3.2 Efficient Use of Frequency Spectrum . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77

A.3.3 8-VSB Similar to Single Sideband Transmission . . . . . . . . . . . . . . . . . . . . . . . 78

A.3.3.1 Single Sideband Waveforms, The Baseband Signal . . . . . . . . . . . . . . . . 78

A.3.3.2 Single Sideband Waveforms, The Modulation Envelope . . . . . . . . . . . . . 79

A.3.3.3 Adding The Pilot . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80

A.3.3.4 Adding Full Amplitude Carrier . . . . . . . . . . . . . . . . . . . . . . . . . . 81

A.4 Fitting Symbols Into a 6-MHz Channel . . . . . . . . . . . . . . . . . . . . . . . . 82

A.4.1 A Simpler Approach . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83

A.4.2 The Raised Cosine Filter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84

A.4.2.1 Raised Cosine Filter Alpha (a) Factor . . . . . . . . . . . . . . . . . . . . . . . 84

A.4.2.1.1 Ringing Period of a Raised Cosine Filter . . . . . . . . . . . . . . 84

A.4.3 Root Raised Cosine filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86

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Page: iv

List of Figures11/19/03

List of FiguresFigure 1-1 8-VSB Spectrums . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .4

Figure 1-2 8-VSB Sideband Response Within +/- 2 kHz of Pilot . . . . . . . . . . . . . . . . . . . . . . . . . .4

Figure 2-1 Simplified Block Diagram of the DTV Transmitter . . . . . . . . . . . . . . . . . . . . . . . . . . .6

Figure 3-1 8-VSB Transmission system block diagram. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .8

Figure 3-2 8-VSB Digital Exciter D/A Converter IF Output . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

Figure 4-1 Symbol Shape and Resulting Modulated Spectral Bandwidth . . . . . . . . . . . . . . . . . . . . . 13

Figure 4-2 Time Response of a Bandwidth limited Pulse . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

Figure 4-3 Symbol Levels for Bandwidth Limited Modulation . . . . . . . . . . . . . . . . . . . . . . . . . . 15

Figure 4-4 8-VSB Eye Pattern (Top) and Constellation (Bottom). . . . . . . . . . . . . . . . . . . . . . . . . 17

Figure 4-5 8-VSB Eye Pattern Arrangement. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

Figure 4-6 Effect of Noise on Eye Pattern . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

Figure 4-7 Effect of Amplitude Response on Eye Pattern . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

Figure 4-8 Effects of Group Delay. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Figure 4-9 Effect of Group delay on Eye Pattern . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

Figure 4-10 Effect of Linearity on the Eye Pattern . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

Figure 4-11 Effects of ICPM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

Figure 4-12 Effect of ICPM on Eye Pattern. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

Figure 5-1 FCC DTV Mask, New vs Old, With Reference Power at 0 dB . . . . . . . . . . . . . . . . . . . . 28

Figure 5-2 Effect of Mask Filter on Transmitter Response . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

Figure 5-3 Transmitter Peak Power Limiting, NTSV vs DTV . . . . . . . . . . . . . . . . . . . . . . . . . . 31

Figure 5-4 Amplifier Precorrection Area and Hard Clipping Level . . . . . . . . . . . . . . . . . . . . . . . . 32

Figure 5-5 PA Peak to Average Ratio Compared to DTV Shoulder Intermod Level . . . . . . . . . . . . . . . 33

Figure 5-6 DTV RF Response Shoulder Level for Various PA Peak to Average Power Ratios . . . . . . . . . 33

Figure 5-7 Effect of Resolution Bandwidth on DTV RF Response . . . . . . . . . . . . . . . . . . . . . . . . 34

Figure 5-8 Standard D-Mask Filter Response and Group Delay Responses. . . . . . . . . . . . . . . . . . . . 36

Figure 5-9 Sharp Tuned Filter and Standard D-Mask Filter Responses . . . . . . . . . . . . . . . . . . . . . . 37

Figure 5-10 Sharp Tuned Filter Group Delay Responses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

Figure 5-11 Effect of Equalization on DTV Mask Filter EVM Performance. . . . . . . . . . . . . . . . . . . . 39

Figure 5-12 Spread Sheet Exhibit, Proving Transmitter FCC Mask Compliance . . . . . . . . . . . . . . . . . 41

Figure 5-13 Graph of Results of Exciter A Spread Sheet (With Standard D-Mask Filter) . . . . . . . . . . . . . 42

Figure 5-14 Graph of Results of Exciter B Spread Sheet (With Sharp Tuned Filter). . . . . . . . . . . . . . . . 42

Figure 5-15 Formulas For FCC Mask Compliance Spread Sheet. . . . . . . . . . . . . . . . . . . . . . . . . . 43

Figure 6-1 An amplitude modulation scheme using a P I N diode. . . . . . . . . . . . . . . . . . . . . . . . . 46

Figure 6-2 Carrier, Upper Sideband, and Lower Sideband Vectors . . . . . . . . . . . . . . . . . . . . . . . . 47

Figure 6-3 Upper and Lower Sidebands Combine to Form Dual Sideband Signal . . . . . . . . . . . . . . . . 48

Page: v

List of Figures11/19/03

Figure 6-4 The sidebands, as viewed on a spectrum analyzer. . . . . . . . . . . . . . . . . . . . . . . . . . . 49

Figure 6-5 A diode ring modulator functioning as a doubly balanced modulator. . . . . . . . . . . . . . . . . 50

Figure 6-6 Modulation Signal Current Path (A) and RF Current Path (B) for Positive Modulation . . . . . . . 52

Figure 6-7 Modulation Signal Current Path (A) and RF Current Path for Negative Modulation . . . . . . . . . 53

Figure 6-8 Waveforms Associated With a Doubly Balanced Modulator (top three waveforms) . . . . . . . . . 54

Figure 6-9 Amplitude Modulator Using A Doubly Balanced Modulator . . . . . . . . . . . . . . . . . . . . . 56

Figure 6-10 Vector Addition In-Phase Carrier and Sidebands (Amplitude Modulation) . . . . . . . . . . . . . . 57

Figure 6-11 Phase Modulator Using Doubly Balanced Modulators . . . . . . . . . . . . . . . . . . . . . . . . 58

Figure 6-12 Vector Addition of Carrier and Sidebands Which Are 90× Out Of Phase (Phase Modulation) . . . . 59

Figure 6-13 Frequency Modulation From a Phase Modulator . . . . . . . . . . . . . . . . . . . . . . . . . . . 60

Figure 6-14 Vector Addition of Carrier and Upper Sideband Only. . . . . . . . . . . . . . . . . . . . . . . . . 61

Figure 6-15 QAM Modulator (Left) and Modulation Vector Diagrams (Right) . . . . . . . . . . . . . . . . . . 62

Figure 6-16 16-QAM Modulator And Modulation Constellation. . . . . . . . . . . . . . . . . . . . . . . . . . 63

Figure 6-17 A Form of a Synchronous Detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 64

Figure 6-18 Synchronous Demodulator Waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65

Figure 6-19 16-QAM Demodulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66

Figure 7-1 A diode ring modulator functioning as a doubly balanced modulator. . . . . . . . . . . . . . . . . 67

Figure 7-2 Positive Carrier Polarity (Top) and Resultant Modulation Current Flow (Bottom) . . . . . . . . . . 69

Figure 7-3 Negative Carrier Polarity (Top) and Resultant Modulation Current Flow (Bottom) . . . . . . . . . 70

Figure A-1 8-VSB Data Frame . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74

Figure A-2 Trellis-Coded 8-VSB Data Segment Format. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75

Figure A-3 Field Sync Segment Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

Figure A-4 8-VSB and NTSC Spectrums . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77

Figure A-5 Baseband Modulating Signal. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78

Figure A-6 Single Sideband Modulation Envelope . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79

Figure A-7 Single Sideband Signal With Pilot Sample Added . . . . . . . . . . . . . . . . . . . . . . . . . . 80

Figure A-8 Adding Full Amplitude Carrier to the Sideband Signal . . . . . . . . . . . . . . . . . . . . . . . . 81

Figure A-9 Symbol Shape and Resulting Modulated Spectral Bandwidth. . . . . . . . . . . . . . . . . . . . . 82

Figure A-10 Pulse shape and Bandwidth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83

Figure A-11 Details About Raised-Cosine Filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85

Figure A-12 Creating a Raised Cosine Filter With Two Root Raised Cosine Filters . . . . . . . . . . . . . . . . 86

Figure A-13 DTV System Raised Cosine Filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86

Page: vi

List of Tables11/19/03

List of TablesTable 1-1 Typical DTV AERP Allocations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

Table 1-2 Transport and Transmission Stream Terminology . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

Table 3-1 Symbol Composition for 2-VSB through 16-VSB. . . . . . . . . . . . . . . . . . . . . . . . . . . .10

Table 3-2 2-, 4-, and 8-VSB Symbol Bits and Levels . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .10

Table 3-3 16-VSB Symbol Bits and Levels. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .10

Table 5-1 Calculating Required Shoulder Levels With Various Offsets . . . . . . . . . . . . . . . . . . . . . .35

Table 5-2 Channel Center Frequency and Pilot Frequency (With No Offset) . . . . . . . . . . . . . . . . . . .44

Table 6-1 Bit Structure and Level Assignment for a 2-Bit Symbol . . . . . . . . . . . . . . . . . . . . . . . .63

Page: vii

List of Tables11/19/03

Page: viii

8-VSB Overview11/19/03

1 8-VSB Overview

1.1 Introduction

The 8-VSB Digital TV transmission system carries the high-definition DTV signal in a 6-MHz channel. This statement seems impossible considering the amount of information that is conveyed and the fact that it offers improved picture, sound and reception quality. The purpose of this paper is to attempt to explain the basics of 8-VSB operation and remove some of the mystery of how DTV can fit into such a small channel.Part of the answer is that the NTSC system is not able to compress the signal as well as DTV, also, NTSC does not achieve efficient use of its broadcast channel frequency spectrum or its transmitting power.

This system is capable of offering the following types of service over a single 6-MHz channel.

• One high definition TV program with 5.1 channel AC-3 sound (or)

• Four to six standard definition programs

• One high definition program and four standard definition programs

• Other types of data services with either of the above.

1.1.1 One-way Modem

The 8-VSB system is a one-way modem from the transmitter to the receiver, therefore, much forward error correction is required. Forward error correction includes:

• Reed-Solomon encoding, which corrects for short duration impulse noise

• Data interleaving, which corrects for long duration impulse noise

• Trellis encoding, which is good in an additive white noise environment.

These processes will be discussed in greater detail later in this report.

1.1.2 Cliff Effect

Analog TV viewers experience graceful degradation as the distance from the transmitter increases. This happens because the signal gets weaker and closer to the noise floor. As this happen, the picture gets more snow until it cannot be viewed.

At the receiver, two possibilities exist for the DTV picture and sound. They are either perfect or they are blanked out by the receiver. The picture and sound are blanked when the byte error rate exceeds 2.4 uncorrected errors per second. This is referred to as the cliff effect, also referred to as the “noise limited coverage contour”. For 8-VSB the cliff effect occurs when the RF signal to noise ratio drops to 15 dB or lower.

The FCC has attempted to make the noise limited coverage contour equal to the NTSC Grade B coverage contour for that station through it allocation of DTV station average effective power levels (AERP).

Note: DTV power levels are measured in average power due to the random nature of the DTV RF signal peaks.

Figure 1-1 lists the typical NTSC effective radiated power (ERP) peak of sync power levels and DTV AERP levels for the various bands.

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Rules adopted by the FCC might allow a UHF DTV station with 50 kW AERP station to increase their AERP to 200 kW, or for any UHF DTV station to increase their AERP up to 1 MW if they increase their beam tilt to meet specified interference requirements and limit their AERP at the horizon to the lower (50 to 200 kW) assigned value.

Some UHF analog to UHF digital stations have been granted AERP levels in the 200kW to 1000kW range without excessive beam tilt.

1.2 MPEG Transport Layer

The MPEG transport layer (also referred to as the transport stream) delivers its data as a series of packets to the input of the 8-VSB exicter. The following is a list of the packet parameters.

• Each packet contains 188 bytes

• One byte consists of eight bits

• Datarate is 19.39 Mbits/s +/-54 Hz

• Packet repetition time is 77.57 microseconds

• A packet may contain video, audio, or ancillary data

• The first byte of each packet is the packet sync (47 hex).

1.3 Transmission Stream

The transport stream enters the exciter and, as a result of the digital processing, becomes the transmission stream. This signal is converted to the analog IF signal, which is up converted to the exciter RF output signal, which is then amplified, filtered, and radiated over the air.

Table 1-1 Typical DTV AERP Allocations

NTSC and DTV Band Allocation NTSC ERP peak sync DTV AERP

VHF low band NTSC to VHF low band DTV 100 kW 6.25 kW

VHF high band NTSC to VHF high band DTV 316 kW 19.75 kW

UHF NTSC to UHF DTV Up to 5000 kW 50 to 200 kW

VHF (low or high band) NTSC to UHF DTV VHF ERPs listed above 1000 kW

Note 1. UHF NTSC to UHF DTV stations can apply for 1000 kW AERP using added beam tilt so that the AERP at the horizon remains at the lower, licensed value.

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1.4 Transport Stream and Transmission Stream Terminology

During the process of converting the transport stream to the transmission stream, a potentially confusing change of terminology occurs. a comparison of some of these terms is given in Table 1-2.

1.5 Efficient Use of Frequency Spectrum

Refer to Figure 1-1. The useful portion of the 6-MHz DTV channel is 5.38 MHz. This amounts to 90% of the allocated channel. 0.31 MHz is used on either end of the channel to provide room to roll off the signal and reduce adjacent channel interference. The frequency efficiency is even greater when the MPEG-2 compression methods are taken into account. In the NTSC system, 4.2 MHz represents the useful video bandwidth and 0.1 MHz is the useful aural bandwidth, which is 72% of the 6 MHz channel.

Table 1-2 Transport and Transmission Stream Terminology

Transport Stream Terminology Transmission Stream Terminology

Bits and bytes Symbols

Packet Segment

Packet sync Segment sync

Transport stream clock Symbol clock

Field sync

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Figure 1-1 8-VSB Spectrums

Figure 1-2 8-VSB Sideband Response Within +/- 2 kHz of Pilot

6.0 MHz

5.38 MHz0.31 MHz 0.31 MHz

Pilot (carrier sample)

8-VSB Spectrum

0.7 (-3 dB)

1.0 (0 dB)

(-50 dB *)

* The IMD level at the edge of the channel (shoulder) is typically -50 dB at the outputof the exciter, and must be at least -37 dB at the output of the RF mask filter.The shoulder is 0.25 MHz into the upper and lower adjacent channels. Set spectrum analyzer for a Resolution Bandwidth and Video Bandwidth of 30 KHz each.

+1 kHz

Pilot

-1 kHz

Within a 2 kHz bandwidth the DTV RF signal has a pilot (carrier sample) and relatively flat upper and lower sidebands.

This helps the narrow band (2 kHz) DTV receiver phase lock loop achieve a pilot lock to recreate the carrier for use in the in-phase and quadrature synchronous detectors.

Carrier to pilot lock is possible down to a 0 db receiver RF signal to noise ratio. For 8-VSB this is 15 db below the receiver cliff effect threshold.

USBLSB

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8-VSB Transmitter Block Diagram Overview11/19/03

2 8-VSB Transmitter Block Diagram Overview

Refer to Figure 2-1. The DTV transmitter consists of the following units.

• The 8-VSB exciter, which consists of:

• The digital exciter chassis

• The RF up/down converter chassis

• The RF amplifier system

• The RF output straddling filter.

2.1 The Digital Exciter

The digital exciter chassis converts the transport stream into a transmission stream, which is a modulated 8-VSB trellis signal with a symbol rate of 10.762 MHz, a usable bandwidth of 5.38 MHz, a total bandwidth of 6 MHz, a center frequency of 10.76 MHz, and a pilot frequency of 8.07 MHz, see Figure 3-2, on page 12.

2.2 The RF Up Converter

The RF up converter heterodynes the IF signal up to the on-channel frequency in two conversions; its on-channel output pilot frequency is locked to an external 10 MHz reference. The output of the first mixer is an IF signal with a 44 MHz center frequency. In the CD-1A exciter, the following corrections are accomplished in the 44 MHz IF section of the RF up converter.

1. All manual precorrection (phase, linearity, response, and group delay) is accomplished in the 44 MHz IF section.

2. The +/- 22.5 Hz variation of the digital exciter RF output signal pilot is tracked out in the phase lock loop of the 54.76 MHz VCO of the 44 MHz IF mixer. This VCO phase lock loop is refer-enced to an external 10 MHz reference.

2.2.1 +/- 54 Hz Transport Stream Bit Rate Variation

The DTV transport stream bit rate of 19.392658 MHz, which is a function of the studio clock, is required to be within +/-54 Hz of its nominal value. It can be shown that this +/-54 Hz maximum variation would cause the symbol rate and channel center frequency to have a +/-30 Hz tolerance, and the pilot to have a +/-22.5 Hz tolerance. When the precise frequency offset is used, an FCC mandated tolerance of +/-3 Hz is required. The +/-22.5 Hz pilot error is tracked out in the IF oscillator phase lock loop in the RF upconverter chassis. From that point and onward, the pilot frequency is controlled by the 10 MHz (internal or external) reference frequency.

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8-VSB Transmitter Block Diagram Overview11/19/03

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2.3 Digital Exciter Adaptive Equalization and the Down Converter

The digital exciter receives an RF-output sample from the down converter, see Figure 2-1. The down converter board in the up/down converter chassis heterodynes the RF-output sample back down to the 10.762 MHz IF signal center frequency. Since the down converter and up converter use the same two local oscillators, the IF sample signal is once again referenced to the 10.76 MHz digital chassis clock.

The digital exciter uses the converted output sample to perform digitally processed phase, linearity, response, and group delay precorrection (adaptive equalization) of the transmitter output signal. Manual precorrection is also available. Manual precorrectors are located in the RF upconverter.

Figure 2-1 Simplified Block Diagram of the DTV Transmitter

2.4 The RF Power Amplifier

The RF power amplifier increases the power output of the RF up converter to the required output power level. Due to its random nature, the 8-VSB output signal does not have a consistent peak power level, but the average level of the signal is very stable. Therefore, DTV transmitter output power is measured by its average RF output level. The peak power level occasionally ranges as high as +10 dB above its average level, but 99.9% of the time the peak power level is less than or equal to +6db (4 x) greater than the average level.

Generally, the peak power rating of an RF PA must be 6 dB (4 x) greater than its average power rating to provide adequate headroom for the signal peaks. For example, a transmitter that will deliver a 25 kw average DTV power level must have a rating of 100 kw peak power.

The PA linearity determines the spurious sideband regeneration at the shoulder (the first 0.5 MHz beyond the upper and lower channel limit) of the DTV channel.

2.5 The DTV Mask Filter

The DTV mask filter will attenuate the spurious sideband signals in the range beyond 0.5 MHz above and below the DTV channel, to make the transmitter compliant with the FCC mask.

8-VSB Exciter

DigitalExciterChassis

Up

DTV Transmitter

RFPowerAmplifier

DTV

Filter

MPEG IITransportStreamPackets(188 Bytes at19.4 MBits/s)

8-VSBRFOutput

Mask

DownConverter

Converter

Chassis

Both signals have10.76 MHz centerfrequency IF

External10 MHzReferenceInput

CD-1 and Apex RTAC Sample

Additional RTAC Samples

IPA

For APEX Exciter

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The 8-VSB Exciter Block Diagram11/19/03

3 The 8-VSB Exciter Block Diagram

This section will involve a more detailed study of the 8-VSB exciter block diagram.

3.1 8-VSB Clock Recovery Block

The block diagram of the 8-VSB transmission system is shown in Figure 3-1, on page 8. The first block of the block diagram recovers the transport stream clock signal and determines the field synchronizer parameters, which are:

• Start of each segment (packet)

• Start of each field

It discards the packet sync byte, and replaces it (later) with the segment sync pulse (which has the same length), thus maintaining the same data efficiency rate. 187 bytes remain in the packet, giving it a data rate of 19.29 Mbits/s. Correct operation of this system requires each data packet to be processed at the same rate (77.57 microseconds). If the packets are transmitted too slow, some of the information must be repeated to maintain the same program reproduction time; if transmitted too fast, some of the information must be dropped. Packets are transmitted at the correct rate by changing the data bit rate as necessary as the signal is processed through the various stages of the exciter.

3.1.1 Related 8-VSB Numbers

All of the numbers listed below end in non-terminating decimals. Since they are all related to the 10.76 MHz clock, that is the only number you need to know down to the Hz.

It is 10.762238MHz

The 43.048 and 10.762 MHz clocks are phase locked to the 19.39 MHz transport stream clock. Therefore, the +/-54 Hz variation of the transport stream clock will cause +/-120 Hz variation to the 43.048 MHz clock and +/-30 Hz variation to the 10.76 MHz clock.

The relationship between the 10.76, 43.048, and 19.39 MHz clocks are as follows.

• 43.048 MHz = 4 x 10.76 MHz clock

• 19.39 MHz = 141 x (43.048/313) MHz

Other specifications are also tied to the 10.76 MHz clock. They include:

• The 5.38 MHz occupied channel bandwidth = 1/2 x 10.76 MHz clock.

• The 310 kHz channel guard bands = 1/2 x (6MHz -5.38MHz)

• The 10.76 Mhz IF center frequency = 10.76 MHz clock

• The 8.069 MHz IF pilot frequency = 3/4 x 10.76 MHz clock

• The 2.69 MHz difference between channel center and pilot frequency (10.762/4).

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Figure 3-1 8-VSB Transmission system block diagram.

3.2 Data Randomizer

With the exception of the segment and field syncs (to be discussed later), the transmitted signal frequency response requires a flat noise-like spectrum in order to use the allotted bandwidth with maximum efficiency. Any recurring patterns would cause the RF energy to concentrate in some parts of the spectrum and leave vacancies in others. This could cause discernible beat patterns in NTSC television receivers. With random noise-like data, the interference to a NTSC signal appears as snow or a uniform grey background.

In the data randomizer, each bit value is changed according to a known pattern of pseudo-random numbers. In the receiver, the process is reversed to recover the original data values.

3.3 Reed-Solomon Encoder

The Reed-Solomon encoder adds 20 parity bytes for every data packet. This encoder provides error correction for up to 10 byte errors per packet, good burst (impulse) noise correction capability, and maintains high data efficiency of 187/207, which is 90.3%. The data rate at this point is 21.35 Mbits/sec (for the 207 byte packets), which translates to a packet repetition time of 77.57 microseconds.

ClockRecovery

Data Randomizer

Reed-SolomonEncoder

DataInterleaver

ClockRate

10.76 MHzClock Output

Control

Segment Sync

Field Sync

RF Up-Converter

MPEGPacketInput(188 Bytes at19.4 Mbit/s)

TrellisEncoder Multiplexer

832 Symbolsin 77.3 usec

43.048 MHz Clock

19.39 MHz Clock

Pilot (dc level)

RF Sample FromRF Mask Filter

Adaptive EqualizationCorrector

D/AConverter

A/DConverter

10.76 MHzClock Input

10 MHz ExternalReference Input

OutputRF Down-Converter

Power FCC MaskFilterAmplifier

NyquistFilter

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3.4 Data Interleaver

The data interleaver convolutionally interleaves the data over many segments (to a depth of approximately 1/6 field) to help protect against the effects of noise bursts. The segment and field syncs are not interleaved; this helps the receiver locate the segment sync and recover the symbol clock independent of data eye opening. Data eye opening is covered later (under Error Vector Magnitude on page 13-16).

3.5 Variations of the DTV System

The DTV system includes five symbol arrangements, each of which offers trade-offs in the incoming data rate and the cliff effect threshold. The symbol rates include 2-VSB, 4-VSB, 8-VSB, 8-VSB-Trellis, and 16 VSB.

The incoming transport stream data rate determines the system capacity in terms of video definition, number of program streams, or data capacity. Higher data rates increase the capacity.

The cliff effect threshold is one determining factor in the location of the noise limited coverage contour.

Example: For a given AERP, a higher cliff effect threshold reduces the distance to the noise limited coverage contour. Higher AERP and/or greater transmitting antenna height are re-quired to recover the lost noise limited coverage area.

Table 3-1 lists the symbol composition for the various VSB levels for DTV along with their data rates and cliff effect thresholds. The symbol rate is for all of the VSB modes is 10.762 MHz.

A symbol can have any of number of levels, the exact number being dependent on the number of bits contained within the symbol. Table 3-2 is the truth table for 2-VSB, 4-VSB, and 8-VSB respectively and Table 3-3 is the truth table for 16-VSB. The relative symbol levels for the various modes in these two tables is assigned by the lookup table

The total number of levels (indicated by the number that precedes VSB) for any VSB mode will constitute 100% modulation. As the number of levels increase, the percent of modulation between the levels decreases. This causes the cliff effect threshold and data capacity to increase as number of levels increases.

8-VSB Trellis is a compromise mode that offers a higher data rate along with a reasonably low cliff threshold.

8-VSB Trellis is recommended for terrestrial broadcasting, where good noise immunity is required, and 16-VSB is recommended for cable TV, where data rate is important and noise is less of a problem.

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Table 3-1 Symbol Composition for 2-VSB through 16-VSB

VSB Mode Description Incoming Data Rate Cliff Threshold Truth Table

2-VSB One symbol contains 1 data bit 9.9 Mbit/second 9.5 dB See Table 3-2

4-VSB One symbol contains 2 data bits 19.39 Mbit/second 16.5 dB See Table 3-2

8-VSB One symbol contains 3 data bits 28 Mbit/second 22.4 dB See Table 3-2

8-VSB-Trellis One symbol contains 2 data bits plus 1 redundant bit

19.39 Mbit/second 15 dB See Table 3-2

16-VSB One symbol contains 4 data bits 38.6 Mbit/second 28.3 dB See Table 3-3

Table 3-2 2-, 4-, and 8-VSB Symbol Bits and Levels

2-VSB 4-VSB 8-VSB

Symbol Data Bits

Relative Symbol Level

Symbol Data Bits


Symbol Data Bits


1

Loo

kup

Tab

le +5 11

Loo

kup

Tab

le

+3 111

Loo

kup

Tab

le

+7

0 -5 10 +1 110 +5

01 -1 101 +3

00 -3 100 +1

011 -1

010 -3

001 -5

000 -7

Table 3-3 16-VSB Symbol Bits and Levels

16-VSB

Symbol Data Bits


Symbol Data Bits


0111

Loo

kup

Tab

le

-1 1111

Loo

kup

Tab

le

+15

0110 -3 1110 +13

0101 -5 1101 +11

0100 -7 1100 +9

0011 -9 1011 +7

0010 -11 1010 +5

0001 -13 1001 +3

0000 -15 1000 +1

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3.6 Trellis Encoder

The trellis encoder creates the data symbols used for 8-VSB-Trellis transmission. The trellis encoder adds additional redundancy to the signal, which allows further error correction in the receiver. It adds one redundant bit to every two data bits provided by the data interleaver. This makes up the three-bit symbol. This type of error correction is good in an additive white noise environment.

3.7 Multiplexer

The Segment and Field syncs are added in the multiplexer. The segment sync is easily recovered by the receiver (since it is independent of data) down to a S/N ratio of 0 dB. The symbol clock is imbedded in the segment sync.

The field sync starts with a segment sync pulse and like a segment, contains 828 symbols. It is received independently of data and can be used as a known reference training signal for the receiver equalizer. It is used as an aid in the receiver data field synchronization.

3.8 Vestigial Sideband Modulation

The Vestigial sideband modulation provides a filtered (root-raised cosine) IF signal, see Figure 3-2 for the signal spectrum. The root-raised cosine filter causes the signal to fit into a 6-MHz bandwidth. The usable response extends between the -3 dB points (0.7 amplitude) on the spectrum. These points are located 0.31-MHz from each end of the 6-MHz channel, making the usable bandwidth 5.38 MHz. The rapid roll off provides maximum usable bandwidth coincident with minimum adjacent channel interference. The pilot is created by adding a dc reference to the modulator input signal. Many of the characteristics of the modulated signal are reminiscent of the single sideband suppressed carrier signals used in radio systems. The modulated RF output signal has a 10.76 MHz center frequency and an 8.07 MHz pilot.

The VSB signal modulator can be either analog or digital, dependent upon the preferences of the design engineer. In Figure 3-1 no specific modulator block is shown because the modulation occurs as a result of the digital processing.

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3.8.1 Nyquist Filter

The nyquist filter block in Figure 3-1 shapes the response of the IF output of the digital to analog (D/A) converter, see in Figure 3-2.

Figure 3-2 8-VSB Digital Exciter D/A Converter IF Output

3.9 8-VSB Similar to Single Sideband Transmission

The 8-VSB system is similar to the single sideband reduced carrier technique used in radio. In both systems, only one sideband is transmitted along with a sample of the carrier (although, some radio transmissions totally eliminate the carrier sample). In the 8-VSB system, the carrier sample is called the pilot. Due to practical filter hardware constraints, a vestige of the lower sideband (0.31 MHz) is transmitted. This is achieved through the use of a 95 pole digital filter.

0.7 (-3 dB)

1.0 (0 dB)

6.0 MHz

5.38 MHz

Pilot

Digital Exciter - D/A Converter IF output

(-50 dB)

8.07 MHz +/-22.5 Hz Pilot Frequency

10.76 MHz +/-30 Hz Center Frequency

0.31 Mhz0.31 Mhz

Note: The center and pilot frequency tolerances result fromthe +/-54 Hz tolerance allowed the transport stream clock.

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Modulation Waveforms11/19/03

4 Modulation Waveforms

4.1 Fitting Symbols Into a 6-MHz Channel

One 832 symbol segment occurs in 77.3 usec, which produces a symbol rate of 10.76 MHz, as shown below.

It seems difficult to fit this symbol rate into a 6 MHz bandwidth, but it can be done. The RF spectrum occupied by the signal depends on the shape of the symbol pulse when it is modulated. Refer to Figure 4-1. If the symbol pulse shape is rectangular, the resulting spectrum is much wider than 6 MHz; but since the symbols take the shape of a ringing sine waves, the modulated signal spectrum is less than 6-MHz.

The modulated signal is passed through a root-raised-cosine filter to create the ringing sine wave-shaped symbols which keeps the bandwidth within 6-MHz.

Figure 4-1 Symbol Shape and Resulting Modulated Spectral Bandwidth

Symboltime77.3µs

832symbols------------------------------- 92.9ns= =

Symbolrate1

symboltime------------------------------- 1

92.9ns---------------- 10.762MHz= = =

DTV RF Envelope Sample (correct approach) DTV spectrum (fits into a 6-MHz channel)

Frequency

Amplitude

Frequency

Amplitude

Spectrum resulting from square pulse modulation(too wide to fit into 6 MHz)

RF envelope with Squarepulses (wrong approach)

Time

Amplitude

Time

Amplitude

6 MHz

12 to 15 pre and postringing cycles per symbol

The peak value representsthe symbol level.

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Each modulation symbol will have between 12 to 15 pre and post ringing cycles along with the modulation pulse; therefore, at any given instant, a modulation symbol pulse plus about 24 to 30 ringing cycles from other symbol pulses will be present in the filtered base band signal and the modulated signal. The question is how the actual modulation level of the signal is maintained in the presence of so much ringing energy? The answer is provided by the pulse timing and the filter characteristic, as shown below. At the instant a given modulation symbol is at its peak, all of the other ringing pulses will be going through their zero axis, see Figure 4-2.

Figure 4-2 Time Response of a Bandwidth limited Pulse

In Figure 4-3, three symbols of levels +5, -3, and +7 along with their ringing pulses are being summed. Notice that the ringing pulses of the other two waveforms are going through the zero axis when the peak amplitude of the symbol pulse is present. Therefore, at that time (vertical line), only one symbol pulse contributes to the total signal amplitude. When observing the composite waveform (on the right side of Figure 4-3), the correct symbol level occurs on the slope of the waveform. The peak of the waveform represents the sum of the ringing voltages present and will, on the average, be 6 dB greater then the RMS level of the waveform.

Since the actual symbol modulation level is on the slope of the waveform and not at the peak (as shown on the waveform on the right side of Figure 4-3), a small symbol-reading (timing or phase) error can result in a large symbol amplitude error. Therefore, an extremely accurate symbol clock is necessary to accurately read the symbol levels.

Notice also, that any signal amplitude or phase error, or any noise riding on the signal will result in a large symbol error. This leads us to a method of evaluating the accuracy of the symbol levels. It is called the “error vector magnitude,” but before that subject is discussed, the 8-VSB Eye pattern must be considered.

Time

Amplitude

Each pulse falls on the zero axis of the ringing waveforms of the other pulses.

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On the left side of Figure 4-3, the symbol levels are stated to be +5, -3, and +7. The amplitude of the symbol pulse tells the value of the symbol, and the polarity of the symbol is determined by the phase of the RF within the symbol envelope compared to the phase of the RF carrier sample (pilot).

Figure 4-3 Symbol Levels for Bandwidth Limited Modulation

4.2 Developing the 8-VSB Eye pattern

The signal level that represents each symbol falls on one of eight horizontal lines, as shown in the waveform on the right side of Figure 4-3. Each of these lines represents one of the eight possible modulation levels.

The eye pattern is created by synchronizing an oscilloscope on the demodulated waveform; with the horizontal trace consisting of two symbol periods. The symbol levels of the various signal waveforms all cross exactly between the eyes, as shown at the top portion of Figure 4-4.

Each vertical column of seven eyes represents the eight modulation levels of the 8-VSB signal. Only the symbol levels are transmitted at this time since all of the ringing cycles from the other symbols are going through their zero axis. The signal levels shown between the eyes represent the sum of the pre- and post-ringing cycles of the several up-coming and past symbols. These levels are random and can add up to various levels, occasionally up to 10 dB above the average signal level.

The time between the columns of eyes is the symbol time (92.92 nsec). Symbol time is the reciprocal of the symbol frequency (10.762 MHz).

=+

+

+5

-3 +7

+7+5+3+1-1-3

-5-7

SymbolClock

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Referring to Figure 4-4, as each oscilloscope sweep occurs, an additional trace is added to the eye pattern. At the symbol time, each trace settles to its appropriate symbol level, and between symbol times the trace represents the instantaneous sum of the ringing voltages. As additional sweeps occur, the repeated traces form the eye pattern.

If the signal is perfect (no noise or distortions present), the size of the eye opening will be maximum; but any distortion or noise causes the individual traces to miss their correct level at symbol time. This closes the eyes. If a trace travels through the center of the eye, half way between two symbol levels, that symbol level cannot be accurately read and an error occurs. If many traces miss their symbol levels by various errors up to or greater than the above described 50%, the eyes close and the DTV receiver may cliff off.

4.2.1 Error Vector Magnitude

Error Vector Magnitude (EVM) is the sum of all components that reduce the open area of the 8-VSB eyes. 12.5% EVM will close the eyes.

These components include IMD, spurious signals, phase noise, gaussian noise, amplitude response errors, incidental carrier phase modulation (ICPM), group delay and indirectly by linearity errors because it can cause increased IMD.

4.3 Constellation Pattern

The constellation pattern is shown at the bottom of Figure 4-4. This pattern results from displaying the outputs of the I (in phase) and Q (quadrature) synchronous demodulators of an 8-VSB receiver or test instrument on an oscilloscope. The output from the I demodulator is applied to the external horizontal input of the oscilloscope and the output from the Q demodulator is applied to the vertical input. The symbol clock from the receiver undergoes pulse shaping and is then applied to Z axis of the scope. This caused the display to intensify when a symbol is present.

Since the I demodulator detects the amplitude modulation of the waveform, the symbols form eight vertical lines on the scope, with each line representing a specific symbol level on the horizontal scale.

The Q demodulator detects the phase modulation component of the RF signal. Since 8-VSB is a single sideband system, the quadrature error of the single sideband modulation causes much phase modulation. This phase modulation causes the symbols to form vertical lines on the scope. If the system had both upper and lower sidebands, no quadrature error would result and the display would consist of a single horizontal row of dots (symbols) with each dot landing on one of the eight symbol levels. Phase distortion of the RF signal causes the vertical lines to curve at the top and bottom of the display.

Page: 16


Figure 4-4 8-VSB Eye Pattern (Top) and Constellation (Bottom)

I Modulation Axis

-7 -5 -3 -1 +1 +3 +5 +7

Q M

odul

atio

n A

xis

+7

+5

+3

-7

-5

-3

-1

+1

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4.3.1 Drawing An Eye Pattern

Use the pattern in Figure 4-5 draw several traces. Allow each of these traces to cross exactly between the eyes. Five dotted traces are shown. If enough traces are added, an eye pattern such as shown in Figure 4-4 will result. Eye patterns of several hundred traces (samples) are typical. The number of samples shown may be selectable on some equipment.

Figure 4-5 8-VSB Eye Pattern Arrangement

-7

-5

-3

-1

+1

+3

+5

+7

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4.4 Noise

Any kind of noise will cause the eye to close from the top and bottom because of the noise riding on the signal. Low frequency noise will make the trace appear fatter, high frequency noise will make it look fuzzy, see Figure 4-6.

Figure 4-6 Effect of Noise on Eye Pattern

-7

-5

-3

-1

+1

+3

+5

+7

High frequency noise on signal.

Low frequency noise on signal.

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4.5 Linear Distortions

Waveform distortions which are frequency dependent are called linear distortions. They result from a system which is unable to uniformly transfer amplitude and phase characteristics at all frequencies. Linear distortions include:

• Amplitude response, which are amplitude verses frequency errors.

• Group delay, which are transit time verses frequency errors.

4.5.1 Amplitude Verses Frequency Response

These errors will cause some of the waveforms to miss the ideal crossing points of the eye pattern. Refer to Figure 4-7. Two low frequency traces (long dashed lines) have the correct amplitude response, but the two short dashed traces represent a high frequency amplitude response rolloff. Draw some traces showing amplitude response problems.

Figure 4-7 Effect of Amplitude Response on Eye Pattern

-7

-5

-3

-1

+1

+3

+5

+7

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4.5.2 Group Delay

Group delay causes groups of frequencies to take different times to get through a system. This will offset waveforms of different frequencies, as shown in Figure 4-8.

Figure 4-8 Effects of Group Delay

Two sine waves of different frequencies in phase, no group delay.

Same two sine waves from above. The timing offset between the waveforms(compare to above) demonstrates the effects of group delay.

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Group delay causes some of the waveforms to move to the left or right, which closes the eye, see Figure 4-9.

Figure 4-9 Effect of Group delay on Eye Pattern

-7

-5

-3

-1

+1

+3

+5

+7

Correct trace with symbols levels of +7, -5, +3, and -3.

Same trace delayed by group delay. Symbol levels are +5, -3, unknown, and -7.

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4.6 Nonlinear Distortions

Waveform distortions which are amplitude level dependent are called nonlinear distortions. They result in amplitude caused changes in amplitude and phase. Typical nonlinear distortions are:

• Linearity (also referred to as AM to AM conversion)

• ICPM (incidental carrier phase modulation), also referred to as AM to PM conversion.

4.6.1 Linearity

Linearity problems change the spacing between the modulation levels of the 8-VSB signal, an example is shown in Figure 4-10. Poor linearity results in the creation of intermodulation products.

Figure 4-10 Effect of Linearity on the Eye Pattern

+1

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4.6.2 ICPM (Incidental Carrier Phase Modulation)

ICPM is a phase shift of the signal as a result of its change in amplitude. Another name for ICPM is differential group delay. It will cause a larger amplitude waveform to be displaced further in time, but as the amplitude of the waveform decreases, it will not be displaced as far, with a small amplitude waveform receiving minimum displacement. See Figure 4-11.

Figure 4-11 Effects of ICPM

Correct timing is shown by the reference waveform.

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ICPM will cause the top or bottom rows of eyes to close from the right or left, but have less effect on the center rows, see Figure 4-12.

In the example shown in Figure 4-12, four similarly shaped traces are drawn. Each successive trace has a larger amplitude and is displaced slightly to the right. This displacement is due to ICPM. The black dots on the larger traces indicate the correct level and location of each symbol.

Figure 4-12 Effect of ICPM on Eye Pattern

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Achieving FCC Mask Specifications11/19/03

5 Achieving FCC Mask Specifications

At the beginning of the DTV rollout, the FCC DTV emissions mask could be met or exceeded through use of a well designed linear RF power amplifier, and the values of the mask could be read on a spectrum analyzer. Then the FCC changed the mask specifications, so that every DTV transmitter required a high power mask filter to meet the new specification, leaving the broadcast engineer in a state of confusion over how to measure his transmitter and prove that it did meet or exceed the new specifications. The purpose of this paper is to attempt answer those questions by covering the following areas.

• Explain terms such as peak to average power specifications of an RF power amplifier, amplifi-er linearity (or non linearity), and how they effect the adjacent channel inter modulation (inter-mod) products.

• Linear distortions (amplitude response and group delay) caused by the mask filter.

• How to measure the transmitter amplitude response and document the results.

5.1 The New (Current) FCC Mask

The following is the text of Part 73.622 (b). It concerns the out of band emissions performance required for an ATSC 8-VSB DTV transmitter.

(h)(1) The power level of emissions on frequencies outside the authorized channel of operation must be attenuated no less than the following amounts below the average transmitted power within the authorized channel. In the first 500 kHz from the channel edge the emissions must be attenuated no less than 47 dB. More than 6 MHz from the channel edge, emissions must be attenuated no less than 110 dB. At any frequency between 0.5 and 6 MHz from the channel edge, emissions must be attenuated no less than the value determined by the following formu-la:

Attenuation in dB = -11.5(∆f + 3.6);

Where: ∆f = frequency difference in MHz from the edge of the channel.

(2) This attenuation is based on a measurement bandwidth of 500 kHz. Other measurement bandwidths may be used as long as appropriate correction factors are applied. Measurements need not be made any closer to the band edge than one half of the resolution bandwidth of the measuring instrument. Emissions include sidebands, spurious emissions and radio frequency harmonics. Attenuation is to be measured at the output terminals of the transmitter (including any filters that may be employed). In the event of interference caused to any service, greater at-tenuation may be required.

Several questions or observations arise from this excerpt of the rules.

1 The statement “more than 6 MHz from the channel edge, emissions must be attenuated no less than 110 dB” is interesting. This statement raises two questions.

A What is the frequency span indicated by this statement? It indicates that the frequency range is dc to light outside the limits of the adjacent channel.

B How many instruments can measure a signal 110 dB below a reference power? Very few. Most instruments can measure signals that are approximately 85 below the refer-ence due to the noise floor of the instrument.

2 The statement “this attenuation is based on a measurement bandwidth of 500 kHz” is ok if the correct test equipment is available. This statement says that we are to measure the average power in a 500 kHz band which centers about the frequency of interest and compare it to the

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average power within the transmitting channel. At this writing, instruments which can perform this type of measurement include the Agilent/HP 89441V vector signal analyzer, the Rohde & Schwarz FSP spectrum analyzer, and the Agilent ESA series spectrum analyzer (E4401B/2B/3B/4B/5B/7B).

A more conventional spectrum analyzer can be used to measure the transmitters response, using an appropriate resolution bandwidth such as 30 kHz, if the proper correction factor is applied. This will be discussed in greater detail later.

Figure 5-1 shows the old and new FCC mask.

Figure 5-1 FCC DTV Mask, New vs Old, With Reference Power at 0 dB

0

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-50

-60

-70

-80

-90

-100

-110

-120

0 2 4 6 8 10-2-4-6-8-10

Your Channel UpperAdjacentChannel

LowerAdjacentChannel

The shoulders are measured 250 kHz intothe adjacent channels,or -3.25 and +3.25 MHz from the centerof the channel.

Frequency MHz

Relative Level, dB

New MaskOld Mask

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5.2 Effects of Mask Filter on Adjacent Channel Intermods

Figure 5-2 shows the response of a transmitter before and after the standard D-Mask filter. The pre filter response shows considerable adjacent channel intermod growth. The filter removes most of these intermods, but notice, the response within the first 0.5 MHz of the adjacent channel is the same with the filter out or in. The mask corrects the adjacent channel response beyond the first 0.5 MHz, but the PA must be linearized to correct the first 0.5 MHz of the adjacent channels. Depending on the choice of DTV exciters, this correction may performed manually or automatically with adaptive precorrection.

The first 0.5 MHz of the adjacent channels are commonly referred to as the shoulders. In this figure the shoulders are -47 dB down, and therefore meet the FCC mask specifications, but on the spectrum analyzer they can only be measured with respect to the center of the in band response. Using this reference, they only measure -37 dB. An offset must be used to prove the response is within the FCC mask specifications. The required offset will be discussed later in this paper.

Note: Refer to Figures 5-1 and 5-2. The shoulder level is measured at a point 250 kHz into each adjacent channel, with respect to the amplitude of the response at the center of the channel. The spectrum analyzer is set up with a 30 kHz RBW (resolution bandwidth), a 10 or 30 kHz video bandwidth, and a vertical sensitivity of 10 dB/cm. The RBW issue will be covered later in this report.

Figure 5-2 shows the adjacent channel intermodulation products attenuated down to -85 dB, not the required -110 dB. This is due to the noise floor limitations of most spectrum analyzers and network analyzers. One method of proving that the response at the output of the mask filter meets the -110 dB specification will be given later in this report.

Figure 5-2 Effect of Mask Filter on Transmitter Response

Before Filter

After Filter

Center 503 MHz Span 30 MHz3 MHz/Div

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-30

-40

-50

-60

-70

-80

-90

-100

Ref Lvl-20 dBm

RBW 30 kHzVBW 30 kHz 10 dB/Div

Shoulders of both responses are at thesame level due to the D-Mask filtercharacteristics

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5.3 Peak to Average Power Rating of DTV Signals and Amplifiers

Both the DTV signal and the linear PA that amplifies the signal have peak to average power ratios. This can be confusing, because both terms are frequently used and often only the context of the surrounding text is the only clue as to which term is being used. The following is an attempt to explain the two terms.

5.3.1 DTV Signal Peak to Average Power Rating

For the DTV signal, this rating compares the average power level of the signal to the instantaneous RMS power level of the signal peaks. The DTV signal peaks are random in nature both in the frequency of their appearance and their power level. Due to this random nature, the DTV signal does not have a consistent peak power level, but the average level of the signal is very stable. Therefore, DTV transmitter output power is measured by its average RF output level. The peak power level occasionally ranges as high as +10 dB above its average level, but 99.9% of the time the peak power level is less than or equal to +6db (4 x) greater than the average level.

5.3.2 Amplifier Peak to Average Power Rating

Linear RF amplifiers have two power ratings, average power and peak power.

The average power rating of an amplifier depends in large measure on its heat build up and cooling. For solid state amplifiers, it depends on the size and shape of the heat sink, the velocity of the cooling air, the volume and density of cooling air that moves passed the heat sink, and the temperature of the cooling air. Other parameters, such as altitude and packaging density will also have an effect on the average power rating. Liquid cooling is frequently used to increase the power rating of high power vacuum tube type amplifying devices.

Note: The heat generated in an RF power amplifier is inversely proportional to its efficiency (whichin part depends on its idle current and power output level), and, for class AB, is directly proportionalto its RF output power level.

The peak power rating of an amplifier relates to how well it can withstand the maximum instanta-neous voltage, current, and power levels associated with its operation. The peak power ratings fortransistors are assigned by their manufacturers and published on the specification sheets for thedevices. For most amplifiers, one of the chief peak power limitations is the output circuit dc supplyvoltage.

5.4 Amplifier Nonlinearity, Chief Cause of Intermod products

Amplifier peak power rating and nonlinearity are the chief causes of adjacent channel intermod products. Proper choice of active device for the amplifier system will eliminate much of the nonlinearity, but major areas of nonlinearity still exist, they are:

1 A curved response when the amplifier just starts to conduct.

This effect is greatly reduced by operating in class A (for low power amplifiers). For high power amplifiers, this nonlinearity is greatly reduced by setting the correct bias and idle current values (the amplifier is operated class AB).

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2 The other nonlinearity in the amplifier response curve occurs when the amplifier is driven into saturation by the signal peaks, this is shown in Figure 5-3. This causes sync compression in an-alog TV and compresses or clips the high amplitude peaks of the DTV waveform.

Amplifier peak power rating is the point where compression and clipping occur. Refer to Figure 5-3. In analog TV, sync compression is cured by applying sync expansion, but when the DTV signal peaks exceed the peak power rating of the amplifier, the resulting compression and clipping of the peaks generate the troublesome intermod products. This gives rise to the peak to average rating for the DTV signal, which is discussed in detail in the next section of this report.

One interesting note is that the peaks of the DTV signal consist mainly of ringing energy, while the actual symbols are at a lower level, see Figure 5-3. If the clipping level is above the symbol level, the resulting intermod products may interfere with other communications and cause the transmitter to fail the FCC Mask requirements, but the few viewers of the station will experience problems due to the clipping.

Figure 5-3 Transmitter Peak Power Limiting, NTSV vs DTV

5.5 Peak To Average Power Ratio vs Shoulder Intermod Levels

When the peak power level of the DTV signal exceeds the peak power rating of the amplifier, the signal peaks are compressed or clipped. This causes adjacent channel intermod products to be formed, and these products are most evident at the shoulders of the DTV spectrum.

RF PA TransferCurve

Compression

CompressionNTSC DTV

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The information in Figures 5-5 and 5-6 was generated by using a linear amplifier of given peak power rating. A DTV signal was applied to the amplifier input and the output terminated in a proper load. Initially, the amplifier was operated at a very low power level so that the peak power level of the amplifier was more than 12 dB above the average power level of the DTV signal. The average DEV power level at the output of the amplifier was increased until it started to approach the peak power rating of the amplifier. The resulting output response was captured for each power level.

Figure 5-5 is a graph of the peak to average power ratio for a typical linear DTV amplifier verses its intermod level at the shoulders of the DTV spectrum. This graph represents an uncorrected amplifier. The vertical axis of the chart is labeled peak to average power ratio, but in this case, it refers to the peak power rating of the amplifier to the average power level of the DTV signal it is amplifying. The horizontal axis represents the shoulder level of the DTV output spectrum compared to its amplitude at the center of the channel.

Refer to Figure 5-6. When the peak to average ratio is high (8dB or better), the intermod (shoulder) level exceeds the FCC mask specification, the low level of intermods being caused mainly by other nonlinearities of the amplifier. At 7 dB or less peak to average ratio, the shoulder level is approaching or is below the FCC mask specification, because of the increased compression or clipping of the signal peaks.

If the peak to average ratio is approximately 6 dB or greater, the PA can be linearized through precorrection, feed forward, minor bias tweaks, or adaptive correction to cause the shoulders to exceed the FCC mask specification. If the sharp tuned mask filter is included, the PA can be linearized to meet or exceed the FCC mask specifications when the peak to average ratio is less than 6 dB, but it becomes extremely difficult to impossible to linearize the PA when the peak to average ratio is less than 5.5 dB.

In Figure 5-4 the dashed line attached to the transfer curve represents the linearity precorrection limit for an amplifier. The transfer curve approaches horizontal mainly because of supply voltage limitation. This area will cause hard clipping and set a limit for peak output power and linearity precorrection.

Figure 5-4 Amplifier Precorrection Area and Hard Clipping Level

RF PA TransferCurve

DTV

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The dashed line on the transfer curverepresents the areasof precorrection.

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Figure 5-5 PA Peak to Average Ratio Compared to DTV Shoulder Intermod Level

Figure 5-6 DTV RF Response Shoulder Level for Various PA Peak to Average Power Ratios

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Adjacent Channel Shoulder Level dB

FCC

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6.7 dB

-30 dB (5.5 dB)

-37 dB (6.7 dB)FCC

-40 dB (7.3 dB)

-50 dB (12 dB)

Center 503 MHz Span 20 MHz2 MHz/Div

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RBW 30 kHzVBW 30 kHz10 dB/Div

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5.6 Resolution Bandwidth and Offsets for FCC Mask Measurements

Since most stations lack the equipment to measure the average power level within a specific bandwidth, DTV RF amplitude response levels and proof of FCC mask compliance exhibits will probably be performed on a spectrum analyzer. This causes some confusion because of the 500 kHz band statement in the FCC rules and the results of a spectrum analyzer measurement with a resolution band width (RBW) of 500 kHz.

The first problems is that very few spectrum analyzers have a 500 kHz RBW setting, and if they did, the resulting response could not be used to accurately evaluate the adjacent channel spectral response, see Figure 5-7. In this figure, it becomes obvious that a 30 kHz RBW (with a 10 or 30 kHz video bandwidth) is practical and provides an excellent view of the adjacent channel spectral response, but the question of how to evaluate the response in terms of the FCC mask specifications arises.

With a 30 kHz RBW, the only practical method of measuring the adjacent channel intermod levels is with respect to the spectral level at the center of the channel, but this requires an offset to convert it to equivalent FCC mask values. Offset calculations are relatively easy at a 500kHz RBW, but what about a 30 KHz RBW? Fortunately, the solution is easy. The 500 kHz offset is a number which is added to the measured values taken with respect to the response at the center of the channel, see the formula below.

Fortunately, the answer to the offset problem is relatively easy. Refer to Figure 5-7. When the RBW is changed from 500 kHz to 30 kHz the entire waveform drops by approximately 12.2 dB. The response level within the adjacent channels appears to drop a greater amount, but this is only due to the greater detail visible at the 30 kHz RBW. Therefore, the offset calculated for a 500 kHz RBW will work for a 30 kHz RBW as well.

Figure 5-7 Effect of Resolution Bandwidth on DTV RF Response

FCC value offset measured value+=

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5.6.1 Calculating the Offset

Several estimations of offset values can be suggested, but so far an exact value has not been agreed on. Five different offset values ranging from -10 to -11 dB will be shown here, each is referenced to a 500 kHz RBW and one or two other stated references.

1 This offset is based on a channel bandwidth of 6 MHz.

2 The next offset is created by rounding the above value to the nearest dB

offset = -11 dB

3 This offset is based on a signal occupied bandwidth of 5.38 MHz

4 The next offset is created by rounding the above value to the nearest dB

offset = -10 dB

5 This offset is created by including the pilot level (0.3 dB) in the -10.3 dB offset.

offset = -10.6 dB

To illustrate the impact of the offsets, the shoulder value will be created for the five offsets, see Table 5-1.

Analysis of the above table reveals that the -11 dB offset provides the easiest shoulder goal and a -10 dB offset provides the most difficult goal. Which one to use? The most conservative value of -10 dB gets my vote.

5.6.1.1 Shoulders Passed, What About the Rest of the Response

If the shoulders of the response pass the FCC mask test, the standard D-Mask filter will ensure that the remainder of the out of band response will also pass the test. The D-Mask filter response is too wide to improve the shoulder response, but the Sharp Tuned Mask filter will also improve the shoulder response. The D-Mask and the Sharp Tuned filters will be discussed later in this report.

Table 5-1 Calculating Required Shoulder Levels With Various Offsets

FCC Mask Spec. Offset Used Shoulder Measurement

-47 dB -10 dB -37 dB

-47 dB -10.3 dB -36.7 dB

-47 dB -10.6 dB -36.4 dB

-47 dB -10.8 dB -36.2 db

-47 dB -11 dB -36 dB

offset 10500kHz6MHz

-------------------- 10.8dB–=log=

offset 10500kHz

5.38MHz----------------------- 10.3dB–=log=

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5.7 Standard D Mask Filter Responses

The standard D-Mask filter was an earlier design. Since the response of this filter has a tendency to drift slightly with temperature, the flat bandwidth of the filter had to be wide enough so that temperature drift would not cause the filter to cut into either side of the signal response. Because of this, the D-Mask filter will not correct the adjacent channel response at the shoulders. Beyond the shoulders, the filter will attenuate the intermods. The skirts of the filter had to be steep enough, so that the transmitter output would exceed FCC specifications in spite of filter temperature drift. The inband amplitude response of the filter is flat within a few tenths of a dB.

5.7.1 Group Delay and Amplitude Response Correction

Due to the nature of a high power filter, the filter introduces group delay to the signal. This group delay, and the fact that the inband amplitude response of the filter is not perfectly flat, increases the EVM (error vector magnitude) of the DTV signal and closes the eyes of the DTV eye pattern. The amplitude and group delay response of the transmitter RF chain before the filter is flat enough so that it does not appreciably increase EVM. Group delay and amplitude response of the transmitter and mask filter is reduced by manual and/or automatic adaptive precorrection.

Figure 5-8 Standard D-Mask Filter Response and Group Delay Responses

Delay

Response

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5.8 Sharp Tuned Filter Responses

A later development of high power filter technology conquered much of the temperature drift of the filters. This allowed the mask filter to have a narrower amplitude response with quicker roll off. This still protects the response of the inband DTV signal (typical in band response variation range is +0.25 to -0.35 dB) but greatly attenuates the adjacent channel shoulder imtermod level. Therefore, when this filter is used, the output of most uncorrected transmitters (with a reasonable peak to average power ratio) would pass the FCC Mask test including the shoulders. This is the sharp tuned filter, sometimes referred to as the “cool fuel” filter. The response of the sharp tuned filter is compared to that or the D-Mask filter in Figure 5-9.

Due to the steep roll off of the sharp tuned filter, it can be used simultaneously to filter the adjacent channel intermods of its DTV transmitter and also used to combine this transmitter with an adjacent channel analog or digital transmitter.

Figure 5-9 Sharp Tuned Filter and Standard D-Mask Filter Responses

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5.8.1 Sharp Tuned Filter Group Delay

The in band group delay of the sharp tuned filter is shown in Figure 5-10. Typical group delay values range from 300 to 400 nanoseconds, Using present technology, the sharp tuned filter in band amplitude response and group delay can be easily corrected by manual or automatic adaptive precorrection.

Figure 5-10 Sharp Tuned Filter Group Delay Responses

5.9 EVM, Eye Patterns, Amplitude Response and Group Delay Interactions

This discussion is intended to reveal which distortions effect which parameters.

5.9.1 PA Nonlinearity vs EVM

PA nonlinear distortions are defined as amplitude level dependent changes in amplitude and phase. Typical nonlinear distortions are:

• Linearity (also referred to as AM to AM conversion)

• ICPM (incidental carrier phase modulation), also referred to as AM to PM conversion.

Nonlinear distortions cause the intermod level of the DTV to increase. This increase is most noticeable in the adjacent channels, particularly at the shoulders. If the PA output equals or exceeds the FCC mask specifications, the shoulder level before the D-Mask filter is at -37 dB with respect to the center of the channel, that level of nonlinearity will have minimum effect on EVM.

The sharp tuned filter will make the intermods of both adjacent channels look better. If the shoulder level before the sharp tuned filter is at -30 dB with respect to the center of the channel, that level of nonlinearity will have a small effect on EVM, it may increase EVM by one or two percent. If the pre-filter shoulder level greater than -30 dB, it will greatly effect EVM.

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5.9.2 Amplitude Response and Group Delay vs EVM

The main linear distortions are amplitude response and group delay. These distortions greatly effect EVM. For most DTV transmitters, the linear distortions before the mask filter are negligible and have minimum effect on EVM, but the linear distortions of the of the mask filter will greatly effect EVM. Since the sharp tuned filter has greater group delay distortions and somewhat greater amplitude distortions than the D-Mask filter, it will have a greater effect on EVM. The linear distortions generated by either filter must be corrected in order to bring the EVM down to 4% or less. This method of correction can be manual or automatic (adaptive equalization).

Figure 5-11 shows the EVM at the output of a D-mask filter. The left side of this figure has no amplitude or delay correction with EVM ranging between 10 to 11 percent. The right side of the figure has manual delay correction. Adaptive correction will typically result in EVM levels of approximately 3.2% or less. Manual correction can often result in better EVM than that shown in the figure if enough care is taken with the adjustments. If both manual and automatic correction is available, a good plan is to adjust the manual correctors for an EVM of approximately 4% (or better) and let the automatic correction do the rest. This way, it the automatic correction is disabled, the transmitter EVM will still be within reason.

One additional note concerning manual amplitude and delay correction. EVM numbers will be lower if the amplitude and/or delay distortions are symmetrical about the DTV channel, and become significantly greater if either or both distortions are non symmetrical. For example, a 1 dB amplitude slope across the channel will increase EVM by approximately 6%.

Figure 5-11 Effect of Equalization on DTV Mask Filter EVM Performance

EVM = 10.583 %rms23.052 % pk. at sym 62

Mag Err = 7.2416 %rms21.706 % pk. at sym 406

Phase Err = 11.442 deg-102.14 deg pk. at sym 319

Freq Err = 147.08 Hz

Pilot Lvl = -0.398 dB

SNR (MER) = 18.486 dB

EVM = 3.5644 %rms8.4655 % pk. at sym 363

Mag Err = 2.3335 %rms8.3027 % pk. at sym 363

Phase Err = 2.6983 deg-15.013 deg pk. at sym 467

Freq Err = -95.482 Hz

Pilot Lvl = -0.12 dB

SNR (MER) = 28.918 dB

Without response and group delay correction with response and group delay correction

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5.10 Creating The FCC Mask Performance Exhibit

Now that the basic theory has been discussed, there still remains the problem of creating an exhibit which proves that the overall transmitter response meets or exceeds the FCC mask specifications in spite of the difficulties encountered in making the measurements with a spectrum analyzer. Fortunately, the nature of the dB and ready availability of spread sheet software greatly simplify the task.

The spectrum analyzer response of the overall transmitter cannot be captured since the noise floor of most spectrum analyzers limit measurement to -85 dB. This difficulty can be overcome by measuring the response of the mask filter and response of the transmitter before the mask filter separately and adding the dB values for the two responses. The results can be verified by comparing them to the FCC mask specifications.

The filter response can be obtained by sweeping the filter with a spectrum analyzer and tracking generator or a network analyzer, or they can be obtained from the filter manufacturer’s test data.

Figure 5-12 is a capture of an Excel spread sheet. The center frequency of the channel is entered in the box at the top of the sheet, and all of the necessary measurement frequencies appear in column A. The filter response and reference values are entered in columns B and C, and the transmitter response and reference values are entered in columns E and F. The net system response appears in column H, and the offset, which was entered in the box at the top of the sheet, is added to the net response and the result appears in column I. The FCC mask specifications have been entered in column J, and the results of the test appear in column K (positive values are passing, negative values are failing).

In this sheet, values have been entered so the results can be viewed. The same pre filter values have been entered for exciters A and B, but the values for a D-Mask filter have been entered for exciter A while the values for a sharp tuned filter (taken from Figure 5-9) have been entered for exciter B.

The results of columns I and J for each exciter has been formed into graphs on the next two sheets of the program. The graph for exciter A (the D-Mask filter) is shown in Figure 5-13 and the graph for exciter B (the sharp tuned filter) is shown in Figure 5-14. Examination of the two graphs show the additional adjacent channel shoulder attenuation provided by the sharp tuned filter.

Figure 5-15 shows the formulas used to create the spread sheet, and a list of channel center frequencies and pilot frequencies are provided in Table 5-2.

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Figure 5-12 Spread Sheet Exhibit, Proving Transmitter FCC Mask Compliance

Note: Both Exciters A and B used the same PA, but for sake of illustration, Exciter A values are for standard D mask filter and Exciter B values are taken from the sharp

tuned filter shown in Figure 5-9.

To use this chart type in center channel frequency and offset below. Then insert data taken into chart, it will fill in automatically.

Channel center freq. in MHz= 719

Enter offset in dB = 10.6

Transmitter FCC Mask Response, Exciter A

Filter Response Transmitter Response before filter Net Resp. FCC Negative #Analyzer Center Freq. Filter Analyzer Center Freq. Transmitter Net Minus Mask Is out of FCC

Frequency Reading Reference Response Reading Reference Response Response Offset Response Specifications710.00 -98.49 -0.80 -97.69 -72.83 -21.58 -51.25 -148.94 -159.54 -110.00 49.54711.00 -64.34 -0.80 -63.54 -71.18 -21.58 -49.60 -113.14 -123.74 -98.90 24.84712.00 -51.26 -0.81 -50.45 -68.66 -21.58 -47.08 -97.53 -108.13 -87.40 20.73713.00 -39.04 -0.81 -38.22 -63.16 -21.64 -41.52 -79.74 -90.34 -75.90 14.44714.00 -25.46 -0.81 -24.65 -62.95 -21.64 -41.31 -65.96 -76.56 -64.40 12.16714.50 -17.50 -0.81 -16.69 -62.51 -21.64 -40.87 -57.56 -68.16 -58.70 9.46715.00 -8.64 -0.81 -7.82 -60.50 -21.56 -38.94 -46.76 -57.36 -52.90 4.46715.50 -2.06 -0.81 -1.25 -62.05 -21.56 -40.49 -41.74 -52.34 -47.00 5.33715.75 -1.20 -0.81 -0.39 -61.37 -21.56 -39.81 -40.20 -50.80 -47.00 3.80

722.25 -1.10 -0.83 -0.27 -61.46 -22.01 -39.45 -39.72 -50.32 -47.00 3.32722.50 -1.70 -0.83 -0.87 -62.03 -22.01 -40.02 -40.89 -51.49 -47.00 4.49723.00 -6.73 -0.83 -5.91 -61.97 -22.01 -39.96 -45.87 -56.47 -52.90 3.57723.50 -14.64 -0.83 -13.82 -63.05 -22.24 -40.81 -54.63 -65.23 -58.70 6.53724.00 -22.10 -0.83 -21.27 -63.56 -22.24 -41.32 -62.59 -73.19 -64.40 8.79725.00 -34.83 -0.83 -34.00 -67.34 -22.24 -45.10 -79.10 -89.70 -75.90 13.80726.00 -45.96 -0.83 -45.13 -69.78 -22.03 -47.75 -92.88 -103.48 -87.40 16.08727.00 -57.04 -0.80 -56.24 -71.97 -22.03 -49.94 -106.18 -116.78 -98.90 17.88728.00 -71.26 -0.80 -70.46 -76.19 -22.03 -54.16 -124.62 -135.22 -110.00 25.22

Transmitter FCC Mask Response, Exciter B

Filter Response Transmitter Response before filter FCC Negative #Analyzer Center Freq. Filter Analyzer Center Freq. Transmitter Net Mask Is out of FCC

Frequency Reading Reference Response Reading Reference Response Response Response Specifications710.00 -69.00 0.00 -69.00 -72.83 -21.58 -51.25 -120.25 -130.85 -110.00 20.85711.00 -65.00 0.00 -65.00 -71.18 -21.58 -49.60 -114.60 -125.20 -98.90 26.30712.00 -63.00 0.00 -63.00 -68.66 -21.58 -47.08 -110.08 -120.68 -87.40 33.28713.00 -68.00 0.00 -68.00 -63.16 -21.64 -41.52 -109.52 -120.12 -75.90 44.22714.00 -50.00 0.00 -50.00 -62.95 -21.64 -41.31 -91.31 -101.91 -64.40 37.51714.50 -40.00 0.00 -40.00 -62.51 -21.64 -40.87 -80.87 -91.47 -58.70 32.77715.00 -30.00 0.00 -30.00 -60.50 -21.56 -38.94 -68.94 -79.54 -52.90 26.64715.50 -20.00 0.00 -20.00 -62.05 -21.56 -40.49 -60.49 -71.09 -47.00 24.09715.75 -34.00 0.00 -34.00 -61.37 -21.56 -39.81 -73.81 -84.41 -47.00 37.41

722.25 -35.000 0.00 -35.00 -61.46 -22.01 -39.45 -74.45 -85.05 -47.00 38.05722.50 -29.000 0.00 -29.00 -62.03 -22.01 -40.02 -69.02 -79.62 -47.00 32.62723.00 -30.000 0.00 -30.00 -61.97 -22.01 -39.96 -69.96 -80.56 -52.90 27.66723.50 -38.000 0.00 -38.00 -63.05 -22.24 -40.81 -78.81 -89.41 -58.70 30.71724.00 -44.000 0.00 -44.00 -63.56 -22.24 -41.32 -85.32 -95.92 -64.40 31.52725.00 -57.000 0.00 -57.00 -67.34 -22.24 -45.10 -102.10 -112.70 -75.90 36.80726.00 -75.000 0.00 -75.00 -69.78 -22.03 -47.75 -122.75 -133.35 -87.40 45.95727.00 -73.000 0.00 -73.00 -71.97 -22.03 -49.94 -122.94 -133.54 -98.90 34.64

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Figure 5-13 Graph of Results of Exciter A Spread Sheet (With Standard D-Mask Filter)

Figure 5-14 Graph of Results of Exciter B Spread Sheet (With Sharp Tuned Filter)


-180.00

-160.00

-140.00

-120.00

-100.00

-80.00

-60.00

-40.00

-20.00

0.00

708.00 710.00 712.00 714.00 716.00 718.00 720.00 722.00 724.00 726.00 728.00 730.00

Frequency MHz

Att

enu

atio

n d

B

FCC Mask

Xmtr Resp.


-180.00

-160.00

-140.00

-120.00

-100.00

-80.00

-60.00

-40.00

-20.00

0.00

708.00 710.00 712.00 714.00 716.00 718.00 720.00 722.00 724.00 726.00 728.00 730.00

Frequency MHz

Att

enu

atio

n d

B

FCC Mask

Xmtr Resp.

Page: 42


Figure 5-15 Formulas For FCC Mask Compliance Spread Sheet

719

10.6

Net Resp. FCC Negative #Analyzer Center Freq. Filter Analyzer Center Freq. Transmitter Net Minus Mask Is out of FCC

Frequency Reading Reference Response Reading Reference Response Response Offset Response Specifications=D5-9 =B14-C14 =E14-F14 =D14+G14 =H14-D7 -110 =J14-I14=D5-8 =B15-C15 =E15-F15 =D15+G15 =H15-D7 -98.9 =J15-I15=D5-7 =B16-C16 =E16-F16 =D16+G16 =H16-D7 -87.4 =J16-I16=D5-6 =B17-C17 =E17-F17 =D17+G17 =H17-D7 -75.9 =J17-I17=D5-5 =B18-C18 =E18-F18 =D18+G18 =H18-D7 -64.4 =J18-I18=D5-4.5 =B19-C19 =E19-F19 =D19+G19 =H19-D7 -58.7 =J19-I19=D5-4 =B20-C20 =E20-F20 =D20+G20 =H20-D7 -52.9 =J20-I20=D5-3.5 =B21-C21 =E21-F21 =D21+G21 =H21-D7 -47 =J21-I21=D5-3.25 =B22-C22 =E22-F22 =D22+G22 =H22-D7 -47 =J22-I22

=D5+3.25 =B24-C24 =E24-F24 =D24+G24 =H24-D7 -47 =J24-I24=D5+3.5 =B25-C25 =E25-F25 =D25+G25 =H25-D7 -47 =J25-I25=D5+4 =B26-C26 =E26-F26 =D26+G26 =H26-D7 -52.9 =J26-I26=D5+4.5 =B27-C27 =E27-F27 =D27+G27 =H27-D7 -58.7 =J27-I27=D5+5 =B28-C28 =E28-F28 =D28+G28 =H28-D7 -64.4 =J28-I28=D5+6 =B29-C29 =E29-F29 =D29+G29 =H29-D7 -75.9 =J29-I29=D5+7 =B30-C30 =E30-F30 =D30+G30 =H30-D7 -87.4 =J30-I30=D5+8 =B31-C31 =E31-F31 =D31+G31 =H31-D7 -98.9 =J31-I31=D5+9 =B32-C32 =E32-F32 =D32+G32 =H32-D7 -110 =J32-I32

Net Resp. FCC Negative #Analyzer Center Freq. Filter Analyzer Center Freq. Transmitter Net Minus Mask Is out of FCC

Frequency Reading Reference Response Reading Reference Response Response Offset Response Specifications=D5-9 =B40-C40 =E40-F40 =D40+G40 =H40-D7 -110 =J40-I40=D5-8 =B41-C41 =E41-F41 =D41+G41 =H41-D7 -98.9 =J41-I41=D5-7 =B42-C42 =E42-F42 =D42+G42 =H42-D7 -87.4 =J42-I42=D5-6 =B43-C43 =E43-F43 =D43+G43 =H43-D7 -75.9 =J43-I43=D5-5 =B44-C44 =E44-F44 =D44+G44 =H44-D7 -64.4 =J44-I44=D5-4.5 =B45-C45 =E45-F45 =D45+G45 =H45-D7 -58.7 =J45-I45=D5-4 =B46-C46 =E46-F46 =D46+G46 =H46-D7 -52.9 =J46-I46=D5-3.5 =B47-C47 =E47-F47 =D47+G47 =H47-D7 -47 =J47-I47=D5-3.25 =B48-C48 =E48-F48 =D48+G48 =H48-D7 -47 =J48-I48

=D5+3.25 =B50-C50 =E50-F50 =D50+G50 =H50-D7 -47 =J50-I50=D5+3.5 =B51-C51 =E51-F51 =D51+G51 =H51-D7 -47 =J51-I51=D5+4 =B52-C52 =E52-F52 =D52+G52 =H52-D7 -52.9 =J52-I52=D5+4.5 =B53-C53 =E53-F53 =D53+G53 =H53-D7 -58.7 =J53-I53=D5+5 =B54-C54 =E54-F54 =D54+G54 =H54-D7 -64.4 =J54-I54=D5+6 =B55-C55 =E55-F55 =D55+G55 =H55-D7 -75.9 =J55-I55=D5+7 =B56-C56 =E56-F56 =D56+G56 =H56-D7 -87.4 =J56-I56=D5+8 =B57-C57 =E57-F57 =D57+G57 =H57-D7 -98.9 =J57-I57=D5+9 =B58-C58 =E58-F58 =D58+G58 =H58-D7 -110 =J58-I58

Channel center freq. in MHz=

Enter offset in dB =

Filter Response Transmitter Response before filter



Transmitter Response before filterFilter Response

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Table 5-2 Channel Center Frequency and Pilot Frequency (With No Offset)
Channel

Center Frequency (MHz)

Pilot and RFA 300 Frequency Channel


Pilot and RFA 300 Frequency Channel


Pilot and RFA 300 Frequency

2 57 54.31 25 539 536.31 48 677 674.31

3 63 60.31 26 545 542.31 49 683 680.31

4 69 66.31 27 551 548.31 50 689 686.31

5 79 76.31 28 557 554.31 51 695 692.31

6 85 82.31 29 563 560.31 52 701 698.31

7 177 174.31 30 569 566.31 53 707 704.31

8 183 180.31 31 575 572.31 54 713 710.31

9 189 186.31 32 581 578.31 55 719 716.31

10 195 192.31 33 587 584.31 56 725 722.31

11 201 198.31 34 593 590.31 57 731 728.31

12 207 204.31 35 599 596.31 58 737 734.31

13 213 210.31 36 605 602.31 59 743 740.31

14 473 470.31 37 611 608.31 60 749 746.31

15 479 476.31 38 617 614.31 61 755 752.31

16 485 482.31 39 623 620.31 62 761 758.31

17 491 488.31 40 629 626.31 63 767 764.31

18 497 494.31 41 635 632.31 64 773 770.31

19 503 500.31 42 641 638.31 65 779 776.31

20 509 506.31 43 647 644.31 66 785 782.31

21 515 512.31 44 653 650.31 67 791 788.31

22 521 518.31 45 659 656.31 68 797 794.31

23 527 524.31 46 665 662.31 69 803 800.31

24 533 530.31 47 671 668.31

Page: 44

Dual Sideband Suppressed Carrier and Quadrature Modulation11/19/03

6 Dual Sideband Suppressed Carrier and Quadrature Modulation

6.1 Introduction

The need to understand the subjects of amplitude modulation, single sideband, dual sideband, reduced carrier operation, and quadrature modulation (QAM) has increased now that digital television is here. This paper contains a review of these subjects starting with amplitude modulation.

6.2 Amplitude Modulation

A P I N diode modulator that will produce an amplitude modulated carrier is shown in Figure 6-1. The diode CR1, differs an impedance (resistive) to RF which is inversely proportional to the D.C. (or audio rate) forward bias current flowing through it.

A high level of forward bias current produces a low RF impedance, and a low forward bias current produces a high RF impedance. The RF signal is not able to make the P I N diode conduct. With no D.C. bias current, the RF impedance of the P I N diode is extremely high.

In Figure 6-1, the RF path is from the generator through C1, R2, C2, and the RF load. Some RF current will also flow through R1, R2, and R3, but most of it will flow through the RF load because its load impedance is much lower than the other resistors in the circuit. The RF impedance of the diode and the load form a voltage divider. The RF developed across the load varies inversely with the RF impedance of the diode.

The D.C. bias current path is the bias source, R1 CR1 and R2; and the audio current (the modulating signal) path is the audio source C3, R3, CR1, and R2, with some audio current lost through R1 and the bias source.

The D.C. bias and audio current (the modulating signal) combine at the junction of R1 and R3 to form a positive D.C. current source that varies at an audio rate from a few micro amps positive to a few milli amps positive.

The P I N diode is always forward biased and this varying bias changes the RF resistance of the diode. The amplitude of the RF carrier in Figure 6-1 varies with the audio (modulating signal).

The amplitude modulated signal voltage changes in magnitude during the audio cycle but the resultant signal is always in phase with the RF carrier.

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Figure 6-1 An amplitude modulation scheme using a P I N diode.

6.2.1 Amplitude Modulation Sideband Frequencies

The A.M. signal contains a carrier, a upper sideband, and a lower sideband. The sidebands carry the intelligence of the modulation (audio). They tell the amplitude, frequency, waveshapes, and the phase of the modulating signal. The intelligence carried by one sideband is identical to that carried by the other.

The sideband frequencies are:

Fusb = Fcarrier + Fmodulation

Flsb = Fcarrier - Fmodulation

Where: Fusb = upper sideband frequencyFlsb = lower sideband frequency.

+

0

-

+

0

-

+

0

-

RFCarrierSource

RF

InputAudio

BiasDC

R3 R1 R2 RFLoad

C1 CR1 C2

C1

RFCarrier

AudioPlusBias

AmplitudeModulatedRF

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For sinewave modulation, the upper and lower sidebands are CW signals with frequencies above and below the carrier. They combine into a single signal which is on the carrier frequency and with an envelope that describes the modulating signal. The combining action of the upper and lower sidebands can be observed by studying the waveforms in Figure 6-3, or by studying the vectors which represent the sidebands.

A vector makes one complete revolution in the counterclockwise direction for each signal cycle. Therefore, a vector rotates faster as the signal frequency increases. Since the upper and lower sidebands are separated from the carrier by the modulating frequency, all three vectors rotate at different speeds. The lower sideband vector rotates slower then the carrier vector and the upper sideband vector rotates faster then the carrier vector.

To make this picture easier to visualize, imagine that a series of stop-action pictures of all three vectors is taken at succeeding times during one modulating cycle, see Figure 6-2. Each time the picture is taken, the carrier vector is at the same angle, and therefore appears motionless. In this series of pictures, the upper and lower sideband vectors appear to be rotating in opposite directions. This rotation is due to the difference in frequencies between the carrier and each sideband. The lower sideband vector appears to be rotating clockwise and the upper sideband vector appears to be rotating counterclockwise.

Refer to Figure 6-3. At any instant, the position of the upper and lower sideband vectors is such that their addition produces a resultant vector that is either in phase or out of phase with the carrier or has zero amplitude.

Figure 6-2 Carrier, Upper Sideband, and Lower Sideband Vectors

Each of the eight drawings shows the carrier vector in the same relative position.The upper and lower sideband vectors appear to rotate in opposite directions, with the upper sideband rotating counterclockwise and the lower sideband rotating clockwise.

USB

CarrierVector

LSBVector Vector

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Figure 6-3 Upper and Lower Sidebands Combine to Form Dual Sideband Signal

The counterclockwise rotating vectors represents the upper sidebandThe clockwise rotating vectors represents the lower sideband

For sinewave modulation, the upper and lower sidebands are CW signals withfrequencies above and below the carrier. They combine into a single signal that is

on the carrier frequency and with an envelope that describes the modulating signal.

ModulatingSignal

Upper

Carrier

ResultantVectors

+

0

-

+

0

-

+

0-Signal

Sideband

1 2 3 4 5 6

7

8

USB and

Lower+

0-Signal

Sideband

VectorsLSB

+

0

-

Dual

SignalSideband

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6.2.2 Power Level of Sidebands and Carrier

The carrier, the upper sideband (usb) and the lower sideband (lsb) are shown in Figure 6-4. The signal voltages represent 100% modulation with a sinewave. At 100% modulation the carrier contains two-thirds of the total power and the power in each sideband is one-sixth of the total power. If the level of modulation drops, the power of the sidebands drop but the carrier power remains constant.

The carrier contains most of the RF power but carries none of the modulation intelligence. If the carrier could be eliminated and reinserted later, the resultant dual sideband suppressed carrier signal would require much less power to do the same work. The required sideband power would be proportional to the power contained by the modulating signal.

Figure 6-4 The sidebands, as viewed on a spectrum analyzer.

999 10011000

Amplitude

Frequency (kHz)

Carrier

UpperSideband

LowerSideband

(voltage)

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6.3 Generation Of Dual Sideband Suppressed Carrier Signals

In the conventional amplitude modulation system the carrier, the sidebands, and the modulating frequency are present in the output signal. The modulating frequency, being much lower then the rest, is easily removed by the RF circuit.

The balanced modulator balances out the carrier leaving only the side-bands and the modulating signal in the output. The modulating signal is easily removed by the output RF tuned circuit.

In a doubly balanced modulator, both the modulating signal and the carrier are balanced out leaving only the sidebands in the output.

In this paper, a diode ring modulator is used to generate the dual sideband suppressed carrier (DSSC) signal. It is a doubly balanced modulator and although many types of doubly balanced modulators exist, this one was chosen for this explanation because of its simplicity of operation.

6.4 Operation Of A Diode Ring Modulator

A schematic of a diode ring modulator is shown in Figure 6-5. There are two ways to explain the operation of a diode ring modulator. They are the PIN Diode Mode of Operation of the Diode Ring Modulator, on page 50, and the Switch Mode of Operation of the Diode Ring Modulator, on page 67.

6.4.1 PIN Diode Mode of Operation of the Diode Ring Modulator

Each diode is a P I N diode. The carrier is applied to the left port, the modulating signal is applied to the X port, and the output signal (DSSC) is taken from the R (right) port.

Figure 6-5 A diode ring modulator functioning as a doubly balanced modulator.

ModulationSource

CarrierGenerator

LeftPort

X Port

RightPort

RF Output

RLT2

CR1

CR2

CR4

CR3T1

Page: 50


To understand the operation of the ring modulator, four equivalent circuits must be observed. These four equivalent circuits are described in the following four paragraphs and shown in Figure 6-6 and Figure 6-7.

When the modulating signal is positive, current flows from ground into the secondary of T1 at the center tap, see upper half of Figure 6-6. The current flows both directions in the secondary of T1 and thus no modu1ation signal feeds back into the carrier generator. The current flows out the upper end of the T1 winding to CR1 and CR2 but it can only flow through CR2 as it is forward biased and CR1 is reverse biased. The current flowing out of the bottom side of T1’s secondary arrives at CR3 and CR4. It flows through the forward biased CR3 but not through the reverse biased CR4. The current now enters the top and bottom of T2’s primary and flows to the center tap and back to the modulation source. Since the modulation current flows in opposite directions in T2’s primary, no modulating signal appears in T2’s secondary (the output).

In lower half of Figure 6-6, the RF current path is shown. Since CR2 and CR3 are forward biased by the modulating signal, they connect the top of T1’s secondary to the top of T2’s primary and the bottom of T1 to the bottom of T2. The RF output is in phase with the carrier. The amplitude of the RF output depends on the amount of modulation signal current since this current determines the RF impedance of the diodes, see waveforms in Figure 6-8.

When the modulating signal is negative, current flows from the modulating source into T2 primary center tap and both directions of T2 primary, see upper half of Figure 6-7. From the top of T2 it flows to the junctions of CR2 and CR4 and from the bottom of T2 to the junctions of CR1 and CR3. CR1 and CR4 are now forward biased so the current flows through them and into the top and bottom of the secondary of T1 and then to ground. Since modulating signal current flows in opposite direction in T1 and T2 no modulating signal gets to the output or the RF carrier source.

In the lower half of Figure 6-7, the RF current path is shown for the negative modulating signal. Since CR1 and CR4 are conducting, the RF path sees the secondary of T2 and the primary of T1 cross connected (the top of T1 to the bottom of T2 and etc). This causes the RF output to be 180° out of phase with the RF carrier. The amplitude of the RF output depends upon the amount of negative modulating signal current that flows through CR1 and CR4.

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Figure 6-6 Modulation Signal Current Path (A) and RF Current Path (B) for Positive Modulation

ModulationSource

X Port

CR2

CR3

CarrierGenerator

LeftPort

RightPort

RF Output

RLT2

CR2

CR3T1

+

-

PositiveModulation

(A) Modulating Signal Current Path

(B) RF Current Path

CR1

CR4

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Figure 6-7 Modulation Signal Current Path (A) and RF Current Path for Negative Modulation

ModulationSource

X Port

CR1

CR4

CarrierGenerator

LeftPort

RightPort

RF Output

RLT2

CR1

CR4

T1

-

+

(A) Modulating Signal Current Path

(B) RF Current Path

NegativeModulation

CR2

CR3

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6.5 Doubly Balanced Modulator Waveforms

The waveforms for the doubly balanced modulator are shown in Figure 6-8. When the modulating signal (waveform A of Figure 6-8) is positive, the RF output sidebands (waveform B) are in phase with the carrier (waveform C). When the modulating signal is negative, the RF output sidebands are 180° out of phase with the carrier. For either polarity of the modulating signal, the amplitude of the sideband output signal is proportional to the absolute amplitude of that portion modulating signal. When the modulating signal voltage is zero, no modulating signal current flows and all four diodes of Figure 6-5 are biased off. The RF output at this instant is zero, as shown in Figure 6-8.

Figure 6-8 Waveforms Associated With a Doubly Balanced Modulator (top three waveforms)

When the modulating signal is positive, the RF carrier and sidebands are in phase.When the modulating signal is negative, the RF carrier and sidebands 180° out of phase

If the carrier and sidebands are added, the original amplitude modulated waveformis produced (bottom waveform).

+

0

-

+

0

-

+

0

-

RFCarrier

Audio

AmplitudeModulatedRF

Waveform A.

Waveform C.

Waveform D.

+

0

-

Upper andLowerSidebands

Waveform B.

Page: 54


6.6 Demodulating The Dual Sideband Suppressed Carrier Signal

Refer to Figure 6-8. If the sideband signal is added to its RF carrier, the result will be an amplitude modulated signal. The in phase component of the sideband signal adds to the carrier and the out of phase component of the sideband signal subtracts from it. The resultant A.M. signal can be detected by any conventional A.M. detector.

The sideband signal and the RF carrier can be mixed directly in a nonlinear device. The output will contain the demodulated intelligence along with RF by-products, which can be easily be removed.

During the demodulation process, the carrier and sidebands are normally in phase or 180° out of phase. If the phase of either the carrier or the sideband signal is reversed (changed 180°) the polarity of the demodulated signal will also be reversed. If the carrier is reinserted at a 90° (or 270°) angle to the sideband signal, there will be little or no output.

6.7 Summary Of Dual Sideband Suppressed Carrier Modulation

The DSSC system does not waste power by creating a carrier. The carrier contains no intelligence and under the best conditions uses 66% of the total A.M. signal power.

The intelligence carrying sidebands are sent with just enough of a carrier sample to lock the demodulator’s carrier reinsertion oscillator to the frequency and phase of the carrier. However, for single or dual sideband communications to occur, it is not necessary to send a carrier sample, but the carrier must still be regenerated in the receiver.

The sidebands carry four important bits of information that describes the modulating signal. They are:

1. The modulating frequency is determined by the frequency difference between the sidebands and the carrier. Thus;

Fmod = Fusb - Fcarrier = Fcarrier - Flsb

Where: Fmod = modulating frequencyFusb = upper sideband frequencyFlsb = lower sideband frequency.

2. The polarity of the modulating signal is determined by the phase of the sidebands with respect to the carrier. If the sidebands and carrier are in phase the modulation polarity is positive, if they are out of phase the modulating signal has a negative polarity.

3. The modulating signal amplitude, measured from zero, is reflected in the amplitude of the side-band signal.

4. The modulating signal waveshape is indicated by the shape of the sideband signal envelope.

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6.8 Amplitude Modulation Produced From A Doubly Balanced Modulator

Figure 6-9 shows amplitude modulation produced from a doubly balanced modulator. The carrier and modulation are applied to the modulator, but do not appear in its output because they are both balanced out in the modulator. The modulator output consists of the upper and lower sidebands only.

The carrier is added to the sidebands in the summing circuit (Σ) and amplitude modulation results. This is shown in waveforms on Figure 6-8, on page 54.

Figure 6-9 Amplitude Modulator Using A Doubly Balanced Modulator

Figure 6-10 shows the waveforms and vector diagrams associated with amplitude modulation. Since both upper and lower sidebands are present, the resultant sideband signal appears to be at the carrier frequency. Its amplitude changes and is either in or 180 degrees out of phase with the carrier. The modulating signal is sampled at eight times (every 45 degrees) during one cycle. At the instant of each sample, the carrier is at 90 degrees, therefore, its vector appears stationary.

Since the upper and lower sideband frequencies are respectively higher and lower than the carrier frequency (by the modulating frequency), their vectors appear to rotate at each sample time. The upper sideband vector rotates counter clockwise and the lower sideband vector rotates clockwise.

At sample times 1 and 5, the sidebands are 180 degrees out of phase and cancel, the resultant vectors represent the carrier amplitude only.

At sample times 3 and 7, the sidebands are in phase. At sample 3, the sidebands and carrier are in phase, with the resultant vector having twice the amplitude of the carrier. At sample 7, the sidebands and carrier are out of phase and cancel, leaving a resultant amplitude of zero.

At times 2, 4, 6, and 8 the sidebands are 90 degrees out of phase and partially add to or cancel some of the carrier.

CarrierOscillator

Doubly

Σ Amplitude

Modulation

Input ModulatorBalanced

ModulationOutput

DSSC

Output

Note: This is also referredto as I (in phase) modulationbecause the carrier and sidebands are in phase or 180 degrees out of phase.

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Figure 6-10 Vector Addition In-Phase Carrier and Sidebands (Amplitude Modulation)

The sideband signal is in phase or 180° out of phase with the carrier.The resultant vector varies in amplitude but maintains a constant phase.

The result is amplitude modulation.

+

0

-

+

0

-

+

0

-

1 2 3 4 5 6 7 8

ModulatingSignal

SidebandSignal

ReinsertedCarrier

ResultantVectors

Carrier,USB, and LSBVectors

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6.9 Phase Modulation Produced From A Doubly Balanced Modulator

Figure 6-11shows phase modulator produced from a doubly balanced modulator. The modulator output consists of the upper and lower sidebands only.

The carrier is added with a +/-90 degree phase relationship to the sidebands in the summing circuit (Σ) and phase modulation results.

Figure 6-11 Phase Modulator Using Doubly Balanced Modulators

Figure 6-12 shows the waveforms and vector diagrams associated with phase modulation. The resultant sideband signal amplitude changes in accordance with the modulating signal, and its phase is +/-90 degrees with respect to the carrier. The modulating signal is sampled at eight times (every 45 degrees) during one cycle. At the instant of each sample, the carrier is at 90 degrees, therefore, its vector appears stationary. The upper and lower sideband frequency vectors appear to rotate at each sample time. The upper sideband vector rotates counter clockwise and the lower sideband vector rotates clockwise.

At sample times 1 and 5, the sidebands are 180 degrees out of phase and cancel, the resultant vectors represent the carrier amplitude only.

At sample times 3 and 7, the sidebands are in phase. At sample 3, the sidebands leads the carrier by 90 degrees, with the resultant vector leading by 45 degrees. At sample 7, the sidebands lag the carrier by 90 degrees, with the resultant vector lagging by 45 degrees.

At times 2, 4, 6, and 8 the sidebands are 90 degrees out of phase, with the resultant vector leading or lagging the carrier by 35 degrees.

The resultant vector varies in phase but maintains relatively constant amplitude. This results in phase modulation with a low modulation index (small amount of deviation). The small amplitude variation is easily removed in subsequent stages through limiting action.

CarrierOscillator

Doubly

90° PhaseShifter Σ Phase

Modulation


ModulationOutput

DSSC

Output

Page: 58


Figure 6-12 Vector Addition of Carrier and Sidebands Which Are 90° Out Of Phase (Phase Modulation)

The carrier is added with a +/-90 degree phase relationship to the sidebands and phase modulation results.

1 2 3 4 5 6 7 8

ModulatingSignal

SidebandSignal

ReinsertedCarrier

ResultantVectors

Carrier,USB, and LSBVectors

+

0

-

+

0

-

+

0

-

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6.10 Frequency Modulation Produced From A Phase Modulator

Refer to Figure 6-13. A low pass filter with a 6 dB per octave slope is added to the phase modulator that was just discussed. This causes the output to be frequency modulated. This conversion is possible because the carrier deviation of phase modulation depends on the modulating signal amplitude as well as its frequency. Since the deviation of a phase modulated signal increases at a rate of 6 dB per octave of the modulating signal frequency, the 6 dB per octave slope of the low pass filter cancels this deviation increase. As a result, the deviation is now dependent on modulating signal amplitude only, and this is the definition of FM.

Figure 6-13 Frequency Modulation From a Phase Modulator

6.10.1 FM and NTSC TV Aural Contain Both FM and PM

Both frequency and phase modulation appear in the output of FM (and NTSC TV aural) transmitters which use pre-emphasis. A 75 microsecond pre-emphasis curve passes modulating frequencies below 2.12 kHz through with a relatively flat response. At these frequencies, the carrier modulation is FM.

The 75 microsecond pre-emphasis curve provides a 6 dB per octave boost to modulating frequencies between 2.12 and 15 kHz. At these frequencies, the carrier modulation is PM.

CarrierOscillator

Doubly

90° PhaseShifter Σ Frequency

Modulation


ModulationOutput

6 dB per OctaveLow Pass Filter

Phase Modulator

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6.11 Single Sideband And Quadrature Distortion

In Figure 6-14, only the upper sideband is added to the carrier. The upper sideband signal has a constant amplitude and is at a higher frequency then the carrier. This results in a continual change of phase with respect to the carrier, which causes the resultant vector to change in both amplitude and phase. This effect is called quadrature distortion.

At samples 1 and 5, the sideband leads or lags the carrier by 90 degrees. The resultant vector leads or lags the carrier by 26.6 degrees, with an amplitude increase of 1.1 times. At times 3 and 7 the sideband signal is in or out of phase with the carrier. The resultant vector is in phase with the carrier and has an amplitude of 1.5 or 0.5 times the carrier.

Figure 6-14 Vector Addition of Carrier and Upper Sideband Only

1 2 3 4 5 6 7 8

ModulatingSignal

Upper

ReinsertedCarrier

ResultantVectors

Carrier andUSB Vectors

+

0

-

+

0

-+

0-Signal

Sideband

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6.12 QAM (Quadrature Amplitude Modulation)

It is possible to modulate, transmit, and separate during demodulation, two separate modulating signals on one carrier. Each modulating signal creates its own set of sidebands in its modulator. This process called quadrature amplitude modulation (QAM) and is accomplished by using the amplitude and phase modulators covered in section 6.8 on page 6-56 and section 6.9 on page 6-58. Demodulation and separation of the two modulation signals is covered in section 6.14 on page 6-66.

A quadrature modulator is shown in Figure 6-15. Two identical modulators, the I and Q balanced modulators, are shown in this diagram. The difference between these modulators is the phase of the carrier applied to them. The I (in phase) modulator receives the reference phase carrier and produces a dual sideband signal that is in phase or 180 degrees out of phase with the carrier. These sidebands fall on the +I or -I axis of the vector diagram on the right side of Figure 6-15.

The Q (quadrature) modulator receives a carrier that is shifted + or -90 degrees from that of the +I modulator. The Q sidebands are in phase or 180 degrees out of phase with the carrier phase applied to the Q modulator. These sidebands fall on the +Q and -Q axis of the vector diagram on the right side of Figure 6-15.

If a carrier sample is sent, it will also fall on the +I axis, although, it is not always necessary to send a carrier sample. It is also possible to send a carrier sample at a different phase angle, for example, in the NTSC chrominance signal, the burst (carrier sample) is sent at zero degrees, the +I axis is at 57 degrees and the +Q axis is 147 degrees.

Figure 6-15 QAM Modulator (Left) and Modulation Vector Diagrams (Right)

0°180°+I axis-I axis

+Q axis

270°

Individual Modulator Outputs (On I and Q axis)

-Q Axis

90°

CarrierOscillator

I BalancedModulator

90° PhaseShifter

Σ

Q BalancedModulator

QAMOutput

QAM Modulator

Attenuator

Modulation B

Input

Modulation A

Input

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6.12.1 16 QAM Modulator

Figure 6-16 shows a quadrature amplitude modulator used to modulate two digital data streams. No carrier sample is sent with this type of modulator, the carrier (amplitude and phase) is reconstructed in the receiver by analyzing the received signal. The modulation signal applied to each modulator consists of a series of symbols, with each symbol containing a specific number of bits.

In the case of the 16 QAM modulator, each symbol contains two bits, therefore, each symbol can have 4 relative voltage levels. These voltage levels are assigned from a lookup table, based on all possible values of the bits that comprises each symbol. The number of possible levels for a symbol are based on the number of bits per symbol, as expressed in the formula: , Where n = number of bits per symbol. The bit structure and level assignments for example are shown in Table 6-1.

The I modulator places these levels on the I axis and the Q modulator levels appear on the Q axis, as shown on the right side of Figure 6-16. Since each modulator has four levels on its axis, and the axis are 90 degrees apart, the result is a lattice of 16 locations (sometimes referred to as a constellation), where each location contains the bit structure of both modulating symbols. Each location on the lattice is described by a vector of given amplitude from the origin, and a phase with respect to the carrier, which is on the +I axis. 64 QAM results when a 3-bit symbol is applied to the input of each modulator.

Figure 6-16 16-QAM Modulator And Modulation Constellation

Table 6-1 Bit Structure and Level Assignment for a 2-Bit Symbol

Bit Combination Assigned Value1,1 +3

1,0 +1

0,1 -1

0,0 -3

levels 2n

=

Modulation AInput

CarrierOscillator

Modulation BInput

I BalancedModulator

90° PhaseShifter Σ

Q BalancedModulator

QAM

Output 0°180°+I axis-I axis

+Q axis

270°-Q Axis

90°

(2-Bits = 1 Symbol)

(2-Bits = 1 Symbol)

+3

+1

-1

-3

+3+1-1-3

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6.13 THE SYNCHRONOUS DETECTOR

A simplified synchronous detector is shown in Figure 6-17. The sideband signal is presented to a switch. The switch is controlled by the carrier signal such that the switch is closed only during the most positive part of the RF cycle of the carrier (60° to 120° for example).

Figure 6-17 A Form of a Synchronous Detector

When the carrier is maximum positive, the switch closes and a sample of the sideband can pass through. In part A of Figure 6-18, the carrier and sideband’s phase is the same as it was when the signal was modulated. Therefore, the polarity of the demodulated signal is the same as it was when it was transmitted.

In part B of Figure 6-18, the carrier phase is shifted 180°. This results in the polarity of the output signal being reversed. This is reasonable because the sideband signal is being sampled at the other side of the envelope.

In part C of Figure 6-18, the carrier is 90° out of phase with the sidebands. The output is zero because the sidebands are sampled as they go through the zero axis.

In part D of Figure 6-18, a phase modulated signal is substituted for the sideband signal. The synchronous detector output is the modulation of the phase modulated signal. This is because the zero crossing point of the phase modulated signal changes with respect to time, therefore, the carrier samples the signal earlier, later, or at the zero crossing point. This detector yields similar results if a frequency modulated signal is substituted for the phase modulated signal.

RFSignalInput

CarrierInput

DemodulatedSignalOutput

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Figure 6-18 Synchronous Demodulator Waveforms

+

0

-

+

0

-

+

0

-

+

0

-

+

0

-

+

0

-

+

0

-

+

0

-

+

0

-

DemodulatedSignal

SidebandSignal

ReinsertedCarrier

DemodulatedSignal

SidebandSignal

ReinsertedCarrier

(A) Carrier and Sideband Phase (B) Carrier Phase Reversed by 180°Same as When Transmitted

(C) Carrier and Sidebands 90° out of Phase (D) Carrier and Phase Modulated Signal

PhaseModulatedSignal +

0

-

+

0

-

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6.14 The Quadrature Demodulator

A quadrature demodulator is shown in Figure 6-19. In this figure, the IF signal is produced by a standard receiver front end, which consists of an RF amplifier, a mixer, and a local oscillator. The IF signal is applied to a phase lock loop, which consists of a phase detector and a carrier oscillator. The phase lock loop locks the carrier oscillator to the carrier sample (the pilot), and the oscillator supplies the recreated carrier to the two synchronous detectors.

6.14.1 The I and Q Synchronous Detectors

Refer to Figure 6-19. The two detector circuits are identical, the only difference being the phase of carrier applied to each detector. The I (in phase) synchronous detector is so named because it gets a carrier which is in phase with the original carrier (and the I sidebands). The result is that it detects the I sidebands and ignores the Q sidebands. The I detector is sometimes referred to as the AM detector, or in the case of the Tecktronix 1450 demodulator, is called the video detector.

The Q (quadrature) synchronous detector gets a carrier that is 90 degrees out of phase with the original carrier, and is in phase with the Q sidebands. The Q detector demodulates the Q sidebands and ignores the I sidebands. This detector is sometimes referred to as the PM detector, or in the case of the Tecktronix 1450 demodulator, is called the quadrature detector.

The output of a synchronous detector depends on the phase of the reinserted carrier. If the carrier is reinserted at a 45 degree angle to both the I and Q axis, its detected signal will consist of the sum of both the I and Q modulation components. IF the carrier is reinserted 180 degrees our of phase with the original carrier, the detector output will be an inverted I-modulating signal.

Figure 6-19 16-QAM Demodulator

I-SynchronousDetector

Q-SynchronousDetector

CarrierOscillator

90° PhaseShifter

PhaseDetector

IF SignalInputReceiver

FrontEnd

I SignalOutput

Q SignalOutput

PilotBandpassFilter

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Switch Mode of Operation of the Diode Ring Modulator11/19/03

7 Switch Mode of Operation of the Diode Ring Modulator

The double-balanced diode ring modulator, shown in Figure 7-1, can best be understood as a switching device which uses the RF carrier input to alternately pass the modulation input current through the upper and the lower half of the output transformer. Thus, the modulation input becomes the modulated RF output.

To produce an amplitude modulated signal, a dc bias is added to the modulating signal so that all parts of the modulating signal are at the same polarity.

For example, a dc bias could be added to a video signal so that the sync tip is at -0.8V, and the white level in the signal is at -0.1V, which is 12.5% of the sync tip voltage. The combined vid-eo and bias are sent to the modulation input through a large resistor (such as 470 ohms) to pre-vent the low impedance at the modulating input from affecting the video and the bias levels. The video level at the modulating input to the device would typically be about 0.07V.

An unmodulated RF carrier is supplied to the RF input of the modulator. The RF input level is adjusted to be high enough to fully turn on two of the series diodes (either CR1 and CR3 or CR2 and CR4) on each half-cycle. If the transformer turns ratios is 1:1 and the diodes are silicone, greater than 1.2Vp is required.

Figure 7-1 A diode ring modulator functioning as a doubly balanced modulator.

ModulationSource

CarrierGenerator

LeftPort

X Port

RightPort

RF Output

RLT2

CR1 CR2

CR4CR3

T1

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7.1 Analysis

Refer to the top half of Figure 7-2. If the instantaneous polarity of the RF carrier input is positive. The resultant secondary voltage forward biases diodes CR2 and CR4. Since each of the forward biased diodes and each half of the secondary of T1 develops 0.7V, the voltage at the junction of CR2 and CR4 is zero volts, making that point virtual ground. This grounds the top side of the primary of T2.

Refer to the bottom half of Figure 7-2. The instantaneous polarity of the modulating signal is shown as positive. Since the upper side of the primary of T2 is at ground, modulating current flows down through the upper half of the primary of T2. As a result, current flows down through the secondary of T2, and up through the load resistor, making the polarity of the load positive during this RF half cycle.

Refer to the top half of Figure 7-3. If the instantaneous polarity of the RF carrier input is negative. The resultant secondary voltage forward biases diodes CR1 and CR3. Since each of the forward biased diodes and each half of the secondary of T1 develops 0.7V, the voltage at the junction of CR1 and CR3 is zero volts, making that point virtual ground. This grounds the bottom side of the primary of T2.

Refer to the bottom half of Figure 7-3. The instantaneous polarity of the modulating signal is shown as positive. Since the lower side of the primary of T2 is at ground, modulating current flows up through the lower half of the primary of T2. As a result, current flows up through the secondary of T2, and down through the load resistor, making the polarity of the load negative during this RF half cycle.

7.2 Conclusion

From the above discussion, it should become apparent that the current flow through the load and the load voltage are proportional to the amplitude of the modulating signal, but the load voltage and current are switching at the RF carrier rate.

Furthermore, it the modulating signal polarity is positive, the RF output signal is in phase with the RF carrier input, and if the modulating signal polarity is negative, the RF output signal is out of phase (by 180 degrees) with the RF carrier input.

If the modulating signal is pure ac, the resulting RF output is a dual sideband suppressed carrier signal. The RF carrier and the modulating signal are not present in the output of this modulator because they are both balanced out due to the modulator design.

If a small dc component is added to the otherwise pure ac modulating signal, the modulator becomes unbalanced and a small amount of carrier appears in the RF output along with the sidebands. This is referred to as a dual sideband reduced carrier signal. As stated earlier, if enough dc bias is added so that the modulating signal is always the same polarity, the RF output will be an amplitude modulated signal.

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Figure 7-2 Positive Carrier Polarity (Top) and Resultant Modulation Current Flow (Bottom)

ModulationSource

CarrierGenerator

LeftPort

X Port

RightPort

RF Output

RLT2

CR1 CR2

CR4CR3

T1

+

-+

-

+

-

+

-

+

-

+

-0.7V

0.7V0.7V

0.7V

= Electron current flow This point is a virtual ground

ModulationSource

X Port

RightPort

RF Output

RLT2

-

+

-

+

-

-

+

-

+

1.4V

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Figure 7-3 Negative Carrier Polarity (Top) and Resultant Modulation Current Flow (Bottom)

ModulationSource

CarrierGenerator

LeftPort

X Port

RightPort

RF Output

RLT2

CR1 CR2

CR4CR3

T1

+

-

+

-+

- +

-

+

-

+

-0.7V

0.7V0.7V

0.7V

= Electron current flow

This point is a virtual ground

ModulationSource

X Port

RightPort

RF Output

RLT2

-

+

-

+

-

-

+

-

+

This point is a virtual ground

1.4V

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7.3 Conclusion

From the above discussion, it should become apparent that the current flow through the load and the load voltage are proportional to the amplitude of the modulating signal, but the load voltage and current are switching at the RF carrier rate.

Furthermore, it the modulating signal polarity is positive, the RF output signal is in phase with the RF carrier input, and if the modulating signal polarity is negative, the RF output signal is out of phase (by 180 degrees) with the RF carrier input.

If the modulating signal is pure ac, the resulting RF output is a dual sideband suppressed carrier signal. The RF carrier and the modulating signal are not present in the output of this modulator because they are both balanced out due to the modulator design.

If a small dc component is added to the otherwise pure ac modulating signal, the modulator becomes unbalanced and a small amount of carrier appears in the RF output along with the sidebands. This is referred to as a dual sideband reduced carrier signal. As stated earlier, if enough dc bias is added so that the modulating signal is always the same polarity, the RF output will be an amplitude modulated signal.

Since the diodes of this modulator are acting as switches, the RF output is free of the nonlinear characteristics of the diodes, resulting in an extremely linear modulator.

7.3.1 Performance Test Results

In test performed on a diode ring modulator, the peak RF output was slightly higher than twice the peak modulating signal input voltage, and the peak amplitude of the RF carrier input was slightly higher than the forward voltage required to cause two silicon diodes to conduct.

It was noted that if the RF carrier drive input was decreased very far from optimum, the modulation input voltage began to rise rapidly, indicating a drop of modulation input current. If the RF drive input was increased above the optimum level, phase distortion appeared at the modulated RF output as the output RF voltage level approached the zero axis. This suggested increased leakage of the input signal through to the output.

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Additional 8-VSB Topics11/19/03

Appendix A Additional 8-VSB Topics

This appendix offers some additional topics which are not covered in the main body of this document.

A.1 Details of the Transport Stream and Symbol Rates

The symbol rate which is supported by the transmitter is lower than the transport stream bit rate, even though the added field sync increases the bit rate by the factor of 313/312 and the Reed Solomon error protection increases the bit rate by 208/188. This causes the bit rate to increases, but due to the 2/3 trellis encoding the symbol rate is 1/2 of the new bit rate.

In the series below, per the ATSC definition, the NTSC horizontal frequency (Fh) is derived first. Then the symbol rate frequency (Fsym) and the transport layer bit rate (Ftp) are derived from Fh.

Fh (NTSC) = 63/4004 MHz = 15734.26573 Hz Fsym = 684 * Fh = 10.762 237 762 237 MHzFtp = 2 * 188/208 * 312/313 * Fsym = 19.392 658 459 9 MHz.

Note: Per FCC rules, the transport stream clock is allow a +/-54 Hz tolerance.

A.2 More About Segments and Fields

Refer to Figure 3-1. The 8-VSB signal consists of a series of segments and fields. One field consists of 313 segments. The segment sync marks the start of each of the 313 segments in a field. The first segment of a field is the field sync, the remaining 312 segments are data segments. Each data segment is comprised of the four-symbol segment sync plus 828 data symbols for a total of 832 symbols. Like the data segments, the field sync is also composed of the four-symbol segment sync plus 828 data symbols. Since 832 symbols occur in 77.3 usec, the symbol rate is 10.762 MHz.

The data in each of the data segments has been randomized and interleaved, but the data in the field sync segment is specified and in a definite order. This makes it easy for the receiver to identify the field sync.

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Figure A-1 8-VSB Data Frame

A.2.1 Data Segment Format

Ignoring the effects of the data interleaver, it is covenant to think of the data segment as consisting of three components, see Figure 3-2.

• The first is the 4-symbol segment sync. Since each symbol represents three bits, with the trellis encoder creating one redundant bit for every two data bits, the segment sync occupies the same length as the MPEG-2 packet sync byte that was discarded earlier.

• The second component is the 187 bytes of the original MPEG-2 packet.

• The third is the 20 Reed-Solomon parity bytes that were added to each packet.

The segment sync is easily recovered by the receiver since it is independent of data, and can be recovered down to a S/N ratio of 0 dB. The segment sync aids in data segment detection, also, the symbol clock is imbedded in the segment sync.

Segm

ent S

ync

Field Sync #1

Field Sync #2

Segm

ent S

ync

Data + FEC

Data + FEC

313

Segm

ents

in 2

4.2

ms

313

Segm

ents

in 2

4.2

ms

1 Segment (832 symbols in 77.3 usec)

One

Fra

me

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A.2.2 Field Sync Segment Format

Refer to Figure 3-3. The binary data field sync segment is 832 symbols long and repeats every 313 segments. It reduces the data efficiency slightly (by 1/313), bringing it down to 90%. Since the data in this pulse is not interleaved, is can be detected independently of data, down to a S/N ratio of 0 dB. It can be used as a known reference training signal for the receiver equalizer, and is used to aid in data field synchronization.

312/313 represents a data efficiency of 99.7% due to the addition of the field sync segment. To maintain proper rate of program reproduction, each data packet must be sent more quickly. The new packet time of 77.3 microseconds is determined by multiplying the input data packet time of 77.57 microseconds by 0.997 (the data efficiency mentioned above).

The 511 symbol PN sequence is used by the receivers in the equalizers to provide linear distortion reduction over a long term. Additional equalization is provided by the three 63-symbol PN sequences. The middle 63-symbol PN sequence is inverted in alternate field syncs to provide a differentiation between field sync 1 and field sync 2.

The 24 symbol level ID is used to identify the VSB mode (4, 8, 8-trellis, or 16) selected for transmission.

The 828 data symbols may be at any of the eight modulation levels,but the four-symbol segment sync always follows the modulation level pattern of +5, -5, -5, and +5.

Figure A-2 Trellis-Coded 8-VSB Data Segment Format

+7+5+3+1-1-3-5-7

+7+5+3+1-1-3-5-7

(111)(110)(101)(110)(011)(010)(001)(000)

Voltage Level

BinaryNumber

Voltage Level BeforePilot Addition (pilotlevel is +1.25)

828 SymbolsData SegmentSync 4 Symbols

Data Segment Sync4 Symbols in 0.37 usec

in 0.37 usec 187 Bytes

188 Byte MPEG-2

Data Segment832 Symbolsin 77.3 usec

Data Packet 20 ByteReed-Solomon Parity

Note: Due to lack of drawing width, the segment sync, data packet, andReed-Solomon parity symbol widths do not have a uniform scale.

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Figure A-3 Field Sync Segment Format

A.3 More About The Spectrum

A.3.1 Efficient Use of Power

In the NTSC system, transmitter output power is measured at the peak of sync. The sync pulses extend from 75% to 100% of modulation. Since power is proportional to the square of modulation, the power used by the sync pulse extends from 56.25% to 100% of the transmitter power. During black picture, which contains no usable content for the viewer, the average power is 59.5% of peak of sync power. When a picture is displayed, the average power ranges from 20% to 50% of peak of sync power. These figures do not take the visual, aural and chroma carriers into consideration, and they contain most of the transmitted power.

Most of the RF power of the DTV 8-VSB system is used to convey useful information rather than to produce high power carriers and sync pulses. The definition of DTV 8-VSB output power is the average RF output power. Due to its random nature, the modulation does not have a consistent peak power level to measure, but its average power is very stable. 99.9% of the time the peak RF power is 6 dB (4 x) greater than the average power.

Power is not wasted in the carrier because it is not transmitted. In place of the full amplitude carrier, a low level pilot is transmitted. Addition of the pilot increases the signal power level by only 0.3 dB. The pilot is a sample of the carrier giving its exact frequency and phase. The receiver uses the pilot to generate the carrier which it uses during the signal demodulation. Reliable carrier recovery can be achieved down to a S/N ratio of 0 dB.

A separate aural carrier is not needed since the aural information is digitally encoded and sent with the video data.

+7+5+3+1-1-3-5-7

Level

Field Sync832 Symbolsin 77.3 usec

Res

erve

d

LevelID

PN511 PN63 PN63(+/-)

PN63

92 12

4 63511 63 63 24 104Number ofSymbols

Repeated from previoussegment

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A.3.2 Efficient Use of Frequency Spectrum

Refer to Figure 4-1. The useful portion of the 6-MHz DTV channel is 5.38 MHz. This amounts to 90% of the allocated channel. 0.31 MHz is used on either end of the channel to provide room to roll off the signal and reduce adjacent channel interference. The frequency efficiency is even greater when the MPEG-2 compression methods are taken into account. In the NTSC system, 4.2 MHz represents the useful video bandwidth and 0.1 MHz is the useful aural bandwidth, which is 72% of the 6 MHz channel.

Figure A-4 8-VSB and NTSC Spectrums

6.0 MHz

5.38 MHz0.31 MHz 0.31 MHz

Pilot

6.0 MHz

Visual Chroma Aural

1.25 MHz 3.58 MHz 0.25 MHz

8-VSB Spectrum

NTSC Spectrum

0.7 (-3 dB)

1.0 (0 dB)

(-50 dB *)

* The IMD level at the edge of the channel (shoulder) is typically -50 dB at the outputof the exciter, and must be at least -37 dB at the output of the RF mask filter.

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A.3.3 8-VSB Similar to Single Sideband Transmission

The 8-VSB system is similar to the single sideband reduced carrier technique used in radio. In both systems, only one sideband is transmitted along with a sample of the carrier (although, some radio transmissions total eliminate the carrier sample). In the 8-VSB system, the carrier sample is called the pilot. Due to practical filter hardware constraints, a vestige of the lower sideband (0.31 MHz) is transmitted. This is achieved through the use of a 95 pole digital filter.

A.3.3.1 Single Sideband Waveforms, The Baseband Signal

A rectangular baseband modulating signal is shown in Figure 4-2. It is used to illustrate the modulating characteristics of the single sideband signal. The use of a rectangular modulating signal is incorrect for DTV, but it is used here for convenience of illustration. The DTV modulation levels, discussed in section 2.5 on page 2-6, are used in these illustrations.

Figure A-5 Baseband Modulating Signal

0-Axis

+7

+5

+3

-7

-5

-3

-1

+1

Baseband signalNote: Use of square shape is incorrect, used here for ease of explanation.

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A.3.3.2 Single Sideband Waveforms, The Modulation Envelope

The single sideband modulation envelope (shown in Figure 4-3) is modulated by the baseband signal shown above. The black dots in the drawing represent the original modulation levels. If the waveform in Figure 4-2 is traced, it will fit over the black dots in this diagram. Three points concerning single sideband modulation is illustrated here, they are:

1. The original modulating envelope (the black dots) and an opposite polarity duplicate (the un-marked levels) are carried by the single sideband signal.

2. The amplitude of the modulating signal is represented by the amplitude of the RF signal.

3. The polarity of the modulating signal (positive or negative) is represented by the phase of the single sideband signal with respect to its carrier.

A. When the black dots are positive, the single sideband signal is in phase with its carrier.

B. When the black dots are negative, the single sideband signal is out of phase with its carri-er.

Figure A-6 Single Sideband Modulation Envelope

0-Axis

+7

+5

+3

-7

-5

-3

-1

+1

Modulated Envelope (without pilot)Note: Use of square shape is incorrect, used here for ease of explanation.

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A.3.3.3 Adding The Pilot

A small percentage of the pilot (carrier) is added to the single sideband signal, as shown in Figure 4-4. This is accomplished by adding a small amount of dc bias to the modulating signal. This throws the doubly balanced modulator slightly out of perfect balance, thereby causing a small amount of carrier to be added to the signal.

Notice that the original modulating envelope is still present (observe the position of the black dots with respect to the modulating signal reproduced to the right). Also, the negative envelope (unmarked levels) still maintains the correct proportions. The effect is as though someone grabbed the envelope be the opposite sides of the envelope and pulled it apart, because all of the dotted levels have moved in the positive direction and the unmarked levels have moved in the negative direction.

There is a good reason for this. Remember, the envelope sections that have the dots on the positive side are in phase with the carrier, and therefore increase in amplitude when carrier is added; likewise, the envelope sections with the negative dots represent the out of phase condition, therefore, their amplitude decreases when the carrier is added.

Figure A-7 Single Sideband Signal With Pilot Sample Added

0-Axis

+8.25

+6.25

+4.25

-5.75

-3.75

-1.75

+0.25

+2.25

Modulated Envelope (with pilot)Note: Use of square shape is incorrect, used here for ease of explanation.

Original modulating envelopewith small dc offset.

Arrows show the direction of envelop movement when pilot (dc offset of +1.25 units) is added.

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A.3.3.4 Adding Full Amplitude Carrier

Adding a full amplitude carrier to the sideband signal causes the waveform to take on the appearance of an amplitude modulated signal, see Figure 4-5. As with the two single sideband waveforms previously shown, both the positive and negative modulation envelopes are still present, but each is on its own side of the zero volt axis.

Some phase error (called quadrature error) is experienced because only one sideband is present, but this does not present a great problem for amplitude modulation reception.

In the receiver, the pilot is used as an input to a phase locked loop to recreate the original carrier, which is used in the signal demodulation process. By using a synchronous detector, the original modulating signal can be accurately recovered.

Figure A-8 Adding Full Amplitude Carrier to the Sideband Signal

+7

+5

+3

-7

-5

-3

-1

+1

+7

+5

+3

-7

-5

-3

-1

+1

ZeroAxis

Original modulating signal

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A.4 Fitting Symbols Into a 6-MHz Channel

One 832 symbol segment occurs in 77.3 usec, which produces a symbol rate of 10.76 MHz, as shown below.

The modulated signal is passed through a root-raised-cosine filter to create the ringing sine wave-shaped symbols which keeps the bandwidth within 6-MHz. The raised and root raised cosine filters are discussed after the next section.

Figure A-9 Symbol Shape and Resulting Modulated Spectral Bandwidth

DTV RF Envelope Sample DTV spectrum (fits into a 6-MHz channel)

Frequency

Amplitude

(correct approach)

Time

Amplitude

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A.4.1 A Simpler Approach

Two approximations will be used to show the effect of pulse shape on bandwidth. Both use the time per symbol (the reciprocal of the symbol rate), which is 92.909 nsec.

The first approach assumes a 92.9 nsec rectangular waveform with a rise time (from 10% to 90%) of 9.29 nsec, see Figure 4-7. Using the approximation for pulse bandwidth (which is the reciprocal of two times the 10% to 90% rise time), the bandwidth is 53.8 MHz.

The second approximation, also shown in Figure 4-7, assumes that the pulse shape is 1/2 of a sine wave with a pulse width of 92.9 nsec. If the other half of the sine wave is drawn, the period of the cycle is 185.8 nsec. Since frequency equals one divided by the period, the frequency of this imaginary sine wave is 5.38 MHz. This is also the usable band width of the 8-VSB signal.

Figure A-10 Pulse shape and Bandwidth

Pulse Width

10%

90%

Rise time = 9.29 nsec

92.9 nsec

185.8 nsec

92.9 nsec Where: BW = bandwidth, and RT = rise time (10% to 90%)

BW1

2RT----------- 53.8MHz= =

If pulse is assumed to be 1/2 sine wave, the period of thesine wave is 185.8 nsec.

Where: BW = bandwidth, and TP

= period of the cycle, and

F BW1

TP------- 1

185.8 109–×

------------------------------- 5.38MHz= = = =

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A.4.2 The Raised Cosine Filter

The raised cosine filter is an integral part of the DTV system because it is able to produce rapid frequency response rolloffs with no group delay. Refer to the right side of Figure 4-8. The defined bandwidth extends from the flat portion of the response to the -6 dB rolloff points. The rolloff skirts extend to a predefined minimum level, which is the total bandwidth of the filter. The difference between the total bandwidth and the -6 dB bandwidth is referred to as the excessive bandwidth.

On the left side of Figure 4-8, the horizontal axis represents time, and on the right side of the drawing it represents frequency. The following facts about this figure are important.

1. The -6 dB bandwidth (right side) for all three filters is the same.

2. The top filter, with the shortest ringing time, has the widest total bandwidth.

3. The middle filter, with a longer ringing time, has a narrower total bandwidth.

4. The bottom filter, with the longest ringing time, has the narrowest total bandwidth (and the steepest skirts). This alpha factor (α) of this filter is low.

5. If the waveform had an infinite ringing time, the total bandwidth would equal the -6 dB band-width. This alpha factor (α) of this filter is large.

6. A filter with a steeper skirt (lower α) requires more poles than one with shallow skirts.

A.4.2.1 Raised Cosine Filter Alpha (α) Factor

The alpha factor of a raised cosine filter is a definition of the amount of excessive bandwidth of the filter. Excessive bandwidth is the difference between the total bandwidth and the -6 dB (useful) bandwidth. It can also be used to describe the steepness factor of the filter skirt response. A filter with α = 0 has a brick wall response (steepest skirts); and a filter with α = infinity would have an infinite excessive bandwidth.

The following formulas show the relationship between total bandwidth (BWTotal), -6 dB bandwidth (BWUseful), excessive bandwidth (BWExcess), symbol frequency (FSymbol), and α.

A.4.2.1.1 Ringing Period of a Raised Cosine Filter

One interesting fact about the raised cosine filter is the period (time) of the ringing cycle and the time of the main pulse (symbol time) are equal (see the left middle waveform of Figure 4-8), and are proportional to the -6 dB bandwidth of the filter.

Where T = symbol time = period of ringing cycle.

This is the feature that makes the narrow bandwidth DTV modulation possible.

αBWTotal

BWUseful------------------------ 1–=

BWTotal BWUseful 1 α+( )×=

BWExcess BWTotal BWUseful– α BWUseful×= =

FSymbol 2 BWUseful× 2BWTotal

1 α+---------------------×= =

T1

2 BWUseful×---------------------------------- 1 α+

2 BWTotal×------------------------------- α

2 BWExcess×----------------------------------= = =

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Figure A-11 Details About Raised-Cosine Filter

The left side of the above drawing is in the time domain.The right side of the above drawing is in the frequency domain.

-6 dBBW

TotalBandwidth

-6 dBBW

TotalBandwidth

-6 dBBW

TotalBandwidth

Least ringing results in most shallowskirts and widest total BW.

Greater ringing results in steeper skirts and narrower total BW.

Even greater ringing results in steeper skirts and narrowest total BW.

Amplitude Amplitude

Amplitude Amplitude

Amplitude Amplitude

T T T

Note: -6 dB bandwidth = 1/(2T).

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A.4.3 Root Raised Cosine filter

The desired raised-cosine filter response can be achieved by connecting two identical filters in series, see Figure 4-9. Each filter will have the defined 3 dB bandwidth and roll-off. The resultant response of both will have the defined bandwidth at the 6 dB points and the roll-off will be twice as steep as either filter. In this case, each individual filter is called a root-raised cosine filter. Such a filter arrangement is used for the filters in the DTV exciter and receiver. The root raised or raised-cosine filters have no group delay across their specified (3 or 6 dB) bandwidth.

In the DTV system, root raised cosine filters are used in the exciter and the receiver, see Figure 4-10. The exciter filter is used to limit the usable bandwidth to 5.38 MHz, remove most of the lower sideband, and reduce the out of band spurious emissions. The receiver filter is used to limit the out of band noise and reduce adjacent channel interference.

Figure A-12 Creating a Raised Cosine Filter With Two Root Raised Cosine Filters

Figure A-13 DTV System Raised Cosine Filter

Root RaisedCosine Filter


Input Output

Raised Cosine Filter



Input Output

DTV Raised Cosine Filter System

In Exciter In Receiver

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