5572 ieee sensors journal, vol. 16, no. 14, july 15, 2016 w-band multichannel fmcw ... · 2019. 5....

5572 IEEE SENSORS JOURNAL, VOL. 16, NO. 14, JULY 15, 2016

W -Band Multichannel FMCW Radar SensorWith Switching-TX Antennas

Seokchul Lee, Joungmyoung Joo, Junhyeok Choi, Wansik Kim, Hosang Kwon,Sangho Lee, Student Member, IEEE, Youngwoo Kwon, Senior Member, IEEE,

and Jinho Jeong, Member, IEEE

Abstract— In this paper, a multichannel frequency-modulatedcontinuous wave (FMCW) radar sensor with switching-transmit (TX) antennas is developed at W -band. To achieve ahigh angular resolution, a uniform linear array consisting of5 switching-TX and 12 receive (RX) antennas is employed withthe digital beamforming technique. The overall radar front-endmodule comprises a W -band transceiver and TX/RX antennas.A multichannel transceiver module consists of 5 up-conversionand 12 down-conversion channels, where one of the TX channelsis sequentially switched ON. The developed radar sensor gener-ates multiple digital beams with high gain, low side lobe level, andnarrow beamwidth. The design of each component in the radarmodule is described with the measurement results. The assembledFMCW radar sensor exhibits a measured angular resolution ofless than 2°, and a field of view of approximately 60°.

Index Terms— Angular resolution, frequency-modulatedcontinuous wave, radar sensor, switching antenna, transceiver,W-band.

I. INTRODUCTION

FREQUENCY-MODULATED continuous wave (FMCW)radars at millimeter-wave frequencies have been widelyused for industrial and military applications. They includeautomotive radars for a driver assistant system at 77/79 GHzand sensors for aircraft landing and obstacle avoidance at94 GHz [1], [2]. In recent years, FMCW radar sensors forunmanned ground vehicles (UGVs) have attracted interestowing to their ability to accurately extract range and velocityinformation [3], [4]. They can also capture images around thevehicle by using angular data. Even in adverse conditionssuch as fog, rain, and dusty environments, FMCW radars

Manuscript received March 22, 2016; accepted May 9, 2016. Date ofpublication May 12, 2016; date of current version June 16, 2016. This workwas supported in part by the Agency for Defense Development through theApplied Research Project and in part by the LIG Nex1 Company Ltd. Theassociate editor coordinating the review of this paper and approving it forpublication was Dr. Lorenzo Lo Monte.

S. Lee was with the Radar Research and Development Center, LIG Nex1Company Ltd., Yongin 446-798, South Korea. He is now with the KoreaElectronics Technology Institute, Seongnam 463-816, South Korea (e-mail:[email protected]).

J. Joo, J. Choi, and W. Kim are with the Radar Research andDevelopment Center, LIG Nex1 Company Ltd., Yongin 446-798,South Korea (e-mail: [email protected]; [email protected]; [email protected]).

H. Kwon is with the Agency for Defense Development, Daejeon 605-600,South Korea (e-mail: [email protected]).

S. Lee and Y. Kwon are with the Department of Electrical and ComputerEngineering, Institute of New Media and Communications, Seoul NationalUniversity, Seoul 151-742, South Korea (e-mail: [email protected];[email protected]).

J. Jeong is with the Department of Electronic Engineering, SogangUniversity, Seoul 121-742, South Korea (e-mail: [email protected]).

Digital Object Identifier 10.1109/JSEN.2016.2567450

perform well compared with optical sensors such as lasers.Furthermore, they can operate at a relatively lower peak powerthan pulse radars.

Radar sensors for UGVs require an improved detectioncapability with a higher angular resolution. It was empiricallydemonstrated that the azimuth angular resolution should bebetter than 2° to successfully distinguish two objects [5].Multichannel radars employing the digital beamform-ing (DBF) technique have been widely used to improve theresolution. Eight or more channels with DBF are typically usedto achieve the required beam-width and angular resolutionfor UGV radars [6]. W-band automotive radars using eightRX channels and DBF showed a typical angular resolutionranging from 2° to 5° and a 3 dB beam-width of approxi-mately 8.5°. It was shown in [7] that parameter estimationmethods can produce angular resolution higher than 4°. Driverassistant systems require higher angular resolution for variousfunctions. The eight-channel radar sensor at W-band in [8]exhibited a 3 dB beam-width of 3.9° and an angular resolutionof 1.7° using the Root-MUSIC algorithm.

Automotive radars minimize the number of RX channelsfor the advantages of low cost and compact size. On thecontrary, the reported FMCW radars for UGV applicationsemployed more than eight RX channels to attain a 3 dBbeam-width of less than 2°. In this study, we present aW-band FMCW radar sensor for UGV applications, where theswitching-antenna technique is adopted in the TX to reducethe number of RX channels. In section II, the fundamentalsof the multichannel uniform linear array (ULA) and theswitching-TX antenna technique are presented to improvethe angular resolution. Section III describes the operatingprinciple of the developed radar. The detailed design andexperimental results of each component are presented insection IV. Finally, the measurement results of each moduleand the performance of the assembled radar sensor are sum-marized in section V.

II. MULTICHANNEL ANTENNA ARRAY

Fig. 1(a) shows a ULA consisting of N identical RXantennas that are equally separated by d with a progressivephase. According to (1), the 3 dB beam-width of the ULA,which is closely related to the angular resolution, is given as

�θ3d B = λd cos θ

√N2 − 1 , (1)

where λ is the wavelength and θ is the steering angle [9].For a large value of N , the aperture size of the ULA is

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LEE et al.: W -BAND MULTICHANNEL FMCW RADAR SENSOR WITH SWITCHING-TX ANTENNAS 5573

Fig. 1. Configuration of multichannel ULA. (a) 1 TX and 60 RXantenna array. (b) 30 sequentially switching TX and 2 RX antenna array.(c) Five sequentially switching TX and 12 RX antenna array.

determined to be

d√

N2 − 1 ≈ d N. (2)According to (1) and (2), the 3 dB beam-width is inverselyproportional to the product of the aperture size and cosine ofthe steering angle. The 3 dB beam-width is minimum at thebroadside (θ = 0°) and smoothly increases until a maximumis reached at an end-fire (θ = 90°) [10].

To remove any grating lobes, the maximum spacing betweenadjacent array elements dmax is determined as

dmax = λ1 + sin |θ | . (3)

The calculated dmax is approximately 0.667λ and 0.54λ forθ = 30° and θ = 60°, respectively.

We can determine the spacing d and the number of arrayelements N by applying Equations (1)–(3) for a requiredangular resolution at a given azimuth angle and frequency. Ourdesign goal is to achieve an angular resolution better than 2°and an azimuth field of view (FoV) of approximately 60° atW-band. Considering the practical implementation of theantenna array, we set d to 0.61λ and N to 60.

However, it is still difficult to implement 60 RX channelsat W-band, because many RX channels require a complicatedlocal oscillator (LO) distribution, their packaging cost ishigh, and RX channel calibration is difficult. In addition,it is not easy to guarantee uniform performance in eachRX channel.

One approach to alleviating this problem is a switching-RXantenna technique, where only one RX channel is switchedon under the radar operation [5], [9]. This method needsthe minimum number of active RF components, but the RXperformance is limited [11]. At millimeter-wave frequency, theRX noise figure (NF) increases owing to the losses of theswitches and corresponding feed lines. Therefore, this methoddegrades the signal-to-noise ratio (SNR) and the detectionrange of the radar.

Another approach is a switching-TX technique based onthe multi-input multi-output (MIMO) radar fundamentals [12].MIMO radars can separate the reflected signals by the target,while the different signals are transmitted through multiple TXantennas. It can be made possible to synthesize virtual antennapositions corresponding to the spatial convolution of TXand RX phase centers [12], [13]. It implies that the same aper-ture size or beam-width can be accomplished using smallernumber of TX and RX antennas, as long as the product of thenumbers of TX and RX antennas (NT X and NR X , respectively)is equal to N .

Fig. 1(b) and (c) demonstrate the switching TX multi-channel radars based on this techniques, where NT X = 30and NR X = 2 in Fig. 1(b), and NT X = 5 and NR X = 12in Fig. 1(c). In both cases, NT X NR X = N = 60. It can beeasily explained as follows that the switching TX radars cansynthesize the virtual antenna positions exhibiting the samephase delay as the ones in Fig. 1(a). For example, the phasedelay from TX antenna #1 to RX antenna #60 is 59dsinθ inFig. 1(a), which is a maximum value that can be obtainedin this configuration. In Fig. 1(b), the same phase delay canbe obtained by TX antenna #30 and RX antenna #2 (totalphase delay = 29dsinθ + 30dsinθ = 59d sin θ). Note that thespacing of TX antenna is d and that of RX antenna 30d. Thephase delay between TX antenna #5 and RX antenna #12 isalso equal to 59dsinθ in Fig. 1(c), where 12 RX is situatedwith a spacing of d and 5 TX antennas with a spacing of 12d.

The RX antenna spacing in Fig. 1(b) is determined by theproduct of the number and the spacing of the TX anten-nas [14], [15]. The configuration in Fig. 1(b) requires longlines for LO distribution due to the RX spacing of 30d.In addition, a complex arrangement of TX switches could limitthe TX output power level and then also degrade the detectionrange of the radar. Therefore, the configuration in Fig. 1(c),which has shorter RX spacing as well as smaller number ofTX and RX antennas, is selected in this study.


Fig. 2. Block diagram of proposed W-band FMCW radar front-end with five sequentially switching-TX and 12 RX array antennas.

III. ARCHITECTURE OF RADAR FRONT-END MODULE

Fig. 2 shows the block diagram of the proposed W-bandmultichannel FMCW radar front-end. It consists of trans-ceiver and antenna modules with 5 switching-TX and 12 RXantennas. TX and RX calibration channels are also includedin the module.

The transceiver module comprises a signal generator,TX up-conversion, W-band TX, and RX down-conversionblocks as shown in Fig. 2. The FMCW chirp signal is gener-ated using a dual-channel direct digital synthesizer (DDS). Thebeat frequency fbeat and range resolution δR are determinedas follows:

fbeat = BWT

2R

c, (4)

δR = c2BW

, (5)

where BW is the bandwidth of the chirp signal, T is themodulation period time, R is the distance of the target, andc is the velocity of light [6], [16]. In this study, the BWand T of chirp signal is determined to be 200 MHz and7 μs, respectively, for the fast modulation. Therefore, fbeat is19 MHz and δR is 0.75 m at R = 100 m.

Fig. 3. Timing diagram of the switching TX multichannel radar.

Fig. 3 shows a timing diagram of the switching TX multi-channel radar. One of five TX channels is sequentially selectedto transmit the chirp signal by the switches, so that RX canreceive the time-divided signals separately. The power ampli-fiers (PAs) in the unused channels are turned off to reducepower consumption and thus heat generation. The switchon-time, Ton , is determined to be larger than T , consideringseveral factors such as the switching speed of the switches


and PA, and the transient signal components caused by adramatic frequency change in the chirp signal when thechannel is switched. In this work, the guard time, tg = Ton –T = 0.2 μs, is used for the stable operation. However,this guard time can cause non-coherency and lead to theincrease of side lobe in performing DBF. This adverse effectis minimized by eliminating this portion of the signal in thesignal processing.

In normal radar operation, the dual-channel DDS (Analogdevice AD9958) generates two identical chirp signals for theTX and RX LO paths. It can also generate two chirp signalswith different frequency slopes, which can be used to calibratethe radar during operation.

The generated chirp signals at 200 MHz by the DDSare frequency-translated to Ku-band by double up-conversionusing two mixers (Mixer 1 and 2), which relaxes the require-ment of high quality (Q)-factor band-pass filters (BPFs) atthe output of the mixers to reject the carrier (LO) signals.Therefore, the compact BPFs (BPF 1 and 2) fabricated onPCB can be used to sufficiently reject carrier signals insteadof high-Q bulky cavity filters. The Ku-band signal is thenup-converted to W-band through the frequency multiplierby 6. The center frequency of the chirp signal can be controlledfrom 76.2 to 76.8 GHz in steps of 0.1 GHz by PLL 2.

In the W-band TX block, a single-pole six-throw (SP6T)switch is realized by combining two SP3Ts and one SPDT,to select one channel from five TX and calibration channels.PAs are added in the last TX stage for each channel to deliveran output power higher than 10 dBm, despite the losses dueto the switches, feed lines, and bonding wires. Microstrip-to-waveguide (MS-to-WG) transitions are used between PAs andantenna modules because they are connected using WGs.

For amplitude and phase calibration of the TX signal, partof the signal is coupled to coupler 1, which is installed in theantenna module. Coupler 2 allows the TX signal to be directlyapplied to RX channels for RX calibration.

The reflected signals from the target reach each of the12 RX channels via the RX antennas. They are first amplifiedby a low-noise amplifier (LNA) and then mixed with theLO signals that are generated by the DDS. For low-noiseand high-linearity performance, the core down-conversioncircuit was designed in a single-chip microwave monolithicintegrated chip (MMIC) by the authors [17]. The dual-channelRX MMIC consists of two RX channels (two LNAs and twomixers) and a single LO buffer amplifier, as shown in Fig. 2.An additional buffer amplifier 2 is used to fully pump theW-band mixers.

IV. DESIGN OF RADAR FRONT-END MODULE

A. Transceiver Module

Fig. 4(a) shows a photograph of the fabricated transceiverand antenna modules, and how they are combined with eachother. The antenna module sits on the transceiver module, andthey are connected by WGs. The total module dimensionsare 225 mm × 100 mm × 57 mm. As shown in Fig. 4(b),the TX/RX blocks and extra blocks such as the control logicand the biasing circuits are placed on the top side of the

Fig. 4. Photograph of antenna and transceiver module. (a) Antennaand transceiver module. (b) Top side of transceiver module without cover.(c) Dual-channel RX carrier and dual-channel RX MMIC.

transceiver module. The W-band TX and RX blocks arerealized on a multilayer stack-up PCB as shown in Fig. 5,which accommodates the IF signal paths of 12 RX channels,bias lines, and a 6-way LO path using six metal layers. Severaldielectric materials are used such as Taconic’s TSM-DS3and FR4.


Fig. 5. Six-layer stack-up PCB for TX and RX blocks.

Fig. 6. Design of element antenna. (a) Photograph of fabricated elementantenna. (b) Power distribution of each element.

TX/RX blocks were implemented by using commer-cially available MMICs such as a frequency multiplier(x6)-integrated medium PA (UMS CHU3377), bare-dieSPDT switch (Hittite HMC-SDD112), flip-chip SP3T switch(TriQuint TGS4305-FC), and LO buffer amplifier 2 (HittiteHMC-ALH509). Owing to the different type of assemblyprocess used, the SPDT and SP3T switches are separated byusing different chip carriers, which results in the additionalloss. To compensate for the switch and feed line losses, each5-TX channel employs PAs (Hittite HMC-AUH318), whichprovide a gain of 24 dB and an output power of 17.5 dBm atthe 1 dB compression point in a bare die.

In the RX down-conversion block, 7 dual-channel single-chip RX MMICs were used for 1 TX calibration and12 RX channels. Each dual-channel RX MMIC was equippedwith an individual carrier for easy replacement as shown inFig. 4(b) and (c). It has a conversion gain of 14.7 dB, a singleside band (SSB) NF of 10.4 dB at an IF of 20 MHz, and aninput 1-dB compression point (P1dB_in) of −25.1 dBm. Thedown-converted IF signal is 36 dB amplified by a three-stageoperational (OP) amplifier and filtered to pass only the desiredbeat frequency components below 20 MHz.

B. Antenna Module

As depicted in Fig. 4(a), 12 RX and 5 TX antennas arepositioned on the left and right parts in the front side ofthe module, respectively. To be specific, 12 RX antennas arelinearly arranged with a spacing of 0.61λ, and 5 TX antennaswith a spacing of 7.32λ.

The element antenna used in the TX and RX array is shownin Fig. 6(a). It consists of 16 MS radiating patches on a

Fig. 7. Isolation between TX and RX channels. (a) Bulkhead and elementantenna. (b) Simulated azimuth radiation pattern of a single element antennadepending on h1 and d1. (c) Measured isolation between the TX #1 andRX #12 antenna of four sample antenna module.

single-layer PCB (5-mil-thick TLY-5) in the form of openstubs. The open stubs are designed to have an orthogonalstructure; that is, the horizontal polarization component isadded to the vertical one in order to ensure 45° polarization.This can reduce the interference between the signals from thedifferent radars in the opposite direction. The 3 dB beam-width was determined to be more than 60° in azimuth and4° in elevation. For a narrow beam-width and low side lobelevel (SLL) in elevation, the target power distribution of16 element antennas is set up as shown in Fig. 6(b). Eachelement was optimized by adjusting the width and length.Furthermore, the spacing between the elements is designedto be in-phase at the operating frequency.

As shown in Fig. 4(a), TX #1 and RX #12 antennaare located in very close proximity, which can causeTX power to leak into RX channel. To minimize RFleakage, a bulkhead was employed between TX and RXantennas as shown in Fig. 4(a) and Fig. 7(a). There are


Fig. 8. Structure of W-band MS-to-WG transition.

TABLE I

DIMENSIONS OF W-BAND MS-TO-WG TRANSITION

Fig. 9. Measured and simulated S11 and S21 of W-band MS-to-WGtransition.

two design parameters in the bulkhead: the height of thebulkhead, h1, and a distance between the bulkhead and anelement antenna, d1. The higher h1 can lead to better isolation,but it can distort azimuth beam pattern of antenna as shownin Fig. 7(b). Therefore, we determined h1 and d1 to be2 and 10 mm based on the simulation results, respectively, con-sidering both RF leakage and antenna beam pattern. Fig. 7(c)shows the measured isolation performance between TX #1 andRX #12 antenna better than 45 dB for all four sample antennamodule, which is sufficient to meet the design goal.

C. MS-to-WG Transition

A MS-to-WG transition is designed to connect the PAwith antennas. There are several techniques that can be usedto design the MS-to-WG transition at millimeter-wave fre-quencies [18]–[20]. A conventional E-plane probe transitionexhibits a broadband characteristic, but it needs a back shortat a precise position from the probe [18]. In this study, thetransition using an MS patch antenna was designed withoutrequiring a precise back short, as shown in Fig. 8 [19], [20].The power from the MS line radiates through the MSpatch antenna to the WG. The dimensions of the designedMS-to-WG transition are shown in Table I.

The simulated and measured performance of the W-bandtransition is compared in Fig. 9. The measured insertion

Fig. 10. Six-way power divider. (a) Layout. (b) Schematic of two-way powerdivider. (c) Simulated S-parameters.

loss per single transition is 1.35 dB at 76.5 GHz, which isapproximately 0.92 dB larger than the simulation result. Themeasured return loss is 28.6 dB at 76.5 GHz, which is betterthan 10 dB across more than 5 GHz.

The PA chip is wire-bonded to the microstrip line on thealumina substrate which is then connected to the MS-to-WGtransition. The parasitic inductances of the bond wires candegrade the performance of the PA especially at frequenciesas high as W-band. Therefore, we tried to minimize theparasitic inductance by using two 1 mil-thick bond wiresbetween the PA and microstrip line, and two 3 mil-thick ribbonbonds between the microstrip line and MS-to-WG transition.The length of wires and ribbons were also minimized andestimated to be less than 150 μm. The fabricated TX mod-ule consisting of the PA chip, microstrip line section, andMS-to-WG transition exhibited an acceptable output powerof 12.6 dBm at W-band.

D. 6-Way Power Divider

A 6-way power divider is used to distribute LO power andpump six RX mixers as shown in Fig. 2. Fig. 10(a) showsthe designed 6-way power divider using five Wilkinson-type2-way power dividers [21], [22]. The schematic of constituting2-way power dividers is illustrated in Fig. 10(b), where twohalf-long transmission lines are added to connect the isola-tion resistor (2Z0) with output ports. These additional lines


TABLE II

TYPICAL MEASURED PERFORMANCE OF TRANSCEIVER MODULE

provide the electrical isolation and physical separation of twooutput ports, so that five 2-way power dividers can be easilyconnected without using sharply-bended lines as shown inFig. 10(a). Fig. 10(c) shows the simulated S-parameters of the6-way power divider. The higher insertion loss is observedalong the paths in LO #3 ∼ #6 compared with the paths inLO #1 and #2, which is caused by the excessive losses inthe additional power divider (circle in Fig. 10(a)) and longerinterconnection lines. To minimize the amplitude mismatchand to provide an equal LO powers to each mixer, gainof LO buffer amplifiers 2 in the path LO #1 and #2 isreduced by controlling the bias voltages. There are also phasemismatches between LO paths, which are compensated for bythe calibration in IF and digital signal processing.

V. PERFORMANCE OF MODULE AND RADAR SENSOR

A. Performance of Transceiver Module

For proper operation of the FMCW radar, the main designgoals of the transceiver at 25 °C are determined as follows:a TX output power greater than 10 dBm, a RX conversiongain (CG) of 46 ± 3 dB, and a RX SSB NF less than 17 dBat an IF of 20 MHz.

Table II summarizes the measured TX and RX performanceof the transceiver module. The output powers at the WGports of 5 TX channels (without the antennas connected) were

Fig. 11. The measured radiation pattern of TX and RX element antennas.(a) Azimuth pattern of TX element antennas. (b) Elevation pattern ofTX element antennas. (c) Azimuth pattern of RX element antennas.(d) Elevation pattern of TX element antennas.

12.6 ± 1 dBm at 76.5 GHz. The output power variation waswithin 0.8 dB from 76.2 to 76.8 GHz. The phase noise atan offset frequency of 100 kHz was typically −81.7 dBc/Hzat 76.5 GHz with a variation of less than ±1.6 dB betweenchannels.


TABLE III

TYPICAL MEASURED PERFORMANCE OF ANTENNA MODULE

Fig. 12. Photograph of the assembled FMCW radar sensor.

The RX performance of the transceiver module was mea-sured at an IF (fIF) of 20 MHz. The measured CG of theRX module was 45 dB. The SSB NF was calculated from thefollowing equation [23]:

N F(d B) = 174d Bm/H z + Nout (d Bm/H z) − CG(d B) (6)In (6), the output noise power Nout was measured with thespectrum analyzer when an input RF port was terminated by50 ohm. The measured SSB NF was 15 dB with a variationof ±1 dB between 12 RX channels, which was caused bythe variations in packaging elements such as MS-to-WGtransitions and bond wires.

The performance of the transceiver module was also verifiedat an ambient temperature from −40 °C to +85 °C. As shownin Table II, the transceiver module exhibited a relatively smallvariation in CG and NF depending on the ambient temperature,owing to the temperature compensation circuit included in thedual-channel RX MMIC.

P1dB_in was also measured to test the linearity performanceof the RX module, which is related to the minimum detectionrange. The measured P1dB_in was typically below −29 dBm.The VSWR of the TX and RX channels was measured tobe less than 2.5:1. The measured isolation between adja-cent RX channels was typically 24.5 dB, the worst valuebeing 21 dB.

The DC power consumption of the transceiver module was28.48 W when all components were switched on. The powerconsumption was reduced to approximately 12 W when severalcomponents were switched off according to the time schedulein the real radar operation.

Fig. 13. Measured beam pattern magnitude of radar signal. (a) Beam patternfor target at bore-sight (θ = 0°). (b) Beam pattern for target at θ = −30°.(c) Beam pattern for target at θ = +30°.

B. Performance of Antenna Module

Fig. 11 shows the radiation patterns of the TX/RX elementantennas for the azimuth and elevation angles. They weremeasured in the electromagnetic anechoic chamber using thecompact range measurement method [24]. The antenna gainwas higher than 13 dBi at 76.5 GHz for all the elementantennas. The gain variation of the RX antennas was approx-imately 3.8 dB, which was somewhat greater than that of theTX antennas. This variation appears to be caused by the lengthdifference between the feed lines of the RX antennas. Themeasured 3-dB beam-width was 66.5° for the azimuth and4.6° for the elevation angle, respectively.

Table III summarizes the typical performance of the antennamodule. The SLL in the elevation was measured to be less than−17 dB, and the VSWR of the overall element antennas wasless than 2.5:1 across the entire band. The isolation betweenthe TX #1 and RX #12 antennas was 45 dB.


TABLE IV

PERFORMANCE COMPARISON OF REPORTED W-BAND FMCW RADAR SENSORS

C. Performance of FMCW Radar Sensor

Fig. 12 shows the assembled FMCW radar sensor. In orderto evaluate the angular resolution of the radar, the target waslocated at the bore-sight of the radar (θ = 0°). The distance Rbetween the radar and the target was 20 m. The received radarsignals were analyzed by performing DBF using a 64-pointfast Fourier transform (FFT). Fig. 13(a) shows the peak valueof the DBF result as a function of the azimuth angle. The peakwas detected at θ = +1.46° in Fig. 13(a). This is very closeto theoretical value of θ = 0°, indicating that the radar canaccurately detect the target location. The magnitude rapidlyreduces as θ deviates from +1.46°. The values at θ = +0°and θ = +2.93° are approximately 10 dB lower than the peak,as shown in Fig. 13(a). This result confirms that a 3 dB beam-width of the radar is less than 2°.

The FoV of the radar was also measured. For this purpose,the target was located at θ = −30° and +30°, and themagnitude of the radar signal was measured as shown inFig. 13(b) and (c), respectively. These figures demonstrate thatthe FoV of the radar is approximately 60°.

The performances of reported W-band FMCW radar sen-sors are summarized and compared in Table IV. This workexhibits the excellent and comparable performance in angu-lar resolution and FoV, especially among the radar sen-sors for automotive vehicle applications. In addition, theradar sensor in this work operates using a very short chirpsignal, which can facilitate the detection of fast-movingtargets.

VI. CONCLUSION

In this paper, a W-band FMCW radar sensor is developedfor UGV applications. The number of RX channels was mini-mized by utilizing the switching-TX antenna technique, while

maintaining the angular resolution. The fabricated transceiverand antenna modules showed the desired performance, satisfy-ing the requirement of FMCW radars. The assembled FMCWradar sensor showed excellent angular resolution. Currently,we are conducting research proving the applicability of thefabricated FMCW radar sensor to UGVs.

ACKNOWLEDGMENT

The authors thank Jeongtae Kim and Dr. Chae for their helpin the design and fabrication of the radar module.

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Seokchul Lee received the B.S. degree in elec-trical and electronics engineering from YonseiUniversity, Seoul, South Korea, in 2010, and theM.S. degree in electrical and computer engineeringfrom Seoul National University, Seoul, in 2012.From 2012 to 2015, he was with the Radar R&DCenter, LIGNex1, Yongin, South Korea, where hewas involved in the development of transceiver sys-tem for FMCW radar. Since 2015, he has beenwith the Korea Electronics Technology Institute,Seongnam, South Korea. His research interests

include RF/millimeter-wave integrated circuits and systems for wirelesscommunication and radar.

Joungmyoung Joo received the B.S. andM.S. degrees in electrical and electronics engineer-ing from Hongik University, Seoul, South Korea,in 2011 and 2013, respectively. Since 2013, he hasbeen with the Radar Research and DevelopmentCenter, LIG Nex1 Company Ltd., South Korea,where he was involved in the development of theantenna system for FMCW radar. His researchinterests include various antenna system of theradar and FMCW radar system architecture.

Junhyeok Choi received the B.S. andM.S. degrees in electrical and electronicsengineering from Kyungpook National University,Daegu, South Korea, in 2007 and 2009, respectively.Since 2009, he has been with the Radar Researchand Development Center, LIG Nex1 CompanyLtd., South Korea, where he was involved in thedevelopment of system for FMCW radar. Hisresearch interests include FMCW radar systemarchitecture.

Wansik Kim received the B.S., M.S., andPh.D. degrees from Konkuk University, Seoul,South Korea, in 1991, 1993, and 2005, respectively,all in electrical and electronics engineering. In 1999,he joined the Institute for Advanced Engineering,Yongin, South Korea, where he was involved in thedevelopment of millimeter-wave transceiver systemfor FMCW radar. Since 2005, he has been with theRadar Research and Development Center, LIG Nex1Company Ltd., South Korea, where he was involvedin the development of transceiver system for FMCW

radar. His research interests include integrated circuits and systems fromX-band to millimeter-wave.

Hosang Kwon received the B.S. and M.S. degreesin electrical and electronics engineering fromKyungpook National University, Daegu,South Korea, in 1997 and 1999, respectively.Since 1999, he has been with the Agency forDefense Development, South Korea, where hewas involved in the development of transceiversystem for FMCW radar. His research interestsinclude monolithic microwave integrated circuitsand transceiver system.

Sangho Lee (S’13) was born in Suwon,South Korea, in 1986. He received the B.S. degreein electrical engineering from Seoul NationalUniversity, Seoul, South Korea, in 2011, where heis currently pursuing the Ph.D. degree in electricaland computer engineering.

His research activities include millimeter-wave/RFintegrated circuits and system design for wirelesscommunication and RADAR especially high-powerand broadband PA design.


Youngwoo Kwon (S’90–M’94–SM’04) was bornin Seoul, South Korea, in 1965. He received theB.S. degree in electronics engineering from SeoulNational University, Seoul, in 1988, and the M.S.and Ph.D. degrees in electrical engineering from theUniversity of Michigan at Ann Arbor, Ann Arbor,MI, USA, in 1990 and 1994, respectively.

He was with the Rockwell Science Center as amember of the Technical Staff from 1994 to 1996,where he was involved in the developmentof millimeter-wave monolithic integrated circuits.

In 1996, he joined as a Faculty Member the School of Electrical Engi-neering, Seoul National University, where he is currently a Professor.He is a co-inventor of the switchless stage-bypass power amplifier architectureCoolPAM. He co-founded Wavics, a power amplifier design company, whichis now fully owned by Avago Technologies. In 1999, he was awarded aCreative Research Initiative Program by the South Korean Ministry of Scienceand Technology to develop new technologies in the interdisciplinary area ofmillimeter-wave electronics, MEMS, and biotechnology. He has authored orco-authored over 150 technical papers in internationally renowned journalsand conferences. He holds over 20 patents on RF MEMS and power amplifiertechnology.

Dr. Kwon has been an Associate Editor of the IEEE TRANSACTIONS ONMICROWAVE THEORY AND TECHNIQUES. He has also served as a Techni-cal Program Committee Member of various microwave and semiconductorconferences, including the IEEE International Microwave Symposium, theRF Integrated Circuit Symposium, and the International Electron DevicesMeeting. Over the past years, he has directed a number of RF research projectsfunded by the South Korean Government and U.S. companies. He was arecipient of a Presidential Young Investigator Award from the South KoreanGovernment in 2006.

Jinho Jeong (S’00–M’05) received the B.S., M.S.,and Ph.D. degrees in electrical engineering fromSeoul National University, Seoul, South Korea,in 1997, 1999, and 2004, respectively. From2004 to 2007, he was with the University ofCalifornia at San Diego, La Jolla, as a Post-DoctoralScholar, where he was involved with the design ofhigh-efficiency and high-linearity RF power ampli-fiers. In 2007, he joined the Department of Electron-ics and Communications Engineering, KwangwoonUniversity, Seoul, South Korea. Since 2010, he has

been with the Department of Electronic Engineering, Sogang University,Seoul. His research interests include monolithic microwave integrated circuits,THz integrated circuits, high-efficiency/high-linearity power amplifiers andoscillators, and wireless power transfers.

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