2808 ieee transactions on microwave … et al.: dual-mode ring resonator bandpass filter 2811 fig....

2808 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 60, NO. 9, SEPTEMBER 2012

Dual-Mode Ring Resonator BandpassFilter With Asymmetric InductiveCoupling and Its Miniaturization

Tsu-Wei Lin, Jen-Tsai Kuo, Senior Member, IEEE, and Shyh-Jong Chung, Senior Member, IEEE

Abstract—Dual-mode ring resonator filters are implementedwith asymmetric inductive perturbation for creating transmissionzeros on both sides of the passband. In analysis, dependence ofthe resonance modes and the zeros on positions and sizes of boththe inductive and capacitive perturbations is investigated. Undercertain conditions, the even- and odd-mode frequencies for acapacitively perturbed ring are the same as the odd and even ones,respectively, for a ring with inductive perturbation. Theoreticalbackground is clearly explained how the two transmission zerosare split up from the center frequency. Two dual-mode ringresonator filters are fabricated and measured for demonstra-tion. To obtain a miniaturized circuit area, the - ring trace isfolded into a double-ring or spiral configuration. The inductiveperturbation is chosen as the crossover and implemented by ashort high-impedance coplanar waveguide interconnection in theground plane of the microstrip. Measurement results show goodagreement with the simulation responses.

Index Terms—Bandpass filter, capacitive coupling, dual-mode,inductive coupling, microstrip, miniaturization, ring resonator.

I. INTRODUCTION

D UAL-MODE ring resonator filters are one of the mostwidely used bandpass filters in microwave frequencies.

It is popular because of its compact size, good frequency selec-tivity, and ease of design. The earliest work of the dual-modering resonator can be traced to 1970s [1]. In the traditional de-sign, the input and output (I/O) ports are separated spatially at90 and a capacitive patch is placed at the end of the symmetricplane. The purpose of the perturbation patch is to split up thetwo degenerate modes so that a nonzero bandwidth can be ob-tained. In addition, transmission zeros can be created on bothsides of the passband, leading to a quasi-elliptic function re-sponse. The transmission zeros can be predicted by the trans-mission-line theory [2]. In [3], given that the bandwidth is con-stant, the attenuation pole frequencies can be controlled by theperturbation size and the angle between the I/O ports. In [4], the

Manuscript received December 21, 2011; revised May 14, 2012; acceptedJune 04, 2012. Date of publication July 05, 2012; date of current version August28, 2012. This work was supported in part by the National Science Council,Taiwan, under Grant NSC 100-2221-E-182-059-MY2, and in part by ChangGung University under Grant UERPD2A0021.T.-W. Lin and S.-J. Chung are with the Institute of Communications Engi-

neering, National Chiao Tung University, Hsinchu 300, Taiwan (e-mail: [email protected]; [email protected]).J.-T. Kuo is with the Department of Electronic Engineering, Chang Gung

University, Taoyaun 333, Taiwan (e-mail: [email protected]).Color versions of one or more of the figures in this paper are available online

at http://ieeexplore.ieee.org.Digital Object Identifier 10.1109/TMTT.2012.2205936

self-coupled ring resonator is used to devise a dual-mode filterwith multiple transmission zeros.Reducing circuit size is always important in implementation

of microwave integrated circuits. The area of a conventionalsquare ring resonator is . Several techniques have beendeveloped for shrinking the ring area [5]–[13]. In [5], the con-ductor trace is folded to form ameander ring. In [6], the periodicstepped-impedance ring is a reduced-size filter. The slow-waveeffect is also a good approach for miniaturizing the ring, e.g.,[7]. The shunt capacitors in [8] and open radial stubs in [9] at-taching to the peripheral of the ring can significantly reduce thecircuit area. In [10], the dual-mode ring is periodically loadedwith butterfly radial slot cells on the ground plane to achieve sizereduction and wide upper rejection band. In [11], three quartersof a - ring is replaced by a microstrip to coplanar wave-guide (CPW) broadside coupled section so that the area is onlyaround 25% of that of a full-length ring. Each of the four sidesof the dual-mode ring in [12] is loaded by a stub with cascadedalternative high- and low-impedance sections for circuit minia-turization and suppression of its second harmonic. In [13], thesize reduction is accomplished by synthesized microstrip linesin an asymmetric form. Good rejection levels are obtained at thesecond, third, and fourth harmonics of the passband.It is shown that if the patch in the symmetric plane is re-

placed by a cut, i.e., capacitive coupling is changed to inductive[14], the passband will be changed from quasi-elliptic functionto a response with two transmission zeros on the real axis ofthe complex -plane [15]. In [16], the attenuation pole frequen-cies for both capacitive and inductive coupling are derived. Theperturbations can be placed in the symmetric axis or in an axisorthogonal to the symmetric axis, forming totally four possibleconfigurations. It is worth mentioning that most of the abovedual-mode rings have their perturbations symmetric about theI/O ports. There have been few dual-mode ring resonator fil-ters with asymmetric perturbation thus far [17]. It is shown thatif the patch or cut perturbation is moved away from the sym-metric plane, the two zeros will change frequencies, showinga flexible technique for realizing various passbands, althoughthe passband becomes asymmetric. The perturbation positionsin [17], however, are limited to the corners of the hexagonalring.In this paper, a dual-mode ring resonator is studied with either

capacitive or inductive perturbation at an arbitrary position. Theinput and output ports can be spatially separated by a non-90angle. The non-90 I/O separation is useful for simultaneousexcitation of multiorder resonance modes in a ring resonator sothat dual-mode dual-band [18] and dual-mode triple-band [19]

0018-9480/$31.00 © 2012 IEEE

LIN et al.: DUAL-MODE RING RESONATOR BANDPASS FILTER 2809

Fig. 1. Dual-mode square ring resonators with a perturbation in symmetricplane. (a) Capacitive. (b) Inductive. Perturbation in an axis perpendicular to thesymmetric axis. (c) Capacitive. (d) Inductive. Dashed lines in (a) and (b) showthe symmetric planes for analysis of resonance modes.

filters can be realized. In Section II, analysis of transmissionpoles and zeros of a perturbed ring resonator is conducted andtheir control is investigated. Based on asymmetric cut perturba-tion, a dual-mode ring resonator filter with quasi-elliptic pass-band is demonstrated. In Section III, a reduced-size circuit isimplemented by routing half of the ring trace inside the otherpart. A CPW high-impedance section, incorporating with twoconducting vias, is implemented to play the role of the crossoveras well as the inductive perturbation, achieving circuit miniatur-ization. Section IV draws a conclusion.

II. ASYMMETRIC DUAL-MODE RING RESONATOR AND FILTER

A. Asymmetric Perturbation

Fig. 1(a) shows the traditional dual-mode ring resonatorincorporated with a capacitive perturbation in the symmetricplane, capable of generating a quasi-elliptic function response[2]. The structure in Fig. 1(b) shows no sharp selectivity in thetransition bands [14]. In Figs. 1(c) and 1(d), the perturbationsare placed in the axis perpendicular to the symmetric axis.These two circuits are simply those in Figs. 1(a) and 1(b)rotated by 90 , while keeping the I/O ports fixed. Let thespatial separation between ports 1 and 2 be , and the tracesbetween the center perturbation and port 1 and port 2 beand , respectively. It is known thatat the center frequency, . Obviously, each resonator in Fig. 1possesses two degenerate modes, and their resonance frequen-cies and are split up by the perturbation, denoted bycharacteristic impedance and electrical length .With weak coupling excitations, the transmission responses

of Fig. 1(c) and (d) are shown in Fig. 2(a) and (b), respectively.The software package IE3D [20] is employed for the simulation.At center frequency GHz, , and the pertur-bation is allocated at , , , and . For the ca-

Fig. 2. Transmission responses of Fig. 1(c) and (d) with , ,when , , , . (a) Capacitive perturbation,. (b) Inductive perturbation, .

pacitive perturbation in Fig. 2(a), the response foris the result of the traditional ring configuration in Fig. 1(a).When the perturbation is placed at , there is no sharpskirt response near the center frequency [14]. Note that when

and , the two peaks have very different magni-tudes, and the bandpass response will be difficult to synthesize,due to the destruction of the zero. For the inductive perturba-tion in Fig. 2(b), on the other hand, the response suitable forsynthesis of a quasi-elliptic passband is . When theinductive segment is allocated at , there is no trans-mission dip.

B. Spacing Between Input and Output Ports

Figs. 3 and 4 investigate the behavior of the zeros when theI/O separation and , respectively. In Fig. 3, theperturbation is placed at , , , and . Com-pared with the results in Fig. 2, both and shift to lowerfrequencies. In Fig. 3(a), when , both of them arein the lower stopband, showing a flexible way of zero control.When , is between the two transmission polesso it is not suitable for passband synthesis. When ,the two zeros are on the different sides of the passband, and thiscircuit is more suitable for passband synthesis than that with

since of the latter is very close to . For the in-


Fig. 3. Transmission responses with , andwhen , , , and . (a) Capacitive perturbation in Fig. 1(c),

. (b) Inductive perturbation in Fig. 1(d), .

ductive perturbations in Fig. 3(b), the responses for ,, , and are similar to those with , ,, and , respectively, in Fig. 3(a).In Fig. 4, the I/O separation is , and the perturbation

is placed at , , , and . It is interesting thatonly one transmission zero can be observed. For the capacitiveperturbation in Fig. 4(a), the zero is on the upper and lowersides of the passband when and , respectively,while in Fig. 4(b) with inductive perturbation, the zeros on theupper and lower sides are resulted from and ,respectively. For both perturbation types, the responses showthat a flat passband supported by and will be difficult tobuild up by using and .

C. Transmission Poles and Zeros

Theoretically, the transmission poles of the perturbed ringsin Fig. 1 are independent of the excitations. Thus, the resonancefrequencies of the rings in Fig. 1(c) and (d) will be the same asthose in Fig. 1(a) and (b), respectively. It can be shown that theresonance conditions are

(odd mode) (1)

(even mode) (2)

where and . Theperturbation is of capacitive and inductive type whenand , respectively. In comparison of (2) and (1), if

(3)

Fig. 4. Transmission responses with , , andwhen , , , . (a) Capacitive perturbation in Fig. 1(c),

. (b) Inductive perturbation in Fig. 1(d), .

holds, then the even- and odd-mode frequencies of the capaci-tive perturbation are the same as the odd- and even-mode fre-quencies of the inductive cut.The zero frequencies of the ring resonator filter will depend

on the positions of the input and output. For both Fig. 1(c) and(d), the transmission zeros can be obtained by enforcing the totalforward transmission admittance of the two signal paths shownin Fig. 5 to zero [2]

(4)

It can be rewritten as

(5a)

(5b)

(5c)

(5d)

The term is resulted from the leading two terms in (4) and rep-resents the sum of two admittances when no perturbation exists.When and , the term indicates asecond-order zero at . The effectiveness of the perturbation onsplit-up or shift of the zeros relies on the second term of (5a).A larger will lead to a larger split-up. It is known that isrelatively small, and . For example, , than

and . For the traditional ca-pacitively perturbed ring in Fig. 1(a), : 1) the


Fig. 5. Two-signal path model for analysis of transmission zeros.

term in (5) can be neglected 2) when frequencies moves notso far away from , and its magnitude decreases. Thus,must be negative to assure the existence of zeros. This reflectsthe fact that if a cut is placed here, there will be no zero since

. When frequency is decreased or increased to the fre-quencies where is satisfied, transmission zeros occur.On the other hand, for the proposed ring with inductive cut inFig. 1(d), and , and .Thus, a patch at this corner will not cause any zero since theterm in (5d) reverses the sign of . Instead, an inductive cut with

is required for the split-up of the two zeros from.Fig. 6(a) plots variations of the transmission zeros and

versus from 90 to 270 , based on , ,, and at GHz. It is

noted that when (4) is used for solving the zeros, all electricallengths are linear functions of frequency. When ,two transmission zeros can be found whenand , but there is no zero otherwise. Thedistance between the two zeros varies when is changed, andthe maximal distance occurs around and . Whenis increased to , i.e., more perturbation is employed,

the zeros move farther away from the center frequency. Whena capacitive patch is applied, e.g., , the two trans-mission zeros can be found only when and

. Again, when and , i.e., theamount of perturbation is increased; the distance between thetwo zeros is increased.Fig. 6(b) and (c) investigates the changes of and

versus for and , respectively. In Fig. 6(b),and variations have smaller ranges than those in Fig. 6(a),and some are close to the center passband. In Fig. 6(c), onlyone zero can be observed, for both capacitive and induc-tive perturbations. A better understanding for existence of zero,one, or two zeros can be referred to the illustrative descriptionfollowing (5d).Fig. 7 shows the variations and with respect to the

continuous changes of from 10 to 150 by solving (2) forthe inductive perturbation with . It is interesting thatboth and have a turning point at around . Whenis decreased from the turning point, decreases while

keeps almost flat. Alternatively, when is above the turningpoint, shows only small changes, but increases. Thetransmission poles can also be plotted along with these zeros[21].

D. Effectiveness of Perturbation Size

As shown in Fig. 6, the frequencies of the two zeros also de-pend on the size of the perturbation. Let and

Fig. 6. Transmission zero versus . , . (a) .(b) . (c) .

Fig. 7. Transmission zero versus continuous changes of for inductive per-turbation. , , .

at , i.e., the structure shown in Fig. 1(d). The peak frequenciesand determine the bandwidth and can be calculated by

using the formula in [2] by changing the type of perturbation.Fig. 8 compares , , , and obtained by the IE3Dwith those obtained by the transmission-line theory for variousnotch sizes. In Fig. 8(a), when is increased, meaning thenotch has less inductive, the resonances and zeros get closer tothe center frequency . In Fig. 8(b), when the length of thehigh-impedance section is increased, and move to lower


Fig. 8. Variations of transmission zeros and resonant modes versus sizes of thecut. (a) mm mm, , mm,

. (b) mm mm, ,mm . , . Dashed lines: full-wave simulation, solid

lines: transmission-line theory.

frequencies, while and shift to higher frequencies, indi-cating the ring resonator filter will have an increasing bandwidthand a wider space between the two zeros. The results based onthe theoretical formulation have goodmatch with those obtainedby the full wave simulation. The maximal relative deviation is0.72%, occurring at the curve.

E. Dual-Mode Ring Resonator Bandpass Filter

The design procedure for the dual-mode ring resonator filteris: 1) determine circuit bandwidth and number and positions oftransmission zeros from the plots in Figs. 2–4; 2) choose typeof perturbation type, i.e., patch or cut, and its position alongthe ring peripheral; and 3) choose the excitation angle and theline-to-ring coupling [22] for I/O matching.The dual-mode ring resonator with asymmetric inductive per-

turbation can be devised to synthesize a quasi-elliptic functionbandpass filter. The circuit layout is shown in Fig. 9(a), andthe photograph of the measured circuit in Fig. 9(b). The designparameters include , , ,

, , and . Fig. 9(c) shows thesimulation and measurement results of Fig. 9(a). It is designedat center frequency GHz with a fractional bandwidth

. The substrate has a dielectric constantand thickness mm. At , the measured insertionloss is 2.37 dB, and the return loss is better than 20 dB. The twotransmission zeros are at 2.23 and 2.72 GHz. Simulation andmeasurement results show good agreement.

Fig. 9. (a) Layout of the dual-mode ring resonator filter. (b) Photograph ofthe measured circuit. (c) Simulation and measurement results of the dual-modering resonator filter. mm, mm, mm,

mm, mm, mm, mm,mm, mm, mm. Substrate: ,

thickness mm.

Fig. 10. Double-ring configuration.

III. CIRCUIT MINIATURIZATION

To obtain a compact circuit area, the ring can be folded to adouble-ring or spiral configuration, as shown in Fig. 10, wherea 3-D structure will be required for connecting points A andB of the inner and outer rings, of which the electrical lengthsare and , respectively. The 3-D implementation can be ashort bonding wire or a short high-impedance section incor-porating with two via-holes. Both of them are equivalent toan inductor. Here, for ease of fabrication, the later is chosen.The high-impedance section is made on the ground plane ofthe microstrip, forming a CPW segment. Such a CPW sectionplays a different role from that in [11] where a quarter-wave


Fig. 11. (a) Layout of miniaturized dual-mode ring resonator filter. (b) Simu-lated and measured results. Photograph of the measured circuit. (c) Top view.(d) Bottom view. mm, mm, mm,

mm, mm, mm, mm,mm, mm, mm, mm,mm. CPW section: , , .

Substrate: , thickness mm.

microstrip-to-CPW broadside coupled line is used to replace atransmission-line section of 270 . For saving the implementa-tion effort, the line-to-ring structure is kept as close as possibleto that in Fig. 9(a). Thus, a certain space between the two loopsis used for preventing coupling between the adjacent traces.Fig. 11(a) shows the top view of the circuit layout. The top metalin black is used for the microstrip, the bottommetal in gray is for

the ground plane, and the CPW high-impedance segment. In de-sign, the coupling between the microstrip and the CPW sectionis neglected since not only the overlapped area is small, but alsoboth signal traces are nearly perpendicular to each other. Thecoupling between parallel microstrip ring traces may change thetotal coupling coefficient, and hence, the filter bandwidth. Toeliminate the unwanted coupling, our simulation data suggestthat the distance between the parallel traces should keep at leastthree times the line width. The radius of the two via-holes is .As a result, the circuit has a size of , and itsarea is about 39.8% of that of a conventional square ring res-onator.Fig. 11(b) compares measurement results with the simula-

tion. The circuit parameters include , ,, , , ,

and GHz. The equivalent inductance is1.83 nH. The measured insertion loss at is 2.52 dB, the frac-tional bandwidth and the return loss is better than15 dB. Two transmission zeros are at 2.235 and 2.74 GHz, re-spectively. Good agreement between simulation and measure-ment results can be observed. Fig. 11(c) and (d) shows the pho-tographs of the top and bottom sizes of the measured circuit.

IV. CONCLUSION

Characteristics of the transmission zeros of dual-mode ringresonator filters with asymmetric and symmetric inductive andcapacitive perturbations are investigated for bandpass filterdesign. Quasi-elliptic function passband can be obtained byplacing the inductive perturbation at a position of 90 awayfrom the end of the symmetric axis, where is the capacitivepatch location of a traditional dual-mode ring resonator filter.When the product of the characteristic impedances of the patchand cut sections equals square of the characteristic impedanceof the uniform trace, the even- and odd-mode frequencies areswitched when the perturbation type is exchanged. Variationsof the transmission zeros versus I/O port separation andperturbation position are investigated, and design curvesplotted. When , the two zeros shift down to lower fre-quencies for both capacitive and inductive perturbations. Thedual-mode ring with inductive cut is folded to a double-ringconfiguration and the inductive perturbation is implemented bya CPW high-impedance section on the ground plane, leadingto the normalized circuit area reduced to 39.8%. Measurementdata show good agreement with simulation results.

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Tsu-Wei Lin was born in Taichung, Taiwan. He re-ceived the B.S. degree from National Chung ChengUniversity, Taiwan, in 2009, the M.S. degree fromNational Chiao Tung University, Taiwan, in 2010,and is currently working toward the Ph.D. degreefrom the Institute of Communications Engineering,National Chiao Tung University, Hsinchu, Taiwan.His research interests include design of planar fil-

ters and associated RF modules for microwave appli-cations.

Jen-Tsai Kuo (S’88–M’92–SM’04) received thePh.D. degree from the Institute of Electronics,National Chiao Tung University (NCTU), Hsinchu,Taiwan, in 1992.From 1984 to 2010, he was with the Department

of Communication Engineering, NCTU. From 1995to 1996, he was a Visiting Scholar with the ElectricalEngineering Department, University of Californiaat Los Angeles (UCLA). He is currently a Professorwith the Department of Electronic Engineering,Chang Gung University, Taoyuan, Taiwan. His

research interests include analysis and design of microwave integrated circuitsand numerical techniques in electromagnetics.Dr. Kuo is a member of the IEEE MTT-8 Subcommittee (Filters and Passive

Components). He is an Editorial Board member for the IEEE TRANSACTIONSON MICROWAVE THEORY AND TECHNIQUES and the IEEE MICROWAVE ANDWIRELESS COMPONENTS LETTERS. He was an associate editor for the IEEETRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES (2008–2010). Hewas a recipient of the Best Paper Award presented at the 2002National Telecom-munication Conference, Taiwan, the 2007 Asia–Pacific Microwave Conference(APMC) Prize, Bangkok, Thailand, and the 2008 APMC Prize, Hong Kong. Hewas the recipient of the 2006 Taiwan Citation Laureate presented by ThomsonScientific and the 2007Distinguished ResearchAward presented by the NationalScience Council, Taiwan.

Shyh-Jong Chung (M’92–SM’06) was born inTaipei, Taiwan. He received the B.S.E.E. and Ph.D.degrees from National Taiwan University, Taipei,Taiwan, in 1984 and 1988, respectively.Since 1988, he has been with the Department of

Communication Engineering and the Departmentof Electrical Engineering, National Chiao TungUniversity (NCTU), Hsinchu, Taiwan, where heis currently a Professor. From 2009 to 2011, hewas the Director of the Institute of CommunicationEngineering, NCTU. From September 1995 to

August 1996, he was a Visiting Scholar with the Department of ElectricalEngineering, Texas A&M University, College Station. His areas of interestinclude the design and applications of active and passive planar antennas,low-temperature co-fired ceramic (LTCC)-based RF components and modules,packaging effects of microwave circuits, vehicle collision warning radars, andcommunications in intelligent transportation systems (ITSs).Dr. Chung was the treasurer of IEEE Taipei Section (2001–2003) and the

chairman of IEEE Microwave Theory and Techniques Society (IEEE MTT-S)Taipei Chapter (2005–2007). He was the recipient of the Outstanding Engi-neering Professor Award of the Chinese Institute of Engineers (2012), the Out-standing Electrical Engineering Professor Award of the Chinese Institute ofElectrical Engineering (2006), and the Teaching Excellence Award of NationalChiao Tung University (2005).

2808 ieee transactions on microwave … et al.: dual-mode ring resonator bandpass filter 2811 fig....

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