1x rf planning guide - nokia

1X RF Planning Guide

68P09316A39-A

Approval Date January 2013

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68P09316A39-A

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JAN 2013 CDMA/CDMA2000 1X RF Planning Guide ii

General Information

Purpose

Nokia Siemens Networks documents provide the information to operate, install, and maintain Nokia Siemens Networks equipment. It is recommended that all personnel engaged in such activities be properly trained by Nokia Siemens Networks.

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iii CDMA/CDMA2000 1X RF Planning Guide JAN 2013

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JAN 2013 CDMA/CDMA2000 1X RF Planning Guide iv

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v CDMA/CDMA2000 1X RF Planning Guide JAN 2013

JAN 2013 CDMA/CDMA2000 1X RF Planning Guide vi

vii CDMA/CDMA2000 1X RF Planning Guide JAN 2013

Table of Contents

CDMA/CDMA2000 1X RF Planning Guide

1 How to Use This Guide

1.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 - 3

1.2 Quick Guide to Contents of Each Section . . . . . . . . . . . . . . . . . . . . . . . 1 - 4

2 Basic CDMA Spectrum Planning

2.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 3

2.2 North American and International Frequency Blocks . . . . . . . . . . . . . . . . . . . . .2 - 3

2.3 CDMA Channel Spacing - General . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 52.3.1 Minimum Spacing Between CDMA Carriers . . . . . . . . . . . . . . . . . . . . . .2 - 52.3.2 Maximum Spacing Between CDMA Carriers. . . . . . . . . . . . . . . . . . . . . .2 - 82.3.3 Multiple Market Spectrum Planning Considerations . . . . . . . . . . . . . . .2 - 112.3.4 Multiple Carrier Overlay Guidelines. . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 11

2.3.4.1 IS-2000 1X New Carrier Overlay . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 142.3.4.2 IS-2000 1X Shared Carrier Overlay . . . . . . . . . . . . . . . . . . . . . . . . 2 - 15

2.3.5 Guard Band Considerations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 152.3.5.1 AMPS Guard Band Recommendation . . . . . . . . . . . . . . . . . . . . . . 2 - 172.3.5.2 2nd CDMA Carrier with AMPS Guard Band. . . . . . . . . . . . . . . . . 2 - 172.3.5.3 Greater Than Two CDMA Carriers with AMPS Guard Band . . . . 2 - 18

2.4 Channel Spacing and Designation - 800 MHz . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 192.4.1 Segregated Spectrum. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 21

2.5 Channel Spacing and Designation - 1900 MHz . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 24

2.6 Dual-Mode vs. Dual-Band. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 26

2.7 Spectrum Clearing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 26

2.8 Background Noise Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 272.8.1 Suggested Measurement Method . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 28

2.8.1.1 Test System Functional Description . . . . . . . . . . . . . . . . . . . . . . . . 2 - 282.8.1.2 Test System Calibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 29

2.8.2 Test Procedures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 302.8.3 Data Analysis. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2 - 31

iCDMA/CDMA2000 1X RF Planning GuideJAN 2013

Table of Contents - continued

2.9 CDMA/LTE Sharing of XMIs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 31

2.10 References. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 33

3 CDMA Capacity

3.1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 5

3.2 Reverse Link Pole Capacity Estimation . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 53.2.1 Data Rates. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 113.2.2 Median Eb/(No+Io) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 113.2.3 Voice or Data Activity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 123.2.4 Cell Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 133.2.5 Sectorization Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 153.2.6 Power Control. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 16

3.3 Reverse Link Soft Blocking Capacity Estimation . . . . . . . . . . . . . . . . . 3 - 183.3.1 Conventional Blocking Analysis . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 183.3.2 CDMA Soft Blocking Capacity Estimation . . . . . . . . . . . . . . . . . 3 - 18

3.3.2.1 Assumptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 193.3.2.2 Theoretical Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 193.3.2.3 Single Cell Case . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 223.3.2.4 Multiple Cell System. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 23

3.4 Reverse Link Noise Rise Capacity Estimation . . . . . . . . . . . . . . . . . . . . 3 - 323.4.1 Reverse Link Noise Rise Capacity Limit . . . . . . . . . . . . . . . . . . . 3 - 323.4.2 Reverse Noise Rise Capacity Estimation . . . . . . . . . . . . . . . . . . . 3 - 333.4.3 Probability Factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 353.4.4 Reverse Link Noise Rise Capacity Estimation Examples . . . . . . 3 - 37

3.4.4.1 Example #1: Voice Only . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 373.4.4.2 Example #2: Voice and Data Users . . . . . . . . . . . . . . . . . . . . 3 - 38

3.4.5 Reverse Link Noise Rise Capacity Estimates for IS-2000 1X. . . 3 - 413.4.5.1 Noise Rise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 413.4.5.2 F-factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 423.4.5.3 Average Eb/No . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 433.4.5.4 Eb/No Standard Deviation . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 433.4.5.5 Processing Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 443.4.5.6 Activity Factor. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 443.4.5.7 Traffic Mix. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 453.4.5.8 Throughput Capacity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 453.4.5.9 IS-2000 1X Reverse Noise Rise Capacity Analysis Results . 3 - 46

3.5 Forward Link Pole Capacity Estimation . . . . . . . . . . . . . . . . . . . . . . . . 3 - 523.5.1 Forward Link Load Factor Estimation . . . . . . . . . . . . . . . . . . . . . 3 - 523.5.2 Forward Link Pole Capacity Estimation . . . . . . . . . . . . . . . . . . . 3 - 53

3.6 Forward Link Fractional Power Capacity Estimation . . . . . . . . . . . . . 3 - 54

3.7 Forward Link Noise Rise Capacity Estimation . . . . . . . . . . . . . . . . . . . 3 - 57

ii CDMA/CDMA2000 1X RF Planning Guide JAN 2013


3.7.1 Forward Link Noise Rise Capacity Limit . . . . . . . . . . . . . . . . . . 3 - 583.7.2 Forward Noise Rise Capacity Estimation. . . . . . . . . . . . . . . . . . . 3 - 593.7.3 Forward Link Noise Rise Capacity Estimation Examples . . . . . . 3 - 60

3.7.3.1 Example #1: Voice Only. . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 613.7.3.2 Example #2: Voice and Data Users . . . . . . . . . . . . . . . . . . . 3 - 62

3.7.4 Forward Link Noise Rise Capacity Estimates for IS-2000 1X . . 3 - 653.7.4.1 Noise Rise. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 653.7.4.2 I-factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 663.7.4.3 Average Eb/No . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 663.7.4.4 Eb/No Standard Deviation . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 673.7.4.5 Processing Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 673.7.4.6 Activity Factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 683.7.4.7 Orthogonality Factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 693.7.4.8 Traffic Mix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 693.7.4.9 Throughput Capacity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 693.7.4.10 IS-2000 1X Forward Noise Rise Capacity Analysis Results 3 - 70

3.8 Forward vs. Reverse Link Capacity Comparison . . . . . . . . . . . . . . . . . 3 - 75

3.9 EIA/TIA Specifications and RF Air Interface Limitations. . . . . . . . . . 3 - 793.9.1 IS-95 Forward Channel Structure. . . . . . . . . . . . . . . . . . . . . . . . . 3 - 793.9.2 IS-95 Reverse Channel Structure . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 803.9.3 IS-2000 1X Forward Channel Structure. . . . . . . . . . . . . . . . . . . . 3 - 81

3.9.3.1 IS-2000 Forward Channels (Nokia Siemens Networks . . . . Implementation) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 823.9.3.2 IS-2000 Forward Link Radio Configurations. . . . . . . . . . . . 3 - 853.9.3.3 IS-2000 Walsh Code Allocation . . . . . . . . . . . . . . . . . . . . . . 3 - 87

3.9.4 IS-2000 Reverse Channel Structure . . . . . . . . . . . . . . . . . . . . . . . 3 - 903.9.4.1 IS-2000 Reverse Channels (Nokia Siemens Networks Implementation) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 903.9.4.2 IS-2000 Reverse Link Radio Configurations . . . . . . . . . . . . 3 - 91

3.10 Handoffs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 933.10.1 Soft Handoff . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 933.10.2 Inter-CBSC Soft Handoff . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 943.10.3 Hard Handoffs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 94

3.10.3.1 Anchor Handoff . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 943.10.3.2 IS-95 to IS-2000 Hand-up . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 943.10.3.3 IS-2000 to IS-95 Hand-down . . . . . . . . . . . . . . . . . . . . . . . . 3 - 953.10.3.4 Packet Data Handoffs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 953.10.3.5 Inter-Carrier Hand-across . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 95

3.11 Budgetary Estimate of Sites for Capacity (Voice Only) . . . . . . . . . . . . 3 - 953.11.1 Required Parameters for Initial System Design . . . . . . . . . . . . . . 3 - 96

3.11.1.1 Busy Hour Call Attempts and Completions . . . . . . . . . . . . . 3 - 963.11.1.2 Average Holding Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 963.11.1.3 Erlangs per Subscriber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 96

3.12 IS-95 and IS-2000 Simulations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 101

3.13 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 103

iiiCDMA/CDMA2000 1X RF Planning GuideJAN 2013


4 Link Budgets and Coverage

4.1 Introduction 4 - 3

4.2 Radio Frequency Link Budget 4 - 44.2.1 Propagation Related Parameters 4 - 6

4.2.1.1 Building Loss . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 64.2.1.2 Vehicle Loss . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 94.2.1.3 Body Loss . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 94.2.1.4 Ambient Noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 94.2.1.5 RF Feeder Losses. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 94.2.1.6 Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 12

4.2.2 CDMA Specific Parameters 4 - 144.2.2.1 Interference Noise Rise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 144.2.2.2 Soft Handoff Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 184.2.2.3 Eb/No . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 19

4.2.3 Product Specific Parameters 4 - 204.2.3.1 Product Transmit Power. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 204.2.3.2 Product Receiver Sensitivity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 23

4.2.4 Reliability (Shadow Fade Margin) 4 - 294.2.5 Example Reverse (Uplink - Subscriber to Base) Link Budget 4 - 364.2.6 RF Link Budget Summary 4 - 40

4.3 Propagation Models 4 - 414.3.1 Free Space Propagation Model 4 - 414.3.2 Hata Propagation Model 4 - 434.3.3 COST-231-Hata Propagation Model 4 - 444.3.4 Additional Propagation Models 4 - 45

4.4 Forward Link Coverage 4 - 464.4.1 BTS Equipment Capabilities 4 - 47

4.4.1.1 4812T-MC BTS Minimum and Maximum Pilot Power . . . . . . . . 4 - 524.4.1.2 Max Pilot Power for M810 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 534.4.1.3 UBS Max Pilot Power for Mix of 1X and DO Carriers (does not apply to M810) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 544.4.1.4 1.9 GHz UBS Sector-Carrier Power Levels . . . . . . . . . . . . . . . . . 4 - 554.4.1.5 Power Out Differences Between UBS-Macro and 4812T/4812T-MC BTSs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 56

4.4.2 CDMA Signal Power Distribution Characteristics and PA Sizing 4 - 564.4.3 General Power Relationships 4 - 574.4.4 Design Guidelines 4 - 58

4.4.4.1 Comparison to Average Rated Power . . . . . . . . . . . . . . . . . . . . . . 4 - 594.4.4.2 Comparison to High Power Alarm Rating. . . . . . . . . . . . . . . . . . . 4 - 594.4.4.3 Comparison to Walsh Code Limit . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 60

4.4.5 General Power Requirements 4 - 604.4.5.1 Minimum ARP Based on LT-AVG Estimate . . . . . . . . . . . . . . . . . 4 - 604.4.5.2 Minimum HPA Based on VST-AVG Estimate . . . . . . . . . . . . . . . 4 - 614.4.5.3 Exceeding the High Power Alarm Rating . . . . . . . . . . . . . . . . . . . 4 - 624.4.5.4 Carrier Load Management Overview . . . . . . . . . . . . . . . . . . . . . . 4 - 62

4.4.6 Power Allocation in Mixed Mode 1X and DO Systems 4 - 64

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4.4.7 Government Regulations 4 - 644.4.7.1 Power Amplifier Operational Measurements (FR9235) . . . . . . . . 4 - 64

4.5 CDMA Repeaters 4 - 664.5.1 CDMA Repeater Design Considerations 4 - 66

4.5.1.1 Coverage Impact . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 664.5.1.2 Cascaded Noise Figure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 694.5.1.3 Interference and Capacity Issues . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 744.5.1.4 Filtering Issues. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 74

4.5.2 CDMA Repeater Installation Considerations 4 - 754.5.2.1 Antenna Isolation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 754.5.2.2 Repeater Antenna Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 774.5.2.3 Repeater Gain Settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 78

4.5.3 CDMA Repeater Optimization Considerations 4 - 804.5.3.1 Timing Impacts . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 804.5.3.2 Optimization Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 82

4.5.4 CDMA Repeater Maintenance Considerations 4 - 824.5.4.1 Future Expansion Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 824.5.4.2 Environmental Changes. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 844.5.4.3 Operations and Maintenance Considerations . . . . . . . . . . . . . . . . . 4 - 84

4.6 Extended Range Cells 4 - 844.6.1 Extended Range Cell RF Planning and Design 4 - 85

4.6.1.1 Tower Top Amplifiers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 854.6.1.2 Reverse Link Budget Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 864.6.1.3 Forward Link Budget Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 884.6.1.4 Extended Range Cell Site Design Limitations . . . . . . . . . . . . . . . . 4 - 904.6.1.5 Site Selection Criteria for Extended Range Cells . . . . . . . . . . . . . . 4 - 99

4.6.2 Extended Range Cell Optimization Considerations 4 - 1014.6.2.1 PN Offset Planning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 1014.6.2.2 Parameter Optimization Impacts . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 102

4.7 Theoretical vs. Simulator 4 - 103

4.8 References 4 - 105

5 PN Offset Planning and Search Windows

5.1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 3

5.2 Number of Pilot Offsets per CDMA Frequency. . . . . . . . . . . . . . . . . . . 5 - 3

5.3 PN Offset Planning - General . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 35.3.1 Consequences and Sources of Offset Interference. . . . . . . . . . . . . . .5 - 35.3.2 PN Offset Planning - Parameters and Terms . . . . . . . . . . . . . . . . . . .5 - 55.3.3 Converting Between Chips and Time or Distance. . . . . . . . . . . . . . .5 - 85.3.4 Search Windows and Geography. . . . . . . . . . . . . . . . . . . . . . . . . . . .5 - 95.3.5 Search Windows and Scan Intervals . . . . . . . . . . . . . . . . . . . . . . . . .5 - 115.3.6 Calculating of Traffic Channel Acquisition Size TchAcqWinSz. . . .5 - 12

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5.4 PN Offset Planning - Solutions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 195.4.1 Mitigating Adjacent Offset Interference - General . . . . . . . . . . . . . .5 - 20

5.4.1.1 Adjacent Offset Interference Protection Based on Timing . . 5 - 205.4.1.2 Adjacent Offset Interference Protection Based on Signal Strength 5 -

215.4.2 Protection Against Co-Offset Interference . . . . . . . . . . . . . . . . . . . .5 - 235.4.3 Incorrect Identification of an Offset by the Base Station. . . . . . . . . .5 - 265.4.4 PILOT_INC and the Scan Rate of Remaining Set Pilots . . . . . . . . .5 - 275.4.5 Summary of Guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5 - 285.4.6 Guidelines for Assigning Offsets. . . . . . . . . . . . . . . . . . . . . . . . . . . .5 - 305.4.7 Guidelines for Planning Inter-CBSC and Intra-CBSC multiple PILOT_INC

Boundaries and Transition Zones. . . . . . . . . . . . . . . . . . . . . . . . .5 - 33

5.5 Reuse Patterns. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 36

5.6 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 37

6 RF Antenna Systems

6.1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 3

6.2 CDMA Cell Site Antenna Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 36.2.1 Antenna Type . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 36.2.2 Antenna Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 46.2.3 Antenna Beamwidth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 66.2.4 Voltage Standing Wave Ratio . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 66.2.5 Return Loss. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 66.2.6 Power Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 76.2.7 Front to Back Ratio . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 76.2.8 Side Lobes & Back Lobes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 76.2.9 Antenna Downtilting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 86.2.10 Antenna Height. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 8

6.3 CDMA Antenna Placement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 96.3.1 Antenna Isolation Considerations. . . . . . . . . . . . . . . . . . . . . . . . . 6 - 9

6.3.1.1 CDMA/AMPS Transmit/Receive Antenna Isolation Requirements 6 - 11

6.3.1.2 Measuring Port-to-Port Antenna Isolation . . . . . . . . . . . . . . 6 - 136.3.1.3 Reducing the Required Antenna Isolation. . . . . . . . . . . . . . . 6 - 136.3.1.4 Typical Antenna Isolation . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 146.3.1.5 CDMA Antenna Placement . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 14

6.3.2 Antenna Diversity (Spacial) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 166.3.2.1 Horizontal Antenna Diversity and Recommended Separation 6 - 166.3.2.2 Vertical Antenna Diversity . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 17

6.4 CDMA Antenna Sharing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 176.4.1 Multiple Frame Antenna Sharing with 800 MHz BTS Products . 6 - 176.4.2 Multiple Carrier Cavity Combining With 1900 MHz BTS Products 6 - 19

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6.4.2.1 Output Power With Combining . . . . . . . . . . . . . . . . . . . . . . 6 - 196.4.2.2 Type of Combining. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 196.4.2.3 Multiple Carrier Scenarios . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 20

6.4.3 Duplexing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 216.4.3.1 Pre-Engineered Kits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 226.4.3.2 Duplexers and Intermodulation . . . . . . . . . . . . . . . . . . . . . . 6 - 226.4.3.3 Proper Installation and Maintenance of Duplexed Antennas 6 - 25

6.5 CDMA Antenna Sharing With Other Technologies . . . . . . . . . . . . . . . 6 - 296.5.1 CDMA/Analog Shared Facilities . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 29

6.5.1.1 Common Transmit Antenna . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 306.5.1.2 Common Receive Antenna(s). . . . . . . . . . . . . . . . . . . . . . . . 6 - 30

6.5.2 Duplexed AMPS/CDMA Antennas . . . . . . . . . . . . . . . . . . . . . . . 6 - 31

6.6 GPS Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 33

6.7 Ancillary Antenna System Components . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 336.7.1 Directional Couplers. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 336.7.2 Surge (Lightning) Protectors . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 336.7.3 Transmission Line . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 34

6.7.3.1 RF Performance of Transmission Lines . . . . . . . . . . . . . . . . 6 - 346.7.3.2 Physical Characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 356.7.3.3 Choice of Transmission Line . . . . . . . . . . . . . . . . . . . . . . . . 6 - 35

6.7.4 Transition Feeder Cables (Jumper Cables). . . . . . . . . . . . . . . . . . 6 - 36

6.8 RF Diagnostic System. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 36

7 RF Antenna Systems - Advanced Topics

7.1 Dual Polarized Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 37.1.1 Fundamental Concepts . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 3

7.1.1.1 Dual Polarization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 37.1.1.2 Diversity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 47.1.1.3 Diversity Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 57.1.1.4 Cross-Correlation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 7

7.1.2 Isolation Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 117.1.3 Performance Impacts - Industry and Motorola Findings . . . . . . . 7 - 137.1.4 Antenna Selection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 15

7.1.4.1 Dual Polarized Antennas versus Singularly Polarized Antennas 7 - 157.1.4.2 Antenna Selection Criteria . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 16

7.1.5 Transmission at 45° . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 167.1.6 Incorporation of Dual Polarized Antennas into a Link Budget . . 7 - 177.1.7 Dual Polarized Antenna Summary . . . . . . . . . . . . . . . . . . . . . . . . 7 - 18

7.2 In-Building Distributed Antenna Systems . . . . . . . . . . . . . . . . . . . . . . . 7 - 197.2.1 In-Building System Architecture Overview. . . . . . . . . . . . . . . . . 7 - 207.2.2 Coaxial Cable System Design Using A Link Budget. . . . . . . . . . 7 - 21

7.2.2.1 Design Procedure Flow Chart . . . . . . . . . . . . . . . . . . . . . . . 7 - 217.2.2.2 Gathering Building Information . . . . . . . . . . . . . . . . . . . . . . 7 - 227.2.2.3 Determining the Base Station Location . . . . . . . . . . . . . . . . 7 - 24

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7.2.2.4 Estimating the Antenna Placement within the Building . . . . 7 - 257.2.2.5 Selecting the Antenna Type: Omni vs. Directional . . . . . . . . 7 - 257.2.2.6 Choosing the Base Station Type . . . . . . . . . . . . . . . . . . . . . . 7 - 267.2.2.7 Choosing the Cable Topology: Splitters, Couplers, and Taps 7 - 267.2.2.8 Estimating Cable Lengths from the Base Station to the Antennas7 - 317.2.2.9 Selecting the Coaxial Cable Type . . . . . . . . . . . . . . . . . . . . . 7 - 317.2.2.10 Link Budgets For In-Building Design . . . . . . . . . . . . . . . . . . 7 - 337.2.2.11 Evaluating the First Pass and Iterating the Design . . . . . . . . 7 - 39

7.2.3 Active Coaxial Cable System Design. . . . . . . . . . . . . . . . . . . . . . 7 - 397.2.3.1 Downlink Amplifier Design Considerations . . . . . . . . . . . . . 7 - 407.2.3.2 Uplink Amplifier Design Considerations . . . . . . . . . . . . . . . 7 - 417.2.3.3 Optimizing Amplifier Placement. . . . . . . . . . . . . . . . . . . . . . 7 - 46

7.2.4 Fiber Optics for In-Building Systems. . . . . . . . . . . . . . . . . . . . . . 7 - 487.2.4.1 Fiber Optic Distribution System Architecture. . . . . . . . . . . . 7 - 487.2.4.2 When To Use Fiber Optics . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 487.2.4.3 Fiber Optic System Design . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 49

7.2.5 In-Building Antenna Systems Summary . . . . . . . . . . . . . . . . . . . 7 - 50

7.3 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 51

8 Synchronization of the CDMA System

8.1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 38.1.1 Synchronization Requirements. . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 3

8.2 The Global Positioning System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 48.2.1 Satellite Constellation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 48.2.2 GPS RF Carrier. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 5

8.3 Typical GPS Antenna Configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 58.3.1 Active GPS Antenna. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 6

8.3.1.1 Antenna Element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 68.3.1.2 Low Noise Amplifier (LNA) / Pre-selector Filter . . . . . . . . . 8 - 78.3.1.3 Overall GPS Antenna RF Requirements . . . . . . . . . . . . . . . . 8 - 7

8.3.2 Antenna Positioning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 88.3.2.1 Required Antenna Visibility . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 88.3.2.2 Antenna Placement Optimization . . . . . . . . . . . . . . . . . . . . . 8 - 98.3.2.3 Lightning Protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 108.3.2.4 Antenna Blockage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 118.3.2.5 RF Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 11

8.3.3 RF Cabling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 118.3.4 Lightning Arrestor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 128.3.5 Signal Splitter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 138.3.6 GPS Antenna System RF Requirements . . . . . . . . . . . . . . . . . . . 8 - 14

8.3.6.1 Antenna System Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 148.3.6.2 Antenna System Noise Figure . . . . . . . . . . . . . . . . . . . . . . . . 8 - 15

8.4 Remote GPS (RGPS) Receiver . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 178.4.1 SC24/48/72xx, SC480 Frame RGPS Operation. . . . . . . . . . . . . . 8 - 178.4.2 UBS / M810 Frame RGPS Operation . . . . . . . . . . . . . . . . . . . . . 8 - 19

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8.4.3 UBS / M810 Synchronization Sharing . . . . . . . . . . . . . . . . . . . . . 8 - 19

8.5 CSM / CSA GPS Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 208.5.1 CSM / CSA Configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 21

8.5.1.1 <CSMRefSrc1> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 218.5.1.2 <CSMRefSrc2> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 228.5.1.3 <BTSLatGps> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 238.5.1.4 <BTSLongGps> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 248.5.1.5 <BTSHeightGps> Parameter. . . . . . . . . . . . . . . . . . . . . . . . . 8 - 248.5.1.6 <LocAccuracy> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 258.5.1.7 <GPSAntDelay> Parameter. . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 258.5.1.8 <HeightMode> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 25

8.6 UBS GPS Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 268.6.1 UBS Configuration. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 26

8.6.1.1 <latitude> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 268.6.1.2 <longitude> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 268.6.1.3 <antheight> Parameter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 278.6.1.4 <locaccuracy> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 278.6.1.5 <gpsantdelay> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 27

8.7 Typical GPS Receiver Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 278.7.1 CSM / CSA GPS Receiver Status . . . . . . . . . . . . . . . . . . . . . . . . 8 - 288.7.2 UBS / M810 GPS Receiver Status . . . . . . . . . . . . . . . . . . . . . . . . 8 - 28

8.8 Cellsite GPS Failures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 298.8.1 Non-Synchronous BTS (Emergency) Operation . . . . . . . . . . . . . 8 - 29

8.9 Appendix A – GPS Antenna Kit Installation Instructions . . . . . . . . . . 8 - 30

9 Inter-System Interference (ISI)

9.1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 3

9.2 Cellular/PCS Inter-System Interference. . . . . . . . . . . . . . . . . . . . . . . . . 9 - 39.2.1 Intra-Band Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 4

9.2.1.1 AMPS Cells to CDMA Subscribers . . . . . . . . . . . . . . . . . . . 9 - 69.2.1.2 AMPS Subscribers to CDMA Cells . . . . . . . . . . . . . . . . . . . 9 - 99.2.1.3 CDMA Cells to AMPS Subscribers . . . . . . . . . . . . . . . . . . . 9 - 99.2.1.4 CDMA Subscribers to AMPS Cells . . . . . . . . . . . . . . . . . . . 9 - 9

9.2.2 Inter-Band Interference. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 109.2.2.1 Preventative Measures: BS-to-BS Interference. . . . . . . . . . . 9 - 139.2.2.2 Preventative Measures: Subscriber-to-Subscriber Interference 9 - 27

9.3 PCS and Microwave Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 289.3.1 PCS to Microwave Interference . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 28

9.3.1.1 Coordination Distances . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 299.3.1.2 Propagation Models. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 309.3.1.3 Power Aggregation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 349.3.1.4 Microwave Receiver Interference Criteria . . . . . . . . . . . . . . 9 - 359.3.1.5 PCS to Microwave Interference Summary . . . . . . . . . . . . . . 9 - 37

ix CDMA/CDMA2000 1X RF Planning Guide JAN 2013


9.3.2 Microwave to PCS Interference . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 389.3.2.1 General Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 389.3.2.2 Calculation of Nominal Noise Floor . . . . . . . . . . . . . . . . . . . 9 - 389.3.2.3 Calculation of Effective Interference Power . . . . . . . . . . . . . 9 - 399.3.2.4 Calculation of Effective Noise Figure . . . . . . . . . . . . . . . . . . 9 - 399.3.2.5 Microwave to PCS Interference Summary . . . . . . . . . . . . . . 9 - 40

9.4 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 40

APPENDICES:

I Terms and Acronyms

I.1 Terms and Acronyms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . I - 3

II Glossary

II.1 Glossary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . II - 3

III Watts to dBm Conversion Table

III.1 Watts to dBm Conversion Table . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . III - 3

IV Complementary Error Function Table

IV.1 Complementary Error Function Table . . . . . . . . . . . . . . . . . . . . . . . . . . IV - 3

x CDMA/CDMA2000 1X RF Planning Guide JAN 2013


NOTES

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xi CDMA/CDMA2000 1X RF Planning Guide JAN 2013

xii CDMA/CDMA2000 1X RF Planning Guide JAN 2013


List of Figures


Figure 1-1: Radio Sub-System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 - 3Figure 2-1: 3G Spectrum Allocations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 5Figure 2-2: Minimum Spacing Between 800 MHz CDMA Channels . . . . . . . 2 - 6Figure 2-3: Minimum Spacing Between 1900 MHz CDMA Channels . . . . . . 2 - 6Figure 2-4: Adjacent Channel Interference Reverse Rise Estimates . . . . . . . . 2 - 7Figure 2-5: Total Channel Numbers Available . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 9Figure 2-6: Assign Guard Band. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 9Figure 2-7: Assign 1st and Last CDMA Carries . . . . . . . . . . . . . . . . . . . . . . . . 2 - 10Figure 2-8: Equally Distribute Remaining CDMA Carriers . . . . . . . . . . . . . . . 2 - 10Figure 2-9: 1-to-1 Overlay Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 12Figure 2-10: Non 1-to-1 Overlay Examples (NOT Recommended). . . . . . . . . . 2 - 12Figure 2-11: Service Acquisition Issues Due To Uneven Carrier Coverage . . . 2 - 13Figure 2-12: New IS-2000 1X Carrier Deployment . . . . . . . . . . . . . . . . . . . . . . 2 - 14Figure 2-13: Second IS-2000 1X Carrier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 14Figure 2-14: IS-2000 1X Shared Carrier Overlay . . . . . . . . . . . . . . . . . . . . . . . . 2 - 15Figure 2-15: Calculation of Spectrum Required for a CDMA Carrier . . . . . . . . 2 - 17Figure 2-16: Calculation of Minimum Spectrum Required for Two CDMA Channels 2 -

17Figure 2-17: 2nd CDMA Carrier. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 18Figure 2-18: 3rd CDMA Carrier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 18Figure 2-19: AMPS Frequency Allocation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 21Figure 2-20: Segregated Spectrum . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 23Figure 2-21: Suggested CDMA Noise Floor Measurement System . . . . . . . . . . 2 - 29Figure 2-22: CDMA/LTE Sharing of XMIs - 2 CDMA carriers, one LTE carrier. .2 - 32Figure 2-23: CDMA/LTE Sharing of XMIs - 1 CDMA carrier, 2 DO carriers . . . .2 - 33Figure 3-1: Impact of Eb/(No+Io) on the Number of Users . . . . . . . . . . . . . . . 3 - 12Figure 3-2: Impact of Voice or Data Activity on the Number of Users . . . . . . 3 - 13Figure 3-3: Impact of Other Cell Interference on the Number of Users . . . . . . 3 - 14Figure 3-4: Impact of Sectorization Gain on the Number of Users (3 Sector) . 3 - 16Figure 3-5: Impact of Imperfect Power Control on the Number of Users . . . . 3 - 17Figure 3-6: Values of the Integral and with Various Path Loss Slope . . . . . . 3 - 26Figure 3-7: Probability of Blocking vs. Erlangs per CDMA Sector with Various Path

Loss Slope Values with Rate Set 1 Vocoder . . . . . . . . . . . . . . . . . 3 - 28Figure 3-8: Probability of Blocking vs. Erlangs per CDMA Sector with Various Power

Control Standard Deviations with Rate Set 1 Vocoder . . . . . . . . . 3 - 29Figure 3-9: Probability of Blocking vs. Erlangs per CDMA Sector with Various Path

Loss Slope Values with Rate Set 2 Vocoder . . . . . . . . . . . . . . . . . 3 - 30

xiiiCDMA/CDMA2000 1X RF Planning GuideJAN 2013

List of Figures - continued

Figure 3-10: Probability of Blocking vs. Erlangs per CDMA Sector with Various Power Control Standard Deviations with Rate Set 2 Vocoder . . . . . . . . .3 - 31

Figure 3-11: Rise versus Percent of Pole Capacity . . . . . . . . . . . . . . . . . . . . . . . 3 - 33Figure 3-12: Standard Normal Distribution. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 35Figure 3-13: Rise and Radius versus Loading Example . . . . . . . . . . . . . . . . . . . 3 - 36Figure 3-14: Reverse Link Rise vs. Throughput . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 49Figure 3-15: Reverse Link Rise vs. Erlangs for Different Data Rates . . . . . . . . 3 - 50Figure 3-16: Reverse Link Total Erlangs & Throughput vs. Data Activity Factor 3 - 51Figure 3-17: Forward Link Rise vs. Throughput. . . . . . . . . . . . . . . . . . . . . . . . . 3 - 72Figure 3-18: Forward Link Rise vs. Erlangs for Different Data Rates . . . . . . . . 3 - 73Figure 3-19: Forward Link Total Erlangs & Throughput vs. Data Activity Factor 3 - 74Figure 3-20: Forward and Reverse Link Rise vs. Throughput - 95% Probability Factor 3

- 75Figure 3-21: Forward and Reverse Link Rise vs. Erlangs for Different Data Rates 3 - 76Figure 3-22: Forward and Reverse Link Erlangs & Thruput vs. Data Activity Factor 3 -

77Figure 3-23: Alternate Forward and Reverse Link Erlangs & Thruput vs. Data Activity

Factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .3 - 78Figure 3-24: Example of IS-95 Forward CDMA Channels. . . . . . . . . . . . . . . . . 3 - 79Figure 3-25: Example of IS-95 Reverse CDMA Channels . . . . . . . . . . . . . . . . . 3 - 81Figure 3-26: Example of IS-2000 Forward CDMA Channels. . . . . . . . . . . . . . . 3 - 82Figure 3-27: QPCH to PCH Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 84Figure 3-28: IS-2000 Walsh Code Tree. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 88Figure 3-29: Walsh Code Allocation Tree. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 89Figure 3-30: Walsh Code Allocation Tree. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 89Figure 3-31: Example of IS-2000 Reverse CDMA Channels . . . . . . . . . . . . . . . 3 - 90Figure 3-32: Subscriber Distribution of Chicago Metropolitan Area . . . . . . . . . 3 - 97Figure 4-1: Percentage of Cells Based on dB Changes to the Link Budget . . . . . . 4 - 4Figure 4-2: RF Link Budget Gains & Losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 5Figure 4-3: In-Building Propagation Scenarios . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 6Figure 4-4: Preferred FWT Locations Without External Antennas. . . . . . . . . . . . . 4 - 8Figure 4-5: Typical Components in the RF Feeder Run . . . . . . . . . . . . . . . . . . . . 4 - 11Figure 4-6: Rise (dB) at the cell of interest versus X (% load) at the cell of interest . 4 -

18Figure 4-7: Example of Two Different Receive Path Configurations . . . . . . . . . 4 - 27Figure 4-8: Impact of Fade Margin on Reliability. . . . . . . . . . . . . . . . . . . . . . . . . 4 - 30Figure 4-9: Edge Reliability vs. Fade Margin . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 32Figure 4-10: Area Reliability vs. Fade Margin . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 33Figure 4-11: Area Reliability as a Function of Shadow Fade Margin. . . . . . . . . . . 4 - 35Figure 4-12: Edge Reliability as a Function of Shadow Fade Margin . . . . . . . . . . 4 - 36Figure 4-13: Impact of dB Trade-off to Number of Sites . . . . . . . . . . . . . . . . . . . . 4 - 41Figure 4-14: Typical Repeater Application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 63Figure 4-15: Repeater Range Analysis Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 64Figure 4-16: Alternate Repeater Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 65

xiv CDMA/CDMA2000 1X RF Planning Guide JAN 2013


Figure 4-17: Cabled Cascaded Noise Figure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 66Figure 4-18: Base Station & Repeater Diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 67Figure 4-19: Repeater Cascaded Noise Figure . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 67Figure 4-20: Multiple Repeater Cascaded Noise Figure . . . . . . . . . . . . . . . . . . . . . 4 - 69Figure 4-21: Alternate Repeater Antenna Configuration . . . . . . . . . . . . . . . . . . . . 4 - 72Figure 4-22: Horizontal Separation Using a Barrier . . . . . . . . . . . . . . . . . . . . . . . . 4 - 72Figure 4-23: Micro-wave Linked Repeater . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 73Figure 4-24: Fiber Linked Repeater . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 73Figure 4-25: Potential Range Reduction Due to Repeaters . . . . . . . . . . . . . . . . . . . 4 - 75Figure 4-26: Available PA Power vs. Pilot/TCH Power . . . . . . . . . . . . . . . . . . . . . 4 - 86Figure 4-27: Antenna Height Calculation Diagram. . . . . . . . . . . . . . . . . . . . . . . . . 4 - 87Figure 4-28: Horizon vs. Antenna Height . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 88Figure 4-29: Propagation Path Loss for Antenna Height of 1130m . . . . . . . . . . . . 4 - 89Figure 4-30: Free Space Path Loss versus Cell Radius . . . . . . . . . . . . . . . . . . . . . . 4 - 90Figure 4-31: Handoff Limitation Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 93Figure 4-32: Chips/Distance Scale for Handoff Limitation Example . . . . . . . . . . . 4 - 94Figure 4-33: SrchWinN Handoff Zone for Previous Example . . . . . . . . . . . . . . . . 4 - 94Figure 4-34: Extended Range Cell Hill/Mountain Top Application . . . . . . . . . . . . 4 - 95Figure 4-35: Extended Range Cell Marine/Coastal Application. . . . . . . . . . . . . . . 4 - 96Figure 4-36: Extended Range Cell Cascaded Repeater Application . . . . . . . . . . . . 4 - 96Figure 5-1: PN Offset Planning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 5Figure 5-2: Short PN Sequence w/PILOT_INC = 2 . . . . . . . . . . . . . . . . . . . . . 5 - 5Figure 5-3: Subscriber Location Relative to Search Window . . . . . . . . . . . . . . 5 - 9Figure 5-4: Search Windows in Space. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 10Figure 5-5: Minimum Distance for Adjacent Offset Interference . . . . . . . . . . . 5 - 20Figure 5-6: Active Window Interference Timing Criteria. . . . . . . . . . . . . . . . . 5 - 23Figure 5-7: Neighbor Window Interference Timing Criteria . . . . . . . . . . . . . . 5 - 24Figure 5-8: Active and Neighbor Areas. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 25Figure 5-9: Phase Measurement Translations . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 26Figure 5-10: Adjacent Sector and Adjacent Site Offset Assignment Approaches 5 - 30Figure 5-11: No Transition Zone - PILOT_INC Boundary. . . . . . . . . . . . . . . . . 5 - 34Figure 5-12: Transition Zone - PILOT_INC Boundary . . . . . . . . . . . . . . . . . . . 5 - 35Figure 5-13: i=3,j=2 Repeat Pattern . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 37Figure 6-1: dBd vs. dBi . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 5Figure 6-2: The Relationship of Antenna Height to Number of Cell Sites. . . . 6 - 9Figure 6-3: Antenna Placement - Shared Platform . . . . . . . . . . . . . . . . . . . . . . 6 - 15Figure 6-4: Antenna Placement - Separate Platforms . . . . . . . . . . . . . . . . . . . . 6 - 15Figure 6-5: SC4812T to SC4812T Expansion Frame . . . . . . . . . . . . . . . . . . . . 6 - 18Figure 6-6: 2 Carrier Configuration. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 20Figure 6-7: 8 Carrier Configuration. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 21Figure 6-8: Duplexer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 22Figure 6-9: Two Tone IM Test Set Up (800 MHz) . . . . . . . . . . . . . . . . . . . . . . 6 - 28Figure 6-10: CDMA Duplexing Options. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 32Figure 7-1: Dual Polarization Antenna Element Configurations . . . . . . . . . . . 7 - 4

xvCDMA/CDMA2000 1X RF Planning GuideJAN 2013


Figure 7-2: Probability Distribution SNR for M-branch Selection Diversity System 7 - 7

Figure 7-3: Rayleigh Probability Density Function. . . . . . . . . . . . . . . . . . . . . . 7 - 8Figure 7-4: Reception of Highly Correlated Signals . . . . . . . . . . . . . . . . . . . . . 7 - 9Figure 7-5: Reception of Uncorrelated Signals . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 10Figure 7-6: Correlated Signal Diversity Gain . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 10Figure 7-7: Uncorrelated Signal Diversity Gain . . . . . . . . . . . . . . . . . . . . . . . . 7 - 11Figure 7-8: Uncorrelated Signal Diversity Gain . . . . . . . . . . . . . . . . . . . . . . . . 7 - 11Figure 7-9: Theoretical Model for Base Station Polarization Diversity . . . . . . 7 - 12Figure 7-10: Tx, Rx and Diversity Rx Antenna Configurations . . . . . . . . . . . . . 7 - 17Figure 7-11: Coaxial Cable Design Approach. . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 20Figure 7-12: Fiber Optic Design Approach . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 21Figure 7-13: Coax Design Flow Chart. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 22Figure 7-14: "Bow Tie" Antenna. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 26Figure 7-15: Schematic Diagram of a Power Tap . . . . . . . . . . . . . . . . . . . . . . . . 7 - 27Figure 7-16: Typical Tap Application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 28Figure 7-17: Diagram of a Power Splitter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 28Figure 7-18: Schematic of a Directional Coupler . . . . . . . . . . . . . . . . . . . . . . . . 7 - 29Figure 7-19: Parallel Power Distribution Using a Power Splitter . . . . . . . . . . . . 7 - 30Figure 7-20: Series Power Distribution Using Directional Couplers . . . . . . . . . 7 - 30Figure 7-21: Radiating Cable . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 32Figure 7-22: Radiating Cable Implementation . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 32Figure 7-23: Radiating Cable Coverage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 33Figure 7-24: Link Budget Block Diagrams . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 34Figure 7-25: Maximum Coverage Distance . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 35Figure 7-26: Multiple Floor Coverage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 35Figure 7-27: Logarithmic Path Loss Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 36Figure 7-28: Linear Path Loss Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 37Figure 7-29: Measurement System Test Setup . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 38Figure 7-30: Bi-Directional Amplifier. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 39Figure 7-31: Uni-Directional Uplink Amplifier . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 40Figure 7-32: Downlink Amplifier Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 41Figure 7-33: Effect of a 10 dB Noise Figure Amplifier . . . . . . . . . . . . . . . . . . . 7 - 42Figure 7-34: Noise Figure of a Lossy Device . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 42Figure 7-35: Cascaded System Noise Figure. . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 43Figure 7-36: Uplink Amplifier Gain Setting . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 44Figure 7-37: Noise Summing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 45Figure 7-38: Amplifier Location . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 46Figure 7-39: Amplifier Performance vs. Location . . . . . . . . . . . . . . . . . . . . . . . 7 - 47Figure 7-40: Fiber Optic Star Architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 48Figure 7-41: Fiber Uplink Noise Summing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 50Figure 8-1: Cellsite GPS Satellite Visibility . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 5Figure 8-2: Typical RF GPS Antenna Configuration Diagram . . . . . . . . . . . . . 8 - 6Figure 8-3: Maximizing GPS Antenna Visibility . . . . . . . . . . . . . . . . . . . . . . . 8 - 9

xvi CDMA/CDMA2000 1X RF Planning Guide JAN 2013


Figure 8-4: GPS Antenna Placement Considerations . . . . . . . . . . . . . . . . . . . . 8 - 10Figure 8-5: GPS Antenna Loss Budget / Noise Figure Calculation . . . . . . . . . 8 - 16Figure 8-6: Single and Multi-Frame Remote GPS Configuration. . . . . . . . . . . 8 - 17Figure 8-7: Single and Multi-Frame Remote GPS Configuration. . . . . . . . . . . 8 - 18Figure 8-8: UBS Synchronization Sharing . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 19Figure 9-1: Intra-Band Interference. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 4Figure 9-2: Example of a (1:3) Overlay . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 5Figure 9-3: AMPS System with a Larger CDMA Site Overlay

(cells marked “a” are potential CDMA sites) . . . . . . . . . . . . . . . . . 9 - 7Figure 9-4: Required CDMA Signal Strength vs. Interfering AMPS Signal Strength 9 -

8Figure 9-5: Inter-Band Interference. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 10Figure 9-6: AMPS/TACS/GSM Spectrum . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 11Figure 9-7: DCS 1800 and PCS 1900 Spectrum . . . . . . . . . . . . . . . . . . . . . . . . 9 - 12Figure 9-8: Transmitter Spectral Mask . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 16Figure 9-9: Interfering Transmit Carrier and Sideband Emission Spectrum. . . 9 - 16Figure 9-10: Transmitter IM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 19Figure 9-11: Interfering Transmit Carriers and Intermodulation Spectrum . . . . 9 - 20Figure 9-12: Receiver IM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 21Figure 9-13: Victim Receiver Out-of-Band Intermodulation . . . . . . . . . . . . . . . 9 - 22Figure 9-14: External IM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 23Figure 9-15: Victim Receiver Out-of-Band Desensitization . . . . . . . . . . . . . . . . 9 - 24Figure 9-16: The PCS Spectrum . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 28Figure 9-17: Example Coordination Distances . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 30Figure 9-18: Propagation Curves for High PCS Antennas . . . . . . . . . . . . . . . . . 9 - 33Figure 9-19: Propagation Curves for Low PCS Antennas. . . . . . . . . . . . . . . . . . 9 - 33Figure 9-20: Example Aggregated Service Area. . . . . . . . . . . . . . . . . . . . . . . . . 9 - 34Figure 9-21: Example C/I Curves for a 10 MHz Microwave Receiver. . . . . . . . 9 - 35

xviiCDMA/CDMA2000 1X RF Planning GuideJAN 2013


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xx CDMA/CDMA2000 1X RF Planning Guide JAN 2013

List of Tables


Table 1-1: Quick Guide . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 - 4Table 2-1: Some Common World-Wide Frequency Bands for Cellular and PCS 2 - 3Table 2-2: CDMA Primary and Secondary Channels . . . . . . . . . . . . . . . . . . . 2 - 19Table 2-3: Channel Numbers and Frequencies for Band Class 0 and Spreading Rate 1

2 - 19Table 2-4: CDMA Channel Number to CDMA Frequency Assignment Correspon-

dence. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 20Table 2-5: 7 Cell (120°), 21 Channel Spacing, "B" Band . . . . . . . . . . . . . . . . 2 - 22Table 2-6: Band Class 1 / 14 System Frequency Correspondence . . . . . . . . . 2 - 24Table 2-7: CDMA Channel Number to CDMA Frequency Assignment. . . . . 2 - 24Table 2-8: Channel Numbers and Frequencies for Band Class 1 / 14 and Spreading

Rate 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 25Table 2-9: Preferred Set of Frequency Assignments for Band Class 1 and Spreading

Rate 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 25Table 3-1: Samples of Various f Factors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 14Table 3-2: Propagation Path Loss in Different Areas . . . . . . . . . . . . . . . . . . . 3 - 24Table 3-3: Probability Factors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 36Table 3-4: Interference Rise Scenarios . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 41Table 3-5: F-factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 42Table 3-6: IS-2000 1X Average Eb/No Values . . . . . . . . . . . . . . . . . . . . . . . . 3 - 44Table 3-7: Traffic Mix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 45Table 3-8: Reverse Capacity per Sector for Various Probabilities of Rise - Pedestrian

3 - 47Table 3-9: Reverse Capacity per Sector for Various Probabilities of Rise - Vehicle 3 -

48Table 3-10: Example of Parameter Values . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 55Table 3-11: Interference Rise Scenarios . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 65Table 3-12: I-factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 66Table 3-13: IS-2000 1X Average Eb/No Values . . . . . . . . . . . . . . . . . . . . . . . . 3 - 68Table 3-14: Traffic Mix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 69Table 3-15: Forward Capacity per Sector for Various Probabilities of Rise - Pedestrian

3 - 70Table 3-16: Forward Capacity per Sector for Various Probabilities of Rise - Vehicle 3 -

71Table 3-17: IS-2000 Forward Link Radio Configurations. . . . . . . . . . . . . . . . . 3 - 85Table 3-18: Forward Link Radio Configuration Support for CBSC Release 16 3 - 86Table 3-19: Forward Link Channel Element Resource Requirement . . . . . . . . 3 - 87

xxiCDMA/CDMA2000 1X RF Planning GuideJAN 2013

List of Tables - continued

Table 3-20: IS-2000 Reverse Link Radio Configurations . . . . . . . . . . . . . . . . . 3 - 92Table 3-21: Reverse Link Radio Configuration Support for CBSC Release 16 3 - 92Table 3-22: Reverse Link Channel Element Resource Requirement. . . . . . . . . 3 - 93Table 3-23: Subscriber Distribution of Chicago Metropolitan Area . . . . . . . . . 3 - 98Table 3-24: Chicago Metropolitan Area Summary . . . . . . . . . . . . . . . . . . . . . . 3 - 100Table 4-1: Example Building Penetration Losses (800 & 1900 MHz) . . . . . . 4 - 7Table 4-2: Example of Main Transmission Line Losses . . . . . . . . . . . . . . . . . 4 - 10Table 4-3: Processing Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 24Table 4-4: Receive Path Noise Figures and Gains . . . . . . . . . . . . . . . . . . . . . . 4 - 27Table 4-5: Link Budget Inputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 28Table 4-6: Example of an IS-95 CDMA Reverse RF Link Budget . . . . . . . . . 4 - 37Table 4-7: Example of an IS-2000 1X CDMA RF Link Budget . . . . . . . . . . . 4 - 39Table 4-8: PA Ratings for Some BTS Products . . . . . . . . . . . . . . . . . . . . . . . . 4 - 47Table 4-9: BTS Pilot Power Adjustment Range . . . . . . . . . . . . . . . . . . . . . . . 4 - 50Table 4-10: SC4812T—MC BTS Maximum Pilot Power (dBm) . . . . . . . . . . . 4 - 51Table 4-11: Relative Tx & Rx Link Difference Example . . . . . . . . . . . . . . . . . 4 - 75Table 4-12: TTA Impacts to Reverse Link Budget Inputs: . . . . . . . . . . . . . . . . 4 - 83Table 4-12: Extended Range Reverse RF Link Budget with TTA Example . . . 4 - 83Table 4-13: Extended Range vs. Normal Range Link Budget Comparison . . . 4 - 84Table 4-14: Extended Range Forward RF Link Budget Example . . . . . . . . . . . 4 - 85Table 4-15: BTS SC4812 Extended Range Cell Feature Limitations . . . . . . . . 4 - 91Table 4-16: BTS SC480 Extended Range Cell Feature Limitations . . . . . . . . . 4 - 91Table 4-17: BTS SC2440/SC4840 Extended Range Cell Feature Limitations . 4 - 92Table 4-18: BTS SC7224 Extended Range Cell Feature Limitations . . . . . . . . 4 - 92Table 5-1: Search Window Size vs. Neighbor Separation . . . . . . . . . . . . . . . . 5 - 11Table 5-2: Traffic Channel Window Size . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 13Table 5-3: Cell Radius vs. TchAcqWinSz . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 16Table 5-4: Distance/Timing Restriction on Adjacent Interference

(assuming SRCH_WIN_N = + 30 chips) . . . . . . . . . . . . . . . . . . . . 5 - 21Table 5-5: Pilot Sequence Offset Index Assignment

(assuming a = 18.0 dB, law = 3.0, k = 2.98). . . . . . . . . . . . . . . . . . 5 - 22Table 5-6: Estimates of Reuse Distance and Cluster Size Based on Timing

(assuming Rcdma = 2Rhex, SNeighbor @ 2Rhex and SActive @ 1Rhex) 5 - 24Table 5-7: Calculation of Reuse Distance

(Assuming SNeighbor @ 2R, SActive @ 1R and Active Area Radius (A) @ 2.2R) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 26

Table 5-8: Summary of PN Offset Planning Guidelines . . . . . . . . . . . . . . . . . 5 - 28Table 5-9: Offset Groupings for PILOT_INC = 2 (also 4, 6, 8, and 12) . . . . . 5 - 31Table 5-10: Offset Groupings for PILOT_INC = 3 (also 6 and 12). . . . . . . . . . 5 - 31Table 5-11: Reuse Pattern Coordinates, i & j,

and Cluster Size, N, and D/R . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 36Table 6-1: CDMA Carrier Frequency Range . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 4Table 6-2: PCS Carrier Frequency Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 4Table 6-3: Degradation to Sensitivity Based on Noise Level Below kTBF . . 6 - 11

xxii CDMA/CDMA2000 1X RF Planning Guide JAN 2013


Table 6-4: Antenna Isolation Requirements. . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 13Table 6-5: Types of Frames Sharing Antennas with Starter Frames. . . . . . . . 6 - 19Table 6-6: Duplexer Frequency Response Characteristics. . . . . . . . . . . . . . . . 6 - 22Table 6-7: Minimum IM Orders. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 23Table 6-8: Frequency Block Pairs with Mobile channel Intermodulation products6 -

24Table 6-9: Unsupported Combinations of commonly used 1.9 GHz CDMA Channels

6 - 25Table 6-10: Possible Duplexed Configurations . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 31Table 6-11: Transmission Line Performance. . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 35Table 6-12: Transition Cable Characteristics. . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 36Table 7-1: Motorola Data Table . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 15Table 7-2: Building Topology Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 24Table 7-3: Estimated Coverage Radius . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 25Table 7-4: Typical Values for Power Splitters . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 29Table 7-5: Path Loss Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 36Table 7-6: Average Floor Loss Attenuation Factors . . . . . . . . . . . . . . . . . . . . 7 - 37Table 8-1: Recommended GPS Antenna Specifications . . . . . . . . . . . . . . . . . 8 - 7Table 8-2: Antenna Cable Loss / Bend Radius Data . . . . . . . . . . . . . . . . . . . . 8 - 12Table 8-3: Approximate CSM / CSA Card Initialization Times . . . . . . . . . . . 8 - 20Table 8-4: <CSMRefSrc 1> Parameter - Valid settings. . . . . . . . . . . . . . . . . . 8 - 22Table 8-5: <CSMRefSrc2> Parameter - Valid Settings . . . . . . . . . . . . . . . . . . 8 - 23Table 9-1: Cellular Spectrum Allocation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 11Table 9-2: Inter-Band Interference Scenarios . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 12Table 9-3: Example IM Products . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 18Table 9-4: Partial Example of Base Station Transmitter Specifications . . . . . 9 - 25Table 9-5: DCS 1800 Base Station Transmitter Specifications (GSM 05.05) . 9 - 25Table 9-6: Partial Example of Base Station Receiver Specifications . . . . . . . 9 - 26Table 9-7: In-Band GSM Base Station Receiver Blocking Specifications (GSM 05.05)

9 - 26Table 9-8: Out-of-Band GSM Base Station Receiver Blocking Specifications (GSM

05.05) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 26Table 9-9: Inter-Band Interference Comparison . . . . . . . . . . . . . . . . . . . . . . . 9 - 27Table III-1: Watts to dBm Conversion Table. . . . . . . . . . . . . . . . . . . . . . . . . . . III - 3Table IV-1: Complementary Error Function, Q(x) . . . . . . . . . . . . . . . . . . . . . . IV - 3

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xxiv CDMA/CDMA2000 1X RF Planning Guide JAN 2013


1 How to Use This
Chapter
1

Table of Contents

Guide

1.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 - 3

1.2 Quick Guide to Contents of Each Section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 - 4

1 - 1JAN 2013 CDMA/CDMA2000 1X RF Planning Guide

1

CDMA/CDMA2000 1X RF Planning Guide1 How to Use This Guide

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1 - 2 CDMA/CDMA2000 1X RF Planning Guide JAN 2013

1


1.1 Introduction

The purpose of this document is to provide systems engineers/planners with a basic set of guidelines required to properly design a high quality Code Division Multiple Access (CDMA) RF System. The demarcation point for this guide is primarily at the antenna connectors of the Base Transceiver Station (BTS) equipment. The CDMA RF Planning Guide (RFPG) commences at these antenna connectors and incorporates the RF antenna system as well as the RF link. In general, most of the content provided in this planning guide can be applied to any CDMA system design. In some instances, specific RF planning information unique to Nokia Siemens Networks’ CDMA BTS product is also provided. The following figure pictorially represents the area within a wireless network that this document is focused.

Figure 1-1: Radio Sub-System

Most of the information in this planning guide can be applied to both the IS-95 and IS-2000 CDMA air interface specifications. Where it is appropriate, IS-95 specific and/or IS-2000 specific information will be provided.

General RF considerations for CDMA system designs are addressed as well as 800 MHz and 1900 MHz specific considerations. Some basic spectrum planning guidelines including channel assignments and designations for both 800 MHz and 1900 MHz are located in Appendix 2. Appendix 6 addresses some RF antenna system issues that differ between 800 MHz and 1900 MHz. Throughout this document the terms 800 MHz and cellular may be used interchangeably, as well as 1900 MHz and PCS may also be used interchangeably.

Terms and acronyms are located in Appendix I. Appendix II is a glossary of terms which are referred to in Chapter 5. An understanding of these terms and acronyms is recommended prior to reading this document.

Radio Sub-System

BTS

CDMA AirInterface

FixedPortable

Mobile

Core Network

1 - 3CDMA/CDMA2000 1X RF Planning GuideJAN 2013

1


1.2 Quick Guide to Contents of Each Section

The CDMA RF Planning Guide is a collection of fairly independent chapters covering various aspects of CDMA system RF design and implementation.

The table below outlines the key features of each Chapter.

Table 1-1: Quick Guide

Chapter Number

Chapter title Use it to

1 How to Use this Guide Understand the contents of this document.2 Basic CDMA Spectrum

PlanningLearn how to allocate spectrum for multiple CDMA carriers including channel spacing and guard band considerations, which bands are used for different technologies (world-wide), and the importance of performing background noise measurements, spectrum clearing, and following Federal Rules and Regulations.

3 CDMA Capacity Learn several different approaches on how to estimate the maximum capacity of a CDMA carrier for the forward or reverse link as a function of system parameters. Understand the importance of performing system simulations. Identify some of the limitations of the air interface. Determine an estimate of the number of CDMA cells required to support a given traffic load.


Understand the parameters that comprise the CDMA RF Link Budget. Learn about some of the basic propagation models. Understand some of the power amplifier considerations as they pertain to forward link coverage. Learn some of the issues and considerations of CDMA repeater usage.


Understand how to perform PN offset planning and how to properly set the search window parameters.

6 RF Antenna Systems Learn some of the basic antenna parameters. Discuss some of the issues involved with antenna placement. Understand how to share antennas with other CDMA equipment as well as with AMPS equipment. Establish guidelines for the installation of CDMA systems antennas.


Discuss some of the issues surrounding the usage of dual polarized antennas. Learn some useful information in the area of in-building antenna system design.


Learn about the strength and weakness of various synchronization strategies. Determine the requirements to provide adequate signals to synchronize the CDMA system.

9 Inter-System Interference

Study interference issues with co-location of CDMA with other technologies.


1


I Terms and Acronyms Learn some of the various terms and acronyms.II Glossary Understand some of the various terms used.III Watts to dBm

Conversion TableConvert from watts to dBm and from dBm to watts.

IV Complimentary Error Function Table

Determine the complimentary error function.

Table 1-1: Quick Guide

Chapter Number

Chapter title Use it to


1


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2 Basic CDMA Spectrum
Chapter
2

Table of Contents

Planning

2.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 3

2.2 North American and International Frequency Blocks. . . . . . . . . . . . . . . . . . . . . 2 - 3

2.3 CDMA Channel Spacing - General . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 52.3.1 Minimum Spacing Between CDMA Carriers . . . . . . . . . . . . . . . . . . . . . 2 - 52.3.2 Maximum Spacing Between CDMA Carriers . . . . . . . . . . . . . . . . . . . . . 2 - 82.3.3 Multiple Market Spectrum Planning Considerations. . . . . . . . . . . . . . . 2 - 112.3.4 Multiple Carrier Overlay Guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 11

2.3.4.1 IS-2000 1X New Carrier Overlay . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 142.3.4.2 IS-2000 1X Shared Carrier Overlay . . . . . . . . . . . . . . . . . . . . . . . . 2 - 15

2.3.5 Guard Band Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 152.3.5.1 AMPS Guard Band Recommendation. . . . . . . . . . . . . . . . . . . . . . . 2 - 172.3.5.2 2nd CDMA Carrier with AMPS Guard Band . . . . . . . . . . . . . . . . . 2 - 172.3.5.3 Greater Than Two CDMA Carriers with AMPS Guard Band . . . . 2 - 18

2.4 Channel Spacing and Designation - 800 MHz. . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 192.4.1 Segregated Spectrum . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 21

2.5 Channel Spacing and Designation - 1900 MHz. . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 24

2.6 Dual-Mode vs. Dual-Band. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 26

2.7 Spectrum Clearing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 26

2.8 Background Noise Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 272.8.1 Suggested Measurement Method. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 28

2.8.1.1 Test System Functional Description . . . . . . . . . . . . . . . . . . . . . . . . 2 - 282.8.1.2 Test System Calibration. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 29

2.8.2 Test Procedures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 302.8.3 Data Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 31

2.9 CDMA/LTE Sharing of XMIs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 31

2.10 References. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 33


2

CDMA/CDMA2000 1X RF Planning Guide2 Basic CDMA Spectrum Planning

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2


2.1 Introduction

This chapter provides a set of general guidelines that can be used to properly allocate spectrum for 1.23 MHz CDMA systems (IS-95A/B and IS-2000 Spreading Rate 1), including issues relating to the co-location of CDMA and AMPS systems. Spectrum planning information for IS-2000 Spreading Rate 3 and for Wideband Code Division Multiple Access (WCDMA) for Universal Mobile Telecommunication System (UMTS) will not be covered in this document. Unless otherwise noted, all references to IS-2000 in this document will imply a Spreading Rate of 1. The information is specific to spectrum allocation based on U.S. and International Standards. Issues regarding technological impacts to capacity will be addressed in Chapter 3. In this chapter, "channels" refer to frequency allocation and not conversation channels. As a result, a CDMA channel reference is the same as a CDMA carrier and the two terms can be interchanged for this chapter.

To design a system adequately, RF system engineers will need to work closely with the customer and carefully follow government codes. To optimize CDMA, the signal to noise ratio must be balanced. The goal is to minimize the noise which will maximize the capacity.

Common world-wide frequency bands for cellular, PCS, and 3G are introduced in the chapter along with a general discussion on CDMA channel spacing, multiple carrier guidelines, and guard band considerations. Specifics are given on CMDA channel designations (North American) for 800 MHz and how to segregate the spectrum with existing 800 MHz technologies. PCS (North American) channel designations are listed, followed by a short discussion of dual-mode and dual-band. The topic of spectrum clearing and background noise measurements appears last; however, it is perhaps one of the most important and challenging aspects to the CDMA system design engineer. References include standards and FCC web page locations.

2.2 North American and International Frequency Blocks

The manner in which the frequency spectrum is allocated in some countries imposes some limitations on where CDMA may be implemented. It is difficult to predict the amount of available spectrum or the frequency band which international operators might be considering for their CDMA systems. With this in mind, prior to designing a CDMA system, the CDMA system design engineer should obtain the frequency spectrum information from the operator and then determine the appropriate BTS products to use based on the desired application and the operating frequency. The table below highlights some of the more common frequency bands which are currently being utilized for cellular, PCS, and other technologies in adjacent spectrum throughout the world.

Table 2-1: Some Common World-Wide Frequency Bands for Cellular and PCS

Block DesignatorTransmit Frequency Band (MHz)

Personal Station Base Station

SMR (US) 816-821 861-866

AMPS / EAMPS 824-849 869-894


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To accommodate the evolution to 3G, the International Telecommunications Union - Radio Communication (ITU-R) standardization sector developed specifications for International Mobile Telecommunications - 2000 (IMT-2000). As an output of the standardization effort, several countries throughout the world have agreed to allocate new spectrum for 3G deployments. The chart in Figure 2-1 highlights some of the common world-wide 3G spectrum allocations.

TACS / ETACS 872-915 917-960

DCS 1800 1710-1785 1805-1880

GSM 890-915 935-960

PCS (Korea) 1750-1780 1840-1870

ARDIS (Pan America) 806-824 851-869

RAM Mobitex(Pan America)

896-901 935-940

PCS(U.S. / Pan America)

1850-1910 1930-1990

PCS BC14(US)

1850-1915 1930-1995

FPLMTS 1885-2025 2110-2200

FPLMTS (satellite) 1980-2010 2170-2200

PDC 900 940-956 810-826

PDC 1500(Malaysia / Moscow)

1477-1501 1429-1453

Japan Marinet 887-889 832-834

Japan Analog 898-901, 915-925 843-846, 860-870

JCDMA (BC0-2) 824-830 869-875

DECT (TDD Systems) 1880-1900 1880-1900

PHS (TDD Systems) 1895-1918 1895-1918

Table 2-1: Some Common World-Wide Frequency Bands for Cellular and PCS




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Figure 2-1: 3G Spectrum Allocations

2.3 CDMA Channel Spacing - General

CDMA (IS-95A/B and IS-2000 Spreading Rate 1) is a broadband technology which utilizes 1.2288 MHz bandwidth per CDMA Channel (this is often rounded off to 1.23 MHz). In order to deploy an initial CDMA channel, spectrum must be allocated for the CDMA channel and the guard bands that are required on each side of the channel. In order to deploy a second CDMA channel, the channel spacing between the CDMA channels must be determined. Prior to deploying the first CDMA channel, long term spectrum planning should be performed in order to maximize the capacity of a multiple carrier CDMA block of spectrum. This section provides information on CDMA channel spacing, multiple carrier guidelines, and guard band considerations.

In this section, "channel" is defined as each 1.2288 MHz carrier and not as a conversation path. For AMPS, each frequency (carrier) corresponds to one conversation path. Therefore, a channel could be used to discuss conversational paths or the number of carriers. For CDMA, each carrier can support many conversation paths and therefore the term "channel" can take on different meanings based upon the context in which it is used.

2.3.1 Minimum Spacing Between CDMA Carriers

As the number of the CDMA subscribers increases, there may be a need to add additional CDMA carrier frequencies to the system. If the first and second carrier frequencies are to be adjacent to one another, then the channel spacing between CDMA carriers (center to center) needs to be determined. For 800 MHz IS-95A/B and IS-2000 based systems with a 30 kHz channel increment, the minimum recommended channel spacing separation between CDMA channels is 1.23 MHz (see Figure 2-2).


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Figure 2-2: Minimum Spacing Between 800 MHz CDMA Channels

Note: For the example in Figure 2-2, the second CDMA channel (whether it is ubiquitous or non-ubiquitous) must be co-located with the first CDMA channel in a 1-to-1 overlay approach throughout the second CDMA channel deployment area (see Section 2.3.4).

For 1900 MHz IS-95A/B and IS-2000 based systems with a 50 kHz channel increment, the minimum recommended channel spacing separation between CDMA channels is 1.25 MHz (see Figure 2-3).

Figure 2-3: Minimum Spacing Between 1900 MHz CDMA Channels

Note: For the example in Figure 2-3, the second CDMA channel (whether it is ubiquitous or non-ubiquitous) must be co-located with the first CDMA channel in a 1-to-1 overlay approach throughout the second CDMA channel deployment area (see Section 2.3.4).

The minimum channel spacing places the broadband carriers adjacent to one another and allows the sidebands of each to intrude into the band of the other. The adjacent channel interference for this minimum channel separation will slightly reduce the capacity of both CDMA carriers. A CDMA channel with adjacent CDMA channels on both sides will have an even greater reduction in capacity. If system noise, non-linearities, or other imperfections increase the energy in the skirts of the carriers, then an increased capacity reduction may be experienced.

A reverse link adjacent channel interference analysis was performed in an attempt to estimate and compare the capacity impact of a 1.26 MHz and a 1.23 MHz channel spacing. The analysis estimates the noise rise for a single carrier configuration (i.e. no adjacent carriers), for the center

1.23 MHz

1st CDMA Channel1.23 MHz

2nd CDMA Channel1.23 MHz

Guard BandGuard Band

1.25 MHz



Guard BandGuard Band


2


carrier of a three carrier configuration with a 1.26 MHz channel separation, and for the center carrier of a three carrier configuration with a 1.23 MHz channel separation. The results of this analysis where all of the carriers are loaded equally is shown in Figure 2-4. (Note: The capacity results shown in Figure 2-4 should not be used to estimate the actual capacity of a CDMA carrier. They are for comparison purposes only.)

Figure 2-4: Adjacent Channel Interference Reverse Rise Estimates

One method of analyzing the impact is to compare the number of users at a fixed maximum noise rise level. Choosing 6 dB to be the maximum noise rise level, the following results can be extrapolated from the chart in Figure 2-4.

• 23.5 Users with 0 Adjacent Carriers• 22.6 Users with 2 Adjacent Carriers @ 1.26 MHz• 21.8 Users with 2 Adjacent Carriers @ 1.23 MHz

The capacity loss from 0 Adjacent Carriers to 2 Adjacent Carriers with 1.26 MHz spacing is approximately 0.9 users. The capacity loss from 2 Adjacent Carriers with 1.26 MHz spacing to 2 Adjacent Carriers with 1.23 MHz spacing is approximately 0.8 users.

Another method of analyzing the impact is to compare the noise rise increase at a fixed maximum number of users. Choosing 23 users to be the maximum number of users, the following noise rise results can be extrapolated from the chart in Figure 2-4.

0

2

4

6

8

10

12

14

0 5 10 15 20 25 30

# of Users

Ris

e -

dB

0 Adjacent Carriers

2 Adjacent Carriers @ 1.23 MHz

2 Adjacent Carriers @ 1.26 MHz


2


• 5.7 dB noise rise with 0 Adjacent Carriers• 6.2 dB noise rise with 2 Adjacent Carriers @ 1.26 MHz• 6.7 dB noise rise with 2 Adjacent Carriers @ 1.23 MHz

The noise rise increase from 0 Adjacent Carriers to 2 Adjacent Carriers with 1.26 MHz spacing is approximately 0.5 dB. The noise rise increase from 2 Adjacent Carriers with 1.26 MHz spacing to 2 Adjacent Carriers with 1.23 MHz spacing is approximately 0.5 dB.

The results of this analysis show a minimal impact going from 1.26 to 1.23 MHz channel spacing. Ultimately, the system operator must decide whether the modest capacity impact of using the minimum channel spacing is worth the marginal gain in frequency spectrum.

2.3.2 Maximum Spacing Between CDMA Carriers

With the allocations of new spectrum for 3G applications through-out the world, a new opportunity for deploying CDMA systems has been created. There are many different considerations that may impact the spectrum planning for a CDMA system (total spectrum available, government rules and regulations, adjacent spectrum guard band requirements, amount of spectrum that is clear and available for use, etc.). For certain applications, there may be some capacity benefits in reducing the adjacent spectrum guard band requirements in order to increase the guard band between the CDMA carriers. This approach will typically be applied towards the deployment of new spectrum allocations (i.e. 3G deployments). An appropriate adjacent spectrum guard band analysis must be performed to justify an adjacent spectrum guard band reduction in order to increase the guard band between the CDMA carriers.

Since the minimum channel spacing recommendation does have some impact on capacity, the optimal channel spacing may not always be the minimum channel spacing recommendation stated in Section 2.3.1. The optimal channel spacing from a CDMA capacity perspective is to maximize the channel spacing within the total contiguous bandwidth available for the CDMA channels (after all of the spectrum planning considerations for guard band and other requirements have been taken into account). For those applications where there is flexibility in performing spectrum planning, the following spectrum planning example of an entire block of spectrum (including guard band requirements) can be performed in order to determine the maximum channel spacing which maximizes capacity. The following multiple carrier, maximum channel spacing example can be applied from a general perspective towards both IS-95A/B and/or IS-2000 1X carrier systems.

Example Assumptions:• 5 MHz "D" block of 1900 MHz full duplexed spectrum (5 MHz for Tx, 5 MHz for Rx)• Channel increment is 50 kHz• Guard band requirements for each end of the spectrum is 290 kHz per side

(Note: The 290 kHz guard band value was arbitrarily chosen for this example. It does not represent an actual guard band recommendation. See Section 2.3.5 for more information regarding a guard band analysis and considerations.)

• Government rules and regulations allow the following spectrum planning assignments


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1. Determine the total number of channel numbers (i.e. 30 kHz channel increments for 800 MHz systems, or 50 kHz channel increments for 1900 MHz systems) that are available within the allocated bandwidth.

Example: 5 MHz / 0.05 MHz = 100 "D" block channel numbers (see Figure 2-5).

Figure 2-5: Total Channel Numbers Available

2. Allocate and assign the guard band channels to each end of the spectrum. Calculate the minimum number of channel numbers to satisfy the guard band requirements by dividing the guard band by the channel increment and rounding up to the nearest integer.

Example: 290 kHz / 50 kHz = 5.8 = 6 channel numbers per side. See Figure 2-6.6 channels x 50 kHz = 300 kHz per side300 kHz x 2 = 600 kHz = 0.6 MHz total guard band

Figure 2-6: Assign Guard Band

3. Use the following equation to calculate the total number of 1.23 MHz CDMA channels (Nc) for the allocated bandwidth.

Nc = [EQ 2-1]

Where:represents the integer value of X (or floor value of X)

BW is the total bandwidth allocated for CDMA channelsGB is the total guard band requirementsFS is the minimum frequency spacing (1.23 for 800 MHz, 1.25 for 1900 MHz)

Example: BW = 5 MHz, GB = 0.6 MHz, FS = 1.25 MHz

Nc = = 3 CDMA channels

100 Channel Increments x 50 kHz = 5 MHz

300-399

Guard

300-305 394-399

BandGuardBand


306-393

BW GB–( ) FS⁄

X

5 0.6–( ) 1.25⁄


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4. Determine the minimum number of channel numbers to allocate for each CDMA channel.

• For 30 kHz channel spacing systems (800 MHz systems) use 41 channel numbers1.23 MHz / 0.03 MHz = 41 channel numbers

• For 50 kHz channel spacing systems (1900 MHz systems) use 25 channel numbers 1.25 MHz / 0.05 MHz = 25 channel numbers

Example: 25 channel numbers for each carrier

5. Assign the minimum number of channels for the 1st and last CDMA carriers next to each of the adjacent spectrum guard bands.

Example: Assign 25 channel numbers for the F1 and F3 CDMA carriers next to each adjacent spectrum guard band. See Figure 2-7.

Figure 2-7: Assign 1st and Last CDMA Carries

6. Equally distribute the remaining CDMA carriers while maximizing the spacing between each carrier.

Example: Assign 25 channel numbers for the single remaining CDMA carrier (F2) as close to the center of the remaining spectrum as possible. See Figure 2-8.

Figure 2-8: Equally Distribute Remaining CDMA Carriers

Note: For the "D" block example shown above, channels 318 and 381 are conditionally valid channel numbers according to the IS-95/IS-2000 standards (see Table 2-8). As stated previously, an appropriate guard band analysis must have been performed to justify a guard band reduction in order to utilize these conditional channel numbers.

Guard

CDMA Carrier F1

300-305 306-330 331-368 369-393 394-399

Band

Center Freq. Channel = 318CDMA Carrier F3

Center Freq. Channel = 381

GuardBand


Guard

CDMA Carrier F1

300-305 306-330 331-337 338-362 363-368 369-393 394-399

Band



Center Freq. Channel = 381

GuardBand

ExcessBand

ExcessBand



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2.3.3 Multiple Market Spectrum Planning Considerations

Prior to finalizing a spectrum planning design for an individual market, there are various inter-system operation aspects between multiple markets which may need to be considered. Inter-system references in this section can be applied towards different systems (or markets) under the control of a single operator (or corporation) or under the control of different operators (or corporations). In either case, a multiple market spectrum planning perspective may need to be considered. There are two major categories of inter-system operation services that will be considered; inter-system handoffs and inter-system automatic roaming.

An inter-system handoff refers to the general provisions by which a call in progress on a traffic channel under the control of one system may be automatically transferred to another traffic channel under the control of a different system without interruption to the ongoing communication. Inter-system handoffs can be inter-vendor (i.e. via IS-41 or GSM MAP) or intra-vendor handoffs. The inter-system intra-vendor handoffs can take the form of soft or hard handoffs. If adjacent markets will need to perform inter-system handoffs to each other, the channel numbers selected between the adjacent markets may need to be coordinated. For example, if inter-system soft handoffs are to be implemented, then the channel numbers between the inter-system handoff boundaries must be the same.

Inter-system automatic roaming refers to the general provisions for automatically providing cellular services to the subscribers which are operating outside their home service area, but within the aggregate service area of all participating systems. Inter-system roaming can be inter-vendor (i.e. via IS-41 or GSM MAP) or intra-vendor automatic roaming. If different systems will need to perform inter-system automatic roaming to each other, the channel numbers selected between the different systems may need to be coordinated. For example, the channel numbers on a preferred roaming list must be coordinated to accommodate all of the roaming markets.

As a result, a spectrum planning design for an individual market may need to be considered from a multiple market spectrum planning perspective depending upon the inter-system services that will be supported.

2.3.4 Multiple Carrier Overlay Guidelines

As the capacity demand of a system increases, the deployment of additional CDMA carriers will eventually be necessary. The capacity demand may or may not require a ubiquitous deployment of a new carrier throughout the underlying carrier region. When a new carrier is deployed (either ubiquitous or non-ubiquitous), the new carrier should be deployed with a 1-to-1 co-location overlay with the underlying carriers (refer to Figure 2-9 for overlay examples) and should also be deployed with the same coverage area as the underlying carrier.


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Figure 2-9: 1-to-1 Overlay Examples

For the examples in Figure 2-9, an F2 carrier must be co-located with every F1 site within the new carrier region. It is important to note that F1 micro-cells located in the new carrier region should also be co-located with F2 micro-cells.

Examples of non 1-to-1 overlays are provided in Figure 2-10. These examples are similar to those provided in Figure 2-9, but are NOT recommended.

Figure 2-10: Non 1-to-1 Overlay Examples (NOT Recommended)

There are two main reasons for requiring a 1-to-1 co-location overlay of a new carrier with the same coverage area.

• To overcome adjacent channel interference causing a near/far interference effect• To overcome a potential service acquisition issue created by uneven coverage between

CDMA carriers

If a 1-to-1 co-location overlay deployment is NOT implemented, a near/far interference effect is created from the adjacent CDMA carriers. This will create coverage holes near the sites that are not co-located with the underlying carriers. See Section 2.3.5 for more details regarding the near/far effect.

F1 & F2 SitesF1 Only Sites

Non-Ubiquitous 1-to-1 Overlay Ubiquitous 1-to-1 Overlay

All Sites have F1 & F2


Non-Ubiquitous Non 1-to-1 Overlay Ubiquitous Non 1-to-1 Overlay



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If a 1-to-1 co-location overlay deployment is NOT implemented, a service acquisition issue may be created by the uneven coverage between the CDMA carriers. A diagram to help explain the service acquisition issue is shown in Figure 2-11.

Figure 2-11: Service Acquisition Issues Due To Uneven Carrier Coverage

There are two different types of service acquisition issues which can be created as a result of uneven carrier coverage as shown in Figure 2-11.

• At point A, the primary carrier (F1) of Cell 1 is transmitting the channel list message containing channel numbers for both F1 and F2. With 2 channels input into the hashing algorithm, half of the subscribers at point A should hash to F2. Since the coverage of F2 is too weak to acquire service, those same subscribers will fall back to the primary carrier and attempt to reread the channel list message. These same subscribers will again try to hash to F2 and again fail to acquire service. This cycle will repeat itself until those subscribers move to a location where both F1 and F2 coverage from Cell 1 is acceptable.

• At point B, the primary carrier (F1) of Cell 1 is transmitting the channel list message containing channel numbers for both F1 and F2. With 2 channels input into the hashing algorithm, half of the subscribers at point B should hash to F2. Since the coverage of F2 is provided by Cell 2 which uses a different PN offset, those subscribers will not be able to decode the synchronization and paging channels and the service acquisition attempt will fail. As a result, those same subscribers will fall back to the primary carrier and attempt to reread the channel list message. These same subscribers will again try to hash to F2 and again fail to acquire service. This cycle will repeat itself until those subscribers move to a location where both F1 and F2 coverage is provided by the same cell.

As a result, a new carrier should always be deployed with a 1-to-1 co-location overlay with the underlying carriers and should also be deployed with the same coverage area as the underlying carriers. Also, if a new cell site is deployed into an existing multiple carrier region, then all of the carriers in this region should be implemented at the new cell site and the coverage area for each carrier should be made the same.

Cell 1

Cell 2B

A

F1&F2

F2

Cell 1 F1Cell 1 - F1 & F2 Coverage

Cell 2 - F2 Coverage

Cell 1 - F1 Coverage

F1 is primary carrier


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2.3.4.1 IS-2000 1X New Carrier Overlay

The multiple carrier overlay guidelines described in Section 2.3.4 apply to both IS-95A/B and IS-2000 1X CDMA carriers. Figure 2-12 shows an example of a new IS-2000 1X carrier being deployed in a system with existing IS-95A/B carriers.

Figure 2-12: New IS-2000 1X Carrier Deployment

A new IS-2000 1X overlay carrier being deployed into an existing IS-95A/B system would have to be implemented in a 1-to-1 co-location overlay with the underlying IS-95A/B carriers and should also be deployed with the same coverage area as the underlying IS-95A/B carriers. For some applications, a new IS-2000 1X carrier may be deployed to support 1X data applications only. Without the burden of the co-existing voice capacity, an IS-2000 1X data only carrier can support higher data rates with improved data capacity. From an overall data performance perspective, a dedicated 1X data only carrier should provide the best data performance results.

With IS-2000 1X, higher data rates can be achieved with smaller radius cell sites. The link budget improvements from a smaller radius cell site can be applied towards producing higher average data rates. As a result, one option is to cell split an area (i.e. deploying more cells in the same area) in order to improve the chances of achieving higher data rates. In a mixed IS-95A/B and IS-2000 system, a new cell site being deployed to improve 1X data performance must also deploy the existing IS-95A/B carriers at the new cell site.

Another method to improve 1X data performance is to deploy a second IS-2000 1X carrier to an area that already has 1X deployed (see Figure 2-13).

Figure 2-13: Second IS-2000 1X Carrier

This is an effective approach to alleviate system loading and also increase end user 1X data

F1 F2 F3 F4

IS-95A/B IS-2000

F1 F2 F3 F4

IS-95A/B IS-2000

F5


2


performance. This approach also offers a simple and cost effective solution to improve 1X data performance, since an additional 1X carrier can be easily implemented by adding extra 1X MCC and BBX cards or licensing and enabling an additional carrier on an XMI/DMI to the existing 1X cell sites (assuming the existing 1X cell sites are not populated to their maximum carrier capacity).

2.3.4.2 IS-2000 1X Shared Carrier Overlay

As an alternate approach to deploying a new CDMA channel frequency, the Walsh code orthogonality between the IS-95A/B and IS-2000 air interfaces will allow a new IS-2000 1X carrier to share the carrier frequency with an existing IS-95A/B carrier (see Figure 2-14).

Figure 2-14: IS-2000 1X Shared Carrier Overlay

For initial 1X deployments with low 1X subscriber penetration rates, this may be a viable option to choose, but it is not recommended if the existing IS-95A/B carrier capacity is already near its maximum limit. With the burden of the co-existing IS-95A/B traffic capacity, an IS-2000 1X carrier will be limited in its data performance. High data rate 1X usage will introduce load that may result in bursty performance degradation of the underlying IS-95A/B voice. On the other hand, the IS-95A/B voice users may end up restricting the high data rate 1X users. To protect the IS-95A/B voice users, it is recommended to limit the high data rate application usage on the 1X carrier for a shared carrier overlay type of deployment. Since one of the main reasons for deploying a 1X carrier is to provide high data rate service, limiting the high data rate usage on the 1X carrier may actually defeat the purpose of deploying the 1X carrier in the first place. As a result, the benefit of using this type of deployment may be somewhat limited.

2.3.5 Guard Band Considerations

General spectrum planning guidelines require the use of a guard band between adjacent spectrum being used for different operator systems or for different air interface technologies. The guard band is required to minimize the intra-band and inter-band interference to and from the adjacent spectrum. The determination of a proper guard band involves a detailed analysis of the forward and reverse links for both systems being analyzed. Guard band planning may need to take into account the adjacent spectrum, which is geographically along the border of the system, as well as that which is geographically co-located with the system. Cooperation between neighboring system operators is essential to minimize interference problems. All of the possible interference scenarios from both systems perspectives must be considered in the analysis. The more common interference scenarios

F1 F2 F3

IS-95A/B IS-2000


2


between two systems are listed below.

• System A subscriber(s) interfering with System B base station• System A base station interfering with System B subscriber(s)• System B subscriber(s) interfering with System A base station• System B base station interfering with System A subscriber(s)

Depending upon the particular interference scenario, there are four predominant interference mechanisms that may need to be analyzed.

• Transmitter sideband emissions interfering with the adjacent band receiver• Transmitter intermodulation (IM) products interfering with the adjacent band receiver• Receiver desensitization from an interfering transmit carrier• Receiver intermodulation from two or more interfering transmit carriers

Additional details regarding the above interference scenarios and interference mechanisms are provided in Chapter 9. A detailed analysis of the guard band requirements may need to take into account the following factors:

Interference Geometries• geographic and/or geometric properties of the interference location• antenna orientation (height, azimuth, downtilt)• total path loss (propagation loss, antenna discrimination, and obstruction losses)

Interference Characteristics (for desired and interference signals)• air interface technologies being used• antenna gain and feeder line losses• transmit power, duty cycle, and power spectral density• transmit and receive frequencies being used• transmit and receive filter characteristics• receiver noise threshold and other receiver performance characteristics

A potential interference problem, known as the near/far effect, is created by the geometric relationship between a subscriber and base station. This effect is produced when a subscriber is located far from its serving base station, but near an interfering base station. Under these circumstances, the strength of the desired signal is low while the strength of the interfering signal is high. A guard band analysis may need to take into account any near/far effects that may be present.

The guard band analysis utilizes all of the relevant parameters from the subscriber/base station geometries and characteristics to calculate the desired signal strength, receiver noise, and the received interfering signal strength. The net value should include all of the relevant effects of transmit powers, transmit power spectral densities, path loss, filtering, duty cycles, and summation over multiple interferers. Depending upon the air interface technology that is being analyzed, a degradation metric is selected (i.e. C/I, noise floor rise, receiver sensitivity, Eb/No, BER, FER, etc.) to determine how these net values will impact the performance of the receiver, and whether


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this impact is acceptable or not. Ultimately, the guard band that is selected should provide an acceptable performance value from the degradation metric. Interference improvement mechanisms (i.e. adjustments to base station transmit powers, adding extra filtering, increasing isolation, etc.) should also be considered in the guard band analysis determination.

2.3.5.1 AMPS Guard Band Recommendation

For an 800 MHz system with a 30 kHz channel spacing, it has been determined through a guard band analysis that the minimum recommended guard band between a CDMA channel and an AMPS channel is 0.27 MHz. The initial introduction of CDMA will require a band segment of 1.77 MHz. The band segment consists of the 1.23 MHz required for the CDMA carrier bandwidth plus 0.27 MHz of AMPS guard band on both sides of the CDMA carrier. The minimum frequency separation required between any CDMA carrier and the nearest AMPS carrier is 900 kHz (center to center).

The CDMA carrier width (1.23 MHz) is the result of the chip rate chosen for the Pseudorandom Noise (PN) spreading sequence. The guard band between CDMA and analog systems is defined as the minimum frequency separation required such that the level of interference caused by one FM subscriber is less than a predetermined threshold. The threshold is taken to be the thermal noise level in each receiver.

Figure 2-15: Calculation of Spectrum Required for a CDMA Carrier

2.3.5.2 2nd CDMA Carrier with AMPS Guard Band

The following figure summarizes the additional and total number of AMPS channels removed to free up spectrum for the second CDMA channel for an 800 MHz system with a 30 kHz channel spacing.

Figure 2-16: Calculation of Minimum Spectrum Required for Two CDMA Channels

CDMA Channel = 1.23 MHz = 1.23MHz / 30kHza = 41 AMPS Channels

CDMA Guard = 0.27 MHz/side = 0.54MHz / 30kHza = 18 AMPS Channels

Totals 1.77 MHz 59 AMPS Channels

a. One AMPS Channel = 30 kHz

CDMA Spacing= 1.23 MHz = 1.23MHz / 30kHza = 41 AMPS Channels

CDMA Channel = 1.23 MHz = 1.23MHz / 30kHza = 41 AMPS Channels

CDMA Guard = 0.27 MHz/side = 0.54MHz / 30kHza = 18 AMPS Channels

Totals 3.00 MHz 101 AMPS Channels

a. One AMPS Channel = 30 kHz


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The difference between the 1st CDMA carrier and the 2nd CDMA carrier is equal to the channel spacing. Minimal channel spacing is 1.23 MHz (41 AMPS channels). The following figure represents the frequency requirements for 2nd carrier implementation.

Figure 2-17: 2nd CDMA Carrier

2.3.5.3 Greater Than Two CDMA Carriers with AMPS Guard Band

Additional carriers can be added as outlined in Section 2.3.4. See Figure 2-18 for a 3-carrier example for an 800 MHz system with a 30 kHz channel spacing. CDMA carriers must be at least 1.23 MHz apart with guard bands on each end. The governing body controlling the frequency allocations will dictate the amount of spectrum available for each operator. This spectrum will limit the number of carriers allowed per block.

Figure 2-18: 3rd CDMA Carrier

1.23 MHz



AMPS GuardAMPS Guard0.27 MHz0.27 MHz

1.23 MHz



AMPS GuardAMPS Guard

1.23 MHz

3rd CDMA Channel1.23 MHz 0.27 MHz0.27 MHz


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2.4 Channel Spacing and Designation - 800 MHz

The Primary and Secondary CDMA Channel will be assigned as indicated in Table 2-2. The information presented in Table 2-3 is taken directly from the IS-95A/B and IS-2000 standards that outline the channel allocations shared by CDMA and AMPS technologies (Note: information provided applies only to Spreading Rate 1 for IS-2000).

Table 2-2: CDMA Primary and Secondary Channels

Table 2-3: Channel Numbers and Frequencies for Band Class 0 and Spreading Rate 1

“A” Band “B” Band BC0-2 Band

Primary 283 384 40

Secondary 691 777a

a. In the United States due to proximity of 800 MHz Air-Ground Radiotelephone Service, channel 777 has interference considerations associated with it. Use of this channel should require determination of sufficient isolation prior to implementation.

1022

SystemDesignator

CDMA Channel Validity

Analog ChannelCount

CDMA Channel Number

Transmitter Frequency Band (MHz) Subscriber Base

A"(1 MHz)

Not Valid 22 991 - 1012 824.040-824.670 869.040-869.670

Valid a 11 1013 - 1023 824.700-825.000 869.700-870.000

A(10 MHz)

Valid a 311 1 - 311 825.030-834.330 870.030-879.330

Not Valid 22 312 - 333 834.360-834.990 879.360-879.990

B(10 MHz)

Not Valid 22 334 - 355 835.020-835.650 880.020-880.650

Valid a 289 356 - 644 835.680-844.320 880.680-889.320

Not Valid 22 645 - 666 844.350-844.980 889.350-889.980

A’(1.5 MHz)

Not Valid 22 667 - 688 845.010-845.640 890.010-890.640

Valid b 6 689 - 694 845.670-845.820 890.670-890.820

Not Valid 22 695 - 716 845.850-846.480 890.850-891.480

B’(2.5 MHz)

Not Valid 22 717 - 738 846.510-847.140 891.510-892.140

Valid a 39 739 - 777 847.170-848.310 892.170-893.310

Not Valid 22 778 - 799 848.340-848.970 893.340-893.970


2


a. The valid channel numbers provided in this table were taken directly from the IS-95 standard. Before using a valid channel number that is near the band edge, an analysis is required to verify proper guard band and FCC emission compliance with the adjacent band.b. The spectrum allocated to the A’ band is not sufficient for a CDMA carrier.

In Table 2-3, the center frequency (in MHz) corresponding to the channel number is calculated as shown in Table 2-4, where N represents the channel number.

Table 2-4: CDMA Channel Number to CDMA Frequency Assignment Correspondence

A visual depiction of the CDMA frequencies is shown in Figure 2-19.

BC0-2Not Valid 22 991 - 1012 824.040-824.670 869.040-869.670

Valid 55 1013 - 144 824.700-829.320 869.700-834.320

Not Valid 22 145 - 166 829.350-829.980 834.350-834.980

Transmitter CDMA Channel Number Center Frequency (MHz)

Subscriber Station 1 < N < 799 0.030 * N + 825.000

991 < N < 1023 0.030 * (N-1023) + 825.000

Base Station 1 < N < 799 0.030 * N + 870.000

991 < N < 1023 0.030 * (N-1023) + 870.000

SystemDesignator

CDMA Channel Validity

Analog ChannelCount

CDMA Channel Number

Transmitter Frequency Band (MHz) Subscriber Base


2


Figure 2-19: AMPS Frequency Allocation

2.4.1 Segregated Spectrum

When the CDMA carrier is deployed where another technology already exists, the system spectrum must be split into two frequency bands. One band is for the existing technology and the other band is for digital frequency bands. This concept is shown in the following “B” band frequency chart (see Table 2-5). Note that the digital band includes a single primary CDMA carrier.

A’A

Wireline

Non-Wireline

AMPSA”

EAMPSB’

EAMPS

691

777384

2831st A Band CDMA

BAMPS

2nd ary A Band CDMA

1st B Band CDMA 2ndary B Band CDMA

666

667

716

717

333

334

991

1023

1 799

EAMPS

1st refers to the primary channel.2nd ary refers to the secondary channel. Not to be confused with a second carrier.


2


Any advanced technology (NAMPS, TDMA or CDMA) that must co-exist with AMPS/EAMPS in the available spectrum requires implementation of segregated spectrum. Transition from AMPS to CDMA consists of effectively replacing AMPS channels with CDMA channels. In such a mixed system, co-channel interference is minimized by dividing the available cellular spectrum into two parts as depicted in Figure 2-20. The segregated spectrum approach also requires the system to be partitioned into three distinct geographic areas. This technique ensures the physical separation needed to permit reuse of AMPS channels from the CDMA band.

There are two benefits to segregated spectrum planning. First, spectrum division reduces concern over introducing interference as each CDMA carrier is implemented. Second, it will allow for independent AMPS and CDMA planning.

The three distinct geographic areas created are identified as follows:

Core Zone - The region in which CDMA carriers are deployed. The core will operate CDMA channels in the CDMA band and AMPS channels in the AMPS band. The existing AMPS frequency plan is modified to delete AMPS channels in the CDMA band.

Table 2-5: 7 Cell (120°), 21 Channel Spacing, "B" Band

A1 B1 C1 D1 E1 F1 G1 A2 B2 C2 D2 E2 F2 G2 A3 B3 C3 D3 E3 F3 G3

334 335 336 337 338 339 340 341 342 343 344 345 346 347 348 349 350 351 352 353 354

355 356 357 358 359 360 361 362 363 364 365 366 367 368 369 370 371 372 373 374 375

376 377 378 379 380 381 382 383 384 385 386 387 388 389 390 391 392 393 394 395 396397 398 399 400 401 402 403 404 405 406 407 408 409 410 411 412 413 414 415 416 417

418 419 420 421 422 423 424 425 426 427 428 429 430 431 432 433 434 435 436 437 438

439 440 441 442 443 444 445 446 447 448 449 450 451 452 453 454 455 456 457 458 459

460 461 462 463 464 465 466 467 468 469 470 471 472 473 474 475 476 477 478 479 480

481 482 483 484 485 486 487 488 489 490 491 492 493 494 495 496 497 498 499 500 501

502 503 504 505 506 507 508 509 510 511 512 513 514 515 516 517 518 519 520 521 522

523 524 525 526 527 528 529 530 531 532 533 534 535 536 537 538 539 540 541 542 543

544 545 546 547 548 549 550 551 552 553 554 555 556 557 558 559 560 561 562 563 564

565 566 567 568 569 570 571 572 573 574 575 576 577 578 579 580 581 582 583 584 585

586 587 588 589 590 591 592 593 594 595 596 597 598 599 600 601 602 603 604 605 606

607 608 609 610 611 612 613 614 615 616 617 618 619 620 621 622 623 624 625 626 627

628 629 630 631 632 633 634 635 636 637 638 639 640 641 642 643 644 645 646 647 648

649 650 651 652 653 654 655 656 657 658 659 660 661 662 663 664 665 666 - - -

- - - - - 717 718 719 720 721 722 723 724 725 726 727 728 729 730 731 732

733 734 735 736 737 738 739 740 741 742 743 744 745 746 747 748 749 750 751 752 753

754 755 756 757 758 759 760 761 762 763 764 765 766 767 768 769 770 771 772 773 774

775 776 777 778 779 780 781 782 783 784 785 786 787 788 789 790 791 792 793 794 795

796 797 798 799

cyan CDMA Channel (364 through 404)yellow CDMA Guard Band (355 through 363 and 405 through 413)


2


Perimeter Zone - The outermost area contains those cells that are located at an adequate distance from the CDMA core such that it is acceptable to assign AMPS channels that are in the CDMA band. This physical separation serves to maintain acceptable interference levels.

Transition Zone - The transition zone (also known as the guard zone) is located between the core and the perimeter. AMPS channels in the CDMA band should not be assigned in the transition zone. This zone should not be confused with the transition cell hand-down capability.

Figure 2-20: Segregated Spectrum

The grade-of-service (blocking) should be checked for all cells to make sure it is acceptable, particularly in the transition zone. In the event that the grade of service is unacceptable and all channels have been assigned, certain design options can be exercised to resolve this problem. The first option that may be considered is to replace the AMPS channels with CDMA channels. The cell would then become a core cell. A second option would be to sectorize or cell split the AMPS cell. A third option would be to reduce the size of the CDMA core to the point that this cell would then be considered a perimeter zone cell.

Segregated spectrum may be implemented in various configurations: uniform, non-uniform and homogenous. Uniform deployment consists of a single core area surrounded by a single transition and perimeter zone. Non-uniform implementation may establish multiple CDMA core and transition zones. A homogeneous implementation occurs when the entire system consists of CDMA and there are no transitions or perimeter zones. Homogeneous system composition may be considered by isolated systems or systems adjacent to another CDMA system operating in the same frequency spectrum.

Perimeter Zone

Core

Transition

CoreCORE

CORE

Option # 3 - Homogeneous

Option # 1 - Uniform Option # 2 - Non-Uniform

Requires Isolated system or adjacent CDMA systems

Zone


2


2.5 Channel Spacing and Designation - 1900 MHz

The block designators for the personal and base station frequencies are as specified in Table 2-6.

Table 2-6: Band Class 1 / 14 System Frequency Correspondence

Band Class 1 includes blocks A-F. Band Class 14 includes blocks A-G.

The channel spacing, CDMA channel designations and transmit center frequencies are specified in Table 2-7.

Table 2-7: CDMA Channel Number to CDMA Frequency Assignment

Transmission on conditionally valid channels is permissible if the adjacent block is allocated to the licensee or if other valid authorization has been obtained. Valid CDMA Channels Numbers are identified in Table 2-8.



A 1850-1865 1930-1945

D 1865-1870 1945-1950

B 1870-1885 1950-1965

E 1885-1890 1965-1970

F 1890-1895 1970-1975

C 1895-1910 1975-1990

G 1910-1915 1990-1995

Transmitter CDMA Channel Number Center Frequency (MHz)

Personal Station (BC1) 0 < N < 1199 1850.000 + 0.050 * N

Base Station (BC1) 0 < N < 1199 1930.000 + 0.050 * N

Personal Station (BC14) 0 < N < 1299 1850.000 + 0.050 * N

Base Station (BC14) 0 < N < 1299 1930.000 + 0.050 * N


2


Table 2-8: Channel Numbers and Frequencies for Band Class 1 / 14 and Spreading Rate 1

Table 2-9: Preferred Set of Frequency Assignments for Band Class 1 and Spreading Rate 1

Block Designator

Valid CDMA Frequency

Assignments

CDMA Channel Number

Transmit Frequency Band (MHz)


A(15 MHz)

Not ValidValid

Cond. Valid

0 - 2425 - 275276 - 299

1850.000 - 1851.2001851.250 - 1863.7501863.800 - 1864.950

1930.000 - 1931.2001931.250 - 1943.7501943.800 - 1944.950

D(5 MHz)

Cond. ValidValid

Cond. Valid

300 - 324325 - 375376 - 399

1865.000 - 1866.2001866.250 - 1868.7501868.800 - 1869.950

1945.000 - 1946.2001946.250 - 1948.7501948.800 - 1949.950

B(15 MHz)

Cond. ValidValid

Cond. Valid

400 - 424425 - 675676 - 699

1870.000 - 1871.2001871.250 - 1883.7501883.800 - 1884.950

1950.000 - 1951.2001951.250 - 1963.7501963.800 - 1964.950

E(5 MHz)

Cond. ValidValid

Cond. Valid

700 - 724725 - 775776 - 799

1885.000 - 1886.2001886.250 - 1888.7501888.800 - 1889.950

1965.000 - 1966.2001966.250 - 1968.7501968.800 - 1969.950

F(5 MHz)

Cond. ValidValid

Cond. Valid

800 - 824825 - 875876 - 899

1890.000 - 1891.2001891.250 - 1893.7501893.800 - 1894.950

1970.000 - 1971.2001971.250 - 1973.7501973.800 - 1974.950

C(15 MHz)

Cond. ValidValid

Cond. Valid

900 - 924925 - 11751176 - 1199

1895.000 - 1896.2001896.250 - 1908.7501908.800 - 1909.950

1975.000 - 1976.2001976.250 - 1988.7501988.800 - 1989.950

G(5 MHz)

Cond. ValidValid

Not Valid

1200-12241225-12751276-1299

1910.000 - 1911.2001911.250 - 1913.7501913.800 - 1914.950

1990.000 - 1991.2001991.250 - 1993.7501993.800 - 1994.950

Block Designator

Preferred Set Channel Numbers

A 25, 50, 75, 100, 125, 150, 175, 200, 225, 250, 275

D 325, 350, 375

B 425, 450, 475, 500, 525, 550, 575, 600, 625, 650, 675

E 725, 750, 775

F 825, 850, 875


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2.6 Dual-Mode vs. Dual-Band

Dual-mode subscriber units can support two air-interfaces using a common frequency band (i.e. CDMA and analog at 800 MHz). In a mixed digital and analog system, normally the registration request will be attempted first to the digital service then to the analog service. Dual-mode allows the digital service provider the option to redirect traffic to a different air-interface where resources are available, for capacity control or emergency hand down. Dual-mode phones also allow the subscriber unit to roam outside of its home network (assuming service is provided).

Dual-band subscriber units are designed to allow a subscriber to utilize two frequency spectrums, such as PCS frequency spectrum and the cellular bands. Handoffs are supported between CDMA providers of different bands (much like dual-mode) and also supported between CDMA, NAMPS and AMPS. With dual-mode phones, the service provider has the option to redirect the subscriber unit to a different air interface; however, dual-band providers redirect the subscriber unit to a different part of the frequency spectrum. An example for dual-mode would be a subscriber unit that is capable of operating on a CDMA 800 MHz system or could be redirected to an AMPS 800 MHz system, assuming resources are available. An example for dual-band operation would be a subscriber unit that is capable of operating on a CDMA PCS (1900 MHz) system and also being able to operate on an AMPS 800 MHz system.

The goal in developing dual-mode and dual-band subscriber units is to ease transition from one technology to a second (such as 800 MHz AMPS to 800 MHz CDMA), allow a single subscriber unit to roam outside of the provider’s service area, and eventually to have a subscriber unit which will work everywhere (domestic and international) thus providing "seamless" coverage. "Seamless" coverage does not necessarily imply a single service provider.

2.7 Spectrum Clearing

Spectrum clearing is a topic which is especially important to CDMA systems. The CDMA technology bases its capacity on a signal to noise balance (uplink and downlink). Adequate spectrum must be cleared to optimize a system to its greatest capacity. Although there are many approaches to testing the airways for clearance, it is advised that drive tests are performed (i.e. with a spectrum analyzer) to verify that the spectrum is clear, and/or locate possible spectrum violators.

For new spectrum allocations or for spectrum that is being reallocated for telecommunication systems (i.e. 3G spectrum allocations), spectrum measurements may be necessary to verify that the spectrum is clear of any previous users of the spectrum (see Section 2.8 for more information).

In the cellular bands, CDMA bandwidth is created by removing the appropriate number of AMPS

C 925, 950, 975, 1000, 1025, 1050, 1075, 1100, 1125, 1150, 1175

G 1225, 1250, 1275

Block Designator

Preferred Set Channel Numbers


2


channels. This should be done in cells within the core and transition zones. For the 1st CDMA carrier, 59 AMPS (30 kHz) channels would need to be cleared.

Cells for the transition (or guard) zone can be identified either by predictive RF propagation or actual noise floor measurements. The coverage area needing spectrum clearing will vary depending upon transmission signal strength, base station height, terrain variation, foliage, and reflection from buildings, hills or the atmosphere. The zone or area of cells to be cleared is related to the reuse distance needed to achieve acceptable C/I levels. The area needing clearing for CDMA may be reduced by controlling interference. Examples of how to control interference include: utilizing directional antennas, increasing or decreasing antenna heights and downtilts, careful adjustment of power applied to pilot and voice channels, or by using geographic elements for isolation.

Because all transmission equipment has the capacity to block or disrupt signalling, each country has laws governing transmission of signals. Many countries have adopted the United States Federal Communications Commission (FCC) standards. However, do NOT assume these standards are international. In the United States, Codes of Federal Regulations must be strictly adhered to. The U.S. government divides these codes into what are called "Titles". Each Title covers a specific topic. For instance, Title 7 covers Agriculture codes, Title 15 covers Commerce and Foreign Trade. The Telecommunication Code of Federal Regulations is listed in Title 47. Title 47 is subdivided into "Volumes" which contain "Parts" or chapters explicitly defining each code. The FCC World-Wide Web Page contains a search engine which can locate specific regulations. For example, regulations governing licensing and use of frequencies in the 806-824, 851-869, 896-901, and 935-940 MHz bands are located under CFR 47, Part 90.

Specific codes for PCS exist under CFR 47, Part 24. Great detail is given to rules and restrictions within CFR 47, Part 24. One rule for example, under paragraph 24.236 gives the field strength limits: "The predicted or measured median field strength at any location on the border of the PCS service area shall not exceed 47 dBuV/m unless the parties agree to a higher field strength."

Rules can be very specific. For instance, regulations are given for items such as antenna mast heights, antenna location, what maximum radiated power is allowed at each frequency, how to divide spectrum, who is responsible for clearing spectrum and what is the allotted time frame. It is important to clearly understand the regulations of the government for which the system will be deployed. Large fines can be assessed to the customer and/or Nokia Siemens Networks.

Although Federal Regulations take priority, each state and town/city may have additional codes or zoning regulations.

For non-U.S. regulations, please contact the governing agency of that country.

2.8 Background Noise Measurements

Capacity and coverage in CDMA systems (IS-95 & IS-2000) are, in part, a function of the background thermal and man-made interference noise levels. For the 1.23 MHz CDMA channel, the background thermal noise is approximately -113 dBm. Man-made interference includes automobile ignition noise and spurious emissions from radio and other electronic equipment.


2


The background man-made noise will vary from site to site depending on the number of interference sources and their proximity to the cell. In order to insure the optimal operation of each of the CDMA cell sites, Nokia Siemens Networks recommends that noise floor measurements are considered as a part of the site selection process for CDMA systems. These noise floor measurements can also be used to make adjustments to the noise margin parameter for a particular link budget analysis (see Section 4.2.1.4).

It is anticipated that CDMA systems may be deployed in the same geographical areas where another technology once occupied the current CDMA system’s spectrum. It is also possible for adjacent band signals from other systems that are in the same geographical areas with the CDMA system to cause interference with the CDMA system. As a result, noise floor measurements are also recommended to be used to identify any in-band or out-of-band interference sources. Once an interference source has been identified, an evaluation of the interference source can be performed to determine the impact to the CDMA system. If the impact is determined to be significant, then proper actions can be taken to reduce the source of interference to an appropriate level.

2.8.1 Suggested Measurement Method

Interference is random in nature, with amplitude and frequency varying over time. Some of the interference sources are thermal noise, environmental noise, and noise from other systems (i.e. AMPS/EAMPS, CDMA, GSM, iDEN, ANSI-136, point-to-point microwave, public safety, land mobile, private mobile, air-to-ground airphone service, etc.). Out of band sources can create interference through intermodulation (IM).

Due to the random nature of the background noise, Nokia Siemens Networks suggests that a data logging system be employed to measure the noise floor over some period of time. Statistical analysis of the collected data can then be performed to determine an average and cumulative distribution function of the noise floor rise. The cumulative distribution function indicates the amount of time the background noise rise exceeds some specified limit.

2.8.1.1 Test System Functional Description

A possible configuration of a noise floor test system is shown in Figure 2-21. The test measurement calibration point (cal point) is at the feedline entrance of a separate antenna or an unused port of the receiver multicoupler. The band-pass filter is used to attenuate out-of-band signals, which otherwise could create in-band intermodulation products. The low noise amplifiers are used to improve the system noise figure and provide enough gain to allow for the measurement of very low level signals. The step attenuator between the amplifiers is used to limit the system gain, again, to reduce the level of possible intermodulation products. The output of the final amplifier is then split using a two-way splitter. The two equal outputs of the splitter are used as inputs to two spectrum analyzers. Spectrum analyzer 1 operates in the manual mode. This spectrum analyzer is equipped with a tracking generator which is used for the system gain calibration. This spectrum analyzer is also used to make noise floor plots and to investigate the nature of interference as it appears on the screen. Spectrum analyzer 2 is under computer control. Measurement traces are collected with this spectrum analyzer and are stored to disk for later processing. Up to two spectrum analyzer traces per second can be recorded for the described system. The noise source is used to measure the


2


system noise figure. The measured system noise figure is used when processing the collected data into the desired cumulative distribution plots.

Figure 2-21: Suggested CDMA Noise Floor Measurement System

2.8.1.2 Test System Calibration

The test system gain and noise figure must be measured before data collection begins. The measured gain and noise figure are used to make adjustments to the collected data during the data analysis operation. The system gain is measured using the tracking generator provided in spectrum analyzer 1. The system noise figure is determined by first measuring the noise floor with the system Calibration Point (input) terminated in 50 ohms and then measuring the noise floor with the system Calibration Point connected to the calibrated noise source. The noise figure is then calculated as follows:

[EQ 2-2]

Where:ENR Equivalent noise ratio of the calibrated noise source (linear ratio)

xCalPoint

BandpassFilter

Plotter

Amplifier

NF = 1 dBG = 15 dBIPi = 4 dBm

Amplifier

NF = 2 dBG = 25 dBIPi = 0 dBm

+28 vdc

+28 vdc +28 vdc

StepAtten.

SpectrumAnalyzer 1

NF = 26 dBIP=16 dBm

w/tracker

SpectrumAnalyzer 2

NF = 26 dBIP = 21 dBm

PC

IEEE 488

ENR = 15 dB

Noise Source50 ohm

termination

NF 10ENR

PonPoff 1–-------------------- -------------------------

log=


2


Pon Noise floor measurement with the noise source connected to the system input (Watts)

Poff Noise floor measurement with the system input terminated in 50 ohms (Watts)

NF System noise figure (dB)

2.8.2 Test Procedures

If the CDMA system is deployed in an area where another technology currently exists, there are two proposed methods of co-existence. One method is to clear all co-channels from the other system within the CDMA band on a system wide basis. Another possibility is to only clear the co-channels from cells which are near the CDMA cells. Co-channels to the CDMA band are then reused at distant cells. Before noise floor testing can begin, co-channel clearing, per the chosen implementation plan, must be completed. This is necessary because co-channels within the CDMA band will appear as interference in the collected data.

After clearing the spectrum, preliminary tests should be run without band select filtering to identify uncleared channels, out of band large signals, and spurious emissions, and to measure any co-located technology antenna isolation. It is best to perform these tests during the busy hour as more uplink and downlink channels will be in use, and recorded by the tests.

Plot the system downlink band to identify possible uncleared co-channels, external sources of downlink interference, and to verify Tx-Rx isolation with any co-located cell sites.

Plot the uplink band to identify receive isolation with any co-located cell sites and to identify any possible sources of uplink interference.

Examine the plot of the adjacent system frequencies for out of band or spurious emissions from the other systems in the adjacent bands.

With a co-located cell site configuration, transmitter IM can be a source of interference with a duplexed antenna. If this configuration exists, all of the channels from the co-located site should be keyed up in the sector, and the spectrum should be scanned for IM and cross modulation products. This can effectively raise the noise floor 10 to 20 dB. It can be caused by connector breakdown in the RF path, and decreased isolation due to the duplexed configuration.

It may also be prudent to perform spot checks to identify possible interference causing conditions. If available, make a call on the competitors system and note the subscriber power level at the CDMA cell site. A maximum subscriber power level near the CDMA cell site may create interference issues.

Once the system has been cleared of analog co-channels, noise floor testing can proceed. For best results, the data should be logged at various times of the day and night at each cell site. This is necessary because varying traffic patterns throughout the day will effect the noise levels present at the cell site. It is recommended that at least 2000 traces be collected in each site.


2


2.8.3 Data Analysis

The collected data must be scaled to account for the measurement system gain, noise figure, and bandwidth before the statistical analysis is performed. Once the data is properly scaled, a statistics software package can be used to calculate the average noise floor rise and cumulative distribution functions. The noise floor rise cumulative distribution plots can then be used to make a judgement on the effect of background interference to CDMA performance at each cell site. Plots can also be produced which show the amplitude and frequency of interferers as a function of time. These plots can be used to help identify the source of interferers, which can lead to methods of interference reduction.

2.9 CDMA/LTE Sharing of XMIs

In Release C24/LTE R2, it will be possible to use XMIs to transmit both LTE and CDMA carriers. The LTE carrier will have a downlink frequency of 865 MHz with a bandwidth of 10 MHz. The CDMA carriers will be in the 870 MHz to 875 MHz range on channels 40 and possibly 81. There will be a third party DO carrier on channel 122 and possibly channel 81. The third party DO carrier transmitter shares the LTE/CDMA diversity antenna using an RF combining network. This requires a vacant CDMA channel between the third party DO and any carriers on the diversity antenna. The operator is assumed to provision carriers such that 1X, EV-DO, and LTE carriers do not occupy the same frequencies.

When an XMI is being used for both CDMA and LTE, the XMI bandwidth capability must be taken into account. The Tx bandwidth is 15 MHz and the Rx bandwidth is 11.4 MHz. For a frame to support 2 CDMA carriers, 1 LTE carrier with 2x2 MIMO, and share antennas with the third party DO carrier, the XMIs must be configured as shown in Figure 2-22.


2


Figure 2-22: CDMA/LTE Sharing of XMIs - 2 CDMA carriers, one LTE carrier

XMI 1 is configured to Transmit LTE and Transmit/Receive the CDMA carrier that is not adjacent to the LTE carrier (channel 81). Since the LTE and CDMA carriers do not fit into the 11.4 MHz bandwidth of the Receiver, the LTE carrier must be configured for "Transmit Only" and only the CDMA carrier will be received by this XMI.

XMI 2 is configured to Transmit/Receive LTE and to Transmit/Receive the CDMA carrier adjacent to the LTE carrier (channel 40). Since the LTE and CDMA carriers both fit into the 11.4 MHz bandwidth of the Receiver, this XMI handles the main and diversity paths for both carriers.

In this manner, a 2x2 LTE carrier and 2 CDMA carriers can be supported.

For the case where 1 CDMA carrier and 2 DO carriers must be supported, the XMIs must be configured as shown in Figure 2-23. XMI 1 must transmit and receive the CDMA carrier on channel 40, since the RF combining network on the diversity antenna must have an unoccupied

XMI 1

XMI 2

LTE Carrier

860 MHz 870 MHzChan 40(1X)

Chan 81 Chan 122(1X or DO) (DO)

Tx

Tx

XMI 1

XMI 2



Rx

Rx

Main & Div

Main & Div

LTE Carrier

LTE Carrier

LTE Carrier

830 MHz

875 MHz

15 MHz Tx Bandwidth

11.4 MHz Rx BW

11.4 MHz Rx BW

(Main ant.)

(Div. ant.)


2


channel between the DO carriers on channels 81 and 122, and the LTE carrier.

Figure 2-23: CDMA/LTE Sharing of XMIs - 1 CDMA carrier, 2 DO carriers

2.10 References

1 TIA/EIA/IS-95-A, Mobile Station - Base Station Compatibility Standard for Dual-Mode Wideband Spread Spectrum Cellular Systems, 1995, Sections 2.1.1.1, 2.2.1.1, 3.1.1.1, 3.2.1.1, 6.1.1.1, 6.2.1.1, 7.1.1.1, Tables 2.1.1.1-1, 6.1.1.1-1, 6.1.1.1-2.

2. ANSI J-STD-008, Personal Station-Base Station Compatibility Requirements for 1.8 to 2.0 GHz Code Division Multiple Access Personal Communications, March 24, 1995, Section 2.1.1.1, Tables 2.1.1.1-1, 2.1.1.1-2, 2.1.1.1-3 and 2.1.1.1-4.

XMI 1

XMI 2

LTE Carrier



Tx

Tx

XMI 1

XMI 2



Rx

Rx

Main & Div

Main & Div

LTE Carrier

LTE Carrier

LTE Carrier

830 MHz

875 MHz

15 MHz Tx Bandwidth

11.4 MHz Rx BW

11.4 MHz Rx BW

(Main ant.)

(Div. ant.)


2


3. CFR 47 (Telecommunications), Office of the Federal Register National Archives and Records Administration, October 1, 1997.

4. FCC Web Page (Wireless Telecommunications Bureau): http://www.fcc.gov/wtb/National Archives and Records Administration (CFR Search Engine): http://www.access.gpo.gov/nara/cfr/index.html

5. TIA/EIA/IS-2000-2, Physical Layer Standard for cdma2000 Spread Spectrum Systems

6. TIA/EIA TSB-84A, Licensed PCS to PCS Interference, Version 1.7, June 9, 1998

7. Dennis Schaeffer (Motorola), "Adjacent Channel Interference Impact In CDMA Systems", August 20, 1999

8. Asia Pacific Telecom Carrier Solutions Group (Motorola), "cdma2000 1X System Planning Guide", Version 0.1, November 7, 2000

9. Motorola, CDMA Uplink Noise Survey Procedure, Version 0.4.

10. Motorola, RF Logger User Guide, January 2000



Table of Contents

Chapter

3 CDMA Capacity
3
3 CDMA Capacity

3.1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 5

3.2 Reverse Link Pole Capacity Estimation . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 53.2.1 Data Rates. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 113.2.2 Median Eb/(No+Io) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 113.2.3 Voice or Data Activity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 123.2.4 Cell Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 133.2.5 Sectorization Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 153.2.6 Power Control. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 16

3.3 Reverse Link Soft Blocking Capacity Estimation . . . . . . . . . . . . . . . . . 3 - 183.3.1 Conventional Blocking Analysis . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 183.3.2 CDMA Soft Blocking Capacity Estimation . . . . . . . . . . . . . . . . . 3 - 18

3.3.2.1 Assumptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 193.3.2.2 Theoretical Analysis. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 193.3.2.3 Single Cell Case . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 223.3.2.4 Multiple Cell System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 23

3.4 Reverse Link Noise Rise Capacity Estimation . . . . . . . . . . . . . . . . . . . . 3 - 323.4.1 Reverse Link Noise Rise Capacity Limit . . . . . . . . . . . . . . . . . . . 3 - 323.4.2 Reverse Noise Rise Capacity Estimation . . . . . . . . . . . . . . . . . . . 3 - 333.4.3 Probability Factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 353.4.4 Reverse Link Noise Rise Capacity Estimation Examples . . . . . . 3 - 37

3.4.4.1 Example #1: Voice Only. . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 373.4.4.2 Example #2: Voice and Data Users . . . . . . . . . . . . . . . . . . . 3 - 38

3.4.5 Reverse Link Noise Rise Capacity Estimates for IS-2000 1X. . . 3 - 413.4.5.1 Noise Rise. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 413.4.5.2 F-factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 423.4.5.3 Average Eb/No . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 433.4.5.4 Eb/No Standard Deviation . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 433.4.5.5 Processing Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 443.4.5.6 Activity Factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 443.4.5.7 Traffic Mix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 453.4.5.8 Throughput Capacity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 453.4.5.9 IS-2000 1X Reverse Noise Rise Capacity Analysis Results 3 - 46

3.5 Forward Link Pole Capacity Estimation . . . . . . . . . . . . . . . . . . . . . . . . 3 - 523.5.1 Forward Link Load Factor Estimation . . . . . . . . . . . . . . . . . . . . . 3 - 52


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CDMA/CDMA2000 1X RF Planning Guide3 CDMA Capacity

3.5.2 Forward Link Pole Capacity Estimation . . . . . . . . . . . . . . . . . . . 3 - 53

3.6 Forward Link Fractional Power Capacity Estimation . . . . . . . . . . . . . 3 - 54

3.7 Forward Link Noise Rise Capacity Estimation . . . . . . . . . . . . . . . . . . . 3 - 573.7.1 Forward Link Noise Rise Capacity Limit . . . . . . . . . . . . . . . . . . 3 - 583.7.2 Forward Noise Rise Capacity Estimation. . . . . . . . . . . . . . . . . . . 3 - 593.7.3 Forward Link Noise Rise Capacity Estimation Examples . . . . . . 3 - 60

3.7.3.1 Example #1: Voice Only . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 613.7.3.2 Example #2: Voice and Data Users . . . . . . . . . . . . . . . . . . . . 3 - 62

3.7.4 Forward Link Noise Rise Capacity Estimates for IS-2000 1X . . 3 - 653.7.4.1 Noise Rise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 653.7.4.2 I-factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 663.7.4.3 Average Eb/No . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 663.7.4.4 Eb/No Standard Deviation . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 673.7.4.5 Processing Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 673.7.4.6 Activity Factor. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 683.7.4.7 Orthogonality Factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 693.7.4.8 Traffic Mix. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 693.7.4.9 Throughput Capacity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 693.7.4.10 IS-2000 1X Forward Noise Rise Capacity Analysis Results 3 - 70

3.8 Forward vs. Reverse Link Capacity Comparison . . . . . . . . . . . . . . . . . 3 - 75

3.9 EIA/TIA Specifications and RF Air Interface Limitations. . . . . . . . . . 3 - 793.9.1 IS-95 Forward Channel Structure. . . . . . . . . . . . . . . . . . . . . . . . . 3 - 793.9.2 IS-95 Reverse Channel Structure . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 803.9.3 IS-2000 1X Forward Channel Structure. . . . . . . . . . . . . . . . . . . . 3 - 81

3.9.3.1 IS-2000 Forward Channels (Nokia Siemens Networks Implementation) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 823.9.3.2 IS-2000 Forward Link Radio Configurations . . . . . . . . . . . . 3 - 853.9.3.3 IS-2000 Walsh Code Allocation . . . . . . . . . . . . . . . . . . . . . . 3 - 87

3.9.4 IS-2000 Reverse Channel Structure . . . . . . . . . . . . . . . . . . . . . . . 3 - 903.9.4.1 IS-2000 Reverse Channels (Nokia Siemens Networks Implementation) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 903.9.4.2 IS-2000 Reverse Link Radio Configurations . . . . . . . . . . . . 3 - 91

3.10 Handoffs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 933.10.1 Soft Handoff . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 933.10.2 Inter-CBSC Soft Handoff . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 943.10.3 Hard Handoffs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 94

3.10.3.1 Anchor Handoff . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 943.10.3.2 IS-95 to IS-2000 Hand-up . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 943.10.3.3 IS-2000 to IS-95 Hand-down . . . . . . . . . . . . . . . . . . . . . . . . 3 - 953.10.3.4 Packet Data Handoffs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 953.10.3.5 Inter-Carrier Hand-across . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 95

3.11 Budgetary Estimate of Sites for Capacity (Voice Only) . . . . . . . . . . . . 3 - 953.11.1 Required Parameters for Initial System Design . . . . . . . . . . . . . . 3 - 96

3.11.1.1 Busy Hour Call Attempts and Completions . . . . . . . . . . . . . 3 - 963.11.1.2 Average Holding Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 96


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3.11.1.3 Erlangs per Subscriber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 96

3.12 IS-95 and IS-2000 Simulations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 101

3.13 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 - 103


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3.1 Introduction

Capacity of a wireless network (for mobile or fixed subscribers) is defined as the number of users that a given cell site can support while maintaining a certain QoS/GOS criteria. With the introduction of various data related services (facilitated by IS-95B or IS-2000), the capacity of a given cell site can also be represented by the number of users along with the associated data throughput and a QoS criteria. The amount of RF spectrum available has a direct relationship on the capacity that can be provided. The air interfaces which make efficient use of the allocated spectrum will offer greater capacity. In AMPS or TDMA systems, blocking occurs when all voice frequencies or time slots are fully occupied by other users in the system. In Code Division Multiple Access (CDMA) systems, all users in the system share a common wideband spectrum over the time they are active.

Capacity of a CDMA system depends upon the amount of interference in the system. Additional users accessing the system will increase the system interference level. In order to maximize the capacity, steps need to be taken to minimize the total power transmitted so as to reduce the total interference in the system. An adjustment to this power will also make an adjustment to the capacity. Blocking in CDMA is defined to occur when the total interference density reaches a predetermined level above the background noise density. This is a soft blocking condition. The blocking probability can be relaxed by allowing the maximum tolerable interference level to increase.

In this chapter, several different capacity equations are provided which can be used to estimate the average cell site capacity under various conditions and assumptions. The capacity of a CDMA system is dependent upon the RF environment (i.e. path loss, delay spread, cell site layout, etc.). There is no single capacity number but a range of values over an environment. With the introduction of various data related services, the capacity will also depend upon the mixture of voice and data traffic models. A capacity equation analysis is a simplistic approach as it assumes uniform loading across all cells. However, in a live network, such a scenario would be rare. Thus, there is no simple formula that can calculate the actual capacity that a live CDMA cell site will be able to support. Though some equations will be provided to allow the approximation of the number of users and data throughput that could be supported, these equations will demonstrate that the capacity of a CDMA carrier varies with many factors. As a result, the capacity equations provided in this chapter should be used for budgetary purposes only. A more sophisticated CDMA simulation program, such as Nokia Siemens Networks’ Intelligent Design and Growth Planning (IDGP) for CDMA tool, should be used for a live CDMA system to model the forward and reverse links for thousands of subscribers in a realistic system environment with different voice and data traffic mixes. The Intelligent Design and Growth Planning(IDGP) for CDMA tool provides detailed simulations of both the forward and reverse links which produces a more accurate and realistic system capacity and coverage prediction.

3.2 Reverse Link Pole Capacity Estimation

In digital systems (i.e. IS-95A/B or IS-2000) the energy per bit needs to be a certain level above the total interference density in order to detect the transmitted bit. (Note: the following section can be applied towards both IS-95 and IS-2000 systems.) This is referred to as Eb/Io. Energy is


3


equivalent to power times time or to power divided by the rate. Therefore, the energy per bit can be expressed as the received power divided by the maximum bit rate:

[EQ 3-1]

Assuming:

• P denotes the received power from each subscriber at the base station antenna

• R denotes the data rate (9600 bps for Rate Set 1, 14400 bps for Rate Set 2)

• Power control is perfect

• Subscribers are transmitting just enough power to be received

• Uniform subscriber distribution

The total interference power density assuming N users, can be expressed as

[EQ 3-2]

Where:Bandwidth of the channel

Using Equation 3-1 and Equation 3-2, the energy per bit to the total interference density can be determined.

[EQ 3-3]

Solving for N yields:

[EQ 3-4]

It should be pointed out that some papers approximate N-1 with N.

The above equation is an ideal case or can be referred to as a first order capacity estimate. The capacity (N) can additionally be impacted by interference from other cell sites, the voice or data activity associated with the users, and the effect of thermal noise. Including these other factors into Equation 3-2 will yield:

EbPR---=

IoN 1–( )P

W---------------------=

W

Eb

Io------ P R⁄

N 1–( )PW

------------------------------------------- W R⁄

N 1–-------------= =

N 1–W R⁄Eb Io⁄-------------- N≈=


3


[EQ 3-5]

Where:Interference power density impacted by other cells, and the number of users with

an average voice or data activity rate

Ratio of out of cell (inter-cell) interference power to in cell (intra-cell) interference power. This factor is used to adjust the capacity of a single cell to account for the interference generated by other users in a multiple cell system.

Average voice or data activity factor

Thermal noise

Using this new value of Io, Equation 3-3 can be rewritten as follows:

[EQ 3-6]

The pole capacity is defined as the maximum capacity that can be achieved under a given set of conditions. At pole capacity, the rise over the thermal noise will approach infinity. This can be calculated from the power rise over thermal rise.

[EQ 3-7]

[EQ 3-8]

As the denominator in Equation 3-8 approaches zero, the power rise over thermal rise will approach infinity. Solving the denominator to be equal to zero will result in the maximum pole capacity.

[EQ 3-9]

Solving for the number of users (N) yields:

Io No+ρ N 1–( )P 1 f+( )

W----------------------------------------- No+=

Io

f

ρ

No

Eb

N( o Io )+----------------------- P R⁄

ρ N 1–( )P 1 f+( )W

----------------------------------------- No+------------------------------------------------------ W

R-----

PNoW-----------

ρ N 1–( ) 1 f+( )PNoW

----------------------------------------- 1+---------------------------------------------------⋅= =

PNoW-----------

Eb

N( o Io )+----------------------- R

W----- ρ N 1–( ) 1 f+( )P

NoW----------------------------------------- 1+⋅ ⋅=

PNoW-----------

Eb

N( o Io )+----------------------- R

W-----⋅

1Eb

N( o Io )+-----------------------–

RW----- ρ N 1–( ) 1 f+( )⋅ ⋅

-----------------------------------------------------------------------------------------=

Eb

N( o Io )+----------------------- R

W----- ρ N 1–( ) 1 f+( )⋅ ⋅ 1=


3


[EQ 3-10]

As mentioned previously, sometimes N-1 is approximated to be only N.

Two additional items can be taken into account to further refine the number of users that can be supported. They are a reduction factor due to imperfect power control and a factor to account for sectorization. In Equation 3-10, the f factor accounts for interference coming from other cell sites. The sectorization factor will account for the impact of interference leakage between sectors.

To approximate the reverse pole capacity point for CDMA (which can be applied to both IS-95 and IS-2000), the following equation can be used.

[EQ 3-11]

Where:Total received signal and noise power spectral density

Thermal noise power spectral density

Energy per bit

Ratio of Signal energy per bit to the sum of interference and noise adjusted for

imperfect power control

Bandwidth of the channel

Data rate

Processing gain

Ratio of out of cell (inter-cell) interference power to in cell (intra-cell) interference power. This factor is used to adjust the capacity of a single cell to account for the interference generated by other users in a multiple cell system.

Average voice or data activity factor

Sectorization gain

N 1–W R⁄

ρ 1 f+( )Eb

N( o Io )+-----------------------⋅

----------------------------------------------- N≈=

ReversePoleCapacity NW R⁄

Eb

No Io+-----------------

adjust

----------------------------------- 11 f+----------- 1

ρ--- Gs⋅ ⋅ ⋅= =

Io

No

Eb

EbN

oIo

+--------------------

adjust

W

R

W R⁄

f

ρ

Gs


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The adjusted Eb/(No+Io) requirement to account for imperfect power control (power control deviation) can be determined by:

[EQ 3-12]

Where:

Signal / (Interference plus noise) ratio requirement under perfect power control

Standard deviation in imperfect power control

Constant value equal to ln(10)/10

Some reverse link pole equations may use the term F, where F is defined as the ratio of in cell (intra-cell) interference power to the sum of out of cell (inter-cell) interference power and in cell (intra-cell) interference power. F is related to f by the following equation.

[EQ 3-13]

Substituting F into Equation 3-11 results in the following equation (which also can be applied to both IS-95 and IS-2000).

[EQ 3-14]

Assuming the following values for the various parameters, the reverse link pole capacity for an IS-95 Rate Set 2 site would be 19 users or roughly 12.3 Erlangs per sector (assuming an Erlang B model with 2% grade of service) for a three sector site (57 users per site). This value represents the pole capacity or the point at which no more users can be added without seriously degrading the quality of the system.

Bandwidth of the channel (only one CDMA Channel) 1228800 Hz

Data rate 14400 bps

Ratio of out of cell (inter-cell) interference power to in cell 0.7

Average voice or data activity factor 0.4

Eb

No Io+-----------------

adjust

Eb

No Io+----------------- e

βσe( )2 2⁄⋅=

Eb

N0 I0+-----------------

σe

β

FInCell

InCell OutCell+-------------------------------------------- 1

1OutCellInCell

---------------------+------------------------------ 1

1 f+-----------= = =

ReversePoleCapacity NW R⁄

Eb

No Io+-----------------

adjust

----------------------------------- F1ρ--- Gs⋅ ⋅ ⋅= =

W

R

f

ρ


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Sectorization gain per sector for a three sector site 2.4/3

Signal / (Interference plus noise) ratio requirement under

perfect power control 6.5 dB

Standard deviation in imperfect power control 2.5

Constant value ln(10)/10

[EQ 3-15]

[EQ 3-16]

Several variations of a reverse link capacity equation exist. The various equations may not be exactly the same as Equation 3-11 or Equation 3-14, but many, if not all, of the items within the equations will be represented: processing gain, Eb/(No+Io) (may also include a factor to account for imperfect power control or power control impact may be its own term), other cell interference, voice or data activity factor, and impact of sectorization. When discussing capacity, it is important to mention all of the factors which are being considered and the assumed value for each factor. For instance, 19 users shown above can easily turn into 32 users, if the calculation does not account for any inter-cell interference (f=0). The capacity results are also highly dependent upon the values that are used for the capacity equation. Even if the equations are similar, the values used may be different which leads to different capacity claims from different sources. Some values are more optimistic, thus leading to more users.

The Eb/(No+Io) performance parameter, used as an input to the equations provided above (Equation 3-11 or Equation 3-14), is usually specified for a particular data rate (along with other assumptions; i.e. flat fading, mobile environment with a 30 kmph worst case speed, 1% FER, diversity, and perfect decorrelation). Although the reverse pole capacity equations can be applied towards both IS-95 and IS-2000 systems, they are typically applied towards analyzing a system utilizing a single data rate. As such, they may be more appropriate in estimating the capacity of an IS-95 system, where it is common to support a single data rate (i.e. Rate Set 1 or Rate Set 2). For IS-2000 systems which utilize multiple data rates, the reverse pole capacity equations can be used to analyze the capacity of each individual data rate. They are not recommended to analyze a mixture of data rates, unless an appropriate average Eb/(No+Io) performance parameter can be produced to correlate with an associated average data rate.

Another point to be made is that these equations are for pole capacity. In designing a CDMA system, the system designer should not assume that the system pole capacity will be achieved. The system designer should plan that the reverse link capacity will not exceed 75% of the pole capacity. From the above example, this would correspond to about 14 users or 8.2 Erlangs. Note that this is

Gs

Eb

No Io+-----------------

σe

β

Eb

No Io+-----------------

adjust10

6.5 10⁄( )e

0.23 2.5⋅( )2 2⁄5.27 7.22dB==⋅=

ReversePoleCapacity N1228800( ) 14400⁄

107.22 10⁄( )

-------------------------------------------- 11 0.7+---------------- 1

0.4------- 2.4

3------- 19≈⋅ ⋅ ⋅= =


3


for the reverse link, the forward link may actually not allow this amount of Erlangs to be provided.

In analyzing Equation 3-11, the following relationships can be observed:

• The reverse pole capacity value is greater for the lower data rate vocoder (i.e. Rate Set 1 at 9600 bps will provide greater reverse link capacity than Rate Set 2 at 14400 bps).

• The reverse pole capacity value is increased if the Eb/(No+Io) requirement is reduced.

• The reverse pole capacity value is increased if the average voice or data activity is reduced.

• The reverse pole capacity value is increased if the inter-cell to intra-cell interference ratio is reduced.

• The reverse pole capacity value is increased if the sectorization gain can be increased (i.e. choosing antennas with better front to back ratios and also antennas that have a quick rolloff from their half power point to the back of the antenna).

• The reverse pole capacity value is increased if the power control standard deviation is reduced.

The following set of graphs demonstrates the six points just made. Only one of the parameter values was varied for each graph with the other parameter values being left to the values given in Equation 3-16. The intent of the graphs is to demonstrate the sensitivity a parameter value has on the capacity of site or system.

3.2.1 Data Rates

The capacity of a CDMA carrier is dependent upon the data rate being used. Referring to Equation 3-11, it can be seen that R (the data rate) has an inverse relationship to the reverse pole capacity. Figure 3-1 through Figure 3-5 will show curves for both Rate Set 1 at 9.6 kbps (which is the air interface data rate used for the 8 kbps vocoder) and Rate Set 2 at 14.4 kbps (which is the air interface data rate used for the 13 kbps vocoder).

3.2.2 Median Eb/(No+Io)

Figure 3-1 shows that lower values for Eb/(No+Io) result in more users being supported. BTS infrastructure enhancements that decrease the required Eb/(No+Io) value is one area Nokia Siemens Networks is researching to improve the capacity of the reverse link.


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Figure 3-1: Impact of Eb/(No+Io) on the Number of Users

For a mobile environment, a 7 to 7.5 dB Eb/(No+Io) value is deemed acceptable. For a fixed system, the Eb/(No+Io) requirement can be as low as 3 to 4 dB for some situations. Fixed units installed indoors with a whip antenna will require Eb/(No+Io) values similar to the mobile environment, whereas fixed units installed with outdoor directional antennas will require lower Eb/(No+Io) values. Further advancements in chipsets and the algorithms employed in those chipset may reduce the Eb/(No+Io) requirement and thus smaller values than these previously listed will be acceptable. For example, the values above are reasonable for an IS-95 site, but new chipsets are being used (i.e IS-95 EMAXX chipset and IS-2000 chipset) which improve upon the Eb/(No+Io) requirement. From the graph above, a 3 dB advantage of a fixed system over a mobile system will yield a pole capacity of approximately twice the number of users (considering just the impact of Eb/(No+Io)).

3.2.3 Voice or Data Activity

As a means to minimize interference, the transmission rate and power can be reduced when the voice or data activity is absent or lessened. This reduction in transmission rate or power reduces the average signal power of all users and thereby reduces the interference seen by each user. This following figure depicts that as the voice or data activity increases, fewer users can be supported.


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Figure 3-2: Impact of Voice or Data Activity on the Number of Users

The typical voice activity factor is 40%.

For some IS-2000 data services applications, a higher data rate coupled with a higher data activity factor may be required. From the results in Figure 3-2, it can be seen that both of these factors will reduce the capacity that can be supported by a CDMA carrier.

3.2.4 Cell Interference

The capacity of a cell depends on the total interference it receives from other cells. The level of power that is received at the base station from different sources is dependent upon the laws of propagation. The following figure shows that when the out of cell interference is increased with respect to the in cell interference that the capacity will degrade.

[EQ 3-17]fOutCellInCell

---------------------=


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Figure 3-3: Impact of Other Cell Interference on the Number of Users

The following table shows several f values that were obtained from simulations assuming a specific propagation model (path loss slope, standard deviation, and correlation).

Table 3-1: Samples of Various f Factors

Note: path loss slope converts to path loss dB/decade by multiplying the slope by a factor of 10

The terms of the propagation model correspond to the path loss slope, the shadowing standard deviation and the site to site correlation value. As shown by the above table, higher propagation exponents (the path loss slope) will reduce the f factor and lower exponents will increase the value of f.

For a system that is only comprised of a single cell (for example a fixed system in a remote area), there will be no out of cell interference and therefore the pole capacity will be higher. Similarly, cell sites positioned along a highway to provide only highway coverage will not see much

Path loss slope

Standard Deviation Correlation f Factor

4.0 6.5 0.9 0.434.0 8.0 0.5 0.553.5 6.5 0.5 0.693.5 8.0 0.5 0.763.5 10 0.1 1.68


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interference from other sites and therefore the f value will be lower for these sites than for a site in the middle of a cluster of sites. In addition the f value will be lower for systems that are only comprised of a few sites than for a system with many sites. As the number of sites increases there is a greater occurrence of interference from other cells which will increase the f value as shown by Equation 3-17.

3.2.5 Sectorization Gain

Sectorization gain can be somewhat of a misleading term. The sectorization gain is actually more of a reduction factor. For an omni site, the sectorization gain is one. For a sector site, one approach may be to multiply the resulting capacity of an omni site (or single sector) by the number of sectors for the sector site (i.e. a three sector site would support three times the number of users than an omni site and a six sector site would support six times the number of users than an omni site). This is not the case though. The additional sectors are considered to be other locations generating interference to the desired sector. The other cell interference factor accounts for just that, interference generated by other sites. The sectorization gain is the adjustment for the other sectors at the local site causing increased levels of interference. The reason it is referred to as a sectorization gain is that for a given physical site location, this site location is able to support many more users when it is sectorized than if it stayed omni.

The sectorization gain can be improved by selecting antennas which have a good front to back ratio and which also exhibit a quick rolloff past the half power points (3 dB down from main lobe). For instance, using a 90° antenna in place of a 120° antenna for a three sector site would decrease the amount of energy (interference) going into adjacent sectors, thus increasing the sectorization gain and thereby improving upon the number of users which could be supported. It is important to note that decreasing the horizontal beamwidth too much can also have a negative impact on the coverage (signal strength) within the cell site’s coverage area. As the sectorization gain increases, the number of users will increase (as seen from the graph in Figure 3-4).

The sectorization gain value which is commonly used is 0.8 per sector or 2.4 for a three sector site (0.8 time 3). This 0.8 sectorization gain can be thought of as a 1 dB impact to the capacity of the site due to other sectors interference.


3


Figure 3-4: Impact of Sectorization Gain on the Number of Users (3 Sector)

The above figure would apply only to a three sector site. The sectorization gain shown is for an entire site. For instance, a sectorization gain of 2.4 corresponds to 0.8 per each sector (= 2.4/3). For an omni site the sectorization gain would be 1. If the sectorization per sector for a six sector site is considered to be similar to that of a three sector site, then the sectorization gain for the site would be 6 times the per sector value (for instance, 6 * 0.8 = 4.8).

3.2.6 Power Control

Traffic capacity of CDMA systems is increased by implementing an appropriate power control scheme to equalize the performance of all subscribers in the system. The appropriate power control scheme reduces the interference to the other adjacent cells. The less interference generated in the spectrum, the more users the CDMA system can support. As previously mentioned, the inaccuracy in power control is roughly a log-normal distributed function. Under different path loss situations, the average required Eb/(No+Io) tends to fluctuate around the mean to maintain a desirable Frame Erasure Rate. The power control standard deviation varies according to the extent of fluctuations.


3


Figure 3-5: Impact of Imperfect Power Control on the Number of Users

This graph shows that improving the accuracy of power control can provide some increase to the number of users.

At relatively slow speeds or in static conditions (fixed), power control is effective in counteracting slow fades, whereas at high speeds, power control is not as effective in counteracting fast fading. At higher speeds, the effects of interleaving become increasingly beneficial.


3


3.3 Reverse Link Soft Blocking Capacity Estimation

3.3.1 Conventional Blocking Analysis

In AMPS and TDMA systems, voice/traffic channels are assigned to users as long as they are available. Given the required offered traffic, the Erlang B model is used to determine the number of traffic channels required to provide a predetermined grade of service. The Erlang B model is based upon a model of serving without queuing. In other words, all blocked calls are cleared. Traffic load is the product of call rate and call holding time. It is a dimensionless quantity measured in Erlangs. One Erlang is the traffic intensity of a traffic channel which is continuously occupied. Grade of service is a term used to quantify the extent to which congestion occurs in any trunking system and is typically expressed as the probability of finding blocking. Blocking in AMPS and TDMA is defined to occur when all voice frequencies (for AMPS) or time slots (for TDMA) have been assigned to other subscriber stations.

The values quoted for traffic load and grade of service for cellular systems are usually taken during the busy hour. Busy hour is defined as the continuous one-hour period in the day during which the highest average traffic density is experienced by the system. The Erlang B formula is given by:

[EQ 3-18]

Where:A is the offered traffic

C is the number of available servers

Assumptions of the Erlang “B” Model:

1. The number of potential users is infinite

2. Intervals between originations are random

3. Call set up time is negligible

3.3.2 CDMA Soft Blocking Capacity Estimation

Unlike the traditional analog design, balanced uplink and downlink cannot be achieved in CDMA because of the differences in waveform design on both links. Originally it was considered that the reverse link (subscriber to base) would usually be the capacity limiting path. However with the Rate Set 2 vocoder and other real world situations, the forward link (base to subscriber) may be the limiting path. With new higher data rate services being introduced (via IS-95B or IS-2000), it is

PBlocking

AC

C!------

AK

K!------

K 0=

C

------------------=


3


expected that the forward link will require higher data downloads than the reverse link. As a result, the forward link is also expected to be the limiting path from a capacity perspective. Even though the forward link may be the limiting factor of capacity for some systems, the reverse link capacity estimates provided in this document can still be used to approximate the capacity under the given assumptions and conditions. In many instances, the capacity analysis results of the reverse link can sometimes provide an adequate estimate. Simulations should be used (i.e. using IDGP) to obtain more accurate capacity estimations. For more detailed results, simulations can take into account many variable elements for which a general reverse or forward link capacity equation cannot adequately model (i.e. non uniform traffic and speed distributions, non uniform cell site layouts, propagation characteristics for a specific area, multiple subscriber classes with various call models, combined forward and reverse link analysis, etc.).

Soft blocking in CDMA systems is defined to occur when the total collection of users both within the serving cell/sector and in other neighbor cells introduce an amount of interference density so great that it exceeds the background noise spectral density by a predefined amount. Under the assumption that the system is not hardware limited, the following analysis applies this soft blocking concept to calculate the Erlang capacity of a CDMA system. The concept of soft blocking will be explained in details in the following paragraphs.

3.3.2.1 Assumptions

1. The number of active calls is a Poisson random variable with mean ( )

2. Each user is active with probability and inactive with probability (1- )

3. Each user’s required energy per bit-to-interference density ratio (Eb/Io) is varied according to propagation conditions to achieve the specified Frame Erasure Rate (FER). The FER is usually taken as 1% (0.01) to provide satisfactory transmission.

4. All the sectors have the same number of users.

5. The users are uniformly distributed over each sector.

3.3.2.2 Theoretical Analysis

In mathematical form, the definition of blocking can be restated as follows:

Interference from the + Interference from + Thermal Noise = Total Interference serving cell other cells

Blocking occurs when

[EQ 3-19]

λμ---

ρ ρ

νi

i 1=

k

EbiR vi j( )Ebi j( )R N0W I0W>+

i 1=

k

j

othercells

+


3


Where:k is the number of simultaneous users per sector. By assumption [1], k is a Poisson

random variable with mean which is the offered traffic

W is the spread spectrum bandwidth allocated to a CDMA channel

R is the data rate

Eb is energy per bit

No is the background thermal noise density

Io is the total allowable interference density

is the voice or data activity and is a binomial random variable with = Pr ( =1), which is the gate on probability.

The voice or data activity factor ( ) is defined as:

= Probability ( =1) [EQ 3-20]

Defining = Eb/Io, which is known as the Bit Energy to Interference Density Ratio, and dividing by IoR, the inequality [Equation 3-19] can be written as follows:

[EQ 3-21]

Where:W/R is known as the processing gain

is the predefined threshold

Hence, the probability of blocking for CDMA is defined as the probability that the above condition holds true.

Pblocking = Probability {Z = } [EQ 3-22]

Notice that the blocking probability for CDMA is determined by the system Eb/Io performance, voice or data activity factor, the spread spectrum bandwidth, the data rate, and the maximum

λμ---

ν ρ ν

ρ

ρ ν

ε

νi

i 1=

k

εi vi j( )εi j( )i 1=

k

j

othercells

+ 1 η–( )> WR-----⋅

ηNo

Io------=

νi

i 1=

k

εi vi j( )εi j( )i 1=

k

j

othercells

+ 1 η–( )> WR-----⋅


3


allowable interference level. The probability of blocking can be relaxed by allowing the maximum tolerable interference level (Io/No) to increase. In this case, the system is forced to accommodate more simultaneous users by degrading its service quality. This phenomenon is called “soft blocking”. The threshold value for the maximum allowable interference shall be defined in the call processing software by the operator.

To evaluate the blocking probability, the distribution of Z has to be determined which, in turn, depends on the following random variables: voice or data activity ( ), bit energy to interference ratio ( ), the total number of users in the sector (Ns), and the number of active users per sector (k).

The voice or data activity ( ), is a binomial random variable with = Pr ( =1), which is the gate on probability. The distribution is given by:

P( =k) = [EQ 3-23]

The distribution of k is Poisson and is given by:

Pk = [EQ 3-24]

Where:

and are the arrival and service rates and is the offered traffic

The distribution of Eb/No depends on the power control mechanism in the system. Power control allows the system to equalize the transmit power of all subscribers within the system. In a trial test, the Eb/No performance was measured with a fixed system Frame Erasure Rate (FER) for a fully loaded CDMA cell. The data showed the overall Eb/No was a log-normal distribution. Hence the distribution of can be written as:

[EQ 3-25]

Where:x is a Gaussian Random Variable with mean m and standard deviation

The first and second moment of are given by:

E( ) = E[ ] = [EQ 3-26]

νε

ν ρ ν

νNs 1–

k ρk

1 ρ–( )Ns k– 1–⋅ ⋅

λμ--- k

k!----------- exp

λ–μ

------ ⋅

λ μ λμ---

ε

ε 10x 10⁄

=

σ

ε

ε exp βx( ) expβσ( )2

2-------------- exp βm( )⋅


3


E( ) = E[ ] = [EQ 3-27]

Where:

β =

3.3.2.3 Single Cell Case

For the single cell case, the second summation term in Equation 3-22 is zero (i.e. no interference for other cells). Since Z is the sum of k random variables, where k is the number of simultaneous users in the system, the Central Limit Theorem can be applied for the approximation for Z. The central limit theorem states that the probability density function for the sum of a number of independent random variables with arbitrary one-dimensional probability density function approaches a Gaussian Distribution. Hence the probability of blocking can be rewritten as:

Probability of Blocking = [EQ 3-28]

Where:E( ) is the expected value

STD( ) is the standard deviation

A = and Q(x) =

The expected value and standard deviation of can be computed as follows. Since Z is the sum of k random variables and k is a Poisson random variable;

Let =

E( )= E(k) E( ) = [EQ 3-29]

ε2 exp 2βx( ) exp 2 βσ( )2[ ] exp 2βm( )⋅

ln 10( )10

----------------

QA E Z( )–

STD Z( )----------------------

ZZ

exp βm( )----------------------=

W R⁄exp βm( )---------------------- 1 η–( )⋅ 1

2π---------- e

x

∞

xpλ2–2

-------- λd⋅

Z

ε εexp βm( )---------------------------

Z γελμ--- ρ exp

βσ( )2

2-------------- ⋅ ⋅


3


VAR( ) = E(k) VAR ( ) + VAR(k) [E( )] 2

= VAR(k) [E( )] 2

= E(k) [E( )] 2

= [E( )] E[ ]

VAR( ) = [EQ 3-30]

STD( ) = [EQ 3-31]

Thus, the probability of blocking for a CDMA single cell system can be formulated as in Equation 3-32.


Although the Single Cell Probability of Blocking equation (Equation 3-32) can be applied towards both IS-95 and IS-2000 systems, it is typically applied towards analyzing a system utilizing a single data rate. As such, it may be more appropriate in estimating the capacity of an IS-95 system, where it is common to support a single data rate (i.e. Rate Set 1 or Rate Set 2). For IS-2000 systems which utilize multiple data rates, the Single Cell Probability of Blocking equation can be used to analyze the capacity of an individual data rate. It is not recommended to analyze a mixture of data rates. Section 3.4 will introduce an analytical approach more suitable for systems serving multiple data rates.

3.3.2.4 Multiple Cell System

In a multiple-cell system the interference created by users in the serving cell and cells other than the serving cell needs to be considered. The path loss characteristics and the overhead capacity for soft handoffs need to be taken into account.

3.3.2.4.1 Path Loss Characteristics

Power control is crucial to CDMA system performance. Assuming that the path loss depends only on the subscriber-to-base distance, the subscribers will be power controlled by the nearest cell. The generally accepted theoretical path loss model is to introduce an attenuation which is the product of, the subscriber-to-base distance to the power , and, a log-normal random variable with zero mean and dB standard deviation.

Z γε γε

γε

γε

λμ--- γ2 ε

2

Zλμ--- ρ exp 2 βσ( )2[ ]⋅ ⋅

Zλμ--- ρ exp 2 βσ( )2[ ]⋅ ⋅

Q

W R⁄exp βm( )---------------------- 1 η–( )⋅ λ

μ--- ρ exp

βσ( )2

2-------------- ⋅ ⋅

–

λμ--- ρ exp 2 βσ( )2[ ]⋅ ⋅

-----------------------------------------------------------------------------------------------------------

αδ


3


In Mathematical form, the path loss between the subscriber and the cell site is proportional to

[EQ 3-33]

Where:r is distance from subscriber to cell site

is a Gaussian random variable with standard deviation and zero mean

The path loss can be expressed as

[EQ 3-34]

Where:r and are the base-subscriber distance and the reference distance respectively

When plotting the signal strengths at a given radio path distance, the deviation from the local mean values is approximately 8 dB. This standard deviation of 8 dB is roughly true in many different areas. The path loss curves can be obtained by collecting data from different drive runs in different environments. As long as the subscriber-to-base distance for each run is the same, the signal strength data measured at that particular subscriber-to-base distance can be used for determining the local mean values for the path loss at that distance.

Measurements of path loss have been made in several major cities. Some of the typical values are tabulated as shown in Table 3-2.

Table 3-2: Propagation Path Loss in Different Areasa

a. William C. Y. Lee, "Mobile Cellular Telecommunications Systems", McGraw-Hill Book Company, Sec-ond Edition 1995, figure 4.3, p. 110.

Propagation Area1 Mile Intercept Point (Po)

in dBmPath Loss Slope (γ)

dB/decade

Free Space -45.0 20.0

Open Area -49.0 43.5

Suburban -61.7 38.4

Philadelphia -70.0 36.8

Newark -64.0 43.1

New York City -77.0 48.0

Tokyo, Japan -84.0 30.5

10

ξ10------

rα–⋅

ξ δ

PL α rr0---- log⋅=

r0


3


Since the main concern about propagation at far distances is for coverage purposes, path loss measurements typically use a 1 mile (or 1 km) intercept point as a starting point for path loss curves. This also tends to eliminate some of the near-field effects of near-by surroundings and vertical beam width shadowing. Although different areas may have different path loss slopes, Table 3-2 also shows that an area-to-area prediction is represented by two parameters, the 1 mile intercept point (Po, the power received at a distance of 1 mile from the transmitter) and the path loss slope (γ). Differences in area-to-area prediction curves are primarily due to the differences in man-made structures. When the base station is located in a city environment, then the 1 mile intercept signal level could be very low, but the slope is flattened out, as shown by the Tokyo data. When the base station is located outside the city, the intercept signal level could be much higher, but the slope is larger, as shown by the Newark data. Due to differences in structure density (average separation between buildings), the 1 mile intercept could be high or low, with the path loss slope still at a typical level of about 40 dB/dec (i.e. compare data of open area to Newark).

3.3.2.4.2 Interference from Other Cells

The normalized interference density from other cells can be written as:

Jo = Ioc / Io = Total Interference from other cells / IoW

Jo = [EQ 3-35]

Where:rm Distance from any subscriber to its own cell not power controlled by the serving

cell

r0 Distance from any subscriber to the serving cell not power controlled by the serving cell

Path loss exponent

Voice or data activity

Ioc Other cells interference density

Io Total allowable interference density

W Spread bandwidth

Eb*R Bit energy * data rate, which is the received power at the base station for any user, assuming power control is applied

Defines the path loss characteristics and is Gaussian random variables with zero means and standard deviation of

User density = 2 * users per sector / *sectorization gain

rm

r0-----

γ10

ξ10------

∅ ξr0

rm-----,

EbRνκIoW

-----------------

dAallcells

γ

ν

ξσ

κ 3


3


= 1, if

= 0, otherwise

By calculating the expected value and standard deviation Jo and , the probability of blocking for a CDMA multiple cell system can be formulated as follows.

E(J0) =

E(J0) = [EQ 3-36]

VAR(J0) =

VAR(J0) = [EQ 3-37]

The following figure provides the values of the numerical integration of the integral and versus various log-normal path loss slopes with a standard deviation of 8 dB.

Figure 3-6: Values of the Integral and with Various Path Loss Slope

∅ ξr0

rm-----,

rm

r0-----

γ10

ξ10------

1≤

z

E ε( ) λμ--- ρ

rm

r0-----

αexp βδ( )2[ ] 1 Q

10α

2δ2-------------

r0

rm----- β 2δ2– log–

dAallcell

E ε( ) λμ--- ρ I α δ,( )[ ]⋅

E ε2( ) λμ--- ρ

rm

r0-----

2αexp βδ( )2[ ] 1 Q

20α

2δ2-------------

r0

rm----- β 2δ2– log–

dAallcell

E ε2( ) λμ--- ρ I 2α δ,( )[ ]⋅

I α δ,( )I 2α δ,( )

I α δ,( ) I 2α δ,( )

50454035300.0

0.5

1.0

1.5

I(alpha,sigma=8dB)I(2alpha,sigma=8dB)

Path Loss (dB/dec)

Val

ues

of

the

Inte

gra

ls


3


Combining [EQ 3-36]and [EQ 3-37] with [EQ 3-29] and [EQ 3-30], the moments of the total normalized interference variable including the interference from outer cells is obtained.

[EQ 3-38]

[EQ 3-39]

Pblocking with outer cell interference = [EQ 3-40]

Where:

[EQ 3-41]

[EQ 3-42]


Note: A Complementary Error Function Q(x) table is provided in Appendix IV.

Using Equation 3-32 and Equation 3-43, the probability of blocking is plotted against the Erlang capacity per CDMA sector in different situations. A list of parameters is included at the bottom of each plot.

Z

E Z( ) λμ--- ρ exp

βσ( )2

2-------------- 1 I α δ r, ,( )+[ ]⋅=

STD Z( ) λμ--- ρ exp 2 βσ( )2[ ] 1 I 2α δ r, ,( )+[ ]⋅=

QA E Z( )–

STD Z( )----------------------

AW R⁄

exp βm( )----------------------= 1 η–( )⋅

Q x( ) 1

2π----------= e

x

∞

xpλ2–2

-------- λd⋅

Q

W R⁄exp βm( )---------------------- 1 η–( )⋅ λ

μ--- ρ exp

βσ( )2

2-------------- 1 I α δ r, ,( )+[ ]⋅

–

λμ--- ρ exp 2 βσ( )2[ ] 1 I 2α δ r, ,( )+[ ]⋅

-------------------------------------------------------------------------------------------------------------------------------------------


3


Figure 3-7: Probability of Blocking vs. Erlangs per CDMA Sector with Various Path Loss Slope Values with Rate Set 1 Vocoder

Note: The figure above is for demonstration purposes, as it is only valid for the assumptions applied and for the following parameters:

Parameters:• Mean Eb/No = 7 dB

• Pwr Ctrl Std Dev = 2.5 dB

• Voice or Data Activity Factor = 0.4

• Spread Bandwidth = 1.23 MHz

• Data Rate = 9600 bps (Rate Set 1)

• Total Interference Density to Background Noise Level (Io/No) = 10 dB

• Shadowing Standard Dev = 8 dB


3


Figure 3-8: Probability of Blocking vs. Erlangs per CDMA Sector with Various Power Control Standard Deviations with Rate Set 1 Vocoder







• Path Loss Slope = 40 dB/dec

• Shadowing Std Dev = 8 dB


3


Figure 3-9: Probability of Blocking vs. Erlangs per CDMA Sector with Various Path Loss Slope Values with Rate Set 2 Vocoder



• Pwr Ctrl Std Dev = 2.5 dB





• Shadowing Standard Dev = 8 dB


3


Figure 3-10: Probability of Blocking vs. Erlangs per CDMA Sector with Various Power Control Standard Deviations with Rate Set 2 Vocoder







• Path Loss Slope = 40 dB/dec

• Shadowing Std Dev = 8 dB

The Multiple Cell Probability of Blocking equation shown in Equation 3-43 can be applied towards both IS-95 and IS-2000 systems. Since, it is typically applied towards analyzing a system utilizing a single data rate, it may be more appropriate in estimating the capacity of an IS-95 system, where it is common to support a single data rate (i.e. Rate Set 1 or Rate Set 2). For IS-2000 systems which utilize multiple data rates, the Multiple Cell Probability of Blocking equation can be used to analyze the capacity of an individual data rate. It is not recommended in analyzing a mixture of data rates. Section 3.4 provides an approach more suitable for systems serving multiple data rates.


3


3.4 Reverse Link Noise Rise Capacity Estimation

The amount of noise rise (interference) that can be tolerated by the CDMA base station will place a limit upon how many users can be supported by the reverse link. As the number of users served by the reverse link is increased, the level of noise rise seen by the base station will also be increased. The cell capacity is determined by calculating the number of users required to produce a maximum accepted noise rise.

This section provides a method of estimating the noise rise for a particular user type. The estimating approach will also allow the calculation of the total noise rise for multiple user types. As a result, the noise rise estimation approach provided in this section is better suited to estimate the capacity of a system which utilizes multiple user types (i.e. multiple data rates). Although this capacity estimation approach can be applied towards both IS-95 and IS-2000 systems, it may be more appropriate in estimating the capacity of an IS-2000 system, where it is more common to support different user type profiles utilizing different data rates.

For IS-2000 systems, it is important to note that the capacity estimation calculation provided in this section does not account for the dynamic resource allocation capabilities of an IS-2000 1X packet data user. Within the IS-2000 1X infrastructure, the subscriber will be assigned supplemental channel resources based upon several criteria (e.g. the demand requirements for the amount of data to be transmitted, RF capacity availability, Walsh code resource availability, etc.). The allocation of these IS-2000 1X supplemental channel resources are also dynamically adjusted throughout the duration of the packet data call. The capacity estimation calculation provided in this section treats a packet data user more like a circuit data user. The capacity formulas provided imply a fixed resource allocation where there are X users at 9.6 kbps, Y users at 19.2 kbps, Z users at 38.4 kbps, etc. As a result, the capacity obtained from the capacity estimation approach will differ from that of an actual IS-2000 1X system. For a more accurate estimation of packet data services, it is recommended to utilize a simulation tool which simulates the dynamic resource allocation capabilities of an IS-2000 1X system. The time-sliced simulation function of the NetPlan tool can be used for this purpose. However, NetPlan has reached end of life and is no longer supported. IDGP also performs capacity analysis using non-time sliced simulation. IDGP's load point analysis feature, shows the increase in noise rise as erlangs are added to the system. See Section 3.12 for more information on the simulation capabilities of the IDGP tool.

3.4.1 Reverse Link Noise Rise Capacity Limit

The reverse link pole capacity is considered to be the point where an additional user will cause the noise rise within the cell to increase exponentially. This will create an unstable situation where user connections may be lost and the network grade of service will be severely degraded. The reverse link noise rise pole capacity can be represented by the following equation:

[EQ 3-44]

Where:X Percent of reverse link pole capacity, traffic loading factor

Z 10 Log10× 11 X–------------ 10– Log10× 1 X–( )==


3


Z Noise Rise (dB)

Refer to Section 4.2.2.1 for a derivation of Equation 3-44. A graph of the reverse noise rise pole capacity equation (Equation 3-44) is shown in Figure 3-11.

Figure 3-11: Rise versus Percent of Pole Capacity

In order to estimate the capacity from a number of users perspective, a reverse noise rise capacity limit must be selected. For CDMA RF system designs (for both IS-95A/B and IS-2000), a peak noise rise of 10 dB is recommended to be the maximum that a system should tolerate. The average noise rise would be several dB below this peak value. It is important to note that the 10 dB noise rise limit is a peak value which is associated with a certain probability factor (see Equation 3-48 and Section 3.4.3). The recommended probability factors associated with the 10 dB peak noise rise recommendation are as follows.

• 10 dB noise rise with a 90% probability factor (for aggressive capacity results)• 10 dB noise rise with a 95% probability factor (for moderate capacity results)• 10 dB noise rise with a 98% probability factor (for conservative capacity results)

Although the above recommendation provides some flexibility in selecting a probability factor, the 10 dB noise rise with a 95% probability factor is the typical limit that is normally recommended.

3.4.2 Reverse Noise Rise Capacity Estimation

To approximate the number of users that could be supported by a site while staying below a desired noise rise limit, the following reverse link capacity equations can be utilized.

A multi-service traffic loading factor, X, can be expressed as follows:

0

5

10

15

20

0% 25% 50% 75% 100%

Loading Factor, X

Inte

rfer

ence

Ris

e, Z

(d

B)


3


[EQ 3-45]

The mean value for the multi-service traffic loading factor, X, is expressed as:

[EQ 3-46]

The variance for the multi-service traffic loading factor, X, is expressed as:

[EQ 3-47]

The following equation provides the distribution of the noise rise, Z, for the multi-service traffic loading factor, X:

[EQ 3-48]

Where:M Number of different service-types

Traffic load of the mth service-type (in Erlangs)

The energy-per-bit to total-interference-density target of the mth service-type

LN(10)/10

Average Eb/No (dB) of the mth service-type

Eb/No standard deviation, in dB of the mth service-type (to account for

inaccuracies in power control)

Activity Factor of the mth service-type

Mean Square of Activity Factor of the mth service-type (variance = 0.1)

F A measure of the in-cell to total interference density (own cell plus other cell)

Processing gain (Bandwidth/Information rate) of the mth service

X L m( )

m 1=

M

ν m( )

F PG m( )×------------------------×

Eb m( )NT

-------------×=

E X[ ] L m( )

m 1=

M

ν m( )

F PG m( )×------------------------× βε m( )

βσ m( )( )2

2---------------------+exp×=

Var X( ) L m( )

m 1=

M

ψ m( ) ν m( )( )+

2

F PG m( )( )2×-----------------------------------× 2βε m( ) 2 βσ m( )( )2

+[ ]exp×=

Z 10– Log10× 1 Pa Var X( ) E X[ ]–×–( )=

L m( )

Eb m( )NT

-------------

β

ε m( )

σ m( )

ν m( )

ψ m( )

PG m( )


3


Z Interference rise (expressed in dB)

Pa Probability factor (inverse of the standard normal cumulative distribution) with a distribution having a mean of 0 and a standard deviation of 1 (see Figure 3-12)

Briefly looking at Equation 3-46 and Equation 3-47, the average and variance of the loading factor will increase as the number of users increases. Additionally, as the average and variance values increase, so does Z, as reflected by Equation 3-48.

In a scenario with multiple services, the equations are a bit more complex than for a single service. Basically, an average and variance needs to be determined for each service offered. The net rise, Z, will need to account for all of the users being handled by each service.

3.4.3 Probability Factor

The probability factor (Pa) in Equation 3-48 is used to calculate a percentile noise rise. The percentile noise rise is used as the interference margin within the RF link budget calculation of cell range. Therefore, scenarios with different traffic mixes and rise probabilities but with a constant percentile noise rise will all maintain the same cell range. However, the mean noise rise and cell capacity (throughput and Erlangs) will vary depending upon the mix of the different services for the given scenario.

The probability factor is calculated as the inverse of the standard normal cumulative distribution with a mean of 0 and a standard deviation of 1. Figure 3-12 shows the relationship of the probability factor with the Probability Density Function (PDF) and the Cumulative Distribution Function (CDF) for a standard normal distribution with a mean of 0 and a standard deviation of 1.

Figure 3-12: Standard Normal Distribution

0

0.2

0.4

0.6

0.8

1

-3 -2 -1 0 1 2 3

Probability Factor (Pa)

Dis

trib

uti

on

50%ile

75%il

85%ile95%ile

98%ile

CDFPDF


3


Table 3-3 provides the probability factor values for some common percentile probability percentages.

The rise curves in Figure 3-13 show the 50th (average) and 95th percentile noise rise against cell loading in terms of the number of users. It can be seen that the 95th percentile noise rise curve rises faster than the 50th percentile (average) noise rise curve and at the 95th percentile noise rise of 10 dB for the example provided in Figure 3-13 below, the 50th percentile (average) noise rise is approximately 5 dB. The relationship between a given percentile rise curve and the average rise curve will be dependent upon what percentile is being represented and also upon the particular call model traffic mix.

Figure 3-13: Rise and Radius versus Loading Example

Note: The figure above is for demonstration purposes, as it is only valid for the assumptions applied for the scenario portrayed (i.e. 30 kmph, 100% voice, etc.).

Figure 3-13 also shows how the relative cell range decreases with the increasing number of users.

Table 3-3: Probability Factors

PercentileProbability

ProbabilityFactor (Pa)

50% 075% 0.674585% 1.036490% 1.281695% 1.644898% 2.0537

0

5

10

15

20

25

0 5 10 15 20 25 30 35

Users

Ris

e (d

B)

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1R

elat

ive

Rad

ius

% of Radius

95%ile Rise

50%ile (Avg.) Rise


3


A relative range impact of 50% corresponding to a 10 dB 95th percentile noise rise can be observed from this figure.

3.4.4 Reverse Link Noise Rise Capacity Estimation Examples

The following section provides two examples of how to use the reverse link noise rise capacity estimation equations provided in Section 3.4.2. The first example estimates the noise rise for a single service type of traffic load of voice users only. The second example provides the calculations required to estimate the noise rise for a multiple service type of traffic load with a mixture of voice and data users.

3.4.4.1 Example #1: Voice Only

The following example calculates the noise rise for, on average, 20 IS-2000 1X voice users at 9600 bps, in a 3-sectored system with a 95% probability factor. Additional assumptions are provided below.

Traffic Load:20 Voice users (average) at 9600 bps

General Assumptions:• 0.45 = F-factor (3 sector cell site assumed)• 1.64 = probability factor for 95% (Pa)• 0.23 = Beta value LN(10)/10 ( )

Traffic Load Assumptions: Voice @ 9600 bps• 20 = number of average users at 9600 bps ( )• 21.1 dB = Processing gain ( ) or 1228800/9600 = 128 linear• 3.6 dB = average Eb/No for 1% FER with vehicular fading at 30 kmph ( )• 2.5 dB = Eb/No standard deviation ( )• 0.713 = voice activity factor ( )• 0.1 = mean square of activity factor ( )

The first step is to calculate the mean value of the traffic loading factor, X, for the 20 average voice users by using Equation 3-46 (repeated below for reference).

Using the input variables from the assumptions above, E[X] is calculated as follows:

β

L m( )PG m( )

ε m( )σ m( )

ν m( )ψ m( )

E X[ ] L m( )

m 1=

M

ν m( )

F PG m( )×------------------------× βε m( )

βσ m( )( )2

2---------------------+exp×=

E X[ ] 200.713

0.45 128×-------------------------× 0.230259( ) 3.6( )⋅ 0.230259( ) 2.5( )⋅( )2

2---------------------------------------------------+exp×=


3


The next step is to calculate the variance for the traffic loading factor, X, for the 20 average voice users by using Equation 3-47 (repeated below for reference).

The final step is to calculate the noise rise, Z, for the 20 average voice users by using Equation 3-48 (repeated below for reference).

dB

3.4.4.2 Example #2: Voice and Data Users

The following example calculates the noise rise for a multiple service type traffic load environment consisting of, on average, 6 IS-2000 1X voice users at 9600 bps, 3 IS-2000 1X data users at 19200 bps, and 1 data user at 38400 bps. Additional assumptions are provided below.

Traffic Load:6 Voice users (average) at 9600 bps3 Data users (average) at 19200 bps1 Data user (average) at 38400 bps

General Assumptions:• 0.45 = F-factor (3 sector cell site assumed)• 1.64 = probability factor for 95% (Pa)• 0.23 = Beta value LN(10)/10 ( )

E X[ ] 0.247569 0.994617[ ]exp× 0.247569 2.703690 0.669351=×==

Var X( ) L m( )

m 1=

M

ψ m( ) ν m( )( )+

2

F PG m( )( )2×-----------------------------------× 2βε m( ) 2 βσ m( )( )2

+[ ]exp×=

Var X( ) 200.1 0.713( )+

2

0.45 128( )2×----------------------------------× 2 0.230259( ) 3.6( )⋅ 2 0.230259( ) 2.5( )⋅( )⋅ 2

+[ ]exp×=

Var X( ) 0.001650 2.320605[ ]exp× 0.001650 10.181831× 0.016803===

Z 10– Log10× 1 Pa Var X( ) E X[ ]–×–( )=

Z 10– Log10× 1 1.644848 0.016803 0.669351–×–( )=

Z 10– Log10× 0.117432( ) 9.30==

β


3


Traffic Load Assumptions: Voice @ 9600 bps• 6 = number of average users at 9600 bps ( )• 21.1 dB = Processing gain ( ) or 1228800/9600 = 128 linear• 3.6 dB = average Eb/No for 1% FER with vehicular fading at 30 kmph ( )• 2.5 dB = Eb/No standard deviation ( )• 0.713 = voice activity factor ( )• 0.1 = mean square of activity factor ( )

Traffic Load Assumptions: Data @ 19200 bps• 3 = number of average users at 19200 bps ( )• 18.1 dB = Processing gain ( ) or 1228800/19200 = 64 linear• 3.0 dB = average Eb/No for 5% FER with vehicular fading at 30 kmph ( )• 2.5 dB = Eb/No standard deviation ( )• 1.0 = data activity factor ( )• 0.1 = mean square of activity factor ( )

Traffic Load Assumptions: Data @ 38400 bps• 1 = number of average users at 38400 bps ( )• 15.1 dB = Processing gain ( ) or 1228800/38400 = 32 linear• 2.4 dB = average Eb/No for 5% FER with vehicular fading at 30 kmph ( )• 2.5 dB = Eb/No standard deviation ( )• 1.0 = data activity factor ( )• 0.1 = mean square of activity factor ( )

The first step is to calculate the mean value of the traffic loading factor for the 6 average voice users at 9600 bps by using Equation 3-46. Using the input variables from the assumptions above, E[X]9600 is calculated as follows:

Now, calculate the mean value of the traffic loading factor for the 3 average data users at 19200 bps by using Equation 3-46. Using the input variables from the assumptions above, E[X]19200 is calculated as follows:

Now, calculate the mean value of the traffic loading factor for the 1 average data user at 38400 bps

L m( )PG m( )

ε m( )σ m( )

ν m( )ψ m( )

L m( )PG m( )

ε m( )σ m( )

ν m( )ψ m( )

L m( )PG m( )

ε m( )σ m( )

ν m( )ψ m( )

E X[ ]9600 60.713

0.45 128×-------------------------× 0.230259( ) 3.6( )⋅ 0.230259( ) 2.5( )⋅( )2

2---------------------------------------------------+exp×=

E X[ ]9600 0.074271 0.994617[ ]exp× 0.074271 2.703690 0.200805=×==

E X[ ]19200 31

0.45 64×----------------------× 0.230259( ) 3.0( )⋅ 0.230259( ) 2.5( )⋅( )2

2---------------------------------------------------+exp×=

E X[ ]19200 0.104167 0.856462[ ]exp× 0.104167 2.354815 0.245293=×==


3


by using Equation 3-46. Using the input variables from the assumptions above, E[X]38400 is calculated as follows:

Finally, calculate the total loading factor E[X]Total for all user types by summing together all of the individual results.

The next step is to calculate the variance for the traffic loading factor for the 6 average voice users at 9600 bps by using Equation 3-47. Using the input variables from the assumptions above, Var(X)9600 is calculated as follows:

Now, calculate the variance for the traffic loading factor for the 3 average data users at 19200 bps by using Equation 3-47. Using the input variables from the assumptions above, Var(X)19200 is calculated as follows:

Now, calculate the variance for the traffic loading factor for the 1 average data user at 38400 bps by using Equation 3-47. Using the input variables from the assumptions above, Var(X)38400 is calculated as follows:

E X[ ]38400 11

0.45 32×----------------------× 0.230259( ) 2.4( )⋅ 0.230259( ) 2.5( )⋅( )2

2---------------------------------------------------+exp×=

E X[ ]38400 0.069444 0.718307[ ]exp× 0.069444 2.050957 0.142428=×==

E X[ ]Total 0.200805 0.245293 0.142428 0.588526=+ +=

Var X( )9600 60.1 0.713( )+

2

0.45 128( )2×----------------------------------× 2 0.230259( ) 3.6( )⋅ 2 0.230259( ) 2.5( )⋅( )⋅ 2

+[ ]exp×=

Var X( )9600 0.000495 2.320605[ ]exp× 0.000495 10.181831× 0.005041===

Var X( )19200 30.1 1( )+

2

0.45 64( )2×------------------------------× 2 0.230259( ) 3.0( )⋅ 2 0.230259( ) 2.5( )⋅( )⋅ 2

+[ ]exp×=

Var X( )19200 0.001790 2.044294[ ]exp× 0.001790 7.723704× 0.013828===

Var X( )38400 10.1 1( )+

2

0.45 32( )2×------------------------------× 2 0.230259( ) 2.4( )⋅ 2 0.230259( ) 2.5( )⋅( )⋅ 2

+[ ]exp×=

Var X( )38400 0.002387 1.767983[ ]exp× 0.002387 5.859025× 0.013986===


3


Finally, calculate the total variance Var(X)Total for all user types by summing together all of the individual results.

The final step is to calculate the noise rise, Z, for the total traffic load using a 95% probability factor by using Equation 3-48 (as shown below).

dB

3.4.5 Reverse Link Noise Rise Capacity Estimates for IS-2000 1X

In order to calculate the capacity supported by the air interface in an IS-2000 1X system, it is important to determine the values of the various factors that affect the capacity. The IS-2000 1X reverse link capacity estimates (throughput and Erlangs) provided in this document are based on the reverse link noise rise capacity estimation equations provided in Section 3.4.2 and utilizing the parameter value assumptions that follow.

The following are the assumptions for the various IS-2000 1X parameter values to be applied to the reverse link noise rise capacity estimation equations.

3.4.5.1 Noise Rise

For the purpose of determining capacity estimates, a 10 dB maximum noise rise value was selected. Additionally, each rise has a probability factor, Pa, associated with it. The following table provides some of the recommended noise rise values and probability factors used for this exercise.

Since the probability factor is associated with a normal distribution, the 50% probability factor implies an average noise rise value. Therefore, for the scenarios where the probability factor is greater than 50%, the average noise rise will be less than the rise value shown. This can be illustrated further through Figure 3-12, where the 50% probability factor is associated with the average point in the normal distribution curve. However, a higher probability factor would be

Table 3-4: Interference Rise Scenarios

PercentileProbability (Pa)

AssociatedNoise Rise (Z)

90% 10 dB95% 10 dB98% 10 dB

Var X( )Total 0.005041 0.013828 0.013986 0.032856=+ +=

Z 10– Log10× 1 Pa Var X( )Total E X[ ]–× Total–( )=

Z 10– Log10× 1 1.644848 0.032856 0.588526–×–( )=

Z 10– Log10× 0.113327( ) 9.46==


3


associated with a value that is greater than the average value.

The capacity tables shown in Section 3.4.5.9 provide both the capacity (in Erlangs and throughput) and the average rise values associated with a 10 dB peak noise rise for the 90%, 95%, and 98% probability factors. Typical RF designs should strive to keep the peak percentile probability reverse noise rise value less than 10 dB.

Various probability factors were used in scenarios to estimate the capacity for aggressive (90%), moderate (95%), or conservative (98%) cases. Additionally, a rise value of less than 10 dB can be used to demonstrate the impact on capacity, in order to trade capacity for increased reverse link coverage.

In all of the test cases, the cell loading is considered uniform in each sector (homogeneous network) and as such the rise is the same across each cell. In practice, the non-homogeneous nature of cell loading will mean that an individual cell may be able to cope with a peak load higher than the homogeneous case.

3.4.5.2 F-factor

F-factor is the ratio of own cell interference to own cell plus other cell interference. Simulations have shown that the F-factor varies with the antenna types and propagation index. For this exercise, the following F-factors have been assumed:

In looking at Equation 3-46 and Equation 3-47, the number of users is proportional to the F-factor in order to maintain the same average and variance load factors. That is, an increase to the F-factor (out of cell interference is reduced compared to own cell interference) will result in an increase in the number of users. A decrease to the F-factor, implying out of cell interference is more prevalent, will result in a decrease to the number of users.1

Table 3-5: F-factor

Site Type F-factorOmni 0.60

3-Sector 0.456-Sector 0.40

1. Additional information showing the relationship of the F-factor to the antenna type and propagation index can be found in the following references.

a. R.H. Owen, Phil Jones, Shirin Dehgan, Dave Lister, "Uplink WCDMA capacity and range as a function of inter-to-intra cell interference: theory and practice", pp. 298-302, VTC 2000.

b. Szu-Wei Wang and Irving Wang, "Effects of Soft Handoff, Frequency Reuse and Non-Ideal Antenna Sec-torization on CDMA System Capacity", pp. 850-854, IEEE 1993.


3


3.4.5.3 Average Eb/No

Eb/No is defined as energy per bit to the noise spectral density. The appropriate value for the required Eb/No is such that the desired bit, block, or frame erasure rate of the received signal is achieved. This gives an indication of the lowest signal strength that the receiver can detect above a certain noise level. Such items as the subscriber speed, the propagation environment, and power control impact the required Eb/No.

The Eb/No numbers used for each data rate in this document are typical numbers that are used for dimensioning purposes. The Eb/No values were obtained from reverse link level simulations. The Eb/No values used for this exercise are shown in Table 3-6.

The link level simulations used to generate the Eb/No values utilized the following assumptions:

• Two receive antennas

• Eb/No values are per antenna

• The power control bit error rate of 4% used

• 1900 MHz

In looking at Equation 3-46 and Equation 3-47 again, the number of users is inversely proportional to the Eb/No in order to maintain the same average and variance load factor. That is, an increase in the Eb/No will result in a decrease in the number of users. A decrease to the Eb/No will result in an increase to the number of users.

3.4.5.4 Eb/No Standard Deviation

A standard deviation of 2.5 dB on the Eb/No is assumed for each rate. This standard deviation for the Eb/No is used to adjust the average Eb/No to compensate for imperfect power control in the real world environment.

The number of users is inversely proportional to the Eb/No standard deviation in order to maintain the same average and variance load factor. That is, an increase in the Eb/No standard deviation will result in a decrease in the number of users. A decrease to the Eb/No standard deviation will result in an increase to the number of users.

The Eb/No standard deviation has been assumed to be the same for each data rate. In a real world situation this may not be the case, but for an estimate of the capacity (as used for this exercise), one value has been assumed for all services.


3


3.4.5.5 Processing Gain

The processing gain is the ratio of the chip rate to the bit rate. For IS-2000 1X, the chip rate is equal to 1.2288 x 106

chips/s. The calculation of the processing gain in linear and in dB units are provided below.

Processing Gain linear =

Processing Gain db =

Where:W Bandwidth (1.2288 Mcps for IS-2000 1X)

R Information rate

The following table provides a summary of the Eb/No, Eb/No Standard Deviation, and the processing gain values used for the various data rates that were analyzed.

Recall that the Eb/No values shown in the above table were obtained from reverse link level simulations and represent typical values that are used for dimensioning purposes. The Eb/No values will vary based on the subscriber speed, propagation conditions, percent FER, etc. For detailed capacity and coverage results, Nokia Siemens Networks recommends using the IDGP simulation tool. This simulation tool incorporates a family of Eb/No curves as opposed to only a few Eb/Novalues.

The bearer rate data in Table 3-6 represents a data link layer rate from the subscriber’s perspective. It does not include any overhead (RLP, framing, etc.). The bearer rates in Table 3-6 are used in the calculation of the throughput capacity (see Section 3.4.5.8).

3.4.5.6 Activity Factor

The activity factor is defined as the percentage of time that a user transmits on an active traffic channel. With IS-95, a typical industry accepted voice activity factor was 40%. This roughly equated to 32% of the time the user was at full rate and 68% of the time the user was at eighth rate.

Table 3-6: IS-2000 1X Average Eb/No Values

Bearer Rate(bits/s)

Data Rate(bits/s)

FER Eb/No (dB) Eb/No Std.Dev. (dB)

Proc. Gain(dB)3 kmph 30 kmph

8600 9600 1% 2.56 3.60 2.5 21.114400 19200 5% 0.76 3.00 2.5 18.132000 38400 5% 0.12 2.40 2.5 15.164000 76800 5% -0.36 2.24 2.5 12.0128800 153600 5% -0.65 1.40 2.5 9.0

W R⁄

10 Log10 W R⁄( )×


3


With IS-2000 1X, the voice activity factor needs to be adjusted to account for the reverse pilot channel and for CRCs being sent at eighth rate. The following calculation provides the adjustment for these factors.

For IS-2000 1X, the extra CRC bits being sent produces an effective eighth rate of 1500 bps. The (0.68/6.4) term accounts for the extra CRC bits (where 9600/1500 = 6.4). The (0.68 x 10(-3.75/10)) term accounts for the reverse pilot overhead channel.

It should be noted that this adjusted activity factor is utilized in the capacity equation as a means to derate the capacity due to the reverse pilot overhead channel and CRC bits. In converting the voice users to an equivalent throughput, the voice activity factor of 40% (0.4) is used.

For the capacity results provided in this section, two different data activity factors (0.9 and 0.2) are assumed to show the impact of a high and low data activity factor user type.

The number of users is inversely proportional to the activity factor in order to maintain the same average and variance load factor. That is, an increase in the activity factor will result in a decrease in the number of users. A decrease to the activity factor will result in an increase to the number of users.

3.4.5.7 Traffic Mix

Four different traffic mix scenarios were analyzed as reflected in the following table.

The percentage of users can be interpreted, for example, as follows. In Scenario A, 100% of the users are voice users at 8.6 kbps. In this scenario, all users in the network are continuously transmitting at the relevant voice activity and at the required power to reach their respective Eb/Novalue. For Scenario C, 50% of the users are voice users at 8.6 kbps, 40% of the users are using 64 kbps, and the remaining 10% of the users are at 128.8 kbps.

3.4.5.8 Throughput Capacity

With multiple rate high-speed data services being introduced into the call model traffic mix, the capacity of a cell/sector should now be quantified with a throughput value in addition to the number

Table 3-7: Traffic Mix

Scenario Bearer ServiceVoice (8.6 kbps) 64 kbps 128.8 kbps

A 100% - -B 80% 20% -C 50% 40% 10%D 10% 60% 30%

0.32 0.68 6.4⁄ 0.68 103.75 10⁄–( )

0.713=⋅+ +


3


of Erlangs. For the capacity analysis results provided below, the estimated throughput capacity is calculated by multiplying the bearer rate, the activity factor, and the number of supported users (continuously transmitting users) together.

For a single data rate user example, consider scenario A with a rise of 10 dB and a probability factor of 95% (see Table 3-8). The voice rate assumed is 8.6 kbps and as such, approximately 27 Erlangs at the pedestrian speed can be supported in a single sector of a 3-sectored cell. This corresponds to a throughput capacity of approximately 93 kbps / sector (8.6 kbps x 0.40 AF x 27 Erlangs). As stated previously (see Section 3.4.5.6), an adjusted activity factor is utilized in the capacity equation as a means to derate the capacity due to the reverse pilot overhead channel and extra CRC bits. In converting the voice users to an equivalent throughput capacity, the non-adjusted voice activity factor of 40% (0.4) is used for the throughput calculation, instead of the adjusted activity factor of approximately 71.3% (as calculated in Section 3.4.5.6).

For a multiple data rate mixture of users, the throughput capacity is calculated for each individual data rate user type and then summed together. For example, consider traffic mix scenario C with a probability of 95% and a data activity factor of 20%. From the results in Table 3-8, an estimated 19.3 Erlangs at the pedestrian speed can be supported in a single sector of a 3-sectored cell with a total throughput of 182 kbps. The traffic distribution for scenario C is 50% for 8.6 kbps voice users, 40% for 64.0 kbps data users, and 10% for 128.8 kbps data users. According to the traffic distribution of scenario C, the throughput capacity is calculated as follows.

8.6 kbps Voice User Thruput = 8.6 kbps x 0.4 AF x (19.3 x 0.5) Erlangs = 33.2 kbps64.0 kbps Data User Thruput = 64.0 kbps x 0.2 AF x (19.3 x 0.4) Erlangs = 98.8 kbps128.8 kbps Data User Thruput = 128.8 kbps x 0.2 AF x (19.3 x 0.1) Erlangs = 49.7 kbps

Total Throughput = 181.7 kbps

3.4.5.9 IS-2000 1X Reverse Noise Rise Capacity Analysis Results

The following two tables provide capacity values (expressed as kbps throughput and Erlangs) per sector for the various scenarios assuming an interference rise limit of 10 dB but with varying levels of probability. For the traffic mix scenarios which include data users (Scenarios B, C, and D), capacity results for two different Data Activity Factors (AF) are provided. A 90% Data AF is used to estimate the results of high data activity factor users such as a File Transfer Protocol (FTP) user. A 20% Data AF is used to estimate the results of lower data activity factor users such as a Low Speed Packet Data (LSPD) or a High Speed Packet Data (HSPD) user. All of the traffic mix scenarios in Table 3-8 below assume pedestrian (3 kmph) Eb/No values.


3


Table 3-8: Reverse Capacity per Sector for Various Probabilities of Rise - Pedestrian

Scenario Rise Probability

DataAF

Avg Rise

(dB)Throughput (kbps/Sector) Erlangs/Sector

Omni 3-Sector 6-Sector Omni 3-Sector 6-Sector

A 98% N/A 4.7 117 88 78 33.6 25.2 22.4

95% N/A 5.3 124 93 83 35.7 26.8 23.8

90% N/A 5.9 131 98 88 37.7 28.3 25.1

B 98% 90% 3.3 209 156 139 14.6 10.9 9.7

98% 20% 4.2 163 122 109 30.5 22.9 20.3

95% 90% 3.8 232 174 154 16.2 12.1 10.8

95% 20% 4.8 175 132 117 32.8 24.6 21.9

90% 90% 4.5 254 191 170 17.8 13.3 11.9

90% 20% 5.4 187 140 125 35.0 26.3 23.4C 98% 90% 2.6 234 176 156 6.4 4.8 4.3

98% 20% 3.4 220 165 147 23.3 17.5 15.5

95% 90% 3.1 268 201 179 7.4 5.5 4.9

95% 20% 3.9 243 182 162 25.8 19.3 17.2

90% 90% 3.8 303 227 202 8.3 6.2 5.5

90% 20% 4.6 266 200 177 28.2 21.1 18.8

D 98% 90% 2.3 241 181 161 3.5 2.6 2.3

98% 20% 2.8 266 200 177 16.9 12.7 11.3

95% 90% 2.9 279 210 186 4.0 3.0 2.7

95% 20% 3.4 301 226 201 19.1 14.3 12.7

90% 90% 3.5 320 240 213 4.6 3.4 3.1

90% 20% 4.0 336 252 224 21.4 16.0 14.2


3


All of the traffic mix scenarios in Table 3-9 below assume vehicular (30 kmph) Eb/No values.

The results in Table 3-8 and Table 3-9 show the capacity estimates for an IS-2000 1X reverse link under the stated configurations, assumptions, and parameter values. As shown above, the capacity estimate can vary greatly depending upon the parameter values that are chosen. Although the stated assumptions and parameter values used for this exercise are deemed to be realistic, the accuracy of the capacity estimate is highly dependent upon the accuracy of the assumptions and parameter values used for the capacity estimate.

With new higher data rate services being introduced (via IS-95B or IS-2000), it is expected that the forward link will require higher data downloads than the reverse link. As a result, the forward link is also expected to be the limiting path from a capacity perspective. Even though the forward link may be the limiting factor of capacity for some systems, it may still be appropriate to use the previous reverse link capacity estimates to approximate the CDMA carrier capacity under the given assumptions and conditions. In many instances, the capacity analysis results of the reverse link can sometimes provide an adequate budgetary estimate for the CDMA carrier. Ultimately, simulations should be used (i.e. using IDGP) to obtain more accurate capacity estimations. Simulations can take into account many variable elements for which a general reverse or forward

Table 3-9: Reverse Capacity per Sector for Various Probabilities of Rise - Vehicle

Scenario RiseProbability

DataAF

Avg. Rise

(dB)Throughput (kbps/Sector) Erlangs/SectorOmni 3-Sector 6-Sector Omni 3-Sector 6-Sector

A 98% N/A 4.4 89 67 59 25.5 19.1 17.0

95% N/A 5.0 95 71 63 27.3 20.4 18.2

90% N/A 5.6 101 76 67 29.0 21.7 19.3

B 98% 90% 2.4 108 81 72 7.5 5.6 5.0

98% 20% 3.5 104 78 69 19.4 14.6 12.9

95% 90% 3.0 124 93 83 8.7 6.5 5.8

95% 20% 4.0 114 86 76 21.4 16.0 14.3

90% 90% 3.6 141 106 94 9.9 7.4 6.6

90% 20% 4.7 125 94 83 23.3 17.5 15.6

C 98% 90% 2.0 113 85 75 3.1 2.3 2.1

98% 20% 2.7 120 90 80 12.8 9.6 8.5

95% 90% 2.5 134 101 89 3.7 2.8 2.5

95% 20% 3.2 137 103 92 14.6 10.9 9.7

90% 90% 3.1 157 118 105 4.3 3.2 2.9

90% 20% 3.8 155 116 103 16.4 12.3 10.9

D 98% 90% 1.8 114 86 76 1.6 1.2 1.1

98% 20% 2.2 132 99 88 8.4 6.3 5.6

95% 90% 2.3 138 103 92 2.0 1.5 1.3

95% 20% 2.7 154 116 103 9.8 7.4 6.5

90% 90% 2.8 163 122 109 2.3 1.8 1.6

90% 20% 3.3 178 133 119 11.3 8.5 7.5


3


link capacity equation cannot adequately model (i.e. non uniform traffic and speed distributions, non uniform cell site layouts, propagation characteristics for a specific area, multiple subscriber classes with various call models, combined forward and reverse link analysis, etc.).

As a point of reference, the CDMA Development Group (CDG) has published a report2 with simulation results for voice users showing 29.9 Erlangs for the reverse link and 23.6 Erlangs for the forward link. These capacity values were based on a generic 37 site system. Furthermore, the sites were three-sector and a vehicular fading model was assumed.

The following figure shows the relationship between the reverse link noise rise and the throughput for several probability curves. The input parameters used to create the figure are shown below. The 50%-ile curve corresponds to the average rise.

Figure 3-14: Reverse Link Rise vs. Throughput


Parameters:• Traffic mix = Scenario B

• Voice activity factor = 57.6%

• Data activity factor = 100%

2. CDG Evolution Study Report, Revision 4.01, January 10,2000

0

2

4

6

8

10

0 50 100 150 200 250

Throughput (Kbps)

No

ise

Ris

e (d

B)

98% 95% 90% 85% 75% 50%


3


• Mean square of activity factor = 0.1 dB

• F-factor = 0.45 (3-sector cell site configuration)

• Vehicular (30 kmph) Eb/No assumptions from Table 3-6 were used

• Eb/No standard deviation = 2.5 dB

The following figure shows the relationship between reverse link noise rise and Erlangs of various data rates. The input parameters used to create the figure are shown below.

Figure 3-15: Reverse Link Rise vs. Erlangs for Different Data Rates


Parameters:• Voice and data activity factor = 57.6%



• Probability factor = 95%



The curves in the figure above show the significant impact that data users can have on the capacity of a system. The voice and data activity factors were purposely set to the same value in order to reflect the capacity impact of just varying the data rate.

0

1

2

3

4

5

6

7

8

9

10

0 5 10 15 20 25 30

Erlangs

No

ise

Ris

e (d

B)

Voice @ 9600 Data @ 19200 Data @ 38400 Data @ 76800 Data @ 153600


3


The following figure shows the relationship between the reverse link total throughput and total Erlangs with respect to the data activity factor. The input parameters used to create the figure are shown below.

Figure 3-16: Reverse Link Total Erlangs & Throughput vs. Data Activity Factor



• Peak noise rise = 10 dB







0

20

40

60

80

100

120

90%80%70%60%50%40%30%20%10%

Data Activity Factor

To

tal

Th

rup

ut

(Kb

ps)

0.0

5.0

10.0

15.0

20.0

25.0

30.0

To

tal

Erl

ang

s

Total Thruput Total Erlangs


3


3.5 Forward Link Pole Capacity Estimation

Forward link (downlink) capacity calculations are similar to the reverse link calculations in that the ratio of energy per bit over the interference density for each subscriber needs to be calculated. The nature of the interference is slightly different in that the pilot, page and synchronization channels need to be considered as interference. Therefore the capacity of the forward link is dependent upon the strength of these channels. Another factor that may need to be considered in the calculation of forward link capacity is the total amount of base station transmission power required. By using the appropriate input parameters, the following approach can be applied towards both types of systems (IS-95 and IS-2000 1X).

3.5.1 Forward Link Load Factor Estimation

A forward link load factor, , can be defined in a similar approach as the reverse link pole

capacity equations, although some of the parameters are slightly different. The following equation can be used to represent the forward link load factor.

[EQ 3-49]

Where:Forward link load factor

Number of connections per cell

Activity factor of user j

Signal energy per bit divided by noise spectral density of user j

Bandwidth of the channel

Data rate of user j

Orthogonality of the channel of user j

Ratio of out of cell to in cell base station power received by user j

When compared to the reverse link equations, the primary new parameter is , which represents

the orthogonality factor for the forward link users. Since the forward link employs orthogonal codes to separate the users, multipath propagation can cause sufficient delay spread in the radio channel which produces interference. Thus, the orthogonality factor is used to estimate the amount of interference created by the multipath propagation environment, where a value of 1 corresponds

ηFL

ηFL νj

j 1=

N

Eb No⁄( )j

W Rj⁄---------------------- 1 αj–( ) ij+[ ]⋅ ⋅=

ηFL

N

νj

Eb No⁄( )j

W

Rj

αj

ij

αj


3


to perfectly orthogonal users and a value of 0 corresponds to no orthogonality. Typically, the orthogonality factor is between 0.4 and 0.9 for multipath channels.

For the forward link, the ratio of out of cell to in cell base station power received, , is dependent

upon the individual user location and is therefore different for each user.

3.5.2 Forward Link Pole Capacity Estimation

When the forward link load factor approaches unity, the system reaches its pole capacity and the noise rise over thermal goes to infinity. The forward link noise rise pole capacity can be represented by the following equation:

[EQ 3-50]

Where:Noise rise (dB)

Forward link load factor (see Equation 3-49)

The forward link noise rise pole capacity equation can be used to estimate the noise rise over thermal noise due to multiple access interference. This is similar to the reverse link equation (see Equation 3-44) and has the same characteristics as shown in Figure 3-11.

For forward link dimensioning, it is important to take into account the total amount of base station transmission power required. The power estimate should be based on the average transmission power for the user and not the maximum transmission power for a user at the cell edge which is typically shown by the link budget. The total base station transmission power for a user at an ‘average’ location within the cell can be mathematically expressed by the following equation.

BS_Tx_Power = [EQ 3-51]

Where:Noise spectral density of the subscriber receiver front-end or ,

where k is Boltzmann’s constant J/K, T is temperature in degrees Kelvin (290 K), and NF is the subscriber station noise figure

Average attenuation between the base station transmitter and the subscriber station receiver

Average load factor using Equation 3-49 with average values for and

ij

Z 10Log10 1 ηFL–( )–=

Z

ηFL

Nrf W L νj

j 1=

N

Eb No⁄( )j

W Rj⁄----------------------⋅ ⋅ ⋅ ⋅

1 ηFL–-----------------------------------------------------------------------

Nrf Nrf k T⋅ NF+=

1.38 1023–⋅( )

L

ηFL αj ij


3


When using Equation 3-51, the power impacts of the forward link common channels (pilot, page, sync, quick paging channel, etc.) and cable losses should be accounted for in the BS_Tx_Powerallocation.

3.6 Forward Link Fractional Power Capacity Estimation

For the forward link of a CDMA cell site, there is a fixed amount of power that is allocated for a CDMA carrier on a per-cell/per-sector basis. Since this is a fixed resource, an alternate method for estimating forward link capacity is to normalize this fixed power resource and estimate the fractional amount of power required for the average user while taking several factors into account (i.e. distribution of users with 1-way, 2-way, & 3-way links, other cell interference, overhead channel power, required Eb/Nt, forward power control error, activity factor, etc.).

The following equation represents a first order approximation of the forward link capacity using a fractional power approach:

[EQ 3-52]

Where:Traffic load supported (in Erlangs)

Effective Voice or Data Activity

Fraction of total cell power for pilot, page, and sync

Fraction of users in i-way handoff

Fraction of allocated cell power for each i-way link

The next step is to provide a more detailed estimate for the fraction of allocated cell power for each i-way link.

[EQ 3-53]

Where:Total normalized interference seen by i-way user

Fraction of recovered power by i-way connection

N1 ζpps–( )

Veff 3S3wayζ3way( 2S2wayζ2way S1wayζ1way )+ +--------------------------------------------------------------------------------------------------------------------<

N

Veff

ζpps

Siway

ζiway

ςi way–

I( on i )( λ i( ) )– 10

Eb

Ntiway

-------------- FPCerror+ 10⁄

⋅

i λ i( )WR-----⋅ ⋅

---------------------------------------------------------------------------------------=

Ion i )(

λ i( )


3


Energy per bit per thermal noise power spectral density per i-way connection

FPCerror Forward power control error in dB

W Bandwidth of channel

R Data rate

The final step is to provide a more detailed estimate for the total normalized interference as seen by each i-way user.

[EQ 3-54]

Where:Adjacent carrier(s) noise factor

Other cell (not including adjacent carrier) normalized interference

For the following examples, the values from Table 3-10 below (0 adjacent carriers is assumed) are entered into Equation 3-52, Equation 3-53, and Equation 3-54 in order to estimate the forward link capacity for a Rate Set 1 and Rate Set 2 system.

Table 3-10: Example of Parameter Values

Parameter 1-way 2-way 3-way0.40 0.35 0.25

0.134 0.30 0.30

0.92 0.92 0.80

for 13 kb 15.5 dB 9 dB 7 dB

for 8 kb 13 dB 7 dB 5 dB

1.2 dB (for 13 kb) or 1.5 dB (for 8 kb)

0.37

85.33 (for 13 kb) or 128 (for 8 kb)

0.48 (for 13 kb) or 0.56 (for 8 kb)

(assume 2% per carrier)

1.00 (for 0 adjacent carrier),1.02 (for 1 adjacent carrier),1.04 (for 2 adjacent carriers)

Eb

Ntiway---------------

Ion i )( i δ Iocn i )(⋅+=

δ

Iocn i )(

Siway

Iocn i )(

λ i( )

Eb Ntiway⁄

Eb Ntiway⁄

FPCerror

ζpps

W R⁄Veff

δ


3


The following examples assume no adjacent carrier interference ( = 1.00).

Example #1: Rate Set 1

1. Estimate for the total normalized interference as seen by each i-way user.

2. Estimate the fraction of allocated cell power for each i-way link.

3. Estimate the first order approximation of the forward link capacity using a fractional power approach.

Erlangs

Example #2: Rate Set 2

1. Estimate for the total normalized interference as seen by each i-way user (same for Rate Set 1).

δ

Ion 1 )( 1 1 0.134 1.134=⋅+=

Ion 2 )( 2 1 0.3 2.3=⋅+=

Ion 3 )( 3 1 0.3 3.3=⋅+=

ς1 way–1.134( 0.92 )– 10

13 1.5+( ) 10⁄⋅1 0.92 128⋅ ⋅

------------------------------------------------------------------------ 0.0512==

ς2 way–2.3( 0.92 )– 10

7 1.5+( ) 10⁄⋅2 0.92 128⋅ ⋅

---------------------------------------------------------------- 0.0415==

ς3 way–3.3( 0.8 )– 10

5 1.5+( ) 10⁄⋅2 0.8 128⋅ ⋅

------------------------------------------------------------- 0.0364==

N1 0.37–( )

0.56 3 0.25 0.0364⋅ ⋅(⋅ 2 0.35 0.0415⋅ ⋅ 0.40 0.0512⋅ )+ +----------------------------------------------------------------------------------------------------------------------------------------------- 14.6=<

Ion 1 )( 1 1 0.134 1.134=⋅+=

Ion 2 )( 2 1 0.3 2.3=⋅+=

Ion 3 )( 3 1 0.3 3.3=⋅+=


3


2. Estimate the fraction of allocated cell power for each i-way link.

3. Estimate the first order approximation of the forward link capacity using a fractional power approach.

Erlangs

3.7 Forward Link Noise Rise Capacity Estimation

The amount of noise rise (interference) that can be tolerated by the CDMA subscriber will place a limit upon how many users can be supported by the forward link. As the number of users served by the forward link is increased, the level of noise rise seen by the subscribers will be increased due to the additional energy being transmitted by the site to support all of the subscribers. The cell capacity is determined by calculating the number of users required to produce a maximum accepted noise rise. This section is similar to Section 3.4 for the reverse link except it is being applied to the forward link.

This section provides a method of estimating the noise rise for a particular user type. The estimating approach will also allow the calculation of the total noise rise for multiple user types. As a result, the noise rise estimation approach provided in this section is better suited to estimate the capacity of a system which utilizes multiple user types (i.e. multiple data rates). Although this capacity estimation approach can be applied towards both IS-95 and IS-2000 systems, it may be more appropriate in estimating the capacity of an IS-2000 system, where it is more common to support different user type profiles utilizing different data rates.

For IS-2000 systems, it is important to note that the capacity estimation calculation provided in this section does not account for the dynamic resource allocation capabilities of an IS-2000 1X packet data system. Within the IS-2000 1X infrastructure, the subscriber will be assigned supplemental channel resources based upon several criteria (e.g. the demand requirements for the amount of data to be transmitted, RF capacity availability, Walsh code resource availability, etc.). The allocation of these IS-2000 1X supplemental channel resources are also dynamically adjusted throughout the duration of the packet data call. The capacity estimation calculation provided in this section treats a packet data user more like a circuit data user. The capacity formulas provided imply a fixed resource allocation where there are X users at 9.6 kbps, Y users at 19.2 kbps, Z users at 38.4 kbps, etc. As a result, the capacity obtained from the capacity estimation approach may differ from that

ς1 way–1.134( 0.92 )– 10

15.5 1.2+( ) 10⁄⋅1 0.92 85.33⋅ ⋅

--------------------------------------------------------------------------- 0.1275==

ς2 way–2.3( 0.92 )– 10

9 1.2+( ) 10⁄⋅2 0.92 85.33⋅ ⋅

---------------------------------------------------------------- 0.0920==

ς3 way–3.3( 0.8 )– 10

7 1.2+( ) 10⁄⋅2 0.8 85.33⋅ ⋅

------------------------------------------------------------- 0.0807==

N1 0.37–( )

0.48 3 0.25 0.0807⋅ ⋅(⋅ 2 0.35 0.0920⋅ ⋅ 0.40 0.1275⋅ )+ +----------------------------------------------------------------------------------------------------------------------------------------------- 7.5=<


3


of an actual IS-2000 1X system. For a more accurate estimation of packet data services, it is recommended to utilize a simulation tool which simulates the dynamic resource allocation capabilities of an IS-2000 1X system.

Another aspect of the forward link capacity is the amount of base station transmission power required. As the subscriber unit experiences more interference, it will request more power from its serving base station to compensate for increased interference. Therefore, the transmission power limitations of the base station may place an upper limit on the forward capacity available.

3.7.1 Forward Link Noise Rise Capacity Limit

The forward link pole capacity is considered to be the point where additional power from the BTSs to support an additional user will cause the noise rise within the subscriber unit to increase exponentially. This will create an unstable situation where user connections may be lost and the network grade of service will be severely degraded.

The forward link noise rise pole capacity can be represented by the same equation that is provided in Equation 3-50. A graph of the forward noise rise pole capacity equation is the same as the one for the reverse noise rise pole capacity equation which is shown in Figure 3-11.

In order to estimate the capacity from a number of users perspective, a forward noise rise capacity limit must be selected. For CDMA RF system designs (for both IS-95A/B and IS-2000), a peak noise rise of 10 dB is recommended to be the maximum that a system should tolerate (which is the same limit for the reverse link). In order to account for the noise rise generated by the pilot, page, and sync overhead channels for the forward link, a de-rating of the noise rise limit is recommended as follows.

Assumptions:Pilot = 20% of total power at maximum capacityPage = 75% of the pilot powerSync = 10% of the pilot powerPPStotal = 20% (for pilot) + 20% x 75% (for paging) + 20% x 10% (for sync) = 37%

Noise Rise De-rating:PPStotal = 37% of total power at maximum capacityTotal User Capacity = 100% - 37% = 63% of total power at maximum capacity10 dB noise rise limit = 10(10/10) = 10 linear unitsDe-rated Noise Rise Limit = 10 x 63% = 6.3 linear = 10log(6.3) = ~8.0 dB

Thus, the recommended de-rated peak noise rise limit is 8 dB. The average noise rise would be several dB below this peak value. It is important to note that the 8 dB noise rise limit is a peak value which is associated with a certain probability factor (see Equation 3-48 and Section 3.4.3). The recommended probability factors associated with the 8 dB peak noise rise recommendation are as follows.

• 8 dB noise rise with a 90% probability factor (for aggressive capacity results)


3


• 8 dB noise rise with a 95% probability factor (for moderate capacity results)• 8 dB noise rise with a 98% probability factor (for conservative capacity results)

Although the above recommendation provides some flexibility in selecting a probability factor, the 8 dB noise rise with a 95% probability factor is the typical limit that is normally recommended.

3.7.2 Forward Noise Rise Capacity Estimation

To approximate the number of users that could be supported by a site while staying below a desired noise rise limit, the following forward link capacity equations can be utilized.

A multi-service traffic loading factor, X, can be expressed as follows:

[EQ 3-55]

The mean value for the multi-service traffic loading factor, X, is expressed as:

[EQ 3-56]

The variance for the multi-service traffic loading factor, X, is expressed as:

[EQ 3-57]

The following equation provides the distribution of the noise rise, Z, for the multi-service traffic loading factor, X (which is the same equation provided for the reverse link, Equation 3-48):

[EQ 3-58]

Where:M Number of different service-types

Traffic load of the mth service-type (in Erlangs)

The energy-per-bit to total-interference-density target of the mth service-type

LN(10)/10

Average Eb/No (dB) of the mth service-type

X L m( )

m 1=

M

ν m( )

PG m( )--------------×

Eb m( )NT

-------------× 1 α m( )–( ) i m( )+[ ]×=

E X[ ] L m( )

m 1=

M

ν m( )

PG m( )--------------× βε m( )

βσ m( )( )2

2---------------------+exp× 1 α m( )–( ) i m( )+[ ]×=

Var X( ) L m( )

m 1=

M

ψ m( ) ν m( )( )+

2

PG m( )( )2-----------------------------------× 2βε m( ) 2 βσ m( )( )2

+[ ]exp× 1 α m( )–( ) i m( )+[ ]×=

Z 10– Log10× 1 Pa Var X( ) E X[ ]–×–( )=

L m( )

Eb m( )NT

-------------

β

ε m( )


3


Eb/No standard deviation, in dB of the mth service-type (to account for

inaccuracies in power control)

Activity Factor of the mth service-type

Mean Square of Activity Factor of the mth service-type (variance = 0.1)

Orthogonality of the channel of the mth service-type

Ratio of out of cell to in cell base station power received by the mth service-type,

where I =

Note: The terms I and i are equivalent to the terms F and f for the reverse link (see Equation 3-13), but from a forward link perspective. For the forward link, the ratio of out of cell to in cell base station power received, , is dependent upon

the individual user location and is therefore different for each user.

Processing gain (Bandwidth/Information rate) of the mth service

Z Interference rise (expressed in dB)

Pa Probability factor (inverse of the standard normal cumulative distribution) with a distribution having a mean of 0 and a standard deviation of 1 (see Section 3.4.3 for more details regarding the probability factor)

In a scenario with multiple services, the equations are a bit more complex than for a single service. Basically, an average and variance needs to be determined for each service offered. The net rise, Z, will need to account for all of the users being handled by each service.

3.7.3 Forward Link Noise Rise Capacity Estimation Examples

The following section provides two examples of how to use the forward link noise rise capacity estimation equations provided in Section 3.7.2. The first example estimates the noise rise for a single service type of traffic load of voice users only. The second example provides the calculations required to estimate the noise rise for a multiple service type of traffic load with a mixture of voice and data users.

3.7.3.1 Example #1: Voice Only

The following example calculates the noise rise for, on average, 16.3 IS-2000 1X voice users at 9600 bps, in a 3-sectored system with a 95% probability factor. Additional assumptions are

σ m( )

ν m( )

ψ m( )

α m( )

i m( )

InCellInCell OutCell+-------------------------------------------- 1

1OutCellInCell

---------------------+------------------------------ 1

1 i+-----------==

i m( )

PG m( )


3


provided below.

Traffic Load:16.3 Voice users (average) at 9600 bps

General Assumptions:• 0.45 = I-factor (3 sector), where

• 1.64 = probability factor for 95% (Pa)• 0.23 = Beta value LN(10)/10 ( )

Traffic Load Assumptions: Voice @ 9600 bps• 16.3 = number of average users at 9600 bps ( )• 21.1 dB = Processing gain ( ) or 1228800/9600 = 128 linear• 6.34 dB = average Eb/No for 1% FER with vehicular fading at 30 kmph ( )• 2.5 dB = Eb/No standard deviation ( )• 0.56 = voice activity factor ( )• 0.1 = mean square of activity factor ( )• 0.6 = Orthogonality factor ( )

The first step is to calculate the mean value of the traffic loading factor, X, for the 16.3 average voice users by using Equation 3-56 (repeated below for reference).

Using the input variables from the assumptions above, E[X] is calculated as follows:

The next step is to calculate the variance for the traffic loading factor, X, for the 16.3 average voice users by using Equation 3-57 (repeated below for reference).

i m( )1I--- 1

10.45---------- 1–=–=

β

L m( )PG m( )

ε m( )σ m( )

ν m( )ψ m( )

α m( )

E X[ ] L m( )

m 1=

M

ν m( )

PG m( )--------------× βε m( )

βσ m( )( )2

2---------------------+exp× 1 α m( )–( ) i m( )+[ ]×=

E X[ ] 16.30.56128----------× 0.23( ) 6.34( )⋅ 0.23( ) 2.5( )⋅( )2

2---------------------------------------+exp× 1 0.6–( ) 1

0.45---------- 1–+×=

E X[ ] 0.115685 1.625527[ ]exp× 0.115685 5.081096 0.587805=×==

Var X( ) L m( )

m 1=

M

ψ m( ) ν m( )( )+

2

PG m( )( )2-----------------------------------× 2βε m( ) 2 βσ m( )( )2

+[ ]exp× 1 α m( )–( ) i m( )+[ ]×=

Var X( ) 16.30.1 0.56( )+

2

128( )2--------------------------------× 2 0.23( ) 6.34( )⋅ 2 0.23( ) 2.5( )⋅( )⋅ 2

+[ ]exp× 1 0.6–( ) 10.45---------- 1–+×=


3


The final step is to calculate the noise rise, Z, for the 16.3 average voice users by using Equation 3-58 (repeated below for reference).

dB

3.7.3.2 Example #2: Voice and Data Users

The following example calculates the noise rise for a multiple service type traffic load environment consisting of, on average, 6 IS-2000 1X voice users at 9600 bps, 1 IS-2000 1X data user at 19200 bps, and 1 data user at 38400 bps. Additional assumptions are provided below.

Traffic Load:6 Voice users (average) at 9600 bps1 Data user (average) at 19200 bps1 Data user (average) at 38400 bps

General Assumptions:• 0.45 = I-factor (3 sector), where

• 1.64 = probability factor for 95% (Pa)• 0.23 = Beta value LN(10)/10 ( )

Traffic Load Assumptions: Voice @ 9600 bps• 6 = number of average users at 9600 bps ( )• 21.1 dB = Processing gain ( ) or 1228800/9600 = 128 linear• 6.34 dB = average Eb/No for 1% FER with vehicular fading at 30 kmph ( )• 2.5 dB = Eb/No standard deviation ( )• 0.56 = voice activity factor ( )• 0.1 = mean square of activity factor ( )• 0.6 = Orthogonality factor ( )

Traffic Load Assumptions: Data @ 19200 bps• 1 = number of average users at 19200 bps ( )• 18.1 dB = Processing gain ( ) or 1228800/19200 = 64 linear• 5.69 dB = average Eb/No for 5% FER with vehicular fading at 30 kmph ( )

Var X( ) 0.000668 3.582424[ ]exp× 0.000668 35.960611× 0.024004===

Z 10– Log10× 1 Pa Var X( ) E X[ ]–×–( )=

Z 10– Log10× 1 1.644848 0.024004 0.587805–×–( )=

Z 10– Log10× 0.157354( ) 8.03==

i m( )1I--- 1

10.45---------- 1–=–=

β

L m( )PG m( )

ε m( )σ m( )

ν m( )ψ m( )

α m( )

L m( )PG m( )

ε m( )


3


• 2.5 dB = Eb/No standard deviation ( )• 0.9 = data activity factor ( )• 0.1 = mean square of activity factor ( )• 0.6 = Orthogonality factor ( )

Traffic Load Assumptions: Data @ 38400 bps• 1 = number of average users at 38400 bps ( )• 15.1 dB = Processing gain ( ) or 1228800/38400 = 32 linear• 4.94 dB = average Eb/No for 5% FER with vehicular fading at 30 kmph ( )• 2.5 dB = Eb/No standard deviation ( )• 0.9 = data activity factor ( )• 0.1 = mean square of activity factor ( )• 0.6 = Orthogonality factor ( )

The first step is to calculate the mean value of the traffic loading factor for the 6 average voice users at 9600 bps by using Equation 3-56. Using the input variables from the assumptions above, E[X]9600 is calculated as follows:

Now, calculate the mean value of the traffic loading factor for the 1 average data user at 19200 bps by using Equation 3-56. Using the input variables from the assumptions above, E[X]19200 is calculated as follows:

Now, calculate the mean value of the traffic loading factor for the 1 average data user at 38400 bps by using Equation 3-56. Using the input variables from the assumptions above, E[X]38400 is calculated as follows:

Finally, calculate the total loading factor E[X]Total for all user types by summing together all of the

σ m( )ν m( )

ψ m( )α m( )

L m( )PG m( )

ε m( )σ m( )

ν m( )ψ m( )

α m( )

E X[ ]9600 60.56128----------× 0.23( ) 6.34( )⋅ 0.23( ) 2.5( )⋅( )2

2---------------------------------------+ 1 0.6–( ) 1

0.45---------- 1–+×exp×=

E X[ ]9600 0.042583 1.625527[ ]exp× 0.042583 5.081096 0.216370=×==

E X[ ]19200 10.964-------× 0.23( ) 5.69( )⋅ 0.23( ) 2.5( )⋅( )2

2---------------------------------------+exp× 1 0.6–( ) 1

0.45---------- 1–+×=

E X[ ]19200 0.022813 1.475859[ ]exp× 0.022813 4.374791 0.099800=×==

E X[ ]38400 10.932-------× 0.23( ) 4.94( )⋅ 0.23( ) 2.5( )⋅( )2

2---------------------------------------+exp× 1 0.6–( ) 1

0.45---------- 1–+×=

E X[ ]38400 0.045625 1.303164[ ]exp× 0.045625 3.680926 0.167942=×==


3


individual results.

The next step is to calculate the variance for the traffic loading factor for the 6 average voice users at 9600 bps by using Equation 3-57. Using the input variables from the assumptions above, Var(X)9600 is calculated as follows:



Finally, calculate the total variance Var(X)Total for all user types by summing together all of the individual results.

The final step is to calculate the noise rise, Z, for the total traffic load using a 95% probability factor by using Equation 3-58 (as shown below).

E X[ ]Total 0.216370 0.099800 0.167942 0.484112=+ +=

Var X( )9600 60.1 0.56( )+

2

128( )2--------------------------------× 2 0.23( ) 6.34( )⋅ 2 0.23( ) 2.5( )⋅( )⋅ 2

+[ ]exp× 1 0.6–( ) 10.45---------- 1–+×=

Var X( )9600 0.000246 3.582424[ ]exp× 0.000246 35.960611× 0.008836===

Var X( )19200 10.1 0.9( )+

2

64( )2-----------------------------× 2 0.23( ) 5.69( )⋅ 2 0.23( ) 2.5( )⋅( )⋅ 2

+[ ]exp× 1 0.6–( ) 10.45---------- 1–+×=

Var X( )19200 0.000360 3.283088[ ]exp× 0.000360 26.657952× 0.009608===

Var X( )38400 10.1 0.9( )+

2

32( )2-----------------------------× 2 0.23( ) 4.94( )⋅ 2 0.23( ) 2.5( )⋅( )⋅ 2

+[ ] 1 0.6–( ) 10.45---------- 1–+×exp×=

Var X( )38400 0.001442 2.937699[ ]exp× 0.001442 18.872371× 0.027207===

Var X( )Total 0.008836 0.009608 0.027207 0.045650=+ +=

Z 10– Log10× 1 Pa Var X( )Total E X[ ]–× Total–( )=


3


dB

3.7.4 Forward Link Noise Rise Capacity Estimates for IS-2000 1X

In order to calculate the capacity supported by the air interface in an IS-2000 1X system, it is important to determine the values of the various factors that affect the capacity. The IS-2000 1X forward link capacity estimates (throughput and Erlangs) provided in this document are based on the forward link noise rise capacity estimation equations provided in Section 3.7.2 and utilizing the parameter value assumptions that follow.

The following are the assumptions for the various IS-2000 1X parameter values to be applied to the forward link noise rise capacity estimation equations.

3.7.4.1 Noise Rise

For the purpose of determining capacity estimates, a 10 dB maximum noise rise value was selected. As shown in Section 3.7.1, this 10 dB limit is de-rated to 8 dB in order to account for the overhead channels. Additionally, each rise has a probability factor, Pa, associated with it. Table 3-11 provides some of the recommended noise rise values and probability factors used for this exercise.

Since the probability factor is associated with a normal distribution, the 50% probability factor implies an average noise rise value. Therefore, for the scenarios where the probability factor is greater than 50%, the average noise rise will be less than the rise value shown. This can be illustrated further through Figure 3-12, where the 50% probability factor is associated with the average point in the normal distribution curve. However, a higher probability factor would be associated with a value that is greater than the average value.

The capacity tables shown in Section 3.7.4.10 provide both the capacity (in Erlangs and throughput) and the average rise values associated with a 8 dB peak noise rise for the 90%, 95%, and 98% probability factors. Typical RF designs should strive to keep the peak percentile probability reverse noise rise value less than 8 dB.

Various probability factors were used in scenarios to estimate the capacity for aggressive (90%), moderate (95%), or conservative (98%) cases. Additionally, a rise value of less than 8 dB can be used to demonstrate the impact on capacity, in order to trade capacity for increased forward link

Table 3-11: Interference Rise Scenarios

PercentileProbability (Pa)

AssociatedNoise Rise (Z)

90% 8 dB95% 8 dB98% 8 dB

Z 10– Log10× 1 1.644848 0.045650 0.484112–×–( )=

Z 10– Log10× 0.164450( ) 7.84==


3


coverage.

In all of the test cases, the cell loading is considered uniform in each sector (homogeneous network) and as such the rise is the same across each cell. In practice, the non-homogeneous nature of cell loading will mean that an individual cell may be able to cope with a peak load higher than the homogeneous case.

3.7.4.2 I-factor

I-factor is the ratio of own cell interference to own cell plus other cell interference from the subscriber perspective. The ratio of out of cell to in cell base station power received by the subscriber is the parameter. The I-factor and parameter have the following relationship.

where I = or

[EQ 3-59]

The terms I and i are equivalent to the terms F and f for the reverse link (see Equation 3-13), but from a forward link subscriber perspective. For the forward link, the ratio of out of cell to in cell base station power received by user m, , is dependent upon the individual user location and is

therefore different for each user.

For this exercise, the following I-factors have been assumed:

3.7.4.3 Average Eb/No

Eb/No is defined as energy per bit to the noise spectral density. The appropriate value for the required Eb/No is such that the desired bit, block, or frame erasure rate of the received signal is achieved. This gives an indication of the lowest signal strength that the subscriber receiver can detect above a certain noise level. Such items as the subscriber speed, the propagation environment, and power control impact the required Eb/No.

The Eb/No numbers used for each data rate in this document are typical numbers that are used for dimensioning purposes. The Eb/No values were obtained from forward link level simulations. The

Table 3-12: I-factor

Site Type I-factorOmni 0.60

3-Sector 0.456-Sector 0.40

i m( ) i m( )

InCellInCell OutCell+-------------------------------------------- 1

1OutCellInCell

---------------------+------------------------------ 1

1 i+-----------==

i1I--- 1–=

i m( )


3


Eb/No values used for this exercise are shown in Table 3-13.

In looking at Equation 3-56 and Equation 3-57 again, the number of users is inversely proportional to the Eb/No in order to maintain the same average and variance load factor. That is, an increase in the Eb/No will result in a decrease in the number of users. A decrease to the Eb/No will result in an increase to the number of users.

3.7.4.4 Eb/No Standard Deviation

A standard deviation of 2.5 dB on the Eb/No is assumed for each rate. This standard deviation for the Eb/No is used to adjust the average Eb/No to compensate for imperfect power control in the real world environment.

The number of users is inversely proportional to the Eb/No standard deviation in order to maintain the same average and variance load factor. That is, an increase in the Eb/No standard deviation will result in a decrease in the number of users. A decrease to the Eb/No standard deviation will result in an increase to the number of users.

The Eb/No standard deviation has been assumed to be the same for each data rate. In a real world situation this may not be the case, but for an estimate of the capacity (as used for this exercise), one value has been assumed for all services.

3.7.4.5 Processing Gain

The processing gain is the ratio of the chip rate to the bit rate. For IS-2000 1X, the chip rate is equal to 1.2288 x 106

chips/s. The calculation of the processing gain in linear and in dB units are provided below.

Processing Gain linear =

Processing Gain db =

Where:W Bandwidth (1.2288 Mcps for IS-2000 1X)

R Information rate

W R⁄

10 Log10 W R⁄( )×


3


The following table provides a summary of the Eb/No, Eb/No Standard Deviation, and the processing gain values for the various data rates that were used in this exercise.

Recall that the Eb/No values shown in the above table were obtained from forward link level simulations and represent typical values that are used for dimensioning purposes. The Eb/No values will vary based on the subscriber speed, propagation conditions, percent FER, etc. For detailed capacity and coverage results, Nokia Siemens Networks recommends using the IDGP simulation tool. This simulation tool incorporates a family of Eb/No curves as opposed to only a few Eb/Novalues.

The bearer rate data in Table 3-13 represents a data link layer rate from the subscriber’s perspective. It does not include any overhead (RLP, framing, etc.). The bearer rates in Table 3-13 are used in the calculation of the throughput capacity (see Section 3.7.4.9).

3.7.4.6 Activity Factor

The activity factor is defined as the percentage of time that a user transmits on an active traffic channel. With IS-95, a typical industry accepted voice activity factor was 40%. This roughly equated to 32% of the time the user was at full rate and 68% of the time the user was at eighth rate. With IS-2000 1X, an adjustment to the voice activity factor of 16% is recommended to account for the impact of the forward power control bits. Thus a 40% voice activity factor is adjusted up to 56%.

It should be noted that this adjusted activity factor (56%) is utilized in the capacity equation as a means to derate the capacity due to the forward power control bits. In converting the voice users to an equivalent throughput, the voice activity factor of 40% (0.4) is used.

For the capacity results provided in this section, two different data activity factors (0.9 and 0.2) are assumed to shown the impact of a high and low data activity factor user type.

The number of users is inversely proportional to the activity factor in order to maintain the same average and variance load factor. That is, an increase in the activity factor will result in a decrease in the number of users. A decrease to the activity factor will result in an increase to the number of users.

Table 3-13: IS-2000 1X Average Eb/No Values

Bearer Rate(bits/s)

Data Rate(bits/s)

FER Eb/No (dB) Eb/No Std.Dev. (dB)

Proc. Gain(dB)3 kmph 30 kmph

8600 9600 1% 7.56 6.34 2.5 21.114400 19200 5% 6.53 5.69 2.5 18.132000 38400 5% 5.65 4.94 2.5 15.164000 76800 5% 4.90 4.53 2.5 12.0128800 153600 5% 5.10 4.86 2.5 9.0


3


3.7.4.7 Orthogonality Factor

CDMA utilizes orthogonal Walsh codes to separate the multiple users or multiple channels in the downlink. In the absence of multipath propagation, the orthogonality of the signal received by the subscriber would be the same as that which is sent by the base station. However, since multipath propagation can produce sufficient delay spread in the radio channel, the subscriber will see part of the base station signal as multiple access interference.

An orthogonality of 1 corresponds to perfectly orthogonal users. Typically, the orthogonality is between 0.4 and 0.9 in multipath channels.

For the capacity analysis provided in this section, an orthogonality factor of 0.6 is used for the vehicular (30 kmph) capacity results and a value of 0.9 is used for the pedestrian (3 kmph) capacity results. These values correspond to the ITU Vehicular A channel and ITU Pedestrian A channel respectively.

3.7.4.8 Traffic Mix

Four different traffic mix scenarios were analyzed as reflected in the following table.

The percentage of users can be interpreted, for example, as follows. In Scenario A, 100% of the users are voice users at 8.6 kbps. In this scenario, all users in the network are continuously receiving the relevant voice activity and at the required signal level to reach their respective Eb/Novalue. For Scenario C, 50% of the users are voice users at 8.6 kbps, 40% of the users are using 64 kbps, and the remaining 10% of the users are at 128.8 kbps.

3.7.4.9 Throughput Capacity

With multiple rate high-speed data services being introduced into the call model traffic mix, the capacity of a cell/sector should now be quantified with a throughput value in addition to the number of Erlangs. For the capacity analysis results provided below, the estimated throughput capacity is calculated by multiplying the bearer rate, the activity factor, and the number of supported users (continuously transmitting users) together.

For a single data rate user example, consider scenario A with a rise of 8 dB and a probability factor of 95% (see Table 3-15). The voice rate assumed is 8.6 kbps and as such, approximately 14.3 Erlangs at the pedestrian speed can be supported in a single sector of a 3-sectored cell. This

Table 3-14: Traffic Mix

Scenario Bearer ServiceVoice (8.6 kbps) 64 kbps 128.8 kbps

A 100% - -B 80% 20% -C 50% 40% 10%D 10% 60% 30%


3


corresponds to a throughput capacity of approximately 49 kbps / sector (8.6 kbps x 0.40 AF x 14.3 Erlangs). As stated previously (see Section 3.7.4.6), an adjusted activity factor is utilized in the capacity equation as a means to derate the capacity due to the forward power control bits. In converting the voice users to an equivalent throughput capacity, the non-adjusted voice activity factor of 40% (0.4) is used for the throughput calculation, instead of the adjusted activity factor of approximately 56% (as calculated in Section 3.7.4.6).

For a multiple data rate mixture of users, the throughput capacity is calculated for each individual data rate user type and then summed together. For example, consider traffic mix scenario C with a probability of 95% and a data activity factor of 20%. From the results in Table 3-15, an estimated 6.6 Erlangs at the pedestrian speed can be supported in a single sector of a 3-sectored cell with a total throughput of 62 kbps. The traffic distribution for scenario C is 50% for 8.6 kbps voice users, 40% for 64.0 kbps data users, and 10% for 128.8 kbps data users. According to the traffic distribution of scenario C, the throughput capacity is calculated as follows.

8.6 kbps Voice User Thruput = 8.6 kbps x 0.4 AF x (6.6 x 0.5) Erlangs = 11.4 kbps64.0 kbps Data User Thruput = 64.0 kbps x 0.2 AF x (6.6 x 0.4) Erlangs = 33.8 kbps128.8 kbps Data User Thruput = 128.8 kbps x 0.2 AF x (6.6 x 0.1) Erlangs = 17.0 kbps

Total Throughput = 62.2 kbps

3.7.4.10 IS-2000 1X Forward Noise Rise Capacity Analysis Results

The following two tables provide capacity values (expressed as kbps throughput and Erlangs) per sector for the various scenarios assuming an interference rise limit of 8 dB but with varying levels of probability. For the traffic mix scenarios which include data users (Scenarios B, C, and D), capacity results for two different Data Activity Factors (AF) are provided. A 90% Data AF is used to estimate the results of high data activity factor users such as a File Transfer Protocol (FTP) user. A 20% Data AF is used to estimate the results of lower data activity factor users such as a Low Speed Packet Data (LSPD) or a High Speed Packet Data (HSPD) user.

All of the traffic mix scenarios in Table 3-15 below assume pedestrian (3 kmph) Eb/No values with an orthogonality factor of 0.9.

Table 3-15: Forward Capacity per Sector for Various Probabilities of Rise - Pedestrian


DataAF

Avg Rise



A 98% N/A 3.0 77 44 37 22.3 12.9 10.7

95% N/A 3.5 85 49 41 24.6 14.3 11.8

90% N/A 4.1 93 54 44 27.0 15.6 12.9

B 98% 90% 1.6 88 51 42 6.2 3.6 3.0

98% 20% 2.4 89 52 43 16.7 9.7 8.0

95% 90% 2.0 106 62 51 7.5 4.3 3.6

95% 20% 2.8 102 59 49 19.1 11.1 9.2

90% 90% 2.5 127 73 61 8.9 5.1 4.3


3


All of the traffic mix scenarios in Table 3-16 below assume vehicular (30 kmph) Eb/No values with an orthogonality factor of 0.6.

90% 20% 3.4 115 67 55 21.6 12.5 10.4

C 98% 90% 1.1 76 44 37 2.1 1.2 1.0

98% 20% 1.6 89 52 43 9.5 5.5 4.5

95% 90% 1.5 97 56 46 2.7 1.5 1.3

95% 20% 2.0 108 62 52 11.4 6.6 5.5

90% 90% 1.9 121 70 58 3.3 1.9 1.6

90% 20% 2.5 128 74 61 13.6 7.9 6.5

D 98% 90% 1.0 71 41 34 1.0 0.6 0.5

98% 20% 1.2 87 51 42 5.5 3.2 2.7

95% 90% 1.3 91 53 44 1.3 0.8 0.6

95% 20% 1.6 109 63 52 6.9 4.0 3.3

90% 90% 1.7 116 67 56 1.7 1.0 0.8

90% 20% 2.1 134 78 64 8.5 4.9 4.1

Table 3-16: Forward Capacity per Sector for Various Probabilities of Rise - Vehicle

Scenario RiseProbability

DataAF

Avg. Rise

(dB)Throughput (kbps/Sector) Erlangs/SectorOmni 3-Sector 6-Sector Omni 3-Sector 6-Sector

A 98% N/A 3.3 78 51 44 22.7 14.9 12.7

95% N/A 3.8 85 56 48 24.8 16.3 13.9

90% N/A 4.4 92 61 52 26.8 17.6 15.0

B 98% 90% 1.6 75 49 42 5.3 3.5 3.0

98% 20% 2.4 82 54 46 15.4 10.1 8.7

95% 90% 2.0 91 60 51 6.4 4.2 3.6

95% 20% 2.9 93 61 52 17.6 11.6 9.9

90% 90% 2.6 108 71 60 7.5 5.0 4.2

90% 20% 3.5 105 69 59 19.8 13.0 11.1

C 98% 90% 1.1 62 41 35 1.7 1.1 1.0

98% 20% 1.6 75 49 42 7.9 5.2 4.5

95% 90% 1.5 78 51 44 2.2 1.4 1.2

95% 20% 2.0 90 59 51 9.6 6.3 5.4

90% 90% 2.0 98 64 55 2.7 1.8 1.5

90% 20% 2.6 107 70 60 11.3 7.5 6.4

D 98% 90% 1.0 57 37 32 0.8 0.5 0.5

98% 20% 1.3 70 46 39 4.4 2.9 2.5

95% 90% 1.3 73 48 41 1.0 0.7 0.6

Table 3-15: Forward Capacity per Sector for Various Probabilities of Rise - Pedestrian


DataAF

Avg Rise




3


The results in Table 3-15 and Table 3-16 show the capacity estimates for an IS-2000 1X forward link under the stated configurations, assumptions, and parameter values. As shown above, the capacity estimate can vary greatly depending upon the parameter values that are chosen. Although the stated assumptions and parameter values used for this exercise are deemed to be realistic, the accuracy of the capacity estimate is highly dependent upon the accuracy of the assumptions and parameter values used for the capacity estimate.

The following figure shows the relationship between the forward link noise rise and the throughput for several probability curves. The input parameters used to create the figure are shown below. The 50%-ile curve corresponds to the average rise.

Figure 3-17: Forward Link Rise vs. Throughput




95% 20% 1.7 87 57 49 5.5 3.6 3.1

90% 90% 1.8 92 61 52 1.3 0.9 0.7

90% 20% 2.2 106 70 60 6.7 4.4 3.8

Table 3-16: Forward Capacity per Sector for Various Probabilities of Rise - Vehicle

0

2

4

6

8

10

0 50 100 150 200 250

Throughput (Kbps)

No

ise

Ris

e (d

B)

98% 95% 90% 85% 75% 50%


3




• I-factor = 0.45 (3-sector cell site configuration)


• Forward link orthogonality factor = 0.6 (30 kmph)


The following figure shows the relationship between forward link noise rise and Erlangs of various data rates. The input parameters used to create the figure are shown below.

Figure 3-18: Forward Link Rise vs. Erlangs for Different Data Rates


Parameters:• Voice and data activity factor = 57.6%







0

1

2

3

4

5

6

7

8

9

10

0 5 10 15 20 25 30

Erlangs

No

ise

Ris

e (d

B)

Voice @ 9600 Data @ 19200 Data @ 38400 Data @ 76800


3


The curves in the figure above show the significant impact that data users can have on the capacity of a system. The voice and data activity factors were purposely set to the same value in order to reflect the capacity impact of just varying the data rate.

The following figure shows the relationship between the forward link total throughput and total Erlangs with respect to the data activity factor. The input parameters used to create the figure are shown below.

Figure 3-19: Forward Link Total Erlangs & Throughput vs. Data Activity Factor



• Peak noise rise = 8 dB








0

10

20

30

40

50

60

70

80

90%80%70%60%50%40%30%20%10%


To

tal

Th

rup

ut

(Kb

ps)

0.0

2.0

4.0

6.0

8.0

10.0

12.0

14.0

16.0

To

tal

Erl

ang

s

Total Thruput Total Erlangs


3


3.8 Forward vs. Reverse Link Capacity Comparison

The Reverse Link Noise Rise Capacity Estimation approach provided in Section 3.4 is almost identical to the Forward Link Noise Rise Capacity Estimation approach provided in Section 3.7. The following section will compare the IS-2000 1X capacity results of the forward and reverse links using the same assumptions and parameter values as stated in the previous sections (refer to the previous sections for specific details regarding the assumptions and parameter values used).

Figure 3-20 shows a comparison of the IS-2000 1X forward and reverse links for the noise rise vs. throughput capacity results for a 95% probability factor capacity estimation.

Figure 3-20: Forward and Reverse Link Rise vs. Throughput - 95% Probability Factor


Parameters: (unless otherwise noted, the parameters below apply to both forward and reverse links)• Traffic mix = Scenario B




• F-factor or I-factor = 0.45 (3-sector cell site configuration)

• Vehicular (30 kmph) Eb/No assumptions from Table 3-6 and Table 3-13 were used



0

2

4

6

8

10

0.0 20.0 40.0 60.0 80.0 100.0

Throughput (Kbps)

No

ise

RIs

e (d

B)

Fwd 95% Rev 95%


3


Figure 3-21 shows a comparison of the IS-2000 1X forward and reverse links for the noise rise vs. Erlangs, capacity results for the 9600 and 19200 bps data rates.

Figure 3-21: Forward and Reverse Link Rise vs. Erlangs for Different Data Rates


Parameters: (unless otherwise noted, the parameters below apply to both forward and reverse links)• Voice and data activity factor = 57.6%







The voice and data activity factors were purposely set to the same value in order to reflect the capacity impact of just varying the data rate.

0

1

2

3

4

5

6

7

8

9

10

0 5 10 15 20 25 30

Erlangs

No

ise

Ris

e (d

B)

Fwd Voice @ 9600 Fwd Data @ 19200 Rev Voice @ 9600 Rev Data @ 19200


3


Figure 3-22 shows a comparison of the IS-2000 1X forward and reverse links for Erlangs and throughput capacity vs. data activity factor.

Figure 3-22: Forward and Reverse Link Erlangs & Thruput vs. Data Activity Factor


Parameters: (unless otherwise noted, the parameters below apply to both forward and reverse links)• Traffic mix = Scenario B

• Forward peak noise rise = 8 dB

• Reverse peak noise rise = 10 dB








The results from all of the figures (Figure 3-20, Figure 3-21, and Figure 3-22) above, show the forward link with less capacity than the reverse link. In a general sense, the forward link may have less capacity than the reverse link, but the difference between the two may not be as wide as depicted in the figures above. Keep in mind that utilizing different assumptions and parameters

0.0

2.0

4.0

6.0

8.0

10.0

12.0

14.0

16.0

18.0

20.0

22.0

24.0

90% 80% 70% 60% 50% 40% 30% 20% 10%


To

tal

Erl

ang

s

0

20

40

60

80

100

120

Th

rou

gh

pu

t (K

bp

s)

Fwd Erlangs Rev Erlangs Fwd Thruput Rev Thruput


3


may be able to close the gap between the forward and reverse links, or even produce results which show the reverse link with less capacity than the forward link.

For example, Figure 3-23 shows a comparison of the IS-2000 1X forward and reverse links for Erlangs and throughput capacity vs. data activity factor using the following parameter changes mentioned below the figure.

Figure 3-23: Alternate Forward and Reverse Link Erlangs & Thruput vs. Data Activity Factor


Parameters: (All of the parameters for Figure 3-22 were used, except for the following changes)• Forward Eb/No @ 9600 = 4.4 dB (Figure 3-22 utilized a value of 6.34 dB)

• Forward Eb/No @ 76800 = 3.3 dB (Figure 3-22 utilized a value of 4.53 dB)

• Forward link orthogonality factor = 0.7 (Figure 3-22 utilized a value of 0.6)

The results in Figure 3-23 show that the forward link capacity is now equal to or slightly better than that of the reverse link. It is important to note that the parameter changes stated above are realistic parameters to use to model certain propagation environments (i.e. depending upon the multipath, ray imbalance, and geometry environment). Although the forward link may have a higher capacity than that of the reverse link (similar to the results in Figure 3-23) in some areas of a system, the general expectation is that the forward link will be the limiting factor from a capacity perspective (similar to the results in Figure 3-22, but maybe not as wide of a gap). Which link will be the limiting factor from a capacity perspective will depend upon the assumptions and parameter values used for a particular system analysis. As stated previously, the accuracy of the capacity estimate is highly dependent upon the accuracy of the assumptions and parameter values used for the capacity estimate.

0.0

2.0

4.0

6.0

8.0

10.0

12.0

14.0

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90% 80% 70% 60% 50% 40% 30% 20% 10%


To

tal

Erl

ang

s

0

20

40

60

80

100

120

Th

rou

gh

pu

t (K

bp

s)

Fwd Erlangs Rev Erlangs Fwd Thruput Rev Thruput


3


It should also be mentioned that these results are what the BTS sector may be able to support. The data applications being used by the subscriber unit may be more demanding on one link over the other. For instance, the user may request a file to be downloaded. To request the file will place a small load on the reverse link, but depending on the size of the file to be downloaded, the load on the forward link may be quite larger. This is commonly referred to as asymmetrical data transfer. This asymmetrical data transfer will be another reason why one of the links will be the limiting link with regards to capacity.

3.9 EIA/TIA Specifications and RF Air Interface Limitations

The CDMA RF Air Interface specifications defines the structure of the Forward and Reverse Channel. These specifications place an upper limitation on the number of channels that can be served by a CDMA frequency. The following sections provide Forward and Reverse Channel structure overviews for both IS-95 and IS-2000 Air Interface specifications.

3.9.1 IS-95 Forward Channel Structure

The following figure shows an example of the code channels transmitted by a base station. Out of the 64 code channels available for use, the example depicts the Pilot Channel (always required), one Sync Channel, seven Paging Channels (the maximum allowed), and fifty-five Traffic Channels.

Figure 3-24: Example of IS-95 Forward CDMA Channels

Code channels on the forward link are addressed by different Walsh codes. Each of these code channels is spread by the appropriate Pseudo-Noise Sequence at a fixed Chip Rate of 1.2288 Mega-Chips per second. The uniqueness of the forward channel structure is the use of the Pilot Channel. It is transmitted by each cell site and is used as a coherent carrier reference for demodulation by all subscriber stations. The pilot signal is unmodulated and uses the zeroth Walsh code which consists of 64 zeros. Hence, the pilot simply contains the I and Q spreading code. The choice of this code allows the subscriber to acquire the system faster. The Walsh codes are generated with a 64 x 64 Hadamard Matrix. Thus, the maximum number of code channels per carrier is 64 which consists of a Pilot Channel, a Sync Channel, a maximum of 7 Paging Channels and a minimum of 55 Traffic Channels (TCH). In view of the channel structure, a 1.23 MHz CDMA carrier can support up to 55 TCHs if the effect of interference is not considered. Another possible configuration could replace

PILOT CH SYNC CHPAGING

CH 1PAGING

CH 7TCH 1 TCH 55

WALSH 0 WALSH 32 WALSH 1 WALSH 63

CDMA FORWARD CHANNEL1.23 MHz

Traffic Power Control Data Sub-Channel (ADDRESSED BY WALSH CODE)

up to


3


Paging Channels and Sync Channels one for one with TCHs to obtain a maximum of 63 TCHs, 1 Pilot Channel, 0 Paging Channel, and 0 Sync Channel. In practice, due to the intense interference in the spectrum, a satisfactory quality of service in terms of voice quality and FER is difficult to maintain if all 55 traffic channels are implemented in the system.

The CDMA equipment requires a carrier frequency, a pilot offset, and a Walsh code to encode/decode the channel. The Base Station System (BSS) allocates a Traffic Channel in response to the Assignment Request message from the MSC. The BSS does not allocate traffic channels unless a request from the MSC is acknowledged. The Traffic Channel will be allocated in the sector with which the call is associated.

The BSS maintains a pool of Traffic Channels and Walsh codes in each sector for new call setups and soft/softer handoffs. Traffic Channel allocation for new originations and soft handoffs require an assignment of a physical Traffic Channel and a Walsh code. Softer handoff requires just the assignment of a Walsh code, no new Traffic Channel element has to be assigned. The assignment of Walsh codes and Traffic Channels is separated to allow the allocation process to adjust for the different needs of soft and softer handoff. In order to reduce the risk of soft/softer handoff assignment failure during the conversation, the BSS denies assignment of Traffic Channels and Walsh codes for new call setups if Traffic Channels or Walsh codes are not available or being used for soft/softer handoffs.

The number of Traffic Channels is defined by the In-Service Hardware in the BSS. It could be less than the number configured if some of the hardware is out of service. The number of Walsh codes assigned to a sector is set to 64 which is the maximum specified by the EIA/TIA standard. Limiting the number of Walsh codes in a sector is a method of controlling service quality. Since Walsh codes are not associated with any hardware, they cannot go out of service. As a result, 64 is the hard limit of the number of code channels per sector according to the protocol specifications.

3.9.2 IS-95 Reverse Channel Structure

The Reverse CDMA Channel is composed of Access Channels and Reverse Traffic Channels. These channels share the same CDMA frequency assignment. Each Traffic Channel is identified by a distinct user long code sequence and each Access Channel is identified by a distinct Access Channel long code sequence. The following figure shows as example of the signals received by a base station on the Reverse CDMA Channel.


3


Figure 3-25: Example of IS-95 Reverse CDMA Channels

The reverse link employs the same 32768 length binary short PN sequences which are used for the forward link. However, unlike on the forward link, a fixed code phase offset is used. A long PN sequence (242-1) with a user-determined time offset is used to identify the subscriber (analogous to ESN in AMPS). The sequence is then modulo-2 added with a 42 bit wide mask.

The subscriber unit convolutionally encodes the data transmitted on the Reverse Traffic Channel and the Access Channel prior to interleaving. The transmitted digital information is convolutional encoded using a rate 1/3 code of constraint length 9 for the Access Channel and for Rate Set 1 of the Reverse Traffic Channel. For Rate Set 2 of the Reverse Traffic Channel, the convolutional code rate is 1/2. The encoded information is then interleaved over a 20 ms interval. The interleaved information is then grouped in code words which consist of 6 symbol groups each. These code words are used to select one of the 64 orthogonal Walsh codes for transmission. On the reverse link, the Walsh codes are used for information transmission. The reverse CDMA frequency channel can support up to 62 TCHs per Paging Channel and 32 Access Channels per Paging Channel.

3.9.3 IS-2000 1X Forward Channel Structure

Support for IS-2000 was first introduced in Release 16. The following figure shows the Forward Channel Structure for IS-2000.

ACCESS ACCESSCH N

TCH 1 TCH M

CDMA REVERSE CHANNEL

1.23 MHz

CH 0

(ADDRESSED BY LONG PSEUDO-NOISE CODE)

(received at the base station)


3


Figure 3-26: Example of IS-2000 Forward CDMA Channels

3.9.3.1 IS-2000 Forward Channels (Nokia Siemens Networks Implementation)

The Common Assignment, Common Power Control, Common Control, and Broadcast Channels are not implemented in current or prior CBSC Releases. In the Common Pilot Channels, only the Forward Pilot Channel is implemented for CBSC Release 16 through the current release. The following sections provide a brief description of the forward channels that are supported for CBSC Release 16 through the current release.

Forward Pilot Channel (F-PICH)The IS-2000 Forward Pilot Channel is identical to the Pilot Channel in IS-95A/B, for backwards compatibility. It is transmitted by each cell site and is used as a coherent carrier reference for demodulation by all subscriber stations. The pilot signal is un-modulated and uses Walsh code 0, which consists of 64 zeros. A Walsh code can be expressed as a Walsh Function Wn

L, where n = Walsh code number and L = Walsh code length. The Walsh code for a F-PICH can be represented as W0

64. The Pilot Channels do not carry any information and essentially consist of Short PN codes. A Short PN code pair is generated by a modified linear feedback shift register. The pilot

Forward CDMA Channels

Quick PagingChannels(0-3)

[F-QPCH]

FundamentalChannel (0 or 1)

[F-FCH]

Supplemental CodeChannels (0-7)

[F-SCCH]

PilotChannel[F-PICH]

Transmit Diversity Pilot Channel[F-TDPICH]

Auxiliary PilotChannel

[F-APICH]

Auxiliary TransmitDiversity Pilot Chan.

[F-ATDPICH]

Common PagingChannels

DedicatedChannels

Dedicated Control Channel (0 or 1)

[F-DCCH]

SupplementalChannels (0-2)

[F-SCH]

PagingChannels (0-7)

[F-PCH]

CommonPilot

Channels

Common PowerControl Channel

[F-CPCCH]

Common ControlChannel

[F-CCCH]

CommonChannels

Common AssignmentChannel

[F-CACH]

SyncChannel

[F-SYNC]

BroadcastChannel[F-BCH]

= Channels NOT implemented in Current or Prior CBSC Releases


3


simply contains the I and Q spreading code.

Forward Sync Channel (F-SYNC)The Forward Sync Channel is used by the subscriber stations operating within the coverage area of the base station to acquire CDMA System time and Long PN code synchronization. It also transmits the system Protocol Revision (P_REV). There is only one Sync Channel per omni-carrier and there is a Sync Channel for each sector-carrier for sectored cells. The Sync Channel is spread by Walsh 32 of length 64 (W32

64), just as in IS-95A/B. The bit rate for the Sync Channel is 1200 bps and the frame is 26.67ms in duration. For CBSC Release 16 and later, the Sync Channel supports new redirection fields which can redirect subscribers to carriers that support Radio Configurations greater than 2 and the Forward Quick Paging Channel (F-QPCH).

Forward Paging Channel (F-PCH)The Forward Paging Channel functionality is basically the same as an IS-95A/B Paging Channel except that there exists new messages specified for IS-2000. The base station uses the Paging Channel to transmit overhead/SMS messages, pages, acknowledgements, channel assignments, and authentications to idle subscribers. IS-2000 supports up to 7 Paging Channels per sector-carrier, but as in earlier releases, CBSC Release 16 through the current release only supports 1 Paging Channel per sector-carrier. The primary Paging Channel number is Paging Channel number 1. This is the mode where the IS-2000 handset emulates an IS-95A/B handset. It is spread by a Walsh i of length 64 (Wi

64), where ‘i’ is the Paging Channel number. The bit rate that a Paging Channel uses is 9600 bps or 4800 bps.

Forward Quick Paging Channel (F-QPCH)The Forward Quick Paging Channel is introduced in IS-2000 to enhance the subscriber’s idle time battery life. It is used by the base station to inform subscriber stations, operating in the slotted mode (where the subscriber only “listens” during an assigned slot), that a page will be transmitted on the next designated slot on the Paging Channel. It is covered by Walsh code 80, 48, or 112 of length 128 (W80

128,W48128,W112

128). The bit rate for a Quick Paging Channel is 4800 or 2400 bps and it is divided into 2048 slots of 80ms duration (the same number of slots as a Paging Channel as determined by the slot cycle index). A subscriber will hash (based upon the IMSI) to 1 of 376 bits (for 4800 bps) or 1 of 188 bits (for 2400 bps) to determine whether it needs to monitor the Paging Channel slot for an impending page message. The slots are sub-divided into Paging Indicators and Configuration Change or Broadcast Indicators. Two Paging Indicators are transmitted in each QPCH slot for each subscriber station that will be paged in the associated Paging Channel slot.

Prior to the occurrence of the Quick Paging/Paging slot, the Access/Paging MCC determines the Paging Indicator bits based on the page messages found which support Quick Paging. It buffers the bits and transmits them when the Quick Paging Channel slot begins. As shown in Figure 3-27, approximately 20ms after the QPCH slot, the associated paging messages are transmitted on the Paging Channel. The Access/Paging MCC schedules only those page messages which have been quick paged on the QPCH slot which occurred 100ms prior to this PCH slot as shown in the figure. Each paging indication is a single bit at a data rate of 4800 bps or 2400 bps. The effective rate is 9600 bps or 4800 bps, respectively as each bit is sent twice (time diversity).

The base station enables the Configuration Change Indicators in each QPCH slot for a period of


3


time following a change in configuration parameters. Configuration Change Indicators are only used on QPCH number 1 and either 4 or 8 of the Paging Indicators are reserved for Configuration Change Indicators depending upon the data rate. Quick Paging capability allows a subscriber to conserve power and hence support extended battery life, by monitoring certain Paging Indicator bits within a Quick Paging slot on a Quick Paging Channel. The structure of the QPCH allows the use of a less complex demodulator which can enhance the battery life even further.

Figure 3-27: QPCH to PCH Timing

Forward Fundamental Channel (F-FCH)The Forward Fundamental Channel, as in IS-95A/B, is used for transmission of user and signaling information to a specific subscriber station for voice or low bit rate data applications during a call. RC 1 and RC 2 channels are backwards compatible to the TCH in IS-95A/B supporting data rates of 9600 or 14400 bps and 20 ms frames. As in IS-95A/B, this channel may be transmitted at a variable rate (on a frame-by-frame basis). New to IS-2000 is that each channel is transmitted on a different variable length Walsh code channel (expressed as Wn

L, where n = Walsh code number and L = Walsh code length). For RC 1 or RC 2 and RC 3 or RC 5, each channel is assigned to code channel Wn

64, where 1 < n < 63. For RC 4, each channel is assigned a code channel Wn128, where

1 < n < 127.

0 1 2 3

Paging Channel

Paging Channel Slot (80 ms)

1 2 3 4 11 2 3 4

Quick Paging Channel Slot (80 ms)

20ms 20ms

1

Quick Paging

Channel

Paging Indicators Paging Indicators

Configuration Change Indicators


3


Forward Dedicated Control Channel (F-DCCH)The Forward Dedicated Control Channel introduced in IS-2000 is used to carry user data as well as signaling and control data while the call is in progress. It does not support voice traffic. The Forward TCH channel may contain one Dedicated Control Channel. For RC 3 or RC 5, each channel is assigned to code channel Wn

64, where 1 < n < 63 and for RC 4, each channel is assigned

a code channel Wn128, where 1 < n < 127. This channel uses a data rate of 9600 bps (for RC 3 and

RC 4) or 14400 bps (for RC 5).

Forward Supplemental Channel (F-SCH)The Forward Supplemental Channel (packet based) introduced in IS-2000 is used for the transmission of user data to a specific subscriber station during a call. It is always accompanied by a dedicated FCH or DCCH. In IS-2000, the Forward Supplemental Channel is designed to reach data rates as high as 1,036,800 bps on a single RF carrier using a Spreading Rate (SR) of 3x. Also with IS-2000, each Forward TCH can have up to 2 Forward Supplemental Channels. For CBSC Release 16 implementation and through the current release, only 1 F-SCH per user with a maximum data rate of 153,600 bps will be supported using a Spreading Factor of 1x. These channels are shared resources which are allocated dynamically in order to meet the required data rate. The resources are scheduled into time slices which leads to a more efficient use of the channel elements. It supports variable data rates with the use of a variable length Walsh code. For RC 3 or RC 4, each channel is assigned a code channel Wn

L, where 1 < n < L-1 [L=4, 8, 16, 32, 64, 128 (where 128 is for RC 4 only)].

Forward Supplemental Code Channel (F-SCCH)The Forward Supplemental Code Channels are used to transmit user’s data from the base station to the subscriber station during a call and are primarily defined for backward compatibility with IS-95B for RC 1 and RC 2 only. The F-SCCH in IS-2000 can simultaneously use up to 7 Supplemental Code Channels in order to enable higher data speeds (for 3G-Type Services) on carriers under RC 1 and RC 2 and each channel is assigned a code channel Wn

64, where 1 < n <63. Nokia Siemens Networks’ implementation of the F-SCCH only supports 5 channels for RS1 and 4 channels for RS2 (similar to Nokia Siemens Networks’ implementation of IS-95B).These channels are dedicated resources which are assigned to a specific user to achieve data rates up to 64 kbps.

3.9.3.2 IS-2000 Forward Link Radio Configurations

The following table briefly explains the Radio Configurations (RC) supported by the forward link in IS-2000 for Spreading Rates (SR) 1 and 3.

Table 3-17: IS-2000 Forward Link Radio Configurations

RC SRData Rates

(kbps)Coding

RateModulation

RC 1 1 1.2, 2.4, 4.8, 9.6 1/2 BPSK

RC 2 1 1.8, 3.6. 7.2, 14.4 1/2 BPSK


3


Nokia Siemens Networks IS-2000 BSS Implementation for CBSC Release 16 Through the Current Release

The following table provides the forward link Radio Configuration and data rates that are supported with CBSC Release 16 through the current release.

RC 3 1 1.5, 2.7, 4.8, 9.6, 19.2, 38.4, 76.8, 153.6 1/4 QPSK

RC 4 1 1.5, 2.7, 4.8, 9.6, 19.2, 38.4, 76.8, 153.6, 307.2 1/2 QPSK

RC 5 1 1.8, 3.6. 7.2, 14.4, 28.8, 57.6, 115.2, 230.4 1/4 QPSK

RC 6 3 1.5, 2.7, 4.8, 9.6, 19.2, 38.4, 76.8, 153.6, 307.2 1/6 QPSK

RC 7 31.5, 2.7, 4.8, 9.6, 19.2, 38.4,

76.8, 153.6, 307.2, 614.41/3 QPSK

RC 8 31.8, 3.6. 7.2, 14.4, 28.8,57.6, 115.2, 230.4, 460.8

1/4 or1/3

QPSK

RC 9 31.8, 3.6. 7.2, 14.4, 28.8, 57.6,

115.2, 230.4, 460.8, 518.4, 1036.81/2 or

1/3QPSK

Table 3-18: Forward Link Radio Configuration Support for CBSC Release 16 Through the Current Release

RC SRData Rates

(kbps)Coding

RateModulation CBSC Notes

RC 1 1 1.2, 2.4, 4.8, 9.6 1/2 BPSKRate Set 1

Backward Compatible

RC 2 1 1.8, 3.6. 7.2, 14.4 1/2 BPSKRate Set 2

Backward Compatible

RC 3 11.5, 2.7, 4.8, 9.6,

19.2, 38.4, 76.8, 153.61/4 QPSK

Supported in 1X Mode only

RC 4 11.5, 2.7, 4.8, 9.6,

19.2, 38.4, 76.8, 153.61/2 QPSK Supported up to 153.6

RC 5 1 1.8, 3.6. 7.2, 14.4 1/4 QPSK Supported up to 14.4

Table 3-17: IS-2000 Forward Link Radio Configurations

RC SRData Rates

(kbps)Coding

RateModulation


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The following table shows the number of channel element (CE) resources that are required for the various data rates for RC 3 and RC 4.

The maximum data rate (153.6 kbps) supported on the forward link is obtained by utilizing RC 3 or RC 4. As shown above, RC 4 requires half as many CE resources compared to RC 3 to support the maximum data rate.

3.9.3.3 IS-2000 Walsh Code Allocation

Unlike IS-95A/B, the number of Walsh codes is not hard limited to 64 in IS-2000. To increase the number of usable Walsh codes, Complex or QPSK modulation is employed where 2 information bits are mapped into a QPSK symbol. Using the same coding rate, this method allows for an increase in the number of Walsh codes by a factor of 2 relative to BPSK, thereby allowing longer Walsh codes (i.e. 128 for RC 4, instead of 64). Implementing QPSK modulation, also allows doubling the original data rate on the same available bandwidth.

A Supplemental Channel in IS-2000 is designed to reach data rates up to 1,036,800 bps on a single RF carrier (refer to Section 3.9.3.2 above for the data rates supported by Nokia Siemens Networks). With the code chip rate fixed at 1228800 chips/sec, the length of the Walsh code spreading must be substantially reduced to achieve the high data rates.

The variable length Walsh code implementation can be visualized as shown in Figure 3-28. As seen in Figure 3-28, codes on different levels of the tree have different Walsh code lengths. The new levels in the tree are constructed by concatenating a root code word with a replica or an inverse of itself generating a long code word. During spreading, each bit is multiplied by an entire code word and longer codes are associated with lower bit rates. The root code word (which is shorter in length) is not guaranteed to be orthogonal to the derived long code words. The short code word is modulated exactly as the long code word is built and hence there is no way to differentiate the signals. Thus, if a root code is assigned to a certain user, then the derivative code words (the branches of the tree structure) should not be used because they are not orthogonal to the root code. Thus assigning a Walsh code at a particular rate will make some higher rate codes and some of the lower rate codes unavailable for assignment.

In this scenario if Walsh code C2,1 is assigned at a particular rate, Walsh codes C4,1 and C4,2 are not orthogonal to C2,1and hence they should not be assigned. At each level, all the code words are the rows of a Hadamard matrix.

Table 3-19: Forward Link Channel Element Resource Requirement

Data Rate(kbps)

Radio Configuration 3CE Resources


9.6 1 119.2 2 138.4 4 276.8 8 4153.6 16 8


3


Figure 3-28: IS-2000 Walsh Code Tree

Nokia Siemens Networks IS-2000 BSS Implementation for Release 16 Through the Current Release

For the multiple data rates achieved with the CBSC Release 16 through the current releaseimplementations, a maximum data rate of 153,600 bps is achieved with one F-SCH. A Walsh code allocation tree with a 153,600 bps maximum data rate is shown in Figure 3-29.

As seen in the figure, assigning a Walsh code at a particular rate would make some higher rate codes as well as lower rate codes unorthogonal and unavailable for assignment. WC0, WC1, and WC32 are reserved for Pilot, Page, and Sync channels respectively. The figure shows the number of Walsh codes available for each of the multiple data rates that CBSC Release 16 through thecurrent release supports. The "X" on some of the higher and lower data rate Walsh codes indicates that they are unavailable or reserved due to the Pilot, Page, and Sync Walsh code allocations.

C1,1=(1,1)

C2,1=(1,1)

C2,2=(1,-1)

C4,2=(1,1,-1,-1)

C4,1=(1,1,1,1)

C4,3=(1,-1,1,-1)

C4,4=(1,-1,-1,1)

1

C8,2=(1,1,1,1,-1,-1,-1,-1)

C8,3=(1,1,-1,-1,1,1,-1,-1)

C8,4=(1,1,-1,-1,-1,-1,1,1)

C8,5=(1,-1,1,-1,1,-1,1,-1)

C8,6=(1,-1,1,-1,-1,1,-1,1)

C8,7=(1,-1,-1,1,1,-1,-1,1)

C8,8=(1,-1,-1,1,-1,1,1,-1)

C8,1=(1,1,1,1,1,1,1,1)


3


Figure 3-29: Walsh Code Allocation Tree

Shorter length Walsh codes limit the number of simultaneous users in the forward link, because of the smaller Walsh code set. If the remaining two high rate (153,600 bps) Walsh codes are also assigned to data users as shown in Figure 3-30, all of the lower rate Walsh codes below those codes become unavailable (shaded Walsh codes). In this scenario, only 29 Walsh codes are available for voice call assignments (9600 bps) as seen in Figure 3-30 below.

Figure 3-30: Walsh Code Allocation Tree

X X

XX

X

9.6 kbps

19.2 kbps

38.4 kbps

153.6 kbps

76.8 kbps

XX

X

WC32 WC0WC1

X X

XX

X

9.6 kbps

19.2 kbps

38.4 kbps

153.6 kbps

76.8 kbps

XX

X

WC32 WC0WC1

X X

X

X

X

9.6 kbps

19.2 kbps

38.4 kbps

153.6 kbps

76.8 kbps

X

X

X

WC32 WC0WC1

X X

X

X

X

9.6 kbps

19.2 kbps

38.4 kbps

153.6 kbps

76.8 kbps

X

X

X

WC32 WC0WC1


3


3.9.4 IS-2000 Reverse Channel Structure

The following figure shows the Reverse Channel Structure for IS-2000.

Figure 3-31: Example of IS-2000 Reverse CDMA Channels

3.9.4.1 IS-2000 Reverse Channels (Nokia Siemens Networks Implementation)

The Reverse link in IS-2000 essentially consists of three new channels. They are Pilot, Supplemental, and Dedicated Control Channels, in addition to the IS-95A/B Access and Fundamental Channels. The following sections provide a brief description of the reverse channels that are supported for CBSC Release 16 through the current release.

Reverse Pilot Channel (R-PICH)The Reverse Pilot Channel introduced in IS-2000 is used to assist the base station in detecting subscriber station transmissions. There exists a Pilot Channel for each subscriber on a TCH in the uplink and it is used for the timing and phase reference to the BTS for coherent demodulation. As in the forward link, the pilot signal is un-modulated and it uses zeroth Walsh code 0 but of length 32 (W0

32). The Pilot Channels do not carry any information and essentially consist of Short PN codes. It allows the use of Walsh code and simultaneous channel transmission on the reverse link. It is only supported on Reverse RCs greater than 2, because RC 1 and RC 2 have to be compatible with IS-95A/B which does not support a Reverse Pilot Channel. The R-PICH also includes a

Reverse CDMA Channels

FundamentalChannel[R-FCH]

SupplementalCode Channels (0-7)

[R-SCCH]

Traffic Channels(RC 1, RC 2)

Enhanced AccessChannel

[R-EACH]+

PilotChannel

[R-PICH]

CommonControl

Channels

PilotChannel

[R-PICH]

Dedicated ControlChannel (0 or 1)

[R-DCCH]

FundamentalChannel (0 or 1)

[R-FCH]

SupplementalChannels (0-2)

[R-SCH]

AccessChannel[R-ACH]

AccessChannels

CommonControl Channel

[R-CCCH]+

PilotChannel

[R-PICH]

Traffic Channels(RC 3-6)

= Channels NOT implemented in CBSC Release 16t through the current release


3


Reverse Power Control Sub-Channel when operating on a TCH with RC 3 and RC 4. It is used by the subscriber station to transmit Forward Power Control commands to the base station.

Reverse Access Channel (R-ACH)The Reverse Access Channel functionality is the same as an IS-95A/B Access Channel, supporting RC 1 and RC 2, in order to allow for the backwards compatibility. It is identified by a Long PN Code offset. There are 32 Access Channels associated to one Paging Channel and the information on the Access Channel is transmitted at data rate of 4800 bps. Nokia Siemens Networks’ implementation supports only 1 Access Channel per Paging Channel prior to Release 21. Begining with Release 21, Nokia Siemens Networks supports 3 Access Channels per Paging Channel.

Reverse Fundamental Channel (R-FCH)For RC 1 and RC 2, the Reverse Fundamental Channel functionality is the same as in IS-95A/B. Only one Reverse Fundamental Channel can be used by the subscriber station during a call. As in IS-95A/B, it supports the basic rates of 9600 bps and 14400 bps. The R-FCH uses Walsh code 4 of length 16 (W4

16) for spreading. It supports orthogonal modulation with RC 1 and RC 2 and orthogonal spreading with RC 3 and RC 4. It performs discontinuous transmission using repetition coding, where a subscriber station operating with RCs 3 through 6 may discontinue transmission of the R-FCH for up to three 5 ms frames in a 20 ms frame.

Reverse Supplemental Channel (R-SCH)The Reverse Supplemental Channel introduced in IS-2000 is used for the transmission of user data to the base station during a call. An R-SCH is always accompanied by a dedicated R-FCH or R-DCCH. They operate with RCs 3 through 6 only (for CBSC Release 16 through the current release,Nokia Siemens Networks only supports RC 3 and RC 4 for the reverse link). There are up to 2 Supplemental Channels. The data rate is selected on a time slice basis and it supports data rates up to 307,200 bps. For spreading, R-SCH uses Walsh code W1

2 or W24. Although IS-2000 supports

up to 2 reverse Supplemental Channels, CBSC Release 16 through the current release supports only 1 R-SCH with a maximum data rate of 153,600 bps. If the second R-SCH were supported, it would use W2

4 or W68 for spreading.

Reverse Dedicated Control Channel (R-DCCH)The Reverse Dedicated Control Channel introduced in IS-2000 is used to carry user data as well as signalling and control information during a call. One Dedicated Control Channel may accompany an R-SCH, but the R-DCCH does not support voice traffic. The subscriber transmits at a fixed data rate of 9600 bps or 14400 bps and it uses Walsh code 8 of length 16 (W8

16) for spreading. It supports orthogonal spreading with RC 3 and RC 4

3.9.4.2 IS-2000 Reverse Link Radio Configurations

The following table briefly explains the Radio Configurations (RC) supported by the reverse link in IS-2000 for Spreading Rates (SR) 1 and 3.


3


Nokia Siemens Networks IS-2000 BSS Implementation for CBSC Release 16 Through the Current Release

The following table provides the reverse link Radio Configuration and data rates that are supported with CBSC Release 16 through the current release.

Table 3-20: IS-2000 Reverse Link Radio Configurations

RC SRData Rates

(kbps)Coding

RateModulation

RC 1 1 1.2, 2.4, 4.8, 9.6 1/3 64-aryOrthogonal

RC 2 1 1.8, 3.6. 7.2, 14.4 1/2 64-aryOrthogonal

RC 3 1

1.2, 1.35, 1.5, 2.4, 2.7, 4.8,9.6, 19.2, 38.4, 76.8, 153.6

1/4 BPSKw/Pilot

307.2 1/2

RC 4 1 1.8, 3.6. 7.2, 14.4, 28.8, 57.6, 115.2, 230.4 1/4BPSKw/Pilot

RC 5 3

1.2, 1.35, 1.5, 2.4, 2.7, 4.8,9.6, 19.2, 38.4, 76.8, 153.6

1/4 BPSKw/Pilot

307.2, 614.4 1/3

RC 6 3

1.8, 3.6. 7.2, 14.4, 28.8,57.6, 115.2, 230.4, 460.8

1/4 BPSKw/Pilot

1036.8 1/2

Table 3-21: Reverse Link Radio Configuration Support for CBSC Release 16 Through the Current Release

RC SRData Rates

(kbps)Coding


RC 1 1 1.2, 2.4, 4.8, 9.6 1/3 64-aryOrthogonal

Rate Set 1Backward Compatible

RC 2 1 1.8, 3.6. 7.2, 14.4 1/2 64-aryOrthogonal

Rate Set 2Backward Compatible

RC 3 11.5, 2.7, 4.8, 9.6,

19.2, 38.4, 76.8, 153.61/4

BPSKw/Pilot

Supported up to 153.6


3


The following table shows the number of channel element resources that are required for the various data rates for RC 3.

The maximum data rate (153.6 kbps) supported on the reverse link is with RC 3.

3.10 Handoffs

The new IS-2000 air interface provides the ability to handoff voice and data calls, as well as other services from an IS-95 system to an IS-2000 system and from an IS-2000 system to an IS-95 system. The following handoff methods are supported in both IS-95 and IS-2000 systems:

• Soft (or Softer) handoff

• Inter-CBSC Soft Handoff

• Hard handoff

3.10.1 Soft Handoff

A soft handoff is a handoff in which a new base transceiver station (BTS) commences communications with the subscriber station without interrupting the communications from the old BTS. The BTS can direct the subscriber station to perform a soft handoff only when all Forward Traffic Channels assigned to the subscriber station have identical frequency assignments. When performing a soft handoff, the subscriber collects the signal-to-noise ratio (pilot Ec/Io) received from each active sector on the downlink along with all candidate sectors. Each active BTS that receives the uplink transmission from the subscriber will relay it to the transcoder (XC). The XC will make the final decision on the eligibility of candidates and the handoff will proceed. While in

RC 4 1 1.8, 3.6, 7.2, 14.4 1/4BPSKw/Pilot

Supported up to 14.4

Table 3-22: Reverse Link Channel Element Resource Requirement

Data Rate(kbps)


9.6 119.2 138.4 276.8 4153.6 8

Table 3-21: Reverse Link Radio Configuration Support for CBSC Release 16 Through the Current Release

RC SRData Rates

(kbps)Coding



3


a soft handoff state, more than 1 TCH is assigned to the subscriber.

The soft handoff factor (SHOF) is used to determine the overhead Erlangs to support different kinds of soft handoffs. The factor is likely to vary from 1.3 to 2.0. It should be noted that the soft handoff factor defined here is a linear scaling factor of the actual usable Erlangs but not the number of traffic channels.

Soft Handoff Factor = 1*(1-a-b) + 2*a + 3*b [EQ 3-60]

where:• 2-way soft handoff fraction, a = Average 2-way soft handoff duration per hold time

• 3-way soft handoff fraction, b = Average 3-way soft handoff duration per hold time

3.10.2 Inter-CBSC Soft Handoff

Inter-CBSC Soft Handoff (ICBSC-SHO) happens when the subscriber communicates with sectors of different BTSs and the BTSs are controlled by different CBSCs. In a Nokia Siemens Networks system, when the subscriber reports a handoff pilot that refers to an external sector database that has inter-CBSC soft handoffs enabled, the call goes into inter-CBSC soft handoff. In this case, the external sector can reside in the source CBSC or can be backhauled from the target CBSC. The source CBSC remains in control of the call until no source handoff legs remain, then control is transferred to the target CBSC by a Anchor Handoff (which is a form of a hard handoff).

3.10.3 Hard Handoffs

Hard Handoffs take place during all "break before make" handoff situations. In an IS-95 and/or IS-2000 system, hard handoffs can be represented by a change from one radio configuration to another, or when a multi-mode subscriber station transitions from CDMA operation to operation on an analog system. In a Nokia Siemens Networks system, hard handoffs which result in the subscriber being supported by a new PDSN will cause the connection to the old PDSN to be dropped. The subscriber must then initiate a new PPP session as well as an IP registration following a hard handoff.

3.10.3.1 Anchor Handoff

Anchor Handoffs are handoffs triggered when a subscriber is in Inter-CBSC soft handoff, and a set of criteria have been met within the database. When the criteria are met (typically no source CBSC handoff legs are active), the target CBSC determines the current strongest Inter-CBSC soft handoff sector and initiates a hard handoff to that sector. The source CBSC maintains control of the call until the criteria is met, then control is transferred to the target CBSC resulting in a change in Walsh codes.

3.10.3.2 IS-95 to IS-2000 Hand-up

Hand-up from IS-95 to IS-2000 happens when an IS-2000 capable subscriber station is directed


3


from an IS-95 channel to an IS-2000 channel. In a Nokia Siemens Networks system, before allocating a channel element for a handoff request, the MM checks the Radio Configuration Class capability of the current sector against the candidate sector. If the candidate sector supports a higher Radio Configuration Class, the MM can pick a channel element with a higher Radio Configuration Class that is supported by the subscriber. For example, if a IS-2000 capable subscriber is on a call using IS-95 with RC 2 radio resources and wishes to add a leg from a BTS that has IS-2000 with RC 3 radio resources available, the MM could decide to perform a hard handoff and hand the subscriber up to the IS-2000 channel with RC 3 radio resources. Increasing the call to a higher Radio Configuration Class is referred to as a hand-up.

3.10.3.3 IS-2000 to IS-95 Hand-down

An IS-2000 to IS-95 hand-down happens when an IS-2000 capable subscriber station is assigned to an IS-2000 channel in the source BTS, and the target BTS has assigned an IS-95 channel. In a Nokia Siemens Networks system, the MM checks the Radio Configuration Class capability of the current sector against the candidate sector. If the candidate sector supports a lower Radio Configuration Class, the MM can pick a channel element with a lower Radio Configuration that is supported by the subscriber. The subscriber would then hand down to the IS-95 channel. An example of this is when the call starts out on an IS-2000 channel with RC 3 radio resources and the subscriber wishes to handoff to a BTS that does not have IS-2000 resources available. The MM could decide to perform a hard handoff and hand the subscriber down from IS-2000 to IS-95. As part of this handoff process, the source radio channel is also handed down to IS-95. Decreasing the call to a lower Radio Configuration Class is referred to as a hand-down.

3.10.3.4 Packet Data Handoffs

In a Nokia Siemens Networks system, when the base station determines that a Hard Handoff is required for a packet data call, the base station will transition a packet data call into dormant mode by initiating a call release. During the release procedure the base station sends the subscriber a Service Option Control message indicating the minimum amount of time the subscriber must wait before trying to transfer the packet data. The subscriber will attempt to access the system again using the best serving cell. Once access has been granted, the subscriber will resume the transfer of the packet data.

3.10.3.5 Inter-Carrier Hand-across

An IS-2000 to IS-2000 inter-carrier hand-across happens when an IS-2000 capable subscriber is assigned to an IS-2000 channel in the source sector, and the target sector can assign an IS-2000 channel. The subscriber would then handoff to the IS-2000 channel. In the case of the hand-across, the source and target sectors are located under two different frequencies, and a hard handoff to the IS-2000 target cell is required. This inter-carrier hand across case can also occur among IS-95 channels.


3


3.11 Budgetary Estimate of Sites for Capacity (Voice Only)

The following section provides a budgetary estimate of sites from a capacity perspective for a Chicago Metropolitan Area example. This example provides a simplified traffic engineering approach to estimating the number of sites required from a capacity perspective for an IS-95 voice only system. If some simplifying assumptions were made towards the voice and data call models, a similar approach could also be performed to estimate the number of sites required from a capacity perspective for a voice and data system as well (IS-95B and/or IS-2000 1X).

It is important to note that the site estimates provided in this section are for budgetary purposes only. Many other issues such as cell coverage, cell location, antenna configurations, unique traffic call models (voice and data), etc. have to be taken into consideration for an actual system design. It is recommended that simulations be performed using a tool like Nokia Siemens Networks’ Intelligent Design and Growth Planning (IDGP) for CDMA tool (see Section 3.12) before finalizing a system design.

This example illustrates the case that the cellular operator decides to deploy a single carrier CDMA system and allocate 1.8 MHz (including the guard band) out of the 12.5 MHz cellular band for CDMA deployment. The system shall be designed to provide service to 40,000 new CDMA subscribers. Prior to the design of the system, information concerning the propagation environment and subscriber distribution has to be gathered for each particular service area.

3.11.1 Required Parameters for Initial System Design

Prior to the design of an IS-95 voice only system, the propagation parameters and the subscriber profile must be available. This section is intended to give an overview of some important parameters and the correct way to apply them to system design. A completed example follows.

3.11.1.1 Busy Hour Call Attempts and Completions

Busy hour is defined as the continuous one hour period in the day during which the highest average traffic density is experienced by the system. Busy Hour Call Attempts (BHCA) is the number of call setup requests during the busy hour. Busy Hour Call Completion (BHCC) is the portion of the requests which succeed in making it to the conversation state.

3.11.1.2 Average Holding Time

Holding time is defined as the average length of time an active user occupies a traffic channel.

3.11.1.3 Erlangs per Subscriber

An Erlang is the traffic intensity of a traffic channel which is continuously occupied. Erlang per subscriber is the product of BHCA per subscriber and the average holding time per access.


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Figure 3-32: Subscriber Distribution of Chicago Metropolitan Area

CHICAGODOWNTOWN

NORTHWESTSUBURBS

UPTOWNAREA

WESTSUBURBS

SOUTHWESTSUBURBS

SOUTHSUBURBS


3


Table 3-23: Subscriber Distribution of Chicago Metropolitan Area

System Parameters:

• Spread Bandwidth = 1.23 MHz• Data Rate = 9600 bps (Rate Set 1)• Median (Eb/Io) = 7 dB• Power Control standard deviation = 2.5 dB• Voice or Data Activity Factor = 0.4• Noise Rise Threshold (Io/No) = 10

Assumptions:

1. Each subscriber’s required energy per bit-to-interference density ratio (Eb/Io) is varied according to propagation conditions to achieve the specified FER of 0.01

2. All the sectors support the same number of subscribers.

3. The subscribers are uniformly distributed over each sector.

4. There is no overflow from the CDMA network to the AMPS network

5. There are 40,000 subscribers distributed across the system as shown in Table 3-23.

6. The Average Hold time per Access is 65 seconds.

7. The path loss slope for a dense urban environment of 32.8 dB/decade is assumed with a shadowing standard deviation of 7.7 dB.

8. The path loss slope for an urban environment of 38.4 dB/decade is assumed with a shadowing standard deviation of 8 dB.

9. 40% of the subscribers will be in soft handoff between two or more sites.

10. The sectorization improvement going from a single sector to three sectors is 2.4 times.

AreaSubscriber

Distribution Environment

Classifications

BHCA per subscriber

1 City core area 50% dense urban 1.40

2 Northwest Suburb 25% suburban 1.40

3 Uptown area 10% dense urban 1.38

4 West Suburb 8% suburban 1.30

5 Southwest Suburb 5% suburban 1.30

6 South Suburb 2% suburban 1.20


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From the Figure 3-7, with fully loaded neighbor cells (worst case), the maximum capacity for 2% probability of blocking is approximately 15 Erlangs per CDMA sector for dense urban areas, and 17.8 Erlangs per CDMA sector for suburban areas.

These results in addition to following are approximations based on the curves and the assumptions which went into generating the curves. Actual system designs will vary from system to system.

For Area 1,

Number of subscribers in the city core = 40,000*50% = 20,000

Required traffic capacity for this area

= BHCA/sub * # of Sub * Average Hold Time per Access(sec) / 3600= 1.4 * 20,000 * 65 / 3600= 505.56 Erlangs (0.0253 Erlang per sub)

Required traffic capacity including soft handoff

= Required traffic capacity * soft handoff factor= 505.56 * 1.4= 707.78 Erlangs

Required number of CDMA sectors

= 707.78 / 15 Erlangs per CDMA sector= 48 CDMA sectors

Required number of CDMA sector cells

= 48 / 2.4 (2.4 is the sectorization gain)= 20 cells

For Area 2,

Number of subscribers in the city core = 40,000*25% = 10,000

Required traffic capacity for this area

= BHCA/sub * # of Sub * Average Hold Time per Access(sec) / 3600= 1.4 * 10,000 * 65 / 3600= 252.78 Erlangs

Required traffic capacity including soft handoff

= Required traffic capacity * soft handoff factor= 252.78 * 1.4= 353.89 Erlangs

Required number of CDMA sectors


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= 353.89 / 17.8 Erlangs per CDMA sector= 20 CDMA sectors

Required Number of CDMA sector cells

= 20 / 2.4= 9 cells

Using a sectorization gain of 2.45 for a three sector CDMA site, a total of 20 sector cells are required for area 1. Propagation studies have to be performed to determine if the system is coverage limited as opposed to capacity limited. If the number of sector cell sites required in this case for coverage is larger than 20 (the system is coverage limited), the system should be designed based on the number of cell sites required for coverage. Propagation studies could be a detailed system wide simulation or a simple link budget analysis based on certain well-known propagation model such as the Okumura Model or the Hata Model (depending on the degree of accuracy required).

By the same method, the calculation of the other areas is summarized in following table:

Table 3-24: Chicago Metropolitan Area Summary

Area BHCA

perSub

Subs.%

Subs. in

Region(k)

Required Traffic

(Erlangs)

SHO Factor

(Erlangs)

Required Traffic w/

SHO (Erlangs)

Max.Traffic

per Sector

RequiredSectorCells

1 1.40 50 20 505.56 1.4 707.78 15.0 20

2 1.40 25 10 252.78 1.4 353.89 17.8 9

3 1.38 10 4 99.67 1.4 139.53 15.0 4

4 1.30 8 3.2 75.11 1.4 105.16 17.8 3

5 1.30 5 2 46.94 1.4 65.72 17.8 2

6 1.20 2 0.8 17.33 1.4 24.27 17.8 1

Total 100 40 997.39 1396.35 39


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3.12 IS-95 and IS-2000 Simulations

Planning a wireless system revolves around three main characteristics: Coverage, Capacity, and Quality. In a CDMA system, these three characteristics must be carefully balanced against one another in order to arrive at the desired level of system performance. If high capacity is desired, there will be some degree of degradation in coverage and/or quality. Likewise, if a better system quality is desired, there will be some degree of degradation in capacity and/or coverage. The important point to realize is that these parameters are intertwined.

It is up to the system designer to determine how to balance these parameters to best serve a particular area. The best balance point will change from cell site to cell site depending on where that cell site is located in the system or the design objectives. Sites in dense downtown areas will trade off coverage for capacity. Conversely, cell sites at the edges of a system could sacrifice capacity for additional coverage.

The capacity of a CDMA site and system is dependent upon many factors which can be unique from one system to the next. Some of these factors that have an impact to both IS-95 and IS-2000 1X systems are:

• Propagation loss (path loss slope, log normal fading, antenna types)• Amount of delay spread in the environment• Terrain and clutter environment• Traffic distribution of the subscribers• Speed distribution of the subscribers• Voice/data call models and activity factors • Soft and softer handoff factors• Channel power settings (Pilot, Page, Sync, FCH, SCH, etc.)• Environmental characteristics (noise, interference from other services, etc.)• Level of reliability• Quantity and placement of sites, in addition to the amount of cell overlap

For IS-2000 1X, the dimensioning of a complex traffic model with variable data rates, which supports both circuit voice call models and packet data call models, creates a new challenge in capacity design. In IS-95 and IS-2000 1X, voice calls are handled by allocating dedicated channels. For IS-95B, data calls use dedicated supplemental code channels, but for IS-2000 1X, data calls employ shared supplemental channels. Therefore, the IS-2000 1X channel structure assures efficient use of the supplemental channels.

Various formulas can be used, dependent upon the level of complexity and accuracy desired, to estimate the capacity of a site. The more accurate calculations will require more time to perform or many simulations executed to obtain results which are statistically and reasonably valid. Due to the variability of the many different factors mentioned above, there is no single capacity number, but a range of values over an environment. The forward and reverse link capacity estimation equations provided in this chapter can only be used as an approximation of capacity of the system and should be used for budgetary purposes only. They do not take into account the size of the cell or the spacing between the sites. These equations do not totally account for the benefits of soft


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handoff, since they assume that all sites serving a given subscriber will experience peak rises at the same time, which in reality is a very small probability. In addition, these budgetary equations assume that the subscriber distribution is uniform, which is not likely. One sector may need to support many users, whereas the nearby sectors may be lightly loaded and therefore result in a lower f-factor which allows for greater capacity. A more accurate estimation can be performed with a more sophisticated CDMA simulation program, such as Nokia Siemens Networks’ Intelligent Design and Growth Planning (IDGP) for CDMA tool. IDGP can be used to model the forward and reverse links for thousands of subscribers in a realistic system environment with different voice and data traffic mixes.

The IDGP CDMA Simulator incorporates IS-2000 1X parameters to perform non time-sliced simulations. Non time-sliced simulations utilize a simulation technique where all of the dropped subscribers are actively bursting (although the power is adjusted according to an activity factor) at simulation time, and data rates assigned according to available capacity. The NetPlan tool, no longer supported, can be used for performing time-sliced simulations. For time-sliced simulations, data subscribers are modeled according to a dynamic source model, which employs a State machine consisting of the Reverse Request State, Server Delay State, Forward Reference or Download State, Think State, and Dormant State.

Various path loss models (statistical and deterministic) may be used by the simulator to aid in defining the CDMA coverage area. Each path loss model has its benefits and disadvantages. While most statistical models, such as Hata, do not consider terrain variation, they do allow for quick budgetary simulations. The Xlos propagation model incorporates terrain variation, antenna pattern, overlay (clutter) data, etc., in an attempt to model actual installations. The location of the CDMA subscriber units within a system will greatly affect total system capacity, coverage and quality, as well as the achieved data rate and distribution of resources. Subscriber positioning may be uniform or may be more accurately modeled with a subscriber traffic map.

In essence, the IDGP CDMA Simulator is a tool to layout a DS-CDMA system resulting in information on predicted capacity, required system parameter values, system quality, predicted coverage and hardware loading information. It permits investigations into real cellular system concerns such as edge effects, propagation anomalies, antenna types, subscriber distribution, call quality, receiver sensitivity impact on capacity, interference mitigation, power control and handoffs. It provides statistical information for the cell, and end-user. Cell statistics include the number of blocked subscribers due to unavailable Walsh codes, good subscriber percentage, total TCH power per data rate, forward and reverse SCH data rate, sector throughput and end user throughput, just to name a few. Because of CDMA system complexity and the inter-dependence between coverage, capacity and quality, it is only when these properties are considered together that a system representation with a higher degree of accuracy can be developed.

3.13 References

1. R.H. Owen, Phil Jones, Shirin Dehgan, Dave Lister, "Uplink WCDMA capacity and range as a function of inter-to-intra cell interference: theory and practice", pp. 298-302, VTC 2000.


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2. Szu-Wei Wang and Irving Wang, "Effects of Soft Handoff, Frequency Reuse and Non-Ideal Antenna Sectorization on CDMA System Capacity", pp. 850-854, IEEE 1993.

3. William C. Y. Lee, "Mobile Cellular Telecommunications Systems", McGraw-Hill Book Company, Second Edition 1995, figure 4.3, p. 110.

4. A. Viterbi & Viterbi, "Erlang Capacity of a Power_Controlled CDMA System", IEEE Selected Areas in Communications, August 1993, pp. 892-900.

5. A. Viterbi, "CDMA Principles of Spread Spectrum Communication", Addison-Wesley Publishing Company, Copyright 1995.

6. R. Padovani, "Reverse Link Performance of IS-95 Based Cellular Systems", IEEE Personal Communications Third Quarter 1994, page 28-34.

7. Charles Noblet, Ray Owen, Simon Burley, Allan Bartlett, “UMTS Network Dimensioning From Theory to Simulations”, version 1.00

8. CDG Evolution Study Report, Revision 4.01, January 10,2000

9. H. Holma & A. Toskala, "WCDMA for UMTS", John Wiley & Sons, Ltd, Copyright 2000, pp. 163-167.


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4 Link Budgets and
Chapter
4

Table of Contents

Coverage


4.1 Introduction 4 - 3

4.2 Radio Frequency Link Budget 4 - 44.2.1 Propagation Related Parameters 4 - 6

4.2.1.1 Building Loss. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 64.2.1.2 Vehicle Loss. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 94.2.1.3 Body Loss . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 94.2.1.4 Ambient Noise. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 94.2.1.5 RF Feeder Losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 94.2.1.6 Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 12

4.2.2 CDMA Specific Parameters 4 - 144.2.2.1 Interference Noise Rise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 144.2.2.2 Soft Handoff Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 184.2.2.3 Eb/No . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 19

4.2.3 Product Specific Parameters 4 - 204.2.3.1 Product Transmit Power . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 204.2.3.2 Product Receiver Sensitivity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 23

4.2.4 Reliability (Shadow Fade Margin) 4 - 294.2.5 Example Reverse (Uplink - Subscriber to Base) Link Budget 4 - 364.2.6 RF Link Budget Summary 4 - 40

4.3 Propagation Models 4 - 414.3.1 Free Space Propagation Model 4 - 414.3.2 Hata Propagation Model 4 - 434.3.3 COST-231-Hata Propagation Model 4 - 444.3.4 Additional Propagation Models 4 - 45

4.4 Forward Link Coverage 4 - 464.4.1 BTS Equipment Capabilities 4 - 47

4.4.1.1 4812T-MC BTS Minimum and Maximum Pilot Power . . . . . . . . . 4 - 524.4.1.2 Max Pilot Power for M810 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 534.4.1.3 UBS Max Pilot Power for Mix of 1X and DO Carriers (does not apply to M810). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 544.4.1.4 UBS Sector-Carrier Power Levels. . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 554.4.1.5 Power Out Differences Between UBS-Macro and 4812T/4812T-MC BTSs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 56

4.4.2 CDMA Signal Power Distribution Characteristics and PA Sizing 4 - 564.4.3 General Power Relationships 4 - 57


4

CDMA/CDMA2000 1X RF Planning Guide4 Link Budgets and Coverage

4.4.4 Design Guidelines 4 - 584.4.4.1 Comparison to Average Rated Power . . . . . . . . . . . . . . . . . . . . . . 4 - 594.4.4.2 Comparison to High Power Alarm Rating. . . . . . . . . . . . . . . . . . . 4 - 594.4.4.3 Comparison to Walsh Code Limit . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 60

4.4.5 General Power Requirements 4 - 604.4.5.1 Minimum ARP Based on LT-AVG Estimate . . . . . . . . . . . . . . . . . 4 - 604.4.5.2 Minimum HPA Based on VST-AVG Estimate . . . . . . . . . . . . . . . 4 - 614.4.5.3 Exceeding the High Power Alarm Rating . . . . . . . . . . . . . . . . . . . 4 - 624.4.5.4 Carrier Load Management Overview . . . . . . . . . . . . . . . . . . . . . . 4 - 62

4.4.6 Power Allocation in Mixed Mode 1X and DO Systems 4 - 644.4.7 Government Regulations 4 - 64

4.4.7.1 Power Amplifier Operational Measurements (FR9235). . . . . . . . 4 - 64

4.5 CDMA Repeaters 4 - 664.5.1 CDMA Repeater Design Considerations 4 - 66

4.5.1.1 Coverage Impact . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 664.5.1.2 Cascaded Noise Figure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 694.5.1.3 Interference and Capacity Issues . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 744.5.1.4 Filtering Issues . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 74

4.5.2 CDMA Repeater Installation Considerations 4 - 754.5.2.1 Antenna Isolation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 754.5.2.2 Repeater Antenna Considerations . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 774.5.2.3 Repeater Gain Settings. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 78

4.5.3 CDMA Repeater Optimization Considerations 4 - 804.5.3.1 Timing Impacts . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 804.5.3.2 Optimization Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 82

4.5.4 CDMA Repeater Maintenance Considerations 4 - 824.5.4.1 Future Expansion Considerations . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 824.5.4.2 Environmental Changes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 844.5.4.3 Operations and Maintenance Considerations. . . . . . . . . . . . . . . . . 4 - 84

4.6 Extended Range Cells 4 - 844.6.1 Extended Range Cell RF Planning and Design 4 - 85

4.6.1.1 Tower Top Amplifiers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 854.6.1.2 Reverse Link Budget Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 864.6.1.3 Forward Link Budget Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 884.6.1.4 Extended Range Cell Site Design Limitations . . . . . . . . . . . . . . . 4 - 904.6.1.5 Site Selection Criteria for Extended Range Cells . . . . . . . . . . . . . 4 - 99

4.6.2 Extended Range Cell Optimization Considerations 4 - 1014.6.2.1 PN Offset Planning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 1014.6.2.2 Parameter Optimization Impacts . . . . . . . . . . . . . . . . . . . . . . . . . . 4 - 102

4.7 Theoretical vs. Simulator 4 - 103

4.8 References 4 - 105


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4.1 Introduction

The RF design of a wireless system revolves around three main principles. Those principles are coverage, capacity and quality. The coverage of a system relates to the area within the system that has sufficient RF signal strength to provide for a quality call. The capacity of a system relates to the ability of the system to support a given number of users. Finally, the quality of the system relates to the ability of being able to adequately reproduce analog voice with a digital system. With CDMA, all three of these quantities are interrelated. To improve quality, some coverage and capacity has to be sacrificed. To improve coverage, capacity and quality would be sacrificed. Finally, to improve capacity, coverage and quality would be sacrificed.

The CDMA system design process consists primarily of three levels or phases. These levels range from an initial budgetary design to a final design used to implement the system. The amount of time and effort required to complete a design increases as the design process moves from a budgetary design to a final design. However, this additional time and effort results in a more accurate system design.

The first level of the design process is a budgetary level. It uses the RF link budget along with a statistical propagation model (such as Hata or COST-231 Hata) to estimate the coverage of the sites and ultimately determine how many sites are required for the particular system. This type of propagation model has a slope and intercept value for each type of environment (Urban, Suburban, Open, etc.) and does not include terrain effects. This relatively simplistic approach allows for a quick analysis of the number of sites that may be required to cover a given area.

The next level of a system design requires a more detailed propagation model. This propagation model takes into account the characteristics of the selected antenna, the terrain, and the land use and land clutter surrounding the site. Since these factors are accounted for, this propagation model will determine a better estimate of the coverage of the sites than the previous statistical propagation model. Thus, its use, in conjunction with the RF link budget, produces a more accurate determination of the number of cells required. This second level of the design process uses the reverse RF link budget to assist in determining the required propagation path loss. Nokia Siemens Networks uses the Intelligent Design and Growth Planning (IDGP) tool for this portion of the design process.

However to complete a system design, the forward link must also be analyzed to determine power settings and pilot coverage. The forward RF link budget consists of many variables including subscriber speed, location, soft handoff, noise figure, voice activity, and pilot range. It is recommended that a simulation be used to analyze the forward link by accounting for the statistical variation in these parameters. Such simulator studies are part of the final design phase.

The final level or phase of the design process incorporates further detail into the design by the use of simulation studies. Nokia Siemens Networks uses the IDGP CDMA Simulator for this analysis. The simulation studies account for subscriber distributions within a coverage area and also for CDMA system and site level parameters. The simulator analyzes both the forward and the reverse links. This final design process is required in the deployment of a system or in determining warranty coverage.

The one element common to all three levels of a system design is the RF link budget. The following


4


section discusses this element in greater detail.

4.2 Radio Frequency Link Budget

There are two main purposes for establishing the RF link budget for CDMA designs. The first purpose is to establish system design assumptions for all of the gains and losses in the RF path (such as vehicle loss, building loss, ambient noise margin, maximum subscriber transmit power, etc.). The second purpose of a link budget is to establish an estimate for maximum allowable path loss. This maximum allowable path loss number is used in conjunction with the propagation model to estimate cell site coverage, which ultimately determines the number of cells required for adequate system RF signal coverage and hence the system cost. Figure 4-1 shows the impact to the quantity of sites required due to changes in the RF link budget. For example, if the RF link budget (maximum allowable path loss) was improved by 5 dB, approximately half the number of sites would be required.

Figure 4-1: Percentage of Cells Based on dB Changes to the Link Budget

The above figure is derived using the COST 231 Hata Suburban propagation model. Other models may differ slightly from this. This figure can be utilized as a quick aid to help quantify the number of sites required based upon a change made to the RF link budget. It should be pointed out that other environmental factors may contribute to the above not holding true. For instance, in a very hilly terrain location, dB improvements may not provide for extra range if the terrain is blocking the propagation.

The system designer will need to determine the specific RF link budget parameters to be used when designing the system. The parameters within the RF link budget can be divided into four major


4


categories. The following lists some of these parameters:

1. Propagation related• Building Loss• Vehicle Loss• Body Loss• Ambient Noise• RF Feeder Losses• Antenna Gain

2. CDMA specific• Interference Noise Rise (other users)• Eb/No• Processing Gain (ratio of bandwidth to data rate)

3. Product specific• Product Transmit Power• Product Receiver Sensitivity

4. Reliability• Shadow Fade Margin

The following figure shows the typical gains and losses that are encountered in the RF link.

Figure 4-2: RF Link Budget Gains & Losses

A RF link budget must be determined for each sector of each site. The RF link budget for each sector must incorporate any specific parameters that have been supplied (such as building

BTS

Sub.

Subscriber Line LossSubscriber Antenna Gain

Body LossVehicle LossBuilding LossMan-made Noise

RF Path Loss

Shadow Fade Margin (Reliability)

BTS Antenna GainTransmission Line LossJumpers & Connector Loss

RFGains

&Losses

Subscriber Tx PowerSubscriber Rx Sensitivity

BTS Tx PowerBTS Rx Sensitivity


4


penetrations, antenna heights, antenna gains, cable losses, coverage criteria, coverage reliability, etc.). It is common that all sectors of a given site may have the same link budget or even that several sites may have the same link budget due to common installation practices being followed. If this is the case, then the same link budget can be used for all of the similarly configured sectors. However, if the parameters change from sector to sector and site to site, then separate link budgets will need to be calculated for each unique sector.

CDMA RF link budgets may make simplifying assumptions regarding noise rise and Eb/Norequirements. For instance, in the RF link budget, the Eb/No value is considered a constant. In actuality, Eb/No is not a constant value but varies with respect to speed, delay spread and other factors. Some of the simplifying assumptions are addressed in the detailed design phase.

4.2.1 Propagation Related Parameters

Propagation related parameters are those gains or losses of a link budget that are constant, independent of the multiple access technology chosen or vendor. The values of these parameters, though, are frequency dependent (i.e. differences would exist between an 800 MHz design and a 1900 MHz design or between a mobile and a fixed environment). These parameters include such factors as: building loss, vehicle loss, body loss, man-made noise margin, RF feeder losses, and antennas. If comparing link budget information between vendors, these propagation related parameters should be set the same so as to obtain a realistic comparison.

4.2.1.1 Building Loss

Building loss is associated with the degradation of the RF signal strength caused by a building structure, when a subscriber handset operating within a building is communicating with a base station. An adequate RF signal strength within a building can be accomplished in one of two ways. One method involves the propagation scenario, where a base station located outdoors communicates with a subscriber unit that is inside a building (see Figure 4-3). The second method involves the propagation scenario, where both the base station and the subscriber unit are within the same building.

Figure 4-3: In-Building Propagation Scenarios

INTOPropagation Scenario where a basestation communicates with a radio

transceiver that is inside a building.

WITHINPropagation Scenario where boththe transmitter and receiver are

within the same building.


4


For this chapter on link budgets, only building losses associated with the building penetration of the RF signal from an outdoor source are considered (refer to Figure 4-3, the diagram labeled INTO). Refer to Section 7.2 of Chapter 7 for further information on in-building designs.

One approach for modeling the “into” building penetration is as an extension of an outdoor propagation model. This method uses a distance-dependent path loss for a subscriber unit that is outside a building, and adds a building loss factor.

This typical approach adds building loss factor to the macro cell link budget. This building loss is highly variable and is a function of such items as: construction material, building layout, user location inside the building, proximity to the base station, and direction from the base station.

Building losses can range anywhere from 5 to 40 dB or more. If actual field data is not available for a given area, a value of building penetration may be assumed. The following table of values can be used for a mobile design as a possible guideline in the absence of field data for the particular environment:

This table of building losses represents the average difference in RF signal strength between the outside environment and numerous points throughout the inside of the building.

Another approach is that radio transmission into buildings should be undertaken separately and not as an extension of the outdoor propagation models plus the building loss factor. Besides the antenna heights and path length, the floor area, number of rooms on the floor, angle of illumination of the building to the base station and the construction of the walls should be considered when trying to determine a new propagation model. This approached is not addressed in this planning guide.

For a fixed system, the subscriber unit is not moving around inside the building but is instead fixed to a position. Since the Fixed Wireless Terminal (FWT) unit is stationary, the installation should be in a position that allows for the best signal to be received from the base station. The preferred installation is to have the FWT with its whip antenna located near a window, preferably on the side of the building closest to the base station. This would minimize the loss required for the signal to penetrate into the building. In addition, the preferred FWT location would have it being mounted above desk height. If this optimum location is achieved, the building loss will be minimized. Careful placement of the fixed wireless terminal’s antenna near a window could reduce the building loss value down to a 3 to 6 dB value. The following figure shows the preferred location of the FWT with whip antennas. Refer to the FWT vendor to determine the recommendations of the FWT placement.

Table 4-1: Example Building Penetration Losses (800 & 1900 MHz)

Environment Penetration LossDense Urban 20 dBUrban 15 dBSuburban 10 dBRural 8 dB


4


Figure 4-4: Preferred FWT Locations Without External Antennas

There are numerous papers that exist which describe building penetration losses. The papers cover many different factors that affect building loss such as: height of base station antennas, angle of illumination to the building, differing heights of buildings, various building constructions, and the impact of frequency on building loss. Some of these papers are contradictory. For example, a paper by Turkmani1 2 concluded that building penetration losses decrease with an increase in frequency. On the other hand, Aguirre3 reached the conclusion that higher penetration losses were experienced at higher frequencies. It should be pointed out that Turkmani’s study had antennas above the rooftop, whereas Aguirre’s study had antennas below the rooftop.

Due to the differences in the papers, an assumption for building penetration loss can be made by utilizing the results that are from a test case more in line with how the operator plans to provide for the building penetration.

As the floors of a building are ascended, the relative signal strength increases. This effect is usually attributed to the increased probability of line of site propagation between the higher floors of the building and the base site. This is commonly referred to as a height gain per floor. This height gain can effectively reduce the building loss by approximately 1.3 to 2 dB per floor. Since the normal design is for a worst case scenario, the height gain would not be considered unless the particular design is to provide coverage only to a given floor(s).

1. Turkmani, Parsons and Lewis, "Measurement of building penetration loss on radio signals at 441, 900 and 1400 MHz", Journal of the Institution of Electronic and Radio Engineers, Vol. 58, No. 6 (Supplement), pp. S169-S174, September-December 1988

2. Turkmani and Toledo, "Modelling of radio transmissions into and within multistory buildings at 900, 1800 and 2300 MHz", IEEE Proceedings-I, Vol. 140, No. 6, December 1993

3. Aguirre, "Radio Propagation Into Buildings at 912, 1920, and 5990 MHz Using Microcells", 0-7803-1823-4/94 IEEE, session 1.6 & 1.7, pp. 129-134

GoodReception

BetterReception

Install FWT near windowthat faces the generaldirection of the cell site.


4


4.2.1.2 Vehicle Loss

Vehicle loss is the degradation of the RF signal strength caused by a vehicular enclosure. A subscriber handset communicating to a cell site from within a vehicle will have a lower signal strength than if that same subscriber unit was operating outside of the vehicle. Vehicle loss has been seen to range from 5 to 12 dB. If the design for a system is to include a vehicle penetration loss, an average range is approximately 5 to 8 dB.

Due to the nature of a fixed system, vehicle loss should not be accounted for.

4.2.1.3 Body Loss

Body loss, also referred to as head loss, is the degradation of the RF signal strength due to the close proximity of the subscriber handset antenna to the person’s body. A 2 dB loss is associated with the antenna in a vertical position; 6 dB is associated with the antenna in a horizontal position. It is assumed that the typical user will rotate the phone or move slightly to help improve the quality of the call. Therefore, a lower body loss of 2 to 3 dB is often used in system designs.

For a fixed system, there will be no body loss since the FWT antenna is either connected directly to the FWT or is installed outdoors.

4.2.1.4 Ambient Noise

The ambient noise defines the environmental noise that is in excess of kTB for the sector. This noise could be generated from automobiles, factories, machinery, and other man-made noise. The ambient noise margin parameter can be added to the link budget to allow for an adjustment to the thermal noise value. Since each environment is unique, a noise floor study should be performed to determine if an adjustment is required to the theoretical thermal noise floor value.

Man-made noise is less significant at 1900 MHz than at 800 MHz. Also, galactic or sky noise is at a minimum.4

4.2.1.5 RF Feeder Losses

RF feeder losses include all of the losses that are encountered between the base station cabinet and the base antenna, or with respect to a subscriber unit, all of the losses between the PA and the antenna. Since a majority of subscriber units for a mobility system being sold to customers are portable, there is minimal feeder loss; therefore, RF feeder loss at the subscriber unit is not considered in the link budget. However, the feeder loss at the base site can account for several dB of loss. The example RF link budgets provided in Table 4-6 and Table 4-7 only reflect the line loss at the base site.

For a fixed system, the Fixed Wireless Terminal (FWT) may have an antenna connected directly to the unit or the antenna may be installed on the outside of the building, thus requiring a

4. Lee, William C.Y. "Mobile Communications Engineering", Copyright 1982, McGraw-Hill Inc. pg. 33-40.


4


transmission run from the FWT to the antenna. For the scenario of an external antenna connected to an FWT, subscriber unit feeder loss needs to be accounted for in the RF link budget. This feeder loss would be the loss encountered from the FWT to the external antenna, which is a function of the size of transmission line and the length of the run. Since this transmission line may need to wind its way from the FWT to the external antenna, the size of the line may be small to allow for better bending radii. A lightning arrestor will also need to be accounted for in this subscriber unit feeder loss.

The base station RF feeder line loss calculations include such losses as: top jumper, main transmission line, bottom jumper, lightning arrestors (surge protector), connectors, duplexers, splitters, combiners, and couplers (see Figure 4-5). The loss associated with the RF feeder system can be minimized by reducing the transmission line run between the base station and its antennas, and/or utilizing lower loss transmission lines. Transmission lines can range from 1/2” to 1-5/8”, or greater, diameter cables. The larger the diameter of the cable, the less lossy the medium, but the sacrifice is more rigid lines, larger bending radius, greater weight, more wind loading and larger area required. Transmission lines are also available with either air or foam dielectrics. The air dielectric cables are more expensive to install and maintain, but are less lossy than the foam lines. Figure 4-5 reflects most of the different components that are encountered between the base site antenna and the base station equipment.

When estimating the amount of transmission line loss, keep in mind that the line loss is frequency dependent. Transmission cables are more lossy at higher frequencies. At 800 MHz, a 7/8” line may suffice, but a 1-5/8” line for 1900 MHz may be required to maintain a similar loss. The following table shows an example of the difference that can exist in transmission line loss as a function of the operating frequency.

Consult the transmission line vendor for the specifications of the installed transmission line or the system operator, if actual field measurements have been made.

Table 4-2: Example of Main Transmission Line Losses

850 MHz 1900 MHz

7/8” Foam Dielectric Coaxial Cable 1.24 dB/100ft.4.07 dB/100meters

1.97 dB/100ft.6.46 dB/100meters

1-5/8” Foam Dielectric Coaxial Cable 0.77 dB/100ft.2.54 dB/100meters

1.25 dB/100ft.4.1 dB/100meters


4


Figure 4-5: Typical Components in the RF Feeder Run

Additionally, the reference point used in the base station specifications should be known. For instance, the duplexer loss and its jumpers/connectors to the base station may already be included in the specifications for the base station’s noise figure and PA output. Typically, the specifications for the base station are at the top of the frame. Therefore, if the duplexer or other components are located within the base station frame, additional loss would not need to be factored in. If, on the other hand, the device is located external to the base station frame, this loss would need to be accounted for.

For sites with multiple CDMA carriers, the Rx signal distribution and the Tx combining schemes are typically addressed within the equipment specifications of the base station frame. If combining or splitting of the RF signal is being performed external to the base station frame, the loss associated with the combining or splitting would need to be added to the link budget.

From a budgetary or approximation viewpoint, one RF feeder loss value could be assumed as the typical value for all of the sites. In real world situations, however, it is rare that one loss value will

Antenna

(A) Top Jumper

(B) Main Transmission Line

(C) Antenna Surge Protector

(D) Jumper to Directional Coupler

(E) Directional Coupler

(F) Jumper to Duplexer

(H) Jumper to Tx and Rx Antenna Port

BTS

Waveguide Entry Port

Note: Each Jumper consists of: Two connectors and One line

(G) Duplexer


4


be common for all of the sites. Some sites (and sectors) may have longer or shorter lengths of transmission line due to being installed with a taller antenna supporting structure or due to the base station equipment being located on the top of a building.

In performing propagation predictions, it is important that each site (sector) is represented as accurately as possible. Therefore, an analysis should be done for each particular sector to determine the RF feeder line loss. This calculation should include all losses between the antenna and the base station such as those components depicted in Figure 4-5. The value of the line loss listed in Table 4-6 is an example which assumes that the base station will be operating at 1900 MHz and the main transmission antenna run is 30 meters (approximately 100 feet). A 1-5/8” heliax cable at 1900 MHz has approximately 4.1 dB loss per 100 meters (1.25 dB loss per 100 feet). Another 0.75 dB was assumed for jumpers and connectors.

Refer to Chapter 6, Section 6.7.3 for additional information on transmission lines.

4.2.1.6 Antennas

Antennas can be either omni or directional. Omni antennas provide approximately the same amount of gain throughout the entire 360° horizontal pattern. Directional antennas, sometimes referred to as sector antennas, have a maximum gain in one direction with the backside being 15 to 25 dB below the maximum gain.

The gain of the antenna is a function of the horizontal pattern, vertical pattern, and number of elements that make up the antenna array. The number of elements will dictate the size of the antenna. The horizontal and vertical beamwidths are referenced as the amount of degrees between the points on the pattern where the gain is down 3 dB from the maximum gain.

The following points should be considered when selecting an antenna:

• The size and weight of the antenna will impact tower loading or the ability to place the antenna in the optimum position.

• Typically, antenna patterns with narrower horizontal and/or vertical beamwidths will result in a higher antenna gain, assuming similar lengths.

• The horizontal and vertical beamwidths will have an impact upon the performance of the site at the locations midway between the sectors. The larger horizontal beamwidths will result in more overlap of signal between sectors and thus increase the amount of softer handoff between sectors and soft handoff with other sites. This impacts the amount of interference seen (thus impacting capacity) and the ability to contain pilot pollution.

• The front to back ratio of the antenna also impacts the amount of interference seen at other sites and the ability to minimize pilot pollution.

The horizontal and vertical patterns provided by the selected antenna should be verified to ensure that there will be coverage in the desired area. For instance, as a means to improve forward gain of the antenna, the vertical beamwidth may be reduced. In some situations, this reduction in the vertical beamwidth may produce unsatisfactory signal strengths near the cell site tower due to the antenna overshooting the area to be covered.


4


Another item to keep in mind is whether the antenna gain is in reference to a dipole or an isotropic antenna. The difference is usually signified by dBd or dBi. A zero dBd gain antenna would correspond to a 2.14 dBi gain antenna. Cellular often referenced antennas in dBd, but PCS RF link budgets normally refer to dBi gain antennas. The important point to be made is that a propagation model may be referenced to an isotropic or dipole antenna. Thus, care should be taken to ensure the correct antenna gain is used with the propagation model, and thereby avoiding a potential error of 4.3 dB.

Refer to Chapter 6 for additional information on antennas.

4.2.1.6.1 Base Station Antenna

The antennas located at the base site can be either omni or directional. In early cellular designs, most sites started out as omni. Fewer antennas were required and the system was lightly loaded. As the traffic requirements grew, sites were required to be sectorized to provide for this additional traffic and to restrict the amount of co-channel and adjacent channel interference.

PCS systems at 1900 MHz initially did not require an abundance of capacity, but utilized directional antennas because of the extra gain associated with a directional antenna as compared to an omni antenna. A 4 dB improvement could easily be achieved by using directional antennas instead of omni antennas. This 4 dB improvement could potentially reduce the quantity of sites required at 1900 MHz by approximately 40%.

It is not mandatory that all sites use the same antenna. The system planner may deploy either omni or directional antennas at a cell site to meet the coverage goals desired.

As mentioned above, the antennas need to be selected to ensure coverage will be provided over the desired area. In addition, antennas need to be selected to minimize the level of interference. Decreasing the level of interference will allow for greater site capacity and improved system performance. Antenna patterns that provide a faster rolloff past the half power points (i.e. 3 dB down from main lobe) will provide for better interference protection. In frequency reuse systems (AMPS, GSM, USDC), improved interference control, such as through the use of sectorized sites, allows for a set of frequencies to be used at closer distances (i.e. tighter reuse pattern), thus providing increased capacity. For CDMA, as mentioned in the chapter on capacity (Chapter 3), interference from other cells and other sectors has an impact on the capacity that can be supported.

4.2.1.6.2 Subscriber Unit Antenna

Our assumptions here are that the portable subscriber unit antenna has a gain of 0 dBi (-2.14 dBd) without factoring in body loss and is an omni antenna. It is possible that differences may exist. The system could be designed for mobile coverage, in which case, the antenna mounted on the external of the vehicle may have higher gain.

Another scenario is a fixed application. An option for the FWT is to have a whip antenna connected directly to the FWT unit. This whip antenna gain may differ based upon product or vendor. Another option is that the FWT installation may utilize yagi or patch antennas with much greater gain and directivity.


4


In some circumstances for a fixed application, particularly for users in fringe coverage areas, external antennas are appropriate alternatives to the simple whip antenna. The vendor of the FWTs should be contacted to determine what antenna options may be available.

4.2.2 CDMA Specific Parameters

CDMA specific parameters are those items in the RF link budget which will have different values based on the technology chosen. CDMA parameters include such factors as: interference margin, soft handoff gain and Eb/No.

4.2.2.1 Interference Noise Rise

In determining RF coverage in CDMA systems, the effect of interference generated from other users on the serving cell as well as the neighboring cells must be considered. As discussed in Chapter 3, this is in contrast to the RF coverage analysis for AMPS cells where interference mainly affects the frequency assignment, but not the coverage.

The interference noise rise margin is dependent upon the amount of loading assumed in the system. Different cell deployment strategies can be modeled by varying the interference margin. CDMA cell deployments could be based on loading individual frequencies one by one, until they achieve the target load (for instance, a 6 dB noise rise). An alternative deployment could utilize more CDMA radio carriers, initially operating at a reduced load, to further extend the range of the cells (for instance, 3 dB noise rise) while trading off capacity (exploiting any immediate spectrum available). This 3 dB system rise improvement would result in approximately 30% fewer CDMA cell sites at system turn-on.

The following equation can be used as a first pass approximation for the amount of interference margin to be added to the reverse RF link budget to account for loading the CDMA system with users.

[EQ 4-1]

Where X is the system load, specified as a fraction of pole capacity. For example, a cell site operating near full capacity has X equal to seventy-five percent (75%). Noise rise varies as a function of propagation, environment, load, user distribution, etc.

The derivation for [EQ 4-1] can be shown as follows.

Assuming a CDMA system with subscribers in the cell of interest and perfect reverse link power control such that the power received at the base site due to each subscriber unit is the same,

, the signal to noise plus total (in-cell and out of cell) inbound

interference ratio on the traffic channel can be defined as:

NoiseRise 101

1 X–------------ log=

N

Pr1Pr2

.... PrNPr= = =


4


[EQ 4-2]

[EQ 4-3]

[EQ 4-4]

Where: is the signal to noise plus total interference ratio

is the power (in Watts) received at the base site from each individual in-cell

subscriber unit. Note that, although the power received at the base site from a particular subscriber unit is a function of several factors (i.e. subscriber unit’s transmit power, subscriber unit antenna gain, base site antenna gain, individual path loss and fading), the reverse link power control ensures that the received power from any subscriber unit in the cell is approximately at the same level .

is the spread bandwidth (in Hz) of the CDMA system

is the thermal noise power spectral density (in Watts/Hz) at the input to the

receiver Low Noise Amplifier (LNA)

, is the interference power spectral density (in Watts/Hz) from all

of the subscriber units within the cell at the input to the receiver LNA. Note that, in the cell of interest, out of a total of subscriber units, only one subscriber unit is the one of interest, hence there are interfering subscriber units.

is the voice activity factor or the fraction of time voice is transmitted during a call

is the interference power spectral density (in Watts/Hz) from all of the subscriber

units in other cells at the input to the receiver LNA and is the function of their respective path loss characteristics, load, size and power control

is the figure of merit for digital systems and is defined as energy per bit to noise

plus total interference power spectral density ratio

, is the Processing Gain of the CDMA system. MHz for

an IS-95 and IS-2000 1X CDMA system and is the bit rate of the traffic

channel (e.g. 9.6 kbps traffic channel or 14.4 kbps traffic channel).

SNRPr

NoW IoW IocW+ +( )---------------------------------------------------=

Eb

Nt------ SNR PG⋅ SNR

WRb------⋅= =

SNREb Nt⁄W Rb⁄---------------=

SNR

Pr

Pr

W

No

Io

N 1–( )αPr

W----------------------------=

NN 1–

α

Ioc

Eb Nt⁄

PG W Rb⁄= W 1.2288=

Rb


4


Furthermore, the frequency reuse factor or the F-factor of a cell is defined as the ratio of inbound interference from subscriber units within the cell (intra-cell) to the total inbound interference from subscriber units in all the cells (including the cell of interest). Since each subscriber unit is a potential interferer, F-factor is given by

[EQ 4-5]

Some references to the frequency reuse factor may be in terms of out of cell interference to in cell interference (f = OutCell/InCell). The frequency reuse factors F and f can be equated with the following equation:

[EQ 4-6]

Substituting the value of into [EQ 4-5] results in:

[EQ 4-7]

Substituting the value of into [EQ 4-2] and dividing both numerator and denominator by ,

can be rewritten as

[EQ 4-8]

[EQ 4-9]

Substituting the value of F-factor from [EQ 4-7] into [EQ 4-9] results in,

[EQ 4-10]

[EQ 4-11]

FInCell

InCell OutCell+--------------------------------------------

Io

Io Ioc+-----------------= =

F1

1 f+-----------=

Io

FN 1–( )αPr

N 1–( )αPr IocW+----------------------------------------------=

Io NoW

SNR

SNRPr

NoW N 1–( )αPr IocW+ +----------------------------------------------------------------

Pr N( oW )⁄

1N 1–( )αPr

NoW----------------------------

IocW

NoW------------+ +

--------------------------------------------------------= =

SNRPr N( oW )⁄

1NαPr IocW+( )

NoW-------------------------------------

αPr

NoW-----------–+

-----------------------------------------------------------------Pr N( oW )⁄

1N 1–( )αPr IocW+

NoW----------------------------------------------+

--------------------------------------------------------= =

SNRPr N( oW )⁄

11

NoW-----------

N 1–( )αPr

F----------------------------⋅+

------------------------------------------------------Pr N( oW )⁄

1α N 1–( )

F----------------------

Pr

NoW----------- +

-------------------------------------------------= =

SNRs

1αF--- N 1–( )s+

----------------------------------=


4


where, .

Solving [EQ 4-11] for s,

[EQ 4-12]

where is defined as the loading factor of the system and is given by

[EQ 4-13]

The upper bound on the number of users or the pole capacity of the cell of interest can be obtained by substituting into [EQ 4-13] and replacing SNR with [EQ 4-4] to yield:

[EQ 4-14]

The system rise is defined as the ratio of thermal noise plus total inbound interference to thermal noise and is given by

[EQ 4-15]

From [EQ 4-2] and [EQ 4-15] yields the following:

[EQ 4-16]

Substituting the value of from [EQ 4-12] in [EQ 4-16] results in:

or [EQ 4-17]

R (dB) is the median rise. In other words, noise rise is above (or below) this level 50% of the time. This is due to the voice activity ( ) term used in the SNR calculation.

Figure 4-6 is a graphical representation of [EQ 4-17] and plots rise versus the loading factor X. From the plot, 50% loading corresponds to a rise of 3 dB and 75% loading corresponds to a 6 dB rise.

s Pr N( oW )⁄=

sSNR

1αF--- N 1–( ) SNR⋅–

----------------------------------------------- SNR1 X–------------= =

X

XαF--- N 1–( ) SNR⋅=

X 1=

Npole 1F

α SNR⋅-------------------+ 1

Fα---

W Rb⁄Eb Nt⁄---------------⋅+= =

RNoW I+ oW IocW+

NoW----------------------------------------------=

RPr N( oW )⁄

SNR--------------------------- s

SNR-----------= =

s

R1

1 X–------------= R (dB) 10 1 X–( )log–=

α


4


Figure 4-6: Rise (dB) at the cell of interest versus X (% load) at the cell of interest

4.2.2.2 Soft Handoff Gain

Soft handoff is the term that is normally associated with the fact that a CDMA system makes a connection to a target cell prior to releasing (breaking) from the source site, commonly referred to as “make-before-break”. A hard handoff, associated with AMPS, GSM, or USDC, requires that the signal strength from the target cell be greater than the signal strength from the source cell by a hysteresis value in order to reduce the number of handoffs per call and the “ping-pong” effect. This hysteresis requires an overlap between the cell coverage areas. The soft handoff gain corresponds to a decreased shadow fade margin required by the CDMA soft handoff over that of a hard handoff system.

Some proponents of CDMA may have a separate entry in the RF link budget for soft handoff gain. The purpose of this is to provide information as to the benefits of CDMA over other technologies. Some system designers believe that the soft handoff gain should be accounted for in the reliability value (shadow fade margin). The example RF link budget provided in a later section incorporates the soft handoff gain in with the shadow fade margin. Refer to the section on Reliability for further discussion on the shadow fade margin.

For a fixed system, the gain offered by soft handoff may or may not be present depending upon the system design. For instance, a single isolated site supporting a fixed system would have no neighboring sites to even allow soft handoff to occur. In this situation, the soft handoff gain would be zero. Another situation is for a fixed system utilizing external FWT antennas. These directional antennas tend to be sited to the best signal source and therefore minimal advantage from soft


4


handoff would be recognized. Even for the situation of a fixed system using the FWT whip antennas, soft handoff gain may be lower than seen in a mobile environment. The FWT installation causes a form of building directionality which may decrease the soft handoff advantage.

4.2.2.3 Eb/No

Eb/No corresponds to energy per bit over interference plus noise density for a given target Frame Erasure Rate (FER, typical voice FER target is 1%). In digital communications, it is customary to designate one-sided noise density with No. In CDMA, interference is dominated by the noise generated due to other users in the system. The notation, No, in this section refers to the total power density due to interference and noise.

Included in the CDMA Eb/No value is diversity improvement arising from performance in Rayleigh fading. This is distinct from the entry “Soft Handoff Gain” which represents an estimate of the performance improvement of soft handoff, relative to hard handoff, when experiencing log normal shadowing.

In general, the required downlink Eb/No, to provide an acceptable audio quality, improves at higher speeds and in soft handoff. In the uplink path, the required Eb/No improves at lower speeds (which is the opposite of the downlink). The worst case Eb/No value for voice communication on the uplink is at about 30 kmph.

The uplink Eb/No value accounts for rake (non-coherent combining) receiver, dual antenna, and interleaving/coding. The downlink Eb/No value accounts for rake (coherent, maximal ratio combining) receiver, and interleaving/coding.

For mobile systems, the Eb/No target varies dynamically as the subscriber moves around. However, FWTs are fixed and the only movement is that of people around the FWT in a building and large vehicles or pedestrians close to an outdoor FWT antenna. Optimized FWT deployment may significantly reduce the Eb/No target by avoiding the fading caused by the surrounding environment.

In a mobile environment, the fading characteristic is Rayleigh. For a fixed system, the fading environment may be more Rician. The Eb/No value assumes a certain type of fading environment. The Eb/No requirement for a fixed system will therefore be different than for a mobile environment. The Eb/No target value may range from 4 dB to 8 dB for CDMA fixed systems. The Eb/No target value should be set to 8 dB for isolated cells using indoor omni FWT antennas or for cells with little SHO benefits in the fringe areas. However, if external directional FWT antennas are used and a Line Of Site (LOS) path exists between the cell site and the FWT antenna, an Eb/No target value of 4 dB may be used.

As improvements are made to the hardware (chip sets) and to the software (how the energy is managed), the Eb/No requirement level may be lessened. Typical Eb/No values used for fixed systems are stated above. The early requirements for a mobile system are approximately 7 to 7.5


4


dB for the 8 kbps and 13 kbps vocoder respectively. With the latest chip sets (e.g. Qualcomm CSM5000, EMAXX), the Eb/No values are approximately 1.5 to 2 dB less for voice communications.

IS-95A and IS-95B assume the same Eb/No values. For the IS-2000 RF reverse link, there are separate Eb/No values provided for the fundamental channel rate and each supplemental channel rate. The Eb/No values for the supplemental channel rates (19.2 kbps and greater) are less than the fundamental Eb/No. Two main factors are contributing to this. A higher FER for the higher data rates may be targeted as compared to lower FER for lower data rates for speech (9600 bps e.g.). This will reduce the required Eb/No. The RF link budget shown in Table 4-7 assumes an FER of 5% for the supplemental channel rates and an FER of 1% for the fundamental channel. It is viewed that the radio link protocols (RLP) will allow for relaxed FER requirements for the supplemental channel. The control channel information carried on the fundamental channel requires the better FER. Turbo coding is the other factor contributing to the lower Eb/No value for the supplemental channels. Turbo coding improves upon the error correction at the higher data rates. The higher the data rate, the larger the benefit from Turbo coding (Turbo coding gain grows as the number of bits sent increases for a given frame size) which results in a lower Eb/No for a given FER target.

From a link budget analysis, only one Eb/No value can be assumed for a given scenario. The appropriate Eb/No value to be used in the RF link budget is based upon the system design assumptions (base station equipment and vocoder rate).

The Nokia Siemens Networks IDGP CDMA Simulator incorporates a family of curves to more accurately account for the Eb/No requirements needed to meet a desired FER for each link that is being analyzed between the user and the site. Refer to Section 4.7 for additional discussion on the simulator.

4.2.3 Product Specific Parameters

Product specific parameters are those items in the RF link budget which can vary based on the product (base station and subscriber) chosen. There may be differences between products within Nokia Siemens Networks’ base station product line, such as differences in PA power. Differences will also exist between different equipment vendors. Each equipment vendor will have their own vision of the type of market their equipment is to satisfy.

4.2.3.1 Product Transmit Power

The transmit power is typically referenced by the power output of the piece of equipment prior to the RF transmission lines and antennas. The point at which the transmit power is being measured needs to be determined to ensure that there are no gains or losses left out of the link budget.

4.2.3.1.1 Subscriber Unit

The IS-95A standard provides the maximum effective radiated power (ERP) for any class of


4


personal station transmitter in Table 6.1.2.1-1. The Class II personal station is not to exceed 2.5 Watts (34 dBm). For the Class III personal station, the minimum ERP is 0.2 Watts (23 dBm) and the maximum ERP is 1 Watt (30 dBm).

The CDMA standard for 1.8 to 2.0 GHz (ANSI J-STD-008) provides the maximum effective isotropic radiated power (EIRP) for any class of personal station transmitter in Section 2.1.2.1. The Class I personal station is not to exceed 2 Watts (33 dBm). For the Class II personal station, the minimum EIRP is 0.2 Watts (23 dBm) and the maximum EIRP is 1 Watt (30 dBm).

There is a slight difference between the PCS and Cellular specifications. Cellular references the output power with respect to a dipole (ERP), whereas PCS makes reference to an isotropic radiator (EIRP). Therefore, there is approximately a 2 dB difference between the specifications given in the standards documents.

The latest version of 3GPP2 C.S0011, Recommended Minimum Performance Standards for cdma2000 Spread Spectrum Mobile Stations, also provides a table of radiated powers for the various band classes that exist.

The typical subscriber value to be used in the reverse link (uplink - subscriber transmit to base receive) is 23 dBm.

With respect to the reverse RF link budget, one parameter could be used for the transmit power of the subscriber unit (the EIRP or ERP value) or it may be desirable to break up this value into three parts. The three parts are: subscriber PA output, transmission line and connector losses, and the antenna gain.

Since the subscriber unit, portable or FWT, can be purchased from different vendors, the specifications for each subscriber unit should be obtained.

With IS-95B, high speed packet data is supported by concatenating multiple RF channels on the forward link (Walsh codes). To enable the concatenation of multiple channels, IS-95B compatible subscriber units are required. IS-95B HSPD was not implemented on the reverse link, thus only one RF channel is supported on the reverse link. It is assumed that the IS-95B subscriber unit’s physical characteristics will be the same as those that were used for IS-95A voice. If a different device is used for data than for voice, the subscriber PA output, transmission line and connector losses, and the antenna gain parameters would need to be determined.

With IS-2000, high speed packet data is supported with the use of supplemental channels. IS-2000 compatible subscriber units are required to support this air interface specification. With IS-2000, the reverse link can support multiple channels (e.g. reverse pilot channel, fundamental channel, supplemental channel). The example IS-2000 reverse RF link budget in Table 4-7 has two additional rows to show the amount of power that would be dedicated to the fundamental or dedicated control channel and to the supplemental channel (for reverse data rates greater than 9.6 kbps). The following definitions were obtained from the IS-2000 specifications.


4


• The Reverse Fundamental Channel (R-FCH) corresponds to a portion of the Reverse Traffic Channel, which carries higher-level data and control information from a subscriber station to a base station.

• The Reverse Supplemental Channel (R-SCH) corresponds to a portion of Radio Configuration 3 through 6 Reverse Traffic Channel, which operates in conjunction with the Reverse Fundamental Channel or the Reverse Dedicated Control Channel. The Reverse Supplemental Channel will provide higher data rate services, and is the channel on which higher-level data is transmitted.

• The Reverse Dedicated Control Channel (R-DCCH) corresponds to the portion of a Radio Configuration 3 through 6 Reverse Traffic Channel used for the transmission of higher-level data and control information from a subscriber station to a base station.

• The Reverse Traffic Channel corresponds to a traffic channel on which data and signaling are transmitted from a subscriber station to a base station. The Reverse Traffic Channel is composed of up to one Reverse Dedicated Control Channel (IS-2000), up to one Reverse Fundamental Channel (IS-95A/B or IS-2000), zero to two Reverse Supplemental Channels (IS-2000), and zero to seven Reverse Supplemental Code Channels (IS-95B).

The subscriber unit transmit power associated with the R-FCH or R-DCCH is dependent upon the processing gain and Eb/No requirements associated with the fundamental channel. The subscriber unit transmit power associated with the R-SCH is dependent upon the processing gain and Eb/Norequirements associated with the data rate of the R-SCH (19.2, 38.4, 76.8 or 153.6 kbps). When a supplemental channel is required, some of the subscriber unit’s transmit power needs to be allocated for the R-FCH or R-DCCH. The remaining transmit power can be utilized for the R-SCH. The difference in the transmit power between the R-SCH and the R-FCH or R-DCCH is based on the difference of the processing gain and Eb/No requirements of the different channels. The following set of equations provide a method to determine the transmit powers for the various reverse traffic channels.

PT = PFCH + PSCH

PSCH = 10^[(Processing_Gain_DeltadB + Eb/No_DeltadB)/10] * PFCH

PT = PFCH + 10^[(Processing_Gain_DeltadB + Eb/No_DeltadB)/10] * PFCH

PFCH = PT / {1 + 10^[(Processing_Gain_DeltadB + Eb/No_DeltadB)/10]}

Where:PT is the total subscriber unit transmit power available (mW)

PFCH is the portion of the total subscriber unit transmit power available for the reverse fundamental channel or reverse dedicated control channel (mW)

PSCH is the portion of the total subscriber unit transmit power available for the reverse supplemental channel (mW)


4


The following set of calculations provide an example of how the subscriber unit transmit powers associated with the R-FCH and R-SCH for the 19.2 kbps data rate, represented in Table 4-7, were obtained. A similar approach would be followed for each of the other supplemental channel rates.

PT = 200 mW

PFCH = 200 /{1+10^[(10*Log(19200/9600)+(3.5-5.6))/10]}

= 89.7 mW

PSCH = 200 - 89.7 = 110.3 mW

= 10 * Log(PSCH) = 20.4 dBm

4.2.3.1.2 Base Station

The CDMA standard for 1.8 to 2.0 GHz (ANSI J-STD-008) in Section 3.1.2 states that the base station shall not transmit more than 1,640 Watts of effective isotropic radiated power (62.1 dBm EIRP) in any direction in a 1.25 MHz band for antenna heights above average terrain less than 300 meters. The base transceiver station power is used in the forward link (downlink - base transmit to subscriber receive).

With respect to the forward RF link budget, one value could be used for the transmit power of the base station (the EIRP value) but typically this value is separated into three parts. The three parts are: base station PA output, transmission line and connector losses, and the antenna gain. The subscriber units are typically more uniform, having similar line losses and antenna gains. The base station, on the other hand, can vary quite a bit from one base station to the next. Based on the configuration of the site, location of antennas with respect to the base station infrastructure, and power out required, it is not as simple to have one EIRP value that is common across the majority of the sites. Since each base station site can be unique, the uniqueness of the site needs to be accounted for to ensure the appropriate EIRP is being designed for. For instance, one site may require a 100 ft. run of main transmission line, whereas another site may only require a 50 ft. run. The additional loss for the longer run would alter the EIRP from the site. Another difference would exist based on differences of antennas and their associated gains.

The power output of the base station is normally assumed to be the power out at the top of the cabinet. It is possible that each vendor will have different transmit powers for their equipment. In addition, one vendor may have different transmit powers for each product in their portfolio of base station products. Obtain the specifications for the particular base station(s) that will be used in the system design. In looking at the specifications, the power amplifiers may be for multiple carriers or for a single tone (carrier). Refer to Section 4.4 for additional information on the Nokia Siemens Networks BTS PAs.

4.2.3.2 Product Receiver Sensitivity

The sensitivity of a radio receiver is a measure of its ability to receive weak signals. The following


4


equation can be utilized in calculating the sensitivity of a radio receiver.

[EQ 4-18]

Where:k Boltzmann’s constant = 1.38x10-23 W/(Hz K)

T Room temperature in degrees Kelvin = 290 K

W Bandwidth of the carrier = 1228800 Hz

NF Noise figure of the equipment

Eb/No Energy bit density over noise

R Information bit rate

[EQ 4-19]

The processing gain, PG, is the result of the bandwidth (W) divided by the data rate (R). For IS-95 Rate Set 1 (8 kbps vocoder), the data rate is 9600 bps. The resulting processing gain for this case is obtained as follows:

PG = W/R = 1228800 / 9600 = 128

PGdB = 10 * Log (128) = 21.1 dB

The following table provides the data rate (R) and the resulting processing gain for various Rate Sets and radio configurations. The data rates provided in the table are those that are supported in CBSC Release 16.0 through the current release. Refer to the latest IS-95A/B and IS-2000 standards for all of the data rates that exist in the air interface standards.

Table 4-3: Processing Gain

Air Interface Reverse Link Radio Configurations Data Rate (bps) Processing Gain (dB)

IS-95A/B Rate Set 1 - Standard 8 kbps Vocoder or EVRC (Enhanced Variable Rate Coder)

9600 21.07

IS-95A/B Rate Set 2 - 13 kbps Vocoder 14400 19.31

IS-2000 1X Reverse Link Radio Configuration 1 9600 21.07




RxSensitivity kT( )dBm Hz⁄ WdB Hz⋅ NF( )dB Eb No⁄( )dB W R⁄( )dB–+ + +=

RxSensitivity 113– dBm NF( )dB Eb No⁄( )dB PGdB–+ +=


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IS-95B supports high speed packet data, but because of the data applications that were being deployed, only the fundamental rate was provided on the reverse link. Therefore, the above table only provides the processing gain for the two different fundamental rates.

Differences in the receive sensitivity will exist between the subscriber unit and base station due to the differences in Eb/No values, as discussed in Section 4.2.2.3, and the noise figure of the equipment. The other parameters in the receive sensitivity calculation will be the same for both ends of the link.

4.2.3.2.1 Base Station

The noise figure, or NF, of a network is a value used to compare the noise in a network with the noise in an ideal or noiseless network. It is a measure of the degradation in signal-to-noise ratio (SNR) between the input and output ports of the network. Noise factor (F) is the numerical ratio of NF, where NF is expressed in dB. The equation for converting noise factor to noise figure is:

[EQ 4-20]

Typically the noise figure value to be used in determining the receiver sensitivity value can be obtained from the specification sheet for the particular product. The noise figure for the base station is approximately 6 to 7 dB maximum with a typical value of approximately 4.5 dB. Consult the base station equipment vendor for the specifics.

In some instances, a tower top amplifier (TTA) may be installed at a site to improve the level of the received signal at the base station. The TTA includes an amplifier and therefore a new noise figure needs to be determined since the configuration now has cascaded amplifiers. A TTA will only benefit the reverse path (subscriber to base station). Since the TTA is only improving the reverse link, the forward link may become more of the limiting path. It may be that a larger power amplifier is needed in the forward link in order to balance both paths.

For a TTA scenario as mentioned above, it will be necessary to calculate the noise figure of a group of amplifiers that are connected in series. This can be accomplished if the noise figure of each




IS-2000 1X Reverse Link Radio Configuration 4 (Only the fundamental is initially supported by Nokia Siemens Networks.)

14400 19.3

Table 4-3: Processing Gain

Air Interface Reverse Link Radio Configurations Data Rate (bps) Processing Gain (dB)

NF dB( ) 10 F( )log=


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individual amplifier is known. The equation for determining the cascaded noise factor is:

[EQ 4-21]

Where:Fn is the noise factor of each stage

Gn is the numerical gain of each stage (not in dB)

The equation for converting Gain dB to linear Gain is:

[EQ 4-22]

One important point to be made with respect to [EQ 4-21] is that if the gain of the first stage G1 is

sufficiently high, the denominators of the subsequent terms will force those terms to be small, leaving only F1. Therefore, the NF of the first stage will typically determine the NF of the cascaded

configuration.

The NF of two or more cascaded lossy networks can be found by simply adding the losses (in dB) of each network element. Examples of a lossy network element are: transmission lines, jumpers, duplexers, filters and mixers. If a duplexer with an insertion loss of 0.5 dB is followed by a main transmission line loss of 3 dB, the combined noise figure of this cascaded network is 3.5 dB.

The following figure shows two different sites. One site has an amplifier located on the top of the tower. The other site is the more conventional site, that has no additional amplification beyond the base station. This diagram will be used to run through an example showing the noise figure improvement with the TTA. In this diagram, stage 2 in the tower top amplifier example and stage 1 of the without tower top amplifier example represent cascaded lossy network elements which are able to be summed together.

FTotal F1

F2 1–

G1---------------

F3 1–

G1G2---------------

F4 1–

G1G2G3---------------------…+ + +=

G dB( ) 10 GLinear( )log=


4


Figure 4-7: Example of Two Different Receive Path Configurations

The following table lists the noise figures, noise factors, and gains for each stage shown above.

Table 4-4: Receive Path Noise Figures and Gains

With Tower Top Amplifier Without Tower Top AmplifierNF1 2.5 dB F1 1.78 NF1 3.0 dB F1 2.0

NF2 3.5 dB F2 2.24 NF2 6.0 dB F2 3.98

NF3 9.5 dB F3 8.91

G1 12.0 dB G1 15.85 G1 -3.0 dB G1 0.5

G2 -3.5 dB G2 0.45

Antenna

Jumper to Antenna

Main Transmission Line

Antenna Surge Protector

Jumper to Directional Coupler

Directional Coupler

Jumper to Duplexer

Jumper to Tx and Rx Antenna Port

BTS

Waveguide Entry Port

Duplexer

Tower Top Amplifier

BTS

Jumper

12 dBd

0.5 dB

3 dB

NF = 2.5 dB, Gain 12 dB

0.5 dB

12 dBd

0.5 dB

3 dB

A

B

C

D

With Tower TopAmplifier

Without TowerTop Amplifier

NF = 9.5 dB NF = 6 dB

Stage1

Stage2

Stage1

Stage2

Stage3


4


Based upon the information in Table 4-4 and [EQ 4-20], [EQ 4-21], and [EQ 4-22], the noise factor at reference point B in Figure 4-7 for the receive path with the TTA can be calculated as follows:

[EQ 4-23]

FB = 2.97

Using [EQ 4-20], the cascaded noise figure would be:

NFB = 4.73 dB

The design without the tower top amplifier would result in the following noise factor at reference point D shown in Figure 4-7:

[EQ 4-24]

FD = 7.96

NFD = 9.0 dB

The noise figure at point D could have also been determined by just adding the noise figure of stage 1 to the noise figure of stage 2 because the elements which made up stage 1 were all lossy.

From the above calculations, the low noise figure and the gain of the TTA produces a cascaded noise figure of 4.73 dB at reference point B. This is a 4.77 dB improvement in the noise figure as compared to the noise figure at point A. Point D, in the non-TTA case, can be compared to point Bto show the improvement in the noise figure and thus the reverse link improvement that can be achieved with the TTA. The reverse link has improved 4.27 dB (9 - 4.73) with the TTA.

If the impact of the TTA is to be applied to a link budget, the following values would be used:

Please note that for the example in Figure 4-7, the base station product which includes a TTA was modified to have a higher noise figure than the typical base station. The higher noise figure for the base station/TTA configuration was implemented so that the gain of the TTA does not overdrive the front-end of the base station. Adding a low noise amplifier to the receive path of a standard BTS (i.e. not modified) will degrade the equivalent 3rd order input intercept point (IIP3) performance

Table 4-5: Link Budget Inputs

Parameter With TTA Without TTABase Rx Feeder Loss 0.5 dB 3.5 dBBase Noise Figure 4.73 dB 6 dBYields Rx Sensitivity @ point B C

FB 1.782.24 1–15.85

------------------- 8.91 1–15.85 0.45×------------------------------+ +=

FD 23.98 1–

0.5-------------------+=


4


of the site. In order to minimize the IIP3 degradation to the cell site, the total gain from the TTA input to the BTS input should be limited to a maximum of 6 dB. Minimizing the total gain will minimize the IIP3 degradation.

Though the above scenario shows a reverse link budget advantage when a TTA is installed, not all aspects of a TTA may be as advantageous. The following lists some of the drawbacks of TTAs:

• Increased susceptibility to reverse interference noise • Since the TTA only improves the reverse link, an increase to the forward power may be

required to maintain a balanced link• Timing concerns (How large can a site be without causing timing issues?)• Active electronics at the top of the antenna structure (more susceptible to lightning,

more difficult for maintenance, etc.)

Due to the increased susceptibility to noise, Nokia Siemens Networks does not typically recommend TTAs. Though in some scenarios (for example in rural applications), TTAs may be beneficial.

4.2.3.2.2 Subscriber Unit

The noise figure for the subscriber unit is approximately 10 dB. The required Eb/No value to provide acceptable audio quality for the subscriber unit is highly dependent on several parameters. These parameters include: the speed, the environmental parameters, multipath and soft handoff of the subscriber unit. This is one of the reasons why it is difficult to determine a forward link budget. It is best left to a CDMA simulator that takes these situations into account.

4.2.4 Reliability (Shadow Fade Margin)

The shadow fade margin (also known as slow or log-normal fading margin) corresponds to the variation in mean signal level caused by the subscriber passing through the shadows of hills or buildings. The log-normal distribution has been found to be a good estimate of the statistical nature of shadowing and is used to calculate the probability of RF coverage at each point in the cell. At points near the base station, the average received signal level and the probability of coverage will be high. At points near the edge of the cell, the average received signal level and probability of coverage will be lower. The total probability of coverage for the entire cell is determined by integrating the point probabilities over the cell area. The desired area coverage (e.g. 90%) is achieved by adjusting the fade margin to the necessary level. A normal distribution of signals can be used in calculating the reliability. The following figure shows that adding a margin to the link budget will increase the reliability (confidence) of achieving the desired signal level.


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Figure 4-8: Impact of Fade Margin on Reliability

The desired level of reliability is used to determine the amount of shadow fade margin that is required, where a 97% design requires several dB more margin than a 95% design. To improve the RF reliability, going further out on the tail of the distribution, additional margin is added to all users. For a fixed system this may not be efficient nor cost effective since subscriber unit placement has a big effect in determining the worst 5% of the users. The cost of increasing the reliability (increasing dB margin that will impact all users) should be replaced with fixing the worst 5% of the users, and thus saving the dB margin for the average users. For a fixed system, the fade margin, building penetration margin, and soft handoff gain should to be considered together to provide for the best achievable link budget.

The fade margin is the amount of margin necessary to achieve the required area reliability (as per Jakes’ equations5) for a given standard deviation. The standard deviation is a measured value that is obtained from various clutter types. It basically represents the variance (log-normally distributed around the mean value) of the measured RF signal strengths at a certain distance from the site.

5. Jakes, W.C., “Microwave Mobile Communications”, IEEE Press Reissue 1993 (Wiley, New York, 1974), pp. 125-127

No Fade Margin

Margin

Edge Reliability at 50%

Edge Reliability at greater than 50%


4


Therefore, the standard deviation would vary by clutter type. Depending on the propagation environment, the log-normal standard deviation can easily vary between 5 and 9 dB or even greater. Assuming flat terrain, rural or open clutter types would typically have lower standard deviation levels than the suburban or urban clutter types. This is due to the highly obstructive properties encountered in an urban environment, that in turn will produce higher standard deviation to mean signal strengths than that experienced in a rural area.

Jakes’ single cell reliability equations (refer to the following equations) that determine the edge and area reliability of a single cell model are commonly used to approximate the reliability of a site.

[EQ 4-25]

Where:Edge reliability

xo Signal threshold level

x Signal mean at edge of the cell

Log normal standard deviation

[EQ 4-26]

[EQ 4-27]

[EQ 4-28]

Where:Fu is the fraction of the total cell area where the signal exceeds a threshold

determined by

Signal mean at edge of the cell

n propagation exponent value

A composite standard deviation can be obtained by the following:

[EQ 4-29]

PxoR( ) 1

2--- 1

2---erf

xo x–

σ 2-------------

–=

PxoR( )

σ

Fu12--- 1 erf a( )–

1 2ab–

b2

------------------ 1 erf

ab 1–b

--------------- +exp+

=

axo α–

σ 2--------------=

b10nLog10 e( )

σ 2--------------------------------=

Pxo

α

σc σ1( )2 σ2( )2… σn( )2+=


4


Where: Log normal standard deviation for environment, n

This composite standard deviation may sometimes be used if there are two or more environments (for instance, outdoors and in-building) which have their own standard deviation. For example if the standard deviation is 6 dB for outdoors and 8 dB for in-building, the composite standard deviation to use in Jake’s equation would be 10 dB.

The following two figures (Figure 4-9 and Figure 4-10) are results from Jake’s single cell model. The edge reliability, Figure 4-9, has been shown for three different standard deviations (6.5, 8, and 10 dB) to demonstrate the impact of the standard deviation.

Figure 4-9: Edge Reliability vs. Fade Margin

Figure 4-9 shows that edge reliability is dependent on the standard deviation and fade margin assumed. The following observations can be seen.

• As the standard deviation increases, the edge reliability is reduced for the same fade margin.

• As the standard deviation increases, a larger fade margin is required to maintain the same edge reliability.

σn

Uplink Shadow Fade Margin (dB)

Edg

e R

elia

bili

ty

40%

50%

60%

70%

80%

90%

100%

0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15

6.5

8.0

10.0


4


The area reliability, Figure 4-10, assumes a standard deviation of 8 dB for the three curves. The difference in the curves is due to three different path loss slopes (32, 35, and 40 dB/decade).

Figure 4-10: Area Reliability vs. Fade Margin

Note: Within the legend of Figure 4-10, the first value corresponds to the propagation loss slope in dB per decade. The second value corresponds to the standard deviation in dB.

Figure 4-10 shows that the area reliability is dependent on the standard deviation, fade margin, and propagation loss slope (the slope is dependent on the height of the antennas). The following observations can be seen.

• As the standard deviation increases, a larger fade margin is required to maintain the same area reliability, assuming the same propagation slope.

• As the level of area reliability increases, a larger fade margin is required, assuming the same standard deviation and propagation slope.

• As the propagation slope (path loss exponent) increases, a smaller fade margin is required to maintain the same area reliability, assuming the same standard deviation.

The preceding information is for a single cell. When multiple cells and soft handoff are accounted for, the probability of meeting a given signal strength is increased. Soft handoff is not an absolute


Are

a R

elia

bili

ty

70%

75%

80%

85%

90%

95%

100%

0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15

40, 6.5

35, 6.5

32, 6.5

40, 8

35, 8

32, 8

40, 10

35, 10

32, 10


4


gain but can be viewed as a reduction in the fade margin requirement needed to meet a desired edge or area reliability goal. For isolated sites, there would be no improvement since there would be no sites to enter into soft handoff with.

Since most systems are comprised of more than a single cell, the benefit of multiple cell effects could be used. Simulations can be performed, given various assumptions (path loss slope, standard deviation, correlation), to determine the appropriate shadow fade margin to be added to the link budget to provide for the reliability desired. This multiple cell effect accounts for the overlap of adjacent cells and the fast handoff capability of the CDMA soft handoff method. As mentioned in the previous soft handoff section, the gain associated with soft handoff can be rolled into one shadow fade margin.

Nokia Siemens Networks has performed various simulations for a multiple cell system and generated some reliability curves. The curves in Figure 4-11 show that 4.7 to 5.6 dB fade margin is required to reach 95% area reliability for a sector site. The curves show that the area reliability is a function of the configuration of the site, as well as the standard deviation and site-to-site correlation assumed. Nokia Siemens Networks typically recommends the 5.6 dB shadow fade margin to design systems with an area reliability of 95% or slightly better.

The following two figures illustrate examples of the required fade margin based on simulations. These simulations account for the soft handoff advantage in a multi-cell system. The two figures illustrate the cell area and edge reliability as a function of shadow fade margin. Note that the required margin varies as a function of the propagation model and sectorization. The notation (x1, x2, x3), in the figures refer to the propagation model, where x1 is the path loss slope, x2 is the lognormal shadow fading standard deviation, and x3 is the site-to-site correlation (Note: path loss slope x1 converts to path loss dB/decade by multiplying x1 by a factor of 10).


4


Figure 4-11: Area Reliability as a Function of Shadow Fade Margin

For the above analysis, the sector sites assumed an antenna with 90° horizontal beamwidth. For a given area reliability, the sector sites required a larger fade margin to account for the reduction of gain experienced between the sectors.

Sector (3.5, 6.5, 0.5)

Sector (4, 8, 0.5)

Omni (3.5, 6.5, 0.5)

Omni (4, 8, 0.5)


Are

a R

elia

bili

ty


4


Figure 4-12: Edge Reliability as a Function of Shadow Fade Margin

As mentioned in the section on soft handoff gain, some RF link budgets may have separate entries for soft handoff gain and shadow fade margin. Typically when this is done, Jakes’ single cell model fade margin is used to obtain the reliability level desired. The CDMA RF link budget, though, still needs to account for the benefit of soft handoff. Therefore, an approximation for the benefit of soft handoff gain is required in the link budget. In the RF link budget spreadsheet analysis, Nokia Siemens Networks typically assumes the benefit for soft handoff in a mobile environment to be approximately 3.5 dB for a cluster of sties. If there is only a single entry in the RF link budget for the fade margin, then the composite fade margin would be the single cell shadow fade margin minus the benefit associated with soft handoff and multiple cells. For example, assuming a 9.1 dB shadow fade margin and 3.5 dB benefit from soft handoff and multiple cells, the composite fade margin would be 5.6 dB (9.1 minus 3.5).This is an approximation based on a single cell model plus an assumed soft handoff benefit.

4.2.5 Example Reverse (Uplink - Subscriber to Base) Link Budget

The following table provides an example of a reverse path RF link budget for both a mobile/portable system and a fixed IS-95 system. This basic RF link budget example could be applied towards an IS-95A or IS-95B system. Antenna gains, feeder losses, noise rise, building losses, vehicle losses, shadow fade margins, etc. will differ from system to system and from site to site

Sector (3.5, 6.5, 0.5)

Sector (4, 8, 0.5)

Omni (3.5, 6.5, 0.5)

Omni (4, 8, 0.5)


Edg

e R

elia

bili

ty


4


(possibly even from sector to sector) based on the design objectives of the system planner.

Table 4-6: Example of an IS-95 CDMA Reverse RF Link Budget

Note: 1. It is assumed that the latest version of chip sets are being utilized.2. Path Loss values shown assume a medium traffic load on the reverse link for the

CDMA system.3. The shadow fade margin assumes the effects of soft handoff and multiple cells.

Where:Sensitivity and path loss are calculated as follows:

Sb = kTB + Nfb + E - PG

Lp = Pp - Lfp + Gp + Gb - Lfb - Sb - Im - Tm - Hm - Vm - Bm - Fm

Li = Lp + (2 * 2.14)

Parameter Unit Reference Mobile13 kbps

Mobile8 kbps

Fixed8 kbps

Subscriber Unit Tx Power dBm Pp Section 4.2.3.1.1 23 23 23

Subscriber Unit Tx Feeder Loss

dB Lfp Section 4.2.1.5 0 0 0

Subscriber Unit Antenna Gain

dBd Gp Section 4.2.1.6 -2.1 -2.1 -1.0

Body Loss dB Hm Section 4.2.1.3 2 2 0

Vehicle Loss dB Vm Section 4.2.1.2 7 7 0

Building Loss dB Bm Section 4.2.1.1 0 0 6

Base Antenna Gain dBd Gb Section 4.2.1.6 14.5 14.5 14.5

Line Loss dB Lfb Section 4.2.1.5 3 3 3

kTB dBm kTB Section 4.2.3.2 -113.1 -113.1 -113.1

Noise Figure dB Nfb Section 4.2.3.2.1 6 6 6

Eb/No (Note: 1) dB E Section 4.2.2.3 6.0 5.6 5.6

Processing Gain dB PG Section 4.2.3.2 19.3 21.1 21.1

Base Rx Sensitivity dBm Sb Section 4.2.3.2 -120.4 -122.6 -122.6

Interference Margin (Note: 2)

dB Im Section 4.2.2.1 3 3 3

Ambient Noise Rise dB Tm Section 4.2.1.4 0 0 0

Shadow Fade Margin (Note: 3)

dB Fm Section 4.2.2.2 & Section 4.2.4

5.6 5.6 5.6

Max. Allowable Path Loss dB Lp 135.2 137.4 141.5

Isotropic Path Loss dB Li 139.5 141.7 145.8


4


In comparing the link budget between mobile (portable) and fixed, there are three main differences. The first being that the fixed link budget has a subscriber antenna gain of 1.1 dB better than the mobile case (assumes FWT has the whip antenna installed, but could be higher with external antennas). It is also assumed that the FWT whip antenna is connected directly to the FWT base unit and therefore there is no line loss between the FWT base and antenna. Other scenarios may require that a line loss be added for antennas not connected directly to the base unit. A second difference is that there is no body loss assumed for the fixed case. The antenna gain and body loss differences give a 3.1 dB link budget advantage of fixed over mobile.

The third difference is with respect to the building/vehicle penetration loss. For the fixed case, a building loss value of 6 dB is shown based upon the assumption that the FWT with whip antenna will be placed close to a window and in a location that will minimize the impact of the building loss. The amount of building penetration will need to be adjusted (could be greater or less than the 6 dB value assumed here) based on the installation location of the FWT antenna and the building characteristics (some buildings may allow RF to pass better than others).

For the mobile case, 7 dB is assumed for a vehicle penetration value. If in-building is desired, then this value would need to be modified accordingly. If it is desired to provide in-building coverage, additional margin would be required.

The fade margin is set the same for fixed and mobile for these link budget examples. One view is that the fade margin should be increased to provide for better reliability for a fixed system. This increased fade margin, though, would apply to all subscribers. Another way to improve the reliability for a fixed system is not by adding margin in the link budget, which effects all users, but to take the worst performing FWT and replace the whip antenna with an external antenna. This will improve its performance, which ultimately improves the overall reliability. Another view is that the reliability for fixed should be higher since fixed is competing with the wireline service. The amount of fade margin is related to the reliability. If the reliability criteria is increased, the fade margin will also need to be increased.

Another value which differs between the fixed and mobile is the subscriber antenna height. This is not part of the link budget above, but would be required in the propagation models. The typical subscriber antenna height assumed for the mobile (portable) case is 1.5 meters. The FWT antenna has the ability of being positioned at various heights (on a desk, on a wall, externally on the roof), therefore the height of the FWT could range from 1 to 3 or more meters.

The following table provides an example of an IS-2000 1X reverse path RF link budget for a mobile/portable system. It represents the reverse Radio Configuration 3. A similar approach can be done for reverse Radio Configuration 4 by replacing the subscriber transmit power, processing gain and Eb/Nos with the appropriate values. Antenna gains, feeder losses, noise rise, building losses, vehicle losses, shadow fade margins, etc. will differ from system to system and from site to site (possibly even from sector to sector) based on the design objectives of the system planner.


4


Table 4-7: Example of an IS-2000 1X CDMA RF Link Budget

Note: 1. Path Loss values shown assume a medium traffic load on the reverse link for the CDMA system.

2. The shadow fade margin assumes the effects of soft handoff and multiple cells.

An observation of the above table shows that the allowable path loss decreases as the data rate increases. This means that a smaller cell radius would be required to support higher data rates. For example, more sites would be required if a system was to be designed based on a reverse link

Parameter Unit Reference 9.6 kbps

9.6 kbps

19.2 kbps

38.4 kbps

76.8 kbps

153.6 kbps

Reverse Traffic Channel FCH SCH SCH SCH SCH SCH

Total Subscriber Unit Tx Power

mW PT Section 4.2.3.1.1 200 200 200 200 200 200

Subscriber Unit R-FCH or R-DCCH Tx Power

mW PFCH Section 4.2.3.1.1 200 111 90 63 41 25

Subscriber Unit R-SCH Tx Power

mW PSCH Section 4.2.3.1.1 - 89 110 137 159 175

Subscriber Unit Tx Power (for the specified reverse traffic channel)

dBm Pp Section 4.2.3.1.1 23 19.5 20.4 21.4 22.0 22.4

Subscriber Unit Tx Feeder Loss

dB Lfp Section 4.2.1.5 0 0 0 0 0 0

Subscriber Unit Antenna Gain

dBd Gp Section 4.2.1.6 -2.1 -2.1 -2.1 -2.1 -2.1 -2.1

Body Loss dB Hm Section 4.2.1.3 2 2 2 2 2 2

Vehicle Loss dB Vm Section 4.2.1.2 6 6 6 6 6 6

Building Loss dB Bm Section 4.2.1.1 0 0 0 0 0 0

Base Antenna Gain dBd Gb Section 4.2.1.6 14.5 14.5 14.5 14.5 14.5 14.5

Line Loss dB Lfb Section 4.2.1.5 3 3 3 3 3 3

kTB dBm kTB Section 4.2.3.2.1 -113.1 -113.1 -113.1 -113.1 -113.1 -113.1

Noise Figure dB Nfb Section 4.2.3.2.1 6 6 6 6 6 6

Eb/No dB E Section 4.2.2.3 5.6 4.6 3.5 3.0 2.5 2.1

Processing Gain dB PG Section 4.2.3.2 21.1 21.1 18.1 15.1 12.0 9.0

Base Rx Sensitivity dBm Sb Section 4.2.3.2 -122.6 -123.6 -121.6 -119.1 -116.6 -114.0

Interference Margin (Note: 1)

dB Im Section 4.2.2.1 3 3 3 3 3 3

Ambient Noise Rise dB Tm Section 4.2.1.4 0 0 0 0 0 0

Shadow Fade Margin (Note: 2)


5.6 5.6 5.6 5.6 5.6 5.6

Max. Allowable Path Loss dB Lp 138.4 135.9 134.8 133.3 131.4 129.2

Isotropic Path Loss dB Li 142.7 140.2 139.1 137.6 135.7 133.5


4


assuming 76.8 kbps than if the system requirement was for 9.6 kbps. Assuming a propagation exponent of 3.5, the 7 dB path loss difference between these two data rates would correspond to the 76.8 kbps scenario requiring approximately 2.5 times the number of sites as the 9.6 kbps scenario.

IS-2000 provides the ability to have asymmetrical data transmission. That is, the data rate on the forward link can be different than the data rate employed on the reverse link. Initial data applications for IS-2000 are assumed to demand more data to be transferred on the forward link than on the reverse link (i.e. the forward link data rate will need to be faster than the reverse link data rate). Additionally, it is viewed that the reverse link will be the limiting link with regards to coverage, whereas the forward link will be the limiting link with regards to capacity. It is possible that an RF reverse link based on a fundamental rate of 9.6 kbps would allow for sufficient path loss so that a forward link of 76.8 kbps could be achieved. This means that the reverse link coverage to support 9.6 kbps may provide for sufficient coverage on the forward link to support a user needing 76.8 kbps. This is not saying that a user rate of 153.6 kbps is not supported. A user, in close proximity to the site, could have a forward and/or reverse supplemental channel at 153.6 kbps, but not at the fringe of the site. Given these views, a system design based on the RF reverse link for reverse data rates above 19.2 kbps may not be necessary. If data applications require a high volume of reverse data, then higher data rates need to be considered.

These link budgets are examples and may need to be modified to accommodate specific design goals for a system. Refer to the previous discussion on each of the parameters to determine if alterations are required for a specific design.

4.2.6 RF Link Budget Summary

The RF link budget propagation related parameters have the most variability. These propagation related parameters are typically vendor and technology independent. The link budget parameters, but not the values, listed above can apply to all technologies and frequencies. For instance, the loss associated with the transmission line is dependent upon the frequency of operation, but not that it will be used for CDMA instead of GSM.

The following figure demonstrates the impact to the quantity of sites required if one assumption is made over another. The figure only shows 5 examples. There are many other combinations that are possible.


4


Figure 4-13: Impact of dB Trade-off to Number of Sites

4.3 Propagation Models

The propagation model is used in conjunction with the RF link budget to obtain an estimate of the cell radius based on the allowable path loss from the link budget. Statistical propagation models are used in budgetary designs to give quick estimates of cell radii within various environments and ultimately to estimate the number of cells required for a system.

There are many RF propagation factors which could extend or restrict the coverage of a site (e.g. proximity to buildings, actual terrain, antenna heights, topology, morphology, etc.). More detailed propagation models, which include some or all of these factors, will produce more accurate predictions of cell radii. The following sections give additional detail concerning statistical propagation models.

4.3.1 Free Space Propagation Model

The free space power received by a receiver antenna, which is at a distance of d from the transmitter antenna, is given by the Friis free space equation.

[EQ 4-30]PR PT G⋅ T GRλ

4πd----------

2

⋅ ⋅=


4


Where:PT is the transmitted power

GT is the transmitting antenna gain

GR is the receiving antenna gain

d is the separation distance between antennas

The path loss, which represents the signal attenuation as a positive quantity, is defined as the difference between the effective transmitted power and the received power. It may or may not include the effects of the antenna gains. The path loss for the free space model, when the antennas are assumed to have unity gain, is provided by the following equation.

[EQ 4-31]

Expressed in dB as:

[EQ 4-32]

Where:d is in meters

f is in Hertz

c is equal to the speed of light (3 x 108 meters per second)

[EQ 4-33]

[EQ 4-34]

[EQ 4-35]

[EQ 4-36]

[EQ 4-37]

The above free space equations show that 6 dB of loss is associated with a doubling of the frequency. This same relationship also holds for the distance, if the distance is doubled, 6 dB of additional loss will be encountered.

PT

PR------ 4πd

λ----------

2 4πdfc

------------

2

==

LFS dB( ) 10PT

PR------ 20

4πc

------ 20 f( ) 20 d( )log+log+log=log=

LFS dB( ) 147.56 20 fHz( ) 20 dmeters( )log+log+–=

LFS dB( ) 32.44 20 f( MHz ) 20 d( km )log+log+=

LFS dB( ) 27.55– 20 f( MHz ) 20 d( meters )log+log+=

LFS dB( ) 36.58 20 f( MHz ) 20 d( miles )log+log+=

LFS dB( ) 37.87– 20 f( MHz ) 20 d( feet )log+log+=


4


4.3.2 Hata Propagation Model

Among the many technical reports that are concerned with propagation prediction methods for mobile radio, Okumura’s6 report is believed to be the most comprehensive one. In his report, many useful curves to predict a median value of the received signal strength are presented based on the data collected in the Tokyo area. The Tokyo urban area was then used as a basic predictor for urban areas. The correction factors for suburban and open areas are determined based on the transmit frequency. Based on Okumura’s prediction curves, empirical formulas for the median path loss, Lp, between two isotropic antennas were obtained by Hata and are known as the Hata empirical

formulas for path loss7. The Hata propagation formulas are used with the link budget calculation to translate a path loss value to a cell radius.

For Urban Area:

[EQ 4-38]

For Suburban Area:

[EQ 4-39]

For Quasi Open Area:

[EQ 4-40]

For Open Rural Area:

[EQ 4-41]

Where:AHm Correction Factor For Vehicular Station Antenna Height

For a medium-small city:[EQ 4-42]

For a large city:

[EQ 4-43]

6. Okumura, Y., Ohmori, E., Kawano, T., Fukada, K.: "Field strength and ITs Variability in VHF and UHF Land-Mobile Radio Service", Rev. Elec. Commun. Lab., 16 (1968), pp. 825-873

7. Hata, M.: "Empirical formula for propagation loss in land mobile radio services", IEEE Trans. on Vehicu-lar and Technology, VT-29 (1980), pp. 317-325

LU 69.55 26.16 fc( )log× 13.82 Hb( )log×– AHm– 44.9 6.55 Hb( )log× ] r( )log×–[+ +=

LS LU 2fc

28------ log

2

×– 5.4–=

Lq LU 4.78 fc( )log[ ]× 2– 18.33 fc( )log× 35.94–+=

Lq LU 4.78 fc( )log[ ]× 2– 18.33 fc( )log× 40.94–+=

AHm 1.1 fc( ) 0.7–log×[ ] Hm× 1.56 fc( ) 0.8–log×[ ]–=

AHm 3.2 11.75 Hm×( ) ]log[× 24.97–=


4


Lu, Ls, Lq Isotropic path loss values

fc Carrier frequency in MHz (valid 150 to 1000 MHz)

Hb Base antenna height in meters (valid 30 to 200 meters)

Hm Subscriber antenna height in meters (valid 1 to 10 meters)

r Radius of site in kilometers (valid 1 to 20 km)

This model is valid for large and small cells (i.e. base station antenna heights above roof-top levels of buildings adjacent to the base station).

4.3.3 COST-231-Hata Propagation Model

The COST 231 Subgroup on Propagation Models proposed an improved propagation model for urban areas to be applied above 1500 MHz8. Like Hata’s model, the COST-231-Hata model is based on the measurements of Okumura. The COST-231-Hata propagation model has been derived by analyzing Okumura’s propagation curves in the upper frequency band. Hata’s analysis was restricted to frequencies below 1000 MHz. The COST-231-Hata propagation model extended the range of parameters to include 1500 to 2000 MHz. Their modified model was based on Hata’s formula for the basic transmission loss in urban areas (see above).

For Urban Area:

[EQ 4-44]

For Suburban Area:

[EQ 4-45]

For Quasi Open Area:

[EQ 4-46]

For Open Rural Area:

[EQ 4-47]

8. COST 231 - UHF Propagation, "Urban transmission loss models for mobile radio in the 900- and 1,800-MHz bands", COST 231 TD (91) 73 The Hagne, September, 1991

LU 46.3 33.9 fc( )log× 13.82 Hb( )log×– AHm– 44.9 6.55 Hb( )log× ] r( )log×–[+ +=

LS LU 2fc

28------ log

2

×– 5.4–=

Lq LU 4.78 fc( )log[ ]× 2– 18.33 fc( )log× 35.94–+=

Lq LU 4.78 fc( )log[ ]× 2– 18.33 fc( )log× 40.94–+=


4


Where:AHm Correction Factor For Vehicular Station Antenna Height

For a medium-small city:[EQ 4-48]

For a metropolitan center:[EQ 4-49]

Lu, Ls, Lq Isotropic path loss values

fc Carrier frequency in MHz (valid 1500 to 2000 MHz)

Hb Base antenna height in meters (valid 30 to 200 meters)

Hm Subscriber antenna height in meters (valid 1 to 10 meters)

r Radius of site in kilometers (valid 1 to 20 km)

This model is valid for large and small cells (i.e. base station antenna heights above roof-top levels of buildings adjacent to the base station).

A comparison between the Hata and COST-231-Hata equations show that they are similar except for the following terms:

Hata yields

COST-231-Hata yields

Measurements which have been taken at 1900 MHz have shown the path loss difference between 800 MHz and 1900 MHz closer to 11 dB. The COST-231-Hata model was developed to account for this difference.

4.3.4 Additional Propagation Models

The above propagation models are widely known and are usually referenced when conversing in more general terms. Numerous books can be referenced for further discussion on these models, such as those listed in references9,10.

9. Parsons, David, "The Mobile Radio Propagation Channel", Copyright 1992, Reprinted 1996 by John Wiley & Sons Ltd.

10. Rappaport, Theodore S., "Wireless Communications Principles & Practices", Copyright 1996 by Prentice Hall PTR

AHm 1.1 fc( ) 0.7–log×[ ] Hm× 1.56 fc( ) 0.8–log×[ ]–=

AHm 1.1 fc( ) 0.7–log×[ ] Hm× 1.56 fc( ) 0.8–log×[ ]– 3–=

69.55 26.16 fc( ) AHm–log+

46.3 33.9 fc( ) AHm–log+


4


These propagation models can be used to obtain an estimate of the expected radius of a site. However, they do not include the effects of the antenna patterns, ground clutter and terrain experienced between the transmitter and receiver. In addition, the Hata and COST-231-Hata model are dependent upon the environment classification. Defining the area types are fairly subjective and the entire cell site is considered to be the defined area type. For instance, if an area is assumed to be urban but is more realistically suburban, a 12 dB impact results (many more sites would be specified than what would really be needed). In addition, these propagation models do not portray ground clutter such as a forested area, though modifications can be made to the propagation model or the link budget to account for loss due to foliage or forest.

One model that does include these effects is the Xlos propagation model in Nokia Siemens Networks’ IDGP propagation analysis tool. This propagation model is based on work from Longley & Rice, Okumura, Bullington and Nokia Siemens Networks’ extensive field measurement data. It takes into account the effects of ground reflections, diffractions and line of sight propagation. It defines the path loss with respect to dipole antennas. Hata or COST-231-Hata propagation models assume path loss is defined with respect to isotropic antennas.

As was mentioned in the introduction, this sophistication in a propagation tool is required to provide a more realistic portrayal of the coverage for a system.

4.4 Forward Link Coverage

In Section 4.2, the CDMA subscriber-to-base link (reverse or uplink) was discussed. This is a many-to-one link, where many subscribers communicate with a single base station (or a fixed number of base stations). Hence, the link can be simply characterized using a link budget with additional margin included for interference. This margin is typically measured in terms of noise rise at the cell, which is specified in terms of the operating point relative to a fixed asymptotic capacity (pole) (e.g. operating at 75% of the pole results in a 6 dB noise rise).

The CDMA base-to-subscriber (forward or downlink) is a one-to-many link, where a single base station (or a fixed number of base stations) communicates with many subscribers. This link is somewhat more complicated to analyze, and it does not lend itself easily to a simple RF link budget method. The reason for the difficulty is:

• In the absence of multipath, the use of orthogonal Walsh codes on the downlink removes the intra-cell interference. With multipath, intra-cell interference causes a reduction in signal-to-noise ratio. However, this is mitigated (in most cases) by the fact that multipath improves the subscriber receiver sensitivity.

• Subscriber receiver sensitivity is characterized in terms of Eb/(Ioc+No), energy-per-bit over other-cell interference (plus noise) power density. It is assumed that there is sufficient power allocated on the downlink such that thermal noise does not significantly affect the performance. It has been determined by link budget analysis that 2.4 Wattspilot power is sufficient to balance the uplink and downlink of an IS-2000 1X system.This analysis assumed CSM5000 demodulator Base Station receiver performance, forward Radio Configuration 3 (RC 3), subscriber noise figure = 10 dB, base noise figure = 5 dB, and subscriber PA power of 200 mW.


4


• The downlink Eb/Ioc varies substantially with multipath (or soft handoff) and subscriber speed. For example, Eb/Ioc in 1-path (i.e. no multipath) Rayleigh fading at slow subscriber speed can be as high as 20-25 dB, whereas with 3-path, Eb/Ioc can be less than 8 dB.

• Soft handoff also complicates the downlink, because typically subscribers in soft handoff require less power (from each cell site). On the other hand, the subscribers at the edge of the soft handoff region experience high interference, and the Eb/Ioc performance (without multipath) is the worst. Thus, for downlink, it is not sufficient to balance the link to the edge of the cell, but it has to be balanced to the edge of the soft handoff region. Note that the soft handoff regions vary dynamically as a function of load in the desired and the surrounding cells, as well as the propagation environment.

Though a forward link budget is not addressed, it is important to account for the power requirements when designing (simulation studies) and optimizing a CDMA system. Forward link power at the base station may limit coverage and capacity. The following sections provide some guidelines to assist the system engineer.

4.4.1 BTS Equipment Capabilities

In these guidelines, two PA parameters are frequently referred to: the Average Rated Power (ARP or Steady State Rating) and the High Power Alarm Rating (HPA). Another PA related term is “Trunk Group”. Trunk Group refers to all the sector-carriers that are amplified by the same PA. For 4812T frames, each carrier is a trunk group. For 4812T-MC frames (not using 9071A separate sub-bands) and single band 7224 frames, all the sector-carriers in a frame are one trunk group. For a 7224 dual band frame, each band is a trunk group. For UBS, coherently combined XMIs and their sector-carriers, belong to the same trunk group. There is a trunk group for each uncombined or non-coherently combined XMI on UBS.

The Flexi CDMA BTS RFM has 3 RF paths per module. Each RF path is a trunk group.

The table below is neither comprehensive nor, necessarily, current; refer to equipment specifications for details on the Base Transmission Station (BTS) product of interest.

Table 4-8: PA Ratings for Some BTS Products

BTS Product Frequency(MHz)

Number of PA

Modulesper Trunk

Group

Average Rated Power

(W)per Trunk

Group

High Power Alarm Rating

(W)per Trunk Group


(dB)a

per Trunk Group

SC300 1X Microcell

800/1900 1 10 31.6 5

SC340 1X Picocell

Japan 800 1 0.2 N/A N/A


4


SC340 1X Microcell

Japan 800 1 5 20 6

SC4812Tb/ET/ET Lite

1900 (4) 22.5/67.5 70.8/107.2 5/2

SC4812 800 2 22.5 36 2

SC4812Tb/ET/ET Lite

800 (4) 22.5/67.5 70.8/107.2 5/2

SC4812Tb-Lite-800

800 3 90 142 2

SC4812Tb-Lite-1.9

1900 3 60 95 2

SC4812Tb-MC-800

800 3 to 16 27 x num-ber of PA modules

27 x number of PA modules x

1.6

2

SC4812Tb-MC-1.9

1900 3 to 16 20 x num-ber of PA modules

20 x number of PA modules x

1.6

2

SC7224e 2100 1 to 4 120 x num-ber of XMI

modules

120 x number of XMI mod-

ules x 1.6

2

SC7224e 800 1 to 4 120 x num-ber of XMI

modules


ules x 1.6

2

SC7224e dual band

2100/800 1 to 2 per band

120 x num-ber of XMI

modules


ules x 1.6

2

SC480 800/1900 1 to 2 20 (CCLPA) 32 2



Number of PA

Modulesper Trunk

Group

Average Rated Power

(W)per Trunk

Group


(W)per Trunk Group


(dB)a

per Trunk Group


4


SC480-800with GSM fil-

ter

800 1 or 2 15 (CCLPA) 24 2

UBS-800h 800 1 151 (3-sector)i

101 (omni)

242 (3-sector)162 (omni)

2

UBS-800f 800 2 269 (3-sector)i

180 (omni)

430 (3-sector) 288 (omni)

2

UBS-800 800 1 to 4 134 x number of

XMI modules (3-

sector)i

90 x number of XMI modules (omni)

215 x number of XMI mod-ules (3-sector)

144 x number of XMI mod-ules (omni)

2

UBS-1.9h 1900 1 118 (3-sector)

80 (omni)


2

UBS-1.9j 1900 1 110 (3-sector)

75 (omni)


2

UBS-1.9h

(C25 andlater)

1900 1 136 (3-sector)

92 (omni)


2

UBS-1.9j

(C25 andlater)

1900 1 124 (3-sector)

84 (omni)


2



Number of PA

Modulesper Trunk

Group

Average Rated Power

(W)per Trunk

Group


(W)per Trunk Group


(dB)a

per Trunk Group


4


a. The High Power Alarm Rating (dB) is represented here in terms of dB above the Average Rated Power. The alarm usually occurs after 2 seconds at the HPA rating.

b. This is a TrunkedPowerTM BTS. It has multiple Trunked LPA modules serving one or more three-sector carriers. Its ARP is shared across all three sectors. (A six-sector carrier is served by two sets of Trunked LPA modules.) The High Power Alarm functions on a total power basis, as opposed to an individual sector-carrier basis as for non-trunked BTSs.

e. These models use a coherent combiner that allows trunking together up to 4 XMI modules.f. These models may or may not use a coherent combiner to trunk together up to 2 XMI modules. If a

coherent combiner is not used, only one XMI can be used.h. This model does not use a coherent combiner. Each XMI is a trunk group.i. If DO carriers are equipped in the trunk group, the average rated power is reduced by 0.7 dB. j. This model uses a cavity combiner. Each XMI is a trunk group.k. Each RFM has 3 trunk groups. Each Flexi CDMA BTS can have 1 or 2 RFMs.

UBS-800Jf 800J 1 151 (3-sector)

101 (omni)


2

UBS-800Jf 800J 2 269 (3-sector)

180 (omni)


2

UBS-2.1f 2100 1 151 (3-sector)

101 (omni)


2

UBS-2.1f 2100 2 269 (3-sector)

180 (omni)


2

M810 2100 1 2 or 20 3.2 or 32 2

M810 800J 1 2 or 20 3.2 or 32 2

M810 800 1 2 or 20 3.2 or 32 2

UBSc with RRH

800J 1 20 32 2

Flexi CDMA BTS with FXCA RFMk

800 1 60 95 2



Number of PA

Modulesper Trunk

Group

Average Rated Power

(W)per Trunk

Group


(W)per Trunk Group


(dB)a

per Trunk Group


4


The following table illustrates the 1X pilot RF power adjustment range capability for several different CDMA BTS products. An external attenuator is required when operating at lower than the minimum specification. The table below is neither comprehensive nor, necessarily, current; refer to equipment specifications for details on the BTS product of interest.

Table 4-9: BTS Pilot Power Adjustment Range

Pilot Power Adjustment RangeBTS Product Frequency (MHz) Minimum Pilot

(dBm)Maximum Pilot

(dBm)SC300 1X Microcell 800/1900 +14.0 +33.0SC340 1X Picocell Japan 800 -2.0 +16.0SC340 1X Microcell Japan 800 +14.0 +33.0SC4812 800 +26.0 +36.0SC4812T/ET/ET Lite 800 +26.0 +36.0SC4812T/ET/ET Lite 1900 +26.0 +36.0SC4840/SC2440 SD 800 (JCDMA) +26.0 +37.0SC4840/SC2440 DD 800 (JCDMA) 29.0 40.0SC4812T-MC 800/1900 See below See belowSC4812MF 2100 -20.5 -10.5SC7224 800/2100 24.8 37.8SC4812T-Lite 3-sector 800 26.0 36.0SC4812T-Lite omni 800 29.5 39.5SC4812T-Lite 1900 26.0 34.4SC480 no PA 800/1900 -32.0 -17.0SC480 with 20 W PA 800/1900 21.0 36.0SC480 with 20 W PA + GSM Filter

800 19.5 34.5

UBSa 800 24.8 34.8

UBS 800 (Japan) 24.8 37.8UBS 1900 24.8 34.8UBS 2100 24.8 37.8UBSc with RRH 800 (Japan) 20 36Flexi CDMA BTS 800 24.8 34.8M810 w/o RFHead 800 (Japan)/2100 12.0 30.0 (single

carrier, see equation below

for multiple carriers)


4


a. To avoid tripping the low power alarm when 3 or 4 XMI’s are configured, a minimum total transmit power of 1.12 W must be maintained. The minimum transmit power level can be attained by various combinations of number of sector-carriers, Pilot, Page and Sync.

The following table illustrates the EV-DO pilot RF power adjustment range capability for several different CDMA BTS products. An external attenuator is required when operating at lower than the minimum specification. The table below is neither comprehensive nor, necessarily, current; refer to equipment specifications for details on the BTS product of interest.

Table 4-10: BTS EV-DO Pilot Power Adjustment Range

M810 with RFHead 800 (Japan)/2100 22.0 40.0 (single carrier, see

equation below for multiple

carriers)M810 w/o RFHead 800 12.0 30.0M810 with RFHead 800 22.0 40.0


(dBm)Maximum Pilot

(dBm)SC4812 800 +36.0 41.8SC4812T/ET/ET Lite 800 +36.0 41.8SC4812T/ET/ET Lite 1900 +36.0 41.8SC4812T-MC 800/1900 See below See belowSC4812T-Lite 3-sector 800 36.0 41.8SC4812T-Lite omni 800 36.0 41.8SC4812T-Lite 1900 36.0 41.8SC480 no PA 800/1900 -28.7 -13.7SC480 with 20 W PA 800/1900 21.0 41.8SC480 with 20 W PA + GSM Filter

800 19.5 34.5

UBS 800 37.1 41.8UBS 1900 37.1 41.8Flexi CDMA BTS 800 37.1 41.8

Table 4-9: BTS Pilot Power Adjustment Range


(dBm)Maximum Pilot

(dBm)


4


4.4.1.1 4812T-MC BTS Minimum and Maximum Pilot Power

The minimum pilot power for 4812T-MC BTS frames is dependent on the number of CLPA modules.

The minimum operating power for a sector-carrier for an 800 MHz or 1.9 GHz SC4812T-MC BTS is 26 dBm. If the number of CLPA modules is greater than the number of sector-carriers, the following equation should be used:

sector-carrier min operating power (dBm) = 26 (dBm)+ 10 log (#CLPA / #sector-carriers)

The sector-carrier minimum operating power may include paging and sync channels in addition to the pilot channel. Minimum sector-carrier power can be calculated as follows:

Minimum sector-carrier power (dBm) = SifPilotPwr (dBm) + 10 log ((PilotGain^2 + PageGain^2 + SyncGain^2) / PilotGain^2)

Note: If the minimum sector-carrier power is used and one or more sector-carriers goes OOS, a low power alarm could be triggered.

For an EV-DO carrier, the sector-carrier minimum operating power is:

sector-carrier min operating power (dBm) = 36 (dBm)+ 10 log (#CLPA / #sector-carriers)

The maximum pilot power for 4812T-MC BTS frames is also dependent on the number of CLPA modules. The maximum pilot power for a 4812T-MC BTS 800 MHz or 4812T-MC BTS 1.9 GHz frame in dBm can be determined by the following equation:

MaxPilotPwr (dBm) = the lesser of FrameTypeMax and 10 log((NPAMod*PAModRating (W) *1000) / NSecCarr) – CarrAdj

Where:

• MaxPilotPwr is the maximum pilot power per sector-carrier

M810 w/o RFHead 800 20.0 33.0 (single carrier)

M810 with RFHead 800 30.0 43.0 (single carrier)


(dBm)Maximum Pilot

(dBm)


4


• FrameTypeMax is from the table (Table 4-11) below:

• NPAMod is the number of installed CLPA modules• PAModRating is 27 W for 800 MHz and 20 W for 1.9 GHz• NSecCarr is the number of sector-carriers in the frame• CarrAdj is 8.6 dB for IS95/cdma2000 carriers and 0 dB for 1xEV-DO carriers (the non-

DO CarrAdj is 10 dB below the maximum sector-carrier plus a 1.4 dB allowance for trunking efficiency)

Power is allocated equally between sector-carriers.

4.4.1.2 Max Pilot Power for M810

Traffic Carriers plus Frequency Hopping Pilot Beacons (M810)

Where:

• RatedPower is 20 Watts with an RF Head and 2 Watts without an RF Head• FHPB_PilotPwr is the Pilot power for the Frequency Hopping Pilot Beacon carriers in

Watts• 1.85 is the ratio of Pilot/Page/Sync power to Pilot Power for Page = 110, Sync = 40. If

Page = 90, use 1.60• #FHPB_Groups is the number of FHPB Groups configured (0, 1 or 2)• #Traffic_Carriers is the number of 1X carriers configured• MaxPwr_to_Pilot_Ratio is operator defined based on expected traffic load (Range of 6

to 10, with 8.3 nominal)

4.4.1.3 UBS Max Pilot Power for Mix of 1X and DO Carriers (does not apply to M810)

Table 4-11: SC4812T—MC BTS Maximum Pilot Power (dBm)

Carrier Type Frame Type

800 MHz 1.9 GHz

IS95 / cdma2000 1x 37.8 dBm 36 dBm

1xEV-DO 41.8 dBm 41.8 dBm

WRatioPilottoMaxPwrCarriersTraffic

GroupsFHPBPilotPwrFHPBRatedPowerrMaxPilotPw

___*_#

_*#85.1*_−=

RatioPilottoMaxPwrCarriersTraffic

CarriersDOPwrDOnsPilotBeacoPilotPwrPBRatedPowerrMaxPilotPw

___*_#

_*#_*#85.1*_ −−=


4


Where:

• RatedPower is based on frame type and # of PA's equipped• PB_PilotPwr is the Pilot power for the Pilot Beacon sector-carriers in Watts (these are

fixed carriers for all frame types except M810, and SC300/SC340)• 1.85 is the ratio of Pilot/Page/Sync power to Pilot Power for Page = 110, Sync = 40. If

Page = 90, use 1.60• #PilotBeacons is the number of Pilot Beacon sector-carriers configured• DO_Pwr is the DO SIFPilotPwr setting in Watts• #DO_Carriers is the number of 1xEV-DO sector-carriers configured• #Traffic_Carriers is the number of 1X sector-carriers configured• MaxPower_To_Pilot_Ratio is 10 for systems that use a Page gain = 110 and 8.3 for

systems that use a Page gain of 90. This represents the ratio of the fully loaded sector-carrier power to the pilot power.

Additional factors that will have an impact on the power amplifier are:

• The use of external duplexers should be accounted for by including an additional 0.5 dB of loss, nominally. For SC4800-series “E” options (i.e. outdoor products), 4812T-Lite, SC7224, SC480 with cCLPA, UBS and the M810, duplexers are included and the specifications will already reflect the duplexer loss.

• Products exploiting PA trunking across sectors (e.g. SC4812T) have both sector-carrier and site-carrier limits of which to be aware. For example, a three-sector SC4812T at either 800 MHz or 1900 MHz can deliver 67.5 Watts total for the site-carrier, but is rated for 22.5 Watts with equal sharing for an individual sector-carrier (not including duplexer loss).

• Verify that the Pilot, Page, Sync, and Traffic Channel power relationships can be established. Although the PA may be rated to deliver the desired total power output, other devices may limit the input signals into the LPA or the ratios among them. For example, there are gain limits on the Paging, Sync, and Traffic channels of 127 (7FHEX). Pilot gain can not be changed and is always be set to 127.

• Account for any thermal limitations. Typically for indoor products, the operating temperature range is 0°C to 50°C. The ARP is expressed in dBm or Watts at 25°C, the midpoint in the temperature range. An allowance for variation due to temperature is provided. For example, the 800 MHz SC4812T specification is as follows: Transmitter Sector Output Power with equal power sharing per sector (non-duplexed):43.5 dBm (22.5 W) @25°C ±2 dB over temperature.

4.4.1.4 UBS Sector-Carrier Power Levels

4.4.1.4.1 1.9 GHz UBS Sector-Carrier Power Levels

The following table shows the allowable 1X/EV-DO carrier combinations and power levels per


4


XMI.

1X Pilot Power assumes Tx Ec/Ior > 0.15 Rated 1X Power assumes 1.5 Trunking factor for 3 sectors.

For C25 and later, Table 4-13 shows allowable 1X/EV-DO carrier combinations and power levels per XMI.

Table 4-13: C25 and Later, Max 1.9-UBS Power Levels in Various Carrier Configurations

Table 4-12: Max 1.9-UBS Power Levels in Various Carrier Configurations

1X Carriers EVDO Carriers# RatedPwr

(W)Pilot Pwr

(W)Pilot Pwr

(dBm)# RatedPwr

(W)Pilot Pwr

(dBm)RatioDO/1X (dB)

Combiner

3 16 2.4 33.8 0 0 0 - W or WO2 16 2.4 33.8 1 15 41.8 -0.28 W or WO

0 0 - - 2 15 41.8 - W or WO0 0 - - 3 13.6 41.4 - WITHOUT

0 0 - - 3 10.7 40.3 - WITH0 0 - - 4 9.0 39.6 - WITHOUT0 0 - - 4 8.1 39.1 - WITH

1 14.1 2.1 33.3 2 13.2 41.2 -0.29 WITHOUT1 12.8 1.9 32.9 2 12 40.8 -0.28 WITH

1 10.4 1.6 32.0 3 9.7 39.9 -0.30 WITHOUT1 9.3 1.4 31.5 3 8.7 39.4 -0.29 WITH

2 11.2 1.7 32.3 2 10.4 40.2 -0.32 WITHOUT2 10.1 1.5 31.8 2 9.4 39.8 -0.31 WITH

3 14 2.1 33.3 1 13.1 41.2 -0.29 WITHOUT3 12.3 1.8 32.7 1 11.5 40.6 -0.29 WITH

4 15.4 2.3 33.7 0 0 - - WITHOUT4 13.7 2.1 33.2 0 0 - - WITH

1X Carriers EVDO Carriers# RatedPwr

(W)Pilot Pwr

(W)Pilot Pwr

(dBm)# RatedPwr

(W)Pilot Pwr

(dBm)RatioDO/1X (dB)

Combiner

0 0 - - 1 to 3 15.0 41.8 - WITHOUT0 0 - - 4 11.3 40.5 - WITHOUT

1 16.0 2.4 33.8 1 to 2 15.0 41.8 -0.28 WITHOUT1 13.0 2.0 32.9 3 12.2 40.9 -0.28 WITHOUT

2 16.0 2.4 33.8 1 15.0 41.8 -0.28 WITHOUT2 14.1 2.1 33.3 2 13.2 41.2 -0.29 WITHOUT

3 15.4 2.3 33.6 1 14.5 41.6 -0.26 WITHOUT1 to 4 16.0 2.4 33.8 0 0 - - WITHOUT

0 0 - - 1 to 2 15.0 41.8 - WITH0 0 - - 3 13.8 41.4 - WITH


4


1X Pilot Power assumes Tx Ec/Ior > 0.15Rated 1X Power assumes 1.5 Trunking factor for 3 sectors.

4.4.1.4.2 800 MHz UBS Sector-Carrier Power Levels

The following table shows allowable 1X/EV-DO carrier combinations and power levels for a single XMI at 800 MHz.

1X Pilot Power assumes Tx Ec/Ior > 0.12Rated 1X Power assumes 1.5 Trunking factor for 3 sectors.

4.4.1.4.3 Flexi CDMA UBS 800 MHz FXCA RFM

The following table shows allowable 1X/EV-DO carrier combinations and power levels

0 0 - - 4 10.3 40.1 - WITH1 16.0 2.4 33.8 1 to 2 15.0 41.8 -0.28 WITH

1 11.9 1.8 32.5 3 11.1 40.5 -0.30 WITH2 16.0 2.4 33.8 1 15.0 41.8 -0.28 WITH

2 12.9 1.9 32.9 2 12.1 40.8 -0.28 WITH3 14.1 2.1 33.2 1 13.2 41.2 -0.29 WITH

1 to 3 16.0 2.4 33.8 0 0 - - WITH4 15.5 2.3 33.7 0 0 - - WITH

Table 4-14: 800 MHz UBS Single XMI Power Levels

1X Carriers EVDO Carriers

# RatedPwr (W)

Pilot Pwr (W)

Pilot Pwr (dBm)

# RatedPwr (W)

Pilot Pwr (dBm)

RatioDO/1X (dB)

Combiner

3 20 2.4 33.8 0 0 - - WITHOUT

2 20 2.4 33.8 1 15 41.8 -1.25 WITHOUT1 19.5 2.3 33.7 2 14.8 41.7 -1.20 WITHOUT0 0 - - 2 15 41.8 - WITHOUT

0 0 - - 3 14.1 41.5 - WITHOUT4 18.75 2.3 33.6 0 0 - - WITHOUT

3 15.2 1.8 32.7 1 11.4 40.6 -1.25 WITHOUT2 14.8 1.8 32.5 2 11.1 40.5 -1.25 WITHOUT

1 14.5 1.7 32.4 3 10.9 40.4 -1.24 WITHOUT0 0 - - 4 10.4 40.2 - WITHOUT

1X Carriers EVDO Carriers


4


for a Flexi CDMA UBS with one 800 MHz FXCA RFM.

4.4.1.5 Power Out Differences Between UBS-Macro and 4812T/4812T-MC BTSs

When replacing a 4812T or a 4812T-MC frame with a UBS-Macro frame, the 1X and DO SifPilotPwr levels will need to be adjusted to obtain the same coverage.

When an operator sets 1X and DO SifPilotPwr levels, the reference point where SifPilotPwr is measured is at the frame output. There is a difference in what is included in the power reference point between 4812T/4812T-MC and UBS-Macro frames.

The 4812T/4812T-MC frames do not include duplexers and interconnecting cables in the power reference point. The UBS-Macro frames include duplexers in the power reference point. Thus, the power supplied to the antenna feed line is reduced by the duplexer and interconnecting cable loss in the 4812T/4812T-MC case. The power supplied to the antenna feed line is NOT reduced by the duplexer and interconnecting cable loss in the UBS case.

If a 4812T/4812T-MC is replaced with a UBS-Macro frame and the SifPilotPwr levels are not changed, the power levels at the antenna will be approximately 0.8 dB higher than the 4812T/4812T-MC case. As a result, coverage of the new UBS frame may be larger than the previous 4812T/4812T-MC frame. So, it is recommended to set all UBS 1X and DO SifPilotPwr levels 0.8 dB lower than the previously installed 4812T/4812T-MC frames.

Note:• For UBS-Macro frames, the SifPilotPwr levels for non-rural BTSs should not be set

higher than 33.0 dBm for 1X carriers. For EV-DO carriers in UBS-Macro frames, the SifPilotPwr levels should never be set higher than 41.8 dBm. If a UBS-Macro frame is replacing a 4812T/4812T-MC frame that had an EV-DO carrier with a SifPilotPwr of 41.8 dBm, the EV-DO SifPilotPwr for the UBS-Macro frame should be set to 41.0 dBm to obtain the same coverage.

• For releases R22.0 and beyond, feature FR8990, Fixed Trunk and Sector Group Limiting, should be enabled to prevent overdriving the XMIs, which would lead to an XMI shutdown. For more information on Fixed Trunk Group Limiting and Fixed

Table 4-15: Flexi CDMA UBS 800 MHz FXCA RFM Power Levels

1X Carriers EVDO Carriers# of

carriersRated Pwr

(W)Pilot Pwr (W) Pilot Pwr

(dBm)# of

carriersRated Pwr

(W)Pilot Pwr (W) Ratio DO/1X

(dB)

3 20 2.4 33.8 0 0 - -2 20 2.4 33.8 1 15 41.8 -1.25

1 20 2.4 33.8 2 15 41.8 -1.250 0 - - 4 15 41.8 -

4 15 1.8 32.6 0 0 - -3 16 1.9 32.9 1 12 40.8 -1.252 17.2 2.1 33.2 2 12.8 41.1 -1.28

1 18.3 2.2 33.5 3 13.9 41.5 -1.19


4


Sector-Carrier Limiting, refer to "Fixed Power Limiting-LPA Overload Protection Optimization" (document number wp316145769) available at https://online.portal.nokiasiemensnetworks.com. Search by the document number to locate the document.

• Calibration of 4812T/4812T-MC BTSs must be done to the top of the frame. Calibration with external duplexers is not recommended, as the output power required could exceed the LPA rating.

• BTS types 4812ET, 4812ET-Lite, and 4812T-Lite have internal duplexers. Therefore, the reference point for these BTSs like the UBS includes the duplexers. Thus, no adjustment in SifPilotPwr is necessary when replacing these BTS types with a UBS frame.

4.4.2 CDMA Signal Power Distribution Characteristics and PA Sizing

There are three characteristics of the CDMA signal power distribution that are useful in discussions on PA requirements, which can be compared to PA equipment capabilities. These include:

1. The Long Term Average (LT-AVG): represents an average over 30 minutes or more. For the PA to be sized correctly, the LT-AVG must be less than or equal to the Average Rated Power (ARP).

2. The Short Term Average (ST-AVG): represents an average over 5 minutes. It may prove useful, as a rule of thumb, to compare the ST-AVG to the ARP. Greater detail on this can be found in the next section.

3. The Very Short Term Average (VST-AVG): represents an average over less than 2 seconds. For the PA to be sized correctly, the VST-AVG must be less than or equal to the High Power Alarm Rating.

Note that any peak excursions significantly higher than the VST-AVG are of very short duration and are managed by PA overload protection mechanisms.

4.4.3 General Power Relationships

As a result of various simulation studies and field data, the following characteristics of a system that is interference limited (i.e. fully loaded) have been derived and may be considered rules of thumb:

1. The LT-AVG is approximately 6.7 times the Pilot power.2. For those systems that use a page gain of 110, the ST-AVG is approximately 10 times

the Pilot power. This is 1.7 dB over the LT-AVG. For those systems that use a page gain of 90, 8.3 times the pilot power is the approximate factor, 0.9 dB over the LT-AVG.

3. The VST-AVG is approximately 15 times the Pilot power. This is ~4.8 dB over the LT-AVG and ~1.8 dB over the ST-AVG.

Given the deviation of the power distribution, the system designer will generally find the indoor products (i.e. SC4812 series, SC7224, and UBS) and the outdoor products with fans (SC4812ET/ET Lite and UBS with an Environmental Enclosure) to be High Power Alarm (HPA) limited. Since


4


the ST-AVG is ~1.8 dB below the VST-AVG and the Average Rated Power (ARP) is 2 dB below the HPA, using a ST-AVG comparison to the ARP can provide a convenient rule of thumb for estimating the PA requirements for these products. Specifically, the ST-AVG should be no greater than the ARP.

Based on analysis of field data, RC3 uses 5.0 to 6.5% pilot power for each traffic channel. This is very typical as a system wide average power. RC3 power requirements depend on where the users are with respect to the BTS. It is best to locate BTSs close to high user densities in order to minimize the power per user, maximize the site user capacity, and minimize the interference to the surrounding sites. RC4 requires an extra 10% power, but has twice the Walsh code capacity. For Rate Set 1 (RC1), ~13.5% of the Pilot power would be consumed on average. For Rate Set 2 (RC2), ~27.8%. IS-2000 forward link RC 3, RC 4, and RC 5 have up to twice the Forward Link capacity of IS-95A/B; therefore, the average TCH powers in these modes are approximately 1/2 the RC1 value. [Greater detail on these estimates can be found in Section 4.4.5.] The number of forward links associated with this estimate is the 98th percentile of forward links and would include soft/softer links (i.e. 2% Erlang B on Walsh code usage). This would also correspond to the ST-AVG.

The RC1 and RC2 traffic channels correspond to the fundamental rates of 9600 bps and 14400 bps modes of RC 1 and RC 2 of IS-2000. The IS-2000 1X BTS will also support RC 3, RC 4, and RC 5. These Radio Configurations employ different error correcting schemes, and offer higher data rates than RC 1 and RC 2 (up to 153,600 bps will be supported in RC 3 and RC 4). In general, data rates higher than 14400 bps will require proportionately higher traffic channel powers (and lower traffic channel capacities) than discussed above.

There is a level of Pilot power which will balance the reverse link. To increase the Pilot power beyond this level will not significantly improve the composite area reliability, since the reverse link becomes limiting. For this reason, it is recommended that the Pilot powers be designed to levels sufficient to balance the reverse link, but not excessively so as to conserve the PA resource. The recommended pilot power is 2.4 W, because it balances the link budget.

The introduction of a tower-top amplifier will improve the reverse link by effectively negating the losses between the antenna and the top of the rack (approximately 3 to 4 dB, refer back to Section 4.2.3.2.1). This improvement would necessitate a compensatory increase in forward power to balance the links. When a TTA is introduced under the assumption of light loading (e.g. “highway site”), it is more likely that the links can be balanced. It is not recommended to use TTAs elsewhere.

4.4.4 Design Guidelines

When initially designing a CDMA system, 2.4 Watts is the recommended pilot power for all BTSs, since it balances the reverse link. Testing shows it is better to run using balanced Forward/Reverse paths in order to maximize in building penetration in dense urban areas and coverage in rural areas.

Nokia Siemens Networks’ IDGP CDMA Simulator (or comparable design tool) can be utilized to generate statistics for a CDMA design. These statistics can be analyzed to determine if any sectors will have a potential PA issue.


4


• Evaluate the coverage/capacity/quality impacts of selected pilot powers.• The confidence level is impacted by the number of Monte Carlo runs performed in

generating the data.• Evaluate the power requirements of each sector-carrier. Outputs from the CDMA

Simulator include statistics on traffic channel (TCH) power and forward links (i.e. Walsh codes). Details on this evaluation can be found in the RF Design Procedure.

For conventionally powered BTS products (i.e. no sharing of PA resources across multiple sectors and/or carriers), it is only necessary to determine the LT-AVG and VST-AVG requirements for the sector-carrier and then compare them with the ARP and HPA ratings, respectively. The ratings must exceed the requirements. For TrunkedPower™ BTS products, there are two steps:

1 Determine the LT-AVG and VST-AVG requirements over the appropriate set of sector-carriers over which the PA resource is shared (i.e the Trunk Group) and then compare them with the ARP and HPA ratings, respectively. The ratings must exceed the requirements.

2 Determine the LT-AVG requirement for each individual sector-carrier and then compare this with the ARP rating for a sector-carrier. The rating must exceed the requirement.

As has been stated earlier, the SC4812T is rated for 22.5 Watts ARP with equal sharing in any individual sector-carrier and 67.5 Watts total for 3 sectors of 1 carrier (not including duplexer loss).

For a 4812T-MC, the ARP per sector-carrier, assuming equal sharing with other sector-carriers in the trunk group, is equal to the max power from Table 4-8 divided by the number of sector-carriers.

4.4.4.1 Comparison to Average Rated Power

The following steps can be performed to obtain the LT-AVG for the sector-carrier(s) which can be compared with the product ARP specification (for many products, these values are provided in Table 4-8).

1. Take the average of the TCH power distribution. For trunked PAs, generate the average for the individual sector-carrier for comparison against sector-carrier ARP limits and then again for all the sector-carriers over which the resource is to be shared for comparison against total ARP limits. For the total ARP comparison, the power statistics must first be summed across the appropriate set of sector-carriers within each Monte Carlo run. Although this will not impact the average, it will impact the deviation.

2. Add in the constant power components associated with the Pilot, Page, and Sync channels.

3. Compare this with the ARP of the PA. It must be lower.

Note: to compare the ST-AVG to the ARP, use the 98th percentile of the TCH power distribution.


4


4.4.4.2 Comparison to High Power Alarm Rating

The following steps can be performed to obtain the VST-AVG for the sector-carrier(s) which can be compared with the product HPA specification (for some products, these values are provided in Table 4-8).

1. Determine the 98th percentile of the TCH power distribution. For trunked PAs, generate the average for all the sector-carriers over which the resource is to be shared for comparison against total HPA limits. The power statistics must first be summed across the appropriate set of sector-carriers within each Monte Carlo run. The 98th percentile is then taken across the summed set of statistics.

2. Scale it up by a factor of 1.5. This compensates for variations in the voice activity factor (up to a level that corresponds to the 98th percentile of the binomial distribution).

3. Add in the constant power components associated with the Pilot, Page, and Sync channels.

4. Compare this with the High Power Alarm Rating. It should be lower.

4.4.4.3 Comparison to Walsh Code Limit

1. Take the average number of forward links. This may be interpreted as Walsh code Erlangs.

2. Calculate a maximum number of forward links based on 2% GOS Erlang B for the number of Walsh code Erlangs derived in step 1.

3. Compare step 2 results to the Walsh code limit. It should be lower.

4.4.5 General Power Requirements

In the absence of more precise simulations, here are some definitions and equations that can be used to provide power requirements as a function of Rate Set, pilot power, and number of forward links.

Definitions:• Ppilot is the Pilot power.• Ppage is the Page power (commonly 75% of P_pilot w/ Pilot gain of 110 or 50% w/ Pilot

gain of 90).• Psync is the Sync power (commonly 10% of P_pilot).• FwdLinks50th-%ile is equivalent to Walsh code Erlangs. It can be derived from the

Effective Traffic Load using the Soft/Softer Handoff Factor.• FwdLinks98th-%ile is equivalent to the number of Walsh codes that result from taking

Walsh code Erlangs at 2% Erlang B.• Veff (Effective Voice Activity Factor) is scaled up from the normal VAF (Voice

Activity Factor) to compensate for Power Control Bit puncturing on the forward link. The PCB bits are transmitted at a constant high power to maintain the integrity of the closed loop power control mechanism. Scaling the VAF is one method of compensating


4


for the effect on forward power output. V_eff is 0.55 and 0.47 for Rate Sets 1 and 2 of IS-95 systems, and Radio Configurations 1 and 2 of IS-2000 1X systems, respectively.

• Vwc represents, for the VAF binomial distribution, a ratio of the 98th percentile to the mean. A value of 1.5 is used.

• Ptch_avg is the Average Traffic Channel Power. As a fraction of Ppilot, these powers are typically 5 to 6.5% of Ppilot for RC3 and RC4 voice, and 13.5% and 27.8% for Rate Sets 1 and 2, respectively with Veff included.

Assume:

[EQ 4-50]

4.4.5.1 Minimum ARP Based on LT-AVG Estimate

The following equations can be used to determine the minimum ARP specification based on the Pilot power and the average number of links.

[EQ 4-51]

RC3:

[EQ 4-52]

RC4:

[EQ 4-53]

Notes:1. To compare the ST-AVG to the ARP, use FwdLinks98th-%ile in place of FwdLinks50th-

%ile2. Formulas for Rate Set 1 and 2 also apply to RC 1 and RC 2 respectively.

4.4.5.2 Minimum HPA Based on VST-AVG Estimate

The following equations can be used to determine the minimum HPA specification based on Ppilotand FwdLinks98th-%ile.

[EQ 4-54]

Ppilot Ppage Psync+ + 1.85 Ppilot× for a Page gain of 110=Ppilot Ppage Psync+ + 1.6 Ppilot× for a Page gain of 90=

AverageRatedPower PPilot PPage PSync FwdLink50th-%ile Ptch_avg× Veff×+ + +=

AverageRatedPower PPilot 1.85[× F( wdLink50th-%ile 0.059 ) ] Page gain =110×+=AverageRatedPower PPilot 1.6[× F( wdLink50th-%ile 0.059 ) ] Page gain =90×+=

AverageRatedPower PPilot 1.85[× F( wdLink50th-%ile 0.065 ) ]× Page gain = 110+=

AverageRatedPower PPilot 1.6[× F( wdLink50th-%ile 0.065 ) ]× Page gain = 90+=

HighPowerAlarmRating PPilot PPage PSync FwdLink98 th-%ile Ptch_avg× Veff Vwc××+ + +=


4


RC3:

[EQ 4-55]

RC4:

[EQ 4-56]

Alternatively, an upper estimate on FwdLinks98th-%ile can be determined based on the HPA rating and Ppilot. This may serve as a Walsh code limit that will block traffic at levels that near the HPA rating.

RC3:

[EQ 4-57]

RC4:

[EQ 4-58]

4.4.5.3 Exceeding the High Power Alarm Rating

On systems where carrier load management features are not enabled, an LPA module which exceeds its High Power Alarm Rating will enter an OOS_RAM maintenance state. The consequences and possible operational response to this event were outlined in FYI No. SCCDM-1997.84 March 20, 1997. LPA modules in systems having these features installed will not enter OOS_RAM.

If OOS_RAM events are occurring, the following design and optimization options could be taken:

• Add more PA power. Depending upon the BTS product and the installed configuration, there may be an ability to add an additional PA module.

• Re-optimize the pilot power to a lower level. Be careful to review the potential consequences on coverage. If the sites involved have the potential for significant overlap, then lowering pilot powers may be the appropriate response.

• Re-optimize the forward power control parameters. For example, reducing the Nominal Traffic Channel Gain can reduce the overall output power and PA requirements.

HighPowerAlarmRating PPilot 1.6[× F( wdLink98th-%ile 0.089 ) ]× Page gain = 90+=




FwdLink98th-%ile HighPowerAlarmRating PPilot )⁄ -1.85 ] 0.089⁄ Page gain = 110([=





4


• A Walsh code limit can be implemented which will maintain traffic on a sector-carrier basis to levels which should not exceed the High Power Alarm Rating of the PA. Determining this threshold can be based on the information provided here. Once Walsh code limits are in place, Walsh code usage and blocking statistics may be monitored and projected against the limit per standard traffic engineering guidelines.

4.4.5.4 Carrier Load Management Overview

With feature 1225B, a Fixed Power Threshold (dBm) sets the maximum output allowed per sector/carrier and will limit the LPAs providing power to that sector/carrier. This parameter is used only when the system has the Activate Fixed Overload Protection parameter enabled. This attribute establishes a high water mark at which the CDMA transceivers will actively reduce gain if this power threshold is exceeded for the given sector/carrier.

With feature 415B, the decision by the mobility manager (MM) to allocate a Walsh code or channel element for subscriber originations and terminations is conditional upon the RF load in the forward and reverse directions on the carrier selected for an allocation attempt.

The Group Line Interface (GLI) or Digital Module Internal (DMI) card at each BTS is responsible for gathering real time forward and reverse link quality data from the traffic channel elements and CDMA transceivers within each sector-carrier under its control. Forward and reverse channel RF quality information is sent to the MM via SCAP (Application Protocol) messaging and used by the MM to make decisions about whether or not to allow new call channel allocation within a sector-carrier and to load balance channel allocation among carriers within a particular sector.

The GLI or DMI will also set a flag in the SCAP measurement report when the sector-carrier's CDMA transceiver exceeds a user defined power output. The MM will deny origination/terminations in the sector-carrier until the flag is cleared in a subsequent SCAP message.

The GLI or DMI will also calculate the actual power being used by each sector-carrier's CDMA transceiver, as well as the total power output of the LPA associated with the sector-carrier, and forward the information to the MM via the periodic SCAP RF metrics reporting messages. This data is for statistics collection and not used by the MM to make channel allocation decisions.

With feature 4472C (available starting with CBSC Release 16.0), in addition to gathering real-time forward and reverse link quality data from the traffic channel elements and CDMA transceivers within each sector-carrier under its control, the RF Load Manager at each BTS is responsible for using the measured forward TCH and SCH power and reverse RNR for each sector-carrier to provide near real-time updates of forward and reverse load conditions to the Time Slice Manager. The Time Slice Manager is a BTS based mechanism to schedule data activity in a series of small periods of time to maximize use of the forward and reverse power capacity.

The RF Load Manager will also inhibit supplemental allocation in the sector-carrier when the sector-carrier's CDMA transceiver exceeds a user-defined fixed limit power output or if the sector-carrier's LPA is in gain limiting mode due to an LPA overload condition.

In R22, FR8990, Fixed Trunk and Sector Group Limiting has been added. This feature provides a


4


method to limit power at the LPA and/or sector level. One of the benefits of FR8990 is that the operator can allocate more power per sector-carrier (higher SIFPilotPwr) than if only FR1225B were used, as the trunk group limiting function will monitor the total power of all sector-carriers in the trunk group to prevent traffic loading from exceeding the PA capability. This can be particularly helpful in cases where the peak traffic load moves between different sectors during the day for BTS's with multiple sectors in a trunk group. For single sector trunk groups, M810 and UBSc with RRH, FR8990 can be used in place of FR1225B to allow full utilization of the PA capability when traffic loading between carriers is not always balanced.

Once enabled, Fixed Trunk Group limiting protects the LPA(s) from being overdriven. Unlike FR1225B, Fixed Sector-Carrier Limiting, FR8990 only limits once the power capability of the LPA or Sector have been reached. There are no thresholds to set. They thresholds are dervied automatically according to the power capability of the LPA. Fixed Sector Group Limiting is only used for SC7224 and UBS Macro frames. Sector Group Limiting has a threshold of 160 W and is designed to protect the IDRF from too much power. Refer to the FR8990 DFD for more information. For more information on Fixed Trunk Group Limiting and Fixed Sector-Carrier Limiting, refer to the "Fixed Power Limiting - LPA Overload Protection Optimization" (document number wp316145769) available at https://online.portal.nokiasiemensnetworks.com. Search by the document number to locate the document.

4.4.6 Power Allocation in Mixed Mode 1X and DO Systems

For SC4812T-MC, SC480, and UBS, it is possible for 1X and DO carriers to share the same LPA i.e. be in the same trunk group. To make sure that the LPA is not overburdened, power allocation needs to be done. The power of a 1X carrier can be approximated by SifPilotPwr in Watts multiplied by a factor that represent the ratio of SifPilotPwr to total sector-carrier power. The power of a DO sector-carrier is just the DOSifPilotPwr. Unlike 1X, DOSifPilotPwr represents the fully loaded sector-carrier power and needs no further adjustment.

The approximate total power of the carriers is as follows:

[EQ 4-59]

Where:

• 1XSifPilotPwrWn is the SifPilotPwr of the n-th 1X sector-carrier in Watts• DOSifPilotPwrWm is the SifPilotPwr of the m-th DO sector-carrier in Watts• MaxPower_To_Pilot_Ratio is 10 for systems that use a Page gain = 110 and 8.3 for

systems that use a Page gain of 90. This represents the ratio of the fully loaded sector-carrier power to the pilot power.

TotalTrunkGroupPower must be less than the ARP of the trunk group.

TotalTrunkGroupPower 1XSifPilotPwrWn*MaxPower _To_Pilot_Ratio

DOSifPilotPwrWm+=


4


4.4.7 Government Regulations

Certain government rules and regulations may exist which prohibit an operator from transmitting an excess of power. For instance, the FCC regulations limit the Base Station output power to 1640 Watts EIRP per carrier for PCS systems. Knowing the maximum power for a sector at the top of the rack, this FCC limit will translate into a limit on antenna gain offset by cable losses. For example, the three-sector SC4812T is rated for 45 Watts maximum for a sector-carrier. Consequently, the maximum gain permitted between the top of the rack and the effective radiated power would be Gmax:

[EQ 4-60]

The RF system designer is advised to determine if any regulations exist in the area of their system.

4.4.7.1 Power Amplifier Operational Measurements (FR9235)

In R19, FR9235, Power Amplifier Operational Measurements, introduced some PM pegs to aid the operator in determining whether or when additional PA resources were required for a BTS.

FR9235 provides peak and average power out readings for each sector-carrier for 1X or DO capable BTSs, and trunk group peak and average power out readings for trunked PA BTSs, at 30 minute intervals. Indicators of the power capacity utilization are also provided. To summarize, here are the power measurements provided:

• Average Sector-Carrier power for each sector-carrier• Peak Sector-Carrier power for each sector-carrier• Average Trunk Group for each trunk group• Peak Trunk Group for each trunk group• Percent average sector-carrier power utilization for each sector-carrier• Percent peak sector-carrier power utilization for each sector-carrier• Percent average trunk group power utilization for each trunk group• Percent peak trunk group power utilization for each trunk group

The peak and average power out readings provided are based on 2 second average power measurements taken at 2 second intervals throughout the 30 minute interval. The reported average power is the average of the 900 sample points, and the peak power is the largest value of the 900 sample points, in the 30 minute interval. For trunk group measurements the results are derived from the sum of the individual sector-carrier measurements.

FR9235 also provides peak and average power utilization. Peak and Average power utilization is calculated by dividing the peak and average powers by the sector-carrier or trunk group rated power. Trunk group power is determined by multiplying the number of PA modules (NUMMODULES) in the trunk group by the rated power (RATEDPWR) of each LPA module. The GLI determines NUMMODULES by looking up the number of LPA modules equipped for a trunk group. RATEDPWR is defaulted at the OMCR based on frame type, bandclass and PA type. In a small number of cases, RATEDPWR may have to be entered manually. For the UBS,

Gmax 10 Pout Pin⁄( )log× 10 1640 45⁄( )log× 15.62 dB= = =


4


RATEDPWR is an internal parameter only and is not at the OMC, the value is determined from the XMI directly. The sector-carrier rated power is determined by dividing the trunk group power by the number of sector-carriers in the trunk group (this assumes equal power division among sector-carriers in the trunk group).

The peak and average power values are in milliwatts and the feature is enabled at the BTS level. The power measurements are only provided for packet backhaul configurations.

The sector-carrier and trunked group average and peak transmit power can be used in the determination of both current utilization and for the forecasting of carrier exhaustion using trending techniques. This allows better utilization of infrastructure investment and can be used for network optimization to identify areas which lag in power utilization. The information could also be used as loading input for system design/simulation software to improve modeling results.

Because Trunked LPAs can share power among the sector-carriers that it amplifies, the LPA capacity is really the total power out the LPA can deliver. Thus, LPA usage is best represented by the sum of all the sector-carrier powers rather than by the power of each sector-carrier. The trunk group peak and average power measurements are a good indication of how much of the LPA capacity is being used.

FR9235 applies to SC™4812- series, SC480, SC™7224, and UBS BTSs. The trunk group PM pegs are only for SC™4812- series, SC™7224, and UBS BTSs.

1 FR9235 will only be implemented in BTSs with 1X or DO capable sector-carriers that support 2 second power reporting to the GLI/DMI.

2 Trunked peak and average power will be available only if all sector-carriers in the trunk group are 1X or DO capable.

3 Power measurements are only provided for packet backhaul configurations.4 All the pegs obtained for FR9235 are based on the fact that the devices reporting are INS

during the whole collection interval.

4.5 CDMA Repeaters

Repeaters have been successfully deployed in CDMA markets. By carefully following the guidelines provided by the repeater vendor, it should be possible to deploy a repeater to enhance system coverage for most repeater applications. The following sections provide considerations regarding the design, installation, optimization, and maintenance of a repeater system. All of the repeater information provided should be evaluated prior to deciding upon a specific repeater application.

4.5.1 CDMA Repeater Design Considerations

The following sections provide useful information that should be considered during the design phase of a repeater deployment.


4


4.5.1.1 Coverage Impact

CDMA system coverage can be traded off for more capacity. This is reflected in the link budget of the reverse link by determining the acceptable interference margin allowed, which will determine the reverse link coverage. By designing the system with a relatively small interference margin, less users can be supported, but a larger coverage area is supported. For a relatively larger interference margin, more users can be supported, but for a smaller coverage area. Similarly on the forward link, it is the required PA power that is used to determine the desired mixture of coverage and capacity. For a given load, a smaller coverage area produces a smaller PA power requirement, while a larger coverage area produces a larger PA power requirement. For a given coverage area, the required PA power is directly proportional to the load. This relationship is maintained up to the point where the system becomes forward link interference limited, such that increasing PA power does not maintain or improve SNR.

4.5.1.1.1 Typical CDMA Repeater Applications

In some cases, it is desirable to use transceivers called repeaters (see Figure 4-14) to boost CDMA signals, which in effect spreads the capacity of the BTS to a larger coverage area. This is especially useful in areas where the signal from the BTS is blocked by some kind of RF obstruction. In this case, a repeater can be used between the donor BTS and the served subscriber to boost the signals. The repeater helps to get both the BTS and subscriber signals around or through such RF obstructions.

Figure 4-14: Typical Repeater Application

Repeaters can typically be used to provide improved coverage for the following applications: terrain limited coverage, in-building coverage, and tunnel/subway/parking garage/underground coverage. Using repeaters in this way maintains the coverage of the donor BTS while eliminating the need for another BTS (assuming the donor BTS has enough capacity availability to accept the additional load from the repeater). This is economical as long as the repeater is significantly cheaper than the type of BTS to be added (in comparison to a macro-cell, micro-cell, or pico-cell) and/or the site costs are less expensive. In the overlap areas of coverage between the donor BTS and the repeater, there is enough delay in the repeater signal path such that the subscriber can

BaseStation Repeater

RepeaterCoverage

BTS CellCoverage

DonorAntenna

SubscriberAntenna


4


resolve the signals between the two sources. The same will be true for the reverse link.

4.5.1.1.2 CDMA Repeaters Used for Range Extension

Another application for repeaters are to use them to extend the range of a CDMA cell site or sector for the case where there is no RF obstruction, such as down a highway. For this type of application, the range extension obtained is largely limited by the following:

• How much the repeater desensitizes the base station (for maximizing range of the repeater, typically a 3 dB desense of the donor BTS allows optimum range of the BTS & repeater combination). Note: maximizing overall coverage of the BTS and repeater will cause a 3 dB desense reduction in the donor BTS’s range.

• The cascaded noise figure at the repeater (determined by the noise figures of the repeater and base station including the transmission gain between them).

• Repeater receiver sensitivity on the reverse link and ability to maintain diversity reception back at the donor base station (repeater with transmit diversity is used for link back to donor base station to compensate for repeater not having diversity reception and rake receiver for subscriber to repeater link).

• The effect of the loss of soft handoff of the donor site at the repeater location.• The size of the repeater PA used on its forward link (typically 6 Watts).

Given these assumptions, it has been determined that approximately 24-26% increase in range extension may be achieved by using existing commercial repeaters (see Figure 4-15).

Figure 4-15: Repeater Range Analysis Results

Figure 4-15 shows the percent improvement in range due to adding a repeater (normalized to the BTS range without the repeater) for different BTS donor configurations. This analysis used a typical noise figure value of 4.5 dB. For a guaranteed coverage calculation or prediction, it may be necessary to use the six sigma value for the noise figure specification which is usually 1.5 to 2.5 dB higher than the typical value. A 20Watt LPA was assumed for all cases above. The dBi numbers

R ange Improv e me nt U sing R e pe ate r

0%

10%

20%

30%

40%

50%

60%

70%

80%

90%

100%

Sys te m C o n fig u r atio n

Per

cen

t In

crea

se in

Ran

ge

Rev Link Inc reas e

Fw d Link Inc reas e

Rev Link Inc reas e 26% 26% 26% 26% 24% 24%

Fw d Link Inc reas e 58% 59% 35% 38% 10% 13%

CSM 17dBi CSM 23dBi EMA XX 17dBi EMA XX 23dBiCSM

TTA .17dBiCSM

TTA .23dBi

BTS NF=4.5 dBRptr NF=7.0dB

NIM=0dB or 3dB des ens e


4


represent antenna gain and TTA indicates a tower top LNA was used at the BTS to reduce the BTS effective noise figure. The range is largely limited by the reverse link allowing about a 25% increase in range. While the forward link range extension can be large (above 50%) for a donor site using a CSM chip set, it quickly drops as the receiver sensitivity is improved by using an EMAXX chip set and then again if tower top low noise amplifiers (LNAs) are used to reduce antenna cable loss. Going from left to right, the CSM to the EMAXX, and then to the CSM w/TTA, each configuration improves the receiver sensitivity of the BTS, which in effect increases the normalized range of the BTS. This also increases the power requirements of the BTS LPAs, which is why the forward link improvement decreases quickly due to the fixed 20Watt LPA assumption. By observing the increase in normalized range with each configuration change, the overall reverse link improvement in range is increasing, but the percentage improvement due to the repeater is still around the 24% range. Figure 4-16 represents an alternate repeater analysis with the following assumptions.

• The total loss/gain is the same between the forward and reverse links• The forward link loss/gain is measured from the Forward Tx output of the base station to

the Forward Tx output of the repeater• The reverse link loss/gain is measured from the Reverse Rx input of the repeater to the

Reverse Rx input of the base station• The base and repeater antennas have the same cable losses and antenna gains serving the

subscribersFigure 4-16: Alternate Repeater Analysis

The Y axis in Figure 4-16 represents the difference in repeater forward Tx power relative to the BTS power plus the difference in the repeater forward Tx gain relative to the repeater reverse Rx gain. This is identical to that of Figure 4-25. In a maximum range extension application, the repeater Tx and Rx gains are typically equal and thus cancel themselves out. As a result, the title in the above figure only mentions the difference in repeater to BTS Tx powers. This alternate analysis also shows a ~26% increase in range. An interesting point to note is that in this type of repeater configuration (maximum range), the donor BTS range is reduced by over 40%, primarily

B TS and R epeater R X R ange3.26 R F prop loss

-20.0

-15.0

-10.0

- 5.0

0.0

0.00 0.10 0.20 0.30 0.40 0.50 0.60 0.70 0.80 0.90 1.00 1.10 1.20 1.30 1.40 1.50 1.60 1.70 1.80 1.90 2.00

N orm alized R X C ell co verage referenced to B T S no ise figure

Rep

eate

r F

orw

ard

TX

po

wer

rel

ativ

e to

BT

S

distance BTS distance R epeater

B T S loses -4 dB soft handoff ga in R epeater loses -1 dB fad ing to B T S - 0 .5 dB E c/Io F ingers

Path LossM obile to Repeater

Path LossM obile to BTS

Link Loss = P ath Loss + C able Losses + Antenna G ains+ R epeater G ainAssum ption

R everse L ink R epeater to B T S = Forward L ink B T S to R epeater


4


due to the lack of soft handoff gain and the repeater desense of the BTS receiver. It also shows the expected reductions in overall range as the relative power levels are changed.

4.5.1.2 Cascaded Noise Figure

The calculation of the cascaded noise figure for multiple amplifiers in a cabled system is different than that for a non-cabled repeater system. The following sections provide an explanation of how to calculate the cascaded noise figures for both cabled and repeater (non-cabled) systems.

4.5.1.2.1 Cascaded Noise Figure for Cabled Systems

In a multiple amplifier cabled system (i.e. only one antenna input), [EQ 4-20] and [EQ 4-21] can be used to calculate the cascaded noise figure, if the noise figure (or noise factor) for each of the individual amplifiers which are connected in series is known.

Figure 4-17: Cabled Cascaded Noise Figure

For the example in Figure 4-17 where the noise figures are illustrated by setting the thermal noise, kTB = 1 (-113 dBm for CDMA), the cascaded noise referenced to the first amplifier input is as follows (note that all values are linear, not dB).

Cascaded Noise @ Input =

To simplify the calculation, let’s assume that the noise figures for F1, F2, and F3 are 3 dB (2.0 linear) and the gain for G1, G2, and G3 are 10 dB (10 linear). For the example in Figure 4-17, the cascaded noise at the input is as follows (assuming no cable loss between the amplifiers):

Cascaded Noise @ Input = = 2.101 = 3.2 dB

4.5.1.2.2 Cascaded Noise Figure for Single Repeater System

The reverse link cascaded noise figure for a BTS repeater system can be easier to comprehend if a few simplifying assumptions are made. First, the total loss/gain is assumed to be the same between the forward and reverse links. Second, the BTS and repeater antennas have the same cable losses and antenna gains serving the subscribers. Using the above assumptions, the forward loss/gain is

G1 G2 G3

F1 - 1 F2 - 1 F3 - 11

Input

kTB=1Output

F1F2 1–

G1---------------- F3 1–

G1 G2•---------------------+ +

22 1–10

------------ 2 1–10 10•-----------------+ +


4


measured as the difference between the Forward Tx output of the BTS and the Forward Tx output of the repeater. Also, the reverse loss/gain is measured as the difference between the Reverse Rx input of the repeater and the Reverse Rx input of the BTS. Using the simplifying assumptions, the cascaded noise figure looking into the repeater Rx will be higher than the cascaded noise figure looking into the BTS Rx by the reverse loss/gain (in dB).

Figure 4-18: Base Station & Repeater Diagram

For the simple example in Figure 4-18, the repeater Tx pilot is 10 dB lower than the BTS Tx pilot. Knowledge of the individual components of the forward loss is not required (i.e. the cable losses, antenna gains, and repeater gain are all hidden to our analysis). Using symmetry between the forward and reverse links, the reverse loss is also 10 dB. A CDMA subscriber received at the repeater at a level of -110 dBm will be presented to the BTS receiver at -120 dBm. Using the simplifying assumptions, the cascaded noise figure looking into the repeater Rx is 10 dB higher than the cascaded noise figure looking into the BTS Rx.

An important point to note is that a cascaded noise figure calculation for a repeater system (non-cabled) is not the same as the cascaded cabled amplifier equation. In a repeater system (non-cabled), [EQ 4-21] cannot be used to calculate the cascaded noise figure. Cascaded amplifiers only have one antenna input. Therefore, thermal noise (kTB) is only injected at the 1st amplifier input. Also, subscribers are only received at the 1st amplifier. A repeater and BTS system has two input antennas. Thermal noise (kTB) and the subscriber signal are injected at both receiver inputs. Figure 4-19 provides an example of a reverse link cascaded noise figure for a simple repeater system.

RepeaterBTS

Tx_BTS

Rx_BTS

Tx_R

Rx_R

Forward Loss = 10 dB

Pilot = 2 watts Pilot = 0.2 watts

Reverse Loss = 10 dB


4


Figure 4-19: Repeater Cascaded Noise Figure

As an alternate approach, the calculation of a cascaded noise figure for a repeater system reverse link can be analyzed as follows (see Figure 4-19). Thermal noise (kTB) is introduced at the repeater Rx by the source impedance of the antenna. The 3 dB noise figure of the repeater doubles the noise by adding another kTB. A reverse loss of 10 dB will lower the repeater noise at the BTS antenna to 0.2(kTB). The BTS receiver antenna and noise figure add another 2(kTB). As a result, the total noise at the BTS receiver is 2.2(kTB). Thus, the cascaded noise figure is 3.4 dB (10log(2.2)) looking into the BTS Rx. A simple equation for the cascaded noise figure at the BTS receiver can be written as follows. All of the variables are in linear units (i.e. 2.2 = 2 + (2 * 0.1)).

Cascaded NF @ BTS = BTS NF + (Repeater NF * reverse loss) [EQ 4-61]

Now, a subscriber looking into the repeater receiver will see a different cascaded noise figure than a subscriber looking into the BTS receiver. Referenced to the repeater receiver input, the 2.2(kTB) noise at the BTS receiver is ten times (10dB) higher at 22(kTB). As a result, the repeater cascaded noise figure is 10 dB higher at 13.4 dB (10log(22)). Notice that the 10 dB difference is exactly the same as the reverse loss. A simple equation for the cascaded noise figure at the repeater receiver can be written as follows. Again, all variables are linear (i.e. 22 = 2.2 / 0.1).

Cascaded NF @ Repeater = Cascaded NF @ BTS / reverse loss [EQ 4-62]

In this example, the repeater is 10 dB less sensitive than the BTS. For a subscriber signal to be received at the BTS at -120 dBm, it must received at the repeater at -110 dBm. A subscriber signal going straight to the BTS would be received at the BTS at -120 dBm.

As a result, the cascaded noise figures for a repeater and base station system are easy to calculate. They are determined by the repeater and BTS noise figures and the ratio of repeater pilot power to BTS pilot power. The simplifying assumptions are that the forward and reverse links are balanced. For unbalanced forward and reverse links or to include the effects of CDMA load, first calculate the simple cascaded noise figure and then add in the other effects.

4.5.1.2.3 Cascaded Noise Figure for Cascaded Repeater Systems

For some highway applications where linear range needs to be maximized, a cascaded repeater system may be a viable choice. Similar to the approach used in Section 4.5.1.1.2, a cascaded

1.0

F1 - 1

0.1

F2 - 1

F2 = 3dBF1 = 3dB

kTBSubscriberkTBkTB

RepeaterBase Station (BTS)

0.2(kTB)2.2(kTB)

kTBSubscriber



4


repeater system will impact the capacity and range of the donor BTS in order to maximize the range of the entire BTS/repeater system. Utilizing the same simplifying assumptions, the same approach to calculating the cascaded noise figure for the single repeater system can be applied to the cascaded repeater system. Figure 4-20 provides an example of a reverse link cascaded noise figure calculation for a cascaded repeater system. (Note: The values used in the following example are not indicative of a cascaded repeater system optimized for maximum range extension. The values are chosen to simplify the calculations.)

Figure 4-20: Multiple Repeater Cascaded Noise Figure

The following calculations are similar to the single repeater example. Thermal noise (kTB) is introduced at Repeater #2 Rx by the source impedance of the antenna. The 3 dB noise figure of the repeater doubles the noise by adding another kTB. A reverse loss of 10 dB will lower the repeater noise at the Repeater #1 antenna to 0.2(kTB). The Repeater #1 receiver antenna and noise figure add another 2(kTB). Another reverse loss of 10 dB will lower the combined repeater noise at the BTS antenna to 0.22(kTB). Finally, the BTS receiver antenna and noise figure add another 2(kTB). As a result, the total noise at the BTS receiver is 2.22(kTB), which produces a cascaded noise figure of 3.46 dB looking into the BTS Rx. A simple equation for the cascaded noise figure at the BTS receiver is as follows. All variables are linear (i.e. 2.22 = 2 + (2 * 0.1) + (2 * 0.01)).

Cascaded NF @ BTS = BTS NF + (Repeater #1 NF * reverse loss to BTS) + (Repeater #2 NF * total reverse loss to BTS) [EQ 4-63]

Similar to the single repeater example, the cascaded noise figure looking into Repeater #1 and Repeater #2 are as follows. Referenced to Repeater #1 receiver input, the 2.22(kTB) noise at the BTS receiver is ten times (10 dB) higher at 22.2(kTB). As a result, the Repeater #1 cascaded noise figure is 10 dB higher at 13.46 dB. Referenced to Repeater #2 receiver input, the 22.2(kTB) noise at the Repeater #1 receiver is ten times (10 dB) higher at 222(kTB). As a result, the Repeater #2 cascaded noise figure is 10 dB higher at 23.46 dB. A simple equation for the cascaded noise figure at the Repeater #1 and #2 receiver is as follows.

Cascaded NF @ Repeater #1 = Cascaded NF @ BTS / Repeater #1 reverse loss [EQ 4-64]

Example. Cascaded NF @ Repeater #1 = 2.22 / 0.1 = 22.2 = 13.46 dB

1.0

F1 - 1

0.1

F2 - 1

F2 = 3dBF1 = 3dB

kTBSubscriberkTBkTB

Repeater #1Base Station (BTS)

2.22(kTB)

kTBSubscriber


0.1

F3 - 1

F3 = 3dB

kTBSubscriber

kTB

Repeater #2



0.22(kTB) 0.2(kTB)


4


Cascaded NF @ Repeater #2 = Cascaded NF @ BTS / Repeater #2 reverse loss [EQ 4-65]

Example. Cascaded NF @ Repeater #2 = 2.22 / 0.01 = 222 = 23.46 dB

It is important to note that the reverse loss for Repeater #2 is the total reverse loss from Repeater #2 to the BTS (which includes the loss from Repeater #1 to the BTS). For the example given in Figure 4-20, the total reverse loss from Repeater #2 to the BTS is 20 dB.

4.5.1.3 Interference and Capacity Issues

The interference and capacity impact of a repeater will most likely depend upon its specific application and installation/optimization. The interference and capacity impact should be minimal for a repeater, that is used for a typical application (i.e. to overcome RF obstructions) and that has been properly installed and optimized. A repeater that has not been properly installed or optimized can have an impact on the interference and capacity of the donor BTS.

A CDMA repeater application that is set up for maximum range extension can have a significant capacity impact upon the donor BTS. Since this repeater application is designed to trade-off capacity for coverage, the donor BTS capacity impact depends upon the amount of interference margin that is traded-off for coverage. Again, a repeater that has not been properly installed or optimized for the range extension desired can have a greater capacity impact on the donor BTS than what it was originally designed for.

In order to reduce the number of BTSs for a new system deployment, a system operator may consider implementing a wide scale repeater deployment. A system with a wide scale deployment of repeaters can create multiple paths of interference (direct path from the subscriber, indirect path through the repeater, and indirect paths through multiple other repeaters). Depending upon the system design, a system of this type may increase the reverse link noise rise which may decrease the system capacity. Reverse link simulations of a couple of wide scale repeater design scenarios have shown a decrease in RF carrier capacity of approximately 9-16%. In order to estimate the capacity impact, simulations are highly recommended for any specific wide scale repeater deployment design.

The probability of interference from IM and spectral regrowth are increased with the use of a repeater. The situation may be worse for repeaters because the repeater receiver will add some additional amount of IM and regrowth to the signal that is transmitted. The receiver absorbing this undesired energy at the end of the chain will need to cope with these increased levels of IM and regrowth.

4.5.1.4 Filtering Issues

Depending upon the specific system design (i.e. repeater application, spectrum planning, adjacent band technology, etc.), additional filtering may be required to minimize the interference between the repeater and the adjacent band technologies that are being used. The Sideband Noise (SBN) performance of the repeater may require additional filters to be installed at the repeater site. A detailed guard band interference analysis should be performed to determine the appropriate guard band and filter requirements to allow the repeater and the adjacent band technologies to co-exist


4


with an acceptable interference impact. An analysis of both repeater links (Rx and Tx) is necessary to determine if filtering is required for either link. Separate filters may be required for each of the repeater links.

If additional filtering is required, the additional space requirements must be taken into account when designing the repeater site. If two separate filters are required, then the amount of space required to house and mount the filter hardware needs to be considered. With the potential use of filters at the CDMA donor BTS, at the receiver input of the repeater, and at the output of the repeater, the total group delay of the filters can become a concern. Too much group delay will distort the CDMA waveform, which may cause unacceptable "rho" performance (a measure of waveform quality). The total maximum group delay must be split between the three filters. Since the group delay for the built-in filters of the donor BTS and of the repeater are already established, a lower group delay specification for the additional repeater filter may be required. It may be difficult to find an economical and compact filter to satisfy the group delay requirements in addition to the other filter requirements determined from the detailed analysis.

If it has been determined that additional filtering is required, then the cost impact of the additional filtering should be taken into consideration when designing a repeater site. Since a repeater does not add any capacity to the system, the additional cost of the filtering should be added to the total cost analysis to determine if a regular BTS (macro-cell, micro-cell, or pico-cell) may be more appropriate for the application.

4.5.2 CDMA Repeater Installation Considerations

When using repeaters for a typical application to overcome an RF obstruction within a BTS’s coverage area or for a highway application to maximize linear range extension, it is important to follow the repeater vendor’s installation engineering guidelines.

4.5.2.1 Antenna Isolation

Antenna isolation is a critical parameter for an over-the-air repeater system. If the repeater’s antennas do not have adequate isolation from each other, the repeater’s amplifiers may start oscillating. Proper donor to subscriber antenna isolation at the repeater may be difficult to achieve for some applications. The amount of antenna isolation that is normally required is equal to 15 dB plus the gain of the repeater (refer to the repeater vendor’s recommendation for the actual value to use). Antenna isolation values of 80 dB (repeater gain = 65 + 15 = 80 dB) or greater are not uncommon. Since the environmental surroundings and the physical construction of the site can have an impact, it is highly recommended to actually measure the antenna isolation for each and every repeater site. The ability to measure the antenna isolation properly and accurately is an important step in the repeater installation. Do not rely on estimated antenna isolation calculations to validate the isolation requirements.

The repeater diagram in Figure 4-14 shows the donor antenna at a higher elevation than the subscriber antenna. This represents a repeater application which takes advantage of vertical separation between the donor and subscriber antenna in order to achieve the isolation requirements. Placing the donor antenna at a higher elevation may also provide a direct line-of-sight path to the


4


donor base station, which is highly recommended for all repeater implementations. In some applications, the subscriber antenna may be mounted at a higher elevation than the donor antenna (see Figure 4-21).

Figure 4-21: Alternate Repeater Antenna Configuration

A viable configuration which utilizes horizontal separation along with a barrier is shown in Figure 4-22. For this application, the building is acting as a physical barrier in order to increase the attenuation between the antennas, which will increase the antenna isolation.

Figure 4-22: Horizontal Separation Using a Barrier

Just as long as the measured isolation and the direct line-of-sight requirements are satisfied, the optimal antenna locations may depend upon the particular application.

In some cases where vertical and/or horizontal separation does not provide enough antenna isolation, it may be possible to install custom RF shielding between the donor and subscriber antennas in order to achieve the desired antenna isolation requirements. RF shields can be constructed with various materials (hardware cloth, cyclone chain-link fence, metal screen, solid metal, etc.) and various types of configurations (flat shield, flat shield with corners, curved shield, etc.). The actual attenuation will depend upon the specific application, but nominal values in the


RepeaterCoverage

BTS CellCoverage

DonorAntenna

SubscriberAntenna

BaseStation

Repeater

RepeaterCoverage

BTS CellCoverage

DonorAntenna

SubscriberAntenna

Building


4


range of 10-30 dB of attenuation may be achievable.

As an alternate solution, a micro-wave or fiber linked repeater may be used instead of an over-the-air type repeater. A linked repeater does not have the same antenna isolation requirements as an over-the-air repeater. An example of a micro-wave linked repeater is shown in Figure 4-23.

Figure 4-23: Micro-wave Linked Repeater

Since the micro-wave link is operating at a different frequency and transmitted in a different format, the isolation between the subscriber antenna and the micro-wave antenna is not as critical as the over-the-air repeater. An example of a fiber linked repeater is shown in Figure 4-24.

Figure 4-24: Fiber Linked Repeater

Since the fiber link is not transmitting over the air, antenna isolation is not even a factor for this repeater application.


RepeaterCoverage

BTS CellCoverage

SubscriberAntenna

Micro-wave Link


RepeaterCoverage

BTS CellCoverage

SubscriberAntenna

Fiber Link


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4.5.2.2 Repeater Antenna Considerations

The following sections provide information regarding the repeater donor and subscriber antennas.

4.5.2.2.1 Repeater Donor Antenna

The repeater donor antenna should have a very narrow beamwidth in order to isolate a single donor BTS. In an area with a dense population of BTSs, isolating a single donor BTS may be difficult. If more than one BTS is seen by the repeater, the performance in the repeater’s coverage area may be degraded. As a result, it is typically recommended to use a highly directional, high gain, high front-to-back ratio (for horizontal separation), and/or high side lobe attenuation (for vertical separation) donor antenna with 15° of horizontal beamwidth or less. Parabolic antennas (solid or grid) are suited very well for this application, which also have an added advantage of high side lobe attenuation, which can help achieve the vertical antenna isolation requirements for the site.

Pilot pollution can be made worse if the repeater donor antenna is not narrow enough and localized to the desired donor base station sector. Since the repeater repeats the entire CDMA carrier (signal plus noise), it is important that the repeater location be line-of-sight to the donor BTS with a dominant PN. It is highly recommended to choose a repeater application that will allow a line-of-sight (LOS) path with a clear Fresnel zone (ideally with 60% of the first Fresnel clearance) between the repeater and the donor BTS. A LOS path will ensure a highly reliable repeater link, which can utilize a smaller fade margin. If a LOS path is not possible, then a path loss measurement is required to estimate the mean path loss of the donor link.

Since a LOS path which isolates a single donor BTS is important, donor antenna alignment is also very critical to the installation of a repeater site. A mis-aligned highly directional donor antenna can also create significant performance issues with the operation of a repeater site.

4.5.2.2.2 Repeater Subscriber Antenna

The subscriber antenna should be chosen (i.e. gain, H/V beamwidth, etc.) to cover the desired area. For over-the-air repeater applications, it is typically recommended to use an antenna with 105° of horizontal beamwidth or less, due to isolation/interference concerns and the unreliability of the beam patterns. It would be very difficult to achieve the antenna isolation requirements using an omni subscriber antenna with an over-the-air repeater application and as such, they are not recommended. On the other hand, micro-wave and fiber linked repeaters do not have the same isolation requirements as the over-the-air repeaters. Thus, the horizontal beamwidth restrictions do not apply towards the micro-wave/fiber linked repeater applications.

For those repeaters which have a diversity receive path capability, two subscriber antennas will be required. The same subscriber antenna restrictions mentioned above would apply for over-the-air diversity receive repeaters. As an alternative, a dual polarized slant 45° antenna may be a logical choice for diversity receive repeaters. Dual pole antennas (see Chapter 7) with the desired horizontal and vertical beamwidths have an advantage of providing two separate antennas in a single housing.


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4.5.2.3 Repeater Gain Settings

The repeater gain settings are a critical component to the successful installation and performance of the repeater. Setting the gain too high for the repeater’s Tx path to the subscriber could cause the repeater Tx PA to be over driven under a loaded condition. Although this may not be a major concern if the repeater PA is designed with gain compression, a significant amount of intermodulation (IM) distortion and spectral regrowth may be generated, which can impact the spectral purity (rho) of the CDMA signal beyond acceptable levels.

Setting the repeater’s Rx path back to the donor BTS too high could cause the BTS receiver to desense. To ensure that the repeater does not desense the donor BTS in a normal application (i.e. the repeater is NOT being used for maximum range extension), the repeater vendors typically recommend that the repeater Rx gain back to the BTS should be set lower (up to 10 dB) than the repeater Tx gain to the subscriber.

It is important to set the repeater gain levels for the Rx & Tx paths properly. Figure 4-25 below shows the potential effects of reducing the range of a donor BTS if the gain settings are not set properly.

Figure 4-25: Potential Range Reduction Due to Repeaters

With the assumption stated in the chart, the Y axis in the figure above represents the difference in repeater forward Tx power relative to the BTS power plus the difference in the repeater forward Tx gain relative to the repeater reverse Rx gain. Table 4-16 provides an example of how to calculate the relative Tx & Rx link difference.

BTS RX Range1, 2, or 4 Repeaters

3.26 RF prop loss

-20.0

-15.0

-10.0

- 5.0

0.0

0.00 0.10 0.20 0.30 0.40 0.50 0.60 0.70 0.80 0.90 1.00

Normalized RX Cell coverage referenced to BTS noise figure

Re

lati

ve

Tx

& R

x L

ink

Dif

fere

nc

es

1 Repeater

2 Repeaters

4 Repeaters

distance BTS with 1 Repeater

BTS

Path LossMobile to BTS

Link Loss = Path Loss + Cable Loss + Antenna Gain+ Repeater GainAssumption

Reverse Link Repeater to BTS = Forward Link BTS to Repeater


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Table 4-16: Relative Tx & Rx Link Difference Example

With this example, the donor BTS’s normalized Rx cell coverage at a -10dB relative Tx & Rx link difference is ~96% of the BTS’s coverage area without the repeater (i.e. the repeater reduced the coverage area by ~4%). Typical settings of the relative Tx & Rx link differences are -15 dB or better which will cause little to no effect on the normal coverage area of the donor BTS.

4.5.3 CDMA Repeater Optimization Considerations

This section discusses some of the optimization considerations around repeater applications.

4.5.3.1 Timing Impacts

The following sections provide some optimization considerations regarding the timing impacts of adding a repeater to a system.

4.5.3.1.1 Search Windows and Parameters

One of the main optimization considerations for the deployment of a repeater is the adjustment of the network parameters associated with search windows and timing. Since the repeater unit itself will add approximately 5-8 micro-seconds (μs) of delay (typically around 6 μs) in both the forward and reverse links, certain timing related parameters need to be expanded in order to handle this extra timing delay. There are four basic timing related considerations for repeaters.

• Access Channel Search Window (Cell Radius - PamSz & AchPamWinSz)• Traffic Channel Search Window (TchAcqWinSz)• Subscriber Search Windows (SrchWinA, SrchWinN, SrchWinR)• PN Offset Interference Protection (Pilot_Inc)

Access Channel Search Window. The access channel search window establishes the maximum round trip propagation delay that the BTS will search for subscriber origination attempts. In effect, it establishes the maximum radius that the BTS will be able to receive an origination attempt. Since a repeater not only increases the radius (distance) of the donor BTS, it also adds delay to the signal which is similar to adding propagation delay. The added delay can be translated back to distance. Thus, the access channel search window of the donor BTS needs to be expanded to compensate for the added distance (repeater coverage plus repeater delay) that the repeater provides. For the Nokia

BTS Tx Pilot Power 30 dBm a

Repeater Tx Pilot Power 25 dBm b

Repeater Tx Path Gain 70 dB c

Repeater Rx Path Gain 65 dB d

Relative Tx & Rx Link Differences -10 dB (b-a) + (d-c)


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Siemens Networks infrastructure, the access channel search window is set by adjusting the Cell Radius parameter (which automatically adjusts the PamSz & AchPamWinSz parameters). Adjustments to the Cell Radius parameter can be calculated as follows:

Cell Radius = Donor BTS Range (km) + Repeater Delay (μs) * 0.299 (km/μs) + Repeater Range (km) [EQ 4-66]

Traffic Channel Search Window. For the Nokia Siemens Networks infrastructure, the traffic channel search window is set by the TchAcqWinSz parameter. This parameter defines the traffic channel acquisition in PN chips, which is used during the handover acquisition of a call. For normal applications (including repeater applications), it should be set at least as large as the AchPamWinSz parameter (which is established by the Cell Radius parameter).

Subscriber Search Windows. The subscriber search window parameters are SrchWinA, SrchWinN, and SrchWinR. SrchWinA is the active/candidate pilot set search window size which should be made large enough to incorporate ~95% of the expected delay spread energy. Since a repeater has an internal delay of 5-8 μs and a subscriber will find itself in places where the BTS and repeater signals are both strong enough to demodulate, a repeater will normally increase the effective delay spread of the donor BTS. The default setting for SrchWinA is 5 which corresponds to 20 PN chips (16 μs or +8 μs from the earliest arriving “usable” delay spread component). The default setting may be adequate for some repeater applications. An evaluation of the specific repeater application is necessary to determine if the SrchWinA parameter for the donor BTS needs to be increased.

The SrchWinN and SrchWinR parameters represent the search window sizes associated with the Neighbor Set and Remaining Set pilots. The size should be made large enough to account for differential time delay between the subscriber and a potential handoff BTS given in the subscriber’s neighbor list. The worst case differential delay would be a scenario where the subscriber is next to a serving site and the subscriber attempts to handoff to a distant site. Since a repeater can increase the differential delay, increasing the SrchWinN and SrchWinR parameters may be necessary for some repeater applications. It is important to note that handoff relationships are symmetrical and reciprocal for the neighboring cells which are candidates for the donor sector. Thus, the SrchWinN and SrchWinR parameters will need to be adjusted for both the donor BTS and the neighbor cells to the donor BTS.

PN Offset Interference Protection. Some level of PN Offset interference protection is provided with the Pilot_Inc parameter. An increase in the Pilot_Inc increases the separation between adjacent PN offset pilots which provides improved adjacent offset interference protection. The increased separation between adjacent PN offsets also reduces the total number of valid PN offsets. A Pilot_Inc of 2 will decrease the total number of valid PN offsets from 512 to 256. Since the cell radius (or propagation delay) is a factor to consider when selecting the appropriate Pilot_Inc setting, adding repeaters to a system may require a re-evaluation of the Pilot_Inc setting. In most cases, adjustments to the Pilot_Inc parameter due to repeater applications will not be necessary, if proper PN offset planning is performed. In some cases, a re-evaluation of the Pilot_Inc setting may be necessary and an adjustment to the setting may be required.

For more detailed information on PN offset planning and search window parameters please refer to Chapter 5.


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4.5.3.1.2 Location Based Services

Another timing related issue to consider is that some implementations of location based services may be affected by the use of repeaters. For a fixed network equipment based solution, Time Difference of Arrival (TDA) measurements are made which will now include both repeater and propagation delays. The repeater delay will add variance to the TDA measurements and may make it difficult to achieve accurate location calculations. There is also a handset based GPS solution which still requires some coordination with the fixed network equipment. Both of these location based service implementations may require some sort of custom solution in order to make the location based feature accurate for repeater applications.

4.5.3.2 Optimization Considerations

Once the repeater site has been fully designed, installed, and verified (i.e. repeater gain settings verification, donor BTS-to-repeater link verification, antenna isolation verification, etc.), the next step is to conduct drive test optimization. After the timing related parameters have been evaluated and adjusted appropriately, there are six drive test areas that need to be analyzed.

• Donor BTS coverage area• Repeater coverage area• Donor BTS to repeater transition zone coverage area• Donor BTS to adjacent cell handoff zones• Repeater to adjacent cell handoff zones• Donor BTS to repeater transition zone to adjacent cell handoff zones

Most of the same basic drive test data collection and optimization techniques used for a normal BTS can also be applied towards a repeater site. Although, the added complexity and functionality of a repeater should be taken in account during the troubleshooting of any performance issues that are identified through the drive test optimization process. Since one PN offset will be transmitted from two separate antennas at two different locations, the optimization engineer needs to be familiar with the donor BTS and repeater antenna configurations, in order to optimize the coverage of the one PN offset.

Since the repeater repeats the entire CDMA carrier (signal plus noise), it is important that the repeater location be line-of-sight to the donor BTS with a dominant PN. Pilot pollution can be made worse if the repeater donor antenna is not narrow enough and localized to the desired donor BTS sector. A repeater deployment should create a dominant pilot area and improve the pilot signal strength coverage.

4.5.4 CDMA Repeater Maintenance Considerations

This section discusses some of the maintenance considerations around repeater applications.


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4.5.4.1 Future Expansion Considerations

As the capacity of a system grows over time, a natural progression is to deploy an additional CDMA carrier to the system. Prior to deploying a repeater for a specific repeater application, the long term expansion planning of the repeater site should be considered. The following sections provide information about two future expansion considerations.

• Multiple Repeater Expansion• Repeater to BTS Conversion

4.5.4.1.1 Multiple Repeater Expansion

The expansion design of a multiple carrier repeater system becomes more complex. Duplication of repeater hardware & installation is required with each additional carrier added to the donor BTS. If a new carrier is added to an area where repeaters are deployed, re-engineering of the repeater site is required to accommodate a multiple repeater configuration. Below are a few design issues to consider when looking at multiple carrier repeater sites.

• Antenna sharing configuration (splitters, combiners, duplexers, etc.)• Separate antennas

If the additional repeater is required to share the antennas of the existing repeater, the antenna sharing combining/splitting/filtering losses for the new antenna configuration will need to be evaluated. Adjustments to the repeater design may be required to overcome the additional combining/splitting/filtering losses of the new antenna sharing configuration. If the additional repeater requires separate antennas, an evaluation of the interference and antenna isolation is still required. For either antenna configuration (antenna sharing or separate antennas), a re-evaluation of the following is required.

• Re-evaluation of interference for additional filtering• Re-evaluation of repeater gain settings• Re-evaluation of repeater antenna isolation requirements• Re-evaluation of donor BTS-to-repeater link engineering

Once the new antenna configuration has been designed and implemented, the new repeater configuration should be reverified (i.e. repeater gain settings verification, donor BTS-to-repeater link verification, antenna isolation verification, etc.). The long term planning and design of a repeater application (i.e. multiple repeaters for multiple carrier support) should be considered during the initial design and deployment of a specific repeater site.

4.5.4.1.2 Repeater to BTS Conversion

Typically, a new carrier is added to expand the capacity of the system. A repeater does not provide any capacity benefit to the system (it only provides expanded coverage). If a new carrier is added to an area where repeaters are deployed, it may make sense to convert the repeater to a regular capacity bearing cell site.


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A significant amount of cell site design, installation, and optimization work at the repeater site is necessary to convert the repeater site to a capacity bearing cell site. All of the initial work in the repeater design, installation, and optimization including RF propagation modeling, antenna isolation measurements, custom shielding, linked network equipment installation, donor antenna alignment, subscriber antenna adjustments, repeater gain settings and verifications, parameter settings, and drive test optimization, will not apply to the capacity bearing cell site. Most of the cell site design, installation, and optimization work required to deploy a new cell site into a system is also required to convert a repeater site to a regular cell site. The long term planning and design of a repeater application (i.e. repeater to BTS conversion) should include a cost analysis of the repeater site which incorporates the cost of all of the rework to convert the repeater to a capacity bearing site.

4.5.4.2 Environmental Changes

Future changes in the environmental conditions surrounding an over-the-air repeater site can have an impact on the performance of the repeater. Changes in the surrounding environment (i.e. changes in the ground clutter such as new buildings, changes to landscaping, seasonal changes to the surrounding foliage, etc.) can have a negative impact on the donor BTS-to-repeater link performance. It may also have a negative impact on the donor-to-subscriber antenna isolation. Both of these conditions can affect the performance of an over-the-air repeater.

4.5.4.3 Operations and Maintenance Considerations

The Operations and Maintenance (O&M) of a repeater network will be different than that of a BTS network. The hardware, software, monitoring access (POTS line w/modem, wireless modem, etc.), configuration management, and alarm monitoring O&M practices and procedures for a repeater network will be different and will require specialized knowledge and skill sets. Different resources or additional training will be required to properly plan, design, install, operate, and maintain a repeater system.

System Capacity planning becomes more complicated with repeaters. Since repeaters connected to one sector will cover more area than sectors without repeaters, the site’s capacity limit will be reached more quickly due to the additional area the sector with the repeater is covering. This may cause a highly imbalanced system where one sector is lightly loaded while another sector is heavily loaded. To overcome capacity loaded donor sectors, a new carrier can be added, the repeater can be replaced with a new cell site, or the repeater can be moved to a lightly loaded donor sector.

4.6 Extended Range Cells

For most IS-95/IS-2000 CDMA systems, the maximum cell site range that is achievable due to limitations within the CDMA technology is approximately 56 kilometers. This technology based range limit is usually not a factor for most applications, but there are some scenarios where exceeding this technology limit is desired. Since the CDMA technology is based upon time synchronization, the technology limitation can be overcome through the development of special time based searching and synchronization techniques. The Extended Range Cell feature within a Nokia Siemens Networks system implements special algorithms and techniques in order to expand


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the maximum cell site range beyond the ~56 km technology limit.

There are two key functional capabilities provided with the Nokia Siemens Networks’ Extended Range Cell feature.

• System support for a maximum cell radius setting of up to 180 km for IS-2000 1X voice calls and up to 144 km for IS-95 voice calls

• Pilot PN Offset Diversification functionality, which allows Pilot PN Offset Increments to be defined at the Sector-Carrier level (this functionality can be applied towards extended range and/or normal range cell sites)

There are many different aspects that need to be considered during the RF planning and design phase for an Extended Range cell site. The following sections discuss some of the major concerns that need to be considered in the RF planning, design, installation and optimization for Extended Range cell sites.

4.6.1 Extended Range Cell RF Planning and Design

The RF planning and design for Extended Range cell sites (i.e. >56 km range) requires some unique considerations as opposed to normal range cell sites. This section discusses the following topics as it relates to Extended Range cell site RF planning and design.

• Tower Top Amplifiers• Reverse Link Budget Analysis• Forward Link Budget Analysis• Extended Range Cell Site Design Limitations• Site Selection Criteria for Extended Range Cells

4.6.1.1 Tower Top Amplifiers

For most Extended Range cell site applications, the limited subscriber power will most likely cause the reverse link to be the limiting factor from an RF link perspective. In order to improve the performance of the reverse link, tower top amplifiers (TTAs) may need to be considered. If the specific design of an Extended Range cell site has a significant amount of attenuation between the Receive antenna and the input to the base transceiver station (BTS), then a TTA should be considered.

There are two performance improvements for the reverse link associated with the implementation of TTAs. The first performance improvement significantly reduces the receive path transmission losses between the antenna and the BTS. The second performance improvement associated with TTAs is a reduction of the cascaded noise figure for the combined TTA/BTS system which results in a better receive sensitivity. Section 4.2.3.2.1 provides a detailed example of the calculations utilized to determine the TTA performance improvements. For this detailed example, Table 4-5 shows a 3.0 dB improvement due to a reduction in transmission path attenuation and a 1.27 dB improvement due to a reduction in the cascaded Noise Figure. The overall benefit provides a 4.27 dB improvement that can be applied directly to the link budget for the reverse link.


4


The reverse link performance improvements for TTAs illustrate the positive aspects of using TTAs, but there are also some issues and concerns with TTA usage. The TTA performance improvements will come at the expense of increased susceptibility to reverse link intermodulation interference. A standard BTS is designed to meet a minimum performance criteria for the 3rd order input intercept point (IIP3). Adding a low noise amplifier to the receive path of a standard BTS will degrade the equivalent IIP3 performance of the site. In order to minimize the IIP3 degradation to the cell site, the total gain from the TTA input to the BTS input should be limited to a maximum of 6 dB. Minimizing the total gain will minimize the IIP3 degradation. An example of the total gain from the TTA input to the BTS input would be from Point B to Point A as shown in Figure 4-7. (Note: The total gain for the example in Figure 4-7 was 12 - 3 - 0.5 = 8.5 dB, which is greater than the 6 dB maximum recommendation. A special BTS with a higher Noise Figure that was designed to minimize the IIP3 performance impact was used for this example. Thus, the 6 dB maximum total gain recommendation would not apply.) The selection of a TTA should take into account the Gain, NF, and IIP3 specifications of the specific TTA model. An analysis of the cascaded Gain, NF, and IIP3 performance impact of the TTA and BTS system should be performed. The performance specifications of the TTA should attempt to maximize the reverse link performance improvements for the combined TTA/BTS system while trying to minimize the IIP3 degradation to an acceptable level.

Adding TTAs to a cell site also places active electronics at the top of the antenna tower. The remote location of the TTA adds some complexity to the installation and maintenance of the TTA. Special filtering, lightning protection, grounding/bonding, and/or shielding requirements may be necessary during the TTA installation. Please refer to the specific TTA vendor’s installation requirements for more details. Maintenance and/or replacement of the TTA is more difficult due to its installation at the top of the tower. All of these factors need to be considered prior to designing a cell site with TTAs.

4.6.1.2 Reverse Link Budget Analysis

Being able to establish and maintain an RF link for an Extended Range cell site is one of the more difficult design objectives. Since the reverse link will most likely be the limiting RF link for an Extended Range cell site, performing a link budget analysis for the reverse link is an important step in the design process. There are two key design configurations that should be considered for all Extended Range cell site designs. The use of TTAs (as discussed in Section 4.6.1.1) and the use of narrow beamwidth high gain antennas should both be considered. Table 4-18 provides an example reverse RF link budget which utilizes both Tower Top Amplifiers and high gain antennas.

Reverse RF Link Budget Assumptions used for Table 4-18• 9.6 kbps Traffic Channel• Antenna Gain = 17 dBd• TTA Specifications: Gain = 9.0 dB (7.943 linear), NF = 2.5 dB (1.778 linear)• Antenna to TTA Jumper Gain = -0.5 dB• TTA output to BTS input Gain = -6.0 dB (0.251 linear), NF = 6 dB (3.981 linear)• BTS NF = 6 dB (3.981 linear)• Total Gain (TTA input to the BTS input) = 9.0 - 6.0 = 3.0 dB


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Cascaded Noise Figure Calculation using [EQ 4-21]:

Table 4-17: TTA Impacts to Reverse Link Budget Inputs:

1. Noise Figure reference point at the input of the TTA2. Noise Figure reference point at the input of the BTS

Table 4-18: Extended Range Reverse RF Link Budget with TTA Example

Note: 1. The shadow fade margin assumes the effects of soft handoff and multiple cells.Where:

Sensitivity and path loss are calculated as follows:

Sb = kTB + Nfb + E - PG

Parameter With TTA Without TTALine Loss 0.5 dB 6.5 dBBase Noise Figure 5.6 dB1 6.0 dB2

Parameter Unit Reference NoTTA

WithTTA

Subscriber Unit Tx Power dBm Pp Section 4.2.3.1.1 23 23

Subscriber Unit Tx Feeder Loss dB Lfp Section 4.2.1.5 0 0

Subscriber Unit Antenna Gain dBd Gp Section 4.2.1.6 -2.1 -2.1

Body Loss dB Hm Section 4.2.1.3 2 2

Vehicle Loss dB Vm Section 4.2.1.2 7 7

Building Loss dB Bm Section 4.2.1.1 0 0

Base Antenna Gain dBd Gb Section 4.2.1.6 17.0 17.0

Line Loss dB Lfb Section 4.2.1.5 6.5 0.5

kTB dBm kTB Section 4.2.3.2.1 -113.1 -113.1

Base Noise Figure(Cascaded NF for TTA example)

dB Nfb Section 4.2.3.2.1 6 5.6

Eb/No dB E Section 4.2.2.3 5.6 5.6

Processing Gain dB PG Section 4.2.3.2 21.1 21.1

Base Rx Sensitivity dBm Sb Section 4.2.3.2 -122.6 -123.0

Interference Margin dB Im Section 4.2.2.1 3 3

Ambient Noise Rise dB Tm Section 4.2.1.4 0 0

Shadow Fade Margin(Note: 1)


5.6 5.6

Max. Allowable Path Loss dB Lp 136.4 142.8

Isotropic Path Loss dB Li 140.7 147.1

NFdB 10 1.7783.981 1–

7.943---------------------- 3.981 1–

7.943 0.251×---------------------------------+ +

log 5.6= =


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Lp = Pp - Lfp + Gp + Gb - Lfb - Sb - Im - Tm - Hm - Vm - Bm - FmLi = Lp + (2 * 2.14)

Table 4-18 provides a side-by-side link budget comparison of a site with and without a TTA installed at the site. The example utilized a line loss of 6.5 dB, which makes it a good candidate for using TTAs. As shown in the link budget analysis, the TTA provided a 6.0 dB improvement in line loss and a 0.4 dB improvement in Base Noise Figure for an overall path loss improvement of 6.4 dB. This example had a total gain from the TTA input to the BTS input of 3.0 dB, which is well within the 6.0 dB guideline to help minimize the IIP3 degradation.

Below is a comparison of the differences between an Extended Range Cell (see Table 4-18) and a normal range cell (see Table 4-7) link budget. From this comparison of typical link budgets, the Extended Range cell has a 5.4 dB advantage over the normal range cell.

4.6.1.3 Forward Link Budget Analysis

Once the link budget for the reverse link has been estimated for the Extended Range cell, the next step is to estimate the link budget for forward link. Since the maximum base station transmit power for the forward link is adjustable, one of the goals for the link budget analysis is to estimate the forward link power requirements to balance the forward and reverse links. Therefore, all of the link budget parameters are entered into the analysis and then the Base Tx Power is adjusted until the maximum allowable path loss for the forward link matches that of the reverse link. Table 4-20 provides an example of a forward link budget example for an Extended Range cell.

Forward RF Link Budget Assumptions used for Table 4-20• Antenna Gain = 17 dBd• Base Line Loss (TTA example) = 6.5 dB• Subscriber NF = 10 dB• Subscriber Eb/No = 7.0 dB

Table 4-19: Extended Range vs. Normal Range Link Budget Comparison

Parameter Normal Range w/o TTAs

Extended Range with TTAs

Extended Range Gain Improvement

Base Antenna Gain (dBd) 14.5 17.0 2.5Line Loss (dB) 3 0.5 2.5Base Noise Figure (dB) 6 5.6 0.4


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Table 4-20: Extended Range Forward RF Link Budget Example

Note: 1. The shadow fade margin assumes the effects of soft handoff and multiple cells.Where:

Sensitivity and path loss are calculated as follows:

Sp = kTB + Nfp + E - PGLp = Pb - Lfp + Gp + Gb - Lfb - Sp - Im - Tm - Hm - Vm - Bm - FmLi = Lp + (2 * 2.14)

From the link budget example shown in Table 4-20, the forward link traffic channel would require about 34.8 dBm or ~3.0 Watts of power to balance the forward and reverse links. If we assume that the pilot and traffic channel gain settings are set to the same value (i.e. 127), then the pilot power setting for this example would also be set at 34.8 dBm or 3.02 Watts. If the pilot and traffic channel gain settings are set to 127 and 110 respectively, then the pilot power setting for this example would be set at 36.0 dBm or ~4.0 Watts (1272 / 1102 * 3.0 = 4.0 Watts).

Parameter Unit Reference ExtendedRange

Base Tx Power dBm Pb Section 4.4.1 34.8

Base Line Loss dB Lfb Section 4.2.1.5 6.5

Base Antenna Gain dBd Gb Section 4.2.1.6 17.0

Body Loss dB Hm Section 4.2.1.3 2

Vehicle Loss dB Vm Section 4.2.1.2 7

Building Loss dB Bm Section 4.2.1.1 0

Subscriber Antenna Gain dBd Gp Section 4.2.1.6 -2.1

Subscriber Line Loss dB Lfp Section 4.2.1.5 0

kTB dBm kTB Section 4.2.3.2.1 -113.1

Subscriber Noise Figure dB Nfp Section 4.2.3.2.2 10.0

Subscriber Eb/No dB E Section 4.2.2.3 7.0

Processing Gain dB PG Section 4.2.3.2 21.1

Subscriber Rx Sensitivity dBm Sp Section 4.2.3.2 -117.2

Interference Margin dB Im Section 4.2.2.1 3

Ambient Noise Rise dB Tm Section 4.2.1.4 0

Shadow Fade Margin(Note: 1)


5.6

Max. Allowable Path Loss dB Lp 142.8

Isotropic Path Loss dB Li 147.1


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4.6.1.4 Extended Range Cell Site Design Limitations

There are many different challenges that need to be addressed when trying to design a cell site with a range that extends beyond 56 km. The following sections discuss many of the design limitations that need to be considered during an Extended Range cell site design process.

4.6.1.4.1 BTS PA Limitations

For some Extended Range cell site examples, the forward link traffic channel power requirements could be as high as 10 Watts. Once the power requirements have been estimated for the forward link traffic channel, an analysis of the PA capacity requirements for the specific BTS should be performed. For Extended Range cell sites, the maximum number of users supported by the site may be limited by the PA capacity. For those cases that have a high forward link traffic channel requirement, it may be necessary to trade out capacity for coverage. For those sites, it is highly recommended that the LPA Overload Protection feature be implemented. Since the typical design criteria for Extended Range cell sites is for rural areas with low capacity requirements, a capacity for coverage trade-off is normal. Figure 4-26 provides an analysis example of the Available PA Power versus Pilot/TCH Power. The assumptions associated with this analysis are provided below.

Assumptions for the Available PA Power vs. Pilot/TCH Power Analysis shown in Figure 4-26• Maximum PA power per carrier = 90 Watts (assumes an SC4812T-Lite)• Maximum traffic channel power = 100% of Pilot• Paging channel power = 75% of Pilot• Sync channel power = 10% of Pilot• Available PA Power provides the maximum trunked power per sector assuming both an

even and uneven load across all of the sectors

Figure 4-26: Available PA Power vs. Pilot/TCH Power

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The results in Figure 4-26 show that a 3 sector site with a 10 Watt Pilot and even loading across all sectors, only has 11.5 Watts of available PA power per sector. Thus, a one or two sectored site may be necessary to provide adequate available PA power for those cell sites that require high Pilot/TCH power requirements. An analysis of the PA requirements and limitations should be performed for all Extended Range cell sites. As stated before, it is highly recommended that the LPA Overload Protection feature be implemented for those sites that have high Pilot/TCH power settings and may not have adequate PA power available for the intended users.

4.6.1.4.2 Earth’s Horizon Limitations

One of the major obstacles to overcome when designing an Extended Range cell site is the curvature of the earth. For most Extended Range cell site designs, a near line-of-sight path between the cell site antenna and the subscriber unit may be necessary to overcome the earth’s horizon limitation and to help support an open or free space path loss model. As a result, the height of the antenna with respect to the desired area of coverage becomes a critical factor. For example, a cell site will typically need an antenna height that is approximately 245 meters above the ground to achieve a line-of-site path of up to 56 km. Thus, an Extended Range cell site which is greater than 56 km would need an antenna height that is greater than 245 meters above the ground level. Figure 4-27 displays a diagram that can be used to calculate the required antenna height as shown in [EQ 4-67] to achieve a line-of-sight path which takes into account the curvature of the earth. Figure 4-28 provides a plot of the line-of-sight distance to the horizon versus antenna height.

Figure 4-27: Antenna Height Calculation Diagram

Antenna Height = h = ((r / cos (360o * d/c)) - r) * 1000 [EQ 4-67]Where:

r = Radius of the Earth = 6,378.16 kmc = Circumference of the Earth = 40,075.16 kmd = Distance from antenna to subscriber (in km)h = Height of antenna above the earth (in meters)


4


Figure 4-28: Horizon vs. Antenna Height

As shown in Figure 4-28, a typical antenna height of 100 meters above the ground produces a line-of-sight path of 35.7 km and a 120 km cell site would require the antenna height to be about 1130 meters above ground level. It is important to note that the antenna height requirements due to the earth’s horizon can also be aided by the elevation of the terrain surrounding the site. A hilltop or mountaintop cell site with a fairly steep decline in elevation can take advantage of the ground elevation at the cell site to help overcome the earth’s horizon antenna height requirements (see Figure 4-34). As a result, the difference in ground elevations between the cell site and the edges of the coverage area may also need to be considered. Due to the earth’s horizon limitations, the antenna height above ground level and a line-of-sight path are some of the critical factors that need to be considered during the design phase of an Extended Range cell site.

4.6.1.4.3 RF Propagation Limitations

As stated in the previous section, a line-of-sight path between the cell site antenna and the subscriber unit may be necessary to overcome the earth’s horizon limitations, but it may also be required to help support an open or free space RF propagation path loss model. In order to achieve RF propagation over very long distances, the RF environment for the Extended Range cell site coverage area will most likely need to approximate the open or free space path loss models.

The next step in validating the Extended Range cell site design is to verify the RF environment for the desired coverage area and determine if the propagation model path loss calculations associated with the RF environment are less than the maximum path loss calculations provided by the link budget estimates. The Free Space, Hata, and COST-231-Hata propagation model calculations provided in Section 4.3.1, Section 4.3.2, and Section 4.3.3 can be used for this step. The Hata propagation model will typically be used for applications that are less than 1000 MHz and the

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Horizon = 35.7 km

Antenna Height = 1130 m

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COST-231-Hata propagation model will typically be used for applications that are between 1500 MHz and 2000 MHz. Figure 4-29 provides a chart of the suburban, rural, and open path loss estimates versus cell radius using COST-231-Hata (@1910 MHz) and Hata (@880 MHz) propagation models with a base antenna height of 1130m.

Figure 4-29: Propagation Path Loss for Antenna Height of 1130m

The results shown in Figure 4-29 show that the COST-231-Hata at 1910 MHz and Hata at 880 MHz using Open and Rural path loss areas with a 1130m antenna height and a horizon distance of 120 km is less than the maximum path loss calculations for the Extended Range link budget estimates from Section 4.6.1.2 and Section 4.6.1.3. The results also show that the Hata at 880 MHz using the Suburban path loss area will be close to meeting the maximum path loss requirements from the link budget estimates. This example shows that selecting an antenna height which satisfies the earth’s horizon limitation has also satisfied the Open and Rural maximum path loss criteria.

Since the Free Space path loss model is based upon a line-of-sight path, the calculation does not need a base and subscriber antenna height. Figure 4-30 provides a chart of the Free Space path loss estimates versus cell radius using an 880 & 1910 MHz transmission frequency.

Path Loss vs Cell Radius(Base Antenna Height = 1130m, Subscriber Antenna Height = 1.5m)

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Horizon = 120 km


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Figure 4-30: Free Space Path Loss versus Cell Radius

The Free Space path loss model will always provide the most optimistic cell radius results. RF measurements of the desired coverage area may be necessary to characterize the RF environment in order to help verify the appropriate RF propagation model to use and the maximum path loss requirements for the area of interest.

4.6.1.4.4 Feature Constraints and Limitations

During the development of the Extended Range Cell feature, several constraints and limitations were implemented as part of the feature. This section provides some of those constraints and limitations that need to be considered during the design phase of an Extended Range cell site.

The following is a list of hardware related constraints that apply to the Extended Range Cell feature.

• The extended cell radius functionality of the Extended Range Cell feature will initially be supported by the MCC 1X, BBX 1X, and JBBX 1X cards. Older models of these cards will not support this functionality.

• The extended cell radius functionality of the Extended Range Cell feature is not supported for EVDO carriers

• The Pilot PN Offset Diversification functionality of the Extended Range Cell feature will initially be supported by the MCC 24 and MCC 1X cards. Older BTS models do not support this functionality.

• The Extended Range Cell feature will initially be supported on the following non-logical BTS frame types with 3 sectors or less: SC4812 series, SC480, SC2440, SC4840, and SC7224. Older BTS models do not support this functionality.

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The maximum cell radius has certain limitations based upon different combinations of frame type, MCC type, air interface, and the status of the Interference Cancellation & Extended Range Cell features. Table 4-21, Table 4-22, Table 4-23, and Table 4-24 provide the Extended Range Cell radius constraints and limitations for the supported SC4812, SC480, SC2440, SC4840, and SC7224 BTS types.

Table 4-21: BTS SC4812 Extended Range Cell Feature Limitations

MCCType

AirInterface

# ofSectors

# ofCarriers

ExtendedRange

Feature

CellRadius

Supported

MaximumTchAcqWinSz

MCC 8/24 IS-95 < 6 < 4 Disabled < 56.0 km 472

MCC 1X IS-95/1X < 6 < 4 Disabled < 56.0 km 472

MCC 1X IS-95/1X < 3 < 4 Enabled < 120.0 km 996

MCC 1X IS-95/1X < 3 < 2 Enabled < 144.0 km 1193

MCC 1X Only 1XRC 3&4

< 3 < 2 Enabled < 180.0 km 1488


MCCType

AirInterface

# ofSectors

# ofCarriers

ExtendedRange

Feature

CellRadius

Supported

MaximumTchAcqWinSz

MCC 8/24 IS-95 < 2 < 2 Disabled < 56.0 km 472

MCC 1X IS-95/1X < 2 < 2 Disabled < 56.0 km 472

MCC 1X IS-95/1X 1 < 2 Enabled < 144.0 km 1193


1 < 2 Enabled < 180.0 km 1488


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Table 4-23: BTS SC2440/SC4840 Extended Range Cell Feature Limitations

MCCType

AirInterface

# ofSectors

# ofCarriers

InterferenceCancellation

Mode

ExtendedRange

Feature

CellRadius

Supported

MaximumTchAcqWinSz

MCC 8/24 IS-95 < 6 < 4 Disabled Disabled < 56.0 km 472

MCC 8/24 IS-95 < 6 < 4 Enabled Disabled < 36.0 km 308

MCC 1X IS-95/1X < 6 < 4 Disabled Disabled < 56.0 km 472

MCC 1X IS-95/1X < 6 < 4 Enabled Disabled < 36.0 km 308

MCC 1X IS-95/1X < 3 < 4 Disabled Enabled < 120.0 km 996

MCC 1X IS-95/1X < 3 < 4 Enabled Enabled < 100.0 km 832




< 3 < 2 Disabled Enabled < 180.0 km 1488


< 3 < 2 Enabled Enabled < 180.0 km 1488


MCCType

AirInterface

# ofSectors

# ofCarriers

InterferenceCancellation

Mode

ExtendedRange

Feature

CellRadius

Supported

MaximumTchAcqWinSz

MCC 1X IS-95/1X < 3 < 4 Disabled Disabled < 56.0 km 472

MCC 1X IS-95/1X < 3 < 4 Enabled Disabled < 36.0 km 308






< 3 < 2 Disabled Enabled < 180.0 km 1488


< 3 < 2 Enabled Enabled < 180.0 km 1488


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4.6.1.4.5 Handoff Limitations

An important aspect that needs to be considered during the design phase of an Extended Range cell site are the handoff limitations to its neighbor cell sites. In order for Extended Range cell sites to handoff to normal range cell sites, the normal range cell site will need to have the Extended Range Cell feature enabled in order to interpret the distant PNs correctly (see Chapter 5 Section 5.3.4 for more details). Another issue that exists with Extended Range handoffs is the subscriber window size limitations. Section 5.3.4 from Chapter 5 provides a general explanation of the mobile search windows and how the search window settings correlate to cell site geography and distance. The range of search window size settings available for SrchWinN is more than adequate for normal range cell sites. For Extended Range cell sites, the maximum SrchWinN window size setting of 15 provides some soft handoff design limitations that need to be considered. The maximum SrchWinN setting creates a subscriber search window of +/- 226 chips which correlates to approximately +/- 55 km. Figure 4-31 Figure 4-32, and Figure 4-33 provide a simple example of how the SrchWinN window size can limit the soft handoff for an Extended Range cell site.

Figure 4-31: Handoff Limitation Example

For the example in Figure 4-31, a subscriber is driving along a linear path from an Extended Range site (Cell A) to a normal range site (Cell B). The subscriber is currently located 100 km from Cell A and 35 km from Cell B and the pilot signal strength from Cell B has just become high enough to be a candidate for the active set. For this example, a subscriber in the active set on Cell A and driving towards Cell B will never be able to perform a soft handoff with Cell B because the signal from Cell B will be outside the SrchWinN window size. Since Cell A & B are both GPS synchronized in time, a Chips/Distance scale (as shown in Figure 4-32) can be used to analyze the situation.

Cell BCell A

100 km 35 km


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Figure 4-32: Chips/Distance Scale for Handoff Limitation Example

As the subscriber approaches the handoff zone between Cell A & B, the subscriber’s time reference is based off of Cell A which is delayed by ~410 chips due to the 100 km distance. Since Cell B’s signal is only 35 km away, the chip delay is only ~143 chips. Since the chip differential between the two cell sites is 410-143=267 chips, which is larger than the maximum SrchWinN window size of 226 chips, the subscriber will never be able to receive the signal from Cell B because it will appear outside the subscriber’s search window. The same situation would occur if the subscriber was active on Cell B and driving towards Cell A.

The differential distance from the subscriber to each BTS trying to perform a soft handoff cannot exceed the 55 km (226 chip) SrchWinN window size limit, otherwise the subscriber will never see the other BTS's pilot. Thus, the soft handoff between an Extended Range cell and a normal range cell must occur before the differential distance between the subscriber and the two BTSs reach the 55 km window size limit. This SrchWinN window size limitation would also apply to soft handoffs between two Extended Range cell sites. Figure 4-33 shows the SrchWinN handoff zone for the previous example which is created as a result of the SrchWinN window size limitation.

Figure 4-33: SrchWinN Handoff Zone for Previous Example

The simplistic example provided in Figure 4-31, Figure 4-32, and Figure 4-33 analyzed a linear path between the two cell sites. Keep in mind that an RF coverage area handoff zone must exist between the two sites for soft handoffs to occur. The design of the RF coverage area handoff zone must also take into account the SrchWinN handoff limitation. Thus, the differential distance to each BTS should be checked for all of the areas that are designated as soft handoff zones.

In some cases, the forward link signal may be adequate to trigger a soft handoff within the handoff zone, but the reverse link may not be adequate enough to maintain the link which can cause a dropped call. If optimization efforts cannot alleviate the situation, PSMM filtering can be used to protect against dropping a leg with significantly better reverse FER. In another case, the forward

Cell A

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Chips0 100 200 300 400 500 600

0 24.4 48.8 73.2 97.6 122 146Distance (km)

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~226 Chips55 km

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55 km 40 km40 kmNo Handoffs No HandoffsHandoffs Allowed

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link pilot signal strength may not be high enough to trigger the soft handoff before the 55 km window size limit is reached. Both of these cases reflect the fact that a handoff zone with adequate coverage overlap must be part of an Extended Range cell site design in order to allow some margin for handoff optimization.

4.6.1.5 Site Selection Criteria for Extended Range Cells

As discussed in the previous sections, there are many different design criteria that need to be considered during the Extended Range cell site planning and design phase. One of the more critical aspects of the Extended Range cell site planning and design is the site selection criteria. There are a few unique site selection applications that tend to promote the previously mentioned Extended Range design criteria. This section will discuss some of the unique site selection design criteria and some unique cell site design applications.

One of the main Extended Range cell site design criteria is to maximize the antenna height in order to overcome the earth’s horizon limitation, which also promotes a line-of-sight open or free space path loss propagation model. For a 120 km Extended Range cell site, the antenna height would need to be approximately 1130 m above ground level. Achieving this type of antenna height is difficult unless the cell site is designed on top of a hill or mountain side. Thus, one of the unique site selection criteria is for a hill and/or mountain top application as shown in Figure 4-34.

Figure 4-34: Extended Range Cell Hill/Mountain Top Application

Since RF propagation travels a lot further over water as opposed to land, another unique site selection application is to design an Extended Range cell site along a marine or coastal border. Figure 4-35 provides an example of an Extended Range cell marine/coastal application.


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Figure 4-35: Extended Range Cell Marine/Coastal Application

Another unique Extended Range cell site application is a cascaded repeater design. Additional information on a cascaded repeater design is available in Section 4.5.1.2.3. The basic premise of a cascaded repeater design is to utilize very narrow beam high gain antennas and cascade multiple repeaters along an open rural highway coverage area with flat terrain (see Figure 4-36 for an example). Since a repeater adds some delay to the transmission back to the BTS, the total distance for the cell site radius setting needs to account for the total distance within the coverage area for cascaded repeater application and the total delay for all of the repeaters that are used within the application. The time delay for the repeater should be available from the repeater vendor and then the time delay can be converted to distance by multiplying by the speed of light (i.e. 299,792.458 km/second or 186,282.397 miles/second).

Figure 4-36: Extended Range Cell Cascaded Repeater Application

It is important to note that the usage of narrow horizontal and/or vertical beamwidth high gain antennas with highly elevated antenna heights can create significant nulls within the coverage area. In most cases, the Extended Range cell site will not be designed to provide ubiquitous coverage around the cell site (as is the case with normal range, urban or suburban cell sites). An Extended Range cell site will typically be designed to provide a unique coverage area application where

BTS Repeater #1Repeater #2BTS Repeater #1Repeater #2


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narrow beamwidth high gain antennas are appropriate. In some cases, the open rural flat terrain RF environment at the edge of coverage may have some small dips and valleys, which can actually create holes within the desired coverage area. A coverage area analysis through simulations and/or drive testing is recommended to verify that adequate coverage is provided in the desired areas for an Extended Range cell site design.

In summary, there are many different criteria that need to be considered during the site selection process for an Extended Range cell site design. The following is a list of some of the previously discussed criteria that influence the site selection process for Extended Range cell sites.

• Highly elevated antenna heights or mountain/hill top sites• Line-of-sight open or free space propagation• Open rural flat terrain• Propagation over water• Narrow beamwidth highway coverage areas

4.6.2 Extended Range Cell Optimization Considerations

For the most part, the same optimization activities utilized for normal range cell sites will also be used for Extended Range cell sites. Since repeaters extend the range of an existing cell site, some of the same optimization considerations that were mentioned for repeaters (see Section 4.5.3) would also apply for Extended Range cell sites. This section provides some of the unique optimization considerations that would apply to Extended Range cell sites.

4.6.2.1 PN Offset Planning

One of the main purposes of PN offset planning within a system is to eliminate the effects of adjacent offset and co-offset interference. One of the key parameters within a PN offset plan is the setting of the Pilot PN Offset Increment (PILOT_INC). The average cell site radius for an area will typically influence the setting of the PILOT_INC parameter. Since the Extended Range cell site significantly increases the cell site radius for an area, a higher setting of the PILOT_INC parameter is typically required for proper PN offset planning.

As stated earlier, there are two key functional capabilities provided with the Nokia Siemens Networks Extended Range Cell feature. The first part of the feature provides system support for a maximum cell radius setting of up to 180 km and the second part enables Pilot PN Offset Increments to be defined at the Sector-Carrier level. With the Nokia Siemens Networks Extended Range Cell feature enabled on a BTS, then this BTS does not have to have the same PILOT_INC as the CBSC. Thus, the PILOT_INC settings are defined at the BTS on a Sector-Carrier level. The information within Chapter 5 can be used to perform the basic PN Offset planning.

The usage of multiple PILOT_INC values within a system produces a situation where PILOT_INC boundaries and transition zones need to be created. PN offset planning for multiple PILOT_INC areas which provides specific guidelines for PILOT_INC boundaries and transition zones are provided in Section 5.4.7.


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4.6.2.2 Parameter Optimization Impacts

There are five basic search window/timing related considerations for Extended Range cell sites.

• PN Offset Interference Protection (Pilot_Inc)• Access Channel Search Window (Cell Radius - PamSz & AchPamWinSz)• Traffic Channel Search Window (TchAcqWinSz)• Subscriber Search Windows (SrchWinA, SrchWinN, SrchWinR)• Edge Sensing Handoffs (RTD DAHO Threshold)

PN Offset Interference Protection. The PN Offset Interference Protection considerations were discussed above and in Chapter 5.

Access Channel Search Window. The access channel search window establishes the maximum round trip propagation delay that the BTS will search for subscriber origination attempts. In effect, it establishes the maximum radius that the BTS will be able to receive an origination attempt. The system will automatically calculate and populate the PamSz & AchPamWinSz parameters based upon the user’s input for the Cell Radius parameter (which is entered in kilometers). When the Extended Range mode is turned ON for a BTS, the range of values for the Cell Radius parameter will be extended to 180 km. It is important to make sure that the Cell Radius parameter setting is within the limitations stated for the specific BTS type as shown in Table 4-21, Table 4-22, Table 4-23, and Table 4-24.

Traffic Channel Search Window. The traffic channel search window (TchAcqWinSz) defines the traffic channel acquisition in PN chips, which is used during the handover acquisition of a call. For normal applications, it should be set at least as large as the AchPamWinSz parameter (which is established by the Cell Radius parameter). For Extended Range cell sites, the guidelines provided in Section 5.3.6 of Chapter 5 should be followed.

Subscriber Search Windows. The subscriber search window parameters are SrchWinA, SrchWinN, and SrchWinR. SrchWinA is the active/candidate pilot set search window size which should be made large enough to incorporate ~95% of the expected delay spread energy. Since the typical application for an Extended Range cell site is for an open or line-of-sight RF environment, the expected delay spread energy should not increase from that of a normal cell site. Thus, the default settings for the SrchWinA parameter should also be adequate for Extended Range cell sites.

The SrchWinN and SrchWinR parameters represent the search window sizes associated with the Neighbor Set and Remaining Set pilots. The size should be made large enough to account for differential time delay between the subscriber and a potential handoff BTS given in the subscriber’s neighbor list. Additional handoff limitation information regarding Extended Range cell sites and the SrchWinN parameter is provided in 1. It is important to note that handoff relationships are symmetrical, so the SrchWinN and SrchWinR parameters will need to be adjusted for both the Extended Range cell site and the neighbor cells to the Extended Range cell site.

Edge Sensing Handoffs. An Extended Range cell site will increase the round-trip delay measurement for Database Assisted Hard Handoffs or Edge Sensing Handoffs. The measurements of the mobile distance from the BTS is impacted only due to the larger valid range of values (PN


4


Phase measurement reports) that may be received from the mobile. In order to accommodate Extended Range cell sites, the maximum RTD DAHO Threshold was increased to 180 km, since the trigger distance for a hard handoff should be greater for Extended Range cell sites.

Another optimization consideration is that the triggers for Edge Sensing handoffs are disabled during the time the subscriber slews to the new time reference from one cell in the active set to another. This time may be considerable if the subscriber is slewing its time reference from a large cell to a small cell. During this slew time, edge sensing handoffs will not be performed.

4.7 Theoretical vs. Simulator

It should be emphasized that a RF link budget and associated statistical propagation model (e.g.Hata), although useful as an analysis technique to evaluate relative differences between radio systems or to obtain a qualitative description of a CDMA system, cannot be used to guarantee capacity or coverage reliability. A detailed system design needs to be completed which takes into account the specific characteristics of the given area. Some of the specific characteristics to be accounted for are: site locations, subscriber distribution, terrain, and ground clutter. The generic assumptions of flat terrain, uniform subscriber distribution, and ideal site locations implied within the propagation and traffic distribution models do not adequately account for specific characteristics of actual systems.

The actual terrain of the area to be covered can greatly influence the range to which a site will propagate. Instead of an ideal line of sight propagation, reflections, diffractions and shadowing of the RF signal are taken into account to adjust the distance that the signal will propagate. In addition to the terrain, what is on the terrain, ground clutter, is quite important. A given RF signal will propagate further in an area that is desolate (little to no buildings or foliage), than in an area which is comprised of many buildings. Also, the placement of the site within this terrain is very important. Simply stated, if the site is surrounded by obstructions, the coverage of the site will be less than if there are no obstructions.

The actual traffic characteristics of systems are non-uniform with large variations possible from sector to sector. The more spectrally efficient a given radio technology is, the more economical it is to maintain the grade of service in these sectors by simply adding additional traffic channels. In less efficient radio systems, cell splitting is the only option available to maintain the grade of service. This often requires the addition of several cells to resolve the blocking problem in a single sector. This characteristic is not accounted for in the RF link budgets.

Many different criteria exist for determining the CDMA coverage area of a system. Among these criteria, differentiation should be made between the forward and reverse links, as well as, between the criteria that can be simulated as opposed to being field test measured. Differentiation of the subscriber unit needs to be considered. Fixed systems need to have different assumptions or considerations applied to the design that will be different from a system being designed to support mobility. Finally, a distinction must be made between coverage area as defined in the loaded system as opposed to the unloaded system. Coverage will change with loading. Any coverage test needs to keep loading in perspective.

Because of the interrelated nature of CDMA coverage, quality and capacity, and all of the issues


4


highlighted above, Nokia Siemens Networks utilizes the IDGP CDMA Simulator to estimate the performance of individual system installations.

The Nokia Siemens Networks IDGP CDMA Simulator may be used for analyzing DS-CDMA performance in proposed and existing systems resulting in predicted capacity, required system parameters and hardware loading information. It provides for a method of understanding the inter-relationship between coverage, capacity, and quality. It permits investigations into real Cellular/PCS system concerns such as edge effects, excess background noise, propagation anomalies, antenna beamwidth, subscriber distribution, receiver sensitivity impact, interference mitigation, power control and handoff. It also provides performance levels and determines required power allocation for page, sync, pilot, forward and reverse traffic channels (TCH) for different channel models, cell loading, and receiver characteristics. Both the reverse and forward link are simulated.

It should be noted that the accuracy of the simulator is dependent on the accuracy of the input it requires (such as path loss, traffic distribution, vehicle speed, etc.).


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4.8 References

1. Turkmani, Parsons and Lewis, "Measurement of building penetration loss on radio signals at 441, 900 and 1400 MHz", Journal of the Institution of Electronic and Radio Engineers, Vol. 58, No. 6 (Supplement), pp. S169-S174, September-December 1988

2. Turkmani and Toledo, "Modelling of radio transmissions into and within multistory buildings at 900, 1800 and 2300 MHz", IEEE Proceedings-I, Vol. 140, No. 6, December 1993

3. Aguirre, "Radio Propagation Into Buildings at 912, 1920, and 5990 MHz Using Microcells", 0-7803-1823-4/94 IEEE, session 1.6 & 1.7, pp. 129-134

4. Lee, William C.Y. "Mobile Communications Engineering", Copyright 1982, McGraw-Hill Inc. pg. 33-40.

5. Jakes, W.C., "Microwave Mobile Communications", IEEE Press Reissue 1993, (Wiley, New York, 1974), pp. 125-127

6. Okumura, Y., Ohmori, E., Kawano, T., Fukada, K.: "Field strength and ITs Variability in VHF and UHF Land-Mobile Radio Service", Rev. Elec. Commun. Lab., 16 (1968), pp. 825-873

7. Hata, M.: "Empirical formula for propagation loss in land mobile radio services", IEEE Trans. on Vehicular and Technology, VT-29 (1980), pp. 317-325

8. COST 231 - UHF Propagation, "Urban transmission loss models for mobile radio in the 900- and 1,800- MHz bands", COST 231 TD (91) 73 The Hagne, September, 1991

9. Parsons, David, "The Mobile Radio Propagation Channel", Copyright 1992, Reprinted 1996 by John Wiley & Sons Ltd.

10. Rappaport, Theodore S., "Wireless Communications Principles & Practices", Copyright 1996 by Prentice Hall PTR

11. Title 47, Part 24, Sub-Part E, Section 24.232.


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5 PN Offset Planning
Chapter
5

Table of Contents

and Search Windows


5.1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 3

5.2 Number of Pilot Offsets per CDMA Frequency. . . . . . . . . . . . . . . . . . . 5 - 3

5.3 PN Offset Planning - General . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 35.3.1 Consequences and Sources of Offset Interference. . . . . . . . . . . . . . .5 - 35.3.2 PN Offset Planning - Parameters and Terms . . . . . . . . . . . . . . . . . . .5 - 55.3.3 Converting Between Chips and Time or Distance. . . . . . . . . . . . . . .5 - 85.3.4 Search Windows and Geography. . . . . . . . . . . . . . . . . . . . . . . . . . . .5 - 95.3.5 Search Windows and Scan Intervals . . . . . . . . . . . . . . . . . . . . . . . . .5 - 115.3.6 Calculating of Traffic Channel Acquisition Size TchAcqWinSz. . . .5 - 12

5.4 PN Offset Planning - Solutions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 195.4.1 Mitigating Adjacent Offset Interference - General . . . . . . . . . . . . . .5 - 20

5.4.1.1 Adjacent Offset Interference Protection Based on Timing. . 5 - 205.4.1.2 Adjacent Offset Interference Protection Based on Signal Strength 5 -

215.4.2 Protection Against Co-Offset Interference . . . . . . . . . . . . . . . . . . . .5 - 235.4.3 Incorrect Identification of an Offset by the Base Station. . . . . . . . . .5 - 265.4.4 PILOT_INC and the Scan Rate of Remaining Set Pilots . . . . . . . . .5 - 275.4.5 Summary of Guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5 - 285.4.6 Guidelines for Assigning Offsets. . . . . . . . . . . . . . . . . . . . . . . . . . . .5 - 305.4.7 Guidelines for Planning Inter-CBSC and Intra-CBSC multiple PILOT_INC

Boundaries and Transition Zones. . . . . . . . . . . . . . . . . . . . . . . . .5 - 33

5.5 Reuse Patterns. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 36

5.6 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 - 37


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CDMA/CDMA2000 1X RF Planning Guide5 PN Offset Planning and Search Windows

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5.1 Introduction

This chapter will discuss the PN Offset Planning. Section 5.3 provides insight into the sources and consequences of offset interference. The definition of important terms and parameters are given. Also, since a knowledge of search windows is considered fundamental, a detailed explanation of this topic is included. Section 5.4 provides the theory that justifies placing certain boundaries on the value of PILOT_INC, which is central to PN Offset Planning. Section 5.4.5 and Section 5.4.6 will prove very useful to the offset planner by providing a summary of the factors pertinent to PILOT_INC selection along with a concise listing of all the planning guidelines. Section 5.4.7 provides guidelines for offset planning at Inter-CBSC and Intra-CBSC boundaries when different PILOT_INC values are involved. Some information has been provided that will benefit system optimizers. This includes information on scanning rates (Section 5.3.5 and Section 5.4.4). Finally, references are provided for further study of this important topic.

Please note that all of the information provided on this topic applies equally to IS-95A, IS-95B, and IS-2000 specifications.

5.2 Number of Pilot Offsets per CDMA Frequency

The Pilot Channel is a spread spectrum signal carrying no data and is always transmitted on a downlink CDMA channel. The subscriber stations use the pilot to acquire the system, and to assist in several signal processing functions such as synchronization, demodulation (phase reference), soft handoff and channel estimation. The uniqueness of the pilot is achieved through time shifts of a basic sequence known as zero shift pilot or short PN sequence. Since sectors are distinguished by time shifts of a given pseudo-noise sequence, enough separation between time offsets must be provided to avoid “mutual pilot interference”. Per TIA/EIA IS-95 Interim Standard, the chosen length for the pilot PN sequences is 32,768 chips (Section 7.1.3.1.9) with a minimum separation of 64 chips (Section 7.1.3.2.1) between adjacent offsets. This leaves a maximum of 512 (32768/64) distinct pilot offsets available for a CDMA frequency.

5.3 PN Offset Planning - General

Before actually doing a PN offset plan, it will be beneficial to have a general understanding of scenarios to avoid when designing the PN offset plan, to learn the general terms and definitions that are associated with PN offset planning, and to gain an understanding to the various search windows.

5.3.1 Consequences and Sources of Offset Interference

The design of a PN offset plan for CDMA is comparable to that of a signalling channel frequency plan in analog. The consequences of poor offset planning include the following:

• Active Set Pilot Interference - This phenomenon would occur in the active area and involve the active search window (SRCH_WIN_A). The interfering signal would need


5


to be strong enough to be processed as an active finger (except in the less likely case where the timing was perfectly coincident with a true active finger).

• Neighbor Set Pilot Falsing - A neighbor set pilot may falsely appear strong enough for the subscriber to promote the pilot to the candidate set and recommend to the base station (BS) to perform a soft handoff ‘add’ via the Pilot Strength Measurement Message (PSMM). This falsing would occur in the neighbor area and involve the neighbor search window (SRCH_WIN_N). The falsing signal strength would need to meet the T-ADD threshold criteria.

The probability for interference or falsing is dependent upon two factors: timing and strength. Time differentials can be translated into geographic regions and have as their threshold the search window size. A detailed discussion of this topic will be found later within this chapter. If a signal falls outside of a search window, its energy becomes nothing more than uncorrelated interference. Note that the term active area is meant to refer to the area in which a signal may be (or is intended to be) actively demodulated. The term neighbor area refers to the area in which a signal will be sought as a candidate. In geographic terms, the neighbor area greatly expands the region where problems may occur since the search for a neighbor signal lies in many areas outside of the active area. The use of large or generous neighbor lists along with the technique of merging neighbor lists when in soft/softer handoff creates further expansion. Mitigating this expansion of the geographic space in which falsing may occur is the heightened signal strength threshold at which interference may occur (a T-ADD of -14dB versus a finger-locking threshold of approximately -24dB).

• Incorrect BS Identification - A signal may travel far enough to be incorrectly identified by the BS when it translates the subscriber reported phase into a PILOT_PN offset index.

In this document, the phrases interference and falsing may be used interchangeably.

In analog systems, ‘co’ and adjacent channel interference are major factors in the system design. The co-channel interference was managed via the antenna configuration and the reuse pattern/distance. The adjacent channel interference was managed through the application of a simple frequency planning rule.

With the CDMA channel, all sites reuse the same frequency. Interference isolation on the forward CDMA channel is obtained via short PN code offsets (inter-sector) and Walsh codes (intra-sector). The possible sources of interference/falsing include ‘co’ and adjacent offsets.

Since CDMA pilots are distinguished through offsets of the same short PN code, adjacent channel interference has its counterpart in CDMA when phase shifts occur caused by propagation delays.Using phase for cell identification may therefore cause falsing problems as depicted in Figure 5-1.


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Figure 5-1: PN Offset Planning

The phase delay used in the figure above need not be so exact to create problems. The falsing of one signal need only fall within the search window of the subscriber.1

The valid set of offsets is limited to multiples of PILOT_INC. In Figure 5-2 below, a PILOT_INC of 2 was chosen. Offset 4 is adjacent to and can interfere with 6 if it arrives ~2 offsets late which implies that 4, the interfering signal, is traversing a significant distance. Conversely, offset 6 may interfere with 4, but 6 would need to arrive ~2 offsets early which implies that the subscriber is acting at a significant distance from the site using offset 4. If the PILOT_INC is chosen carefully, there should be little concern with 2 interfering with 6 or 6 with 2.

Figure 5-2: Short PN Sequence w/PILOT_INC = 2

As with analog, a reuse distance must be maintained between sectors implementing the same PN offset to avoid interference. Since the pilot signal is integral to the operation of a CDMA system, careful PN offset planning should be performed to mitigate interference between sites using the same offset and falsing between adjacent PN codes which result from phase delay.

5.3.2 PN Offset Planning - Parameters and Terms

There are various parameters and terms which come into play when discussing PN offsets and their function in CDMA.

1. Note also how time, distance, and chips are all related. Refer to Section 5.3.3.

PN 0

PN 1

t0 = 102 μsec

t1 = 50 μsec

Avoid ambiguity which could result from phase delay.

Δt = t0 - t1 = 102 μsec - 50 μsec = 52 μsec PN 1 - PN 0 = 64 chips = 52 μsec = 9.6 miles Traversing the additional distance of 9.6 miles, the PN 0 signal has phase shifted sufficiently so as to be received by the subscriber with essentially the same phase as PN 1.

2 4 6 8 10


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System Time

All base station digital transmissions are referenced to a common CDMA system-wide time scale that uses the Global Positioning System (GPS) time scale, which is traceable to and synchronous with Universal Coordinated Time (UTC).2

Time Reference

The subscriber establishes a time reference which is used to derive system time. This time reference will be the earliest arriving multipath component being used for demodulation.3 This reflects the assumption that the subscriber’s fix on system time is always skewed by delay associated with the shortest active link.

PILOT_PN

The Pilot PN sequence offset (index), in units of 64 PN chips. It ranges from 0 to 511. Every transmit sector will have an offset assigned to it.

Active Set

The pilots associated with the Forward Traffic Channels assigned to the subscriber.4 It is the base station that assigns all active set pilots to subscribers.

Candidate Set

The pilots that are not currently in the Active Set but have been received by the subscriber with sufficient strength to indicate that the associated Forward Traffic Channels could be successfully demodulated. As a property of the Mobile Assisted HandOff (MAHO), the subscriber promotes a Neighbor Set or Remaining Set pilot to the Candidate Set when certain pilot strength criteria are met and then recommends the pilot to the base station for inclusion in the Active Set.

Neighbor Set

The pilots that are not currently in the Active Set or the Candidate Set and are likely candidates for handoff. Neighbor Set pilots are identified by the base station via Neighbor List and Neighbor List Update messages.

Remaining Set

The set of all possible pilots in the current system on the current CDMA frequency assignment, excluding pilots in the other sets. These pilots must be integer multiples of PILOT_INC (defined below).

2. EIA/TIA/IS-95A, Mobile Station - Base Station Compatibility Standard for Dual-Mode Wideband Spread Spectrum Cellular System, §1.2.3. Ibid., §6.1.5.1.4. Ibid., §6.6.6.1.2.


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SRCH_WIN_N, SRCH_WIN_R

These parameters represent the search window sizes associated with Neighbor Set and Remaining Set pilots.5 The subscriber centers the search window for each pilot around the pilot’s PN sequence offset using timing defined by the subscriber’s time reference.

In general, a neighbor search window, SRCH_WIN_N, will be sized so as to encompass the geographic area in which the neighbor may be added (a soft handoff “add” zone or “initial detection area”). The largest a neighbor search window need be is such that it is sufficient to cover the distance between the neighbors, , plus an accommodation of the time-of-flight delay (approx. 3 chips).

SRCH_WIN_A

This parameter represents the search window size associated with the Active Set and Candidate Set pilots.6 The subscriber centers the search window for each pilot around the earliest arriving usable multipath component of the pilot. Note that in contrast to the neighbor or remaining set search windows, the active/candidate search windows "float" with the desired signals. That is to say that the center position of the search window is updated every scan to track the new location of the earliest arriving multipath component.

To better illustrate the relationships between search windows, consider the following scenario:

A subscriber monitors a neighbor pilot. The neighbor search window is centered on the neighbor pilot offset. This centering is relative based on timing derived from the time reference. When the pilot strength of a neighbor pilot recommends promotion to the candidate set, then the search window will be tightened to the active search window size. The active search window is sized to compensate for delay spread only and is, therefore, smaller than the neighbor search window. In addition, the active search window locks onto and tracks the candidate pilot.

PILOT_ARRIVAL

The pilot arrival time is the time of occurrence of the earliest arriving usable multipath component of a pilot relative to the subscriber’s time reference.7

PILOT_PN_PHASE

The subscriber reports pilot strength and phase measurements for each active and candidate pilot in the Pilot Strength Measurement Message when recommending a change in the handoff status (i.e. mobile assisted handoff). The subscriber computes the reported PILOT_PN_PHASE as a function of the PILOT_ARRIVAL and the PILOT_PN.8 The pilot arrival component represents the time delay of the pilot relative to the time reference or, in other words, how skewed the pilot is

5. Ibid., §6.6.6.2.1.6. Ibid.7. Ibid., §6.6.6.2.4.8. Ibid.

3R


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from the subscriber’s concept of system time. Both the PILOT_ARRIVAL and PILOT_PN_PHASE measurements are in chips (15 bits, 0 to 32,767 or 215-1) while the PILOT_PN is in offsets (9 bits, 0 to 511). The difference (6 bits) corresponds to the 64 chip interval between successive PN offsets.

Note also that the subscriber does not identify pilots by their offset index directly, but by their phase measurement. If the pilot arrival was larger than 32 chips (1/2 of a pilot offset or 4.8 miles), then this could undermine the ability of the base station to properly translate pilot phase into pilot offset index (given a PILOT_INC of 1).

PILOT_INC

The pilot PN sequence offset index increment is the interval between pilots, in increments of 64 chips. Its valid range is from 1 to 15. The subscriber uses this parameter in only one manner, to determine which pilots to scan from among the Remaining set. Only valid pilots (i.e. those pilots that are multiples of PILOT_INC) will be scanned. For the subscriber, PILOT_INC impacts only the scanning rate applied to Remaining pilots. It accomplishes this by reducing the number of Remaining pilots that need to be scanned.

For the base station, the effect of the PILOT_INC is different. In the base station, it is used in properly translating pilot phase back into pilot offset index. The consequence is that the operator may artificially increase the separation between valid time offsets. By selecting a PILOT_INC of 2, for instance, an operator chooses to limit the number of valid offsets to 256 (i.e. 0, 2, 4,..., 508, 510) instead of 512. The increased separation means that the pilot arrival must be larger before adjacent offset ambiguity is possible and consequently the likelihood of a strong adjacent interferer is reduced.

5.3.3 Converting Between Chips and Time or Distance

Chips are related to time by the following relationship:

[EQ 5-1]

Chips are related to distance by the following relationship:

[EQ 5-2]

Or, in kilometers:

[EQ 5-3]

Note that the chip rate (1.2288 Mcps) and the speed of light (186,000 miles/sec) are fundamental to these conversions.

Time (us)Chips

1.2288 Mcps------------------------------- Chips 0.8138 us/chip×= =

Distance (miles) Chips 0.8138 us/chip 186,000 miles/1,000,000 us×× Chips 0.1514 miles/chip×= =

Distance (km) Chips 0.8138 us/chip 299,311 km 1,000,000 us⁄×× Chips 0.244 km/chip×= =


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5.3.4 Search Windows and Geography

Before discussing offset planning in any detail, a brief discussion of search windows and their spatial relationships to cell sites and subscribers is needed. Base Stations, by virtue of their GPS tracking, have an exact concept of system time. This, in turn, means that signals leaving these sites have precise offsets and identities. On the other hand, subscribers derive their timing from a time reference. Their concept of system time is skewed late by the time-of-flight delay associated with this time reference signal. The greater the distance between the subscriber and the time reference site, the greater the skewing.

Consider the diagram below:

Figure 5-3: Subscriber Location Relative to Search Window

Let subscriber A, Site 1 and Site 2 be co-linear with subscriber A positioned exactly between Sites 1 and 2 and with Site 1 active. The subscriber’s concept of system time is skewed from real system time by X, the distance between the subscriber’s concept of time and its time reference. When the subscriber searches for a neighbor, it will center the search window on the offset associated with the neighbor, but based on its own system time (which, of course, is a little late compared with real system time). Assuming Site 2 to be a neighbor of interest, its signal traverses a distance to subscriber A that is exactly as late as the subscriber’s time reference. Under these circumstances, the time differential between the two signals is zero (i.e. X-X = 0) and the signal from Site 2 will fall directly in the center of the neighbor search window in which the subscriber is searching for Site 2.

Now, consider subscribers B and B’. Subscriber B is located 1 chip closer to Site 1 with Site 1 active; therefore, subscriber B’s system time is skewed by only X-1. The signal from Site 2 traverses X+1 and the time differential between the two signals is (X-1) - (X+1) = -2; consequently, the signal from Site 2 is arriving 2 chips late and will appear 2 chips off center in the neighbor search window. Please note that a 1 chip shift in spatial location has had a 2 chip impact on the location within the search window. Conversely, subscriber B’ has timing skewed by X+1 while Site 2’s signal traverses only X-1 chips, leading to a time differential of (X+1) - (X-1) or 2 chips. Site 2’s signal is arriving early by 2 chips. To design a search window large enough to encompass locations B and B’, a search window of at least 4 chips or + 2 chips wide would be required.

The worst case time differential is when the subscriber is located directly adjacent to one site while trying to detect or demodulate the signal from the other site. For example, subscriber C effectively has timing that is coincident with system time (i.e. its skewing is 0). Site 2’s signal is arriving D

A

B’C

2 1

B

D = distance between Site 1 and Site 2X = D/2

1 chip


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chips late. For this signal to fall into the search window, it must be sized + D chips or 2D chips wide. Since this is the worst case scenario, the following should be understood: if a search window is sized large enough to compensate for the distance between the two sites (i.e. 2D), then there is no location where a subscriber would have one site as its time reference and not see the other site in its search window.

Here is a more generalized depiction of search windows in space:

Figure 5-4: Search Windows in Space

The two sites are located at (0,0) and (10,0) and are 10 units apart. The curves represent constant time differentials between the two sites and will correspond to the edges of certain search window sizes. Search windows will be centered on the perpendicular line half-way between the sites. The width of the search window in space will correspond to half of the search window size in chips. For example, the two lines corresponding to time differentials of -4 and +4 demarcate an area that corresponds to a search window that is + 4 units or 8 units in width. In geographic space, the width of the area on the line between the two sites will only be 4 units wide or 1/2 of the search window size. Between the curves, a subscriber tied to one site will see the other site fall within its search window. Conversely, no matter how strong a neighbor signal may be, if the subscriber is located outside of the search window area, it will not detect the signal.

Note how the curves bend as the search window is enlarged. When the search window is made large enough to compensate for the distance between the two sites, the curves collapse upon themselves indicating that there is no longer any region in space where the signal will not fall within the search window. In general, a generous attitude toward search window sizing should exist. The ability to demodulate a signal depends on being able to see it. The table below correlates distance between neighbors to search window sizes.

Diff = -8

Diff = -6Diff = -4

Diff = -2 Diff = 2 Diff = 6Diff = 4

Diff = 8

Diff = -10 Diff = 10


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The SrchWin sizes come from their definition in IS-95A/J-STD-8. The equation correlating Window Size (in chips) to distance between neighbors (in miles) is:

[EQ 5-4]

The two chips removed from the Window Size compensate for time-of-flight (i.e. real world) delays. If starting with a distance between sites to calculate a window size, two chips would need to be added.

This discussion on search windows was designed to help the system engineer visualize the spatial relationship of search windows to cell sites. An individual out in the field can estimate how large a search window would need to be for a particular location by estimating the time differential between the two sites of interest (use the absolute value only), adding 1 chip (to compensate for time-of-flight delays), and multiplying by 2.

5.3.5 Search Windows and Scan Intervals

The following information is provided to give insight to system optimizers and is based on Nokis Siemens Networks’ general understanding of subscriber vendor pilot scan algorithms. It is important to note that such algorithms are not specified through IS-95A/J-STD-008 and are, therefore, manufacturer specific. Also, pilot scanning rates/intervals are a function of many variables.

In general, active and candidate pilots are scanned at a rate of 50 times/second or better. This would be valid for up to a total of 6 pilots and is not impacted by the number of neighbors or remaining set pilots.

Neighbor set pilots are scanned anywhere between 2 to 40 times/second with a common range being 4 to 15 times/second. The rate is dependent on the number of actives/candidates and neighbors.

Remaining set pilots are scanned on the order of seconds. The remaining set pilots will be scanned NR times slower than the neighbors (where NR represents the number of remaining set pilots, a function of PILOT_INC).

Table 5-1: Search Window Size vs. Neighbor Separation

SrchWin 0 1 2 3 4 5 6 7Window Size (chips) 4 6 8 10 14 20 28 40Delay (μs) 1.6 3.3 4.9 6.5 9.8 14.6 21.2 30.9Neighbor Separation (mi) 0.2 0.3 0.5 0.6 0.9 1.4 2.0 2.9SrchWin 8 9 10 11 12 13 14 15Window Size (chips) 60 80 100 130 160 226 320 452Delay (μs) 47.2 63.5 79.8 104.2 128.6 182.3 258.8 366.2Neighbor Separation (mi) 4.4 5.9 7.5 9.7 12.0 17.0 24.2 34.2

distance miles( )Window Size 2–( )

2---------------------------------------------- 0.1516×=


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5.3.6 Calculating of Traffic Channel Acquisition Size TchAcqWinSz

In CDMA system, the parameter TchAcqWinSz is used to specify the traffic channel acquisition search window size. This parameter is used in soft handoff and its range is calculated based on the following factors: Cell radius of the current sector-carrier, cell radius of neighboring cells (including itself), and internal hardware delays.

Starting from R18.0, the calculation of TchAcqWinSz has changed for the following reasons:

• The hardware adjustment needed to account for the delay in the BTS has been elimi-nated.

• The Extended Range Cell feature 8059 allows the radius of CDMA cells to be expanded from the current 56 km limit to a maximum of 180 km. The increase in cell radius size needs to be considered for proper setting of TchAcqWinSz.

• The OMC-R does not restrict operators from entering TchAcqWinSz value outside the allowed range that the BTS hardware can handle. The OMC-R does not check for input value of TchAcqWinSz based on MCC hardware. The MCC-1X allowed value is between 100 and 1500 PN Chips. Whereas the other devices (MCC8/MCC24 card, SC3XX and SC3XX-1X frames) allowed value is between 100 and 510 PN Chips. The improper setting of TchAcqWinSz would cause drop calls during soft handoff or adversely affect the operation of BTS MCCs.

If the Extended Range Cell feature is used, a MCC-1X card is required when:

• The range of the cell is beyond 56 km and the interference cancellation is disabled.• The range of the cell is beyond 36 km and the interference cancellation is enabled.

The TchAcqWinSz needs to be set correctly by release and MCC hardware type. The formula is different between Pre-Release 18.0 and Release 18.0 onwards. The steps used to determine proper TchAcqWinSz are given below.


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1 To determine TchAcqWinSz from OMC-R CLI, the command DISPLAY BTS-BTS# TCHGEN is used to retrieve the traffic channel acquisition search window size. The default value is 100 chips.

Traffic Channel Acquisition

Search Window Size

(ACQ WINSZ) (PN Chips)

-------------------------|-------|--------|-----------|--------|-----------|----------|--------|-----------|-----------|---------|-----------

CARRIER-209-1-76 3.8 0.400 25 5.58 6 25 5.27 6 100 6.66 6

CARRIER-209-1-292 3.8 0.400 25 5.58 6 25 5.27 6 100 6.66 6

CARRIER-209-2-76 3.8 0.400 25 5.58 6 25 5.27 6 100 6.66 6

CARRIER-209-2-292 3.8 0.400 25 5.58 6 25 5.27 6 100 6.66 6

CARRIER-209-3-76 3.8 0.400 25 5.58 6 25 5.27 6 100 6.66 6

CARRIER-209-3-292 3.8 0.400 25 5.58 6 25 5.27 6 100 6.66 6

2 The calculation for TchAcqWinSz in Release 17.0 and earlier releases is as follows:

TchAcqWinSz would fall with the range given below:

Table 5-2: Traffic Channel Window Size

TchAcqWinSz for sector-carrier = Ceiling ((X + Y) * OneWayDelay + HwDelay)

Where: X = the cell radius in km of its sector-carrier Y= the highest cell radius in km of neighboring cells (including itself X) OneWayDelay = 4.096 chips/km (one way delay)


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3 In Release 18.0, if non-MCC-1X card, SC3XX or SC3XX-1X frames are used, the R17.0 equation is still applicable for calculating TchAcqWinSz. This equation also applies to BTS frames that are equipped with a mixture of both MCC-1X and non-MCC-1X cards. The allowed value for TchAcqWinSz is between 100 and 510 PN chips. The equation change in R18.0 specified in next paragraph is used only if the BTS frame contains solely MCC-1X cards. In other words, if Feature 8059 is enabled or SC7224 frame is used, the new equation will be used to calculate TchAcqWinSz since both Feature 8059 and SC7224 frames would require BTS to be equipped with MCC-1X cards only.

4 In Release 18.0, TchAcqWinSz range is calculated as follows. Internal hardware delays has been incorporated as part of Feature 9161 changes. Based on this equation, the effective range for TchAcqWinSz is affected by the BTS type, the Interference Cancellation functionality, and the expanded Cell Radius size in Feature 8059.

Example #1: TchAcqWinSz calculation in R17.0 and earlier release:

Assumptions:* X and Y are neighbor cells. * Cell X has a radius of 40 km. * Cell Y has a radius of 30 km (with or without Interference Cancellation enabled).

Cell X TchAcqWinSz CalculationAs stated in the equation, Y is selected from the highest cell radius of any neighboringcells of its own cell. In this case, since cell X has a radius greater than cell Y, cell X radiusis used for both X and Y distance values.

Cell X TchAcqWinSz = ((40 + 40 km) * 4.096 chip/km) + 38.5 chips

= 341 chips

Cell Y TchAcqWinSz CalculationIn this case, since the neighboring cell X has a radius greater than cell Y, cell radiusdistances X and Y are used in the formula. This example also demonstrates the maximumchip size that can be used by a cell that has interference cancellation enabled.

Cell Y TchAcqWinSz = ((30 + 40 km) * 4.096 chip/km) + 38.5 chips = 326


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The TchAcqWinSz range is computed as follows. With Feature 8059, the recommended range increases proportionally to the cell radius. The allowed range for MCC-1X is between 100 and 1500 PN chips, while the range for MCC24 cards and SC3xx series BTS is between 100 and 510 PN chips. This rule is enforced in Release 19.0 where preconditions are checked at OMC-R client to ensure the input value for TchAcqWinSz is within the recommended range.

TchAcqWinSz for sector-carrier = Ceiling ((X + Y) * OneWayDelay + PathDelay)

Where: X = the cell radius in km of its sector-carrier Y = the highest cell radius in km of neighboring cells (including itself X) OneWayDelay = 4.096 chips/km (one way delay) PathDelay = 13 chips (multipath delay)


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Table 5-3: Cell Radius vs. TchAcqWinSz


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Example #2: TchAcqWinSz calculation in R18.0 and later releases:

Cell X TchAcqWinSz = ((40 + 40 km) * 4.096 chip/km) + 13 chips

Cell Y TchAcqWinSz = ((10 + 40 km) * 4.096 chip/km) + 13 chips = 218 chips

Assumptions: * X and Y are neighbor cells. * Cell X has a radius of 40 km. * Cell Y has a radius of 10 km. * Interference Cancellation disabled for both cells. * Feature 8059 disabled for both cells.

Cell X TchAcqWinSz Calculation Similarly, cell X has a radius greater than the neighboring cell Y. Hence its cell radius is used for both X and Y distance values.

= 341 chips

Cell Y TchAcqWinSz Calculation The neighboring cell X has a radius greater than cell Y. Hence cell radius distances X and Y are used in formula.


5



Assumptions:* X and Y are neighbor cells. * Cell X has a radius of 80 km with FR8059 enabled. * Cell Y has a radius of 45 km. * Interference Cancellation disabled for both cells.

Cell X TchAcqWinSz Calculation Cell X has a radius greater than the neighboring cell Y. Hence its cell radius is used for bothX and Y distance values. Cell X would have to be equipped only with MCC-1X cards perFeature 8059 requirements.

Cell X TchAcqWinSz = ((80 + 80 km) * 4.096 chip/km) + 13 chips = 669 chips Cell Y TchAcqWinSz Calculation

The neighboring cell X has a radius greater than cell Y. Hence cell radius distances X and Yare used in the formula. This configuration should be avoided as it will not be possible tofully support soft handoffs to cell Y. This is because the sum of both cells exceeds the 112 kmlimit allowed for cells with a cell radius between 0 and 56 km, and the required chips tofacilitate soft handoff is greater than 472 chips (refer to table above). There are two possibleoptions to resolve this conflict: 1) Increase the cell radius Y to the next range (56.1 km)allowing the TchAcqWinSz limit to be larger, or 2) Decrease the cell radius X such that thesum of the two cell radii is smaller than 112 km. With the first option, the extended rangefunctionality of Feature 8059 would be used requiring all MCCs within the BTS to bereplaced with MCC-1X cards.



5


5.4 PN Offset Planning - Solutions

Current concepts for PN offset planning generally center on finding an appropriate value for PILOT_INC. A large value would provide good protection against adjacent offset interference since the pilot needs to travel a greater distance before potentially falsing (since signal attenuation is highly correlated with propagation distance). However, too large a value implies too few valid PN offsets and too small a reuse distance, thereby increasing the likelihood of co-offset interference. Conversely, a small value of PILOT_INC delivers a large set of valid PN offsets, a large reuse pattern and reuse distance, thereby reducing the likelihood of any co-offset interference. However, too small a value will not provide good isolation against adjacent offset interference or ambiguity. With the implementation of the Extended Range Cell feature, it is possible to set the PN offsets on a per sector/carrier basis.

Prior to discussing in detail the planning limits for PILOT_INC, it is important to note the following concerning R, the radius of the cell site. CDMA’s use of soft handoff makes the radius of the active area significantly larger than that which is accustomed with analog and which is associated with a hexagonal grid. Speaking of the radius of a site conveys significant information since both reuse distance, D, and cluster size, N, are related as follows:


Cell X TchAcqWinSz = ((60 + 60 km) * 4.096 chip/km) + 13 chips


Assumptions: * X and Y are neighbor cells. * Cell X has a radius of 60 km with Extended Range Cell feature enabled. * Cell Y has a radius of 10 km with Interference Cancellation enabled.

Cell X TchAcqWinSz Calculation

= 505 chips

Cell Y TchAcqWinSz Calculation

Cell X has a radius greater than the neighboring cell Y. Hence its cell radius is used for both X and Y distance values. Cell X would have to be equipped only with MCC-1X cards per Feature 8059 requirements.

The neighboring cell X has a radius greater than cell Y. Hence cell radius distances X and Y are used in formula. The resultant 300 chips is less than the 308 chips allowed when the cell radius is 0-36 km with Interference Cancellation enabled (refer to table above), so this is an acceptable configuration. Had the value been greater than 308 chips the cell radii would need to be reconsidered in order to fully support soft handoffs.


5


[EQ 5-5]

However, with CDMA and soft handoff there is significantly greater overlap between sites. If the hexagon/analog oriented radius is labeled as Rhex and the CDMA active area radius is labeled as Rcdma, then it needs to be understood that Rcdma can easily be twice as large as Rhex, perhaps slightly larger. Many discussions of offset planning have failed to characterize this difference and consequently lead to faulty conclusions. Specifically, consider a recommendation that suggests that 5R is sufficient separation for reusing sites. If the R is taken to be Rhex, then D/R would be 5 and the cluster size would be 9. However, if it is understood that R is Rcdma, then D/R would be more on the order of 10 and the cluster size would be 36, which is a significant difference.

5.4.1 Mitigating Adjacent Offset Interference - General

The following explanations, which define the limits of adjacent offset interference based on timing and signal strength considerations, are not impacted by antenna configuration (whether the sites are omni, 3-sector, or 6-sector). This attribute simplifies the discussion.

5.4.1.1 Adjacent Offset Interference Protection Based on Timing

For an adjacent offset to have the potential of falsing, it must meet a timing criteria. That is to say that it must fall into the search window. This is depicted below:

Figure 5-5: Minimum Distance for Adjacent Offset Interference

A signal from a potential adjacent interferer must traverse a minimum distance to be able to fall into the search window of the adjacent offset.

[EQ 5-6]

In this equation, S is 1/2 of the search window size. In cases where a boundary exists between regions using different Pilot Increments, use the larger of the two Pilot Increment values to perform the calculation. For example, with a PILOT_INC = 3 and SRCH_WIN_N = + 30 chips, this minimum distance corresponds to 3 x 64 - 30 = 162 chips = 39.5 Km = 24.6 miles. A larger PILOT_INC provides greater isolation; conversely, larger SRCH_WIN_N values mitigate the

DR---- 3 N×=

3 6

PILOT_INC

SRCH_WIN_X = + S

Minimum Distance PILOT_INC S–=


5


isolation.

Of course, the value of 60 chips for SRCH_WIN_N is a recommended starting value and will take on larger or smaller values. Since SRCH_WIN_A is always smaller than SRCH_WIN_N, an adjacent offset interferer must always travel a greater distance to potentially interfere in the active search window.

Due to this timing requirement, a general rule can be established concerning placement of an adjacent offset and its neighbors. They should be located under the PILOT_INC - S umbrella (Equation 5-6) within the cluster. To the degree that this criteria is met, it eliminates the potential for adjacent interference within the cluster. The limit of this example is to place adjacents with sectors that are co-located. Under these conditions, there is no time differential between signals leaving the site and only distant reflections can possibly achieve the time constraints of interference, which is highly unlikely.

5.4.1.2 Adjacent Offset Interference Protection Based on Signal Strength

The timing discussion can be expanded by taking into account signal strength considerations. The lower bound on PILOT_INC is identified and will correlate to an acceptable C/I threshold. Consider this equation which seeks to guarantee a bounded interference between correlated pilots, effectively yielding the PILOT_INC.9

[EQ 5-7]

Table 5-4: Distance/Timing Restriction on Adjacent Interference(assuming SRCH_WIN_N = + 30 chips)a

a. For ease of performing mental math, note that each offset of 64 chips contributes a little less than ~10 miles (9.7) or a little more than ~15 km (15.6). The 30 chip search window accounts for a 7.3 km or 4.5 mile reduction.

PILOT_INC (offsets)

PILOT_INC (chips)

Minimum Distance (chips)

Minimum Distance

(km)

Minimum Distance (miles)

1 64 34 8.3 5.2

2 128 98 23.9 14.9

3 192 162 39.5 24.6

4 256 226 55.1 34.3

5 320 290 70.8 44.0

6 384 354 86.4 53.7

9. Qualcomm, “The CDMA Network Engineering Handbook”, March 1, 1993, §9.4.2.

m 10a law 10×( )⁄

1–( ) R× S+≥ k R× S+=


5


In this equation, R is the radius of the cell in chips, S is 1/2 of the search window size, a is the desired C/I in dB, and law represents the propagation exponent. The result, m, represents the required offset, in chips, between any two pilots so that the desired C/I can be achieved. The relationship can be interpreted as recommending that for each chip of R, there should be k chips of separation for an adjacent offset so that a minimum C/I threshold is achieved. In this equation, the presence of S reflects the fact that the correlation need not be perfect for interference to exist. The adjacent signal need only fall into the search window (a less stringent timing criteria).10 Note also that Equation 5-6 and Equation 5-7 are identical in form. Equation 5-7 is stating that at a distance of PILOT_INC - S (or m - S), the C/I threshold will be achieved. The following table shows a few different examples of the calculation:

A conservative propagation exponent was chosen to compensate for the simplicity of the approach (for example, the assumption of uniform power at both sites). The C/I threshold was set at 18.0 dB to correspond to a 12 dB C/I threshold (6 dB fade margin, 90% area reliability w/8dB deviation) for a 2 cell system. This 12 dB imbalance seems sufficient to predict that the searcher will not select the interfering energy within the active window. Under unloaded conditions (worst case), this threshold corresponds to an interferer Ec/Io of -14.9 dB which is below the normal range for the T-ADD setting; therefore, neighbor window falsing is unlikely. Additionally, to generate the table values, neighbor search window sizes, which vary with cell radius, were used.

Although these table values seem fairly generous, there is one element mitigating the results. An appropriate value for R must take into account two factors. First, the R is Rcdma. Additionally, since path loss is not isotropic and systems are not ideally laid out on grids (i.e. are non-uniform) the selection of R should reflect a limiting case. Since a system-wide value of PILOT_INC is being determined, the value of R should more closely represent the 90th percentile rather than the mean. The radius of highway sites and other larger radius sites that are not clustered need not dominate the analysis since spatial separation may be used to mitigate interference in those cases.

10. An earlier, more conservative version of this relationship had S also scaled by k.

Table 5-5: Pilot Sequence Offset Index Assignment(assuming a = 18.0 dB, law = 3.0, k = 2.98)

R (km)

R(miles)

R(chips)

S(chips)

m(chips)

PILOT_INC (offsets)

Number of Valid Offsets

Cluster Size

(3-sector)

24.9 15.5 102 80 384 6 85 28

20.9 13.0 85.5 65 320 5 102 34

16.9 10.5 69.1 50 256 4 128 42

12.4 7.7 51.0 40 192 3 170 57

8.0 5.0 32.9 30 128 2 256 85

4.1 2.5 16.8 14 64 1 512 170


5


5.4.2 Protection Against Co-Offset Interference

The following explanations, which define the limits of co-offset interference based on timing and signal strength considerations are impacted by both the antenna configuration (i.e. omni or sector) and whether the subscriber is in the active area or in the larger neighbor area. As such, they will need to be more extensive.

The study of co-offset interference is started by looking at the timing considerations involved in interfering within the active search window. Consider the following diagrams:

Figure 5-6: Active Window Interference Timing Criteria

It has been stated elsewhere11 that if two users of the same offset where positioned 2R + S away from each other (where S is 1/2 of the search window size), then the potential for co-offset interference is avoided due to the timing criteria not being met. From the discussion on search windows in Section 5.3.4, it can be seen that if two sites met this criteria for separation, then the search window would spatially fall completely outside of R. For the sectorized case, the requirement was modified to R+S.

While meeting this criteria is sufficient to protect against interference within the active search window, it does not protect against falsing within the neighbor search window. From a timing perspective, neighbor falsing will be limiting. Consider the following diagrams:

11. Qualcomm, “The CDMA Network Engineering Handbook”, March 1, 1993, §9.4.2.

A

B

SActive

R

R

OMNI SECTOR

A

B

SActive

R


5


Figure 5-7: Neighbor Window Interference Timing Criteria

Here are some guidelines used in generating these approximations:

• There can be no common neighbors among users with the same offset, no sector may share an offset assignment with one of its neighbors nor may any of its neighbors share the same offset assignment.

• The distance 2R + SActive is sufficient to define non-neighbors.

• A’s Neighbor Area is limited to 3R + SActive for omni and 2R + SActive for sector.

• For omni systems, B must be separated by SNeighbor from A’s Neighbor Area to avoid neighbor falsing.

• For sector systems, B possesses back-side neighbors (i.e. the co-located sectors) which must be separated by SActive from A’s Neighbor Area to avoid sharing common neighbors.

The conclusions from this exercise are summarized in the following table:

The previous analysis, though simple, can help establish a safe margin easily. A somewhat more detailed analysis below may help determine an absolute minimum reuse distance based on timing.

Table 5-6: Estimates of Reuse Distance and Cluster Size Based on Timing(assuming Rcdma = 2Rhex, SNeighbor ≅ 2Rhex and SActive ≅ 1Rhex)

Antenna Configuration Reuse Distance Equation Reuse Distance Cluster SizeOmni 4Rcdma + SActive + SNeighbor 11Rhex 43

Sector 3Rcdma + 2 x SActive 8Rhex ~21

A

B

C

SActive

R

R

R

R

SNeighbor

OMNI

Neighbor Area Radius A

B

SActive

R

R

SActive

SECTOR

Neighbor Area

Radius

R


5


Figure 5-8: Active and Neighbor Areas

To help visualize the true requirements of the situation, consider Figure 5-8. The sector labelled with 0 represents the sector of interest. The active area for this sector is depicted in yellow. Depicted in blue is all of the active area pertaining to the top 10 neighbors. (As with search window sizing, it is also recommended to be generous with neighbor lists.) Keep in mind that the blue area represents the neighbor area to which is being referenced. That is to say, areas where a subscriber might be looking for the offset of sector 0 even though it is well outside of the area where sector 0 is actively demodulated. By this means alone, the neighbor area represents an expansion of greater than 300% over the active area. If the next six most significant neighbors (sectors labelled 2) were included as neighbors, the neighbor area expands even further (area depicted in cyan). Note how both the front and back of sector 0 have neighbor search areas. The front is more pronounced while the back is affected mostly by the co-located sectors. (These neighbor relationships and subscriber locations are based on soft handoff relationships identified through CDMA static simulations for an ideal grid and uniform distribution.)

Estimates based on this perspective will prove more optimistic than those derived earlier since they account for the overlapping of cells and they better estimate the true neighbor area size.

01

1

1

1

1

1

1

11

12

2

2 22

2

Sector 0Top 10 Neighbors11 - 16 Neighbors


5


Note: To take advantage of sectorization, the planner must reuse offsets with the same orientation.

5.4.3 Incorrect Identification of an Offset by the Base Station

The CBSC (i.e. the XC subsystem) translates phase measurements to offsets by pooling them to the nearest valid offset based on its knowledge of PILOT_INC. The value of PILOT_INC can be set either at the CBSC level (system wide) or on a per sector/carrier level (as when using the Extended Range Cell feature). For correct identification, this process assumes that the PILOT_ARRIVAL component of the phase measurement never exceeds 1/2 of PILOT_INC. As a check on the selection of PILOT_INC, planners should ask whether or not locations exist within the system where subscribers may be active with a site at a distance greater than 1/2 of PILOT_INC. [Note: the process by which phase measurements are translated to offset indices is not specified by IS-95A/J-STD-008.

Figure 5-9: Phase Measurement Translations

Now, compare the relationship between SRCH_WIN_N and PILOT_INC. It is a rule that SRCH_WIN_N (and SRCH_WIN_R) never exceed PILOT_INC. The consequences of doing so are that the two adjacent windows would overlap. The BS may incorrectly identify the offset and

Table 5-7: Calculation of Reuse Distance(Assuming SNeighbor ≅ 2R, SActive ≅ 1R and Active Area Radius (A) ≅ 2.2R)

Front(F)

Back(B)

ReuseEquationa

a. The reuse equation is based on spatial relationships depicted in Figure 5-7. The Front range corresponds to the Neighbor Area Radius.

Reuse Distance

ClusterSize

Top 10 Neighbors - Sector 3.1 R 2.2 R F + SActive + Bb

b. Under these conditions, the back-side requirement for 2A + SNeighbor ≅ 6.4R would become limiting.

6.3 R 13

Expanded Neighbor List - Sector 4.3 R 2.2 R F + SActive + B 7.5 R 19

1 Tier - Omni 2.7 R - F + SNeighbor + A 6.9 R 16

2 Tier - Omni 4.4 R - F + SNeighbor + A 8.6 R 25

0 3 6 9 12

PILOT_INC = 3 = spacing between ‘valid’ offsets

0 3 6 9 12

pilot phase reported by subscriber in PSMM

SRCH_WIN_N


5


the subscriber may report multiple signals where only one is present. This guideline, easy to express and understand, is frequently the truly limiting factor on the lower bound for PILOT_INC (and conversely, the upper bound on cluster size). When situations arise where an area of the system requires very large search windows, so as to permit soft handoff between distant neighbors (as with extended range cells), the PILOT_INC must be resized large enough to accommodate the search window.

5.4.4 PILOT_INC and the Scan Rate of Remaining Set Pilots

The following information is provided to give insight to system optimizers and is based on Nokia Siemens Networks’ general understanding of subscriber vendor pilot scan algorithms. It is important to note that such algorithms are not specified through IS-95A/J-STD-008 and are, therefore, manufacturer specific.

As was noted in the definition of PILOT_INC, according to IS-95A/J-STD-008, the only impact of PILOT_INC on the subscriber is to influence the scanning rate of remaining set pilots. Please note that for optimum system performance, the scanning rate of remaining set pilots is notconsidered a dominant factor in determining the size of PILOT_INC. Remaining set pilots are at a distinct disadvantage over neighbor set pilots due to the scanning prioritization of pilot sets. For example, all active and candidate set pilots are scanned between scans of individual neighbor or remaining set pilots. All neighbor set pilots are scanned between scans of individual remaining set pilots. The scanning order is represented as follows for 3 active set pilots and 1 candidate set pilot [please remember that the actual scanning order is subscriber manufacturer specific]:

A1A2A3C1N1A1A2A3C1N2A1A2A3C1N3...A1A2A3C1NNA1A2A3C1R1A1A2A3C1N1...A1A2A3C1NNA1A2A3C1RNBegin again from the top.

A remaining set pilot is scanned N times slower than a neighbor set pilot where N is the number of remaining set pilots. In addition to their low scanning priority, IS-98 specifies no performance criteria for remaining set pilots.

Any remaining set pilot that appears strong enough (and long enough) to recommend promotion to the active set needs to be analyzed as part of the optimization process. Perhaps, it should be added to the neighbor list (or have its coverage adjusted). Feedback on these events can be derived from callproc logs in the pre-commercial phase and Call Detail Logs (CDL and vCDL) in the commercial phase.

Note: Since remaining set pilots are prioritized low and, currently, Nokia Siemens Networks does not honor requests to enter into soft handoff with a remaining set pilot, some operators have considered reducing SRCH_WIN_R to a minimum (i.e. 4 chips) and trading off the remaining set scan time for improved scan time on actives, candidates and neighbors. This is not recommended. The most significant reason is that the remaining set search window provides a means by which “truncated” neighbors can be recognized by the system. When in soft/softer handoff, a merging of neighbor lists take place. If the merge yields more than 20 neighbors, the subscriber limit of 20 neighbors requires that the list be truncated to only higher prioritized neighbors. Although these


5


neighbors may not be identified to the subscriber as neighbors, they nevertheless may be detected through a remaining set scan. The system will recognize and honor these remaining set pilot requests. A secondary motivation for permitting the windows to stay “open” is that they provide a means for optimizing neighbors lists by recognizing those sites which should be neighbors, but are not on the neighbor list. On the other hand, the improvement in the scan interval will only be modest on average (~6%).

5.4.5 Summary of Guidelines

The table below and the following text provide a summary of the PN offset planning guidelines.

Table 5-8: Summary of PN Offset Planning Guidelines

PILOT_INCComments

8a

a. The maximum Pilot Increment value is 15. Pilot Increment values larger than 8 may be required for some cells if the Extended Range Cell feature is used.

6 4 3 2 1

Cluster Size (3-sector) 19 25 37 52 76 148 co-offset

D/R (3-sector) 7.5 8.7 10.5 12.5 15.1 21.1 co-offset

Extra Sites (3-sector) 2 3 5 4 8 20 insurance

Cluster Size (6-sector) 9 13 19 25 37 76 co-offset

D/R (6-sector) 5.2 6.2 7.5 8.7 10.5 15.1 co-offset

Extra Sites (6-sector) 1 1 2 3 5 8 insurance

C/I (5km/10km)b

b. Refer to Section 5.4.1.2.

40.3/31.9

36.6/28.3

31.3/23.4

27.8/20.2

20.4/15.9

16.1/10.4

adjacent offset - PCS

C/I (8km/16km) 34.5/26.4

30.9/23.0

25.9/18.5

22.5/15.6

18.0/11.9

12.1/7.4

adjacent offset - 800

S (chips) 80 65 50 40 30 14 varies w/cell radius

PILOT_INC - S (chips)

432 319 206 152 98 50 adjacent offset

PILOT_INC - S (km) 105.4 77.8 50.2 37.1 23.9 12.2 adjacent offset

PILOT_INC (chips) 512 384 256 192 128 64 compare w/SRCH_WIN_N

PILOT_INC/2 (chips) 256 192 128 96 64 32 Neighbor Proximity Check?

PILOT_INC/2 (km) 62.5 46.8 31.2 23.4 15.6 7.8 Neighbor Proximity Check?

C/I 30 m S–( ) R⁄ 1+( )log=


5


To summarize the key guidelines for sectorized systems on sizing PILOT_INC are:

1. Minimum cluster size is 19 for 3-sector or 6-sector systems. Refer to Section 5.4.2 for details.

2. Maximum PILOT_INC is 8 for 3-sector and 4 for 6-sector. This correlates to the minimum cluster size.

3. For Suburban environments at 1900 MHz, minimum PILOT_INC is 3 (based on a minimum C/I threshold of 18.5 dB and unloaded carriers). This will serve the Urban/Dense Urban areas as well.

Note: Due to the approximate 9 dB difference between path loss at 1900 MHz and 800 MHz, PCS systems have smaller sites and consequently lower minimum PILOT_INC values.

4. For Suburban environments at 800 MHz, minimum PILOT_INC is 4 (based on a minimum C/I threshold of 18.5 dB and unloaded carriers). This will serve the Urban/Dense Urban areas as well.

5. PILOT_INC must be larger than the Neighbor and Remaining Set search windows, SRCH_WIN_N and SRCH_WIN_R. All timing differentials must be less than PILOT_INC/2. Carefully review the system design for any neighbors that are separated by more than PILOT_INC/2 since potentially these neighbors can generate sufficiently large timing differentials to cause translation errors (i.e. Neighbor Proximity Check). Refer to Section 5.4.3 for details.

6. To eliminate the potential for adjacent interference within a cluster, an adjacent offset and its neighbors should be separated from the potential interferer by a distance no greater than PILOT_INC - S. The distance PILOT_INC/2 is a safer limit (since S is a variable with an upper limit of PILOT_INC/2). This criteria can best be met by either co-locating the adjacent offsets within the same site or by assigning them to 1st tier neighbors. Refer to Section 5.4.1.1 for details.

7. If the system is truly characterized by Urban/Dense Urban environments, then smaller PILOT_INC values may be justified. If an entire CBSC is characterized by smaller radii, then that CBSC may have its PILOT_INC set lower.

8. Small sized trials are very easy to plan for. The largest PILOT_INC which will not require the trial system to have any reuse at all is suggested. Under these conditions, co-offset interference is non-existent and adjacent interference protection is maximized. If the PILOT_INC is selected to be a multiple of that which will ultimately be migrated to, then implementing changes in PILOT_INC later will not force a change to the sector level PN offset assignments.

9. Multiple carriers in a sector are all assigned the same PN offset.10. The implementation of CDMA at 1900 MHz is, generally, not tied to an already existing

analog base with its locations and antennas where significant cell splitting has taken place. The site grid should be more uniform than the mature analog counterpart. This should lend itself to a simpler repeat pattern implementation.

11. From a practical perspective, it should be understood that the majority of Nokia Siemens Networks systems that are commercial use a PILOT_INC in the range of 2 to 4. The


5


systems using a PILOT_INC of 2 can be characterized as possessing small radius sites. The systems employing a PILOT_INC of 4 can be characterized as possessing some areas of extensive propagation (water, mountains) that have required resizing SRCH_WIN_N, and consequently PILOT_INC, larger. With the introduction of the Extended Range Cell feature the maximum PILOT_INC supported is now 15.

5.4.6 Guidelines for Assigning Offsets

It has already been explained that there should be a goal for locating adjacent offsets close to each other. In the figure below, the Adjacent Sectors configuration shows co-located sectors containing adjacent offsets. This represents the absolute limit on how close adjacent offsets can be located. Under these conditions, two-thirds of all adjacent assignments (for 3 sector sites) will have reduced the time differential to zero. For the remaining third, the adjacent offset is located in an adjacent site. This approach also has the benefit of easy recognition of co-located sectors during system optimization.

Figure 5-10: Adjacent Sector and Adjacent Site Offset Assignment Approaches

Previously, this has been the only recommendation. There is now an alternative recommendation, Adjacent Sites, which locates all adjacent offset assignments within adjacent sites (and not within adjacent sectors of the same site). The Adjacent Sites approach has Offset Groupings associated with it that are found in Table 5-9 and Table 5-10. Although this represents a slight compromise with respect to the timing margin of the Adjacent Sectors configuration, there are several characteristics with this approach that make it worth recommending:

• Virtually all adjacent offsets possess the same antenna orientation (as co-offsets normally do). This provides an additional measure of interference protection and simplifies system optimization.

• A uniform increment of 168 exists between co-located sectors regardless of the PILOT_INC in use. This will help optimization through easier recognition of co-site offsets. (The Adjacent Sectors approach also benefits from easy recognition of co-site offsets.)

ADJACENT SECTORS ADJACENT SITES

6

3

9

12

15

18

6174

342

9

177

345


5


• A 3-sector site uses one group while a 6-sector site uses 2 groups. (The Adjacent Sectorsapproach possesses this benefit as well.)

• Table 5-9 contains 84 groupings for a PILOT_INC of 2. Subsets of this table apply to PILOT_INC values of 4 (42 sets), 6 (28 sets), 8 (21 sets), and 12 (14 sets). These groupings will prove useful in any transition or migration between different PILOT_INCs.

• Table 5-10 contains 56 groupings for a PILOT_INC of 3. Subsets of this table apply to PILOT_INC values of 6 (28 sets) and 12 (14 sets). These groupings will prove useful in any transition or migration between different PILOT_INCs.

Generic information on reuse patterns can be found in Section 5.5. Here are some possible cluster configurations:

Table 5-9: Offset Groupings for PILOT_INC = 2 (also 4, 6, 8, and 12)

Alpha 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 32 34 36 38 40 42

Beta 170 172 174 176 178 180 182 184 186 188 190 192 194 196 198 200 202 204 206 208 210

Gamma 338 340 342 344 346 348 350 352 354 356 358 360 362 364 366 368 370 372 374 376 378

Alpha 44 46 48 50 52 54 56 58 60 62 64 66 68 70 72 74 76 78 80 82 84

Beta 212 214 216 218 220 222 224 226 228 230 232 234 236 238 240 242 244 246 248 250 252

Gamma 380 382 384 386 388 390 392 394 396 398 400 402 404 406 408 410 412 414 416 418 420

Alpha 86 88 90 92 94 96 98 100 102 104 106 108 110 112 114 116 118 120 122 124 126

Beta 254 256 258 260 262 264 266 268 270 272 274 276 278 280 282 284 286 288 290 292 294

Gamma 422 424 426 428 430 432 434 436 438 440 442 444 446 448 450 452 454 456 458 460 462

Alpha 128 130 132 134 136 138 140 142 144 146 148 150 152 154 156 158 160 162 164 166 168

Beta 296 298 300 302 304 306 308 310 312 314 316 318 320 322 324 326 328 330 332 334 336

Gamma 464 466 468 470 472 474 476 478 480 482 484 486 488 490 492 494 496 498 500 502 504

Table 5-10: Offset Groupings for PILOT_INC = 3 (also 6 and 12)

Alpha 3 6 9 12 15 18 21 24 27 30 33 36 39 42 45 48 51 54 57 60 63

Beta 171 174 177 180 183 186 189 192 195 198 201 204 207 210 213 216 219 222 225 228 231

Gamma 339 342 345 348 351 354 357 360 363 366 369 372 375 378 381 384 387 390 393 396 399

Alpha 66 69 72 75 78 81 84 87 90 93 96 99 102 105 108 111 114 117 120 123 126

Beta 234 237 240 243 246 249 252 255 258 261 264 267 270 273 276 279 282 285 288 291 294

Gamma 402 405 408 411 414 417 420 423 426 429 432 435 438 441 444 447 450 453 456 459 462

Alpha 129 132 135 138 141 144 147 150 153 156 159 162 165 168

Beta 297 300 303 306 309 312 315 318 321 324 327 330 333 336

Gamma 465 468 471 474 477 480 483 486 489 492 495 498 501 504


5


The 19-cell repeat pattern is easy to use. A site number, N, within the pattern can easily be translated into a PN offset assignment for a particular sector.

For Adjacent Sectors, 6-sector, and PILOT_INC of 4: SECTOR x OFFSET= ((N-1) * 6 + x) * 4

For Adjacent Sites, 6-sector, and PILOT_INC of 4: SECTOR x OFFSET= N*4 + (x-1)*168; (x = 1,2,3) SECTOR x OFFSET= (N+21)*4 + (x-4)*168; (x = 4,5,6)



For Adjacent Sites, 6-sector, and PILOT_INC of 3: SECTOR x OFFSET= N*3 + (x-1)*168; (x = 1,2,3) SECTOR x OFFSET= (N+28)*3 + (x-4)*168; (x = 4,5,6)



For Adjacent Sites, 3-sector, and PILOT_INC of 4: SECTOR x OFFSET= N*4 + (x-1)*168; (x = 1,2,3)

17 9 1 12 4

10 2 13 5

16 8 19 11

3 14 6

15 7 18

3

10

17614

4 8 13 19 22

7 14 18 23

3 9 12 20

15 17 24

2 10 11

1 6 15

11 16 25

24

25 16

21 21

22

5

6

4

5

1

2

23 12 1 27 16

13 2 28 17

22 11 37 26

3 29 18

21 10 36

20 9 35

4 30 19

31

32 25

33 15

34

24

14

5

6

7

8


5




For Adjacent Sites, 3-sector, and PILOT_INC of 3: SECTOR x OFFSET= N*3 + (x-1)*168; (x = 1,2,3)

Note - Larger values of PILOT_INC (up to 15) may be deployed in support of Extended Range Cells. In these situations, the number of PN values available for reuse patterns may be restricted (see Section 5.4.7 below).

5.4.7 Guidelines for Planning Inter-CBSC and Intra-CBSC multiple PILOT_INC Boundaries and Transition Zones.

CDMA sites of similar coverage radius that are in close proximity to each other are normally assigned a common PILOT_INC. This allows the application of PN Offset planning rules given above to be used for an orderly assignment of PN Offsets to each sector. Maintaining one PILOT_INC for the sites associated with a CBSC used to be mandatory and is still a common implementation. Intra-CBSC multiple PILOT_INC boundaries only exist when the Extended Range Cell feature is utilized.

For a given system there may exist situations where maintaining the same PILOT_INC for all sites may not be optimum. One situation that may be encountered is when two adjacent CBSCs that share a coverage boundary are deployed using different PILOT_INC values.

A second situation encountered is when the coverage area of a CBSC requires more than one PILOT_INC. This is done to accommodate an outlying coverage region that has sites (such as Extended Range Cells) that provide significantly greater cell coverage radius than the inner sites. The multiple PILOT_INC is based on the Extended Range Cell feature introduced in R18. With Extended Range Cells, certain BTS hardware needs to be in place (e.g. BBX1X, MCC24 or MCC1X)). The inner sites in this situation would be assigned smaller PILOT_INC values than the outlying coverage region sites.

21 47 1 27 7

48 2 28 8

20 46 26 52

3 29 9

19 45 25

18 44 24

4 30 10

38

39 51

40 6

41

22

49

33

34

35

36

37 17 4311

12

23

42

16

15

14

13

32

315

50

5

10

12

16

23

31

36

50

42


5


Both of these situations result in a boundary between coverage regions that are functioning with different PILOT_INC values. The CBSC translates the reported PN phases to PN offsets using the Pilot Increment of sectors in the active set. The actual PILOT_INC applied is based on the strongest reported pilot in the most recent Pilot Strength Measurement Message.

The translation of PN phases to PN offsets will have the following assumptions:

1. The neighbors of the active sectors have PN offsets in multiples of Pilot Increment associated with active sectors.

2. The sectors in the active set and all their respective neighbors are placed such that delay of Pilot Increment (of the active sector) /2 chips between active set sectors and their neighbors is not exceeded.

Figure 5-11: No Transition Zone - PILOT_INC Boundary

The challenge in transitioning the boundary between two coverage areas with differing PILOT_INC is characterized by subscribers in the coverage region with the larger PILOT_INC seeing a site from the coverage region with the smaller PILOT_INC. The CBSC may not interpret the phase correctly for the reported site because a different PILOT_INC is in use. For the example in Figure 5-11, Coverage Region A is using a PILOT_INC of 3 and Coverage Region B is using a PILOT_INC of 6. A subscriber (X) tied to Coverage Region B sees a Coverage Region A site using offset 39 and reports it in a PSMM. Coverage Region B will interpret the offset as either 36 or 42, because it does not recognize 39. This problem does not manifest itself in the other direction since all multiples of 6 are already multiples of 3.

Coverage Region A

PILOT_INC = 3

Coverage Region B

PILOT_INC = 6

multiple of 6multiple of 3

90

108

10296

120114

126138

132

3

69

237

240243

39

4245

1218

15

246252

249

4854

51

X

Y


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Due to the Pilot Increment setting at the sector-carrier level, there will be areas of transition between one region/group of sectors and next having different Pilot Increment values. This will require special PN offset planning and TCH window size calculations by the operator.

The following guidelines should be followed in selecting the PN Offset values along the boundaries of such regions:

Figure 5-12: Transition Zone - PILOT_INC Boundary

• A transition zone should be planned on the boundary between the region with the smaller Pilot_Inc and the region with the larger Pilot_Inc. This is done to facilitate the transition by the mobile from one Pilot_Inc setting to the other.

• Within this transition zone the Pilot Offsets must be selected so that they can be resolved correctly by the CBSC regardless of which of the two Pilot_Inc values is/are in use. This can be achieved by ensuring that each Pilot Offset in the transition zone is divisible by the lowest common multiple of the two Pilot_Inc values.

• The spacing of the sites within the transition zone must be such that the maximum dis-tance of each sector from its neighbors does not exceed a delay of Pilot_Inc / 2 chips, calculated using the lower of the two Pilot_Inc values present along the boundary

• The same PN offset planning rules apply for both inter and intra CBSC handoffs.


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The Pilot Increment parameter is also used by the mobile, to determine which pilots to scan from among the remaining set. Only valid pilots (i.e. those pilots that are multiples of Pilot Increment) will be scanned by the mobile. For the mobile, Pilot Increment impacts only the scanning rate applied to remaining pilots. It accomplishes this by reducing the number of remaining pilots that need to be scanned.

5.5 Reuse Patterns

This table can help in defining reuse patterns through use of i & j coordinates. For example, to create a normal analog 7 cell reuse pattern, follow along the i axis for 2 cells and then follow the j axis (either clockwise or counter-clockwise, but be consistent) for 1 cell. The grayed out table elements pertain to cluster sizes not likely to be used in CDMA.

Table 5-11: Reuse Pattern Coordinates, i & j,and Cluster Size, N, and D/R

i j N D/R i j N D/R1 0 1 1.73 7 1 57 13.08

1 1 3 3.00 5 4 61(4 ring)

13.53

2 0 4 3.46 6 3 63 13.75

2 1 7(1 ring)

4.58 8 0 64 13.86

3 0 9 5.20 7 2 67 14.18

2 2 12 6.00 8 1 73 14.80

3 1 13 6.24 5 5 75 15.00

4 0 16 6.93 6 4 76 15.10

3 2 19(2 ring)

7.55 7 3 79 15.39

4 1 21 7.94 8 2 84 15.87

5 0 25 8.66 6 5 91(5 ring)

16.52

3 3 27 9.00 7 4 93 16.70

4 2 28 9.17 8 3 97 17.06

5 1 31 9.64 6 6 108 18.00

6 0 36 10.39 7 5 109 18.08

4 3 37(3 ring)

10.54 8 4 112 18.33

5 2 39 10.82 7 6 127(6 ring)

19.52

6 1 43 11.36 8 5 129 19.67

4 4 48 12.00 7 7 147 21.00

5 3 49 12.12 8 6 148 21.07

7 0 49 12.12 8 7 169(7 ring)

22.52

6 2 52 12.49 8 8 192 24.00


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The following illustrates the i & j coordinates for a 19-cell repeat pattern (e.g. N = 19) with i = 3 and j = 2 with a clockwise rotation.

Figure 5-13: i=3,j=2 Repeat Pattern

5.6 References

Prior discussions of topics significant to PN Offset Planning which are useful references include the following:

1. TIA/EIA/IS-95A, Mobile Station-Base Station Compatibility Standard for Dual-Mode Wideband Spread Spectrum Cellular System, version 0.07, §6.1.5.1, §6.6.6.1.2, §6.6.6.2.1, §6.6.6.2.4.

2. Qualcomm, “The CDMA Network Engineering Handbook”, March 1, 1993, §9.1.1, §9.2.3, §9.4.

3. Scott M. Hall (Motorola), “Simple CDMA PN Search Windows”, January 5, 1995.

IEEE Conference Papers on this topic include:

4. Chu Rui Chang, Jane Zhen Wan and Meng F. Lee (NORTEL Wireless Engineering Services), “PN offset planning strategies for non-uniform CDMA networks”, 1997 IEEE 47th Vehicular Technology Conference, May 4-7, 1997.

5. Jin Yang, Derek Bao and Mo Ali (Airtouch Cellular), “PN offset planning in IS-95 based CDMA systems”, 1997 IEEE 47th Vehicular Technology Conference, May 4-7, 1997.


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6 RF Antenna
Chapter
6

Table of Contents

Systems

6 RF Antenna Systems

6.1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 3

6.2 CDMA Cell Site Antenna Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 36.2.1 Antenna Type . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 36.2.2 Antenna Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 46.2.3 Antenna Beamwidth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 66.2.4 Voltage Standing Wave Ratio . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 66.2.5 Return Loss. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 66.2.6 Power Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 76.2.7 Front to Back Ratio . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 76.2.8 Side Lobes & Back Lobes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 76.2.9 Antenna Downtilting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 86.2.10 Antenna Height. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 8

6.3 CDMA Antenna Placement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 96.3.1 Antenna Isolation Considerations. . . . . . . . . . . . . . . . . . . . . . . . . 6 - 9

6.3.1.1 CDMA/AMPS Transmit/Receive Antenna Isolation Requirements 6 - 11

6.3.1.2 Measuring Port-to-Port Antenna Isolation . . . . . . . . . . . . . . 6 - 136.3.1.3 Reducing the Required Antenna Isolation . . . . . . . . . . . . . . 6 - 136.3.1.4 Typical Antenna Isolation. . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 146.3.1.5 CDMA Antenna Placement . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 14

6.3.2 Antenna Diversity (Spacial) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 166.3.2.1 Horizontal Antenna Diversity and Recommended Separation 6 - 166.3.2.2 Vertical Antenna Diversity . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 17

6.4 CDMA Antenna Sharing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 176.4.1 Multiple Frame Antenna Sharing with 800 MHz BTS Products . 6 - 17

6.4.1.1 SC4812T/ET Antenna Sharing . . . . . . . . . . . . . . . . . . . . . . . 6 - 17 6.4.1.2 UBS-Macro Antenna Sharing . . . . . . . . . . . . . . . . . . . . . . . . 6 - 20

6.4.2 Multiple Carrier Cavity Combining With 1900 MHz BTS Products 6 - 196.4.2.1 Output Power With Combining . . . . . . . . . . . . . . . . . . . . . . 6 - 196.4.2.2 Type of Combining. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 196.4.2.3 Multiple Carrier Scenarios . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 20

6.4.3 Duplexing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 216.4.3.1 Pre-Engineered Kits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 226.4.3.2 Duplexers and Intermodulation . . . . . . . . . . . . . . . . . . . . . . 6 - 226.4.3.3 Proper Installation and Maintenance of Duplexed Antennas 6 - 25


6

CDMA/CDMA2000 1X RF Planning Guide6 RF Antenna Systems

6.5 CDMA Antenna Sharing With Other Technologies . . . . . . . . . . . . . . . 6 - 296.5.1 CDMA/Analog Shared Facilities . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 29

6.5.1.1 Common Transmit Antenna . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 306.5.1.2 Common Receive Antenna(s) . . . . . . . . . . . . . . . . . . . . . . . . 6 - 30

6.5.2 Duplexed AMPS/CDMA Antennas . . . . . . . . . . . . . . . . . . . . . . . 6 - 31

6.6 GPS Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 33

6.7 Ancillary Antenna System Components . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 336.7.1 Directional Couplers. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 336.7.2 Surge (Lightning) Protectors . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 336.7.3 Transmission Line . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 34

6.7.3.1 RF Performance of Transmission Lines . . . . . . . . . . . . . . . . 6 - 346.7.3.2 Physical Characteristics. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 356.7.3.3 Choice of Transmission Line. . . . . . . . . . . . . . . . . . . . . . . . . 6 - 35

6.7.4 Transition Feeder Cables (Jumper Cables). . . . . . . . . . . . . . . . . . 6 - 36

6.8 RF Diagnostic System. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 - 36


6


6.1 Introduction

This chapter will outline RF engineering considerations that should be incorporated into the design of CDMA "antenna systems". The antenna system is defined as those elements between the BTS equipment cabinet (top of rack) and the Tx or Rx antenna. A detailed discussion of the various available equipment and antenna sharing configurations and requirements are discussed, including those involving co-location with other technologies, duplexing, and multiple carrier combining.

The guidelines below are intended to assure the most efficient implementation of Nokia Siemens Networks’ CDMA system while minimizing the risk to other fixed and mobile radio operators.

6.2 CDMA Cell Site Antenna Parameters

This section of the document will outline the main antenna parameters that the system engineer should consider when choosing the optimum antenna to be used in a CDMA system. Guidelines are provided where possible, although it is recognized that a number of issues are beyond the scope of this document and may require site specific engineering.

6.2.1 Antenna Type

If separate omni-directional type transmit antennas are to be used for the CDMA system (e.g. no antenna sharing), a type similar to those used for other cellular technologies, such as AMPS or GSM, can be used, obviously dependent on the required antenna operating frequency specifications.

The same convention is basically held for sector type directional CDMA antennas, with the exception of the consideration of desired beamwidths. Typically, antennas with narrower horizontal beamwidths than their AMPS or GSM supporting counterparts are used for CDMA to help limit noise contribution to adjacent sectors. As a result, suitable antenna types should be chosen if the CDMA system being installed is not to share antennas currently existing at the site.

Sufficient isolation between CDMA antennas and other existing antennas at the site should be readily obtained. Considering that physical separation between co-located antennas may be required to assist in achieving this isolation, physically smaller antenna types may be required to allow for proper installation on the tower.

In general, the log-periodic reflector type directional antennas have smaller height and width dimensions for the same forward gain than dipole panel antennas or collinear dipole reflector type antennas. They, of course, have a larger dimension in the direction of maximum gain due to the length of the log-periodic array(s) which form the overall antenna system. Because of the smaller area occupied on the face of the tower or its platform, it should be possible to fit at least seven of these antennas in the same space originally allocated for the AMPS sector antennas.

Log-periodic reflector type antennas also appear to have excellent front-to-back and front-to-side ratios. It appears that the isolation between adjacent antennas is significantly higher than for dipole type directional antennas. This is based on measured data taken by Allgon System AB on their line


6


of log-periodic reflector antennas. This provides the same isolation with closer spacing than for comparable gain panel antennas or greater isolation for the same spacing.

Special consideration should be given to the antenna bandwidth. If the use of duplexers is required then a wideband antenna capable of supporting the primary and the secondary CDMA carriers should be selected (see tables below)

Refer to Chapter 2 concerning other frequency bands that might be utilized.

6.2.2 Antenna Gain

This is often referred to as "power gain" and is the ratio of the maximum radiation in a given direction to that of a reference antenna in the same direction for equal power input. Usually this gain is referenced to either an isotropic antenna or a half wave dipole in free space at 0° elevation.

Table 6-1: CDMA Carrier Frequency Range

Frequency Band

Primary CDMA Carrier - Center Channel

(& Broadband Channel Range)

Frequency Range in MHz (Base Rx/Tx)

A 283 (263-303) 832.89-834.09 / 877.89-879.09

B 384 (364-404) 835.92-837.12 / 880.92-882.12

Frequency Band

Secondary CDMA Carrier - Center Channel

(& Broadband Channel Range)

Frequency Range in MHz (Base Rx/Tx)

A 691 (671-711) 845.13-846.33 / 890.13-891.33

B 777 (757-797) 847.71-848.91 / 892.71-893.91

Table 6-2: PCS Carrier Frequency Range

Frequency BandFrequency Range in MHz

(Base Rx/Tx)

A 1850-1865/1930-1945

D 1865-1870/1945-1950

B 1870-1885/1950-1965

E 1885-1890/1965-1970

F 1890-1895/1970-1975

C 1895-1910/1975-1990

G 1910-1915/1990-1995


6


An isotropic reference (dBi) generally pertains to a theoretical antenna having a spherical radiation pattern with equal gain in all directions. When used as a gain reference, the isotropic antenna has a power of 0 dBi. The halfwave dipole (dBd) is an antenna which is center fed as to have equal current distribution in both halves. When used as a theoretical reference antenna it has a power gain of 0 dBd, which equates to a 2.14 dB difference compared to an Isotropic antenna. For a graphical representation of the different antenna patterns, please refer to the following figure.

dBi = dBd + 2.14 dBd = dBi - 2.14

Figure 6-1: dBd vs. dBi

The gain of the antenna will impact other antenna characteristics such as: size, weight, horizontal beamwidth, vertical beamwidth, cost. The RF Engineer will need to select the appropriate antenna for the particular situation. A trade-off will need to be made by the RF Engineer as to whether a higher gain or lower gain antenna should be chosen. The higher gain antenna typically is physically larger, more expensive and has a narrower vertical beamwidth than would a lower gain antenna.

The gain of an antenna has a direct interaction with other antenna parameters, (the technical depth of which is beyond the scope of this document). The following paragraphs will provide the system engineer with general guidelines:

Vertical Beamwidth - Generally, the greater the gain of the antenna, the narrower the vertical beamwidth. The vertical beam can be used to focus coverage in some circumstances, but the engineer should ensure that the optimum vertical beamwidth is used to prevent the creation of "nulls" or coverage holes near to the site.

Physical Size - The size of an antenna will generally be greater as an antenna gain increases. This is due to the greater number of dipole array and electrical elements required to reach the desired gain. The system engineer should remember that PCS frequencies are approximately half the wavelength of 800 MHz and therefore the antennas will typically be smaller for a common gain.


6


6.2.3 Antenna Beamwidth

Antenna beamwidth is measured in degrees between the half power points (3 dB) of the major lobe of the antenna. Beamwidth can be expressed in terms of azimuth (horizontal or H-plane) and elevation (vertical or E-plane).

The predominant type of antenna configuration within urban areas will be three sectored. This implies that each sector should utilize an antenna with 120° horizontal beamwidth; however, it has been found through simulation that the use of 120° antennas provide too much overlap. As the coverage of any sector within a CDMA system is directly affected by the noise generated by its neighboring sectors and traffic within those sectors, the use of 120° can lead to reduced coverage area through the rise in system noise. The excessive overlap of sectors can also lead to increased softer handoff and therefore the reduction of call processing capability.

If narrow horizontal beamwidth antennas are used, for example 60°, simulation has shown that insufficient coverage (i.e. coverage holes) can exist between adjacent sectors. The use of 60° high gain antennas can also restrict the vertical beamwidth and can lead to coverage nulls close to the cell site.

From current simulation, the optimum horizontal antenna beamwidth for PCS systems has been found to be between 90° and 100°. This beamwidth has been proven to minimize softer handoff while providing adequate coverage. However, before choosing an antenna of this beamwidth, the system engineer should ensure that all factors outlined within this "Antenna Parameters" section have been identified.

6.2.4 Voltage Standing Wave Ratio

Voltage Standing Wave Ratio (VSWR) is another parameter used to describe an antenna performance. It deals with the impedance match of the antenna feed point to the feed or transmission line. The antenna input impedance establishes a load on the transmission line as well as on the radio link transmitter and receiver. To have RF energy produced by the transmitter radiated with minimum loss or the energy picked up by the antenna passed to the receiver with minimum loss, the input or base impedance of the antenna must be matched to the characteristics of the transmission line. The VSWR of a CDMA antenna should be less than 1.5:1.

6.2.5 Return Loss

Return Loss (RL) is the decibel difference between the power incident upon a mismatched continuity and the power reflected from that discontinuity. Return loss is related to the reflection coefficient (p) and VSWR as follows;

RLdB = 20 log (1/p)Where p = (VSWR-1)/(VSWR+1)VSWR = Vmax/Vmin


6


In other words, the return loss of an antenna can be considered as the difference in power in the forward and reverse directions due to impedance mismatches in the antenna design.

All other things being equal, the higher the antenna return loss, the better the antenna. The system engineer should choose an antenna with a return loss of 14 dB or better. Note that 14 dB corresponds to a VSWR of 1.5:1 as per the following example;

VSWR = 1.5/1 = 1.5p = (1.5-1)/(1.5+1) = 0.5/2.5 = 0.2RLdB = 20log (1/0.2)RLdB = 13.979 dB

6.2.6 Power Rating

The Power Rating of an antenna is the maximum power, normally expressed in Watts that the antenna will pass without degraded performance. Typical values for the power rating of an antenna are between 300 and 500 Watts. As CDMA will employ a smaller number of carriers and due to the losses associated with combining, the power rating of an antenna is not expected to be a limiting factor for antenna choice. Even so, when choosing an antenna, the system engineer should consider system expansion and the theoretical maximum configuration of carriers that could be placed onto a single antenna (please refer to Section 6.4.2).

6.2.7 Front to Back Ratio

The front to back ratio of an antenna is an important measure of performance. It is the ratio of the power radiated from the main ray beam forward to that radiated from the back lobe behind the antenna. Front to back ratio is normally expressed in terms of dB. This means that a signal at the back of the antenna should be X dB down on a signal at a mirror angle in front of the antenna. The front to back ratio for a typical CDMA antenna should be in the region of 25 dB.

6.2.8 Side Lobes & Back Lobes

Side and Back lobes are those undesirable directions where the chosen "directional" antenna may present gain. The system engineer should pay particular attention to these characteristics when downtilting an antenna, the mechanical downtilting of an antenna will directly affect the radiation of both side and back lobes. The mounting of panel antennas on buildings or the use of antenna with electronic down/up tilt are two possible ways to limit back lobe interference.

The system engineer should choose the optimum directivity and gain of an antenna while limiting the number of side lobes and the strength of the back lobe (refer to previous paragraph - front to back ratio).


6


6.2.9 Antenna Downtilting

Downtilting is the method of effectively adjusting the vertical radiation pattern of the antenna to direct the main energy more downwards and reduce the energy directed towards the horizon. Downtilting can be used to increase the amount of coverage close to the site where "nulls" (holes) may exist due to the effective height of the antenna. Downtilting can also be used to reduce "pilot pollution" caused by reflections or undesired RF propagation beyond a predetermined footprint. There are principally two types of antenna downtilting possible, mechanical and electronic.

Mechanical downtilting can be achieved through the mechanical adjustment of an antenna’s physical position. The main advantage of the mechanical type of downtilting is the ease (dependent upon location) of mechanically adjusting the antenna’s direction following system optimization. Note that any CDMA network will require some degree of system optimization based upon site specific variables. The adjustment of antenna downtilt has historically been one of the principle methods of tuning system performance, therefore the system engineer should consider if the chosen antenna can be downtilted and if so, by how much?

The second method of downtilting that can be used is electronic downtilt. This is the only way to implement downtilt for an omni directional antenna. The level of electronic downtilt for an antenna can be preset and ordered directly from the antenna manufacturer. The system engineer should be aware that electronic antenna downtilt is preset. Thus, the field adjustment of downtilt and therefore vertical radiation can not normally be reduced. There are antenna suppliers that provide the capability of being able to alter the downtilt characteristics of the antenna from the base of the cell site. This may take the form of motors to perform the physical downtilt or electronics used to alter the electrical characteristics of the antenna. Refer to the numerous antenna vendors for the various antennas that they supply.

The system engineer should also remember that the amount of gain in the antenna will also have a direct affect both on the physical size of the antenna and the vertical beamwidth. If a low gain antenna is utilized, the vertical beamwidth will be relatively broad and therefore the benefits of downtilting will be minimal.

6.2.10 Antenna Height

In general the 6 dB per octave rule will apply to the cell site antenna height in a flat terrain, that is doubling the antenna height causes a gain increase of 6 dB. The system engineer should compare this possible gain height increase with the effects of doubling the transmission line loss and the possible appearance of nulls close to the site.

Figure 6-2 shows the comparative number of cell sites required for a given area based upon differing base station antenna heights and the Cost-231 Hata propagation model (i.e. flat terrain only). If 100 ft. (30 m) is considered as the reference point, the system engineer should note that by doubling the antenna height to 200 ft., there is a reduction of 50% in cell sites required.


6


Figure 6-2: The Relationship of Antenna Height to Number of Cell Sites.

6.3 CDMA Antenna Placement

The placement of required CDMA antennas will typically depend on two main factors:

• the isolation required between the CDMA antennas to be installed and other antennas existing at the site

• the amount of spatial diversity provided between CDMA Rx antennas.

It is important that enough physical separation be used between affected antennas to ensure the best possible performance of the CDMA BTS while minimizing the threat of interference to/from other co-located technologies. The following sections discuss the above considerations in more detail.

6.3.1 Antenna Isolation Considerations

The following recommendations are general guidelines on the base station antenna isolation required between two or more of the following radio systems:

• 800 MHz AMPS• 800 MHz CDMA• 1900 MHz CDMA PCS


6


Typical examples of site sharing are an 800 MHz CDMA system overlayed on an existing 800 MHz AMPS system, or a 1900 MHz CDMA PCS system sharing the same tower/rooftop with an existing 800 MHz AMPS/CDMA system. This section describes the RF isolation requirements between the various transmit and receive antennas of two or more of the above radio systems which share a common tower/platform/rooftop location. The following antenna isolation scenarios need to be considered.

Tx to Tx Antenna Isolation: There must be sufficient isolation between any two transmit antennas to attenuate the signals from one antenna sufficiently before they enter another transmit antenna and create transmitter IM products in the associated transmitters that are strong enough to cause a problem for the system.

Rx to Rx Antenna Isolation: For adequate receive diversity performance there must be sufficient spacing between the two antennas to achieve the desired degree of de-correlation of the two receiver feeds for the signals being received.

Tx to Rx Antenna Isolation: The isolation between the transmit and receive antennas at a cell site must be high enough to provide sufficient attenuation to eliminate the following three potential problems:

1. Receiver overload caused by the high level transmit carriers being picked up by the receive antennas and causing receiver desensitization and/or generating IM or cross- modulation products within the receiver which interfere with the reception of the desired signals.

2. Interference with the reception of the desired signals caused by transmitter sideband noise and/or spurious signals generated in the transmitter which fall in the receive band and whose energy is radiated from the transmit antennas and picked up by the receive antennas.

3. Interference with the reception of the desired signals caused by transmit IM products falling in the receive band that are generated in the transmit antenna systems consisting of feed line and jumper connectors and/or the transmit antennas themselves. These IM products are produced after the transmitter output filtering and therefore cannot be eliminated by any transmitter filtering. These IM products will be radiated by the transmit antennas and picked up by the receive antennas.

Also included in this section are several antenna placement examples as well as a discussion of some typical isolations that can be expected between various combinations of 800 MHz and 1900 MHz antennas.

Additional base station antenna isolation requirements, involving scenarios such as the co-location of 800 MHz CDMA and TACS antennas, the co-location of DCS 1800 and PCS 1900 CDMA antenna and the co-location of PCS 1900 CDMA and microwave antennas, are considered in Chapter 9.


6


6.3.1.1 CDMA/AMPS Transmit/Receive Antenna Isolation Requirements

The following sections provide the calculations for antenna isolation requirements.

800/1900 MHz Tx-Tx ANTENNA ISOLATION

CDMA Tx - CDMA TxThe maximum Tx reverse signal that can be applied to a BTS Tx port is +30 dBm (1 Watt). A typical high power LPA can deliver +50 dBm (100 Watts) to the antenna system. Taking into consideration the coupling from the adjacent sectors, the minimum antenna-to-antenna isolation should be:

50 dBm + 3 dB - 30 dBm = 23 dB

Since the minimum AMPS transmit antenna-to-antenna isolation is typically 20 dB, the worst case antenna isolation required between any AMPS and CDMA transmit antenna combination will be chosen to be 23 dB. (This applies to both 800 and 1900 MHz transmit antennas.)

800/1900 MHz Rx-Rx ANTENNA ISOLATION

A minimum isolation of 20 dB is desired between any two antennas. This would apply to separate AMPS and CDMA receive antennas mounted in close proximity to each other. When evaluating two receive antennas connected to the same BTS for diversity reception, a more important factor is the spatial separation of the two antennas. If their responses are uncorrelated to fading, good diversity reception is assured. (According to Lee, William C.Y. in “Mobile Cellular Telecommunications Systems”, uncorrelated antennas require from 8 to 14 wavelengths of horizontal separation. This equates to about 3 to 5 meters at 800 MHz or about half that much at 1900 MHz.) The internal requirement of the BTS is 20 dB isolation, so the antenna system need only be 20 dB also. The physical spacing required for spatial separation greatly exceeds 20 dB of isolation between the two receive antennas.

800/1900 MHz Tx-Rx ANTENNA ISOLATION

In Cases 1 through 3 below, Transmit to Receive Antenna Isolation requirements are estimated based on reducing transmitter noise and spurs in the receive band to the point where only 0.5 dB of receiver noise floor rise or receiver threshold sensitivity is produced. If either more or less degradation is tolerable, the information given in Table 6-3 can be used to modify them as desired. Similarly, if specific information as to the transmitter noise and spurious signal levels for a particular Base Station model of interest is known, Cases 1 through 3 can be used as a guide.

Table 6-3: Degradation to Sensitivity Based on Noise Level Below kTBF

Noise level below kTBF Degradation to sensitivity16 dB 0.1 dB13 dB 0.2 dB9 dB 0.5 dB6 dB 1.0 dB


6


a: The added noise at this level is equal to kTBF

Case 1: CDMA Tx - CDMA Rx

From Table 6-3, a 0.5 dB sensitivity degradation occurs when the transmitter noise is at a level of 9 dB below kTBF. For a CDMA receiver with a Noise Figure of 4 dB, kTBF is -109 dBm. This results in a maximum acceptable interference power of -118 dBm.

Typical CDMA Tx noise level due to CDMA spurs (CDMA Tx IM) in the receive band is less than -85 dBm in a 1 MHz bandwidth. In the CDMA receiver bandwidth of 1.2288 MHz this is -84 dBm. The resulting antenna-to-antenna isolation requirement for 0.5 dB sensitivity degradation is:

-84 dBm - (-118 dBm) = 34 dB

Case 2: AMPS Tx - CDMA Rx

The AMPS Tx specification requires the AMPS Rx band spurs to be at a maximum level of -90 dBm/30 kHz. The total Tx SBN and spurs in the CDMA Rx band is maintained at -85 dBm/1 MHz with proper frequency planning (no 3rd order IM inside CDMA Rx). The resulting antenna-to-antenna isolation requirement for a 0.5 dB degradation is:

-84 dBm - (-118 dBm) = 34 dB

For a multitone LPA application, the worst case Tx SBN measured in the Rx band should be less than -85 dBm/1 MHz.

Case 3: CDMA Tx - AMPS Rx

Typical CDMA Tx noise level due to CDMA spurs (CDMA Tx IM) in the receive band is less than -85 dBm in a bandwidth of 1 MHz. This is -100 dBm in the AMPS receiver bandwidth of 30 kHz. The kTBF for a typical AMPS receiver is -123 dBm. Using Table 6-3, 0.5 dB sensitivity degradation occurs when the Transmitter noise is 9 dB below kTBF, which is -132 dBm in this case. The resulting antenna-to-antenna isolation requirement for 0.5 dB sensitivity degradation is:

-100 dBm - (-132 dBm) = 32 dB

The worst case AMPS or CDMA transmit antenna to AMPS or CDMA receive antenna isolation will be chosen to be 34 dB. (This also holds for any combination of

3 dB 1.8 dB

0 dB a 3.0 dB

Table 6-3: Degradation to Sensitivity Based on Noise Level Below kTBF

Noise level below kTBF Degradation to sensitivity


6


800 and 1900 MHz antennas.)

Since the required isolation between the Tx-Tx, Rx-Rx, and Tx-Rx pairs of antennas is for the most part identical for all of the combinations of both 800 MHz AMPS/CDMA and 1900 MHz CDMA PCS systems, it is appropriate that a single set of isolation requirements should be adopted. Table 6-4 summarizes the isolation requirements between two transmit antennas, two receive antennas, or a transmit and receive antenna pair which share a common location and are operating in the 800 MHz Cellular and/or 1900 MHz PCS bands and utilizing analog or CDMA technology.

Table 6-4: Antenna Isolation Requirements

The antenna isolation requirements in Table 6-4 represent the port-to-port isolation between the equipment end of the bottom jumper of one antenna system to the equipment end of the bottom jumper of the other antenna system. Therefore, if the combined jumper and main transmission line losses of the transmit and receive antenna systems are say 5 dB then the required isolation between the two antennas themselves would only have to be 29 dB to achieve the required 34 dB port-to-port isolation listed in Table 6-4.

6.3.1.2 Measuring Port-to-Port Antenna Isolation

The Tx-Rx isolation can be measured by feeding a test signal into the transmit antenna bottom jumper input (normally connected to the transmitter output port) and measuring the level of the signal at the output end of the receive antenna bottom jumper (normally connected to the receiver input port).

A typical measurement setup for port-to-port isolation between two antennas is a signal generator feeding the desired transmit frequency (at a level of about -20 dBm) into the transmit antenna bottom jumper and a spectrum analyzer or calibrated test receiver (adjusted to measure the level of the transmit test signal) connected to the receive antenna bottom jumper. The difference between the received level and signal generator test level is the port-to-port isolation. For example, if the level of the received signal is -60 dBm for a signal generator output level of -20 dBm, the port-to-port isolation would be 40 dB.

6.3.1.3 Reducing the Required Antenna Isolation

Except for overload of the victim receiver front ends by interfering transmit carriers, which require a minimum isolation between the transmit and receive antennas of 20 dB, all of the isolation requirements above 20 dB outlined above are due to the effects of either the noise energy or IM

Cellular Band (824-894 MHz) PCS Band (1.85-1.995 GHz)

Tx-Tx Rx-Rx Tx-Rx Tx-Tx Rx-Rx Tx-Rx

Cellular 23 dB 20 dB 34 dB 23 dB 20 dB 34 dB

PCS 23 dB 20 dB 34 dB 23 dB 20 dB 34 dB


6


products that are produced in the interfering base station PAs/LPAs and which fall in the receiver band.

If the receive band attenuation of the bandpass filter in the output of an interfering LPA is increased (or additional external receiver band filtering is added), the required antenna isolation may be reduced. However, transmitter IM products generated by hardware in the RF path following the bandpass or an added external filter may limit the amount of improvement that can be achieved.

6.3.1.4 Typical Antenna Isolation

For 800 MHz directional panel antennas it should be possible to achieve 25-30 dB of isolation with 0.45-0.6 meters of spacing and 35 dB or so at 1 meter of horizontal spacing. However, reflections from the tower structure and coupling effects from other antennas may reduce the isolation obtainable. This is especially true for the advertised front-to-back ratios for many directional antennas which do not have metal reflector panels on the back sides of the panel structures.

1900 MHz PCS directional panel antennas should be able to achieve isolation levels comparable to similar 800 MHz types at spacings approaching half of the 800 MHz spacings. Because of this the tower platform sizes at 1900 MHz can be significantly smaller than those at 800 MHz.

On the basis of limited testing by several of the antenna vendors it would appear that the cross band isolation between 800 MHz and 1900 MHz antennas in close proximity can run 10-15 dB better than the same band isolation would be for similar physical spacings. Because of differences between various antenna types, the actual antenna isolation of a proposed site sharing configuration should be measured using the techniques in Section 6.3.1.2.

6.3.1.5 CDMA Antenna Placement

In consideration of the above isolation requirements, Nokia Siemens Networks recommends that any required CDMA antennas be mounted on the tower above or below any existing antennas being used by other wireless technologies such that superior isolation provided by vertical spacing is obtained while at the same time providing the required CDMA coverage to the surrounding area.

The goal of this approach is to leave any existing antennas untouched. If, however, CDMA antennas are to be installed on a tower platform that is already supporting antennas from other technologies (provided that enough isolation is provided), it may be necessary to replace the existing antennas with smaller antennas to physically accommodate the newly-added CDMA


6


antennas. Figure 6-3 provides an antenna placement example using a “shared” platform approach.

Figure 6-3: Antenna Placement - Shared Platform

Figure 6-4 provides an antenna placement example using a “separate” platform approach.

Figure 6-4: Antenna Placement - Separate Platforms

With reference to Figure 6-3, the shared platform approach can be readily utilized for an 800 MHz AMPS/CDMA configuration with shared receive antennas and one or two sets of separate transmit antennas. An eight antenna configuration involving two receive and two transmit antennas for each

AMPSRx

(Main)

CDMARx

(Main)

CDMARx

(Diversity)

AMPSRx

(Diversity)

CDMATx

20 dB of isolationdesirable



34 dB of isolationrequired


AMPSTx

Notes: 1. Only 1 face of a 120° S/S implementation is shown here.

AMPSRx

(Main)

CDMARx

(Main)

CDMARx

(Diversity)

AMPSRx

(Diversity)

CDMATx

AMPSTx

Notes: 1. Only 1 face of a 120° S/S implementation is shown here.



1 m min.verticalseparation


6


of the AMPS and CDMA systems, can get rather unwieldy, and the separate platform approach in Figure 6-4 might be more appropriate.

For 800 MHz and 1900 MHz shared sites, the separate platform approach would appear to be the better choice, not that sufficient isolation could not be obtained with the single platform but because of the potential for conflicts should either of the systems want to change existing antennas or add additional antennas. Any physical changes in the antennas for one system could impact the other system because of a reduction in antenna isolations on the same platform. Separate platforms will normally provide a higher degree of isolation between the two systems which reduces the possibility of "political problems" between the two systems when either system desires changes in their antennas.

6.3.2 Antenna Diversity (Spacial)

The CDMA system employs time, space and frequency diversity. Spatial diversity is implemented through the use of two receive antennas at the base station, commonly called "Antenna Diversity". Receive antenna diversity is employed at the base site to improve the uplink by approximately 3 to 5 dB. The gain obtained by spatial diversity is based on the assumption that the signals received by the two separated antennas are not correlated or have a low degree of correlation, the affects of fading on one path will therefore be independent from the second. The 3 to 5 dB improvement is already incorporated into the equipment Eb/No receiver sensitivity specification. Note that if horizontal diversity is not utilized, the equipment performance may degrade.

6.3.2.1 Horizontal Antenna Diversity and Recommended Separation

The conventional method for determining the minimum separation for horizontal antennas to achieve non correlation is normally expressed as a factor of the wavelength (equal to the speed of light/frequency). The recommendation for standard cellular implementation (800 MHz) has generally been accepted as 10 times the wavelength (lambda). This figure should only be considered as an average distance as the level of correlation for horizontal diversity can also be affected by a number of variables, for example; the height of the antennas, the type of surrounding clutter (i.e. the level of multipath) and the typical angular arrival of the signals (i.e. are the antennas mounted perpendicular to a highway). See IEEE reference paper.

CELLULAR AND PCS PROPAGATION MEASUREMENTSAND STATISTICAL MODELS FOR URBAN MULTIPATHON AN ANTENNA ARRAY

Catherine M. Keller and Daniel W Bliss IEEE date: 2000

http://ieeexplore.ieee.org/iel5/7043/18961/00877962.pdf

As the wavelength of PCS frequencies is approximately half that of conventional cellular, the diversity antenna separation for PCS will effectively be half that of 800 MHz systems. The antenna separation of 10 lambda at the base site is considered sufficient for the non correlation of uplink signals within an urban environment (obviously greater than 10 lambda will provide even less


6


correlation). Also see the IEEE paper referenced above.

Note that Lee’s equation utilizes the antenna height in addition to frequency to determine the minimum horizontal diversity separation. This equation can be used as a more accurate planning guideline where the antenna height is known.

Frequency: 1850 MHz Wavelength: 16 cm Diversity distance (x10): 1.6 m (5.3 ft.)

Lee’s Equation: d = 77.27*h/f Where d = Rx antenna separation, h = Rx antenna height (ft.), f = frequency (MHz)

Example (1850 MHz @ 100 ft.)d = 77.27*100/1850 d = 4.2 ft.

It is believed that the horizontal separation of 5.3 (ft.) is an achievable separation distance for PCS cell site installations. Field trials and performance tests on PCS systems will determine if this minimum separation can be reduced under certain conditions.

6.3.2.2 Vertical Antenna Diversity

The vertical separation of two diversity antennas could be an appealing alternative for CDMA operators where the location of two horizontally separated antennas is hard to achieve. Unfortunately, the system engineer should be aware that the vertical separation of antennas provides poor diversity performance. This is due to a higher degree of correlation for a given distance compared to horizontal separation. In other words, the vertical separation distance required between two base site antennas is much larger than the horizontal separation required to gain the same correlation coefficient of two received branches.

The preferred method of implementing diversity at a base site is horizontal diversity. While vertical separation of receive antennas will provide a degree of non correlation, the performance of vertical diversity is not considered as effective as horizontal diversity.

6.4 CDMA Antenna Sharing

The following section discusses the various antenna sharing strategies that are currently available with respect to the Nokia Siemens Networks CDMA BTS.

6.4.1 Multiple Frame Antenna Sharing with 800 MHz BTS Products

This section provides some of the multiple frame antenna sharing configurations for the Nokia Siemens Networks BTS product lines at 800 MHz that are currently supported.

6.4.1.1 SC4812T/ET Antenna Sharing

Each 800 MHz SC4812T frame is capable of supporting up to two IS-95A/B or IS-2000 1X six-


6


sector carriers or up to four IS-95A/B or IS-2000 1X three-sector carriers. The SC4812T starter frame can currently support one (BC0) or two (BC1) SC4812T expansion frames, depending on the frequency of operation. External low-loss cavity combining for transmit antenna sharing is not supported. An optional duplexer can be used to share Tx and Rx antennas (see Figure 6-5). The SC4812T differs from the earlier SC4812 in that it contains Trunked LPAs in place of the dedicated per-sector LPAs. The Trunked LPA contains 3 or 4 LPA modules and supports 1 CDMA RF carrier for all sectors. Its power output capacity is shared between all sectors proportional to the traffic on each sector. Internal 2:1 or 4:1 cavity combiners are used to combine the Trunked LPAs to increase the number of CDMA RF carriers available.

Figure 6-5: SC4812T to SC4812T Expansion Frame

Note: m = main, div = diversity, exp = expansion

There are three versions of the SC4812T frame, a starter frame, an expansion frame, and a modem frame. The general differences between the three different versions are as follows. A starter frame is a standard stand-alone BTS frame which is designed to amplify the Rx & Tx signals while connected directly to the antenna feed line jumpers. An expansion frame shares the Rx signals from a starter frame and thus it is designed with a lower Rx gain in the front end, since the starter frame provides the first stage of amplification. The Tx signals of an expansion frame are independent from that of the starter frame and are typically connected to their own antenna (unless some sort of external combining technique is used). An SC4812T modem frame shares the Rx signals from another frame as well as providing a low level Tx output signal which requires further amplification from yet another frame.

For expansion kit ordering information refer to the latest version of the equipment planning guide or contact the Product Management group for more information.

Other frames have similar methods of sharing antennas. The following table shows which types of

SC4812TExp. FrameSC4812T

Rx Exp.

Rx-m Tx

D

Tx

OptionalDuplexer

Rx-divD


6


frames can share antennas with various starter frames.

6.4.1.2 UBS-Macro Antenna Sharing

Two UBS Macro 800 MHz frames that each use only a single trunk group (XMI modules all co-herently combined) can be configured to share receive antenna signals as shown in Figure 6-6. Each frame provides a single Tx/Rx path to one antenna for each sector. The Rx Expansion path from each frame is connected through attenuators to the Diversity Rx input of the other frame. This configuration can provide up to 16 carriers at 800 MHz per site. Each frame has its own BTS ID.

Table 6-5: Types of Frames Sharing Antennas with Starter Frames.

Starter Frame Type Expansion Frame Types

4812T 4812T, 4812T-Lite, 4812T-MC

4812ET 4812ET, 4812ET-Lite

4812T-Lite 4812T-Lite

4812ET-Lite 4812ET-Lite

4812T-MC 4812T, 4812T-Lite, 4812T-MC

SC480 SC480

SC2440 SC2440

SC4840 SC4840

SC7224 none

UBS-Macro 800 MHz

UBS-Macro 800 MHz

M810 none


6


Figure 6-6: UBS Macro to UBS Macro Antenna Sharing

The gain provided by the other BTS will cause the default RSSI value reported for the Diversity branch to be higher than that of the main branch for the same input signal level. Although this does not impact BTS operation, if the operator desires to correct the RSSI values, Rx path calibration should be performed on both BTSs using winLMF.

The Rx Expansion path in this configuration is provided after the LNA of the receive path which is provided by XMI 1 of the respective BTS. Power cycling or resetting XMI 1 will cause the LNA signal to the Expansion Rx path to be disabled, which in turn may impact the Reverse Noise Rise (RNR) reported by the 1x sector-carriers on the other BTS due to the nominal gain of the Rx Ex-

IDR

F**

Tx Coherent Combiner

XM

I

XM

I

XM

I

XM

I

RX Splitter

IDR

F**

IDR

F**

IDR

F**

IDR

F**

IDR

F**

30010145001JumperCables

EX

P R

X 4

A

EX

P R

X 5

A

EX

P R

X 6

A

AN

T 1

A

AN

T 2

A

AN

T 3

A

AN

T 1

B

AN

T 2

B

AN

T 3

B

15 dBAttenuators@ 800 MHz

SGLN6426RX ExpaninCable

EXPANSION /3RD PARTY port

UBS Frame #1

IDR

F**

Tx Coherent Combiner

XM

I

XM

I

XM

I

XM

IRX Splitter

IDR

F**

IDR

F**

IDR

F**

IDR

F**

IDR

F**

30010145001JumperCables

EX

P R

X 4

A

EX

P R

X 5

A

EX

P R

X 6

A

AN

T 1

A

AN

T 2

A

AN

T 3

A

AN

T 1

B

AN

T 2

B

AN

T 3

B

15 dBAttenuators@ 800 MHz

SGLN6426RX ExpansionCable

EXPANSION /3RD PARTY port

UBS Frame #2

**All IDRFs in both frames must have the same part number, thus requiring that all CDMAcarriers in both frames must fall within the IDRF passband.

Lightning arresterson all RF antennaconnections

Site"Main"

AntennasS1 S2 S3

"Diversity"AntennasS1 S2 S3

Site


6


pansion path. If the sector-carriers on the other BTS have been operational for more than 24 hours, the impact on RNR is negligible and may be ignored. However, if the 1x sector-carriers have been in operation for less than 24 hours, locking and unlocking the 1x sector-carriers is recommended to prevent the RNR from being increased, which may reduce the 1x SCH throughput capacity. This recommendation only applies to the antenna share configuration, as the two BTSs are not aware of the state of XMI 1 in the other BTS.

EV-DO sector-carrier operation is not affected by the resetting or power cycling of XMI 1, as the EV-DO carrier monitors the RSSI value during a periodic "quiet time" interval that is part of the reverse link signal processing.

800 MHz UBS Macro frames configured with two trunk groups and 1.9 GHz UBS Macro frame configurations require two Tx/Rx antennas per sector, so there is no advantage to antenna sharing unless external equipment is provided to combine Tx antenna paths. Customers should consult with BTS RF engineering if such configurations are desired.

6.4.2 Multiple Carrier Cavity Combining With 1900 MHz BTS Products

Combining is considered desirable by PCS operators in order to support multiple carriers at cell sites with a minimum number of antennas. It is important to remember that the function of combining will inherently add loss to the forward link. The following section will therefore provide the system engineer with general guidelines on how combining is implemented within the Nokia Siemens Networks BTS architecture (at 1900 MHz).

6.4.2.1 Output Power With Combining

The SC4812T will provide 22.4 Watts “top of cabinet” output power assuming that the RF power delivered to each sector is equal. For PCS applications, Link Budget analysis shows that a pilot power of 2.4 Watts is needed to balance the forward with the reverse link. Nokia Siemens Networks assumes a nominal 8.3 Maximum Power to Pilot ratio for a fully loaded carrier. This means that 20 Watts is sufficient to balance the uplink and downlink paths. The combining is internal to the frame and is accounted for in the 22.4 Watts "top of cabinet" power. This leaves 0.5 dB of loss for the duplexer. Note that 22.4 Watts “top of cabinet” does not include the 0.5 dB loss of the duplexer, which is external to the cabinet.

6.4.2.2 Type of Combining

Nokia Siemens Networks will provide multiple pole cavity filter combiners, utilizing conventional phased transmission line combining techniques, which are self contained within a “cast” housing. A maximum of 4 branch combining will be supported allowing up to a maximum of 4 alternate carrier channels to be combined per antenna/per sector (with duplexers).

The following table lists the band designators of the various cavity combiners. Not all of the combiner combinations are orderable. (The ordering guide has a list of combiner combinations.)


6


1. Band Edge, limit low end to 25

2. Band Edge, limit upper end to 1275

3. In C24, H is also used for any new designator range. H Block is defined for future usage. Cannot be supported on current UBS

6.4.2.3 Multiple Carrier Scenarios

The SC4812T will support a maximum of 12 sector-carriers per site (i.e. 3 sectors with 4 RF carrier, or 6 sectors with 2 RF carriers). The 22.4 Watts at the "top of the cabinet" for SC4812T

Table 6-6: Band Designators for Cavity Combiners

Designator Min. Chan. Max. Chan. Designator Min. Chan. Max. Chan.A3 251 100 EV 650 750

A2 25 125 EM 675 775A3P 50 150 E 700 800A4M 75 175 EP 725 825A4 100 200 EFV 750 850A4P 125 225 FM 775 875AS 150 250 F 800 900A1 175 275 FP 825 925A5 200 300 FV 850 950A5P 225 325 C3M 875 975A1P 250 350 C3 900 1000DM 275 375 C2 925 1025D 300 400 C4V 950 1050DP 325 425 C4M 975 1075DS 350 450 C4 1000 1100B3M 375 475 C4P 1025 1125B3 400 500 C5M 1050 1150B2 425 525 C1 1075 1175B3P 450 550 C5 1100 1200B4M 475 575 C5P 1125 1225B4 500 600 C5V 1150 1250B4P 525 625 GM 1175 1275B5M 550 650 G 1200 12752

B1 575 675 GP 1225 12752

B5 600 700 GV 1250 12752

B5P 625 725 H 1300 13753


6


includes the combining loss with the cabinet for either configuration, therefore 20 Watts of output power per each RF carrier on each sector can be provided through up to 0.5 dB of external duplexer, and cable loss.

The SC300 1X supports one carrier per FRU. Up to four carriers can be supported by interconnecting four FRUs. Each FRU supports 10 Watts per carrier RF output power.

The following figures provide a high level outline of the combining required to support two and eight carriers with the SC4812T (note that a only a single sector is shown as all 3 sectors are identical).

Assuming that the maximum number of antennas allowed at a cell site is 6 (2 per sector), Figure 6-7 shows that combining is not required for a two adjacent carrier configuration. If 6 duplexers are utilized, each antenna within each sector can be duplexed to either carrier 1 or carrier 2. This configuration will allow for balanced receive paths (i.e. no need for pads) and will allow for sufficient power (20 Watts) to balance the uplink. Provided that both carriers are duplexed in every sector, only 6 antennas will be required for a 3 sector site.

Figure 6-7: 2 Carrier Configuration

Alternatively, the SC4812T frame may include either 2:1 or 4:1 Tx cavity combiners. Adjacent RF carriers cannot be combined using cavity combiners. Alternate adjacent carriers can be combined with the cavity combiners. In the single frame 3-sector 6 antenna case, only 1 duplexer per sector is needed for the 2 carrier non-adjacent channel case. This configuration will also allow for balanced receive paths (i.e. no need for pads) and will allow for sufficient power (at least 20 Watts) to balance the uplink.

Figure 6-8 shows how the configuration of 8 carriers for the SC4812T may be combined onto 6 antennas. Note the following applies to SC4812T:

1. Only a single stage of 4 branch cavity combining is required.2. The use of alternate (non adjacent) frequencies is required.3. Duplexers for each antenna are required.

The configuration of 8 carriers will require (2) SC4812T cabinets. The cavity combiners are

Tx1 Tx2

Antenna 1(Sector 1)

Antenna 2(Sector 1)

Duplexers

2 Carriers with 2 Duplexers (no combining)

Rx-m Rx-div


6


contained within the SC4812T cabinets, thus helping to minimize the cable lengths.

An 8 carrier configuration is also possible with two SC4812T-MC frames. For the SC4812T-MC, combining is before the PA. This means no cavity combiners are required and the alternate channel restrictions are removed.

An SC7224 frame also uses pre-PA combining. There is just one Tx antenna connection per sector for a single band SC7224. There are two antenna connections per sector for dual band SC7224 - one per band.

UBS frames with coherent combiners will use a single Tx antenna per sector. UBS frames without combiners will have a Tx antenna for each XMI. UBS frames with cavity combiners will combine pairs of XMIs, so the number of Tx antennas per frame will be one for each pair of cavity combined XMIs and one for each uncombined XMI per sector.

Figure 6-8: 8 Carrier Configuration

6.4.3 Duplexing

Duplexing is one of the options that can be used to reduce the number of antennas required to support a CDMA base station. The duplexer for the SC4812T, for example, is a standard, three-port device, which allows for the combination of transmit and receive signals onto one antenna.

Tx1 Tx2Tx3 Tx4

Tx5 Tx6Tx7 Tx8

Antenna 1(Sector 1)

Antenna 2(Sector 1)

DuplexerDuplexer

To Rx A To Rx B

4 BranchCavity Combiners

4 Branch Cavity Combining for 8 Carriers


6


Figure 6-9: Duplexer

The duplexer does not incorporate a circulator. Therefore, port isolation is achieved through the phasing and stop band attenuation of the two bandpass filters. The following table outlines the frequency response characteristics for a 1900 MHz duplexer.

Table 6-7: Duplexer Frequency Response Characteristics

The duplexer 3rd order intermodulation (IM) products between the Tx port and Rx port, for two (10) Watt carriers in the transmit band (1930 - 1990 MHz) will be below -100 dBm and the fifth order (and higher) IM products will be below -120 dBm. Note: Duplexers that include G-block will have somewhat different characteristics.

The duplexer is physically included within the SC4812ET/ET Lite (outdoor products) and the SC300 1X cabinet, but is not located within the SC4812T (indoor) product. Please refer to the current “B1” document for full equipment specifications.

6.4.3.1 Pre-Engineered Kits

Note that Nokia Siemens Networks offers pre-engineered RF kits as part of its equipment offering for the SC4812T, these kits include items such as duplexers and directional couplers.

6.4.3.2 Duplexers and Intermodulation

The use of duplexed antennas will allow the combination of transmit and receive signals onto a single antenna via a duplexer. This solution may be considered desirable by a number of PCS operators in order to reduce the total number of antennas required per site. The Nokia Siemens

Antenna Port to Receive Port

Transmit Portto Antenna Port

Transmit Port to Receive Port

Pass Band 1850 - 1910 MHz 1930 - 1990 MHz -Stop Band DC to 1770 MHz

& 1990-4000 MHz3860 - 5970 MHz 1850 - 1910 MHz

1930 - 1990 MHzPass Band Insertion Loss 0.5 dB max 0.5 dB max -Stop Band Isolation 30 dB 30 dB 40 dB minimum

ANT PORT

Rx PORT Tx PORT

Path 1 Path 2

Path 3


6


Networks PCS infrastructure will be capable of supporting duplexed antenna configurations. The SC4812ET/ET Lite (outdoor products) and the SC300 1X include internal duplexing equipment.

The use of duplexers implies zero isolation at the antenna port between transmit and receive carriers. Under these conditions any transmit IM spurs created by non-linearities, in active or passive components, in the common path, might produce significant interferers if they fall within the receive carrier band. Duplexers can be made to work, in some applications, under ideal conditions; but any imperfections introduced by aging, lightning, thermal cycling, bi-metallic interaction or other common stresses can reduce system performance to below acceptable levels.

With regard to duplexing at 1900 MHz, it is useful to look at the potential for Transmitter Intermodulation (IM) in duplexer equipped installations and to compare it to some of the existing cellular technology systems. The following table examines the operation of AMPS/GSM/CDMA and outlines the minimum Transmitter IM order required to generate IM products in the Rx band of each technology. The minimum is calculated since the power generated by IM tends to fall off fairly quickly with increasing IM order. Therefore, the majority of interference is generated by the lowest order products.

Note that the IM orders presented in the following table for 1900 MHz refer to a single PCS band case, operation within multiple PCS bands at the same site may require further investigation. Certain combinations of sector-carriers can produce 3rd and 5th order intermodulation products that are within one of the receive bands, and so should be avoided. In general, 11th order frequency separation is sufficient to maintain control of transmitter passive intermodulation in duplexed systems if all equipment recommendations are followed.

Table 6-8: Minimum IM Orders.

Nokia Siemens Networks believes that duplexers are a viable solution for PCS systems due to the fact that for many configurations only high order IM products will fall within the PCS band. However, the following outlines some of the 1900 MHz sub-band combinations that could create IM issues.

While there are no known issues in the Nokia Siemens Networks BTS products with receiver desensitization due to transmit intermodulation, a BTS serving non-adjacent sub-bands should be operated to avoid CDMA channel combinations where a 3rd order product of the transmitted carriers falls in a receive channel. The IM products of concern would be generated between carriers in widely separated sub-bands. Coupling between transmit antennas is the most likely path for the intermodulation to occur within components of the dual trunk group BTS. A minimum Tx to Tx

SystemOperator

BandwidthTx-Rx

SpacingMin IM Order

AMPS A side 22.5 MHz 45 MHz 5thAMPS B side 14.0 MHz 45 MHz 7thGSM (best case) 12.5 MHz 45 MHz 7th1900 MHz CDMA 15.0 MHz 80 MHz 11th1900 MHz CDMA 5.0 MHz 80 MHz > 30th


6


isolation of 23 dB (as noted in Table 6-4) between the separated sub-bands is required to achieveadequate IM performance, even with, the following frequency planning restrictions. The relatively few affected CDMA channel combinations have been identified in this section.

For 3rd order IM the transmit carriers to avoid are around 40 MHz apart, resulting in a product in the receive channel 80 MHz below the higher transmit carrier. Transmit carriers can be around 40 MHz apart only in these combinations of frequency blocks: A with F, A with C and D with C. The involved combinations of sub-bands and CDMA channels can be summarized for the dual trunkgroup starter frame as follows:

• Intermodulation between block A and block F could degrade block F reception. CDMA channels 25 to 95 in A-block (in sub-blocks A-2 & A-3) IM with F-block.

• Intermodulation between block C and block D could degrade block C reception. CDMA channels 1105 to 1175 in C-block (sub-blocks C-1 & C-5) IM with D-block.

• Intermodulation between block A and block C could degrade block C reception. CDMA channels 105 to 275 in block A IM with channels 925 to 1095 in block C.

• Intermodulation between block G and block B could degrade block G reception. CDMA channels 1205 to 1275 in G-block IM with B-block.

Table 6-9 below shows the block and sub-block pairs that contain one or more IM channel combi-nation. Most of the pairs are between A-block and C-block, 15 MHz wide sub-bands that is less likely to be encountered together in the field. There are a number of usable specific channel com-binations within the identified IM sub-block pairs and expansion frames are a special case, so the actual channel combinations should be examined as outlined below.

Table 6-9: Frequency Block Pairs with Mobile channel Intermodulation products

Table 6-10: Unsupported Combinations of commonly used CDMA Channels provides another lay-er of detail below the sub-block level. For a dual trunk group starter frame, it shows an adjacent channel pair in Block-A or Block D, and the associated individual upper frequency CDMA chan-nels of concern. Use of the carrier frequency combinations shown should be avoided if practical. Unique site engineering should be applied to determine duplexer and other requirements if use of

F-block C-block G-blockBlock /

sub-blockF C-1 C-2 C-3 C-4 C-5 G

CDMA Ch. #

825-875 1075-1175

925-1025 925-975 1025-1075

1125-1175

1225-1275

A-1 175-275 IM IM IM IMA-2 25-125 IM IM IMA-3 25-75 IMA-4 125-175 IM IMA-5 225-275 IM IM IMD 325-375 IM IM

B-3 425-475 IM


6


these restricted combinations is anticipated.

If an expansion frame is used so that there are more than the two carriers in the lower frequency sub-band, or CDMA channel assignments are other than the normal ones shown, the receive chan-nels where third order IM products can fall in the upper receive sub-band are calculated as follows:

Lowest CDMA channel number affected = 780 plus the CDMA channel number of the lowest car-rier in the lower sub-band.

Highest CDMA channel number affected = 820 plus the CDMA channel number of the highest car-rier in the lower sub-band.

Table 6-10: Unsupported Combinations of commonly used 1.9 GHz CDMA Channels

6.4.3.3 Proper Installation and Maintenance of Duplexed Antennas

The comments below are intended to show proper installation and component selection in systems where duplexer use cannot be avoided.

Lower Sub-band Designators

Channel 1 CDMA Ch. #

Adjacent Ch 3 CDMA Ch. #

3rd Order IM CDMA Ch. #s

Upper Sub-band Designators

Block Sub-blocks Sub-blocks BlockA A-2, A-3 25 50 825, 850 FA A-2, A-3 50 75 850, 875 FA A-2 75 100 875 FA A-2 100 125 925 C-2, C-3 CA A-4 125 150 925, 950 C-2, C-3 CA A-4 150 175 950, 975 C-2, C-3 CA A-1 175 200 975 C-2, C-3 C

1000 C-2 CA A-1 200 225 1000 C-2 C

1025 C-2, C-4 CA A-1, A-5 225 250 1025 C-2, C-4 C

1050 C-4 CA A-1, A-5 250 275 1050 C-4 C

1075 C-1, C-4 CD 325 350 1125, 1150 C-1, C-5 CD 350 375 1150, 1175 C-1, C-5 CB B-3 425 450 1225, 1250 GB B-3 450 475 1250, 1275 G


6


6.4.3.3.1 Equipment Recommendations

All RF components in the cell site common receive/transmit path must be certified by the equipment manufacturer for IM performance. A typical (derived from GSM) IM specification is that all transmit IM products appearing in the receive band should be less than -110 dBm for two input transmit carriers, at a power level of 25 Watts per carrier. In addition, a regularly scheduled Preventative Maintenance Inspection (PMI) plan should be developed to verify that system IM performance has not been degraded and to ensure component integrity. Typical requirements for a PMI plan are described below.

The following components at the site would require IM certification:

Coax - Standard “Heliax” type coax is considered to have acceptable IM performance if undamaged and unkinked. Other types of coax would have to be individually tested and certified. Cable installation should include visual inspections for cable damage and electrical measurements to verify performance. Provisions for strain relief to minimize stress on cables and maintain proper bend radii should be made. Cables should be mounted securely so as to prevent vibration and movement per vendor specifications.

Connectors - The connectors in the common transmit/receive path are the most likely cause of system IM problems. System planning should attempt to minimize the number of connections in this path in order to prevent IM problems from occurring. Connectors with good IM properties have silver plating and mechanical rigidity. 7/16 type connectors have been optimized for IM performance and should be used, if possible, in all paths with potential for IM problems. Assembly and installation instructions should be provided by the manufacturer and should include torque specifications. All connectors should be thoroughly cleaned, prior to installation, and waterproofed, if exposed to outdoor elements. Care should be taken when mating and unmating connectors to prevent contamination and to maintain plating integrity. Connectors should be regularly inspected for damage and proper torque.

Lightning Arrestors - Certification of lightning arrestors is the same as that of connectors. In addition, lightning arrestor performance will degrade if a lightning strike has been taken by the antenna. Verification of component performance should be made regularly.

Duplexers - Considerable effort has been made by duplexer manufacturers to improve IM performance of duplexers. A duplexer that has been certified for its IM performance should include adequate silver plating of components and 7/16 type connectors. Accelerated life testing should be performed as part of the certification process. Only IM certified duplexers should be used in a duplexed system.

Antennas - Each antenna installed in a cell site should be tested and certified for IM performance. This is due to the additional potential IM risk of contamination of the material used for the radiating elements (no ferromagnetic materials). Proper care in installation should be used to prevent antenna damage and to verify that there are no metallic objects in the radiation paths close enough to reradiate back into the receiver (the “rusty bolt effect”). Mechanical stability should be provided to protect from exposure and wind effects. Inspection and electrical verification should be made on a regular basis, especially after a lightning strike or other unusual weather occurrence.


6


6.4.3.3.2 Installation Recommendations

Antennas - Care should be taken in installation to maintain proper distances from any other radiators or other obstruction on the same tower.

Cable Lashing - All cables should be prevented from movement. A major source of IM is the movement of the cable at any connector. In addition, damage may result to the cable at a connector from continued movement.

Cable Bends - Care should be taken to prevent any excessive bends in cabling. Slack and service loops should be provided in cable runs to prevent stress to cables.

Water Proofing - All external connectors should be waterproofed and regularly inspected for hermeticism. External components should be installed to prevent internal water capture. Components should be removed from any areas with potential standing water.

6.4.3.3.3 Maintenance

A Preventative Maintenance Inspection (PMI) plan should be developed and followed in order to maintain the IM performance of a cell site. A PMI should include a complete visual inspection of the cell site for obvious component damage or misapplication and an RF two tone test to verify system performance is satisfactory. Figure 6-10 is a diagram of the two tone test setup and is shown below.

The low noise amplifiers combined with the spectrum analyzer in the above diagram should be sensitive enough to measure IM products at -120 dBm or lower. The frequencies of the CW tones should be such that the spurious product of interest should fall within the passband of the receive path. All measured IM products should be below -116 dBm (for 0.5 dB typical sensitivity degradation).

If any anomalies are observed, a sweep of the transmit path using a Time Domain Reflectometer (TDR) or equivalent should be performed. A TDR will identify the existence and location of significant RF discontinuity in the signal path.

Monitoring cell site received signal strength indicator statistics for consistent foreign carriers is also a good indication of IM problems and should be part of a PMI plan. Monitoring the receiver port in the cell site with a spectrum analyzer for foreign carriers should also be performed. The port should be monitored with the transmit carriers keyed and unkeyed to verify whether interference is internally or externally generated.


6


Figure 6-10: Two Tone IM Test Set Up (800 MHz)

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6


6.5 CDMA Antenna Sharing With Other Technologies

The following section discusses topics associated with the sharing of antennas which is sometimes required to support both CDMA and AMPS technologies in the 800 MHz spectrum. Various methodologies for implementing co-located AMPS and CDMA cell sites are provided. Issues of mutual system interference and cell site equipment sharing are considered. Where appropriate, this discussion could be extended to include other antenna sharing configurations, provided that minimum isolation requirements are met.

While Nokia Siemens Networks recommends that CDMA implementations not share equipment with existing AMPS systems, it is understood that zoning restrictions and other hard realities might make the sharing of some equipment a virtual requirement from the customer's point of view. The guidelines below are intended to assure the most efficient implementation of the CDMA system while minimizing the risk to operation of the host AMPS system.

For this version of the CDMA RF Planning Guide it is assumed that the CDMA antennas will be co-located with existing AMPS antennas and will be sharing the same tower or roof top location.

For sites where the AMPS and CDMA systems are both omni-directional, it is assumed that the receive antennas will be shared between the two systems. Nokia Siemens Networks recommends that a separate transmit antenna be installed for the CDMA system to simplify the system design. Nokia Siemens Networks does not generally recommend the use of duplexers to allow the AMPS and CDMA systems to share common antennas. Please refer to Section 6.4.3 and Section 6.5.2 for more details on the subject of duplexed antennas.

For sites where the AMPS and CDMA systems will both be sectorized, Nokia Siemens Networks recommends that whenever possible the CDMA system should have separate antennas from the AMPS system. In fact, there are several CDMA system requirements which can only be satisfied by the use of separate CDMA and AMPS antenna systems. For example, the coverages of the AMPS and CDMA systems at the site require different downtilt angles for their respective antennas, or the CDMA softer handoff considerations require a narrower horizontal beamwidth for the CDMA sector antennas than for the AMPS sector antennas. Refer to the tower specifications to balance the weight to height ratio (tower loading).

It should be understood that in order to even allow for the possibility of sharing, the antenna will need to be able to operate in both of the frequency bands to be shared. For instance, an antenna that operates in the AMPS frequency band would not be acceptable to also share carriers assigned for the PCS band. Another instance to consider is if the antennas are only specified for operation in the transmit or receive portion of the band. An antenna of this type would not be acceptable to support both transmit and receive bands.

6.5.1 CDMA/Analog Shared Facilities

Sharing equipment virtually always implies sharing antennas. Three likely conditions for antenna sharing might exist:


6


• Common transmit antenna• Common receive antenna(s) • Duplexed antennas

In all cases where equipment is shared between a CDMA BTS and an analog BTS, a site-by-site evaluation of the changes to basic parameters (receive noise figure, receive input intercept point - IPi, receive sensitivity, transmit maximum power, transmit IM spur potential of the site, etc.) is required (in most cases).

6.5.1.1 Common Transmit Antenna

To share a transmit antenna, either a wide-band hybrid combiner or a shared PA is necessary. In the case of hybrid combining, it must be determined that the >3 dB loss is acceptable. Nokia Siemens Networks no longer supports PAs that can share CDMA and analog carriers. An external 3rd party PA may be possible.

6.5.1.1.1 Unapproved Combining Configurations

Ring Combiners

Combining of CDMA transmit signals with AMPS signals using ring combiners is not recommended. The constraints on the passband amplitude and phase characteristics for the wide bandwidth CDMA signal, and the narrow transition region between the CDMA carrier and the AMPS carriers, results in a filter design that would be undesirable because of high insertion loss. Such a filter would have to be tuned for a specific frequency plan, and would change as additional CDMA carriers are added. A wideband hybrid combiner (3 dB) would be smaller and less expensive, while still lossy.

Pseudo-Omni Cell Using Splitters/Combiners

It is possible to construct a unique AMPS cell site configuration using panel antennas with passive Tx splitters and Rx combiners to achieve a pseudo-omni pattern using an omni configuration BTS. While such a configuration would function for CDMA, the risk of performance degradation is significant. The deliberate creation of a deeply faded field in the antenna overlap areas, without the benefit of softer handoff, is likely to require increased average power per subscriber. The delay spread between these simulcast signals from each antenna can be less than 1 chip time. Forward and reverse power control operation in this situation would be more highly taxed. How much degradation occurs would depend on the amount of multipath present. This configuration is not recommended.

6.5.1.2 Common Receive Antenna(s)

There are two basic approaches to sharing receive antennas: 1) use an output of a non-CDMA multicoupler to feed a CDMA frame and 2) use the expansion frame outputs from a CDMA frame to feed the analog receiver. In either case, it will be necessary to do a system engineering analysis


6


of the noise figure (NF), gain, and intercept points (IP3). Pads may be necessary to balance the NF and IP3 considerations.

6.5.2 Duplexed AMPS/CDMA Antennas

The use of duplexers implies zero isolation between a family of transmit carriers and a family of receive carriers. Under these conditions, any transmit IM spurs created by non-linearities, in active or passive components, in the common path might produce significant interferers in the receive band. Duplexers can be made to work in some applications under ideal conditions; but any imperfections introduced by aging, lightning, thermal cycling, bi-metallic interaction or other common stresses can reduce system performance to below acceptable levels.

Nokia Siemens Networks does not recommend the use of duplexers for AMPS/CDMA systems at 800 MHz; however, certain situations may require their use. Intermodulation products introduced by the duplexed antenna system may degrade either the CDMA or the analog system depending upon the duplexing scheme implemented. For further clarification, refer to Figure 6-11 and the accompanying text.

Duplexing a 800 MHz CDMA system has been broken down into three options. These are the only options that are considered to be acceptable at this time. Any duplexing configurations that are different from what is shown below would require evaluation of its acceptability. The following table and figure illustrate three possible configurations where duplexers could be used with CDMA and AMPS carriers and the acceptability of each:

Table 6-11: Possible Duplexed Configurations

CDMA Tx AMPS Tx CDMA & AMPS Tx

CDMA&

AMP-SRx

Option #1: Unconditionally acceptable for one CDMA carrier. Conditionally acceptable for multiple CDMA carriers.

Option #2: Unconditionally acceptable for SIG only. Conditionally acceptable for multiple AMPS carriers. NOT acceptable for multiple AMPS carriers including SIG.

Option #3: Conditionally acceptable for one or multiple CDMA and AMPS SIG only. NOT acceptable for CDMA and multiple AMPS voice carriers.


6


Figure 6-11: CDMA Duplexing Options

Option 1: Duplexing One CDMA Transmit Carrier with CDMA and/or AMPS Receive.

This is the recommended implementation. Duplexing multiple CDMA transmit carriers with CDMA and/or AMPS receive may be acceptable if the proper IM prevention site engineering, frequency planning, and maintenance techniques are employed.

Option 2: Duplexing AMPS Voice or One AMPS SIG Channel (control channel) with CDMA and/or AMPS Receive.

This is an acceptable configuration. Duplexing multiple AMPS voice transmit carriers with CDMA and/or AMPS receive may be acceptable with proper IM prevention site engineering, frequency planning, and maintenance techniques. This is the least desired option due to the complexity of implementing and maintaining the proper IM frequency planning techniques for the multiple AMPS carriers. Duplexing multiple AMPS voice and SIG carriers with CDMA and/or AMPS receive is not acceptable.

Option 3: Duplexing One or Multiple CDMA and AMPS SIG Carriers with CDMA and/or AMPS Receive

This may be an acceptable configuration if the proper IM prevention site engineering, frequency planning, and maintenance techniques are employed. Duplexing one or multiple CDMA and multiple AMPS voice carriers with CDMA and/or AMPS receive is not acceptable.

The only inherently acceptable application of a duplexed CDMA system is to duplex the Tx of one CDMA carrier or one AMPS SIG carrier with the Rx of CDMA and/or AMPS. This is always acceptable because there is no transmitter generated receive band IM for one carrier.

Configurations that are inherently not acceptable are multiple AMPS carriers, including signalling channels, combined with CDMA carriers. These configurations are considered unacceptable because there is a potential problem of in-band intermodulation generation with difficult spurious frequency location prediction. The IM frequency planning mentioned above refers to planning the transmit frequencies into the duplexer such that high energy, low order IM products, do not

Option #1 Option #2 Option #3

CDMATx

Tx

Rx

Tx

Rx

Tx

Rx

CDMA& AMPS SIG

CDMA&

RxAMPS

CDMA&

RxAMPS

CDMA&

RxAMPS

AMPSVoice or SIG


6


interfere with the planned receive frequencies of the duplexer. The potential for interference and difficulty in spurious location prediction increases significantly when using EAMPS and NAMPS channels due to the increased number of carriers used in such configurations. The increase of frequency spacing of EAMPS channels also allows IM products, as low as fifth order for non-wireline systems, and seventh order for wireline systems to potentially exist (non-expanded AMPS systems only had potential for eleventh order IM products and higher).

Combined analog and CDMA systems, that are considered conditionally acceptable, require site engineering and preventative maintenance in order to provide acceptable system performance. Some of the guidelines for site engineering and preventative maintenance are presented in Section 6.4.3.3.

6.6 GPS AntennasThe installation of a GPS antenna and associated cabling is discussed in Chapter 8. As the recommendations for GPS antenna mounting (etc.) are common for both 800 MHz and 1900 MHz no further guidelines will be proposed here.

6.7 Ancillary Antenna System Components

In addition to a duplexer, there are other RF components that are considered part of the antenna system. Some of the more common components will be highlighted next.

6.7.1 Directional Couplers

A directional coupler is a power "sampler" with selective directivity. It is a relatively simple waveguide device that is used to sample the power on a transmission line, both in the forward and reverse directions. The sampling (or coupling) performed by the directional coupler is attenuated at a level (typically 30 dB) as to not affect the power on the transmission line, (i.e) it is sampling rather than splitting. In the Nokia Siemens Networks antenna system, directional couplers are used for the connection of the RFDS/CRMS (see Section 6.8 for more information). For PCS applications the directional couplers are connected in line with the transmission coax and may be mounted either at the waveguide (cable entry window) or within a 19" rack. Many of Nokia Siemens Networks’ BTSs incorporate directional couplers as part of the Rx and Tx filters.

6.7.2 Surge (Lightning) Protectors

To complement the existing internal and external grounding system (Please Reference: "Motorola’s Grounding Guideline for Cellular Radio Installations" - 68P81150E62), all transmission cables entering the cell site must be protected by devices such as "grounding kits" and tube or MOV protectors, commonly called "Surge or Lightning Protectors". Surge protectors are required in order to dissipate surge energy that can be generated from a local lightning strikes or other energy sources on the transmission lines.

A single surge protection unit is required (in addition to sufficient grounding equipment) for every transmission cable entering the site (Tx/Rx/GPS). The following description outlines the Nokia


6


Siemens Networks recommended surge protection unit. Please contact Nokia Siemens Networks ancillary (dropship) for specific product part numbers.

The Huber and Suhner 3400 Series protector consists of a coaxial transmission line and an optimized 1/4-wave shorting stub which is located between the center conductor and outer conductor. These protectors are designed as coaxial feedthroughs. A V-groove washer made of soft copper ensures that a low contact resistance between protector body and the mounting wall is achieved.

6.7.3 Transmission Line

The standard type of transmission line used for antenna systems is coaxial cable. There are a number of factors that must be considered in the choice of coaxial cable both in terms of RF performance and physical application.

6.7.3.1 RF Performance of Transmission Lines

For RF performance, the most important parameters in the choice of coaxial cable include, attenuation loss for a given frequency/ambient temperature, the VSWR (Voltage Standing Wave Ratio), return loss, power rating and insulation properties of the cable.

The loss of a coaxial cable will vary with frequency. Generally, the higher the frequency, the greater the loss for a fixed distance. Transmission line losses are incorporated into link budget calculations to determine the total loss of a RF transmission path. As this "path loss" will impact cell radius, the loss associated with the transmission cable should be kept to a minimum. Different types of coaxial cable are available and those with superior electrical properties (lower loss) are normally both larger (thickness) and more expensive (per meter).

The VSWR rating of a cable is the additional load allowed due to the mismatch of impedance. The system engineer should ensure that a cable with a VSWR rating between 1.01:1 and 1.15:1 is ordered. A cable which allows higher VSWR and hence load (due to reflected power) will increase the attenuation of the transmission line. Note that a VSWR of 1.15:1 equates to 23 dB return loss.

The return loss of a cable can be directly related to the VSWR rating. The return loss of a transmission cable can be considered as the difference in power in the forward and reverse directions when measured into a well matched load. All other things being equal, the higher the return loss the better the cable. The system engineer should choose a transmission cable with a return loss of 23 dB or better.

Please refer to the antenna parameter Section 6.2.5 for an explanation on how to convert VSWR to return loss.

The peak power rating of a coaxial cable refers to the maximum amount of power that can be safely sent over the coax. The power rating is determined by the type of insulation material and the structure between the inner and outer conductors of the cable (dielectric).


6


Power rating is not expected to be a problem for low powered CDMA PCS applications, as standard cable power ratings are rarely reached even for multiple carrier cellular configurations.

6.7.3.2 Physical Characteristics

The physical characteristics of coaxial cable should not be overlooked in the choice of transmission line. Although from a system perspective, the goal may be to limit loss, site specific installation criteria may limit the type of coaxial cable that can be used. The system engineer should consider; the cable length required, minimum bending radius allowed, the weight of multiple cables, the effects of wind loading, the ability to correctly mount/ground the cables and the cost of installation and expansion.

Generally, thicker cables allow less loss over a given distance but require more substantial hardware for mounting and grounding. The system engineer should plan for an achievable transmission line loss during initial system planning, bearing in mind both the optimum cable performance and the physical limitations of the cell site. During preliminary planning, it is recommended that the system engineer plans for approximately 2-3 dB total transmission line loss (including transition cables).

6.7.3.3 Choice of Transmission Line

The recommended type of transmission line in terms of performance versus cost, is foam dielectric coaxial cable. The dielectric material used is a closed-cell, low density polyethylene foam which prevents water penetration and allows for repeated bending. A solid corrugated outer conductor results in low loss, high power handling and continuous RF/EMI shielding. The combination of both a solid inner and outer conductor minimizes the potential for intermodulation generation. The following table gives an example of typical foam dielectric cables and their respective attenuation per 100 ft. at an operating frequency of 1850 MHz. At lower operating frequencies the attenuation values would be lower.

Table 6-12: Transmission Line Performance.

CharacteristicAndrews

LDF5-50A (7/8")

Andrews LDF6-50 (1-1/4")

AndrewsLDF7-50(1-5/8")

Attenuation dB/100ft @ 1850 MHz 1.88 dB 1.38 dB 1.19 dBImpedance (Ohms) 50 50 50Peak Power Rating @ 1850 MHz (kW) 1.45 2.20 2.96DC Breakdown volts 6000 9000 11000Diameter over jacket (mm) 28 39.4 50Minimum bending radius (mm) 250 380 510Cable Weight (kg/m) 0.49 0.98 1.36


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6.7.4 Transition Feeder Cables (Jumper Cables)

While the system engineer is considering the transmission line loss within the link budget, the loss of transition cables or “Jumpers” that may be required both at the antenna and equipment hardware also need to be included. These jumpers will generally be required due to the physical limitations of low loss thicker cable (i.e. the bending radius). The length of these jumper cables should be kept to within a few meters and the associated loss of both the cable and connectors should be calculated. The following table outlines a typical jumper cable type, Andrews 1/2" Superflex operating at 1850 MHz. Different characteristics would result if the operating frequency was changed.

Table 6-13: Transition Cable Characteristics.

6.8 RF Diagnostic System

The Nokia Siemens Networks RF Diagnostic System (RFDS) or Cellular Remote Monitoring System (CRMS) is a self contained unit within the BTS architecture that monitors and tests the RF paths of the BTS site. The aim of the unit is to identify faults or deteriorated conditions that are sufficient to impair the performance of the cell site RF channels. For the SC4812T (indoor) product, the RFDS/CRMS is 19" rack mountable. The RFDS/CRMS for the SC4812ET/ET Lite is incorporated into the product (within the cabinet).

The RFDS/CRMS connects to the RF paths of the cell site via pre-installed directional couplers (see above) and the RFDS/CRMS itself is comprised of the following equipment:

• Directional Couplers• Controller Card• A Test Subscriber Unit• Up to 2 Antenna Selector Units

The RFDS/CRMS measurement unit consists of directional couplers which sense and couple test signals to and from the RF system, an RF switch that connects "test equipment" to the RF path under test and a controller which is used to setup/execute tests. Access points are provided to allow external measuring instruments to be connected, this means that tests not performed by the RFDS/CRMS may be conducted. Examples are transmitter frequency, in-band transmit spurious output, transmit occupied bandwidth and adjacent channel leakage.

Characteristic Andrews FSJ4-50B

Attenuation dB/100ft @ 1850 MHz 5.17 dBImpedance (Ohms) 50Peak Power Rating @ 1850 MHz (kW) 0.625DC Breakdown volts 2500Diameter over jacket (mm) 13.2Minimum bending radius (mm) 32Cable Weight (kg/m) 0.21


6


The RFDS/CRMS will allow remote testing through interface connections to the Operations and Maintenance Centre-Radio (OMC-R) and/or the Local Maintenance Facility (LMF). RFDS/CRMS (with initial BTS installations) supports call termination loopback, call origination with subscriber status reports, forward (Tx) pilot channel power, Tx/Rx antenna VSWR, and forward FER rate.

The RFDS/CRMS can improve system performance by providing a quick and efficient method of detecting faults and it will provide the operator with the earliest notification of degraded equipment performance.


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7 RF Antenna Systems -
Chapter
7

Table of Contents

Advanced Topics


7.1 Dual Polarized Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 37.1.1 Fundamental Concepts . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 3

7.1.1.1 Dual Polarization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 37.1.1.2 Diversity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 47.1.1.3 Diversity Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 57.1.1.4 Cross-Correlation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 7

7.1.2 Isolation Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 117.1.3 Performance Impacts - Industry and Motorola Findings . . . . . . . 7 - 137.1.4 Antenna Selection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 15

7.1.4.1 Dual Polarized Antennas versus Singularly Polarized Antennas 7 - 157.1.4.2 Antenna Selection Criteria . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 16

7.1.5 Transmission at 45° . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 167.1.6 Incorporation of Dual Polarized Antennas into a Link Budget . . 7 - 177.1.7 Dual Polarized Antenna Summary . . . . . . . . . . . . . . . . . . . . . . . . 7 - 18

7.2 In-Building Distributed Antenna Systems . . . . . . . . . . . . . . . . . . . . . . . 7 - 197.2.1 In-Building System Architecture Overview. . . . . . . . . . . . . . . . . 7 - 207.2.2 Coaxial Cable System Design Using A Link Budget. . . . . . . . . . 7 - 21

7.2.2.1 Design Procedure Flow Chart . . . . . . . . . . . . . . . . . . . . . . . 7 - 217.2.2.2 Gathering Building Information . . . . . . . . . . . . . . . . . . . . . . 7 - 227.2.2.3 Determining the Base Station Location . . . . . . . . . . . . . . . . 7 - 247.2.2.4 Estimating the Antenna Placement within the Building. . . . 7 - 257.2.2.5 Selecting the Antenna Type: Omni vs. Directional . . . . . . . 7 - 257.2.2.6 Choosing the Base Station Type. . . . . . . . . . . . . . . . . . . . . . 7 - 267.2.2.7 Choosing the Cable Topology: Splitters, Couplers, and Taps 7 - 267.2.2.8 Estimating Cable Lengths from the Base Station to the Antennas7 - 317.2.2.9 Selecting the Coaxial Cable Type. . . . . . . . . . . . . . . . . . . . . 7 - 317.2.2.10 Link Budgets For In-Building Design . . . . . . . . . . . . . . . . . 7 - 337.2.2.11 Evaluating the First Pass and Iterating the Design. . . . . . . . 7 - 39

7.2.3 Active Coaxial Cable System Design. . . . . . . . . . . . . . . . . . . . . . 7 - 397.2.3.1 Downlink Amplifier Design Considerations . . . . . . . . . . . . 7 - 407.2.3.2 Uplink Amplifier Design Considerations. . . . . . . . . . . . . . . 7 - 417.2.3.3 Optimizing Amplifier Placement . . . . . . . . . . . . . . . . . . . . . 7 - 46

7.2.4 Fiber Optics for In-Building Systems. . . . . . . . . . . . . . . . . . . . . . 7 - 487.2.4.1 Fiber Optic Distribution System Architecture . . . . . . . . . . . 7 - 487.2.4.2 When To Use Fiber Optics . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 487.2.4.3 Fiber Optic System Design. . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 49

7.2.5 In-Building Antenna Systems Summary . . . . . . . . . . . . . . . . . . . 7 - 50

7.3 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 - 51


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CDMA/CDMA2000 1X RF Planning Guide7 RF Antenna Systems - Advanced Topics

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7


7.1 Dual Polarized Antennas

The availability of sufficient antenna tower/platform space to house required cellular/PCS antennas is becoming more and more of an issue in recent years with the addition of new wireless technologies, both cellular and non-cellular. As a result, operators are looking for ways to reduce the amount of physical equipment that is required to be mounted on the antenna tower or platform. The use of dual polarized antennas offers one such solution, provided that the technology being supported by them does not suffer a significant impact in performance as a result.

It is the goal of this section to present the fundamental concepts associated with dual polarized antennas, discuss any potential performance impacts and provide guidelines that can be used to assist the system engineer in deciding which dual polarization antenna design is optimum, if any, for a particular CDMA application. The performance impacts provided in this section were made from general observations taken from several different industry and Nokia Siemens Networks studies that were found on this subject.

7.1.1 Fundamental Concepts

In order to be able to make an educated decision as to which base station antenna polarization scheme to use (single vs. dual, horizontal/vertical vs. slant 45°, etc.), it is important to understand the various fundamental concepts associated with polarization diversity. Some key concepts are discussed below.

7.1.1.1 Dual Polarization

Conventional cellular and PCS antennas are typically 1/2 wavelength dipoles designed for vertical (usually) or horizontal polarization. Recall that a dipole produces a linearly polarized signal. The polarization itself is achieved by the specific placement of the elements within the antenna stack. If the alignment produces an E vector (electric field vector) which is vertical with respect to the earth, the antenna is considered vertically polarized. In contrast, if the alignment produces an E vector which is horizontal with respect to the earth, the antenna is considered horizontally polarized.

In a dual polarized antenna, the elements within the antenna housing are alternately placed. As depicted in Figure 7-1, some antenna models alternate the polarization from horizontal to vertical, others set the elements such that the polarization is crossed at 45° (sometimes referred to as slant polarization).


7


Figure 7-1: Dual Polarization Antenna Element Configurations

PCS and cellular dual polarized antennas are orthogonally polarized (horizontal/vertical or slant 45°). As will be discussed in more detail below, the antenna isolation and antenna cross polarization suppression (antenna coupling effects) need to be considered. Orthogonally polarized antennas have their polarizations ideally isolated and the cross polarization suppression is most distant.

Much like a singularly polarized antenna, a dual polarized antenna is capable of handling multiple frequencies. If so desired, a duplexer can be used with the dual polarized antenna to combine transmit and receive signals onto one set of elements, although there are issues associated with this configuration, as discussed in Section 7.1.2 and Section 7.1.5.

7.1.1.2 Diversity

In communication systems, diversity is used to increase the probability of receiving a given signal (message), which improves the ability of interpreting that signal (message). ‘Distinct parts’ are needed so that if one ‘part’ alone fails to deliver the message, perhaps a second ‘part’ will succeed. However, diversity is not simply a backup. Diversity is used to increase the probability of receiving a good signal, whether two signal components are combined or the stronger of the two signal components is selected. The use of the phrase ‘signal components’ here is meant to emphasize that one message or signal is transmitted, then split into separate components by various means (such as reflection, refraction, scattering, etc.). The components of the message are then used individually, or combined, to recompose the original message.

Examples of diversity being utilized in CDMA can be seen throughout the infrastructure. The following is a brief list, differentiated by the type of diversity that is offered.

Y

X

V1V2

x-y plane

Y

X

V1

V2

(front view)

Slant 45°

x-y plane

(front view)

Horizontal/Vertical


7


• Time Diversity: Error Control Coding, Data Repeat Schemes, Interleaving

• Frequency Diversity: Carrier Bandwidth versus Coherence Bandwidth of the Channel

• Path Diversity: Soft Handoff/Multipath Diversity

• Spacial Diversity: Receive Antenna Diversity (as provided by the physical separation of antennas)

• Polarization Diversity: Receive Antenna Diversity (proposed topic of this section)

Spacial and polarization diversity are techniques used in what is commonly referred to as ‘antenna diversity’. This section focuses on base station receiver antenna diversity, specifically that which can be provided by dual polarization. Antenna diversity is approached with the hope that if one radio path experiences deep fading, then a second independent path may have a signal with a reasonable probability of not being in a fade at the same time.

Presently, commercial CDMA systems typically use two antennas at the base station for diversity on the reverse link (subscriber to base station signals). As was mentioned in Chapter 6, the two antennas are separated (normally a horizontal separation) by at least 10 wavelengths at 800 MHz and at least 20 wavelengths at 1800 MHz. In this situation, the engineer assumes the signal components into each antenna will have a polarization identical to the receive polarization1. However, if a transmitted signal scatters, and one of the scatter components undergoes additional scattering, eventually some signal components may change polarization. Polarization diversity could then take advantage of this change. A system engineer could use a diversity antenna which has a polarization which is unique as compared to the primary antenna. A dual polarized antenna is, in fact, two antennas in a single housing with one antenna polarized orthogonal to the second.

7.1.1.3 Diversity Gain

Diversity gain measures the improvement in signal reception due to the utilization of a diversity path. It is the difference in signal level between one reference signal and the signal received at the output of the diversity combiner for a given probability or signal reliability. Signal reliability is the probability that the signal is adequate for a given period of time under the conditions encountered (usually measured between a 90% and 99% level)2.

Diversity gain can be measured as improvement in the signal-to-noise ratio (SNR) or Eb/No in CDMA. It is not a difference in SNRs, but rather a comparison between the final received SNR and what that SNR would have been without diversity. In real-world conditions, it may be difficult to measure the SNR, so measurements are typically taken of the signal plus the noise.

When a system engineer chooses to use antenna diversity, the type of diversity selected is based on

1. In CDMA, the base station transmit and receive antennas normally utilize linear vertical polarization.

2. Wahlberg, Ulrik. 1997. “Polarization Diversity for Cellular Base Stations at 1800 MHz.” Revision 1.0. Allgon.


7


the probability of receiving an uncorrelated3 signal. Generally speaking, in clean, high power, line-of-sight paths, diversity may be unnecessary. However, in CDMA systems, receive diversity is highly recommended to achieve desired capacity and performance. If the probability of the signal to undergo a fade is high, then some type of diversity is normally used. [Note: In CDMA, even if the signals are correlated, the diversity gain has been found to improve uplink capacity since the subscriber’s transmit power required is reduced (see Section 7.1.1.4.1). CDMA also uses other techniques such as convolutional encoding to help capture enough information from the signal to understand the message.]

Diversity gain is affected directly by the correlation of the signal envelopes, branch imbalance and also by the combining technique4. The greatest gain is achieved when two uncorrelated signals are received with equivalent energy (balanced branches) and combined.

Two branches (of a dual polarized or spacial diversity antenna system) can be individually selected or combined to improve the single branch performance. In a two branch selection diversity system, it has been found that the potential savings in power offered is equal to approximately 10 dB (at 99% reliability, see Figure 7-2) as compared to a single branch. In a two branch combining diversity system, the power savings (at 99% reliability) is equal to approximately 11.5 dB as compared to a single branch, or a 1.5 dB improvement over selection diversity5. In CDMA, the combining diversity method is used.

3. Correlation of signals literally means the “same-ness” of those signals.

4. There are four general methods used in selecting or combining signals in a diversity system:- Selection Diversity- Maximum-Ratio Combining (a.k.a. Maximal Ratio Combining)- Equal-Gain Combining- Switched Combining

5. Jakes, William C. 1974. “Microwave Mobile Communications.” New York. American Telephone and Telegraph Company. Reissued in Cooperation with IEEE Communications Society. pp. 309-324.


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Figure 7-2: Probability Distribution SNR for M-branch Selection Diversity System

7.1.1.4 Cross-Correlation

Cross-correlation is used to measure the correlation between two signal envelopes. In general, the literature shows that diversity gain is best realized when the signal envelope cross-correlation coefficient is under 0.7. If the signal envelope cross-correlation coefficient, on the other hand, is equal to 1.0, then the system is identical to a system without diversity. This is a key concept to remember. For CDMA systems utilizing spacial diversity, (at 1800 MHz) 10 wavelengths separation is used between the main Rx and the spacial diversity Rx antennas. This separation is necessary to achieve a signal envelope cross-correlation coefficient of less than or equal to 0.7.

(Note: the separation required to achieve a cross-correlation coefficient of less than 0.7 varies with antenna height, surrounding clutter, and typical angle of arrival. See IEEE reference paper at: http://ieeexplore.ieee.org/iel5/7043/18961/00877962.pdf for more details.

CELLULAR AND PCS PROPAGATION MEASUREMENTSAND STATISTICAL MODELS FOR URBAN MULTIPATH ON AN ANTENNA ARRAY

Catherine M. Keller and Daniel W Bliss IEEE date: 2000)

10 dB

Probability Distribution SNR γs for M-BranchSelection Diversity System.Γ=SNR on one Branch.


7


Signal correlation is dependent on the path loss and fading conditions encountered as the signal transverses space. The type of fading depends on the environment and has both small scale and large scale characteristics.

Small scale fading (or fast fading) is that which creates deep and rapid amplitude fluctuations. These deep fades are created by summing multiple signals, with random phases and amplitudes, in a highly reflective environment. Normally, fast fading implies a Rayleigh fading distribution.

The Rayleigh probability density function (pdf) is shown in Figure 7-3. If statistically independent samples are considered, the reception of a weak signal from a Rayleigh pdf infers that the probability of receiving a stronger signal (shaded area) with the next sample is quite good.

Figure 7-3: Rayleigh Probability Density Function

Large scale fading (slow fading) is used to describe fading which occurs over long distances (several hundred or even thousands of meters apart). Large scale fading is normally due to shadowing in both the terrain profile and the nature of the surroundings. This type of fading is log-normally distributed and in urban environments has a standard deviation of approximately 10 dB.

While signals normally undergo large scale fading, it is in the small scale fading environments where signal components are found to be uncorrelated, and therefore diversity combining makes a significant impact.

The following examples show the difference in diversity gain if two correlated signals are received as opposed to two uncorrelated signals.

7.1.1.4.1 Reception of Highly Correlated Signals and Their Effect on Diversity Gain

Consider a system with a spatially diversified pair of antennas, each of which is vertically polarized. If a subscriber transmits a signal in an unobstructed environment (line-of sight) towards the antenna pair, then the signal undergoes free-space path loss (assuming no reflection) and is received, highly correlated, at each antenna branch (see Figure 7-4).

Received signal envelope voltage r (volts)

Weak RxSignal

Probability of Receivinga Stronger Signal


7


Figure 7-4: Reception of Highly Correlated Signals

If the system used a selection method, the gain achieved by utilizing diversity is zero since the branches have identical received signals.

If the system combines the identical received signal energy using a maximum ratio combining method, then the maximum gain achieved is 3 dB (i.e. doubling the power of a single branch). The gain achieved will never exceed 3 dB for correlated signals. This also applies to equal gain combining.

7.1.1.4.2 Reception of Uncorrelated Signals and Their Effect on Diversity Gain

Diversity gain is at its highest value when the signals received at the spatially separated Rx antennas are uncorrelated. In this case, the received signals have different amounts of fading (see Figure 7-5). If the system uses selection or sampling methods, then the signal with the greatest SNR (lowest FER) is chosen. For example, assume the signal energy in branch A is four times as strong as that of branch B. In this example, diversity provides a 6 dB gain over a system without the diversity antenna. Although in this example the mean signal levels of each branch are not balanced, it is important to know that diversity gain is greatest when the mean signal levels of the two branches are balanced.

If the system receives uncorrelated signals using a combining method, the maximum gain achieved can vary significantly.

Two Vertically Polarized Antennas(front view)

Original Signal Faded Received SignalHighly Correlated at Each Branch


7


Figure 7-5: Reception of Uncorrelated Signals

To emphasize the importance of uncorrelated signals, assume the minimum receive level required is set at Level RCV as shown in Figure 7-6. If the signals are correlated and a diversity combining method is assumed, then the greatest improvement would be a 3 dB signal gain. If the original signal was received below the minimum requirements, it is possible that the improvement due to diversity combining may not be sufficient to provide a minimum number of good frames. In this case, the frame erasure rate (FER), or SNR would be poor.6

Figure 7-6: Correlated Signal Diversity Gain

In Figure 7-7, two uncorrelated signals are combined. In this example, the system samples each

6. Here the assumption is that the signal portion doubles, but it is important to note that the noise also doubles.

Two Vertically Polarized Antennas(front view)

Original SignalFaded Received Signal

Uncorrelated at Each Branch

A B

Level RCV (minimum)

Original Signal Correlated Signal Diversity Gain

{

{ {

Bad Frames

3 dB


7


signal and the combined signal exceeds the minimum required receive level.

Figure 7-7: Uncorrelated Signal Diversity Gain

As a subscriber travels in dense clutter, such as urban and suburban environments, its transmitted signal is reflected and undergoes various degrees of fading. Sometimes the base station antenna is line-of-sight with the subscriber and sometimes it is shadowed. It is in these fading conditions that the benefit of the diversity gain is intended to capture enough energy to interpret the message (see Figure 7-8).

Figure 7-8: Uncorrelated Signal Diversity Gain

7.1.2 Isolation Considerations

As was discussed in Section 6.3.1, there are isolation requirements that exist between both Tx and Rx antennas and between main and diversity Rx antennas that are primarily provided by the physical separation. However, since both Rx antenna elements and possibly a third Tx antenna element occupy the same dual polarized antenna housing, physical isolation requirements are

Level RCV

Original Signal Uncorrelated Signal Diversity GainUncorrelated Signal

(minimum)

Signal A

Signal A’

Fade Regionof

Signal A

As the subscriber travels, its transmitted signal is received with varying fades. If a signal is undergoing a deep fade, thenan uncorrelated signal can save the call (even if its mean energy level is weaker). Intuitively, when the signal is in a fade, the slower the subscriber speed, the higher the probability of continued signal fading.

Minimum RequiredRx Signal Strength


7


replaced by port-to-port isolation requirements. Port-to-port isolation is measured at the input of the base station equipment between the bottom jumpers of the antenna.

As with physically separated base station antennas, if the isolation requirements are not met within the antenna specifications, then the system engineer will need to consider using external band pass filters (for the Rx band at the Rx ports) or duplexers.

Note that antenna isolation is not dependent on the angles α and β, as shown in Figure 7-9.7

Figure 7-9: Theoretical Model for Base Station Polarization Diversity

However, cross polarization suppression, or cross coupling, is dependent on α and β. It can be shown8 that because of this dependence, orthogonally polarized dual pole antennas are ideally isolated. This is why dual polarized antennas are designed with orthogonal branches.

The orthogonality of a dual polarized antenna is specified by the antenna cross polarization discrimination. This is the ratio of the outputs from the co-polarized and cross-polarized ports when an antenna receives a signal from one plane (i.e. co-planar). As an example, if a horizontal/vertical dual polarized antenna receives a signal which is vertically polarized, then the antenna is considered to have good orthogonality if a very small portion of the signal is received at the horizontal port. Likewise, a large portion of the signal should be received at the vertical port. Good orthogonality has an antenna cross polarization discrimination value of about +20 dB (or greater). Poor orthogonality can push the antenna cross polarization discrimination value down to 0 dB (or less). Antenna cross polarization discrimination is generally required to be greater than +15 dB9. [Note: The term antenna cross polarization discrimination (AXPD) differs from the term cross

7. Kozono, S. 1985. “Base Station Polarization Diversity Reception for Mobile Radio.” IEEE Transactions on Vehicular Technology. Vol. VT-33. No. 4. pp. 301-306.

8. Wahlberg, Ulrik. 1997. "Polarization Diversity for Cellular Base Stations at 1800 MHz." Revision 1.0. Allgon.

9. This number may vary per antenna manufacturer. Nokia Siemens Networks recommends a minimum 34 dB isolation requirement between the Tx and Rx branches, therefore if the separation is rated to be 15 dB, the system engineer will need to insert a duplexer to ensure at least 34 dB separation.

Y

X

V1V2 α

β

Z

X

x-y plane x-z plane

Multipath

Main Beam

(front view) (top view)


7


polarization discrimination (XPD). AXPD is the measure of orthogonality of the antenna. XPD is the power ratio of the antenna branches.]

7.1.3 Performance Impacts - Industry and Motorola Findings

Many studies on the performance of a polarization diversity system utilizing dual polarized antennas have been performed throughout the industry. A consolidation of various findings is provided below. For more detailed information, consult the provided references (see Section 7.3).

NOTE: Most of the studies observed were completed on systems other than CDMA. All studies not done on CDMA systems focused on the signal envelope and therefore, focused on the signal decorrelation and received signal strength.10 In a cellular CDMA system, power control will directly effect the received signal strength, making it a nearly impossible task to measure any change from one diversity scheme to another. It is precisely because of the power control issues in CDMA that Motorola chose to study the received Eb/No requirements. Branch imbalance and signal correlation were included in the study. Branch imbalance, rather than the signal correlation, showed a greater impact to the quality of the diversity scheme (the greater the branch imbalance, the smaller the diversity gain). It is unclear at this time as to the impact polarization diversity may have on CDMA specific issues such as power control. The power control is both an open and closed loop process which relies on measured signal strength and Eb/No (for correction). If the loop becomes imbalanced, the impact on capacity or quality could be significant.

The effect of surrounding clutter type:

It has been shown11 that clutter type greatly effects the ability of the signal to change polarization sufficiently enough to be received decorrelated at the base station antennas. The denser the clutter, the higher the probability of receiving a decorrelated signal at each polarization. This finding was consistent throughout the studies. However, there appeared to be a greater branch imbalance if horizontal/vertical polarizations were used as opposed to slant 45° polarizations, and it follows that due to the large branch imbalance, diversity gain degrades.

The effect of subscriber unit antenna tilting:

Several studies transmitted signals utilizing varying degrees of subscriber transmit antenna inclination. Some studies tested performance at several different angles; however, only tests at 0° and 45° subscriber inclinations were common in each study. Reviewing this aspect of the data, it appeared that larger values in diversity gain were achieved when the subscriber was inclined at 45°. For example, a study provided in Vaughan’s paper showed that a subscriber transmitting at 45° had a 1.7 dB improvement in diversity gain over a subscriber transmitting at a 0° inclination. In this study, a horizontal/vertical polarized base station antenna was used and the data was collected from

10. This is often referred to as continuous wave (CW) testing.

11. Vaughan, Rodney G. 1990. “Polarization Diversity in Mobile Communications.” IEEE Transactions on Vehicular Technology. Vol. 39. No. 3. (August): 177-186.


7


an urban clutter environment. Similarly, if a slant 45° dual polarized base station antenna was used, then a 0.3 dB improvement was shown for the subscriber transmitting at an inclination of 45° versus 0°. The results from the latter study also shows that the slant 45° dual polarized base station antenna performs more consistently than the horizontal/vertical polarization.

It is important to note that in all of the studies, the improvement of using polarization diversity versus spacial diversity NEVER exceeded a 1 dB improvement (base station received power). In most cases, the polarization diversity performed worse than spacial diversity. Although the worst case results showed a 2.7 dB degradation, the polarization diversity was normally within 1 dB of the spacial diversity results (base station received power).

The effect of branch imbalance and correlation:

The Motorola lab study12 examined Rayleigh distributed signal envelopes. The test verified that branch imbalance decreased diversity gain. The test also seemed to show that correlated signals achieved greater gain than uncorrelated signals. This result seemed to counter the common finding in diversity systems; however, the data was calculated using a combined-minus-maximum-received-signal-strength equation and not the standard diversity gain equation. Diversity gain was measured as the combined Eb/No into the system, less the maximum of the Eb/No received in either branch. The maximum 3 dB gain achieved agrees with the maximum 3 dB gain of combining two identical (correlated) signals (discussed in Section 7.1.1.4). The greatest input from the Motorola studies was the observation that Eb/No and power control issues also needed to be addressed when studying antenna diversity systems for CDMA technology.

Table 7-1 shows data Motorola collected in a field trial test in Israel13. It confirms that the branch imbalance was a greater issue than the correlation. Remember, branch imbalance refers to the amount of energy received at each branch, where correlation refers to the “sameness” of the signal components received. This table goes on to show that the Eb/No requirement was larger for the dual polarized antenna system (albeit very slightly), which also translates into a slight degradation of performance. And finally, Motorola shows little change in diversity gain between the two diversity

12. Tobin, Joe, Rob Nikides, Devesh Patel, Edward Golovin. 1997. “CDMA Dual Pole Antenna Testing - Arlington Heights, IL.” Version 1.0. Motorola.

13. Golovin, Edward. 1998. “A Comparison of CDMA Reverse Link Performance with Base Station Spatial and Polarization Diversity Reception (Motorola Israel Measurement Campaign in Urban Area at 900 MHz)” Version 2.0. Motorola.


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techniques. This data was collected in an urban clutter environment.

7.1.4 Antenna Selection

The following sections provide additional information to consider before selecting a dual polarized antenna.

7.1.4.1 Dual Polarized Antennas versus Singularly Polarized Antennas

The most obvious advantage with using a dual polarized antenna, is the elimination of a second receive antenna unit (and possibly a third transmit antenna unit, if a duplexer or 3-port dual polarized antenna is used). This saves on real estate and mounting hardware.

Due to the fact that elements are alternated, and the number of elements per pole are reduced, antenna gain is typically decreased in a dual polarized antenna (or the length of the antenna is increased to accommodate the extra elements). Therefore, improved diversity gain may be achieved at the expense of antenna gain (for like-sized units). This may be an acceptable trade off, if the diversity gain is sufficient and range is not an issue. Otherwise, the loss of signal due to antenna gain may be intolerable. Therefore, dual polarized antennas should not be utilized to solve range problems. If the longer unit is selected for improved gain, tower loading issues will need to be readdressed.

The front-to-back ratio is typically decreased in a dual polarized antenna as compared to an optimized vertical antenna (this varies with antenna vendor). For CDMA systems, this translates into a slight reduction in capacity due to an increase of interference seen from the adjacent sectors.

Some manufacturers have solved the size, gain, and front-to-back ratio issues by layering the antenna elements on top of one another. This keeps the antenna parameters consistent with singularly polarized antennas.

Due to the fact that multiple antenna elements share the same antenna housing, dual polarized

Table 7-1: Motorola Data Table

Parameters (average over all locations)

Spacial Diversity (two vertically polarized antennas)

Polarization Diversity (dual polarized antenna)

Diversity Scheme & CDMA Reverse Link Degradation

Branches Imbalance & XPD (median)

1.15 dB 2.16 dB 1.01 dB

Cross Polarization Correlation (XPC) (median)

0.19 0.25 0.06

Mean Eb/No

Measured Results

8.22 dB 9.08 dB 0.86 dB

Diversity Gain (median)

4.89 dB 4.68 dB 0.21 dB


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antennas are also particularly susceptible to intermodulation distortion.

If the antenna is experiencing problems, with only one antenna at a given site, there is no opportunity to “hot swap” a line (i.e. no backup).

Dual polarized antennas, though seen as one antenna, will still require two separate transmission lines (one for each polarization).

7.1.4.2 Antenna Selection Criteria

It is recommended that the system engineer consider three main polarization specifications when choosing an antenna14:

• Tracking of the radiation patterns radiated through two polarizations. Verifying that the antenna radiation patterns (relative amplitudes) of the two branches are similar. Unbalanced branches can impede diversity performance and create unequal coverage footprints.

• Antenna cross-polarization discrimination (XPD). In dealing with 3-sector sites, orthogonality needs to be controlled over an angle of +60° off of bore sight. Cross-polarization discrimination is generally required to be higher than +15dB. Due to isolation requirements, Nokia Siemens Networks recommends port-to-port isolation of at least 34 dB (Tx-Rx).

• Isolation. For details on antenna isolation requirements, please refer to Section 7.1.2.

7.1.5 Transmission at 45°

Currently, the effect of transmitting the base station signal from an antenna which is polarized 45° from vertical needs further analysis. The emphasis on dual polarization antenna studies has been on the receive signal only; however, Nokia Siemens Networks recognizes that in CDMA, the open and closed loop power control are of key importance. Figure 7-10 provides various examples of the transmitting techniques involved with the different diversity antenna configurations.

14. Xiang, Jun. 1996. “Diversity Antenna Systems for GSM900/GSM1800/PCS1900 Networks.” Issue A. Motorola.


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Figure 7-10: Tx, Rx and Diversity Rx Antenna Configurations

If a horizontal/vertical polarized antenna is employed, then the vertical polarization element can be used for both transmit and receive, and the horizontal polarization element can be used for diversity receive. However, if a slant 45° polarized antenna is used, then regardless of which polarization element is chosen, the signal will be transmitted at 45° from the vertical.

In many cases, the maximum capacity is limited by the forward link. Therefore, any degradation to the forward link will typically impact the performance (i.e. coverage, capacity, and quality) of the entire site. An analysis of how large an impact to system performance is introduced by the forward link transmission of a 45° polarized signal is needed. Some believe if the clutter is sufficient to induce scattering such that the reverse link variance in polarization is adequate to utilize a dual polarized diversity scheme, then the forward link should also be sufficient. This assumption cannot be readily made without testing, since path fading characteristics are normally determined by the near field clutter.

It has been shown that in given situations (see Section 7.1.3), slant 45° polarization diversity is superior to horizontal/vertical polarization diversity (although neither is as good as spacial diversity). In order to leverage the advantages of slant 45° polarization and also minimize the risks of transmitting with a 45° polarization, an alternate solution is to use an antenna designed with three separate polarizations. An example of this type of antenna is shown in Figure 7-10. With this type of 3-port polarization antenna, one port is polarized at +45° from the vertical, a second port is polarized at -45° from the vertical, and a third port is vertically polarized.

Other considerations include the size of the antenna unit and the gain. A 3-port polarization antenna would be even longer than a 2-port slant 45° antenna.

7.1.6 Incorporation of Dual Polarized Antennas into a Link Budget

Utilizing dual polarized antennas as a means of diversity may have an impact on the CDMA RF link budget. The CDMA base station receiver sensitivity is comprised of several components, one of which is the required Eb/No to meet a specified performance (FER). The benefit of diversity gain is typically accounted for within the Eb/No value. Therefore, if a different diversity gain value is

Note: The arrows represent the direction of polarization.

Tx/Rx (main) Rx (diversity)

Spacial Diversity Slant 45° Horizontal/

Slant 45° &vertical in asingle unit

Dual Polarization Diversity

Tx/Rx (main)

Rx (diversity)

(either branch)

(either branch)

Tx/Rx (main)

Rx (diversity)

Tx

Rx (main)Rx (diversity)

(either branch)

Vertical


7


obtained from the various antenna diversity schemes, then a different Eb/No value will be required. This change in Eb/No will thereby impact the base receiver sensitivity and ultimately the maximum reverse link path loss. One of the Motorola field tests (discussed in Section 7.1.3) shows that the reverse Eb/No requirements are increased when using a polarization diversity scheme.

For the downlink, should a provider choose to use a 2-port slant 45° dual polarized antenna, the base station transmit antenna would radiate with a 45° polarization. As stated in Section 7.1.5, further studies are needed which analyze the effect that transmit inclination at 45° could have on the forward link, and (specifically from a CDMA perspective) how this may effect the Eb/Noperformance.

When inputting the value for antenna gain, the system engineer should use the gain value given by the antenna vendor.

7.1.7 Dual Polarized Antenna Summary

Unfortunately, the majority of the case data analyzed for this document was derived from field experiments and not from well controlled laboratory settings. Consequently, conclusions drawn from this data should be treated with a degree of skepticism, knowing that the environment in which each test was conducted and the test performed, may have had an impact on the result.

With the exception of the Motorola test cases, data was collected with continuous wave (CW) testing. For a CDMA system, CW testing may not be sufficient. Due to the power control in CDMA, Eb/No measurements are preferred. This makes data comparison between studies (CDMA vs. CW) extremely difficult.

In the past, most of the subscriber unit antennas were mounted onto the vehicle and their transmission was assumed to be vertically polarized. Today, most of the subscriber units are handheld and are subjected to “hand-tilting”. With the introduction of non-vertically polarized signals, polarization diversity is assumed to be a potential option to improve signal reception. In fact, it has been found that the tilt of the subscriber has less to do with the effectiveness of the diversity scheme than the environment in which the subscriber and base station are located.

It is understood that scattering is required to change the polarization. Thus, dense urban environments lead to more scattering and a higher probability of creating decorrelated signals with respect to polarization (as seen by the base station antenna system).

In the proper environment (dense urban), polarization diversity performed well. Surprisingly, it did not perform as well as a spacial diversity scheme, but was normally within 1 dB. (The worst case shown was 2.7 dB which was seen in a suburban environment.) Taking into consideration the losses incurred in transmission lines, connectors, duplexers and combiners, the loss in diversity gain may be offset dependent on the quality of the antenna system. Decisions would need to be made as to whether or not an estimated 1 dB degradation would be acceptable.

Diversity gain could be offset in a CDMA system by a capacity degradation taken by utilizing a dual polarized antenna. There are two main issues around which dual polarized antennas may


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effect capacity. The first is the effect it may have on the Eb/No performance. Reverse pole capacity equations depend directly on target Eb/No values. Secondly, dual polarized antennas have smaller front-to-back ratios than singularly polarized antennas which means they tend to introduce more noise into the system (reducing capacity).15 How significant a degradation on capacity is yet to be determined.

Although the findings show that a dual polarized antenna with a slant 45° configuration performs better than a horizontal/vertical configuration, a significant concern lies in what may happen to the forward link and power control by transmitting with a 45° polarization. The horizontal/vertical polarization configuration would at least leave one variable constant as compared to the present spacial diversity scheme (the base station transmit antenna would continue to be vertically, linearly, polarized). However, to achieve the same level of decorrelation in branches would require a more dense clutter environment, and even then, studies show that horizontal/vertical polarized antennas tend to have a large branch imbalance with the vertical branch being most dominant. Ideally the choice would be to use the 3-port antenna model presented earlier which uses three polarizations (vertical and slant 45°). The cost and extra weight added (for identical antenna gain) would need to be determined, and factored into the decision process.

Finally, the Motorola field tests found that the greatest factor to impact the diversity gain was branch imbalance (more so than signal correlation). In measuring Eb/No requirements of a polarization diversity system as compared to a spacial diversity system, it was found that there was little change to the gain between the systems. The polarization diversity scheme showed a slight (less than 1 dB) degradation in performance. This study was performed in an urban environment.

Whether or not to use polarization diversity is left to the system engineer. The recommendation is to use this scheme if real estate is not available for spacial diversity, and the environment clutter type is urban or dense urban. Isolation between ports needs to be strictly adhered to and Nokia Siemens Networks recommends a minimum of 34 dB between Tx and Rx ports. It is unknown how transmission at 45° may effect the forward link, and until further studies are performed, the system engineer should be cautious in using this approach.

7.2 In-Building Distributed Antenna Systems

With the proliferation of portable cellular phones, wireless subscribers not only require service on roadways, they also desire CDMA cellular service within buildings. Typical in-building applications include:

• Office Buildings• Airports• Hospitals• Shopping Malls• Hotels & Convention Centers• Sports Arenas• Colleges & Universities

15. This is dependent on antenna manufacturer.


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Improving indoor coverage is an important step towards meeting the customer’s expectation. By choosing the proper design philosophy, an indoor system will increase system capacity and improve call quality. One approach to meeting the customer requirement for in-building coverage is to increase the ambient power of the outdoor macro-cellular system, allowing signals to penetrate outer walls and provide coverage within buildings. This method is used with limited success due to the wide variation of building materials and their respective penetration loss. In order to provide high quality in-building cellular service, it may be necessary to place the coverage within the building through the use of Micro sites and distributed antenna systems.

Several methods of in-building coverage solutions exist including passive coax, active coax, fiber optics, leaky feeder, Micro RF heads and hybrid combinations of these types. Each approach has a unique set of attributes, which makes it most suited for a particular application.

7.2.1 In-Building System Architecture Overview

The goal of in-building system design is to distribute the RF signal uniformly throughout all of the areas to be covered. The system should be easy to install, inexpensive, unobtrusive, and highly reliable. Distributing antennas within the building, using coaxial cable, fiber optics cable or Pico RF heads, can meet these requirements. Figure 7-11 illustrates a typical coaxial cable design approach.

Figure 7-11: Coaxial Cable Design Approach

The coaxial cable approach uses splitters, directional couplers, or taps to direct the RF signal to various locations within the building.

Current fiber optic distribution systems employ a star architecture. In large buildings, the fiber runs may be reduced by distributing the fiber control units as in Figure 7-12.


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Figure 7-12: Fiber Optic Design Approach

Passive coaxial cable systems typically offer the most cost effective implementation solution for small building applications. The term passive coax is used to describe a coaxial cable system that does not have any active devices, such as in-line amplifiers. Coaxial cable systems that do employ amplifiers are referred to as active coax systems. Because of their low cost, passive coax systems should be used to distribute the RF signal whenever practical. The limitation of passive coax is that the cable loss increases as the cable run length increases. Higher cable loss results in lower downlink power and degrades the system uplink performance. For larger buildings, it may not be possible to meet the coverage goals using only passive coax. For these larger applications fiber optic distribution systems, active coaxial cable or Pico-Cell systems can be employed.

7.2.2 Coaxial Cable System Design Using A Link Budget

The following section (Section 7.2.2.1) provides a flow chart with the steps involved with designing an in-building antenna system using a passive coaxial cable system design. The remaining sections provide a brief description of the various steps provided in the flow chart.

7.2.2.1 Design Procedure Flow Chart

The flow chart shown in Figure 7-13 describes a process that can be used for estimating the design of an in-building RF distribution system.

Details on each of the flow chart steps are given in the following sections.

Fiber Optic Cable

RF Base Station

Ceiling

Fiber Antenna Unit

Fiber Base Unit


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Figure 7-13: Coax Design Flow Chart

7.2.2.2 Gathering Building Information

Before the design of the distributed antenna system can begin, some basic information about the building, equipment locations and areas requiring coverage must be obtained. With this information, the system planner can begin to construct the details of the design, such as the number of antennas required throughout the building.

Phase I: Education

Nokia Siemens Networks recommends training be provided for as many key people as possible, especially the individuals who will be participating in the installation, optimization and trial stages.

Gather The Building Information

Determine The Base Station Location

Estimate The Antenna Placement

Select The Antenna Types

Choose The Base Station Type

Choose The Cable Topology

Estimate The Cable Run Lengths

Select The Coaxial Cable Type

Link Budget Design

Meet Coverage Requirements

Estimated Design Is Complete

NO

YES


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Phase II: System Requirements

Clear specifications and requirements are a key to any project. In-building antenna systems are no exception. Nokia Siemens Networks recommends extensive training and design sessions due to the special requirements of in-building systems.

Phase III: Building Design Information

Building design information in the area of traffic requirements, coverage area, and building details are needed.

Traffic requirements:

• The number of subscribers for the system (this will be necessary to size the final equipment)

• The Average Holding time including Erlang studies of the call duration on the PBX or landline

• Peak traffic periods during the day

• Desired Grade Of Service (GOS)

• The type of back-haul spans should also be determined (i.e. Microwave, T1 or E1, etc.)

Coverage Area:

Floor plans, including a scale, are required. The plan dimensions should be clearly legible and should detail the layout of the floor. The height of each floor and clearance above the ceilings should also be detailed. In consultation with the customer and using the floor plans, the number of floors and areas within the building requiring coverage can be determined. This decision can be made by examining the probability that a call will be made (or received) in a particular area. Locations that can be expected to see activity, such as an office space or conference room, should be adequately covered. Locations where call activity will be minimal, such as storage rooms and mechanical sectors, may not need to be covered. There will be a trade-off when determining the coverage requirement. As an example, a design for 90% area coverage may be significantly less expensive than a design for 100% area coverage. The coverage goal should be established prior to beginning the system design and implementation.

Building Details:

Details on the building construction will help with the system design. An arrangement to ensure complete building access must be agreed upon with the customer. Table 7-2 provides examples of building topology that should be discussed to give a more detailed description of the areas to be covered:


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A second marked-up copy of the floor plans can be used to illustrate and describe where these materials or obstacles are located.

The vertical elevator shafts, stairwells, fire escapes and any vertical duct or passageway should be illustrated on the plan or cross-sectional diagram. Any in-building parking facilities should also be described.

Photographs and/or video would also be helpful for more complex implementations.

A brief review should be done to see if the size of the passageways will be sufficient to transfer the equipment from the delivery truck to the installation area. The equipment may need to be un-crated before moving it into the building. The Telco rooms and PBX rooms should be clearly marked as well. Considerations should be given to the provisioning of back-haul transmission connections.

A list of contacts from the customer, designating key individuals to support the project, must also be defined. Individuals to address building code, electrical, plumbing, duct work, and structural questions should also be identified. These individuals should be readily available to answer any on-site questions, especially questions pertaining to cable runs and locations.

With the above information, a spreadsheet design procedure can be used to determine the amount of equipment, cable and antennas required for the desired coverage area. Additionally, preliminary plans for cable runs, equipment locations and antenna placement can be formulated.

7.2.2.3 Determining the Base Station Location

A survey of the building should be carried out to determine the equipment room location and check the cable routing options. Using the floor plan drawings and inputs from the customer, an estimate of the base station location(s) can be made. The base station should be located as centrally as possible within the building. This will minimize the coaxial cable lengths and cable losses thereby

Table 7-2: Building Topology Examples

• glass content • open areas• re-bar • sky lights• metal • atriums• struts • mezzanines• wire mesh • fiberglass insulation• metal-skin walls • steel beams• partial walls • asbestos• full walls • cubicles• floor thickness • basements• floor materials • tunnels• duct work • ceiling plumbing


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providing the best possible downlink and uplink performance. If there are multiple floors to be covered, the base station should be located on the middle floor if possible. All design information should be drawn on the building floor plans.

7.2.2.4 Estimating the Antenna Placement within the Building

The coverage (ERP) from an antenna is dependent on the cable loss from the base station to the antenna. Therefore, RF distribution design is an iterative process. In order to start the process, an estimate of the number of antennas and antenna placement within the building is required. Table 7-3 can be used to determine a first pass estimated coverage radius for each antenna.

Using this estimated coverage radius as a starting point, first pass antenna locations can be derived and drawn on the building floor plans. In a future step, the drawing is used to calculate the cable runs.

A concern with antenna placement is the maximum allowable received power at the subscriber. As with any active device, the subscriber receiver is only designed to operate over a range of input power levels. If the maximum input power level is exceeded, distortion will occur. The input power level received at the subscriber is a function of the distributed antenna system ERP and the minimum distance between the subscriber and the distributed antenna. As a rule of thumb, for low gain, ceiling mounted antennas, the power into the antenna should not exceed 10 dBm. In general, the antenna ERP should be set to a level that will result in no higher than -30 dBm at the subscriber. If these design guidelines are not followed, distortion may occur when a subscriber is used in close proximity to an antenna. This distortion will rapidly decrease as the subscriber moves away from the antenna.

7.2.2.5 Selecting the Antenna Type: Omni vs. Directional

After estimating the antenna placement, the type of antenna(s) to be used must be selected. In general, there are two types of antennas to choose from: omni-directional (omni) and directional antennas. Omni antennas provide a uniform field pattern in 360° in the horizontal.

Directional antennas have increased gain in one or more directions at the expense of reducing the gain in other directions. There are a number of directional antennas available for indoor use with a variety of gain patterns.

Omni antennas can be employed in most cases. Directional antennas are useful when covering an area that is shaped similar to the antenna gain pattern. For example, a long hallway might best be covered by a "bow tie" antenna as in Figure 7-14.

Table 7-3: Estimated Coverage Radius

Area Type Estimated Coverage Radius (Feet)Office 200Factory 350Store 350


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Figure 7-14: "Bow Tie" Antenna

7.2.2.6 Choosing the Base Station Type

There are several parameters that need to be considered when choosing the base station type to be used. The primary variables are:

• Channel capacity

• Maximum downlink power

• Physical size

• Antenna system complexity

• Installation and maintenance

• Cost

For applications requiring only a few traffic channels and minimal forward power, a small microcell product (limited capacity) may prove to be both economical and easy to install.

For applications requiring greater capacity or higher RF penetration, a larger BTS product, for instance a macro site BTS, can be used.

The system designer will need to weigh the attributes of each BTS product to determine the best BTS for their design. For instance, if passive coaxial cable is used to distribute the RF to antennas, it may be preferable to use more BTS products to limit the length of transmission run and thereby minimize cable loss (BTS is placed closer to area to be covered). If fiber optic transmission is being used to distribute the RF, line loss is not as much an issue and therefore, the BTS can be located further from the area to be covered.

7.2.2.7 Choosing the Cable Topology: Splitters, Couplers, and Taps

There will inevitably be a need to split a single coaxial cable branch into multiple branches. For example, a main feeder run may have to be split into two branches to feed two separate antennas.

Bow Tie Antenna


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There are several different approaches to accomplish this signal division:

• RF taps

• Power splitters or dividers

• Directional couplers

Each method has benefits and limitations that must be considered.

7.2.2.7.1 RF Tap

An RF tap acts like a pin hole in a water hose. As the water (RF) flows past the hole, some leaks out. The RF tap is basically a small antenna that is inserted into the main coaxial line which drains a small portion of energy from the tapped branch into the new branch (see Figure 7-15). The drained or coupled energy propagates down the new line.

Figure 7-15: Schematic Diagram of a Power Tap

Standard taps are available from commercial sources and provide a relatively inexpensive way to branch from a main feeder.

The coupling loss indicates how much of the signal will enter the new branch. For example, if the source line is at 0 dBm and the tap has a coupling loss of 12 dB, then the power in the new branch will be -12 dBm. The majority of the energy continues to propagate down the main branch. A typical application for an RF tap would be to branch several antennas off of a main branch. For instance, RF taps could be used to provide an antenna for a meeting room with its own local branch (see Figure 7-16).


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Figure 7-16: Typical Tap Application

7.2.2.7.2 Power Splitter or Divider

The power divider uses a resistive network (or similar approach) to break one input into two or more outputs of equal power. For instance, a two-way splitter may typically have a loss of 3.5 dB, so a 0 dBm signal entering a 2-way splitter will exit as two -3.5 dBm signals (see Figure 7-17).

Figure 7-17: Diagram of a Power Splitter

There are two components to the splitter loss. The first is the loss associated with actually splitting the signal into multiple outputs, and the second is the insertion loss due to resistive loss. This is why a two way splitter has 3.5 dB of loss rather than 3 dB. There are usually a large selection of commercial power splitters available for use. A brief sample of various output ports and their associated loss values are presented in Table 7-4.


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7.2.2.7.3 Directional Coupler

With a directional coupler, most of the signal is transmitted to the through port while a small portion of the signal is diverted to the coupled port. This is similar to a tap; however, the method used to couple the signal is different and, in general, more efficient (see Figure 7-18).

Figure 7-18: Schematic of a Directional Coupler

The directional coupler can be used in situations where a small amount of power needs to be drawn off of a main branch with minimal disturbance. Directional couplers also come in a multitude of values.

7.2.2.7.4 Cable System Distribution Examples

When selecting splitters, taps or directional couplers, there is a choice between parallel or series power distribution. A parallel method would use a splitter to branch the main run into local runs. A series method would use directional couplers or RF taps to divert power from the main cable run to local runs. Both methods work equally well for short runs. However, as cable runs, antennas and branches increase, the series method can provide increased power levels at antennas located furthest from the base station. The benefit of series distribution can be seen in Figure 7-19 and Figure 7-20.

Table 7-4: Typical Values for Power Splitters

# of Output ports Total Loss (dB)

2 3.53 5.84 7.08 10.0


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Figure 7-19: Parallel Power Distribution Using a Power Splitter

In the case of parallel power distribution, the power reaching the antenna closest to the insertion point (250 ft.) has 22 dB more power than the antenna furthest (1000 ft.) from the insertion point. The non-uniform distribution of power will cause an increased coverage area for the antenna closest to the insertion point and a decreased coverage area for the antenna farthest from the insertion point.

Figure 7-20: Series Power Distribution Using Directional Couplers

In the case of series power distribution using directional couplers, three directional couplers will be required: 15 dB, 10 dB and 6 dB. With directional couplers, the power delivered to each antenna is more uniform than with splitters. In addition, there is an 8.5 dB improvement at the final antenna using the series method as opposed to the parallel method. Overall, series distribution may be used when a power improvement is required at a distant antenna, or when multiple cable runs becomes cost prohibitive.

Often, a combination of parallel and series power distribution methods may be used. For example, a power splitter can be used to divide a main branch into several sub-branches; then, directional couplers or RF taps can be used to distribute power from the sub-branch to the antennas.


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7.2.2.8 Estimating Cable Lengths from the Base Station to the Antennas

Once the base station and preliminary antenna locations are marked on the floor plans, estimates for the cable lengths, number of directional couplers, splitters, and taps can be made. The cable runs should be located in standard cable areas within the building. If the standard cable locations are not known, a good rule of thumb is to assume that the cable will run down the hallways. The estimated cable runs and network components (couplers, splitters, and taps) should be drawn on the building floor plans.

7.2.2.9 Selecting the Coaxial Cable Type

There are several alternatives to be considered when selecting the media for delivering the RF signal from the base station to the antenna, and vice versa. Ideally the distribution media should have the following characteristics:

• Low loss

• Flexible

• Durable

• Light weight

• Fire resistant

• Low cost

• Minimum space requirement

There are several varieties of coaxial cable that can meet the above requirements for distributed antenna applications. Each variety of cable has its own advantages. However, there are trade-offs involved in selecting a cable type. For example, cable runs that do not require many turns and bends can utilize typical foam dielectric coaxial cable. This type of cable has low loss, light weight and excellent durability. The cable comes in a variety of sizes, with loss decreasing as diameter increases. However, the larger sizes are less flexible, cost more and suffer from increased weight and space requirements.

A large building with minimal turns and bends can use a larger diameter cable with lower loss. In some cases, the standard low loss cable may not have enough flexibility for the particular application. If the system has numerous turns, or sharp bends, a super-flexible cable may be required. This type of cable trades increased loss for increased flexibility.

If the cable is to be placed in or near air handling spaces, the use of plenum rated cable may be required. The plenum rating specifies that the cable meets certain fire resistance and smoke producing specifications. Although most coaxial cables have a fire retardant option, a plenum rating may be necessary (check local code requirements).

Low loss 7/8" coax may be a good choice for in-building applications. However, because of the higher price of 7/8" coax, 1/2" coax may be desired.


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7.2.2.9.1 Designing With Radiating (Leaky) Coax

Radiating (also known as leaky) cable is a type of coaxial cable which has holes milled in the outer conductor as shown in Figure 7-21.

Figure 7-21: Radiating Cable

A small portion of the RF energy that is transmitted down the radiating cable leaks out from the holes, hence the term "leaky" coax.

Radiating cable can be used in place of point source antennas to provide coverage within buildings as shown in Figure 7-22.

Figure 7-22: Radiating Cable Implementation

Some of the advantages of radiating coax are that the coverage is more uniform and the radiated power levels are low, which improves signal containment and reduces the risk of overloading the subscriber unit. Although radiating cable can be used most anywhere, typical applications have been for elevator shafts, long tunnels and for hallways.

A radiating coax system design is similar to a conventional coaxial cable design. A link budget can be used to tabulate all of the system losses up to the radiating cable. Coupling loss and cable loss per unit distance factors, which can be obtained from vendor data sheets, are used to determine the power level radiated by the cable. The coupling factors are typically specified for a 20 foot distance on either side of the cable as shown in Figure 7-23.


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Figure 7-23: Radiating Cable Coverage

In this example, the input power is 10 dBm and the radiating cable loss per 100 feet is assumed to be 5.0 dB. The power remaining at the end of the cable is 0 dBm. The power received at a distance of 20 feet from the radiating cable is the power remaining at the end of the cable less the coupling loss, which is assumed to be 66 dB for this type of cable.

An indoor propagation model can be used to estimate the path loss between the 20’ mark and the edge of the building to determine the worst case receive signal level.

If the cable is to be placed in or near air handling spaces, the use of plenum rated cable may be required. Although most coaxial cables have a fire retardant option, a plenum rating may be necessary (check local code requirements).

7.2.2.10 Link Budgets For In-Building Design

The link budget provides a means to determine the maximum allowable path loss between the base station and the subscriber unit. The path loss ultimately determines the coverage area, which equates to the amount of equipment necessary to meet the system performance goals. The actual environment of the area to be covered can greatly influence the range to which a site will propagate. The link budget analysis technique takes these environmental characteristics into account. The link budget is an important part of the detailed design which must be done to ensure coverage quality and reliability.

The link budget can be used for passive coax systems or active coax systems. The active design parameters can be included in a spreadsheet tool, although it is more complex than a simple passive design. However, with a combined passive and active design, "what-if" scenarios can be tested to see if using amplifiers will improve system performance.

Nokia Siemens Networks recommends that a passive design be considered first, due to its lower cost.

Figure 7-24 shows a block diagram of the components that enter into the link budget calculation for both the passive and active cases.


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Figure 7-24: Link Budget Block Diagrams

For the passive coax system, the downlink received signal strength at the subscriber is calculated by subtracting the network and propagation losses from the base station transmit power. The uplink received signal strength at the base station is calculated by subtracting the propagation and network losses from the subscriber transmit power. In both directions, the received signal level should be at or above the receive threshold for satisfactory system performance.

With active systems, the amplifier gain must also be included in the link budget. The uplink noise power changes due to the active components and must also be considered when analyzing the system.

The propagation path loss is used to determine the maximum coverage radius from each antenna as shown in Figure 7-25.

AmplifierLossy

Network

Base

Propagation

Loss

Lossy

Network

Active Coax

BasePortable

LossyNetwork

Passive Coax

PropagationLoss

Portable

Downlink

Base TxPower

+Amplifier

Gain-Network

Loss-Propagation

Loss

>= Portable

Rx Threshold-Network

Loss

Uplink

+Amplifier

Gain-Network

Loss-Propagation

Loss

-NetworkLoss

PortableTx Power

>=Base RxThreshold

Uplink

Downlink

Base TxPower

-NetworkLoss

-PropagationLoss

>= PortableRx Threshold

Portable

Tx Power-Propagation

Loss

-Network

Loss

>=Base RxThreshold

Base Station

Base Station


7


Figure 7-25: Maximum Coverage Distance

A floor penetration loss component may also be included as part of the propagation loss within the link budget analysis. Depending on the floor construction materials, it may be possible to cover several floors using one antenna as shown in Figure 7-26. The additional propagation loss due to penetrating the floors must be included in the link budget calculation when using this approach. Suggested floor penetration loss factors are presented in Section 7.2.2.10.1.

Figure 7-26: Multiple Floor Coverage

7.2.2.10.1 Estimating In-Building Path Loss Using Statistical Models

Due to the existence of many variables in an indoor propagation environment, accurate path loss prediction becomes difficult. These variables include floor/ceiling materials and various wall construction materials and geometry, in addition to numerous obstacles between the transmitter and receiver. Presently, there are several methods for predicting path loss for indoor environments. Among these methods are deterministic models, such as ray tracing, site specific diffraction, and wall/material loss models. All of these methods describe the path loss for a variety of circumstances, with a fair amount of accuracy. Their major drawback is computational complexity since they need to account for a large number of variables (such as wall size and material, location of furniture, light fixtures, etc.). In addition, these methods are time consuming and costly.

Base

Floor Loss

Floor Loss


7


Another method is statistical modeling, which has proven to be effective in predicting path loss for indoor environments. Statistical models are based on measurements recorded in various different building types. The main advantage of using statistical models is the simplicity of representing the path loss between the transmitter and a receiver. These path loss calculations are easily implemented in a spreadsheet design tool. All equations in Table 7-5 express path loss as a function of distance only. The path loss equations express the path loss in dB in terms of distance in feet. Included in these models are linear regression methods based on measurements taken in two Motorola facilities16.

Figure 7-27 shows the plots of the logarithmic models in Table 7-5.

Figure 7-27: Logarithmic Path Loss Models

Table 7-5: Path Loss Models

Model Name Path Loss Equation (dB)

Retail Store PL(d) = 22 Log(d) + 20.1

Suburban Office Bldg.open plan

PL(d) = 24 Log(d) + 19.1

Suburban Office Bldg.soft partition

PL(d) = 28 Log(d) + 17.0

Suburban Office Bldg.hard partition

PL(d) = 30 Log (d) + 16.0

Motorola Cluttered PL(d) = 0.18(d) +71

Motorola Uncluttered PL(d) = 0.11(d) +55

University PL(d) = 0.19(d) + 63

Free Space @ 894 MHz PL(d) = 20 Log(d) + 21.1

Free Space @ 1900 MHz PL(d) = 20 Log(d) + 27.7

16. These measurements, except for the Free Space models, were made at cellular frequencies.


7


Figure 7-28 shows the plots of the linear models, based on measurements at Motorola facilities and at a college university. Free space path loss is included in both figures as a reference. The Motorola cluttered model is based on measured data from a Motorola office facility. The office area was comprised of both hard metallic walls and soft walled cubes. The Motorola uncluttered model is based on path loss measurements that were taken in open factory and distribution areas.

Figure 7-28: Linear Path Loss Models

The path loss curves represent the average path loss as a function of distance for a large number of data points. Some areas within the building will have higher path loss and some areas will have lower path loss than the average. The distribution of path loss values around the mean has been found to approximate a log-normal (bell shaped) curve with a standard deviation in the range of about 5 to 10 dB. A fade margin can be added in the link budget, if desired, to account for this log-normal variation of received signal level. The fade margin can be adjusted to achieve the desired percent area coverage.

In addition to these path loss models, floor attenuation factors (FAFs) have been developed by S.Y. Seidel and T. S. Rappaport based on thousands of signal strength measurements taken in two multiple floor buildings. A summary of their experimental results is listed in Table 7-6. The values in Table 7-6 are the average floor attenuation factors with their respective standard deviations.

Table 7-6: Average Floor Loss Attenuation Factors

Location FAF(dB) σ (dB)Office Building 1Through 1 floor 12.9 7.0Through 2 floors 18.7 2.8Through 3 floors 24.4 1.7Through 4 floors 27.0 1.5Office building 2Through 1 floor 16.2 2.9Through 2 floors 27.5 5.4Through 3 floors 31.6 7.2


7


The model that most closely resembles the particular area to be covered should be used. Each antenna can have a different path loss model associated with it, depending on the area type that it will cover.

Rather than using one of the statistical path loss models, it is possible to make on-site measurements for the purpose of determining the path loss characteristics of a particular environment.

7.2.2.10.2 Measuring In-building Path Loss

Signal strength measurements can be performed on-site, so that the transmit power requirements can be evaluated and established. This information can be used to decide very accurately how many antennas will be needed to provide adequate coverage throughout the building.

A test transmitter is set up in the area to be covered. An antenna is connected to the test transmitter and signal strength measurements are recorded systematically within the expected coverage area. These measurements may be taken manually with a measuring receiver or automatically with a commercially available data collection system.

The data collection system produces coverage plots to determine the signal strengths within the building. Since this is a downlink coverage measurement, care must be taken to insure that the uplink coverage will also be adequate. A link budget tool can be used in conjunction with the measured coverage data to insure that both the downlink and uplink coverage requirements are met.

Figure 7-29: Measurement System Test Setup

This design approach is more costly due to the required on-site visit. Approximately one day is required at the building location to view the equipment room, likely cable routes, and antenna locations. A test transmitter can be set up in representative areas and coverage data can be collected using a portable data collection system, or other test receiver. Using the on-site test results, an accurate system design can be developed to meet the coverage requirements.


7


7.2.2.11 Evaluating the First Pass and Iterating the Design

After all of the antenna link budget information has been calculated and the propagation path loss determined using a path loss model or by actual measurements, the results can be compared to the coverage goals. If the design does not meet the coverage requirements, a second pass should be completed. The design may be improved by adding antennas, using lower loss cable and/or changing the cable topology. Several iterations may be necessary to reach a point where all of the coverage objectives are met.

If the analysis shows that the coverage margins are excessive, some antennas should be removed to reduce the system cost. Depending on the implementation method and building structure, it may only be necessary to use one antenna for every two or three floors of coverage.

If after several iterations, the coverage objectives cannot be reached using a simple passive coax design, two alternatives can be investigated. These alternatives are:

• Fiber optic distribution

• Active coax distribution using bi-directional or uni-directional amplifiers to overcome cable losses

The use of in-line amplifiers must be considered carefully because of the higher cost and implementation and maintenance complexity. Employing lower loss coaxial cable and locating the base station as near as possible to the center of the coverage area is recommended. If the system cannot be designed using passive coax, then in-line amplifiers or fiber optics must be considered.

7.2.3 Active Coaxial Cable System Design

The following sections discuss the technical issues and design alternatives for active coaxial cable system planning.

Bi-directional or uni-directional amplifiers can be used to overcome the cable and network losses in an RF distribution system. Bi-directional amplifiers provide amplification in both the uplink and downlink direction as shown in Figure 7-30.

Figure 7-30: Bi-Directional Amplifier


7


This is in contrast to a uni-directional amplifier that only provides amplification in one direction. A possible implementation for an uplink uni-directional amplifier is shown in Figure 7-31.

Figure 7-31: Uni-Directional Uplink Amplifier

In general, the bi-directional amplifier can be used when the uplink and downlink are nearly balanced. A uni-directional amplifier can be used to improve performance when the system is uplink limited, possibly due to the use of high power LPAs on the downlink.

7.2.3.1 Downlink Amplifier Design Considerations

The main concerns when using downlink amplification are the amplifier’s maximum composite output power, gain, and intermodulation performance. The amplifiers that are used in RF distribution systems have a maximum composite output power that must be shared among all of the carriers. In order to keep the intermodulation product levels within specification, the power output per carrier must be limited to a maximum value based on the amplifier’s specifications.

The downlink parameters related to an active system can be described as follows:

In-Line Amplifier Gain: The amplifier gain is usually adjustable within a given range.

If the calculated required gain is lower than the minimum amplifier gain setting, then an attenuator must be used in front of the amplifier, as shown in Figure 7-32, to reduce the input signal level.

Uni-Directional Uplink Amplifier

Duplexer


7


Figure 7-32: Downlink Amplifier Gain

Power Out of In-line Amplifier: The power out of the in-line amplifier is calculated as the input power plus the amplifier gain. The calculated output power of the amplifier should not exceed the manufacturer's specification for the amplifier.

The amplifier gain is typically adjustable, thus the gain of the amplifier must be set at a level that will insure that the maximum composite output power specification is not exceeded.

7.2.3.2 Uplink Amplifier Design Considerations

In order to fully understand the uplink performance characteristics of an active coaxial cable system, it is necessary to understand some of the fundamentals of receiver system design. These receiver system design basics are discussed as follows.

7.2.3.2.1 Receiver System Fundamentals

Noise Figure: By definition, noise figure (NF) is the difference between the input signal-to-noise ratio and the output signal-to-noise ratio in dB.

NF = (S/N)in - (S/N)out (dB)

Minimum Amplifier Gain = 30 dB Maximum Amplifier Output = 17 dBm Per Channel

Base- 20 dB

27 dBm 7 dBm 37 dBm

+30 dB

Power =

Network Loss

Base- 20 dB

27 dBm 7 dBm 17 dBm

+30 dB

Power =

Network Loss

-20 dB

Pad

Too High

In Spec.-13 dBm


7


Figure 7-33 illustrates the effect of a 10 dB noise figure amplifier with 10 dB Gain.

Figure 7-33: Effect of a 10 dB Noise Figure Amplifier

The input signal and noise is amplified by 10 dB and the output noise is also increased by the noise figure of 10 dB. Therefore, the total increase in the noise floor is 20 dB.

Noise Figure of a Lossy Device: The noise figure of a lossy device, such as a length of a coaxial cable, filter, splitter, or attenuator is equal to the loss of the device. Figure 7-34 illustrates this concept.

Figure 7-34: Noise Figure of a Lossy Device

The term kTB in this figure is used to represent the thermal noise in dBm. The thermal noise level is the same at the input as it is at the output of a lossy device. However, the signal level has dropped by the amount equal to the device loss. Therefore, the signal-to-noise ratio at the output of the lossy device is lower than that at the input by an amount equal to the device loss. Hence, the noise figure is equal to the device loss.

Cascaded Noise Figure: When two or more system blocks are cascaded together as in Figure 3.20, the cascaded noise figure formula can be used to determine the total system noise figure.


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System NF(dB) =

Where F1, F2, F3, F4.... are the stage noise figures in linear terms, and G1, G2, G3, G4.... are the stage gains (the gains will be less than one for lossy system blocks), also in linear terms.

Figure 7-35: Cascaded System Noise Figure

As seen in the network drawing, the blocks that are cascaded can be active devices or lossy network devices such as coaxial cable, splitters, couplers, attenuators, or filters.

Sensitivity: The receiver sensitivity is defined as the minimum allowable receive signal level that will result in a given audio quality, as specified by audio signal-to-noise ratio or audio SINAD. SINAD is similar to signal-to-noise, and is defined as the ratio of the Signal plus Noise plus Distortion to Noise plus Distortion. This can be related to a statistical number called the Bit Error Rate or BER for digital systems.

SINAD = (S + N + D)/(N + D)

System sensitivity can be calculated as follows:

Where:k Boltzmann’s constant = 1.38 x 10-23 W/(Hz K)

T Room temperature of 290° Kelvin

B Bandwidth of the carrier in Hz

NF Noise figure of the equipment

Eb/No Energy bit density over noise

R Information bit rate

10Log10 F1F2 1–

G1---------------- F3 1–

G1 G2⋅------------------- F4 1–

G1 G2 G3⋅ ⋅-------------------------------- …+ + + +

Base

NetworkLoss

NetworkLossAmplifier

F4 G4 F3

G3F1 G1

F2 G2

System NF

SensitivitydBm kTB( )dBm NF( )dB Eb No⁄( )dBBR---

dB–+ +=


7


The value kTB is the noise power at the receiver input due to thermal noise in dBm. For CDMA, the thermal noise power is -113 dBm.

As a point of reference, the SINAD for land line call quality ranges between 25 dB and 40 dB. In order to have acceptable call quality in a fading environment, a higher minimum signal strength is required.

When amplifiers are used, the uplink noise figure is increased and therefore the receive threshold must also be increased by the same amount to maintain the same call quality.

7.2.3.2.2 Uplink Design Parameters

The main parameters of concern for an uplink amplifier are noise figure, gain and 3rd order intermodulation performance.

The amplifier gain is typically adjustable within a specified window. Since the amplifier gain enters into the cascaded noise figure it must be set as part of the design procedure. It is customary to set the amplifier gain equal to the cable and network losses between the amplifier and the base station as shown in Figure 7-36.

Figure 7-36: Uplink Amplifier Gain Setting

There is no advantage to increasing the amplifier gain above the level of the network losses. In fact, raising the gain will degrade the system intermodulation performance because both the received signal and the input noise are amplified equally. As such, there is no improvement in the output signal-to-noise ratio when increasing the amplifier gain above the network losses.

7.2.3.2.3 Uplink Link Budgets For Active Coax Systems

The uplink parameters related to an active coax system can be described as follows:

Amplifier Noise Figure: The amplifier noise figure, which can be obtained from the manufacturer's data sheet is entered in dB.

Amplifier Gain: The amplifier gain is set equal to the network losses between the amplifier and the base station.

Base Station Noise Figure: The noise figure for the Nokia Siemens Networks base station is


7


entered in dB. A maximum value of 6 to 7 dB is normally used for the estimated design. Typical base station noise figure values are approximately 4.5 dB.

Noise Summing Degradation: When two or more amplifiers are used in parallel within a network, the noise power from each amplifier adds together. The end result is to raise the system noise floor. Figure 7-37 illustrates this concept.

Figure 7-37: Noise Summing

Receiver Noise Rise: To determine the noise rise, which is the amount by which the uplink receive threshold should be increased, both the cascaded noise figure and the noise summing degradations are taken into account.

System Noise Figure Without Amplifiers: It is important to determine if the addition of uplink amplifiers is actually improving uplink performance. The uplink system noise figure is calculated for the case where amplifiers are not used and compared to when amplifiers are used. The noise figure for the system, excluding amplifiers, is simply the sum of all of the uplink network losses and the base station noise figure.

System Noise Figure With Amplifiers: The cascaded noise figure equation is used to determine the system noise figure with in-line amplifiers. The noise summing degradation is also added in the system noise figure calculation.

Amplifier Uplink Improvement: The amplifier uplink improvement is the difference between the system noise figure without amplifiers (passive system) and the system noise figure with amplifiers. Since uplink amplifiers can only overcome losses between the amplifier and the base station, the addition of an in-line amplifier may actually degrade system performance. If the result of an amplifier uplink improvement calculation is negative, then the amplifier has actually degraded the system uplink performance.

System Level Receive Threshold: Assume that the uplink amplifier gain has been set to be equal to the loss between the amplifier and the base station. The receive threshold is being increased to overcome the noise added by the amplifiers. At first glance, it may appear that the amplifiers are not improving system performance since the receiver noise is increased. However, the receiver noise is likely increased by a few dB, while the uplink amplifier can overcome 30 to 40 dB of cable/network loss, when properly located in the system.


7


7.2.3.3 Optimizing Amplifier Placement

Once it has been determined that an amplifier is necessary, the next step is to decide where the amplifier should be located in the network. In general, the most improvement in coverage will be obtained by placing the amplifier as near to the antenna as possible. On the downlink side, this will reduce the line loss between the amplifier output and the antenna. On the uplink side, this will provide the best improvement in system noise figure and sensitivity. There will usually be a trade-off between how close the amplifiers are located to an antenna and the number of amplifiers needed in the system. Figure 7-38 illustrates this idea.

Figure 7-38: Amplifier Location

In Configuration 1, the amplifier is placed after the splitter so that only one amplifier is required for two antennas. In Configuration 2 the maximum improvement in system performance is achieved by placing two amplifiers before the splitter/combiner.

Splitter

Splitter

Configuration 1

Configuration 2

Uplink Signal Direction


7


Figure 7-39 illustrates the performance trade-off associated with moving the amplifier further away from the antenna.

Figure 7-39: Amplifier Performance vs. Location


7


For Scenario 1: There is no amplifier and the system noise figure is simply the base station noise figure plus the cable loss (NF = 52 dB).

For Scenario 2: An uplink amplifier is placed relatively near the antenna. This improves the uplink noise figure by 36 dB (from 52 dB to 16 dB). The improvement is nearly equal to the loss between the amplifier and the base station (Loss = 40 dB).

For Scenario 3: The amplifier has been located too close to the base station producing a system noise figure of 51 dB. Only a 1 dB improvement in the system noise figure and sensitivity is provided for this configuration. Scenario 3 demonstrates that there is no advantage to using an uplink amplifier close to the base station.

In summary, placing the uplink amplifier close to the antenna is analogous to using tower mounted amplifiers in a macro-cellular system. The amplifier gain compensates for the coaxial cable line loss, thereby increasing performance.

7.2.4 Fiber Optics for In-Building Systems

The following sections provide information regarding fiber optic system architecture and design.

7.2.4.1 Fiber Optic Distribution System Architecture

A fiber optic distribution system employs a fiber optic base unit along with a number of fiber optic antenna units to distribute RF throughout a building. Figure 7-40 illustrates the star architecture.

Figure 7-40: Fiber Optic Star Architecture

7.2.4.2 When To Use Fiber Optics

Fiber optic systems for distributing RF in buildings offer a number of advantages over coaxial cable as follows:


7


Low Cable loss: The attenuation of fiber cable is on the order of 1.5 dB per mile. For in-building applications, the cable loss is negligible. This significantly eases the system design and implementation tasks.

Installation flexibility: Since the fiber optic cable loss is negligible, deviations from the planned cable route during the installation process will not affect the system performance. In coax systems deviations from the designed cable route can result in more cable loss and degraded system performance. Deviations from the planned cable route are common because the building drawings used to lay out the cable runs are not always complete or up to date.

Reduced Interference: Optical cable does not radiate, which eliminates any electromagnetic interference concerns for the optical cables.

Installation Ease: Optical cable is flexible and light weight, which simplifies the system installation.

The main drawback to fiber optics is the relative expense. The fiber optic cable itself is generally less expensive than coaxial cable; however, the cost of the active fiber base unit and the active fiber antennas add to the system expense.

Fiber optic systems in general will not make good economic sense for smaller implementations, where low cost coaxial cable can be employed to provide good system performance. For larger facilities, where very long cable lengths may be required, either active coax systems or fiber optics may need to be employed. For larger implementations, the cost of a fiber optic system is approximately the same as an equivalent active coaxial cable system. Fiber optic systems have several advantages over active coaxial systems:

• Easier to engineer

• Less sensitive to installation variations

• Easier to install

• Easier to maintain

7.2.4.3 Fiber Optic System Design

Fiber optic distribution systems are less complicated to design than active coax systems. The downlink ERP and uplink receive threshold are essentially the same for every antenna unit. It doesn’t matter whether the antenna is located near to the base unit or at a great distance from the base unit. If the propagation environment within the building is somewhat uniform, then the coverage radius for each fiber antenna will be the same. If there are different propagation environments, such as factory and office areas, then a different coverage radius would be expected for each area type; however, within a given area type, the coverage radius from each antenna unit would be approximately the same.


7


Estimated Number of Antenna Units

With the assumption that the uplink coverage is noise limited, the number of antenna units used will govern the uplink coverage. This is because each fiber optic antenna unit and base transceiver pair have active devices that contribute noise in the uplink direction as depicted in Figure 7-41.

Figure 7-41: Fiber Uplink Noise Summing

The noise sums together at the RF combiner. The total noise power increases and the coverage area decreases as more antenna units are used. The reduced coverage radius is due to the increase in uplink noise associated with adding more fiber links.

A good starting point for the fiber optic design is to assume that at least ten fiber antenna units will be employed. Between one and ten antennas, the uplink coverage radius decreases very rapidly. Above ten antennas, the uplink coverage radius decreases at a much more gradual rate. Even if only a few antennas are expected for the initial implementation, it is preferable to design the system using the coverage radius associated with ten antennas. In this way, there will be little impact on system performance if additional antennas are required at a later date to expand the system.

7.2.5 In-Building Antenna Systems Summary

A number of in-building distribution system alternatives have been presented. Because the cost of passive coaxial RF distribution systems is typically lower, it should be used whenever practical. For larger facilities, fiber optics can be used to distribute the RF signal. Fiber offers the key advantage of negligibly low cable loss, which eases system design and implementation. Another alternative for larger buildings is to employ active coaxial cable systems. Care must be taken when using in-line amplifiers to insure that amplifier gain and noise are properly accounted for.

The design process discussed here can be used to obtain an estimate of the in-building system requirements. For larger, more complex buildings, or when a firm quotation is required, a more accurate site survey method should be used.


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7.3 References

1. Wahlberg, Ulrik. 1997. “Polarization Diversity for Cellular Base Stations at 1800 MHz.” Revision 1.0. Allgon.

2. Jakes, William C. 1974. “Microwave Mobile Communications.” New York. American Telephone and Telegraph Company. Reissued in Cooperation with IEEE Communications Society. pp. 309-324.

3. Kozono, S. 1985. “Base Station Polarization Diversity Reception for Mobile Radio.” IEEE Transactions on Vehicular Technology. Vol. VT-33. No. 4. pp. 301-306.

4. Vaughan, Rodney G. 1990. “Polarization Diversity in Mobile Communications.” IEEE Transactions on Vehicular Technology. Vol. 39. No. 3. (August): 177-186.

5. Tobin, Joe, Rob Nikides, Devesh Patel, Edward Golovin. 1997. “CDMA Dual Pole Antenna Testing - Arlington Heights, IL.” Version 1.0. Motorola.

6. Golovin, Edward. 1998. “A Comparison of CDMA Reverse Link Performance with Base Station Spatial and Polarization Diversity Reception (Motorola Israel Measurement Campaign in Urban Area at 900 MHz)” Version 2.0. Motorola.

7. Xiang, Jun. 1996. “Diversity Antenna Systems for GSM900/GSM1800/PCS1900 Networks.” Issue A. Motorola.

8. Rappaport, Theodore S., and Sandhu, Sandip, "Radio-Wave Propagation for Emerging Wireless Personal-Communication Systems". IEEE Antennas and Propagation Magazine, Vol. 36, No. 5, October 1994.

9. Fennick, John, Quality Measures and the Design of Telecommunications Systems. Artech House, Inc., 1988.


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8 Synchronization of the
Chapter
8

Table of Contents

CDMA System


8.1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 38.1.1 Synchronization Requirements. . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 3

8.2 The Global Positioning System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 48.2.1 Satellite Constellation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 48.2.2 GPS RF Carrier. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 5

8.3 Typical GPS Antenna Configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 58.3.1 Active GPS Antenna. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 6

8.3.1.1 Antenna Element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 68.3.1.2 Low Noise Amplifier (LNA) / Pre-selector Filter . . . . . . . . 8 - 78.3.1.3 Overall GPS Antenna RF Requirements . . . . . . . . . . . . . . . 8 - 7

8.3.2 Antenna Positioning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 88.3.2.1 Required Antenna Visibility . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 88.3.2.2 Antenna Placement Optimization. . . . . . . . . . . . . . . . . . . . . 8 - 98.3.2.3 Lightning Protection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 108.3.2.4 Antenna Blockage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 118.3.2.5 RF Interference. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 11

8.3.3 RF Cabling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 118.3.4 Lightning Arrestor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 128.3.5 Signal Splitter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 138.3.6 GPS Antenna System RF Requirements . . . . . . . . . . . . . . . . . . . 8 - 14

8.3.6.1 Antenna System Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 148.3.6.2 Antenna System Noise Figure . . . . . . . . . . . . . . . . . . . . . . . 8 - 15

8.4 Remote GPS (RGPS) Receiver . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 178.4.1 SC24/48/72xx, SC480 Frame RGPS Operation. . . . . . . . . . . . . . 8 - 178.4.2 UBS / M810 Frame RGPS Operation . . . . . . . . . . . . . . . . . . . . . 8 - 198.4.3 UBS / M810 Synchronization Sharing . . . . . . . . . . . . . . . . . . . . . 8 - 19

8.5 CSM / CSA GPS Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 208.5.1 CSM / CSA Configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 21

8.5.1.1 <CSMRefSrc1> Parameter. . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 218.5.1.2 <CSMRefSrc2> Parameter. . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 228.5.1.3 <BTSLatGps> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 238.5.1.4 <BTSLongGps> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 248.5.1.5 <BTSHeightGps> Parameter . . . . . . . . . . . . . . . . . . . . . . . . 8 - 248.5.1.6 <LocAccuracy> Parameter. . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 258.5.1.7 <GPSAntDelay> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 258.5.1.8 <HeightMode> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 25

8.6 UBS GPS Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 26


8

CDMA/CDMA2000 1X RF Planning Guide8 Synchronization of the CDMA System

8.6.1 UBS Configuration. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 268.6.1.1 <latitude> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 268.6.1.2 <longitude> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 268.6.1.3 <antheight> Parameter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 278.6.1.4 <locaccuracy> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 278.6.1.5 <gpsantdelay> Parameter . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 27

8.7 Typical GPS Receiver Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 278.7.1 CSM / CSA GPS Receiver Status . . . . . . . . . . . . . . . . . . . . . . . . 8 - 288.7.2 UBS / M810 GPS Receiver Status . . . . . . . . . . . . . . . . . . . . . . . . 8 - 28

8.8 Cellsite GPS Failures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 - 298.8.1 Non-Synchronous BTS (Emergency) Operation . . . . . . . . . . . . . 8 - 29

8.9 Appendix A – GPS Antenna Kit Installation Instructions. . . . . . . . . . . . . . . . . 8 - 30


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8.1 Introduction

A CDMA spread spectrum digital cellular system requires a degree of synchronization between base stations not required by other cellular technologies. As defined in the Physical Layer Standard for cdma2000 Spread Spectrum Systems standard (TIA/EIA/IS-2000), CDMA base stations must contain a time base reference from which all time critical transmissions, including pilot PN sequences, frames, and Walsh functions, must be derived. This time base reference must be aligned to CDMA system time and provide a means to maintain time alignment in the event that the external source of system time is lost. CDMA system time is defined to begin on January 6, 1980 00:00:00 Universal Coordinated Time (UTC), which coincides with the start of GPS time, but does not incorporate UTC leap second adjustments to system time clocks (i.e. GPS time).

The synchronization of Nokia Siemens Networks cdmaOne/cdma2000 standard CDMA base station transmissions is primarily provided by a Global Positioning System (GPS) timing receiver. A high degree of reliability can be expected from the Base Transceiver Station (BTS) GPS timing receiver but the overall receiver performance is heavily dependant on the GPS antenna installation. To realize the expected GPS performance the GPS antenna installation must satisfy minimum requirements regarding antenna placement, cabling, lightning protection and interference suppression.

During any interruptions in the availability of the primary GPS reference, BTS synchronization can be maintained for a minimum time interval by an MSO (Medium Stability Oscillator), HSO (High Stability Oscillator) or QHSO (Quartz High Stability Oscillator) backup oscillator.

This chapter provides a brief overview of the GPS followed by a summary of CDMA base station GPS antenna installation requirements and guidelines for optimizing antenna installations.

8.1.1 Synchronization Requirements

The following synchronization requirements defined in the TIA/EIA-97E “Recommended Minimum Performance Standards for Base Stations Supporting Dual Mode Spread Spectrum Systems” standard are necessary for proper cdma2000 system operation.

Frequency Tolerance

• Origin: TIA/EIA-97E Specification, Section 4.1.2.3

• Requirement: For all operating temperatures specified by the manufacturer, the average frequency difference between the actual CDMA transmit carrier frequency and specified CDMA transmit frequency assignment shall be less than ±5x10-8 of the frequency assignment (±0.05 ppm).


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Pilot Time Tolerance

8.2 The Global Positioning System

The C/A (Coarse Acquisition) Global Positioning System (GPS) is a space based radio–navigation system that provides precise three dimensional (3D) positioning and time of day information to civilian users.

8.2.1 Satellite Constellation

The GPS is designed to operate with a minimum constellation of 21 operational and 3 active spare (24 total) satellites. The GPS satellite constellation is organized with 6 orbital planes each with an inclination of 55 degrees to the equator and an altitude of 20,200 kilometers. GPS satellite orbits are non-geosynchronous and complete one orbit in approximately 12 hours. It therefore is

• Origin: TIA/EIA-97E Specification, Section 4.2.1.1

• Requirement: Each base station shall use a time base reference from which all time-critical CDMA transmissions, including pilot PN sequences, frames and Walsh functions, shall be derived. The time base reference shall be time-aligned to CDMA System Time, as described in section 1.2 of the “3GPP2 C.S0002-B, Physical Layer Standard for cdma 2000 Spread Spectrum Systems” (TIA/IEA/IS-2000.1-B/C). Reliable external means should be provided at each base station to synchronize each base station time base reference to CDMA System Time. Each base station should use a frequency reference with sufficient accuracy to maintain time alignment to CDMA System Time. With the external source of CDMA System Time disconnected, the base station shall maintain transmit timing to within ±10 S of CDMA System Time for a period of not less than 8 hours.

• Minimum Standard: For all base stations the pilot time alignment error should be less than ±3 S and shall be less than ±10 S. For base stations supporting multiple simultaneous CDMA channels, the pilot time tolerance of all CDMA channels radiated by a base station shall be within ±1 S of each other.

μ

μμ

μ


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important to understand that a GPS receiver will be exposed to a varying (but predictable) constellation of satellites which are constantly moving across the sky.

Figure 8-1: Cellsite GPS Satellite Visibility

8.2.2 GPS RF Carrier

All C/A GPS satellites employ 1.023MHz wide spread spectrum modulation on a common carrier frequency of 1.57542GHz. GPS transmissions received at the surface of the Earth should have a minimum RF signal strength of -160dBW.

8.3 Typical GPS Antenna Configuration

As shown in Figure 8-2, Typical RF GPS antenna Configuration Diagram a typical cellsite installation consists of an active GPS antenna, low loss RF cabling, lightning arrestor and optional signal splitter (for use in multi-frame logical BTS configurations). A description of the GPS antenna system elements along with relevant installation requirements will be presented in following subsections of this document.

A complete GPS antenna kit that includes an active GPS antenna, antenna mast, mounting

GPS Antenna

CDMA Cellsite

GPS Satellite2

Carrier Frequency = 1.57542G

Hz +1.023M

Hz

GPS Satellite1

GPS Satellite3

GPS SatelliteN

The fully operational GPS constellationconsists of a minimum of 24 satellitesarranged in 6 orbital planes. GPS satellitesoperate in a non-geosynchronous orbit whereone orbit is completed in approximatelytwelve hours. The minimum GPSconstellation is arranged such that aminimum of 4 satellites are in view anywhereon the Earth with an unobstructed view.Typically 6 to 8 satellites are in view with thecurrent GPS constellation.

The GPS Constellation


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hardware, grounding accessories, lightning arrestor, weatherproofing materials and a 9.1m length of FSJ4-50B 12.7mm diameter Superflex cable is available as ancillary equipment under the following part number: CGDSAWGPSKIT12. An RF GPS antenna is the recommended solution for most cellsite installations for distances up to ~100 meters. For these applications it provides the lowest overall cost solution, simplest installation / maintenance yet provides excellent ruggedness when properly installed.

Figure 8-2: Typical RF GPS Antenna Configuration Diagram

8.3.1 Active GPS Antenna

The GPS antenna is used to capture and amplify transmissions from multiple in view GPS satellites while providing adequate rejection of out of band signals. The GPS antenna consists of an antenna element, pre-selector filter and Low Noise Amplifier (LNA). The Motorola Timing-2000 (no longer in production), Andrew Corporation GPS-QBW-20N and Synergy-Systems Timing-3000 antennas have been certified for use with Nokia Siemens Networks CDMA base station products.

8.3.1.1 Antenna Element

The antenna element must be capable of receiving transmissions from multiple GPS satellites while in view throughout their orbit. A suitable GPS antenna will have a nearly uniform gain throughout a 360 azimuth and to within 20 above the horizon. Extending antenna gain to elevations of less than 20 offers little benefit in terms of satellite tracking and can actually make the GPS receiver more vulnerable to terrestrial interference sources.

° °°

Optional Equipment(For use in multi-frame

installations)

LightningArrestor

GPS Antenna

Low Loss RF Cabling

Cellsite Cable"Entry Point"

CellsiteSingle Point

Ground

Earth Ground

To

Exp

ansion Fra

mes

GPS SignalSplitter

SC24/48xx SC7224 UBS Macro

SC480

M810

INOutA

OutC

OutB

OutD

To RF GPS AntConnection

To E-GPS AntConnection

To RF GPS AntConnection

CDMA MODEM Frame RF GPS Antenna Connection

ExpansionFrame


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8.3.1.2 Low Noise Amplifier (LNA) / Pre-selector Filter

An active antenna is utilized to minimize antenna system noise figure and maximize usable antenna cable lengths. A double or triple filtered antenna is recommended to provide a high level of LNA overload and interference rejection.

The active antenna LNA is powered by a +5VDC ±0.5V bias provided from the CDMA BTS through the GPS antenna cabling. The CDMA BTS can source a maximum antenna supply current of 80mA to power the antenna LNA and an active signal splitter (if used). A current draw of greater than 80mA from the GPS receiver antenna port may result in a “GPS Receiver Failed Self-Test Failure” (28-10050) or “Source Unavailable” (28-29052) alarm.

8.3.1.3 Overall GPS Antenna RF Requirements

The Nokia Siemens Networks recommended GPS antennas meet the requirements outlined in Table 8-1, Recommended GPS Antenna Specifications and have proven quality with good field performance. While it is possible to use other antennas, considerable care must be exercised in selecting alternatives to insure full BTS compatibility for all conditions.

Table 8-1: Recommended GPS Antenna Specifications

Requirement / Recommendations

Recommended Antenna Specifications

Motorola Timing-2000 / Synergy-Systems

Timing-3000

Andrew GPS-QBW-20N

Gain +25dB (recommended) 25dB 24.5dB ±3dB

Noise Figure <2.5dB (recommended) 1.5dB (Typical) <1.8dB

Operating Frequency

1575.42 MHz with less than 3dB attenuation at a

±1.2MHz offset from center (Required)

1575.42 MHz ±2 MHz 1575.42 MHz ±1.2 MHz

Filtering >40dB attenuation at ±50 MHz offset from L1

center (Recommended)

>40dB at ±50 MHz off-set from L1 center

>60dB at ±50 MHz offset from L1 center

LNA Supply Voltage

Operation with a supply voltage of +5VDC ±0.25V (Required)

5VDC ±0.25V 5VDC ±0.5V

LNA Supply Current

<30mA (Required) 26mA 25mA


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8.3.2 Antenna Positioning

GPS antenna placement can be one of the most critical factors in realizing reliable BTS synchronization. Considerations of several (often conflicting) factors are required in determining the optimal GPS antenna position. The GPS antenna position should ideally be chosen to provide a maximum view to the horizon in all directions. However, sacrifices in antenna visibility are often necessary when considering other factors such as RF interference from nearby transmit antennas or potential antenna damage due to impact from falling ice or lightning strikes. For example the placement of a GPS antenna atop a cellular tower would likely provide a maximum satellite visibility but would be a poor positioning choice due to potential lightning damage or interference from collocated cellsite transmit antennas.

Consideration must also be given to changing environmental conditions when selecting a GPS antenna location. Dense foliage can significantly attenuate GPS transmissions and may not cause problems during fall or winter months but could impact GPS reception during other seasons or with continued tree growth.

In most cases suitable compromises can be made to obtain the necessary mix of GPS satellite visibility and protection of antenna hardware.

8.3.2.1 Required Antenna Visibility

A line of sight view is required between the GPS antenna and any satellites used in determining a timing solution. Before the GPS receiver can provide accurate timing information it must first determine the location of its antenna in 3D space. Information from a minimum of four (4) satellites is required to determine the 3D antenna location. After the GPS receiver has determined its antenna location it can continue to provide accurate timing information with as few as one (1) satellite in view.

Because GPS satellites operate in non-geosynchronous orbital patterns the GPS antenna must be prepared to receive GPS signals from most any direction. The minimum GPS constellation guarantees that a minimum of four (4) GPS satellites will be in view from any location on the Earth

Azimuth Coverage

360 (Required) 360 360

Elevation Coverage

10 to 90 (Required) 10 to 90 10 to 90

Operating Temperature

-40 C to +85 C (Recommended)

-40 C to +85 C -40 C to +85 C

Requirement / Recommendations

Recommended Antenna Specifications

Motorola Timing-2000 / Synergy-Systems

Timing-3000

Andrew GPS-QBW-20N

° ° °

° ° ° ° ° °

° ° ° ° ° °


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with an un-obstructed view to the horizon. Typically however, six (6) or more satellites are visible allowing for some obstruction of the GPS antenna visibility if necessary.

8.3.2.2 Antenna Placement Optimization

While the ideal GPS antenna installation provides an unobstructed view in all directions to within 20 of the horizon, practical limitations exist at some cellsite locations that afford only a fraction of the desired visibility. Reductions in GPS antenna visibility can result in increased BTS initialization times or in the reporting of numerous “GPS/Primary Reference Source Failure” (28-10040) or “Source Unavailable” (28-29052) alarms from affected BTS’s. Care should be taken in selecting the antenna location to maximize the available visibility. Even small improvements in visibility can have a large impact in receiver performance as can be seen in the example presented in Figure 8-3, Maximizing GPS Antenna Visibility.

Cellsites that must operate in environments that do not allow for the reception of the minimum four (4) GPS satellites required for cellsite initialization such as in “urban canyons” of dense cities should consider the use of the Remote GPS (RGPS) or to operate in the GPS “surveyed” mode. Information regarding the RGPS receiver and “surveyed” mode operation are presented in Section 8.4 and Section 8.5.1.5 respectively of this document.

°


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Figure 8-3: Maximizing GPS Antenna Visibility

8.3.2.3 Lightning Protection

The GPS antenna position should be chosen to minimize the potential of damage due to a direct or nearby lightning strike. In cellsite installations employing an antenna tower it is recommended that the GPS antenna be mounted near the tower base (but not where it may be impacted by falling ice) to allow the tower to act as a lightning rod. Many different approaches have been used to define the protective area provided by a nearby tower or lightning rod such as the 45° protection zone or the rolling ball method. While the accuracy of these methods has recently come into question the general rule holds where higher antenna elevations yield a higher probability of damage due to lightning. The GPS antenna should never be at or near the highest elevation of its surroundings.

The GPS antenna cabling must be grounded at the cellsite master ground point. If the GPS antenna must be positioned on or close to the cellsite tower it will necessary to bond the antenna and cabling shield to the tower as well to avoid lightning flashover.

GPSSatellite

AntennaLocation "B"

Obstructed GPS signal

GPSSatellite

GPSSatellite

GPSSatellite

GPSSatellite

Obstructed G

PS

signalO

bstructed GP

S sign

al

AntennaLocation "A"

Antenna position should be chosen tomaximize visibility to horizon. Antennalocation "B" minimizes shadowing bynearby building making it possible toreceive more GPS satellites. ImprovedGPS antenna visibility will reduce numberof GPS alarms and BTS initialization time.


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Figure 8-4: GPS Antenna Placement Considerations

8.3.2.4 Antenna Blockage

While Nokia Siemens Networks recommended GPS antennas utilize a conically shaped radome to minimize the likelihood of blockage due to debris or snow buildup care should still be taken in positioning the antenna to minimize any such accumulation on the antenna surface. Mounting the GPS antenna on a pole or mast at an elevation above any potential snow or debris buildup is a common solution to this problem. Pole or mast mounting also allows any temporary ice accumulation to quickly dissipate after severe weather conditions subside.

8.3.2.5 RF Interference

While the BTS GPS receiver and recommended antennas provide a high level of out of band interference rejection care should be taken to avoid placing the GPS antenna in the direct radiation path of cellular or other transmit antennas. To minimize interference potential the GPS antenna should be positioned at a different elevation and as far as possible from nearby transmit antennas.

8.3.3 RF Cabling

A wide range of coaxial cable types may be used between the GPS antenna and CDMA BTS connection. In choosing the GPS antenna cable all RF, physical installation and environmental

45°

Lightning "Cone of Protection"

Not Recommended due to:- High risk of direct lightning strike damage- Interference for closely located transmit antennas.

Tower Top Mounting

Not Recommended due to:- Potential damage due to falling ice- Potential antenna shawdowing

Tower Base Mounting

Provides adequate visibility with freedom from:- Falling ice- RF Interference from cellsite transmissions- Direct lightning strikes

Recommended Mounting Location

Not Recommended due to:- Potential damage due to lightning strike- Potential damage due to falling ice- Potential antenna shadowing unless long outrigger is used

Outrigger Mounting45°

Light

ning

"Con

e of

Pro

tect

ion"

"Rol

ling

Bal

l" P

rote

ctio

n A

rea "R

olling Ball" Protection A

rea


8


requirements must be considered. The GPS antenna cable must have a characteristic impedance of 50 and must satisfy the signal loss requirement outlined in Section 8.3.6 of this document for the installation length.

The chosen antenna cable must support all installation requirements such as the minimum cable bend radius, operating temperature range any special insulation requirements such as ultra-violet light resistance, armor jacketing for rodent proofing, plenum ratings, etc. Signal loss and minimum bend radius information for commonly used GPS antenna cable types is presented in Table 8-2 antenna cable loss / bend radius data.

Table 8-2: Antenna Cable Loss / Bend Radius Data

Care must be exercised in properly terminating and weather-proofing all cable connections. A common problem experienced with GPS antenna installations involves poor cable terminations or weather-proofing that result in degraded performance over time and eventual failures. Please refer to the GPS Antenna Kit installation and cable termination instructions contained in appendix A of this document for proper weather-proofing and cable termination guidelines.

8.3.4 Lightning Arrestor

A surge arrestor should always be employed at the GPS antenna cable building or outdoor BTS enclosure entry point to protect cellsite equipment and for operator safety. The surge arrestor should be connected to the cellsite single point ground using a low impedance conductor as shown in Figure 8-2. The surge arrestor should have a low insertion loss within the GPS L1 band and must be capable of passing the antenna LNA 5VDC supply voltage. The Polyphaser DGXZ+06NFNF-A (Part Number: 8089211C01) surge arrestor is Nokia Siemens Networks recommended due to its low clamping voltage and high surge handling capabilities. The DGXZ+06NFNF-A has the following performance specifications:

Cable TypeLoss / length @

1575MHzMinimum Bend Radius Cable Diameter

RG-142 0.56 dB/m 51mm 4.95mm

LMR-400 0.17 dB/m 25.4mm 10.3mm

FSJ4-50 0.15 dB/m 32mm 13.2mm

LDF4-50 0.091 dB/m 125mm 16mm

Parameter Specification

Frequency Range 800MHz to 2500MHz

VSWR <1.1:1

Insertion Loss < 0.1dB over frequency range

Ω


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8.3.5 Signal Splitter

A GPS signal splitter may be used in multi-frame installations to share one GPS antenna with the GPS receiver in each connected frame. The Symetricom 58536A (Part Number: CGDSHP58536A) GPS 4-output active signal splitter is Nokia Siemens Networks recommended due to its ease of use and reliable design. The 58536A contains an integrated LNA to compensate for splitting losses that is powered by the GPS antenna supply voltage from any connected frame while also properly biasing the antenna port. The 58536A has the following performance specifications:

Turn-on Voltage +6.5VDC

Turn-on Time 4 S for 2kV/ S

Maximum Surge 20kA per IEC 61000-4-5 8/20 S Waveform

Throughput Energy < 175 J for 3kA, 8/20 S waveform

User Voltage +6.0VDC MAX

Operating Temperature Range -50°C to +85°C


Frequency Range 1575.42MHz ±10MHz

Gain 0dB ±3dB

Noise Figure 5dB (typical) @ 25°C

Port Isolation -26dB (typical) @ 1575.42 MHz (L1) -50dB (typical) @ L1+40MHz

DC Power+4.5VDC to +13VDC @ 23mA to 48mA

RF Input Level (Antenna Input) Maximum Operating: <-25dBm

Damage Level: +17dBm @ L1

Input / Output Impedance 50

Connectors Type-N Female

Operating Temperature Range -35°C to +75°C


η η

μ

μ μ

Ω


8


When using an active signal splitter care must be exercised in insuring that the maximum GPS receiver antenna current level (80mA) is not exceeded. The total GPS receiver antenna current draw (IGPS Antenna) can be calculated by summing the GPS antenna and signal splitter LNA currents using the following equation:

The total GPS receiver antenna current when using the Andrew GPS-QBW-20N antenna and Symmetricom 58536A signal splitter would be:

IGPSAntenna = 25mA + 48mA

IGPSAntenna = 73mA

8.3.6 GPS Antenna System RF Requirements

To realize optimal GPS receiver performance the total GPS antenna system gain and noise figure at the CDMA BTS GPS antenna input must meet the following requirements:

Operation outside of these limits can negatively impact receiver operation. Potential negative impacts include increased cellsite initialization times, the reporting of “GPS/Primary Reference Source Failure” alarms and a reduction in receiver interference immunity.

8.3.6.1 Antenna System Gain

The total antenna system gain can be calculated by summing the gains and losses (in decibels) of all antenna system elements using the following equation:

ParameterFrame Type

SC24/48/72xx, SC480 UBS Macro, M810

Total Antenna System Gain (G): (Within the GPS L1 band (1575.42 MHz +1.023MHz))

+10dB < G < +26dB +14dB < G < +32dB

Total Antenna System Noise Figure:

<4.0 dB <4.0dB

IGPSAntenna IAntennaLNA ISplitterLNA+=


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As an example the antenna system gain of a hypothetical SC48xx cellsite GPS installation utilizing the following elements (pictured in Figure 8-5, GPS antenna Loss Budget / Noise Figure Calculation) is calculated as follows:

GAntenna= +25dB (Timing-3000 Antenna)

GCable 1= -6.2dB (40m FSJ4-50 cable (0.15dB/m * 40m + 0.2dBconnector loss)

GLightning Arrestor= -0.1dB (Polyphaser DGXZ+06NFNF-A)


GSplitter= -3.0dB (Symmetricom 58536A minimum gain)


G = +25dB – 6.2dB -0.1dB -0.5dB -3.0dB -0.5dB

G = 14.7dB

The calculated gain of this hypothetical GPS antenna system is +14.7dB which is within the required gain range of +10dB to +26dB. If the resulting gain were less than +10dB modifications to the antenna system such as using a lower loss cable to increase the overall gain would be required.

8.3.6.2 Antenna System Noise Figure

The GPS antenna system noise figure requirement is seldom violated unless a non recommended GPS antenna is used or additional signal amplifiers are employed to compensate for large cable losses. The total antenna system noise figure can be calculated using the following equation:

Where:f = the total antenna system noise figure

f1 = noise figure of stage 1

g1 = gain of stage 1

G GAntenna GCable1 GLightning Arrestor GCable2 GSplitter GCable3+ + + + +=

f f1

f2 1–

g1-------------

f3 1–

g1 g2×----------------- …+ + +=


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Note: all gain and noise figure values are absolute and not in decibels

The noise figure of the hypothetical GPS antenna system would be calculated as follows:

f = 1.62

NF(dB) = 10 * Log(1.62) = 2.09 dB

The calculated noise figure of this hypothetical GPS antenna system is 2.09dB which is below the +4.0dB maximum as required. Violations in antenna system noise figure can be addressed by utilizing lower loss antenna cabling, signal amplifiers with lower noise figures and reducing the signal losses prior to any in-line signal amplifiers. Difficulties in satisfying antenna system noise figure requirements are usually due to need to utilize long cable runs. The use of an RGPS receiver is recommended for installations that require lengthy cable runs to the point that system gain and noise figure requirements are difficult to satisfy.

f 1.414.17 1–

316------------------- 1.02 1–

316 0.240×---------------------------- 1.12 1–

316 0.240× 0.977×------------------------------------------------ 3.16 1–

3.16 0.240× 0.977× 0.891×--------------------------------------------------------------------- 0+ + + + +=

1.22 1–316 0.240× 0.977× 0.891× 0.501×---------------------------------------------------------------------------------------- 5.62 1–

316 0.240× 0.977× 0.891× 0.501× 0.891×------------------------------------------------------------------------------------------------------------+


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Figure 8-5: GPS Antenna Loss Budget / Noise Figure Calculation

8.4 Remote GPS (RGPS) Receiver

The RGPS module consists of an integrated GPS antenna, receiver and power supply / interface designed to support longer cabling lengths than are possible with the RF GPS antenna thereby providing more options in terms of receiver / antenna positioning. The normal GPS antenna positioning guidelines apply when using an RGPS receiver while the RF loss and noise figure requirements are not applicable.

8.4.1 SC24/48/72xx, SC480 Frame RGPS Operation

The RGPS receiver is only supported with CSM-II (Kit Number: SGLN4132ED or later hardware) and all CSA card types. A typical legacy frame RGPS installation is presented in Figure 8-6, Single and Multi-Frame Remote GPS Configuration.

The digital interface supported by legacy frames allows for cable distances of up to 1km between the RGPS receiver and base station. Signal delays through all RGPS cabling and hardware are automatically measured and compensated for by BTS hardware and software. An optional Remote


8


GPS Distribution (RGD) card (Kit Number: SGLN5864) can be used to share one (1) RGPS receiver with up to four (4) legacy CDMA MODEM frames.

Please refer to the appropriate frame type Hardware Installation manual for additional details regarding RGPS installation.

Figure 8-6: Single and Multi-Frame Remote GPS Configuration

Expansion Frames

Starter Frame

Lightning Arrestor(See Note 2)

(CGDS0971017AA1)

CellsiteGround

30-86433H02

30-86433H02

30-86433H02

30-86433H02

RGPS InterfaceCable (See Note 1)

(01-86012H0x)

RGPSReceiver

RGPS Receiver Cable (See Note 1)

RGPS Connetor(See Note 3)

Notes: 1. One (1) of two approaches can be used to make connections between the RGPS receiver, Lightning Arrestor and RGPS connection on the Modem Frame or RGD card. A single 30-87465C0x cable which is terminated with an RGPS "Deutsch" connector on one end and a 15 pin subminature D connector can be used by cutting the cable where the lightning arrestor is to be installed (any excess cable can also be removed at this point). The other approach is to use a 30-86039H between the RGPS receiver and Lightning Arrestor and an 30-86433H07 cable between the lightning arrestor and RGPS MODEM Frame connector. 2. The pictured lightning arrestor is recommended for use at the cable building entry point. Outdoor cellsites may include a punchblock type lightning arrestor / cable termination point. Please refer to the appropriate frame type Hardware Installation manual for detailed RGPS connection information. 3. An optional Remote GPS Distribution (RGD) card can be utilized to share one (1) RGPS receiver with multiple co-located CDMA frames.

Optional Equipment(For use in multi-frame installations)


RG

D

RGPS IN

EXP 1

Main Out

EXP 2

EXP 3

(See Note 3)


8


8.4.2 UBS / M810 Frame RGPS Operation

A typical UBS frame RGPS installation is presented in Figure 8-7, UBS Remote GPS Configuration. UBS frames support operation with either a Trimble Accutime-Gold (Kit Number: STLN6594) or ONCORE based (Part Number: 0186012H03 / H04) RGPS receiver.

Figure 8-7: Single and Multi-Frame Remote GPS Configuration

The digital interface supported by UBS frames allows for cable distances of up to 304m between the RGPS receiver and base station. Signal delays through all RGPS cabling and hardware are automatically measured and compensated for by BTS hardware and software.

Please refer to the appropriate frame type Hardware Installation manual for additional details regarding RGPS installation.

8.4.3 UBS / M810 Synchronization Sharing

UBS frames support a synchronization sharing interface that allows a single GPS receiver to provide synchronization to multiple frames connected in a daisy-chain configuration. The synchronization sharing interface can support the operation of four (4) frames with a maximum total cable length of 915m between the RGPS receiver or starter frame and last frame and individual cable not exceeding 304m in length as shown in Figure 8-8, UBS Synchronization Sharing.

Lightning Arrestor(See Note 2)

(CGDS0971017AA1)

CellsiteGround

RGPS InterfaceCable 3086433H14(15.2m (50ft) max.)

RGPS Receiver Cable 3086039Hxx(See Note 4)

RGPS Connetor


01-86012H03/H04(Legacy)

RGPS

TrimbleAccutime-Gold

Notes: 4. A maximum cable length of 304.8m (1000ft) is supported by the UBS RGPS interface. The 3086039Hxx cable is avaliable in the following 4 lengths:

P/NSuffix

Cable Lengthm ft

38.1H11 12576.2H12 250152.4H13 500304.8H14 1000

5. The Yellow and Yellow/Black color coded conductors used in legacy (SC24/48/72xx and SC3xx) installations must be disconnected from the lightning arrestor if the Trimble Accutime-Gold RGPS receiver is used or damage to the RGPS receiver may result. Care must be exercised when making connections to the lightning arrestor to insure proper connectivity. Wiring errors can lead to improper operation or permanent RGPS receiver damage. Please refer to the 1X UBS Macro BTS Hardware Installation or 1X M810 Picocell BTS Hardware Installation manual for additional information regarding RGPS connectivity requirements.

Note 5

UBS Macro

M810


8


Figure 8-8: UBS Synchronization Sharing

8.5 CSM / CSA GPS Operation

CSM / CSA initialization times are dependant on GPS receiver operation. A properly installed GPS antenna and configured CSM / CSA card will result in minimal CSM initialization times. Table 8-3, Approximate CSM / CSA Card Initialization Times presents the expected CSM / CSA initialization times for a properly configured CSM / CSA card operating with a fully compliant GPS antenna system.

Table 8-3: Approximate CSM / CSA Card Initialization Times

Reset Reason Initialization Time Description

Process Total

CSM Card Replacement

12 minutes (code load) Note 6 20 minutes (satellite acquisition) 2 minutes (synchronization)

34 minutes

Requires complete code load and GPS receiver download of almanac / satellite ephemeris information from tracked satellites prior to enable.

CSM Cold Start Initialization Note 7

20 minutes (satellite acquisition) 2 minutes (synchronization)

22 minutes

Requires GPS receiver download of almanac / satellite ephemeris information from tracked satellites prior to enable.

CSM Warm Start Initialization Note 8

1.5 minutes (satellite acquisition) 2 minutes (synchronization)

3.5 minutes

Satellite almanac / ephemeris information maintained in GPS receiver memory allowing for fast satellite acquisition.

RGPS

Maximum Total Cable Length BetweenRGPS and all Connected Frame (915m)

Maximum Cable LengthBetween BTS's (304m)

M810 M810 M810 M810

Lightning Arrestor(Required when sync-sharing cable is installed in

environments subject to electrical surges)


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Note 6 – Code load times may vary depending upon image size in a given software release. SC480 CSA card code load times are significantly shorter than CSM code load times.

Note 7 – A cold start initialization involves a power interruption to the CSM card.

Note 8 – A warm start initialization involves a CSM software reset.

It is recommended that an examination be performed with any cellsite that requires significantly longer initialization periods than those listed in Table 8-3. Long initialization periods could be due to a degrading antenna system problem that could eventually lead to a complete GPS failure and the inability to initialize the cellsite.

8.5.1 CSM / CSA Configuration

A total of seven (7) configuration parameters contained within a Circuit BTS (cBTS) Configuration Data File (CDF) or Packet BTS (pBTS) Network Element Configuration File (NECF) configuration file can impact GPS receiver operation. Each of these elements must contain accurate information to insure proper GPS receiver and cellsite operation. A description of each parameter and their required settings are contained in the following sections of this document.

Please refer to the CDMA2000 1X “System Commands Reference” manual for additional information regarding CSM / CSA card configuration parameters.

8.5.1.1 <CSMRefSrc1> Parameter

The <CSMRefSrc 1> parameter is used to select the cellsite primary reference source type. Valid settings include:

CSM Code Upgrade (outage)

12 minutes (code load) Note 6 1.5 minutes (satellite acquisition) 2 minutes (synchronization)

15.5 minutes

Complete code load and fast satellite acquisition due to almanac / ephemeris information being maintained in GPS receiver memory.

Reset Reason Initialization Time Description

Process Total


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Table 8-4: <CSMRefSrc 1> Parameter - Valid settings

Note 9 – The “Mate GPS” reference source type is not supported by the SC480 CSA card.

An improper configuration of the <CSMRefSrc 1> parameter can prevent either or both CSM cards from having access to a primary timing reference source. Such an error would likely prevent the affected CSM from being brought into service.

The <CSMRefSrc 1> parameter configuration can be displayed or modified using the OMCR “DISPLAY BTS CSMGEN” or “EDIT CSM CSMGEN” commands respectively.

8.5.1.2 <CSMRefSrc2> Parameter

The <CSMRefSrc2> parameter is used to select the cellsite backup synchronization reference type. Valid settings include:

Data Domain Usage

Craft Person<CDF> / <NECB>

Configuration

GPS 0 Used with primary CSM (CSM-<bts>-1) when an internal GPS receiver is employed. The primary CSM must be an SGLN1145 (CSM with GPS receiver) kit type.

MATEGPS 1 Used with secondary CSM (CSM-<bts>-2) when an internal GPS receiver is employed. The secondary CSM should be an SGLN4132 (CSM without GPS receiver) kit type. Note 9

REMOTEGPS 12 Used with both the primary and secondary CSM when an external (Remote GPS) receiver is employed. The primary and secondary CSM cards should be CSM-II (SGLN4132ED or later) type hardware type.


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Table 8-5: <CSMRefSrc2> Parameter - Valid Settings

Note 10 – A CSA/CSM-II card configured for operation with an HSO will automatically detect and compensate for operation with an installed HSO, MSO or QHSO card. CSM-II hardware is required in the frame primary and secondary CSM slots for operation with an MSO or QHSO backup.

8.5.1.3 <BTSLatGps> Parameter

The <BtsLatGps> parameter reflects the GPS antenna location in the milliseconds format. Valid values can range from -324000000 (which corresponds to -90°) to +324000000 (which corresponds to +90°).

Positive latitude values represent locations in the northern hemisphere and negative latitude values correspond to locations in the southern hemisphere.

The following equation is used to convert from the <Deg:Min:Sec> format to the milliseconds format:

The <BTSLatGps> parameter is typically used by the GPS receiver as a seed in determining the actual GPS antenna location (when the <LocAccuracy> flag is set to the “estimated” mode). When the <LocAccuracy> flag is set to the “surveyed” mode the GPS receiver assumes that the supplied location data is the current GPS antenna location and no determination is performed by the GPS receiver. The <BtsLatGps> parameter should be accurate to within +1 minute (±1850 meters) when operating in the “estimated” mode and within ±1.6 seconds (±50 meters) when operating in

Data Domain Usage

Craft Person<CDF> / <NECB>

Configuration

None 16 Used when a backup reference is not employed (Not Recommended)

HSO 18 Used when either an HSO, MSO or QHSO backup reference is employed. Note 10

HSOX 20 Used when an HSOX (HSO eXpander) card is employed. An HSOX card is typically used in the HSO card slot of logical BTS expansion frames to allow for the distribution of the master frame HSO, MSO or QHSO clock.

BtsLatGps SIGN LatDegrees( ) ABS LatDegrees( ) 3600×( ) LatMinutes 60×( )+ LatSeconds+( )× 1000×=


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the “surveyed” mode (See Caution in Section 8.5.1.6).

The <BTSLatGps> parameter configuration can be modified using the OMCR “EDIT TBTS SITELOCATION” command.

8.5.1.4 <BTSLongGps> Parameter

The <BtsLongGps> parameter reflects the GPS antenna location in the milliseconds format. Valid values can range from -648000000 (which corresponds to -180°) to +648000000 (which corresponds to +180°). Positive longitude values represent locations in the eastern hemisphere and negative longitude values correspond to locations in the western hemisphere.

The following equation is used to convert from the <Deg:Min:Sec> format to the milliseconds format:

The <BTSLongGps> parameter is typically used by the GPS receiver as a seed in determining the actual GPS antenna location (when the <LocAccuracy> flag is set to the “estimated” mode). When the <LocAccuracy> flag is set to the “surveyed” mode the GPS receiver assumes that the supplied location data is the current GPS antenna location and no determination is performed by the GPS receiver. The <BtsLongGps> parameter should be accurate to within +1 minute (±1850 meters) when operating in the “estimated” mode and within ±1.6 seconds (±50 meters) when operating in the “surveyed” mode (See Caution in Section 8.5.1.6).

The <BTSLongGps> parameter configuration can be modified using the OMCR “EDIT TBTS SITELOCATION” command.

8.5.1.5 <BTSHeightGps> Parameter

The <BtsHeightGps> parameter reflects the GPS antenna height in centimeters using a WGS-84 GPS ellipsoid. Valid values can range from -100000cm to +1800000cm. The <BtsHeightGps> parameter is typically used by the GPS receiver as a seed in determining the actual GPS antenna height (when the <LocAccuracy> flag is set to the “estimated” mode). When the <LocAccuracy> flag is set to the “surveyed” mode the GPS receiver assumes that the supplied location data represents the current exact GPS antenna location and no location determination is performed by the GPS receiver. The <BtsHeightGps> parameter should be accurate to within ±100000 centimeters when operating in the “estimated” mode and within ±5000 centimeters when operating in the “surveyed” mode (See Caution in Section 8.5.1.6).

The <BTSHeightGps> parameter configuration can be displayed or modified using the OMCR “DISPLAY BTS/FRAME GPSANT” or “EDIT FRAME GPSANT” commands respectively.

BtsLongGps SIGN LongDegrees( ) ABS LongDegrees( ) 3600×( ) LongMinutes 60×( )+ LongSeconds+( )× 1000×=


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8.5.1.6 <LocAccuracy> Parameter

The <LocAccuracy> parameter is a flag to set the GPS receiver navigation mode. When set to the “estimated” mode (LocAccuracy = 0) the GPS receiver assumes that any supplied location coordinates (BtsLatGps, BtsLongGps, BtsHeightGps) are approximate and that the GPS receiver should determine (navigate) its actual antenna location. When set to the “surveyed” mode (LocAccuracy = 1) the GPS receiver assumes that the supplied location coordinates represent the actual GPS antenna position which time solutions will be based on.

CAUTION: The “surveyed” mode should be used with extreme care and only in situations where it is impossible or impractical to install the GPS antenna in a position where it can typically receiver four (4) or more satellites for a significant length of time. The “surveyed” mode does not reduce satellite acquisition time and should not be used for this purpose. The use of the surveyed mode needs to be used with caution as any errors in the supplied location data beyond the specified limits can result in significant timing errors or erratic GPS receiver operation.

It is recommended that guidance from the Nokia Siemens Networks GCC office be obtained before configuring any cellsite to operate in the GPS “surveyed” mode.

The <LocAccuracy> parameter configuration can be modified using the OMCR “EDIT TBTS SITELOCATION” command.

8.5.1.7 <GPSAntDelay> Parameter

The <GPSAntDelay> parameter is used to compensate for timing delays in cellsite installations that require long GPS antenna cable runs. The <GPSAntDelay> parameter is typically used in cellsites that employ an RF GPS receiver (an internal GPS receiver with an RF connection to an external RF GPS antenna) with cable runs in excess of 100 meters. Cellsites employing antenna cable lengths of less than 100 meters generally set the <GPSAntDelay> parameter to a value of zero (0) due to the minor introduced cable delay. Valid values can range from 0 S to 999999 S.

The <GPSAntDelay> parameter configuration can be displayed or modified using the OMCR “DISPLAY BTS/FRAME GPSANT” or “EDIT FRAME GPSANT” commands respectively.

8.5.1.7.1 Remote GPS (RGPS) Receiver

In systems employing a Remote GPS (RGPS) receiver the <GPSAntDelay> parameter should be set to zero (0) as the RGPS cable delay is automatically measured and compensated for. If a non-zero value is supplied to an RGPS receiver, the supplied configuration value will be used instead of the automatically measured value.

8.5.1.8 <HeightMode> Parameter

The <HeightMode> parameter should always be set to the “ELLIP” (0) setting as the “SEA_LEV” (1) setting is no longer supported. The <HeightMode> parameter selects the height reference used by the BTS GPS receiver in interpreting the GPS antenna height information supplied in the

η η


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<BTSHeightGps> parameter.

CAUTION: Setting the <HeightMode> parameter to the “SEA_LEV” value may prevent an affected BTS from being brought into service.

The <HeightMode> parameter configuration can be displayed or modified using the OMCR “DISPLAY BTS/FRAME GPSANT” or “EDIT FRAME GPSANT” commands respectively.

8.6 UBS GPS Operation

8.6.1 UBS Configuration

The UBS GPS receiver configuration can be displayed or modified using the OMCR ”DISPLAY GPS_SRC GPSCONF” or “ADD GPS_SRC DEVICE” commands respectively. The “ADD GPS_SRC DEVICE” command format and parameters are as follows:

add gps_src-bts#-ssi#-gps_src# device LATITUDE=<latitude>LONGITUDE=<longitude> ANTHEIGHT=<antheight>LOCACCURACY=<locaccuracy>GPSANTDELAY=<gpsantdelay>

Please refer to the CDMA2000 1X “System Commands Reference” manual for additional information regarding the “DISPLAY GPS_SRC GPSCONF” and “ADD GPS_SRC DEVICE” commands.

8.6.1.1 <latitude> Parameter

The <latitude> parameter is typically used by the GPS receiver as a seed in determining the actual GPS antenna location (when the <locaccuracy> flag is set to the “estimated” mode). When the <locaccuracy> flag is set to the “surveyed” mode the GPS receiver assumes that the supplied location data is the current GPS antenna location and no determination is performed by the GPS receiver. The <latitude> parameter should be accurate to within +1 minute (±1850 meters) when operating in the “estimated” mode and within ±1.6 seconds (±50 meters) when operating in the “surveyed” mode (See Caution in Section 8.6.1.4).

8.6.1.2 <longitude> Parameter

The <longitude> parameter is typically used by the GPS receiver as a seed in determining the actual GPS antenna location (when the <locaccuracy> flag is set to the “estimated” mode). When the <locaccuracy> flag is set to the “surveyed” mode the GPS receiver assumes that the supplied location data is the current GPS antenna location and no determination is performed by the GPS receiver. The <longitude> parameter should be accurate to within +1 minute (±1850 meters) when operating in the “estimated” mode and within ±1.6 seconds (±50 meters) when operating in the “surveyed” mode (See Caution in Section 8.6.1.4).


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8.6.1.3 <antheight> Parameter

The <antheight> parameter reflects the GPS antenna height in centimeters using a WGS-84 GPS ellipsoid. Valid values can range from -100000cm to +1800000cm. The <antheight> parameter is typically used by the GPS receiver as a seed in determining the actual GPS antenna location (when the <locaccuracy> flag is set to the “estimated” mode). When the <locaccuracy> flag is set to the “surveyed” mode the GPS receiver assumes that the supplied location data represents the current exact GPS antenna location and no location determination is performed by the GPS receiver. The <antheight> parameter should be accurate to within ±100000 centimeters when operating in the “estimated” mode and within ±5000 centimeters when operating in the “surveyed” mode (See Caution in Section 8.6.1.4).

8.6.1.4 <locaccuracy> Parameter

The <locaccuracy> parameter is a flag to set the GPS receiver navigation mode. When set to the “estimated” mode the GPS receiver assumes that any supplied location coordinates (latitude, longitude, antheight) are approximate and that the GPS receiver should determine (navigate) its actual antenna location. When set to the “surveyed” mode the GPS receiver assumes that the supplied location coordinates represent the actual GPS antenna position which time solutions will be based on.

CAUTION: The “surveyed” mode should be used with extreme care and only in situations where it is impossible or impractical to install the GPS antenna in a position where it can typically receiver four (4) or more satellites for a significant length of time. The “surveyed” mode does not reduce satellite acquisition time and should not be used for this purpose. The use of the surveyed mode needs to be used with caution as any errors in the supplied location data beyond the specified limits can result in significant timing errors or erratic GPS receiver operation.

It is recommended that guidance from the Nokia Siemens Networks GCC office be obtained before configuring any cellsite to operate in the GPS “surveyed” mode.

8.6.1.5 <gpsantdelay> Parameter

The <gpsantdelay> parameter is used to compensate for timing delays in cellsite installations that require long GPS antenna cable runs. The <gpsantdelay> parameter is typically used in cellsites that employ an E-GPS receiver (an internal GPS receiver with an RF connection to an external RF GPS antenna) with cable runs in excess of 100 meters. Cellsites employing antenna cable lengths of less than 100 meters generally set the <gpsantdelay> parameter to a value of zero (0) due to the minor introduced cable delay. Valid values can range from 0 S to 999999 S.

8.7 Typical GPS Receiver Operation

A properly installed and functioning GPS antenna system should result in the ability to typically track four (4) or more satellites with normal GPS satellite acquisition periods and a minimal of GPS related alarms. Occasional, brief GPS alarms can be triggered by conditions external to the BTS that are not indicative of a GPS antenna system or receiver problem. Cellsites that report numerous

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or lengthy GPS alarms or that require longer than normal CSM initialization periods should be scheduled for an inspection and possible maintenance of their antenna system and receiver hardware.

The current status of a cellsite GPS receiver can be obtained through OMCR command.

8.7.1 CSM / CSA GPS Receiver Status

The currently active CSM / CSA reference source, number of GPS satellites being tracked and the determined cellsite GPS antenna location can be obtained through the OMCR <status csm-bts#-csm#> command as in the following example:

OMC-0000>status csm-153-1

CSM-153-1 07-10-01 11:37:18 omc-0000 BTS-153 M000000.02872 926319/508122 INFO:55 "Network Element Status Response" UNKNOWN_STATUS=FALSE DEVICE_ID TELEPHONY ADMIN OPERATIONAL USAGE CONTROL AVAILABILITY --------- ---------- ------ ----------- ------ ------- ------------ CSM-153-1 INS_ACTIVE UNLOCKED ENABLED ACTIVE NONE NONE PROTECTION_ATTRIBUTE=FALSE CLOCK_SRC=GPS LATITUDE="42:08:01.451" LONGITUDE="-87:59:56.711" ALTITUDE="20647" NUM_SATELLITES=8 REASON_CODE="No Reason"

8.7.2 UBS / M810 GPS Receiver Status

Detailed GPS receiver status including the current receiver operating mode, GPS time, number of satellites tracked, cellsite GPS antenna location for UBS / M810 BTS’s can be obtained through the OMCR <status GPS_SRC-BTS#-2-1> for an M810 and <status GPS_SRC-BTS#-1-1> for a UBS Macro type frame. An example of a GPS status command and response from an M810 BTS is as follows:

OMC-0000>status gps_src-1906-2-1

GPS_SRC-1906-2-1 07-09-26 11:13:42 inca9 OMC-0000 M00802.0009 06881/11765 INFO:3 "Command in Progress" STATUS=STARTED

GPS_SRC-1906-2-1 07-09-26 11:13:42 inca9 BTS-1906 M00802.0009 01815/11767 INFO:72 "Network Element Status Response"

Device : GPS_SRC-2 GPS TOD Status : Good -------------------------- GPS TIME/POSITION --------------------------- Date : 26-09-2007 Latitude : 42:08:01.451 N GPS Time : 18:13:42 Longitude : -87:59:56.711 W UTC Offset : +14 Height : 206.47 Meters Heigth Mode : ELLIP ----------------------- GPS TRACKING INFORMATION -----------------------


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Tracked Sats : 9 Survey Progress : 100%

ID Status ID Status ID Status ID Status ID Status -- ------- -- ------- -- ------- -- ------ -- ------ 10 Track 13 Track 27 Track 19 Track 25 Track 3 Track 23 Track 8 Track 28 Track

--------------------- GPS RECEIVER INFORMATION ------------------------- Receiver Type : Trimble ResolutionT SW Major ver : 0x0E SW Minor ver : 0x01 Receiver Status --------------- Receiver Mode : Overdetermined Clock Questionable Accuracy : no Antenna Open : no Antenna Shorted : no

Measured GPS Digital Cable Delay : 22nS Configured GPS Antenna Cable Delay : 0nS

8.8 Cellsite GPS Failures

Nokia Siemens Networks CDMA base stations provide protection against temporary GPS interruptions through the use of a backup clock reference source. The Medium Stability Oscillator (MSO) can maintain BTS synchronization for a minimum of 8 hours and the High Stability Oscillator (HSO) or Quartz High Stability Oscillator (QHSO) can maintain BTS synchronization for a minimum of 24 hours during the absence of the GPS timing reference. Soft-handoff failures may be experienced for cellsites that have been operating without a GPS timing reference for periods greater than the backup clock holdover capabilities. The “Reference Source Unreliable” (28-10111) or “Synchronization Degraded” (28-29051) alarm will be reported by any cellsite to indicate a GPS outage longer that has persisted beyond the minimum backup clock holdover period. It should be realized that while soft-hand-off failures may occur call originations/terminations and softer hand-off will still be possible with this BTS. Since it is not possible to re-initialize a BTS with a non-functional GPS sub-system it is important to know if the GPS antenna is functioning properly prior to removing the primary CSM/CSA from service.

8.8.1 Non-Synchronous BTS (Emergency) Operation

Feature 9336 provides a means to enable CDMA BTS’s in cases where the GPS has become temporarily unavailable. Under normal circumstances the availability of the GPS is necessary to synchronize CDMA BTS clocks to GPS time prior to being enabled. BTS’s operating with FEATURE 9336 enabled (non-synchronous mode) will demonstrate a reduced level of call quality compared to those that are synchronized to GPS time. Impacts to call quality when feature 9336 is active will include (but are not limited to): the inability to support soft and hard handoff’s,increased system noise near the handoff region (reducing maximum number of supported calls), rogue pilots, etc. Because of the problems associated with operating a BTS in the non-synchronous mode, feature 9336 use should be reserved for emergency situations as a means to provide a limited


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level of call support that otherwise would not be possible. Feature 9336 is a packet BTS and UBS only feature. Please refer to the “Feature 9336 – Non-Synchronous BTS Enable During Emergency GPS outage Detailed Functional Description” for additional information regarding non-synchronous BTS operation.

8.9 Appendix A – GPS Antenna Kit Installation Instructions

(Reprinted with permission from Andrew Corporation)


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9 Inter-System
Chapter
9

Table of Contents

Interference (ISI)

9 Inter-System Interference (ISI)

9.1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 3

9.2 Cellular/PCS Inter-System Interference. . . . . . . . . . . . . . . . . . . . . . . . . 9 - 39.2.1 Intra-Band Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 4

9.2.1.1 AMPS Cells to CDMA Subscribers . . . . . . . . . . . . . . . . . . . 9 - 69.2.1.2 AMPS Subscribers to CDMA Cells . . . . . . . . . . . . . . . . . . . 9 - 99.2.1.3 CDMA Cells to AMPS Subscribers . . . . . . . . . . . . . . . . . . . 9 - 99.2.1.4 CDMA Subscribers to AMPS Cells . . . . . . . . . . . . . . . . . . . 9 - 9

9.2.2 Inter-Band Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 109.2.2.1 Preventative Measures: BS-to-BS Interference . . . . . . . . . . 9 - 139.2.2.2 Preventative Measures: Subscriber-to-Subscriber Interference 9 - 27

9.3 PCS and Microwave Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 289.3.1 PCS to Microwave Interference . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 28

9.3.1.1 Coordination Distances . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 299.3.1.2 Propagation Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 309.3.1.3 Power Aggregation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 349.3.1.4 Microwave Receiver Interference Criteria . . . . . . . . . . . . . . 9 - 359.3.1.5 PCS to Microwave Interference Summary. . . . . . . . . . . . . . 9 - 37

9.3.2 Microwave to PCS Interference . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 389.3.2.1 General Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 389.3.2.2 Calculation of Nominal Noise Floor . . . . . . . . . . . . . . . . . . 9 - 389.3.2.3 Calculation of Effective Interference Power . . . . . . . . . . . . 9 - 399.3.2.4 Calculation of Effective Noise Figure . . . . . . . . . . . . . . . . . 9 - 399.3.2.5 Microwave to PCS Interference Summary. . . . . . . . . . . . . . 9 - 40

9.4 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 - 40


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CDMA/CDMA2000 1X RF Planning Guide9 Inter-System Interference (ISI)

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9.1 Introduction

The purpose of this chapter is to provide systems engineers/planners with a basic understanding of several inter-system interference issues that can adversely affect CDMA system deployments. In this chapter, CDMA is defined as being a general term that applies not only to 2nd generation (2G) digital cellular, as defined in IS-95A/B, but also to 3G digital cellular, as defined in IS-2000. Currently covered in this chapter are cellular/PCS inter-system interference, as well as 1900 MHz CDMA and Microwave interference. For this discussion, there are no material differences between 2G and 3G-1X that would have to be dealt with for Intra-band and Inter-band interference issues. In the future, any additional inter-system interference scenarios that arise will be addressed in later versions of this document, as necessary.

9.2 Cellular/PCS Inter-System Interference

In real world situations, frequency spectrum is the most limited resource for implementing or expanding cellular radio telecommunications systems. As cellular service continues to migrate from the use of analog technologies to digital technologies such as CDMA, operators are often faced with choosing one of two options:

• Spectrum clearing, when deploying a CDMA system into an existing frequency band, by clearing spectrum that was formerly used by other cellular technologies. Examples of such deployments could include the clearance of AMPS analog spectrum for use with co-existing 800 MHz CDMA systems and the clearance of TACS analog spectrum for use with co-existing 900 MHz CDMA systems.

• Spectrum Reassignment, when deploying a CDMA system in an alternate frequency band previously unallocated for cellular use. Examples of this deployment strategy could include the use of the AMPS band for CDMA in an area already using TACS and/or GSM spectrum, and the use of the PCS 1900 MHz band for CDMA in an area already using DCS 1800 MHz spectrum.

Associated with each of the above deployment options is the potential for interference between the system being introduced and the currently existing, co-located cellular system(s). The severity of this interference, and its impact, will depend mainly on how frequency spectrum is assigned to all cellular systems that are required to co-exist in a given coverage area. The interference can be divided basically into two categories that will be referred to here as intra-band and inter-band. Intra-band interference corresponds to interference between co-existing systems that share the same cellular frequency band allocations, such as AMPS and TACS. Inter-band interference corresponds to interference between co-existing systems that utilize multiple cellular frequency band allocations, such as:

• AMPS with TACS and GSM, or• DCS 1800 with PCS 1900


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9.2.1 Intra-Band Interference

Many cellular operators are installing, or have already installed, CDMA digital technology that allows them to continue the process of expanding capacity in their currently existing AMPS, or TACS, analog markets. These cellular operators may now be at the point where they want to install, or are installing, the next phase of CDMA, namely IS-2000 1X technology having the same spectral bandwidth of 1.23/1.25 MHz, but twice as many Walsh codes (128). For purposes of this next step in technology migration, which enables greater throughput and data services, there are no material differences between 2G and 3G-1X that would have to be dealt with for Intra-band and Inter-band interference issues. In these markets, both CDMA and the currently operating analog system must exist simultaneously and in some cases even share the same spectrum. As a result, in addition to the possibility of increased blocking on the existing analog cellular system (due to spectrum clearing), there exists the potential threat of inter-system interference between the co-existing, co-frequency-band-allocation systems. This interference, referred to as intra-band inter-system interference, exists typically between the base stations of one system and the subscriber stations of the other, co-existing system (Figure 9-1).

Figure 9-1: Intra-Band Interference

Here, the interference arises as a consequence of the near-far effect, an example being created when a nearby base station transmitter, serving one system, captures the receiver of a subscriber unit being served by another system base station that is significantly farther away. The closer, interfering base station transmitter is able to capture the victim subscriber unit receiver because of the small propagation path loss between them. This interference phenomenon can have a significantly greater effect on a new system being deployed with fewer cell sites than the other pre-existing, co-band system. This is because the new system, with its fewer cell sites, creates greater differences in the signal levels seen either in the Forward (downlink), or Reverse (uplink), RF channels. Practically, this situation can be avoided by system planners if they strive to keep the cell site ratio (B:A) as close to (1:1) as possible, where "B" indicates the number of new cell sites, for system B, relative to "A," the number of old cell sites already existing in system A. Another important aspect regarding the relationship between B and A is that the cell site base stations for system A and B should be located very near each other, or co-located, as close as possible.

An example of an unbalanced situation, reflecting a cell site (B:A) ratio of (1:3), is depicted in Figure 9-2. Here, a subscriber being served by system B could potentially be threatened with intra-band interference to its receiver from cell site transmitters in the co-existing system A. System B subscribers could potentially experience their worst operating performance at the edges of cells in system B that lie close to the centers of the cells from system A.

System A System B System BCell Subscriber Cell


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Similarly, if the system A subscriber is close enough to a non-co-located system B cell site, the system A subscriber’s transmitter could also potentially cause interference to the system B cell site receiver if the subscriber is transmitting at a high enough power level.

Figure 9-2: Example of a (1:3) Overlay

Depending on the actual overlay of the two co-existing systems, there exists the potential for four different interference scenarios:

• System A subscriber(s) interfering with System B base station• System A base station interfering with System B subscriber(s)• System B subscriber(s) interfering with System A base station• System B base station interfering with System A subscriber(s)

The above four scenarios are discussed in more detail in the following four sub-sections, using a co-located AMPS and 800 MHz CDMA system as an example. Note that intra-band interference is not a problem unique to CDMA, as it is a radio-systems issue. The same issues will occur with a GSM system if overlaid on a TACS system in the same frequency band. All technologies have the same set of contributing factors. Some key variables for the interfering transmitter are: ERP (Effective Radiated Power, which is RF power directed towards the receive antenna), transmit nominal power, and sideband emissions. A few key variables for a potential victim receiver are: IM (Intermodulation) intercept point of the receiver, filter protection available, and gain of the receive antenna system.

After the potential for interference has been assessed, corrective action can then be taken, if required. Corrective action can be in the form of improving the filtering at the site. Or, it can be related to any of the other variables noted above: improving Tx sideband emissions, adjusting ERP, doing frequency planning, etc. In all cases, the potential for interference, and the best corrective action, is site specific. There is no generic solution, so site engineering is required. Recommendations for corrective action are addressed, where appropriate, in later paragraphs of this section.

System B

System A

X

XX

X

XX

X - Potential InterferenceAreas

X

X

X

X

X

X

XX

X

X

XX

X

XX

X

X

X


9


One additional note that cannot be overlooked has to do with Rogue transmitters, which are unauthorized and illegal transmitting units. Even though the existence of Rogue transmitters are rare, the operation of just a single Rogue transmitter can cause problems for one or more sectors of a CDMA system, if this Rogue unit has a relatively high RF power output. A common symptom that is observed when a Rogue unit is transmitting is that the impacted CDMA cells will exhibit a decrease in coverage area due to the elevated interference noise rise caused by the Rogue unit.

9.2.1.1 AMPS Cells to CDMA Subscribers

There are several potential inter-system interference mechanisms, but the dominant problem is an interference product resulting from strong AMPS base station signals mixing in the front end of a subscriber receiver, thereby creating unwanted signals that land inside the CDMA subscriber receiver passband. The subscriber receiver intermodulation (IM) performance is essentially identical for all technologies.

In order for this phenomena to occur, three things must happen simultaneously. First, the CDMA subscriber unit must be physically close to the AMPS base transmitter site. Second, the AMPS transmitter frequencies must create a third order mix. Third, the desired CDMA received signal must be relatively weak.

Anything that can be done to overcome or inhibit any combination of the above mechanisms will help in preventing an interference problem. For example, one of the easiest ways to prevent this particular interference problem is to make sure that there is a CDMA base station located at each one of the AMPS transmitter sites. Such a configuration was described earlier and is termed a (1:1) one-to-one overlay. In this arrangement, the undesired mix products will still occur, but because the desired CDMA signal from the local transmitter is always stronger than the mix products, the problem is prevented. Hence, the first and third mechanisms are no longer contributors.

Usually, the problem will appear when an operator tries to deploy CDMA at fewer sites than every AMPS site (usually during the initial phase of introducing CDMA into a market). The operator may try to put CDMA base stations into fewer AMPS sites, to save initial system deployment costs. If this ratio is about one-third of the AMPS sites, then it would be called a (1:3) overlay, reflecting the (B:A) ratio. (See Figure 9-3).


9


Figure 9-3: AMPS System with a Larger CDMA Site Overlay(cells marked “a” are potential CDMA sites)

In this case, it would be possible for the CDMA subscribers to be exposed to strong local AMPS signals while trying to receive a weak CDMA signal from a great distance, because two-thirds of the AMPS sites would lack a co-located CDMA transmitter. In fact, if the system is laid out on a regular grid, with a (1:3) overlay, the AMPS base sites that lack co-located CDMA transmitters will be exactly halfway between the CDMA base sites, thereby allowing the areas having the weakest CDMA signals to have AMPS base sites located there.

It should be noted that a (1:1) deployment is a “fix” for reducing the interference between an operator’s AMPS base stations and his own CDMA system, but this does not totally eliminate the interference. There is still the possibility that a CDMA subscriber could experience interference when in the vicinity of the other operator’s cell site in an area having a weak CDMA desired signal. As a practical matter, high traffic areas will attract cells from both operators. As a result, the high traffic areas will usually have CDMA base stations deployed in the same area which will produce a strong CDMA signal in order to overcome these problems.

When there is an AMPS site without a co-located CDMA site, a subscriber may or may not experience noticeable interference, depending on the number, level, and frequency of the AMPS carriers, and the CDMA signal strength itself. Using a few simplifying assumptions, Figure 9-4 shows the relationship between the signal levels at which the interference will appear for an un-modified IS-98 subscriber receiver, and for a subscriber receiver having either of two proposed subscriber changes, namely a switchable attenuator, or a continuously variable attenuator.


9


Figure 9-4: Required CDMA Signal Strength vs. Interfering AMPS Signal Strength

As can be seen from the graph, the interference can be mitigated by reducing the AMPS signal level, or by raising the CDMA signal level. The most likely way of increasing CDMA signal levels would be to add one or more CDMA transmitter sites in the immediate vicinity of any potential interfering AMPS transmitters.

Recently, a specification change for IS-98A has been proposed that addresses the need for improved IM performance in the subscriber receiver. As the graph above clearly shows, the proposed change allows for a wider front end dynamic range in the subscriber, either by utilizing a variable attenuator, or a switchable attenuator, in the front end. The use of either of these attenuators introduces approximately 20 dB of loss so that the subscriber can operate in a strong signal environment (for example, if the received CDMA signal strength were greater than -79 dBm). The operator will still have to manage a minimum CDMA signal strength in accordance with the anticipated interference levels that may be potentially encountered.

-120

-100

-80

-60

-40

-20

0

-45 -40 -35 -30 -25 -20 -15 -10 -5

IS-98 spec (1% FER)

IS-98A spec (1% FER) w/ variable attenuator

IS-98A spec w/ step attenuator

MA signal strength

Interfering AMPS signal Strength (per carrier for 2 carriers)

JSR 2/7/96

IS-98A; Step Attenuator

IS-98 Spec Level

IS-98A; Variable attenuator

Req

uir

ed C

DM

A S

ign

al S

tren

gth

Interfering AMPS Signal Strength(per carrier for 2 carriers)


9


To facilitate assessing the impact of any potential inter-system interference, the Nokia Siemens Networks CDMA Simulator has been modified to include an interference zone simulation and prediction.

As for the potential for inter-system interference at 1900 MHz, this type of interference is not expected to be a serious issue because the power levels are generally lower, the path losses are higher, and the environment will not be one of an unbalanced overlay. Still, there may have to be some engineering to provide interference control at the 1900 MHz band edges, where two different operators meet on different site grids. This situation will not be unique to CDMA, as the subscriber receiver intermodulation performance is essentially identical for all technologies.

In summary, the most desirable way to design a CDMA overlay is as a (1:1) deployment, although it will still be necessary to review AMPS site placement in weaker CDMA coverage areas. If the operator chooses to initially implement a lower density deployment, with something less than a (1:1) deployment, then the design of both the A and B sides will need to be very carefully engineered for interference control. If a lower initial cost is desired, then a system utilizing a (1:1) deployment with omni cells is preferred over a system using (1:3) or higher deployments. This would result in a system with the same number of sectors deployed, but not susceptible to the same amount of system interference.

9.2.1.2 AMPS Subscribers to CDMA Cells

Narrowband AMPS subscribers are not viewed as posing a problem to CDMA cells. Out-of-band AMPS subscriber Tx sideband emissions are not significant across the recommended 9 AMPS channels comprising the CDMA guard band. In-band AMPS subscribers must be geographically separated by a guard zone of sufficient path loss.

9.2.1.3 CDMA Cells to AMPS Subscribers

Although CDMA cells will have a lower Tx ERP, CDMA cells may still interfere with AMPS subscribers that are far from an AMPS cell. The interference is caused by CDMA cell Tx sideband emissions, which do not roll off as fast as those associated with a narrow band AMPS transmitter. Note that this should not cause the same amount of system interference since the CDMA sites will be co-located with the same AMPS system sites.

9.2.1.4 CDMA Subscribers to AMPS Cells

The CDMA subscriber Tx power is typically low, so low sideband emission power results. However, a (1:3) overlay will significantly increase the probability of interference from CDMA subscribers because all AMPS-only cells are located near the edge of the CDMA cells. CDMA users near the AMPS sites will be at the higher power levels and offset in frequency by as little as 900 kHz from the center of the CDMA channel. Depending on the path loss from the CDMA subscribers to the AMPS Rx cell sites, the CDMA subscribers might cause interference to the AMPS receive signal.


9


9.2.2 Inter-Band Interference

Some cellular system operators who want to introduce digital cellular technologies into their existing systems may prefer, instead, to deploy such digital systems like CDMA in an alternate frequency band previously not allocated for cellular use in a given country. Examples of this type of deployment strategy can involve the following:

• Use of the AMPS band for CDMA, in an area already using TACS or GSM spectrum• Use of the TACS band for CDMA, in an area already using AMPS spectrum• Use of the PCS 1900 band for CDMA, in an area already using DCS 1800 spectrum

While implementing any one of the above examples might allow a new cellular system to be deployed more readily (i.e. with little to no effect on the traffic performance of the existing analog systems), there may be an increased threat of inter-system interference depending on what operating spectrum is being used for the new and the existing cellular systems. To say this another way, inter-band interference typically occurs between the base stations and/or between the subscriber stations of two or more co-existing systems (see Figure 9-5), unlike the aforementioned intra-band interference.

Figure 9-5: Inter-Band Interference

Thus, while inter-band interference is a radio-systems issue that is not unique to a particular cellular technology and has been dealt with previously, what may be different with current deployments is how eager some operators are in trying to co-locate multiple-band cellular technologies affected by these issues. Such ambitious system deployments result in:

1. Less guard band than is recommended between two systems that must co-exist2. Smaller antenna separation (along with less isolation between systems), due to high

system densities3. More aggressive antenna sharing requirements between different technologies through

the use of combiners, duplexers, etc.

As a result, it is imperative to give proper consideration to the threat of these interference phenomena and to take proper measures to prevent any potential system performance degradation, such as a reduction in system capacity or RF link quality/reliability. It is the goal of this section to provide such consideration.

System B System A System B System ACellSubscriber SubscriberCell


9


As was mentioned previously, the threat and severity of inter-band interference between two or more co-existing cellular systems using multiple frequency bands will depend on what spectrum is assigned to each system. Table 9-1 shows how the AMPS/TACS/GSM spectrum has been assigned by the EIA/TIA/ANSI standards organization, for AMPS spectrum, and by the ETSI standards organization, for TACS/GSM spectrum.

In the 800/900 MHz band, extended bands EAMPS and ETACS overlap by as much as 22 MHz. The end of the standard AMPS band at 890 MHz is also the beginning of the standard TACS band (Figure 9-6).

Figure 9-6: AMPS/TACS/GSM Spectrum

Likewise, the DCS 1800 band overlaps the PCS 1900 spectrum by as much as 30 MHz (see Figure 9-7).

Table 9-1: Cellular Spectrum Allocation

StandardsBody

Cellular BandBS Tx / Sub. Rx Operating Band

(MHz)

Sub. Tx / BS Rx Operating Band

(MHz)

ANSI/ AMPS 869-894 824-849

EIA/TIA PCS 1900 1930-1995 1850-1915

TACS/ETACS 917-960 872-915

ETSI GSM 935-960 890-915

DCS 1800 1805-1880 1710-1785

869

870

880

890

891.

589

4

917

925

935

942.

5

950

960

872

880

890

897.

5

905

915

824

825

835

845

846.

584

9

B’A’A”

AM

PS

A

AM

PS

B

ET

AC

S B

TA

CS

B/

TA

CS

A/

ET

AC

S A

AM

PS

A

AM

PS

B

B’A’A”

TA

CS

Res

erve

d/

GS

M

GS

M

GS

M

ET

AC

S B

TA

CS

B/

TA

CS

A/

ET

AC

S A

TA

CS

Res

erve

d/

GS

M

GS

M

GS

M

BASE Rx/Subscriber Tx

BASE Tx/Subscriber Rx


9


Figure 9-7: DCS 1800 and PCS 1900 Spectrum

Table 9-2 provides a summary of the various interference scenarios that can result when attempting to utilize these different spectrum allocations.

While the use of overlapping operating bands in co-existing systems would be unacceptable due to the threat of co-channel interference, use of adjacent operating bands has already been implemented in or is being considered for some markets. Due to the typical wide band nature of cellular base station and subscriber station receivers, inter-system interference is also a threat in this scenario.

Table 9-2: Inter-Band Interference Scenarios

Interferer Victim

AMPS-Band Base Station

TACS/GSM-Band Base Station

TACS/GSM-Band Subscriber Station

AMPS-Band Subscriber Station

DCS 1800-Band Base Station

PCS 1900-Band Base Station

PCS 1900-Band Subscriber Station

DCS 1800-Band Subscriber Station

D EF

1930

1850

A B C

A B C

1880

1805

1785

1710

D EF

DCS 1800

DCS 1800

PCS 1900

PCS 1900 BASE Rx/Subscriber Tx

BASE Tx/Subscriber Rx

G

G

1995

1915


9


There are four predominant inter-band interference mechanisms:

• Interfering transmitter sideband emissions landing on-channel in a victim receiver’s Rx frequency band.

• Interfering transmitter intermodulation (IM) products landing in a victim receiver’s Rx frequency band.

• Victim receiver desensitization from an interfering transmit carrier.• Victim receiver intermodulation from two or more interfering transmit carriers.

9.2.2.1 Preventative Measures: BS-to-BS Interference

There are several options available to help prevent the occurrence of inter-band interference between base stations. Some examples include:

1) Providing ample guard band between the co-existing systems. In this case, base station transmitter equipment specifications for the interfering system and base station receiver equipment specifications for the victim system would provide enough protection from potential interference.

2) Separating interfering and victim base station antennas as much as possible, both horizontally and vertically, to provide the necessary isolation.

3) Providing adequate filtering of the interfering base station transmitter and/or the victim base station receiver to achieve additional isolation.

• Tx Filters would aid in attenuating transmitter intermodulation and/or sideband emissions to levels low enough so that they would not cause interference to, and/or desensitization of, the victim receiver.

• Rx Filters would aid in attenuating off-channel signals that pose a threat of either receiver desensitization or receiver intermodulation.

4) Modifying the frequency plan of either the interfering system or the affected system, on a site-by-site basis, to minimize the possibility of interference.

5) Reducing interfering base station RF power.

9.2.2.1.1 BS-to-BS Interference Analysis Procedure

There are two steps involved in a BS-to-BS interference analysis procedure.

Step 1.

The first step in the analysis procedure is to determine the minimum isolation required between co-existing base stations when just considering the relevant equipment specifications. The minimum required isolation between an interfering base station Tx antenna and a victim base station Rx antenna can be approximated by using some simple calculations that take into account various transmitter and receiver specifications (as provided in Section 9.2.2.1.3), antenna gains, free space


9


path loss, etc. Typically, up to four such calculations are required, one with respect to each of the aforementioned potential interference scenarios: interfering transmitter sideband emissions, interfering transmitter IM, receiver desensitization, and receiver IM. See Section 9.2.2.1.2 for further information regarding radio equipment interference mechanisms.

Which calculations to use for a given interference analysis will depend on what type of interference is possible and where the potential interference may fall with respect to the victim base station receiver’s operating spectrum. For example, if it is determined that no interfering Tx carrier frequencies fall within or near the wide passband of a victim cellular receiver, then the receiver desensitization calculation may not be required. Furthermore;

• IF an interfering system utilizes just a single Tx carrier (as is possible with CDMA), AND

• IF there are no other interfering Tx carriers present to mix with it to create IM products potentially falling within a victim receiver’s passband

• THEN the transmitter IM and the receiver IM calculations would NOT be required1

Step 2.

The second step in the analysis procedure is to take the largest isolation requirement and determine if it is reasonable to achieve it solely through antenna separation. Required antenna separation for a given isolation value can be approximated using free space path loss equations:

SH =[10 ((PLmin - 32.44-20*(log(f)))/20)]*1000 [EQ 9-1]

SV =10 (PLmin - 28)/40 * 300/f [EQ 9-2]

Where:SH Minimum horizontal antenna separation, in meters, for use with non-co-located

sites

SV Minimum vertical antenna separation, in meters, for use with co-located sites2

PLmin Minimum required isolation

f Interfering base station transmit frequency, in MHz

1. Where appropriate, it is recommended that consideration be given to the possibility for future expansion of the interfering system (resulting in the allocation of additional Tx carriers) when determining isolation requirements in order to prevent any future interference scenarios.

2. NOTE: The vertical spacing decoupling equation (Equation 9-2) provides a rough estimate of required antenna separation and does not consider near-field effects that can alter the actual isolation provided. It is strongly recommended that appropriate on-site testing be completed to verify the actual isolation achieved by vertical antenna separation.


9


Note that while other path loss models can be used to approximate antenna separation (such as Hata, Okumura, etc.), it is recommended to use the above path loss equations as a worst-case scenario.

If it turns out that the required antenna separation requirements are not reasonable between the co-existing systems (e.g. too large), then appropriate filtering may be considered to provide the remaining isolation. The amount of isolation provided from filtering will depend on the amount of guard band available between interfering systems and the amount of attenuation needed in the filter’s stop-band. The transmitter sideband emission and transmitter IM isolation calculations are to be used with respect to any Tx filter requirements. The receiver desensitization and receiver IM isolation calculations are to be used with respect to any Rx filter requirements.

If required, filter quantities should be ordered as follows. As a guide, order one set of Tx filters per interfering base station, where the quantity of filters in a set would depend on the number of interfering base station antennas present at the site. If Rx filters are required, order one set of Rx filters for each affected base station that is either co-located or directly adjacent to an interfering base station. Note that the need for Rx filters at a given affected base station may need to be determined on a site-by-site basis considering actual antenna separation distances and the amount of path loss between them.

9.2.2.1.2 Radio Equipment Interference Mechanisms

Inter-system interference scenarios addressed in this section are the result of several common interference mechanisms that can occur either in the transmitter or receiver of a radio communications system. The following radio equipment interference mechanisms are discussed below:

• Transmitter Sideband Emissions• Intermodulation (Tx IM, Rx IM, External IM)• Receiver Desensitization

1. Transmitter Sideband Emissions

Transmitter sideband emissions occur primarily in either the speech amplifier, oscillator and/or modulator of the transmitter. Sideband emissions are created by the infinite bandwidth characteristics of white noise modulating the Tx carrier. Most transmitter equipment specifications require a minimum of 60 dB attenuation of sideband emissions with respect the mean power level of the transmitter (see Figure 9-8).


9


Figure 9-8: Transmitter Spectral Mask

When sideband emissions fall within the passband of a sensitive communications receiver, it creates interference. This can happen when transmitters operate near receivers with adjacent passbands. The effect on the victim receiver is that of a reduction to the usable sensitivity for desired channel performance. With this type of interference, no particular “sound” is created at the receiver, just receiver noise (see Figure 9-9).

Figure 9-9: Interfering Transmit Carrier and Sideband Emission Spectrum

Emission profiles vary between different transmitter designs but, in general, have an energy (depicted in the above figure as Isideband), that is some specified level below the carrier’s power level, (Iout). Table 9-4 and Table 9-5 in the following section provide Tx sideband emission specifications for relevant technologies, listed both as defined values and as a function of Iout. The transmit sideband emission level must be received at the victim base station receiver below a maximum allowable interference level (VINT), which results in a certain tolerable degradation in receiver sensitivity. For example, most cellular/PCS receiver equipment specifications allow for a maximum degradation in receiver sensitivity of 3 dB, which corresponds to a maximum on-channel interference level equal to that of the receiver’s thermal noise floor (kTBF). In this case, interfering transmit sideband emission levels would then need to be received by a victim receiver

-10

-70

-20

-30

-40

-50

-60

0

f

( )

60 dB

Iout = Interfering BS RF equipment Tx

Interfering BS Tx Sideband Emissions Performance (dBc)

Isideband (dBm)

FTx FRx

output power level (top of frame) in dBm


9


at a level below its thermal noise floor (e.g. VINT = kTBF). For example, tolerating a 3 dB sensitivity degradation, a GSM receiver having a noise figure (F) of 4 dB and a channel bandwidth (B) of 200 kHz would have a maximum tolerable interference level of -117 dBm:

VINT,GSM = GSM Rx thermal noise floor = kTBFGSM

= -174 dBm/Hz + 10 * log(200 * 103) dB-Hz + 4 dB = -117 dBm

In addition to Isideband and VINT, the following must also be accounted for in order to determine the isolation required to avoid sideband emissions interference to a victim base station receiver:

• Interfering base station feeder loss (Ifeeder) and antenna gain (Iant, which is equal to 0 dBi if co-located w/victim base station antenna)

• Victim base station antenna gain (Vant, which is equal to 0 dBi if co-located with interfering base station antenna), feeder loss (Vfeeder), receiver multicoupler/preselector loss/gain3 (VRMC), receiver bandwidth adjustment factor4 (VBWA), and receiver sensitivity (Vsens)

The following relationship shows the minimum isolation, PLmin,Sideband, required between an interfering base station transmit antenna and a victim base station receive antenna to prevent sideband emission interference:

[EQ 9-3]

2. Intermodulation (IM)

Intermodulation, or IM, can occur anywhere in the transmission path from the transmitter to the receiver. IM is caused by non-linearities in transmitter circuitry, receiver circuitry, and/or along the RF path from the transmitter to the receiver. Severity of the IM process will depend on the number of IM products involved, their signal strengths, and bandwidths.

IM can be detected as either a distinctive sound or as noise. For example, with 3rd-order, FM-modulated IM, an analog receiver hears two voices, one loud and distorted and the other normal. On the other hand, IM produced by two or more signals, where at least one of them is a CDMA signal, would be detected by the user of analog receiver as noise.

3. The amount of available multicoupler/preselector loss (VRMC) will depend on the amount of guardband between the two systems. It is expected that for most cases this loss will be minimal considering the very gradual roll-off attributed to these normally wideband filters. In fact, some multicouplers and preselectors contain LNA’s that may have gain rather than loss in-band. In those cases, VRMC would have a negative value.

4. The VBWA term is necessary to adjust the sideband emission power specification, Isideband (listed in Table 9-4 and Table 9-5 in units of dBm/30 kHz), to that of the channel bandwidth of the victim receiver. For example, VBWA for a CDMA receiver would be equal to 10*log(1228800/30000) = 16.12 dB.

PLmin Sideband dB( ), Isideband dBm 30kHz⁄( ) Ifeeder dB( )– Iant dBi( ) Vant dBi( )+ +=

Vfeeder dB( )– VRMC dB( )– VBWA dB( ) VINT dBm( )–+


9


For this discussion, the IM types will be divided into three categories:

• Transmitter IM• Receiver IM• External IM

While the three categories of IM are distinctly separate matters, which are subject to different engineering considerations, the frequency relationships applying to IM products are common. Frequencies of IM products can be defined in the following manner:

• Fundamental Frequencies - referring to the center frequencies of the signals from which IM products are derived.

• Harmonics - corresponding to the whole number multiples of a fundamental frequency.• Order - corresponding to the classification of IM products according to the sum of the

harmonics of the constituent frequencies (e.g. 2nd, 3rd, 4th,... Nth).

For example, a 3rd order IM signal centered at frequency C could result from the combination of the 2nd harmonic of a signal whose fundamental center frequency is A and a second signal whose fundamental center frequency is B:

C = 2A + (1)B (where order = 2 + 1 = 3)

Some examples of 2nd through 5th order intermodulation products are provided in Table 9-3.

Some generalizations can be made concerning IM products. First, the signal strength level of harmonic decreases rapidly with its order (e.g. 3A would be weaker than 2A). Second, higher order IM products may require too many different transmitters to be keyed simultaneously (e.g. A+B+C+2D+2E) for the IM to occur. Lastly, even order IM products almost always fall out of the local systems’ operating bands. For these reasons, the third and fifth order intermodulation products are the more prevalent and therefore more prone to cause IM interference.

Table 9-3: Example IM Products

Order Intermodulation Products

2nd A+B, A-B

3rd 2A+B, 2A-B, 2B+A, 2B-A, A+B+C

4th 2A+2B, 2A-2B, 3A+B, 3A-B

5th A+4B, A-4B, 4A+B, 4A-B, 2A+3B, 2A-3B...


9


The following sections discuss the three categories of IM.

Transmitter Intermodulation (IM)

There are at least two distinctive types of Transmitter IM:

• Multi-carrier LPA IM• Transmitter-to-Transmitter IM

Multi-carrier LPA IM can occur as a result of the amplification of different RF carriers by a common linear power amplifier. In this case, any resulting transmitter IM products that fall inside of the Tx frequency band or close to it cannot be attenuated by RF filtering, and thus tend to all be of approximately the same power level. Any transmitter IM products that fall well outside of the Tx frequency band could be attenuated by RF filtering.

Transmitter-to-Transmitter IM can occur inside the transmitter circuitry if two or more transmitters are installed closely together (and thus offering low isolation). Conducted transmit intermodulation is the effect of frequency mixing in the final amplifier stage of one interfering carrier transmitter with the outputs of others. The non-linear final amplifier circuit generates the IM and the antenna radiates it. The result is that unwanted channel power may be generated in the interfering transmitter and land in the victim receiver’s Rx band (see Figure 9-10).

Figure 9-10: Transmitter IM

When transmitter IM products fall within the passband of a sensitive communications receiver, it creates interference. The effect on the victim receiver is that of a reduction to the usable sensitivity for desired channel performance.

Reradiated signals are subject to a mixing loss in the IM-producing transmitter, which can be defined by the dB difference between the power of the incoming signal and outgoing

Tx AFreq.870.57MHz

Tx BFreq.892.89MHz

870.57 and also 848.25

892.89 and also 915.21

MHz radiated

* IM Products are formed in

transmitter final amplifier and

* ReceiverFreq. = 848.25or 915.21 MHz

MHz radiated

are radiated.

LOWISOLATION


9


intermodulation. A typical value for this loss is 60 dB. Most transmitter equipment specifications require a minimum of 60 dB attenuation of IM signals with respect to the mean power level of either transmitter, equivalent to this mixing loss.

Power levels of potential transmitter IM products vary between different transmitter designs but, in general, have an energy (depicted in Figure 9-11 as IIM), that is some specified level below the Tx carrier’s power level (Iout).

Figure 9-11: Interfering Transmit Carriers and Intermodulation Spectrum

As with transmitter sideband emissions, all transmitter IM products falling within a victim base station receiver’s passband must be received at a level below a maximum allowable interference level (VINT), which results in a certain tolerable degradation in receiver sensitivity. Table 9-4 and Table 9-5 in the following section provide Tx IM specifications for relevant technologies, listed both as defined values and as a function of Iout.

In addition to IIM and VINT, the following must also be accounted for in the calculation:

• Interfering base station feeder loss (Ifeeder) and antenna gain (Iant, which is equal to 0 dBi if co-located w/victim base station antenna)

• Victim base station antenna gain (Vant, which is equal to 0 dBi if co-located with interfering base station antenna), feeder loss (Vfeeder), and receiver multicoupler/preselector loss/gain (VRMC)

The following relationship shows the minimum isolation, PLmin,TxIM, required between an interfering base station transmit antenna and a victim base station receive antenna to prevent Tx IM interference:

[EQ 9-4]

Receiver Intermodulation (IM)

Receiver IM occurs when two or more off-channel signals enter and drive a receiver’s RF amplifier or 1st mixer stage. The nonlinear nature of the electronic devices commonly used in receiver

IIM (dBm)

F3Tx

X X =

F1Tx F2Tx FRx

Iout = Interfering BS RF equipment Tx output power level (top of frame) in dBm

Interfering BS Tx IM Performance (dBc)

PLmin TxIM dB( ), IIM dBm( ) Ifeeder dB( )– Iant dBi( ) Vant dBi( ) Vfeeder dB( )– VRMC dB( )– VINT dBm( )–+ +=


9


amplification and mixing circuits leads to the production of undesired responses, such as IM, in addition to the desired response (see Figure 9-12). The closer to saturation that an amplifier or stage is driven, the worse (higher in level) the IM products become.

Figure 9-12: Receiver IM

If one or more of the victim receiver-produced IM products falls on or near a frequency to which the victim receiver is tuned, the effect is that the product will be an interferer to the desired receive channel. Since the receiver is most sensitive to this in-band product, the IM must be reduced at this point by signal level reduction of one or more of the mixing frequencies.

Tolerance to receiver IM will vary between different receiver designs. In general, performance will be limited by a maximum allowable interfering (e.g. receiver IM-producing) signal level as received at the victim receiver (depicted in Figure 9-13 as VIMR), which is some specified level above the receiver’s Rx sensitivity, Vsens.

A given receiver’s ability to combat receiver IM, is quantified by its intermodulation rejection specification, or IMR. To prevent receiver IM, interfering signal(s) must be received at a signal strength lower than a level as determined by the receiver’s reference sensitivity and IMR specifications:

FRx < (Vsens + VIMR)

Tx AFreq

870.57MHz

Tx BFreq

892.89MHz

870.57 MHz

892.89 MHz

only radiated

* IM Products are formed in

receiver amplifier or mixer.

* ReceiverFreq. = 848.25or 915.21 MHz

only radiated

HIGHISOLATION


9


Figure 9-13: Victim Receiver Out-of-Band Intermodulation

Table 9-6, Table 9-7 and Table 9-8 in the following section provide receiver IM specifications for relevant technologies, listed both as a defined value and as a function of Vsens.

In addition to VIMR and Vsens, the following must be taken into account in the calculation:

• Interfering BS RF equipment Tx output (top of frame) power level (Iout), feeder loss (Ifeeder), and antenna gain (Iant, which is equal to 0 dBi if co-located w/victim base station antenna)


The following relationship shows the minimum isolation, PLmin,RxIM, required between an interfering base station transmit antenna and a victim base station receive antenna to prevent receiver IM:

[EQ 9-5]

The resulting isolation requirements can be achieved through both antenna separation and filtering of the interfering transmitter(s) and affected receiver(s).

External Intermodulation (IM)

External IM is created by passive, non-linear elements in the transmission path from transmitter to receiver such as antennas, combiners, duplexers, cables, connectors, etc. and other elements in the immediate vicinity of the transmission line, such as guy wires, tower joints, anchor rods, etc. Here, signals are picked up by these elements and reradiated as IM products (see Figure 9-14).

X =

F1Tx F2Tx FRx

VIMR = BS Max Allowable Out-of-band, Receiver

Victim BS Receiver IM Rejection Performance (dB)

Vsens = Victim BS Receiver Sensitivity (dBm)

= kTBF + S/N - Processing Gain*

* if applicable

IM-Producing Rx Signal Level (dBm)

PLmin RxIM, dB( ) Iout dBm( ) Ifeeder dB( ) I+ant dBi( )

– Vant dBi( ) Vfeeder dB( )– VRMC dB( )– VIMR dBm( )–+=


9


Figure 9-14: External IM

Locating the actual source of external IM may be very difficult. There are really no preventative measures with respect to external IM other than to conduct thorough periodic maintenance of elements in and around the transmission path. However, while low-order external IM created in the antenna path system can easily cause interference to base station receivers that share the same antenna, resulting IM signal levels are usually low enough that they won’t create interference to subscribers or other base stations.

3. Receiver Desensitization

Receiver desensitization, also known as receiver blocking, is usually caused by strong off-channel interfering signals that fall within or just outside the often wide passband of the receiver. If the interference is strong enough, bias conditions can be changed on certain receiver stages, causing them to lose gain. This makes the receiver less sensitive to any desired signals. The ability of a receiver to receive an intended signal in the presence of these interfering signals is measured by its desensitization or blocking level specification. Associated with this level, is an allowable degradation in Rx sensitivity, usually 3 dB.

While its negative effects might not be immediately noticeable in a desired received signal, receiver desensitization could result in an increased susceptibility to fading and a reduction in channel capacity. Example causes of receiver desensitization are interfering Tx carrier power level, transmitter sideband emissions and transmitter IM products.

Desensitization levels vary between different receiver designs but, in general, have an energy (depicted in Figure 9-15 as Vblock), that is some specified level above the receiver’s Rx sensitivity, Vsens. Off-channel interfering signal(s) must be received at the victim base station receiver at a level below Vblock.

AN

ELECTRICALLY

NON-LINEAR

OBJECT Rx

Tx1

Tx2

IM INTERFERENCE


9


Figure 9-15: Victim Receiver Out-of-Band Desensitization

The Rx sensitivity level, Vsens, is a certain number of dB above or below the receiver’s thermal noise floor (kTBF) and is a function of the receiver’s required S/N ratio (Eb/No, C/I, etc.) and processing gain (CDMA only). Processing gain is equivalent to the receiver’s channel bandwidth in Hz (B) divided by the Rx data rate in Hz (R). An example calculation for 8 kbps CDMA with an Eb/No (S/N) of 7 dB and a receiver noise figure (F) of 6 dB is provided below:

Vsens,CDMA = kT (dBm/Hz) + 10*log(BCDMA) (dB) + F (dB) + S/NCDMA (dB) - 10*log(BCDMA/RCDMA) (dB)

= -174 + 10*log(1.2288*106) + 6 + 7 -10*log(1.2288*106/9600) = -121.2 dBm

Table 9-6, Table 9-7 and Table 9-8 in the following section provide desensitization specifications for relevant technologies, listed both as defined values and as a function of Vsens.

In addition to Vblock and Vsens, the following must also be accounted for in the calculation:

• Interfering BS RF equipment Tx output (top of frame) power level (Iout), feeder loss (Ifeeder), and antenna gain (Iant, which is equal to 0 dBi if co-located w/victim base station antenna)


The following relationship shows the minimum isolation, PLmin,Desense, required between an interfering base station transmit antenna and a victim base station receive antenna to prevent receiver desensitization as a result of the presence of strong off-channel signals:

[EQ 9-6]

Vblock = Victim Max Allowable BS Receiver

Vsens = Victim BS Receiver Sensitivity (dBm)

FTx FRx

Blocking/Desense Level (dBm)

= kTBF + S/N - Processing Gain*

* if applicable

PLmin Desense dB( ), Iout dBm( ) Ifeeder dB( ) I+ant dBi( )

– Vant dBi( ) Vfeeder dB( )– VRMC dB( )– Vblock dBm( )–+=


9


9.2.2.1.3 Equipment Specifications

Isideband, IIM, Vblock and VIMR will vary with equipment type. Typical values, based on standard equipment specifications are provided in Table 9-4 through Table 9-8 below.

Table 9-4: Partial Example of Base Station Transmitter Specifications

Technology (Specification)

Maximum Transmitter Sideband Emission Level, Isideband

(dBm/30 kHz)

Maximum Transmitter IM Power Level, IIM

(dBm)

AMPS(IS-20A)

< Larger of -31 or (Iout - 78) @ |f-fc| > 90 kHz

< (Iout - 60 dB)

800 MHz CDMA

(IS-97A)

< (Iout - 45)@ 1.98 > |f-fc| > 0.75 MHz

< (Iout - 60) @ |f - fc| > 1.98 MHz

< (Iout - 45)@ 1.98 > |f-fc| > 0.75 MHz

< (Iout - 60) @ |f - fc| > 1.98 MHz

Table 9-5: DCS 1800 Base Station Transmitter Specifications (GSM 05.05)

Offset Range From Tx Carrier

(kHz)

Maximum Transmitter Sideband Emission Level, Isideband

(dBm/30 kHz)

Maximum Transmitter IM Power Level, IIM

(dBm)

200 < (Iout - 30) < (Iout - 30)

250 < (Iout - 33) < (Iout - 33)

400 < (Iout - 60) < (Iout - 60)

600 to <1200 < -27 <-27

1200 to <1800 < -30 <-30

1800 to <6000 < -37.2a

a. AMPS IS-20A lists a sideband emission level in dBm/300 Hz. GSM 05.05 lists sideband emission levels in dBm/100 kHz for frequency offset ranges > 1800 kHz. A conversion to dBm/30 kHz is used here to be consistent with units used for other specification values.

<-32

> 6000 (Iout - 85.2)a < Larger of -36 or (Iout - 70)


9


Table 9-6: Partial Example of Base Station Receiver Specifications

Technology(Specification)

Maximum Receiver IM Level, VIMR

(dBm)

Maximum Receiver Desense/Blocking Level, Vblock

(dBm)

TACS > (Vsens + 65)

> -50 (for signals falling within TACS A band)

> -23 (for signals falling within TACS B band)

900 MHz GSM(GSM 5.05)

> - 43 (See Tables 3-6 and 3-7)

900 MHz CDMA(China IS-97A) > (Vsens + 72)

> (Vsens + 50)@ 0.9 > |f-fc| > 0.75 MHz

> (Vsens + 87)@ |f-fc| > 0.9 MHz

PCS 1900 CDMA(J-STD-019) > (Vsens + 72)

> (Vsens + 50)@ 0.9 > |f-fc| > 0.75 MHz

> (Vsens + 87)@ |f-fc| > 0.9 MHz

Table 9-7: In-Band GSM Base Station Receiver Blocking Specifications (GSM 05.05)

Offset Range From Intended Rx

Carrier(kHz)


(dBm)

600 to <800 > -26

800 to <3000 > -16

Table 9-8: Out-of-Band GSM Base Station Receiver Blocking Specifications (GSM 05.05)

Frequency Band(MHz)


(dBm)

0.1 to <915 > 8

980 to <12750 > 8


9


Note that the values in the previous tables are worst case and are based solely on the methods of measurement as outlined in each technology’s specification documentation. Actual values may vary according to both base station equipment manufacturer and desired quality of service (SINAD, Eb/No, BER, FER, data rate, etc.). With this in mind, vendor-specific documentation and system design constraints should be obtained to determine more accurate and/or appropriate data for a given interference analysis.

9.2.2.2 Preventative Measures: Subscriber-to-Subscriber Interference

The severity of interference between subscribers will depend on the subscriber densities of each system involved and the distances between them, because both the interfering transmitters and the affected receivers are moving with respect to one another and are in random positions relative to one another. The likelihood of experiencing interference will also depend on the power control capabilities of the interfering subscriber transmitter(s) and the affected subscriber station receiver(s).

As Table 9-9 illustrates, interference between subscriber stations is generally less of a problem than between base stations.

Since intermodulation requires two or more interfering signals at precise frequency spacing, the probability of any resulting product causing interference could therefore be very low. Therefore, it

Table 9-9: Inter-Band Interference Comparison

Subscriber Station-to-Subscriber Station

Base Station-to-Base Station

Antenna Separation Distance

Variable Fixed

Path Loss Variable Fixed

# of Distinct Tx Frequencies-

FDMA/TDMA Systems

1 per subscriber (narrow-band carrier)

> 1 per Base Station

# of Distinct Tx Frequencies-

CDMA Systems

1 per subscriber (wide-band carrier)

> 1 per Base Station

Power/channel Low High

Antenna Cross Polarization Loss

Variable Fixed


9


is reasonable to conclude that most Subscriber-to-Subscriber interference is typically caused by Tx sideband emissions. In areas subject to a high density of pedestrian subscriber traffic (for example, in a shopping mall, subway station, etc.), this interference could be significant enough to affect call quality or cause a dropped call.

Unfortunately, there is little that can be done to prevent Subscriber-to-Subscriber interference other than to address the potential for interference in the actual physical design of both the interfering and victim subscriber units so that sufficient isolation is provided. This, however, seems to be an unlikely possibility as subscriber performance requirements (again, generated by distinctly different standards bodies: ANSI/EIA/TIA and ETSI) typically do not address inter-band interference issues of this nature.

Note that frequency plans could also be modified to help prevent interference in certain areas, depending on the technologies involved. However, in high-traffic areas which are of the most concern, frequency plan flexibility may be limited.

9.3 PCS and Microwave Interference

Within the US 1900 MHz band, there are over 4,500 microwave links, the majority of which are 5 MHz (300 channel) or 10 MHz (600 channel) analog FM-FDM systems. The chart below illustrates where these links are centered with respect to the PCS MTA and BTA license bands.

Figure 9-16: The PCS Spectrum

9.3.1 PCS to Microwave Interference

PCS license requirements essentially dictate that any PCS system may not cause any harmful interference into incumbent microwave systems. Detailed interference analysis is needed to determine the interference potential of PCS into microwave systems.


9


Recommendations and guidelines for analyzing potential interference into microwave systems are provided in the Telecommunications Industry Association’s (TIA) Bulletin, TSB-10-F. The four main considerations detailed in Bulletin TSB-10-F are:

• Coordination Distances• Propagation Models• Power Aggregation• Microwave Receiver Interference Criteria

These considerations must incorporate interference from all system sources and subscriber units. As a result, the term PCS transmitter can refer to a base station or a subscriber unit. Please refer to the Bulletin for more information. The following sections will summarize the four main considerations.

9.3.1.1 Coordination Distances

It is necessary to determine a search area around each PCS transmitter within which the process of interference analysis needs to be undertaken. This is known as the Coordination Distance. The primary factors governing the coordination distance for a PCS transmitter are its antenna height and EIRP. In general, the PCS base station transmitter will define the coordination distance. The minimum Coordination Distance is calculated by using the following formula set:

; [EQ 9-7]

[EQ 9-8]

[EQ 9-9]

[EQ 9-10]

[EQ 9-11]

Where:D Coordination distance

P EIRP (dBm)

HT Transmitting antenna height above average terrain (m)

DL Free Space distance (km)

DD Diffraction distance (km)

DLT Distance to horizon (km)

DLT 2.56 HT( )=

DL 1051.87 P+

20----------------------- =

DD65 1.85DLT P+ +

0.106 DLT 33.6+( ) 0.899+log------------------------------------------------------------------------=

DS19.9– 0.12 DLT P+×+

0.1156 5.65–×10 DLT×–

-----------------------------------------------------------=

D min DL max D DS,( ),( )=


9


For example, a typical PCS BTS with an antenna height of 30 meters and an EIRP of 100 Watts requires a coordination distance of 275 km.

The following graph shows the coordination distances for a PCS transmitter with 30 and 90 meter antenna heights, over a wide range of EIRP values.

Figure 9-17: Example Coordination Distances

As can be seen, the distances involved are substantial and may even extend beyond the MTA/BTA (Major Trading Area / Basic Trading Area) license boundary.

9.3.1.2 Propagation Models

To determine the level of interference into a microwave system, it is necessary to calculate the signal strength of the PCS signal at the microwave receiver. In traditional microwave systems, the free space path loss calculation is used in link planning. However, with the lower antenna heights of PCS transmitters, the effects of local clutter must be considered. For this reason, the Hata model with suburban correction is used as the base propagation model. In addition, because of the large coordination distances, propagation beyond the transhorizon (the point at which line of sight communications between two fixed antennas is no longer possible) must also be considered. The forward scatter loss model is used for propagation beyond the horizon.

The Hata and forward scatter loss models are used for both the subscriber unit and the base station path loss calculations. However, different correction factors are set to account for differences in antenna heights.


9


9.3.1.2.1 Basic Propagation Models

Free Space Path Loss

The free space path loss calculation is represented by the following equation:

[EQ 9-12]

Where:d distance (km)

f frequency (MHz)

Hata Model

The Hata based propagation model (suburban area) is represented by the following equation:

[EQ 9-13]

Where:Lpcs Loss between PCS and MW antennas using the modified Hata model.

PCS antenna height correction factor

Forward Scatter Loss (Troposcatter) Model

The actual distance to the transhorizon is calculated by using the smooth earth transition method, which specifies the receiver and transmitter antenna heights above the average elevation along the path. Assuming no clutter or terrain obstacles, the smooth earth transition distance (transhorizon) is represented by the following formula:

[EQ 9-14]

Where:dh Transition distance (km)

hpcs PCS antenna height above average terrain (m)

hmw Microwave antenna height above average terrain (m)

The recommended equation for forward scatter loss, adjusted for hourly median loss, is as follows:

[EQ 9-15]

Lfs 32.44 20 d( )log 20 f( )log+ +=

Lpcs 69.5 26.16 f( ) 13.82 hmw( ) 44.9 6.55 hmw( )log–[ ] d( ) α hpcs( )– 2f

28------ log

2– 5.4–log+log–log+=

α hpcs( )

dh 4.123 hpcs hmw+( )×=

L50 29.73 30 f( ) 10 d( ) 30 θ( ) N H h,( )+log+log+log+=


9


Where:L50 Hourly median transmission loss 50% of the time (dB)

f Frequency (MHz)

d Path length (km)

dh Smooth Earth Transition Distance

[EQ 9-16]

Where:

H

h

9.3.1.2.2 PCS Base Station Correction Factors

The same basic Hata model is used for path loss calculations for both subscriber unit and base station sources. However, a correction set is applied to account for differences in antenna heights. The Hata model from above and the following correction factors should be used for microwave antennas below 180 m and PCS antennas below 60 m.

For PCS antennas below 9 m (ground level subscriber unit sources), the following Hata suburban correction factor equation is used:

[EQ 9-17]

For PCS antennas between 9 m and 60 m (base station sources), the Hata large city correction factor equation is as follows:

[EQ 9-18]

Outside of these ranges, the free space path loss formula should be used to predict the propagation loss to the transhorizon. The following graph shows the relationship between the three propagation models at both PCS downlink frequencies. As previously mentioned, the local clutter has the effect of increasing the propagation loss above that of free space path loss. This in turn results in the transition from the Hata model to the troposcatter model occurring further out than the transhorizon distance.

θ d dh–( )8.5

------------------- milliradian( )

N H h,( ) 20 5 γh+( ) 4.343γh+log=

θd4000------------

1.0633–×10 θ2×

γ 0.27

α hpcs( ) 1.1 f( )log 0.7–[ ] hpcs 1.56 f( )log 0.8–[ ]–×=

α hpcs( ) 3.2 11.75 hpcs×( )log[ ]2× 4.97–=


9


Figure 9-18: Propagation Curves for High PCS Antennas

Downlink PCS Frequency = 1960 MHz

9.3.1.2.3 PCS Subscriber Unit Correction Factors

Subscriber transmissions will be a significant factor in the interference analysis as, unlike the base station, the subscriber unit is not fixed and will try to access the PCS system in various locations. For PCS subscriber units on the street, the recommended loss model is the mean Hata suburban model, Equation 9-7, with the suburban correction factor as stated in Equation 9-15. Refer to the following figure.

Figure 9-19: Propagation Curves for Low PCS Antennas

Uplink PCS Frequency = 1880 MHz

Transition Point

Transition Point


9


One of the most significant issues of interference into microwave systems is line of sight situations. The most common occurrence of this will be from a subscriber unit located in a high rise building or on a balcony. In this case, path loss figures approaching free space loss may be experienced between the subscriber unit and microwave antennas.

It is possible for this situation that the subscriber unit’s interfering signal will be stronger than the aggregated powers of many base station transmissions at the microwave receiver. In urban environments, the probability of an elevated subscriber unit is greater. Thus, the impact of the subscriber unit interference sources on the microwave receiver will be more substantial than in residential areas.

TSB-10-F, Section F-4.4.1.1 provides statistical adjustments to the mean Hata suburban model to account for the above effects.

9.3.1.3 Power Aggregation

When considering the interference level into a microwave receiver, the combined effect of all the PCS transmitters in a service area must be considered. The aggregated power will be a function of the total number of PCS transmitters (both base station and subscriber units) included within the service area.

Figure 9-20: Example Aggregated Service Area

Statistical methods for aggregating the PCS transmitter powers may be used to determine the expected spatial PCS distribution within the service area. As a default, uniform distribution of powers should be assumed. From the specified distribution, the aggregated interference signal can be determined by using either analytical techniques or Monte Carlo simulation methods.

Micr

owav

e Main

Bea

m

Microwave Site

Angle Theta


9


9.3.1.4 Microwave Receiver Interference Criteria

Three interference criteria are used to determine if a PCS system will interfere with microwave:

1. Carrier to Interference2. Threshold Degradation3. Reliability

All three forms of interference criteria should be assessed utilizing the analysis procedure in order to determine which microwave systems require relocation because they are vulnerable to interference, as well as to demonstrate non-interference into other microwave systems situated within the coordination distance.

9.3.1.4.1 Carrier to Interference

The Carrier to Interference criteria is used to specify the threshold at which an unwanted signal will cause harmful interference upon the wanted signal. For single frequency transmission systems, a single C/I ratio may be quoted for the receiver. However, with multi-channel microwave systems, the C/I criteria is expressed in terms of a curve representing the allowable C/I ratio at a specific frequency separation of an unwanted signal from the center of the microwave carrier.

The C/I curves are calculated based on the transmit power spectral densities of both the microwave and PCS systems, as well as the receiver selectivity of the microwave system. The power spectral density, the number of channels, the modulation type, and the bandwidth play an important role in determining the shape of the curve. The following graph is an example of such a curve for both GSM and CDMA carriers as interferers.

Figure 9-21: Example C/I Curves for a 10 MHz Microwave Receiver


9


9.3.1.4.2 Threshold Degradation

Threshold degradation is the reduction in the microwave receiver sensitivity caused by an interfering PCS signal. Bulletin TSB-10-F states that the maximum interfering signal level for analog receiver threshold degradation in Bulletin TSB-10-F links can be represented by the following equation:

[EQ 9-19]

Where:Imax Maximum interfering signal level, dBm

Rt Receiver threshold, dBm

Fα The difference between the operating fade margin and that required to meet the outage objective, dB

Se Effective selectivity of the victim receiver to the interfering signal, dB

(fs) Interfering signal frequency at which Se is defined, MHz

For example, if the microwave receiver sensitivity is -80 dBm, then the co-channel interfering PCS signal must be -90 dBm or less to avoid degrading the sensitivity of the receiver (assuming no degradation due to fade margin).

9.3.1.4.3 Reliability

Reliability is an all-encompassing term that describes how well the microwave link guarantees communications. In general, because link failures are mostly attributable to fading having very short durations, the microwave link reliability measures can be expressed in either of two main forms:

• Availability, quoted as a percentile, such as 5 nines (99.999%) or 6 nines (99.9999%)• Annualized outage time (seconds)

For most microwave links, the operator defines a minimum required reliability. Reliability within a microwave link is, in fact, a function of the fade margin allocated for the link. A reduction in the fade margin will reduce the availability and increase the outage time per year. It is not uncommon for the microwave link to have been over-engineered, which means that the fade margin allocated is in excess of the reliability required. Thus, a 1 dB degradation or more caused by a PCS interferer may not always compromise the minimum required reliability.

Imax Rt Fα Se fs( ) 10–+ +=


9


Calculating Outage Time:

[EQ 9-20]

Where:T Annual outage time (seconds)

r Fade occurrence factor

To (t/50)(8*106) = length of fade season (seconds)

t Average annual temperature in oF

CFM Fade Margin (dB)

Io Space Diversity Improvement Factor = 1 for non-diversity, > 1 for diversity

Calculating Availability:

[EQ 9-21]

Where:A Annual availability (%)

T Annual outage time (seconds)

9.3.1.5 PCS to Microwave Interference Summary

The methods and procedures required to perform microwave interference analysis are complex. Thus, this section serves to demonstrate the fundamental aspects of the process. A full guide detailing all scenarios is beyond the scope of this document. Therefore, it is recommended that Bulletin TSB-10-F be used as a reference when considering any in-depth PCS to Microwave interference analysis.

Note: Detailed analysis is best performed with the use of an automated microwave interference analysis tool.

TrTo

CFM10

------------- –

×10Io

-----------------------------------=

A31.4496

6×10 T–

31.44966×10

----------------------------------------

100×=


9


9.3.2 Microwave to PCS Interference

In contrast to PCS to microwave interference, there are no recommendations or guidelines presented by the TIA for the calculation of microwave interference into PCS systems. The PCS system supplier must therefore determine the appropriate method and levels.

The relocation of microwave links degraded by the PCS systems will naturally remove the majority of sources of microwave to PCS interference. However, it should not be assumed that no interference will occur.

9.3.2.1 General Considerations

Interference to PCS base stations is best characterized as a degradation to the receiver noise figure. The degradation to the noise figure produces an effective noise figure, which must then be used in the link budget for the affected cell or sector. The reason interference can be treated as noise is that the de-spreading following the receiver filtering will result in widening of the interferer’s spectrum such that it is seen as a noise rise.

The procedure for calculating the effects of microwave interference on the PCS base stations (BTSs) can similarly be applied in calculating the effects of microwave interference on the PCS subscriber units. However, it must be remembered that the subscriber unit receiver has a higher noise figure, and its selectivity is different from that of the base station. Hence, the calculations should reflect these differences.

9.3.2.2 Calculation of Nominal Noise Floor

The nominal noise floor is set by the bandwidth of the receiver and its noise figure. For example, the noise figure of some base station receivers is designed to be 6 dB. The corresponding noise bandwidth of these base station receivers is approximately 1.25 MHz.

Given:Nominal Noise = Nnom

Thermal Noise = Nth(dB) = -174 dBm/Hz

Noise Figure = NF(dB) = 6 dB

Noise Bandwidth = fnb = 1.25 MHz

The nominal noise is the linear sum of these three parameters, in dB:

Nnom(dB) = Nth(dB) + NF(dB) + 10 Log (fnb)

= -174 dBm/Hz + 6 dB + 61 dB Hz

= -107 dBm


9


To calculate the nominal noise floor in a subscriber unit receiver, the appropriate noise figure for the particular unit type must be substituted for the noise figure quoted for the base station receiver. All compliant subscriber units guarantee a noise figure of 10 dB, which compares to a 6 dB noise figure for the base station. The net result is that the subscriber unit receiver has a noise floor that is 4 dB higher than that for the base station.

9.3.2.3 Calculation of Effective Interference Power

The receiver filtering and the spectrum of the interferer establishes the effective interference power. The receiver filtering can be determined by the receiver desense curve. Since the desense specification includes the effects of processing gain and receiver Eb/No, these must be removed as the first step in the process. This is easy to do, as their effect is equal to the desense at 0 Hz channel offset, which is nominally 14 dB. Define the following:

Receiver Filtering = |H(f)| = - (Desense(f) - Desense(0)),

Where:f Frequency offset from the carrier frequency in Hz

Note, the minus sign in the above equation is due to the fact that desense is a positive quantity. The spectrum of the interfering signal must also be known,

Interferer Power = G(f)

Effective interference power is calculated as the integrated product:

Effective Interference Power = Ieff

=

Performing rectangular integration is adequate in most cases, allowing the calculation to be completed by using a summation of the products.

9.3.2.4 Calculation of Effective Noise Figure

The effective noise in the receiver's bandwidth is the sum of the nominal noise power and the effective interference power, i.e.

Effective Noise = N eff

= Nnom + Ieff

H f( )2G f( ) fd


9


Expressed in dB:

Effective Noise (dB)= Neff (dB)

= 10 log (Nnom + Ieff)

The ratio of the effective noise and the nominal noise is the effective noise figure:

Effective NF = NFeff

= Neff /Nnom

Expressed in dB:

Effective NF (dB) = 10 log [(Nnom + Ieff) / Nnom]

If Ieff = Nnom, the Effective NF is increased by 3 dB.

9.3.2.5 Microwave to PCS Interference Summary

Any interference in-band to the 1.25 MHz channel will directly add to the nominal noise power of either the base station or the subscriber unit. Therefore, with a 4 dB higher receiver noise floor, the subscriber unit is less sensitive than the base station. If it is assumed that an interfering signal 10 dB below the receiver sensitivity will cause a 1 dB increase in the signal to noise ratio, then the interfering signal at the subscriber unit must be 4 dB greater than that at the base station to cause an equivalent effect.

Whether either the base station receiver or the subscriber unit receiver is more severely degraded by a microwave interferer is determined on an individual basis. This determination includes dependencies such as the location of the microwave transmitter relative to the PCS system coverage area, and also the heights of both the base station antenna and the subscriber unit antenna.

9.4 References

1. ANSI IS-20A, Recommended Minimum Standards for 800-MHz Cellular Land Stations, May, 1988, Sections 3.4.1 and 3.4.4.

2. ANSI J-STD-019, Base Station Compatibility Requirements for 1.8 to 2.0 Ghz Code Division Multiple Access (CDMA) Personal Communications Systems, August, 1995.

3. Clapp, Scott (Motorola), “Inter-band Interference Control”, August 15, 1998.

4. EIA/TIA IS-97-A, Recommended Minimum Performance Standards for Base Stations Supporting Dual-Mode Wideband Spread Spectrum Cellular Mobile Stations, June, 1996, Sections 9.4.3, 9.4.4 and 10.5.1.


9


5. ETSI/GSM 05.05, Digital Cellular Telecommunications System Radio Transmission and Reception, July, 1996, Sections 2, 4.2.1, 4.7.2, 5.3 and 6.2.

6. Leonard, Terry (Motorola), “CDMA to GSM Base Station Interference Control”, May 5, 1997.

7. Tajaddini, Mohammad (Motorola), “Analysis of AMPS B Band and GSM Systems Interference in Co-Located Sites”, December 15, 1993.

8. United Kingdom Total Access Communication System Mobile Station - Land Station Compatibility Specification, Issue 4, Amendment 2, February, 1995, Sections A.7 and A.8.

9. Wilcox, Gordon (Motorola), “Radio Frequency Interference in Two-Way Radio Systems”, November, 1975.

10. TIA/EIA IS-97-D, Recommended Minimum Performance Standards for cdma2000 Spread Spectrum Base Stations, March 30, 2001. (See http://www.3gpp2.org/Public_html/specs/index.cfm for more information.)


9


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I Terms and Acronyms
Appendix
I

Table of Contents

I.1 Terms and Acronyms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . I - 3

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CDMA/CDMA2000 1X RF Planning GuideI Terms and Acronyms

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I


I.1 Terms and Acronyms• AGC - Automatic Gain Control

• AMR - Alarm, Monitoring and Reporting Card

• AMPS - Advanced Mobile Phone System/Service

• ARP - Average Rated Power

• ATCH - Actual Traffic CHannels (including SHO)

• BBX - Broad Band Transceiver card

• BBX-R - BBX Redundant

• BBX I/O - BBX Input Output card

• BDC - Baseband Distribution Card

• BHCA - Busy Hour Call Attempts, the number of call attempts during the busiest hour of the day

• BSC - See CBSC

• BSS - The Base Station System consists of one BSC and its associated BTSs

• BTA - Basic Trading Area

• BTS - The Base Transceiver Sub-System includes the equipment necessary to implement a CDMA Digital Cellular Base Station

• BTS Cluster - A group of BTSs controlled by a single BSC

• BTS Site - The location where a particular BTS resides

• CCP - CDMA Channel Processor

• CDMA - Code Division Multiple Access

• CBSC - The Centralized Base Site Controller consists of the Mobility Manager and Transcoder

• CCITT - International Consultative Committee for Telegraph and Telephony is a standards committee that recommends specific implementations of various communication protocols

• CPU - Central Processing Unit

• CSM - Clock Synchronization Module

• C7 - CCITT #7 or Signaling System #7 (SS7)

• CW - Continuous Wave

• dBc - Decibels below carrier

• dBd - Decibels referenced to a half wave dipole

• dBi - Decibels referenced to an Isotropic radiator

• dBm - Decibels referenced to a milliWatt

• DDC - Duplexer with integrated Directional Coupler

• DoD - Department of Defense

• EAMPS - Extended Advanced Mobile Phone System/Service

• Eb - Energy per Bit

• Ec - Energy per Chip


I


• EIRP - Effective Isotropic Radiated Power

• ELPA - Enhanced Linear Power Amplifier

• EMI - Electro-Magnetic Interference

• ERP - Effective Radiated Power

• ETCH - Effective Traffic CHannels

• F - Noise Factor

• FER - Frame Erasure Rate

• Frame - an enclosed rack of equipment

• FWT - Fixed Wireless Terminal

• GHz - Giga-Hertz (109 Hz)

• GLI - Group Line Interface

• GOS - Grade Of Service, the blocking probability.

• GPS - Global Positioning Satellite system used to synchronize Sites around the System

• GSM - Global System for Mobile communications (at 900 MHz) - Previously known as Groupe Special Mobile (Pan-European digital cellular standard). GSM900 is used only when necessary to differentiate it from DCS1800.

• HDII - High Density Analog Base Station

• HSO - High Stability Oscillator

• IM - InterModulation

• IPi - InPut intercept point

• ISI - Inter-System Interference

• ISO - International Standards Organization

• Io - Total interference density

• kbps - kilo bits per second

• kHz - kilo-Hertz (103 Hz)

• km - kilometers

• kTB - Thermal noise calculated from the product k x T x B, where k = Boltzmann’s constant (1.38x10-23

W/HzK), T = room temperature in degrees Kelvin (290 K), and B = bandwidth (in Hz)

• LFR - Loran Frequency Receiver card

• LMF - Local Maintenance Facility

• LNA - Low Noise Amplifier

• LORAN-C - LOng RAnge Navigation Low Frequency Broadcast

• LOS - Line-Of-Sight

• LPA - Linear Power Amplifier amplifies multiple carriers

• LTMS - Laboratory Test-oriented Mobile Station

• MAWI - Motorola Advanced Wideband Interface


I


• Mbps - Mega bits per second

• MCC - Multiple Channel CDMA card

• MF - 1. Multifrequency 2. Modulated Frequency 3. Modem Frame

• MHz - Mega-Hertz (106 Hz)

• MPC - Multicoupler Preselector Card

• MS - Mobile Station

• MSC - Mobile Switching Center

• MSF - European Low Frequency Broadcast of Standard Time

• MSI/O - The physical termination card for the RF Modem Frame (similar to the BIB)

• MTA - Major Trading Area

• NAMPS - Narrowband Advanced Mobile Phone Service

• NF - Noise Figure

• OH - OverHead Channels

• O & M - Operations and Maintenance

• PA - Power Amplifier

• PCM - Pulse Code Modulation

• PCS - Personal Communication System

• POTS - Plain Old Telephone Service

• PSTN - Public Switched Telephone Network

• PTCH - Physical Traffic CHannels (including SHO+OH)

• PN - Pseudo-random Noise spreading sequence

• QoS - Quality of Service

• RF - Radio Frequency

• RFMF - RF Modem Frame

• RFDS - The Radio Frequency Diagnostic Sub-system monitors the performance of the BTS

• RGD - Remote GPS Distribution box

• RGPS - Remote Global Positioning Satellite

• RL - Return Loss

• Rx or RX - Receive

• RXDC - Receiver Distribution Card

• SBN - Side Band Noise

• Sector - An RF coverage area segment

• Shelf - Generic name used to describe a mechanical enclosure, included in several types of BSS Frames

• SHO - Soft HandOff

• SIF - Site Interface Frame


I


• SNR - Signal to Noise Ratio

• Span Line - A T-1 (1.544 Mbps) or E-1 (2.048 Mbps) transmission link

• TCH - A Traffic CHannel is a single voice or data channel. Normally considered to be on the BTS side of the BSC and/or on the air interface.

• TDA - Time Difference of Arrival

• TDMA - Time Division Multiple Access

• TDR - Time Domain Reflectometer

• TIB - Telco Interconnect Board

• Trunk - A Trunk is a single 64 kbps voice or data channel (DS-0) on a given span line between the BSC and MSC.

• Tx or TX - Transmit

• TRX - Transceiver

• TTA - Tower Top Amplifier

• USDC - United States Digital Cellular, based upon the IS-136 specification

• UTC - Universal Coordinated Time

• VSWR - Voltage Standing Wave Ratio

• WiLL - Wireless Local Loop



II Glossary
Appendix
II

Table of Contents

II.1 Glossary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . II - 3

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CDMA/CDMA2000 1X RF Planning GuideII Glossary

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II


II.1 Glossary

Active SetThe pilots associated with the Forward Traffic Channels assigned to the subscriber. It is the base station that assigns all active set pilots to the subscribers.

AttenuatorA device for reducing the energy level of a signal without introducing distortion. Also called a pad.

BlockingThe inability of the calling subscriber to be connected to the called subscriber because either all paths are busy, or because idle paths in the calling group cannot be accessed by idle paths in the called group.

Candidate SetThe pilots that are not currently in the Active Set but have been received by the subscriber with sufficient strength to indicate that the associated Forward Traffic Channels could be successfully demodulated. As a property of the Mobile Assisted HandOff (MAHO), the subscriber promotes a Neighbor Set or Remaining Set pilot to the Candidate Set when certain pilot strength criteria are met, and then recommends the pilot to the base station for inclusion in the Active Set.

Channel1) A particular member of a group, that is associated with a unique time slot. Each member is associated with one port in the switch; either an RF channel, a land trunk, a three-party conference circuit, or a tone signalling port. 2) A particular member of an RF group that has a unique frequency. 3) For a TDMA air interface, it describes the unique frequency and time slot allocation for a single call. 4) For a CDMA air interface, it describes the Walsh code assignment allowed for the subscriber unit.

Directional CouplerA bi-directional coupler carrying Tx and Rx RF signals to and from the antennas. It includes a port which allows the signals to be routed to the RFDS for direct measurement of in-band forward (Tx) signals without service interruption.

ErlangA measure of telephone traffic intensity equivalent to the average number of simultaneous calls. Alternatively, it is the total circuit usage in an interval of time divided by that interval. Thus, 1 Erlang equals 3600 call seconds per hour or 36 CCS per hour.

EAMPSExtended Advanced Mobile Phone System - Refers to additional voice channels defined as an extension to the AMPS systems. Analogous to ETACS in TACS systems.


II


Neighbor SetThe pilots that are not currently in the Active Set or the Candidate Set, but are likely candidates for handoff. Neighbor Set pilots are identified by the base station via Neighbor List and Neighbor List Update messages.

PILOT_ARRIVALThe pilot arrival time is the time of occurrence of the earliest arriving usable multipath component of a pilot relative to the subscriber’s time reference.

PILOT_INCThe pilot PN sequence offset index increment is the interval between pilots, in increments of 64 chips. Its valid range is from 1 to 15. The subscriber uses this parameter in only one manner, to determine which pilots to scan from among the Remaining set. Only valid pilots (i.e. those pilots that are multiples of PILOT_INC) will be scanned. For the subscriber, PILOT_INC impacts only the scanning rate applied to Remaining pilots. It accomplishes this by reducing the number of Remaining pilots that need to be scanned.

For the base station, the effect of the PILOT_INC is different. In the base station, it is used in properly translating pilot phase back into pilot offset index. The consequence is that the operator may artificially increase the separation between valid time offsets. By selecting a PILOT_INC of 2, for instance, an operator chooses to limit the number of valid offsets to 256 (i.e. 0, 2, 4,..., 508, 510) instead of 512. The increased separation means that the pilot arrival must be larger before adjacent offset ambiguity is possible and consequently the likelihood of a strong adjacent interferer is reduced.

PILOT_PNThe pilot PN sequence offset (index), in units of 64 PN chips. It ranges from 0 to 511. Every transmit sector will have an offset assigned to it. This parameter is set for each sector.

PILOT_PN_PHASEThe subscriber reports pilot strength and phase measurements for each active and candidate pilot in the Pilot Strength Measurement Message (PSMM) when recommending a change in the handoff status (i.e. mobile assisted handoff). The subscriber computes the reported PILOT_PN_PHASE as a function of the PILOT_ARRIVAL and the PILOT_PN. The pilot arrival component represents the time delay of the pilot relative to the time reference or, in other words, how skewed the pilot is from the subscriber’s concept of system time. Note also that the subscriber does not identify pilots by their offset index directly, but by their phase measurement. If the pilot arrival was larger than 32 chips (1/2 of a pilot offset or 4.8 miles), then this could undermine the ability of the base station to properly translate pilot phase into pilot offset index (given a PILOT_INC of 1).

Remaining SetThe set of all possible pilots in the current system on the current CDMA frequency assignment, excluding pilots in the other sets. These pilots must be integer multiples of PILOT_INC (defined above).


II


Reuse PatternThe minimum number of cells required in a pattern before channel frequencies are reused, to prevent interference. Varies between cell configuration type (omni, sector, etc.) and channel type (traffic, control). The pattern shows assignments of adjacent channels to minimize interference between cells and sectors within the pattern area. In CDMA, reuse pattern refers mainly to the pattern of the pilot assigned to each sector in the system.

SRCH_WIN_AThis parameter represents the search window size associated with the Active Set and Candidate Set pilots. The subscriber centers the search window for each pilot around the earliest arriving usable multipath component of the pilot. Note that in contrast to the neighbor or remaining set search windows, the active/candidate search windows "float" with the desired signals. That is to say that the center position of the search window is updated every scan to track the new location of the earliest arriving multipath component.

SRCH_WIN_N, SRCH_WIN_RThese parameters represent the search window sizes associated with Neighbor Set and Remaining Set pilots. The subscriber centers the search window for each pilot around the pilot’s PN sequence offset using timing defined by the subscriber’s time reference.

In general, a neighbor search window, SRCH_WIN_N, will be sized so as to encompass the geographic area in which the neighbor may be added (a soft handoff “add” zone or “initial detection area”). The largest a neighbor search window need be is such that it is sufficient to cover the distance between the neighbors, , plus an accommodation of the time-of-flight delay (approx. 3 chips).

To illustrate these relationships better, consider the following scenario. A subscriber monitors a neighbor pilot. The neighbor search window is centered on the neighbor pilot offset. This centering is relative based on timing derived from the time reference. When the pilot strength of a neighbor pilot recommends promotion to the candidate set, then the search window will be tightened to the active search window size. The active search window is sized to compensate for delay spread only and is therefore smaller than the neighbor search window. In addition, the active search window locks onto and tracks the candidate pilot.

System TimeAll base station digital transmissions are referenced to a common CDMA system-wide time scale that uses the Global Positioning System (GPS) time scale, which is traceable to and synchronous with Universal Coordinated Time (UTC).

Time ReferenceThe subscriber establishes a time reference which is used to derive system time. This time reference will be the earliest arriving multipath component being used for demodulation. This reflects the assumption that the subscriber’s fix on system time is always skewed by delay associated with the shortest active link.

3R


II


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III Watts to dBm
Appendix
III

Table of Contents

Conversion Table

III.1 Watts to dBm Conversion Table . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . III - 3

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III

CDMA/CDMA2000 1X RF Planning GuideIII Watts to dBm Conversion Table

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III - 2 CDMA/CDMA2000 1X RF Planning Guide JAN 2013

III


III.1 Watts to dBm Conversion Table

The following table provides a conversion from Watts to dBm.

Table III-1: Watts to dBm Conversion Table

Watts dBm Watts dBm Watts dBm Watts dBm Watts dBm

200 53.010 174 52.405 148 51.703 122 50.864 96 49.823

199 52.989 173 52.380 147 51.673 121 50.828 95 49.777

198 52.967 172 52.355 146 51.644 120 50.792 94 49.731

197 52.945 171 52.330 145 51.614 119 50.755 93 49.685

196 52.923 170 52.304 144 51.584 118 50.719 92 49.638

195 52.900 169 52.279 143 51.553 117 50.682 91 49.590

194 52.878 168 52.253 142 51.523 116 50.645 90 49.542

193 52.856 167 52.227 141 51.492 115 50.607 89 49.494

192 52.833 166 52.201 140 51.461 114 50.569 88 49.445

191 52.810 165 52.175 139 51.430 113 50.531 87 49.395

190 52.788 164 52.148 138 51.399 112 50.492 86 49.345

189 52.765 163 52.122 137 51.367 111 50.453 85 49.294

188 52.742 162 52.095 136 51.335 110 50.414 84 49.243

187 52.718 161 52.068 135 51.303 109 50.374 83 49.191

186 52.695 160 52.041 134 51.271 108 50.334 82 49.138

185 52.672 159 52.014 133 51.239 107 50.294 81 49.085

184 52.648 158 51.987 132 51.206 106 50.253 80 49.031

183 52.625 157 51.959 131 51.173 105 50.212 79 48.976

182 52.601 156 51.931 130 51.139 104 50.170 78 48.921

181 52.577 155 51.903 129 51.106 103 50.128 77 48.865

180 52.553 154 51.875 128 51.072 102 50.086 76 48.808

179 52.529 153 51.847 127 51.038 101 50.043 75 48.751

178 52.504 152 51.818 126 51.004 100 50.000 74 48.692

177 52.480 151 51.790 125 50.969 99 49.956 73 48.633

176 52.455 150 51.761 124 50.934 98 49.912 72 48.573

175 52.430 149 51.732 123 50.899 97 49.868 71 48.513

III - 3CDMA/CDMA2000 1X RF Planning GuideJAN 2013

III


Watts = [10(dBm/10)] / 1000

dBm = 10 Log(Watts * 1000)

dBm = dB + 30

70 48.451 46 46.628 22 43.424 0.94 29.731 0.46 26.628

69 48.388 45 46.532 21 43.222 0.92 29.638 0.44 26.435

68 48.325 44 46.435 20 43.010 0.90 29.542 0.42 26.232

67 48.261 43 46.335 19 42.788 0.88 29.445 0.40 26.021

66 48.195 42 46.232 18 42.553 0.86 29.345 0.38 25.798

65 48.129 41 46.128 17 42.304 0.84 29.243 0.36 25.563

64 48.062 40 46.021 16 42.041 0.82 29.138 0.34 25.315

63 47.993 39 45.911 15 41.761 0.80 29.031 0.32 25.051

62 47.924 38 45.798 14 41.461 0.78 28.921 0.30 24.771

61 47.853 37 45.682 13 41.139 0.76 28.808 0.28 24.472

60 47.782 36 45.563 12 40.792 0.74 28.692 0.26 24.150

59 47.709 35 45.441 11 40.414 0.72 28.573 0.24 23.802

58 47.634 34 45.315 10 40.000 0.70 28.451 0.22 23.424

57 47.559 33 45.185 9 39.542 0.68 28.325 0.20 23.010

56 47.482 32 45.051 8 39.031 0.66 28.195 0.18 22.553

55 47.404 31 44.914 7 38.451 0.64 28.062 0.16 22.041

54 47.324 30 44.771 6 37.782 0.62 27.924 0.14 21.461

53 47.243 29 44.624 5 36.990 0.60 27.782 0.12 20.792

52 47.160 28 44.472 4 36.021 0.58 27.634 0.10 20.000

51 47.076 27 44.314 3 34.771 0.56 27.482 0.08 19.031

50 46.990 26 44.150 2 33.010 0.54 27.324 0.06 17.782

49 46.902 25 43.979 1 30.000 0.52 27.160 0.04 16.021

48 46.812 24 43.802 0.98 29.912 0.50 26.990 0.02 13.010

47 46.721 23 43.617 0.96 29.823 0.48 26.812 0.01 10.000

Table III-1: Watts to dBm Conversion Table

Watts dBm Watts dBm Watts dBm Watts dBm Watts dBm

III - 4 CDMA/CDMA2000 1X RF Planning Guide JAN 2013


IV Complementary Error
Appendix
IV

Table of Contents

Function Table

IV.1 Complementary Error Function Table . . . . . . . . . . . . . . . . . . . . . . . . . . IV - 3

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IV

CDMA/CDMA2000 1X RF Planning GuideIV Complementary Error Function Table

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IV


IV.1 Complementary Error Function Table

The following Complementary Error Function Table is supplied for the reader’s reference. Note that the value of x within Q(x) is the sum of value in the first column of a specific row plus the value given in the top row. For example, Q(0.76) corresponds to 0.2236 and Q(2.42) corresponds to 0.0078.

Table IV-1: Complementary Error Function, Q(x)

Q(x)

x 0.00 0.01 0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09

0.0 0.5000 0.4960 0.4920 0.4880 0.4840 0.4801 0.4761 0.4721 0.4681 0.4641

0.1 0.4602 0.4562 0.4522 0.4483 0.4443 0.4404 0.4364 0.4325 0.4286 0.4246

0.2 0.4207 0.4168 0.4129 0.4090 0.4052 0.4013 0.3974 0.3936 0.3897 0.3859

0.3 0.3821 0.3783 0.3745 0.3707 0.3669 0.3632 0.3594 0.3557 0.3520 0.3483

0.4 0.3446 0.3409 0.3372 0.3336 0.3300 0.3264 0.3228 0.3192 0.3156 0.3121

0.5 0.3085 0.3050 0.3015 0.2981 0.2946 0.2912 0.2877 0.2843 0.2810 0.2776

0.6 0.2743 0.2709 0.2676 0.2644 0.2611 0.2579 0.2546 0.2514 0.2483 0.2451

0.7 0.2420 0.2389 0.2358 0.2327 0.2297 0.2266 0.2236 0.2207 0.2177 0.2148

0.8 0.2119 0.2090 0.2061 0.2033 0.2005 0.1977 0.1949 0.1921 0.1894 0.1867

0.9 0.1841 0.1814 0.1788 0.1762 0.1736 0.1710 0.1685 0.1660 0.1635 0.1611

1.0 0.1586 0.1562 0.1539 0.1515 0.1492 0.1468 0.1446 0.1423 0.1401 0.1378

1.1 0.1357 0.1335 0.1313 0.1292 0.1271 0.1251 0.1230 0.1210 0.1190 0.1170

1.2 0.1151 0.1131 0.1112 0.1093 0.1075 0.1056 0.1038 0.1020 0.1003 0.0985

1.3 0.0968 0.0951 0.0934 0.0917 0.0901 0.0885 0.0869 0.0853 0.0838 0.0823

1.4 0.0807 0.0793 0.0778 0.0764 0.0749 0.0735 0.0721 0.0708 0.0694 0.0681

1.5 0.0668 0.0655 0.0643 0.0630 0.0618 0.0606 0.0594 0.0582 0.0570 0.0559

1.6 0.0548 0.0537 0.0526 0.0515 0.0505 0.0495 0.0485 0.0475 0.0465 0.0455

1.7 0.0446 0.0436 0.0427 0.0418 0.0409 0.0401 0.0392 0.0384 0.0375 0.0367

1.8 0.0359 0.0352 0.0344 0.0336 0.0329 0.0322 0.0314 0.0307 0.0301 0.0294

1.9 0.0287 0.0281 0.0274 0.0268 0.0262 0.0256 0.0250 0.0244 0.0239 0.0233

2.0 0.0228 0.0222 0.0217 0.0212 0.0207 0.0202 0.0197 0.0192 0.0188 0.0183

2.1 0.0179 0.0174 0.0170 0.0166 0.0162 0.0158 0.0154 0.0150 0.0146 0.0143

2.2 0.0139 0.0136 0.0132 0.0129 0.0126 0.0122 0.0119 0.0116 0.0113 0.0110

2.3 0.0107 0.0105 0.0102 0.0099 0.0097 0.0094 0.0091 0.0089 0.0087 0.0084

IV - 3CDMA/CDMA2000 1X RF Planning GuideJAN 2013

IV


2.4 0.0082 0.0080 0.0078 0.0076 0.0074 0.0072 0.0070 0.0068 0.0066 0.0064

2.5 0.0062 0.0060 0.0059 0.0057 0.0056 0.0054 0.0052 0.0051 0.0049 0.0048

2.6 0.0047 0.0045 0.0044 0.0043 0.0042 0.0040 0.0039 0.0038 0.0037 0.0036

2.7 0.0035 0.0034 0.0033 0.0032 0.0031 0.0030 0.0029 0.0028 0.0027 0.0026

2.8 0.0026 0.0025 0.0024 0.0023 0.0023 0.0022 0.0021 0.0021 0.0020 0.0019

2.9 0.0019 0.0018 0.0018 0.0017 0.0016 0.0016 0.0015 0.0015 0.0014 0.0014

3.0 0.0014 0.0013 0.0013 0.0012 0.0012 0.0011 0.0011 0.0011 0.0010 0.0010

3.1 0.0010 0.0009 0.0009 0.0009 0.0008 0.0008 0.0008 0.0008 0.0007 0.0007

3.2 0.0007 0.0007 0.0006 0.0006 0.0006 0.0006 0.0006 0.0005 0.0005 0.0005

3.3 0.0005 0.0005 0.0005 0.0004 0.0004 0.0004 0.0004 0.0004 0.0004 0.0004

3.4 0.0003 0.0003 0.0003 0.0003 0.0003 0.0003 0.0003 0.0003 0.0003 0.0002

3.5 0.0002 0.0002 0.0002 0.0002 0.0002 0.0002 0.0002 0.0002 0.0002 0.0002

3.6 0.0002 0.0002 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001

3.7 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001

3.8 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001 0.0001

3.9 0.0000 0.0000 0.0000 0.0000 0.0000 0.0000 0.0000 0.0000 0.0000 0.0000

Table IV-1: Complementary Error Function, Q(x)

Q(x)

x 0.00 0.01 0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09

IV - 4 CDMA/CDMA2000 1X RF Planning Guide JAN 2013

1x rf planning guide - nokia

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