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Intensity-Modulation Full-Field DetectionOptical Fast OFDM

Jian Zhao

AbstractWe propose and investigate intensity-modulation (IM) full-field detection (FD) optical fastorthogonal frequency division multiplexing (F-OFDM).

A16 .8 Gbits binary phase-shift keying (BPSK) FD F-OFDMsystem over 480 km fiber transmission is experimentallydemonstrated to validate the feasibility. BPSK and four am-plitude phase-shift keying (4ASK) FD optical F-OFDM withoptimized system parameters are numerically investigatedand compared with IM direct-detection (DD) F-OFDM andIM FD conventional OFDM at the same spectral efficiency.It is shown that the proposed scheme, while avoiding the

use of a coherent receiver, exhibits greatly improveddispersion tolerance over DD optical F-OFDM. It is robustto the phase noise induced in the full-field reconstruction.

As a result, significant performance advantages over FDconventional OFDM is obtained.

Index TermsChromatic dispersion; Detection;Modulation; Optical fast OFDM.

I. INTRODUCTION

O

ptical fast orthogonal frequency division multiplex-

ing (F-OFDM) [1

14], with subcarrier spacingreduced to half of that in conventional OFDM, is a prom-ising multicarrier scheme. The basic concept of F-OFDMwas first proposed in wireless communications [1517]and has been studied thereafter [1823]. Recently, thisconcept was introduced to optical communications [1] andworld-first experiments were performed to verify its imple-mentation feasibility at both 1.55 [2] and 2 m opticalwavelengths [3]. In coherent detection, optical F-OFDM ex-hibits greatly improved performance over conventionalOFDM in frequency offset compensation [4]. In thisscheme, the subcarrier multiplexing and demultiplexingare commonly implemented using a discrete cosine trans-

form (DCT) pair. The designs of precise symbol synchroni-zation [5] and guard interval (GI) [6] specific to opticalF-OFDM were proposed. It was shown that, in contrastto conventional OFDM, a symmetric extension (SE) ratherthan cyclic-extension-based GI was required to enablechromatic dispersion (CD) compensation using one-tapequalizers. By using this design, transmission of coherentoptical F-OFDM over 840 km single-mode fiber (SMF) was

demonstrated. However, coherent detection is expensiveand is not suitable for cost-sensitive applications, suchas 40/100/400G Ethernet, and access and short metronetworks.

On the other hand, direct-detection (DD) conventionalOFDM has been extensively investigated. Both double-sideband (DSB) and single-sideband (SSB) DD OFDMsystems have been proposed [2429]. DSB DD OFDMemploying intensity modulation (IM) has the simplestimplementation [2427]. However, it has significantlydegraded CD tolerance. IM and DD were also applied inoptical F-OFDM recently over either single-mode [810]or multimode fiber [1113] for cost-sensitive applications.SSB OFDM exhibits better performance than DSB OFDM;it is implemented either by inserting a spectral gap be-tween the optical carrier and the signal [28] or by applyingthe information to the envelope of the SSB signal [29]. Theformer exhibits high performance approaching that usingcoherent detection, but increases the specification require-ments in the sampling rate and component bandwidth. Incertain cases, additional components, such as an electricalI-Qmodulator, are needed. The latter scheme still exhibitsreduced CD tolerance and requires an additional Hilberttransform. Furthermore, an I-Q optical modulator isrequired for the SSB systems unless an optical filter withsharp roll-off is used.

Full-field detection (FD) [3036], which can extract thefull optical field using an asymmetric MachZehnder inter-ferometer (AMZI) and two photodiodes, is a promisingscheme to balance the performance and the implementa-tion complexity. This scheme was recently demonstratedin both onoff keying (OOK) [32] and offset differentialquadrature phase-shift keying [33]. The recovered opticalfield allows subsequent near-ideal CD compensation.

However, to the best of our knowledge, FD multicarriersystems (including both conventional OFDM and F-OFDM)have not been investigated yet. In this paper, we proposefor the first time IM FD optical F-OFDM, and compareit with DD F-OFDM and FD conventional OFDM. We showthat the proposed scheme, while avoiding the use of acoherent receiver, exhibits greatly improved CD toleranceover DD optical F-OFDM. It is also shown that opticalF-OFDM using single-quadrature formats is more robustto the phase noise induced in full-field reconstruction,and so shows great performance advantages when com-pared to FD conventional OFDM at the same spectralefficiency. These advantages make the proposed scheme

very promising for applications that demand low costhttp://dx.doi.org/10.1364/JOCN.5.000465

Manuscript received December 18, 2012; revised February 13, 2013;

accepted March 26, 2013; published April 18, 2013 (Doc. ID 181999).

Jian Zhao is with Photonic Systems Group, Tyndall National Institute

and Department of Physics, University College Cork, Lee Malting, Prospect

Row, Cork, Ireland (e-mail: [email protected]).

Jian Zhao VOL. 5, NO. 5/MAY 2013/J. OPT. COMMUN. NETW. 465

1943-0620/13/050465-10$15.00/0 2013 Optical Society of America

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but higher performance beyond the conventional IM DDsolution, including long-reach 40/100/400G Ethernet(80 km), long-reach access networks (100 km), and shortmetro networks (300 km).

The paper is organized as follows: we present the principleof the proposed scheme in SectionII. Its feasibility is exper-imentally validated in Section III. In SectionIV, extensive

simulations are performed to study the performance limitsand the advantages over other schemes. Finally, SectionVsummarizes the results.

II. PRINCIPLE

The decoding principle of the proposed scheme is illus-trated in Fig. 1. The optical signal is first processedby an AMZI with 2differential phase shift andtdiffer-ential time delay (DTD). The value oft is chosen to bal-ance the precision of phase reconstruction and the noiseimpact. The optimized t value also depends on thedispersion. The two outputs of the AMZI are detected bytwo photodiodes, sampled by analog-to-digital converters(ADCs). In practical implementation, these samples are in-terpolated, allowing a high resolution for the identificationof the start-of-frame (SOF) symbol and the determinationof the DCT window. Additionally, the 2differential phaseshift of the AMZI results in similar received power on thetwo photodiodes. Consequently, the retiming stage can beapplied to the two received signals separately, and the rel-ative timing delay between them can also be automaticallyrecovered. In the case where the receivers are AC coupled,biases are added before the full-field reconstruction. As-suming that the baseband optical field is jEtj expjt,where Et and t are the optical field and its phase,respectively, Vt and Vt in Fig.1are

Vt jEtj2 jEt tj2 2jEtj jEt tj

cost t t AMZI; (1)

where AMZIis the differential phase of the AMZI 2.If the t value is not large, jEt tj jEtj. We canderive

VftVt Vt

Vt Vt sint t t: (2)

In the case that t t t is small, sintt t t t t. By recovering the phase differ-ence in t, the full optical field can be reconstructed by

Vfullt Vt Vt12 exp

j

Z asinVftt dt

.

(3)

Ideally, Vfullt is perfectly obtained given a sufficientlysmall t. However, in practice, similar to the FD OOK for-mat [32], low-frequency components of the noise accumu-late during the integration, and so have to besuppressed by the use of a high-pass electrical filter.

The recovered signal consists of amplitude and phase, sothe F-OFDM decoding is similar to that in coherent detec-tion. By using the SE- instead of CP-based GI, the opticalF-OFDM subcarriers can be demultiplexed without inter-carrier interference [6]. CD results only in different con-stants multiplied to different subcarriers, and so can becompensated using one-tap equalizers, as shown in Fig.1.

III. EXPERIMENTAL STUDY

An experiment was demonstrated to verify the feasibil-ity of the proposed scheme. Figure2shows the experimen-tal setup for 16.8 Gbits binary phase-shift keying (BPSK)optical F-OFDM. The data was encoded in Matlab. The in-verse-DCT (IDCT) and DCT used 128 points, of which 108subcarriers (Nos. 3110) were used for data transmission.The first subcarrier was not modulated, allowing for theDC bias for IM. The last 18 subcarriers were zero-paddedto avoid aliasing. After IDCT and parallel-to-serial (P/S)conversion, 12 samples were added to each symbol as aSE-based GI. The peak-to-average power ratio (PAPR)was controlled to be 8 dB by clipping. The generatedF-OFDM signal was downloaded to a 20 GSs arbitrarywaveform generator with a resolution of 8 bits. The signalrate including the GI and forward error correction was16.8Gbits. A distributed feedback laser with an 1MHzlinewidth was used to generate the optical carrier. A MachZehnder modulator (MZM) was used for modulation with

Fig. 1. Decoding principle of the proposed scheme. Fig. 2. Experimental setup for FD optical F-OFDM.

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a peak-to-peak voltage of 3.6 V. This was equivalent to0.65V. The bias was set to be1.4V away from the nullpoint of the modulator, unless otherwise stated.

The modulated optical signal was then amplified andtransmitted over a recirculating loop comprising 60 kmSMF with 14 dB fiber loss. The noise figure of the loopamplifier was5 dB and another 0.8 nm optical bandpass

filter (OBPF) was used in the loop to suppress the amplifiedspontaneous emission noise. The launch power per spanwas around 5 dBm. At the receiver, the optical signalwas detected with an optically preamplified receiver anda variable optical attenuator (VOA) was used to vary theinput power to an erbium-doped fiber amplifier (EDFA).The preamplifier was followed by an OBPF with a 3 dBbandwidth of 0.64 nm, a second EDFA, and another opticalfilter with a 3 dB bandwidth of 0.8 nm. Then the opticalsignal passed through a Kylia AMZI with 20 ps DTD and2 differential phase shift. The two outputs of the AMZIwere detected by two 7 GHz photodiodes. Both detected sig-nals were sampled by a real-time oscilloscope at 25GSs.

In off-line processing (implemented in Matlab), the signalswere upsampled. An algorithm similar to that in [5] wasused for automatic symbol synchronization. The algorithmconsisted of two stages. The first stage identified the SOFsymbol while the second one precisely determined the DCTwindow. This method was dispersion transparent, and theperformance was comparable to that in coherent detectionwhere the standard variation of timing errors was less than2.2 ps for OSNRs as low as 3 dB [ 5]. Note that, differentfrom coherent detection, the output power of the two photo-diodes was approximately the same. Consequently, thealgorithm could be applied to the two received signals sep-arately, and could simultaneously remove any temporal

misalignment between the signals due to the imbalancedpath lengths of the two receiver chains. Because of theAC-coupled receivers, bias was added to the two signalsbefore they were added and subtracted for full-fieldreconstruction. The decoding algorithms after full-fieldreconstruction included DCT and channel equalization.Conventional time-domain averaging over multiple train-ing symbols (TSs) [37] was used for channel estimation,and the number of the TSs was 20 unless otherwisestated. By using SE-based GI, F-OFDM symbols couldbe demultiplexed by DCT without intercarrier interference[6]. Consequently, similar to conventional OFDM, one-tapequalizers could be used to compensate the dispersion.Note that frequency-domain equalization as used in FDOOK could be used to avoid the GI and reduce the over-head. However, this method increased the implementationcomplexity of CD compensation when compared to one-tapequalizers, and so was not considered in this paper. Foreach experimental curve in the following investigation,three sets of data were extracted independently, with eachconsisting of 1200 measured optical F-OFDM symbols,giving a total number of measured bits of108 1200 3 388;800. The results indicated that the bit-error-rate (BER)variation was not significant for these three sets of data,so the recirculating loop, as well as the symbol synchroni-zation and channel estimation algorithms, was relativelystable during the measurements. Note that the use of asynchronous polarization scrambler would further stabi-lize the BER measurements.

Figure3depicts the performance of FD optical F-OFDMafter 240 (triangles), 360 (squares), and 480 km (dia-monds), in comparison with the back-to-back case (circles).The insets of Fig. 3illustrate the recovered constellation

14 16 18 20 22 24

-2

-3

-4

-2 -1 0 1 2-2

-1

0

1

2

-2 -1 0 1 2-2

-1

0

1

2

-2 -1 0 1 2-2

-1

0

1

2

Received OSNR (dB)

Log10(BER)

0 km

240 km

360 km

480 km

Real component (a.u.)

0 km

Imaginary

component(a.u.)

360 km 480 km

Fig. 3. Performance versus OSNR for BPSK FD F-OFDM. Insets are the recovered constellation diagrams at21dB OSNR.


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diagrams at around 21 dB OSNR. The investigated systemrequired 18 dB OSNR for a BER of 103. We expect animproved back-to-back performance by using a narrowerOBPF after the preamplifier (as investigated in the nextsection) and reduced PAPR with pre-coding. It is alsoshown that the penalties after 360 and 480 km were lessthan 1 and 2 dB, respectively. These penalties might havebeen caused by the imperfectly recovered phase duringfull-field reconstruction, and the insufficient GI length.At 0 km, the signal phase was close to zero, and so wouldnot have a significant impact on the signal performance.However, as the fiber length increased (240360 km), therole of the reconstructed phase became important and therequirement for precise phase estimation was morestringent. In the experiment, imperfect bias value, t ofthe AMZI, thermal noise, and limited bandwidth ofphotodiodes could be potential sources to reduce theestimation precision. For 480 km, the CD value was larger,and the additional 1 dB penalty was likely induced by theinsufficient GI length.

Figure 4 gives the BER measurements for more fiberlengths. The OSNR values for 60, 120, 180, 240, 300,360, 420, and 480 km are around 22.6, 22.3, 22, 21.6,21.4, 21.1, 20.8, and 20.7 dB, respectively. As expected,the BER degraded gradually as the transmission distanceincreased, due to the degraded OSNR and transmissionpenalty. In particular, the insufficient GI length might re-sult in the BER degradation for 420 and 480 km. However,a BER of 6 104 was still achieved after 480 km at20.7 dB OSNR.

In the experiment, a bias of1.4V with respect to thenull point of the modulator and a 1.5 GHz high-pass filterafter the integrator in phase reconstruction were used. Theeffect of this bias on the performance is shown in Fig. 5. Inprinciple, a bias closer to the null point results in signaldistortion arising from clipping after square-law detection,while a bias farther away from the null point wouldincrease the DC component and the associated power con-sumption. It is observed from Fig.5 that the performancewas optimized for a bias range of1.5 to 1V for both 0and 360 km. The optimal bias for 360 km was slightlycloser to the null point when compared to the 0 km casedue to the dispersed/broadened signal.

As described in SectionII, a high-pass filter is requiredto suppress the low-frequency noise amplification.Figures 6(a) and 6(b) depict the spectra of the reconstructedVf and the output of the integrator (before the electrical

high-pass filter). Statistically, Vf should have very fewlow-frequency signal components, as can been seen inFig.6(a). The DC component in the figure might have beeninduced by the imperfect bias added to V

t and Vt.

However, Fig. 6(b) shows that the low-frequency compo-nents of the reconstructed phase were very large. In fact,the integrator has a frequency transfer function of12jf,where f is the frequency. It is clear that the componentswith small f values dominate during integration, with ascaling factor of1j2fj. Therefore, any noise or inaccuracyin the low-frequency components would accumulate andeventually limit the system performance.

Figure7 depicts the performance versus the bandwidth

of the high-pass electrical filter to suppress the low-frequency components. The figure confirms the necessityof the high-pass filter. In the absence of the high-pass filteror when the filter bandwidth was not sufficiently large, thelow-frequency components of the noise accumulated duringintegration, which severely degraded the performance. Onthe other hand, the filter also removed the low-frequencycomponents of the signal, which resulted in signal distor-tion when the filter bandwidth was too large. The optimalfilter bandwidths for both distances were 1.5 GHz.

Figure 8investigates the fiber nonlinearity effect whenthe transmission distance is 360 km. It can be seen that

signal power (into the fiber) of1

dBm resulted in similarperformance as that of5 dBm. It also suggests that, at5 dBm as used in the experiment, the nonlinearity effectwas negligible. When the input power was increased beyond3 dBm, the performance started to be degraded prominently.

In the experiment, the coefficients of the one-tap equal-izers were obtained by time-domain averaging using TSs,the number of which should be sufficient to mitigate thenoise effect. Figure9 depicts the BER versus the numberof TSs after 360 and 480 km. The performance was near-optimal when the number of TSs was larger than fivefor both distances. It was equivalent to 35 ns adaptationtime. In practice, additional channel tracking might also be

needed after the initial training.

0 100 200 300 400 500

-3

-4

-5

Fiber length (km)

Log10(BER)

Fig. 4. Performance as a function of transmission distance.

-3 -2.5 -2 -1.5 -1 -0.5

-2

-3

-4

Log10(BER)

Bias voltage with respect to the null point (V)

0 km; 21.1dB OSNR

360 km; 21.1dB OSNR

Fig. 5. Performance sensitivity to the MZM bias at 0 and 360 km.


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IV. SIMULATION INVESTIGATION

Section III verifies the feasibility of the proposedscheme. However, the experimental parameters were notoptimized. In this section, we present results from numeri-cal simulations to investigate the performance limit, and tocompare it with other schemes to illustrate its performanceadvantages. To facilitate the comparison with the experi-ment, the simulation setup, ADC sampling rate, data rate,etc., are similar to those in Section III. However, the datarate/distance can be readily rescaled to provide guidelinesfor practical applications, such as 56Gbits 2 for 100GE

using commercially available 2556 GSs ADCs. TheIDCT/DCT used 128 points, of which 108 subcarriers(Nos. 3110) were used for data transmission. The sam-pling rate of the digital-to-analog converter (DAC)/ADCwas 20 GSs. A 9 GHz third-order Gaussian-shaped low-pass filter was used after the DAC to remove the aliasing.The laser linewidth was 1 MHz, unless otherwise stated.

The peak-to-peak voltage of the signal was 0.65V. Thefiber had a CD value of 16 psnmkm and a loss of0.2dBkm. The bandwidth of the third-order Gaussian-shaped OBPF at the receiver was optimized and was foundto be around 20 GHz. The DTD of the AMZI was 20 ps andthe receiver electrical filter bandwidth was 10 GHz. Both16.8Gbits BPSK and 33.6 Gbits four amplitude phase-shift keying (4ASK) FD F-OFDM systems were simulated.

For comparison, BPSK/4ASK DD optical F-OFDM andQPSK/16 quadrature amplitude modulation (16QAM) FDconventional OFDM were also simulated with the sameparameters as FD F-OFDM. In the first case, the signalat the receiver was directly detected. In the digital domain,

the square root was applied to the detected signal. The sig-nal was then serial-to-parallel converted, demultiplexed byDCT, equalized by one-tap equalizers, and decoded. In thesecond case, as F-OFDM had a subcarrier spacing reducedto half of that in conventional OFDM, QPSK/16QAM wasemployed in the conventional OFDM to obtain the samespectral efficiency [1]. This can also be understood as fol-lows: the 128-point DCT in F-OFDM could directly gener-ate a real time-domain signal with 108 subcarrierscarrying BPSK /4ASK data. On the other hand, in order

-20 -10 0 10 20

-60

-40

-20

0

20

-20 -10 0 10 20

40

60

80

100

120

Spectrumo

fVf(t)(dB)

Spectrumo

fVp

(t)(dB)

Frequency (GHz) Frequency (GHz)

(b)(a)

Fig. 6. Spectra of (a) the recoveredVft and (b) its integration, Vpt, versus the frequency at 480 km.

0 1 2 3 4

-2

-3

-4

360 km; 21.1dB OSNR

480 km; 20.7dB OSNR

Filter bandwidth (GHz)

L

og10(BER)

Fig. 7. Performance versus the bandwidth of the high-pass filterin full-field reconstruction for noise suppression.

16 18 20 22 24 26

-2

-3

-4

-5 dBm

-1 dBm

3 dBm

5 dBm

Received OSNR (dB)

Log10(BER)

Fig. 8. BER versus OSNR for different input power at

360 km.

0 10 20 30 40

-2

-3

-4

360 km

480 km

Number of training symbols

Log10(BER)

Fig.9. BER versusthe number of TSs at 360 km(OSNR: 21.1dB)

and 480 km (OSNR: 20.7 dB).


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to produce a real time-domain signal for IM, the subcar-riers in conventional OFDM had to be Hermitian symmet-ric. For a fixed number of samples (or time period) persymbol, the numbers of subcarriers for DC, data, and zero-padding in conventional OFDM were only 1, 54, and 9,respectively. Consequently, QPSK/16QAM should be usedin conventional OFDM to obtain the same spectral effi-ciency as BPSK/4ASK optical F-OFDM.

Figure10shows the required OSNR to achieve a BER of103 versus transmission distance for (a) BPSK and(b) 4ASK F-OFDM with different GI lengths. The perfor-mance of DD optical F-OFDM with the GI length of 12is also illustrated for comparison (circles). In Fig.10(a), thecurve for FD F-OFDM with system parameters the same asthose in the experiment is also plotted. It can be seen fromFig. 10(a) that, with the experimental parameters, thesimulated back-to-back receiver sensitivity at 103 wasaround 17 dB. This was close to the experimental result(18 dB). The additional 1 dB penalty might be due tothe imperfect differential phase shift of the AMZI, the bias

of the MZM, and timing error arising from symbol synchro-nization. In the simulation, the transmission reach at 2 dBpenalty was 600 km, which was also close to that in theexperiment (480 km). When the system parameters wereoptimized, 3.5dB improvement in the back-to-backreceiver sensitivity was observed in BPSK FD F-OFDM.

From Figs. 10(a) and 10(b), the required OSNRs at aBER of 103 for BPSK and 4ASK were around 13.5 and21.5 dB, respectively. It is also seen that the supportedtransmission reach increased as the length of the GI in-creased. With the GI length of 12, the transmission reachesat 3 dB OSNR penalty were 600 and 480 km for BPSK and4ASK, respectively. In contrast, DD optical F-OFDM exhib-ited limited dispersion tolerance. At 3 dB OSNR penalty,only 30 km could be supported for both 16.8 Gbits BPSKand 33.6 Gbits 4ASK.

To further illustrate the advantage of the proposedscheme, Fig. 11 compares the performance of FD opticalF-OFDM and FD conventional OFDM for (a) BPSKF-OFDM and QPSK OFDM and (b) 4ASK F-OFDM and16QAM OFDM. Figures 12(a)12(d)depict the recoveredconstellation diagrams of (a) BPSK F-OFDM, (b) QPSKOFDM, (c) 4ASK F-OFDM, and (d) 16QAM OFDM at 0 km.From these figures, it can be clearly seen that, at the samespectral efficiency, F-OFDM outperformed conventionalOFDM, and around 3.5 dB penalties were observed for

FD conventional OFDM in Figs.11(a)and 11(b). The per-formance advantage of F-OFDM in FD may be due to thefact that F-OFDM with single-quadrature formats is morerobust to the phase noise induced in the full-fieldreconstruction. Note that this phase noise might not beslow-varying, so would introduce intercarrier interferenceas well as phase rotation.

0 200 400 60020

24

28

32

Fiber length (km)

RequiredOSN

RatBERof10-3 DD; 12 GI

FD; 2 GI

FD; 6 GI

FD;12 GI

(b) 4ASK F-OFDM

0 200 400 600 800 100012

16

20

24

RequiredOSNRatBERof10-3

Fiber length (km)

DD; 12 GI

FD; 2 GI

FD; 6 GI

FD; 12 GI

Exp. parameters

(a) BPSK F-OFDM

Fig. 10. Required OSNR versus transmission distance for DDF-OFDM with GI length of 12 (circles) and FD F-OFDM withGI lengths of 2 (triangles), 6 (squares), and 12 (diamonds).(a) and (b) represent 16.8Gbits BPSK and 33.6Gbits 4ASKF-OFDM, respectively. In (a), pluses represent FD F-OFDM withthe GI length of 12 using system parameters the same as those in

the experiment.

0 200 400 60020

24

28

32

Fiber length (km)

RequiredOSNRat

BERof10-3 FD 16QAM OFDM; 12 GI

FD 4ASK F-OFDM; 12 GI

(b)

0 200 400 600 800 100012

16

20

24

Fiber length (km)

RequiredOSNRatBERof10-3 FD QPSK OFDM; 12 GI

FD BPSK F-OFDM; 12 GI

(a)

Fig. 11. Required OSNR versus fiber length for FD F-OFDM andFD conventional OFDM at the same spectral efficiency.(a) 16.8Gbits BPSK F-OFDM and QPSK conventional OFDMand (b) 33.6Gbits 4ASK F-OFDM and 16QAM conventional

OFDM.


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Figure 13 depicts the OSNR penalty versus the band-width of the high-pass electrical filter used for noisesuppression. The OSNR penalty is with respect to theback-to-back receiver sensitivity of FD 4ASK F-OFDMat 103. At back-to-back, the system performance wasinsensitive to the filter bandwidth provided that the

low-frequency noise amplification was suppressed. This isbecause the received signal was real (i.e., the phase wasclose to zero) and any distortion on the reconstructed phasedue to a wider filter bandwidth would not result indegraded performance. At 360 km, as the received signalwas complex after dispersion, optimal filter bandwidthwas required to balance the noise suppression and the dis-tortion arising from the filtered signal phase components.

The optimal filter bandwidth was around 11.5 GHz,matching that in the experiment (Fig. 7). It is also seenthat 16QAM conventional OFDM was more sensitive tothe filter bandwidth, especially when the low-frequencynoise amplification dominated (low filter bandwidth re-gion). This confirms the benefit of optical F-OFDM over

conventional OFDM when FD is applied.In contrast to coherent detection, the proposed scheme

does not require a high-specification laser. Figure14showsthe OSNR penalty versus the laser linewidth. As expected,FD was insensitive to the laser linewidth. Lasers withlinewidth of less than 2 MHz resulted in negligible penalty,indicating the possible use of a cost-effective laser.

-2 -1 0 1 2-2

-1

0

1

2

-2 -1 0 1 2-2

-1

0

1

2

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-2

0

2

4

-4 -2 0 2 4-4

-2

0

2

4

Imaginary

component(a.u.)


(a) BPSK F-OFDM (b) QPSK OFDM



(c) 4ASK F-OFDM (d) 16QAM OFDM


Imaginary

component(a.u.)

Fig. 12. Constellation diagrams of FD F-OFDM and conventional OFDM at 0 km. (a) and (b) 20 dB OSNR. (c) and (d) 25 dB OSNR.

0 0.5 1 1.5 2 2.5

0

2

4

6

8

10

Filter bandwidth (GHz)

OSNRpenaltyat10

-3(

dB)

4ASK F-OFDM; 0 km

4ASK F-OFDM; 360 km

16QAM OFDM; 360 km

Fig. 13. OSNR penalty versus the bandwidth of the high-pass fil-ter for noise suppression. Circles, triangles, and squares representFD 4ASK F-OFDM at 0 and 360 km, and FD 16QAM conventional

OFDM at 360 km, respectively.

10-1

100

101

-2

0

2

4

6

8

10

Laser linewidth (MHz)

4ASK F-OFDM; 0 km

4ASK F-OFDM; 360 km

16QAM OFDM; 360 km

OSNRpenaltyat10-3(

dB)

Fig. 14. OSNR penalty versus the laser linewidth for FD 4ASKF-OFDM at 0 (circles) and 360 km (triangles), and FD 16QAM

conventional OFDM at 360 km (squares).


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We have also investigated the flexibility to choose theAMZI DTD. Figure15illustrates that the optimal range ofthe AMZI DTD is large (20 ps for both 0 and 360 km). Inprinciple [Eqs. (2) an d(3)], a smaller DTD can provide moreaccurate estimation. However, when the DTD is small, thevalue ofVf is small such that the effect of thermal noiseincreases. The optimal DTD should balance the precisionof phase reconstruction and the noise impact. Additionally,it is shown from Fig. 15 that the optimal value alsodepended on the fiber length. In the presence of CD, thesignal was broadened, so the optimal DTD increasedaccordingly.

Figure16further investigates the performance sensitiv-ity to the AMZI differential phase. Ideally, the requiredAMZI phase should be 2. However, this phase cannotbe perfectly obtained in practice and may also drift over

time. The figure shows that, at 2 dB OSNR penalty, the tol-erance range was as wide as35 35and 45 45for 0and 360 km, respectively. This requirement can be readilyachieved by commercial optical devices.

The principle of FD [Eqs. (1)(3)] is based on singlepolarization without polarization mode dispersion (PMD),so the PMD tolerance of the proposed scheme is investi-gated in Fig.17. The figure indicates that, at 1 dB OSNRpenalty with respect to 0 ps differential group delay (DGD),the scheme could tolerate 20 ps DGD at both 0 and 360 km

for 33.6 Gbits 4-ASK F-OFDM. When rescaled to56Gbits, the DGD tolerance would still be larger than10 ps. This is sufficient to most applications that the pro-posed scheme targets, including long-reach 100/400GE,

long-reach access networks (100 km), and short metro net-works (300 km). Note that the proposed scheme may pos-sibly be extended to support two polarizations togetherwith enhanced PMD tolerance, based on recently reportedwork for polarization multiplexed FD QPSK [36].

Finally, although all the experimental and numericalresults are based on the single-channel investigation, webelieve that the proposed scheme can be readily appliedto wavelength division multiplexing configurations. Theconclusions should be still valid unless the channel spacingis close to the symbol rate per channel (commonly calledoptical superchannel). In that case, the channel orthogon-ality in the optical domain should be considered. However,it is beyond the scope of this paper.

V. CONCLUSION

We have proposed for the first time IM FD opticalF-OFDM and investigated its performance advantagesover DD optical F-OFDM and FD conventional OFDM. A16.8Gbits BPSK FD optical F-OFDM system over480 km without optical dispersion compensation has beenexperimentally demonstrated to verify the implementationfeasibility. BPSK/4ASK FD optical F-OFDM systems with

optimized system parameters have been numerically in-vestigated, and compared with BPSK/4ASK DD F-OFDM,and QPSK/16QAM FD conventional OFDM at the samespectral efficiency. It is shown that the proposed scheme,by recovering the full optical field without a coherentreceiver, exhibits greatly improved CD tolerance over DDF-OFDM. This scheme is also more robust to the phasenoise induced in full-field reconstruction, and so shows sig-nificant performance advantages over FD conventionalOFDM. These advantages make the proposed scheme verypromising for applications that demand low cost but higherperformance beyond the conventional IM DD solution, suchas high-speed long-reach 40/100/400G Ethernet, long-reach

access networks, and short metro networks.

0 10 20 30 40 50-1

0

1

2

3

4

AMZI differential time delay (ps)

4ASK F-OFDM; 0 km

4ASK F-OFDM; 360

OSNRpenaltyat10-3(

dB)

Fig. 15. OSNR penalty versus the AMZI DTD for 33.6Gbits FD4ASK F-OFDM at 0 (circles) and 360 km (triangles).

30 60 90 120 150

0

2

4

6

8

AMZI phase (degree)

4ASK F-OFDM; 0 km

4ASK F-OFDM; 360

OSNRpenaltyat10-3(

dB

)

Fig.16. OSNR penalty versus the AMZI phase for33.6Gbits FD

4ASK F-OFDM at 0 (circles) and 360 km (triangles).

0 10 20 30 40

0

2

4

6

DGD (ps)

OSN

Rpenaltyat10-3(

dB) 4ASK F-OFDM; 0 km

4ASK F-OFDM; 360

Fig. 17. OSNR penalty versus DGD for FD 4ASK F-OFDM.


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ACKNOWLEDGMENTS

This work was supported by Science Foundation Irelandunder grant nos. 11/SIRG/I2124 and 06/IN/I969 and theEU Seventh Framework Program under grant agreement318415 (FOX-C).

REFERENCES

[1] J. Zhao and A. D. Ellis, A novel optical fast OFDM withreduced channel spacing equal to half of the symbol rateper carrier, in Optical Fiber Communication Conf., 2010,paper OMR1.

[2] S. K. Ibrahim, J. Zhao, D. Rafique, J. ODowd, and A. D. Ellis,Demonstration of world-first experimental optical fastOFDM system at 7.174Gbits and 14.348Gbits, in

European Conf. Optical Communication, 2010, paper PDP3.4.

[3] N. MacSuibhne, Z. Li, B. Baeuerle, J. Zhao, J. P. Wooler, S. U.Alam, F. Poletti, M. N. Petrovich, A. M. Heidt, I. P. Giles,D. J. Giles, B. Palsdottir, L. Gruner-Nielsen, R. Phelan, J.OCarroll, B. Kelly, D. Murphy, A. D. Ellis, D. J. Richardson,

and F. C. G. Gunning, Wavelength division multiplexing at2 m, in European Conf. Optical Communication, 2012,post-deadline paper Th.3.A.3.

[4] J. Zhao and A. D. Ellis,Advantage of optical fast OFDM overOFDM in residual frequency offset compensation, IEEE

Photon. Technol. Lett., vol. 24, pp. 22842287, 2012.

[5] J. Zhao, S. K. Ibrahim, D. Rafique, P. Gunning, and A. D. Ellis,Symbol synchronization exploiting the symmetric propertyin optical fast OFDM, IEEE Photon. Technol. Lett., vol. 23,pp. 594596, 2011.

[6] J. Zhao and A. D. Ellis, Transmission of 4-ASK optical fastOFDM with chromatic dispersion compensation, IEEE


[7] J. Zhao and H. Shams, Fast dispersion estimation in coher-

ent optical 16QAM fast OFDM systems, Opt. Express, vol.21,pp. 25002505, 2013.

[8] C. Lei, H. Chen, M. Chen, and S. Xie, A high spectral effi-ciency optical OFDM scheme based on interleaved multiplex-ing, Opt. Express, vol. 18, pp. 2614926154, 2010.

[9] C. Lei, H. Chen, M. Chen, and S. Xie,16 10 Gbs symmet-ric WDM-FOFDM-PON realization with colorless ONUs,Opt. Express, vol. 19, pp. 1527515280, 2011.

[10] E. Giacoumidis, S. K. Ibrahim, J. Zhao, J. M. Tang, A. D. Ellis,and I. Tomkos,Experimental demonstration of cost-effectiveintensity-modulation and direct-detection optical fast-OFDMover 40 km SMF transmission, in Proc. Optical Fiber Com-munication Conf. and Expo., 2012, paper JW2A.65.

[11] E. Giacoumidis, I. Tomkos, and J. M. Tang, Performance ofoptical fast-OFDM in MMF-based links, in Optical FiberCommunication Conf., San Diego, CA, 2011, paper OWU3.

[12] E. Giacoumidis, S. K. Ibrahim, J. Zhao, J. M. Tang, A. D. Ellis,and I. Tomkos, Experimental and theoretical investigationsof intensity-modulation and direct-detection optical fast-OFDM over MMF-links,IEEE Photon. Technol. Lett., vol. 24,pp. 5254, 2012.

[13] E. Giacoumidis, A. Tsokanos, C. Mouchos, G. Zardas, C. Alves,J. L. Wei, J. M. Tang, C. Gosset, Y. Jaouen, and I. Tomkos,Extensive comparison of optical fast OFDM and conven-tional OFDM for local and access networks,J. Opt. Commun.

Netw., vol. 4, pp. 724733, 2012.

[14] W. Long, J. Zhang, D. Wang, J. Han, S. Chen, A. M. Han, G.

Gao, H. Leng, L. Liu, W. Zhu, Y. Zhao, and W. Gu, Mitigation

of the interference between odd and even terms in optical fastOFDM scheme based on interleaved multiplexing, IEEE


[15] M. R. D. Rodrigues and I. Darwazeh,Fast OFDM: a proposalfor doubling the data rate of OFDM schemes, in Int. Conf.Telecommunications, 2002, pp. 484487.

[16] F. Xiong, M-ary amplitude shift keying OFDM system,IEEE Trans. Commun., vol. 51, pp. 16381642, 2003.

[17] G. Mandyam, On the discrete cosine transform and OFDMsystems,in Int. Conf. Acoustics, Speech, and Signal Process-ing, 2003, pp. 544547.

[18] Y. Yeh and S. Chen, Efficient channel estimation based ondiscrete cosine transform, in Int. Conf. Acoustics, Speech,and Signal Processing, 2003, pp. 676679.

[19] P. Tan and N. C. Beaulieu, A comparison of DCT-basedOFDM and DFT-based OFDM in frequency offset and fadingchannels, IEEE Trans. Commun., vol. 54, pp. 21132125,2006.

[20] N. Al-Dhahir, H. Minn, and S. Satish, Optimum DCT-basedmulticarrier transceivers for frequency-selective channels,

IEEE Trans. Commun., vol. 54, pp. 911921, 2006.

[21] F. Gao, T. Cui, A. Nallanathan, and C. Tellambura, Maxi-mum likelihood based estimation of frequency and phaseoffset in DCT OFDM systems under non-circular transmis-sions: algorithms, analysis, and comparisons, IEEE Trans.Commun., vol. 56, pp. 14251429, 2008.

[22] R. G. Clegg, S.Isam, I. Kanaras, and I. Darwazeh, A practicalsystem for improved efficiency in frequency division multi-plexed wireless networks,IET Commun., vol.6, pp. 449457,2012.

[23] P. N. Whatmough, M. R. Perrett, S. Isam, and I. Darwazeh,VLSI architecture for a reconfigurable spectrally efficientFDM baseband transmitter, IEEE Trans. Circuits Syst.,

vol. 59, pp. 11071118, 2012.

[24] J. L. Wei, X. Q. Jin, and J. M. Tang, Influence of directly

modulated DFB lasers on the transmission performance ofcarrier-suppressed single-sideband optical OFDM signalsover IMDD SMF systems, J. Lightwave Technol., vol. 27,pp. 24122419, 2009.

[25] J. Xiao, J. Yu, X. Li, Q. Tang, H. Chen, F. Li, Z. Cao, and L.Chen, Hadamard transform combined with compandingtransform techniques for PAPR reduction in an opticaldirect-detection OFDM system, J. Opt. Commun. Netw.,

vol. 4, pp. 709714, 2012.

[26] L. Chen, Y. Qiao, Y. Zhao, and Y. Ji, Wide range frequencyoffset estimation method for a DD-OFDM-PON down-stream system,J. Opt. Commun. Netw., vol. 4, pp. 565570,2012.

[27] E. Giacoumidis, A. Kavatzikidis, A. Tsokanos, J. M. Tang, andI. Tomkos, Adaptive loading algorithms for IMDD opticalOFDM PON systems using directly modulated lasers, J.Opt. Commun. Netw., vol. 4, pp. 769778, 2012.

[28] B. J. C. Schmidt, A. J. Lowery, and J. Armstrong,Experimen-tal demonstrations of electronic dispersion compensation forlong-haul transmission using direct-detection opticalOFDM, J. Lightwave Technol., vol. 26, pp. 196203, 2008.

[29] M. Schuster, S. Randel, C. A. Bunge, S. C. J. Lee, F. Breyer, B.Spinnler, and K. Petermann, Spectrally efficient compatiblesingle-sideband modulation for OFDM transmission withdirect detection, IEEE Photon. Technol. Lett., vol. 20,pp. 670672, 2008.

[30] A. D. Ellis and M. E. McCarthy, Receiver-side electronic

dispersion compensation using passive optical field detection


8/13/2019 06515890

10/10

for low cost 10Gbits 600 km-reach applications, in Proc.Optical Fiber Communication Conf., 2006, paper OTuE4.

[31] N. Kikuchi and S. Sasaki,Highly sensitive optical multileveltransmission of arbitrary quadrature-amplitude modulationsignals with direct detection,J. Lightwave Technol., vol. 28,pp. 123130, 2010.

[32] J. Zhao, M. E. McCarthy, and A. D. Ellis, Electronicdispersion compensation using full optical field recon-

struction in 10Gbits OOK based systems, Opt. Express,vol. 16, pp. 1535315365, 2008.

[33] J. Zhao and A. D. Ellis,Full-field detection based multi-chipMLSE for offset-DQPSK modulation format, in EuropeanConf. Optical Communication, 2011, paper Tu.5.A.2.

[34] X. Liu, S. Chandrasekhar, and A. Leven, Digital self-coherent detection,Opt. Express, vol. 16, pp. 792803, 2008.

[35] I. Tselniker, M. Nazarathy, S. B. Ezra, J. Li, and J. Leuthold,Self-coherent complex field reconstruction with in-phase andquadrature delay detection without a direct-detectionbranch, Opt. Express, vol. 20, pp. 1545215473, 2012.

[36] J. Li, R. Schmogrow, D. Hillerkuss, P. C. Schindler, M.Nazerathy, C. S. Langhorst, S. B. Ezra, I. Tselniker, C. Koos,W. Freude, and J. Leuthold, A self-coherent receiver for de-

tection of PolMUX coherent signals, Opt. Express, vol. 20,pp. 2141321433, 2012.

[37] B. Inan, S. Adhikari, O. Karakaya, P. Kainzmaier, M. Mocker,H. V. Kirchbauer, N. Hanik, and S. L. Jansen, Real-time

93.8Gbs polarization-multiplexed OFDM transmitter with1024-point IFFT, Opt. Express, vol. 19, pp. B64B68, 2011.

Jian Zhao(M08) received a B.Eng. degree from the University ofScience and Technology of China in 2002, and M.Phil. and Ph.D.degrees from the Chinese University of Hong Kong in 2004 and2007, respectively. He joined the Photonic Systems Group at the

Tyndall National Institute, Cork, Ireland, as a PostdoctoralResearcher in August 2007, and is currently a Staff Researcherand Research Fellow. Hiscurrent research interests include opticalOFDM, advanced modulation and detection schemes, communica-tion and information theory, and digital signal processing. Heproposed, and together with his colleagues, demonstrated theworld-first optical F-OFDM at 1550 and 2000 nm optical wave-lengths. He led a team that developed a 10 G real-time electronic-dispersion-compensation integrated receiver prototype supporting900 km transmission and was the runner-up of the Alcatel-Lucent(UK & Ireland) Innovation Competition. Since 2009, he has been aPrincipal Investigator or a Partner with more than 1M fundingsupport from Enterprise Ireland, Science Foundation Ireland, andEU FP7. He has published more than 75 papers (around 50 ofwhich are senior authored) in peer-reviewed journals and confer-ences, and two book chapters. He serves as a reviewer for various

journals (Journal of Lightwave Technology, IEEE PhotonicsTechnology Letters, IEEE Transactions on Vehicular Technology,Optics Express, Optics Letters, Electronics Letters, etc.), andas a session cochair or TPC member for several IEEE/IETinternational conferences. He is a member of IEEE.


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