0071453474 ar037bridge converters

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2.103

CHAPTER 13

BRIDGE CONVERTERS

13.1 INTRODUCTION

The full-bridge push-pull converter requires four power transistors and extra drivecomponents. This tends to make it more expensive than the flyback or half-bridgeconverter, and so it is normally reserved for higher-power applications.

The technique has a number of useful features; in particular, a single primarywinding is required on the main transformer, and this is driven to the full supplyvoltage in both directions. This, together with full-wave output rectification,provides an excellent utility factor for the transformer core and windings, andhighly efficient transformer designs are possible.

A second advantage is that the power switches operate under extremely welldefined conditions. The maximum stress voltage will not exceed the supply linevoltage under any conditions. Positive clamping by four energy recovery diodeseliminates any voltage transients that normally would have been generated by theleakage inductances.

To its disadvantage, four switching transistors are required, and since twotransistors operate in series, the effective saturated on-state power loss issomewhat greater than in the two-transistor push-pull case. However, in high-voltage off-line switching systems, these losses are acceptably small.

Finally, the topology provides flyback energy recovery via the four recovery

diodes without needing an energy recovery winding.

13.2 OPERATING PRINCIPLES

13.2.1 General Conditions

Figure 2.13.1 shows the power section of a typical off-line bridge converter. Diagonalpairs of switching devices are operated simultaneously and in alternate sequence. For

example, Q1 and Q3 would both be on at the same time, followed by Q2 and Q4.In a pulse-width-controlled system, there will be a period when all four devices willbe off. It should be noted that when Q2 and Q4 are on, the voltage across theprimary winding has been reversed from that when Q1 and Q3 were on.

In this example, a proportional base drive circuit has been used; this makes thebase drive current proportional to the collector current at all times. This technique is

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Source: SWITCHMODE POWER SUPPLY HANDBOOK


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2.104

FIG.

2.1

3.1

Fu

ll-bridgeforwardpush-pullcon

verter,showinginrushlimiting

circuitandinputfilter.



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2.10513. BRIDGE CONVERTERS

particularly suitable for high-power applications and is more fully described in Part1, Chap. 16.

During the off period, under steady-state conditions, a current will have been

established in L1, and the output rectifier diodes D5 and D6 will be acting asflywheel diodes. Under the forcing action of L1, both diodes conduct an equal shareof the inductor current during this off period (except for a small magnetizingcurrent). Provided that balanced diodes are used, the voltage across the secondarywindings will be zero, and hence the primary voltage will also be zero (after a shortperiod of damped oscillatory conditions caused by the primary leakage inductance).Typical collector voltage waveforms are shown in Fig. 2.13.2.

FIG. 2.13.2 Primary voltage and current waveforms for full-bridge converter.



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2.106 PART 2

13.2.2 Cycle of Operations

Consider a cycle of operation under steady-state conditions in Fig. 2.13.1. Assumethat the drive circuit initiates a turn-on pulse for Q1 and Q3. These two deviceswill start to turn on. Collector current will now flow via the primary winding ofT1P through the primary of the proportional drive transformers T2A and T2B. Bypositive regenerative feedback, the turn-on action of the two transistors is en-hanced, and this results in rapid switching to the fully on state, with Q1 and Q3fully saturated.

As soon as Q1 and Q3 turn on, current will start to build up in the primarywinding of T1P at a rate defined by the primary leakage inductance. This current ismade up of the reflected load current and a small proportion of magnetizing currentfor the transformer magnetic field, as shown in Fig. 2.13.2.

Simultaneously, during the turn-on edge, the current in the secondary rectifier

diode D5 will increase and that in D6 will decrease at a rate defined by the totalsecondary leakage inductance and the external loop wiring inductance through D5and D6. For low-voltage, high-current outputs, the external loop wiring inductancecan be the predominant effect. When the secondary current has increased to thevalue which was flowing in L1 prior to the switch-on of Q1 and Q3, D6 will becomereverse-biased, and the voltage at the input of L1 will now increase to the secondaryvoltage Vs less the drop in diode D5.

The voltage across L1 will be (Vs-Vout) in the forward direction, and the current inL1 will ramp up during this period. This current is transferred to the primary asshown in Fig. 2.13.2.

After an on period defined by the control circuit, the base drive current will be

diverted away from the power transistors by the drive transformer, and Q1 and Q3will turn off. However, a magnetizing current has now been established in thetransformer primary, and leakage inductances and the ampere-turns will remainconstant, the current transferring to the secondary. Therefore, by flyback action, thevoltages on all windings will reverse. If sufficient energy has been stored in theleakage inductances, the primary voltage will fly back to a point at which diodes D2and D4 conduct, and the excess flyback energy will be returned to the supply lines.If the leakage inductance is very low, the snubber capacitor C5, R5, and the outputdiodes D5 and D6 will provide effective clamping. D5 and D6 will divert themajority of the flyback energy to the output. Because of the hard clamping action ofprimary diodes D1 through D4 and secondary diodes D5 and D6, the voltage across

the switching transistors cannot exceed the supply line voltage by more than a diodedrop at any time.

13.2.3 Snubber Components

During the turn-off transient, the snubber components R5, C5 will reduce the turn-off stresses on the power devices by providing an alternative path for the collectorcurrent during the turn-off transient. It is possible to replace the four snubbernetworks by a single RC network across the transformer primary, but better

common-mode control of the primary voltage is given by the arrangement shownwhen all power transistors are off. This action is more fully described in Part 1,Chap. 17.

The flywheel action provided by the output diodes is an important feature ofthis type of push-pull circuit. Figure 2.9.2a and b shows the working range for the



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core flux swing in both push-pull and single-ended operation. The range is muchwider in the push-pull case, as the core will not restore to zero, even when alltransistors are turned off, because of the flywheel conduction of D5 and D6 as

follows.Because D5 and D6 remain conducting during the off period, the voltageacross the secondary, and hence across all windings, is zero when the switchingtransistors are turned off. As a result, the core will not restore to B

rduring the off

period but will be held at +Bor B. Hence, when the following diagonal pair ofinput transistors are turned on, the full flux density range of 2B (from Bto +B)isavailable for use, allowing the transformer to be designed for a lower number ofprimary turns. The secondary voltage waveform is shown in Fig. 2.13.2.

This secondary diode clamping effect is lost when the load falls below themagnetization current (as referred to the secondary). However, this would notnormally be a problem, as under these conditions the on pulse would be very

short and B small.

13.2.4 Transient Flux Doubling Effect

Under transient loading conditions, a problem can sometimes occur if the full rangeof the B/Hcharacteristic is used. If the supply has been running under light load, thepulse width will be narrow, and the core will be working near B=0. If a suddenincrease in load drives the unit to full pulse width, only half the range ofB is nowavailable for this transient change, and the core may saturate. Be careful to consider

transient conditions and either allow sufficient flux density margin to cope with thiscondition or limit the control slew rate to allow the core to establish a new workingcondition. (This effect is sometimes referred to as flux doubling.)

13.3 TRANSFORMER DESIGN (FULL BRIDGE)

The design approach for the push-pull transformer is relatively straightforward. The

single primary winding is used for both half cycles of operation, providing extremelygood utility of core and windings.To minimize magnetization currents, the maximum primary inductance

consistent with minimum turns is required. Consequently, a high-permeabilitymaterial will be selected, and the core will not be gapped. (A small core gap issometimes introduced if there is a chance of a DC current component in thetransformer, as the onset of saturation is more controllable with a core gap.)

13.4 TRANSFORMER DESIGN EXAMPLE

Assume that a transformer is to be designed on a ferrite core to meet the followingrequirements:



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2.108 PART 2

Input voltage 90137 or 180264 (by link change)Frequency 40 kHzOutput power 500 W

Output voltage 5 VOutput current 100 A

13.4.1 Step 1, Select Core Size

Assume an initial efficiency for the transformer and secondary rectifier circuit of75%. The transmitted power for the transformer will then be 500/0.75=666 W.

From Fig. 2.13.3, for push-pull operation at this power level, an EE 555521

FIG. 2.13.3 Core selection chart for balanced push-pull operation, showing throughputpower as a function of frequency with core size as a parameter. (Courtesy of Mullard Ltd.)



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core is indicated for a temperature rise of 40C under convected air-coolingconditions. Hence this core will be used in the following example.

13.4.2 Step 2, Select Optimum Flux Density

For push-pull operation, the full B/Hloop may be used. (See Fig. 2.9.2.) A large fluxdensity excursion gives fewer primary turns and lower copper losses, but increasedcore losses.

Normally, it is assumed that minimum loss (maximum efficiency) will be foundnear the point where the copper and core losses are equal, and this would be thenormal design aim in the selection of working flux density.

Figure 2.13.4 shows how core losses increase for A16 ferrite core material as thenumber of turns is reduced and the peak flux density is increased from 25 to 200 m

T, (At the same time, of course, the copper losses will decrease, but this is notshown here.)

Figure 2.13.5 shows, for a pair of EE555521 cores in A16 ferrite at 40 kHz,how the core, copper, and total losses change as the number of turns is changed andthe peak flux density is increased toward 200 mT. It will be seen that a minimumtotal loss occurs near 70 mT. (For each turns example, optimum use of the corewindow area and wire gauge is assumed.)

FIG. 2.13.4 Core loss per gram of A16 ferrite as a function of fre-quency, with peak flux density as a parameter. (Note: Graph isplotted for peak flux density B ; flux density sweep B is 2B.)(Courtesy of Mullard Ltd.)



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2.110 PART 2

In this example, the minimum loss (maximum efficiency) occurs when the core lossis 44% of the total loss at 70 mT. However, the minimum loss condition has a relativelywide base, and the choice of peak flux density for optimum efficiency is not critical inthe 50- to 100-mT range. The normal assumed optimum choice (where the copper andcore losses are equal) would be 80 mT, which is not very far from optimum.

For each design there is an optimal flux density swing, depending upon theoperating frequency, the core loss, the topology, and the winding utilization factors.Figure 2.13.6a, b, and c shows the manufacturers peak and optimum flux

density recommendations for optimum transformer designs using the EE555521and other cores in forward and push-pull applications. From Fig. 2.13.6c (at 40kHz), the manufacturers recommended peak flux density is 100 mT, which is notfar from optimum, and this higher value will be used in this example to reduce thenumber of turns.

13.4.3 Step 3, Calculate Primary Voltage (Vcc)

As the peak flux density was chosen for near optimum efficiency and is well clear ofsaturation, the design approach used here will be to calculate the primary turns forthe maximum on period (50% duty ratio) with the input voltage at minimum.This will occur at the minimum line input of 90 V rms, and the DC voltage for thevoltage doubler connection will be

FIG. 2.13.5 A16 ferrite core loss, copper loss, and total loss for a pair of EE55/55/21 cores,when wound for optimum performance in a typical switchmode transformer. Loss is shown asa function of the peak flux density. Note that the minimum total loss occurs when the trans-former induction (turns) is optimized so that the core loss is 44% of the total loss.



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2.111

FIG. 2.13.6 (a) Magnetization curve for N27 ferrite material at 25C and 100C.(Courtesy of Siemens AG.) (b), (c) Optimum peak flux density as a function offrequency, with core size as a parameter. (Courtesy of Mullard Ltd.)



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2.112 PART 2

where Vin=ac input voltage, rms

Note: The derivation of the DC voltage and ripple component is more fully coveredin Part 1, Chap. 6. Hence, at 90 V input, using the voltage doubler connection,

13.4.4 Step 4, Calculate Maximum On Period

If cross conduction (two series transistors on at the same time) is to be avoided,

then the maximum on time cannot exceed 50% of the total period.Hence

At 40 kHz,

13.4.5 Step 5, Calculate Primary Turns

The voltage waveform applied to the transformer primary during an on period isrectangular, and the turns can be calculated using the volt-seconds (Faradays law)approach.

In the push-pull transformer, both quadrants of the BHloop are used, and theflux density swing, under steady-state balanced conditions, will go from -B to +B fora positive half cycle.

It should be noted that Fig. 2.13.4 shows the peak flux density B, but assumeslosses for a peak-to-peak swing . For optimum efficiency, B wasselected at 100 mT. Therefore, the peak-to-peak change (flux density swing)

, or=200 mT for this example.The core area for the EE555521 is 354 mm2, and the primary turns may now

be calculated as follows:

where Vcc = minimum DC header voltageton = maximum on period, sB = total flux density swing, TAm = minimum pole area, mm2



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Hence

13.4.6 Step 6, Calculate Secondary Turns

When the bridge converter is operating at full conduction angle (maximum output),the primary waveform tends to a square wave. Consequently, the rectified outputtends to DC, and the output voltage will be the secondary voltage less rectifier, choke,and wiring losses.

Allowing 1 V for all losses, the transformer secondary voltage Vs will be 6 V.Therefore, the turns for each half of the secondary winding will be

Note: The secondary is normalized to one turn and the primary to 37 turns, giving apeak flux density slightly higher than 100 mT in this example.

The secondary voltage Vs used for this calculation is the voltage produced at theminimum line input of 90 V. At this voltage the pulse width is at maximum. At

higher input voltages, the pulse width will be reduced by the control circuit tomaintain output voltage regulation.To minimize the copper losses and leakage inductance, it is important to choose

the optimum gauge and shape of transformer wire and to arrange the makeup forminimum leakage inductance between primary and secondary windings. A split-layer winding technique will be used in this example.

The high-current secondary winding should use a copper strip spanning the fullwidth of the bobbin (less creepage distance). Suitable winding techniques andmethods of optimizing wire shapes and gauges are more fully covered in Part 3,Chap. 4.

13.5 STAIRCASE SATURATION

There will, inevitably, be some imbalance in the forward and reverse volt-secondsconditions applied to the transformer. This may be caused by differences in storagetimes in the transistors or by some imbalance in the forward voltage of the outputrectifier diodes. However it is caused, this imbalance results in the flux density in thetransformer core staircasing toward saturation with each cycle of operation.

Restoration of the core during the off period cannot occur, because during thisperiod the secondary is effectively short-circuited by the clamping action of diodesD5 and D6, which will both be conducting, under the forcing action of L1 (providedthat the load is above the critical current value for L1).

When the core reaches saturation, there is a compensation effect, as thetransistors which are conducting the higher current on the saturating cycle will havetheir storage times reduced, and so a measure of balance will be restored. However,



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2.114 PART 2

a problem still exists for transient operation, and this is discussed in the followingsection.

13.6 TRANSIENT SATURATION EFFECTS

Assume that the power supply has been operating for a period under light loadingconditions, staircase saturation has occurred, and one pair of transistors is operatingnear the saturated point. If a transient increase in load is now applied, the controlcircuit will demand a rapid increase in pulse width to compensate for losses and toincrease the current flow in L1. The core will immediately saturate in one direction,and one pair of transistors will take an excessive current, with possible catastrophicresults.

If the power transistors have independent fast-acting current limits, then theon pulse will be terminated before excessive current can flow, and failure of thepower devices can be avoided. This is not an ideal solution, since the transientresponse will now be degraded.

Alternatively, the slew rate of the control amplifier can be reduced so that theincrease in pulse width is, say, less than 0.2 s on each cycle. Under these conditions,the storage self-compensation effect of the power transistors will normally be able toprevent excessive saturation. However, the transient response will be very muchdegraded, once again. Nevertheless, these two techniques are commonly used.

13.7 FORCED FLUX DENSITY BALANCING

A much better solution to the staircase saturation problem may be applied to thepush-pull bridge circuit shown in Fig. 2.13.1.

If two identical current transformers are fitted in the emitters of Q3 and Q4, thepeak values of the currents flowing in alternate pairs of transistors (and, therefore,in the primary winding) can be compared alternately on each half cycle.

If any unbalance in the two currents is detected, it acts upon the rampcomparator to adjust differentially the width of the drive pulses to the power

transistors. This can maintain the transformers average working flux density nearthe center of the B/Hcharacteristic, detecting any DC offset and adjusting the drivepulses differentially to maintain balance.

It should be noted that this technique can work only if there is a DC paththrough the transformer winding. A capacitor is sometimes fitted in series with theprimary winding to block any DC current; DC transformer saturation is thusavoided. However, under unbalanced conditions this capacitor will take up a netcharge, and so alternate primary voltage pulses will not have the same voltageamplitude. This results in a loss of efficiency and subharmonic ripple in the outputfilter; further, maintaining balanced transformer currents and maintaining capacitorcharge are divergent requirements, leading to runaway condition. Therefore, this

DC blocking arrangement using a capacitor is not recommended.The series capacitor Cx must notbe fitted if the forced current balancing system is

used, as it will eliminate the detectable DC component, and the circuit will not beable to operate. This is more fully explained in Part 3, Sec. 6.3.



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If the transformers working point can be maintained close to its center point, fulladvantage can be taken of the working flux density range, giving improved transientcapability without the possibility of transformer saturation and power device

failure.When current-mode control is used for the primary pulse-width modulation, fluxbalancing happens automatically, provided Cx is not fitted. See Part 3, Chap. 10.

13.8 PROBLEMS

1. Why is the full-bridge converter usually reserved for high-power applications?2. What is the major advantage of the full-bridge converter?3. Why is the proportional drive circuit favored in the bridge converter?

4. Why is it particularly important to prevent staircase saturation in the fulgeconverter?5. What measures are usually taken to prevent staircase saturation in the bridge

converter

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