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PRINCIPLES OF QUASI- AND MULTI-RESONANT POWER CONVERSION TECHNIQUES

Wojciech A . Tabisz and F r e d C . L e e

Vi r g i n i a P o w e r E l e c t r o n i c s C e n t e rT h e B r a d l e y D e p a r t m e n t of E l e c t r i c a l E n g i n e e r i n gVi r g i n i a P o l y t e c h n i c I n s t i t u t e a n d S t a t e U n i v e r s i t y

B l a c k s b u r g , Vi r g i n i a 24061

ABSTRACT

Recent ly developed quasi - resonant and mul t i - resonantdc-dc po wer conv ers ion techniques are reviewed. Quasi -resonant conver ter QRC) and mul t i - resonant conver ter MRC)topologies are der ived f rom the con vent ional square-wave,pulse-width modulated PWM) conver ters by adding resonantcompon ents to achieve zero-current sw i tching ZCS) or zero-voltage switching ZVS) of the semic ondu ctor devices. ZCS andZVS reduce swi tching losses caused by the paras i t ic leakage

inductance of the pow er t ransformer and junct ion capaci tancesof s emiconductor devices . A s a result, QRCs and MRCs canoperate a t h igher sw i tching f requencies and achieve higherpower densi ty than PWM conver ters . Explanation of operat ionof zero-current-switched and zero-voltage-switched QRCs andMRCs is provided, and thei r re lat ive advantages and l imi ta t ionsare d iscussed.

1 INTRODUCTION

Most of present ly used swi tching dc/dc power conver ters usethe pulse-width modulat ion (PWM) technique. High eff ic iency,design simplicity, and ava ilability o f standard integrated PWMcontrol c i rcui ts have a l l contr ibuted to the popular i ty of t t i istechnique. PWM converters have been widely used in everyfacet o f e lec t ronics indust ries , such as comp uters , communi-cation, and transporta tion. The continuous evolutiori of the vel-ylarge-scale integration (VLSI) ci rcui t technology has resul ted inmodern computer and communicat ion equipment wi th a vast

amount of data s torage and high process ing capabi l it ies , and yetever shrinking size and weight. Switching pow er supply (SPS)technology must be compat ible wi th th is technology t rend.

High operat ing f requency is the key to reducing the s ize andweight , and increas ing the performance of SPS equipmentHigher swi tching f requency a l lows s ize reduct ion of the powert ransformer and the energy s torage compo nents ( inductors andcapacitors). In addition, the quality of the pow er supplied to I heload can be improved , s ince higher sw i tching f requency a l lowshigher con rol - loop bandwidth and bet ter d yn a m c r e g u at on.

Switching frequencies of PWM converters are restricted byperformance l imi ta t ions of semiconductor and magnet ic com po-nents. The major limiting factoi-s are switching losses and os-ci l la t ions caused by the paras i t ic reactances of po wer s tage .Output capaci tance o f the po wer MOSFET and junct ioncapaci tance of the Schot tky diode are the tw o m ajor paras i t iccapaci tances . The m ajor source of paras i t ic inductance is theleakage inductance of the powe r t ransformer.

The detrimental effects of the parasitic reactances can be ex-pla ined us ing a s imple buck conver ter, shown with the paras i t iccomponents and i t s waveforms in Figure 1 . Switching MOSFET,S , is periodically on and off, resulting in a square-wave voltageappl ied to the low-pass filter LF CF. The dc output voltage isregula ted by changing the duty ra t io of S. The paras i t ic compo-nents are represented in Fig. 1 by C O ,C and Llh. Diode Ds r e -presents the body diode of the MOSFET.

Ideally, the switch current and voltage l s and vs , and thecurrent a+ vol tage of the rec t i f ier, I D and vD,should be squarewaves , peak values equal to lo and V,,, respectively Inpract ice , how ever, the waveforms are severely dis tor ted byparas i t ic osci l la t ions induced in the resonant c i rcui ts formed byL/k CO and C

At to S is turned-off, and the energy stored in L,I, induces avoltage spike across the MOSFET and the subsequent oscil-la t ions in the c i rcui t formed by and O The uncontrolled volt-age spike across the MOSFET may cause a breakdown of thedevice while the oscillations are a source of high-frequency

noise Typically thes e oscillatio ns decay completely before S isis turned on, and the ent i re energy s tored in Llkpr ior to to isdiss ipated

During the off-time to - , S blocks the input voltage, andcer ta in amount of energy is s tored in O When S i s turned onat tl O s rapidly discharged f rom VI, to 0 while C, is chargedf rom 0 to VI,,, There is energy loss associated with dischargingCO and charging C determined by values of CO and C and themag nitude of the switched voltage Additional turn-on loss in theMOSFET is caused by the finite switc hing speed of the MOSFETwhich resul ts in over lap of dra in current and vol tage I] heswitching power diss ipat ion increases in propor t ion to theswitching f requency A s a result PWM converters cannot oper-ate efficiently a t elevated frequencies

+ vs

VGs

IS

I I I

I I's I .

i-Y

4 N

I l0 I t 1

Fig. 1. PWM buck converter an d its waveforms.

CH 0064/91/0000 1053 1.00 0 I

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LF4 LRNY h

(a) (b)

Fig. 4. Quasi-resonant switc hing cells: (a) ZCS. (b) AIS.

In practice, this is not possible due to the presence of C j . In-stead, a series-resonant circuit is formed by LR and C,, and diodevoltage is oscillatory with a peak value equal to 2VlN. I f the os-cillation decays prior to turning-off of the MOSFET, the energyassociated with charging of C will be lost, just as in a PWMconverter. If the oscillation does not decay before the MOSFETis turned off, the conversion-ratio characteristics will be ad-versely affected, resulting in difficulties in co ntrolling the con-verter [6]

An important drawback of ZVS-QRCs is an extensive voltagestress on the switching transistor. This stress is proportional tothe load range. A s a result, ZVS-QRCs are not suitable for ap-plications with wide load variations.

The families of ZCS- and ZVS-QRCs contain a large numberof topologies, including the basic topologies: buck, boost, buck-boost, and numerous isolated topologies, both bridge-type andsingle-ended. All topologies of ZCS-QRCs, however, share acommon structure of switching network, shown in Fig. 4(a).Likewise, all topologies of ZVS-QRCs share the s witching struc-ture of Fig. 4(b). The structures shown in Fig. 4, are called res-onant swi tch cells and are extracted from QRCs by replacingvoltage sources and filter c apacitors with short circuits, and filterinductors with open circuits.

It can be observed in Fig. 4 that in ZCS-QRCs, the active switchS s in series with the resonant inductor, while the diode is inparallel with the res onant capacitor. In ZVS-QRCs, the activeswitch is in parallel with the capacitor, and the diode is in serieswith the inductor. The arrangement of the resonant componentswith respect to the switching devices determines wh ich parasiticreactances will be absorbed by the resonant circuit. A s a result,ZCS-QRCs are insensitive to the transformer leak age inductanceand the junction cap acitance of the rectifier diode, whileZVS-QRCs are insensitive to transformer leakage inductanceand the MOSFET output capacitance. It is clear, however, thatneither ZCS-QRCs nor ZVS-QRCs are capable of utilizing a//major parasitic reactances.

IV. MULTI-RESONANT TECHNIQUE

The underlying principle of the multi-resonant technique is toutilize all essential parasitic elements in a conve rter circuit. Toachieve this objective it necessary to employ a mul t i -elementresonant network with no less than three energy storage com-ponents. Figure 5 shows the ZCS and ZVS multi-resonant switchcells. Multi-resonant converters (MRCs) [5-91 are generated byincorporating these switching networks into basic PWMtopologies. In ZCS-MRCs, the resonan t circuit is formed in aT-network, with reso nant inductors in series with the s witchingdevices, as shown in Fig. 5(a). In ZVS-MRCs, the res onant circuiti s formed in a U-netw ork with resonant capacitors connected inparallel with the switches, as shown i n Fig. 5(b). During one cy-c le of operation of an MRC, three different reso nant circuits canbe formed, depending on whether the active switch and diodeare open or closed. This results in operation of the converterwilh three different resonant stages in one cycle of operation(hence the term "multi-resonant").

Figure 6 shows a circuit diagram of a buck ZCS-MRC. Reso-nant inductors Ls and L D are formed by the inherent parasiticinductances, LIX.and LIX.,, nd the external inductances, Lsx andLo,. The multiple resonances result in zero-current sw itching ofboth the MOSFET and the rectifying diode. For optimal ZCS-MRCoperation, the parasitic capacitances, CO nd C,, ideally shouldbe zero. Howeve r, in practical higli-frequency applications,

S ) ? T I D s )FtnD(a) b)

Fig. 5. Multi-resonantswitc hing cells: (a) ZCS. b) ZVS.

these capacitances cannot be reduced to negligible values. A sa result, parasitic oscillations are present in the waveforms.

The ZCS-MRC technique is not suitable for high-frequencyapplications since it cannot accomm odate the parasiticcapacitances of the semicondu ctor devices. It could be useful,however, in high-power high-current converters using SCRs orGTOs, where inductive turn-off is of primary conc ern.

The ZVS-MRC technique is much more suitable for h igh-frequency operation. Figure 6 shows buck ZVS-MRC and itswaveform s. The resonant network consists of the resonant ele-ments LR Cs, and Co. Each of the resonant elements is formedby a combination of the externally added components L R ~ ,s X ,and C D x and the parasitic elements Llk, CO,and Cj). If the con-verter is operated at a sufficiently high switching frequency, theresonant elements can be formed exclusively by the parasiticcomponents i . e . , LR,= 0, C s x = 0, and CD, = 0).

Prior to to, he MOSFET conducts, and a res onant circuit isformed by LR and CD. Current s and voltage vD are sinusoidal.When the MOSFET is turned off at to a resonant circuit is formedby LR,CD nd Cs. During the resonance, VD falls to zero, andconsequently, diode D turns on at t , . This results in yet anotherresonant circuit formed by LR and Cs. Resonanc e in this circultreduces vs to zero, atlowing the MOSFET to turn on at t2 withoutturn-on switching loss. Subsequently, both S and D are on, VINis applied to LR, and iLR increases linearly. As a result, iD de-creases linearly until i t drops to zero at ts forcing diode D toturn off. At this instance, a resonant circuit is formed by LR andCo. This resonant stage is terminated at f when the MOSFETis turned off.

+ vs

1 1 - I

vGs

s

VD

I I 1 It

' 0 t l t f3 14

Fig. 6. Buck zero-current-switchedmulti-resonant converter.

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111 QUASI RESONANT TECHNIQUE

The quasi-resonant technique [2-51 i s a ge neral method forimproving swi tching waveforms of dc/dc conver ters in the pres-ence of significant parasitic reactances. This is achiev ed bymeans of a resonant c i rcui t added to a bas ic PWM conver tertopology. When a quasi-resonant topology is generated, thebasic topological character is t ics of i t s "parent" PWM topoloyyrema in unaltered, i.e., the top ologica l interrelations of the non-l inear components (ac t ive and pass ive swi tches) and the low-pass filter components in a resulting QRC topology are the same

as in the or ig inal PWM topology. There are two famil ies ofQRCs: zero-c urrent-s witch ed (ZCS) QRCs and zero-voltage-switched (ZVS) QFC:; each family is character ized by a uniquearrangement of the resonant components .

T'le basic operation al principle s of ZCS-QRCs can be ex-pla ins4 us ing the buck topology shown in Fig. 2. This topologyis generated f rom a PWM buck topology by adding an externalresonant inductor, LRx, n series with the MOSFET. and an ex-ternal resonant capacitor, C D x , n para l le l wi th the diode. Theeffective resonant inductance, LR, i s the sum of LRx and L l k .Similar ly, the effect ive resonant c-paci tance C D i s t he sum o fCox and C For s impl ic i ty, the low-pass output f i l ter and loadres istance are represented by a constant current source , lo.

When the MOSFET is off to - , ,output current i s f reewheel ingthrough D . Ideally, the voltage across the MOSFET would beequal to VI, dur ing th is t ime, as indicated by the dashed l ine . npract ice , however, the output capacitance o f the MOSFET form.a parasitic resonant circuit with LR,and paras i t ic osci l la t ions areinduced in vs .

At t , , he MOSFET is turned on. Since diode D continues toconduct the load current , V ~ Ns appl ied di rect ly to the resonantinductance. Subsequent ly, the current through LR increases l in-ear ly, an d the load current gradual ly shi f ts f rom the f reewheel ingdiode to the MOSFET. At t2 , D reaches zero , forc ing D to turn off.Consequently, CD and L R form a resonant circuit. If the charac-ter is t ic impedance of th is c i rcui t , ZD = a s chosen so thatV/,/Z, 2 lo, the resonan ce wi l l force the act ive swi tch current tozero. The MOSFET is then turned off at f O with zero-currentcondition, and the turn-off switching l o s s associated wi th over-lapping of the dra in current and vol tage is pract ica l ly e l iminated.Furthermore, the energy stored in LR at turn-off is zero, and thelosses and oscillations associated with an inductive turn-off arealso e l iminated.

The resonant frequency of LR and C D dictates the duration ofthe on-time A s a result, ZCS-QRCs operat e with a fixed on-tim eand the output vol tage is regula ted by varying the off -t ime (var i -able frequency control)

The virtual elimination of turn-off losses is an obvious advan-tage of ZCS-QRCs, when compared to PWM converters Theover lap of MOSFET drain vol tage and c urrent i s e l iminated andno leakage-inductance energ y is dissipa ted at turn off In addi-tion, the junction capacitance of the rectifying diode is chargedand discharged in a resonant fashion, thus , there i s no energydiss ipat ion re la ted to charging of C ZCS-QRCs have the a bility

to ut i l ize the paras i t ic leakage inductance of the t ransformer andthe junct ion capaci tance of the rec t i f ier It can be said thatZCS-QRCs are InsensItwe to L l k and C because these parasiticcomponents are abs orbed by the resonant components LR andCO, and do not cause any detrimental effects

Although the ZCS-QRC technique dramatically improvesswitching waveforms and e l iminates turn-off losses in the po werMOSFET and losses associa ted wi th charging of C,, it fails to al-levia te the problem o f the turn-on diss ipat ion of the energystored in the MOSFET output capacitance

V e p rob l em o f c apac i t i i - l o s se s a t t urn -on i s so lved i nZVS-QRCS by ' ice the drain to-source voltage to - ro pr ior O turn-on of the MDSFET Figure 3shows a buck ZVS-QRC This topology is derived fiom its PWIAcounterpart by adding an external resonant capacitor C s x inparallel with the MOSFET an d an external resonant inductor,Lq, in ser ies wi th Llh The equivalent resonant capacitance, CSi s formed by CO and C s x while the equivalent resonantinductance, LR IS formed by L l k and L R ~

Piior to to the MOSFET is on, and the load current flows

through LR At to the MOSFET is turned off , and the lsad current19 diver ted in to C S Subsequently, vs increases l inear ly, whi leI/ decreases linearly

At t , , vD s reduced to ze ro and D turns on This starts a res-onance in the LR - CS ci rcui t I f the character is t ic impedanceZS = is chosen so that l o Z s 2 V IN ,the resonance willf o i ce vs to ze ro A s a result the MOSFET can be turned on at17 unde r a zero-voltage cond ition, thus eliminating the capacitiveturn-on loss

During the inductor-charging stage f 2 3 .both S and D areo n A s a resul t , VI, i s appl ied to LR, and ILR increases l inear lyA t f 3 I ~ R eaches the magni tude of lo and diode D turns offUndei ideal conditions C, = 0), vD would instantaneously in-c r ea se f rom ze ro to V),,,, as indicated by the dashed waveform

1n a resonant ci--uit to

+ V r

v G s I

vs

i D

I 1 I 1 I

t 0 t l t t 3 t 4

Fig. 2. Buck zero-current-switched quasi-resonant converter.

tot1 t 2 t 3 t 4 'Fig. 3. Buck zero-voltage-switched quasi-resonant converter

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t vs -

Fig. 7. Buck zero-voltage-switched multi-resonant converter.

The ZVS-MRC technology is very practical for high-frequencyconverter applications. The resonant circuit in ZVS-MRCs isdesigned to absorb all the major paras i t ic components : leakageinductance of the power t ransformer, output capaci tance of thepower MOSFET, and junction capacitance of the rectifier.Single-ended ZVS-MRCs also have a m uch reduced voltagestress on the switching transistor when compared to that of

An example of successful application of the multi-resonantconcept is the forward ZVS-MRC [8]. shown in Fig. 8. In the for-ward ZVS-MRC, as in any other ZVS-MRC, the re sonant com po-nents are placed in the circuit in such a m anne r that the paras itic

compon ents are absorbed in the resonant circuit. The prima ry-side resonant capacitance C s x s in para l le l wi th the outputcapacitance of the MOSFET. The resonant inductan ce is formedin part by an external inductance, L R x and in par t by the pr imaryand secondary leakage inductances of the transformer. Thesecondary-side resonant capacitance is formed in part by theexternal capacitance, CD and in part by the junctioncapacitances of the rectifying diodes.

ZVS-QRCS.

C i i

Fig. 0. Forward ZVS-MRC.

Due to the presence of D nd the resonant operation, voltageacross C D ~an be e i ther posi t ive or negat ive, thus an automat iccore resetting mechanism is provided without resorting to t h euse of any additional windings or components.

A hybridized prototype of a board-mount 50 W forwardZVS-MRC was designed for VHSIC applications [SI. The con-ver ter operated at 3 MHz wit h an inpu t voltage of 50 V and outputvoltage of 5 V. The complete packaged converter had powerdensity of 50 W/in3

VI. CONCLUSIONSThe switching losses and parasitic oscillations found in PWM

topologies can be minimized us ing the quasi - resonant andmulti-resonant conversion techniques. QRCs use a two-elementresonant circuit to achieve e i ther zero-current o r zero-voltageswitching of the pow er MOSFET. The ZCS-QRC technique mini-mize s turn-off losses in the pow er MOSFET. Howeve r, it does notreduce the turn-on losses caused by the discharge of theMOSFET's output capacitance. In ZVS-QRCs, these capac itiveturn-on losses are minimized, but the single-ended ZVS-QRCtopologies have excessive voltage stress on the switching tran-sistor. Also, the junction capacitance of the rectifying diode isnot absorbed int o the m ain resonant circuit, and consequen tly,causes severe parasitic oscillations and difficulty in control.

The multi-resonant conve rsion technique uses a three-elementresonant circuit to achieve ZCS or ZVS o f the semiconductordevices. The ZCS-MRC technique does not have a potential forvery high-frequency operation; however, ZCS-MRC topologies[night find application in high-power high-current circuits usingminority-carrier de vices such as BJT, SRC or GTO.

ZVS-MRCs are capa ble of operating at tens of megahertz, andprovide a means of further reduction of the size of power con-verters in the future. it will be a particularly pow erful techniquewhen some other technological limitations, particularly in thearea of high -frey enc y magnet ic materia ls , are overcorne.

REFERENCES

I I j M chlecht, L. Casey, 'Com pariso n of the square-wave andquasi-resonant topologies," IEEE Trans. Power Electronics,Vol. 3, no. 1, pp. 83-92, Jan. 1988.

[ % ] E. Buchanan, E. J. Miller , "Resonant s witching po wer con-version technique," IEEE Power Electronics Specialists Conf.Rec ord, p p. 188-193, 1975.

[3] K.Liu, F.C. Lee, "Resonant switches - A unified approach toimpro ve performance o f switching converters, ' ' IEEE Int.Telecomm unications Energy Conf. P roceedings, pp. 334-341,1984.

[4] K.H. Liu, F.C. Lee, "Zero-voltage switching technique inDClDC converters, ' ' IEEE Power Electronics Spec ialists Conf.Rec ord, pp. 58-70, 1986.

[5] F.C. Lee, W.A. Tabisz, M.M. Jovan ovid , "Recent develop-ments in high-frequency quasi-resonant converter technolo-gies, '' European Power Electronics Conf., Aachen, Germany.

IF] W.A. Tabisz, F.C. Lee, "Zero-voltage-switching multi-resonant technique - A novel approach to improve perform-ance of high-frequency quasi-resonant converters," IEEETrans. Power Electronics, Vol. 4, no. 4, pp. 450-458, 1989.

171 W.A. Tabisz, F.C. Lee, "DC analysis and design of zero-voltage-switched multi-resonant converters," IEEE PowerElectronics Specialists Conf. Record, pp. 243-251, 1989.

[ R I W.A. Tabisz a nd F.C. Lee, "A no vel, zero-voltage-switchedmulti-resonant forward converter," High Frequency PowerCon vers ion Conf., pp . 309-318, 1988.

[Y] W.A. Tabisz and F.C. Lee, "Design of high-density on-boardsingle- and multiple-output multi-resonant converters," HighFrequency Power Conversion Conf., pp. 45-57, 1990.

[IO] W. A . Tabisz, M.M. Jovanovid , F.C. Lee, "High-frequencymulti-resonant converter technology and its applications,"Int. Conf. on Power Electronics and Variabk Speed Drives,pp. 1 -9, 1990. ~

PP. 401-410. 1989.

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